Emitter Switching for Power BJT Based LED Drivers

Home , Inductor

Yue Ma

Student number: 4329732

A DISSERATION PRESENTED

YUE MA

IN PARTIAL FULFUILMENT OF THE REQUIREMENTS

FOR THE DEGREE OF

MASTER OF SCIENCE

TRACK ELECTRICAL SUSTAINABLE ENERGY

ELECTRICAL ENGINEERING, MATHEMATICS AND COMPUTER SCIENCE

DELFT UNIVERSITY OF TECHNOLOGY

DELFT, THE NETHERLANDS

JUNE 2015

The thesis of Yue Ma was approved by the following:

Responsible professor: Prof. J.A. Ferreira

Daily supervisor: Dr. Jelena Popovic Dr. Yi Wang

MSc Thesis Committee:

Dr. Arno Smets Photovoltaic Materials and Devices, EEMSC, TU Delft

Dr. Jelena Popovic Electrical Power Processing, EEMCS, TU Delft

Dr. Jose Rueda Torres Intelligent Electrical Power Grids, EEMCS, TU Delft

Dr. Yi Wang LED Platform Development, Philips Lighting

TRACK ELECTRICAL SUSTAINABLE ENERGY

FACULTY OF ELECTRICAL ENGINEERING, MATHEMATICS AND COMPUTER SCIENCE

DELFT UNIVERSITY OF TECHNOLOGY

LED PLATFORM DEVELOPMENT, PHILIPS LIGHTING B.V.

An electronic version of this dissertation is available at: http://repository.tudelft.nl/ ii

Abstract

In recent years, the LED lighting developed rapidly because of its superior efficiency, eco-friendliness, and longer lifetime compared to the fluorescent and its incandescent counterparts. As the core of LED systems, LED drivers need to be well designed in order to properly drive the LEDs with high efficiency, high reliability, long lifetime, and last but not least low cost.

The major objective of this thesis project was to design an emitter switching power Bipolar Junction Transistor (BJT) based LED driver using the low-cost switch mode driving topology that can achieve high efficiency and low power dissipation. Using power BJT instead of power MOSFETs was because of the low cost feature. At the same time, there was a topology investigation on the low-cost high-quality driver. BIFRED was chosen due to the single switch structure.

An average model of the BIFRED was built to facilitate the analysis and design. In order to make the power BJT has the same switching performance as power MOSFETs, there was a literature study of the characteristic of the power BJT. Aiming at overcoming the disadvantage of power BJTs, the emitter switching technology was investigated. Next was the design of the emitter switching power BJT-based LED driver. The hardware of the LED driver was experimentally tested, and it showed a good performance with high efficiency of the driver and low power dissipation in power BJT.

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Table of contents

Abstract ...... iii

Table of contents ...... iv

List of figures ...... vii

List of tables...... ix

Acronyms ...... x

Acknowledgements ...... xi

Chapter 1 Introduction ...... 1

1.1 Background information ...... 1

1.2 LED driver: driving options ...... 1 1.2.1 Unregulated linear drivers ...... 2 1.2.2 Regulated linear driver ...... 3 1.2.3 Switch mode drivers ...... 4 1.2.4 Comparison ...... 5

1.3 Problem statement ...... 6

1.4 Research objective ...... 7

1.5 Research questions ...... 7

1.6 Approach and outline ...... 8 1.6.1 Research approach ...... 8 1.6.2 Thesis outline...... 9

Chapter 2 Characteristics of Power BJT ...... 10

2.1 Introduction ...... 10

2.2 Structure...... 11

2.3 풊 − 풗 characteristics ...... 12

2.4 Breakdown mechanism ...... 13 2.4.1 Primary breakdown ...... 13 2.4.2 Second breakdown ...... 15

2.5 Charge carriers distribution ...... 16 2.5.1 Active region ...... 17 2.5.2 Quasi-saturation region ...... 19 2.5.3 Hard saturation region ...... 21 2.5.4 Cutoff situation ...... 21

2.6 Switching characteristics ...... 22 2.6.1 Switching on ...... 22 2.6.2 Switching off ...... 26

2.7 Emitter switching technology ...... 33

2.8 Summary ...... 35

Chapter 3 Investigation of Switch Mode LED Driver Topology ...... 36

3.1 Introduction ...... 36

3.2 Typical LED driver topology ...... 36

3.3 RCC ...... 38 3.3.1 Structure ...... 39 3.3.2 Operation principle ...... 39

3.4 BIFRED ...... 48 3.4.1 Different operating modes ...... 49 3.4.2 Operation principle of the DCM-BCM BIFRED ...... 51

3.5 Summary ...... 53

Chapter 4 Average modeling of DCM-BCM BIFRED ...... 54

4.1 Introduction ...... 54

4.2 Average model of the ideal components BIFRED ...... 54 4.2.1 Consider the Constant DC input ...... 54 4.2.2 Analysis of the operation principle with rectified mains input ...... 59 4.2.3 Average model in LTspice ...... 61 4.2.4 Model verification ...... 63

4.3 Average model concerning the practical factors ...... 65 4.3.1 Identify key practical factors ...... 65 4.3.2 Storage time of power BJT ...... 67 4.3.3 Resonance effect ...... 69 4.3.4 Analysis of the operation principle including the practical factors ...... 75 4.3.5 Average model in LTspice ...... 79 4.3.6 Model verification ...... 81

4.4 Summary ...... 84

Chapter 5 Emitter switching LED driver design ...... 85

5.1 Introduction ...... 85

5.2 Power BJT-based LED driver ...... 85 5.2.1 Operation principle of the self-oscillating BIFRED ...... 87 5.2.2 LTspice simulation ...... 91

5.3 Emitter switching BIFRED LED driver design ...... 92

5.3.1 Operation principle of the emitter switching self-oscillating BIFRED LED driver ... 94 5.3.2 LTspice simulation ...... 99

5.4 Experimental result analysis ...... 100 5.4.1 Experimental waveform comparison ...... 100 5.4.2 Result analysis ...... 101

5.5 Summary ...... 103

Chapter 6 Conclusions and recommendations ...... 104

6.1 Conclusions ...... 104

6.2 Recommendations ...... 104

Reference ...... 106

List of figures

Figure 1-1 Factors affect the LED design ...... 2 Figure 1-2 Typical LED’s driving options ...... 2 Figure 1-3 Discrete component based drive ...... 3 Figure 1-4 Linear voltage regulator driver ...... 4 Figure 1-5 Linear current regulator driver ...... 4 Figure 1-6 A switch mode driver example (buck based LED driver) ...... 4 Figure 1-7 Thesis project research approach flow chart ...... 8 Figure 2-1 The circuit symbol of power BJT ...... 10 Figure 2-2 Structure of power BJT ...... 12 Figure 2-3 BJT operating 𝑖 − 푣 characteristics ...... 12 Figure 2-4 Experimental measurement of breakdown voltage (sample 1) ...... 14 Figure 2-5 Experimental measurement of breakdown voltage (sample 2) ...... 15 Figure 2-6 Current and voltage relationship of switch BJT ...... 16 Figure 2-7 Different regions of BJT ...... 17 Figure 2-8 Depletion region and minority charge carriers of BJT at active region ...... 18 Figure 2-9 Darlington transistor ...... 19 Figure 2-10 Formation of the quasi-saturation ...... 20 Figure 2-11 Depletion region and minority charge carriers of BJT at the edge of quasi- saturation region ...... 20 Figure 2-12 The depletion region and minority charge carriers of BJT at hard saturation region ...... 21 Figure 2-13 depletion region and minority charge carriers of BJT at cutoff situation ...... 22 Figure 2-14 Growth of the storage charge during switch-on transient ...... 23 Figure 2-15 Power BJT current and voltage waveforms during the turn-on process ...... 24 Figure 2-16 2D charge distribution during switch-on process ...... 25 Figure 2-17 Switch-on base drive effect ...... 25 Figure 2-18 Power BJT current and voltage waveforms during the turn-off process (Gradually change of 𝑖퐵) ...... 27 Figure 2-19 Decrease of the storage charge during switch-off transient (Gradually change of 𝑖퐵) ...... 28 Figure 2-20 Power BJT current and voltage waveforms during the turn-off process (Abruptly change of 𝑖퐵) ...... 29 Figure 2-21 Decrease of the storage charge during switch-off transient (Abruptly change of 𝑖퐵) ...... 30 Figure 2-22 2D charge distribution during switch-off process ...... 31 Figure 2-23 Switch-off base drive effect ...... 32 Figure 2-24 Emitter switching bipolar transistor ...... 34 Figure 2-25 Basic operation principle of emitter switching theory ...... 34 Figure 3-1 Major switch mode LED driver topologies ...... 37 Figure 3-2 Ringing Choke Converter (RCC) ...... 39 Figure 3-3 Start-up process...... 40 vii

Figure 3-4 On-state ...... 40 Figure 3-5 Turn-off process ...... 41 Figure 3-6 Off-state ...... 41 Figure 3-7 Current waveform of the turn-off process ...... 42 Figure 3-8 Circuit of the buck-boost converter ...... 43 Figure 3-9 Off-state of the buck-boost converter ...... 43 Figure 3-10 Buck-boost converter with the parasitic capacitor...... 44 Figure 3-11 LC resonance of Buck-boost converter (푉𝑖 = 푉표) ...... 45 Figure 3-12 Parasitic capacitor of the switch ...... 46 Figure 3-13 LC resonance (푉𝑖 < 푉표)...... 46 Figure 3-14 LC resonance (푉𝑖 > 푉표)...... 47 Figure 3-15 Circuit diagram of BIFRED ...... 48 Figure 3-16 Inductor current of BIFRED ...... 49 Figure 3-17 Waveforms of different operation mode ...... 50 Figure 3-18 Current waveform of DCM-BCM mode ...... 51 Figure 3-19 On-state of BIFRED ...... 52 Figure 3-20 Off-state 1 of BIFRED ...... 52 Figure 3-21 Off-state 2 of BIFRED ...... 53 Figure 4-1 Circuit diagram of BIFRED ...... 54 Figure 4-2 Waveform of a DCM-BCM BIFRED...... 55 Figure 4-3 Average model of BIFRED...... 58 Figure 4-4 average model ...... 62 Figure 4-5 BIFRED converter model with ideal switch ...... 63 Figure 4-6 3rd order RC low pass filter ...... 64 Figure 4-7 Waveform comparison ...... 64 Figure 4-8 Non-ideal factors of BIFRED ...... 65 Figure 4-9 Sensitivity curves of the input diode forward voltage ...... 66 Figure 4-10 Sensitivity curves of the output diode forward voltage ...... 66 Figure 4-11 Storage time measure waveform ...... 67 Figure 4-12 Storage time measured results ...... 68 Figure 4-13 storage time/ on-time characteristics ...... 68 Figure 4-14 Different periods of the rectified mains...... 70 Figure 4-15 Resonance waveform when input voltage is low ...... 71 Figure 4-16 Resonance waveform when input voltage is high ...... 72 Figure 4-17 Two-phase resonance ...... 73 Figure 4-18 Simplified LC resonant circuit ...... 74 Figure 4-19 Voltage waveform of the resonance ...... 74 Figure 4-20 Waveform of BIFRED include the practical factor ...... 76 Figure 4-21 Current and voltage waveform of power BJT ...... 77 Figure 4-22 Average model of the non-ideal situation ...... 80 Figure 4-23 Verification model of the non-ideal average model ...... 81 Figure 4-24 Digital function of the storage time delay ...... 82 Figure 4-25 Digital function of the resonance delay ...... 82 Figure 4-26 Waveform including the switching delay ...... 83 viii

Figure 4-27 Waveform comparison ...... 84 Figure 5-1 Circuit diagram of the BIFRED driver ...... 86 Figure 5-2 Structure of the self-oscillating BIFRED ...... 87 Figure 5-3 Start-up process...... 88 Figure 5-4 On-state ...... 88 Figure 5-5 Switch-off process ...... 89 Figure 5-6 Off-state 1 ...... 90 Figure 5-7 Off-state 2 ...... 90 Figure 5-8 Switch-on process ...... 91 Figure 5-9 Waveform of power BJT-based BIFRED (rectified mains cycle) ...... 91 Figure 5-10 Waveform of power BJT-based BIFRED (switching cycle) ...... 92 Figure 5-11 Circuit diagram of emitter switching BIFRED LED driver...... 94 Figure 5-12 Structure of the emitter switching BIFRED ...... 94 Figure 5-13 start-up process of ES BIFRED ...... 95 Figure 5-14 On state of ES BIFRED ...... 95 Figure 5-15 Turn off process of ES BIFRED ...... 96 Figure 5-16 Flow chart of the turn-off process ...... 96 Figure 5-17 Off state 1 (ES BIFRED) ...... 97 Figure 5-18 Off state 2 (ES BIFRED) ...... 97 Figure 5-19 Turn on process (ES BIFRED) ...... 98 Figure 5-20 Waveform of the ES BIFRED (rectified mains cycle) ...... 99 Figure 5-21 Waveform of the ES BIFRED (switching cycle) ...... 99 Figure 5-22 Waveform comparison ...... 100 Figure 5-23 Waveform comparison over one switching cycle ...... 100 Figure 5-24 Efficiency comparison ...... 101 Figure 5-25 Temperature comparison...... 102

List of tables

Table 1-1 Comparison between different driving options ...... 5 Table 4-1 Parameters in the waveform of DCM-BCM BIFRED ...... 56 Table 4-2 List of the controlled source in average model ...... 59 Table 4-3 List of the voltage ...... 72 Table 5-1 Efficiency comparison ...... 101 Table 5-2 Temperature comparison ...... 102

Acronyms

BJT Bipolar Junction Transistor

LED Light-Emitting Diode

MOSFET Metal–Oxide–Semiconductor Field-Effect Transistor

BIFRED Boost Integrated with Flyback/ Energy Storage/ DC-DC Converter

AC Alternating Current

DC Direct Current

2D Two Dimensional

DCM Discontinuous Conduction Mode

CCM Continuous Conduction Mode

BCM Boundary Conduction Mode

B/C/E of BJT Base/ Collector/ Emitter of BJT

SMPS Switch Mode Power Supply

ES Emitter Switching

RCC Ringing Choke Converter

PWM Pulse Width Modulation

IC Integrated Circuit

THD Total Harmonics Distortion

ZVS Zero Voltage Switching

IHQRR Integrated High Quality Rectifier Regulators

HV High Voltage

LV Low Voltage

Acknowledgements

Time flies. Now I still clearly remember the time when I started to do the thesis project last year. During this time, there are many people who helped me, supported me and encouraged me. I would like to take this opportunity to express my gratitude to them. First of all, I wish to express my sincere thanks to Dr. Jelena Popovic, my supervisor from TU Delft. There is a saying “Well begin, half done”, so I really appreciate her for giving me some guidance at the beginning of the thesis project that helped me to be clear of the whole process of the project. We had the monthly progress update meeting, she is always very patient listening to my presentation and giving me lots of advice. After I finished the draft thesis report, she gave me some feedbacks that were so exquisite that I was deeply impressed. I am really grateful to have such a good supervisor. I am also grateful to Dr. Yi Wang, my supervisor from Philips Lighting. He offered me the opportunity to do my thesis project at Philips, I am very appreciated to have this chance to work with all the great colleagues in this group. Yi Wang and Christiaan van Dijk joined weekly progress update meeting with me every week, they gave me advice and sometimes helped me to overcome difficulties. Special thanks to Dik Pardijs, Marcel Beij and all the other colleagues in LPD group, because they make me feel like working in a big family with a cozy atmosphere. I would like to thank my new friends who I met in Netherlands, and my old friends who are studying and working at different places in the world. Friendship is priceless, I really appreciate their support. I want to thank my parents who always concern about me for living and studying in a country that is thousands miles away from home with 6-hour time difference. I am so grateful for their unceasing encouragement, attention and support.

Chapter 1 Introduction

1.1 Background information

LED, a cost effective, energy efficient solution for lighting, is a key component of today’s technology. It is being considered as a promising illumination solution because of its superior efficiency, eco-friendliness, and longer lifetime compared to the fluorescent and the incandescent lightings [1].

It is predicted that over the next decade the value of the burgeoning LED market will exceed 15 billion US dollars per year [2]. With the development of technology and the innovation in manufacturing industries, the price of LED lightings keeps decreasing.

Compared with the conventional light sources, for example incandescent or fluorescent lamps, LED has some advantages. The most significant advantages are faster turn-on, lower heat generation, lower power consumption, higher operating life, etc. However LED does have some disadvantages, one of them is that LED requires electronic drive circuit for operation.

LED driver is the power supply for an LED system, which formed the core of the LED lighting system [3]. Finding the best solution for the low cost LED driver is one of the chief objectives now.

1.2 LED driver: driving options

LEDs need to be driven properly for good system efficiency, acceptable reliability, and lifetime. Designing and implementing an effective driver is the key to obtain all the benefits of LEDs [4].

When choosing the suitable driving method for a LED application, it is not sufficient to consider electrical characteristics only. 1

Thermal characteristics

Electrical Optical characteristics characteristics

LEDs

Figure 1-1 Factors affect the LED design

There are several methods to drive LEDs. The commonly used methods are: unregulated linear drivers based drive, regulated linear drivers based drive and switch mode drivers based drive. Different driving options are given in Figure 1-2.

Unregulated •Resistor based linear drivers •Transistor based

Regulated •Fixed voltage linear drivers •Constant current

Switch mode •Buck model •Boost/ SEPIC mode drivers •Buck-boost/ Flyback

Figure 1-2 Typical LED’s driving options

1.2.1 Unregulated linear drivers

The simplest method of the discrete component based drive is using a resistor in series with the supply voltage to limit the current, which shows in Figure 1-3. This method is

cheap and simple but the disadvantages are obvious. The most significant weakness is that if the input voltage changes there will be a variation in current flow through LED, which leads to poor illumination quality. For the situation of high line-voltage, it might results in the damage of LED device because of the overheating caused by the elevated current. It might degrade the LED’s quality and lifetime. And it has continuous power dissipation in the resistor, which leads to low efficiency.

Figure 1-3 Discrete component based drive

1.2.2 Regulated linear driver

Regulated linear driver based drive is another commonly used driving option, the circuit concept is shown in Figure 1-4 and Figure 1-5. It uses a linear regulator and a series resistor. The LED’s current is determined by the regulated output voltage of the linear regulator (e.g. voltage regulator) and the resistance of 푅1. So the current flow through LED is dependent on the performance of the regulator.

This type of driving method can provide good performance of LED. However the main disadvantage of this driver is low efficiency. It is because the regulated linear driver regulates the voltage by dissipating excess power as heat.

Figure 1-4 Linear voltage regulator driver Figure 1-5 Linear current regulator driver

1.2.3 Switch mode drivers

Figure 1-6 A switch mode driver example (buck based LED driver)

Figure 1-6 illustrates the basic structure of one type of switch mode driver: buck based LED driver.

Switch mode regulators offer the advantage of high efficient power conversion, especially in high power applications. It is because the switch of the switch mode regulator only operates in four different states: low-dissipation switch-on transition, full-on states, full-off state, and a very short high-dissipation switch-off transition. For the ideal situation, during the full-on state, the voltage across the switch is zero; while during the full-off state, there is no current flow through the switch. By optimizing SMPS design, the amount of power loss and heat can be minimized. 4

Typically, there are several switch mode driver topologies: buck, boost, buck-boost, flyback, etc. More information of these driver topologies will be given in Chapter 3.

1.2.4 Comparison

Each method has its own advantages and disadvantages. Table 1.1 [5] clearly illustrate the characteristics of each method from different aspects.

Unregulated Regulated linear Switch Mode

linear driver driver Regulator

푽 mismatch between LED 풇 NO YES YES addressed

푽 change due to 풇 NO YES YES temperature addressed

Source voltage variation NO YES YES addressed

Tight current regulation NO YES YES

Simple solution YES YES NO

YES (compared to Costly solution NO YES unregulated driver)

Efficient solution NO NO YES

Stable over wide-range of NO YES YES temperature

Table 1-1 Comparison between different driving options

The biggest advantages of a discrete based drive are low cost and simplicity, while the major disadvantages of a switch driver are the complexity of the design and the cost associated with it.

1.3 Problem statement

Optimized driver design is to meet the system specifications (efficiency, thermal characteristics, the size, EMI, lifetime) with low cost. Regardless of the complexity of the design and the cost, the switch mode regulator is the best choice.

However, in the design of LED driver, the cost cannot be ignored. There is a pressing need for low cost LED driver.

Over the past decades, bipolar power switching transistors have increasingly been replaced by power MOSFET transistors in switching power supplies. New designs in coming years will most frequently be conducted with MOSFETs [6]. The power bipolar junction transistor (BJT) is generally considered an outdated device due to its poor switching performance.

Recently, however, electronics market leaders in low-power AC/DC off-line power- supply designs have adopted BJT control ICs in applications such as high-volume cellular-phone chargers [7]. The key reason is because of the low cost. The power BJT is a technology that became mature in the mid-1960s, while the MOSFET did not become practical until the late 1970s. Fundamentally, BJTs cost less than power MOSFETs because their fabrication involves fewer layers and simpler processes than the power MOSFET.

For the LED system, application of power BJT helps to realize the goal of low cost compared to power MOSFET. However, for a good LED driver, high performance is also one of the key factors which need to be taken into consideration.

In summary, the problem statement can be defined to:

“Using power BJT as the switch of driver can realize a lower cost objective, however it tends to deteriorate the system efficiency due to its limited switching speed.”

1.4 Research objective

Following the problem statement in the previous section, the research objective is to:

“Find and investigate a low cost switch mode driving topology, and design a LED driver which can realize a good switching performance based on low-cost power BJT.”

1.5 Research questions

Several questions are raised below in order to achieve the research objective. This research will focus on the following questions:

1 What methods can be used to improve the switching performance of power BJT?

2 What kind of topology can reach the optimal trade-off between low-cost and high-efficiency?

3 What kind models are needed facilitate the important design parameters and optimize the driver operation?

4 If the solutions to overcome the problems of BJT are applied to the LED driver, how much improvement of efficiency could it reach compared to the topology only use BJT as the switch?

1.6 Approach and outline

1.6.1 Research approach

Topology Literature study Average Model investigation

Conclusions & Emitter Experimental recommendatio Switching evaluation ns driver design

Figure 1-7 Thesis project research approach flow chart

This research consists of several steps, the flow chart of the research approach is given in Figure 1-7.

First step is the literature study, which reviews the characteristics of power BJT and emitter switching technique.

Next is the LED driver topology study, which investigated the low cost driving topology and different operating modes. This step aims at finding the suitable driving solution for low cost driver.

The third one is the average modeling of Discontinuous-Boundary condition mode (DCM-BCM) BIFRED. This modeling is for analyzing of the switching characteristics of BIFRED driver.

After the modeling step comes the design part, which covers the SPICE simulation and experiment validations.

Next is the analysis of the results from the previous steps, then draws the conclusions and gives recommendations.

1.6.2 Thesis outline

Chapter 1 is the introduction chapter. It begins with the background information of the LED driver. Next section is the brief introduction of different driving methods. The problem statement and research objective are given, followed by the specified research questions. The flow chart of the research approach and thesis outline are given as well.

Chapter 2 reviews the characteristics of power BJT, which explains the poor switching performance of power BJT by examining its structure and dynamic charge distribution during operations.

Chapter 3 investigates switch-mode LED driver topologies. In this chapter, LED driver which can realize low-cost and high efficiency is identified. And operation principle is analyzed in details.

Chapter 4 presents the modeling approach of the BIFRED which operates in DCM- BCM. The modeling starts with the ideal situation, and then includes the key practical factors which cannot be ignored in the practical situation.

Chapter 5 elaborates the design of the DCM-BCM BIFRED driver using power BJT as the switch. Then this design is modified by applying the emitter switching technology and is followed with experimental results.

At last is Chapter 6 with the conclusions and recommendations.

Chapter 2 Characteristics of Power BJT

2.1 Introduction

In recent years, bipolar power transistors have increasingly been replaced by MOSFET transistors in switching power supplies. However, with the development of new technologies, the LED price drops spectacularly in the past years. There is a specific demand from the market for lower price drivers with comparable performance. BJT could better meet this demand than MOSFET since BJT has a significant advantage: low cost.

Figure 2-1 is the circuit symbol of power BJT, which is the same as the symbols of logic-level counterpart. However, in order to develop the ability of blocking high voltage in off-state and carrying high current in the on-state, the power BJT has some differences from the logical BJT, which will be explained in the following sections.

a) npn type b) pnp type

Figure 2-1 The circuit symbol of power BJT

In this chapter, the characteristics of the power BJT will be given in detail from different respects. It first starts with the structure of the power BJT, and explained why power BJT is so different with the logic level BJT. Then introduce the 𝑖 − 푣 characteristic of power BJT, which gives a brief idea of the different operating region. It follows with the explanation of different region: breakdown region, active region, quasi-saturation region, hard saturation region and the cut off region. After analyzing how the charge carrier distributes in different operating region, the switching characteristics are

explained.

2.2 Structure

The major difference of power BJT from the logic-level BJT is the need for large blocking voltage and high current-carrying capability. This means the power BJT must have a different structure.

The structure of the power BJT is shown in Figure 2-2. It has a four layer structure, which has different doping type and doping level at different layers.

The doping levels and layer thickness has a huge impact on the characteristics of the power BJT. Let’s take the npn BJT as an example. As can be seen from the figure, the doping level of emitter layer and collector layer are high (typically 1019푐푚−3). While the base layer only has moderate doping level (typically 1016푐푚−3 ). The layer between the base and collector is called drift region, which is with low doping level (typically 1014푐푚−3).

The thickness of the drift region is an important element which determines the breakdown voltage of the transistors.

The thickness of the base layer is a tradeoff between the amplification capability and breakdown voltage capability. In order to get a good amplification capability, the base layer should be as small as possible. However the layer thickness is too small, the capability of the breakdown voltage will be compromised. So the design of the base layer should get the optimal performance considering both of the two capabilities. Typically several micrometers to a few tens of micrometers [8].

Figure 2-2 Structure of power BJT

2.3 풊 − 풗 characteristics

Figure 2-3 shows the operating 𝑖 − 푣 characteristics of power BJT. The orange- colored line is the idealized operating characteristics. The idealized characteristics are the voltage is zero when BJT is at on-state and the current is zero when the BJT is at off-state. This means there is no power dissipation.

Figure 2-3 BJT operating 𝑖 − 푣 characteristics

However, in practice, the characteristics are more complex than the ideal one. The

practical characteristics are those black curves in Figure 2-3 with different base current. There are several different regions: active region, quasi-saturation, hard saturation region, cutoff region and the region for primary breakdown and secondary breakdown.

In Figure 2-3, there are some featured values that should be noted.

1) 퐵푉푆푈푆 is the maximum 푉퐶퐸 that can be sustained when the transistor is

carrying essential 퐼퐶.

2) 퐵푉퐶퐸푂 is the collector-emitter breakdown voltage when the base is open circuit. This breakdown voltage is always used to judge the transistor’s voltage standoff capability.

3) 퐵푉퐶퐵푂, similar to 퐵푉퐶퐸푂, is the collector-base breakdown voltage when the

emitter is open circuit. There is a fact that 퐵푉퐶퐵푂 is larger than 퐵푉퐶퐸푂. Using this advantage, the circuit could be modified as the open-emitter transistor turn-off circuit, which has higher breakdown voltage.

The details of these different operation regions are given in the following sections.

2.4 Breakdown mechanism

2.4.1 Primary breakdown

The region which is labeled primary breakdown is due to conventional avalanche breakdown of C-B junction. The primary breakdown will lead to large flow of current, which should be avoided because of the accompanied large power dissipation.

The primary breakdown voltage also depends on the chosen circuit configuration:

(1) In an open base mode;

(2) In an open emitter mode.

The breakdown voltage of the open base mode is 퐵푉퐶퐸푂, and the breakdown voltage of the open emitter mode is 퐵푉퐶퐵푂.

As it is been indicated in section 2.3, there is a fact that 퐵푉퐶퐵푂 is larger than 퐵푉퐶퐸푂. 13

This relationship can be found in Figure 2-3. There is a semi-empirical relationship between these two parameters [9]:

퐵푉 퐵푉 = 퐶퐵푂 (2-1) 퐶퐸푂 (훽 + 1)1/푛

Where 훽 is the current gain of the transistor. 푛 = 4 for npn transistors and 푛 = 6 for pnp transistors.

For the npn type of the power transistor, 훽 is typically between 10 and 20. So the value of 퐵푉퐶퐸푂 could be around a half of 퐵푉퐶퐵푂.

This semi-empirical equation can also be proven by experiments. The breakdown voltages are measured by the curve tracer. In the experiments, I tested the breakdown voltages 퐵푉퐶퐸푂 and 퐵푉퐶퐵푂 of two different types of power BJT.

The first type of power BJT is 3DD4242DM. For this type, 6 different samples are been measured.

3DD4242DM 1000 900 퐵푉 800 퐶퐵푂 700 600 500 퐵푉퐶퐸푂 400 300

200 Breakdown Breakdown voltage [V] 100 0 0 20 40 60 80 100 120 140 160 BJT temperature [C]

Figure 2-4 Experimental measurement of breakdown voltage (sample 1)

Sample 2 is 3DD4520A6. For this type, 4 different samples are been measured.

3DD4520A6 1000 900 퐵푉퐶퐵푂 800 700 600 퐵푉 500 퐶퐸푂 400 300

200 Breakdown Breakdown voltage [V] 100 0 0 20 40 60 80 100 120 140 160 BJT temperature [C]

Figure 2-5 Experimental measurement of breakdown voltage (sample 2)

From Figure 2-4 and Figure 2-5, it can be concluded that 퐵푉퐶퐵푂 is higher than 퐵푉퐶퐸푂.

This characteristic of 퐵푉퐶퐵푂 will be used in the emitter switching circuit, which is also called the open-emitter transistor turn-off circuits. More detailed information about emitter switching will be given in section 2.7.

2.4.2 Second breakdown

Power BJT has a potential failure mode, normally termed second breakdown. It usually appears during the switching process. For example, during the BJT turn on, there is a drop in collector-emitter voltage with an increase in collector current. So there will be a substantial increase in the power dissipation. For the switch off process is vice versa, which can be seen from Figure 2-6.

푡 Switch-on On-state Switch-off Off-state transition transition Figure 2-6 Current and voltage relationship of switch BJT

What makes the power dissipation particularly dangerous is because the dissipation is not uniformly spread over the BJT device, which means it will concentrate in localized region. This will lead to the rise of temperature in local area to a very high value. This may even result in damage of the power BJT device [10].

There are two distinct second breakdown modes are observed in BJT: thermal mode second breakdown and current mode second breakdown [11]. Both of the two modes of the second breakdown need to be avoided. BJT can be protected by the rapid diversion of power flow away from the point of local thermal concentration or local current concentration.

In summary, there are mainly two ways to avoid second breakdown:

(1) Keep the total power dissipation under control;

(2) Avoid any current density concentration.

It can be seen that for the design of the LED driver, the second breakdown of BJT is a major issue that need to be avoided. The key to solve the problem is to design a driver circuit that has the ability to control the power dissipation of BJT.

2.5 Charge carriers distribution

Because of the different doping type and doping level of different layers, depletion

regions are formed. In order to understand the difference of active region, quasi- saturation region and hard saturation region, knowing the distribution of charge carriers is of great importance. Figure 2-7 shows the E-B depletion region and C-B depletion region without external voltage, which is the intrinsic situation. For simple analysis, the lightly doped drift region can be seen as part of the collector. The depletion region of C-B is assumed to be across the whole lightly doped drift region from collector to base [8].

Figure 2-7 Different regions of BJT

2.5.1 Active region

In the active region, the E-B junction is forward biased, while the C-B junction is reversed biased. Electrons from the emitter are injected into the base, at the same time holes from the base are injected to the emitter.it can be seen from Figure 2-8, the depletion region of E-B is thinner, whilst the depletion region of C-B is wider. The distribution of the minority charge carriers is also given in Figure 2-8.

Figure 2-8 Depletion region and minority charge carriers of BJT at active region

Those electrons, which are injected to the base from the emitter, are more likely to escape from the base to collector instead of escaping from the base terminal. There are three reasons for this phenomenon:

a) The thickness of the base is rather small compared to the electron diffusion length. It means there are few opportunities for electrons to recombine in this region; b) The large dimension of the collector, as well as the narrow base region, make the possibility is higher that the electrons reach the collector area; c) The electrons in the base have large gradient, which can be seen from Figure 2.3.2. The density gradient carries most of the injected electrons, and very few of them escape through the base terminal.

Because few electrons are exiting via the base terminal, the emitter current is much larger than the base current. So a small amount of base current can cause a large current which flow between emitter and collector. If the base current 퐼퐵 is chosen to be the input current and the collector current 퐼퐶 as output current, there will be a current gain:

퐼퐶 = 훽퐼퐵 where 훽 is the current gain of BJT.

Now the principle of current gain is clear, so the methods to increase the current gain 훽 can be easily concluded:

a) Increase the doping level of the emitter; b) Extend the minority carrier life time in the base; c) Make the base thickness smaller.

As mentioned in previous section, increase of the current gain will conflict with other wanted characteristics, for instance the ability to withstand high breakdown voltage. So the compromise between these two aspects is made: larger base thickness than the logic- level counterpart. Due to the small typical 훽 value, Darlington pairs are commonly used, which is given in Figure 2-9.

Figure 2-9 Darlington transistor

2.5.2 Quasi-saturation region

To analyze the quasi-saturation, the active region is taken as the initial condition, which begins with the base current increases. The formation of quasi-saturation is given as a flow chart in Figure 2-10.

𝑖퐵 ↑ voltage drop in drift region↑ (ohmic resistance)

𝑖퐶 ↑ reverse biased C-B junction↓

voltage drop across the collector load↑ holes from B begins to be injected to C

푣 ↓ electroncs begin to be injected to the 퐶퐸 drift region

Figure 2-10 Formation of the quasi-saturation

When the excess carriers begin to be built up in the drift region, the quasi-saturation region is entered. Figure 2-11 shows the charge carrier distribution when the BJT is entering the quasi-saturation region, which is the edge between the active region and the quasi-saturation region. Briefly speaking, in quasi-saturation region, E-B junction is forward biased, and C-B junction begins to be forward biased.

Figure 2-11 Depletion region and minority charge carriers of BJT at the edge of quasi-saturation region

The boundary between the quasi-saturation region and the active region can be given by:

푣퐶퐸 𝑖퐶 = 푅푑

where 푅푑 is the ohmic resistance of the drift region.

2.5.3 Hard saturation region

When the excess carriers reaches the edge between the drift region and the collector region (the boundary of 푛− and 푛+), the hard saturation occurs. Figure 2-12 shows the minority carrier distribution at the hard saturation region.

Figure 2-12 The depletion region and minority charge carriers of BJT at hard saturation region

푄1 in Figure 2-12 is the minimum stored charge of hard saturation. The additional stored charge 푄2 will push BJT deeper into hard saturation, which causes overdrive.

2.5.4 Cutoff situation

Cutoff is the case when E-B junction and C-B junction are both reverse biased. If the transistor enters this region, it means the transistor is in off-state. This situation is easier than the previous ones. The depletion region and minority charge carrier distribution of cutoff are given in Figure 2-13.

Figure 2-13 depletion region and minority charge carriers of BJT at cutoff situation

2.6 Switching characteristics

Normally, BJT can be used in many circuit configurations. It can be an amplifier, or a switch. In section 2.5, the different operating regions have been explained. If the BJT operates in the active region, it is been used as an amplifier; if the BJT operates in the saturation region and the cut-off region, it is been used as a switch.

Since the LED driver is a switch mode power supply (SMPS). The use of power BJT is as a switch. It is of great importance to understand the switching characteristics of it. The details of the characteristics of switching on and switching off will be given in the following part.

2.6.1 Switching on

From the description in section 2.5, it can be seen that in order to switch the BJT from off state to on state, there need to be sufficient charge supply that can make the BJT enter the quasi-saturation region and hard-saturation region.

The approximate pattern of the storage charge distribution grows during the switch-on process is given in Figure 2-14. There is a forward-biased base current applied at 푡 = 0.

Figure 2-14 Growth of the storage charge during switch-on transient

First the charge established in emitter and base but not in collector, which is because of the discharge of the B-E space negative charge. During this time interval, only the base current flows and only the B-E voltage changes. This time interval is called the turn-on delay time 푡푑(표푛), of which there is no built-up of storage charge. This time interval is given in Figure 2-15.

Figure 2-15 Power BJT current and voltage waveforms during the turn-on process

After the turn-on delay time, the B-E junction is forward biased. Then the growth of the storage charge proceeds. And the collector current begins to rise until it reaches its on-state value. The time interval of the rise of collector current is called the current rise time 푡푟푖.

After the current rise time, the collector-emitter voltage falls. From Figure 2-15, it can be found that the drop of C-E voltage has two different phase. In the 푡푓푣1 interval, the transistor is still in active region, the C-E voltage drops fast. After this interval, the transistor enters the quasi-saturation region. Because of the reduction of 훽, the slope of the C-E voltage decrease is lower. Then at the end of the 푡푓푣2 interval, the hard- saturation region is entered, which means the transistor is in on-state.

For a more clear understanding of the charge carrier distribution during the switch-on process, the following 2D figures are given in Figure 2-16 [12].

B E B B E B B E B

C C C a. Active region b. Quasi-saturation c. Hard saturation

Figure 2-16 2D charge distribution during switch-on process

Different base drive leads to different switch-on process.

𝑖퐵 푡

푣퐶퐸 푡

a. Gradually increase of base current (slow turn-on)

𝑖퐵 푡

푣퐶퐸 푡

b. Ramped increase of base current (medium turn-on)

𝑖퐵 푡

푣퐶퐸 푡

c. Abruptly increase of base current (fast turn-on)

Figure 2-17 Switch-on base drive effect 25

Figure 2-17 shows three possible base drive current waveforms and resulting collector voltage waveforms associated with each [13]. 2-17-a is an inefficient base drive where the current rise is relatively slow, which cause power dissipation during switch-on. 2- 17-b shows a better drive current that the base current increase faster, which allows the C-E voltage drop more rapidly toward saturation. 2-17-c shows the best method for achieving the rapid turn-on of BJT, which is overdrive the base during the switch-on interval. It has the lowest power dissipation compared to the previous ones.

2.6.2 Switching off

The switch-off process is the opposite of the switch-on process. It means during switching off, the storage charge in the transistor need to be removed. Normally it is applied by a negative base current. It is because if reduce the base current to zero, only the internal recombination process occurs in the transistor, which will take a really long time to remove all the storage charge.

There are two ways to drive the base current to negative value:

(1) Abruptly change to negative (e.g. step function);

(2) Gradually change to negative (e.g. ramped with a controlled 푑𝑖퐵⁄푑푡).

The waveforms of current and voltage during the switch-off process are given in Figure 2-18 and Figure 2-20. These two figures indicate the waveform response to the different base drive.

Figure 2-18 Power BJT current and voltage waveforms during the turn-off process (Gradually change of 𝑖퐵)

In Figure 2-18, the switch-off process starts with the change of the controlled base current from positive to negative value. First, there is a time interval called storage time, which is labeled 푡푠. During this interval, the collector current keeps the on-state value. It is because the transistor still operates in the hard-saturation region, and the excess storage charge 푄2 (refer to Figure 2-12) is being removed.

After this storage time interval, the transistor enters the quasi-saturation. Because of the low value of 훽, the C-E voltage begins to increase with a shallow slope. After the time interval 푡푟푣1, the transistor leaves the quasi-saturation region and enters the active region. Since the value of 훽 is high in the active region, the C-E voltage increases with a much steeper slope. After the 푡푟푣2 interval, the C-E voltage stops increase, while the collector current begins to decrease. After the time interval of 푡푓푖, the rest of the storage charge is been removed and the collector current becomes zero. Then the transistor enters the cut-off, which is the off-state of the transistor.

The storage charge distribution is the same as Figure 2-14, only need to change the 27

direction of time, which is given in Figure 2-19.

Figure 2-19 Decrease of the storage charge during switch-off transient (Gradually change of 𝑖퐵)

The switching-off process given above is when the base current is ramped with a controlled 푑𝑖퐵⁄푑푡. If the base current is given with a very fast transition, there will be a significant difference in the collector current response from the gradually changed base current situation. This case is given in Figure 2-20.

Figure 2-20 Power BJT current and voltage waveforms during the turn-off process (Abruptly change of 𝑖퐵)

In this case, the storage time is shortened. And the time intervals 푡푟푣1 and 푡푟푣2 also shortened because of the large negative base current, which is given in Figure 2-21. There will be more storage charge in base region being removed. However, the storage charge in the drift region will not be removed with same proportion. It is because the storage charge in the drift region is mostly removed by the collector current. Since the collector current begins to decrease very early, it will lead to the storage charge left in the drift region. And the only ways to remove these left storage charge is the internal recombination and by the negative base current, of which the value is very small compared to the collector current. This produces the current tail in the interval 푡푓푖2. This tail will lead to power dissipation during switching off, which is undesirable and need to be avoided.

Compare the different cases with the switch of process. We want to shorten the storage time, but if the storage time is too short, it will lead to the current tail. Just like a Chinese

saying “Things will develop in the opposite direction when they become extreme”. We need to know there is a tradeoff between the situations, and we can find the optimal base drive method.

Figure 2-21 Decrease of the storage charge during switch-off transient (Abruptly change of 𝑖퐵)

Figure 2-21 shows the decrease of the storage charge during the switch-off transient. In order to make this process more clearly, the 2D charge distribution is given in Figure 2-22.

B E B B E B

C C a. Hard-saturation b. Quasi-saturation B E B B E B

C C c. Active region with remained charge d. Remained storage charge in drift in drift region region

Figure 2-22 2D charge distribution during switch-off process

𝑖퐵 푡 𝑖퐶 푣퐶퐸

푡

a. Long storage time

𝑖퐵 푡 𝑖퐶 푣퐶퐸

푡

b. Optimum

𝑖퐵 푡 𝑖퐶 푣퐶퐸

푡

c. Long current tail

Figure 2-23 Switch-off base drive effect

Figure 2-23 shows the effect of different base drive to the switch-off process. It can be concluded from the figures, increasing reverse base drive reduces the storage time. However the optimal drive is not the one with very high reverse base drive, but the intermediate between low and high reverse drive [14]. 2-21-a is the situation that the base current decrease with a relatively low slope. There is no current tail in this situation

but the storage time is long, which results in power dissipation. 2-21-c is the situation that the base current decrease abruptly that the storage time is short. However, this leads to long current tail. The best situation is 2-21-b, there is no current tail and has a relatively short storage time.

In can be concluded from section 2.6 that: in order to minimize the power dissipation of power BJT, the switching transition of turn-on and turn-off need to be minimized. For the turn-on transition, better to apply the overdrive base current; while for the turn- off transition, find the optimal reverse drive of the base current is the key.

2.7 Emitter switching technology

As mentioned in section 2.4.1, there is an open-emitter turn-off technology called emitter switching.

The “emitter switching” concept was widely investigated a few decades ago with the aim to improve the trade-off among the forward voltage drop, the breakdown voltage, and the switching speed of a power [15]. The emitter switching BJT having high breakdown voltage, low conduction drop and high frequency switching capability, represents the only solution to increase the efficiency and showing the a competitive cost at the same time [16].

In general, it is a combination of high voltage BJT and low voltage switching device. There is concept of “cascode connection” [17] meaning placing a low voltage fast switching MOSFET or BJT at the emitter of the high voltage BJT. This configuration has the advantages of both devices, which can provide great characteristics that are not able to reach by either one of itself [18]. The symbols of this cascode structure are given in Figure 2-24.

a) cascode with low voltage MOSFET b) cascode with low voltage BJT

Figure 2-24 Emitter switching bipolar transistor

The basic operation principle of the emitter switching bipolar transistor is given in Figure 2-25 [19]. For easier understanding, the low voltage switching device, either MOSFET or BJT, is represented by the symbol of a switch.

Collector Collector 𝑖퐵 ≈ 𝑖푐 𝑖푐

Base Base 𝑖퐵 Emitter Emitter

𝑖퐸

a. On-state b. Off-state

Figure 2-25 Basic operation principle of emitter switching theory

If the emitter switching BJT operates at on-state, which is the condition that both the power BJT and the low voltage switch is conduction. The behavior of the emitter

switching BJT is similar to that of a power BJT. With the drive of the base current, current flows from collector to emitter, with the current relation shows in Figure 2-25- a:

𝑖퐶 = 훽𝑖퐵 (2-2)

When the low voltage switch is switched off, the emitter current is been cut off. At this moment, all the collector current diverted to the base terminal. Just as it shows in Figure 2-25-b, the current relation is:

𝑖퐵 ≈ 𝑖푐 (2-3)

Because of this floating emitter configuration, the high value of the reversed base current results in the very fast removing of the storage charge. This achieves a very fast turn-off process. What is more important is this configuration results free from the current tail that characterize the power BJT device (refers to section 2.6.2). So this configuration provides an extra safety margin in reverse safe operating area, by increasing the ruggedness versus the secondary breakdown.

As mention in section 2.4.1, this configuration is called the open-emitter transistor turn- off circuits. The primary breakdown voltage of this configuration is higher, which also provides an extra safety margin in reverse safe operating area.

2.8 Summary

In summary, the special structure of the power BJT makes it has the potential failure mechanism of breakdown. An effective way to avoid the breakdown of BJT is keeping the total power dissipation under control. After analyzing the switching characteristics of BJT from the perspective of the charge carrier distribution, the proper driving of BJT can be concluded to minimize the switching transition, and reach the goal of minimizing the power dissipation.

Chapter 3 Investigation of Switch Mode LED Driver Topology

3.1 Introduction

Switch mode drivers offer the advantages of more efficient power conversion. So the

LED driver in this thesis project will use this switch mode power supply.

There are some typical LED driver topology, for instance Buck, Boost, Flyack, etc.

These topologies will be described briefly in section 3.2.

In order to achieve the objective of the low cost, a simple driver topology called Ringing

Choke Converter (RCC) is been introduced. RCC is a robust, low component-count circuit that has been widely used in low power off-line applications. The detailed operation principle will be explained in section 3.3.

Next a low cost topology named BIFRED is been introduced. BIFRED refers to the

Boost Integrated with Flyback Rectifier/ Energy storage/ DC-DC converter. As the name suggests, the BIFRED converter consists of a boost circuit and a flyback circuit with only one switch is required. It offers a low cost alternative to the conventional methods of the LED driver. A brief introduction of BIFRED will be given in section

3.4. More detailed research of the BIFRED will be given in the following chapters.

3.2 Typical LED driver topology

There are several major switch mode driver topologies that are commonly used in LED systems:

(1) Buck. These are step down regulators; the output voltage is lower than the input

voltage.

(2) Boost. These are step up regulators; the output voltage is higher than the input. 36

(3) Buck-boost. These are step down/step up regulators; the output voltage is

inverted.

(4) Flyback. These are step up or step down regulators with a transformer instead

of an inductor.

In the following section, the introduction of each driver topology will be given briefly.

a. Buck b. Boost

c. Buck-boost d. Flyback

Figure 3-1 Major switch mode LED driver topologies

Figure 3-1-a is buck converter. It is also named the step-down converter, is a regulator that has the lower output voltage than the input voltage.

Buck converter can achieve the voltage regulation from high voltage to low voltage, while the boost converter works in the opposite. Boost converter, which is given in

Figure 3-1-b, has the output voltage which has higher value than the input voltage. It is also called the “step up” converter.

Now both “step up” and “step down” converters are introduced. However, these converter have the limit of the input/output voltage ratio, the ratio can only be higher than one or lower than one. Buck-boost is a converter that has the output voltage can be either greater or less than the input voltage. The circuit diagram is shown in Figure

3-1-c. 37

There is one thing that need to note, the buck-boost is an inverting topology, which means the output voltage has the opposite polarity against the input voltage. Whether the value of the output voltage is higher or lower than the input voltage, it depends on the duty ratio and the operation mode.

Figure 3-1-d shows the flyback converter. Flyback is derived from the buck-boost converter. If compare the circuit of the buck-boost and the flyback in Figure 3-1, the only difference is the flyback use a transformer instead of an inductor. And the transformer is an inverse coupled structure. So the output voltage of the flyback converter has the same polarity as the input voltage

Most of the switch mode LED driver is based on these four basic topologies

3.3 RCC

RCC (Ringing Chock Converter) is a self-oscillating flyback converter [20]. As the name suggests, RCC is a flyback-based converter. What makes it very popular is mainly because the overall RCC cost is relatively low compared to the conventional Pulse

Width Modulation (PWM) IC flyback converter [21].

What makes the cost low is because the RCC control is achieved by using discrete components to control the peak current mode, which results in low component count and low cost [22]. As a result, RCC is widely used for cost-sensitive applications as a simple and cost-effective solution [23].

3.3.1 Structure

Start-up section Flyback converter section Self-oscillating section

Figure 3-2 Ringing Choke Converter (RCC)

The circuit diagram of RCC is given in Figure 3-2. This circuit has three sections: the start-up section, the flyback converter section and the self-oscillating section.

The core of the start-up section is the start-up resistor 푅1. The flyback converter section is the main body of RCC. The self-oscillating section controls the switch 푄1.

The detailed operation principle will be given in the following section.

3.3.2 Operation principle

The circuit structure is simple, but the operation principle is complex that is generally not well understood.

The operation of RCC is composed of three processes: start up, turn-off and turn-on.

After the start of RCC, the circuit has the ability to complete the self-oscillating process.

3.3.2.1 Start up

Before the start operation of RCC, the transistor is at off state. When the input voltage is applied, first the capacitor 퐶1 is being charged, so the base voltage increases. When 39

the voltage at the base of BJT is higher than the threshold voltage of BE junction, the

BE junction is forward biased and transistor is at the active region. When there is enough stored charge in the transistor, the BJT is on.

Figure 3-3 Start-up process

Figure 3-4 On-state

When BJT enters the on state means the end of the startup process. After the startup process, the start resistor 푅1 is with less influence on the circuit. 푅1 has a really high value, normally hundreds of kilo ohm to several mega ohm. One of the reasons is the base current should be controlled within a proper value in case of overdrive. Another reason is after the start process, the high value resistor can be regarded as open-circuit, which has little effect on the circuit.

3.3.2.2 Turn-off process

Figure 3-5 Turn-off process

Figure 3-6 Off-state 41

The moment that the switch is on, the collector current 𝑖푐 increases linearly. When

𝑖푐 = 𝑖퐵 ∙ ℎ퐸퐹, it means the transistor has reached the point of the expected collector current. The current 𝑖푐 has a trend of continuing increase with the same slope, but the transistor itself limits the collector current, and make 𝑖푐 constant.

푑푖 The voltage of the inductor 퐿 is 푉 = 퐿 퐿1. When the switch is on, as can be seen 1 퐿1 1 푑푡 from the circuit, the collector current 𝑖푐 is equal to the current flows through the inductor, so the equation can be written as:

푑𝑖 푉 = 퐿 푐 퐿1 푑푡

So once the current becomes constant, from the equation above we can see that 푉퐿1 =

0. So there is a steep decrease of 푉퐿1. Because the inductor 퐿2 is in coupled with 퐿1, so there is also a steep decrease of 푉퐿2. This means the base voltage will decrease very fast. This will result in the turn-off of the transistor.

At the same time, because the collector current 𝑖푐 want to keep constant instead of increasing linearly, there is a need of decreasing the base current 𝑖퐵 . When 𝑖퐵 decreases, there is a positive feedback on these currents. This lead to the steep decrease of both 𝑖퐵 and 𝑖푐. At last, the switch will turn off after 𝑖퐵 = 0.

Figure 3-7 Current waveform of the turn-off process

3.3.2.3 Turn-on process

In order to understand the turn-on principle of RCC, first start with the analysis of the

BCM buck-boost converter. Figure 3-8 is the circuit of the buck-boost converter.

+ + − 푉 퐿 푉표 푉푖 − + −

Figure 3-8 Circuit of the buck-boost converter

From the circuit, we can see that the voltage of the inductor 푉퐿 is

푉푖 푠푤𝑖푡푐ℎ 푂푁 푉퐿 = { −푉표 푠푤𝑖푡푐ℎ 푂퐹퐹 푉 퐷 표 = 푉푖 1 − 퐷

Choose 퐷 = 0.5 for simple analysis, it can be concluded that 푉표 = 푉푖.

When the switch is off, the inductor is still conducting, and release energy to the load.

Then the current will gradually decrease to zero.

Figure 3-9 Off-state of the buck-boost converter

When the current 𝑖퐿 = 0, however, the voltage 푉퐿 ≠ 0. Because of the existence of the parasitic capacitance, the inductance and the parasitic capacitance of the switch form a

basic LC resonant circuit. The capacitor 퐶2 in Figure 3-10 is the parasitic capacitor, together with the inductor 퐿1 form the LC resonant circuit.

LC resonant circuit

Figure 3-10 Buck-boost converter with the parasitic capacitor

For a practical model, LC circuit always has energy loss because of the resistance within the component and connecting wires. For simple analysis, it can be considered as the idealized model, of which there is no dissipation of energy. The principle of the LC resonant circuit is using the storage characteristics of the inductor and capacitor; make the energy transfer between L and C and keep oscillating without damping. The waveforms of the inductor voltage 푉퐿 and inductance current 𝑖퐿 are both sinusoidal, and there is a phase difference of 90°.

It can be concluded that after 𝑖퐿 reaching zero, the circuit can oscillates.

Figure 3-11 LC resonance of Buck-boost converter (푉푖 = 푉표)

If the transistor turn-on again at the moment 푣퐿 = 푉푖, the voltage across the switch is zero. This is called the zero-voltage switching (ZVS) [24]. In this situation, the switching loss during the switch-on process is zero. So for the design of the LED driver, this ZVS is desired.

However in some situation, it is not possible to achieve the zero-voltage switching.

Then the valley switching is required. This will be explained in the following section.

Now we look back to the RCC. For the RCC circuit, this LC oscillating also exits.

LC resonant circuit

Figure 3-12 Parasitic capacitor of the switch

Figure 3-13 LC resonance (푉푖 < 푉표)

Figure 3-14 LC resonance (푉푖 > 푉표)

During the resonance process, the voltage of the inductor 퐿2 will increase from negative to positive. So the voltage of the coupled inductor will also increase to positive.

The base voltage increases with the change of 푣퐿2. Once the base current is higher than the threshold voltage of BE junction, which is the same process given in section 3.3.2.1, the transistor will turn on again. Then the transistor will achieve the self-oscillating operation mode which makes this ringing choke converter operates.

Figure 3-13 and Figure 3-14 show the different situation of the resonance. If Vi < Vo, the ideal switching moment is when 푣퐿1 = Vi. It is the same as the case in Figure 3-11, which is to reach the zero-voltage switching. If Vi > Vo, it will never have the moment of 푣퐿1 = Vi. In order to minimize the switching loss, the best switching moment in this case is when the difference between 푣퐿1 and Vi is the smallest. It is called the valley switching. It is the moment that 푣퐿1 = Vo in Figure 3-14.

The valley switching point is the desired switching-on moment. However, in practice, it is not I want the valley switching point then it can switch at that point. It need to be tuned with the parameters of the circuit components.

3.4 BIFRED

BIFRED refers to Boost Integrated with Flyback Rectifier/ Energy storage/ DC-DC converter. Owing to the growing concern regarding harmonic pollution of the power distribution system, and the adoption of standards, there is a need for single-phase power supplies with ac line currents that are low in harmonic content and have power factor close to unity [25]. Harmonic pollution of power distribution systems and harmonic standards make BIFRED converters attract more attention. It has been shown that BIFRED converters is suitable for power factor correction. Also from the cost point of view, BIFRED seems particularly attractive among the single-step approaches.

Figure 3-15shows the circuit diagram of BIFRED. There is only one switch 푆1, which is the switch for both boost converter and the flyback converter.

The inductor 퐿1, the switch 푆1, the output diode 퐷2, the capacitor 퐶1 and the LED load 퐷3 together compose the boost power stage. The switch 푆1, the energy storage capacitor 퐶2, the coupled inductor windings 퐿1 and 퐿2, and the output part 퐷2, 퐶1,

퐷3 together compose the flyback power stage.

Storage capacitor Flyback converter section

Boost conveter section

Figure 3-15 Circuit diagram of BIFRED

3.4.1 Different operating modes

풊푳ퟏ 풊푳ퟐ

Figure 3-16 Inductor current of BIFRED

Since BIFRED is composed of two converters, each converter has an inductor. Each inductor can operates in CCM (Continuous Conduction Mode) or in DCM

(Discontinuous Conduction Mode). So there are four different operating mode combination in total: DCM-CCM, DCM-DCM, CCM-DCM and CCM-CCM. The different operation mode are given in Figure 3-17 [26].

a. CCM-CCM b. CCM-DCM

c. DCM-CCM d. DCM-DCM

Figure 3-17 Waveforms of different operation mode

However, the BIFRED converter should be designed such that the input inductor current 𝑖퐿1 becomes discontinuous before that of the transformer magnetizing inductor

𝑖퐿2. If this condition is not satisfied, 𝑖퐿1 becomes continuous at the peak of the input voltage, resulting in a surge of current, which leading to an increase in the input current total harmonic distortion (THD) [27].

The discontinuous-conduction-mode (DCM) of the current 𝑖퐿1 avoids the light-load high-voltage stress problem associated with the continuous-conduction-mode design, while still achieving the combined advantages of a low-cost single-stage topology with high displacement factor and low total harmonic distortion [28]. To achieve low harmonic distortion in the input current, 퐿1 must be operated in the DCM. It means the input current 𝑖퐿1 must reach zero before the switch 푆1 is turned on again.

Generally, 퐿2 can be operated in the DCM or the CCM.

The control of the switch is the key for the operation mode. As mentioned in section

3.3, there is a control method called self-oscillating in RCC. This self-oscillating can also be applied to BIFRED. Like RCC, the self-oscillating BIFRED has the advantage of low cost.

Most of the research of BIFRED is focus on the DCM-DCM mode or the DCM-CCM mode [29]. While this thesis research is focused on the mode that 퐿1 operates in the

DCM and 퐿2 operates in the BCM (DCM-BCM mode) because of applying the self- oscillating control method.

3.4.2 Operation principle of the DCM-BCM BIFRED

Figure 3-18 Current waveform of DCM-BCM mode

Figure 3-18 shows the current waveform of DCM-BCM BIFRED. The detailed operation principle is given in the following section.

3.4.2.1 On-state

Figure 3-19 On-state of BIFRED

When the switch is on, both inductor 퐿1 and 퐿2 conduct. The current of the switch is the sum of the current of the inductors. The voltage polarity of 퐿2 and 퐿3 is given in

Figure 3-19. Due to the polarity, the output diode 퐷2 in reserve biased. But the energy stored in the output capacitor during off-state can offer energy to the load.

3.4.2.2 Off-state

Since the operating mode is DCM-BCM, there are two different phases of the off-state.

The first phase is given in Figure 3-20, it is the time interval that both 퐿1 and 퐿2 conduct. In this interval, the voltage polarity of the 퐿2 reversed. So the output diode

퐷2 conducts.

Figure 3-20 Off-state 1 of BIFRED

The second phase of off-state is given in Figure 3-21. In this interval, the current of 퐿1 52

decreased to zero, only 퐿2 and 퐿3 are still conducting. When the current of 퐿2 reaches zero, the switch will turn on again.

Figure 3-21 Off-state 2 of BIFRED

3.5 Summary

From the cost point of view, RCC is a good choice as the LED driver. It is because RCC use discrete components control the peak current mode, which results in low component count, thus low cost.

However, owing to the growing concern regarding harmonic pollution of the power distribution system, BIFRED attracts more attention. Because BIFRED I suitable for power factor correction. Also from the cost point of view, it has a big advantage. Since it uses only one switch control both the boost section and the flyback section.

The analysis of the BIFRED’s operation principle is given. The boost section of

BIFRED need to operate in DCM to avoid the input current total harmonic distortion.

While the flyback section of BIFRED can operate in BCM. Apply the self-oscillating control method of RCC to BIFRED can achieve the DCM-BCM operation mode. The self-oscillating BIFRED offers great advantage of low cost.

Chapter 4 Average modeling of DCM-BCM BIFRED

4.1 Introduction

Equivalent circuit modeling is an essential tool for design, worst-case analysis, and simulation of switching converters. The use of averaging has been well accepted as a way to model the low-frequency components in a switching converter [30].

In this chapter, the average model of the self-oscillating DCM-BCM BIFRED will be built. Not only consider the ideal situation, but also includes the practical factor that may influence the operation of the BIFRED.

4.2 Average model of the ideal components BIFRED

Figure 4-1 Circuit diagram of BIFRED

4.2.1 Consider the Constant DC input

The modeling first start with the most simple situation: the input voltage is constant DC voltage. In this situation, the BIFRED converter operate in the way which explained in section 3.4.2. The major waveform of the DCM-BCM BIFRED is shown in Figure 4-2.

푣푖푛 푣퐿1

푁1 푣푖푛 − 푣표푢푡 − 푣퐶 푁2

푣퐶 푣퐿2

푁1 − 푣표푢푡 1 1 푁 푁2 푣 (푣 − 1 푣 − 푣 ) 𝑖푛 퐿 푖푛 푁 표푢푡 퐶 퐿1 1 2 𝑖푝푒푎푘1

𝑖퐿1

1 1 푁1 푣퐶 (− 푣 ) 퐿 표푢푡 𝑖푝푒푎푘 2 퐿2 푁2 2

𝑖퐿2

𝑖푆 𝑖푝푒푎푘푆 = 𝑖푝푒푎푘1 + 𝑖푝푒푎푘2

𝑖푝푒푎푘1 𝑖퐶

푡표푓푓2 −𝑖푝푒푎푘2 푡 푡 표푛 표푓푓1

Figure 4-2 Waveform of a DCM-BCM BIFRED

In Figure 4-2, the labeled parameters are explained in Table 4-1:

풗푳ퟏ Voltage across the boost inductor 퐿1

풗푳ퟐ Voltage across the flyback inductor 퐿2

풗풊풏 Input voltage

풗풐풖풕 Output voltage

풗푪 Voltage of the energy storage capacitor 퐶2

풊푳ퟏ Current through 퐿1

풊푳ퟐ Current through 퐿2

풊푺 Current through 푆1

풊푪 Current through 퐶2

풊풑풆풂풌ퟏ Peak current of the boost inductor 퐿1

풊풑풆풂풌ퟐ Peak current of the flyback inductor 퐿2

풊풑풆풂풌푺 Peak current of the switch inductor 푆1

풕풐풏 On-time

풕풐풇풇ퟏ Off-time of 𝑖퐿1

풕풐풇풇ퟐ Off-time of 𝑖퐿2

Table 4-1 Parameters in the waveform of DCM-BCM BIFRED

In Figure 4-2, the first four waveforms are the voltage and the current of 퐿1 and 퐿2. From the waveforms, it is clear to see that the BIFRED converter operates in DCM-

BCM mode, because of the discontinuous 𝑖퐿1 and the boundary condition 𝑖퐿2. And the slope of 𝑖퐿1 is the derivative of 푣퐿1, also the slope of 𝑖퐿2 is the derivative of 푣퐿2.

For this DCM-BCM BIFRED, the control method is the peak current control. And the

control parameter is the peak current of the switch 𝑖푆. So the value of 𝑖푝푒푎푘푆 is one of 56

the key parameters.

For the BIFRED converter, one unique component is the energy storage capacitor. So the current of the energy storage capacitor 𝑖퐶 is also an important concern to the analysis.

From Figure 4-2, the following equations can be derived.

푁1 푣푖푛푡표푛 = ( 푣표푢푡 + 푣퐶 − 푣푖푛) 푡표푓푓1 (4-1) 푁2

푁1 푣퐶푡표푛 = 푣표푢푡푡표푓푓2 (4-2) 푁2

1 1 𝑖 푡 = 𝑖 푡 (4-3) 2 푝푒푎푘2 표푛 2 푝푒푎푘1 표푓푓1

1 𝑖푝푒푎푘1 = 푣푖푛푡표푛 (4-4) 퐿1

1 (4-5) 𝑖푝푒푎푘2 = 푣퐶푡표푛 퐿2

𝑖푝푒푎푘_푆 = 𝑖푝푒푎푘1 + 𝑖푝푒푎푘2 (4-6)

In the situation of the constant DC input, input voltage 푣푖푛 and the capacitor voltage

푣퐶 are both constant.

Assume the following parameters are known: 푣푖푛 , 푣표푢푡 , 퐿1 , 퐿2 , 𝑖푝푒푎푘_푆 , the unknown parameters are: 푡표푛, 푡표푓푓1, 푡표푓푓2, 𝑖푝푒푎푘1, 𝑖푝푒푎푘2, 푣퐶.

With the equations from (4-1) to (4-6), the unknown parameters can be found.

With the calculated equations, the average value of the required current and voltage can be found, which is given in follow:

1 푡표푛 + 푡표푓푓1 푖̅̅퐿̅1̅ = 𝑖푝푒푎푘1 (4-7) 2 푡표푛 + 푡표푓푓2

1 푖̅̅̅̅ = 𝑖 (4-8) 퐿2 2 푝푒푎푘2

1 푡표푓푓1 푖̅̅퐶푖푛̅̅̅ = 𝑖푝푒푎푘1 (4-9) 2 푡표푛 + 푡표푓푓2

1 푡표푛 푖̅̅퐶표푢푡̅̅̅̅ = 𝑖푝푒푎푘2 (4-10) 2 푡표푛 + 푡표푓푓2

1 𝑖푝푒푎푘1푡표푓푓1 + 𝑖푝푒푎푘2푡표푓푓2 푁1 푖̅̅표푢푡̅̅̅ = ∙ (4-11) 2 푡표푛 + 푡표푓푓2 푁2

With the equations, the average value model can be built. The basic idea of the average model is to replace the non-linear components (for example switch and diode) by the controlled voltage and current generators, which represent the relations between average voltage and currents.

푖̅̅퐿̅1̅

푖̅̅̅̅̅ 푖̅̅̅̅̅̅ 푣푖푛 퐶푖푛 퐶표푢푡

a. Boost inductor section b. Boost output section

푖̅̅퐿̅2̅

푖̅̅표푢푡̅̅̅

c. Flyback inductor section d. Load section

Figure 4-3 Average model of BIFRED

The average model is given in Figure 4-3 indicates the idea of the average model. The switching model in Figure 4-1 is been transferred to the average model, which has 4 different sections: the boost inductor section, the boost output section, the flyback 58

section and the load section.

Each controlled source is explained in Table 4-2.

Name Meaning

풗풊풏 Input voltage

̅풊̅푳ퟏ̅̅ Average current of the boost inductor

풊̅̅푪풊풏̅̅̅ Average of the current in-to the capacitor

풊̅̅푪풐풖풕̅̅̅̅ Average of the current out-of the capacitor

̅풊̅푳ퟐ̅̅ Average current of the flyback inductor

풊̅̅풐풖풕̅̅̅ Average of the output current

Table 4-2 List of the controlled source in average model

4.2.2 Analysis of the operation principle with rectified mains input

If the input voltage is applied with a rectified mains, there are some changes in the equations given above.

For the rectified mains input, during every switching cycle, the input voltage can be seen as the constant value. Equations from (4-1) to (4-5) are still valid for the rectified mains situation. However, the equation (4-6) is no longer valid for every switching cycle.

If the input voltage is constant DC, the current flows through the capacitor will be zero over every HF (high frequency) cycle, which means the equation (4-6) is valid.

However, if the input voltage is rectified mains, equation (4-6) will be no longer valid for every switching cycle because the input and output current of the storage capacitor is not equal due to the constantly changing input voltage. However, the current flow through the capacitor will be zero over the half mains cycle: 59

푇 푇 2 1 2 1 (4-12) ∫ 𝑖푝푒푎푘2푡표푛 푑푡 = ∫ 𝑖푝푒푎푘1푡표푓푓1 푑푡 0 2 0 2

Assume the voltage of the energy storage capacitor is constant. So with the given equations from (4-1) to (4-5), five parameters can be found:

퐿2 ∙ 𝑖푝푒푎푘_푆 ∙ 푣푖푛 𝑖푝푒푎푘1 = (4-13) 퐿1 ∙ 푣푐 + 퐿2 ∙ 푣푖푛

퐿1 ∙ 𝑖푝푒푎푘_푆 ∙ 푣푐 𝑖푝푒푎푘2 = (4-14) 퐿1 ∙ 푣푐 + 퐿2 ∙ 푣푖푛

퐿1 ∙ 퐿2 ∙ 𝑖푝푒푎푘_푆 푡표푛 = (4-15) 퐿1 ∙ 푣푐 + 퐿2 ∙ 푣푖푛

−퐿1 ∙ 𝑖푝푒푎푘_푆 ∙ 푣푖푛 푡표푓푓1 = 퐿1 푁1 (4-16) (푣푖푛 + 푣푐) ∙ [푣푖푛 − (푣푐 + 푣표푢푡)] 퐿2 푁2

퐿1 ∙ 퐿2 ∙ 𝑖푝푒푎푘_푆 ∙ 푣푐 푡표푓푓2 = (4-17) (퐿1 ∙ 푣푐 + 퐿2 ∙ 푣푖푛) ∙ 푣표푢푡

So the average value can be calculated:

1 푡표푛 + 푡표푓푓1 푖̅̅퐿̅1̅ = 𝑖푝푒푎푘1 (4-18) 2 푡표푛 + 푡표푓푓2

1 푖̅̅̅̅ = 𝑖 (4-19) 퐿2 2 푝푒푎푘2

1 푡표푓푓1 푖̅̅퐶푖푛̅̅̅ = 𝑖푝푒푎푘1 (4-20) 2 푡표푛 + 푡표푓푓2

1 푡표푛 푖̅̅퐶표푢푡̅̅̅̅ = 𝑖푝푒푎푘2 (4-21) 2 푡표푛 + 푡표푓푓2

1 𝑖푝푒푎푘1푡표푓푓1 + 𝑖푝푒푎푘2푡표푓푓2 푁1 푖̅̅표푢푡̅̅̅ = ∙ (4-22) 2 푡표푛 + 푡표푓푓2 푁2

For the simplification of equations from (4-18) to (4-22), the on-time and off-time can

be replaced with the duty ratio. With equation (4-15) and (4-17), the switching cycle can be calculated:

푁1 퐿1 ∙ 퐿2 ∙ 𝑖푝푒푎푘 ∙ (푣푐 + 푣표푢푡) 푆 푁2 푡푠푤 = (4-23) 푁1 (퐿1 ∙ 푣푐 + 퐿2 ∙ 푣푖푛) ∙ 푣표푢푡 푁2

The duty ratio are given:

푁1 푣표푢푡 푡표푛 푁2 퐷표푛 = = (4-24) 푡푠푤 푁1 푣푐 + 푣표푢푡 푁2

푁1 푣푖푛 ∙ 푣표푢푡 푁2 퐷표푓푓1 = (4-25) 푁1 푁1 (푣푐 + 푣표푢푡) ∙ (푣푐 − 푣푖푛 + 푣표푢푡) 푁2 푁2

퐷표푓푓2 = 1 − 퐷표푛 (4-26)

With the average equations from (4-18) to (4-22) and the controlled signal equations from (4-13) to (4-17), the average model can be built. The average model in LTspice is given in the following section.

4.2.3 Average model in LTspice

The average model in LTspice [31] is composed of different sections [32]: boost inductor section, storage capacitor section, flyback inductor section and output section.

1 푖̅̅̅̅ = 𝑖 (퐷 + 퐷 ) 퐿1 2 푝푒푎푘1 표푛 표푓푓1

푣푖푛 = |푣푖푛_푝푒푎푘 ∙ sin(2휋푓푡)|

a. Boost inductor section

1 1 푡표푛 푖̅̅̅̅̅̅ = 𝑖 푖̅̅퐶푖푛̅̅̅ = 𝑖푝푒푎푘1퐷표푓푓1 퐶표푢푡 푝푒푎푘2 2 2 푡표푛 + 푡표푓푓2

b. Storage capacitor section

1 푖̅̅̅̅ = 𝑖 퐿2 2 푝푒푎푘2

c. Flyback inductor section

1 푁1 푖̅̅표푢푡̅̅̅ = (𝑖푝푒푎푘1퐷표푓푓1 + 𝑖푝푒푎푘2퐷표푓푓2) ∙ 2 푁2 d. Output section

Figure 4-4 average model

4.2.4 Model verification

Since the average model is been built, it need to be verified. The verified approach is by building an ideal switch model [33]. The model is given in Figure 4-5.

This model is composed of ideal components. The control of the switch is achieved by

the digital function in LTspice.

)

휋푓푡

(

sin

∙

put

푝푒푎푘

푖푛

푣

on control on in

푖푛

푣

off input control off

up control input control up

A1: Switch A1: control voltage B1: Input Switch B2: Switch B5, B6: B3, Start B4:

Figure 4-5 BIFRED converter model with ideal switch

Since the signal in the switch model is with high switching frequency, it is difficult to verify the average signal. One solution is using the RC low pass filter. Because the RC circuits can be used to filter a signal by blocking certain frequencies and passing others.

Figure 4-6 3rd order RC low pass filter

𝑖푝푒푎푘2

𝑖푝푒푎푘1

푣̅̅퐶̅

푖̅̅표푢푡̅̅̅

푖̅̅퐿̅2̅

푖 퐶

푖̅̅퐿̅1̅ a. Waveform of the average model

𝑖퐿2

𝑖퐿1

푣̅̅퐶̅

푖̅̅표푢푡̅̅̅

푖̅̅퐿̅2̅

푖 퐶

푖̅̅̅̅ 퐿1 b. Waveform of the verification model

Figure 4-7 Waveform comparison

From Figure 4-7, it can be seen that the waveform of the average model and the verification model are well matched. 64

4.3 Average model concerning the practical factors

4.3.1 Identify key practical factors

Figure 4-8 Non-ideal factors of BIFRED

The key for the average model is the non-linear components. Section 4.2 explained the average model with the ideal components. In this section, the non-ideal factor will be included.

First step is to identify the key non-ideal factors, so the uninfluential factor can be ignored.

Input diode 퐷1, output diode 퐷2 and the switch 푆1 all have some properties:

(1) Forward voltage of the non-ideal diode

(2) Storage time of the switch

(3) Parasitic capacitor of the switch

First consider the effect of the diode’s forward voltage.

If the forward voltage of the input diode is included in the model, how it will affect the result. The measured sensitivity curves with the influence of the input diode forward voltage are given in Figure 4-9.

a. Input current vs. diode forward voltage b. Output current vs. diode forward voltage

c. Capacitor voltage vs. diode forward voltage d. Frequency vs. diode forward voltage

Figure 4-9 Sensitivity curves of the input diode forward voltage

The forward voltage of the output diode also have the sensitivity curves, which is given in Figure 4-10.

a. Input current vs. diode forward voltage b. Output current vs. diode forward voltage

c. Capacitor voltage vs. diode forward voltage d. Frequency vs. diode forward voltage

Figure 4-10 Sensitivity curves of the output diode forward voltage 66

It can be seen from the figures above, the forward voltage of the input diode has little influence on the current, voltage and frequency, because there is not obvious trend of changing with the various diode forward voltage. This factor can be ignored. However, for the forward voltage of the output diode, the influence cannot be ignored. It shows an obvious influence of the forward voltage in Figure 4-10.

In the average model, the output voltage part should add the forward voltage of the output diode. The non-ideal model will be explained in details in the following section.

4.3.2 Storage time of power BJT

For the power BJT, one of the most important problem is the storage time. The storage time can be affected by many practical factors, for example the base drive current, the temperature, etc. It is difficult to calculate the exact equation of the storage time. So the proper solution is to measure the storage time by experiment.

The storage time measurement is to observe the waveform of the base current and the collector current. There is an example in Figure 4-11.

푣푖푛

𝑖퐵 = 0 𝑖퐶

𝑖퐵 1 𝑖퐶 = 𝑖푝푒푎푘 푣퐶퐸 2 퐶

푡푆

Figure 4-11 Storage time measure waveform

The start of the storage time is defined by the moment that the base current changes to

zero from positive value. And the end of the storage time is the middle point of the collector current drop. The cursors in Figure 4-11 is an example of measuring the storage time, the measured result in the figure is 푡푠 = 826 푛푠.

There are 4 power BJT samples are been tested. The results are given in Figure 4-12.

storage time 850 800 750 700 650 sample 1 600 sample 2

550 sample 3 storage storage [ns] time 500 sample 4 450 400 0 100 200 300 400 input voltage [V]

Figure 4-12 Storage time measured results

Four different BJT samples shows the same characteristics. It has an approximate linear relation with the input voltage.

Storage time-On time 850 800 750 700 650 600

storage storage [ns] time 550 500 450 4.5 5.5 6.5 7.5 8.5 9.5 10.5 on time [us]

Figure 4-13 storage time/ on-time characteristics

Actually the input voltage is not the direct influence of the storage time. The storage time is proportional to on-time 푡표푛. For the BIFRED control method is by the peak current control. The control current is the peak current of switch, and it is a constant value 𝑖푝푒푎푘_푆 = 1. So it means the on-time is inversely proportional to the slope of the switch current. The equation can be derived from equation (4-15) as:

퐿1 ∙ 퐿2 푡표푛 = (4-27) 퐿1 ∙ 푣푐 + 퐿2 ∙ 푣푖푛

The slope of the switch current is proportional to the input voltage. So the storage time is inversely proportional to the input voltage. This relation proves the result in Figure 4-12.

From Figure 4-12 we can see the trend line of the storage time. It has a linear relation with the input voltage. The equation of the linear relationship can be derived from Figure 4-12:

1 5 푡 = 푡 + × 10−7[s] (4-28) 푠푡 15 표푛 3

The detailed approach of the modeling will be given in the following section.

1 퐿 ∙ 퐿 5 1 2 −7 (4-29) 푡푠푡 = + × 10 [s] 15 퐿1 ∙ 푣푐 + 퐿2 ∙ 푣푖푛 3

4.3.3 Resonance effect

In section 3.3.2.3, it is been explained that the resonance is really important for the self- oscillating process of RCC driver. The resonance principle is a bit different from the one from the RCC driver. Then first there will be an explanation of the BIFRED resonance principle.

Unlike the basic topologies given in section 3.2, the BIFRED has two inductors: the boost inductor and the flyback inductor. This topology doesn’t change the order of the circuit, but the two inductor make the operation principle different from the ones of the single-inductor converter. Because the two inductor may both cause the LC resonant 69

circuit with the parasitic capacitor.

This make the input diode of very importance. The voltage at the anode of the input

diode is the input voltage 푣푖푛, and the voltage at the cathode of the diode is the voltage

across the switch 푣퐶퐸.

푣푖푛 푣퐶퐸

During the resonant period, there is a drop in 푣퐶퐸 . The amplitude of the voltage

푁1 resonance is 푣표푢푡. So the possible minimal voltage during resonant period is: 푁2

푁1 푣퐶퐸푟푒푠_푚푖푛 = 푣퐶퐸표푓푓 − 2 푣표푢푡 (4-30) 푁2

If the input voltage is higher than 푣퐶퐸푟푒푠_푚푖푛 , the input diode may conduct during the

resonant period. This makes the resonant situation more complex.

So the rectified mains can be divided into two different period, which shows in Figure 4-14. The detail analysis is in the following sections.

푣푖푛 푁1 푣퐶퐸푟푒푠_푚푖푛 = 푣퐶퐸표푓푓 − 2 푣표푢푡 푁2

푡1 푡2

Figure 4-14 Different periods of the rectified mains

4.3.3.1 Input voltage is low (풗풊풏 ≤ 풗푪푬풓풆풔_풎풊풏)

When the input voltage is low, the voltage at the collector of the switch is relatively high. It is the period 푡1 in Figure 4-14. It means during the resonant time, the input diode keeps reversed biased and no current flow. Only the flyback inductor 퐿2 and the parasitic capacitor form the LC resonant circuit. This situation is relatively simple. The complex situation is explained in the following part.

푡푟푒푠

Figure 4-15 Resonance waveform when input voltage is low

푽(풊풏) Input voltage

푽(풏ퟎퟏퟏ) Capacitor voltage

푽(풍ퟐ) Collector voltage

Table 4-3 List of the voltage

The resonant frequency is

1 푓푟푒푠 = (4-31) 2휋√퐿2퐶푝

Because the inductor value and the parasitic capacitor value don’t change with the rectified mains, the resonant frequency is a constant value during 푡1.

4.3.3.2 Input voltage is high (풗풊풏 > 풗푪푬풓풆풔_풎풊풏)

푡푟푒푠

Figure 4-16 Resonance waveform when input voltage is high 72

When the input voltage is very high, the resonant region is composed of two different phase. It is the period 푡2 in Figure 4-14.

First phase is when the input voltage is higher than the collector voltage. This situation is same as the one explained in the previous section. Only the flyback inductor 퐿2 and the parasitic capacitor form the LC resonant circuit.

Second phase is when the input voltage is higher than the collector voltage. It is because during the resonant time, there is a decrease of the collector voltage. At the same time, the input voltage is high, so there is a change that the collector voltage is lower than the input voltage. Because of this, the input diode is forward biased, so the boost inductor

퐿2 is conducting again. So the resonance is formed by the parasitic capacitor together with 퐿1 and 퐿2. Figure 4-17 shows the two phases of the resonance clearly.

푡푟푒푠1 푡푟푒푠2

Figure 4-17 Two-phase resonance

A simple explanation is given by the example of the simple LC resonant circuits, using the superposition idea. The three circuits are given in Figure 4-18. 73

a. Simplified BIFRED resonance b. Diode conducting c. Diode open circuit

Figure 4-18 Simplified LC resonant circuit

The first circuit is the simplified circuit of the BIFRED driver. In this case, the resonance is first caused by the 퐿1퐶1 oscillation. After the diode conducting, the resonance is because of the oscillation by 퐶1 with 퐿1 and 퐿2.

The second circuit is the one without the input diode. It is the situation when the diode is conducting.

The third circuit is the one without the inductor 퐿2. It is the situation when the diode is open circuit.

The simulation result is given in Figure 4-19. It can be seen that, before the diode current become positive, the voltage 푣푠푤 = 푣푠푤1. After the moment that the diode conducting, 푣푠푤 ≠ 푣푠푤1, but the waveform follows the 푣푠푤2.

푣푆푊1

푣푆푊

푣푆푊2

Figure 4-19 Voltage waveform of the resonance

With this simple simulation, the two-phase resonance is clearly explained.

Now the principle of the switch-on process is very clear. Next is to find the optimal switch-on point. In section 3.3.2.3, two optimal switch-on process is explained: zero voltage switching and valley switching. In the case of the BIFRED, the zero voltage

switching cannot be achieved. So the objective is to make the BIFRED switch on at the valley switching point. At this moment, voltage at the minimal value. And due to the BCM of the inductor current, only small value of the resonant current exists. The minimal switch-on loss can be achieved. However, it is impossible to make sure in every switching cycle reaches the valley switching for BIFRED because the resonant frequency are different of the two cases that explained in section 4.3.3.1 and 4.3.3.2. So the optimal solution is to make the one of the case reaches the valley switching.

4.3.4 Analysis of the operation principle including the practical

factors

Consider all the important practical factors, the current and voltage waveform of the BIFRED is different from the ones in Figure 4-2. The waveform which includes the forward voltage of the output diode, the storage time of the power BJT, the resonant effect during switching-on process are given in Figure 4-20.

4.3.4.1 Forward voltage of the output diode Considering the forward voltage of the output diode, the voltage of secondary winding of the flyback inductor is 푣표푢푡 + 푣퐷. So the primary winding of the flyback inductor is

푁1 (푣표푢푡 + 푣퐷). This change in the voltage of the inductor leads to the change in the slope 푁2 of the inductor current. Values are labeled in Figure 4-20. 4.3.4.2 Storage time of the power BJT Storage time is a characteristic of the power BJT. It is a time delay during switching- off process. This lead to two effects: 1. Increase of the inductor peak current; 2. Power dissipation of the power BJT during switching-off process.

The peak current of the two inductor windings are:

1 𝑖푝푒푎푘1 = 푣푖푛(푡표푛 + 푡푠푡) (4-32) 퐿1 1 𝑖푝푒푎푘2 = 푣퐶(푡표푛 + 푡푠푡) (4-33) 퐿2 75

푣푖푛 푣퐿1

푁1 푣푖푛 − (푣표푢푡 + 푣퐷) − 푣퐶 푁2 푣퐶 푣퐿2

푁1 1 1 푁1 − (푣표푢푡 + 푣퐷) 푣 푁2 𝑖푛 푣𝑖푛 − (푣표푢푡 + 푣퐷) − 푣퐶 퐿1 퐿1 푁2 𝑖푝푒푎푘1

𝑖퐿1

1 1 푁1 푣퐶 − (푣 + 푣 ) 표푢푡 퐷 𝑖푝푒푎푘 퐿2 퐿2 푁2 2

𝑖퐿2

𝑖 = 𝑖 + 𝑖 𝑖푆 푝푒푎푘푆 푝푒푎푘1 푝푒푎푘2

푣퐶퐸 푣 퐶퐸표푓푓

𝑖 𝑖푐 푐푝푒푎푘

푡표푛 푡표푓푓2 푡 푡푠푡 표푓푓1 푡푟푒푠

Figure 4-20 Waveform of BIFRED include the practical factor

Figure 4-21 Current and voltage waveform of power BJT

Figure 4-21 shows the current and voltage waveform of the power BJT including the storage time. For simplified analysis, the current and voltage change linearly during the storage time.

The power dissipation of the switch is:

푝푆 = 푣퐶퐸𝑖퐶 (4-34)

So it can be seen from Figure 4-21 that during the on-time and off-time, there is no power dissipation of BJT. However during the storage time interval, the power dissipation is significant. It can be concluded that the shorter the storage time the less power dissipation in switch.

4.3.4.3 Resonant effect during the switching-on process

During the resonant part, the voltage and current equations are given below:

푁1 1 푣퐿2_푟푒푠 = − 푣표푢푡 푐표푠 [ 푡 − (푡표푛 + 푡푠푡표푟푎푔푒 + 푡표푓푓2)] (4-35) 푁2 √퐿2퐶푝

1 푁1 1 𝑖퐿2_푟푒푠 = − 푣표푢푡√퐿2퐶푝 푠𝑖푛 [ 푡 − (푡표푛 + 푡푠푡표푟푎푔푒 + 푡표푓푓2)] (4-36) 퐿2 푁2 √퐿2퐶푝

The average value of the resonant period is the average value of the sinusoidal wave.

Since the optimal point is the valley switching point, the resonant period is half of the resonant cycle.

푣푟푒푠_푎푣 = 0 (4-37)

2 1 푁1 𝑖푟푒푠_푎푣 = − 푣표푢푡√퐿2퐶푝 (4-38) 휋 퐿2 푁2

Similar with the derivative of the equations in section 4.2.2, the equations of the practical model is given in

푁1 푣푖푛(푡표푛 + 푡푠푡) = − 푣푖푛 − (푣표푢푡 + 푣퐷) − 푣퐶 푡표푓푓1 (4-39) 푁2

푁1 푣퐶(푡표푛 + 푡푠푡) = (푣표푢푡 + 푣퐷)푡표푓푓2 (4-40) 푁2

1 𝑖푝푒푎푘1 = 푣푖푛(푡표푛 + 푡푠푡) (4-41) 퐿1

1 𝑖푝푒푎푘2 = 푣퐶(푡표푛 + 푡푠푡) (4-42) 퐿2

𝑖푝푒푎푘_푆 = 𝑖푝푒푎푘1 + 𝑖푝푒푎푘2 (4-43)

From equation (4-28) in section 4.3.2, we know that the storage time has the linear relationship with the on time. So the sum of the on-time and storage time can be rewritten as:

푡표푛 + 푡푠푡 = 퐴푡표푛 + 퐵 (4-44)

16 5 Where 퐴 = , 퐵 = × 10−7. 15 3

So the key parameters can be found:

1 퐿1퐿2𝑖푝푒푎푘_푆 푡표푛 = ( − 퐵) (4-45) 퐴 퐿1푣퐶 + 퐿2푣푖푛

퐿1퐿2𝑖푝푒푎푘_푆푣푖푛 푡표푓푓1 = 푁1 (4-46) (퐿1푣퐶 + 퐿2푣푖푛) [푣퐶 − 푣푖푛 + (푣표푢푡 + 푣퐷)] 푁2

퐿1퐿2𝑖푝푒푎푘_푆푣퐶 푡표푓푓2 = 푁1 (4-47) (푣표푢푡 + 푣퐷)(퐿1푣퐶 + 퐿2푣𝑖푛) 푁2

퐿2𝑖푝푒푎푘_푆푣푖푛 𝑖푝푒푎푘1 = (4-48) 퐿1푣퐶 + 퐿2푣푖푛

퐿1𝑖푝푒푎푘_푆푣퐶 𝑖푝푒푎푘2 = (4-49) 퐿1푣퐶 + 퐿2푣푖푛

With equation (4-45), (4-29) and (4-47), the switching cycle can be calculated:

푡푠푤 = 푡표푛 + 푡푠푡 + 푡표푓푓2 + 푡푟푒푠 (4-50)

4.3.5 Average model in LTspice

1 푡표푛 + 푡푠푡 + 푡표푓푓1 푖̅̅퐿̅1̅ = 𝑖푝푒푎푘1 (4-51) 2 푡푠푤

1 2 1 푁1 𝑖푝푒푎푘2(푡표푛 + 푡푠푡 + 푡표푓푓2) + (− 푣표푢푡√퐿2퐶푝) 푡푟푒푠 2 휋 퐿2 푁2 (4-52) 푖̅̅퐿̅2̅ = 푡푠푤

1 푡표푓푓1 푖̅̅퐶푖푛̅̅̅ = 𝑖푝푒푎푘1 (4-53) 2 푡푠푤

1 푡표푛 + 푡푠푡 푖̅̅퐶표푢푡̅̅̅̅ = 𝑖푝푒푎푘2 (4-54) 2 푡푠푤

1 2 1 푁1 (𝑖푝푒푎푘1푡표푓푓1 + 𝑖푝푒푎푘2푡표푓푓2) + (− 푣표푢푡√퐿2퐶푝) 푡푟푒푠 2 휋 퐿2 푁2 푁1 (4-55) 푖̅̅표푢푡̅̅̅ = ∙ 푡푠푤 푁2

1 푡표푛 + 푡푠푡 + 푡표푓푓1 푖̅̅퐿̅1̅ = 𝑖푝푒푎푘1 2 푡푠푤

푣푖푛 = |푣푖푛_푝푒푎푘 ∙ sin(2휋푓푡)|

a. Boost inductor section

1 푡표푓푓1 푖̅̅퐶푖푛̅̅̅ = 𝑖푝푒푎푘1 1 푡 + 푡 2 푡푠푤 표푛 푠푡 푖̅퐶표푢푡̅̅̅̅̅ = 𝑖푝푒푎푘2 2 푡푠푤

b. Storage capacitor section

1 𝑖 푡 + 푡 + 푡 + 푖̅̅̅̅̅ ∙ 푡 2 푝푒푎푘2( 표푛 푠푡 표푓푓2) 푟푒푠 푟푒푠 푖̅̅퐿̅2̅ = 푡푠푤

c. Flyback inductor section

1 (𝑖 푡 + 𝑖 푡 ) + ̅푖̅̅̅̅ ∙ 푡 2 푝푒푎푘1 표푓푓1 푝푒푎푘2 표푓푓2 푟푒푠 푟푒푠 푁1 푖̅̅표푢푡̅̅̅ = ∙ 푡푠푤 푁2 d. Output section

Figure 4-22 Average model of the non-ideal situation 80

4.3.6 Model verification

)

휋푓푡

(

sin

∙

푝푒푎푘

푖푛

푣

on control on input

푖푛

푣

on resonance delay resonance on

off delay off

off input control off

up control input control up

put voltage voltage put

B4: Start B4: A1: Switch A1: control B1: In Switch B2: Switch B5, B6: B3, Switch A6: A4, Switch A7:

Figure 4-23 Verification model of the non-ideal average model 81

The verification is similar to the one in section 4.2.4, while including the non-ideal factors.

4.3.6.1 Diode forward voltage

The forward voltage is rather simple to apply in the LTspice verification model by editing the diode model as:

. 푚표푑푒푙 퐷표푢푡 퐷(푉푓푤푑 = 1 푉푟푒푣 = 1000)

4.3.6.2 Storage time

Storage time is not a constant value, but a value that has a linear relation with the input voltage. Adding this linear time delay need some strategy.

The linear time delay can be achieved by SCHMTBUF with a RC filter. The resistor has a linear variable resistance.

Figure 4-24 Digital function of the storage time delay

4.3.6.3 Resonance effect

The resonance effect can be applied with a capacitor across the switch, and a switch-on delay. The switch-on delay is a constant value, can be applied with a BUF function or a SCHMTBUF function.

Figure 4-25 Digital function of the resonance delay 82

Figure 4-26 shows the waveform including the switch-on and switch-off delay.

푡표푛1 푡표푛2 푡표푓푓1 푡표푓푓2 Figure 4-26 Waveform including the switching delay

In Figure 4-26, there are four important time point: 푡표푛1, 푡표푛2, 푡표푓푓1, 푡표푓푓2. Each represents:

풕풐풏ퟏ Switch on moment before delay

풕풐풏ퟐ Switch on moment after delay

풕풐풇풇ퟏ Switch off moment before delay

풕풐풇풇ퟐ Switch off moment after delay

In can be seen from the waveform of 𝑖퐿1, that the switch turns on and off after the expected time delay.

𝑖푝푒푎푘2

𝑖푝푒푎푘1

̅푣̅퐶̅

푖̅̅표푢푡̅̅̅

푖̅̅퐿̅2̅

푖 퐶

푖̅̅퐿̅1̅

a. Waveform of the average model

𝑖퐿2

𝑖퐿1

푣̅̅퐶̅

푖̅̅표푢푡̅̅̅

푖̅̅퐿̅2̅

푖 퐶

푖̅̅퐿̅1̅

b. Waveform of the verification model

Figure 4-27 Waveform comparison

As can be seen from Figure 4-27 that the waveform of the average model are well matched with the verification model. So the average model is verified.

4.4 Summary

The ideal components average model and the average model including practical factors of DCM-BCM BIFRED are presented in this chapter. And the average model is verified by the switching model. These models will be used in the next chapter for the LED driver design.

Chapter 5 Emitter switching LED driver design

5.1 Introduction

The objective of this thesis project is to design a low-cost LED driver that can reach a n optimal efficiency. The low-cost topology we choose is self-oscillating BIFRED. The average model of BIFRED is built and verified. So the next step is to design the LED driver with the topology of self-oscillating BIFRED.

The design approach starts with analyzing the operation principle of the power BJT- based DCM-BCM self-oscillating BIFRED. The next step is applying the emitter switching configuration based on the power BJT-based LED driver. At last the experimental comparison between the BJT-based LED driver and Emitter switching BJT-based LED driver are given.

5.2 Power BJT-based LED driver

There are several papers on the BIFRED design, but using power BJT as the switch and applying the self-oscillating switch control is very new in this field.

The circuit diagram of the state-of-art design is given in Figure 5-1.

In the following section, an introduction of the operation principle is given.

Figure 5-1 Circuit diagram of the BIFRED driver

5.2.1 Operation principle of the self-oscillating BIFRED

Start-up resistor Self-oscillating section Snubber section BIFRED section

Figure 5-2 Structure of the self-oscillating BIFRED

From Figure 5-2, it can be seen that the self-oscillating BIFRED is mainly composed of four sections: start-up section, self-oscillating section, snubber section and the most important BIFRED section.

Here is a brief introduction of different sections:

a. Start-up section: key for the start-up process, which will be detailed explained in section 5.2.1.1.

b. Self-oscillating section: control of the switch. It makes the circuit achieve the self-switch-on transient, which will be explained in section 5.2.1.5.

c. Snubber section:

d. BIFRED section: the main body of the LED driver. Introduction and analysis are given in section 3.4 and Chapter 4.

5.2.1.1 Start up

The start-up process is similar with the start-up of RCC circuit, which has been explained in section 3.3.2.1. The key is the start-up resistance. Before the switch turned 87

on, the current flow through the start-up resistor, and the capacitor being charged. Once the voltage across the capacitor is higher than the B-E voltage, switch turns on.

Figure 5-3 Start-up process

5.2.1.2 On-state

During on-state, the power BJT is conducting. Due to the polarity of the voltage across the flyback winding, the output diode is reversed biased.

Figure 5-4 On-state

5.2.1.3 Turn off process

At the beginning of the turn-off process, similar with the process explained in section 3.3.2.2, because of the increase of the collector current and the limit of the BJT device itself, the switch can be turned off.

The switch-off process has two difference phase of removing the storage charge:

When the collector current increases, the same current flows through the sense resistor. The voltage across the resistor increase, also the base voltage increases. Once the voltage across the zener diode reaches its breakdown voltage, the zener diode starts conducting (the green path in the figure below). Then the voltage across the resistor starts increase, once the voltage is higher than the BE voltage of the feedback transistor, the transistor starts to conduct. Both of the transistors forms the positive feedback to remove the storage charge in power BJT very fast.

Collector current increase Base voltage increase Zener diode breakdown 푄 and 푄 activated 2 3

Figure 5-5 Switch-off process

5.2.1.4 Off-state

During off-state, the voltage polarity of the flyback winding make the output diode conducting.

The off-state has two different phases. The first phase is the current of 퐿1 and 퐿2 are

both flowing. The second phase is 𝑖퐿1 = 0 while 퐿2 still has current flowing. The two different phases are shown in Figure 5-6 and Figure 5-7.

Figure 5-6 Off-state 1

Figure 5-7 Off-state 2

5.2.1.5 Turn on process

Since BIFRED is a converter that integrate boost and flyback, the turn-on process is slightly different from the RCC. This special process has been discussed in Chapter 4. The switch-on process is because of the resonance, so the detailed switch-on explained in section 4.3.2. 90

Figure 5-8 Switch-on process

The switch-on process is because the LC resonance. And the resonance results in the

voltage increase of inductor 퐿4. Similar to the start-up process, the voltage of the BJT base increase then switch turn on.

5.2.2 LTspice simulation

푣푖푛

푣퐿1

푣퐿2

𝑖퐿1

𝑖퐿2

푣퐶퐸

𝑖푐표푙푙푒푐푡표푟

𝑖푐푎푝푎푐푖푡표푟

Figure 5-9 Waveform of power BJT-based BIFRED (rectified mains cycle)

푣푖푛

푣퐿1

푣퐿2

𝑖퐿1

𝑖퐿2

푣퐶퐸

𝑖푐표푙푙푒푐푡표푟

𝑖푐푎푝푎푐푖푡표푟

Figure 5-10 Waveform of power BJT-based BIFRED (switching cycle)

Figure 5-9 and Figure 5-10 show the key waveform result of the simulation.

5.3 Emitter switching BIFRED LED driver design

Applying the emitter switching technology to the BIFRED driver is the objective. Successfully apply the emitter switching with the least change in the old topology is also wanted.

In this design, the low voltage switch is a low voltage BJT.

The basic idea of design, is to add a low voltage BJT. Drive the low voltage BJT with the RCC circuit. And give sufficient energy supply to the power BJT.

The base of the low voltage BJT is connected to the old base drive circuit. Since it is a low voltage BJT, the base current need be to lower than the one for the power BJT. So the resistor need to change to higher value. From the 3rd winding of the flyback transformer, it connect to the base of the power BJT with a lower value resistor. Also because of the base current change to a lower value, in order to make it switch-off fast,

the resistor need to change to a higher value. And the external capacitor of the switch need to change value, and make it connect across the whole emitter switching device, which means connect with the collector of the high voltage BJT and the emitter of the low voltage BJT. The design is given in Figure 5-11.

Figure 5-11 Circuit diagram of emitter switching BIFRED LED driver

5.3.1 Operation principle of the emitter switching self-oscillating

BIFRED LED driver

Start-up resistor Self-oscillating section BIFRED section Emitter switching BJT

Figure 5-12 Structure of the emitter switching BIFRED

The structure of the emitter switching BJT based BIFRED is given in Figure 5-12. Similar to the self-oscillating BIFRED, it is mainly composed of three different parts: start-up section, self-oscillating section, BIFRED section. But what makes it different is the emitter switching BJT and along with the different control of the high voltage BJT and low voltage BJT. The operation principle will be explained as follows.

5.3.1.1 Start up

Figure 5-13 start-up process of ES BIFRED

There are two start-up resistor. 푅14 together with 푅14 are the start-up resistor for the high voltage (HV) BJT, 푅1 is the start-up resistor for the low voltage (LV) BJT.

5.3.1.2 On-state

Figure 5-14 On state of ES BIFRED

During on-state, both of the power BJT and the low voltage BJT are conducting. Due to the polarity of the voltage across the flyback winding, the output diode is reversed biased.

5.3.1.3 Turn off process

𝑖퐶

푣퐵−퐿푉 − +

푣푅 − Figure 5-15 Turn off process of ES BIFRED

The turn-off process can be explained by the flow chart in Figure 5-16.

charge LV BJT in 푄 𝑖 ↑ 3 푄 turn 퐶 fast 3 off removed

푄 , 푄 푣 ↑ 2 3 𝑖퐵−퐻푉 = 𝑖퐶 퐵−퐿푉 activated

Zener HV BJT 푄 diode 1 푣푅 ↑ fast turn breakd off own

Figure 5-16 Flow chart of the turn-off process

It can be briefly described as: first the peak current control make the LV BJT switch off very fast. After the LV BJT turning off, the reversed base current of the HV BJT is equal to the collector current, which has high value. The storage charge in the HV BJT can be removed very fast, then the HV BJT can be turned off without current tail.

5.3.1.4 Off-state

Figure 5-17 Off state 1 (ES BIFRED)

Figure 5-18 Off state 2 (ES BIFRED)

Similar to the off-sate that explained in section 5.2.1.4. There are two different phases of off-state.

5.3.1.5 Turn on process

Figure 5-19 Turn on process (ES BIFRED)

The turn on process is also similar to the one that explained in section 5.2.1.5. Because of the parasitic capacitor, there will be resonance after 𝑖퐿2 reaching zero. The resonance will lead to increase of the base voltage of both switches, then the switches are turned on again.

5.3.2 LTspice simulation

푣푖푛

푣퐿1

푣퐿2

𝑖퐿1

𝑖퐿2

푣퐶퐸

𝑖푐표푙푙푒푐푡표푟

𝑖푐푎푝푎푐푖푡표푟

Figure 5-20 Waveform of the ES BIFRED (rectified mains cycle)

푣푖푛

푣퐿1

푣퐿2

𝑖퐿1

𝑖퐿2

푣퐶퐸

𝑖푐표푙푙푒푐푡표푟

𝑖푐푎푝푎푐푖푡표푟

Figure 5-21 Waveform of the ES BIFRED (switching cycle)

5.4 Experimental result analysis

The ES BIFRED can reach a lower switching loss in the high voltage bipolar transistor:

 For the turn-on process, ES BIFRED reaches an improved valley switching.

 For the turn-off process, ES BIFRED has shorter current tail.

The system efficiency has improved over 0.9%, where the BJT loss is halved approximately.

The detailed experiment results (waveforms and data) and analysis are in the following sections.

5.4.1 Experimental waveform comparison

𝑖퐵

푣퐶퐸

𝑖퐶

a. BIFRED b. Emitter switching BIFRED

Figure 5-22 Waveform comparison

𝑖퐵 푣퐶퐸

𝑖퐶

a. BIFRED b. Emitter switching BIFRED

Figure 5-23 Waveform comparison over one switching cycle

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It can be seen from Figure 5-22 that both BIFRED and ES BIFRED operate at the same operating point, to make sure the validity of comparison. If observing the waveform in one switching cycle in Figure 5-23, there are two conclusions: (1) During turn-on process, ES BIFRED reaches a valley switching; (2) During turn-off process, ES has shorter current tail. These make lower switching loss in the HV BIJ.

5.4.2 Result analysis

5.4.2.1 Efficiency

BIFRED ES BIFRED Efficiency Input voltage [V] Efficiency [%] Efficiency [%] difference [%]

200 85.298 86.052 0.754

210 85.346 86.082 0.736

220 85.223 86.058 0.835

230 85.070 85.947 0.877

240 84.910 85.862 0.952

250 84.658 85.69 1.032

260 84.592 85.523 0.931

Table 5-1 Efficiency comparison

Efficiecncy comparison 86.200 86.000 85.800 85.600 85.400 85.200 BIFRED 85.000

efficiency[%] 84.800 ES BIFRED 84.600 84.400 180 200 220 240 260 280 input voltage[V

Figure 5-24 Efficiency comparison

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The rated voltage of the BIFRED driver is 220-240V. The testing voltage range is ±10% of the rated voltage, which is 200-260V.

The efficiency has improved about 0.9%. With the input power of 40W, the emitter switched BIFRED driver can reduce around 0.36W power loss, which is around 50% of the power dissipation in BJT of the traditional BIFRED drive circuit.

5.4.2.2 Temperature

BIFRED ES BIFRED Input Percentage voltage Temperature Temperature Temperature Temperature [%] [V] [℃] increase [℃] [℃] increase [℃]

200 54.1 30.8 44.1 20.8 67.53 210 58.1 34.8 43.9 20.6 59.20 220 60.4 37.1 45.3 22 59.30 230 63.1 39.8 45.9 22.6 56.78 240 66.4 43.1 47.1 23.8 55.22 250 69.7 46.4 48.9 25.6 55.17 260 72.3 49 51.2 27.9 56.94

Table 5-2 Temperature comparison

Temperature comparison 75

70 ]

℃ 65 60 ES BIFRED 55 temperature 50 BIFRED

Temperature Temperature [ temperature 45 40 180 230 280 Input voltage [V]

Figure 5-25 Temperature comparison

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푡푒푚푝푒푟푎푡푢푟푒 푖푛푐푟푒푎푠푒 표푓 퐵퐽푇 푖푛 푬푺 푩푰푭푹푬푫 The ‘Percentage’ in the table above is × 100% 푡푒푚푝푒푟푎푡푢푟푒 푖푛푐푟푒푎푠푒 표푓 퐵퐽푇 푖푛 푩푰푭푹푬푫 The ambient temperature is 23.3℃, the temperature increase is calculated by the measured temperature. The temperature increase represent the power dissipation in the HV BJT It can be concluded that the power dissipation in BJT of the ES BIFRED is about 55% of the BIFRED. Around 45% of the power dissipation in HV BJT is reduced.

5.5 Summary

The design of emitter switching BIFRED LED driver is presented in this chapter. The design focuses on the switching process: turn-on and turn-off transients. During turn- on process, the valley switching point is desired. By choosing the proper value of the capacitor of switch, the valley switching can be achieved. This results in low power dissipation during turn-on transients.

What is more important is the turn-off transient. It is because the majority of the power dissipation of the switch is during turn-off transient. If only using power BJT as the main switch, there is a need to find the optimal point of the trade-off between storage time and current tail. This can be improved by the emitter switching technology.

The emitter switching BIFRED design modifies both turn-on and turn-off process to minimize the power dissipation, and improve the efficiency. The experimental results have demonstrated that the emitter switching is a good solution to reduce power dissipation and enhance efficiency.

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Chapter 6 Conclusions and recommendations

6.1 Conclusions

The major objective of this thesis project was to design an emitter switching power BJT-based LED driver using the low-cost switch mode driving topology that can achieve high efficiency and low power dissipation.

In order to achieve this goal, the project was divided in several research questions. Some conclusion are drawn below.

 Emitter switching is a good solution to overcome the switching problems of power BJTs. It speeds up the turn-off process by utilizing the huge collector current to quickly sweep out the stored junction charge. On the other hand, the

opened emitter enables 푉퐶퐵푂 reverse voltage rating which is about double of

푉퐶퐸푂rating.

 The self-oscillating BIFRED is a low-cost topology for the LED driver due to its self-oscillating feature and boost-flyback-integrated structure.

 An average model of DCM-BCM BIFRED has been built to facilitate the important design parameters and optimize the driver operation.

 Experimental results have demonstrated that compared with normal self- oscillating BIFRED, the emitter switching driver reaches a lower turn-off switching loss in power BJT.

 The system efficiency has improved over 0.9% with the power level of 40푊, where the BJT loss is halved approximately. With this lower loss consequence, the driver can use cheaper package for further cost reduction.

6.2 Recommendations

Due to time restraints, there are still some space for the driver improvement driver. So

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further study is still necessary.

 Optimize the transformer. This thesis project focus on the optimization of the switching loss. When measure the temperature of the power BJT, the temperature of the transformer is also very high. If higher efficiency is required, the transformer optimization should be a key step.

 Transformer coupling factor. In the average model, the coupling factor is not included. For the further detailed analysis of the driver, the research on the influence of the coupling factor can be an interesting subject.

 Frequency issue. Consider the influence of the switching frequency to the switching loss of the LED driver. Find the optimal switching frequency for this LED driver in terms of optimize the efficiency.

 Sensitivity to the power BJT spread. Low cost power BJTs have wide parameters spread. What is the consequence of these spreads on component and system levels is an interesting topic.

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