Development of Power Converter for High Power Piezoelectric Motors

DEVELOPMENT OF POWER CONVERTER FOR HIGH POWER PIEZOELECTRIC MOTORS

Thomas SCHULTE, Hugues D. NJIENDE, Norbert FRÖHLEKE Institute of Power Electronics and Electrical Drives University of Paderborn, FB14/LEA 33095 Paderborn

Abstract quency is about 20kHz and the motors voltage ampli- tudes are in a range of up to 540Vp . Additionally the Several types of piezoelectric motors are known to HPM should be fed with almost unipolar voltages to deliver few watts of mechanical output power. This avoid depolarisation of the piezoelectric ceramic. paper deals with the design and development of a Therefore a DC-offset voltage must be superimposed LLCC-resonant converter for a novel type of high on the supplied AC-operating voltage. power piezoelectric motor of up to 4kW mechanical Currently, adequate power supplies are not commer- power being used in avionics. The development of a cially available. This paper deals with the develop- laboratory power supply became necessary, since suit- ment of a provisional power supply for testing the able power supplies for testing the novel piezoelectric HPM under laboratory conditions. In chapter 2 the motor during its breadboard stage are not available on general operation of LLCC resonant converters is out- the market. lined. The realisation problems including the inductor The general function of the LLCC-resonant converter design using a CAE tool are discussed in chapter 3, which also provides a DC-offset voltage for avoiding while in chapter 4 experimental results are presented. depolarisation problems is described, implementation highlights and especially the inductor design are out- 2 LLCC-Resonant converter lined. Different types of power converters have been dis- 1 Introduction cussed for feeding piezoelectric motors. Voltage fed resonant converters with half- or full-bridge topology Several types of piezoelectric motors are known, like turned out as most suitable. In [2] LLCC-resonant cir- traveling wave type ultrasonic motor or inchworm cuits are proposed for shaping the transfer function so motor, etc., see [1]. To the authors best knowledge the that almost fixed output voltages are generated and in most powerful up to date piezoelectric motors deliver [5] the LLCC-resonant converter is discussed with about 20 Watt of mechanical output power at maxi- respect to its advantages in conjunction with a control mum efficiency of about 20%. system. Within the scope of the EUREKA-project PAMELA Fig. 1 depicts the general topology (a) (full-bridge) and (Piezo Active Motor for more ELectrical Aircraft) the the frequency response of a LLCC-resonant converter development of a high power piezoelectric motor (c), which shows a gain of almost 0dB and a phase of (HPM) is aimed by company SAGEM SA. The HPM is almost 0deg around the nominal operating frequency designed for replacing hydraulic actuators for operat- ω . C is the nominal capacitance of the piezoelec- ing secondary flight control surfaces in aircrafts. Since sN pN tric actuator. Since piezoelectric motors are operated piezoelectric actuators (motors) combine such features within a relative small bandwidth (20kHz ±1kHz for as high power and torque (or force) density especially HPM), the LLCC-resonant technique provides almost at low speed and low electromagnetic interferences, impressed motor voltages in a sufficient wide fre- they are suitable for being used in avionics. quency range. In Fig. 1c the load parameters C and The novel HPM is designed to generate a mechanical p R , representing a simple passive test load, are varied output power of about 4kW . The total electrical input p in order to simulate equivalent behaviour of the HPM power is about 18kW active power1 and from 50 to with temperature (C ) or load (R ) variation. 80kVAr capacitive power. This data reflects a new p p In Chapter 1 it is mentioned, that the motor must be fed class of piezoelectric actuators. The operating fre- with an additional DC-offset voltage. One simple and reliable measure for providing a DC-offset voltage in a 1. The efficiency of piezoelectric motors is usually less than 25%. range of the AC-voltage amplitude is to use the DC- 20 a) im c) Ls 10 TA+ TB+ Cs in dB C R 0 p p inv V V vinv L vp / d p p V -10

T TB- A- gain of -20

-30 b) Z Ls/2 DC 150 im 100 . nominal values TA+ TB+ 2 Cs in deg 50 variation of Cp (+ 35%) inv variation of Rp (+ 50%) V piezo- / 0 Lp p

v V Vd vinv p electric actuator -50 Coff -100 . T TB- 2 C phase of A- s -150

10 60kHz 100 ZDC Ls/2 19kHz 20kHz 21kHz frequency in kHz Fig. 1: Topology of a conventional LLCC-resonant converter (a), topology of a LLCC-resonant converter with DC-offset voltage (b) and frequency responses of LLCC resonant tanks (c). bus voltage directly linked to the output (see Fig. 1b) mon inverter stage via two independent resonant cir- via resistors RDC and/or inductors LDC (marked by cuits, see Fig. 3. The DC bus is realized by direct 2 ZDC in Fig. 1b) for decoupling the AC part. The con- rectification of the 3 phases 400VRMS , 50Hz main . verter stage is decoupled from the DC-offset voltage Relative high capacitance value for Cd and input by series capacitors Cs distributed on both lines of the inductors Ld for ensuring a low ripple on the DC bus LLCC-resonant tanks, while additional capacitor Coff voltage Vd . Due to the input inductors Vd becomes is blocking the DC-voltage from parallel coil Lp . In load dependent and ranges from 520V to 565V . The order to minimize the influence of the blocking capaci- amplitude of vpi is calculated by tor capacitance Coff must by relatively high in com- β parison to the other components. Since only leakage vˆ ⋅⋅4 i pi = Vd π--- sin---- .(1) current must be fed to the resonant circuits by the DC 2 vˆ ⁄ π ⋅ link, the realisation of RDC (LDC resp.) is no prob- Thus, the maximum amplitude pmax = 4 Vd is lem. In the framework of this project resistive decou- about 720V and for limiting the motors voltages vpi β ° pling by RDC is used only. i must be limited to about 120 .

V 1a Ts β 3 Power converter for HPM 360 1 Vp1a Vinv1a Vd t Ts β 3.1 Topology and Total System 360 1 TS=1/fS v ϕ Fig. 3 shows the total topology of the system. The V1b p1 el HPM has two times two phases (1a, 1b and 2a, 2b); the t Vinv1b phases with index 1 belong to the main mode and Vp1b v phases with index 2 belong to the auxiliary mode. Each p2 V V 2 p2b Ts β phase of the resonant converter feeds two series con- 360 2 Vd t nected sets of piezoelectric actuators. Ts β V Vinv2 360 2 The AC-voltages of phases with index a and index b of p2a each mode show temporal phase shifts of π , see Fig. 2. Fig. 2: Principle waveforms of motor and inverter The control of the HPM relies on a variation of fre- voltages. Due to the behaviour of the LLCC- ϕ quency fs , temporal phase angle el (between 1a and resonant circuits the fundamental components 2a, 1b and 2b respectively) and duty cycles β (for 1a 1 of the inverters output voltages vinv and cor- and 1b) and β (for 2a and 2b). 2 responding motors voltages vp are equal. Phases 1a and 1b of the main mode are fed by two independent full bridge inverter stages, while phase 2a and 2b of the auxiliary mode are driven by one com- 2. In aircrafts the main supply is via 3 phases with 115VRMS , 300Hz, affected by heavy voltage and frequency variations. cabinet motor box im1a +L TA1+ TB1+ HPM Cd Cd ip1a V /2 is1a d Coff1 +L 2C v 0 0 S1a vinv1a S1 LS1/2 p1a 1a -L Lp1 Vd/2 T T 2C Cd Cd A1- B1- S1 LS1/2 -L main imt2 mode

ip1b T T A2+ B2+ Coff1 +L 0 is1b vp1b 1b B6 -L 2C Lp1 S1b vinv1b S1 LS1/2

TA2- TB2- 2CS1 L /2 L S1 d im2a 2a Coff2 +L 2C is2a S1 /2 v 0 LS1 p2a L EMI TC+ TD+ p2 -L 2CS1 L /2 i axilary S1 m2b mode v 3 S2 inv2 Coff2 -L 2C is2b S1 /2 v 0 LSn p2b L 3 phases p2 +L TC- TD- 400VRMS 2C 2b S1 LSn/2

Fig. 3: Topology of two times two phases 80kVAr LLCC-resonant converter

3.2 Power Converter for HPM ±35% ). Additionally the operating frequency is varied in a range of ±1kHz . While the frequency response of In Table 1 the parameters of both phases are listed. the system in Fig. 1c shows almost fixed gain and The realisation of the auxiliary mode circuit is no prob- phase values independent from frequency, Cp and Rp , lem but of main mode circuit. One problem is the large we observe heavy variation of the admittance curves in variation of the load parameters. The active power is Fig. 4 (calculated for the main mode with passive Cp - R test load). phase 1 phase 2 p At the nominal operating point A the admittance value (each a and b) (each a and b) is low and the phase angle is 0° . This means active P 9kW 160W 10 nominal values variation of Cp (+ 35%) µ + CpN 1.75 F75nF 0 variation of Rp ( 50%)

µ µ in V/A dB L 38.8 H 908 H inv -10

p V / s I B C µ -20 Cs 1.75 F75nF A gain of µ µ -30 Ls 31.7 H 792 H 100 µ µ Coff 24 F1F C 50 in deg

Table 1: Nominal parameters of resonant circuits. inv U / 0 s I A varied while different mechanical load is applied to the -50

phase of B system. Reactive power is varied since Cp depends on the operating temperature. The HPM is operated in a -100 10 19kHz 20kHz 21kHz 50 wide temperature range and therefore the power con- frequency in kHz verter must match a wide variation of Cp (about Fig. 4: Admittance curves of LLCC resonant tanks. 1200 • Since the inductance values of the main mode are

vp1a very low (s. Table 1) it is important to ensure that 600 vp1a inductivities of cables etc. do not disturb the opera-

in V tion of the LLCC resonant circuits. v 0 vinv1a • Insulation capability of components, cables, struc- -600 ture, etc. must be considered carefully. 200 • Semiconductors (half bridge modules) must be over- i m1a i 100 p1a sized with respect to the nominal current due to the

0 unusual operating conditions and the resulting heat in A is1a i dissipation. -100

-200 3.3 Development of magnetic components 0 25 50 75 100 t in µs Magnetic components are key components in the Fig. 5: Simulation of main mode 1a in worst LLCC-resonant converter, due to their complicated case operation (operation point C). design at high currents and frequency. In order to reduce volume and losses, the behavior of magnetic power feeding by the inverter stage only and a relative components has to be accurately predicted and some low current through the series components Cs and Ls . geometrical parameters be optimized. Therefore the But in worst case operating points B and C the admit- design and optimization tool (CAEOMAG), was used tance values are more than doubled and the phase angle which is integrated in a circuit simulator SIM- is about ±60° implying that high reactive power must PLORER. The optimization parameters are usually the be fed by the inverter stage. core geometry, the air gap length or the layer thick- nesses. Using a new approach of optimization: the dis- series current is1a,eff 55.3A tance between the air gap and the inner layer is optimized for reducing the losses caused by the air gap current through parallel coil i 84.0A p1a,eff fringing and for decreasing hot spot temperatures.

motor current im1a,eff 129.6A The structure and data flow of CAEOMAG is described in [3]. The design of magnetic components 398V AC motor voltage vp1a,AC,eff is performed within two steps. Fig. 6 shows the set up DC motor voltage v 561V of CAEOMAG including iterations loops. Starting p1a,DC from the circuit model a simulation is performed at Table 2: Load values of main mode 1a at worst steady state, generating approximate stress quantities. case operation (operation point C). The pre-optimization algorithm provides data about the expected volume, losses and temperature rise of a Fig. 5 shows simulation results of main mode 1a in fixed core and winding set-up of inductors and trans- operation point C, which turned out as worst case with formers and are inputted via a graphical user interface. respect to total losses. Corresponding load values cal- Thus initial values for a subsequent optional parameter culated from simulation results are stated in Table 2. optimization of these components are derived. The The AC part of the motor voltages v is almost sinu- p1a component geometry is optimized with respect to soidal and in phase with the fundamental component of desired design objectives which are defined by a inverter voltage v . It has 398V which results in inv1a eff weighting vector. This main optimization is repre- 563V for the AC-amplitude. The DC part is 561V p sented in the left loop. which results in almost unipolar voltage feeding. The total voltage within the system is up to 1100V . While Within high current high frequency inductors 2D-air- gap effects are significant. So far winding thickness current ip1a is almost sinusoidal (due to sinusoidal d and the airgap length l are optimized only by vp1a ) the motor current im1a and current is1a from the cu G inverter stage shows small harmonic distortion. CAEOMAG for reducing winding losses. In order to further reduce the winding losses and particularly the In summary from above investigation the following hot spot temperature the distance between air gap and points must be taken into account for realization of the adjacent layer h (underlay) is optimized, too, see provisional power converter for the HPM: un Fig. 7. All three parameters have to be optimized • The design of the highly stressed resonant inductors simultaneously. is difficult. This problem is discussed in more detail In this contribution, the discussion is constrained to the in chapter 3.3. design of the parallel inductors Lp of the main mode, since they are most critical. Initially, previous to the The temperature (hot spot) of the layer adjacent to the main (winding) optimization, an adequate core is airgap is still too high. Separating the layers from the selected. It is retained that 4 EE80 cores staked in par- airgap is one way to minimize this temperature by allel correspond closest to the calculated optimal core reducing the field strength in the conductors and there- configuration. On the one hand their area product as fore decreasing the 2D-losses, [4]. Fig. 8 illustrates the well as effective cross section and volume are next to decrease of hot spot temperature ratio FT (temperature the calculated optimized values, on the other hand their with underlay divided by temperature without under- winding window is large enough and offers a high lay) with increasing the underlay hun . potential for further winding and underlay optimiza- tion. Air gap length 4.7mm It is then proceeded to a precise parameter optimization Material litz; 102 strands; 0.1mm; 10 in parallel for layer configuration, conductor thickness and air gap length, without consideration of underlay. The results Turns per Layer 3;3;3;2 (11 turns in total) are stated in Table 3. Total Losses 181.86 W

Air gap length 6.6mm Hot spot 273.86°C

Material litz; 124 strands; 0.1mm; 7 in parallel hun ~5mm Turns per Layer 4;4;4;1 (13 turns in total) Table 4:Results of inductor optimization with underlay Total Losses 202.26 W Another optimization, using the iteration by underlay hun, yields the results in Table 4. It depicts the Hot spot 390.47°C enhancement of the inductor with adequate winding Table 3: Results of inductor optimization. configuration and thickness. The underlay clearly leads to a decrease of hot spot temperature. Note, that due to the optimization procedure the layer configuration of

minimisation of F(X) simulation inductors in Table 3 and 4 differ though inductance and until f(t)=f(t+T)

starting current are equal. F(X) parametervector x values For the technical implementation some practical

boundary conditions cost function stress aspects are to be taken into consideration: g(x) quantities • The 10 parallel litz wires, longitudinally wound on temperature rise the core center leg, should not be intermeshed, other- losses circuit model (coupling)capacitance (short circuit inductance) wise it will lead to additional power losses due more mass / volume FFT pronounced skin and proximity effects. Z(ω) calculation • Conductor layers should be accurately isolated from model core 1. geometric dimensions 2. mass each other. 3. volume model 4. flux density winding 5. core losses • For appropriate cooling an air channel is built. 6. winding losses thermal 7. magnetic permeance Impedance measurements of the inductors L are car- models 8. capacitances p 9. impedances ried out. Fig. 10 depicts the AC/DC resistance ratio 10. temperature rise FAC . Up to 25 kHz measured and calculated resistance Fig. 6: Setup and iteration loops of CAEOMAG. ratio match very well. Above 25 kHz there is a diver- gence which can partly be allocated to the differences

1.0 lG

dcu 0.8 T F

hun

0.6

hun in mm Fig. 7: Geometry of magnetic components. Fig. 8: Hot spot temperature ratio of inductor. between modelling assumptions and practical layout of a)1000 500 the physical winding with 10 parallel wound litz wires. 800 400 . vp1a 600 300 400 200 in A in

2D in V 200 100 p1a m1a i meas v 0 0 i -200 m1a -100 2Du -400 -200 AC

F b) 600 300 vp1a 400 200

200 100

in V 0 0 in A in

i p1a Freq. Hz v -200 -100 i ig. 9: Measured AC/DC resistance ratio of inductor ir1a m1a -400 -200

-600 -300 4 Measuring results 0 20 40 60 80 100 t in µs First measurements are performed by use of a Rp -Cp - test load (without HPM). Fig. 10a shows a measure- Fig. 10:Measuring results. ment with applied DC-voltage. Therefore Cp -test load can be used, only. Since there is no transformer in the 6 Acknowledgement system the output voltages are not independent and can The development of the high power LLCC-resonant only be determined by subtraction of measuring signals converter is a project in cooperation between SAGEM from two probes. The maximum permittable voltage SA (France) and the institute mentioned above in the for the probes is 1000V . Thus, the AC-amplitude must scope of the EUREKA-project PAMELA which is an not be higher than about 440V . The voltage wave- p European project with partners from industries and uni- forms shows relatively heavy distortion from sinusoi- versities. dal waveform. This distortions can not be observed when measuring the voltage by means of a special References independent scope. Hence, it results from the measur- ing method and common mode disturbances. [1] T. Sashida, T. Kenjo. An Introduction to Ultra- Fig. 10b shows a measurement without applied DC- sonic Motors, Clarendon Press, Oxford 1993. voltage. The full AC-amplitude for vp1a of 540Vp is [2] F.-J. Lin, R.-Y. Duan, H.-H. Lin. An Ultrasonic driven and load resistor Rp is introduced. Since no low Motor Drive Using LLCC Resonant Technique. Ω inductive resistor (Rp = 16.2 , P = 9kW ) was Proc. of PESC’99, Vol. 2, pp. 947-952. available an additional rectifier is introduced. Due to [3] P. Wallmeier, N. Fröhleke, D. Hahm, H. Mundi- the inherent inductance of the resistor the waveform of ger. Integrating magnetic component design and current ir1a measured at the rectifier before Rp is more optimization into circuit simulator SIMPLORER. rectangular than sinusoidal. Proc. of Power Convertion and Intelligent Motion Conf. (PCIM) 2000, Nürnberg, June 2000, pp 5 Conclusion 533-538. This paper deals with the development of a LLCC-res- [4] N. H. Kutkut. Minimizing winding losses in foil onant converter for high power piezoelectric motors. winding using field shaping techniques. Proc. of The general concept, the topology and realisation prob- PESC’97, Vol. 5, pp. 634-640. lems and the design of the magnetic components using [5] T. Schulte, N. Fröhleke, H. Grotstollen. Control a special CAE tool are outlined. Simulation and meas- for Ultrasonic Motors with LLCC-Resonant Con- urement results are presented. verter. Proc. of Conf. on New Actuators 2000, Commissioning of the power converter in conjuction Bremen, Germany, June, 2000, pp. 411-414. with the HPM will be performed soon.