High Step-Up Boost Converter Integrated with Voltage-Doubler

High Step-up Boost Converter Integrated with Voltage-Doubler

Ki-Bum Park, Gun-Woo Moon, and Myung-Joong Youn Department of Electrical Engineering, KAIST, Daejeon, Republic of Korea E-mail: [email protected]

Abstract -- The voltage-doubler provides an additional step- up gain on top of that of the boost converter, while distributing voltage stresses on devices as well. The interface between the boost converter and the voltage-doubler is accomplished by a transformer and a balancing capacitor, which also constitute a resonant tank. Since this resonant operation shapes the current sinusoidal, a switch turn-off loss and a reverse recovery on diode can be reduced. Therefore, the proposed converter is promising for high step-up applications with high efficiency Fig. 1. Boost converter with auxiliary step-up circuit. Index Terms— Boost converter and voltage-doubler.

I. INTRODUCTION For a battery powered system, electric vehicles, fuel cell system, and photovoltaic systems, where low voltage sources need to be converted to a high voltage of output, non-isolated high step-up conversion techniques find increasing necessities. A classical boost converter is generally used for its simple structure and continuous input current. However, it is hard to satisfy both high voltage conversion ratio and high efficiency at the same time with a plain boost converter. In high output voltage applications, moreover, high voltage stress on switch Fig. 2. Proposed boost converter integrated with voltage-doubler. and diode degrades the performance of devices, causing a severe hard switching loss, a conduction loss, and a reverse recovery problem [1]-[8]. To relieve abovementioned limitations in high step-up applications, various types of step-up techniques can be applied [4]-[14]. A coupled-inductor boost converter is a favorable candidate for its simple structure, however an input current ripple is large and an auxiliary circuit is required to suppress the switch voltage spike [5]-[9]. A voltage multiplier cell or a switch-capacitor circuit can be useful to raise a step- up gain in collaboration with classical topologies [10]-[12]. As the output voltage is increased, however, the number of stage is increased, requiring more capacitors and diodes. Besides, a current snubber is required to reduce the reverse recovery on diode. To raise step-up gain of a boost converter further in non- isolated applications, the alternative structure, which combines a boost converter with an auxiliary circuit in series, can be considered as shown in Fig. 1 [15],[16]. Proper selection of an auxiliary module can give many advantages such as high step-up capability, design flexibility, and distributed voltage stress. In this paper, as an auxiliary step- up circuit, the voltage-doubler is integrated with a boost converter as shown in Fig. 2 [17]. The voltage stresses on the Fig. 3. Key waveforms in BR region. diodes Do2 and Do3 in the volage-doubler are clamped to its

(a) (b) (c)

(d) (e) (f)

Fig. 4. Topological states in BR region. (a) Mode 1 [ t0 ~ t1 ]. (b) Mode 2 [ t1 ~ t2 ]. (c) Mode 3 [ t2 ~ t3 ]. (d) Mode 4 [ t3 ~ t4 ]. (e) Mode 5 [ t4 ~ t5 ]. (f)

Mode 6 [ t5 ~ t6 ].

output, Vo2 + Vo3, therefore it is inherently suitable for high in a similar way to conventional resonant converters. The voltage application. detailed operation is presented as follows. The interface between the boost converter and the voltage- TLC= 2p (1) doubler is accomplished by the transformer, which also R lkg R contribute to a step-up gain by the turn ratio n. Since a square A. BR region ( TR/2 < DTS ) voltage waveform, i.e., AC voltage, is applied across the switch Q, the transformer can be inserted in parallel with Q. The key waveform and the topological states in BR region are shown in Fig. 3 and 4, respectively. Then, CR is inserted into the primary side of the transformer to make up for a flux-balance of the transformer. Thereby, the Mode 1 [ t0 ~ t1 ] : Q is on-state and VS is applied to the voltage-doubler is coupled with the boost converter by boost inductor LB. The boost inductor current ILb flows sharing the common switch. That is, with a switching action through Q and is increased linearly. At the same time, the of Q, both the boost converter and the voltage-doubler are voltage-doubler is operated with a common switching action operated. Moreover, the leakage inductance of the of Q. The powering path from CR to the lower output Vo2 of transformer Llkg and CR constitute a resonant tank. Since the the voltage-doubler is formed through the transformer, Q, and resonant operation between Llkg and CR shapes the current Do2, as presented by the dotted line. Llkg and CR constitute a sinusoidal during Q on-state, a switch turn-off loss and a resonant tank and derive a powering current with a sinusoidal reverse recovery on diode can be reduced. As a result, the shape. The resonant capacitor voltage VCr is decreased. The proposed converter is promising for high step-up isolated switch current IQ comprises ILb and the resonant current of the applications with high efficiency. voltage-doubler as well. Since Do2 is turned-off with very slow slope of IDo2, a reverse recovery would be minimized. II. OPERATION PRINCIPLES Do3 is blocked by Vo2 + Vo3.

The proposed converter combines the boost converter and Mode 2 [ t1 ~ t2 ] : Since TR/2 is shorter than switch on- the voltage-doubler, with a common switching function of Q, interval DTS, the resonant operation is finished at t1 before Q employing a pulse-width modulation (PWM). Since the is turn-off. Therefore, only ILb flows through Q and the voltage-douber utilize the resonant operation between Llkg and resonant current does not increases a turn-off current. That is, CR, its operation can be divided into two regions according to it does not affect turn-off loss at all. Since no current flows the relationship between the resonant period TR in (1) and through CR, VCr keeps its value. duty cycle D. That is, the above-resonant (AR) region [ TR/2 Mode 3 [ t2 ~ t3 ] : Q is turned-off at t2 and ILb flow > DTS ] and the below-resonant (BR) region [ TR/2 < DTS ], through Do1. Meanwhile, the voltage-doubler is starting to be conducted in the opposite direction. The resonant operation

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o i t a r

n r u t

r e m r o f s n a r T

Fig. 7. Transformer turn ratio n according to a variation of M.

voltage-doubler. The ILb charges CR, increasing VCr linearly. Do1 is blocked.

Mode 5 [ t4 ~ t5 ] : At t4, VCr is increased enough to conduct Do1 again. In this mode, as contrary to mode 3, the powering path from Vo3 to Vo1, is formed through Do3, the transformer and Do1. Therefore, IDo1 is increased and IDo3 is Fig. 5. Key waveforms in AR region. decreased by the resonant operation between Llkg and CR. Here, a reverser recovery of Do3 can be reduced because of the slow slope of IDo3.

Mode 6 [ t5 ~ t6 ] : IDo3 reaches zero at t5 and the entire of ILb flow through Do1. Since no current flows through CR, VCr keeps its value. As D is increased, mode 5 and mode 6 fade gradually and disappear finally.

B. AR region ( T /2 > DT ) R S (a) The operation in AR region is similar to that of BR region except that mode 2 of BR region, where ILb flows solely through Q, is ignored since TR/2 is longer than DTS. The key waveform and the topological states in AR region are shown in Fig. 5 and 6, respectively. Since some topological states are the same to those of BR region, only different topological states, the intervals t0 ~ t1 and t2 ~ t3, are presented in Fig. 6. The topological states of the intervals t1 ~ t2, t3 ~ t4, and t4 ~ t5 in AR region are corresponding to Figs. 4(a), 4(c), and 4(d),

(b) respectively.

Fig. 6. Topological states in AR region. (a) t0 ~ t1. (b) t2 ~ t3. In AR region, a switch current at the turn-off instant of t2 comprises ILb and Ilkg as well, therefore a turn-off loss can be increased compared with that in BR region where only ILb path from the output of the boost converter, Vo1, to the upper flows through at the switch turn-off instant. output of the voltage-doubler, Vo3, is formed through Do1, the transformer and Do3, as presented by the dotted line. III. ANALYSIS AND CHARACTERISTICS Therefore, by the resonant operation between L and C , I lkg R Do3 A. Input-Output Voltage Gain is increased. Since the resonant current flows through Do1 in For the sake of analysis, assuming the ripple of V is the opposite direction to ILb, IDo1 is decreased accordingly. Cr ignored and using a flux-balance on the boost inductor and Do2 is blocked by Vo2 + Vo3. the transformer, several voltage equations are obtained as Mode 4 [ t3 ~ t4 ] : IDo1 reaches zero at t3, therefore the follows. entire of ILb flows through the transformer and Do3 of the

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Fig. 9. Transformer turn ratio n according to a variation of M. (a)

o 20 I

b 18 M = 12 n = 3.5

e 16 voltage stress on the voltage-doubler is n times higher z i l M = 10 a 14 compared with that on the boost converter. Since the voltage- m r

o 12 M = 8.3 doubler provides n/(1+n) of the output voltage, the n

n 10 M = 7 transformer handles n/(1+n) of the total power accordingly. e r r

u 8 Since Co1, Co2, and Co3 are connected in series, an average c

M = 5 s 6

m current of each IDo1, IDo2, and IDo3 is the same to IO. The peak r M = 3

h 4

c current of IDo1 is the same to the turn-off current of IQ. The t i 2 M = 2 w peak current of IDo3 is similar to that of ILb reflected to the S 0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 transformer secondary. Assuming DTS ≈ TR/2, the peak Duty Ratio (D) current stress on Do2 and Q can be expressed as in (7) and (8), (b) respectively. Fig. 8 shows the peak current stress and rms Fig. 8. (a) Switch peak current stress and (b) switch rms current current of Q according to the variation of D and M. In the according to a function of M. same M, as D is increased, the current stress in decreased.

pTII p I »»SOO (7) 1 (2) Do2_ peak VV= TDR 2 o1 1- D S np T I ìMDDp+2 - p - p ü V= nV (3) SO ( ) (8) o2 S IIIQ__ peak» in avg + » í ý O TDR î2 þ nD (4) VVo3 = S 2 1- D TR æ ö 1 2 np TSO I IQ__ rms»çsin (w R t) + I in avg ÷ dt 1+ n TTò0 (9) VV= (5) SRè ø OS1- D 1 »IMMMMDMD2 -2 + 1 + 16 2 - 14 - 9 2 2 (6 O { ( ) } VVCr_ avg= S ) 8D

The V is the same to the output voltage of a classical o1 C. ZCS on Diodes boost converter and the voltage-doubler provides n times The diode currents in the voltage-doubler always flow higher voltage, nVo1 ( = Vo2 + Vo3 ). Fig. 7 shows required turn ratio n according to the variation of D and the input- through Llkg, which provides a current snubbing effect, output voltage conversion ratio M. When D becomes zero, therefore a reverse recovery on Do2 and Do3 can always be reduced. Especially in BR region, where the half-period the voltage-doubler does not operated and ILb flows through resonant operation between L and C is ensured, a zero- Do1, Do2, and Do3. That is, Vo2 and Vo3 become zero and VO lkg R current-switching (ZCS) is achieved on D minimizing a follows VS like that of a conventional boost converter. o2 reverse recovery. B. Voltage and Current Stress on Device In the mode analysis of BR region, during the switch off- state, I is decreased to zero and then is increased again. In the boost converter, voltage stresses on Q and D are Do1 o1 This operation is also caused by the resonant operation Vo1, i.e., VS/(1-D). In the voltage-doubler, the voltage stresses between Llkg and CR, therefore it depends on TR. As TR and on Do2 and Do3 are Vo2+Vo3, i.e., nVS/(1-D). That is, the DTS get smaller, in both BR and AR regions, there is more

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To utilize a 100 V switch with a sufficient margin, Vo1 is selected about to 50 V and the output of the voltage-doubler, Vo2 + Vo3, is selected about to 150 V accordingly, which allows a use of 200 V rating diodes for the voltage-doubler. Therefore, n is designed to 3.5, which make the operating duty cycle change from 0.33 to 0.6 corresponding to 18 ~ 30 V input variation as shown in Fig. 9.

B. Inductor and Transformer The design of the boost inductor is the same as those of conventional ones. Considering a current ripple to be 15 % of the input current 6.7 A, LB is designed as 120 uH [18]. Fig. 10. Area-product AP of transformer according to variation of VS. Normally, the area-product AP method can be used to predict the size of the magnetic core [18]. The AP represents the product between a cross-section area and the window area of the magnetic core. In the case of the proposed converter, the AP of one transformer can be obtained as in (10), where Ku : the window utilization factor, J : the current density, and Bmax : the maximum flux density. Considering Ku to be 0.3, J 2 to be 300 A/cm , Bmax to be 0.1 T, IO to be 0.8 A, and FS to be 100 kHz, the AP of the transformer according to the function of VS is illustrated in Fig. 10, where the dot represents the Fig. 11. Resonant current waveforms according to a variation of T . R case of n = 3.5. The AP is varied according to a change in the 4 VS and the maximum AP is 0.73 cm in case of VS = 24 V. chance to re-increase of IDo1. In this case, when Q is turned-on, DVDVI(1-- ) 2 ( OSO) p 1 (10) an abrupt change in IDo1 is occurred to cause a reverse AP = + BFKJDDmax S u 8 1- recovery. Unless IDo1 is increased again, a reverse recovery on

Do1 would not be occurred. C. Resonant Tank

IV. DESIGN CONSIDERATIONS Fig. 11 shows the current waveform according to TR. In AR To illustrate the design procedure, 18 ~ 30 V input, 200 V region, the switch turn-off loss is increased. On the other hand, output, 160 W prototype converter is presented. The required in BR region, the switch turn-off loss is reduced and Do2 input-output voltage gain M is varied from 6.7 ( = 200/30 ), achieves a zero-current-switching (ZCS) turn-off that for the maximum input 30 V, to 11.1 ( = 200/18 ) for the minimizes the reverse recovery of the diode. However, the minimum input 18V. The nominal input voltage is 24 V, of current stress and conduction loss of the devices are rather which the required gain M is 8.3 ( = 200/24 ). increased. Therefore, TR can be designed around the midpoint, TR/2 = DTS, to achieve a ZCS of the diode while minimizing A. Transformer Turn Ratio n and Duty Cycle the switch turn-off loss. Therefore, once Llkg is determined from the fabricated transformer, L is set as it is and C can In the proposed converter design, the selection of a switch, lkg R be selected as in (11). which is rather burdened by the sum of the boost inductor current and the resonant current, is primarily considered in DT2 2 C = S (11) terms of cost and an efficiency. As presented in (9) and Fig. R 2 p Llkg 8(b), for the same M, an rms value of switch current is slowly decreased as a duty cycle is increased. On the other hand, a V. EXPERIMENTAL RESULTS switch voltage stress, Vo1 = VS/(1-D), is decreased with a decrease of the duty cycle, which lead to a use of lower To verify the proposed converter, the prototype is voltage switch having a smaller on-resistance. However, a implemented. The specification and design parameters smaller duty cycle results in a larger turn ratio n as show in obtained from the design example are presented in Table I. Fig. 7, which increases a voltage stress of the voltage-doubler. Fig. 12 shows the experimental waveforms at the nominal Therefore, a duty cycle should be selected to accommodate as input 24 V with full load condition. The duty cycle is about 0.5 and is similar to T /2. The resonant operation between low a voltage stress of switch as possible while not increasing R L and C shapes I sinusoidal, resulting in a reduced a burden of the voltage-doubler too much. lkg R lkg switch turn-off current. That is, although the peak switch

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(a) (b) (c)

Fig. 12. Experimental waveforms at VS = 24 V with full load condition.

(a) (b) (c)

Fig. 13. Experimental waveforms at VS = 18 V with full load condition.

(a) (b) (c)

Fig. 14. Experimental waveforms at VS = 30 V with full load condition.

current exceed 16 A, the turn-off current is only under 9 A. Fig. 13 and 14 show the experimental waveforms at VS = Moreover, both Do2 and Do3 achieve ZCS turn-off, which 18 V and VS = 30 V, respectively. In case of VS = 18 V, the alleviate the reverser recovery. The boost converter output duty cycle is increased and the circuit is operated in BR Vo1 is about 50 V, therefore the voltage stresses on Q and Do1 region, i.e., DTS > TR/2. In case of VS = 30 V, the circuit is is under 100 V including voltage spike by parasitic operated in AR region, i.e., DTS < TR/2. In both cases, the inductances, which allows the use of a Schottky diode for Do1. reverse recoveries on Do2 and Do3 are sufficiently suppressed The voltage stresses on Do2 and Do3 are clamped to Vo2+Vo3, by the current snubbing effect of Llkg. about 150 V. Fig. 14 shows the efficiency curves with respect to the variation of VS. Since the resonant tank is designed to satisfy

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the condition DTS = TR/2 at VS = 24 V, the proposed circuit shows the high efficiency over 93 % at this point along a wide load range. In the case of VS = 18 V, an increased conduction loss degrades the efficiency. On the other hand, in the case of VS = 30 V, the operation in AR region increases the switch turn-off loss, though a conduction loss is decreased. Consequently, it is noted that the efficiency is highly affected by the resonant tank design.

Table I Experimental parameters

Part Value

Input voltage VS 18 V ~ 30 V ( nominal input : 24 V ) Fig. 15. Measured efficiency. Output voltage VO 200 V . Output Power PO 160 W ( IO = 0.8 A ) Switching frequency FS 100 kHz overview,” IEEE Trans. Industrial Electronics, vol. 52, no. 3, pp. Inductance = 120 uH, high flux, Boost inductor LB 701-708, Jun. 2005. outer diameter = 27 mm, μ = 125 [4] F. L. Luo and H. Ye, “Positive output cascade boost converters,” 4 Transformer turn ratio n 3.5, core : EER28/16/11, AP = 0.91cm IEE Proc. Electr. Power Appl., vol. 151, no. 5, sep. 2004. Transformer leakage [5] T.-F. Wu, Y.-S. Lai, J.-C. Hung, and Y.-M. Chen, “Boost converter 2 uH inductance Llkg with coupled inductors and buck-boost type of active clamp.” IEEE Balance capacitance CR 1 uF Trans. Industrial Electronics, vol. 55, no. 1, pp. 154-162, Jan. 2008. IRF540A [6] Q. Zhao and F. C. Lee, “High-efficiency, high step-up dc-dc Switch Q ( R = 0.052 Ω, V = 100 V ) converters.” IEEE Trans. Power Electronics, vol. 18, no. 1, pp. 65- ds DSS 73, Jan. 2003. Do1: 16CTQ100 [7] K. C. Tseng and T. J. Liang, “Novel high-efficiency step-up ( V = 0.58 V, V = 100 V ), Diodes F RRM converter,” IEE Proc. Electr. Power Appl., vol. 151, no. 2, pp. 182- Do2, Do3: 10ETF02 190, Mar. 2004. ( VF = 1.2 V, VRRM = 200 V ) [8] R.-J Wai and R.-Y. Duan, “High step-up converter with coupled- inductor,” IEEE Trans. Power Electronics, vol. 20, no. 5, pp. 1025- 1035, Sep. 2005 [9] W. Li and X. He, “A family of interleaved DC-DC converters VI. CONCLUSION deduced from a basic cell winding-cross-coupled inductors To raise step-up gain of a boost converter further in non- (WCCIs) for high step-up or step-down converters,” IEEE Trans. Power Electronics, vol. 23, no. 4, pp. 1791-1801, Jul. 2008. isolated applications, a voltage-doubler, which is inherently [10] H. Ye and F. L. Luo, “Positive output super-lift converters,” IEEE suitable for high voltage applications, is integrated with a Trans. Power Electronics, vol. 18, no. 1, pp. 105-113, Jan. 2003. boost converter in series as an auxiliary step-up circuit. The [11] E. H. Ismail, M. A. Al-Saffar, A. J. Sabzali, and A. A. Fardoun, “A family of single-switch PWM converters with high step-up voltage-doubler provides an additional step-up ratio on top of conversion ratio,” IEEE Trans. Circuit and System I, vol. 55, no. 4, the gain of the boost converter and distributes voltage stresses pp. 1159-1171, May 2008. on devices as well. The interface between the boost converter [12] M. Prudente et al, “Voltage multiplier cells applied to non-isolated and the voltage-doubler is accomplished by the transformer, DC-DC converters,” IEEE Trans. Power Electronics, vol. 23, no. 2, pp. 871-887, Mar. 2008. which also contribute to a step-up gain by the turn ratio n. [13] W. C. P. de Aragao Filho and I. Barbi, “A comparison between two The transformer leakage inductance and the balancing current-fed push-pull DC-DC converters – analysis, design and capacitor constitute the resonant tank and its resonant experimentation,” INTELEC, 1996, pp. 313-320. [14] Y. Jang and M. M. Javanovic, “New two-inductor boost converter operation shapes the current sinusoidal, resulting in a reduced with auxiliary transformer,” IEEE Trans. Power Electronics, vol. 19, switch turn-off loss and a reverse recovery on diode. no. 1, pp. 169-175, Jan. 2004. Therefore, the proposed converter is promising for high step- [15] K.-B. Park, H.-W. Seong, H.-S. Kim, G.-W. Moon, and M.-J. Youn, "Integrated boost-sepic converter for high step-up applications," in up applications with high efficiency. It is also noted that other Proc. IEEE PESC, 2008, pp. 944-950. type of rectifier can also be integrated with a boost converter, [16] K.-B. Park, C.-E. Kim, G.-W. Moon, and M.-J. Youn, “"Non- being interfaced by a transformer and a balancing capacitor, isolated high step-up converter based on boost integrated half- bridge converter," in Proc. INTELEC, 2009, PC13-3. in the same way to the voltage-doubler. [17] K.-B. Park, C.-E. Kim, G.-W. Moon, and M.-J. Youn, “PWM resonant single-switch isolated converter,” IEEE Trans. Power REFERENCES Electronics, vol. 24, no. 8, pp. 1876-1886, Aug. 2009. [18] L. H. Dixon, “Transformer and inductor design for optimum circuit [1] R. W. Erickson and D. Maksimovic, Fundamentals of Power nd performance,” in Proc. Unitrode Power Supply Design Seminar, electronics, 2 Ed., John Wiley, New York, USA, 1950, pp. 39-55. 2002. [2] K. M. Smith and K. M. Smedly, “Properties and systhesis of passive lossless soft-switching PWM converters,” IEEE Trans. Power Electronics, vol. 14, no. 5, pp. 890-899, Sep. 1999. [3] M. M. Jovanovic and Y. Jang, “State-of-the-art, single-phase, active power-factor-correction techniques for high-power applications – an

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