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Simple and Inexpensive High-Efficiency Power Amplifier Using New APT Mosfets

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Simple and Inexpensive High-Efficiency Power Amplifier Using New APT Mosfets APT9403 By: Kenneth Dierberger Simple and Inexpensive High-Efficiency Power Amplifier Using New APT MOSFETs Presented at RF Expo East November 1994 Simple and Inexpensive High-Efficiency Power Amplifier Using New APT MOSFETs Kenneth Dierberger Applications Engineering Manager Advanced Power Technology Inc. 405 SW Columbia St. Bend, Oregon 97702 USA Frederick H. Raab, Ph.D. Lee Max Green Mountain Radio Research Co. Independent Consultant 50 Vermont Ave., Fort Ethan Allan 6284 Squiredell Dr. Colchester, Vermont 05446 San Jose, California 95129 ABSTRACT 2. BASIC DESIGN CONSIDERATIONS This note describes a simple and inexpensive high- Class-D power amplifiers employ two transistors in efficiency RF power amplifier based upon the new Ad- a push-pull configuration. The transistors are driven to vanced Power Technology (APT) RF-power MOSFETs. act as switches and generate a square-wave voltage. Its key features are a transformerless driver, low-cost The fundamental-frequency component of the square RF-power MOSFETs, a simple output transformer, and wave is passed to the load through a filter. Power out- Class-D operation. The PA and driver operate from put is controlled by varying the supply voltage. +12 and +50V and require an RF input of only 10mW. They produce an output of 250W or more at frequen- A class-D PA is ideally 100-percent efficient at all cies from 1.8 to 13.56MHz. amplitudes and in spite of load reactance. In practice, it is significantly more efficient than a similar class-B PA, especially for lower amplitudes and reactive loads. 1. INTRODUCTION Drain-Load Line and Turns Ratio This note describes a simple and inexpensive high- efficiency RF power amplifier based upon the new APT The power output of a class-D PA [1] [2] is: RF-power MOSFETs. Its key features are: 2 8 v eff • A transformerless driver, PO = 2 (1) • Low-cost RF-power MOSFETs, p 2R • Simple output transformer, and where the effective supply voltage (for MOSFETs) is: • Class-D operation. R The PA and driver operate from +12 and +50V and re- V = V (2) quire an RF input of only 10mW. They produce an out- eff DD R + R put of 250W or more at frequencies from 1.8 to ON 13.56MHz. Above, R is the drain-load line (seen by one drain with the other open) and R is the on-state drain-source The basic concept ensures an amplifier that is in- ON herently inexpensive and easy to manufacture. In the resistance. For the ARF440 and ARF441, RON»0.8W. prototype described here, however, no attempt has To obtain a 250W output with a 50V supply thus re- been made to minimize the number of components and quires R£6.4W. every attempt has been made to ensure good RF grounds and bypasses. For a simple transformer, the drain-load line is re- lated to the load Ro by: 2 3. CIRCUIT R m R (3) = ()n o where m and n are the numbers of turns in the primary The circuit of the power amplifier and driver is and secondary windings, respectively. For an RF shown in Figure 1; parts are identified in Tables 1 and transformer, m and n must be small integers. For a 2. All stages are ac-coupled. Consequently, the unex- 50W load, a bit of iteration yields m=1, n=3, and pected absence of an input signal results only in an in- R=5.56W. active circuit with minimal current flow and power con- sumption. Ignoring transformer, switching, and capacitance losses, the efficiency of the PA is: The total cost of all of the parts in the prototype breadboard is about $190, based upon quantities V ranging from 1 to 100. h= eff = 0.874 (4) V DD Pre driver The effects of switching and drain capacitance can be predicted by the equations given in [2]. These calcula- The pre driver uses a pair of low-cost ICs (U1 and tions yield a 76.2 percent efficiency for operation at U2 in Figure 1) rather than the conventional RF trans- 13.56MHz, excluding losses in the transformer and former to provide out-of-phase driving signals for the output filter. With typical losses in those components, two final MOSFETs. It also provides hard limiting (sine- an efficiency of 70 percent is expected. wave to square-wave conversion) of the input signal. The Elantec EL7144C is intended for use as a gate Gate Voltage and Current driver. The internal Schmitt trigger allows it to serve as hard limiter, and the presence of both inverting and The peak drain current [1] [2] is: non-inverting inputs allows a pair to serve as a phase splitter. 4 veff iDmax = = 10.01A (5) The RF input is ac-coupled to the non-inverting in- p R put of U1 and the inverting input of U2. Adjustment of the biases via R6 and R8 allows the transition points to which corresponds to Idc=6.37A. The data sheets for be selected to produce the desired duty ratio. the ARF440/ARF441 show that a gate-source voltage of 9 to 10V should be sufficient for minimum RON at For operation at the lower frequencies (£4MHz), the iDmax=10A. Since the threshold voltage is about 3.5V, duty ratio can be set to 50:50. In this case, the phase the RF drive voltage must be about 6.5V or slightly error between the two EL7144s is about 0.5ns. more. At higher frequencies (³10MHz), the difference in The data sheets give typical small-signal active-re- the turn-on and turn-off delay times of the ARF440 and gion gate-source and gate-drain capacitances of 700 ARF441 can cause both to be on at the same time and 55pF, respectively. The input capacitance is, how- (Appendix A). The associated increase in current con- ever, the sum of small-signal capacitances only while sumption is eliminated by adjusting the duty ratios of the MOSFET is in the cut-off region. As the MOSFET U1 and U2 so that the final MOSFETs operate with enters the active (velocity-saturation) region, the gate- 50:50 duty ratios. This requires adjusting R6 and R8 so drain capacitance is Miller-multiplied by the voltage that the low-state pulse width is about 12ns shorter gain (plus one). Finally, as the MOSFET enters the than the high-state pulse width. Some phasing error linear region (resistive region), the gate-drain (up to 25°) is unfortunately introduced by this adjust- capacitance inflates by a factor of five or so. As a ment. result, the effective gate-drain capacitance is close to 2600pF during the switching process [3]. The EL7144s have high input impedances, so R5 provides a 50W input impedance for the signal source. The total gate charge is 37nC at 10V. Transitioning Input signals in the range of 10 to 100mW are satisfac- the gate voltage in one tenth of the RF period (73.7ns tory, allowing this PA to be driven directly from a labo- at 13.56MHz) thus requires an average of 5A of gate ratory signal generator. current. A low driving resistance and low inductance are clearly required. The best switching speed is obtained with VDD1=12V. J3 J4 (VDD1 ) J5J6 (VDD2 ) J7 J8 (VDD3 ) D1 R1 C3 C C1 D2 C2 D3 R3 R2 L1 AD B E C R6 B R10 ECR12 CR18 R4 C7 C14 C15 C24 C16 A B R7 C5 C6 R11 R19 Q1 C4 1 8 2 7 Q5 3 U1 6 C12 4 5 R13 C22 J1 Q2 RF INPUT J2 R5 C13 RF OUTPUT C26 T1 L2 C28 C21 A B C9 C10 C27 C17 Q3 C8 1 8 C23 2 7 Q6 3 U2 6 R15 4 5 R9 Q4 R21 C18 R17 C11 C19 C20 C25 D E DD R8B R14 R16 R20 Figure 1. Circuit of power Amplifier and driver REFERENCE DESIGNATOR PART DESCRIPTION C1,C2 33mF, 50V electrolytic, Mallory SKR330M1HE11V C3 20mF, 250V electrolytic, Mallory TT250M20A C4-C27 0.1mF, 50WV chip, ATC 200B104NP50X C28 See Table 2. D1,D2,D3 5.1V, 0.25W Zener, 1N751A J1,J2 BNC female bulkhead Recommended: Amphenol 31-5538 Used: RF Industries RFB-1116S/UG J3-J8 European-style binding post, Johnson 111-0104 L1 3.5mH, 7 turns #24 enameled wire on Ferroxcube 768XT188 4C4 toroid L2 See Table 2. Q1,Q3 p-channel MOSFET, Siliconix 2N7016 Q2,Q4 n-channel MOSFET, Siliconix 2N7012 Q5 n-channel MOSFET, APT ARF440 Q6 n-channel MOSFET, APT ARF441 R1,R2 330W RC07 R3 220W RC07 R4 10KW RC07 R5 51W RC07 R6,R8,R10,R12,R14,R16,R18,R20 1KW trimpot, Bournes 3299X-1-102 R7,R9,R11,R13,R15,R17,R19,R21 4.7kW RC07 T1 One Ceramic Magnetics 3000-4-CMD5005 block of CMD5005 ferrite wired per Figure 3 U1,U2 Schmitt trigger/limiter, Elantec EL7144C Heat sink for DIP IC Aavid 5802 clip-on Heat sink for Q5 and Q6 Aavid 61475, 6.5-in wide by 4-in long IC socket, 8-pin DIP (4) Augat 208-AG190C PC board Approximately 6.5-in wide by 8-in long Plastic bracket for L2 Cut from plastic L Plastic bracket for C28 Cut from plastic L Feet (6) Aluminum threaded spacer, 4-40 x 2 in (Keystone 2205) Table 1. Components other than tuning. 1.8MHz L2: 22mH, Micrometals T200-6 toroid (2-in O.D.) with 52 turns of 24-AWG enameled wire C28: 354pF, 2.5kV padder, FW Capacitors AP091HV 3.5MHz L2: 11.4mH, Micrometals T200-6 toroid (2-in O.D.) with 32 turns of 24-AWG enameled wire C28: 180pF, 2.5-kV padder, FW Capacitors AP051HV 7MHz L2: 5.7mH, Micrometals T200-2 toroid (2-in O.D.) with 20 turns of 24-AWG enameled wire C28: 90pF, 2.5kV padder, FW Capacitors AP031HV 10MHz L2: 2.93mH, Micrometals T200-2 toroid (2-in O.D.) with 14 turns of 20-AWG enameled wire C28: 86pF, 2.5kV padder, FW Capacitors AP031HV 12MHz L2: 2.93mH, Micrometals T200-2 toroid (2-in O.D.) with 14 turns of 20-AWG enameled wire C28: 60pF, 2.5kV padder, FW Capacitors AP031HV 13.56MHz L2: 2.93mH, Micrometals T200-2 toroid (2-in O.D.) with 14 turns of 20-AWG enameled wire C28: 47pF, 2.5kV padder, FW Capacitors AP031HV Table 2.
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