University of Calgary PRISM: University of Calgary's Digital Repository

Graduate Studies The Vault: Electronic Theses and Dissertations

2019-05-15 A Novel, 2.4 and 5.8GHz Dual-band, 2-Stage RF-DC Charge-pump with Load-tuned Transmission Lines for DC Output Boost

Li, Sichong

Li, S. (2019). A Novel, 2.4 and 5.8GHz Dual-band, 2-Stage RF-DC Charge-pump with Load-tuned Transmission Lines for DC Output Boost (Unpublished master's thesis). University of Calgary, Calgary, AB. http://hdl.handle.net/1880/110361 master thesis

University of Calgary graduate students retain copyright ownership and moral rights for their thesis. You may use this material in any way that is permitted by the Copyright Act or through licensing that has been assigned to the document. For uses that are not allowable under copyright legislation or licensing, you are required to seek permission. Downloaded from PRISM: https://prism.ucalgary.ca UNIVERSITY OF CALGARY

A Novel, 2.4 and 5.8GHz Dual-band, 2-Stage RF-DC Charge-pump with Load-tuned

Transmission Lines for DC Output Boost

by

Sichong Li

A THESIS

SUBMITTED TO THE FACULTY OF GRADUATE STUDIES

IN PARTIAL FULFILMENT OF THE REQUIREMENTS FOR THE

DEGREE OF MASTER OF SCIENCE

GRADUATE PROGRAM IN ELECTRICAL ENGINEERING

CALGARY, ALBERTA

MAY, 2019

© Sichong Li 2019

Abstract

A dual-band, RF-DC charge pump without external matching network for wireless energy harvesting in 2.4 and 5.8 GHz bands is proposed. The first novelty here is in the use of optimal length transmission lines on the load side of the 4 half-wave rectifying stages that make up the 2- stage RFCP topology. Simulations and measurements show that doing so boosts the RFCP’s output voltage due to an induced standing wave at each diode’s input and gives the RFCP a 50Ω input impedance without needing an external matching network. The second novelty of this RFCP is the tuned secondary feed that connects the RFCP input to its 2nd stage to give dual-band performance.

By tuning this feed such that the RFCP’s 2nd stage and 1st stage impedances in parallel have 0 reactance at 5.8 GHz, return loss resonance in the 5.8 GHz band is achieved in addition to the 2.4

GHz band.

ii

Acknowledgements

iii

Table of Contents

Abstract...... ii

Acknowledgements ...... iii

Table of Contents...... iv

List of Tables...... vi

List of Figures and Illustrations...... vii

List of Abbreviations and Symbols...... xi

Chapter 1. Introduction ...... 1

1.1 Specific Aims and hypothesis ...... 1

1.2 Organization of Thesis ...... 2

Chapter 2. Background ...... 3

2.1 Battery Issues ...... 3

2.2 Radio Frequency Power Harvesting Sources ...... 5

2.3 Wireless power harvesting ...... 7

2.3.1 Wireless Power Harvesting System ...... 7

2.3.2 Antenna Design ...... 13

2.3.3 RF-DC Charge Pump Design ...... 14

Chapter 3. Antenna Design ...... 19

3.1 Vivaldi Antenna Design Overview ...... 19

3.2 Wide Band Medium Gain Antenna Design ...... 21

Chapter 4. Optimized Transmission Line Turned 2.4 GHz Rectifier ...... 27

iv

4.1 Working Principle and Analysis ...... 27

4.2 Simulation and Measurement Results ...... 35

4.3 Conclusion and Performance Comparison ...... 43

Chapter 5. 2.4 GHz and 5.8 GHz Dual Band Rectifier ...... 44

5.1 Introduction of A Novel Dual Band Working Mechanism ...... 44

5.2 Working Principle and Analysis ...... 47

5.3 Simulation and Measurement Results ...... 50

5.4 Conclusion and Performance Comparison ...... 56

Chapter 6. 2.4 GHz and 5.8 GHz Dual Band Rectenna ...... 57

6.1 Overview of the Rectenna ...... 57

6.2 Simulation and Measurement Results ...... 59

Chapter 7. Summary and Future Directions ...... 62

References ...... 64

v

List of Tables

TABLE I. Antenna Comparison………………………………………………………………….13

TABLE II. Measured and Simulated Antenna Gain……………………………………………...26

TABLE III. Transmission line (TL) lengths (βl) including pad and via lengths………………….27

TABLE IV. Performance Comparison. (* includes antenna gain)………………………………..43

TABLE V. Performance Comparison. (* includes antenna gain)………………………………...56

vi

List of Figures and Illustrations

Figure 2.3. 1. A dual band single stage rectifier designed in [21]...... 7

Figure 2.3. 2. A CMOS multi-stage charge pump designed in [26]...... 9

Figure 2.3. 3. System diagram of dual-band charge-pump. The design of the optimized charge- pump eliminates the need of a separate input matching network...... 10

Figure 2.3. 4. Proposed RF-DC charge-pump schematic and layout...... 11

Figure 2.3. 5. Fabricated dual-band RF-DC charge-pump with optimized transmission lines used to feed primary and secondary stages...... 12

Figure 2.3. 6. Simulated input impedance vs. load resistance...... 17

Figure 2.3. 7. Simulated input impedance vs. RF input power...... 17

Figure 3.1. 1. Vivaldi antenna...... 19

Figure 3.2. 1. Structure of microstrip line to slot line transition...... 21

Figure 3.2. 2. Simulated directivity vs. frequency...... 22

Figure 3.2. 3. Measured and simulated reflection coefficient of proposed Vivaldi antenna...... 23

Figure 3.2. 4. Simulated 3D radiation pattern at 2.4 GHz...... 24

Figure 3.2. 5. Simulated 3D radiation pattern at 5.8 GHz...... 24

Figure 3.2. 6. Simulated and measured normalized radiation pattern. (i) 2.4 GHz. (ii) 5.8 GHz. 25

vii

Figure 4.1. 1. Proposed RF-DC charge pump Circuit (RFCP) layout & prototype designed on 0.02 inch FR4. L1-L9 are optimized transmission line sections. C1-C3= 2pF; Cp=47µF; D1-

D4:BAT1503W; Rleak=1MΩ...... 27

Figure 4.1. 2. 4 Half-wave rectifying stages making up proposed RFCP...... 27

Figure 4.1. 3. Each half-wave rectifying stage as a rectifying 2-port network (RTPN)...... 27

Figure 4.1. 4. (i) Simulated output voltages (VDb and Vdc) and input reflection coefficient (|Γin|) of RTPN vs. TL line length (βl). (ii) Input reflection coefficient (Γin) of RTPN vs. TL electrical length (βl)...... 31

Figure 4.1. 5. Simulated voltage and current of RTPN vs. TL line length (βl)...... 32

Figure 4.1. 6. Simulated voltage (VDb) of RTPN vs. TL line length (βl) at different RF input power levels...... 33

Figure 4.1. 7. Simulated voltage cross diode-capacitor pair of RTPN vs. TL line length (βl) at different RF input power levels...... 33

Figure 4.1. 8. Simulated DC output voltage of RTPN vs. TL line length (βl) at different RF input power levels...... 34

Figure 4.2. 1. Design flow chart to optimize RF-DC Charge pump transmission line lengths. ... 35

Figure 4.2. 2. Simulated DC output voltage of RFCP vs. TL electrical length at 2.4 GHz...... 36

Figure 4.2. 3. Simulated magnitudes of DC and RF components after 1st stage...... 38

Figure 4.2. 4. Simulated magnitudes of DC and RF components after 2nd stage...... 38

Figure 4.2. 5. Simulated and measured DC output voltage and reflection coefficient vs RF input frequency...... 39

viii

Figure 4.2. 6. Simulated and Measured RFCP input impedance for RF input power levels of between -9 and 3dBm and during charge-up...... 41

Figure 4.2. 7. Simulated and measured RFCP DC output voltage and energy efficiency vs. RF input power...... 41

Figure 4.2. 8. Normalized electrical power density vs. transmission line electrical length...... 42

Figure 5.1. 1. Dual-band RF-DC charge pump (RFCP) topologies, (i) Parallel single-band half or

1-stage RFCPs and matching networks, (ii) Single half-stage RFCP with dual-band matching network, (iii) Single 2-stage RFCP with no matching network (Proposed work)...... 44

Figure 5.2. 1. (i). Proposed dual-band, 50Ω RF-DC Charge pump design (RFCP) on 0.02 inch

FR4; C1-C3= 2pF; Cp=47µF; D1-D4:BAT1503W; Rleak=1M; L3= optimized 2nd stage input feed for dual-band performance. (ii) 1st and 2nd stages of the RFCP...... 47

Figure 5.2. 2. Calculated effect of L10 to RFCP’s impedance and reflection coefficient with optimal L3 length...... 49

Figure 5.3. 1. Simulated and Measured impedances of RFCP’s 1st stage, 2nd stage and overall

RFCP as a function of L3 line length between 0 and 20 mm...... 50

Figure 5.3. 2. Simulated RFCP reflection co-eff. (Γt) and DC output (Vout) versus (i) L3 length in the 5.8 GHz band. (ii) Frequency. PRF = -6dBm...... 51

Figure 5.3. 3. RFCP measured and simulated reflection co-eff. (Γt) and DC output voltage (Vout) vs. Frequency for PRF=-6dBm...... 52

Figure 5.3. 4. Measured RFCP reflection co-eff. (Γt) vs. Frequency...... 52

ix

Figure 5.3. 5. Simulated and measured RFCP input impedance for RF input power of between -9 and 3 dBm for the following bands (i) 2.4 GHz (ii) 5.8 GHz ...... 53

Figure 5.3. 6. (i) Simulated and Measured DC output vs RF input power. (ii) Measured Energy efficiency vs RF input power. Cp=47µF, Rleak=1MΩ...... 54

Figure 5.3. 7. Simulated DC output voltage vs. RF input power for two-tone signal...... 55

Figure 6.1. 1 Dual Band Rectenna...... 57

Figure 6.2. 1. Test Setup for Rectenna Characterization. i...... 59

Figure 6.2. 2. Test Setup for Rectenna Characterization. ii...... 59

Figure 6.2. 3. DC output voltage vs. incident power density, performance at 2.4 GHz is compared with [48]...... 60

Figure 6.2. 4. Measured DC output voltage vs. PAPR...... 61

x

List of Abbreviations and Symbols

Abbreviations Definition

ZnC Zinc Carbon

NiCd Nickel Cadmium

NiMh Nickel Metal Hydride

Li-ion Lithium-ion

AD Anno Domini

Hz Hertz

RF Radio frequency

DC Direct current

RFCP Radio frequency charge pump

MCU Microcontroller unit

RFID Radio frequency identification

ISM Industrial, scientific and medical radio band

PWM Pulse width modulation

AC Alternating current

CMOS Complementary metal oxide semiconductor

PCB Printed circuit board

PMU Power management unit

WPH Wireless power harvester

WEH Wireless energy harvester

xi

ADS Advanced Design System

TEM Transverse electromagnetic mode

TE Transverse electric mode

FR4 Flame retardant 4

RTPN Rectifying 2-port network

TL Transmission line

EMN External matching network

VNA Vector network analyzer

PCE Power-conversion efficiency

ECE Energy-conversion efficiency

IC Integrated circuit

FET Field-effect transistor

RFSG RF signal generators

PA Power amplifier

PAPR Peak-to-average-power-ratio

xii

Symbol Definition

ηP Power efficiency

Ro Output resistive load

Cp Output capacitor for charge storage

η Energy efficiency of charge-pump

PRF RF input power

Vout.half Half of maximum DC voltage across output

capacitor Cp when fully charged tcap Time taken by Cp to charge from 0 volt to Vout.half

VO Voltage output of the capacitor in the RTPN

Vdc DC component of VO

VDb Output voltage of RTPN

Vin Input voltage of RTPN

Vg Source voltage

S11 Scattering parameters, S11

S12 Scattering parameters, S12

S21 Scattering parameters, S21

S22 Scattering parameters, S22

ΓL Output reflection coefficient of RTPN

Γin Input reflection coefficient of RTPN

ωc Fundamental frequency

βl Transmission line electrical length

xiii

Vout DC output voltage of RF charge pump

Rleak Output leak resister

RL Load resistance

Γt Input reflection coefficient f1 2.4 GHz band f2 5.8 GHz band

Zt-f1 Input impedance of the RFCP at f1

Zt-f2 Input impedance of the RFCP at f2

Z1-f2 Input impedance of the first stage at f2

Z2-f2 Input impedance of the second stage at f2

L1, L2, L3, L4, L5, L6, L7, L8, L9, L10 Transmission line 1, 2, 3, 4, 5, 6, 7, 8, 9, 10

D1, D2, D3, D4 Diode 1, 2, 3, 4

C1, C2, C3 Capacitor 1, 2, 3

xiv

Chapter 1. Introduction

1.1 Specific Aims and hypothesis

The aim of this research is to design a RF rectifier for ambient RF power harvesting with improved

DC output voltage that could be used to turn on and run end-applications such as a microcontroller unit (MCU) and sensors in a duty cycle mode without the need of a DC to DC boost converter.

Typical RF-DC rectifiers suffer from low power efficiency and low DC output voltages at low wireless input power levels. To overcome this issue, a novel RF-DC charge pump topology that uses optimized load-side transmission lines to improve DC output voltage at sub-milliwatt input

RF power levels is hypothesized in the 2.4 GHz band. In addition, dual-band wireless power harvesting capability at 5.8 GHz has also been bestowed on to the proposed design. The eventual goal of this work is the design of a novel RF-DC charge pump topology with superior RF-DC conversion characteristics in the 2.4 and 5.8 GHz bands without the need for an external matching network.

1

1.2 Organization of Thesis

Chapter 2 provides the background of battery issues, radio frequency harvesting sources as well as an overview of wireless power harvesting. Chapter 3 presents the design of the proposed Vivaldi antenna, followed by the simulated and measured results. Chapter 4 and 5 present the theory analysis of the dual band rectifier design tuned by optimized transmission lines. Next, design procedures are explained in detail, then simulated and measured results are given. Finally, performance comparison charts are provided. As proposed, advantage in DC output voltage with high impedance load over typical designs is proofed. Chapter 6 starts from an overview of the rectenna and ends up with the simulation and measurement results. Chapter 7 gives a summary of the designed dual band rectenna, and the future directions of this work.

2

Chapter 2. Background

2.1 Battery Issues

Of the 22,907 tons of disposable batteries used in Canada, 78% are non-rechargeable while the remaining 22% are rechargeable as of 2011 [1]. The most prevalent batteries are the non- rechargeable Alkaline batteries making up 58% of the total followed by Zinc Carbon (ZnC 18%),

Nickel Cadmium-(NiCd 14%), Nickel Metal Hydride (NiMh 4%) and Lithium-ion (Li-ion 1%) types [1]. All batteries use electrodes immersed in an electrolytic solution the ionic transfer through which constitutes current flow. Extreme temperatures adversely affect this electrode-electrolyte junction in batteries, which limits their operating temperature. Popular alkaline and ZnC batteries lose up to seven-eighths of their regular capacity when used in sub-zero temperatures whereas most of rechargeable batteries stop charging below 0 degrees Celsius [2]. Batteries normally also cannot operate above 55 to 60 degrees Celsius beyond which unsupervised Li-ions can explode

[3]. At high temperatures, alkaline batteries also rupture and leak potassium hydroxide, which causes eye and skin irritations, and is corrosive to exposed circuit traces and surfaces [4].

Commercial NiCd, NiMH battery electrodes used in portable electronics also contain nickel, which is known to cause skin allergies and rashes in 10-20% of the population [5] [6]. Lithium metal in

Li-ion batteries can explode in moist conditions causing injury [2] [3]. Batteries also are a major source of toxic mercury, lead and cadmium that can leach the soil, ground water and marine life in its vicinity, and cause neuro-toxicity in exposed humans [1] [7].These bottlenecks limit the use of batteries in many outdoor and wearable applications.

3

Unlike batteries, capacitors store energy in the form of a static electric field through a dielectric, which make them safer, environmentally cleaner and operable between -40 to 125 degrees Celsius without loss of operating or charging capacity. Capacitors can also be recharged over a million times, which is over 500 times that of Li-ions. On the flip side, capacitors only have energy densities between 0.01 and 15 watt-hour per kilogram, which is one tenth of that of Li-ion batteries at best case, and current leakage rates of between 0.001 and a couple of mill ampere. However, capacitors can sink and source large currents quickly with over 90% efficiency. By themselves, capacitors cannot supply power over prolonged periods like batteries but capacitors recharged with power harvesting devices can overcome limitations of batteries.

The use of power harvesting techniques predates batteries and power grids by over a millennia with windmills invented in 1AD being the first power harvesters for milling grains and irrigation

[8]. Recently, more compact harvesters to harvest power from motion, temperature and sunlight have been developed with outputs between tens of microwatts to a watt for a number of applications [9][10][11][12]. In kinetic and thermal energy harvesters, the power source is largely localized to a host that is either vibrating at low frequencies between 80 and 1500 Hz or at a temperature difference of over 85 Kelvin to generate power. By comparison power harvesting using solar or ambient wireless energy is more feasible, and does not require moving or extremely hot surfaces. Solar cells offer the highest power density of 10mW/cm2 which drops to below 10 microwatts per square centimeter or zero within indoors or opaque objects [13]. Wireless power has much lower power density compared to solar power, and decreases as a function of the distance from the source. But wireless power can penetrate opaque and non-metallic objects depending on the frequency of transmission [14].

4

2.2 Radio Frequency Power Harvesting Sources

Everyday radio frequency (RF) energy is broadcasting from billions of RF transmitters all over the world, most of them are personal devices, i.e. cell phone, laptop, Wi-Fi router, Bluetooth devices and handheld radios. Harvesting RF energy from ambient sources, will enable continuous charging of low-power devices and could eliminate the need for a battery[15]. Low-power battery-free devices can be designed to operate on demand or whenever output capacitor fully charged. In both cases, these battery-free devices can be implemented in harsh environment without the need of battery related maintenance while charging and in use.

With particular interest of the fast increasing of wireless mobile subscriptions, cell phones represent a large source of transmitters from which to harvest RF energy and will potentially provide on-demand power for a variety of close range sensing applications. With additional of Wi-

Fi routers and other wireless end devices such as tablets and laptops, in some urban environments, one can detect tens of Wi-Fi transmitters from a single location[16]. At close range, as is the case with indoor Wi-Fi router, it is possible to harvest energy from a typical Wi-Fi router transmitting at a power level of 100 mW, such power level could be even higher with additional laptops and cell phones are around. Such ambient RF energy can be used to charge or operate a wide range of low-power devices. At close range to a typical Wi-Fi router, this energy can be used to charge a number of devices including radio frequency identification (RFID) tags, low power sensors and lower power consumer electronics [17]. What makes ambient Wi-Fi energy harvesting even more appealing is it operates in the 2.45 GHz ISM band (the same as Bluetooth, ZigBee, RFID, cordless

5

phone etc.), meaning more RF sources rather than just Wi-Fi devices available to the same ambient

Wi-Fi energy harvesters.

6

2.3 Wireless power harvesting

2.3.1 Wireless Power Harvesting System

The last decade has seen a huge increase in the number of wireless devices that exchange information over the air. A number of researchers have focused on harvesting power from ambient wireless signals primarily from Wi-Fi, Cellular and TV broadcasts over the air [18] [19] [20]. Main issues in designing such harvesters have been RF-DC conversion efficiency and input return losses at different RF input power levels coming into mainly capacitive loads of RF-DC voltage converters. On the application side, main issues have been voltage stepping-up and cold-start issues to turn on and run end-applications such as a microcontroller unit (MCU) and sensors that typically require a stable 1.8V supply or higher from the microwatts of power harvested from ambient wireless signals.

Figure 2.3. 1. A dual band single stage rectifier designed in [21].

7

The primary mechanism to convert RF signals into DC has been a single-wave/stage or full-wave rectifier as shown in Figure 2.3.5 as a typical example, an optimized microstrip matching network is used to minimize energy reflection and improve conversion efficiency, the dc-pass filter is realized by a microstrip line low-pass filer, followed by a resistor to extract the DC power[21].

Those harvesters have focused on improving the conversion efficiency of single stage RF-DC rectifiers through the use of output DC filters, transmission lines and lumped elements to suppress the RF components and maximize the DC output across the output load. While achieving high efficiencies of 54-65%, these harvesters yield low output voltages of 0.224-1.2V at RF power levels of -6 dBm at 2.4 GHz [22] [21] [23].

In order to stepping up these low voltage levels in excess of 1.8V, these rectifiers further require passive secondary stages that act as charge-pumps or active boost-converters. Active boost converters do step up the rectifier’s voltage to usable 1.8V or higher forms and also perform an optimal impedance match with high efficiencies at high RF input power levels. However, they require additional reserve power between 0.3 and 144 microwatts to run the power management system and oscillator for active PWM duty cycle control for the DC-DC conversion, which eats into the overall efficiency. As a result, they prove to be usable only at higher power levels of well- over 100µW with the single diode rectifier included or with a lower clamped down output DC voltage charge. [18] [24] [25]

Purely passive RF-DC solutions include RF-DC charge-pump circuits, which use additional diode capacitor pairs stacked up to boost voltage during each consecutive half RF/AC in addition to rectifying it. Based on the original Cockcroft–Walton (CW) multiplier, charge pumps are the

8

primary voltage boosting mechanism from a fixed RF input power in far-field RFIDs operating in the 900 MHz bands as shown in Figure 2.3.2, in this design, multi-stage charge pump topology is used to boost DC output voltage at low RF power input power [26]. In recent years, a number of charge-pump based topologies with high conversion efficiencies of over 50% and output voltages of up to 3.2V (-15dBm input) have also been implemented in the 868-900 MHz bands on CMOS in which lower parasitics allow for more number of stages [26]–[29].

Figure 2.3. 2. A CMOS multi-stage charge pump designed in [26].

9

However, DC output voltage and efficiency of RF-DC charge-pumps tends to drop to 0.1 to 1.5V and 8 to 65% in the 2.4 GHz band [30][31][17]. The voltage and efficiency drop even further in the higher 5.8 GHz band to 0.161V and under 30% for RF inputs around -6dBm [32][33][34]. A common issue with charge-pumps also is reflection loss or power transfer efficiency resulting from impedance mismatches between a standard 50 ohm antenna and the charge-pump’s capacitive impedance. Most of the previous works have addressed this power transfer issue by using a lumped or passive transmission line based matching network [23][21][33][35][36].

Figure 2.3. 3. System diagram of dual-band charge-pump. The design of the optimized charge-pump eliminates the need of a separate input matching network.

As a proposed design, we introduce an optimized dual-band transmission line tuned RF-DC charge-pump topology with high RF-DC voltage conversion and energy transfer efficiencies in the

2.4 and 5.8 GHz Wi-Fi bands. The system diagram of the proposed charge-pump is shown in

Figure 2.3.3. The optimized layout and prototype of this charge-pump are shown in Figure 2.3.4 and Figure 2.3.5. A novel aspect of this charge-pump design is that it uses transmission lines L1

10

through L9 at different stages of the charge-pump’s topology to maximize the DC voltage at the rectified output by boosting up the voltage across the diode by introducing standing waves and keeping a near 50-ohm matched input impedance during charging. This optimized layout eliminates the need of an external input matching circuit while still achieving a high DC output voltage of 2.12V with an energy-conversion efficiency of 18% in the 2.4 GHz band, and 1.26V with an efficiency of 9% in the 5.8 GHz band with just -6 dBm of RF input power.

Figure 2.3. 4. Proposed RF-DC charge-pump schematic and layout.

11

Figure 2.3. 5. Fabricated dual-band RF-DC charge-pump with optimized transmission lines used to feed primary and secondary stages.

12

2.3.2 Antenna Design

To harvest ambient RF signal from indoor Wi-Fi devices and routers as proposed, a medium gain antenna that could cover both of interested bands was considered. In addition, to let the antenna integrate with the rectifier circuit easily and to keep the cost low, compatibility with the fabrication process of printed circuit board (PCB) is preferred. In the end, wide band Vivaldi antenna was chosen among planar bow-tie antenna, dual-band patch antenna, Vivaldi antenna, planar Log-

Periodic antenna and spiral antenna for the proposed application, comparisons in detail were made as shown in TABLE I.

TABLE I. Antenna Comparison.

Pattern Gain Bandwidth Comments

Planar Bow-Tie Broadside Low/Medium Wide Difficulty in integrating

with rectifier.

Dual-Band Patch Broadside Medium Dual-Band Narrow bandwidth.

Vivaldi End-Firing Medium/High Wide Larger in size, higher in

gain.

Planar Log-Periodic End-Firing Medium Wide Comparable with Vivaldi,

but lower in gain.

Difficulty in integrating

with rectifier.

Spiral Broadside Low/Medium Wide Difficulty in integrating

with rectifier.

13

2.3.3 RF-DC Charge Pump Design

The design goal of this RF-DC charge-pump is to convert approximately -6dBm of ambient RF input power from a Wi-Fi router into usable energy as shown in Figure 2.3.2. The converted energy is required to be in the form of a DC voltage of over 1.8V stored in a large enough output capacitor to power on and run low-power MCUs and sensor applications without the aid of batteries. Since most electronic application cannot be continuously powered using -6dBm (250 microwatts) of power, the harvested wireless power can only be used over limited active duty cycles (active mode) only after the output capacitor has been sufficiently charged first (charge mode) as has been shown in [19][27]. To enable this charge and active mode schemes requires a power management unit

(PMU) to maintain as a large load, typically 1 MΩ or higher, in between the capacitor and end- application during charge modes, and to switch into a low-impedance state typically close to 0 ohm in active modes. To distinguish this dual mode working rectifier with Wireless Power

Harvesting circuits (WPH), the proposed rectifier is named Wireless Energy Harvesting circuits

(WEH). The WEH is supposed to work with high efficiency primarily during the charge mode while charging the output capacitor from incoming RF input power. Since in charge modes, the charge-pump is not supplying the incoming RF power to a load continuously but rather storing it as energy in the output capacitor first over the charge time, it makes more sense to look at the overall energy efficiency rather than power efficiency ηP given by (1) of the WPH which has a constant consumption by an output resistive load (Ro), which is typical just 1 kΩ for better ηP. The energy efficiency of WEH can be determined using (2). To minimize error and maintain consistency during energy efficiency measurement, half of maximum DC voltage across output

14

capacitor Cp when fully charged which is quarter energy charged instead of fully charged of capacitor was taken with corresponding time interval to calculate energy efficiency.

2 푉표푢푡 ⁄푅표 휂푃 = 푃퐷퐶⁄푃푅퐹 = (1) 푃푅퐹

1 2 퐸푛푒푟푔푦 ⁄ ∙퐶푝∙ V η = 표푢푡 = 2 표푢푡.ℎ푎푙푓 × 100 (2) 퐸푛푒푟푔푦𝑖푛 푃푅퐹∙푡푐푎푝 where:

η = Energy efficiency of charge-pump.

PRF = RF input power from antenna.

Cp = Capacitance of output capacitor for charge storage.

Vout.half = Half of maximum DC voltage across output capacitor Cp when fully charged. tcap = Time taken by Cp to charge from 0 volt to Vout.half.

It is important to note that the DC voltage across output capacitor Cp encompasses losses due to the forward-junction voltage drop across the diodes used in the charge-pump; any leakages in the output capacitor and PMU circuit tied across it; and reflection losses at the charge-pump input

‘S11’. Higher junction loss in the diodes, leakages across the output capacitor, and input reflection losses will result in a lower DC voltage across output capacitor Cp and overall energy efficiency as a result. The overall energy efficiency of the WEP is dependent on its power transfer efficiency and power conversion efficiency. The power transfer efficiency depends on the charge-pump’s ability to minimize reflection or return losses at its input i.e. lower ‘S11’ over the entire charging period. The conversion efficiency depends on the charge-pump’s ability to minimize internal losses through the diodes and capacitors. Design and leakage issues of the PMU are not subject of

15

this work. The PMU is modeled as a 1 MΩ resister Rleak in charge modes based on 1 µA current leakage of commercial power supervisors [37].

Typically, during each negative half AC cycle, capacitor C1 gets charged through diode D1 via transmission line L1 while charge in C3 gets pumped into C2 through diode D3 via lines L2, L8,

L4, L7 and L3 as shown in Figure 2.3.2. During the next consecutive positive half cycle, the incoming RF voltage signal pushes this stored charge into capacitors C3 and Cp via diodes D2, lines L6, L8 and L2; and diode D4 and lines L3, L5 and L9 respectively. At low RF power levels down in the microwatts, the incoming RF power is lost in reflection losses at the input port, which lowers the incoming signal’s voltage part of which is further lost in overcoming p-n junction barrier in the diodes of the charge pump circuit. While low p-n barrier Schottky diodes do help, the losses can still be high enough to lower efficiencies and output voltages. A typical WPH which has a constant load overcome this reflection issue by adopting an input matching network optimized to a certain frequency, load resistance and input power, but the same method cannot be implemented to the WEH optimally. Diodes in the harvester have strong non-linearity and as the load effect from the final output capacitor in the WEH is changing during charging mode, the reflection contributed by diodes could only be minimized optimally at a portion time of the charging period. The overall energy efficiency improvement from the input matching network of

WEH is limited by this non-linearity. The effect of diode’s non-linearity was verified using

Keysight Technologies’ Advanced Design System (ADS). The simulation was carried out by sweeping load resistance as well as the input power level. Comparing two ideally typical WPH

RFCP design which has a lumped elements and passive transmission lines based matching network with proposed load-side tuned transmission line WEH RFCP, the matching network of the 1st

16

comparison is optimized for load resistance of 1 MΩ and RF input power of -6 dBm, the matching network of the 2nd comparison is optimized for load resistance of 10 kΩ and RF input power of -6 dBm. Load resistance sweeps are given by Figure 2.3.6 and input RF power level sweeps are given by Figure 2.3.7, input impedance of the proposed design remains much more stable at both of load and power sweeps comparing to two typical WPH RFCP designs.

Figure 2.3. 6. Simulated input impedance vs. Figure 2.3. 7. Simulated input impedance load resistance. vs. RF input power.

As proposed design, for a terminated port at the end of a long enough transmission line, the RF signal on the line tends to be a voltage standing wave with zero voltage at the node and maximum voltage at the anti-node along the transmission line. Furthermore p-n junction losses in diodes decrease with increasing forward voltage across the p-n junction. By optimizing the transmission line lengths L1, L2 and L9 feeding the rectifying diodes D1, D2, D3 and D4 in both stages of the charge-pump such that the maximum standing wave voltage occurs at the input of these diodes

17

during the positive and negative cycles, the DC output voltage and overall power conversion efficiency of the charge-pump can be maximized without external matching network.

18

Chapter 3. Antenna Design

3.1 Vivaldi Antenna Design Overview

Figure 3.1. 1. Vivaldi antenna.

Vivaldi antenna was first invented by Gibson in 1979 [38]. As it is a surface-wave traveling-wave antenna with exponentially tapered slot, it has broadband characteristics and some other excellent properties such as end-firing, low profile, light weight, high efficiency and compatibility with low cost printed circuit board (PCB). The Vivaldi antenna has many applications, such as imaging system, radar, satellite communication, see-through wall, phase array and UWB system. There are 19

three types of Vivaldi antenna, the coplanar Vivaldi antenna, the antipodal Vivaldi antenna and the balanced antipodal Vivaldi antenna. The coplanar Vivaldi antenna has the tapered slot on one side of the substrate, aperture coupling is used as the feeding mechanism on the other side [39].

The antipodal Vivaldi antenna has two layers tapered slots on the opposite sides of substrate [40].

The balanced antipodal Vivaldi antenna can improve the cross-polarized radiation [41], as the balanced antipodal Vivaldi antenna structure, another dielectric substrate is added on the top of antipodal Vivaldi antenna structure, and another layer of tapered slot as the same direction of the bottom dielectric substrate is printed on the top of the newly added substrate [42]. The main, non- resonant, travelling-wave mechanism of radiation is generated by waves travelling down a curved path along the antenna. The energy in the travelling wave is tightly bound to the conductors when the separation is very small compared to the free space wavelength and becomes progressively weaker and more radiative as the separation is increased [38]. Lower radiation frequency is determined by the aperture width of the Vivaldi antenna and higher radiation frequency is determined by the minimum width of the exponentially tapered slot, in addition, the overall frequency response is limited by the feeding structure of the Vivaldi antenna. In common with other travelling-wave antennas the gain is proportional to the overall length. Furthermore, antenna gain could be improved by adopting corrugated edge on tapered slot [41][43][44].

20

3.2 Wide Band Medium Gain Antenna Design

Figure 3.2. 1. Structure of microstrip line to slot line transition.

A medium gain coplanar Vivaldi antenna with corrugated edge is proposed for this dual band ambient energy harvesting application as shown in Figure 3.1.1. As a typical travelling wave antenna, a proper sized Vivaldi antenna inherently has the ultra-wide band capability that could cover both of interested bands. Maintaining the optimal return loss over the interested bands, wide band aperture coupling is used for microstrip line quasi-TEM mode to slot line TE mode transition as shown in Figure 3.2.1, width of the microstrip line(d2) has been chosen as 0.92 mm to realize the characteristic impedance of 50 ohm, width of slot line(d1) has been chosen as 0.3 mm by trading off between the characteristic impedance and fabrication tolerances, then the diameter(D)

21

and radius(R) of the aperture coupling are tuned in the CST Microwave Studio to achieve minimum return loss over the interested bands. Aperture width and antenna length were set to 150 mm and 210 mm respectively to realize medium gain from 2.4 GHz. Then CST Microwave Studio and ANSYS HFSS were used to precisely tune the exponentially tapered transition from the slot line to antenna aperture for stable medium gain coverage from 2.4 GHz up till 5.8 GHz, simulated directivity versus frequency is given by Figure 3.2.2, gain variation is within 1 dB on interested bands.

Figure 3.2. 2. Simulated directivity vs. frequency.

In addition, corrugated edges are adopted on the design as current chokes to maximum the radiation efficiency. Simulated 3D radiation pattern at 2.4 GHz and 5.8 GHz are given by Figure 3.2.4 and

Figure 3.2.5 respectively. The main beam at 5.8 GHz starts to steer from end-firing, which reveals the phase velocity starts to be faster than light speed and slow wave condition starts to vanish [45],

22

slightly decreasing in gain at 5.8 GHz as expected. Antenna is fabricated using a double-sided copper FR4 substrate (εr = 3.9, tan(δ) = 0.019, h = 20 mil). Simulated and measured return loss is given by Figure 3.2.3. Reflection coefficient maintains below -10 dB within 2 GHz and 6 GHz ensures antenna`s return loss is minimum for the proposed dual band application. Comparing simulated return loss using different simulators with measured results, CST Microwave Studio simulated return loss has more agreement with measurement than ANSYS HFSS. Radiation characteristic was tested in an anechoic chamber. Simulated and measured normalized radiation pattern at 2.4 GHz and 5.8 GHz are given by Figure 3.2.6 (i) and Figure 3.2.6 (ii) respectively, simulated and measured realized gain are shown in TABLE II, CST Microwave Studio simulated radiation characteristic has more agreement with measured results than ANSYS HFSS.

0

S11_Measured S11_HFSS S11_CST

-10

-20 Reflection Coefficient (dB) Coefficient Reflection

-30 1 2 3 4 5 6 7 8 Frequency (GHz)

Figure 3.2. 3. Measured and simulated reflection coefficient of proposed Vivaldi antenna.

23

Figure 3.2. 4. Simulated 3D radiation pattern at 2.4 GHz.

Figure 3.2. 5. Simulated 3D radiation pattern at 5.8 GHz.

24

0

-10

-20

-30 (dB) Normalized

-40 -180 -120 -60 0 60 120 180 Angle (degree)

Measured H Measured E HFSS H HFSS E CST H CST E

(i)

0

-10

-20

-30 (dB) Normalized

-40 -180 -120 -60 0 60 120 180 Angle (degree) Measured H Measured E HFSS H HFSS E CST H CST E

Figure 3.2. 6. Simulated and measured normalized radiation pattern. (i) 2.4 GHz. (ii) 5.8 GHz. (ii)

25

TABLE II. Measured and Simulated Antenna Gain.

Measured HFSS CST

Gain at 2.4 GHz 10.5 dB 9.6 dB 10.9 dB

Gain at 5.8 GHz 10.7 dB 7.6 dB 10.2 dB

26

Chapter 4. Optimized Transmission Line Turned 2.4 GHz Rectifier

4.1 Working Principle and Analysis

Figure 4.1. 1. Proposed RF-DC charge pump Circuit Figure 4.1. 2. 4 Half-wave (RFCP) layout & prototype designed on 0.02 inch FR4. rectifying stages making up L1-L9 are optimized transmission line sections. C1-C3= proposed RFCP. 2pF; Cp=47µF; D1-D4:BAT1503W; Rleak=1MΩ. TABLE III. Transmission line

(TL) lengths (βl) including pad

and via lengths.

βl deg Meas. mm

L1 80.4 14.8

L2 71.7 13.2

Figure 4.1. 3. Each half-wave rectifying stage as a L3 96.7 17.8 rectifying 2-port network (RTPN). L5 21.2 3.9

L9 50.5 9.3

27

The proposed WEH RFCP topology is a 2-stage charge pump with tuned transmission lines placed along the load side of each of its 4 half stages in order to boost Vout while achieving a low return loss as shown in Figure 4.1.1. The RFCP output voltage (Vout) is the cumulative sum of the voltage output of each of its 4 half cycles or half-wave rectifiers i.e. DC voltages across C1-C3 and CP as shown in Figure 4.1.2. To understand the effect of optimally placed transmission at increasing

RFCP’s overall output Vout, consider the performance of each of its half-wave rectifiers. The rectifier can be modeled as a rectifying 2-port network (RTPN) that is terminated with an optimally sized transmission line (TL) as shown in Figure 4.1.3. The voltage output of the capacitor in the

RTPN (VO) is the sum of its DC (Vdc) and RF components (at frequencies ωc, 2ωc, 3ωc) as in (3), where Vdc depends on the voltage difference between output (VDb) and input (Vin) of the RTPN as in (4). During each half cycle, maximizing Vdc thereby requires maximizing VDb and lowering Vin, which depend on the source voltage (Vg), 2 port S-parameters of just the RTPN (i.e. S21 and S22), output and input reflection coefficients (ΓL and Γin ) as in (5) and (6).

|푉퐷푏−푉𝑖푛| |푉퐷푏−푉𝑖푛| 푉푂 = 푉푑푐 + sin( 휔퐶푡) + sin( 2휔퐶푡) + ⋯ (3) 퐾1 퐾2

1 휃 푉 = ∫ 표푛|푉 − 푉 | ∙ sin(휔 푡) 푑(휔 푡) (4) 푑푐 2휋 0 퐷푏 푖푛 퐶 퐶

푉푔 푆21(Γ퐿+1) 푉퐷푏 = ∙ (5) 2 (1−S22Γ퐿)

푉푖푛 = 푉푔⁄2 ∙ (1 + 훤푖푛) (6)

28

For the RFCP and its RTPNs, an Infineon BAT15-03W Schottky diodes with low forward voltage along with a 2pF capacitor with a self-resonating frequency higher than 2.4 GHz were used. Using the manufacturer-provided diode model and Agilent’s ADS software, the 2-port S-parameters for the network shown in Figure 4.1.3 was determined to be S11= S22=0.94∠-19.63°; S21= S12 =

0.34∠70.08° for input power of -6dBm on a slightly lossy FR4 substrate. It can be seen from (5) that ideally VDb is maximum when ΓL = 1/S22, which gives 1.07∠21.42 or 1.0∠21.42 for an ideal lossless passive transmission line load (TL), however on a low-cost FR-4 substrate, VDb is maximum when ΓL = 0.97∠20.18° for a via shorted passive transmission line (TL) load. Having

ΓL = 0.97∠20.18° i.e. a complete impedance mismatch produces a maximum standing wave at port 2 that results in peak VDb being higher than the source voltage Vg. For a reasonably matched input, an RF input power of -6dBm, and ΓL =0.97∠20.18° i.e. via shorted transmission line on a slightly lossy FR4 yields a Vin of only 0.158V, but produces a VDb of 1.23V using (5). Since the

RTPN’s capacitor nearly charges to this peak voltage minus the diode’s forward voltage, Vdc is also maximized as a result.

On the input side of the RTPN, the input voltage Vin is lowered by reducing input reflections or

Γin as in (6). Typical forward input impedances of diode-capacitor pairs tend to be capacitive with a low resistance due to the diode junction loss. In typical WPHs, an external matching network

(EMN) along with the constant resistive output load to a forward diode-capacitor pair helps in producing a near 50Ω input impedance thereby lowering Γin with respect to a 50Ω source antenna.

By comparison, WEHs use large energy-storing output capacitors along with large PMU impedances in charge modes with input impedance that vary with time, which make WEHs harder to match to a 50Ω source. For the proposed WEH RFCP, the input impedance of each of its RTPN

29

in Figure 4.1.3 equals the diode-capacitor pair’s impedance in series with the lossy via-shorted TL on FR4. With S21= S12 = 0.34∠70.08° at -6dBm, the RTPN can be thought of as a lossy matching network between the 50Ω source (Zg) and a lossy FR4 via shorted transmission line load (ΓL) with an input return loss Γin given by (7). The RTPN’s Γin is dependent on its S-parameters and the output return loss of the TL load (ΓL) as in (7). An analysis of (7) shows that for an ideal ΓL = S22*,

Γin and Vin are minimized, it can be seen that tuning the length of the TL (∠ΓL or βl) to be as close to S22*= 0.94∠19.63° moves its input impedance closer to 50Ω for better input match (low Γin).

As the ΓL = 0.97∠20.18° which is chosen to give VDb to its maximum is close to S22*=

0.94∠19.63° which would contribute a minimum return loss to RTPN, the by-product of improved return loss (a near 50 ohm input impedance) from tuning VDb to its maximum using via-shorted

TL with optimal length could be expected and explained.

푆12푆21Γ퐿 Γ푖푛 = 푆11 + (7) 1−푆22Γ퐿

The effect of the via-shorted TL length (∠ΓL or βl) on VDb, Vdc and Γin was studied by simulating the RTPN in figure 4.1.3 on a double-sided copper FR4 substrate (εr = 4.7, tan(δ) = 0.022, h = 20 mil) using Agilent ADS software. Simulation shows a minimum return loss (Γin=0.1-j0.26 or |Γin|

° ° = -11.52dB) for ΓL=0.97∠20.18 , which translates to a TL line length (βl) of 80 at 2.4 GHz as shown in Figure 4.1.4 (i) and (ii). Simulations also show that for a matched RTPN, an input power of -6dBm yields an input voltage of only 0.158V but a maximum output capacitor voltage of Vdc

° =1.18V (VDb = 1.23V as theory expected) for a close ΓL=0.97∠20.18 (TL length βl = 80°) as shown in Figure 4.1.4 (i). It should be noted that such an optimal TL load line works only with a

WEH in which the input power is converted to energy stored in the capacitor and loss through the

30

diode and optimized TL, which gives the RTPN a high capacitor DC voltage (Vdc) and a near 50Ω input impedance in steady state (fully charged up).

Figure 4.1. 4. (i) Simulated output voltages (VDb and Vdc) and input reflection coefficient (|Γin|) of RTPN vs. TL line length (βl). (ii) Input reflection coefficient (Γin) of RTPN vs. TL electrical length (βl).

As the transferred power is increasing as the increment of VDb, the current is expected to be increasing as well. Current simulation is given as below in Figure 4.1.5, the magnitude of current verses transmission line length has the same changing and is maximized at the same transmission line length as VDb as expected. A transient simulation in ADS was also carried out to study the phase difference (time difference) between the standing wave voltage (VDb) and current peaks (I).

However transient simulations in ADS were found to not include voltage standing wave unlike harmonic balance simulations and so the phase or time difference between the 2 could not be seen.

31

Figure 4.1. 5. Simulated voltage and current of RTPN vs. TL line length (βl).

Nonlinearity of the RTPN was studied using harmonic balance simulator in ADS, the simulated results are given in Figure 4.1.6, 4.1.7 and 4.1.8. Although the optimal electrical length for maximized VDb do shift few degrees when the incident voltage of diode is higher than diode’s breakage voltage, the optimal electrical length for maximized voltage cross the diode-capacitor pair as well as the optimal electrical length for maximized DC output voltage remains stable from

-24 dBm to 9 dBm and even not at the linear region of the diode. Very little shift in the optimum

TL line length is observed for between -24 and -3 dBm between which most of the ambient wireless signal power is expected to be present.

32

Figure 4.1. 6. Simulated voltage (VDb) of RTPN vs. TL line length (βl) at different RF input power levels.

Figure 4.1. 7. Simulated voltage cross diode-capacitor pair of RTPN vs. TL line length (βl) at different RF input power levels.

33

Figure 4.1. 8. Simulated DC output voltage of RTPN vs. TL line length (βl) at different RF input power levels.

34

4.2 Simulation and Measurement Results

Figure 4.2. 1. Design flow chart to optimize RF-DC Charge pump transmission line lengths.

The proposed WEH design is implemented with 4 RTPNs containing diode-capacitor-TL pairs

D1-C1-L1, D2-C3-L2, D3-C2-L2 and D4-Cp-L9 form the 4 RTPNs in the RFCP within a RF-DC charge pump topology as shown in Figure 4.1.1 and Figure 4.1.2. The incoming RF signal is rectified by RTPNs 1 and 3 only during negative RF cycles, and by RTPNs 2 and 4 only during 35

positive RF cycles. As a result, the 50Ω RTPNs 1 and 2 in stage 1, and RTPNs 3 and 4 in stage 2 of the RFCP act as a near 50Ω impedance together during a full RF cycle. It can also be seen from

Figure 4.1.1 and Figure 4.1.2 that the total RFCP input impedance is the parallel combination of stage 1 and stage 2 through a common L2 and C3, but which get used during alternate half RF cycles. As a result, the RFCP with its 2 stages is expected to have an overall impedance of near

25Ω with little reactive part, and a return loss of 10dB. The RFCP with 4 RTPNs was designed using Agilent ADS with TL lines L1, L2 and L5+L9 designed with a characteristic impedance of

50Ω and an initial length of 80° (14.7mm). To separate Cp and its output probe, L5 and L9 are set to 3.9mm and 10.8mm. The TL lengths were then iteratively tuned as show in Figure 4.2.1 in ADS software to compensate for parasitic parameters and get the optimal DC output voltage.

3.5

3 L1 L2 L9 L3

2.5

2 Vout (V) Vout 1.5

1

0.5 40 50 60 70 80 90 100 Transmission Line Electrical Length at 2.4 GHz (degrees)

Figure 4.2. 2. Simulated DC output voltage of RFCP vs. TL electrical length at 2.4 GHz.

From the iterative optimization process, a number of traits were observed for the optimized 2-stage

RF-DC charge pump design. For all transmission lines used in the design (L1-L9), at their optimum lengths, the DC output voltage converges to a common maximum value of close to 2.3V for an RF

36

input power of -6dBm at 2.4 GHz as shown in individual sweeps of transmission line lengths L1,

L2, L9 and L3 in Figure 4.2.2. The optimization process also shows that the charge pump’s DC output voltage is more sensitive to transmission line lengths L1 and L2 used in placements of diodes D1, D2 and D3 in both stages of the charge pump. As seen from sweeps in Figure 4.2.2, the optimum lengths for L1 (13.7mm) and L2 (12.3mm) with the diode/capacitor pads and ground via lengths added are found to be 14.5 mm (78.8°) and 13.2 mm (71.7°) for maximum DC output with -6dBm of RF input in the 2.4 GHz band, L1 is identical as theory expected, L2 was observed to be lower than a single RTPN’s due to parallel connections within the RFCP. However, measured optimum length for L1 was found to be 14.8 mm and Vout looks more sensitive to L1 than simulation which were due to tolerance from assembling and inaccuracy of simulation tools. Vout was also observed to be more sensitive to the L1 and L2 lengths than L9 since harmonic balance simulation in ADS show little RF component after the initial 2 RTPNs. This rectification can be seen from the harmonic balance simulations of the output voltage of the charge pump’s 1st and 2nd stages in Figure 4.2.3 and Figure 4.2.4 that show a sharp increase in the DC (1.3 to 2.3V) and decrease in the 2.4 GHz RF component (0.75 to 0.2V). L3 was found to not have a significant effect on Vout in the 2.4 GHz band as shown in Figure 4.2.2. However, L3 can be tuned as a parallel stub at the RFCP’s input to give it dual-band characteristics. Work on designing RFCPs with dual- band characteristics is covered in Chapter 5. With the standing wave occurring along L1, L2, L5 and L9, these optimal L1, L2 and L9 lengths ensure that the maximum voltage occur just at the input of diodes D1, D2, D3 and D4 shown in Figure 4.1.1 thereby boosting the DC output voltage and power conversion efficiency as verified by simulation and measurement Vout as shown in

Figure 4.2.5. Simulation and measurements of the optimized RFCP’s input impedance are shown in Figure 4.2.6.

37

Figure 4.2. 3. Simulated magnitudes of DC and RF components after 1st stage.

Figure 4.2. 4. Simulated magnitudes of DC and RF components after 2nd stage.

38

The optimized RFCP design was prototyped on a double-sided copper FR4 substrate (εr = 4.7, tan(δ) = 0.022, h = 20 mil) due to its low cost and fast turn-around time as shown in Figure 2.3.3 and Figure 4.1.1. The RFCP’s RF input return loss, DC output voltage and energy efficiency were characterized using a NI PXI-E5632 Vector network analyzer (VNA) and Tektronix TDS 2024B

Oscilloscope with a 1 MΩ output resistor across the output capacitor Cp to emulate a typical PMU with 1µA leakage current [37]. The proposed RFCP charges Cp to a measured peak DC output of

2.1V (simulated 2.27V) from just -6dBm of RF input power in the 2.4 GHz band with measured

S11= -11dB (simulated -9.6dB) as shown in Figure 4.2.5. The measured return loss was observed to vary as a function of Cp’s charge up voltage and varied between -6 dB and -13 dB between output voltage of 0 V and 2.1V as shown by the measurements in Figure 4.2.6.

0 4

-5 3 -10 S11 Sim. S11 Meas. -15 2

Vout (V) Vout Vout Meas.

-20 Reflection Coefficient (dB) Coefficient Reflection 1 Vout Sim. -25

-30 0 2 2.5 3 3.5 4 Frequency (GHz)

Figure 4.2. 5. Simulated and measured DC output voltage and reflection coefficient vs RF input frequency.

39

Diode non-linearity is a known cause for impedance mismatches at low RF power levels. In order to study these effects due to load-side TLs, the RFCP’s input return loss and DC output were measured at RF input levels of between -9 and 3 dBm. The RFCP showed a near 50Ω input impedance (return loss between -7.4 and -15.8 dB), and DC output voltage of between 1.36 and

6.32V for between -9 and 0 dBm of RF input power as shown in Figure 4.2.6 and Figure 4.2.7 respectively, the phase shift of input impedance between measured and simulated is caused by the slight shift of resonating frequency of the fabricated RFCP as input impedance and DC output voltage were measured at the resonating frequency. Since the bulk of the RFCP’s RF input is converted to energy across Cp as in a WEH, the RFCP’s overall energy efficiency was determined using (2) as a function of RF input levels and is shown in Figure 4.2.7. The energy efficiency was found to increase from 8.3 to 60% for RF input levels of between -9 and 3dBm. Standing wave effect was observed using an e-field probe to measure the electrical field power density along the transmission line L1 and L2 as shown in Figure 4.2.8, the e-probe measured power density ripple is caused by the 90 degrees turn of L1 as shown in Figure 2.3.5 and Figure 4.1.1.

40

Figure 4.2. 6. Simulated and Measured RFCP input impedance for RF input power levels of between -9 and 3dBm and during charge-up.

10 100 9 Vout (V) - Sim. 90 8 80 7 Vout (V) - Meas. 70 6 60 5 50

Vout (V) Vout Energy Eff. (Meas) 4 40 3 30 2 20 (%) Eff. Energy 1 10 0 0 -20 -15 -10 -5 0 5 10 Input Power (dBm)

Figure 4.2. 7. Simulated and measured RFCP DC output voltage and energy efficiency vs. RF input power.

41

0 -2 -4 -6 -8

-10 TL L1 -12 TL L2

Normalized power density (dB) density power Normalized -14 0 22.5 45 67.5 90 Transmission Line Electrical Length at 2.4GHz ( degrees )

Figure 4.2. 8. Normalized electrical power density vs. transmission line electrical length.

42

4.3 Conclusion and Performance Comparison

A novel 2-stage RF-DC charge pump circuit design with load-side tuned rectifier stages to passively boost DC output and get reflection co-efficient to under -10 dB without a matching network is presented. Comparisons with other charge pumps with similar input power, frequency bands and output loads are shown in TABLE IV, and shows the proposed design achieving a higher

DC output with a near 50Ω input.

TABLE IV. Performance Comparison. (* includes antenna gain)

[46] [47] [48]* [24] This work

Freq.(GHz) 2.4 2.4 2.4 2.4 2.4

PRF (dBm) -9 to 3 -9 to 3 -9 to 2.5 -9 to 3 -9 to 3

RL (Ω) 1M 13k 10M 10M 1M

Vout (V) 0.1 to 1.2 0.76 to 4.3 0.9 to 3.8 1.2 to 1.9 1.4 to 6.5

43

Chapter 5. 2.4 GHz and 5.8 GHz Dual Band Rectifier

5.1 Introduction of A Novel Dual Band Working Mechanism

Figure 5.1. 1. Dual-band RF-DC charge pump (RFCP) topologies, (i) Parallel single-band half or 1-stage RFCPs and matching networks, (ii) Single half-stage RFCP with dual-band matching network, (iii) Single 2-stage RFCP with no matching network (Proposed work).

RF-DC charge pump circuits (RFCPs) are used for converting RF signals into usable DC voltage

(Vout) for either powering a resistive load in a wireless power harvester (WPH) or storing energy in a battery or capacitor in a wireless energy harvester (WEH). RFCPs are made up of one or more diode-capacitor pairs configured as voltage multipliers, and offer a passive method for converting

RF signals into DC form without start-up power. RFCP suffer from low power-conversion (PCE) or energy-conversion efficiency (ECE) due to diode losses at low RF inputs (PRF) and input reflections (Γt) due to the RFCP’s capacitive impedance. These losses limit the number of voltage- multiplying stages and output loads. Recently, benefits of harvesting power in multi-band RF signals in getting higher Vout have been reported [49]. But capacitive impedances of RFCPs makes it hard to have low input reflections in 2 or more bands.

44

Solutions to this problem have been proposed as shown in Figure 5.1.1 (i) and (ii). A common method is using dual-band matching networks with half or 1-stage RFCPs that are terminated with optimal output loads (RL) of between 1 and 13kΩ as shown in Figure 5.1.1 (ii). Tuned matching networks with a suitable load helps improve return loss, and produce a PCE of between 50 and

60% for input power of 0dBm and higher [50] [51]. A 2nd method is connecting 2 or more single- band matching networks and half or 1-stage RFCP’s tuned to different frequencies (0.9-2.4 GHz) in parallel as shown in Figure 5.1.1 (i) [52] [49]. In [51], a topology like Figure 5.1.1 (ii) is used to get resonances at higher 2.4 and 5.8 GHz with peak PCE of 62% but at a higher 10dBm of RF input into a 1kΩ output load. A common issue with half or 1-stage RFCPs is the low amount of

Vout for PRF below 0dBm. In the 2.4 and 5.8 GHz bands, maximum Vout of 1.65 and 0.31V respectively have been reported for PRF of -6dBm [53][24]. Getting a Vout of 1.8V with topologies in Figure 5.1.1 (i) and (ii) require power-consuming boost converters.

Another method used by 900 MHz RFID ICs is RFCPs with more stages and use of synchronized

FET switches as diodes with a large output energy storing capacitor in parallel with a large voltage- supervising resistive load (RL ≥ 1MΩ). RFID ICs get Vout of 1V or higher for input PRF of less than

-6dBm. At this point the RFID’s logic and modulator can be powered for limited duty cycles without Boost converters. However RFCPs with 2 or more stages in RFIDs require inductive antennas to match conjugately with RFCP’s capacitive impedance [27]. In [17], [48] and [46] higher output loads have shown to increase Vout at 2.4 GHz albeit at the cost of input match and

PCE.

45

In this work, a novel 2-stage RFCP structure that achieves dual-band Γt resonance in the 2.4 and

5.8 GHz bands without an external matching network by tuning the secondary feed is shown as in

Figure 5.1.1 (iii). Simulations and measurements show this RFCP needing RF power of just -7.3 and -3.7 dBm at 2.4 and 5.8 GHz individually to generate 1.8V across an output energy output capacitor (Cp) of 47µF and a 1MΩ voltage supervisor (Rleak) load in parallel [37]. And a 3.5V DC output with combined -3 dBm in both bands. The RFCP’s higher DC output with a large output capacitor and PMU allow heavy loads like a processor to run for limited duty cycles without Boost converters.

46

5.2 Working Principle and Analysis

Figure 5.2. 1. (i). Proposed dual-band, 50Ω RF-DC Charge pump design (RFCP) on 0.02 inch FR4; C1-C3= 2pF; Cp=47µF; D1-D4:BAT1503W; Rleak=1M; L3= optimized 2nd stage input feed for dual-band performance. (ii) 1st and 2nd stages of the RFCP.

The RFCP output voltage (Vout) is the cumulative sum of the DC output of each of its 4 half-wave rectifiers (RTPNs) in each of its 2 stages i.e. DC voltages across C1-C3 and CP that charge up during consecutive negative and positive RF cycles as shown in figure 5.2.1 (i). It has been shown in chapter 4 that tuning load-side transmission lines (TL) L1, L2 and L9 in the RFCP’s 1st and 2nd stages give each of its 4 RTPNs a higher DC output, and a near 50Ω impedance in the 2.4 GHz band without an external matching network (EMN). Further analysis covered in this chapter show

47

that dual band characteristics can be designed into the RFCP structure by tuning the input feed L3 and L10 as seen in Figure 5.2.1 (i).

In the 2.4 GHz band (f1), the two 50Ω RTPNs in each of the RFCP’s 2 stages conduct during each half of an RF cycle thereby giving each stage a 50Ω impedance over a full cycle. With the 2 stages combined in parallel as shown in Figure 5.2.1 (ii) and re-tuned, the overall RFCP impedance is matched (Measured Γt= -11dB as seen in Figure 4.2.5). As shown in Figure 5.3.1 and Figure 4.2.2,

L3 has less of an effect on input impedance and DC output voltage (Vout) of the RFCP.

However, in the 5.8 GHz band (f2), the RFCP’s 4 RTPNs in the 2 stages shown in Figure 5.2.1 (ii) are expected to be non 50Ω and of a capacitive nature. The total RFCP input impedance in the 5.8

GHz band (Zt-f2) is therefore approximately equal to the parallel combination of the impedance of its 2-stages i.e. (Z1-f2) and (Z2-f2) through a common L2 and C3 that get used during alternate half

RF cycles as in (8). The goal of this work is to study the effect of the RFCP’s 2nd stage feed L3 as well as 1st stage feed L10 on the impedance of the two stage RFCP with the aim of cancelling out the reactive part of RFCP’s 1st stage by L3 and tuning the real part of the impedance towards 50 ohm by L10. Doing so would make the RFCP’s overall impedance real and close to 50 ohm, thereby helping improve the RFCP’s overall reflection co-eff. in the 5.8 GHz band.

(푍1−푓2∙ 푍2−푓2) 푍푡−푓2 = (8) (푍1−푓2+ 푍2−푓2)

Starting the study of this dual band mechanism from the length of L10. The calculated effect of

L10 to RFCP’s input impedance and reflection coefficient with the assumed accordingly optimal

48

L3 length ( cancel out any reactive part of input impedance ideally ) is given in Figure 5.2.2, it is obvious L10 has the capability with the help of L3 to tune the impedance from 0 ohm to over 600 ohm as proposed, hence ensure a 50 ohm real impedance and improve the reflection co-eff.

Considering the possible additional phase delay in the fabricated board from footprints of capacitors and diodes, 0 mm was used for L10 instead of calculated optimal length which is 1 mm.

Figure 5.2. 2. Calculated effect of L10 to RFCP’s impedance and reflection coefficient with optimal L3 length.

49

5.3 Simulation and Measurement Results

Figure 5.3. 1. Simulated and Measured impedances of RFCP’s 1st stage, 2nd stage and overall RFCP as a function of L3 line length between 0 and 20 mm.

Large Signal S-Parameter (LSSP) and Harmonic Balance (HB) simulations in ADS were carried out on the proposed RFCP design to study the effects of tuning the 2nd stage input feed (L3) on input reflection co-eff. (Γt) and DC output (Vout) for different frequencies and RF input power

(PRF). The RFCP structure was designed with a 50Ω line for L3, and components as shown in

Figure 5.2.1 (i). An output resistor (Rleak) of 1 MΩ across Cp was used to emulate the PMU for duty cycle control in Wireless energy harvesting (WEH) [37]. Load side lines L1, L2, L5 and L9 were designed with optimal lengths of 14.5, 13.2 ,3.9 and 9.3 mm respectively to achieve a maximum DC output voltage in the 2.4 GHz band as shown in Chapter 4.

The simulated effect of tuning L3 on the impedances of the RFCP and its 2 stages in the 5.8 GHz band is shown in Figure 5.3.1. For L3=0mm and similar RTPNs in both of 2 stages, the simulated

50

impedances of RFCP’s 1st and 2nd stages were determined to be similar and capacitive as expected with Z1-f2 = 1.3 – j20.2Ω at 5.8 GHz. The effect of varying L3 between lengths of 0 and 20 mm

nd (254°) on the RFCP’s 2 stage is shown in Figure 5.3.1, and shows Z2-f2 vary between a mostly reactive 1.2 – j24.7 and 10.7 + j81.6 Ω. At 5.8 GHz both Z1-f2 and Z2-f2 are mostly reactive and away from 50Ω. However the parallel combination of Z1-f2 and Z2-f2 yields a mostly resistive Zt-f2 of 165.8 + j10 and 95– j4 Ω for L3= 3.6 and 17.2 mm respectively in the 5.8 GHz band as seen in

Figure 5.3.1. However L3=3.6mm is too short to connect the input port to C2 in the RFCP’s 2nd stage as seen in Figure 5.2.1 (i). Starting with an initial length of L3=17.2mm in the overall RFCP design, L3 was then optimized for maximizing Vout in 5.8 GHz band. The simulated effect of L3 on the RFCP’s input reflection (Γt) and DC output (Vout) is shown in Figure 5.3.2 (i) and shows a maximum Vout of 1.11V and Γt of -8dB for an optimal L3=17.8mm at PRF = -6dBm. The effect of varying L3 between 0 and 20 mm (254°) on the RFCP’s total impedance at 2.4 GHz (Zt-f1) is also shown in Figure 5.3.1 and shows Zt-f1 stay close to the 50Ω region.

Figure 5.3. 2. Simulated RFCP reflection co-eff. (Γt) and DC output (Vout) versus (i) L3 length in the 5.8 GHz band. (ii) Frequency. PRF = -6dBm.

51

The effect of L3 lengths on the RFCP’s Γt and Vout as a function of frequency is shown in Figure

5.3.2 (ii). It can be seen that increasing L3 from its optimal length of L3= 17.8 mm has no effect on the reflection co-eff. and DC output voltage in the 2.4 GHz band but produces a downward shift around the 5.8 GHz band justifying the dual band design principle of the proposed RFCP. The final RFCP was designed on FR4 with an L3 = 17.8mm for dual band resonance in the 5.8 GHz band as seen in Figure 5.2.1 (i).

0 4 -5 t (dB) t RL Sim.

Γ 3 -10 RL Meas. Vout Sim. -15 2 Vout Meas. -20 1 -25 (V) Vout -30 0 2 3 4 5 6 Frequency (GHz)

Figure 5.3. 3. RFCP measured and simulated reflection co-eff. (Γt) and DC output voltage (Vout) vs. Frequency for PRF=-6dBm.

0

-5 Prf=-9dBm -10

t (dB) t Prf=-3dBm Γ

-15 Prf=3dBm

-20 2 2.5 3 3.5 4 4.5 5 5.5 6

Frequency (GHz)

Figure 5.3. 4. Measured RFCP reflection co-eff. (Γt) vs. Frequency.

52

Figure 5.3. 5. Simulated and measured RFCP input impedance for RF input power of between -9 and 3 dBm for the following bands (i) 2.4 GHz (ii) 5.8 GHz

The reflection co-eff. and DC output of the RFCP versus frequency were measured using a NI

PXI-E VNA and Tektronics oscilloscope, and are shown along with the RFCP’s simulated results in Figure 5.3.3 for comparison. The dual band RFCP gives Vout of 2.12 and 1.26V, which agree with the simulated values of 2.265 and 1.11 V in the 2.4 and 5.8 GHz bands respectively for PRF=-

6dBm. The measured resonance of the RFCP was measured at a slightly lower 5.77 GHz compared to Simulation, which was due to the unstable permittivity of FR4 at higher frequency and the inaccuracy of simulation tool. The measured RFCP reflection co-efficients vs frequency are plotted in Figure 5.3.4 and Figure 5.3.5, and show stable return loss resonances in both 2.4 and 5.8 GHz bands for RF inputs of between -9 and 3 dBm. The RFCP also measures a high DC output of between 0.86 and 4.22V for RF input of between -9 and 3 dBm in the 5.8 GHz band plotted alongside the RFCP’s performance in the 2.4 GHz band as shown in Figure 5.3.6 (i). Since most of the harvested RF input power is used to charge the 47µF output capacitor (Cp), the RFCP’s

53

energy conversion efficiency (ECE) was gauged using the relationship from Chapter 4, and determined to be between 3.7 and 34.9 % as shown in Figure 5.3.6 (ii).

Figure 5.3. 6. (i) Simulated and Measured DC output vs RF input power. (ii) Measured Energy efficiency vs RF input power. Cp=47µF, Rleak=1MΩ.

Two-tone simulation was carried out in ADS and the result is shown in Figure 5.3.7. Before the input power level reaching diode’s forward break down voltage, adding the second tone at the same time does not deteriorate the DC output performance, so the standing wave introduced by the 2.4 GHz tone is not canceled or affected by adding the 5.8 GHz tone. The responding mechanism of 2.4 GHz is utilizing standing waves but the responding mechanism of 5.8 GHz is reply on self-reactance cancellation, they are working independently in RF and add up each of their DC output voltage linearly in DC as a result.

54

Figure 5.3. 7. Simulated DC output voltage vs. RF input power for two-tone signal.

55

5.4 Conclusion and Performance Comparison

This chapter shows a novel 2-stage RF-DC charge pump topology with a tuned secondary feed that incorporate dual-band return loss resonance in the 2.4 and 5.8 GHz bands without an external matching network. A comparison of this design with other RFCPs with comparable output loads and RF input power in TABLE V show the proposed design requiring a much lower RF input of -

7.3 and -3.7 dBm to generate 1.8V individually in the 2.4 and 5.8 GHz bands.

TABLE V. Performance Comparison. (* includes antenna gain)

[46] [17] [48]* [24] [53] [53] This work

Freq. (GHz) 2.4 2.4 2.4 2.4 5.78 CW 5.78 OOK 2.4 5.77

PRF (dBm) -9 to -9 to -9 to -9 to 3 -9 to -4 -9 to -4 -9 to 3 -9 to 3

3 3 3

RL (Ω) 1M 13k 10M 10M 150K 150K 1M 1M

Vout (V) 0.1 0.76 0.5 1.2 to 0.08 to 0.15 to 1.4 to 0.86 to

to to to 1.9 0.27 0.46 6.5 4.2

1.2 4.3 2.4

Vout at 0.35 1.36 0.7 1.65 0.18 0.31 2.14 1.3

-6 dBm(V)

PRF for Vout=1.8 6.5 -4 0.8 -3.5 NA NA -7.3 -3.7

V (dBm)

56

Chapter 6. 2.4 GHz and 5.8 GHz Dual Band Rectenna

6.1 Overview of the Rectenna

Figure 6.1. 1 Dual Band Rectenna.

A typical rectenna consists of three main sections: the antenna, used for receiving free space propagating electromagnetic wave, a rectifier which converts the radio frequency alternating signal received by the antenna into DC form, and a matching network that matches the rectifier with the antenna. A medium gain wide band Vivaldi antenna is designed for this application which is mentioned in Chapter 3, while the rectifier is a dual-band two-stage charge pump with optimized transmission line according to Chapter 4 and Chapter 5, additional matching network is eliminated in the proposed rectenna as both of antenna and rectifier have a near 50 ohm impedance at both of

57

interested bands. A photograph of the proposed rectenna is given in Figure 6.1.1. Both of antenna and rectifier are on a double-sided copper FR4 substrate (εr = 3.9, tan(δ) = 0.019, h = 20 mil).

58

6.2 Simulation and Measurement Results

Figure 6.2. 1. Test Setup for Rectenna Characterization. i.

Figure 6.2. 2. Test Setup for Rectenna Characterization. ii.

59

Individual characteristic of proposed antenna and rectifier are covered in Chapter 3-5, as for the characterization of the rectenna, measurements were performed by using a setup consists of an oscilloscope and two RF signal generators(RFSG) with power combiner and power amplifier(PA) as shown in Figure 6.2.1 and Figure 6.2.2. RFSG1 was used to generating 2.4G band signal meanwhile 5.8 G band signal was generated by RFSG2, dual-band signal was then combined through a two-way power combiner. Finally, transmitting antenna was fed by a PA to ensure the power level of individual channel could reach to 17 dBm. DC output voltage of the rectenna was sampled by an oscilloscope.

8 7 2.4 GHz Band Sim. 6 2.4 GHz Band Meas. 5 4 5.8 GHz Band Sim. 3 5.8 GHz Band Meas. 2

DC Output Voltage (V) Voltage Output DC 1 2.4 GHz Comparison. 0 0 20 40 60 80 100 Incident Power Density (µW/cm^2)

Figure 6.2. 3. DC output voltage vs. incident power density, performance at 2.4 GHz is compared with [48].

The measured and simulated DC output voltage verse incident power density at 2.4 G band single tone and 5.8 G band single tone are given in Figure 6.2.3. The power sensitivity of rectifier is known to be in favour of higher peak-to-average-power-ratio (PAPR) of the incident signal as higher transient peak incident voltage associated with higher PAPR [54]. As a result, DC output voltage vs. PAPR of the input RF signal at 2.4 G band and 5.8 G band for the proposed dual band 60

rectifier were carried out as seen in Figure 6.2.4. A typically designed 2.45 GHz rectenna [48] consists of a Vivaldi antenna with comparable gain as well as a high impedance load of 10 Mohm resister was compared with proposed dual band rectenna as shown in Figure 6.2.3, huge advantage of proposed topology in DC output voltage was proofed.

5 4 3 2 1 0 DC Output Voltage (V) Voltage Output DC 0 3 6 9 12 15 18 PAPR (dB)

2.4G Band Measured 5.8G Band Measured

Figure 6.2. 4. Measured DC output voltage vs. PAPR.

61

Chapter 7. Summary and Future Directions

A novel 2-stage RF-DC charge pump circuit design with load-side tuned rectifier stages to passively boost DC output and get reflection co-efficient to under -10 dB without a matching network is presented in chapter 4. In addition, keeping the same 2-stage RF-DC charge pump topology but with a tuned secondary feed that could incorporate dual-band return loss resonance in the 2.4 and 5.8 GHz bands without an external matching network is illustrated in chapter 5. For the proposed end application, a medium gain wide band Vivaldi antenna was designed for the rectenna and is shown in chapter 3, simulated as well as measured performance with multi-tone signals of the dual-band rectenna is in chapter 6. DC output is higher compared to other RF-DC charge pump circuits with comparable operating frequency, RF input power and output load. Input impedance is observed to be more immune to different RF input levels, load’s resistances and so more stable during changing up, however, the explanation of improved immunity of the proposed charge pump topology is not clear, studying of the mathematical model to explain overall improved immunity over the traditional matching network matched design during charging up would be a vital step, understanding of this principle could lead us to a revised design with even more improved sensitivity.

For other future directions, frequency shift issue of fabricated rectifier circuit could be solved by fabrication tolerance analysis to enable the proposed design topology working with in-door Wi-Fi routers. Amount of room could be saved by integrating the rectifier circuit, PMU and MCU on the proposed antenna. Furthermore, wider bandwidth that at least covers the entire Wi-Fi band could be achieved by adopting phase compensation structure to the load-side tuning transmission line

62

and higher boosted DC output voltage can be realized using less lossy microwave substrate

(Rogers Duroid). At the same time with all the above revised designs, a revised less gain but wider beamwidth antenna can be fitted to expand the coverage to harvesting from more surrounding Wi-

Fi devices. Therefore, charging time could be minimized and longer duty cycle is available for the end application.

63

References

[1] CM Consulting Inc, “Managing Canada’s Waste Batteries,” Peterborough, ON, 2012.

[2] T. R. Crompton, Battery Reference Book, 3rd ed. Newnes, 2000.

[3] Q. Wang, P. Ping, X. Zhao, G. Chu, J. Sun, and C. Chen, “Thermal runaway caused fire and

explosion of lithium ion battery,” J. Power Sources, vol. 208, pp. 210–224, Jun. 2012.

[4] Energizer Battery Manufacturing Inc., Alkaline Manganese Dioxide Handbook and

Application Manual. Energizer Battery Manufacturing Inc.

[5] Energizer Battery Manufacturing Inc., Nickel Metal Hydride (NiMH) Handbook and

Application Manual. Energizer Battery Manufacturing Inc., 2010.

[6] W. S. Loh, J.-L. Leong, and H. K. K. Tan, “Hazardous Foreign Bodies: Complications and

Management of Button Batteries in Nose,” Ann. Otol. Rhinol. Laryngol., vol. 112, no. 4, pp.

379–383, Apr. 2003.

[7] L. Järup, “Hazards of heavy metal contamination.,” British Medical Bulletin, vol. 68, pp.

167–82, 2003.

[8] A.G.Drachmann, “Heron’s Windmill,” Centaurus, 1961.

[9] J. Kymissis, C. Kendall, J. Paradiso, and N. Gershenfeld, “Parasitic power harvesting in

shoes,” in Digest of Papers. Second International Symposium on Wearable Computers (Cat.

No.98EX215), 1998, pp. 132–139.

[10] L. V. Minh, M. Hara, and H. Kuwano, “Miniaturized multi-beam (KNa)NbO 3 -based

energy harvester,” PowerMEMS, pp. 34–37, 2012.

[11] N. F. Besse, N., Quintero, A. V., Briand, D., Janphuang, P., Lockhart, R., Ruan, J. J., & de

Rooij, “Pzt-based energy harvesters on plastic foil optimized through theoretical modeling

and fabrication improvements,” in Proc. PowerMEMS, 2012, pp. 42–45.

64

[12] S. K. T. Ravindran, M. Kroener, and P. Woias, “A standalone pyroelectric harvester for

thermal energy harvesting,” PowerMEMS, pp. 2–5, 2012.

[13] M. G. and M. Raju, “ULP Meets Energy Harvesting: A game-changing combination for

Designers.” Texas Instruments, 2010.

[14] H.W.Ott, Noise Reductaion Techniques in Electronic Systems, 2nd ed. New York, US,

1988.

[15] J. M. Gilbert and F. Balouchi, “Comparison of energy harvesting systems for wireless

sensor networks,” Int. J. Autom. Comput., vol. 5, no. 4, pp. 334–347, 2008.

[16] Y. C. Cheng, Y. Chawathe, A. LaMarca, and J. Krumm, “Accuracy characterization for

metropolitan-scale Wi-Fi localization,” Proc. 3rd Int. Conf. Mob. Syst. Appl. Serv., pp. 233–

245, 2005.

[17] J. L. Volakis, U. Olgun, and C.-C. Chen, “Design of an efficient ambient WiFi energy

harvesting system,” IET Microwaves, Antennas Propag., vol. 6, no. 11, pp. 1200–1206,

2012.

[18] A. Dolgov, R. Zane, and Z. Popovic, “Power Management System for Online Low Power

RF Energy Harvesting Optimization,” IEEE Trans. Circuits Syst., vol. 57, no. 7, pp. 1802–

1811, 2010.

[19] R. J. Vyas, B. B. Cook, Y. Kawahara, and M. M. Tentzeris, “E-WEHP: A batteryless

embedded sensor-platform wirelessly powered from ambient digital-TV signals,” IEEE

Trans. Microw. Theory Tech., vol. 61, no. 6, pp. 2491–2505, 2013.

[20] V. Liu, A. Parks, V. Talla, S. Gollakota, D. Wetherall, and J. R. Smith, “Ambient

backscatter: wireless communication out of thin air,” Acm Sigcomm, vol. 43, no. 4, pp. 39–

50, 2013.

65

[21] H. Sun, Y. X. Guo, M. He, and Z. Zhong, “A dual-band rectenna using broadband yagi

antenna array for ambient rf power harvesting,” IEEE Antennas Wirel. Propag. Lett., vol.

12, pp. 918–921, 2013.

[22] P. Nintanavongsa, S. Member, U. Muncuk, and D. R. Lewis, “Design Optimization and

Implementation for RF Energy Harvesting Circuits Design Optimization and

Implementation for RF Energy Harvesting Circuits,” vol. 2, no. MARCH, pp. 24–33, 2012.

[23] B. R. Franciscatto, V. Freitas, and J. Duchamp, “High-Efficiency Rectifier Circuit at 2.45

GHz for Low-Input-Power RF Energy Harvesting,” Proc. 43rd Eur. Microw. Conf., pp.

507–510, 2013.

[24] J. Masuch, M. Delgado-Restituto, D. Milosevic, and P. Baltus, “Co-integration of an RF

energy harvester into a 2.4 GHz transceiver,” IEEE J. Solid-State Circuits, vol. 48, no. 7,

pp. 1565–1574, 2013.

[25] Z. Popovi et al., “Scalable RF Energy Harvesting,” IEEE Trans. Microw. Theory Tech., vol.

62, no. 4, pp. 1046–1056, 2014.

[26] M. A. Abouzied and E. Sanchez-Sinencio, “Low-Input Power-Level CMOS RF Energy-

Harvesting Front End,” IEEE Trans. Microw. Theory Tech., vol. 63, no. 11, pp. 3794–3805,

Nov. 2015.

[27] Z. Hameed and K. Moez, “A 3.2 v -15 dBm Adaptive Threshold-Voltage Compensated RF

Energy Harvester in 130 nm CMOS,” IEEE Trans. Circuits Syst. I Regul. Pap., vol. 62, no.

4, pp. 948–956, 2015.

[28] Z. Hameed and K. Moez, “Hybrid forward and backward threshold-compensated RF-DC

power converter for RF energy harvesting,” IEEE J. Emerg. Sel. Top. Circuits Syst., vol. 4,

no. 3, pp. 335–343, 2014.

66

[29] M. A. Abouzied, K. Ravichandran, and E. Sanchez-Sinencio, “A fully integrated

reconfigurable self-startup RF energy-harvesting system with storage capability,” IEEE J.

Solid-State Circuits, vol. 52, no. 3, pp. 704–719, 2017.

[30] J. Masuch, M. Delgado-Restituto, D. Milosevic, and P. Baltus, “Co-integration of an RF

energy harvester into a 2.4 GHz transceiver,” IEEE J. Solid-State Circuits, vol. 48, no. 7,

pp. 1565–1574, 2013.

[31] W. Lerdsitsomboon and K. K. O, “Technique for integration of a wireless switch in a 2.4

GHz single chip radio,” IEEE J. Solid-State Circuits, vol. 46, no. 2, pp. 368–377, 2011.

[32] T. M. Chiam, L. C. Ong, M. F. Karim, and Y. X. Guo, “5.8GHz circularly polarized

rectennas using schottky diode and LTC5535 rectifier for RF energy harvesting,” Asia

Pacific Microw. Conf., pp. 32–35, 2009.

[33] K. Nishida et al., “5.8 GHz high sensitivity rectenna array,” 2011 IEEE MTT-S Int. Microw.

Work. Ser. Innov. Wirel. Power Transm. Technol. Syst. Appl. IMWS-IWPT 2011 - Proc., no.

1, pp. 19–22, 2011.

[34] C. R. Valenta and G. D. Durgin, “Harvesting Wireless Power: Survey of Energy-Harvester

Conversion Efficiency in Far-Field, Wireless Power Transfer Systems,” IEEE Microw.

Mag., no. May, pp. 108–120, 2014.

[35] J. O. McSpadden, Lu Fan, and Kai Chang, “Design and experiments of a high-conversion-

efficiency 5.8-GHz rectenna,” IEEE Trans. Microw. Theory Tech., vol. 46, no. 12, pp.

2053–2060, 1998.

[36] D. Wang and R. Negra, “Design of a dual-band rectifier for wireless power transmission,”

in 2013 IEEE Wireless Power Transfer (WPT), 2013, pp. 127–130.

[37] Maxim 6461-6466, “Ultra-Low-Power Voltage Detectors and μP Supervisory Circuits.” pp.

67

1–10.

[38] P. J. Gibson, “The Vivaldi Aerial,” in 1979 9th European Microwave Conference, 1979,

pp. 101–105.

[39] M. C. Greenberg, K. L. Virga, and C. L. Hammond, “Performance characteristics of the

dual exponentially tapered slot antenna (detsa) for wireless communications applications,”

IEEE Trans. Veh. Technol., vol. 52, no. 2, pp. 305–312, Mar. 2003.

[40] A. Z. Hood, T. Karacolak, and E. Topsakal, “A Small Antipodal Vivaldi Antenna for

Ultrawide-Band Applications,” IEEE Antennas Wirel. Propag. Lett., vol. 7, pp. 656–660,

2008.

[41] F. Oktafiani, Y. S. Amrullah, Y. P. Saputera, Y. Wahyu, and Y. N. Wijayanto, “Analysis of

corrugated edge variations on balanced antipodal Vivaldi antennas,” in 2015 International

Conference on Radar, Antenna, Microwave, Electronics and Telecommunications

(ICRAMET), 2015, pp. 1–5.

[42] J. D. S. Langley, P. S. Hall, and P. Newham, “Balanced antipodal Vivaldi antenna for wide

bandwidth phased arrays,” IEE Proc. - Microwaves, Antennas Propag., vol. 143, no. 2, p.

97, 1996.

[43] J. B. Rizk and G. M. Rebeiz, “Millimeter-wave Fermi tapered slot antennas on

micromachined silicon substrates,” IEEE Trans. Antennas Propag., vol. 50, no. 3, pp. 379–

383, Mar. 2002.

[44] S. Sugawara, Y. Maita, K. Adachi, K. Mori, and K. Mizuno, “Characteristics of a MM-

wave tapered slot antenna with corrugated edges,” in 1998 IEEE MTT-S International

Microwave Symposium Digest (Cat. No.98CH36192), vol. 2, pp. 533–536.

[45] D. Schaubert, E. Kollberg, T. Korzeniowski, T. Thungren, J. Johansson, and K. Yngvesson,

68

“Endfire tapered slot antennas on dielectric substrates,” IEEE Trans. Antennas Propag., vol.

33, no. 12, pp. 1392–1400, Dec. 1985.

[46] H. J. Visser, V. Pop, J. M. Vullers, H. Centre, and C. Eindhoven, “Remote Rf Battery

Charging,” pp. 1–4, 2011.

[47] U. Olgun, C. C. Chen, and J. L. Volakis, “Wireless power harvesting with planar rectennas

for 2.45 GHz RFIDs,” Symp. Dig. - 20th URSI Int. Symp. Electromagn. Theory, EMTS 2010,

pp. 329–331, 2010.

[48] F. Congedo, G. Monti, L. Tarricone, and V. Bella, “A 2.45-GHz vivaldi rectenna for the

remote activation of an end device radio node,” IEEE Sens. J., vol. 13, no. 9, pp. 3454–

3461, 2013.

[49] V. Kuhn, C. Lahuec, F. Seguin, and C. Person, “A multi-band stacked RF energy harvester

with RF-to-DC efficiency up to 84%,” IEEE Trans. Microw. Theory Tech., vol. 63, no. 5,

pp. 1768–1778, 2015.

[50] Z. Liu, Z. Zhong, Y. Guo, and S. Member, “Enhanced Dual-Band Ambient RF Energy,”

IEEE Microwave Wirel. Components Lett., vol. 25, no. 9, pp. 630–632, 2015.

[51] K. Hamano, R. Tanaka, S. Yoshida, A. Miyachi, K. Nishikawa, and S. Kawasaki, “Design

of Concurrent Dual-Band Rectifier with Harmonic Signal Control,” p. 2017, 2017.

[52] K. Niotaki, A. Georgiadis, A. Collado, and J. S. Vardakas, “Dual-band resistance

compression networks for improved rectifier performance,” IEEE Trans. Microw. Theory

Tech., vol. 62, no. 12, pp. 3512–3521, 2014.

[53] P. Kuhn, P. Schmidt, F. Meyer, B. Gerd, A. Grabmaier, and A. Motivation, “Comparison

of Energy Harvesting via Modulation Schemes for Passive Sensor RFID,” pp. 63–68, 2016.

[54] C. R. Valenta, M. M. Morys, and G. D. Durgin, “Theoretical energy-conversion efficiency

69

for energy-harvesting circuits under power-optimized waveform excitation,” IEEE Trans.

Microw. Theory Tech., vol. 63, no. 5, pp. 1758–1767, 2015.

70