The Cascode Amplifier by Glenn B Coughlan a THESIS Submitted To
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Chapter 7: AC Transistor Amplifiers
Chapter 7: Transistors, part 2 Chapter 7: AC Transistor Amplifiers The transistor amplifiers that we studied in the last chapter have some serious problems for use in AC signals. Their most serious shortcoming is that there is a “dead region” where small signals do not turn on the transistor. So, if your signal is smaller than 0.6 V, or if it is negative, the transistor does not conduct and the amplifier does not work. Design goals for an AC amplifier Before moving on to making a better AC amplifier, let’s define some useful terms. We define the output range to be the range of possible output voltages. We refer to the maximum and minimum output voltages as the rail voltages and the output swing is the difference between the rail voltages. The input range is the range of input voltages that produce outputs which are not at either rail voltage. Our goal in designing an AC amplifier is to get an input range and output range which is symmetric around zero and ensure that there is not a dead region. To do this we need make sure that the transistor is in conduction for all of our input range. How does this work? We do it by adding an offset voltage to the input to make sure the voltage presented to the transistor’s base with no input signal, the resting or quiescent voltage , is well above ground. In lab 6, the function generator provided the offset, in this chapter we will show how to design an amplifier which provides its own offset. -
Power Electronics
Diodes and Transistors Semiconductors • Semiconductor devices are made of alternating layers of positively doped material (P) and negatively doped material (N). • Diodes are PN or NP, BJTs are PNP or NPN, IGBTs are PNPN. Other devices are more complex Diodes • A diode is a device which allows flow in one direction but not the other. • When conducting, the diodes create a voltage drop, kind of acting like a resistor • There are three main types of power diodes – Power Diode – Fast recovery diode – Schottky Diodes Power Diodes • Max properties: 1500V, 400A, 1kHz • Forward voltage drop of 0.7 V when on Diode circuit voltage measurements: (a) Forward biased. (b) Reverse biased. Fast Recovery Diodes • Max properties: similar to regular power diodes but recover time as low as 50ns • The following is a graph of a diode’s recovery time. trr is shorter for fast recovery diodes Schottky Diodes • Max properties: 400V, 400A • Very fast recovery time • Lower voltage drop when conducting than regular diodes • Ideal for high current low voltage applications Current vs Voltage Characteristics • All diodes have two main weaknesses – Leakage current when the diode is off. This is power loss – Voltage drop when the diode is conducting. This is directly converted to heat, i.e. power loss • Other problems to watch for: – Notice the reverse current in the recovery time graph. This can be limited through certain circuits. Ways Around Maximum Properties • To overcome maximum voltage, we can use the diodes in series. Here is a voltage sharing circuit • To overcome maximum current, we can use the diodes in parallel. -
Cascode Amplifiers by Dennis L. Feucht Two-Transistor Combinations
Cascode Amplifiers by Dennis L. Feucht Two-transistor combinations, such as the Darlington configuration, provide advantages over single-transistor amplifier stages. Another two-transistor combination in the analog designer's circuit library combines a common-emitter (CE) input configuration with a common-base (CB) output. This article presents the design equations for the basic cascode amplifier and then offers other useful variations. (FETs instead of BJTs can also be used to form cascode amplifiers.) Together, the two transistors overcome some of the performance limitations of either the CE or CB configurations. Basic Cascode Stage The basic cascode amplifier consists of an input common-emitter (CE) configuration driving an output common-base (CB), as shown above. The voltage gain is, by the transresistance method, the ratio of the resistance across which the output voltage is developed by the common input-output loop current over the resistance across which the input voltage generates that current, modified by the α current losses in the transistors: v R A = out = −α ⋅α ⋅ L v 1 2 β + + + vin RB /( 1 1) re1 RE where re1 is Q1 dynamic emitter resistance. This gain is identical for a CE amplifier except for the additional α2 loss of Q2. The advantage of the cascode is that when the output resistance, ro, of Q2 is included, the CB incremental output resistance is higher than for the CE. For a bipolar junction transistor (BJT), this may be insignificant at low frequencies. The CB isolates the collector-base capacitance, Cbc (or Cµ of the hybrid-π BJT model), from the input by returning it to a dynamic ground at VB. -
Notes for Lab 1 (Bipolar (Junction) Transistor Lab)
ECE 327: Electronic Devices and Circuits Laboratory I Notes for Lab 1 (Bipolar (Junction) Transistor Lab) 1. Introduce bipolar junction transistors • “Transistor man” (from The Art of Electronics (2nd edition) by Horowitz and Hill) – Transistors are not “switches” – Base–emitter diode current sets collector–emitter resistance – Transistors are “dynamic resistors” (i.e., “transfer resistor”) – Act like closed switch in “saturation” mode – Act like open switch in “cutoff” mode – Act like current amplifier in “active” mode • Active-mode BJT model – Collector resistance is dynamically set so that collector current is β times base current – β is assumed to be very high (β ≈ 100–200 in this laboratory) – Under most conditions, base current is negligible, so collector and emitter current are equal – β ≈ hfe ≈ hFE – Good designs only depend on β being large – The active-mode model: ∗ Assumptions: · Must have vEC > 0.2 V (otherwise, in saturation) · Must have very low input impedance compared to βRE ∗ Consequences: · iB ≈ 0 · vE = vB ± 0.7 V · iC ≈ iE – Typically, use base and emitter voltages to find emitter current. Finish analysis by setting collector current equal to emitter current. • Symbols – Arrow represents base–emitter diode (i.e., emitter always has arrow) – npn transistor: Base–emitter diode is “not pointing in” – pnp transistor: Emitter–base diode “points in proudly” – See part pin-outs for easy wiring key • “Common” configurations: hold one terminal constant, vary a second, and use the third as output – common-collector ties collector -
ECE 255, MOSFET Basic Configurations
ECE 255, MOSFET Basic Configurations 8 March 2018 In this lecture, we will go back to Section 7.3, and the basic configurations of MOSFET amplifiers will be studied similar to that of BJT. Previously, it has been shown that with the transistor DC biased at the appropriate point (Q point or operating point), linear relations can be derived between the small voltage signal and current signal. We will continue this analysis with MOSFETs, starting with the common-source amplifier. 1 Common-Source (CS) Amplifier The common-source (CS) amplifier for MOSFET is the analogue of the common- emitter amplifier for BJT. Its popularity arises from its high gain, and that by cascading a number of them, larger amplification of the signal can be achieved. 1.1 Chararacteristic Parameters of the CS Amplifier Figure 1(a) shows the small-signal model for the common-source amplifier. Here, RD is considered part of the amplifier and is the resistance that one measures between the drain and the ground. The small-signal model can be replaced by its hybrid-π model as shown in Figure 1(b). Then the current induced in the output port is i = −gmvgs as indicated by the current source. Thus vo = −gmvgsRD (1.1) By inspection, one sees that Rin = 1; vi = vsig; vgs = vi (1.2) Thus the open-circuit voltage gain is vo Avo = = −gmRD (1.3) vi Printed on March 14, 2018 at 10 : 48: W.C. Chew and S.K. Gupta. 1 One can replace a linear circuit driven by a source by its Th´evenin equivalence. -
Cascode Techniques
Analysis and Design of Analog Integrated Circuits Lecture 8 Cascode Techniques Michael H. Perrott February 15, 2012 Copyright © 2012 by Michael H. Perrott All rights reserved. M.H. Perrott Review of Large Signal Analysis of Current Mirrors Vdd Δ V2 I I 1 2 1 μ W2 2 λ nCox (VGS2-VTH) (1+ 2Vds2) I2 2 L = 2 I 1 W 2 1 μ C 1(V -V ) (1+λ V ) 2 n ox GS1 TH 1 ds1 M L1 2 ΔV M1 Vds2 > Vdsat2 1 V +ΔV V +ΔV Δ Δ Δ Δ Vss=0 TH 1 TH 2 But, VTH+ V1=VTH+ V2 V1 = V2 λ I2 W2 L1 (1+ 2Vds2) I2 = λ I1 W1 L2 (1+ 1Vds1) M2 in Saturation Current Mismatch setting due to Vds based on difference M2 in Triode geometry Note: for accurate ratio, set L1 = L2 Vds2 Vdsat2 M.H. Perrott 2 The Issue of Vds Mismatch in Current Mirrors V λ dd I2 W2 (1+ 2Vds2) = λ I1 W1 (1+ 1Vds1) I1 I2 Current Mismatch setting due to Vds based on difference geometry V V ds1 ds2 Note: we are assuming L = L M1 M2 1 2 . Issue: Current I2 can vary significantly as a function of the drain voltage of M2 - We often want a tightly controlled current set only by I1 and transistor sizes . How do we improve the current mirror matching performance? M.H. Perrott 3 Cascoded Current Source I Rth R ref d3 thd3 Ibias Vbias V M bias 3 M3 M ro1 2 M1 . Offers increased output resistance - Reduces small signal dependence of output current on the output voltage of the current source - From Lecture 6, we derived: . -
6 Insulated-Gate Field-Effect Transistors
Chapter 6 INSULATED-GATE FIELD-EFFECT TRANSISTORS Contents 6.1 Introduction ......................................301 6.2 Depletion-type IGFETs ...............................302 6.3 Enhancement-type IGFETs – PENDING .....................311 6.4 Active-mode operation – PENDING .......................311 6.5 The common-source amplifier – PENDING ...................312 6.6 The common-drain amplifier – PENDING ....................312 6.7 The common-gate amplifier – PENDING ....................312 6.8 Biasing techniques – PENDING ..........................312 6.9 Transistor ratings and packages – PENDING .................312 6.10 IGFET quirks – PENDING .............................313 6.11 MESFETs – PENDING ................................313 6.12 IGBTs ..........................................313 *** INCOMPLETE *** 6.1 Introduction As was stated in the last chapter, there is more than one type of field-effect transistor. The junction field-effect transistor, or JFET, uses voltage applied across a reverse-biased PN junc- tion to control the width of that junction’s depletion region, which then controls the conduc- tivity of a semiconductor channel through which the controlled current moves. Another type of field-effect device – the insulated gate field-effect transistor, or IGFET – exploits a similar principle of a depletion region controlling conductivity through a semiconductor channel, but it differs primarily from the JFET in that there is no direct connection between the gate lead 301 302 CHAPTER 6. INSULATED-GATE FIELD-EFFECT TRANSISTORS and the semiconductor material itself. Rather, the gate lead is insulated from the transistor body by a thin barrier, hence the term insulated gate. This insulating barrier acts like the di- electric layer of a capacitor, and allows gate-to-source voltage to influence the depletion region electrostatically rather than by direct connection. In addition to a choice of N-channel versus P-channel design, IGFETs come in two major types: enhancement and depletion. -
CHAPTER 3 Frequency Response of Basic BJT and MOSFET Amplifiers (Review Materials in Appendices III and V)
CHAPTER 3 Frequency Response of Basic BJT and MOSFET Amplifiers (Review materials in Appendices III and V) In this chapter you will learn about the general form of the frequency domain transfer function of an amplifier. You will learn to analyze the amplifier equivalent circuit and determine the critical frequencies that limit the response at low and high frequencies. You will learn some special techniques to determine these frequencies. BJT and MOSFET amplifiers will be considered. You will also learn the concepts that are pursued to design a wide band width amplifier. Following topics will be considered. Review of Bode plot technique. Ways to write the transfer (i.e., gain) functions to show frequency dependence. Band-width limiting at low frequencies (i.e., DC to fL). Determination of lower band cut-off frequency for a single-stage amplifier – short circuit time constant technique. Band-width limiting at high frequencies for a single-stage amplifier. Determination of upper band cut-off frequency- several alternative techniques. Frequency response of a single device (BJT, MOSFET). Concepts related to wide-band amplifier design – BJT and MOSFET examples. 3.1 A short review on Bode plot technique Example: Produce the Bode plots for the magnitude and phase of the transfer function 10s Ts() , for frequencies between 1 rad/sec to 106 rad/sec. (1ss / 1025 )(1 / 10 ) We first observe that the function has zeros and poles in the numerical sequence 0 (zero), 2 5 2 10 (pole), and 10 (pole). Further at ω=1 rad/sec i.e., lot less than the first pole (at ω=10 rad/sec), Ts() 10 s. -
Optimal High Performance Self Cascode CMOS Current Mirror
View metadata, citation and similar papers at core.ac.uk brought to you by CORE provided by Global Journal of Computer Science and Technology (GJCST) Global Journal of Computer Science and Technology Volume 11 Issue 15 Version 1.0 September 2011 Type: Double Blind Peer Reviewed International Research Journal Publisher: Global Journals Inc. (USA) Online ISSN: 0975-4172 & Print ISSN: 0975-4350 Optimal High Performance Self Cascode CMOS Current Mirror By Vivek Pant, Shweta Khurana Kurukshetra University Kurukshetra Abstract - In this paper the current mirror presented, having low voltage and mixed mode structure has been proposed. The performance of self cascade MOSFET current mirror is optimized with high output impedance and can operate at 1 V or below. Simulation results conform to Analog Mentor tools having Design Architect for schematics and Eldonet for SPICE simulation, with input reference current of 20μA. This review paper presents a comparative performance study of self cascode current mirror with other current mirrors. Keywords : current mirrors, cascode current mirror, low voltage analog circuit. GJCST Classification : I.2.9 Optimal High Performance Self Cascode CMOS Current Mirror Strictly as per the compliance and regulations of: © 2011. Vivek Pant, Shweta Khurana.This is a research/review paper, distributed under the terms of the Creative Commons Attribution-Noncommercial 3.0 Unported License http://creativecommons.org/licenses/by-nc/3.0/), permitting all non commercial use, distribution, and reproduction in any medium, provided the original work is properly cited. Optimal High Performance Self Cascode CMOS Current Mirror Vivek Pantα, Shweta KhuranaΩ Abstract - In this paper the current mirror presented, having I = I (W/L) 2 (1+λVds2) (3) out ref low voltage and mixed mode structure has been proposed. -
Differential Amplifiers
www.getmyuni.com Operational Amplifiers: The operational amplifier is a direct-coupled high gain amplifier usable from 0 to over 1MH Z to which feedback is added to control its overall response characteristic i.e. gain and bandwidth. The op-amp exhibits the gain down to zero frequency. Such direct coupled (dc) amplifiers do not use blocking (coupling and by pass) capacitors since these would reduce the amplification to zero at zero frequency. Large by pass capacitors may be used but it is not possible to fabricate large capacitors on a IC chip. The capacitors fabricated are usually less than 20 pf. Transistor, diodes and resistors are also fabricated on the same chip. Differential Amplifiers: Differential amplifier is a basic building block of an op-amp. The function of a differential amplifier is to amplify the difference between two input signals. How the differential amplifier is developed? Let us consider two emitter-biased circuits as shown in fig. 1. Fig. 1 The two transistors Q1 and Q2 have identical characteristics. The resistances of the circuits are equal, i.e. RE1 = R E2, RC1 = R C2 and the magnitude of +VCC is equal to the magnitude of �VEE. These voltages are measured with respect to ground. To make a differential amplifier, the two circuits are connected as shown in fig. 1. The two +VCC and �VEE supply terminals are made common because they are same. The two emitters are also connected and the parallel combination of RE1 and RE2 is replaced by a resistance RE. The two input signals v1 & v2 are applied at the base of Q1 and at the base of Q2. -
INA106: Precision Gain = 10 Differential Amplifier Datasheet
INA106 IN A1 06 IN A106 SBOS152A – AUGUST 1987 – REVISED OCTOBER 2003 Precision Gain = 10 DIFFERENTIAL AMPLIFIER FEATURES APPLICATIONS ● ACCURATE GAIN: ±0.025% max ● G = 10 DIFFERENTIAL AMPLIFIER ● HIGH COMMON-MODE REJECTION: 86dB min ● G = +10 AMPLIFIER ● NONLINEARITY: 0.001% max ● G = –10 AMPLIFIER ● EASY TO USE ● G = +11 AMPLIFIER ● PLASTIC 8-PIN DIP, SO-8 SOIC ● INSTRUMENTATION AMPLIFIER PACKAGES DESCRIPTION R1 R2 10kΩ 100kΩ 2 5 The INA106 is a monolithic Gain = 10 differential amplifier –In Sense consisting of a precision op amp and on-chip metal film 7 resistors. The resistors are laser trimmed for accurate gain V+ and high common-mode rejection. Excellent TCR tracking 6 of the resistors maintains gain accuracy and common-mode Output rejection over temperature. 4 V– The differential amplifier is the foundation of many com- R3 R4 10kΩ 100kΩ monly used circuits. The INA106 provides this precision 3 1 circuit function without using an expensive resistor network. +In Reference The INA106 is available in 8-pin plastic DIP and SO-8 surface-mount packages. Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet. All trademarks are the property of their respective owners. PRODUCTION DATA information is current as of publication date. Copyright © 1987-2003, Texas Instruments Incorporated Products conform to specifications per the terms of Texas Instruments standard warranty. Production processing does not necessarily include testing of all parameters. www.ti.com SPECIFICATIONS ELECTRICAL ° ± At +25 C, VS = 15V, unless otherwise specified. -
Building a Vacuum Tube Amplifier
Brian Large Physics 398 EMI Final Report Building a Vacuum Tube Amplifier First, let me preface this paper by saying that I will try to write it for the idiot, because I knew absolutely nothing about amplifiers before I started. Project Selection I had tried putting together a stomp box once before (in high school), with little success, so the idea of building something as a project for this course intrigued me. I also was intrigued with building an amplifier because it is the part of the sound creation process that I understand the least. Maybe that’s why I’m not an electrical engineer. Anyways, I already own a solid-state amplifier (a Fender Deluxe 112 Plus), so I decided that it would be nice to add a tube amp. I drew up a list of “Things I wanted in an Amplifier” and began to discuss the matters with the course instructor, Professor Steve Errede, and my lab TA, Dan Finkenstadt. I had decided I wanted a class AB push-pull amp putting out 10-20 Watts. I had decided against a reverb unit because it made the schematic much more complicated and would also cost more. With these decisions in mind, we decided that a Fender Tweed Deluxe would be the easiest to build that fit my needs. The Deluxe schematic that I built my amp off of was a 5E3, built by Fender from 1955-1960 (Figure 1). The reason we chose such an old amp is because they sound great and their schematics are considerably easier to build.