Application Note 32 March 1989

High Effi ciency Linear Regulators Jim Williams

Introduction Linear voltage regulators continue to enjoy widespread use switching supply output. Figure 1 shows such an arrange- despite the increasing popularity of switching approaches. ment. The main output (“A”) is stabilized by feedback to Linear regulators are easily implemented, and have much the switching regulator. Usually, this output supplies most better noise and drift characteristics than switchers. Ad- of the power taken from the circuit. Because of this, the ditionally, they do not radiate RF, function with standard amount of energy in the is relatively unaf- magnetics, are easily frequency compensated, and have fected by power demands at the “B” and “C” outputs. fast response. Their largest disadvantage is ineffi ciency. This results in relatively constant “B” and “C” regulator Excess energy is dissipated as heat. This elegantly sim- input voltages. Judicious design allows the regulators to plistic regulation mechanism pays dearly in terms of lost run at or near their dropout voltage, regardless of loading power. Because of this, linear regulators are associated or switcher input voltage. Low dropout regulators thus with excessive dissipation, ineffi ciency, high operating save considerable power and dissipation. temperatures and large heat sinks. While linears cannot L, LT, LTC, LTM, Linear Technology and the Linear logo are registered trademarks of Linear compete with switchers in these areas they can achieve Technology Corporation. All other trademarks are the property of their respective owners. signifi cantly better results than generally supposed. New $V components and some design techniques permit reten- +VIN tion of linear regulator’s advantages while improving VREG “C” effi ciency. One way towards improved effi ciency is to minimize the input-to-output voltage across the regulator. The smaller $V this term is, the lower the power loss. The minimum input/ VREG “B” output voltage required to support regulation is referred to as the “dropout voltage.” Various design techniques and technologies offer different performance capabilities. Appendix A, “Achieving Low Dropout,” compares some approaches. Conventional three terminal linear regulators “A” have a 3V dropout, while newer devices feature 1.5V drop- out (see Appendix B, “A Low Dropout Regulator Family”) SWITCHING at 7.5A, decreasing to 0.05V at 100μA. REGULATOR

Regulation from Stable Inputs AN32 F01 Lower dropout voltage results in signifi cant power sav- ings where input voltage is relatively constant. This is Figure 1. Typical Switching Supply Arrangement Showing Linear normally the case where a linear regulator post-regulates a Post-Regulators

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AN32-1 Application Note 32

Regulation from Unstable Input—AC Line Derived Case Unfortunately, not all applications furnish a stable input the output voltage. The 15V output comparison still favors voltage. One of the most common and important situations the low dropout regulator, although effi ciency benefi t is is also one of the most diffi cult. Figure 2 diagrams a classic somewhat reduced. Figure 4 derives resultant regulator situation where the linear regulator is driven from the AC power dissipation from Figure 3’s data. These plots show line via a step-down transformer. A 90VAC (brownout) to that the LT1086 requires less heat sink area to maintain 140VAC (high line) line swing causes the regulator to see the same die temperature as the LM317. a proportionate input voltage change. Figure 3 details ef- Both curves show the deleterious effects of poorly con- fi ciency under these conditions for standard (LM317) and trolled input voltages. The low dropout device clearly cuts low dropout (LT®1086) type devices. The LT1086’s lower losses, but input voltage variation degrades obtainable dropout improves effi ciency. This is particularly evident effi ciency. at 5V output, where dropout is a signifi cant percentage of

15 14 13 12 LM317 VO = 15V, 1A 11 10 LT1086 VO = 15V, 1A 9 VIN—MOVES WITH 8 LM317 VO = 5V, 1A AC LINE VOLTAGE 7 LINEAR 6 V VOLTAGE OUT 5 REGULATOR 90 TO 140 4 AN32 F02 3

VAC LINE IN WATTS POWER DISSIPATION 2 LT1086 VO = 5V, 1A 1 + 0 80 90 100 110120 130 140 AC LINE VOLTAGE

AN32 F04

Figure 2. Typical AC Line Driven Linear Regulator Figure 4. Power Dissipation for Different Regulators vs AC Line Voltage. Rectifi er Losses are not Included

120 110 LT1086 WITH SCR PRE-REGULATOR 100 V = 15V, 1A V = 16.5V VOUT = 15V, 1A OUT 90 IN V = 18V 80 IN VIN = 6.5V LM317 70 LT1086 60 VIN = 8V VIN = 25.6V VIN = 28V 50 LT1086 WITH VIN = 10.1V

EFFICIENCY (%) 40 SCR PRE-REGULATOR VOUT = 5V, 1A LT1086 LM317 VIN = 12.4V V = 5V, 1A 30 OUT 20 DATA DOES NOT INCLUDE 10 DIODE LOSSES. SCR PRE-REGULATOR LM317 DROPOUT = 3V INCLUDES SCR LOSSES LT1086 DROPOUT = 1.5V 0 80 90 100 110 120 130 140 150 AC LINE VOLTAGE

AN32 F03

Figure 3. Effi ciency vs AC Line Voltage for LT1086 and LM317 Regulators

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AN32-2 Application Note 32

SCR Pre-Regulator Figure 5 shows a way to eliminate regulator input variations, SCR and a path from the main transformer to L1 (Trace D) even with wide AC line swings. This circuit, combined with occurs. The resultant current fl ow (Trace E), limited by L1, a low dropout regulator, provides high effi ciency while charges the 4700μF . When the transformer out- retaining all the linear regulators desirable characteristics. put drops low enough the SCR commutates and charging This design servo controls the fi ring point of the SCRs to ceases. On the next half-cycle the process repeats, except stabilize the LT1086 input voltage. A1 compares a portion that the alternate SCR does the work (Traces F and G are of the LT1086’s input voltage to the LT1004 reference. The the individual SCR currents). The loop phase modulates amplifi ed difference voltage is applied to C1B’s negative the SCR’s fi ring point to maintain a constant LT1086 input input. C1B compares this to a line synchronous ramp voltage. A1’s 1μF capacitor compensates the loop and its (Trace B, Figure 6) derived by C1A from the output 10kΩ-diode network ensures start-up. The three rectifi ed secondary (Trace A is the “sync” point in the terminal regulator’s current limit protects the circuit from fi gure). C1B’s pulse output (Trace C) fi res the appropriate overloads.

L1 500μH ≈16.5V TO 17V 15V L2 LT1086 OUT 1N4148 + 1A + 121Ω* 20VAC 4700μF 158k* 20μF AT 115VIN (TRIM) 90VAC TO 1k 140VAC 1.37k* 20VAC AT 115VIN 12k*

1N4002 1N4002

“SYNC” +V

1N4002 10k ≈30V TO ALL + 200k “+V” POINTS 1N4148 + C1A 22μF 750Ω 1/2 LT1018 – 0.1μF

+V

0.001 2.4k + C1B 1/2 LT1018 1N4148 + – A1 LT1006 10k 10k 10k – +V 1μF +V

L1 = PULSE ENGINEERING, INC. #PE-50503 LT1004 L2 = TRIAD F-271U (TYPICAL) 1.2V

* = 1% FILM AN32 F05 = G.E. C-106B

Figure 5. SCR Pre-Regulator

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AN32-3 Application Note 32

This circuit has a dramatic impact on LT1086 effi ciency DC Input Pre-Regulator versus AC line swing*. Referring back to Figure 3, the Figure 8a’s circuit is useful where the input is DC, such data shows good effi ciency with no change for 90VAC as an unregulated (or regulated) supply or a battery. This to 140VAC input variations. This circuit’s slow switching circuit is designed for low losses at high currents. The preserves the linear regulators low noise. Figure 7 shows LT1083 functions in conventional fashion, supplying a slight 120Hz residue with no wideband components. regulated output at 7.5A capacity. The remaining com- *The transformer used in a pre-regulator can signifi cantly infl uence overall ponents form a switched-mode dissipation regulator. effi ciency. One way to evaluate power consumption is to measure the actual power taken from the 115VAC line. See Appendix C, “Measuring This regulator maintains the LT1083 input just above the Power Consumption.” dropout voltage under all conditions. When the LT1083 input (Trace A, Figure 9) decays far enough, C1A goes high,

A = 50V/DIV B = 10V/DIV

C = 50V/DIV

D = 20V/DIV A = 5mV/DIV E = 10A/DIV (AC-COUPLED)

F = 10A/DIV G = 10A/DIV

HORIZ = 5ms/DIV AN32 F06 HORIZ = 5ms/DIV AN32 F07

Figure 6. Waveforms for the SCR Pre-Regulator Figure 7. Output Noise for the SCR Pre-Regulated Circuit

Q1 L2 P50N05E 335μH ≈6.75V 12V DS 5V IN LT1083-5 OUT (7V TO 20V) + + 1N966 + + 7.5A 10μF 330Ω 47μF 16V 1000μF 10μF

L1 MBR1060 820μH 100Ω 1N4148 5.1k 56k* (TRIM) 1N4148 270k ≈30V 12V + 22μF 100k D – Q2 C1B VN2222 0.01 S

1/2 LT1018 + – 1N4148 C1A

10k 1/2 LT1018 12k* + 20k 12k 12V 10k AN32 F08a 12V LT1004 47pF 1.2V 2k 1M L1 = PULSE ENGINEERING, INC. #PE-52630 L2 = PULSE ENGINEERING, INC. #PE-51518 P50N05E = MOTOROLA

Figure 8a. Pre-Regulated Low Dropout Regulator an32f

AN32-4 Application Note 32

FROM LT1083 V L2 + OUT 120 470Ω 10μF

12V 1N4148 1k

V ≈ 1.8V

4N28 REF – 10k AN32 F08b

TO REMAINING

1/2 LT1018

CIRCUITRY + 20k 12k 12V 1N4148 47pF 1M

Figure 8b. Differential Sensing for the Pre-Regulator Allows Variable Outputs

100 VIN = 12V 90 80 70 • • • 60 • 50 A = 100mV/DIV (AC-COUPLED) 40 EFFICIENCY (%) 30 B = 20V/DIV 20 10 C = 20V/DIV 0 0 132 4 5 6 7 8 D = 4A/DIV OUTPUT CURRENT (A)

AN32 F10 HORIZ = 100μs/DIV AN32 F09 Figure 9. Pre-Regulator Waveforms Figure 10.Effi ciency vs Output for Figure 8a allowing Q1’s gate (Trace B) to rise. This turns on Q1, and overdrives. These measures are required because alterna- its source (Trace C) drives current (Trace D) into L2 and the tives are unattractive. Low loss P-channel devices are not 1000μF capacitor, raising regulator input voltage. When currently available, and bipolar approaches require large the regulator input rises far enough C1A returns low, Q1 drive currents or have poor saturation. As before, the cuts off and capacitor charging ceases. The MBR1060 linear regulator’s current limit protects against overloads. damps L2’s fl yback spike and the 1M-47pF combination Figure 10 plots effi ciency for the pre-regulated LT1083 over sets loop hysteresis at about 100mV. a range of currents. Results are favorable, and the linear Q1, an N-channel MOSFET, has only 0.028Ω of satura- regulator’s noise and response advantages are retained. tion loss but requires 10V of gate-source turn-on bias. Figure 8b shows an alternate feedback connection which C1B, set up as a simple fl yback voltage booster, provides maintains a fi xed small voltage across the LT1083 in ap- about 30V DC boost to Q2. Q2, serving as a high voltage plications where variable output is desired. This scheme pull-up for C1A, provides voltage overdrive to Q1’s gate. maintains effi ciency as the LT1083’s output voltage is This ensures Q1 saturation, despite its source follower varied. connection. The clamps excessive gate-source

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AN32-5 Application Note 32

Q1 5V P50N05E 10A 0.01Ω** DS VIN 1N966 + 10k* 10k* 16V 22μF 100k*

180Ω

– 1N4148 38k* A1B

1/2 LT1013 + 0.001

100k* –

VIN L1

A1A 12k*

150μH 1/2 LT1013 + 10k + VIN AN32 F11 4.7μF V LT1004 IN 1.2V VSW 1N4148 LT1072 ≈30V 28k + L1 = PULSE ENGINEERING, INC. #PE-52645 47μF P50N05E = MOTOROLA FB * = 1% FILM RESISTOR GND VC 1.2k ** = DALE RH-10

1k + 1μF

Figure 11. 10A Regulator with 400mV Dropout

10A Regulator with 400mV Dropout across the 0.01Ω , provides by forcing In some circumstances an extremely low dropout regulator A1A to swing negatively. The low resistance shunt limits may be required. Figure 11 is substantially more complex loss to only 100mV at 10A output. Figure 12 plots current than a three terminal regulator, but offers 400mV dropout limit performance for the regulator. Roll-off is smooth, with at 10A output. This design borrows Figure 8A’s overdriven no oscillation or undesirable characteristics. source follower technique to obtain extremely low satura- 6 tion resistance. The gate boost voltage is generated by the 5 • • LT1072 switching regulator, set up as a fl yback converter.* • • • This confi guration’s 30V output powers A1, a dual op MAXIMUM RATED 4 OUTPUT CURRENT amp. A1A compares the regulators output to the LT1004 FOR 5VOUT reference and servo controls Q1’s gate to close the loop. 3

The gate voltage overdrive allows Q1 to attain its 0.028Ω 2 saturation, permitting the extremely low dropout noted. The OUTPUT VOLTAGE (V) • Zener diode clamps excessive gate-source voltage and the 1 0.001μF capacitor stabilizes the loop. A1B, sensing current • 0 8 9101112 13 14 15 *If boost voltage is already present in the system, signifi cant circuit OUTPUT CURRENT (A) simplifi cation is possible. See LTC Design Note 32, “A Simple Ultra-Low AN32 F12 Dropout Regulator.” Figure 12. Current Limit Characteristics for the Discrete Regulator

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AN32-6 Application Note 32 22μF + 38k* 12k* OUT 10A 5V – + AN32 F13 0.001 1/2 LT1013 18V 1N967A 180Ω P50N05E 1N4148 100k* 1/2 LT1013 + – 10k* 0.01Ω 10k* 100k* 42.5k* (TRIM) 12k* ≈5.45V 12V LT1004 1.2V 12k 1000μF 20k + – + L2 MBR1060 335μH 47pF 1/2 LT1018 1M 18V 1N967A 1N4148 100Ω P50N05E Figure 13. Ultralow Dropout Linear Regulator with Pre-Regulator VN2222 47μF 100k + ≈30V 22μF 330Ω + 10μF L1 820μH 1N4148 12V + IN 1N4148 10k 10k 12V 12V 270k 1/2 LT1018 + – 2k 5.1k (5.9V TO 20V) L1 = PULSE ENGINEERING, INC. #PE-52630 L2 = PULSE ENGINEERING, INC. #PE-51518 * = 1% FILM RESISTOR 0.01Ω = DALE RH-10 0.01

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AN32-7 Application Note 32

Ultrahigh Effi ciency Linear Regulator 100 VIN = 12V 90 Figure 13 combines the preceding discrete circuits to • • 80 • • • • achieve highly effi cient linear regulation at high power. 70 This circuit combines Figure 8a’s pre-regulator with 60 Figure 11’s discrete low dropout design. Modifi cations 50 include deletion of the linear regulators boost supply and 40 EFFICIENCY (%) slight adjustment of the gate-source Zener diode values. 30 Similarly, a single 1.2V reference serves both pre-regula- 20 tor and linear output regulator. The upward adjustment in 10 0 the Zener clamp values ensures adequate boost voltage 0 1 2 3 4 5 678 910 under low voltage input conditions. The pre-regulator’s OUTPUT CURRENT (A) AN32 F14 feedback set the linear regulators input voltage Figure 14. Effi ciency vs Output Current for Figure 13 just above its 400mV dropout. This circuit is complex, but performance is impressive. Micropower Pre-Regulated Linear Regulator Figure 14 shows effi ciency of 86% at 1A output, decreasing Power linear regulators are not the only types which to 76% at full load. The losses are approximately evenly can benefi t from the above techniques. Figure 15’s pre- distributed between the and the MBR1060 catch regulated micropower linear regulator provides excellent diode. Replacing the catch diode with a synchronously effi ciency and low noise. The pre-regulator is similar to switched FET (see Linear Technology AN29, Figure 32) and Figure 8a. A drop at the pre-regulator’s output (Pin 3 of the trimming the linear regulator input to the lowest possible LT1020 regulator, Trace A, Figure 16) causes the LT1020’s value could improve effi ciency by 3% to 5%. comparator to go high. The 74C04 inverter chain ,

100mH DALE TE-5Q4-TA +V IRFD9120 5.2V 3 2 V V 5V 6V TO 10V IN OUT OUT LT1020 S D + 2.5V + 1N5817 220μF 680pF0.001μF 10μF 1M* REF GND FB 1M* 4911

825k* 1M 909k* 200k 200k

PRE-REG

OUTPUT TRIM – TRIM 220k 8 74C04 6 LT1020

COMP + 7

270pF * = 1% METAL FILM RESISTOR UNUSED 74C04 INPUTS

HP5082-2810

AN32 F15

Figure 15. Micropower Pre-Regulated Linear Regulator

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AN32-8 Application Note 32

100 OUT OF 90 VDIFF LT1020 = 0.2V REGULATION 80 VDIFF LT1020 = 0.5V 70 60 A = 50mV/DIV 50 (AC-COUPLED) 40

B = 10V/DIV EFFICIENCY (%) 30 20 C = 10V/DIV 10 0 D = 100mA/DIV 0 5 10 15 20 25 30 3540 45 50 OUTPUT CURRENT (mA) HORIZ = 500μs/DIV AN32 F16 AN32 F17 Figure 16. Figure 15’s Waveforms Figure 17. Effi ciency vs Output Current for Figure 15 biasing the P-channel MOSFET ’s grid (Trace B). The circuit’s low 40μA quiescent current is due to the low The MOSFET comes on (Trace C), delivering current to LT1020 drain and the MOS elements. Figure 17 plots ef- the (Trace D). When the voltage at the inductor- fi ciency versus output current for two LT1020 input-output 220μF junction goes high enough (Trace A), the compara- differential voltages. Effi ciency exceeding 80% is possible, tor switches high, turning off MOSFET current fl ow. This with outputs to 50mA available. loop regulates the LT1020’s input pin at a value set by the resistor divider in the comparator’s negative input and the References LT1020’s 2.5V reference. The 680pF capacitor stabilizes 1. Lambda , Model LK-343A-FM Manual the loop and the 1N5817 is the catch diode. The 270pF capacitor aids comparator switching and the 2810 diode 2. Grafham, D.R., “Using Low Current SCRs,” General prevents negative overdrives. Electric AN200.19. Jan. 1967 The low dropout LT1020 linear regulator smooths the 3. Williams, J., “Performance Enhancement Techniques switched output. Output voltage is set with the feedback for Three-Terminal Regulators,” Linear Technology pin associated divider. A potential problem with this circuit Corporation. AN2 is start-up. The pre-regulator supplies the LT1020’s input 4. Williams, J., “Micropower Circuits for Signal Condition- but relies on the LT1020’s internal comparator to function. ing,” Linear Technology Corporation. AN23 Because of this, the circuit needs a start-up mechanism. The 74C04 inverters serve this function. When power is 5. Williams, J. and Huffman, B., “Some Thoughts on DC-DC applied, the LT1020 sees no input, but the inverters do. The Converters,” Linear Technology Corporation. AN29 200k path lifts the fi rst inverter high, causing the chain to 6. Analog Devices, Inc, “Multiplier Application Guide” switch, biasing the MOSFET and starting the circuit. The inverter’s rail-to-rail swing also provides good MOSFET grid drive.

Note: This application note was derived from a manuscript originally prepared for publication in EDN magazine.

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AN32-9 Application Note 32

APPENDIX A

Achieving Low Dropout Linear regulators almost always use Figure A1a’s basic on-resistance varies considerably under these conditions, regulating loop. Dropout limitations are set by the pass although bipolar losses are more predictable. Note that elements on-impedance limits. The ideal pass element voltage losses in driver stages (Darlington, etc.) add di- has zero impedance capability between input and output rectly to the dropout voltage. The follower output used in and consumes no drive energy. conventional three terminal IC regulators combines with drive stage losses to set dropout at 3V.

PASS INPUT OUTPUT Common emitter/source is another pass element option. ELEMENT REGULATING This confi guration removes the Vbe loss in the bipolar LOOP DRIVE case. The PNP version is easily fully saturated, even in IC

– form. The trade-off is that the base current never arrives at the load, wasting substantial power. At higher cur-

+ AN32 FA1a rents, base drive losses can negate a common emitter’s VREF saturation advantage. This is particularly the case in IC designs, where high beta, high current PNP Figure A1a. Basic Regulating Loop are not practical. As in the follower example, Darlington connections exacerbate the problem. At moderate currents Common FollowersEmitter/Source Compound PNP common emitters are practical for IC construction. VIN VOUT VIN VOUT VIN VOUT The LT1020/LT1120 uses this approach. Common source connected P-channel MOSFETs are VIN VOUT VIN VOUT also candidates. They do not suffer the drive losses of bipolars, but typically require 10V of gate-channel bias

VIN to fully saturate. In low voltage applications this usually VOUT V requires generation of negative potentials. Additionally, IN V OUT P-channel devices have poorer saturation than equivalent size N-channel devices. –V AN32 F1b Figure A1b. Linear Regulator with Some Pass Element The voltage gain of common emitter and source confi gura- Candidates tions is a loop stability concern, but is manageable. Compound connections using a PNP driven NPN are a A number of design and technology options offer various reasonable compromise, particularly for high power (be- trade-offs and advantages. Figure A1b lists some pass yond 250mA) IC construction. The trade-off between the element candidates. Followers offer current gain, ease of PNP Vce saturation term and reduced drive losses over a loop compensation (voltage gain is below unity) and the straight PNP is favorable. Also, the major current fl ow is drive current ends up going to the load. Unfortunately, through a power NPN, easily realized in monolithic form. saturating a follower requires voltage overdriving the This connection has voltage gain, necessitating attention input (e.g., base, gate). Since drive is usually derived to loop frequency compensation. The LT1083-6 regulators directly from V this is diffi cult. Practical circuits must IN use this pass scheme with an output capacitor providing either generate the overdrive or obtain it elsewhere. This compensation. is not easily done in IC power regulators, but is realiz- able in discrete circuits (e.g., Figure 11). Without voltage Readers are invited to submit results obtained with our overdrive the saturation loss is set by Vbe in the bipolar emeritus thermionic friends, shown out of respectful case and channel on-resistance for MOS. MOS channel courtesy.

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AN32-10 Application Note 32

APPENDIX B

A Low Dropout Regulator Family The LT1083-6 series regulators detailed in Figure B1 In contrast, the LT1020/LT1120 series is optimized for feature maximum dropout below 1.5V. Output currents lower power applications. Dropout voltage is about 0.05V range from 1.5A to 7.5A. The curves show dropout is at 100μA, rising to only 400mV at 100mA output. Quiescent signifi cantly lower at junction temperatures above 25°C. current is 40μA. The NPN pass based devices require only 10mA load current for operation, eliminating the large base drive loss characteristic of PNP approaches (see Appendix A for discussion).

LT1083 Dropout Voltage vs Output Current LT1084 Dropout Voltage vs Output Current LT1085 Dropout Voltage vs Output Current

2 2 2 • INDICATES GUARANTEED TEST POINT • INDICATES GUARANTEED TEST POINT • INDICATES GUARANTEED TEST POINT

0°C ≤ TJ ≤ 125°C • 0°C ≤ TJ ≤ 125°C • 0°C ≤ TJ ≤ 125°C • –55°C ≤ TJ ≤ 150°C • –55°C ≤ T ≤ 150°C • • –55°C ≤ TJ ≤ 150°C • • J • • • • • • • 1 1 1 TJ = –55°C TJ = 150°C TJ = 150°C T = –55°C T = –55°C TJ = 25°C J J TJ = 150°C TJ = 25°C TJ = 25°C

MINIMUM INPUT/OUTPUT DIFFERENTIAL (V) 0 MINIMUM INPUT/OUTPUT DIFFERENTIAL (V) 0 MINIMUM INPUT/OUTPUT DIFFERENTIAL (V) 0 0 1 2 3 4 5 678 910 0 1 2 3 4 5 6 0 1 2 3 4 OUTPUT CURRENT (A) OUTPUT CURRENT (A) OUTPUT CURRENT (A)

AN32 FB1a AN32 FB1b AN32 FB1c

LT1086 Dropout Voltage vs Output Current LT1020/LT1120 Dropout Voltage and Supply Current

2 1.00 10 • INDICATES GUARANTEED TEST POINT SUPPLY CURRENT (mA) 0°C ≤ TJ ≤ 125°C • –55°C ≤ TJ ≤ 150°C • • 1 0.10 1 TJ = 150°C TJ = –55°C

TJ = 25°C (V) DROPOUT VOLTAGE

MINIMUM INPUT/OUTPUT DIFFERENTIAL (V) 0 0.01 0.1 0 0.5 1.0 1.5 0.1 1 10100 1000 OUTPUT CURRENT (A) OUTPUT CURRENT (mA)

AN32 FB1d AN32 FB1e

Figure B1. Characteristics of Low Dropout IC Regulators

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Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa- tion that the interconnection of its circuits as described herein will not infringe on existing patent rights. AN32-11 Application Note 32

APPENDIX C overloads. Load voltage is derived from the 100k-4k divider. The shunts low value minimizes voltage burden error. Measuring Power Consumption The voltage and current signals are multiplied by a 4-quad- Accurately determining power consumption often neces- rant analog multiplier (AD534) to produce the power prod- sitates measurement. This is particularly so in AC line uct. All of this circuitry fl oats at AC line potential, making driven circuits, where transformer uncertainties or lack of direct monitoring of the multipliers output potentially lethal. manufacturer’s data precludes meaningful estimates. One Providing a safe, usable output requires a galvanically way to measure AC line originated input power (Watts) is isolated way to measure the multiplier output. The 286J a true, real time computation of E-I product. Figure C1’s isolation amplifi er does this, and may be considered as circuit does this and provides a safe, usable output. a unity-gain amplifi er with inputs fully isolated from its BEFORE PROCEEDING ANY FURTHER, THE READER output. The 286J also supplies the fl oating ±15V power IS WARNED THAT CAUTION MUST BE USED IN THE required for A1 and the AD534. The 286J’s output is re- CONSTRUCTION, TESTING AND USE OF THIS CIRCUIT. ferred to circuit common ( ). The 281 oscillator/driver is HIGH VOLTAGE, AC LINE-CONNECTED POTENTIALS necessary to operate the 286J (see Analog Devices data ARE PRESENT IN THIS CIRCUIT. EXTREME CAUTION sheet for details). The LT1012 and associated components MUST BE USED IN WORKING WITH AND MAKING CON- provide a fi ltered and scaled output. A1B’s gain switching NECTIONS TO THIS CIRCUIT. REPEAT: THIS CIRCUIT provides decade ranging from 20W to 2000W full scale. CONTAINS DANGEROUS, AC LINE-CONNECTED HIGH The signal path’s bandwidth permits accurate results, even VOLTAGE POTENTIALS. USE CAUTION. for nonlinear or discontinuous loads (e.g. SCR choppers). To calibrate this circuit install a known full-scale load, set The AC load to be measured is plugged into the test socket. A1B to the appropriate range, and adjust the trimpot for Current is measured across the 0.01Ω shunt by A1A with a correct reading. Typical accuracy is 1%. additional gain and scaling provided by A1B. The and protect the shunt and amplifi er against severe

DANGER! LETHAL POTENTIALS 15V PRESENT IN SCREENED AREA! ISO 15V DO NOT CONNECT GROUNDED TEST EQUIPMENT—SEE TEXT FLOATING –15V COMMON ISO 281 OSCILLATOR/DRIVER A = 1 Y AD534 Z + 15V X 286J + Z = X • Y – 56.2k WATTS OUT 10 LT1012 1M 0V TO 2V 100k – 0.47 150k –15V

20A 100k

10k 4k 2kW 665k 100k 200W TEST LOAD 1M

SOCKET 20W 115VAC LINE 10k – ALL RESISTORS = 1% FILM RESISTORS

* = DALE RH-25 1N1183 1N916 A1A 15.8k –

= FLOATING COMMON—AC LINE COMMON

s2 s2 1/2 LT1057 0.01Ω* + A1B

1/2 LT1057 = CIRCUIT COMMON +

USE A LINE ISOLATED ±15V SUPPLY TO POWER COMPONENTS OUTSIDE SCREENED AREA. DO NOT TIE AND POINTS TOGETHER AN32 FC1

Figure C1. AC Wattmeter DANGER! Lethal Potentials Present—See Text

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