C-Band Microwave Oscillator

DEPARTMENT OF TECHNOLOGY AND BUILT ENVIRONMENT

C-BAND MICROWAVE OSCILLATOR

Maaz Rasheed

September 2008

Master's Thesis in Electronics/Telecommunications

Master's Program in Electronics/Telecommunications Supervisor: Steffen Kirknes Examiner: Olof Bengtsson

Abstract

The work focuses on the study, design and implementation of a C band Microwave oscillator using coaxial resonators, for the transceiver used in wave radar. It involves a literature study discussing different aspects of microwave oscillators, mainly the shielding of the oscillators, frequency pulling due to load and supply pulling, tuning range and the temperature performance of the oscillator. The study of the shielding resulted in proposing a high quality metallic shield with high elastic modulus, high strength and high density, as the wave radar will be a stationary, standalone system and the weight of the shield is not a limiting factor. The metallic shield provides better EMI and EMP performance than the carbon ferrites. The characterization of the resonator is critical as a small mistake pulled the frequency about 300 MHz. This can be achieved by careful design and measuring the resonator test circuits for one port. The tuning range of the oscillator is important as the temperature, bias, and load mismatches can increase or decrease the frequency of the oscillator. The varactor in combination with a capacitor increases the tuning range to about 10 times. The high reverse isolation of 47 dB is achieved by a passive attenuator and a buffer amplifier. The temperature performance is also important and there was a 30 MHz variation in frequency from 0 − 60표 퐶 , and the output power was between 3-4 dBm. The Load puling was 1 MHz with a 12 dB return loss test setup for a phase change of 0 − 180표. The phase noise was −98 푑퐵푐/퐻푧 at 100 푘퐻푧 offset. Overall the coaxial resonator oscillator proves to be a very good stable oscillator suitable for aerospace and ground based industry.

Acknowledgements

This Master’s thesis in Electronics/Telecommunications was conducted at Norbit AS, Trondheim, Norway, which represents one of the largest RF and microwave competence centre, in Nordic countries. The duration of this project is six months from 1st March to 31st August 2008.

First of all, I would like to thank my Supervisor, Steffen Kirknes for his guidance, encouragement and support during this project. I would also appreciate all my colleagues at Norbit for their continuous help and encouragement. Apart from guidance, I was also given access to all the equipment and facilities at Norbit AS that were essential for my thesis.

Special thanks go to the staff of ITB/Electronics, University of Gavle, Sweden with my gratitude to Prof. Claes Beckman, Prof. Niclas Björsell, Olof Bengtsson, Per Ängskog, Magnus Isaksson and Prof. Edvard Nordlander for their support during the period of studies. I would like to appreciate all my good and kind friends in Gavle, Sweden.

Finally I would like to express my gratitude to my parents, who have financed, encouraged and helped me in my studies. Without their support, it was impossible for me to come to Sweden and study this prestigious Master’s degree program.

To my motherland, Pakistan!

To my Parents!

To my Wife!

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Table of Contents

Page No. Chapter 1 Introduction………………………………………………. 1

1.1 Introduction……………...... 1 1.2 Susceptibility to Electromagnetic Fields and Shielding………...... 2 1.3 Tuning Range……………………………………………………… 5 1.4 Temperature Stability……………………………………………… 5 1.5 Frequency Pulling due to Load…………………………………….. 7 1.6 Frequency Pulling due to Supply…………………………………... 8 1.7 Thesis Outline……………………………………………………… 10

Chapter 2 Theory ……………………………………………………. 11

2.1 Theory of Oscillators…………………………………………….. 11 2.2 Microwave Oscillator Configurations…………………………… 11 2.3 Voltage Controlled Oscillators………………………………….. 12 2.4 Considerations for Oscillators…………………………………… 12 2.4.1 Phase Noise…………………………………………………… 12 2.4.2 Harmonic Suppression……………………………………..… 12 2.4.3 Spurious signals in the Oscillator Output……………………. 13 2.4.4 Post-tuning Drift……………………………………………... 13 2.4.5 Blocking or Reciprocal Mixing……………………………… 13 2.4.6 Linear Tuning characteristics………………………………... 14 2.5 Choice of the Active Device and Modeling …………………….. 14 2.6 Loaded Q…………………………………………………………. 15

Chapter 3 Method…………………………………………………….. 16

3.1 Passive Attenuator Design……………………………………….. 16 3.1.1 T- Attenuator…………………………………………………. 17 3.1.2 Pi Attenuator…………………………………………………. 18 3.2 Resistive Elements Models ……………………………………… 18 3.3 Buffer Amplifier………………………………………………….. 19

3.4 Resonator Network……………………………………………………. 20 3.4.1 Modeling the Resonator…………………………………………… 24 3.4.2 Coaxial Resonator Tuning………………………………………… 26 3.5 Varactor Diode Tuning………………………………………………… 26 3.5.1 Mathematical model of a Varactor………………………………… 28 3.5.2 Tuning Ratio of the Varicap……………………………………….. 29 3.5.3 Varactor in VCOs…………………………………………………… 29 3.6 Directional Coupler…………………………………………………….. 29 3.7 Bias Networks and Tees………………………………………………... 30 3.8 Shielding ………………………………………………………………. 31

Chapter 5 Process and Results…………………………………………….. 32

5.1 VCO’s Transistor Measurements……………………………………… 32 5.2 Attenuator Results……………………………………………………... 32 5.3 Buffer Amplifier…………………………………………………..…… 34 5.4 Power Supply Bias…………………………………………………….. 36 5.5 Coupler Design………………………………………………………… 38 5.6 Voltage Controlled Oscillator (VCO) Circuit………………………….. 41 5.7 Fabrications and Measurements ……………………………………….. 44

Chapter 6 Discussions & Conclusions……………………………………… 48 Chapter 7 Future Work……………………………………………………... 54 References……………………………………………………………………. 55 Appendices……………………………………………………………………. 57

List of Figures Page No Fig. 1.1: Far Field shielding as a function of frequency for different materials…. 4 Fig. 1.2: DRO response with temperature control …………………………….. 6 Fig. 1.3: Temperature Performance of a 9 GHz DRO…………………………. 7 Fig. 1.4 The transistor oscillator ……………………………………………… 8 Fig. 1.5: Low Frequency current noise at 100 Hz and oscillator’s phase noise at offset of 100Hz …………………………………………….. 9 Fig. 2.1: Reciprocal Mixing …………………………………………………… 13 Fig. 3.1: Tee attenuator …………………………………………………..…… 17 Fig. 3.2: Pi attenuator…………………………………………………………. 18 Fig. 3.3: Equivalent Circuit of a Resistor at high frequencies……………. …. 19 Fig. 3.4: Typical coaxial ceramic Resonator…………………………………. 21 Fig. 3.5: Equivalent circuit of a Coaxial Resonator………………………… .. 22 Fig. 3.6: Self Resonant Frequency of Coaxial Resonator………………… …. 23 Fig. 3.7: Parallel RLC Model for the Resonator……………………………… 24 Fig. 3.8: Impedance Response of the resonator model……………………….. 25 Fig. 3.9: Equivalent Model of the coaxial resonator with pads………………. 25 Fig. 3.10: Impedance vs. Frequency response for the model with pads……… 26 Fig. 3.11: Varactor Diode Equivalent Model…………………………………. 27 Fig. 3.12: Junction Capacitance as a function of Bias Voltage + Contact Potential 27 Fig. 3.13: Equivalent Model for a varactor…………………………………… 28 Fig. 3.14: Coupled lines Directional coupler…………………………………. 30 Fig. 4.1: VCO’s transistor measurements…………………………………….. 32 Fig. 4.2: Schematics of the 10 dB Pi Attenuator……………………………… 33 Fig. 4.3: S parameter simulations of the 10 dB attenuator……………………. 33 Fig. 4.4: Input and Output Impedance of the Attenuator……………………… 34 Fig. 4.5: Buffer Amplifier Schematics with 3V, 21mA bias point……………. 35 Fig. 4.6: Response of the Buffer amplifier……………………………………. 35 Fig. 4.7: Input and output impedance match of the buffer amplifier………….. 36 Fig. 4.8: Schematics of a bias Tee…………………………………………….. 37 Fig. 4.9: Layout of the bias Tee……………………………………………….. 37 Fig. 4.10: Simulations results of the Bias Tee………………………………… 38 Fig. 4.11: Schematics of 12 dB Coupler………………………………. …….. 39

Fig. 4.12: Response of the 12 dB Coupler……………………………………. 40 Fig. 4.13: Schematics of the Voltage Controlled Oscillator …………………. 41 Fig. 4.14: Layout of the Voltage Controlled Oscillator………………………. 41 Fig. 4.15: OscTest Response of the Oscillator………………………………... 42 Fig. 4.16: PCB of the VCO…………………………………………...... 44 Fig. 4.17: VCO measurements through Spectrum Analyzer…………………. 45 Fig.4.18: Load Pull measurements …………………………………………... 47 Fig. 5.1: Schematics for Matching Network of the Buffer Amplifier………... 49 Fig. 5.2: Results of the Matching Network of Buffer Amplifier……………... 50 Fig. 5.3: Recommended pad for the Resonator………………………………. 50 Fig. 5.4: Resonator with Pads, One Port Test circuit………………………… 51 Fig. 5.5: One Port Measurements of the Resonator………………………….. 51

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List of Tables Page No

Table 1.1: Phase Locked DRO/CRO Performance ………………………………… 2 Table 1.2: Properties of Shielding Materials …………………………………….. 3 Table 2.1: Center Frequency, Tuning range (absolute and relative) for different standards ……………………………………………….. 14 Table 4.1: Oscillation Frequency of VCO as a function of varactor bias voltage … 43 Table 4.2: Results of the VCO for the Varactor tuning………………………….. 45 Table 4.3: Results of the VCO over a Temperature Range……………………… 46 Table 5.1: Tuning range results with varying capacitor values………………….. 48

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Chapter 1 Introduction

1.1 Introduction

This Masters Degree Project is a study, design and implementation of a C band microwave oscillator for the C band transceiver, used in wave radar. The most important parameters affecting the microwave oscillator design are studied which include the tuning range, shielding of the oscillator circuits, temperature performance and frequency instability due to load and supply pulling.

A state of the art phase locked DRO/CRO is described, mainly used in space applications [1]. Previously the implementation of oscillators was realized using crystal oscillator followed by a chain of multipliers for good stability and phase noise. The new approach is to use a Dielectric Resonator Oscillator (DRO) or a Coaxial Resonator Oscillator (CRO). This approach provides very good long term stability, better temperature performance and reduced sizes. The DRO or CRO is phase locked to a temperature compensated crystal oscillator (TCXO). The main advantage of the phase locked loop is that the crystal oscillator’s long term stability is inherited to the DRO or CRO. The offset frequencies within the loop have the same phase noise and bandwidth as that of the crystal oscillator. This eliminates the use of filters for removing spurious harmonics. The resonators used, are made of dielectric materials. These have a very high unloaded Q at microwave frequencies and are insensitive to radiations which make them ideal for aerospace applications. The tuning of these resonators is very simple as well. Overall the entire phase locked loop DROs and CROs have shown good spurious and phase noise performance and are less complicated. Table 1 shows some of the characteristic results of the B. Hitch and T. Holden’s DRO and CRO implementations.

푷풆풓풇풐풓풎풂풏풄풆 푷풂풓풂풎풆풕풆풓 푷풉풂풔풆 푳풐풄풌풆풅 푪푹푶 푷풉풂풔풆 푳풐풄풌풆풅 푫푹푶 퐹푟푒푞푢푒푛푐푦 푆푡푎푏푖푙푖푡푦 ± 1.5 푝푝푚 푎푙푙 푐푎푢푠푒푠 ± 1.5 푝푝푚 푎푙푙 푐푎푢푠푒푠 푃푕푎푠푒 푁표푖푠푒 10 퐻푧 −63 푑퐵푐/퐻푧 −63 푑퐵푐/퐻푧 100 퐻푧 −94 푑퐵푐/퐻푧 −94 푑퐵푐/퐻푧 1 퐾퐻푧 −113 푑퐵푐/퐻푧 −113 푑퐵푐/퐻푧 10 퐾퐻푧 −117 푑퐵푐/퐻푧 −117 푑퐵푐/퐻푧 100 퐾퐻푧 −120 푑퐵푐/퐻푧 −128 푑퐵푐/퐻푧

Table 1.1: Phase Locked DRO and CRO performance [1]

The stability of signal source depends on maintaining the phase locked loop conditions. The bandwidth requirements may force to use multiple narrowband ceramic CROs. The spectral purity of these ceramic CROs is very good but they have several disadvantages. These include limited temperature and tuning ranges. Another disadvantage is that they are not suitable for integrated circuits (ICs) fabrication at the present. Further more the ceramic resonators are sensitive to phase hits due to tension in the crystal structure [2].

1.2 Susceptibility to Electromagnetic Fields and Shielding

Electronic circuits may be classified as shielding circuits, non-shielding circuits and semi-shielding circuits each having different requirements for shielding in a multilayer PCB. The three-dimensional configuration of the shielding circuits plays a vital role in achieving a stable shielded structure [3]. The shielding may be electric, magnetic or electromagnetic in nature.

The electromagnetic shield takes the form of an enclosure which consists of metal plates and foils usually. The electromagnetic shield construction on the actual printed board also determines its effectiveness. The two factors which increase the efficiency of the electromagnetic shields are the electrical contact in the shield layer and making the path longer for the leakage of the electromagnetic fields. This increases the efficiency of the contact joints in electromagnetic shields [4].

The aerospace structures require a high degree of electromagnetic interference (EMI) shielding, compared to other electronics structures. The weight of these shields is a limiting factor, thus a high strength, high elastic modulus and light weight shields are required with better EMI shielding abilities. Metallic shields are used currently which have higher densities.

푴풂풕풆풓풊풂풍 푫풆풏풔풊풕풚 푹풆풔풊풔풕풊풗풊풕풚 푺풕풓풆풏품풕풉 푴풐풅풖풍풖풔 ퟑ 품/풄풎 흁훀 풄풎 푴푷풂 푮푷풂 Copper 8.96 1.78 420 110 Iron 7.86 10 200 200 Aluminum Alloy 2.80 10 520 71 Aluminum 2.70 2.82 210 60 Beryllium 1.85 4.0 620 290 P-100 + Br/epoxy 1.78 90 840 430 P-100/epoxy 1.72 460 840 430 T-300/epoxy 1.51 5000 3200 228

Table 1.2: Properties of shielding materials [5]

Table 1.2 describes some of the properties of the metallic and composite shielding materials. The most commonly used and most effective shielding material is aluminum, but it has a very high density. Carbon fibers are light weight, but not having enough conductivity, which results in insufficient shielding. Paints, plating and foils on the carbon composites may provide needed shielding, but there are issues with reliability, scratching, low adhesion and they may oxidize in air.

The Fig. 1.1 describes the total far field shielding at different frequencies for different materials. At low frequencies the attenuation is almost frequency independent, until up to some characteristic value after which it increases sharply. The metals Cu and Al provide the best shielding as the attenuation rises sharply after 105 Hz. The other composites have low shielding at high frequencies, PAN having the least.

Fig. 1.1: Far Field shielding as a function of frequency for different materials [5]

The Al/Cu foil is very effective EMI shield. It is low cost and may be used with high conductive adhesive. The EMI conductive coatings or adhesives on light weight plastics is also becoming in use. These conductive paints are of many types such as silver-copper paints, copper, nickel or silver paints etc [6].

The phase cancellation principle works in uniform magnetic materials at front face providing up to 40 dB absorption per ounce. For 50-800 MHz frequency range, spinal ferrites are used while for 800 MHz-2 GHz frequencies, ferroxplana is common. On the contrary thick EMI shielding materials with a uniform dielectric does not follow the phase cancellation principle due to their lossy conductor fibers where the absorption is equivalent to the power loss. These have low densities and permittivities. Comparing with ferrites, metals have higher permittivities at microwave frequencies and are better EMI absorbers. The losses of the magnetic absorbers are dependant on their magnetic fields so a non-conducting material is desired. Dielectric absorbers require thick layers, where the relative permittivity is desired close to one. Practically these absorbers have higher permittivities, so are not appropriate in many cases [7].

Another problem in circuits is the low frequency Electromagnetic Pulse (EMP). Filtering this EMP is usually achieved through proper isolation which is provided by highly permeable ferrite shields. But in cases where there are large distances between the wave impedance of the shield and the intrinsic impedance of the metal, a higher reflection occurs for the impinging wave. Thus copper and aluminum are preferred choices compared with ferrites in such conditions. It can be concluded that with low frequencies and large separation between source of the signal and the shields, higher conducting materials provides better shielding abilities [8].

1.3 Tuning Range

The tuning network plays a key role in defining the frequency range of an oscillator. The tuning network consists mainly of the resonator whose type determines the frequency of oscillation. The voltage controlled oscillator may have frequency output at some specified frequency range. In general, higher the center frequency of the oscillator, the more difficult the design is. Oscillators are analyzed usually over a frequency range around the center frequency.

The MMIC varactor diode has a specific tuning range provided by the manufacturer. The introduction of negative resistance connected to the varactor diode using an active circuit increases the varactor’s tuning range more than 10 times. In wide band 푓 VCOs, the frequency range relation 푚푎푥 is proportional to the tuning range of the 푓푚푖푛 퐶 varactor diode 푚푎푥 . The negative capacitance can be created by two common 퐶푚푖푛 source transistors loaded with inductor [9].

1.4 Temperature Performance

The temperature variations affect the performance of the active devices and oscillator as a whole. For oscillators, a working temperature range is defined and the change in the output power of the oscillator is specified in that temperature range. The simulations usually imply that the junction temperature is constant at 25표퐶. For this oscillator design, a range of 0표 − 60표 C is specified.

The temperature variations also affect the operation of the oscillators causing a change in the output power and frequency. The temperature stability is improved by having a temperature controlled resonator in a DRO based on simple loop design. The dynamic temperature sensitivity depends on the ambient temperature variations. The copper may expand and contract relative to ambient temperature quicker than the dielectric puck which has a poor thermal conductivity. The static temperature conductivity describes the equilibrium between the initial and final ambient temperatures and is a function of temperature coefficients, thermal expansion of the dielectric material and cavity. A temperature control circuit is used consisting of a resistive heater mat, and NTC thermister to sense the temperature [10]. The Fig. 1.2 describes the temperature sensitivity of the 1.3 GHz DRO. The temperature control enabled is represented by the red line and show a stable frequency offset with temperature while the temperature disabled control show large variations in frequency offset.

Fig. 1.2: DRO response with temperature control [10]

Fig. 1.3 shows the temperature response of a 9 GHz DRO, with a CW output power of 2.5 watts at room temperature using high power GaAs MESFET. The frequency stability is 130 푝푝푚 without any temperature compensation from −50표퐶 푡표 + +50표 퐶. The variation in output power at −50표 퐶 is +35 푑퐵푚 (3.2 푊) and at +50표 퐶 was 33 푑퐵푚 (2 푊) which describes output power as a function of the temperature [11].

Fig. 1.3: Temperature performance of a 9 GHz DRO [11]

1.5 Frequency Pulling due to Load

The load variations affect the frequency of the voltage controlled oscillator (VCO). The variation in the output impedance affects the DC voltages of the VCO’s transistor. The BJT (Bipolar Junction Transistor) has supply voltages for different junctions that must be constant throughout the operation of the oscillator. A change in the Base-Collector voltage (푉퐶퐵) is caused by the change in the VCO’s output Base- Collector capacitance. This varies the frequency of oscillation to a large value. Load pulling is minimized by using a very high isolation stage between the load and VCO. Frequency Pulling is a measure of the shift in frequency for a unity VSWR load (usually 50 ohms) to the non-unity VSWR due to change in the load.

The load pulling measurements are done using variable transmission line and a load impedance which is not matched. The VCO and load are connected together. The transmission line is used for varying the phase angle between the load and VCO between 0표 − 360표. In this way the frequency variations due to the load variations are measured.

A very linear MIC bipolar VCO with 100 MHz FM rate is described where the frequency pulling is ± 1 푀퐻푧 into a 2:1 mismatched load. The power flatness and immunity from load pulling is achieved by using a two stage GaAs FET buffer amplifier. The total load isolation achieved is more than 45 dB. The FET amplifier provides 35 dB of isolation while 12 dB of isolation is provided by a thin film pi attenuator [12].

1.6 Frequency Pulling due to Supply

The frequency of the VCO is affected by the supply voltage to some extent. In transceiver systems, power amplifier uses a lot of the power and turning it on, will increase the current drastically, which may affect the LO power supply and the VCO frequency will increase. This can be minimized by isolating the VCO unit from the power amplifier unit. The battery supply voltage drops to a certain minimum during its life time and the oscillator design should work in a certain range of voltage. This may be indicated in terms of tolerance as well for example ±10 % of 3 푉 power supply. Temperature also affects the supply voltage bias. Noise on the supply voltage lines causes noise and spurious on the oscillator.

The external environmental conditions like temperature or supply voltage variations affect the active elements of the microwave oscillators thus inducing frequency instability. The two reasons for this instability are variations in the cut off frequency and collector capacitance. For the analysis, a simple tuned circuit is chosen as a passive frequency determining network while a common base configuration is selected for the active element as seen in Fig. 1.4.

Fig. 1.4 The transistor oscillator [13]

The high frequency analysis shows that the two most important parameters of the reactive elements of the transistor are the collector capacitance and the frequency dependence of the current gain. Variations in these factors give rise to variations in the oscillation frequency. The collector capacitance is mainly a function of the collector voltage. The relation between the changes in collector capacitance 퐶푐 and

Collector voltage 푉푐 for a typical p-n-p transistor is given in eq. 1.1. 휕 퐶 1 휕 푉 푐 = − 푐 (1.1) 퐶푐 2 푉푐 The effective base width of the transistor depends on the collector voltage and hence frequency of oscillation. The Ebers and Miller [14] equation for the relation between the 푓푎 and 푉푐 is approximated as

푓푎 = 1 + 퐾푉퐶 푓푎표 (1.2) 푓푎표 where

푓푎표 = 푇푕푒 푣푎푙푢푒 표푓 푓푎 푓표푟 푧푒푟표 푉푐

The low frequency (LF) noise up conversion in HBT transistors significantly contributes to the close in carrier phase noise of the transistor based microwave oscillators [15]. High Q resonator networks are used with HBT transistors having LF noise, to reduce close in carrier phase noise. The noise up conversion factor is adjusted to reduce further this close in carrier noise. The investigations showed that there was a 15 dB reduction in phase noise by maintaining the bias point where the small transistor phase sensitivity is observed to the transistor bias current.

Fig. 1.5: Low Frequency current noise at 100 Hz and oscillator’s phase noise at offset of 100 Hz [15].

The Fig. 1.5 shows the measured and predicted results and are calculated based on curve fitted LF noise. The experimental results predicts that the residual phase noise of the HBT based oscillators is a function of the bias dependant LF noise up conversion factor of the device. This concludes that for a low phase noise, the two important decisions are device selection and matching network design.

The variations in supply voltage could cause a change in the output power of the microwave oscillator. The output power can be controlled for microwave oscillators by varying the bias voltage of the active element while using the scheme with a synchronized oscillator. The results from the theoretical and experimental investigations proves that a change of 0.2 V or 2.7 % of bias voltage change may cause an output power change of up to 40 dB, with frequency being the constant factor [16].

1.7 Thesis Outline

Chapter 2 describes the theory of the oscillator design. The different types of oscillator topologies are described. There is a discussion of the oscillator properties, characteristics and limitations.

Chapter 3 describes the design strategies of the VCO and the sub circuits used in it. These sub circuits include the attenuator, buffer amplifier, resonator, and varactor tuning network, coupler etc.

Chapter 4 consists of the simulations, results and fabrication of the VCO. The expected results are analyzed to the measured results.

Chapter 5 describes the conclusions, discussions, week points and probable solutions to these problems. It also includes a comparative study of this project results with the state of the art work done by the others.

Chapter 6 gives an idea of the future work to be done in this design.

Chapter 2 Theory

2.1 Theory of Oscillators

An oscillator is a non linear circuit which transforms DC power to an AC waveform. This circuit usually consists of an amplifier, a resonator and a feedback network. The feedback may be internal i.e. a part of energy from the active device is fed back to the resonator or it may be an external feedback circuit.

The active device in the oscillator takes in DC power from a regulated supply and for an input power gives a specific output power which is several times higher in magnitude. It can be a bipolar junction transistor (BJT), or a field effect transistor (FET) or a Gain block which is usually wideband.

The specifications affecting the quality of operation of the whole system depends on the cleanliness of the oscillator signal i.e. low phase noise and low spurious, which constitute noise in systems. The desired characteristics for oscillators are sufficient output RF power level, low phase noise, efficiency and stability of the signal etc. Several noises contribute to the total noise of the oscillator. These include losses in the resonator, transistor noise, noises modulated in power supply and noise due to the varactor diode tuning.

2.2 Microwave Oscillator Configurations

There are three types of approaches for designing oscillators.

In one port oscillators, the transistor and the feedback network is replaced by a negative resistance. For example, in Colpitt and Clapp oscillators, the capacitive feedback creates a negative resistance along the tuning network.

In two port oscillators, the transistor acts as a two port device with its third terminal grounded. The tuning network is used for the feedback which determines the frequency. Such oscillators have a specific gain, phase shift, resonator network and matching network.

Three port oscillators have an inductor at the base with some capacitance at the output port. This feedback configuration generates negative resistance at input and output port.

2.3 Voltage Controlled Oscillators

Voltage controlled oscillators or simply VCOs are the class of oscillators in which the frequency determining reactance is varied by voltage. For high frequency applications, the voltage controlled element is a typical varactor diode.

The VCOs find its use in many important applications such as function generators, transmitters of every kind, frequency synthesizers and almost every type of wireless communication equipment.

2.4 Considerations in Oscillators

2.4.1 Phase Noise Phase noise is measured as dBc in a bandwidth of 1 Hz at an offset of a specific frequency. There are several types of noises and spurious which modulate the output signal of the VCO on either sides of the carrier.

Phase noise is short term phenomenon and some of the possible causes of phase noise due to improper isolation are

 Variations in the impedance of load  Mismatch reflections back to the VCO’s output  Rise in ground current  Coupling of radiations from nearby layout circuits  Changes in the bias supply of the VCO transistor due to load variations

The loaded Q is mainly responsible for the phase noise performance [17].

2.4.2 Harmonic Suppression The typical harmonic suppression of a voltage controlled oscillator is about 15 dB, but for certain systems, a very low harmonic content is desired. This can be achieved by placing a microstrip low pass filter at the output.

2.4.3 Spurious signals in the Oscillator Output Apart from harmonics, unwanted signals found on the sides of the carrier of an oscillator are called spurious. A spurious free range is usually specified in terms of dB. The synthesizer signals may be a cause for the generation of these spurious.

2.4.4 Post-tuning Drift The tuning network which is a varactor diode is biased with supply voltage. After the supply voltage has been applied, the frequency of the oscillator still drifts for some time. This drift can affect the voltage controlled oscillators tuning speed.

2.4.5 Blocking or Reciprocal Mixing The mixing of the local oscillator noise sidebands with the incoming strong signals is called reciprocal mixing or blocking. This mixing produces unwanted noise at the intermediate frequency. Fig. 2.1 shows the carrier signal A´ of the oscillator mixes with the wanted signal A. The side bands of the oscillator B´, C´, D´ mixes with the undesired signals A, B, C and creates interference in the intermediate frequency IF. This affects considerably the receiver selectivity of increasing the noise floor.

Fig. 2.1 Reciprocal Mixing [18]

2.4.6 Linear Tuning characteristics A linear relationship is desired for the variations in the frequency of the oscillator, caused by varying the tuning voltage. This is an important factor for the stability of synthesizers. Table 2.1 shows some standards for wireless communications. The center frequency and tuning range (absolute and relative) is given for each standard. The absolute tuning range shows the minimum and maximum frequencies while relative tuning describes the percentage of the center frequency. The relative tuning ranges for TV receiver, Satellite TV front end and DVB-T and FM radio front end are very high. The GSM and UMTS relative ranges are narrow band with just around 3%. The SONET standards are for fixed bit rates and only the center frequency is indicated.

Table 2.1 Center Frequency, Tuning range (absolute and relative) for different Standards [19]

The oscillator design requires a tuning range of about 80-100 MHz around the center frequency.

2.5 Choice of the Active Device and Modeling

The two main choices available are FET and bipolar transistors. The transition frequency and flicker noise corner frequency are important while choosing the active device for the oscillator.

The frequency of transition 푓푇 of a transistor plays an important role in defining its frequency of oscillation. The bipolar transistors have a 푓푇 of up to 25 GHz where as

SiGe based transistors have higher values of 푓푇 upto 100 GHz. The improved GaAs based bipolar transistors called Heterojunction bipolar transistors (HBTs) have 푓푇 of about 100 GHz also but it costs much more than silicon based transistors.

The flicker noise frequency of HBTs is higher than the SiGe transistors but it is not prominent in practical circuits which have high lossy transmission media. Thus the total oscillator noise in SiGe transistors and HBTs remains almost the same.

2.6 Loaded Q The Loaded Quality factor of an oscillator is

푇표푡푎푙 푒푛푒푟푔푦 푠푡표푟푒푑 푖푛 푡푕푒 푠푦푠푡푒푚(푖푛 표푛푒 푓푢푙푙 푐푦푐푙푒) 푄 = 2휋 퐿 퐸푛푒푟푔푦 푙표푠푡 푖푛 푡푕푒 푠푦푠푡푒푚 푖푛 푒푎푐푕 푐푦푐푙푒

In steady state, the external source supplies the energy to be lost.

The high loaded Q has certain advantages.

 The high loaded Q reduces the frequency drift as resonator becomes the sole frequency determining component.  The isolation of the resonator from the active device reactance minimizes the effect of temperature.  The long term stability and phase noise performance is improved.

Chapter 3 Method

3.1 Passive Attenuator Design

Attenuator is a circuit which reduces the power of an incoming signal without the significant distortion of the signal waveform and attenuation is expressed in 푑퐵. The main purpose of the attenuator, in combination with a buffer amplifier, is that the proper output signal level is maintained at the VCO’s output and thus valuable reverse isolation is achieved.

The VSWR values are of critical importance in the design of attenuators with resistive elements. The attenuator is also used to improve the input match of the amplifier. This may decrease the VSWR ratio at the input of the buffer amplifier and improve gain. On the other side, every dB of attenuation at the input of the buffer amplifier increases the Noise Figure (NF) of the amplifier.

The desirable characteristics of attenuators are reliability at the frequency of operation and power applied to it. The attenuator design is usually achieved with the help of resistors. The resistors used are usually surface mount. Since the output power from the VCO transistor is few milliwatts, thus it doesn’t affect the device performance a lot due to heating considerations.

Low Standing Wave Ratio (SWR) at input and output of the attenuator is always desired. The low SWR at input and output can be achieved by careful design of the attenuator circuit usually by having a symmetrical network. The SWR of the attenuator and the input and output networks will contribute to the mismatch. This variation is frequency dependant and it may degrade the flatness response of the attenuator. The SWR at the input of attenuator may not be very important as VCO transistor’s output is loaded with the attenuator to maintain the negative resistance region. But at the output, a stable SWR is required to have the required reverse isolation and matching. [20]

There are several topologies for achieving the attenuation in circuits. The passive techniques i.e. Tee and Pi configurations are described here.

3.1.1 Tee Attenuator

Fig. 3.1 shows the Tee attenuator configuration and consists of two resistors of the same value which are the series R1 & R2 resistors.

R 1 R 2

Z in Z out

R 3

Fig. 3.1: Tee attenuator

The resistances R1, R2 and R3 are calculated as

퐿 1 푍퐼푁 푍푂푈푇 10 푅3 = 10 − 1 퐿 (3.1) 2 10 10

1 푅2 = 퐿 (3.2)

10 10 + 1 1 퐿 − 푅3 푍푂푈푇 (10 10 − 1)

1 푅1 = 퐿 (3.3)

10 10 + 1 1 퐿 − 푅3 푍퐼푁 (10 10 − 1) where

L = desired attenuation expressed in dB

풁푰푵 = desired Input impedance expressed in ohms

풁푶푼푻 = desired Output impedance expressed in ohms

3.1.2 Pi Attenuator

Fig. 3.2 shows the Pi attenuator configuration consisting of two resistors of the same value i.e. R1and R2.

R 3 Z in Z out

R 1 R 2

Fig. 3.2: Pi attenuator

The resistances R1, R2 and R3 are calculated as

퐿

2 푍퐼푁 푍푂푈푇 10 10 푅3 = 퐿 (3.4)

10 10 − 1

퐿

10 10 + 1 푅2 = 퐿 푍푂푈푇 − 푅3 (3.5)

10 10 − 1

퐿

10 10 + 1 푅2 = 퐿 푍퐼푁 − 푅3 (3.6)

10 10 − 1

In terms of transducer gain which is equal to the attenuation of the attenuator 2 퐴푡푡푒푛푢푎푡푖표푛 = 푆21

3.2 Resistive Elements Models The attenuator at low frequency can be designed using simple formulas as the resistors act as lumped components. The resisters at high frequency have different behavior at different frequencies and depend on a number of processes like the type of resistors selected, accurate models for those resistors, accurate pads etc.

The factors affecting the modeling of the resistors at high frequencies are [21]

 Dimensions of the resistor (Length, width, height etc.)  Skin Effect  Ground plan layout  The electrical length of the resistor relative to quarter wavelength (approximately 1/10)

Components at high frequencies behave as non-ideal. For example the capacitor leads have significant inductance while inductors have some self-capacitance. The Q of the inductors is also considerable along with the parasitics due to coupling between inductors. These parasitics should be considered in design and simulations. There are parasitics associated with amplifiers as well. Thus the real world resistor acts as a distributed component at high frequencies. Proper modeling of the resistors and other components make them behave as lumped even at very high frequencies.

Fig. 3.3 shows the equivalent circuit of a resistor at high frequency, with a parasitic inductance 퐿푆 in series and parallel capacitance 퐶푃. The parasitic reactance is dependant upon the dimensions and mounting techniques of the resistors. The inductance is usually just a few 푛퐻 while the capacitance is a fraction of a pF.

Fig. 3.3 Equivalent Circuit of a Resistor at high frequencies

3.3 Buffer Amplifier

The component selected for providing the reverse isolation and amplification, in the oscillator, is a low noise amplifier. It is Avago’s MGA-665P8 GaAs MMIC. It has a unique active power down function. The features of this gain block are high gain, low noise Figure and very high reverse isolation. It is incorporated in LPCC package suitable for surface mounting. This improved performance is based on the Avago’s

19 state of the art E-HEMT (Enhancement Mode Pseudomorphic High Electron Mobility Transistor).

There is a very good isolation between the output and input of the two stage MGA-

665P8 amplifier. It has a very low 푆22 i.e. −21.83 푑퐵 while 푆12 is −37.72 푑퐵, at 5.8

GHz. Thus there is a very good output match at the output. At the input, 푆21 is

15.99 푑퐵 while 푆11 is −5.5 푑퐵, at 5.8 GHz. Since the gain of the amplifier is very high and a very high reverse isolation is present, thus no impedance matching is used in the design. The buffer amplifier consists of two stages of amplifiers, each requiring a separate bias. The pin 6 is supplied directly with the bias while the pin 7 is supplied with a bias fed through a Bias Tee.

3.4 Resonator Network

The frequency of resonance determines the type of resonator used. For low frequencies, lumped resonators are used while at higher frequencies, coaxial, ceramic, microstrip resonators are commonly used. It also determines the phase noise performance of the oscillator.

Fig. 3.4 shows a ceramic resonator with outer square cross section and inner cylindrical shape. The W and 푙 are the width and length of the outer conductor while d is the inner conductor diameter.

The approximate characteristic impedance of the coaxial line resonator is 60 푊 푍표 = ln 1.08 (3.7) 휀푟 푑

The 푅표 is usually between 5- 15 ohms while 휀푟 is from 10 to 100. This reduces the size of the length of these resonators and is suitable for many applications including spacecrafts.

휆푓푟푒푒 휆푒푓푓 = (3.8) 휀푟

Fig. 3.4 Typical coaxial ceramic Resonator [22]

The calculation of length of the quarter wave shorted coaxial resonator length is 1 푅푒푠표푛푎푡표푟 퐿푒푛푔푡푕 = 휆 (3.9) 4 푐표푎푥 The unloaded Q factor of the coaxial resonator is 1 1 1 = + (3.10) 푄푈 푄퐶 푄퐷 where

푄퐶 = 푄 푑푢푒 푡표 퐶표푛푑푢푐푡표푟 퐿표푠푠푒푠

푄퐷 = 푄 푑푢푒 푡표 퐷푖푒푙푒푐푡푟푖푐 퐿표푠푠푒푠

The 푄퐶 is due to the conductor current flow and is given as

2 휋 푓 휇휍 푏 푄 = ln (3.11) 퐶 1 1 + 푎 푎 푏 where 휇 = 푃푒푟푚푒푎푏푖푙푖푡푦 표푓 푡푕푒 푐표푛푑푢푡표푟 휍 = 퐶표푛푑푢푐푡푖푣푖푡푦 표푓 푡푕푒 푐표푛푑푢푡표푟

The 푄퐷 is due to the dielectric material that is in between the two conductors lines a and b in the coaxial resonator. 휍 푄퐷 = tan 훿 = (3.12) 2 휋 푓 휀1휀푟 where

1 휍 = = 퐶표푛푑푢푐푡푖푣푖푡푦 표푓 푡푕푒 퐷푖푒푙푒푐푡푟푖푐 휌

휀푟 = 푅푒푙푎푡푖푣푒 푃푒푟푚푖푡푡푖푣푖푡푦 −12 −1 휀표 = 푃푒푟푚푖푡푡푖푣푖푡푦 표푓 푡푕푒 푓푟푒푒 푠푝푎푐푒 = 8.854 × 10 퐹 푚

A cylindrical coaxial resonator with inner conductor diameter d and outer conductor diameter W. The equivalent circuit of this coaxial resonator is a parallel RLC circuit as given in Fig. 3.5. The coaxial resonator is plated with silver plating.

R C L

Fig. 3.5 Equivalent circuit of a Coaxial Resonator

The unloaded Q for this coaxial resonator is

푊 ln⁡(1.079 ) 푄 = 푘 푓 푑 (3.13) 푈 표 1 1 25.4( + ) 푊 푑 where 푊 = 푂푢푡푒푟 푐표푛푑푢푐푡표푟 푑푖푎푚푒푡푒푟 푑 = 퐼푛푛푒푟 푐표푛푑푢푐푡표푟 푑푖푎푚푒푡푒푟

푘 = 240 푓표푟 푠푖푙푣푒푟푒푑 푑푖푒푙푒푐푡푟푖푐 푤푖푡푕 휀푟 = 38.6

푘 = 200 푓표푟 푠푖푙푣푒푟푒푑 푑푖푒푙푒푐푡푟푖푐 푤푖푡푕 휀푟 = 88.5

The input impedance of the resonator is

60 푊 푍퐼푁 = ln 1.079 (3.14) 휀푟 푑 The inductance L is

푙 푍표 8 휀푟 퐿 = (3.15) 25.4 휋2 3 × 108 where 푙 = 푝푕푦푠푖푐푎푙 푙푒푛푔푡푕 표푓 푡푕푒 푐표푎푥푖푎푙 푟푒푠표푛푎푡표푟 푖푛 푚푚

The Capacitance is

푙 휀푟 퐶 = 8 (3.16) 25.4 × 2 × 3 × 10 푍표 The resistance is 4 푍 푄 푅 = 표 (3.17) 휋

The characteristics of coaxial line resonators below resonance are, they are high Q components and temperature stable and act as ideal inductor elements over a narrow range of frequencies as seen in Figure 3.6. The coaxial resonator acts as a distributed structure having capacitance and inductance. The line has a specific frequency at which it resonate called Self Resonant Frequency (SRF). The line exhibits inductive reactance when it is operated below the SRF while acts as capacitive if operated above SRF.

Fig. 3.6 Self Resonant Frequency of Coaxial Resonator [23]

Compared to traditional coil resonators, the ceramic coaxial resonators have a number of advantages  Their ruggedness makes it suitable for incorporating in PCBs.  They have a very good Q value  They have a better temparture stability perforamce  Immunity to microphonics

3.4.1 Modeling the Resonator The CAD (Computer Aided Design) software used is Agilent ADS (Advanced Design System). Since the manufacturer didn’t provide a specific CAD model for the resonator, a model is established for this simulation purposes.

The quarter wave shorted coaxial resonator model can be realized by using the parallel RLC circuit as below. The R, L and C values supplied by the manufacturer are simulated in parallel RLC circuit as in Fig. 3.7.

Fig. 3.7 Parallel RLC Model for the Resonator

Fig. 3.8 shows the impedance response of the above model and it shows that the impedance is highest at the 6.2 GHz frequency which is the frequency of oscillation. Thus this model acts as specifications.

m1 freq=6.200GHz mag(Meas1)=25641.000

3.0E4 m1 2.5E4

2.0E4

1.5E4

1.0E4 mag(Meas1)

5.0E3

0.0 5.8 5.9 6.0 6.1 6.2 6.3 6.4 6.5 6.6 freq, GHz

Fig. 3.8 Impedance Response of the resonator model

The manufacturer has its own recommended layout for the resonator pads as given in Appendix B. The recommendations are followed and pads are added to the model as shown in Fig. 3.9.

Fig. 3.9 Equivalent Model of the coaxial resonator with pads

This effectively changes the model supplied by the manufacturer. Fig. 3.10 shows that the frequency is shifted and the new model oscillates at 5.137 GHz which is more than one GHz below the frequency of the original model.

Readout m1 freq=5.137GHz mag(Meas1)=17048.076

2.0E4 m1

1.5E4

1.0E4

mag(Meas1) 5.0E3

0.0 4.4 4.6 4.8 5.0 5.2 5.4 5.6 freq, GHz

Fig. 3.10 Impedance vs. Frequency response for the model with pads

3.4.2 Coaxial Resonator Tuning

The prototype test circuit requires tuning for test purposes. This is due to the fact that the resonator is coupled with the stray capacitances in the vicinity which decreases the resonant frequency (SRF). There are three common methods for varying the SRF of the resonator in oscillator. These methods are mechanical in nature.

The SRF can be increased by removing the silver metallization from the resonator’s open end. This can be done from the top and also the bottom of the resonator. This can increase the SRF about 10-20 % without degrading the unloaded Q factor. Similarly the SRF can be increased by removing the silver metallization from the shorted end but with some degradation of the Q value. The change of the position of the ground plane can also vary the SRF [24].

3.5 Varactor Diode Tuning

Varactor diode or Voltage variable capacitor or simply varicap is a type of diode whose capacitance is a function of its applied voltage. The varactor diode which acts as capacitor in the presence of a shorted quarter wave resonator and is used to vary the operating frequency of the oscillator.

Readout

Fig. 3.11 Varactor Diode Equivalent Model

Fig. 3.11 shows the varactor diode which is reverse biased for proper operation. A small capacitor is also used in series with the varactor. This capacitor helps in increasing the tuning range of the varactor.

The varactors may be built from Silicon and Gallium Arsenide (GaAs). Si is economical for large scale production while GaAs provides higher Q values suitable for high frequency applications. The advances in material gradient doping have paved the way for improved processes. The abrupt junction is formed by the uniform doping and is commonly used method. This provides an inverse square root relationship which is considered as non linear.

The hyper abrupt junction provides a linear response over the frequency range by varying the control voltage. These are narrowband in nature for linear region and the Q is reduced, which imply that it can be used in lower frequency applications.

Fig. 3.12 Junction Capacitance as a function of Bias Voltage + Contact Potential [25].

Fig. 3.12 shows the relationship of the junction capacitance and bias voltage along with contact potential. With increase the control voltage of the varactor diode, the capacitance of the varactor diode decreases and this shifts the oscillator’s frequency.

퐶푗 표 퐴푏푟푢푝푡 퐽푢푛푐푡푖표푛 퐶 푉 = 퐴 = (3.18) 퐽 푅 푉 훾 1 + 푅 휙

퐶푗 표 퐹푟푒푞푢푒푛푐푦 퐿푖푛푒푎푟 퐶퐽 푉푅 = 퐹 = 훾 (3.19) 1 + 푉푅

퐶푗 표 퐻푦푝푒푟 퐴푏푟푢푝푡 퐽푢푛푐푡푖표푛 퐶 푉 = 퐻 = (3.20) 퐽 푅 푉 훾 1 + 푅 휙 where 훾 = 0.5 푓표푟 푎푏푟푢푝푡 푗푢푛푐푡푖표푛 훾 > 0.5 푓표푟 퐻푦푝푒푟 푎푏푟푢푝푡 푗푢푛푐푡푖표푛 휙 = 퐽푢푛푐푡푖표푛 퐶표푛푡푎푐푡′푠 푃표푡푒푛푡푖푎푙 = 0.7 푣 푓표푟 푆푖 & 1.1 푣 푓표푟 퐺푎퐴푠

퐶푗 표 = 퐷푖표푑푒 푐푎푝푎푐푖푡푎푛푐푒 푎푡 푧푒푟표 푣표푙푡푠 푏푖푎푠 푣표푙푡푎푔푒

푉푅 = 퐴푝푝푙푖푒푑 푟푒푣푒푟푠푒 푏푖푎푠 푣표푙푡푎푔푒

3.5.1 Mathematical Model for a varactor The mathematical model of a varactor diode is described in Fig. 3.13. The variable capacitance of the junction is 퐶퐽 (푉) at applied voltage, while 푅푆(푉) is the series resistance of the varactor diode. The diode has some constant parasitic capacitance 퐶푃 as well due to packaging, dimensions of the diode and wiring. A parasitic inductance is 퐿푝 also present. The varactor diode is supplied with reverse bias voltage which changes its capacitance and series inductance. This in turn changes the frequency and/or phase of the electrical network.

Fig. 3.13 Equivalent Model of a Varactor Diode

3.5.2 Tuning Ratio of the Varicap The tuning ratio is defined as the ratio in the capacitance between two values of the applied reverse bias voltage.

퐶푗 (푉2) 푉 + 휙 푇푢푛푖푛푔 푅푎푡푖표 = = ( 1 )훾 (3.21) 퐶푗 (푉1) 푉2 + 휙 where

퐶푗 푉1 = 퐽푢푛푐푡푖표푛 퐶푎푝푎푐푖푡푎푛푐푒 푎푡 푣표푙푡푎푔푒 푉1

퐶푗 푉2 = 퐽푢푛푐푡푖표푛 퐶푎푝푎푐푖푡푎푛푐푒 푎푡 푣표푙푡푎푔푒 푉2

Provided that

푉1 > 푉2

3.5.3 Varactor in VCOs The varactors are mainly used in VCOs for frequency tuning. The desirable characteristics of varactors are  Minimum series resistance so that it does not affect the resonator Q factor.  Less noise as its noise is added to the over all VCO noise  The appropriate tuning range and 퐶 − 푉 characteristics [22]

3.6 Directional Coupler

The single section microstrip coupled line Directional Coupler consist of two microstrip transmission lines close together. Due to this close proximity, the electromagnetic energy or power is coupled between the lines.

Fig. 3.14 shows a microstrip directional coupler with four ports i.e. Input, transmitted, coupled and isolated. One of the lines is termed as the “main line” which is between port 1 and 2. This is the transmitted part where most of the power flows. The other arm is called the coupled arm where a fraction of the input power is coupled. The isolated port is terminated with matched impedance. The need for a coupler is to divert some fraction of a power i.e. −10 푑퐵 coupling in case of this oscillator, to the synthesizer without interrupting much the main line power transfer.

Fig. 3.14 Coupled lines Directional coupler [26]

The microstrip coupler design and size depends on the choice of the substrate, height of the substrate, dielectric permittivity, spacing between the coupled lines 푆, width of the lines 푤 and length of the lines.

The directional coupler is linear and symmetrical in nature with the same width and length of both the coupled lines so any port can be taken as the input port while the port on the other side of the same arm is automatically the output. Similarly the adjacent arm has the coupled and isolated ports. The coupled port provides the frequency as well as a fraction of the power from the main line.

3.7 Bias Networks and Tees

There are three active semiconductor devices in the VCO’s design. These are the VCO’s transistor, Buffer amplifier gain block and the varactor diode. All these require a supply voltage which is fed through bias Tees.

The Bias Tee consists of a quarter wavelength transmission line which is about 7.8 푚푚 for the selected Rogers RO4003 substrate. Thus adding quarter wavelength of line which is a short at the active device input will be open at quarter wave length away. This is due to the fact that the quarter wavelength will move the short in is Smith chart to open, 180 표clockwise around the center of the smith chart. This transforms the RF short circuit to RF open circuit at the other end of the quarter wave line. The radial stub acts as a RF short and its DC open. The radial stub offers good bandwidth compared to other stubs. Its isolates the RF from the bias tee and inducts the DC freely.

3.8 Shielding

The requirements of shielding depend mainly on the use of the oscillator in a specific application. For aerospace applications, the weight of the shield may be 10-15 % of the total mass of the systems; a lighter shield is desired with good EMI shielding abilities, high strength and high elasticity. In such cases metallic shields may not be used and instead carbon composites/epoxies with some painting.

For this wave radar, the weight of the shield is not problem, as the radar is stationary at some specific location, and the best shield may be selected which may be some metallic shielding, like Aluminum, Copper etc. These Cu and Al have a very good attenuation properties starting from 1 MHz and onward frequencies as in Fig. 1.1.

The metallic shielding is very efficient even in low frequency EMP. Since the shield may be around the resonator and the oscillator circuit, so the distance is not very large and ferrites may not be a better option than metallic shields as they are more conductive.

Chapter 4 Process and Results

4.1 VCO’s Transistor Measurements

Fig. 4.1 displays the VCO’s transistor measurement test board. The transistor is NEC’s NE685M03 and the configuration used is common base for the design of the oscillator. The manufacturer does not provide a good model for up to 6 GHz frequency with common base configurations, thus the measurements for the transistor are carried out.

There is a radial stub at the base of the transistor for grounding. The measurements are carried out through Vector Network Analyzer (VNA). The VNA is first calibrated with Through Reflect Line (TRL) calibration method (See appendix C). The measurement of the S parameters for the transistor with a radial stub at its base is better characterized in this way. The transistor is supplied with a proper bias voltage.

Fig. 4.1 VCO’s transistor measurements

4.2 Attenuator Results

Fig. 4.2 shows the attenuator selected for the oscillator which is a symmetrical Pi Network of resistors. The design is supplemented by the introduction of transmission lines in between the resisters which are used to cancel out the reactance of the resistors as the resisters are not perfect lumped components at very high frequencies.

Fig. 4.2 Schematics of the 10 dB Pi Attenuator

0 m1 m1 freq=5.870GHz -10 dB(S(1,2))=-9.642

-20 m3

dB(S(2,2)) dB(S(1,1)) dB(S(2,1)) dB(S(1,2)) freq=5.870GHz -30 dB(S(1,1))=-37.311 m3

-40 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 freq, GHz

Fig. 4.3 S parameter simulations of the 10 dB attenuator

Fig. 4.3 shows the S parameter simulations of the 10 dB attenuator. The response is over a frequency range of 4.8-6.8 GHz. The attenuation is with very good flatness of a fraction of a dB over the entire frequency range. The attenuation of the attenuator i.e. S(2,1) is 9.642 푑퐵 which is close to 10 푑퐵. Since the specifications of this attenuation is very flexible, the difference of 0.4 푑퐵 is not a problem at all. The S(1,2) shows the reverse isolation of the attenuator and it is the same as S(2,1). This is due to the fact that this design is a symmetrical Pi network.

Readout

The reflections at the input and output of the attenuator are minimum as shown by very low S(1,1) and S(2,2), which are −37.3 푑퐵. This means that the returns loss is very high. Again the S(1,1) and S(2,2) are both equal due to the symmetrical design of the attenuator.

m2 m2 freq=5.870GHz

S(1,1)=0.014 / 21.340 S(2,2) S(1,1) impedance = Z0 * (1.026 + j0.010)

freq (4.800GHz to 6.800GHz)

Fig. 4.4: Input and Output Impedance of the Attenuator

Fig. 4.4 shows the input and output impedance of the attenuator to be almost 50 ohms and thus the VSWR is close to 1. This minimizes the reflections at the input and output of the attenuator.

4.3 Buffer Amplifier

Fig. 4.5 shows the buffer amplifier schematics. The amplifier is a gain block thus it is used directly at the output of the attenuator. The amplifier is supplied with a bias supply of 3 푉 푎푛푑 21 푚퐴. This gain block is not supplied with any input or output matching.

Readout

Fig. 4.5: Buffer Amplifier Schematics with 3V, 21mA bias point.

m1 freq=5.870GHz 20 m1 dB(S(2,1))=15.985

m2 10 freq=5.870GHz m7 dB(S(1,1))=-5.261 0 m2 m3 freq=5.870GHz -10

dB(S(2,2))=-22.136 nf(2)

dB(S(2,2)) dB(S(2,1)) dB(S(1,2)) dB(S(1,1)) m4 -20 m3 freq=5.870GHz dB(S(1,2))=-37.721 -30 m7 m4 freq=5.876GHz nf(2)=1.688 -40 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 7.0 f req, GHz

Fig. 4.6: Response of the Buffer amplifier

The Buffer amplifier response is shown in Fig. 4.6. The gain of the amplifier without any input or output matching is about 16 dB. The most important parameter is the reverse isolation of the amplifier S(1,2) which is −37.72 푑퐵. This is the most significant factor and very important for the oscillator along with the attenuator’s reverse isolation. The others parameter S(2,2) is −22.14 푑퐵 which is sufficient as more than 20 푑퐵 of return loss is always desirable. But the S(1,1) is just about −5.2 푑퐵 which is a bit too low.

5.867G5.852GReadout5.874G -5.271-37.72-22.15 m6 freq=5.870GHz S(1,1)=0.546 / -125.040 impedance = Z0 * (0.365 - j0.464)

S(2,2) S(1,1) m5 freq=5.870GHz S(2,2)=0.078 / -140.330 m6 impedance = Z0 * (0.882 - j0.089)

freq (4.870GHz to 6.870GHz)

Fig. 4.7: Input and output impedance match of the buffer amplifier

The Input and output impedance match for the buffer amplifier is shown in Fig. 4.7. The output impedance match is very good nearly 50 ohms but the input impedance match is very poor. For the initial design, the Buffer Amplifier Gain block is used without any input and output impedance match. This is due to the fact that the Buffer amplifier has a very good gain of 16 dB without any matching circuit at the input or output.

4.4 Power Supply Bias

The three active devices i.e. VCO’s transistor, varactor diode and the Buffer amplifier is supplied with the proper bias voltage. The bias is supplied through bias tees and Fig. 4.8 shows the schematics of bias tee. The bias tee consists of a quarter wave transmission line which is about 7.8 mm on the Rogers 4003 substrate and a radial stub. The layout of the bias tee is shown in Fig. 4.8.

Readout

Fig. 4.8: Schematics of a Bias Tee

Fig. 4.9: Layout of the Bias Tee.

m1 freq=5.780GHz S(1,1)=0.008 / 95.719 0.00 impedance = Z0 * (0.998 + j0.015)

-0.02

-0.04

-0.06

dB(S(3,3)) dB(S(2,1))

-0.08 m1

-0.10

S(3,1) S(3,3) 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 S(1,1) freq, GHz

-20 m2 freq=5.870GHz

-40 dB(S(3,1))=-79.907 freq (4.800GHz to 6.800GHz) dB(S(3,1)) dB(S(1,1)) -60

m2 -80 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 m3 freq, GHz freq=5.870GHz phase(S(3,1))=-179.901

200

100

phase(S(3,1)) -100 m3

-200 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 freq, GHz

Fig. 4.10: Simulations results of the Bias Tee

The response of the bias tee is shown in Fig. 4.10. The S(2,1) is almost 0 dB which implies that the high frequency signal is not affected by the bias Tee thus it passes from port 1 to port 2 without any loss. The S(3,3) is also close to 0 dB which is port 3 is an RF Open and thus DC Short. Thus the RF signal will not enter the bias supply. The value of -80 dB of the S(3,1) also confirms this. The reflections at port 1 are minimum as S(1,1) is about -60 dB. Also at port S(1,1), the impedance is almost 50 ohms. This bias Tee is although not ideal but very much optimized and closer to ideal.

4.5 Coupler Design

The oscillator output is coupled with the synthesizer and the coupler is designed as in Fig. 4.10. The coupler has four ports which are input, output, coupled and isolated. The design is based on two microstrip lines coupled to each other. The isolated port is terminated with matched impedance.

Readout

Fig. 4.11: Schematics of 12 dB Coupler

The response of this coupler is given in Fig. 4.11. The coupled port has a coupling of about 11.8 dB which is almost equal to 12 dB. The thru is -0.345 dB which depicts the fact that most of the power is transmitted to the output port and only a small fraction is sent to the coupled port. The directivity is -24.17 dB which is fairly good. The Return loss is -29.9 dB thus there is very little reflections and the input port acts as a matched port.

Thru(dB) Coupled Port(dB) -0.20 m1 -11.0 freq= 5.870GHz dB(S(2,1))=-0.345 -0.25 -11.5 m2 m2 freq= 5.870GHz -12.0 -0.30 dB(S(3,1))=-11.802

m1 -12.5 dB(S(3,1)) dB(S(2,1)) -0.35 -13.0

-0.40 -13.5

-14.0 -0.45 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0 freq, GHz freq, GHz

Directvity(dB) Return Loss(dB) -10 -19 m3 freq= 5.870GHz -20 -15 dB(S(3,2))=-24.170

-21 m4 -20 freq= 5.870GHz

-22 dB(S(2,2))=-29.972

dB(S(3,3)) dB(S(2,2))

dB(S(1,1)) -25 dB(S(3,2)) -23 m4 -30 m3 -24

-35 -25 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0 freq, GHz freq, GHz

m5 freq= 5.870GHz S(1,1)=0.031 / -141.241

m5 impedance = Z0 * (0.952 - j0.037) S(1,1)

freq (3.800GHz to 7.800GHz)

Fig. 4.12: Response of the 12 dB Coupler

Fig. 4.12 displays the voltage controlled coaxial resonator oscillator. The VCO’s transistor is connected to the tuning network. The tuning network consists of the coaxial resonator with one end shorted to ground. A varactor diode is also present in the tuning network. The varactor and the resonator are connected through a tee and are fed into the input of the VCO’s transistor. A 0.3 pF capacitor is also added along the varactor diode.

Readout Readout

Readout 4.6 Voltage Controlled Oscillator (VCO) Circuit

The VCO circuit is shown in Fig. 4.13. A varactor is supplied with a variable bias voltage which is used to tune the circuit. The VCO’s transistor is followed by an attenuator of Pi shape. The attenuator is followed by the buffer amplifier and the output is fed into the coupler. Fig. 4.14 displays the layout of the voltage controlled oscillator. This layout is generated in Agilent ADS and does not include the voltage bias circuits. This layout was imported in layout software “Mentor” and a PCB layout was generated which contain all other components like the bias networks, connectors, transmission lines etc.

Fig. 4.13: Schematics of the Voltage Controlled Oscillator

Fig. 4.14: Layout of the Voltage Controlled Oscillator

m1 freq=5.570GHz dB(S(1,1))=3.716

5 200 m1

4 phase(S(1,1)) 100 m2 3 m2 freq=5.570GHz 0 phase(S(1,1))=-0.008

2 dB(S(1,1)) -100 1

0 -200 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 freq, GHz

-2.0 -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 2.0 S(1,1) m3 freq=5.570GHz S(1,1)=1.534 / -0.008

freq (4.800GHz to 6.800GHz)

Fig. 4.15: OscTest Response of the Oscillator

These simulations are based on the OscTest component of Agilent ADS which provides the S parameter analysis of the small signal gain (closed loop) for this oscillator. The loop gain is represented by S(1,1) while it also measures the closed loop phase. The oscillation is indicated by a loop gain of more than unity and the phase at zero, while the frequency increase brings about a decrease in the phase.

Readout

Readout The Agilent ADS simulations of the VCO are given in Fig. 4.15. The oscillator has S(1,1) greater than unity at the 5.57 GHz while the phase at this frequency is almost zero. Thus this is the oscillation frequency with a varactor bias voltage of 4 volts.

The Table 4.1 displays the results of simulations of the VCO. The varactor diode bias voltage is selected from 4-18 volts with increments of 2 volts, which is the standard operating voltage for Microsemi MPV2100 varactor diodes. The oscillation frequency starts at 5.57 GHz at 4 volts of varactor voltage and reaches 5.636 GHz at a voltage of 18 volts. This gives a total tuning voltage of 80 volts.

Serial Number Varactor bias voltage Oscillation frequency (volts) (GHz) 1 0 5.538 2 4 5.557 3 6 5.582 4 8 5.587 5 10 5.601 6 12 5.610 7 14 5.619 8 16 5.628 9 18 5.636

Table 4.1: Oscillation Frequency of VCO as a function of varactor bias voltage

4.7 Fabrications and Measurements

Fig. 4.16: PCB of the VCO

Fig. 4.16 shows the fabricated coaxial resonator oscillator. The results show that the oscillator oscillates at 5.54 GHz in the absence of the 0.3 pF capacitor and the varactor diode. This is almost very close to the designed frequency of oscillation. The output power is 3.83 dBm measured though a spectrum Analyzer as shown in Fig. 4.17. It’s a bit lower than the expected 7 dBm for the specifications.

When the 0.3 pF capacitor is added with the resonator, the frequency of oscillation of the VCO drops to 5.45 GHz while the output power is 4 dBm. Finally the varactor diode is soldered and the results are obtained as in the Table 4.2.

Fig. 4.17: VCO measurements through Spectrum Analyzer

Serial Number Varactor bias Oscillation Output Power voltage frequency (dBm) (volts) (GHz) 1 0 5.230 4 2 4 5.270 4 3 6 5.290 3.83 4 8 5.300 3.83 5 10 5.314 3.83 6 12 5.324 3.67 7 14 5.333 3.83 8 16 5.340 3.83 9 18 5.345 3.50

Table 4.2: Results of the VCO for the Varactor tuning

Table 4.2 show that the VCO has almost constant power of around 4 dBm. The tuning range is 75 MHz which is good enough for the design.

푇푒푚푝 = 0표 퐶 푇푒푚푝 = 20표 퐶 푇푒푚푝 = 40표 푇푒푚푝 = 60표

S 푉푣푎푟 푓표 푃푂푈푇 푓표 푃푂푈푇 푓표 푃푂푈푇 푓표 푃푂푈푇 No (푣표푙푡푠) (퐺퐻푧) (푑퐵푚) (퐺퐻푧) (푑퐵푚) (퐺퐻푧) (푑퐵푚) (퐺퐻푧) (푑퐵푚) . 1 4 5.285 4.17 5.279 3.67 5.271 3.67 5.254 3.13 2 6 5.300 4.33 5.292 3.67 5.283 3.67 5.274 3.33 3 8 5.313 4.17 5.305 3.83 5.295 4.00 5.284 3.13 4 10 5.327 4.33 5.317 4.17 5.305 3.5 5.293 3.13 5 12 5.337 4.17 5.328 4.17 5.314 3.67 5.305 3.5 6 14 5.346 4.00 5.337 4.17 5.325 3.84 5.312 3.5 7 16 5.354 3.53 5.345 4.00 5.334 3.83 5.319 3.5 8 18 5.357 4.00 5.347 3.83 5.337 3.83 5.340 3.67

Table 4.3 Results of the VCO over a Temperature Range where

푓표 = 푂푠푐푖푙푙푎푡푖표푛 퐹푟푒푞푢푒푐푛푦 표푓 푡푕푒 푉퐶푂 푒푥푝푟푒푠푠푒푑 푖푛 퐺퐻푧

푉푣푎푟 = 퐵푖푎푠 푣표푙푎푡푔푒 푎푝푝푙푖푒푑 푡표 푡푕푒 푣푎푟푎푐푡표푟 푑푖표푑푒 푒푥푝푟푒푠푠푒푑 푖푛 푣표푙푡푠

푃푂푈푇 = 푂푢푡푝푢푡 푝표푤푒푟 표푓 푡푕푒 푉퐶푂 푒푥푝푟푒푠푠푒푑 푖푛 푑퐵푚

Table 4.3 displays the temperature performance of the oscillator. The temperature is selected from 0표 to 60표 which induces a shift of the frequency about 30 MHz.

Phase Noise Measurements

The phase noise of this oscillator is −88 푑퐵푚/퐻푧 푎푡 50 푘퐻푧 offset from 4 푑퐵푚 carrier or −92 푑퐵푐/퐻푧. At 100 kHz offset the phase noise is −94 푑퐵푚/퐻푧 ( −98 푑퐵푐/퐻푧).

Load Pull Measurements

Figure 4.1 shows the Load Pull measurement setup, with the DUT attached to the cable having 1.5 dB loss, and a 3 dB attenuator. The attenuator is connected to -10 dB approximately. The coupled port is connected to the Spectrum Analyzer while the thru port is connected to the Line Stretcher which is open circuit at other end.

Fig. 4.18: Load Pull measurements

The measurement shows that with a return loss of the setup equal to almost 12 dB, and with a change of phase from 0 − 180표 , a 1MHz of frequency pulling is calculated.

The results were a bit different from the expected simulations. The biggest cause was that the resonator layout was incorrectly perceived, and it pulled the frequency about 300 MHz down from its target of 5.87 MHz. This was due to incorrect layout realization due to which the exact frequency of 6.2 GHz was not achieved. This calls for a need to make test circuits for the resonator and the test it using one port VNA measurements. Thus the S Parameters obtained this way can be used in the simulations.

Chapter 5 Discussions & Conclusions

Table 4.1 shows simulations of the varactor diode tuning and the resulting frequency response in GHz. The varactor diode has an operating voltage range from 4 to 18 volts. This will give a tuning range of 푇푢푛푖푛푔 푅푎푛푔푒 = 5.557 퐺퐻푧 − 5.636 퐺퐻푧 = 79 푀퐻푧

The printed board measurements through a spectrum Analyzer showed the tuning range as

푇푢푛푖푛푔 푅푎푛푔푒 = 5.270 퐺퐻푧 − 5.345 퐺퐻푧 = 75 푀퐻푧

This shows a very good result of the tuning range as the 75-80 MHz tuning is sufficient for the products at the company.

The capacitor in series with the varactor diode has a value of 0.3 pF in the VCO design. This value is varied and some new results have been obtained as given in Table 5.1. If this capacitor is not included, the OscTest S(1,1) phase is at zero but the S(1,1) is not at maximum. In some cases the tuning range is very low if this capacitor is not added. Thus the capacitor is added to the design. The value of the capacitor is varied and results obtained in Table 5.1. These results show that the increase in the capacitance increases the tuning range.

S. No Capacitor Frequency Frequency Tuning (pF) (at 4 volts) (at 18 volts) frequency range 1 0.1 5.765 5.775 10 MHz 2 0.2 5.664 5.694 30 MHz 3 0.3 5.574 5.634 60 MHz 4 0.4 5.493 5.574 81 MHz 5 0.5 5.433 5.544 111 MHz

Table 5.1 Tuning range results with varying capacitor values

There is another problem with this capacitor, as its value is too low i.e. 0.3 pF. The high tolerance of this capacitor will increase the uncertainty. The parasitic reactance could also be appreciable due to the manufacturing, soldering and mounting of the components. Simulations have shown that increasing this capacitor value can increase the tuning range but then a very high value capacitor value in the tuning network will shift the oscillation frequency down which is not desirable. Thus a trade off was made and a capacitor value was suggested and used.

The design of oscillator gave a low output power of around 3.5-4 dBm which is lower than the expectations of the 7 dBm. In the previous design, no matching network was used for the buffer amplifier although its input had a very bad match. Thus the input of the buffer amplifier has to be matched with a matching network, to achieve a higher gain.

Fig. 5.1 shows that the input is matched and the gain of the amplifier is 17.5 dB which is 1.5 dB higher. Also the Smith chart shows the input and output match to be around 50 ohms.

Fig. 5.1 Schematics for Matching Network of the Buffer Amplifier

20 m1 m1 freq=5.870GHz 10 dB(S(2,1))=17.503 0 m2 freq=5.870GHz dB(S(1,1))=-30.917 -10 m3

-20

dB(S(2,2)) dB(S(2,1)) dB(S(1,2)) m3 dB(S(1,1)) freq=5.870GHz m2 -30 dB(S(2,2))=-19.050 m4

m4 -40 freq=5.870GHz dB(S(1,2))=-36.203 -50 4.8 5.0 5.2 5.4 5.6 5.8 6.0 6.2 6.4 6.6 6.8 7.0 f req, GHz

m6 freq=5.870GHz S(1,1)=0.028 / 146.630 m6

impedance = Z0 * (0.953 + j0.030) m5

S(2,2) S(1,1)

m5 freq=5.870GHz S(2,2)=0.112 / -104.515 impedance = Z0 * (0.924 - j0.202)

f req (4.870GHz to 6.870GHz)

Fig. 5.2 Results of the Matching Network of Buffer Amplifier

There is an uncertainty with the resonator modeling and characterization as well. The resonator data was taken from the manufacturer datasheets and a parallel RLC model was created as shown in Fig. 3.7 and 3.8. The model showed perfect match with the manufacturer data sheet values.

The manufacturer provided the resonator’s recommended pad for mounting on the PCB as seen in Fig. 5.3. One end is shorted and the other end is connected to the varactor tuning network through a Tee that leads into the transistor.

Fig. 5.3 Recommended pad for the Resonator

Readout

Readout Simulating again the model as in Fig. 3.9 and 3.10 shows that the addition of the pads pulled the resonator frequency down to 5.137 GHz, which is about 1 GHz less than the resonator original frequency.

Another approach to characterize the resonator network is to take one port measurements of the resonator and then use these measurements in the simulations. S parameters were extracted and simulated as in Fig. 5.4 and 5.5. This showed that the resonance frequency at 6.63 GHz. This method characterizes the resonator along with the pads.

Fig. 5.4 Resonator with Pads, One Port Test circuit

m1 freq=6.630GHz dB(S(1,1))=-5.204 1

-1

-2

-3

dB(S(1,1)) -4 m1 -5

-6 4.5 5.0 5.5 6.0 6.5 7.0 7.5 8.0 freq, GHz

Fig. 5.5 One Port Measurements of the Resonator

6.630G -5.204 The manufacturer provided 10 samples and they were about at maximum 30 MHz offset from the resonator’s 6.2 GHz frequency. So this tolerance has to be taken into consideration as well.

The attenuator design was according to specifications. The input and output match were almost 50 ohms thus the VSWR is close to 1. The output of the attenuator is fed to the input of buffer amplifier, thus the matching is preserved and the buffer amplifier provides very good gain and high reverse isolation. The flatness is just a fraction of a dB for a 2 GHz frequency range which is perfect.

The comparative study of this oscillator design with the state of the art studies is described. Comparing with [1], where a state of the art phase locked DRO/CRO is presented, with a phase noise 표푓 − 120 푑퐵푐/퐻푧 푎푡 100 퐾퐻푧 offset from carrier. The phase noise of our oscillator at 100 KHz offset is 94 푑퐵푚/퐻푧 (−98 푑퐵푐/ 퐻푧 푎푠 푐푎푟푟푖푒푟 푝표푤푒푟 푖푠 4 푑퐵푚). There is a difference of 22 dB and this may be due to several facts. One of the reasons is that it is only the VCO design, and it’s not phase locked CRO. The 2nd reason is that the resonator layout pad was not designed as recommended by the manufacturer and this pulled the frequency down. The bias Tees, attenuator, and all other circuits were optimized for the 5.87 GHz, which may have contributed to this phase noise. Thus approximating, this VCO will provide almost the same response of the phase noise as in [1], when the resonator layout pads is correctly designed and measured to provide the 6.2 퐺퐻푧 as specified by the manufacturer.

In [9], where the range of a varactor diode is increased by more than 10 times, by the introduction of the negative capacitance connected to the varactor. The negative capacitance is created by two common source transistors, loaded with inductors. The table 5.1 describes the tuning characteristics of the varactor in the presence of a small capacitor connected to the varactor. The increase in the capacitance increases the tuning range by 10 times by increasing the value of this capacitance.

In [11], the frequency stability is 130 ppm without any temperature compensation from −50표퐶 푡표 +50표 퐶. The variation in output power at −50표퐶 is +35 푑퐵푚 (3.2 푊) and at +50표 퐶 was 33 푑퐵푚 (2 푊) which shows the temperature stability of the DROs. Comparing, the Table 1.3, shows that from 0 − 60표 퐶 , the change in output power is about 0.5-1 dBm, which is relatively stable with the change of temperature.

In [12], a very linear MIC bipolar VCO with 100 MHz FM rate s described with frequency pulling is ± 1 푀퐻푧 into a 2:1 mismatched load. The total load isolation achieved is more than 45 dB. The FET amplifier provides 35 dB of isolation while 12 dB of isolation is provided by a thin film pi attenuator. Comparatively, this oscillator achieves almost the same results with a total isolation of about 47 dB, and a 1 MHz frequency pull due to load and with 10 푑퐵 coupler and a variation of 0 − 180표 phase change.

In [15], he experimental results predicts that the residual phase noise of the HBT based oscillators is a function of the bias dependant LF noise up conversion factor of the device. This concludes that for a low phase noise, the two important decisions are device selection and matching network design. Since the resonator layout was mistakenly incorrect, and shifted the frequency of the resonator to about 350 MHz, the Bias Tees and other circuits designed for the original frequency or 5.87 MHz, were not matched, thus it is may have contributed to the phase noise also.

Chapter 6 Future Work

The Agilent ADS simulations, as seen in Fig. 4.15 show that the oscillator is oscillating at 5.57 GHz of frequency. This is the 1st test board where there were some uncertainties. Thus the next version of this oscillator will be designed to oscillate at 5.87 GHz as the product specifications. This would mean that the uncertainty of the resonator layout will be removed by careful design and testing.

The capacitor in series with the varactor diode should have an optimized value that could give us a very good tuning frequency range. The uncertainties regarding the low value of the capacitance, component tolerances and manufacturing parasitic reactances are to be removed. This can be done by a design which uses a capacitor with 0.5-1 pF capacitor. This increase in bandwidth could also make it useful for other frequencies within the band.

The next version of the oscillator will have the Phase Locked Loops (PLL) by the introduction of the synthesizer and microcontroller. The size of the microstrip lines will be reduced by some adjustments to the resonator network. This could improve the power out of the VCO transistor.

At last, the simple bias will be replaced by a bias which produces minimum spurious and phase noise. This can improve the phase noise performance of the oscillator. This can be realized by having very high capacitance, de-coupling capacitors in the supply to filter out the noise.

References

[1] B. Hitch and T. Holden, “Phase Locked DRO/CRO for Space use”, IEEE International Frequency Control Symposium, pp. 1015-1023. May 1997. [2] U. L. Rohde and A. K. Poddar, “Novel Multi-Coupled Line Resonators Replace Traditional Ceramic Resonators in Oscillators/VCOs”, IEEE International Frequency Control Symposium and Exposition, pp. 432-442, June 2006. [3] X. Wang and H.Wang, “The Innovative Research of Integrating Electromagnetic Shield into Three- Dimensional Circuit”, in Seventh International Conference on Electronic Packaging Technology, pp. 1-7, Aug. 2006. [4] S. Miyake, Y. Umezu, Y. Sagawa, T. Morita, and R. Yoshino, “Investigation related to Construction Method and Performance of an Electromagnetic Shielded Enclosure”, IEEE International Symposium on Electromagnetic Compatibility, pp. 120-125, Aug. 1991. [5] J. R. Gajer, “Intercalated Graphite Fiber Composites as EMI Shields in Aerospace Structures” IEEE Transactions on Electromagnetic Compatibility, vol 34, no. 3, part 1, pp. 351-356, Aug. 1992. [6] S. R. Ramasamy and Devender, “A Review of EMI Shielding and Suppression Materials” In the Proceedings of the International Conference on electromagnetic Interference and Compatibility, pp. 459-466, Dec. 2007. [7] C. A. Grimes and D. M. Grimes, “A Brief Discussion of EMI Shielding Materials”, IEEE Aerospace Applications Conference, Dig. , pp. 217-226, 1993. [8] F. Lawrence, Babcock, “Shielding Circuits from EMP” IEEE Transactions on Electromagnetic Compatibility, vol. 9, no. 2, pp. 45-48, Sept. 1967. [9] S. Koley, B. Delacressonniere, J.-L. Gautier, “Using a Negative capacitance to increase the tuning range of the varactor diode in MMIC Technology”, IEEE Transactions on Microwave Theory and Techniques, vol. 49, no. 12, Dec. 2001. [10] P. Stockwell, D. Green, C. McNeilage and J. H. Searls, “A Low Phase Noise 1.3 GHz Dielectric Resonator Oscillator”, IEEE International Frequency Control Symposium and Exposition, pp. 882-885, June 2006. [11] M. Mizan, D. Sturzebecher, T. Higgins and A. Paolella, “An X- Band, High Power Dielectric Resonator Oscillator for Future Military Systems”, IEEE Transactions on Ferroelectrics and Frequency Control, vol. 40, no. 5, pp. 483-487, Sept. 993.

[12] R. G. Winch and J. L. Matson, “Very Linear X-Band MIC Bipolar VCO with 100 MHz Rate”, MTT-S International Microwave Symposium, Dig. vol. 80, pp. 499-500, May 1980. [13] R. Spence, “A Transistor Oscillator Frequency Stability Study”, IRE Transactions on Circuit Theory, vol. 9, no. 2, pp. 110-115, June 1962. [14] J. J. Ebers and S. L. Miller, “Design of Alloyed Junction Germanium Transistors for High-Speed Switching”, Bell Sys. Tech. J., vol. 34 761-781, July 1955. [15] X. Zhang and A. S Daryoush, “Bias Dependant Noise UP-Conversion Factor in HBT Oscillator”, IEEE Microwave and Guided Wave Letters, vol. 4, no. 12, pp. 423-425, Dec. 1994. [16] D. Usanov, A. Skripal, A. Abramov and V. Pozdnyakov, “ Semiconductor Microwave Oscillators Controlled by the Bias Point Voltage”, pp. 13-17, Sept. 2004. [17] RFIC Theory Tutorials, August 2008, www.rfic.co.uk [18] U. L. Rhode, A. K. Poddar and G. Bock, “The Design of Modern Microwave Oscillators for Wireless Applications”, John Wiley & Sons,Inc. New Jersey, 2005. [19] J. Tang and D. Kasperkovitz, “High Frequency Oscillator Design for Integrated Transceivers”, Kluwer Academic Publishers, New York. [20] Agilent Technologies, “Attenuator Overview”, April 2008. [21] H. Kinley, “Demystifying RF Attenuators” Urgent Communications, Nov 1, 2004. [22] High Frequency Electronics, “Basic Data on High Q Ceramic Coaxial Resonators” Summit Technical Media, LLC, November 2002. [23] Skyworks, Appli. Note No 1008, “Coaxial Resonator for VCO Applications”. [24] Skyworks, Appli. Note No 1010, “Frequency Tuning of Coaxial Resonators”. March 9, 2007. [25] MicroMetrics, Appl. Note, “Tuning Varactors” May 2008. http://www.micrometrics.com/pdfs/TV_AppNotes.pdf [26] Directional Coupler, July 2008, www.ee.bilkent.edu.tr/~microwave/programs/magnetic/dcoupler/theory.htm

Appendix A IMC Resonators

The resonators selected for the oscillator are coaxial resonators from Integrated Microwaves Corporation (IMC). The selected resonator for the design is about 4 푚푚2 in size. Fig. below shows the resonator dimensions where the inner diameter is 1.14 mm. The length of the resonator is 4.27 mm.

Fig. 4.16 Dimensions of the IMC resonator 4 풎풎ퟐ coaxial resonators.

The software provided by IMC simulates the 4 푚푚2 quarter wave shorted resonators. The impedance response vs. frequency is given as in Fig. 4.17. The impedance is highest at the 6.2 GHz.

Fig. 4.17 Impedance vs. frequency response of the coaxial resonator

The phase response is given in Fig. 4.18.

Fig. 4.18 Phase response of the IMC resonators

The tolerance of the resonator is± 1 %. The 10 sample resonators acquired from the IMC were having about ± 0.5 − 0.7 % of tolerance which is inside the manufacturer specifications. The resonator selected was a without the tab. Further the resonators are ceramic in nature.

The following characteristics are for the resonator, supplied by manufacturer.

퐷푖푒푙푒푐푡푟푖푐 퐶표푛푠푡푎푛푡 = 휀 = 8 푄푢푎푙푖푡푦 퐹푎푐푡표푟 = 푄 = 720

푍표 = 28.03 Ω 푅 = 25696.01 Ω 퐿 = 0.9161 푛퐻 퐶 = 0.7193 푝퐹

Appendix B Recommended Layout for Resonator

These Figures show the recommended pads layout for the dielectric coaxial resonator supplied by IMC.

Appendix C TRL Calibrations

These Figures describe the Through Reflect Line test circuits that were used to calibrate the VNA.

Through

Reflect

Line