Chapter 4. Switch Realization

Home , Buck converter, Power electronics

4.1. Switch applications Single-, two-, and four-quadrant switches. Synchronous rectifiers 4.2. A brief survey of power semiconductor devices Power diodes, MOSFETs, BJTs, IGBTs, and thyristors 4.3. Switching loss Transistor switching with clamped inductive load. Diode recovered charge. Stray capacitances and inductances, and ringing. Efficiency vs. switching frequency. 4.4. Summary of key points

Fundamentals of Power Electronics1 Chapter 4: Switch realization SPST (single-pole single-throw) switches

Buck converter SPST switch, with voltage and current with SPDT switch: L polarities defined 1 iL(t) + 1 2 + i Vg CRV + – v – – with two SPST switches: 0 i A L All power semiconductor A iL(t) + devices function as SPST + vA – – switches. v V + B B CRV g – + iB –

Fundamentals of Power Electronics2 Chapter 4: Switch realization Realization of SPDT switch using two SPST switches

G A nontrivial step: two SPST switches are not exactly equivalent to one SPDT switch G It is possible for both SPST switches to be simultaneously ON or OFF G Behavior of converter is then significantly modified —discontinuous conduction modes (chapter 5) G Conducting state of SPST switch may depend on applied voltage or current —for example: diode

Fundamentals of Power Electronics3 Chapter 4: Switch realization Quadrants of SPST switch operation

1 Switch i on state A single-quadrant + current switch example: v ON-state: i > 0 – OFF-state: v > 0 0 Switch off state voltage

Fundamentals of Power Electronics4 Chapter 4: Switch realization Some basic switch applications

switch switch on-state on-state Single- current current quadrant switch switch switch off-state voltage off-state Current- voltage bidirectional two-quadrant switch

switch switch on-state on-state current current

Voltage- bidirectional switch switch two-quadrant off-state off-state voltage voltage switch Four- quadrant Fundamentals of Power Electronics5 switch Chapter 4: Switch realization 4.1.1. Single-quadrant switches

1 Active switch: Switch state is controlled exclusively i + by a third terminal (control terminal). v Passive switch: Switch state is controlled by the applied current and/or voltage at terminals 1 and 2. – SCR: A special case — turn-on transition is active, 0 while turn-off transition is passive. Single-quadrant switch: on-state i(t) and off-state v(t) are unipolar.

Fundamentals of Power Electronics6 Chapter 4: Switch realization The diode

• A passive switch i • Single-quadrant switch:

1 on • can conduct positive on- + i state current off v v • can block negative off- state voltage – • provided that the intended 0 on-state and off-state operating points lie on the i-v thendiode switch characteristic, can be i-v Symbol characteristic realized using a diode instantaneous

Fundamentals of Power Electronics7 Chapter 4: Switch realization The Bipolar Junction Transistor (BJT) and the Insulated Gate Bipolar Transistor (IGBT)

1 • An active switch, controlled BJT i by terminal C i + C • Single-quadrant switch: v on – • can conduct positive on- off v 0 state current • can block positive off-state 1 IGBT voltage i + C • provided that the intended v on-state and off-state – operating points lie on the 0 i-v characteristic thentransistor switchi-v cancharacteristic, be realized using a BJT or IGBT instantaneous

Fundamentals of Power Electronics8 Chapter 4: Switch realization The Metal-Oxide Semiconductor Field Effect Transistor (MOSFET)

• An active switch, controlled by i terminal C • Normally operated as single- 1 on quadrant switch: i + C off v • can conduct positive on-state v current (can also conduct negative current in some – on 0 (reverse conduction) circumstances) • can block positive off-state voltage

Symbol i-v characteristic • provided that the intended on- instantaneous state and off-state operating points lie on the MOSFETi-v characteristic, then switch can be realized using a MOSFET Fundamentals of Power Electronics9 Chapter 4: Switch realization Realization of switch using transistors and diodes

Buck converter example i A L A iL(t) + + vA – – v V + B B CRV g – + iB –

iA iB Switch A Switch B iL on iL SPST switch on Switch A: transistor operating points Switch A Switch B off off Switch B: diode

Vg vA –Vg vB

Switch A

Fundamentals of Power Electronics10 Chapter 4: Switch realization

Switch B Realization of buck converter using single-quadrant switches

i vA L A +– iL(t) +– vL(t) – V + v g – B +

iA iB Switch A Switch B on iL on iL

Switch A Switch B off off

Vg vA –Vg vB

Fundamentals of Power Electronics11 Chapter 4: Switch realization 4.1.2. Current-bidirectional two-quadrant switches

• Usually an active switch, controlled by terminal C i 1 on • Normally operated as two- i (transistor conducts) quadrant switch: + C off v • can conduct positive or v negative on-state current – on • can block positive off-state 0 (diode conducts) voltage • provided that the intended on- state and off-state operating i-v BJT / anti-parallel i-v points lie on the composite characteristic, then switch can diode realization characteristic instantaneous be realized as shown

Fundamentals of Power Electronics12 Chapter 4: Switch realization Two quadrant switches

switch i on-state on current 1 (transistor conducts) i + v v off switch – off-state voltage 0 on (diode conducts)

Fundamentals of Power Electronics13 Chapter 4: Switch realization MOSFET body diode

i 1 on (transistor conducts) i + v C off v – on (diode conducts) 0 Power MOSFET characteristics Power MOSFET, and its integral body diode Use of external diodes to prevent conduction

Fundamentals of Power Electronics14 Chapterof body 4: Switch diode realization A simple inverter

iA +

Q1 V + D vA v0(t)=(2D –1)V g g – 1 – L iL

+ +

V + D CRv g – 2 vB 0 Q2 – –

Fundamentals of Power Electronics15 Chapter 4: Switch realization Inverter: sinusoidal modulation of D

v0(t)=(2D –1)V g Sinusoidal modulation to v0 produce ac output: V g D(t)=0.5 + D m sin (ωt)

D The resulting inductor 0 0.5 1 current variation is also sinusoidal:

–V v (t) Vg g i (t)= 0 =(2D –1) L R R

Hence, current-bidirectional two-quadrant switches are required.

Fundamentals of Power Electronics16 Chapter 4: Switch realization The dc-3øac voltage source inverter (VSI)

ia + Vg – ib

Switches must block dc input voltage, and conduct ac load current.

Fundamentals of Power Electronics17 Chapter 4: Switch realization Bidirectional battery charger/discharger

D1 L

+ + Q vbus 1 D vbatt spacecraft 2 main power bus Q2 – –

vbus > vbatt

A dc-dc converter with bidirectional power flow.

Fundamentals of Power Electronics18 Chapter 4: Switch realization 4.1.3. Voltage-bidirectional two-quadrant switches

• Usually an active switch, controlled by terminal C

1 i • Normally operated as two- i + on quadrant switch: • can conduct positive on-state v v current C off off (diode (transistor • can block positive or negative blocks voltage) blocks voltage) – off-state voltage 0 • provided that the intended on- state and off-state operating points lie on the compositei-v BJT / series i-v characteristic, then switch can diode realization characteristic be realized as shown instantaneous • The SCR is such a device, without controlled turn-off

Fundamentals of Power Electronics19 Chapter 4: Switch realization Two-quadrant switches

1 i switch + i on-state current v on –

0 v switch 1 off off off-state (diode (transistor voltage i + blocks voltage) blocks voltage)

v C

– 0

Fundamentals of Power Electronics20 Chapter 4: Switch realization A dc-3øac buck-boost inverter

φ iL a +

vab(t) –

φb + V + g – vbc(t) –

φc

Requires voltage-bidirectional two-quadrant switches. Another example: boost-type inverter, or current-source inverter (CSI).

Fundamentals of Power Electronics21 Chapter 4: Switch realization 4.1.4. Four-quadrant switches

switch on-state current • Usually an active switch, controlled by terminal C • can conduct positive or negative on-state current switch off-state • can block positive or negative voltage off-state voltage

Fundamentals of Power Electronics22 Chapter 4: Switch realization Three ways to realize a four-quadrant switch

1 1 1 i i i 1 + + + i + v v v v

– – – – 0 0 0 0

Fundamentals of Power Electronics23 Chapter 4: Switch realization A 3øac-3øac matrix converter

3øac input 3øac output

+ van(t) – vbn(t)

i +

– b

– vcn(t) +

•All voltages and currents are ac; hence, four-quadrant switches are required. • Requires nine four-quadrant switches

Fundamentals of Power Electronics24 Chapter 4: Switch realization 4.1.5. Synchronous rectifiers

Replacement of diode with a backwards-connected MOSFET, to obtain reduced conduction loss

i 1 1 1 on i i (reverse conduction) + + i + C off v v v v – – – on 0 0 0

ideal switch conventional i-v characteristic

diode rectifier MOSFET as synchronous 25 instantaneous Fundamentals of Power Electronicsrectifier Chapter 4: Switch realization Buck converter with synchronous rectifier

•MOSFET Q2 is controlled to turn on i vA L A +– iL(t) when diode would normally conduct Q1 – • Semiconductor + C Vg vB conduction loss can – C + be made arbitrarily Q 2 iB small, by reduction of MOSFET on- resistances •Useful in low-voltage high-current applications

Fundamentals of Power Electronics26 Chapter 4: Switch realization 4.2. A brief survey of power semiconductor devices

G Power diodes G Power MOSFETs G Bipolar Junction Transistors (BJTs) G Insulated Gate Bipolar Transistors (IGBTs) G Thyristors (SCR, GTO, MCT)

G On resistance vs. breakdown voltage vs. switching times G Minority carrier and majority carrier devices

Fundamentals of Power Electronics27 Chapter 4: Switch realization 4.3.1. Transistor switching with clamped inductive load

transistor waveforms v L V i A i (t) g A +– L vA(t) iL physical i (t) MOSFET – A + ideal Vg vB

– 00 – + diode gate + t i DT T driver i L s s B diode waveforms i (t) Buck converter example 00B t vB(t) vB(t)=vA(t)–Vg

i (t)+i (t)=i –Vg A B L transistor turn-off

transition p (t) A Vg iL = vA iA area W 1 off Woff = 2 VgiL (t 2 – t 0)

t t0 t1 t2 Fundamentals of Power Electronics28 Chapter 4: Switch realization Switching loss induced by transistor turn-off transition

Energy lost during transistor turn-off transition:

1 Woff = 2 VgiL (t 2 – t 0)

Similar result during transistor turn-on transition. Average power loss: 1 Psw = pA(t) dt =(Won + Woff ) fs Ts switching transitions

Fundamentals of Power Electronics29 Chapter 4: Switch realization 4.2.1. Power diodes

A power diode, under reverse-biased conditions:

v + –

low doping concentration {

pn- n

+ –

+ – E + – – v + { depletion region, reverse-biased

Fundamentals of Power Electronics30 Chapter 4: Switch realization Typical diode switching waveforms

v(t)

i(t)

tr 0 t di dt

area –Qr (1) (2) (3) (4) (5) (6)

Fundamentals of Power Electronics31 Chapter 4: Switch realization Forward-biased power diode

i – +

conductivity{ modulation

pn- n

+ + – + + – + + –

minority carrier injection

Fundamentals of Power Electronics32 Chapter 4: Switch realization Charge-controlled behavior of the diode

i – The diode equation: +

λv(t) q(t)=Q0 e –1 pn- n Charge control equation: + + dq(t) q(t) + + + = i(t)– + dt τL + +

With: } Total stored minority charge q λ= 1/(26 mV) at 300 K In equilibrium: dq/dt = 0, and hence τL = minority carrier lifetime

(above equations don’t q(t) Q0 λv(t) λv(t) i(t)= = e –1 = I 0 e –1 include current that τL τL charges depletion region capacitance) Fundamentals of Power Electronics33 Chapter 4: Switch realization Charge-control in the diode: Discussion

•The familiar i–v curve of the diode is an equilibrium relationship that can be violated during transient conditions •During the turn-on and turn-off switching transients, the current deviates substantially from the equilibrium i–v curve, because of change in the stored charge and change in the charge within the reverse-bias depletion region • Under forward-biased conditions, the stored minority charge causes “conductivity modulation” of the resistance of the lightly-doped n– region, reducing the device on-resistance

Fundamentals of Power Electronics34 Chapter 4: Switch realization Diode in OFF state: reversed-biased, blocking voltage

v(t) v + – t

pn– n

E i(t) – v + 0 { Depletion region, reverse-biased t

• Diode is reverse-biased • No stored minority charge: q = 0 (1) • Depletion region blocks applied reverse voltage; charge is stored in capacitance of depletion region

Fundamentals of Power Electronics35 Chapter 4: Switch realization Turn-on transient

v(t) The current i(t) is determined by the converter circuit. This t current supplies: Diode conducts with low on-resistance •charge to increase Diode is forward-biased. Supply minority charge to n– region to reduce on-resistance voltage across depletion region Charge depletion region •charge needed to i(t) support the on-state On-state current determined by converter circuit current •charge to reduce t on-resistance of n– (1) (2) region

Fundamentals of Power Electronics36 Chapter 4: Switch realization Turn-off transient

i (< 0) – +

pn- n

+ + + + +

+ + +

}

Removal of stored minority charge q

Fundamentals of Power Electronics37 Chapter 4: Switch realization Diode turn-off transient continued

v(t)

(4) Diode remains forward-biased. Remove stored charge in n– region (5) Diode is reverse-biased. i(t) Charge depletion region capacitance.

tr 0 t di dt

Area –Qr (1) (2) (3) (4) (5) (6)

Fundamentals of Power Electronics38 Chapter 4: Switch realization The diode switching transients induce switching loss in the transistor

v L i (t) see Section 4.3.2 i A i (t) A A +– L transistor waveforms Q fast r V transistor – g + silicon i Vg vB v (t) L

– A – + diode + 0 0 i B t

i (t) diode B waveforms iL vB(t) • Diode recovered stored charge 0 0 Qr flows through transistor t area –Q during transistor turn-on r –Vg transition, inducing switching loss

tr • Qr depends on diode on-state pA(t) forward current, and on the = vA iA rate-of-change of diode current area ~QrVg during diode turn-off transition area ~iLVgtr t t t t Fundamentals of Power Electronics39 0 1 2 Chapter 4: Switch realization Switching loss calculation

iA(t) Energy lost in transistor: transistor waveforms Q r Soft-recovery V g diode: iL vA(t) WD = vA(t) iA(t) dt (t – t ) >> (t – t ) 0 0 2 1 1 0 switching transition t Abrupt-recovery i (t) diode B diode: With abrupt-recovery diode: waveforms iL vB(t) 0 0 (t2 – t1) << (t1 – t0) t WD ≈ Vg (iL – iB(t)) dt area –Q –V switching r g transition

= Vg iL tr + Vg Qr tr

pA(t) = vA iA • Often, this is the largest area ~Q V component of switching loss r g area ~iLVgtr t t0 t1 t2 Fundamentals of Power Electronics40 Chapter 4: Switch realization Types of power diodes

Standard recovery Reverse recovery time not specified, intended for 50/60Hz Fast recovery and ultra-fast recovery Reverse recovery time and recovered charge specified Intended for converter applications Schottky diode A majority carrier device Essentially no recovered charge Model with equilibrium i-v characteristic, in parallel with depletion region capacitance Restricted to low voltage (few devices can block 100V or more)

Fundamentals of Power Electronics41 Chapter 4: Switch realization Characteristics of several commercial power rectifier diodes

Part number Rated max voltage Rated avg current V F (typical) tr (max) Fast recovery rectifiers 1N3913 400V 30A 1.1V 400ns SD453N25S20PC 2500V 400A 2.2V 2µs Ultra-fast recovery rectifiers MUR815 150V 8A 0.975V 35ns MUR1560 600V 15A 1.2V 60ns RHRU100120 1200V 100A 2.6V 60ns Schottky rectifiers MBR6030L 30V 60A 0.48V 444CNQ045 45V 440A 0.69V 30CPQ150 150V 30A 1.19V

Fundamentals of Power Electronics42 Chapter 4: Switch realization Paralleling diodes

Attempts to parallel diodes, and share the i current so that i1 = i2 = i/2, generally don’t work. i1 i2 + + Reason: thermal instability caused by temperature dependence of the diode equation. v1 v2 Increased temperature leads to increased current, or reduced voltage. – – One diode will hog the current. To get the diodes to share the current, heroic measures are required: •Select matched devices •Package on common thermal substrate •Build external circuitry that forces the currents to balance

Fundamentals of Power Electronics43 Chapter 4: Switch realization Ringing induced by diode stored charge see Section 4.3.3

L vi(t) iL(t) V1 +– t v (t) iB(t) L + 0 –V v (t) + silicon v (t) 2 i – diode B C – iL(t)

•Diode is forward-biased while iL(t) > 0 0 • Negative inductor current removes diode area t – Qr stored charge Qr v (t) •When diode becomes reverse-biased, B t negative inductor current flows through 0 capacitor C. –V2 • Ringing of L-C network is damped by t t t parasitic losses. Ringing energy is lost. 1 2 3

Fundamentals of Power Electronics44 Chapter 4: Switch realization Energy associated with ringing

t3 vi(t) V1 Recovered charge is Qr =– i L(t) dt t 2 t 0

Energy stored in inductor during interval –V2 t ≤ t ≤ t 2 3 : t3 W = v (t) i (t) dt L L L iL(t) t2 Applied inductor voltage during interval t ≤ t ≤ t 2 3 : 0 diL(t) area t vL(t)=L =–V 2 dt – Qr Hence, v (t) t3 t3 B diL(t) t WL = L iL(t) dt =(–V2) iL(t) dt 0 t2 dt t2

–V 1 2 2 W L = 2 LiL(t 3)=V 2 Qr

t1 t2 t3

Fundamentals of Power Electronics45 Chapter 4: Switch realization 4.2.2. The Power MOSFET

Source •Gate lengths Gate approaching one micron • Consists of many n nnn small enhancement- ppmode parallel- connected MOSFET cells, covering the n- surface of the silicon wafer n •Vertical current flow •n-channel device is shown Drain

Fundamentals of Power Electronics46 Chapter 4: Switch realization MOSFET: Off state

source – • p-n- junction is reverse-biased •off-state voltage n nnn appears across n- ppregion

depletion region n-

drain +

Fundamentals of Power Electronics47 Chapter 4: Switch realization MOSFET: on state

- source • p-n junction is slightly reverse- biased • positive gate voltage n nnn induces conducting ppchannel •drain current flows channel through n- region n- and conducting channel n • on resistance = total resistances of n- region, conducting drain drain current channel, source and drain contacts, etc.

Fundamentals of Power Electronics48 Chapter 4: Switch realization MOSFET body diode

- source • p-n junction forms an effective diode, in parallel with the channel n p nnn • negative drain-to- p source voltage can forward-bias the Body diode body diode n- •diode can conduct the full MOSFET rated current n • diode switching speed not optimized drain —body diode is slow, Qr is large

Fundamentals of Power Electronics49 Chapter 4: Switch realization Typical MOSFET characteristics

• Off state: VGS < Vth V >> V 10A = 10V •On state: GS th = 200V

DS DS V V = 2V V DS •MOSFET can ID conduct peak currents well in on state excess of average 5A current rating = 1V VDS —characteristics are unchanged off V = 0.5V state DS • on-resistance has positive temperature 0A coefficient, hence 0V 5V 10V 15V easy to parallel VGS

Fundamentals of Power Electronics50 Chapter 4: Switch realization A simple MOSFET equivalent circuit

D • Cgs : large, essentially constant

• Cgd : small, highly nonlinear Cgd • Cds : intermediate in value, highly G nonlinear Cds •switching times determined by rate Cgs at which gate driver charges/discharges Cgs and Cgd

' C0 V0 C 0 Cds(vds)= Cds(vds) ≈ C 0 = v vds v 1+ ds ds V0

Fundamentals of Power Electronics51 Chapter 4: Switch realization Switching loss caused by semiconductor output capacitances

Buck converter example

Cds

+ Vg Cj –

– +

Energy lost during MOSFET turn-on transition (assuming linear capacitances):

1 2 W C = 2 (C ds + C j) V g

Fundamentals of Power Electronics52 Chapter 4: Switch realization MOSFET nonlinear Cds

Approximate dependence of incremental Cds on vds :

' V0 C 0 Cds(vds) ≈ C 0 = vds vds

Energy stored in Cds at vds = VDS :

VDS WCds = vds iC dt = vds C ds(vds) dvds 0

VDS ' 2 2 WCds = C 0(vds) vds dvds = 3 Cds(VDS) V DS 0

4 — same energy loss as linear capacitor having value 3 Cds(VDS)

Fundamentals of Power Electronics53 Chapter 4: Switch realization Characteristics of several commercial power MOSFETs

Part number Rated max voltage Rated avg current R on Qg (typical) IRFZ48 60V 50A 0.018Ω 110nC IRF510 100V 5.6A 0.54Ω 8.3nC IRF540 100V 28A 0.077Ω 72nC APT10M25BNR 100V 75A 0.025Ω 171nC IRF740 400V 10A 0.55Ω 63nC MTM15N40E 400V 15A 0.3Ω 110nC APT5025BN 500V 23A 0.25Ω 83nC APT1001RBNR 1000V 11A 1.0Ω 150nC

Fundamentals of Power Electronics54 Chapter 4: Switch realization MOSFET: conclusions

G A majority-carrier device: fast switching speed G Typical switching frequencies: tens and hundreds of kHz G On-resistance increases rapidly with rated blocking voltage G Easy to drive G The device of choice for blocking voltages less than 500V G 1000V devices are available, but are useful only at low power levels (100W) G Part number is selected on the basis of on-resistance rather than current rating

Fundamentals of Power Electronics55 Chapter 4: Switch realization 4.2.3. Bipolar Junction Transistor (BJT)

Base Emitter •Interdigitated base and emitter contacts •Vertical current flow • npn device is shown n n n p •minority carrier device

n- • on-state: base-emitter and collector-base junctions are both n forward-biased • on-state: substantial minority charge in p and n- regions, conductivity Collector modulation

Fundamentals of Power Electronics56 Chapter 4: Switch realization BJT switching times

vs(t) Vs2

–Vs1

VCC vBE(t)

0.7V

RL –Vs1 iC(t) + iB(t) RB iB(t) IB1 vCE(t) + 0 – vBE(t) –IB2 + vs(t) – – vCE(t)

VCC

IConRon

iC(t) ICon

0 t (1) (2) (3) (4) (5) (6) (7) (8) (9)

Fundamentals of Power Electronics57 Chapter 4: Switch realization Ideal base current waveform

iB(t) IB1

IBon 0 t

–IB2

Fundamentals of Power Electronics58 Chapter 4: Switch realization Current crowding due to excessive IB2

Base Emitter

–IB2

n can lead to p – – p + + – – – – formation of hot spots and device n- failure

Collector

Fundamentals of Power Electronics59 Chapter 4: Switch realization BJT characteristics

• Off state: IB = 0 IC V = 200V CE I I β V = 20V •On state: B > C / 10A active region CE = 5V • Current gain β decreases quasi-saturation VCE rapidly at high current. Device should not be operated at slope saturation region = β instantaneous currents V = 0.5V 5A CE exceeding the rated value

cutoff VCE = 0.2V

0A 0V 5V 10V 15V

Fundamentals of Power Electronics60 Chapter 4: Switch realization Breakdown voltages

BV I CBO: avalanche breakdown C voltage of base-collector increasing I B junction, with the emitter open-circuited

BVCEO: collector-emitter breakdown voltage with zero

IB = 0 base current

open emitter BVsus: breakdown voltage observed with positive base

BVsus BVCEO BVCBO VCE current In most applications, the off- state transistor voltage must not exceed BVCEO.

Fundamentals of Power Electronics61 Chapter 4: Switch realization Darlington-connected BJT

•Increased current gain, for high-voltage applications Q1 •In a monolithic Darlington device, Q Q Q transistors 1 and 2 are integrated on the 2 same silicon wafer

• Diode D1 speeds up the turn-off process, by allowing the base driver to actively D1 remove the stored charge of both Q1 and Q2 during the turn-off transition

Fundamentals of Power Electronics62 Chapter 4: Switch realization Conclusions: BJT

G BJT has been replaced by MOSFET in low-voltage (<500V) applications G BJT is being replaced by IGBT in applications at voltages above 500V G A minority-carrier device: compared with MOSFET, the BJT exhibits slower switching, but lower on-resistance at high voltages

Fundamentals of Power Electronics63 Chapter 4: Switch realization 4.2.4. The Insulated Gate Bipolar Transistor (IGBT)

Emitter •A four-layer device

Gate •Similar in construction to MOSFET, except extra p region

n nnn •On-state: minority carriers ppare injected into n- region, leading to conductivity minority carrier modulation n- injection •compared with MOSFET: p slower switching times, lower on-resistance, useful at higher voltages (up to Collector 1700V)

Fundamentals of Power Electronics64 Chapter 4: Switch realization The IGBT

Symbol collector

gate

emitter Location of equivalent devices C n p nnn Equivalent p circuit i i2 1

n- G p i1 i2

E Fundamentals of Power Electronics65 Chapter 4: Switch realization Current tailing in IGBTs

IGBT waveforms Vg vA(t) iL iA(t) C current tail

00} t iL diode waveforms i (t) 00B t v (t) G B

–Vg i1 i2

p (t) A Vg iL = v i E A A

area Woff t t0 t1 t2 t3

Fundamentals of Power Electronics66 Chapter 4: Switch realization Switching loss due to current-tailing in IGBT

IGBT waveforms i vA L Vg A iL(t) +– vA(t) iL physical iA(t) IGBT – current tail + ideal Vg vB –

– + } diode 00 gate + t DTs Ts driver iB iL diode waveforms Example: buck converter with IGBT i (t) 00B t vB(t)

–Vg transistor turn-off

p (t) transition A Vg iL = v i 1 A A Psw = pA(t) dt =(Won + Woff ) fs Ts switching transitions area Woff t t0 t1 t2 t3 Fundamentals of Power Electronics67 Chapter 4: Switch realization Characteristics of several commercial devices

Part number Rated max voltage Rated avg current V F (typical) t f (typical)

Single-chip dev ices HGTG32N60E2 600V 32A 2.4V 0.62µs HGTG30N120D2 1200V 30A 3.2A 0.58µs

Multiple-chip power modules CM400HA-12E 600V 400A 2.7V 0.3µs CM300HA-24E 1200V 300A 2.7V 0.3µs

Fundamentals of Power Electronics68 Chapter 4: Switch realization Conclusions: IGBT

G Becoming the device of choice in 500 to 1700V+ applications, at power levels of 1-1000kW G Positive temperature coefficient at high current —easy to parallel and construct modules G Forward voltage drop: diode in series with on-resistance. 2-4V typical G Easy to drive —similar to MOSFET G Slower than MOSFET, but faster than Darlington, GTO, SCR G Typical switching frequencies: 3-30kHz G IGBT technology is rapidly advancing:

G 3300 V devices: HVIGBTs

G 150 kHz switching frequencies in 600 V devices

Fundamentals of Power Electronics69 Chapter 4: Switch realization 4.2.5. Thyristors (SCR, GTO, MCT)

The SCR

symbol equiv circuit K G K Anode (A) Anode construction n n p Q 2 Q Gate (G) 1 - n Q2 Gate Cathode (K) Q1 p

Cathode A

Fundamentals of Power Electronics70 Chapter 4: Switch realization The Silicon Controlled Rectifier (SCR)

•Positive feedback —a latching device iA forward •A minority carrier device conducting • Double injection leads to very low on-resistance, hence low forward iG = 0 voltage drops attainable in very high increasing iG voltage devices reverse forward vAK •Simple construction, with large blocking blocking feature size • Cannot be actively turned off reverse breakdown •A voltage-bidirectional two-quadrant switch • 5000-6000V, 1000-2000A devices

Fundamentals of Power Electronics71 Chapter 4: Switch realization Why the conventional SCR cannot be turned off via gate control

K G K • Large feature size –iG • Negative gate current

induces lateral voltage n – n

– – +

drop along gate-cathode – – – p junction n- •Gate-cathode junction becomes reverse-biased p only in vicinity of gate contact iA

Fundamentals of Power Electronics72 Chapter 4: Switch realization The Gate Turn-Off Thyristor (GTO)

•An SCR fabricated using modern techniques —small feature size •Gate and cathode contacts are highly interdigitated • Negative gate current is able to completely reverse-bias the gate- cathode junction

Turn-off transition: •Turn-off current gain: typically 2-5 •Maximum controllable on-state current: maximum anode current that can be turned off via gate control. GTO can conduct peak currents well in excess of average current rating, but cannot switch off

Fundamentals of Power Electronics73 Chapter 4: Switch realization The MOS-Controlled Thyristor (MCT)

Anode •Still an emerging device, but some Gate devices are commercially available

•p-type device n p n •A latching SCR, with n Q channel added built-in 3 p- MOSFETs to assist the Q4 channel turn-on and turn-off processes n •Small feature size, highly interdigitated, Cathode modern fabrication

Fundamentals of Power Electronics74 Chapter 4: Switch realization The MCT: equivalent circuit

Cathode • Negative gate-anode voltage turns p-channel MOSFET Q3 on, causing Q1 and Q2 to latch ON Q1 •Positive gate-anode voltage turns n-channel Q4 Q3 MOSFET Q4 on, reverse- Q Gate 2 biasing the base-emitter junction of Q2 and turning off the device •Maximum current that can be interrupted is limited by Anode the on-resistance of Q4

Fundamentals of Power Electronics75 Chapter 4: Switch realization Summary: Thyristors

•The thyristor family: double injection yields lowest forward voltage drop in high voltage devices. More difficult to parallel than MOSFETs and IGBTs •The SCR: highest voltage and current ratings, low cost, passive turn-off transition •The GTO: intermediate ratings (less than SCR, somewhat more than IGBT). Slower than IGBT. Slower than MCT. Difficult to drive. •The MCT: So far, ratings lower than IGBT. Slower than IGBT. Easy to drive. Second breakdown problems? Still an emerging device.

Fundamentals of Power Electronics76 Chapter 4: Switch realization 4.3. Switching loss

•Energy is lost during the semiconductor switching transitions, via several mechanisms: •Transistor switching times • Diode stored charge •Energy stored in device capacitances and parasitic inductances •Semiconductor devices are charge controlled •Time required to insert or remove the controlling charge determines switching times

Fundamentals of Power Electronics77 Chapter 4: Switch realization 4.3.1. Transistor switching with clamped inductive load

transistor waveforms v L V i A i (t) g A +– L vA(t) iL physical i (t) MOSFET – A + ideal Vg vB

– 00 – + diode gate + t i DT T driver i L s s B diode waveforms i (t) Buck converter example 00B t vB(t) vB(t)=vA(t)–Vg

i (t)+i (t)=i –Vg A B L transistor turn-off

transition p (t) A Vg iL = vA iA area W 1 off Woff = 2 VgiL (t 2 – t 0)

t t0 t1 t2 Fundamentals of Power Electronics78 Chapter 4: Switch realization Switching loss induced by transistor turn-off transition

Energy lost during transistor turn-off transition:

1 Woff = 2 VgiL (t 2 – t 0)

Similar result during transistor turn-on transition. Average power loss: 1 Psw = pA(t) dt =(Won + Woff ) fs Ts switching transitions

Fundamentals of Power Electronics79 Chapter 4: Switch realization Switching loss due to current-tailing in IGBT

IGBT waveforms i vA L Vg A iL(t) +– vA(t) iL physical iA(t) IGBT – current tail + ideal Vg vB –

– + } diode 00 gate + t DTs Ts driver iB iL diode waveforms Example: buck converter with IGBT i (t) 00B t vB(t)

–Vg transistor turn-off

p (t) transition A Vg iL = v i 1 A A Psw = pA(t) dt =(Won + Woff ) fs Ts switching transitions area Woff t t0 t1 t2 t3 Fundamentals of Power Electronics80 Chapter 4: Switch realization 4.3.2. Diode recovered charge

v L i (t) i A i (t) A A +– L transistor waveforms Q fast r V transistor – g + silicon i Vg vB v (t) L

– A – + diode + 0 0 i B t

i (t) diode B waveforms iL vB(t) • Diode recovered stored charge 0 0 Qr flows through transistor t area during transistor turn-on –Q r –Vg transition, inducing switching loss

tr • Qr depends on diode on-state pA(t) forward current, and on the = vA iA rate-of-change of diode current area ~QrVg during diode turn-off transition area ~iLVgtr t t t t Fundamentals of Power Electronics81 0 1 2 Chapter 4: Switch realization Switching loss calculation

= Vg iL tr + Vg Qr tr

pA(t) = vA iA • Often, this is the largest area ~Q V component of switching loss r g area ~iLVgtr t t0 t1 t2 Fundamentals of Power Electronics82 Chapter 4: Switch realization 4.3.3. Device capacitances, and leakage, package, and stray inductances

•Capacitances that appear effectively in parallel with switch elements are shorted when the switch turns on. Their stored energy is lost during the switch turn-on transition. •Inductances that appear effectively in series with switch elements are open-circuited when the switch turns off. Their stored energy is lost during the switch turn-off transition. Total energy stored in linear capacitive and inductive elements:

1 2 1 2 WC = CiV i WL = L jI j capacitiveΣ 2 inductiveΣ 2 elements elements

Fundamentals of Power Electronics83 Chapter 4: Switch realization Example: semiconductor output capacitances

Buck converter example

Cds

+ Vg Cj –

– +

Energy lost during MOSFET turn-on transition (assuming linear capacitances):

1 2 W C = 2 (C ds + C j) V g

Fundamentals of Power Electronics84 Chapter 4: Switch realization MOSFET nonlinear Cds

Approximate dependence of incremental Cds on vds :

' V0 C 0 Cds(vds) ≈ C 0 = vds vds

Energy stored in Cds at vds = VDS :

VDS WCds = vds iC dt = vds C ds(vds) dvds 0

VDS ' 2 2 WCds = C 0(vds) vds dvds = 3 Cds(VDS) V DS 0

4 — same energy loss as linear capacitor having value 3 Cds(VDS)

Fundamentals of Power Electronics85 Chapter 4: Switch realization Some other sources of this type of switching loss

Schottky diode • Essentially no stored charge •Significant reverse-biased junction capacitance Transformer leakage inductance •Effective inductances in series with windings •A significant loss when windings are not tightly coupled Interconnection and package inductances • Diodes •Transistors •A significant loss in high current applications

Fundamentals of Power Electronics86 Chapter 4: Switch realization Ringing induced by diode stored charge

L vi(t) iL(t) V1 +– t v (t) iB(t) L + 0 –V v (t) + silicon v (t) 2 i – diode B C – iL(t)

Fundamentals of Power Electronics87 Chapter 4: Switch realization Energy associated with ringing

t3 vi(t) V1 Recovered charge is Qr =– i L(t) dt t 2 t 0

–V 1 2 2 W L = 2 LiL(t 3)=V 2 Qr

t1 t2 t3

Fundamentals of Power Electronics88 Chapter 4: Switch realization 4.3.4. Efficiency vs. switching frequency

Add up all of the energies lost during the switching transitions of one switching period:

W tot = W on + W off + W D + W C + W L + ...

Average switching power loss is

Psw = W tot fsw

Total converter loss can be expressed as

Ploss = Pcond + P fixed + W tot fsw

where Pfixed = fixed losses (independent of load and fsw) Pcond = conduction losses

Fundamentals of Power Electronics89 Chapter 4: Switch realization Efficiency vs. switching frequency

Ploss = Pcond + P fixed + W tot fsw Switching losses are equal to 100% the other converter losses at the dc asymptote critical frequency f 90% crit Pcond + Pfixed fcrit = Wtot 80%

η This can be taken as a rough 70% upper limit on the switching frequency of a practical converter. For fsw > fcrit, the 60% efficiency decreases rapidly with frequency. 50% 10kHz 100kHz 1MHz

fsw Fundamentals of Power Electronics90 Chapter 4: Switch realization Summary of chapter 4

1. How an SPST ideal switch can be realized using semiconductor devices depends on the polarity of the voltage which the devices must block in the off-state, and on the polarity of the current which the devices must conduct in the on-state. 2. Single-quadrant SPST switches can be realized using a single transistor or a single diode, depending on the relative polarities of the off-state voltage and on-state current. 3. Two-quadrant SPST switches can be realized using a transistor and diode, connected in series (bidirectional-voltage) or in anti-parallel (bidirectional- current). Several four-quadrant schemes are also listed here. 4. A “synchronous rectifier” is a MOSFET connected to conduct reverse current, with gate drive control as necessary. This device can be used where a diode would otherwise be required. If a MOSFET with sufficiently low Ron is used, reduced conduction loss is obtained.

Fundamentals of Power Electronics91 Chapter 4: Switch realization Summary of chapter 4

5. Majority carrier devices, including the MOSFET and Schottky diode, exhibit very fast switching times, controlled essentially by the charging of the device capacitances. However, the forward voltage drops of these devices increases quickly with increasing breakdown voltage. 6. Minority carrier devices, including the BJT, IGBT, and thyristor family, can exhibit high breakdown voltages with relatively low forward voltage drop. However, the switching times of these devices are longer, and are controlled by the times needed to insert or remove stored minority charge. 7. Energy is lost during switching transitions, due to a variety of mechanisms. The resulting average power loss, or switching loss, is equal to this energy loss multiplied by the switching frequency. Switching loss imposes an upper limit on the switching frequencies of practical converters.

Fundamentals of Power Electronics92 Chapter 4: Switch realization Summary of chapter 4

8. The diode and inductor present a “clamped inductive load” to the transistor. When a transistor drives such a load, it experiences high instantaneous power loss during the switching transitions. An example where this leads to significant switching loss is the IGBT and the “current tail” observed during its turn-off transition. 9. Other significant sources of switching loss include diode stored charge and energy stored in certain parasitic capacitances and inductances. Parasitic ringing also indicates the presence of switching loss.

Fundamentals of Power Electronics93 Chapter 4: Switch realization