Investigation of Digital Terrestrial Television Receiver Architectures for DVB-T2 Standard

Warsaw University of Technology Faculty of Electronics and Information Technology

Ph. D. Thesis

Marcin Dąbrowski

Supervisor: Professor dr hab. inż. Józef Modelski

Warsaw, 2013

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I would like to express my deepest grati- tude to Professor Józef Modelski for for- mulation of the subject of this thesis, in- spiration, encouragement, help, amend- ments, many stimulating discussions, and patience.

Marcin Dabrowski ˛

Abbreviations 9

Designations 17

Abstract 21

1 Subject, scope, aims, and scientific thesis 23 1.1 Introduction ...... 23 1.2 Subject, scope, and significance of the work ...... 24 1.3 Aims and scientific thesis ...... 25 1.4 Text structure ...... 25

2 Digital terrestrial television 27 2.1 DTT concept ...... 27 2.2 General DTT transmission scheme ...... 29 2.3 DTT Recommendations ...... 31 2.3.1 Video coding ...... 31 2.3.2 Audio coding ...... 32 2.3.3 Transport layer ...... 33 2.3.4 Physical layer ...... 34 2.3.5 Uniﬁed receiver architecture ...... 36

3 European DVB-T system 37 3.1 DVB history and future trends ...... 37 3.2 DVB-T basics ...... 40 3.3 Source coding and MPEG-2 multiplexing ...... 41 3.4 DVB transmission ...... 42 3.5 Encryption and metadata ...... 42 3.6 DVB Multimedia Home Platform ...... 43

4 Orthogonal frequency division multiplexing 45 4.1 Mathematical relationships ...... 46 4.2 OFDM features ...... 47 4.3 OFDM receiver architecture ...... 48 4.4 Error correction coding ...... 49 4.5 Adaptive OFDM modulation ...... 49 4.6 OFDM based wide area broadcasting ...... 50 4.7 Multiple access OFDM ...... 50

5 Single frequency network technology 51 5.1 SFN principle ...... 51 5.2 Wideband digital broadcasting ...... 52 5.3 SFN gain and self-interference ...... 55

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6 Overview and diﬀerences between DVB-T and DVB-T2 59 6.1 DVB-T2 overview ...... 59 6.2 DVB-T and DVB-T2 technical comparison ...... 61 6.2.1 Transport layer capabilities and DVB-T2 gateways . . 61 6.2.2 Null packet deletion mechanism ...... 63 6.2.3 Modulation ...... 63 6.2.4 Constellation rotation ...... 64 6.2.5 PAPR reduction ...... 64 6.2.6 Pilot sub-carriers ...... 67 6.2.7 Signal bandwidth ...... 67 6.2.8 MISO ...... 68 6.2.9 Time-Frequency Slicing ...... 68 6.2.10 Channel coding ...... 69 6.2.11 Symbol organization, framing and signaling ...... 70 6.3 Beneﬁts of using DVB-T2 for broadcasters and network operators ...... 70 6.3.1 Higher bandwidth ...... 70 6.3.2 Flexibility ...... 71 6.3.3 Usage of available channels ...... 71 6.4 Next steps of DVB-T2 standard implementation ...... 71

7 Propagation models and coverage calculations 73 7.1 Deﬁnitions and assumptions ...... 73 7.1.1 Mobile channels ...... 75 7.1.2 Log-normal distribution ...... 76 7.1.3 Thermal noise and its inﬂuence on the receiver . . . . 77 7.2 Field strength summation procedures ...... 78 7.2.1 Simple power-sum method ...... 78 7.2.2 t-LNM method ...... 79 7.3 ITU-R P.370 and ITU-R P.1546 models ...... 79 7.4 Other models ...... 82

8 Specifications for receivers 83 8.1 Role and need of specifications ...... 83 8.2 Common guidelines of specifications ...... 83 8.2.1 Receiver performance ...... 83 8.2.2 Common receiver model ...... 85 8.2.3 Propagation channel assumptions ...... 86 8.2.4 Image channel issue ...... 86 8.3 ETSI EN 300 744 norm ...... 87 8.4 Chester’97 Coordination Agreement ...... 88 8.5 Final acts off RRC’06, GE’06 coordination agreement . . . . 89 8.6 IEC 62216 norm ...... 89 8.7 NorDIG specification ...... 92 8.8 DGTVi D-Book specification and AGCOM 216/00 Resolution 95

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8.9 PolSpec ...... 95

9 Receiver architectures 97 9.1 Digital TV receiver elements and interfaces ...... 97 9.2 Tuners ...... 98 9.2.1 Adjacent channel signals rejection ...... 101 9.2.2 Image channel signals rejection ...... 101 9.2.3 Direct conversion tuners ...... 103 9.2.4 Local oscillators ...... 104 9.2.5 Surface acoustic wave filters ...... 104 9.2.6 Tuner parameters ...... 105 9.3 Demodulators ...... 106 9.3.1 Channel estimation and equalization ...... 106 9.4 Diversity, MISO, and MIMO ...... 107 9.5 SFN influence on receiver operation and FFT window positioning ...... 108 9.5.1 Symbol Timing Recovery (STR) and Symbol Timing Offset (STO) ...... 111 9.5.2 Impairments within receivers ...... 112 9.5.3 LDPC decoding ...... 112 9.6 CD3 algorithm ...... 113 9.7 Diversity receivers ...... 113 9.8 Recommended DVB-T2 receiver architecture ...... 114

10 Digital terrestrial television in Poland — current state and future trends 115 10.1 Beginning of DVB-T in Poland ...... 115 10.2 DVB-T in progress ...... 115 10.3 DVB-T measurements ...... 116 10.4 DVB-T future trends ...... 119

11 Conclusions 121

Bibliography 125

128-QAM 128-state Quadrature Amplitude Modulation 16-APSK 16-state Amplitude and Phase-Shift Keying 16-PSK 16-state Phase-Shift Keying 16-QAM 16-state Quadrature Amplitude Modulation 256-QAM 256-state Quadrature Amplitude Modulation 2D Two-dimensional picture quality 32-APSK 32-state Amplitude and Phase-Shift Keying 32-QAM 32-state Quadrature Amplitude Modulation 3D Three-dimensional picture quality 3DTV Three-dimensional Television 64-QAM 64-state Quadrature Amplitude Modulation 8-PSK 8-state Phase-Shift Keying 8-VSB 8-level Vestigial Sideband Modulation ABC American Broadcasting Company AC-3 Audio Codec Dolby Digital 3rd generation ACE Active Constellation Extension AD Analog to Digital (conversion) ADSL Asymmetric Digital Subscriber Line AGC Automatic Gain Control AM Amplitude Modulation APSK Amplitude and Phase-Shift Keying ASI Asynchronous Serial Interface ASO Analog Switch-Oﬀ ATSC Advanced Television Standards Committee AVC Advanced Video Coding AWGN Additive White Gaussian Noise channel BBC British Broadcasting Corporation BBFRAME Baseband Frame BCH Bose-Chaudhuri cyclic error-correcting codes BER Bit Error Rate BiCMOS Bipolar Complementary Metal-Oxide-Semiconductor C Central audio channel CBC Canadian Broadcasting Corporation CCIR Comité Consultatif International des Radiocommunications or Comité Consultatif International pour la Radio CCITT Consultative Committee for International Telephony and Telegraphy

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CD3 (or CDDD) Coded Decision-Directed Demodulation CDMA Code Division Multiple Access CENELEC European Committee for Electrotechnical Standardization CFO Carrier Frequency Oﬀset CFR Channel Frequency Response CIR Channel Impulse Response CM-3DTV Commercial Module for Three-dimensional Television CMOS Complementary Metal-Oxide-Semiconductor COFDM Coded Orthogonal Frequency Division Multiplexing COST European Cooperation in Science and Technology DA Digital to Analog (conversion) DAB Digital Audio Broadcasting DCA Dynamic Channel Allocation DCII DigiCipher 2 DECT Digital Enhanced Cordless Telecommunications DGTVi Associazione per la Televisione Digitale Terrestre Interattiva DFT Discrete Fourier Transformation DMB-T/H Digital Multimedia Broadcast – Terrestrial / Handheld DMT Discrete Multitone modulation DNP Deleted Null Packet DSFN Dynamic Single Frequency Network DSL Digital Subscriber Line DSP Digital Signal Processor DTMB Digital Terrestrial Multimedia Broadcast DTG Digital TV Group DTT (DTTV) Digital Terrestrial Television DTTB Digital Terrestrial Television Broadcasting DVB Digital Video Broadcasting DVB-BCG Digital Video Broadcasting – Broadband Content Guide DVB-C Digital Video Broadcasting – Cable DVB-C2 Digital Video Broadcasting – Cable (second generation) DVB-CA Digital Video Broadcasting – Conditional Access DVB-CI Digital Video Broadcasting – Common Interface DVB-CPCM Digital Video Broadcasting – Content Protection and Copy Management DVB-CSA Digital Video Broadcasting – Common Scrambling Algorithm DVB-DATA Digital Video Broadcasting – Data

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DVB-H Digital Video Broadcasting – Handheld DVB-IPI Digital Video Broadcasting – Internet Protocol Interface DVB-IPTV Digital Video Broadcasting – Internet Protocol Television DVB-MC Digital Video Broadcasting – Microwave Cable DVB-MHP Digital Video Broadcasting – Multimedia Home Platform DVB-MS Digital Video Broadcasting – Microwave Satellite DVB-MT Digital Video Broadcasting – Microwave Terrestrial DVB-NPI Digital Video Broadcasting – Network Protocol Independent DVB-RCS Digital Video Broadcasting – Return Channel Satellite DVB-RC Digital Video Broadcasting – Return Channel DVB-RCT Digital Video Broadcasting – Return Channel Terrestrial DVB-S Digital Video Broadcasting – Satellite DVB-S2 Digital Video Broadcasting – Satellite (second generation) DVB-SI Digital Video Broadcasting – Service Information DVB-SH Digital Video Broadcasting – Satellite services to Handhelds DVB-SMATV Digital Video Broadcasting – Satellite Master-Antenna Television DVB-SUB Digital Video Broadcasting – Subtitling DVB-T Digital Video Broadcasting – Terrestrial DVB-T2 Digital Video Broadcasting – Terrestrial (second generation) DVB-TVA Digital Video Broadcasting – TV-Anytime DVB-TXT Digital Video Broadcasting – Teletext DVB-VBI Digital Video Broadcasting – Vertical Blanking Interval DVD Digital Video Disc or Digital Versatile Disc DVR Digital Video Recorder EBU European Broadcasting Union EIT Event Information Table EPG Electronic Program Guide EPT Eﬀective Protection Target

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ETSI European Telecommunications Standards Institute EVM Error Vector Magnitude FCA Fixed Channel Allocation FCC Federal Communications Commission FDMA Frequency Division Multiple Access FEC Forward Error Correction FEF Future Extension Frame FFT Fast Fourier Transformation FIR Finite Impulse Response FM Frequency Modulation FTA Free-To-Air GCS Generic Continuous Stream GFPS Generic Fixed-length Packetized Stream GI Guard Interval GPS Global Positioning System GS Generic Stream GSE Generic Stream Encapsulated GSM Groupe Spécial Mobile (primarily) Global System for Mobile Communications (nowadays) H.264 MPEG-4 part 10 or AVC standard HD(TV) High Deﬁnition (Television) HP (TS) High Priority (transport stream) I In-phase IBC International Broadcasting Convention ICI Inter-Carrier Interference IDFT Inverse Discrete Fourier Transformation IEC International Electrotechnical Commission IF Intermediate Frequency IFFT Inverse Fast Fourier Transformation (conceptually equivalent to FFT) IIP3 Third order Input Intercept Point IL Implementation Loss IP Internet Protocol IPTV Internet Protocol Television IP3 Third order Intercept Point IRD Integrated Receiver / Decoder ISDB-T Integrated Services Digital Broadcasting — Terrestrial ISDN Integrated Services Digital Network ISI Inter-Symbol Interference ITU International Telecommunications Union ITU-R International Telecommunication Union – Radiocommunication Sector ITV Interactive Television JTC Joint Technical Committee

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KIGEiT Krajowa Izba Gospodarcza Elektroniki i Telekomunikacji L Left audio channel LDPC Low Density Parity Check codes LFE Low Frequency Enhancement LNA Low Noise Ampliﬁer LNM Log-Normal Method LO Local Oscillator LP (TS) Low Priority (transport stream) LUT Look-Up Table MAC Media Access Control MC-CDMA Multi-Carrier Code Division Multiple Access MER Modulation Error Ratio MFN Multi-Frequency broadcast Network MHP Multimedia Home Platform MIMO Multiple Input Multiple Output MIP Mega-frame Initialization Packet MISO Multiple Input Single Output MLE Maximum Likelihood Estimation MMDS Multichannel Multipoint Distribution Service MPEG Moving Pictures Expert Group MPEG-1 MPEG 1st generation standard MPEG-2 MPEG 2nd generation standard MPEG-4 MPEG 4th generation standard MPEG-TS MPEG – Transport Stream MRC Maximal-Ratio Combining MUSICAM Masking pattern adapted Universal Subband Integrated Coding And Multiplexing audio codec MUX Multiplexer MVDS Multipoint Video Distribution System NBC National Broadcasting Company NF Noise Figure NorDig Nordic DVB-T2 system NTSC National Television Standards Committee OIP3 Third order Output Intercept Point OFDM Orthogonal Frequency Division Multiplexing OFDMA Orthogonal Frequency Division Multiple Access PAL Phase Alternating Line PAPR Peak-to-Average Power Ratio PC Personal Computer PDF Probability Density Function PER Packet Error Ratio PID Packet ID PLL Phase-Locked Loop PLP Physical Layer Pipe

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PP Pilot Pattern PPF Polyphase Filter PPS Pulse Per Second PRBS Pseudo Random Binary Sequence PS Program Stream PSI Program Specific Information PSK Phase-Shift Keying PSN Polskie Sieci Nadawcze PSTN Public Switched Telephone Network Q Quadrature QAM Quadrature Amplitude Modulation QEF Quasi Error Free reception QPSK Quadrature Phase Shift Keying R Right audio channel RA6 Typical Rural reception with 6 paths RC Return Channel RCT Reserved Carrier Technique or Return Channel Terrestrial RCS Return Channel Satellite RF Radio Frequency RPC1, RPC2, RPC3 Reference Planning Configurations RS Reed-Solomon decoder S Single Surround channel SAW Surface Acoustic Wave SD(TV) Standard Definition (Television) SECAM Séquentiel Couleur avec Mémoire SFN Single Frequency Network SHF Super High Frequency SI Service Information SISO Single Input Single Output SL Surround Left channel SMATV Satellite Master-Antenna Television SMPTE Society of Motion Picture and Television Engineers SNR Signal-to-Noise Ratio SPI Synchronous Parallel Interface SR Surround Right channel SSI Signal Strength Indicator or Synchronous Serial Interface STB Set-Top Box STO Symbol Timing Offset STR Symbol Timing Recovery T2-MI DVB-T2 Modulator Interface T2-MIP DVB-T2 Mega-frame Initialization Packet TCA Terrain Clearance Angle TDMA Time Division Multiple Access T-DMB Terrestrial Digital Multimedia Broadcast

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TFS Time Frequency Slicing t-LNM trilinear Log-Normal Method TM-T2 Technical Module on Next Generation DVB-T TPS Transmission Parameters Signaling TR Tone Reservation TS Transport Stream TTL Transistor-Transistor logic TU6 Typical Urban reception with 6 paths TV Television TVP Telewizja Polska S. A. UHF Ultra High Frequency, 470–862 MHz UKE Urzad ˛ Komunikacji Elektronicznej VDSL Very-high-bit-rate Digital Subscriber Line VHF Very High Frequency, 174–230 MHz VSB Vestigial Sideband Modulation WCP-OFDM Weighted Cyclic Preﬁx – Orthogonal Frequency-Division Multiplexing XML Extensible Markup Language ZIF Zero Intermediate Frequency

1TR Vector with all entries equal to 0 except for some ones put under the indexes where the reserved tones are located 5.1 5 standard + 1 LFE channel surround sound A0 Additional protection ratio ai Protection ratio against signal from i-th transmitter Ai i-th interferer protection ratio B Bandwidth B3dB 3 dB filter bandwidth bi Receiving antenna discrimination for the proper angle of arrival c Speed of light in vacuum C 4–8 GHz band satellite channel C/ I Carrier to interferer ratio C/ N Carrier to noise ratio d Propagation path length δn Kronecker delta Δ Duration of OFDM guard intervals Δfn Doppler shift of n-th wave in Rayleigh channel Δf Frequency offset Δt sampling temporal offset Ei Field generated by the i-th transmitter Emin Minimum required field strength Eu Resultant usable fields strength f Frequency f0 Resonant frequency fc Carrier frequency fd Doppler frequency shift fD Maximum Doppler frequency shift fIF Intermediate frequency fim Image channel frequency fLO Frequency of local oscillator Fs Sampling rate in the frequency domain G Power gain h Planck’s constant H Screen height H Parity check matrix h1s Transmitter antenna height above sea level h2s Receiver antenna height above sea level heff Effective antenna height above sea level

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I Cumulated signal of all interferers I/ C Interferer to carrier ratio Ii i-th interferer Ii + Ai Nuisance field ik k-th bit in a 360-bit group {z} Imaginary part of complex number z j Imaginary unit k Boltzmann’s constant Ka 26.5–40 GHz band satellite channel Kmax Maximum boundary sub-carrier of DVB-T signals Kmin Minimum boundary sub-carrier of DVB-T signals Ku 12–18 GHz band satellite channel mod m Modulo m operation μ Mean (expected) value NFFT Number of sub-carriers NLDPC LDPC code wordlength NTR Number of reserved tones ν(t) Complex-valued baseband signal ω 0 Resonant frequency (pulsation) ωIF Intermediate frequency (pulsation) ωLO Local oscillator frequency (pulsation) ωu Frequency (pulsation) of desired signal p Matched kernel vector P Total power in bandwidth B pi Parity bit P(f ) Power density of the thermal noise as a function of frequency Px Power of additional noise source φ 0 Phase offset Q Quality of a resonant circuit R Code rate S(f ) Normalized power spectrum of the received signal σ Standard deviation s(t) Physical (real-valued) transmitted signal System A ATSC System B DVB {z} Real part of complex number z t Continuous time T Temperature in Kelvins Td Symbol temporal offset θr TCA reference angle Ts overall OFDM symbol duration time TU Duration of useful parts of OFDM symbols u Field strength variable u mean value of variable u v Speed of electromagnetic wave

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W Screen width W:H Screen aspect ratio wi Useful signal versus interference weight x datum, signal value x[n] digital signal x(t) signal in time domain X Random variable Xk k-th imaginary DFT component z Complex number

A subject of this thesis in analysis of the second generation European Digital Video Broadcasting – Terrestrial system (i.e., DVB-T2) published in 2009 by the European Telecommunications Standards Institute (ETSI) under the number EN 302 755. This new system is a successor of the elder DVB-T system, which is also considered in this thesis, including its implementation in Poland. Digital Terrestrial Television (DTT) is still the most important platform for delivering television to households, despite the increasing importance of non-linear and on-demand media consuming models and the already observed, related to this trend, some fall of the need for traditionally organized broadcasting, i.e., the linearly transmitted programs. In result, DTT remains continuously important to television broadcasters as a platform that conserves traditional market division into big players. DTT is also important to viewers, nowadays, mainly thanks to many popular high quality live or near-live transmissions as well as due to rich and interesting film content in the high definition (HD) video standard. With the matter of DTT transmission, the receiver design is a closely related problem. However, most set-top box (STB) and television (TV) set manufacturers do not pay too much attention to the proper design of the radio frequency (RF) front-ends they mount on-board. It is thus an important issue to investigate how the DVB-T and DVB-T2 RF front-ends should be built properly in order to achieve robustness against various effects, e.g. signal fading and multipath propagation, and in order to achieve universal receivers capable to cope with different standards even with a transparent, i.e., “plug and play” manner. This is just one of the main subjects of this thesis. Although a careful analysis and evaluation of the DVB-T2 receiver architectures is the central problem, the author, who is an emission engineer working for a television company, namely for the Polish National Televi- sion (Telewizja Polska S. A.), decided to enlarge the scope of this thesis and include analyzes of the process of the television digitization (especially in Poland) and its impact on decisions about the broadcast modernization in future, taking the impact of the respective decisions on appropriate organization of the whole system and on architectures the respective DVB-T and DVB-T2 receivers. A scientific aim of this thesis is the analysis of the process of the digitization of the terrestrial television and estimation of future trends of replacement of the contemporary DVB-T system by the second generation DVB-T2 standard taking evaluation and proper choice of new receiver architectures into account. The aim of this work is also to investigate various architectures of DTT receivers that may be adopted to the new DVB-T2 standard and to indicate the most convenient architectures for the DVB-T and DVB-T2

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fixed reception. The receiver is understood in this work mainly as the radio frequency (RF) front-end, which consists of the tuner and the demodulator. Various tuner and demodulator architectures were investigated. Adequate architectures for the DVB-T2 fixed reception scenario were indicated. A precise analysis of the process of television digitization in Poland (especially taking into account the experiments and measurements prepared by the Author of this thesis and efforts and achievements of the Polish National Television Company) together with the estimation of the future trends in this process is an additional, practical, and technical aim of the carried research.

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1.1 Introduction

In January 1992 the International Telecommunications Union (ITU), which is a special agency of the United Nations, responsible for developing international standards on wired and wireless communications (typically referred to as agreements), formed a special task group (named Task Group 11/3) to develop recommendations for advanced terrestrial television broadcasting services [1]. The efforts were directed towards a completely new family of digital standards named Digital Terrestrial Television Broad- casting (DTTB) [2]. After four years of intensive works at the meeting held in Sydney, Australia in November 1996, the complete recommendation was announced. The proposed DTTB system was adapted to both 50 Hz (European) and 60 Hz (American) environments and could be applied to broadcasting within all typical television channels, i.e., those of 6, 7, or 8 MHz bandwidth (Chapter 2). This achievement can be considered as the beginning of a new era of digital terrestrial (and satellite) television. In Europe the European Broadcasting Union (EBU) and the Europe concentrated consortium named Digital Video Broadcasting (DVB) developed a digital broadcast transmission standard for the terrestrial television that was first published in 1997 [3]. The first digital terrestrial broadcast took place in the United Kingdom in 1998. The developed DVB-T system was not only able to transmit compressed digital video but also digital audio and other data in the MPEG transport stream format (Chapter 3) using the so-called coded orthogonal frequency-division multiplexing (COFDM known also as OFDM) modulation (Chapter 4) and single frequency network (SFN) technique (Chapter 5). In 2009 the second generation of the DVB-T, i.e., DVB-T2 standard was published by the European Telecommunications Standards Institute (ETSI) under the number EN 302 755 [4]. Analysis of this standard and comparison to its predecessor DVB-T,

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taking special attention to the impact on the receiver architectures together with considerations about the choice of the most preferable architecture, is just the subject of this thesis. Among new DVB-T2 features over and above the DVB-T are:

higher modulation orders

application of low density parity check codes (LDPC) [5] and Bose- Chaudhuri (BCH) cyclic error-correcting codes [6] for channel coding instead of convolution and Reed-Solomon codes [7]

higher number of transmission variants, i.e, more ﬂexible choice of protection intervals and carriers

a possibility for immediate use of multiple radio channels for a single transmission

modulation constellation rotation option

a possibility for modiﬁcation of emission parameters during emission

a possibility of arbitrary multimedia data streams transmission instead of MPEG-2 transport streams only

sending from two antennas using the Alamouti scheme [8, 9]

reduction of peak-to-average power ratio (PAPR).

For all these reasons the new DVB-T2 standard should in future replace the elder, less eﬃcient, and less convenient DVB-T standard.

1.2 Subject, scope, and signiﬁcance of the work

Digital Terrestrial Television (DTT), despite the increasing importance of non-linear and on-demand media consuming model and related fall of the need for traditionally organized linear programs, still remains the most important platform for delivering television to households. It also remains important to television broadcasters as a platform that conserves traditional market division into big players. DTT is also important for viewers, nowadays, mainly thanks to high quality live or near-live transmissions as well as popular film content in high definition (HD). With the matter of DTT transmission, the receiver design is closely related. Most set-top box (STB) and television (TV) set manufacturers do not pay too much attention to the design of RF front-ends they mount on- board. Indeed, many popular front-ends are substantially simplified and usually do not take advantage of the state of the art in the standardization and technology. It is important to know, how to build the receiver front-ends properly in order to achieve robustness against various effects, e.g. multipath

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propagation and signal fading. It frequently happens that signal power indicator says it is enough “power” in the air, however reception is hardly possible due to low signal quality [10]. That is why the careful analysis and evaluation of the DVB-T and DVB-T2 receiver architectures is the main subject of this thesis. However, during the research works, the Author, who is an emission engineer in a television company, namely in the Polish National Television (Telewizja Polska S. A.), decided to enlarge the scope of this thesis and include analyzes of the process of the television digitization (especially in Poland), including his own experiments and measurements and the impact on foreseeable decisions about the broadcast modernization in future, taking the impact of the respective decisions on the appropriate system structure and organization and on the architectures of the DVB-T and DVB-T2 receivers [11, 12, 13, 14].

1.3 Aims and scientiﬁc thesis

A scientific aim of this thesis is the analysis of the process and estimation of future trends of the replacement of the DVB-T (Digital Video Broadcasting Terrestrial) system by the second generation DVB-T2 (Digital Video Broad- casting Terrestrial 2nd generation) standard taking evaluation and proper choice of new receiver architectures into account. The aim of this work is also to investigate various architectures of DTT (Digital Terrestrial Television) receivers that may be adopted to the new DVB-T2 standard and to indicate the most convenient architectures for the DVB-T and DVB-T2 fixed reception. The receiver in a narrow sense is mainly understood in this work as the RF (radio frequency) front-end of the whole receiver, which consists of the tuner and demodulator. Various tuner and demodulator architectures were investigated. Adequate architectures for the DVB-T2 fixed reception scenario were indicated. A precise analysis of the television digitization in Poland (especially taking the Polish National Television Company into account) together with the estimation of the trends of this process in future is an additional practical and technical aim of the carried research. The scientific thesis can be formulated as follows: proper choice of the receiver architectures of the DVB-T2 receivers together with the choice of the technical / technological parameters and offered services is crucial for the success of introduction of the new DVB-T2 standard into practice.

1.4 Text structure

This thesis begins with a description of the history and an analysis of the future of the digital terrestrial television broadcasting in Chapter 2. Then

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the European DVB-T system is presented in detail in Chapter 3. The main technical and technological DVB-T details, namely the OFDM technology and the SFN concept are introduced in Chapters 4 and 5, respectively. A detailed analysis of similarities and differences between the DVB-T and DBB-T2 standards is given in Chapter 6. Next Chapter 7 is devoted to the description and analysis of the respective propagation models. Results of calculations are also given. The next and main part of this thesis starts with Chapter 8, which contains the carefully prepared presentation and analysis of specifications for the receivers. In the Author’s opinion, these are the most important sources of requirements for the receiver front-end designers, as recommendations included in the DVB-T and DVB-T2 standards define only the transmitted signal and say nothing about how the receiver shall be built and what parameters it shall achieve. It has to be stressed that also propagation calculations take into account parameters defined in specifications in order to assume and determine such critical constants as the minimum signal level or the required carrier to noise ratio C/ N. It is worth mentioning that engineers from seemingly unrelated fields as radio frequency (RF) designers and propagation models specialists, including the regulatory issues specialists, must rely on the receiver requirements and actual receiver architectures. After the previously mentioned important chapter, the leading and most weighty part comes, namely evaluation of the receiver architectures in Chapter 9. This is in fact one of the main parts of this thesis, in which various designs and ideas considered, analyzed, and simulated by the Author, are described and evaluated. The last part of this work is Chapter 10 containing presentation of the Author’s experiments and measurements supplemented with the description of the television digitization process in Poland together with the forecast of the future trends and their impact on the technological decisions, including receiver architectures and planned services. Thus the author presents in this Chapter his practical and technical considerations based on the results of his experiments and measurements. The text is closed with the conclusions in Chapter 11 summarizing the Author’s achievement and with the list of references including the author’s publications on the subject of this thesis.

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2.1 DTT concept

It was already mentioned in Chapter 1 that the first stable standardized concept of the contemporary digital terrestrial television (DTT or DTTV) systems was announced in November 1996 during the ITU Task Group 11/3 meeting in Sydney, Australia. The main idea of the new terrestrial television broadcasting technology was, on one hand, to use conventional television transmission media including channel definitions (i.e., conventional television antennas on both transmission link sides) but, on the other hand, to win much more capacity and flexibility for the information content (by means of very efficient data compression and substantially more efficient use of spectrum), in addition to win better video and audio quality and many new quite innovative services as e.g. electronic program guide (EPG) or captions (including special services for hearing and viewing im- paired people) [15]. Moreover, all these wonderful features were achieved with lower emission and transmission costs for broadcasters (certainly after the initial investments for the new technology) [2, 16]. The mentioned better video quality does not only mean larger image resolution, i.e., high definition television (HDTV), or lower noise level even for the conventional standard definition television (SDTV) present in parallel with HDTV, nor the 16:9 panoramic screen aspect ratio (instead of the conventional 4:3) — the so-called “screen aspect ratio” is the ratio of picture width W to height H, expressed as W:H)), but also a quite new feature of visual spatial reality, namely the three-dimensional (3D) view option in parallel to the conventional two-dimensional (2D) view. Historically, the 4:3 (or 12:9) picture aspect ratio of the conventional television systems was chosen from a collection of many various aspect ratios used in the film industry already in the 1930’s. There were attempts to modify it, e.g., the Japanese 5:3 (or 15:9) proposal for a wide screen HDTV service, but only the 16:9 standard has really become popular. High popularity of the new 16:9 wide (panoramic) screen aspect ratio adopted by SMPTE and the ITU began in 1984 as a response to requests

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formulated by the film industry (especially by the Hollywood community) to adequately standardize the just bearing technology of electronic film production. On one hand, over decades of photo-chemical film history, the film industry had become accustomed to use numerous incompatible picture aspect ratios. On the contrary, the electronic production community pursued to restrict to a few and preferably to only one properly chosen aspect ratio. Thus a problem arose what single aspect ratio would be the best for the needs of the electronic film producers. In this way the panoramic 16:9 aspect ratio was chosen. As a compromise between 4 : 3 12 : 9 and 16:9 aspect ratios their arithmetic mean equal to 14:9 was proposed in order create compromise quality pictures using television sets with both 4:3 and 16:9 screen types. Transmissions with 14:9 aspect ratio were quite popular in Great Britain, Ireland, France, and Spain. Better audio quality means broader bandwidth (according to the human hearing range approximately from 20 Hz up to 20 kHz), larger dynamic range, and better spatial reality, i.e., not only stereo sound but also surround sound due to the possibility of multichannel audio reception. The worldwide DTT competition resulted in various standards, which now have been spread all over the world and divided it into disjoint DTT areas. Nowadays the following DTT systems exist simultaneously in different countries:

European DVB-T standard is used in Europe, Greenland, Russia, New Zealand, Australia, Colombia, and in some parts of Africa and Asia [17]

ATSC standard, developed in USA by the Advanced Television Stan- dards Committee (ATSC), the successor of the National Television Standards Committee (NTSC), is used in North and Central Amer- ica (the United States, Canada, Mexico, El Salvador, Honduras) and South Korea

Integrated Services Digital Broadcasting — Terrestrial (ISDB-T) standard developed in Japan is used in Japan and Brazil

DTMB (Digital Terrestrial Multimedia Broadcast), the China’s own standard, is used in the People’s Republic of China, Hong Kong, Macau, and Cuba; this standard was ﬁrst known as DMB-T/H (Digital Multimedia Broadcast – Terrestrial/Handheld).

All DTT standardization activities of the ITU Task Group 11/3 were based on an assumption that a real, practical success of the new technology is directly related to inexpensive personal appliances that should merge many functions. They shell namely not only be the television receivers but also communication devices and even computers. Such receivers should have clear, universal, and for many years stable architecture in order to

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guarantee for a particular receiver to cope with many standards (possibly with all existing ones and also with future DTT standards). Therefore the Task Group 11/3 recommended to design and apply standard receiver (decoder) architectures, which would make it possible to operate with entire set of services defined in the ITU-R recommendation BS.1196 [18]. By this means the important request for low cost and universal consumer appliances for all DTTB systems was fully met [19]. As already mentioned above, the main concept of the DTT transmission is to apply the same radio frequencies (in both UHF or VHF ranges) and exactly the same terrestrial channel definitions as those used before for the analog television transmission but to exploit them much more efficiently. The respective technology, referred to as the “TV program multiplexing”, consists in the transmission of multiple television programs (sub-channels) in a single standard TV channel. The number of sub-channels that can be multiplexed in a single channel depends on the channel capacity (i.e., on the amount of digital data that can be transmitted in such channel) as well as on the technology used for the sub-channel data compression. Thus the channel capacity is defined, first, by the digital data modulation scheme and, second, by the radio signal modulation scheme. The European DVB-T channels have capacity of 24 Mbit/ s. Data are sent using either 16 or 64-state Quadrature Amplitude Modulation (QAM) and the radio signals are formed according the modulation method called the Coded Orthogonal Frequency Division Multiplexing (COFDM).

2.2 General DTT transmission scheme

A general DTT transmission scheme is shown in Fig. 2.1. It contains the following four subsystems:

Source coding and compression

Service multiplexing and transport

Physical layer containing the channel coding and the modulation scheme

Planning factors, which include transmission and receiver environments together with implementation strategies [20].

The source coding and compression subsystem refers to the data compression and coding mechanisms designed to reduce large streams of data created when particular images (frames) are represented in a natural way, i.e., by individually quantized picture elements: pels (subpixels) and pixels and when sound is represented by a stream of individually digitized samples. Source coding should also include error protection (detection and correction) techniques that are appropriate for application to video, audio, and ancillary digital data. “Ancillary data” are: system control data,

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Figure 2.1: General DTT transmission scheme

conditional access control data, and data associated with additional program services such as captions. Ancillary data can also be associated with entirely new, innovative program services as e.g. the electronic program guide (EPG). The proper compression techniques of video, audio, and ancillary data should be efficient, i.e., on one hand, should minimize the number of bits needed to represent the relevant information but, on the other hand, should be able to precisely recreate the multimedia information. Thus they can or even should be lossy (especially for video) but transparent for human perception in a sense that the perceived video and audio quality degradation is either completely not noticeable by humans or is at least fully acceptable. The service multiplexing and transport subsystem first divides streams of individual sub-channels (television programs) and additional services into information packets containing unique identifiers of each packet and packet type, then it merges (multiplexes) these video, audio, and ancillary data stream packets into a single program (sub-channel) data stream and finally combines particular program data streams into a single broadcast channel stream for simultaneous emission. Multiplexing also provides the transport mechanism, which is appropriate for the interoperability between digital media such as the terrestrial or satellite broadcasting and the cable distribution, including interfaces with the receivers, computers, and recording devices [21]. The physical layer module contains channel coding and modulation schemes. The channel coding consists in adding supplementary bits to the compressed data in order to detect and correct transmission errors. The coding mechanisms have to be adjusted to the modulation scheme, which is in turn adopted to the transmission medium. The modulation scheme converts the error protected data stream into a modulated signal on one or more carriers for final physical transmission. Thus the transmission system

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is referred to as the single-carrier or multiple-carrier system, respectively. Planning factors together with the implementation strategies include organization of services and the whole broadcasting system (taking the already existing broadcast services into account) together with considerations about the appropriate transmission media and receiver features (including appropriate receiver architectures) in order to maximize the consumer sat- isfaction and to reduce the consumer costs.

2.3 DTT Recommendations

The DTT recommendations published by the Task Group 11/3 were divided into two subsets according to two standards, namely to: System A (ATSC) for the use with 60 Hz environment and 6 MHz channels and System B (DVB) for the use with 50 Hz environment and 7 or 8 MHz channels. The necessary diﬀerences between these two subsets were minimized with respect to the video and audio coding and transport mechanisms and harmonized in such a way that uniﬁed, low cost receivers with the same architecture can operate in a “plug and play” manner and decode signals from both systems.

2.3.1 Video coding For broadcasting with both HDTV and SDTV qualities the MPEG-2 transport stream is used [22, 23]. Although the video source coding is based on the MPEG-2 standard, MPEG-4 coding mechanisms are possible. This option has been chosen in Poland. The applicable coding variants are highly reduced as compared to the full standards. For instance, the originally expected spatial and temporal video scaling has finally been dropped out as in the result of careful investigations and experiments performed by EBU it occurred that this functionality is too complex with respect to the necessary receiver architectures, thus inefficient for the terrestrial broadcasting. However, a number of video coding profiles and levels has been included, despite a relatively high impact on the cost and architecture complexity of the consumer receivers and the costs of the broadcasting equipment. The DTTB video subsystem tools are defined in the recommendation ITU-R BT.1208 [24], which for the content producers makes it possible to provide programs with both conventional and wide screen aspect ratios as well as HDTV and SDTV quality formats. Despite this variety of options, single decoders with a unified architecture, capable of decoding all options in the above recommendation are possible. According to the Au- thor’s investigations and experience inexpensive and universal consumer appliances are plausible and should be reality in near future. The main standardized and internationally accepted HDTV format is the 1080 line system. Other HDTV vertical resolutions, e.g., 1152 lines / 50 Hz (in Europe) and 1035 lines / 60 Hz (in Japan) are specific to

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certain nations or regions. The 720 lines HDTV standard (USA) is usually considered as too close in performance to the existing progressive scanning versions of the 625 lines conventional broadcasting standard. Only the 1080 lines standard existing in 50 Hz, 60 Hz, and 24 frame per second ﬁlm compatible versions is seen as providing suﬃcient improvement over all existing, conventional formats to form the basis for the international agreement. The same approach can be implemented in the transmission of multi-program signals or stereoscopic television services over existing digital satellite or terrestrial links or cable television networks.

2.3.2 Audio coding

Selection of an appropriate audio source coding standard was a difficult and controversial problem. There were two primarily considered candidates: the European MUSICAM system and the United States supported Dolby Digital AC-3 system. MUSICAM is a part of the MPEG-2 specification and is backwards compatible with MPEG-1. AC-3 was already used in other motion picture standards, e.g., on DVDs. Thus, both candidate standards had already been popular and applied with wide acceptance n many media. In the surround sound tests conducted by EBU to evaluate the performance of these two systems, the AC-3 system was found to be more efficient — the AC-3 operating at 600 kbit/ s occurred to be equivalent in performance to the MUSICAM system operating at 900 kbit/ s. Nevertheless, the final decision was to preserve both standards in the receiver architecture in a form of appropriate arithmetic units and instruction sets placed in memory. In result, an appropriate instruction set is to be selected for the desired level of audio performance: monaural, stereo, or surround sound. Thus, from the receiver architecture point of view each decoder can be considered as a series of resources with three resident instruction sets (for monaural, stereo, and surround sound). The appropriate instruction set has to be selected for the desired level of service. Furthermore, since the resources required to implement both standards (MUSICAM and AC-3) are very similar, a dual decoder would not require two separate decoder modules, but can be implemented with the same architecture and with six resident instruction sets: three for MUSICAM and three for AC-3. After discussing this architecture with manufacturers of audio equipment, it was concluded that the additional cost to the consumer purchasing a digital television receiver with a dual decoder would be less than a 0.25 % increase over the cost of a receiver with a single system decoder capability only. Thus this effect is negligible and the ITU- R recommendation BS.1196 includes both MUSICAM and AC-3 standards [18]. Again an existence of a single decoder architecture, capable of decoding the entire set of options and tools defined in the mentioned recommendation is feasible. This fully meets the requirement of the World Broadcasting

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Union for unique, inexpensive, and universal consumer appliances. Among various audio transmission possibilities are: one (monaural), two independent (stereo) sound channels, as well as matrix services, which may include L, C, R (Left, Center, Right) channel, and the surround sound. The surround sound options include a single surround channel (S) or a surround sound pair: Surround Left (SL) and Surround Right (SR). In addition, a low frequency enhancement (LFE) or a sub woofer channel capability may be added to any of the matrix services. The LFE channel is deﬁned as having a limited frequency range (20 Hz to 120 Hz) and allows the listener to extend the low frequency content of the sound format in terms of both frequency and level. It essentially duplicates the sub woofer channel used in digital ﬁlm sound formats. Since the LFE channel is coded at a lower bit rate, it does not constitute an additional rightful sixth channel but the somehow fractional channel, indicated as “.1” channel in the “5.1” notation. Therefore a commonly accepted agreement was reached that any new television service should provide a full range of multi-channel audio options from a single (monaural) channel to 5.1 channels for each service. In order to accommodate existing services to this requirement, they are com- prised of some (from 1 up to 5.1) audio channels, depending on the data capacity of the multiplexed bit stream. In result, the digital television systems provide audio compression mechanisms that produce bit streams as low as 32 kbit/ s for voice / dialog services up to 384 kbit/ s for 5.1 channel surround sound reproduction.

2.3.3 Transport layer

The so-called transport layer is based on the service multiplex and the MPEG-2 transport stream. Since the MPEG-2 standard had originally been developed for video recording and storage applications such as DVD, it required some minor simplifications and additions to allow for its use in the television broadcast environments. The respectively modified and constrained subset of the MPEG-2 standard tool set is defined in the ITU -R recommendation BT.1299 [25]. Due to the use of the Descriptor Tags, the Table ID assignments, and the assignment of the Packet Identification (in a harmonized way between Systems A and B) a single device with a unified architecture can be capable of decoding the entire set of the defined tools. As it was already stressed an existence of such a single decoder architecture meets the main request of the World Broadcasting Union for definition of globally harmonized broadcasting systems leading to single universal “plug and play” appliances for the use by consumers without any need to consider the specific situation.

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2.3.4 Physical layer Recommendations concerning the DTTB physical layer (channel coding and modulation scheme) are defined in the ITU-R recommendation BT.1305 [26] and take into consideration the existing 6, 7, and 8 MHz allocation of channel assignments and the need to accommodate differing environments and planning factors. The major obstacle to agree on a unique worldwide standard for the unified modulation system is the lack of uniformity in the use of the broadcast spectrum throughout the world. Countries that had adopted the NTSC system developed spectrum plans employing 6 MHz wide channels. Countries that had adopted the PAL and SECAM systems developed spectrum plans with channel bandwidths that range from 6 MHz to 8 MHz (in Poland 8 MHz channels are used only). Therefore, the finally standardized systems and the available bit capacity vary geographically. Some countries: United States, Canada, and Australia developed national broadcasting systems that are focused on local broadcasting. In these environments, one or more national broadcasters produce and distribute programs nationally to local service providers, but allow for local insertion of commercials and locally generated programming such as news throughout the broadcast day. Other countries developed their national broadcasting systems that are truly national in their content. In these environments, one or more national broadcasters produce and distribute programs that are distributed nationally without any local modification. The “local” broadcaster is simply a re-transmission tower. Such national broadcasters are interested in the efficient use of the spectrum in the national basis. Therefore a new technique named SFN was introduced that allows for the use the single frequency network (SFN) throughout the whole country. SFN is insensitive to reflections and de- ployment of receivers that respond to the strongest SFN signal in the local environment. Thus, it was impossible to agree on a single approach to the transmission standard and two different approaches were documented:

Coded Orthogonal Frequency Division Multiplexing (COFDM) in Eu- rope (Fig. 2.2)

8-level vestigial sideband modulation (8-VSB) in the United States (Fig. 2.3).

COFDM modulation is less susceptible to interference and is capable to accommodate national single frequency networks. Thus, as already mentioned above, it is used in Europe. On the contrary 8-VSB is slightly more bit eﬃcient (carried more bits per MHz) and was designed for use with 6 MHz bandwidth. Thus 8-VSB modulation was adopted for use in the North American System A (ATSC), where SFN concept is irrelevant.

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Figure 2.2: COFDM modulation concept

Figure 2.3: 8-VSB modulation concept

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Although a set of COFDM parameters was also proposed for use in System A by a group of broadcasters including ABC and NBC in the United States and the CBC in Canada, the FCC Advisory Committee deemed the proposal as coming too late in the testing schedule, and the system was never accepted for testing.

2.3.5 Unified receiver architecture The Task Group 11/3 paid particular attention to constructing a digital receiver architecture that could accommodate both HDTV and SDTV services in the terrestrial broadcasting environment and that could also be inter- operable with cable delivery, satellite broadcasting, and recording media. The approach taken provides harmonization between services by using a unified, common method of video and audio source coding and a unified, common service multiplex and transport techniques. As noted above, two different subsets of this unified set are defined; System A (ATSC) and Sys- tem B (DVB). The two subsets are compatible and single decoders can be provided that can extract either subset from the data stream. This unified transport data stream is then provided with a framing structure, error protection mechanism, and modulation scheme appropriate to the distribution media. The common transport is seen as a “container” and facilitates the interoperability of the signal through different delivery media. This results in a common data stream after demodulation in the receiver, which sim- plifies the receiver architecture. Thus reduces complexity and costs of the consumer receiver appliances.

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3.1 DVB history and future trends

As it was already mentioned Digital Video Broadcasting (DVB) is a suite of internationally accepted but Europe oriented open standards for digital television. These standards are maintained by the DVB project, an international industry consortium with more than 270 members. The standards are published by the Joint Technical Committee (JTC) of the European Telecommunications Standards Institute (ETSI), European Committee for Electrotechnical Standardization (CENELEC), and European Broadcasting Union (EBU). The interaction of the DVB sub-standards is described in the DVB Cookbook [27]. Many aspects of DVB are patented, including elements of the MPEG video coding and audio coding. DVB-T is an abbreviation for “Digital Video Broadcasting – Terres- trial”. It is the DVB European-based consortium for standardization of the broadcast transmission of digital terrestrial television. It was first published in 1997 [3]. This system transmits compressed digital video, digital audio, and other data in the MPEG-2 transport stream, using coded orthogonal frequency- division multiplexing (COFDM or OFDM for short) modulation scheme and the single frequency network (SFN) [28, 29]. The DVB-T standard is published as specification EN 300 744 (Framing structure, channel coding, and modulation for digital terrestrial television) [3]. This is available from the ETSI website as ETSI TS 101 154 specification [30] for the use of video and audio coding in broadcasting applications based on the MPEG-2 transport stream. Many countries that have adopted DVB-T system have published standards for their implementation. These include the D-book in Great Britain [31], the DGTVi in Italy [32], the ETSI E-Book [33], and the Scandinavian NorDig [34, 35, 36]. DVB-T has been adopted for digital television broadcasting by many countries, using mainly VHF 7 MHz and UHF 8 MHz channels whereas Taiwan, Colombia, Panama, Trinidad, Tobago, and Philippines use 6 MHz channels.

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DVB-T has been further developed into such standards as DVB-S, DVB-C, and DVB-H (handheld). The latter was a commercial failure and now is practically no longer offered. DVB-S and DVB-C were ratified in 1994. DVB-T was ratified in early 1997. The first commercial DVB-T broadcasts were performed by the United Kingdom’s Digital TV Group in late 1998. In 2003 Berlin, Ger- many was the first area, in which analog TV broadcasting was completely stopped. Most European countries are fully covered by digital television and many have switched off PAL / SECAM services. In result the DVB is used in Europe, as well as in Australia, South Africa, and India. This is also true for cable and satellite transmission in most Asian, African, and many South American countries. However, many of these countries have not yet selected the format for the digital terrestrial broadcast (DTTB). Some countries including the United States, Canada, Mexico, El Salvador, Honduras, South Korea have already chosen ATSC instead of DVB-T. In Japan and Brazil ISDB-T system is used. China, Hong Kong, Macau, and Cuba use DTMB system. In March 2006 the decision was made by the DVB group to study a concept of a new DVB-T standard. In June 2006 a formal study group named TM-T2 (Technical Module on Next Generation DVB-T) was established and the works in order to develop the new generation of the digital terrestrial television, namely the DVB-T2, were started. According to the commercial requirements for a low cost end user equipment (receivers with unified, optimized, and universal architectures), the call for respective technologies was issued in April 2007. The first phase of the DVB-T2 works was devoted to provide the op- timum reception for stationary (fixed) receivers (i.e., those, which in principle can move but slowly, i.e., rather with a walking person speed than with a speed of a moving car). The next phases delivered methods for higher payloads (with new aerials) and the mobile reception issues. An assumption was made that the novel system should provide a minimum 30 % increase in payload under similar channel conditions already used for the DVB-T. Thus, the DVB-T2 standard gives more robust TV reception and increases the possible bit rate by over 30 % for single transmitters (as in the UK) and should increase the maximum bit rate by over 50 % in large single-frequency networks (as in Germany and Sweden). The DVB-T2 draft standard was ratified by the DVB Steering Board on June 26, 2008 and published on the DVB homepage as the DVB-T2 standard BlueBooks [37, 38]. The DVB-T2 specification was approved by the DVB Steering Board and sent to ETSI for adoption as a formal standard. ETSI adopted the standard on September 9, 2009. It was handed over to the European Telecommunications Standards Institute (ETSI) on June 20, 2008. The ETSI adopted the DVB-T2 standard on September 9, 2009. NorDig published a DVB-T2 receiver specification and performance

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requirement on the July 2009 [36]. In March 2009 the Digital TV Group (DTG), the industry association for digital TV in Great Britain, published the technical specification for high definition services on digital terrestrial television (Freeview) using the new DVB-T2 standard in D-book, 6th edition [31]. The DVB-T2 standard was finalized in August 2011. This system also transmits compressed digital video, audio, and other data in but in so- called “physical layer pipes” (PLPs), also using OFDM modulation with concatenated channel coding and interleaving. The higher offered bit rate, with respect to its predecessor DVB-T, makes it a suited system for carrying HDTV signals on the terrestrial TV channel (though many broadcasters including Poland still use elder DVB-T for this purpose) [39]. PLP’s are very convenient for broadcasters, who want to form a joint signal in a shared frequency band, as by this means it is possible to organize a number of individual statistical groups within one DVB-T2 signal. The first DVB-T2 test modulator was developed by the BBC Research and Innovation Center using 8 MHz Channel. BBC had developed and built the modulator / demodulator prototype in parallel with the DVB-T2 standard being drafted. The first test from a real TV transmitter was performed by the BBC Research – Innovation in the last weeks of June 2008 using channel 53 from the Guildford transmitter (southwest of London). The BBC, ITV, Channel 4, and Channel 5 decided to convert one of the UK multiplexes (namely B or PSB3) to DVB-T2 to increase capacity for HDTV. The first TV region to use the new standard was Granada in November 2009. The expectations are that over time there will be enough DVB-T2 receivers among the end users to switch all DTT transmissions to DVB-T2 (thus to H.264 although with the MPEG-2 transport stream). Up to this time both systems will be used in parallel. DVB-T2 was also tested in October 2010 in Geneva region using the UHF band and Channel 36. A mobile van was testing BER, strength,and quality reception, with special PCs used as spectrum analyzers and constellation testers. This van was moving in Canton Geneva (Switzerland), and France (Annemasse, Pays de Gex). Many test broadcast transmissions using DVB-T2 standard are being in process in France near Rennes. Prototype receivers were shown in September 2008 and more recent version at the IBC 2009 in Amsterdam. A number of other manufacturers demonstrated DVB-T2 at IBC 2009 including Albis Technologies, Arqiva, DekTec, Enensys Technologies, Harris, Pace, Rohde&Schwarz, Tandberg, Thomson Broadcast and TeamCast. Since 2012 Appear TV also produce DVB-T2 receivers, DVB-T2 modulators and DVB-T2 gateways. Other manufacturers that plan DVB-T2 equipment production are Alitronika, CellMetric, Cisco, Digital TV Labs, Humax, NXP Semiconductors, Pana- sonic, ProTelevision Technologies, Screen Service, SIDSA, Sony, ST Micro- electronics, and T-VIPS. Other companies like ENKOM or IfN develop software (processor) based decoding [40].

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Lately, the DVB has established a 3D TV group (CM-3DTV) to identify what kind of 3D-TV solution does the market want and need, and how can DVB play an active role in the creation of such solution. The CM-3DTV group held a DVB 3D-TV Kick-off Workshop in Geneva on January 25, 2010, followed by the first CM-3DTV meeting the next day. Just now the DVB is working on a definition of a new standard for 3D video broadcast, namely the DVB 3D-TV. Besides digital audio and digital video transmission, DVB also defines data connections (DVB-DATA – EN 301 192 specification) with return channels (DVB-RC) for several media (DECT, GSM, PSTN/ISDN, satellite, etc.) and protocols (DVB-IPTV: Internet Protocol; DVB-NPI: network protocol independent) [41]. Elder technologies such as teletext (DVB-TXT) and vertical blanking interval data (DVB-VBI) are also supported by the standards to ease conversion. However, for many applications more advanced alternatives like DVB-SUB for subtitling are available.

3.2 DVB-T basics

DVB-T offers three different modulation schemes (QPSK, 16-QAM, 64- QAM) [42, 43]. Rather than carrying one data carrier on a single radio frequency (RF) channel, COFDM works by splitting the digital data stream into a large number of slower digital streams, each of which digitally modulate a set of closely spaced adjacent sub-carrier frequencies. In the case of DVB-T, there are two choices for the number of carriers known as 2 K or 8 K modes. These are actually 1705 or 6817 sub-carriers that are approximately 4 kHz or 1 kHz apart, respectively. DVB-T as a digital transmission delivers data in a series of discrete blocks at the symbol rate. DVB-T is a COFDM transmission technique, which includes the use of the so called guard intervals. They allow the receiver to cope with strong multipath transmission circumstances. Within a geographical area, DVB-T also allows single-frequency network (SFN) operation, where two (in practice many more) transmitters carrying the same data operate on the same frequency. In such cases the signals from each transmitter in the SFN need to be accurately time-aligned, which is done by synchronization of stream timings at each transmitter with the reference to GPS. The length of the guard interval can be controlled (i.e. changed and chosen). It is a trade off between data rate and SFN capability. The longer the guard interval the larger is the potential SFN area without creating an intersymbol interference (ISI). It is possible to operate SFNs, which do not fulfill the guard interval condition if the self-interference is properly planned and monitored.

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3.3 Source coding and MPEG-2 multiplexing

The compressed video, compressed audio, and data streams are multiplexed into MPEG program streams (PSs). One or more PSs are joined together into a transport stream (TS). This is the basic digital stream, which is being transmitted and received at homes by TV sets or so-called set top boxes (STBs). Allowed bitrates for the transported data depend on a number of coding and modulation parameters (they can range from about 5 to about 32 Mbit/ s). Two different MPEG-TSs can be transmitted at the same time, using a technique called hierarchical transmission. It can, e.g, be used to transmit an SDTV signal and a HDTV signal on the same carrier. Generally, the SDTV signal is more robust than the HDTV one. At the receiver, depending on the quality of the received signal, the STB may be able to decode the HDTV stream or, if the signal strength lacks, it can switch to the SDTV one. By this means, all receivers that are in proximity of the transmission site can lock the HDTV signal, whereas all others, even the farthest ones, can still be able to receive and decode the SDTV signal. The MPEG-TS is a sequence of data packets of the fixed length of 188 bytes. This sequence is decorrelated by a technique called the energy dispersal. Then the error correction method is applied to the transmitted data, using the Reed-Solomon (RS) code. This makes it possible to correct of up to a maximum of 8 wrong bytes for each 188-byte packet. Moreover, the convolutional interleaving is used to rearrange the transmitted data sequence, in such a way that it becomes more rugged to long sequences of errors. Thus a second level of the error correction is given by a punctured convolutional code, which is often denoted in STBs menus as the forward error correction (FEC). There are five valid coding rates: 1/2, 2/3, 3/4, 5/6, and 7/8. Then, the data sequence is rearranged again, aiming to reduce the influence of burst errors. This time, a block interleaving technique is adopted, with a pseudo-random assignment scheme (this is really done by two separate interleaving processes, one operating on bits and another on groups of bits). Next, the resulting digital bit sequence is mapped into a baseband modulated sequence of complex symbols (in fact represented with harmonic signals). There are three valid modulation schemes: QPSK, 16-QAM, 64-QAM. In a process named frame adaptation, these complex symbols are grouped into blocks of constant length (1512, 3024, or 6048 symbols per block). In this way a 68 blocks long frame is generated and a superframe is built with 4 frames. Pilot and TPS (Transmission Parameters Signaling) information is added in order to simplify the reception of the signal being transmitted on the terrestrial radio channel. Pilot signals are used during the synchronization and equalization phase, while TPS parameters are transmitted to unequiv- ocally identify the transmission cell. The receiver must be able to synchro-

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nize, equalize, and decode the signal to gain access to the information held by the TPS pilots. Thus, the receiver must know this information before- hand, and the TPS data is only used in special cases, such as changes in the parameters, resynchronizations, etc. [44].

3.4 DVB transmission

DVB systems distribute data using a variety of approaches, including:

Satellite: DVB-S, DVB-S2 and DVB-SH [4, 45, 46], DVB-SMATV for distribution via SMATV [47]

Cable: DVB-C, DVB-C2 [48]

Terrestrial television: DVB-T, DVB-T2 [3, 49, 50]

Digital terrestrial television for handhelds: DVB-H, DVB-SH [4, 45, 51, 52, 53]

Microwave: using DTT (DVB-MT), the MMDS (DVB-MC), and/or MVDS standards (DVB-MS).

These standards define the physical layer and data link layer of the distribution system. Devices interact with the physical layer via a synchronous parallel interface (SPI), synchronous serial interface (SSI), or asynchronous serial interface (ASI). All data is transmitted in MPEG transport streams with some additional constraints (DVB-MPEG). A standard for temporally-compressed distribution to mobile devices (DVB-H) was published in November 2004 [54]. These distribution systems differ mainly in the modulation schemes used and error correcting codes used, due to the different technical constraints. DVB-S (SHF) uses QPSK, 8-PSK or 16-QAM. DVB-S2 uses QPSK, 8-PSK, 16-APSK or 32-APSK, at the broadcasters decision. QPSK and 8-PSK are the only versions regularly used. DVB-C (VHF/UHF) uses QAM: 16-QAM, 32-QAM, 64-QAM, 128-QAM or 256-QAM. Lastly, DVB- T (VHF/UHF) uses 16-QAM or 64-QAM (or QPSK) in combination with (C)OFDM and can support hierarchical modulation [55]. A resulting, characteristic flat-top DVB-T signal power spectrum is discussed in Chapter 4 [56].

3.5 Encryption and metadata

The conditional access system (DVB-CA) deﬁnes a Common Scrambling Algorithm (DVB-CSA) and a physical Common Interface (DVB-CI) for accessing scrambled content. DVB-CA providers develop the conditional access systems with reference to these speciﬁcations. Multiple simultaneous

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DVB-CA systems can be assigned to a scrambled DVB program stream providing operational and commercial flexibility for the service provider. DVB has also developed a Content Protection and Copy Management (DVB-CPCM) system for protecting content after it has been received. It is intended to allow flexible use of the recorded content at home and beyond, while preventing unconstrained sharing if using the Internet. DVB-CPCM has been evaluated as controversial, however, in the technical literature it is usually considered as the European answer to failures in this matter in the rashly organized American broadcast. DVB transport includes metadata called Service Information (DVB-SI), described in recommendations ETSI TR 101 211, ETSI TS 101 211, and ETSI EN 300 468 [57, 58], that links various elementary streams into co- herent programs and provides human-readable descriptions for electronic program guides as well as for automatic searching and filtering. The dat- ing system used with this metadata suffers from a year 2038 problem, in which due to the limited 16-bit words and modified Julian day offset used, will cause an overflow issue similar to the year 2000 problem. By comparison, in the American rival DigiCipher 2 (DCII), based on MPEG-2 transmission in the ATSC system, this problem will not occur until the year 2048 due to the 32-bit format used there. Recently, the DVB has adopted a profile of the metadata defined by the TV-Anytime Forum (DVB-TVA, ETSI TS 102 323) [59] to control digital video recorders (DVRs). This is an XML based technology and the DVB profile is tailored for enhanced personal digital recorders. DVB lately also started an activity to develop a service for Internet Protocol Television (IPTV), i.e., DVB-IPI (ETSI TR 102 033, ETSI TS 102 034, ETSI TS 102 814) [60, 61, 62], which also includes metadata definitions for a Broadband Content Guide (DVB-BCG), ETSI TS 102 539 [63].

3.6 DVB Multimedia Home Platform

The DVB Multimedia Home Platform (DVB-MHP), which is a Java-based environment, has been designed to deﬁne and realize many innovative consumer applications. Among them are return channels, developed in order to create bi-directional communication, which makes the so called “interactive television” service possible. DVB has standardized a number of return channels that work together with DVB-S, DVB-T, or DVB-C technologies, e.g.: Return Channel Satellite (DVB-RCS) in C (4–8 GHz), Ku (12–18 GHz), and Ka (26.5–40 GHz) frequency bands with return bandwidth of up to 2 Mbit/ s and Return Channel Terrestrial (DVB-RCT) speciﬁed by ETSI EN 301 958 [64]. DVB-MHP realizes also many other DVB and MPEG-2 concepts and provides interfaces for other features like network card control, application download, and layered graphics.

Orthogonal frequency-division multiplexing (OFDM), or more strictly, the coded orthogonal frequency-division multiplexing (COFDM) due to forward error correction codes, which are typically added to the scheme in order to protect data [65], is a channel modulation technique applied in the DVB-T system but also in many other wide-band communication standards [66, 67].

Figure 4.1: OFDM modulation scheme

Figure 4.2: OFDM demodulation scheme

OFDM is a multi-carrier, or in or in other words, a multi-tone modulation method. Its principle is illustrated in Fig. 4.1. First, a unique high rate data stream (or more precisely, a stream of symbols) is divided (decimated) into multiple, parallel low rate symbol streams. Next, to each

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of these low rate streams a typical digital modulation technique [10] is applied (a 64-state or 16-state quadrature amplitude modulation (QAM) in the case of the DVB-T). Next, to each of the QAM constellations of the low symbol rate streams a unique frequency (the so-called sub-carrier) from the uniformly distributed baseband sub-carrier frequencies is assigned by connection to consecutive inputs of the inverse discrete Fourier transformation (IDFT) block [68, 69]. Finally, the multifrequency digital complex valued signal occurring at the outputs of the IDFT block is converted to the analog form in order to modulate the analog sinusoidal carrier signal of, say, frequency fc, according to the respective television channel used e.g., for the DVB-T transmission [70]. The corresponding receiver scheme, which is just inverse of the OFDM transmitter from Fig. 4.1, is shown in Fig. 4.2.

4.1 Mathematical relationships

Let us assign the number of sub-carriers by N. If each sub-carrier is modulated with a constellation of M symbols, then the OFDM low rate streams at the IDFT block inputs represent an alphabet of MN symbols. The hypothetical analog (continuous-time) OFDM sub-carrier signals, i.e., components (harmonics including the DC component) of the IDFT block output signal after the AD converter but before the carrier (of frequency fc) modulation, can be expressed as π / ej2 kt TU k 0, 1,...,N − 1 , (4.1)

where TU is the OFDM useful symbol duration. These hypothetical complex- valued analog sub-carrier signals, spaced in frequency by multiples of Δ / f 1TU, are the AD converted versions of the DFT discrete-time components and are therefore orthogonal, i.e. T U − π / π / e j2 kt TU · ej2 lt TUdt 0 (4.2) 0 for k l, k, l 0, 1,...,N − 1 and T 1 U − π / π / e j2 kt TU · ej2 lt TUdt 1 (4.3) TU 0 for k l, k, l 0, 1,...,N − 1, i.e., T 1 U − π / π / j2 kt TU · j2 lt TU δ e e dt k−l (4.4) TU 0 , , ,..., − δ where k l 0 1 N 1, and k−l is the Kronecker delta deﬁned as 0 for n 0 δ . (4.5) n 1 for n 0

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Figure 4.3: DVB-T and DVB-T2 OFDM symbol structure

The hypothetical analog baseband complex-valued OFDM signal ν(t) can then be expressed as

N−1 π / ν j2 kt TU, ≤ ≤ . (t) Xke 0 t TU (4.6) k0 To avoid intersymbol interference in multipath fading channels, a guard interval of length Δ has to be inserted prior to the just described theoretical OFDM block [71]. During this interval, a cyclic preﬁx is transmitted such that the signal in the interval −Δ ≤ t ≤ 0 equals the signal in the interval −Δ ≤ ≤ TU t TU. The practical OFDM signal with cyclic preﬁx is thus given by N−1 π / ν j2 kt TU, −Δ ≤ ≤ (t) Xke t TU (4.7) k0 Δ The overall symbol duration is thus equal to Ts TU + (Fig. 4.3). Finally, the complex-valued baseband signal ν(t) is converted into the transmitted signal s(t) (Fig. 4.1) with the carrier frequency fc. The physical (real-valued) transmitted signal s(t) is given by π s(t) ν(t)ej2 fct N−1 | | π / . Xk cos 2 (fc + k TU)t + arg Xk (4.8) k0

4.2 OFDM features

The most important advantage of the OFDM technique, which makes it very practical for the DVB application over and above single-carrier modulation schemes, is insensitivity to narrowband interference and frequency- selective fading caused by the unavoidable multipath transmission phenomenon. Due to many narrowband and thus slowly modulated OFDM signal components instead of a single quickly modulated wideband signal of a single-carrier modulation technique, complex time-domain adaptive equalization filters are not necessary. Instead of them a simple technique of cyclic prefixes and guard intervals between symbols is sufficient to eliminate the intersymbol interference [72]. This makes it possible to achieve a diversity gain, i.e. a large signal-to-noise ratio [73]. This feature makes

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design and realization of the very spectrum efficient single frequency network (SFN) feasible as several adjacent transmitters producing a large bunch of slowly modulated narrowband signals with the same frequencies interfere constructively, rather than destructively as it occurs in the case of the fast-modulated single-carrier system [74, 75]. Although during the guard intervals redundant data are transmitted only, which causes reduction of the system capacity, in the OFDM-based broadcasting systems, such as in the DVB long guard intervals are used in order to allow the transmitters to be spaced a long way apart in the SFN. The longer guard intervals the larger the SFN cell-sizes. The maximum distance between transmitters in the SFN is equal to the distance the electromagnetic signal travels during the guard interval [76]. Summarizing, the sub-carrier orthogonality allows for high spectral ef- ficiency, with a total symbol rate near to the theoretically maximal Nyquist rate. Moreover, the OFDM spectrum is nearly white. This feature reduces the negative electromagnetic co-channel interference. However, the OFDM requires very accurate frequency synchronization between the receiver and the transmitter. Otherwise, with some frequency deviation the sub-carriers will no longer be orthogonal, causing inter-carrier interference (ICI) (i.e., cross-talk between signals among the sub-carriers). Such a frequency deviation can be caused by mismatched transmitter and receiver oscillators, or by Doppler shift due to moving objects. The Doppler shift is typically combined with a multipath transmission phenomenon. Thus, various signal interferences and reflections can appear with various frequency offsets, which can be quite difficult to correct. This limits the use of the OFDM in vehicles moving with high- speed [77].

4.3 OFDM receiver architecture

On one hand, the described principle of the OFDM scheme is advantageous for the DVB transmitter and receiver architectures as the IDFT and DFT blocks are very efficiently realized with fast and computationally lossless (i.e., computationally very precise) IFFT and FFT algorithms. The respective optimized architectures are based on the reverse addressing technique and are typical building blocks of many digital signal processors (DSPs). On the other hand, the Doppler shift together with the multipath transmission result in the interference (overlapping) of shifted, i.e., non- orthogonal sub-carriers. A solution can be the scheme referred to as WCP- OFDM (Weighted Cyclic Prefix Orthogonal Frequency-Division Multiplex- ing). However, it complicates the receiver architecture, because it consists in using short FIR filters at the transmitter output in order to get weighted (non-rectangular) signal windows [78]. Such signal shaping results in a merely near perfect reconstruction, using a single-tap per sub-carrier equalization. More sophisticated ICI suppression techniques increase drastically

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the receiver architecture complexity.

4.4 Error correction coding

A classical type of error correction coding used with OFDM-based systems is convolutional coding, often concatenated with Reed-Solomon coding. Usually, additional interleaving (on top of the time and frequency interleaving mentioned above) in between the two layers of coding is implemented. The choice for Reed-Solomon coding as the outer error correction code is based on the observation that the Viterbi decoder used for inner convolutional decoding produces short errors bursts when there is a high concentration of errors, and Reed-Solomon codes are inherently well-suited to correcting bursts of errors. Newer systems, however, usually adopt near-optimal types of error correction codes that use the turbo decoding principle, where the decoder iterates towards the desired solution. Examples of such error correction coding types include turbo codes and LDPC codes, which perform close to the Shannon limit for the Additive White Gaussian Noise (AWGN) channel. Some systems that have implemented these codes have concatenated them with either Reed-Solomon (for example on the MediaFLO system) or BCH codes (on the DVB-S2 system) to improve upon an error ﬂoor inherent to these codes at high signal-to-noise ratios.

4.5 Adaptive OFDM modulation

Due to division the transmission to many sub-carriers, the OFDM technique is robust against disadvantageous communication conditions. How- ever, this feature can be further improved using a return (or in other words a feedback) channel with the so-called adaptive modulation. This adaptivity consists in the following mechanism. In diﬃcult channel conditions, the relevant information is transmitted back over a return- channel. It causes appropriate changes in the channel coding and power allocation across all sub-carriers or to individually chosen sub-carriers. If in particular instants particular ranges of frequencies suﬀer from interferences or large attenuation, the sub-carriers within these ranges can be disabled or be slower or even more robustly modulated by means of additional error correcting coding. For example, the so-called bitrate adaptive DSLs (ADSLs and VDSLs) are based on the real time bitrate control in such a way that larger bandwidth is dynamically allocated to those subscribers, who momentary need it most. This is achieved by means of the so-called discrete multitone modulation (DMT), i.e., the adaptive OFDM, which individually controls each sub-carrier according to the channel conditions and reacts with the respective bit loading technique. Thus, the upstream and the downstream

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speeds can be continuously varied according to the needs by allocating either more or fewer carriers for each of these streams.

4.6 OFDM based wide area broadcasting

Due to the structure of the OFDM symbols (Fig. 4.3) it is possible to realize DTTB by covering a large area (even an area of a whole country) with many spatially dispersed transmitters sending simultaneously the same but suitably delayed signal. This is the so-called single frequency network (SFN), which is beneﬁcial with respect to conventional multi-frequency broadcast network (MFN) as it:

ﬁrst, utilizes the available spectrum much more eﬀectively

and, second, the fading probability is much less in comparison to that obtainable with the MFN.

The SFN technology is described in detail in Chapter 5.

4.7 Multiple access OFDM

OFDM is typically considered as a single channel modulation scheme for transmission of one bit stream over a communication (television) channel with a sequence of OFDM symbols shown in Fig. 4.3. However, it can be extended to to a multi-user technology by combining it with some time, frequency, or even code division multiple access techniques. The frequency division multiple access (FDMA) can be realized by assigning different OFDM sub-channels to different users. This is the so- called orthogonal frequency-division multiple access (OFDMA). OFDMA supports differentiated quality of service by assigning different number of sub-carriers to different users in a similar way as in CDMA (code division multiple access). Thus sophisticated packet scheduling or media access control (MAC) algorithms can be avoided. In the technique called OFDM-CDMA or MC-CDMA (multi-carrier code division multiple access) OFDM is combined with CDMA and the spread spectrum transmission schemes in order to separate different users. By this means, complex dynamic channel allocation (DCA) algorithms are not necessary. Moreover, the neighbor channel interference can be reduced with the optimally chosen fixed channel allocation (FCA) technique.

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5.1 SFN principle

A single-frequency network or SFN is a broadcast network where several transmitters simultaneously send the same signal over the same frequency channel [79, 80]. Analog AM and FM radio broadcast networks as well as digital broadcast networks can operate in this manner. SFNs are not generally compatible with analog television transmission, since the SFN results in ghosting due to echoes of the same signal [81]. A simplified form of SFN can be achieved by a low power co-channel repeater, booster or broadcast translator, which is utilized as gap filler transmitter [82, 83, 84]. The aim of SFNs is efficient utilization of the radio spectrum, allowing a higher number of radio and TV programs in comparison to traditional multi-frequency network (MFN) transmission. An SFN may also increase the coverage area and decrease the outage probability in comparison to an MFN, since the total received signal strength may increase to positions midway between the transmitters [85, 86]. SFN schemes are somewhat analogous to what in non-broadcast wireless communication, for example cellular networks and wireless computer networks, is called transmitter macrodiversity, CDMA soft handoff and Dynamic Single Frequency Networks (DSFN) [87]. SFN transmission can be considered as a severe form of multipath propagation. The radio receiver receives several echoes of the same signal, and the constructive or destructive interference among these echoes (also known as self-interference) may result in fading. This is problematic especially in wideband communication and high-data rate digital communications, since the fading in that case is frequency-selective (as opposed to flat fading), and since the time spreading of the echoes may result in intersymbol interference (ISI) [88]. Fading and ISI can be avoided by

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means of diversity schemes and equalization ﬁlters [89].

5.2 Wideband digital broadcasting

In wideband digital broadcasting, self-interference cancellation is facilitated by the OFDM or COFDM modulation method. OFDM uses a large number of slow low-bandwidth modulators instead of one fast wide-band modulator. Each modulator has its own frequency sub-channel and sub-carrier frequency. Since each modulator is very slow, we can afford to insert a guard interval between the symbols, and thus eliminate the ISI [90]. Al- though the fading is frequency-selective over the whole frequency channel, it can be considered as flat within the narrowband sub-channel. Thus, advanced equalization filters can be avoided [91]. A forward error correction code (FEC) can counteract that a certain portion of the sub-carriers are exposed to too much fading to be correctly demodulated [92, 93]. OFDM is utilized in the terrestrial digital TV broadcasting systems DVB-T and DVB-T2 (used in Europe and many other areas) as well as ISDB-T (used in Japan and Brazil). OFDM is also widely used in digital radio systems, including DAB, HD Radio, and T-DMB. Therefore these systems are well-suited to SFN operation [94]. In DVB-T a SFN functionality is described as a system in the implementation guide. It allows for re-transmitters, gap-filler transmitters (essentially a low-power synchronous transmitter) and use of SFN between main transmitter towers [95]. The DVB-T SFN uses the fact that the guard interval of the COFDM signal allows for various length of path echoes to occur is not different from that of multiple transmitters transmitting the same signal onto the same frequency. The critical parameters is that it needs to occur about in the same time and at the same frequency. The versatility of time-transfer systems such as GPS receivers (here assumed to provide PPS and 10 MHz signals) as well as other similar systems allows for phase and frequency coordination among the transmitters [96]. The guard interval allows for a timing budget, of which several microseconds may be allocated to time errors of the time-transfer system used. A GPS receiver worst case scenario is able to provide ±1 µs time, well within the system needs of DVB-T SFN in typical configuration [97, 98]. In order to achieve the same transmission time on all transmitters, the transmission delay in the network providing the transport to the transmitters needs to be considered. Since the delay from the originating site to the transmitter varies, a system is needed to add delay on the output side such that the signal reaches the transmitters at the same time. This is achieved by the use of a special information inserted into the data stream called the Mega-frame Initialization Packet (MIP) which is inserted using a special marker in the MPEG-2 Transport Stream forming a mega-frame [90]. The structure of DVB-T frames, super-frames and mega-frames is

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Figure 5.1: Structure of DVB-T frames, super-frames and mega-frames

shown in Fig. 5.1. The MIP is time-stamped in the SFN adapter, as measured relative the PPS signal and counted in 100 ns steps (period time of 10 MHz) with the maximum delay (programmed into the SFN adapter) alongside. The SFN adapter measures the MIP packet against its local variant of PPS using the 10 MHz to measure the actual network delay and then withholding the packets until the maximum delay is achieved [99]. In DVB-T MIP packets consist of 4 bytes of header and 184 bytes of the useful part. Each mega-frame has one MIP. The PID value of MIP in the transmitted Transport Stream equals 0x15 (decimal 21). The useful part of MIP consists of ﬁelds. The most important ﬁelds in MIP are:

pointer, length: 16 bits, deﬁnes the number packets between MIP and the ﬁrst packet of the next megaframe, it helps to precisely locate MIPs in the stream

maximum_delay, length: 24 bits, multiples of 100 ns, deﬁnes maximum time in SFN between the beginning of the megaframe in the SFN controller (in the head-end) and the beginning of the megaframe in the SFN adapter (at transmitter site); this value is set up manually by the administrator of the broadcasting network, the default value is around 900 ms and in most cases is suﬃcient, i.e. usually the maximum delay in the whole network is much lower

synchronization_time_stamp, length: 24 bits, multiples of 100 ns, for m−th MIP packet it speciﬁes the time between the last 1PPS pulse and

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Figure 5.2: 1PPS signal

the beginning (the first bit) of the m−th + 1 MIP. Besides synchronization purposes, MIPs may be also used for transmitter configuration, e.g. one may change modulation variant, obviously if this option of remote configuration is enabled in the transmitter. A typical GPS receiver produces at its output the following signals: sinusoid of frequency 10 MHz (100 ns period) and RMS 0.5V 1PPS (Pulse Per Second) The 1PPS (Pulse Per Second) signal is shown in Fig. 5.2. In a standard GPS receiver the PPS signal has the following characteristics: signal level: TTL slew time: 10 ns pulse width: 5ms accuracy: ±100 ns, understood as accuracy of the rising edge instant synchronization to the beginning of the UTC second It should be understood that the resolution of the mega-frame format is being in steps of 100 ns, whereas the real accuracy needs can be in the range of 1–5 s. The resolution is sufficient for the needed accuracy. There is no strict need for an accuracy limit as this is a network planning aspect, in which the guard-interval is being separated into system time error and path time-error. A 100 ns step represents a 30 m difference, while 1 s represents a 300 m difference. These distances needs to be compared with the worst case distance between transmitter towers and reflections. Also, the time accuracy relates to near-by towers in a SFN domain, since a receiver is not expected to see the signal from transmission towers being geographically far apart, so there is no accuracy requirements between these towers. The so-called GPS-free solutions exists, which essentially replaces GPS as the timing distribution system. Such system may provide benefit in integration with transmission system for the MPEG-2 Transport Stream. It does not changes any other aspect of the SFN system as the basic requirements can be met.

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Figure 5.3: An example of a measured channel impulse response in SFN with the guard interval shown below

Reflected paths as well as signals coming from different transmitters in SFN are shown in Fig. 5.3). If the delayed paths are within the guard interval, they are correlated on each sub-carrier and they interfere. For some sub-carriers the interference is positive and the power rises and form some there are selective fades in the spectrum. En example of this phenomenon for two paths is depicted in Fig. 5.4. If two paths of arrival time difference ΔT interfere, the resultant spectrum is a multiplication the OFDM rectangle with a periodical fade pattern of period in frequency domain ΔF 1/ΔT. A simulated example for influence of delayed paths on a DVB-T2 signal spectrum is shown in Fig. 5.5.

5.3 SFN gain and self-interference

SFN gain is observed when signals coming from a number of transmitters add up in the receiver. SFN gain is one of the most important reasons why SFNs are built. Assume there are two transmitters of equal power operating in MFN mode. Assume also that somewhere between there is an uncovered area because power level from both transmitters is not sufficient for the receiver. The receiver can make use just of one of the transmitted signals at the same time as both transmitter produce their signals in different channels. Now if assume they can transmit identical and synchronized signals in the same channel, the receiver will “see” the power gain provided. This may change the the are out of coverage to a covered region. Obviously provided the time differences between the incoming signals are not exceeding a given limit, which depends on the length of the guard interval.

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Figure 5.4: An example of a measured channel impulse response in SFN with the guard interval shown below

Figure 5.5: An example of a simulated DVB-T2 signal, unaﬀected signal on the left side and inﬂuence of SFN on the spectrum on the right side

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Figure 5.6: Large SFN for theoretical calculations

The other side of the coin is the effect of self-interference. As due to SFN gain some regions get higher signal level, some other regions observe higher noise level due to self-interference, which emerges when incoming signals time of arrival difference is high. In such a case for OFDM modulation, another uncorrelated OFDM signal behaves like additive Gaussian noise. In terms of a single sub-carrier phase-shifted sinusoid still interfere, however if you take into consideration thousands of sinusoid of different amplitudes and phases, you may treat it as Gaussian noise. In Fig. 5.6 a theoretical SFN network is shown with a number of transmitters distributed over hexagonal cells. The maximum SFN gain observed in such a network is reaching 5 dB in the points where signals coming from three transmitters arrive at the same time (as shown in Fig. 5.7). At the same time at such points self interference is low because signal levels from far transmitters are relatively low compared to the closest three with zero relative delay (Fig 5.8).

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Figure 5.7: SFN gain for transmitter network from Fig. 5.6

Figure 5.8: SFN self interference level for a transmitter network from Fig. 5.6

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6.1 DVB-T2 overview

The DVB-T standard (numbered ETSI EN 300 744 [3]) along the past decade has shown thaT it may be successful and thanks to itS SD, HD, and interactive services appeared in many households throughout Europe. DVB-T has been widely adopted in Europe as well in Poland in particular. Over recent years it has been replacing the analog PAL / SECAM transmission. As it was already mentioned in Chapters 2 and 3 the first edition of this standard was published as early as in 1997. Thus DVB-T does not cover the latest achievements of in the field of digital transmission, namely in space-time coding, channel coding, and others [100, 101]. Indeed, from the oday’s perspective the physical and the transport layers defined in the DVB-T standard seem to be rather outdated [102]. DVB-T has also turned out to be insufficiently flexible for a number of different use cases and transmission variants. This is why the DVB project consortium in cooperation with EBU and ETSI proposed the new standard DVB-T2 (numbered ETSI EN 302 755 [49]), which was issued in 2009. As it was already mentioned in Chapters 2 and 3 the DVB forum has approved the DVB-T2 standard proposal as a DVB BlueBook [37, 38], which was published in June 2008. After that this document has been sent to ETSI in order to be approved and published as a European standard. This workflow enabled development of broadcast and receiving equipment, which was even being done before issuing the formal standard [103, 11, 104]. DVB-T and DVB-T2 are physical and transmission layer standards designed for carrying television streams in terrestrial environment [105]. This assumption however does not prevent from using these standard, e.g., in cable television or to other than broadcast purposes. Note, e.g., that DVB-T is used also for wireless cameras links because of proved robustness against multipath propagation. Both standards, i.e. DVB-T and DVB-T2,

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use OFDM modulation, which is suitable for terrestrial channels, in which deep selective fades are observed because of multipath propagation and the single-frequency networks (SFNs) operation [106, 107]. Both standards use two stages of channel coding [108]. DVB-T2 introduces many changes relative to its previous generation DVB-T. It is more flexible and has much more transmission modes. These changes concern modulation variants, channel coding, utilization of space- time codes, signaling of transmission parameters, and many more. The price for them is much more complex signaling and much more complicated implementation. This is why many receivers implement only portions of the new standard. In order to provide more flexibility and more options for trade-offs between throughput and robustness, new constellation 256-QAM and new numbers of sub-carriers of 1 K, 16 K and 32 K, so called FFT sizes, have been specified in the DVB-T2. Note that in order to preserve throughput, the bigger the number of sub-carriers the shorter the frame duration. Thus the new FFT sizes allow longer channel impulse response, thus more robust single frequency networks (SFNs) [12]. The DVB-T option of hierarchical transmission of independent high- priority and low-priority TSs is no longer supported in the DVB-T2. Thus non-uniform constellations of DVB-T are no longer an option in the DVB- T2. Note that in the newer standard we may use different PLPs to transmit separate TSs. Those PLPs may be transmitted using different constellations and coding schemes, which is signaled in the base-band headers [71]. As it was already mentioned in the Introduction the DVB-T2 standard differs from the DVB-T by the following features:

higher modulation orders

LDPC and BCH channel coding instead of convolutional code and Reed-Solomon code in DVB-T

operation in frequency blocks, even discontinuous

constellation rotation

possibility of changing modulation parameters during transmission

carrying generic streams, not only MPEG-2 Transport Stream

broadcasting from two antennas using a space-time code called the Alamouti scheme

reduction of PAPR (peak to average power ratio).

All the above mentioned features are optional and are activated by proper signaling parameters [109]. There are more changes introduced in the DVB-T2. Only the most signiﬁcant have been described in this chapter.

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6.2 DVB-T and DVB-T2 technical comparison

6.2.1 Transport layer capabilities and DVB-T2 gateways The most important functional diﬀerence between the DVB-T and DVB-T2 standards is that in the DVB-T one signal carries only one transport stream (TS) whereas a single DVB-T2 signal may transport a number of transport streams or generic streams. In order to preserve DVB-T2 transparency to upper layers, Physical Layer Pipe (PLP) concept has been introduced. Note that through DVB-T we can transmit one TS in non-hierarchical mode and HP TS and LP TS in the hierarchical mode [110]. A new layer above TS and Generic Streams was introduced. Each PLP may take a form of:

transport stream

GSE (Generic Encapsulated Stream) for transport of generic packetized streams

GCS (Generic Continuous Stream) where packet boundaries are not known by the modulator

GFPS (Generic Fixed-length Packetized Stream), another option for packetized streams, deﬁned to preserve compatibility with DVB-S2.

The signaling of stream type is deﬁned in base-band headers. This PLP mechanism introduced a new layer of multiplexing. In addition to multiplexing services into TS in DVB-T, one may multiplex entire transport streams or generic streams into DVB-T2 signals using PLPs. Modulation and FEC mode is conﬁgurable for each PLP. Thanks to this the modulation and FEC may change over time in DVB-T2 signals as many PLPs may be transmitted. In the DVB-T the user could use statically only one system variant. In the multiple PLP scheme, PLPs are divided into their separate and common parts, called PLP(n) and Common PLP, respectively. The Com- mon PLP may be useful for PSI/SI transport of multiple transport streams in order to prevent repetition of e.g. electronic program guide (EPG) transmitted in EIT tables. In consequence of introducing the PLPs a new multiplexing layer emerged. Note that for traditional TS-carrying systems, like DVB-T, multiplexing is performed in the TS layer only. With PLPs we may decide which stream shall go through which PLP, thus a new layer of multiplexing emerges. Note that the above-mentioned multiplexing capabilities may be implemented in transmitters with multiple TS and IP inputs. Today Common PLP concept is not widely used as the throughput gain is low. Instead of this, in a PLP the network operator may transmit PSI/SI tables relevant for TS inside the PLP.

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Organization of the OFDM framing resembles that from the DVB-T. Again we have frames consisting a specified number of OFDM symbols. A number of frames constitutes a superframe. OFDM symbols contain useful Δ parts of duration TU and guard intervals of duration , which are cyclic continuations of useful parts and are placed before the useful part. How- ever, in the DVB-T2, some combinations of FFT sizes and guard interval durations are not allowed, whereas in DVB-T all combinations are possible. In completely non-multipath propagation environments there is no point of using guard intervals, which causes throughput losses. Thus the DVB-T2 broadcasting network may be precisely tailored to the assumed propagation conditions. In the DVB-T the only signaling data is transmitted within the dedicated signaling sub-carriers TPS (Transmission Parameters Signaling). In the DVB-T2 the signaling mechanism is much more complex as there are more transmission parameters. The new feature of DVB-T2 is the need of inserting so called DVB-T2 gateway into the transmission chain. This kind of equipment is absent in DVB-T because the Transport Stream generated by the multiplexer in the head-end is directly passed to transmitters, in the case of SFNs through SFN controllers and adapters. Thus in DVB-T the transmitter interface is the Transport Stream. DVB-T2 introduced a new layer or sublayer in the physical layer. Transport Streams or generic streams are wrapped in PLPs (Physical Layer Pipes). In the simplified variant A, there is just one PLP in one DVB-T2 signal. Thus this is a “legacy” mode resembling the traditional DVB-T case. In variant B there are possibly multiple PLPs with a special Common PLP within one broadcast. In DVB-T setting up a bouquet of services could only be done using multiplexers whereas in DVB-T2 it may done by multiplexers as well as by gateways. Gateways play the following roles: the interface head-ends with transmitter through T2-MI (DVB-T2 Modulator Interface) they insert synchronization packets for SFNs called T2-MIP they encapsulate Transport Streams and map them into suitable PLPs they generate signaling for modulator set-up Signal in T2-MI format after being generated by gateway is then transmitted over the distribution network instead of Transport Stream. In DVB-T2 different PLPs may be transmitted in different system variants, including different code rates and modulations. It is one of the most important differences between DVB-T2 and DVB-T in which system variant is assumed constant in one radio channel. Although one may take advantage of hierarchical modulation of DVB-T, where constellation point subsets constitute signal of higher robustness against noise, however hierarchical mode is rarely used in practical situations.

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Figure 6.1: Laboratory set-up showing diﬀerent PLP coverages

Modulator present on the market support T2-MI with all modulation variants including number of sub-carriers and guard intervals. Most of them support MISO as well. It is worth mentioning that different PLPs in one DVB-T2 signal may have different coverages. A laboratory setp-up showing this feature shown in figure 6.1.

6.2.2 Null packet deletion mechanism

The DVB-T carries transport streams without analyzing, nor changing its contents. Whereas in the DVB-T2 the null packets may be deleted to improve the bandwidth eﬃciency. Clearly the deleted null packets shall be recovered in the receiver in order to preserve original throughputs of the transmitted services. Null packets may be recovered by the receiver thanks to a 1-byte counter of null packets called DNP (Deleted Null Packet), value of which is transmitted in the DVB-T2 signal. The counter data is added after every useful packet informing how many null packets were deleted after it.

6.2.3 Modulation

In order to provide more flexibility and more options for trade-offs between throughput and robustness, new constellation 256-QAM and new numbers of sub-carriers of 1K, 16K and 32K, so called FFT sizes, have been specified in DVB-T2. Note that in order to preserve throughput, the bigger the number of sub-carriers, the shorter frame duration. Thus the new FFT sizes allow longer channel impulse response, thus more robust SFN networks. The DVB-T option of hierarchical transmission of independent high-priority and low-priority TSs is no longer supported in DVB-T2. Thus non-uniform constellations of DVB-T are no longer an option in DVB-T2. Note that in the newer standard we may use different PLPs to transmit separate TSs. Those PLPs may be transmitted using different constellations and coding, which is signaled in base-band headers.

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Figure 6.2: Constellation rotation in a QPSK example. Here the Q channel is faded badly and using constellation rotation the Euclidean distance between points in the rotated case (b) is bigger than in the non-rotated case (a)

6.2.4 Constellation rotation Constellation rotation by a constant angle is a new feature in DVB-T2. Ro- tation enlarges Euclidean distance between constellation points for faded symbols, as shown in ﬁgure 6.2. Assuming that I and Q channels experience independent fades, usually only one channel is faded badly within one symbol. In order to achieve such independence, data for I and Q paths are additionally interleaved. It has been shown that constellation rotation provides gain for channels with fades. Assuming a Gaussian channel, is has been shown, that constellation rotation gives no gain.

6.2.5 PAPR reduction OFDM signals suﬀer from high Peak to Average Power Ratio (PAPR). For a signal x(t) PAPR is deﬁned by equation 6.1.

∗ ∗ PAPR max[x(t)x (t)]/ E[x(t)x (t)] (6.1) in dB scale it is given by 6.2

PAPR dB 10 log PAPR (6.2) For a signal x(t) sin(2πft) within period T 1/ f , PAPR 2 whereas j2πft for x(t)e withing period T 1/ f , PAPR 1. Mean PAPR dB values for single carrier QPSK modulation is 3.5 dB and for 64-QAM it equals 7.7 dB. In the worst case PAPR of the OFDM signal with N uniformly powered sub-carriers equals N. PAPR is an unwanted eﬀect that causes uncontrolled non-linear distortions of the signal, which manifest as higher out-of-band power, which must be ﬁltered out. This distortion may be modeled as a higher noise

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Figure 6.3: Typical DVB-T signal in time domain, 8 K mode, 64-QAM; timeline includes successive RF signal samples

level. A signal of high PAPR may cause clipping in modulators and also may be affected by high power amplifiers in transmitters. In figure 6.3 a couples of frames of ETSI EN 300 744 compliant signal in 8K mode and in 64-QAM variant is shown. Nothing special is visible in figure 6.3 until we zoom it around the maximum value, which is shown in figure 6.4. Distinctive peaks in figure 6.4 may be observed. The reason for this is that many sub-carriers may occur in phase within a given symbol. In the DVB-T there are no mechanisms protecting from high PAPR, which is a drawback of this standard. In DVB-T2 four modes of operation have been standardized (three of them cause the PAPR reduction): no PAPR reduction

ACE (Active Constellation Extension), designed for lower modulation orders

TR (Tone Reservation) or RCT (Reserved Carrier Technique), designed for higher modulation orders

both above methods at the same time. The ACE consists in recursive distorting constellation of useful symbols until the desired level of PAPR is reached. This algorithm is described in Section 9.6.1. of [22]. The time domain OFDM signal obtained by modulation of the base-band symbols by IFFT operator is oversampled, then low-pass ﬁltered and then thresholded. After thresholding it is low-pass

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Figure 6.4: Typical DVB-T signal of high PAPR zoomed in time domain, 8 K mode, 64-QAM; timeline includes successive RF signal samples

filtered again and then downsampled. As the ACE method is distorting constellations, it shall not be used to pilot sub-carriers, nor to the reserved carriers. From the receiver point of view it adds additional noise, however the benefit is that the noise level has lower peaks and is better distributed over time. If this method is used, all modulators within an SFN must use exactly the same constellation modifications all SFN sited shall transmit identical signals. The ACE is usually not implemented in devices present on the market.

Reserved Carrier Technique In RCT special PAPR reducing sub-carriers called “dummy carriers” are inserted. Their values are calculated in a way that minimizes PAPR. They do not carry any useful information and may be omitted by the receiver. This PAPR reduction is not affecting receiver operation except for the fact that the signal may be amplified without serious non-linear distortion. Obviously this worsens throughput because reserved tones do not carry useful information. In this method a special kernel is defined as in (6.3) √ NFFT p IFFT(1TR) (6.3) NTR

where NFFT is the number of sub-carriers, NTR is the number of tones reserved and 1TR is a vector with all entries equal to 0 except for some

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ones put under the indexes where the reserved tones are located. The modulator searches for a peak in the signal in time domain x(t), which is produced by the IFFT block of the modulator. The kernel vector is then shifted and ﬁt in amplitude and phase to the found peak in x(t). Then the matched kernel is subtracted from x(t) and PAPR for such diﬀerence is calculated. One above mentioned iteration can reduce one peak. However thanks to repeating iteration and linear combinations of kernels, a number of peaks can be decimated using this method.

6.2.6 Pilot sub-carriers Due to multipath propagation selective fading occurs. In order to estimate channel frequency characteristics, pilot carriers are inserted. In DVB-T continual pilots are transmitted on fixed sub-carrier positions and scattered pilots are transmitted on different sub-carriers according to the defined pattern. In DVB-T there is only one pattern for scattered pilots and the pattern repeats every four symbols [3]. In DVB-T2 [49], there are a lot pilot patterns from which the broadcasting network operator can choose. The pilot patterns are named PP1, PP2, PP3, PP4, PP5, PP6, PP7, and PP8. They differ for SISO and MISO modes. In MISO, inverted scattered pilots are inserted. DVB-T pilots seem to take too much overhead. An algorithm facing this problem was presented in [15]. It was based on a feedback in channel equalization. In this approach every symbol constellation on every sub-carrier is regenerated in the receiver, after the channel decoder block. After the FEC we may reconstruct constellation points that have been likely transmitted. Comparing those with received constellation points, we may estimate channel characteristics for every sub-carrier. In [84] a method for reduction number of pilots was proposed. In DVB-T2 the CD3 algorithm has been applied leading to reduction of number of pilots. What is more, contrary to DVB-T, many pilot patterns with different pilot density may be used. The currently used pilot pattern is signaled in DVB-T2.

6.2.7 Signal bandwidth The nominal bandwidth, which is the frequency diﬀerence between the boundary sub-carriers Kmax and Kmin for DVB-T signals equals 7.61 MHz in 8 MHz channels and 6.66 MHz in 7 MHz channels. In DVB-T2 the set of possible bandwidth options is wider: 1.7 MHz, 5 MHz, 6 MHz, 7 MHz, 8 MHz and 10 MHz. Focusing on the most popular 8 MHz bandwidth, the nominal bandwidth of so called normal carrier mode of DVB-T2 signal is the same as in the DVB-T case, i.e. 7.61 MHz for all FFT modes. In extended carrier mode, which is available for 8 K, 16 K and 32 K modes, the nominal bandwidth is 7.71 MHz or 7.77 MHz.

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Figure 6.5: Power spectral densities of DVB-T (a) and DVB-T2 (b) signals in 8 MHz channels all possible FFT sizes

Figure 6.6: Modiﬁed Alamouti scheme in DVB-T2

Comparison between DVB-T and DVB-T2 spectrum masks are shown in Fig. 6.5.

6.2.8 MISO In DVB-T2 a MISO method based on Alamouti scheme has been applied. In this mode there are two outputs from the modulator and they have to be passed to two transmitters. There are several diﬀerences to the original Alamouti scheme. The signal passed to the ﬁrst transmitter remains unchanged and instead of processing two symbols adjacent in time, the DVB-T2 MISO processes two adjacent sub-carrier cells of a single OFDM symbol. This operation is shown in Fig. 6.6.

6.2.9 Time-Frequency Slicing In DVB-T only one RF channel could be reserved for one broadcast and that channel must be a continuous segment of spectrum. In DVB-T2 quite a revolutionary idea of Time-Frequency-Slicing has been standardized. It is a new to the broadcast technology method of frequency hoping and reserving more channels for a single TV broadcast. In TFS the reserved channels may not be adjacent. In this mode PLPs are divided into subslices and those are sent over a number of radio channels. Thus one input stream is interleaved both over time, OFDM sub-carriers and now additionally over RF channels. Minimum frequency hoping time must be preserved in

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order to get AGC and PLL locks in the receiver. If the number of tuners in the receiver equals the number of frequencies used, no such restrictions are necessary. However it is expected that practical tuners will have two tuners when the manufacturers decide to build devices supporting this mode. One shall notice the today there is no device on the market that im- plements full DVB-T2 standard, especially the TFS mode. This is due to complexity of the norm and because some mechanisms were not important from a practical point of view, which was quick rollout of HD channels in the UK. TFS mode requires reserving at least two coverages when the regulatory practice is to reserve only one per multiplex, as OFCOM did for Freeview.

6.2.10 Channel coding In DVB-T2 as well as in DVB-T two-stage inner and outer channel coding is used. In DVB-T the Reed-Solomon code protects a 188-byte TS packet producing a 204-byte RS codeword and then the punctured convolutional coding is applied. By standardization of puncturing of a basic code of R 1/ 2, other values of R may be achieved. In DVB-H, an extension of DVB-T for mobile receivers, additional layer of coding was employed the MPE-FEC. The idea of inner and outer coding is continued in DVB-T2 but in this case BCH and LDPC are the outer and inner codes respectively. The new codes in DVB-T2 resulted in much longer codewords comparing with DVB-T, equaling either NLDPC 16200 or 64800. A systematic LDPC code is used in DVB-T2 both for preamble protection as well as for useful frame. LDPC codes belong to a class of Shannon-limit approaching codes. In [65] the deﬁnition of LDPC codes has been given in a form of simple explanation:

the parity check matrix H has constant row and column weights,

row and column weights are small compared with the length of the code,

the number of 1’s common between any two columns is no greater than 1.

LDPC code defined in DVB-T2 is identical to that in DVB-S2 standard. The code is defined in such a way, that the information bits are divided into groups of 360 bits. For every first bit in such 360-bit groups, i.e., for , ,... i0 i360 there are parity bit equations defined in the form of table. ··· ··· For all other bits in the 360-bit groups i.e. i1, i2, , i361, i362, , the same equations are used but with shifted indexes by a specified value given in the norm. After applying the above mentioned rules, the final values of parity bits are obtained by pi pi ⊕ pi − 1. This kind of parity bit accumulation equations regularity enabled compact LDPC code definition.

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Without such a property, huge matrices or huge number of equations must have been published as a part of the standard. The code was designed in such a way that the parity check matrix is lower triangular, which makes encoding easier.

6.2.11 Symbol organization, framing and signaling Organization of OFDM framing resembles that from DVB-T. Again we have frames consisting a speciﬁed number of OFDM symbols. A number of frames constitute a superframe. OFDM symbols contain useful parts Δ of duration TU and guard intervals of duration , which are cyclic continuations of useful parts and are placed before the useful part. However in DVB-T2, some combinations of FFT size and guard interval durations are not allowed, whereas in DVB-T all combinations are possible. In completely non-multipath propagation environments there is no point of using guard intervals, which causes throughput losses. Thus. A DVB-T2 broadcasting network may be precisely tailored to assumed propagation conditions. In DVB-T the only signaling data is transmitted within dedicated signaling sub-carriers TPS (Transmission Parameters Signaling). In DVB-T2 the signaling mechanism is much more complex as there are more transmission parameters.

6.3 Beneﬁts of using DVB-T2 for broadcasters and network operators

6.3.1 Higher bandwidth It is estimated that in an average case, the throughput gain of DVB-T2 over DVB-T will equal around 30 %. However in [15] it is expected that a DVB-T single-frequency network transmitting a 19.91 Mbit/ s stream with Δ/ / , / 64-QAM, 8 K, TU 1 4 R 2 3 may be replaced by a DVB-T2 broad- . / / ,Δ/ / cast of a 33 3 Mbit s stream with 256-QAM, 32 K, R 35 TU 116. In this case the throughput improvement would equal 67 %. However network planning criteria, including the required ﬁeld strengths, are still not known for DVB-T2. For that we have to wait until consumer tuner and demodulator chips show up in the market. The motivation for introducing DVB-T2 for broadcasters in the countries where the test or regular transmissions take place are such factors as:

need for transmitting FTA HD channels in DTT

former usage of MPEG-2 in DVB-T, which caused lack of free spectrum for the needed number of HD channels or unacceptable potential costs of HD transmission.

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6.3.2 Flexibility Assume a broadcaster produces two types of television channels: general channels to cover a wide target group and a number of thematic channels for small and atomized groups of consumers. Usually the general channels for historical reasons are distributed free of charge with as wide coverage as possible. Whereas thematic channels are targeting specific groups usually located in vicinity of agglomerations. In order to fulfill requirements of such a scenario a broadcaster may choose a robust transmission variant for general channels and a high-bandwidth but exposed to propagation phenomena variant for a number of thematic channels. In DVB-T in order to transmit signals of different modulations schemes, two multiplexes with separate distribution networks must be built. It means that costs of separate broadcasting networks are doubled. Among these costs we find frequency reservation fee paid to the regulator and costs of distribution network and transmitter sites.

6.3.3 Usage of available channels In DVB-T2 broadcaster may also use discontinuous bundles of channels (TFS mode), however this feature will be rather used in the future, when VHF and UHF band will be utilized by non-broadcasting services and this spectrum will become valuable. See the section on Time-Frequency Slicing for detailed information.

6.4 Next steps of DVB-T2 standard implementation

Time Frequency Slicing (TFS) method of transmission consists in combining a number of radio channel in order to make a single virtual channel. In this mode sub-slices of PLPs of one T2 frame are sent over multiple radio channels. The DVB-T2 standard defines how PLPs are distributed over a number of RF channels in parallel using TFS. The norm limits the number frequencies and it can take the values from 2 to 6. Through TFS technique also frequency diversity has been introduced in DVB-T2. The signal is transmitted using several frequency channels thus it is spread over a wide spectrum that may be affected by frequency-selective fading. In this sense it is an extension of OFDM concept where sub-carrier interleaving is performed. TFS mode requires at least two tuners for frequency hoping without signal interruptions. The main advantage is more efficient statistical multiplexing as the statistical group is wider. The average gain is 20 % increase in number television services and 4 dB average gain in C/ I, which improves the link budget. There are two receivers profiles standarized in DVB-T2 for TFS support. In so called “single profile” only one tuner is mandatory.

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Most RF front-ends manufacturers considered TFS mode as too complicated and unnecessary. Therefore in the standard it was moved to Annex E as a feature for future implementations in multi-tuner front-ends. It is not required that a receiver with a single tuner shall support TFS. Note that there is no commercially available TFS supporting tuner on the market.

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7.1 Deﬁnitions and assumptions

In the field of radio planning for DVB-T2 [49] many planning parameters were taken from DVB-T [3], particularly from GE’06 [111] coordination agreement. The required C/ N values were determined theoretically for the new standard and the rest of assumptions were preserved. Not only field strengths themselves are influencing reception quality. The following elements of the receiver influence signal coverage and thus radio planning methods in DTT: tuner, which selects the desired channel and converts it to the standard intermediate frequency in a way it rejects unwanted signals, especially in adjacent and image channels

demodulator, where the intermediate frequency signal is sampled, OFDM symbols are synchronized, demodulated and the convolutional code is decoded By choosing in the wealth of modulation, code rate, PLP and other transmission variants, one may get a wide spectrum of network designs. As- suming that the density of transmitter sites and their powers are hardly changeable, the price for high throughputs is low coverage and the price for good coverage is low throughput. It is assumed is most cases, that the wanted coverage of new DVB-T2 multiplexes for fixed reception shall be the same as DVB-T. Thus usually the design rule a trade-off between obtaining coverage similar to previous DVB-T coverages with a moderate growth of throughput at the same time. Propagation models are crucial in DTT for network planning and coverage calculations. Formulas and curves defined in such models determine the results of coverage calculations and because of fundamental differences in the models, the results of calculations may differ highly. E.g. the model ITU-R P.1546 [112, 111] is known to smoothen the border of the coverage area and it hardly shows

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gaps inside the coverage area. In extreme cases the diﬀerence between simulated ﬁeld strength and measured can exceed 15 dB. Propagation models can be divided into three groups:

physical, which use physical principles and equations only

empirical, which compute the ﬁeld strength using data derived from measurements in the form of curves or formulas

mixed, which use physical equations as well as corrections based on measurements.

We can also divide popular channel models into two groups:

taking into account the Doppler eﬀect

neglecting the Doppler eﬀect.

We shall bear in mind, that there are two channel types referred to as the “Rayleigh channel”. In Jake’s approach, the Rayleigh channel is also incorporating the Doppler shift. Jakes assumes that n-th wave is shifted in frequency by Δfn fd cos φn . In COST 207 [113] the Rayleigh channel is introducing the Doppler shift as well. On the contrary, the Rayleigh channel defined in ETSI 300 744 [3] neglects the Doppler shift. We observe Rayleigh fading when there is no dominant path. If the Doppler effect is omitted, the fading does not change with time. Doppler shift causes inter-carrier interference (ICI). When Doppler shift increases above the Doppler limit, demodulation is impossible [114]. In the so-called Doppler channel, we assume that reflected waves arrive from different directions and each of them has a different Doppler shift. We further assume that angles of arrival are uniformly distributed and we have infinite number of incoming waves. This results in a characteristic shape of sinusoid after the Doppler channel in the frequency domain. The maximum Doppler / frequency shift is equal to fD fc(v c). For a particular incoming wave Δ / φ φ the shift equals f fc(v c) cos fD cos . The Doppler channel is causing fades and frequency shifts. The shifted components likely fall close to fc + fd and fc − fd, however no components are of shift greater than fd [71]. Assuming that a sinusoidal wave is transmitted, reflected waves propagate horizontally, angle of arrival distribution is uniform and the receiving antenna is omnidirectional, the normalized power spectrum of the received signal equals

1 S(f ) , |f |≤fd (7.1) 2 πfd 1 − (f / fd)

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7.1.1 Mobile channels

P1 and F1 channels In ETSI 300 744 [3], a channel with 20 paths has been deﬁned in order to describe portable indoor and outdoor reception. This model does not include the Doppler shift. This model because of its complexity is often approximated by only 6 paths calculated as a result of low-pass ﬁltering of the original channel and selecting dominant paths. Path amplitudes, delays, and phases are shown in table 7.1. Both the original 20-path and 6-path versions preserve signal power, which means that the signal power before and after the channel is the same.

Table 7.1: F1 channel approximation Path number Delay Amplitude Level Phase µs dB rad 10.050 0.36 −8.87 −2.875 20.479 1 0 0 30.621 0.787 −2.09 2.182 41.907 0.587 −4.63 −0.460 52.764 0.482 −6.34 −2.616 63.193 0.451 −6.92 2.863

Typical Urban (TU6) channel The COST 207 [113] specification described channels of bandwidths from 10 MHz to 20 MHz around 900 MHz center frequency. Channel profiles defined in COST 207 were adopted to mobile DVB-T reception within the Motivate project. Urban reception is usually modeled using the so called Typical Urban (TU6) COST 207 channel profile with 6 paths. This severe channel have paths of relatively high power are relatively high time dispersion. The respective model for the transmission in typical rural areas

Table 7.2: TU6 channel deﬁnition Path number Delay Level µsdB 10.0 −3 20.20 30.5 −2 41.6 −6 52.3 −8 65.0 −10

also assumes six paths and is deﬁned by the Typical Rural (RA6) COST

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207 proﬁle. The so called 0 dB echo with Doppler is proﬁle which consists of two paths of the same power and the delay equal to the half of the Guard Interval (GI).

7.1.2 Log-normal distribution Log-normal distribution is a distribution of a random variable, logarithm of which is normally distributed. This means that if random variable X is

log-normally distributed, then loga X is normally distributed. Certainly, the logarithm base a is not important for the above deﬁnition. The probability density function (PDF) is given by formula (7.2)

2 1 − (ln x−µ) f (x) √ e 2σ2 , x > 0 (7.2) xσ 2π

In television broadcasting network planning it is assumed that ﬁeld strengths are log-normally distributed. In the domain of propagation models and coverage calculations a couple of deﬁnitions were agreed:

nuisance ﬁeld strength — ﬁeld strength of an unwanted signal (interference) with the adequate protection ratio added,

usable ﬁeld strength — ﬁeld strength required to serve a given reception quality in the presence of noise, which comes from natural thermal noise as well as man-made interference,

minimum median ﬁeld strength — ﬁeld strength required to obtain reception for 50 % locations and 50 % time 10 m above ground level.

It is always assumed that the receiving antenna is directed toward the wanted transmitter. This antenna pattern for DVB-T or DVB-T2 services is defined e.g. in recommendation ITU-R BT.419 [115] or in GE’06 [111]. However the consumer antenna gain and directional pattern in the above- given recommendation is rather poor and usually viewers install better antennas when located close to the coverage area border. In order to determine reception probability or the minimum required field strength, a procedure of adding all interferences in the point of reception must be performed. The wanted signal is the field strength of the wanted transmitter. I is the cumulated signal of all interferers Ii. A proper protection ratio Ai is determined for each interferer. A0 is an additional protection ratio. C C (7.3) I A0 + SiIi + Ai

where Si(x) is a function representing a summation method of interfering signals. The term Ii + Ai is called the nuisance ﬁeld and the cumulated interference level is called the usable ﬁeld strength.

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The majority of models assume that field strength is a random variable of a log-normal distribution. It means that the result of propagation calculations just give mean values. Deviations and reception probabilities may be derived. E.g. in ITU-R P.1546 [112] propagation curves define field for 50 % of area and 50 % of time for a given point. There are then special correction factors defined to get other probability of the field strength exceeding a given threshold. In order to determine whether reception is possible in a given point in space, interference levels from other transmitters and other services must be done. Every source of interfering field must be taken into account as well as protection ratios from those interfering transmissions and noise. There is a number of ways how the accumulated effect of interfering fields can be determined. Some of the methods use simplifications in order to minimize computational complexity. Obviously those simplifications cause loosing accuracy. However there are so many simplifications and assumptions made in the whole process of radio planning and simulations that the power summing method is not introducing relevant inaccuracy [116].

7.1.3 Thermal noise and its inﬂuence on the receiver According to John Bertrand Johnson’s and Harry Nyquist’s works of 1928 ([117, 118]), thermal noise, or Johnson-Nyquist noise, is produced by all conductors regardless of their resistance. This noise is a manifestation of blackbody radiation of the conductor [119]. The power density of the thermal noise as a function of frequency is given by formula

hf P(f ) hf df (7.4) e kT − 1 where k is the Boltzmann’s constant, T is temperature in Kelvins, h is the Planck’s constant. Formula (7.4) can be expanded into a series, which gives hf P(f ) df (7.5) 2 3 hf hf 1 hf 1 ··· kT + kT 2! + kT 3! + If we keep the ﬁrst order term only we obtain

P(f ) ≈ kT · df (7.6)

thus the total power in a bandwidth B equals

P ≈ kTB. (7.7)

Assuming the noise temperature T0 290 K we can calculate thermal noise level in DVB-T and DVB-T2 tuners. Receiver noise bandwidth B can take the following values for DVB-T:

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6.66 MHz for a 7 MHz channel

7.61 MHz for a 8 MHz channel.

For DVB-T2 B can take one of the following values:

1.54 MHz for a 1.7 MHz channel in normal carrier mode

1.57 MHz for a 1.7 MHz channel in extended carrier mode

6.66 MHz for a 7 MHz channel in normal carrier mode

6.80 MHz for a 7 MHz channel in extended carrier mode

7.61 MHz for a 8 MHz channel in normal carrier mode

7.71 MHz for a 8 MHz channel in extended carrier mode.

7.2 Field strength summation procedures

It is important to sum up fields impinging the receiver coming from different transmitters. Both wanted and unwanted levels must be taken into consideration with proper protection ratios, depending e.g. on frequency or type of interference. In propagation calculations it is assumed that not only each nuisance field has log-normal distribution, but the resultant sum distribution of the wanted and unwanted fields are log-normally distributed too. Note that the noise generated in the receiver may be seen as yet another source of interference with protection ratio equal to the required C/ N. As it is not trivial to obtain sums of a number of log- normally distributed random variables, a couple of simplifying methods were developed. For example the simplest is to take into account just the strongest interferer. In this method the minimum required field strength is also taken as an interfering field due to noise. This method however does not offer sufficient accuracy and would give too optimistic results. In the following subsections some more sophisticated methods will be discussed.

7.2.1 Simple power-sum method This simple method is based on the sum of squares of nuisance fields and the minimum required field strength. Then the usable fields strength is approximated by the square of the square sums as given in equation (7.8) N 2 2 Eu Emin + (aibiEi) (7.8) i1

where Eu is the resultant usable ﬁelds strength, Emin is the minimum required ﬁeld strength, ai is the protection ratio against signal from i-th transmitter, bi is the receiving antenna discrimination for the proper angle

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of arrival, Ei is the field generated by the i-th transmitter. The drawback of this method is that it neglects the fact, that field strengths are random variables of log-normal distribution. Usually the interference level is lower than in reality and the usable field strength is underestimated.

7.2.2 t-LNM method The t-LNM stands for trilinear log-normal method. It is an approximation for simplified calculation of a sum of a number of log-normally distributed random variables. It is more accurate than its predecessors (LNM and k-LNM) and it is able to take different standard deviations into account for individual fields. In this method noise is treated as interference of 0 dB standard deviation. The summation process may be made cumulatively in pairs as a sum of two log-normally distributed variables remains a log- normal variable. Assuming u1 and u2 are such variables, the sum field is given by 7.9.

− − 1 u1 u2 − u1 u2 u lneu1 +eu2 (u + u )+ln e 2 +e 2 (7.9) 2 1 2 The mean value of the new variable u equals

− − 1 u1 u2 − u1 u2 u (u + u )+ ln e 2 +e 2 (7.10) 2 1 2 and the standard deviation of f equals σ2 2− 2 u u u (7.11) Now an approximation can be made based on equation (7.12)

ξ ξ − 1 − |ξ|− ξ2 ln e 2 +e 2 ≈ |ξ| + Ce A B (7.12) 2 With properly calculated A, B, and C the mean value and variance can be approximated in a simpliﬁed way.

7.3 ITU-R P.370 and ITU-R P.1546 models

The field strength prediction of this family of models is based on sets of curves derived from measurements. Additional correction factors are included and they differentiate the versions of the recommendation. In ITU-R P.370 [120] propagation curves were given for frequency ranges 30 − 250 MHz and 450 − 1000 MHz and they are valid for distances grater than 10 km. This was improved by ITU-R P.1546 [112], were the curves are given for three frequencies and shall be interpolated for other frequencies. In ITU-R P.370, because ranges of frequencies were defined, no interpolation is used. In ITU-R P.370 values were not given in tabular form, whereas in the ITU-R P.1546 model beginning from version 2 contains tabulated values

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of propagation curves in the form of predicted field strength expressed in dBµV/ m as a function of frequency and effective antenna height. These curves are defined assuming the transmitter power ERP 1 kW, the receiving antenna height is 10 m over terrain level and the field strength exceeds the given value for 50 % of locations for 50 %, 10 % or 1 % of time. ITU-R P.370 takes only the effective antenna height heff neglecting the absolute height of the antenna, whereas in ITU-R P.1546 the higher the absolute antenna height, the better the field level. The ITU-R P.1546 recommendation is valid in the range of frequencies from 30 MHz to 3000 MHz, however curves were given just for 100 MHz, 600 MHz, and 2000 MHz and values from curves shall be interpolated as well. The recommendation gives propagation curves in form of numerical tables as were as plots. Curves were defined for distances from 1 km to 1000 km. For distances out of this range, interpolations are done. Curves are defined for a number of possible transmitter antenna heights h1: 10, 20, 37.5, 75, 150, 300, 600, and 1200 m, also are extrapolated for real values. The higher the antenna, the higher the field strength. En example of selected curves from ITU-R P.1546 for climatic Zone 1 (temperate and sub- tropical) are shown in Fig. 7.1. In the range 1 km–15 km, Okumura-Hata curves from ITU-R P.529 are used. For 15 km–1000 km scope the curves are based on ITU-R P.370. The ITU-R P.1546 model give more realistic results than ITU-R P.370, which, e.g., for mixed land and sea paths could cause a weird and incorrect recovery effect (field strength rises if signal is propagated over land and then over sea). Compared to the ITU-R P.370, the land-sea path method of ITU-R P.1546 predicts less field strength. The recommendation is valid for distances from the transmitter greater than 1 km. For smaller distances usually propagation software extrapolates the curves, however limited to the free space loss. The idea of Terrain Clear- ance Angle (TCA) was introduced in ITU-R P.370 and was improved in ITU-R P.1546. The latter utilizes terrain data more precisely than ITU-R P.370. TCA is a correction using digital maps for detecting obstacles near the receiver that signal from the transmitter encounters. In order to calculate TCA, an angle θ must be calculated first. It is measured relative to the line from the receiving antenna that clears all terrain obstacles in direction of the transmitter up to the distance of 16 km, but not further than the transmitter is located. This idea is shown in Fig. 7.2. Having θ, the terrain clearance angle may obtained using 7.13 h [m] − h [m] TCA θ − θ θ − tan 1s 2s (7.13) r 1000d [km]

where θr is the so called reference angle, h1s and h2s are the transmitter and receiver antenna height above the sea level, respectively; d is the propagation path length. Influence of TCA on field strength is shown in Fig. 7.3. The most field strength spoiling case is when TCA is negative, i.e. the receiver is located higher than the transmitter. ITU-R P.370 model gave wrong results, e.g. for mixed land and sea paths could cause a weird

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Figure 7.1: Example of ITU-R P.1546 curves for 100, 600, and 2000 MHz in climatic zone 1

Figure 7.2: Procedure for calculating θ angle

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Figure 7.3: TCA correction

and incorrect recovery effect - field strength rises if signal is propagated over land and then over sea. ITU-R P.1546 is precise enough to detect the coverage boundary, however it gives too optimistic results withing the calculated coverage area as such reasons grow too smooth. This is because the model is based only on statistical curves and is not using besides calculating heff and TCA.

7.4 Other models

There is a number of models implemented in commercially available software, which were not adopted in the form of recommendation for official use between regulators, nonetheless some of them give most realistic results. Some of them implement diffraction and reflections from terrain obstacles, even using a method similar to ray-tracing in 3D field simulation technique.

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8.1 Role and need of speciﬁcations

Standards such as DVB-T and DVB-T2 give only definitions of transmitted signals. They do not tell the designers how to build a receiver and what architecture shall be implemented. However the receiver architecture influences the receiver parameters such as the noise figure (NF), sensitivity, robustness against interference, etc. In order to design a receiver, some requirements must be defined to fulfill. Such requirements are written in a form of specifications. Thanks to specifications, a designer is therefore able to assess whether the design is fulfilling certain requirements or not. Specifications usually define receiver characteristics in all layers — from physical and transport layer to application layer and human factors such as the remote control layout. For the purpose the of this work the physical and transport layer requirements are relevant [121]. In most catalog cards of commercial RF front-ends, NorDig compliance is declared. However receiver parameters are also defined in coordination agreements but these touch only radio parameters and neglect upper layer behavior. It shall be mentioned that Chester’97 and RRC’06 aimed at describing typical receiver parameters in order to make assumptions for radio planning, whereas NorDig [34, 35, 36] or IEC 62216 (called the E-Book) defined minimum requirements . Most network planning is done now under RRC’06 and until 2006 it was done using Chester’97. Most important specification requirements are gathered in Table 8.1. Detailed explanations on the cited values are given in the following sections of this chapter.

8.2 Common guidelines of speciﬁcations

8.2.1 Receiver performance The most important parameter of the receiver operation quality is BER (bit error rate), in DVB-T measured before the RS decoder. BER is the

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Table 8.1: Comparison of cited speciﬁcations

Minimum Permissible and I/ C [dB] [dB] [dB]

maximum[dBc] in a channel x [dBm] requiredIL type: NF P

∗ Maximum Maximum input level C/ N [dB] oﬀset [kHz] adjacent other image ETSI EN 3.5–20.2 — — — — — — — — 300 744 Chester’97 3.1–20.1 7 — — — — 40 40 40 RRC’06 4.9–22.3 7 — — — — 30 — — IEC 62216 5.6–23.0 8 2.5 −33 166.67 −35 27–29∗∗ 40 29–39∗∗ VHF: — NorDig 2.0 5.1–22.5 7 — — 50 −35 28 38 UHF: 28 DGTV D-Book EN 300 8 3 — 166.67 −28 25 50 30 and 744+IL AGCOM 5.6 KIGEiT lack for 8 2.5 −33 50 −35 30 40 30 project R 7/ 8 ∗ in Gauss channel the data are valid for QPSK R 1/ 2 and for 64-QAM R 7/ 8 ∗∗ according to the system variant

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primary parameter, which describes the quality of the digital transmission link. The BER is defined as the ratio between erroneous bits and the total number of the transmitted bits. All of the presented specifications and other documents refer to the same definitions of quasi error-free reception (QEF) from EN 300 744 [3]. For DVB-T Quasi Error Free (QEF) reception is defined as occurrence of less than one uncorrected bit error per hour in the entire Transport Stream or in the elementary stream of a service. It corresponds to BER 10−11 after Reed-Solomon decoder at the input of the MPEG-2 Transport Stream demultiplexer. This leads to BER 10−4 after Viterbi decoder, which is so called reference BER and it is the basis for network planning and coverage calculations. The target performance for DVB-T2 is defined as less than one uncorrected error per hour at the level of 5 Mbit/ s single TV service decoder. According to the standard, the new DVB-T2 QEF corresponds to a Trans- port Stream Packet Error Ratio PER < 10−7 before the demultiplexer. Some other performance criterion is picture failure point. It is defined in specifications as the minimum C/ I value causing more than 1 TS erroneous packet in a 10 s period, with a correction component added to it, value of which is depending on measurement conditions [39]. QEF criterion is more restrictive than picture failure point and the required C/ N is higher for QEF than for the picture failure point. In all the specifications described below, the front-end shall be able to tune to center frequencies given by formulas (8.1) and (8.2).

fc 114 MHz + K · 8 MHz, K ∈{0, 1,... ,93} (8.1)

fc 107.5 MHz + L· 7 MHz, L ∈{0, 1,... ,27} (8.2)

8.2.2 Common receiver model Most speciﬁcations assume that the receiver consists of the following components:

the RF front-end includes the tuner and demodulator

Tuner thanks to ideal automatic gain control (AGC) provides constant power level at its output, the tuner noise ﬁgure NF is the noise ﬁgure of the entire front-end and for simplicity it is assumed that it is independent from frequency; one shall bear in mind that in real front-ends NF depends on frequency.

Demodulator includes a channel corrector, which compared to an ideal demodulator has an implementation loss in dB denoted IL.

There is an additional noise source in the receiver; the level of this noise is denoted Px and the level proportional to the wanted signal C.

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Figure 8.1: Typical DTT RF front-end assumed in speciﬁcations

This is why the level of this additional noise is given in relative units dBc. This kind of noise models e.g. the phase noise of the tuner’s LO, quantization noises of AD converters, etc.

A typical front-end model with the above-mentioned features is shown in ﬁgure 8.1 At the demodulator output the signal to noise takes the form of (8.3).

C C/ N (8.3) kTBF + CPx ≈ . · −23 J where k 1 38 10 K is the Boltzmann constant, T 290 K is the reference temperature. Chester’97 and RRC’06 are similar, however they neglect the source of the additional noise of the receiver Px. What is more, the RRC’06 neglects the implementation loss. It is assumed in all the speciﬁcations that the input impedance of the receiver is 75 Ω. It is usually required that the input RF connector shall comply IEC 60169-2.

8.2.3 Propagation channel assumptions Most speciﬁcations look into three types of propagation channels: Gaussian, Rician and Rayleigh. Gaussian channel is characterized by adding white Gaussian noise to the wanted signal. The Rician channel is the one with dominant direct path and a number of reﬂected signals. Rayleigh channel is a channel with the direct path absent.

8.2.4 Image channel issue

Most tuners work on the basis of single conversion from fc to fIF, usually fIF ≈ 36 MHz. Shifting the signal toward IF requires a local oscillator working at fLO fc + fIF. In the process of mixing, apart from the wanted signal of frequency fc, which is converted into fLO, also signals from so called image channel add to IF signal. The image channel frequency equals fim fLO+fIF fc+2fIF. The image channel signal shall be ﬁltered out by the tracking ﬁlter, however in real receivers robustness against image channel remains around 10 dB worse than against other unwanted channels.

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8.3 ETSI EN 300 744 norm

Although ETSI EN 300 744 [3] as the DVB-T norms itself, defines generally the transmitted signal only, there are some assumptions made about the receiver parameters and theoretical performance is given. In the norm tables were given including theoretically calculated C/ N for Gaussian, Rician and Rayleigh channels 8.2. It is assumed in the norm, that the receiver performs ideal channel estimation and phase noise is neglected. Rician and Rayleigh channels were defined by giving equations for signals in time domain for each channel type. Those equations use common for both channels echo profile consisting of 20 paths. The norm defines relative attenuation and phase shift for each path. Due to such a high number of paths, models from [3] are not practical for testing real receivers. This is why in some other specifications, e.g. NorDig, echo models were simplified.

Table 8.2: Required C/ N for non-hierarchical transmission in DVB-T 8 K mode System variant Required C/ N [dB] in a channel of type: Modulation R Gauss Rice Rayleigh QPSK 1/2 3.1 4.1 5.9 QPSK 2/3 5.3 6.1 9.6 QPSK 3/4 6.3 7.2 12.4 QPSK 5/6 7.3 8.5 15.6 QPSK 7/8 7.9 9.2 17.5 16-QAM 1/2 9.3 9.8 11.8 16-QAM 2/3 11.4 12.1 15.3 16-QAM 3/4 12.6 13.4 18.1 16-QAM 5/6 13.8 14.8 21.3 16-QAM 7/8 14.4 15.7 23.6 64-QAM 1/2 13.8 14.3 16.4 64-QAM 2/3 16.7 17.3 20.3 64-QAM 3/4 18.2 18.9 23.0 64-QAM 5/6 19.4 20.4 26.2 64-QAM 7/8 20.2 21.3 28.6 source: EN 300 744 [3]

Values from Table 8.2 were derived theoretically and shall be corrected according to the realistic receiver model; e.g. AGCOM 216/00 [122] there is a reference to Table 8.2 with implementation margin added.

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8.4 Chester’97 Coordination Agreement

The receiver noise ﬁgure in any case equals NF 7 dB and the implementation margin always equals IL 3 dB. Protection ratio for DVB-T broadcasts in adjacent and image channels were assumed 40 dB, however it was admitted that the value was a result of lack of experimental data. The required C/ N in Chester’97 is given it Table 8.3. The Table 8.4 shows co-channel protection ratios for DVB-T interfered by DVB-T in 2K mode.

Table 8.3: Required C/ N

System variant Required C/ N [dB] in a channel of type: Modulation R Gauss Rice Rayleigh QPSK 1/2 3.1 3.6 5.4 QPSK 2/3 4.9 5.7 8.4 QPSK 3/4 5.9 6.8 10.7 QPSK 5/6 6.9 8.0 13.1 QPSK 7/8 7.7 8.7 16.3 16-QAM 1/2 8.8 9.6 11.2 16-QAM 2/3 11.1 11.6 14.2 16-QAM 3/4 12.5 13.0 16.7 16-QAM 5/6 13.5 14.4 19.3 16-QAM 7/8 13.9 15.0 22.8 64-QAM 1/2 14.4 14.7 16.0 64-QAM 2/3 16.5 17.1 19.3 64-QAM 3/4 18.0 18.6 21.7 64-QAM 5/6 19.3 20.0 25.3 64-QAM 7/8 20.1 21.0 27.9 source: Chester’97 [123]

Table 8.4: Co-channel protection ratios for DVB-T interfered by DVB-T in 2 K mode System variant Protection ratio [dB] in a channel of type: Modulation R Gauss Rice Rayleigh QPSK 1/2 5 7 8 16-QAM 1/2 13 14 16-QAM 3/4 14 16 20 64-QAM 1/2 18 19 64-QAM 2/3 19 20 22 source: Chester’97 [123]

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8.5 Final acts off RRC’06, GE’06 coordination agreement

As a result of the coordination conference in Geneva in 2006, known as GE’06, a coordination agreement was signed between countries of ITU region 2. The Final Acts [111] were released comprising propagation model and its curves, protection ratios, etc. The purpose of this conference was to prepare a frequency plan and coordination methodology for implementation of DVB-T in ITU region 2. Four reference networks (RN) have been defined to cover possible applications of DVB-T networks. Reference Planning Configurations have been defined to cover possible modes of reception: RPC1, RPC2, , and RPC3. Frequency planning configurations may be grouped with respect to reception mode and frequency. The reception modes have been divided into three categories:

RPC1 - ﬁxed reception

RPC2 - porTable outdoor reception, mobile reception and lower coverage quality porTable indoor reception

RPC3 - porTable indoor reception of higher coverage.

Besides reference planning, also different channels models were proposed: PO - porTable outdoor channel, PI - porTable indoor channel and MO - mobile channel. Two reference frequencies have been selected: 200 MHz from the VHF band and 650 MHz from the UHF band. In annex 3.2 to [111] there were C/ N values defined as well as minimum median field strengths of electric field for all DVB-T variants. The C/ N values are cited in Table 8.5. In annex 3.4 formulas were defined for calculating required electric field strengths based on receiver architectures and assumptions made on the receiving installation. The noise figure was assumed as always equal NF 7 dB. In adjacent channels the protection ratio is assumed 30 dB. Also co-channel protection ratios were given, in Table 8.6 protection ratios for DVB-T service interfering another DVB-T service is given.

8.6 IEC 62216 norm

Implementation margin is deﬁned as IL 2.5 dB, additional noise introduced by the receiver equals Px −33 dBc. Noise ﬁgure of the receiver is assumed NF > 8dB. According to this norm the receiver shall deliver the reference BER for signal levels greater then Pmin where

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Table 8.5: Required C/ N for diﬀerent DVB-T variants

System variant Required C/ N [dB] in a channel of type: Rice Rayleigh Modulation R Gauss FX PO PI MO QPSK 1/2 4,9 5,9 8,1 8,1 11,1 QPSK 2/3 6,8 7,9 10,2 10,2 13,2 QPSK 3/4 7,9 9,1 11,5 11,5 14,5 QPSK 5/6 9,0 10,3 12,8 12,8 15,8 QPSK 7/8 9,9 11,3 13,9 13,9 16,9 16-QAM 1/2 10,6 11,6 13,8 13,8 16,8 16-QAM 2/3 13,0 14,1 16,4 16,4 19,4 16-QAM 3/4 14,5 15,7 18,1 18,1 21,1 16-QAM 5/6 15,6 16,9 19,4 19,4 22,4 16-QAM 7/8 16,1 17,5 20,1 20,1 23,1 64-QAM 1/2 16,2 17,2 19,4 19,4 22,4 64-QAM 2/3 18,4 19,5 21,8 21,8 24,8 64-QAM 3/4 20,0 21,2 23,6 23,6 26,6 64-QAM 5/6 21,4 22,7 25,2 25,2 28,2 64-QAM 7/8 22,3 23,7 26,3 26,3 29,3 source: RRC’06 [111]

Table 8.6: Co-channel protection ratios for DVB-T interfered by DVB-T

System variant Protection ratio [dB] in a channel of type: Rice Rayleigh Modulation R FX PO PI MO QPSK 1/2 6.0 8.0 8.0 11.0 QPSK 2/3 8.0 11.0 11.0 14.0 QPSK 3/4 9.3 11.7 11.7 14.7 QPSK 5/6 10.5 13.0 13.0 16.0 QPSK 7/8 11.5 14.1 14.1 17.1 16-QAM 1/2 11.0 13.0 13.0 16.0 16-QAM 2/3 14.0 16.0 16.0 19.0 16-QAM 3/4 15.0 18.0 18.0 21.0 16-QAM 5/6 16.9 19.4 19.4 22.4 16-QAM 7/8 17.5 20.1 20.1 23.1 64-QAM 1/2 17.0 19.0 19.0 22.0 64-QAM 2/3 20.0 23.0 23.0 26.0 64-QAM 3/4 21.0 25.0 25.0 28.0 64-QAM 5/6 23.3 25.8 25.8 28.8 64-QAM 7/8 24.3 26.9 26.9 29.9 source: RRC’06 [111]

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Pmin −97.2 dBm + C/ N[ dB] for 8 MHz channels (8.4)

Pmin −97.8 dBm + C/ N[ dB] for 7 MHz channels (8.5)

and C/ N values are given in Table 8.7.

Table 8.7: Required C/ N for for reaching reference BER

System variant Required C/ N [dB] in a channel of type: Modulation R Gauss Rice Rayleigh QPSK 1/2 5.6 6.1 7.9 QPSK 2/3 7.4 8.2 10.9 QPSK 3/4 8.4 9.3 13.2 QPSK 5/6 9.4 10.5 15.7 QPSK 7/8 10.2 11.2 19.0 16-QAM 1/2 11.3 12.1 13.8 16-QAM 2/3 13.7 14.2 16.8 16-QAM 3/4 15.1 15.6 19.4 16-QAM 5/6 16.1 17.0 22.1 16-QAM 7/8 16.5 17.6 26.1 64-QAM 1/2 17.0 17.3 18.7 64-QAM 2/3 19.2 19.8 22.1 64-QAM 3/4 20.8 21.4 24.8 64-QAM 5/6 22.1 22.9 29.4 64-QAM 7/8 23.0 24.0 33.9 source: IEC 61126 [33]

Picture failure point correction component is deﬁned equal to 0.5dB. According to the norm the receiver shall guarantee reception up to the maximum input level −35 dBm. Protection ratios against interfering signals were given as in Table 8.8.

Table 8.8: Receiver immunity against interfering DVB-T signals (values are valid for 2 K and 8 K modes and all guard intervals)

Maximum I/ C [dB] in a channel of type: System variant adjacent other image 16-QAM, R 1/ 2294039 16-QAM, R 2/ 3294036 16-QAM, R 3/ 4294035 64-QAM, R 2/ 3274031 64-QAM, R 3/ 4274029 source: IEC 62216 [33]

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Receiver shall enable reception of reference BER in existence of two paths of relative delays from 0.2 µsto0.9 µs regardless of amplitudes and phases of such paths. In multipath environment in MFNs so called long echo proﬁle is deﬁned, which is shown i Table 8.9. Receiver shall achieve the reference BER if C/ N ≥ 24.2 dB in the system variant agreed as a / Δ/ / representative variant, namely 64-QAM, 2K, R 2 3, TU 1 32.

Table 8.9: Long echo proﬁle

Path Delay Attenuation index [ µs ] [dB] 10 0 25 9 31422 43525 55427 67528 source: IEC 62216 [33]

For SFNs a short echo proﬁle is deﬁned and it is cited in Table 8.10.

Table 8.10: Short echo proﬁle

Path Delay Attenuation index [ µs ] [dB] 1 0 2.8 2 0.05 0 3 0.4 3.8 4 1.45 0.1 5 2.3 2.6 6 2.8 1.3 source: IEC 62216 [33]

8.7 NorDIG speciﬁcation

In [34, 35] receiver requirements are deﬁned in chapter 3.4 “Terrestrial Tuner and Demodulator”. Also an addendum to version 2.1 was issued where changes compared to DVB-T were marked out [36]. The new NorDig receiver [35] shall include at least one tuner and demodulator for reception of signals from terrestrial transmitters broadcasting a signal compatible with the DVB-T or DVB-T2. It means that a DVB-T2 receiver shall also be backward compatible with DVB-T. The 8 MHz and 7 MHz channel widths are mandatory in the UHF band. The 8 MHz raster is optional in VHF.

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The support for 1.7 MHz channels in VHF is optional for receivers issued before 2012. In 2012 and later, the NorDig compatible receiver shall also support the 1.7 MHz raster. If the receiver supports 8 MHz as well as 7 MHz rasters in the VHF band, the receiver shall automatically detect, which bandwidth is used. Reception of both hierarchical and non-hierarchical modes is required. The receiver shall support both the normal and extended carrier modes.

Pmin −105.7 dBm + NF[ dB] + C/ N[ dB] for 7 MHz channels (8.6)

Pmin −105.2 dBm + NF[ dB] + C/ N[ dB] for 8 MHz channels (8.7)

The required C/ N (cited in Table 8.11) was deﬁned for two interference proﬁles:

Proﬁle 1: the interfering signal is a Gaussian noise

Proﬁle 2: receivers get the signal from direct path as well as from a delayed path of delay from 1.95 µsto0.95 of the Δ. The delayed path is of equal power of the main path and there is no phase of frequency shift.

Table 8.11: Required C/ N for achieving reference BER

System variant Required C/ N [dB] Modulation R Proﬁle 1 Proﬁle 2 QPSK 1/2 5.1 8.8 QPSK 2/3 6.9 13.7 QPSK 3/4 7.9 17.4 QPSK 5/6 8.9 — QPSK 7/8 9.7 — 16-QAM 1/2 10.8 13.3 16-QAM 2/3 13.1 17.9 16-QAM 3/4 14.6 22.1 16-QAM 5/6 15.6 — 16-QAM 7/8 16.0 — 64-QAM 1/2 16.5 19.0 64-QAM 2/3 18.7 19.0 64-QAM 3/4 20.2 27.6 64-QAM 5/6 21.6 — 64-QAM 7/8 22.5 — source: NorDig [36]

The required C/ N from Table 8.11 in order to achieve QEF are given Δ/ / assuming TU 1 4 and the 8K mode.

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NorDig compliant receiver shall perform correct channel estimation / with path delays up to 7 24TU regardless of the echo profile. With respect to variants defined as reference (8K, 64-QAM, R 2/ 3, Δ/ / / Δ/ / / TU 18; 8K, 64-QAM, R 23, TU 14; 8K, 64-QAM, R 34, Δ/ / TU 14), the receiver shall always achieve QEF in a channel with two paths of relative delay from 1.95 µsto0.95Δ regardless of relative amplitudes and phases. The receiver must also achieve QEF in the case when the delayed signals arrive outside the guard interval, They must however fulfill parameters shown in Table 8.12, the example is cited for 8 MHz channels.

Table 8.12: Delays and relative attenuations of signals outside guard interval Relative delay −260 −230 −200 −150 −120 120 150 200 230 260 [ µs] System variant Relative attenuation [dB] 8 K 64-QAM 15 — 13 10 5 5 10 13 — 15 / Δ/ / R 2 3, TU 1 8 8 K 64-QAM 10 5 — — — — — — 5 10 / Δ/ / R 2 3, TU 1 4 8 K 64-QAM 12 6 — — — — — — 6 12 / Δ/ / R 3 4, TU 1 4 source: NorDig [36]

According to [36], the receiver shall be able to automatically detect network parameter change from normal to extended carrier mode and back. It is interesting that NorDig required support for TFS before 2012, however this was not followed by the manufacturers. Single PLP and multiple PLP mode support is required with the maximum allowed number of PLPs equal to 255. NorDig requires a DVB-T2 IRD to show DVB-T2 system ID as well as PLP ID. In [36] a couple of features were marked optional, such as:

the receiver is not required to support generic streams (GS or GSE), however the existence of them shall not cause faulty operation

carrier modes 16 K and 32 K are not required for 1.7 MHz channels.

Receivers shall receive signals with an oﬀset up to 50 kHz. RF input is an IEC female connector with input impedance of 75 Ω. The required maximum Noise Figure (NF) depends on frequency band. Selected NF requirements from [34], with additional S Band requirements from [34] omitted, are presented in Table 8.13.

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Table 8.13: Maximum receiver NF from [34] Band Noise Figure (NF) VHF band III 5 dB UHF band IV and V 8 dB

Besides standard DVB-T transmission parameters recovered by the IRD from transport stream PSI/SI tables, namely: DVB triplet

– Original Network ID – Transport Stream ID – Service ID

Network ID.

8.8 DGTVi D-Book speciﬁcation and AGCOM 216/00 Resolution

In DGTVi D-Book [32] speciﬁcation and AGCOM 216/00 Resolution [122] the required noise ﬁgure equals NF 8 dB and the implementation margin is IL 3 dB. Maximum input signal not destroying QEF reception is −28 dBm at input impedance 75 Ω. Receiver shall enable QEF reception in the presence of two paths of relative delays from 0.2 µsto0.9Δ, regardless of phase or amplitude differences between the paths. The required C/ N are identical to those in EN 300 744 [3]. Immunity against interference was taken from [122]. In the case of 0 dB echo with the noise and interferers absent, the receiver must enable QEF when echo paths arriving with delays not longer that any length of Δ.

8.9 PolSpec

For the purpose of regulating the Polish market of DTT receivers, a speciﬁ- cation under the umbrella of then Ministry of Infrastructure was developed by representatives of STB and TV manufacturers, broadcasters, regulators and other professional organizations. The leading role for editorial works was taken by KIGEiT organization, which also issued the document [124] parallel to the regulation of the Ministry of 2009 [125]. Both documents are similar, however in [125] some requirements were mitigated compared to [124], however changes concerned not the physical and transmission layers.

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PolSpec is in principle based on IEC 62216 [33], from which the receiver model was taken as well as the required C/ N, correction tables, and other. The implementation margin is assumed IL 2.5 dB and additional noise source power is assumed Px −33 dBc. Minimum required C/ N are identical to those in [33] except for the variants with R 7/ 8. The STB shall receive signals in 7 MHz channels in VHF band and 8 MHz in UHF band. The demodulator shall automatically detect change of modulation, which is signaled on TPS sub-carriers in DVB-T. The recon- ﬁguration time after parameter change shall be less than 1 s. The receiver shall display C/ N and BER after Viterbi decoder. The input impedance shall equal 75 Ω. The input connector of IEC type connector shall support antenna power supply of 5 V voltage and 30 mA with short circuit protection. The unique feature is requirement that the reﬂected signal shall be attenuated by at least 6 dB. The RF front-end shall properly demodulate all transmission variants described in [3] including the hierarchical mode.

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In this Chapter architectures and parameters of digital terrestrial television receivers (tuners and demodulators) are analyzed and evaluated in order to find the right architecture for universal (backward or down compatible), low cost DVB-T2 consumer appliances as well as the right model for coverage calculations. The receiver models, that are currently used in the radio planning considerations, are far too simplified and thus not adequate [126, 127]. Therefore, in this work the tuner and demodulator structures are discussed together, which means the analysis of interfaces from N to Z [55] with all elements between (cf., Fig. 10.1). Receivers shall automatically detect, which transmission mode is used. This is possible using transmission parameter signaling, which is transmitted using carriers of known parameters such as modulation type and constellation. The most important parameter indicating the receiver operation quality is the bit error rate (BER) before the Reed-Solomon (RS) decoder (module X in Fig. 10.1). BER is defined as the ratio between erroneous bits and the total number of transmitted bits [128]. Thus, it is the primary parameter, which describes the quality of the digital transmission link.

9.1 Digital TV receiver elements and interfaces

According to ETSI TR 101 290 [55], the DTT receiver may be decomposed into elements shown in Fig. 10.1 and the following elements inﬂuence signal coverage and thus the radio planning methods in DTT:

tuner, which selects the desired channel and converts it to the standard intermediate frequency in a way it rejects unwanted signals, especially in adjacent and image channels

demodulator, where the intermediate frequency signal is sampled, OFDM symbols are synchronized, demodulated and the convolutional code is decoded; the resulting MPEG-2 transport stream (TS) is sent to the demodulator output.

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In diversity reception the receiver itself may have several tuners and demodulators connected with each other. Diversity data containing symbol reliabilities are transmitted between tuners via a dedicated interface. One of the tuner and demodulator pairs takes the role of the master unit, in which the MRC algorithm is implemented. The output MPEG-2 TS is produced in this case at the output of the master unit. Most of tuner and demodulator modules, usually called RF front-ends in the field of digital television, send additional information beyond the demodulated transport stream. Among these the most important is the signal strength indicator. DTT receivers may have a couple of automatic gain control (AGC) points where first the AGC loop measures the wideband RF signal power. Some receivers enable using an external AGC that uses its own AD converter [129]. In DVB-T tuners, power measurement for AGC loop is performed in several points, i.e.: before Viterbi decoder, after it, as well as after the Reed-Solomon decoder. Seven bits AD converters are often used for this purpose. Some elements of the receiver do not effect quality of reception, nor radio planning and belong to such fields as: coding and compression of video and audio components, conditional access systems, middleware and a several other fields.

9.2 Tuners

Tuner’s on-chip architectures are typically built in CMOS (or BiCOMS) technology are integrated into a single circuit with 2.7 V or 3.3 V power supply, part of which is compatible with the TTL voltage level. Current architectures are implemented on the main stream 0.35 µmor0.18 µm CMOS technologies. As a part of the project described in [130], a prototype in the 0.35 µm CMOS technology was built. Power consumption of the overall device is close to 1 W, the maximum value does not exceed 1.5W. The output peak signal is close to 1 Vpp, with 500 Ω output impedance. Tuner variable gain falls between 65 dB and 90 dB [33, 126, 127, 131, 132, 129, 133, 134, 135, 136, 130, 137]. Typically tuners accept signals in the range as in [128], -91.6–35 dBm. The RF front-end NF is typically 8 dB Most tuners use the super-heterodyne architecture. In the first stage the input signal is amplified by LNA. Placing LNA at the beginning of the chain maintains noise figure on the acceptable level. Then the LO, mixer, and PLL downconvert the RF signal into IF. The requirement for LO is relatively low phase noise. The LO is a voltage controlled oscillator providing a clock signal for the demodulator. The main tuner architectures are:

single IF conversion (Fig. 9.3)

double IF conversion (Fig. 9.4)

quadrature conversion with a polyphase ﬁlter (PPF) (Fig. 9.5)

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Figure 9.1: Image channel problem

Figure 9.2: A classical single conversion tuner architecture with tracking ﬁlter and SAW ﬁlter

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Figure 9.3: A classical single conversion tuner architecture with tracking ﬁlter and SAW ﬁlter

Figure 9.4: Double conversion tuner architecture with two intermediate frequencies

Figure 9.5: Quadrature conversion with a polyphase ﬁlter (PPF)

Figure 9.6: Double IF conversion with I and Q streams

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double IF conversion with I and Q streams (Fig. 9.6)

Nowadays, most typical consumer tuners are based on the so-called single conversion scheme (Fig. 9.3) and contain tracking filters [138, 139]. Note that the performance of this kind of tuners depends on the quality of the tracking filter [126, 127]. In the classical design, after the tracking filter, a selective surface acoustic wave (SAW) filter is placed. This kind of filter because of its dimensions cannot be implemented on-chip.

9.2.1 Adjacent channel signals rejection Filters with the Q parameter higher than 300 are used for final filtering of IF signals before passing them to AD converters. It should be mentioned that the Q parameter, the 3 dB filter width B3dB and the resonance ω frequency 0 are connected with the following equation ω B 0 . (9.1) 3dB Q . ω / π ≈ . Assuming B3dB 761 MHz and Q 300 we get f0 0 (2 ) 36 3 MHz, which is the most often used value for intermediate frequency. Surface acoustic wave (SAW) filters are usually used in order to achieve sufficient Q parameter. Typical SAW filters are not integrated on-chip and are adjusted for the specific channel width. Compatibility with 6, 7 and 8 MHz channel widths is realized using three switched SAW filters or with one SAW filter with 8 MHz passband but with additional digital filter after the AD converter.

9.2.2 Image channel signals rejection As it was already stressed above, the single conversion architecture suffers from necessity of using a tracking filter for image channel rejection. The most popular intermediate frequency for European systems DVB-T and DVB-T2 is 36 MHz whereas for ATSC it is usually 44 MHz. In Europe such IF means that image channels are 72 MHz away from the wanted channel, which is the distance of 9 channels for 8 MHz channel width (Fig. 9.1). This is why receiver robustness against interference is worse for N +9 channels than other channel, e.g. N+ 4, where N is the wanted channel. In Fig. 9.2 an average single-conversion receiver C/ I after the tuner is shown for DVB-T variant C2 (64-QAM, R 2/ 3) for interfering DVB-T signals in differ-ent channels. Notice vulnerability to image channel shown in Fig. 9.2. For this reason double conversion architecture is employed with the first intermediate frequency over 1 GHz and down-conversion into the second intermediate frequency equal for instance 36.167, 44, or 4.57 MHz (Fig. 9.4). In this case the image channel falls outside the television

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band. The first IF above the television band is used also in order to solve the problem of local oscillator (LO) phase noise, which is hard to reduce when applying monolithic design. The receiver architecture proposed in [140] uses a fixed first intermediate frequency of 1.2 GHz that moves the first local oscillator range from 1.37 to 2.06 GHz. The second IF equals 4.57 MHz and for that reason the second local oscillator frequency is fixed at 1.20457 GHz. This approach is the so-called low-IF solution. In result, two SAW filters need to be used in this double conversion architecture. Intermediate frequency signals need to be amplified for nominal AD converter signal level and full dynamic converter range should be used. Signals from I and Q paths are sampled with 18.286 MHz frequency in typical AD converters. However, there are solutions with I and Q signals generated after tuner, namely in the demodulator. Image channels may be efficiently rejected by polyphase filters (PPFs). This kind of filters can distinguish between positive and negative frequency in the input signals. This feature is possible because harmonic components of the input signal are given in several phases at the same time, as shown in Fig. 9.7.

Figure 9.7: Polyphase ﬁlter input and output signals

Multiplication is carried out for frequency conversion in mixers. Let us assume that local oscillator angular frequency equals

ωLO ωIF − ωu (9.2)

where ωIF is the IF and ωu is the desired signal frequency. Mixing of the LO signal with the desired one may be expressed as

cos (ωut + θ) · cos ωLOt cos (ωut + θ) · cos ((ωIF − ωu)t) 1 1 cos (ω t + θ)+ cos ((2ω − ω )t + θ) . (9.3) 2 IF 2 u IF

Now let the image signal has frequency ωim ωu − 2ωIF. Now the result

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of mixing equals

cos (ωimt + θ) · cos ωLOt cos ((ωu − 2ωIF)t + θ) · cos ((ωIF − ωu)t) 1 1 cos (−ω t + θ)+ cos ((2ω − 3ω )t + θ) . 2 IF 2 u IF (9.4)

According to the last equation, the image channel signal has negative frequency, whereas the desired signal is of positive frequency. Passive polyphase ﬁlters may be used to reject the unwanted signals of negative frequencies (Fig. 9.8).

Figure 9.8: Passive polyphase ﬁlter architecture, used in [130] for image channel rejection

9.2.3 Direct conversion tuners

In direct conversion, the LO frequency equals the center frequency of the wanted channel. Thank to this the IF is reduced to zero and this method is also known as zero-IF (ZIF) [128]. The IF signal is obtained in the form of baseband I and Q paths. In order to gain this, the quadrature LO with two outputs is used. In this architecture, the phase diﬀerence between LO paths is 90° and any phase inaccuracies between I and Q paths increase IQ crosstalks and noise level. The practical problem of such architectures is reﬂecting LO signal back to the tuner input. This signal may be radiated back by the receiving antenna. Another problem is that during conversion of a VHF signal, the harmonics of the LO may introduce interference from the UHF channel and mix with the wanted signal in the VHF band.

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9.2.4 Local oscillators The double IF conversion architecture DVB-T receivers (Fig. 9.4) usually use one local oscillator with one external crystal. A signal from the oscillator is passed to two-channel frequency synthesizer and further those two signals are used by two independent VCOs or PLLs. Outputs of those are used as LO signals in mixers as it is shown in Fig. 9.9.a. On the contrary, phase shifters are used in the single IF conversion architecture with I and Q streams as shown in Fig. 9.9.b.

Figure 9.9: Local oscillators: a) double IF conversion architecture, b) single IF conversion architecture with I and Q streams

9.2.5 Surface acoustic wave filters The SAW filter operates in IF and rejects signals in adjacent channels. SAW filters have fixed bandwidth, dedicated for a specified system variant, e.g. 8 MHz in UHF band. Tuners designed for receiving signals of variable bandwidths have a number of SAW filters and switch between them. For example, a tuner may have three SAW filters for 6 MHz, 7 MHz, and 8 MHz signals. In older tuners SAW (Surface Acoustic Wave) filters were used both in the single and double conversion architectures. They were used as tuners used to be pure analog devices and SAW filters were the only that could provide a high Q parameter. As complexity of the RF front-end grew up in order to support DVB-T2, this drawback of usage of an analog filter was clear. In modern front-ends, tuner and demodulator are blocked and use digital filters of high Q. In the Author’s opinion, today, SAW filters should no longer be used in modern receiver designs, as sufficiently quick,

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Figure 9.10: IP3, IIP3 and OPI3 deﬁnition illustration

cheaper, more reliable, and more accurate digital ﬁlters can be applied instead [126, 127, 141].

9.2.6 Tuner parameters Noise figure Noise figure (NF) is defined as quotient of input to output circuit signal to noise, and defines noise added by elements of the circuitry. For a number of cascaded blocks of a receiver, where each has noise figure NF and power gain G, the total noise figure is defined as: − − − NF2 1 NF3 1 ··· NFN 1 NF1 + + · + + · · ...· (9.5) G1 G1 G2 G1 G2 GN−1 The first stages in the chain affect the NF most, thus low noise amplifiers are used on the tuner input. A low noise amplifier is characterized the Noise Figure value close to 4 dB, and gain reinforcement from 20 to 50 dB.

Linearity Receiver linearity is described by the third order Input Intercept Point (IIP3). In order to deﬁne IIP3 later on, let us deﬁne the third order Intercept Point (IP3): it is the point, in which the extrapolated power level of the third order intermodulation product intercepts with the extrapolated level of the two-tone desired signal. Extrapolations are calculated from the point, above which the intermodulation product power level increase is proportional to the input signal level. In the region above that point, every 1 dB increase of the input level results in the 3 dB increase of the third

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order intermodulation products. Deﬁnition explained above is illustrated in ﬁgure 9.10. Observing input power level for the IP3 point, IIP3 parameter can be calculated. If output signal power level is considered, OIP3 parameter is obtained. IIP3 and OIP3 are bound with the following equation:

− , IIP3 OIP3 G (9.6)

where G is the element gain. The total linearity of the receiver could be expressed by IIP3 of its N blocks by the equation

2 2 · 2 2 · 2 · ...· 2 1 1 A A A A A A − + 1 + 1 2 + ···+ 1 2 N 1 , (9.7) 2 2 2 2 2 IIP3 IIP31 IIP32 IIP33 IIP3N

where Ai is voltage gain of the i-th block out of N. Contrary to NF, IIP3 depends mainly on the linearity of the last element of a circuit. In work [140] it has been proved that IIP3 could be determined with the use of only 40 sub-carriers instead of simulating e.g. full 6817 DVB-T sub-carrier proﬁle in 8 K mode. Using this simpliﬁcation, the IIP3 can be determined with error not exceeding 0.1dB.

9.3 Demodulators

9.3.1 Channel estimation and equalization There are two types of channel estimation for OFDM: parametric and non- parametric. The parametric method uses a deterministic channel model, which assumes a ﬁnite number of delaying paths. In order to estimate the entire channel in parametric manner, the phases and gains of every delaying path shall be found. On the other hand, the non-parametric method use some simplifying channel assumptions. Such non-parametric method is used in the case of DVB-T and DVB-T2 receivers. It consists in estimat- ing the pilot signals and interpolating the channel. In order to estimate the CFR (Channel Frequency Response) as well as CIR (Channel Impulse Response), both in DVB-T and DVB-T2, pilots are used. The channel equalizer block in a typical DVB-T receiver estimates CFR by measuring of amplitudes and phases of pilot sub-carriers. In most designs channel equalizer estimates the CFR by a division of incoming symbols from pilot sub-carriers by symbols stored in a look-up table (LUT). For data sub- carriers between pilots, CFR is interpolated. In SFNs with an increase of distances between transmitters, aliasing phenomenon may occur due to channel transfer function sampling and restricted number of pilot sub- carriers. Assuming 8 K mode with 8 MHz channel width and supposing that TPS sub-carriers do not take part in the channel transfer function estimation, we obtain the average number of 701.25 pilot sub-carriers per

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one OFDM symbol. Precisely there are 701 sub-carriers in symbols with l mod 4 0, 1, 2 and 702 sub-carriers in symbols with l mod 4 3 where l is the symbol index in frame and the sets of continual and scattered pilots indexes have from 43 to 45 common elements. According to Nyquist theorem it is possible to capture components of the transfer function with periods over 21.7 kHz, which corresponds to the signal arrival time difference of 1/ (27.2 kHz) 0.046 ms and the distance between transmitters equal to 13.8 km. A case with two transmitters is shown in Fig. 9.11. It can be proved that some aliasing can occur even with low distances between transmitters and it can aﬀect the receiver synchronization.

Figure 9.11: SFN with two transmitters, impulse response, and channel transfer function at receiver site

In DVB-T2 there is a number of pilot patterns out of which one may choose a trade-oﬀ between accuracy of channel estimation and loss of throughput. E.g. in PP1 the pilot pattern for the SISO mode of DVB-T2, if you assume that the channel is not variable over four symbols, you sample the channel every fourth sub-carrier.

9.4 Diversity, MISO, and MIMO

According to [77] and [52], diversity reception improves C/ N by 7 dB. Test diversity receivers for DVB-T used Maximum Ratio Combining (MRC) method. In the MRC algorithm, signals from multiple antennas are combined. In the project described in [100], MRC was implemented using multiple demodulator chips. The solution of simple connecting demodulator chips, e.g. DIB3000-M, has been patented, according to [100]. It connects chips in a cascade. Every chip gets a separate IF signal from a dedicated antenna. The first chip operates as a regular, non-diversity receiver. The second one, besides its IF signal, receives signal quality information from the first chip, the third from the second and so on. Finally, the last chip in the cascade produces the final MRC output based on the

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information from all chips in the chain. In the DIB3000-MC demodulator described in [52], in the diversity mode with two antennas the required C/ N for the TU6 proﬁle is up to 7 dB better compared to the single reception mode. The DIB3000-MC chips operates up to maximum Doppler frequency fd 120 Hz. In DVB-T2 special pilot sub-carrier pattern were designed for MISO. Two groups of transmitters broadcast inverted pilots. This information about phase inversion of selected pilots can be used for better channel estimation. The channel coeﬃcients H11,k, H12,k, H21,k, H22,k can be calculated from the following equations:

a a a a · , · , r1,k x1,k h11 k + x1,k h12 k + n1,k a a a a · , · , r2,k x1,k h21 k + x1,k h22 k + n2,k b b b b (9.8) · , − · r1,k x1,k h11 k x1,k h12,k + n1,k b b b b · , − · , r2,k x1,k h21 k x1,k h22 k + n2,k where

a b a b r1,k, r1,k, r2,k, r2,k are received symbols for the cases a and b from antennas 1 and 2 on k-th sub-carrier, respectively

a a b b n1,k, n2,k, n1,k, n2,k is noise for the cases a and b from antennas 1 and 2onk-th sub-carrier,

a b x1,k, x1,k are pilot sub-carriers for the cases a and b on k-th sub-carrier, there are no x with subscript 2 as the sub-carriers are dependent.

For MIMO demodulators, symbol synchronization is done using only one channel by correlating the guard interval with the symbol. Then the guard interval is removed and the FFT block is used as the demodulator itself. In 2×2 MIMO the transmittance matrix is of size 2×2. It is assumed that the radio channel is stationary over four subsequent symbols — thanks to this the sampling density gain of the channel characteristic in the frequency domain. After channel equalization, the C/ N is calculated for each data sub-carrier in order to determine its reliability for Viterbi decoder. The carrier C in the formula is the square of symbol magnitude. The noise N is calculated as interpolation of noise levels estimated using pilot sub-carriers.

9.5 SFN inﬂuence on receiver operation and FFT window positioning

Single Frequency Networks (SFNs) are networks of synchronized transmitters radiating identical signals in the same channel. In case of multipath propagation or single-frequency operation, a number of copies of a single signal reach the receiver with varying delays [142]. The weighting function tells us how much of a signal contribution from a given transmitter in a

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SFN is the wanted part and how much shall be considered as interference. The weighting function for the DVB-T and DVB-T2 shows the following attributes:

it is not symmetric

it has a cut-oﬀ time after which the arriving signal makes pure interference.

The longer the echoes in time domain, the denser the frequency notches in the frequency domain. Depending on the instant of arrival, those copies are useful signals or interference. To determine the ratio of those two components, i.e. useful signal versus interference, in radio planning the following formulas were derived: ⎧ ⎪ 0 for τ ∈ (−∞,Δ− T ] ⎪ p ⎪ τ 2 ⎨⎪ Tu+ τ ∈ Δ − , ⎪ T for [ Tp 0] wi ⎪ u (9.9) ⎪ 1 for τ ∈ [0,Δ] ⎩⎪ 3 for 3

The quadratic dependence may be explained as follows: the receiver decision of the most probable transmitted symbol is made based on the correlator output. In the OFDM modulation, the role of parallel correlators for each sub-carrier is played by the FFT block. The useful part weight of the signal, that arrived earlier before or later after the reference signal, shall be interpreted as the ratio deﬁning what would be the equivalent power of that signal if it arrived at the same instant as the reference signal, to which the receiver has synchronized. In coverage calculations there are diﬀerent methods to simulate the behavior of the receiver. For the purpose of simulation of an SFN operations, the receiver synchronization method must be assumed. The following methods of receiver locking are possible to choose from:

minimum delay or ﬁrst above threshold called the correlation-based synchronization (Fig. 9.12)

maximum level (Fig. 9.13)

maximum power (Fig. 9.14)

theoretically optimal, accumulating maximum useful signal power using equation 9.9, a theoretical model assumed in some propagation simulations

The minimum delay method (a.k.a. ﬁrst above threshold) synchronizes to the ﬁrst signal exceeding a given threshold. It is popular in many demodulators as it is relatively easy to implement using comparing the correlator output with a given triggering threshold. The maximum level method consists in searching the peak power value of the incoming signals.

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Figure 9.12: Minimum delay method for FFT window positioning

Figure 9.13: Maximum level method for FFT window positioning

Figure 9.14: Maximum power method for FFT window positioning

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The maximum power method means that the receiver searches for such positions of the window, that it uses maximum of incoming signal power. Using the last, optimal method in the receiver places the FFT window in such a way that the highest wanted ﬁeld strength is acquired. This method is rather complex to implement in real demodulators and is used most commonly in simulations.

9.5.1 Symbol Timing Recovery (STR) and Symbol Timing Offset (STO) Channel impulse response (CIR) may be estimated through correlation between the received signals and the pilots. It may be also done efficiently using the IFFT block, however if the STO is estimated with insufficient accuracy, ambiguity of CIR estimation emerges, which makes further fine STO estimation impossible. Also fine STR depends on the quality of the initial STO estimation. Unfortunately, conventional STR may recognize the final echo a circularly shifted pre-echo, which would find wrong FFT window position. Initial circular shift of the FFT window caused the ambiguity of CIR. To estimate accurate STO, the ambiguity effect must be reduced. The greater STO, the worse the found position of the FFT window. Such dislocation of the window causes ISI (Inter-Symbol Interference) and prevents from proper estimation of the CFR (Channel Frequency Re- sponse). It turned out that the best method is two-stage STR calculation for receivers working in SISO mode: coarse and fine. As mentioned in [114], the fine STO estimation depends on the quality of coarse STR. In a classical approach, coarse STR may be done using the MLE (Maximum Likelihood Estimation) method to find initial FFT window. Fine STR may be performed using a number of methods:

analyzing slope angle of phases of pilot sub-carriers, this method, although accurate, remains ineﬀective for fast varying channels in the frequency domain as it requires to analyze the phase of thousands of pilot sub-carriers

CIR (Channel Impulse Response) calculation using IFFT - rather in- eﬀective real in hardware implementations

suboptimal combination of diﬀerent steps:

– predeﬁning a set of possible STR times, calculation of SNR and than selecting the best variant – analyzing the inﬂuence of timing on constellation.

In MISO mode, the conventional STR is outperformed by the method described in [114]. Using the MISO channel, CIR was estimated ﬁrst by observing special pilot patterns.

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9.5.2 Impairments within receivers Carrier Frequency Oﬀset (CFO) causes reception quality degradation since: symbol on sub-carrier k will appear on some other sub-carrier k + l

phase shift will occur, growing with every OFDM symbol. There are a number of CFO estimators, e.g. Schmidt estimator, which finds offset for the sub-carrier, for which the correlation with pilots is maximal. In OFDM systems, CFO can spoil orthogonality. CFO is caused by the local oscillator inaccuracy in the receiver as well as the Doppler shift. The instability of the sampling clock in the receivers causes sampling frequency offset. The DIB3000-MC demodulator chip recovers ±300 kHz frequency offset [52] if such offset is used by the broadcaster for interference mitiga- tion purposes. In a real receiver, various implementation impairments are observed. Among these the most important error sources are:

symbol timing offset Td sampling frequency or timing offset Δt φ phase offset 0 carrier frequency offset Δf ICI (Inter-Carrier interference) for k-th carrier Ik. A received symbol on k-th sub-carrier with above-mentioned errors included may take the form of [142] equation

π Td π Δt(n) φ π Δ j(2 k T )+2 k T + 0+2 k fTs) . Rk HkXke u u + Ik + Nk (9.10)

9.5.3 LDPC decoding LDPC codes can be decoded using the Sum-Product Algorithm (SPA) based on message passing method. The LDPC codes selected for DVB-S2 and DVB-T2 are irregular type. The parity check matrix is a sparse matrix and | it takes the form H [H1 H2] where H2 is a staircase (lower triangular) matrix and H1 is a random sparse matrix with column weight from 3 to 13 depending on the code rate. The LDPC decoder consists of bit nodes and check nodes, where bit nodes are in groups of amount equal M. The submatrix H1 is periodic in DVB-T2 in order to reduce memory requirements M times in the receiver, where M 360 in DVB-T2. It is suﬃcient to specify check nodes for the ﬁrst bit node in a group while the check nodes for the rest of bits can be calculated. The Sum-Product Algorithm is the following: initialization

iterative process with bit and check node updates

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decision

Updates of the nodes may be found by using formulas for log-likelihood ratio (LLR) of messages that check nodes send to bit nodes. Diﬀerent suboptimal decoding methods were developed for receivers: piecewise linear (PW), Look-up-Table (LUT), Jacobian and Min-Sum. The LUT approach does not require any operations during the iterative phase as logarithmic function values for calculating LLR are stored in the receiver memory.

9.6 CD3 algorithm

To improve SFN operation and temper aliasing phenomenon the CD3 algorithm was proposed. The abbreviation CD3 (or CDDD) comes from Coded Decision-Directed Demodulation. CD3 is a method of demodulation with the feedback loop proposed in [131, 84] in order to improve operation in large single frequency networks (SFNs). In a classical demodulation, pilot sub-carriers are used for channel estimation. Contrary to this, in the CD3 method, the channel is estimated for every FFT position. Thus the channel transfer function may be calculated for every data sub-carrier. This is possible thanks to taking information after the FEC (forward error correction) decoder and comparing with the original received symbol. Signal after FEC decoder is reencoded, reinterleaved, remapped and then remodu- lated. However, some reference symbol is needed to lock-in the algorithm. In the DVB-T standard, the drawback is the frequency domain subsam- pling resulting in rapid degradation of the reception quality if echoes are outside the guard interval (GI). Due to aliasing, the receiver estimates the channel incorrectly. Using CD3, gradual degradation of reception capability is observed with the increase of delayed paths exceeding the GI. In the ﬁrst step of the algorithm, it operates with the feedback loop open. In this approach transmitting scattered pilots is not necessary and throughput may be increased by about 8 % for DVB-T. Typically this method is not implemented. Implementation of CD3 required some hardware modiﬁca- tions to the DVB-T receiver architecture. CD3 algorithm developed for the DVB-T was adopted as standard to DVB-T2. In DVB-T2 CD3 is a native mode for some pilot patterns.

9.7 Diversity receivers

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a dedicated antenna. The first chip operates as a regular, non-diversity receiver. The second one, besides its IF signal, receives signal quality information from the first chip, the third from the second and so on. Finally, the last chip in the cascade produces the final MRC output based on the information from all chips in the chain. In the DIB3000-MC demodulator described in [52], in the diversity mode with two antennas the required C/ N for the TU6 profile is up to 7 dB better compared to the single reception mode. The DIB3000-MC chips operates up to maximum Doppler frequency fd 120 Hz.

9.8 Recommended DVB-T2 receiver architecture

Note, that because of application of analog SAW filters, the interface between the tuner and the demodulator is analog. Thus, after the digital processing in the tuner, the signal is converted back to the analog domain in order to be passed to the demodulator [143, 144]. In modern tuners low intermediate frequency (IF) around 5 MHz is used [126, 127, 140]. In the Author’s opinion, from the perspective of the receiver architecture, both DVB-T and DVB-T2 standards are similar [145]. This is a very interesting and optimistic observation, as it makes it possible to chose a unique receiver architecture for both standards and achieve the so-called backward compatibility and even the “plug-and-play” like behavior, i.e., the universal future receivers. There are, however, some differences between the standards that may to some extent affect the receiver architecture. Among them are:

MISO (multiple input single output) and MIMO (multiple input multiple output) in DVB-T2 — in DVB-T the MRC (maximal-ratio combining) method for the diversity receivers is used

TFS (time frequency slicing) — still not implemented in commercially available front-ends.

Nevertheless, these factors do not cause essential diﬀerences and thus do not mean signiﬁcant costs to get a universal, i.e., backward (or down) compatible DVB-T2 receivers. Thus, the most convenient modern receiver architecture for both DVB- T2 and DVB-T standards seems to be the double frequency conversion architecture, i.e., this with two intermediate frequencies, as it is robust to noise in the image channel. In this receiver the coded decision-directed demodulation (CD3 or CDDD for short) should be used.

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10.1 Beginning of DVB-T in Poland

The Polish public service broadcaster, Telewizja Polska S. A., hereafter denoted as TVP, received in 2009 a frequency reservation for a nationwide coverage from the regulator UKE (Urzad ˛ Komunikacji Elektron- icznej) [124, 125]. This is called MUX-3. The respective set of the reserved frequencies consisted of final as well as temporary frequencies. Temporary frequencies were included due to still existing analog broadcasts until July 2013. Full migration toward the final frequency will take place in April 2014 and is possible thanks to the analog switch-off in Poland (ASO), which already took place on July 23rd, 2013 [146]. In 2009 a public tender was announced by TVP [147], which was won by EmiTel, the dominant broadcasting network operator in Poland [147]. The basic agreement obliged the contractor to launch transmission in four stages between October 2010 and April 2014. In MUX-3 a variant carrying 24.88 Mbit/s with 8K sub-carrier mode is used. Most transmitters have 64-QAM, R 3/ 4, Δ/ Tu 1/ 8 [3], however for some cases of large SFN cells, R 5/ 6 and Δ/ Tu 1/ 8 are used [3]. Note that those two variants have equal throughputs, which is convenient from the head-end and TS shaping point of view. In such a case throughput thresholds configured in all the multiplexers within the broadcasting network are the same.

10.2 DVB-T in progress

Because of settlements between TVP and the contractor, it was important to determine coverage of the service after launching. In the agreement

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between TVP and EmiTel, a defect was deﬁned as occurrence of at least one the following:

drop of the transmitted power higher than 2 dB

BER after the Viterbi decoder higher than QEF criterion

bad packet order (Continuity_count_error)

loss of TS synchronization (TS_sync_loss).

10.3 DVB-T measurements

The normative basis for the measurements was ETSI TR 101 290 [55, 148, 149]. Measurement campaign of MUX-3 was carried on in 2010 and 2011. In order to ensure disinterested results, they were performed by two teams and the results were compared. One measurement team was organized by EmiTel and the other was formed by Polskie Sieci Nadawcze (PSN), EmiTel’s competitor. Both teams measured signals independently, however in possibly similar conditions [12]. This measurements were based on a predefined set of points. These were selected in such a way that they were located near urban agglomerations, they laid inside the contracted coverage, and the points were easily available in large public areas, such as supermarket parking lots, etc. It was important to select places were there were no terrain obstacles near the reception points in order to get the measurement results repeatable and plausible. However, in some cases finding such points was impossible. In some cases dense urban environments were selected for the purpose to investigate differences between two teams. Such situations sometimes caused significant differences in the measurement results [150]. Both teams were fit with equipment that allows to measure or visualize such parameters as: MER (Modulation Error Ratio), EVM (Error Vector Magnitude), SNR (Signal-to-Noise Ratio), spectral density as a function of frequency, channel impulse response with Δ/ Tu overlaid, BER (Bit Error Rate) before and after the Viterbi decoder. Thus, majority of significant physical layer parameters were measured. Out of these the most important were RF frequency accuracy, RF channel width, and Symbol length. In the transport layer all measurements were carried out from first_priority, second_priority and third_priority tables from [55]. The PSN team could only measure signals from the air, thus on the interface N (Fig. 10.1) [55]. The procedure for both teams was the following:

ﬁnding out the azimuth of the transmitter site

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Figure 10.1: Interfaces of the transmitter and receiver deﬁned for measurements in [55]

searching for a place least obscured in the direction of the transmitter site

pulling out the antenna and rotating until the azimuth of the strongest signal is reached

measuring physical and transmission layers according to [55] and recording the stream at TVP’s head-end at the same time.

In the case of SFNs, the antennas were directed into the direction of individual transmitters. EmiTel, besides field measurements, also carried out on-site measurements. Transport streams were recorded on the spot as well as at the TVP head-end at the same time. Thank to this, it was possible to find weather the possible errors were due to the broadcasting network or whether they originated from the multiplexing process at the TVP’s head-and [14, 151]. As both teams worked independently, precisely described locations, read from the GPS (Global Positioning System) receivers, differ. The Emi- Tel team used a car equipped with a pneumatic 5 m high mast with a calibrated wideband log-periodic antenna (gain 4.7dBd at 600 MHz) and the calibrated cable. PSN used a similar car equipped with a pneumatic 10 m high mast with a calibrated wideband log-periodic antenna (gain 9.2 dBd at 600 MHz) and also the calibrated cable [152]. The from the air reached MER values were between 21 dB and 34, 3dB whereas the MER values measured directly after the transmitter and before

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Figure 10.2: Measured MUX-3 constellation, week signal of MER 18.1dB and SNR 20.5 dB, decoding impossible due to low ﬁeld strength

Figure 10.3: Measured MUX-3 constellation, strong signal of MER 37 dB, SNR 40.5dB

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the channel filter were in the range 29 dB and 41.8 dB with 36 dB as a typical, median value. In the specification of the MUX-3 tender [147], the minimum required field strength was 57 dBµV/ m. The measurements confirmed that signals below this level caused violating the QEF (quasi error free) reception. Interesting data were collected during EmiTel’s and PSN’s campaigns. The measured field strengths differed in 16 dB at the maximum and these could not be explained at the stage of drawing the reports up. Due to such inconsistency, the measurements in the most arguable points were repeated. To compare results as precisely as possible, also both cars were placed exactly at the same point. As a part of this experi- ment, signals from EmiTel’s antenna were directed to the PSN’s equipment and vice-versa. It turned out that in dense urban areas signal level measurements are strongly dependent on the reception point and the signal level differences reached even 16 dB. This means that measuring in urban area within a discrete set of points will not give satisfactory precision and the obtained values are scattered [153]. Due to such effects, in order to find coverage over a given region, the ITU-R SM.1875 [150] shall be used, in which it is recommended to measure signal levels in a grid of 500 m × 500 m size. If the distribution of the measured signal levels is found uniform, the grid element size shall be divided by 2 and the measurements shall be repeated. The results are recognized as correct when there is a clear median value in the distribution of signal levels [40].

10.4 DVB-T future trends

Poland started regular DTT broadcasts in October 2010 (MUX-2 and MUX-3), switched off analog TV in July 2013 and will complete the process of launching DVB-T multiplexes in April 2014. It is significantly late compared to the countries that now are in the process of the DVB-T2 roll out. The DVB-T standard was chosen in 2008 for the first Polish multiplexes. At that time the discussion about leaping DVB-T and taking DVB-T2 at once, as it was done in 2007 with respect to H.264 instead of MPEG-2, was quite intensive. However, DVB-T was chosen. Choosing DVB-T2 then would involve the following risks:

there were no RF front-ends supporting the draft of DVB-T2, thus not STBs were available, only manufacturers road-maps to DVB-T2 were announced, prices of the ﬁrst batches of STBs would be 3–4 times higher than DVB-T and H.264 compliant STBs

there was no ETSI standard at that time, only the draft A122 [22] existed

there were no DVB-T2 gateways in production at that time

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choosing a new standard would prolong huge delay of launching DTT in Poland, which had become at last an urgent issue; no- one would launch such a big project without gathering experience through, e.g., experimental transmissions, laboratory tests, re-planning of the network designs, etc.

In countries such as Great Britain introducing DVB-T2 was bound to offering nation wide FTA HD (free-to-air high definition) broadcasts. There is no such strong motivation to activate DVB-T2 in Poland because TVP already launched HD in DVB-T and the viewers would not accept another switch-off that would not bring any added value. On the other hand, Polish commercial broadcasters are not interested in offering HD content for free, in the contrary, in their business models they assume subscription-based HD packages in cable and satellite TV. It is however very plausible that as a part of the digital dividend a need for the DVB-T2 system would emerge in Poland in the VHF band. It was primarily dedicated for digital radio in DAB / DAB+ standard, however reserving one nationwide coverage channel for a new TV multiplex is feasible. Indeed, the Office of Electronic Communication in Poland (Urzad ˛ Ko- munikacji Elektronicznej or UKE for short) has just announced accessibility for the whole country of the band 174–230 MHz. The new multiplex is already roughly named the MUX-8. It should start after the year 2015. It is probable that this new MUX-8 will be shared by a number of providers and they will be highly interested to use individual physical layer pipes (PLPs) — a possibility offered only in the DVB-T2 standard. Thus activation of the DVB-T2 system in Poland seems to be realistic in a near future.

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In Chapters 7, 8 and 9 it has been shown that the considered models, specifications, and receiver architectures are closely interrelated. The receiver designs shall take into account how the RF signal arrives from transmitters. When designing a network using radio planning software, which imple- ments propagation models, it is also important to know how the receiver is built and what is its architecture. DVB-T2 introduces many changes relative to its predecessor DVB-T, i.e., the first generation DTTB system. These changes concern modulation variants, channel coding, utilization of space-time codes, signaling of transmission parameters, and many more. The price for the new standard is a much more complex signaling and more complicated implementation. Up to now there is no commercially available time frequency slicing (TFS) supporting tuner on the market. Although the differences are significant, from the Author’s investigations it turned out that the optimal architecture is interchangeable between the DVB-T and DVB-T2 standards. It means that the most suitable architecture for the DVB-T shall also be the best for the DVB-T2 standard. This observation can be considered as the main result obtained by the Author of this thesis on the basis of the carried investigations. Due to this statement, each new DVB-T2 receiver can be designed, with practically no additional costs, as the backward compatible device with the elder DVB-T standard. This means a substantial reduction of costs necessary for the equipment modernization (receiver and possibly the antenna system) for the consumers who would in future potentially be interested to switch to the new digital television standard, i.e., to DVB-T2 instead of DVB-T. This will in turn stimulate a commercial success of the whole process not only in the national but also in the global scale. From the Author’s considerations, it follows that the most convenient modern receiver architecture for both DVB-T2 and DVB-T standards is that with double frequency conversion, i.e., with two intermediate frequencies, as it provides more robustness to noise in the image channel. In such a receiver the coded decision-directed demodulation (CD3 or CDDD for short) should be used.

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Assuming an antenna system composed of a number of antennas, the receiver is composed of many tuners. In this case, in the Author’s opinion, the maximum ratio combining (MRC) algorithm should be implemented. On the other hand, application of the TFS mode is not necessary as the commercially tuners will not support it in a foreseeable future. Furthermore, the Author showed that the best solution consists in res- ignation from the intermediate frequency and the analog interface between the tuner(s) and the demodulator. As long as the tuner had been a purely analog device, production of the analog intermediate frequency signal would have been justified. Indeed, nowadays and in future, if we replace SAW filters with digital ones, conversion back to the analog domain and then again to the digital domain in the demodulator, will be irrelevant. Moreover, th Author stated that it is important to know how to build radio frequency (RF) front-ends properly in order to achieve robustness against various important effects, e.g. multipath propagation. Recapitulating:

most DTT tuners should be built with BiCMOS technology, which means that bipolar junction transistors and CMOS technology are integrated; in elder tuners selective analog filters were used both in the single and double conversion architectures; in modern front-ends, tuner and demodulator should be blocked and digital filters should be used (only when tuner and demodulator are separate blocks and SAW filters are present, as in classic solutions, interface between them must be the analog IF signal)

among single front-end tuners, the double conversion architecture tuner with the maximum power method symbol synchronization and the CD3 algorithm implemented, turned out to be theoretically the best for both DVB-T and DVB-T2 standards

obviously, the demodulator part is far more complicated in the case of the DVB-T2, which shall additionally be backward compatible with the DVB-T; however, the diﬀerences between the DVB-T and DVB- T2 are not fundamental to such extent that diﬀerent architectures shall be used; therefore, universal, low cost, and “plug and play” receivers, which are very important for the market success of the new DVB-T2 standard, are feasible in the near future

in demodulator, which is a digital signal processing device, the important issues in terms of reception quality are: symbol timing recovery (STR), symbol timing offset (STO), and FFT window positioning for SFN operation; using the main focus method, the receiver places the FFT window in such a way that the highest wanted field strength is acquired; this method of finding the FFT window position is rather complex to implement in real demodulators, however it gives the best results; moreover, the author wants to stress that the FFT algorithms

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The results obtained in this thesis, concerning features and design of the DVB-T2 receivers, occur to be particularly important taking perspectives of the DVB-T2 activation in Poland into account. Indeed, the Office of Electronic Communication in Poland (Urzad ˛ Komunikacji Elektronicznej or UKE for short) has just announced accessibility for the whole country of the band 174–230 MHz. As it was already mentioned in Chapter 10, this new VHF band placed multiplex, which is popularly referred to as the MUX-8, should start after the year 2015. It will very probably be shared with multiple providers. Thus a possibility offered by the DVB-T2 standard, namely organization of the individual physical layer pipes (PLPs) (i.e., the provider individual statistical groups) is very tempting and DVB-T2 will in near future be brought into practice in Poland. The problem of the choice of the proper DVB-T2 receiver architecture, considered in this thesis, will quite soon occur to be of primary technical importance in Poland and with the min- imization of the costs of the equipment modernization for the mass users will stimulate success of this process. With this statement the Author would finally like to conclude that the aims of his research presented in this work, formulated in Chapter 1, have been fulfilled, and the scientific thesis has been proved.

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