Conformal Body-Worn Smart Antenna System for Wideband UHF Operation

Dissertation

Presented in Partial Fulﬁllment of the Requirements for the Degree Doctor of Philosophy in the Graduate School of The Ohio State University

Gil Young Lee, M.S., B.S.

Graduate Program in Electrical and Computer Engineering

The Ohio State University

2012

Dissertation Committee:

John L. Volakis, Advisor Chi-Chih Chen, Co-Advisor Fernando L. Teixeira Dimitris Psychoudakis c Copyright by

Gil Young Lee

2012 Abstract

There is much interest in developing body-centric wireless communication systems

(BWCS) for mobile health care systems. However, the realization of a BWCS is challenging due to the body’s interference with the antenna’s operation. More specifically, body-worn antennas suffer from impedance detuning, pattern deformation, and gain reduction caused by the body. Therefore, it is important to consider these effects in evaluating body-worn antennas. In this regard, a diversity technique is proposed to improve body-worn antenna performance.

More speciﬁcally, a channel decomposition method (CDM) is proposed and used to evaluate body-worn antenna systems. The CDM signiﬁcantly reduces computation time when evaluate body-worn antennas and is applicable to various surrounding environments without recalculation of the more complex interaction. A second contribution of this dissertation is design of a diversity systems which automatically determines the minimum number of antennas while maximizing performance. This approach is employed to design body-worn antenna diversity systems for given communication scenarios. The results obtained via this process demonstrated that this simple method can substantially reduced computation time in designing body-worn antenna diversity system. As a demonstration of the proposed methodology, a vest- mounted UHF body-worn antenna diversity system (BWADS) is developed using 4 light-weight antennas. The proposed BWADS is transparent and unobtrusive to the

ii users but provides performance superior to commercial antennas. A variety of tests were performed to validate the proposed BWADS. It was found that the proposed

BWADS provided 7 dB (outdoor) to 16.5 dB (indoor) of higher gain as compared to commercial antennas. The dissertation concludes by proposing other applications of the developed body-worn antennas and design methods.

iii To my wife, daughters, and son

iv Acknowledgments

First and the foremost, I would like to thank Almighty God for bestowing His blessings upon me and giving me the strength to carry out and complete this work.

I would especially like to thank my advisor Professor John L. Volakis for all of the support and invaluable guidance that he provided in completing this work and in pursuing my Ph.D. degree. I cannot help admiring him for his integrity and diligence.

It is an honor for me to be his student.

I am also extremely grateful to my co-advisor Professor Chi-Chih Chen for all his support and advice in the technical details of my dorctoral work. He encouraged me to overcome many issues I have faced during my research and I really enjoyed academic discussion with him. He is like a brother to me and I look forward to working with him in future.

I would also like to express my deep gratitude to my good friend and colleague,

Dr. Dimitris. Psychoudakis, who has been very supportive and generous in sharing his knowledge. My gratitude is extended to Professor Fernando L. Teixeira and

Professor Philip Schniter for serving my committee and providing insightful ideas and suggestions. Their lectures also were always great and I learned a lot of valuable knowledges in electromagentics and signal processing from them.

v I offer my blessings to all the staff and fellow students of the ElectroScience Lab- oratory. I really want to thank for their valuable discussion and friendship. Spe- cial thanks to officemates - Jae-Young Chung, Yijun Zhou, Jing Zhao, Ming Chen,

Mustafa Kuloglu, Ugur Olgun, Erdinc Irci, Ioannis Tzanidis, and Tao Peng. My gratitude is also extended for other friends at ESL - Chun-Sik Chae, James Park, Jun

Seok Lee, Keum-Su Song, Kyung-young Jung, Haksu Moon, Pil Sung Park, Youngseo

Ko, Justin Kasemodel, Praphun Naenna, and William Moulder.

Finally, I would like to thank my family. The unconditional love and support of my parents and parents-in-law have been the greatest motivation during the graduate study. And most importantly, I would like to express the deepest gratitude to my wife, In Kyung, who always loved, encouraged, and supported me. I also thank to my two daughters and a son - Ye Eun, Ye Won, and Jum beom. You are all of my pleasures and hopes. I love you!

vi Vita

January 11, 1975 ...... Born - Kwangchon, Chungnam-Do, Korea March, 1997 ...... B.E., Electronics Eng., Korea Air-Force Academy, Cheongju, Chungbuk-Do, Korea February, 2001 ...... B.S., Electrical Eng. & Computer Sci., Seoul National University, Seoul, Korea February, 2004 ...... M.S., Electrical Eng. & Computer Sci., Seoul National University, Seoul, Korea June, 2011 ...... M.S., Electrical & Computer Eng., The Ohio State University, Columbus, Ohio, USA

Publications

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “MIMO for Body-Worn Antennas: Approach and Measurements,” IEEE International Symposium on An- tennas and Propagation & USNC/URSI National Radio Science Meeting, San Diego, CA, USA, Jul. 5–11, 2008.

D. Psychoudakis, G.-Y. Lee, C.-C Chen, and J. L. Volakis, “Body-worn diversity antennas for Squad Area Networks (SAN),” XXIXth General Assembly of Union Radio Science International (URSI), Chicago, IL, USA, Aug. 07-16, 2008.

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “Diversity Evaluation for Multiple Body-Worn Antennas,” URSI-National Radio Science Meeting, Boulder, CO, USA, Jan. 5–8, 2009.

vii J. L. Volakis, G.-Y. Lee, D. Psychoudakis, and C.-C Chen, “Multiple Body-Worn Antenna Diversity,” IEEE International Workshop on Antenna Technology, Santa Monica, CA, USA, Mar. 2–4, 2009.

D. Psychoudakis, G.-Y. Lee, C.-C Chen, and J. L. Volakis, “Diversity Evaluation for Multiple Body-Worn Antennas,” 3rd European Conference on Antennas and Propa- gation, Berlin, Germany, Mar. 23–27, 2009.

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “A Novel Evaluation Method for Body-Worn Radio Systems,” IEEE International Symposium on Anten- nas and Propagation & USNC/URSI National Radio Science Meeting, Charleston, SC, USA, Jun. 1–5, 2009.

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “Systematic Design Approach for Diversity Antenna Systems,” 31st Antenna Measurement Techniques Association (AMTA) Symposium, Salt Lake City, UT, USA, Nov. 1–6, 2009.

D. Psychoudakis, G.-Y. Lee, C.-C Chen, and J. L. Volakis, “Military UHF Body- Worn Antennas for Armored Vests,” 4th European Conference on Antennas and Propagation, Barcelona, Spain, Apr. 12–16, 2010.

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “A Computationally Eﬃcient Method for Body-Worn Antenna Diversity Design,” 26th Applied Computa- tional Electromagnetics Society (ACES) conference, Tampere, Finland, Apr. 26–29, 2010.

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “Body-Worn Antenna Diversity Design using FEKO,” 26th Applied Computational Electromagnetics Society (ACES) conference, Tampere, Finland, Apr. 26–29, 2010.

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “Multiple Antenna Design Method for Mobile Platform Diversity Systems,” IEEE International Symposium on Antennas and Propagation & USNC/URSI National Radio Science Meeting, Toronto, Canada, Jul. 11–17, 2010.

G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “Channel Decomposition Method for Designing Body-Worn Antenna Diversity Systems,” IEEE Transactions on Antennas and Propagation, vol. 59, no. 1, pp. 254–262, Jan. 2011.

viii G.-Y. Lee, C.-C Chen, D. Psychoudakis, and J. L. Volakis, “Omnidirectional Vest- Mounted Body-Worn Antenna System for UHF Operation,” IEEE Antennas and Wireless Propagation Letters, vol. 10, pp. 581–583, Jun. 2011.

Fields of Study

Major Field: Electrical and Computer Engineering

Studies in: Electromagnetics and Antenna Design Communications and Signal Processing

ix Table of Contents

Page

Abstract...... ii

Dedication...... iv

Acknowledgments...... v

Vita ...... vii

ListofTables...... xiii

ListofFigures ...... xiv

1. Introduction...... 1

2. Human Body Model for Body-Centric Wireless Communication . .... 9

2.1 Introduction ...... 9 2.2 Review of Electromagnetic Properties of the Human Body . . . . . 10 2.2.1 Numerical Expression of the Dielectric Properties for Human Tissues ...... 10 2.2.2 Dielectric Properties of Human Tissue ...... 11 2.3 Propagation Modes for Body-Centric Wireless Communications . . 15 2.3.1 Penetrating and Reflecting Wave Analysis ...... 17 2.3.2 Creeping Wave Analysis ...... 20 2.3.3 Space Wave Analysis for Off-Body Communication . . . . . 22 2.4 EM Human Torso Model for Body-Centric Wireless Communication 25 2.4.1 Human Torso Model for In-Body Communication ...... 25 2.4.2 Human Torso Model for On-Body Communication . . . . . 28 2.4.3 Human Torso Model for Off-Body Communication . . . . . 30 2.5 Summary ...... 31

x 3. Body-WornAntennaEvaluationMethod ...... 33

3.1 Introduction ...... 33 3.2 Body-WornAntennaElement ...... 34 3.3 Evaluation Method for Diversity System ...... 35 3.3.1 Channel Decomposition Method for Diversity Evaluation . . 39 3.3.2 CDM Validation ...... 42 3.4 Multiple Body-Worn Antenna Diversity System ...... 47 3.4.1 SingleBody-WornAntennaStudy ...... 48 3.4.2 DualBody-WornAntennaStudy ...... 51 3.4.3 Multiple Body-Worn Antenna Diversity ...... 52 3.5 Diversity Module Realization and Measurements ...... 54 3.5.1 Diversity Module Realization ...... 54 3.5.2 IndoorMeasurement ...... 56 3.6 Conclusion ...... 59

4. Systematic Design Method for Body-Worn Antenna Diversity Systems . 60

4.1 Introduction ...... 60 4.2 Systematic Design Method for Body-Worn Diversity Systems . . . 62 4.2.1 Deﬁnition of the Design Parameters ...... 62 4.2.2 Systematic Design Process for Antenna Diversity ...... 64 4.3 Diversity Design Examples for Body-Worn Antenna ...... 66 4.3.1 Communication Scenario I (Ground to Ground Communica- tion)...... 68 4.3.2 Communication Scenario II (All-Purpose Communication) . 71 4.4 Measurement and Validation ...... 74 4.5 Conclusions...... 77

5. Vest-mounted Body-Worn Antenna Diversity System for Wideband UHF Operation ...... 78

5.1 Introduction ...... 78 5.2 Conformal Wideband Body-Worn Antenna Design for UHF Operation 79 5.2.1 Antenna Elements and Diversity Conﬁguration ...... 80 5.2.2 Antenna Performance on Body ...... 83 5.3 Compact Diversity Module Realization ...... 88 5.3.1 Operation of the Diversity Module ...... 88 5.3.2 Diversity Module Fabrication using COTS Components . . 90 5.4 Field Test for Body-Worn Antenna Diversity System ...... 93 5.4.1 Summary of the Reference Antenna ...... 93

xi 5.4.2 TestSetup ...... 94 5.4.3 OutdoorTests ...... 96 5.4.4 IndoorTests ...... 103 5.4.5 FieldTestSummary ...... 105 5.5 Conclusions...... 105

6. FutureWorks ...... 108

6.1 Introduction ...... 108 6.2 Vehicular Antenna Diversity System ...... 109 6.3 Vital Sign Monitoring System using Body-Worn Antennas . . . . . 112 6.4 Body-Worn Antenna System for RF Power Harvest ...... 115

7. Conclusions ...... 121

Bibliography ...... 125

xii List of Tables

Table Page

2.1 Parameters required to ﬁnd human tissue dielectric constant and conductivity[23]...... 12

2.2 Electromagnetic properties of skin (dry), muscle, and fat at selected frequencies (30 MHz, 433 MHz, 915 MHz, 2.45 GHz, and 12 GHz). . . 16

2.3 Calculated SWR, reflection coefficient, and transmitted E-field at each frequency...... 18

4.1 Comparison of selected antenna positions, pattern coverage, and correlation coeﬃcients for n-antenna diversity between the proposed method and full evaluation method for scenario I (Abbreviations for antenna positions; 1st letter- B: back, F: front, S: shoulder, T: Thigh, 2nd letter- C: center, H: Horizontal, L: left, R: right)...... 70

4.2 Comparison of selected antenna positions, pattern coverage, and correlation coeﬃcients for n-antenna diversity between the proposed method and full evaluation method for scenario II...... 73

5.1 Summary of calculated SAR values and limitations...... 88

5.2 Summary of the technical speciﬁcations of the diversity module. . . . 92

5.3 Summary of the test results in outdoor and indoor environments. . . 107

6.1 Electromagnetic properties of deﬂated lung and inﬂated lung at selected frequencies (30 MHz, 433 MHz, 915 MHz, 2.45 GHz, and 12 GHz)...... 113

xiii List of Figures

Figure Page

1.1 Brief illustration of the works in the dissertation: (a) concept of channel decomposition method (CDM), (b) best body-worn antenna diversity for omni-directional pattern, (c) vest-mounted body-worn antenna diversity system for wideband UHF operation ...... 5

2.1 Electromagnetic properties of skin, fat, muscle, and lung (deﬂated); (a) relative permittivity (real value), (b) conductivity, and (c) penetration depthinmm...... 13

2.2 Electromagnetic properties of heart, liver, lung (deﬂated), and lung (inﬂated); (a) relative permittivity (real value), (b) conductivity, and (c)penetrationdepthinmm...... 14

2.3 Inﬁnite human tissue layers (skin - fat - muscle - fat - skin) and radiating sources near the body surface; (a) plane wave source, (b) vertical dipole antenna, and (c) horizontal dipole antenna...... 17

2.4 Penetrating and reﬂecting wave on the boundary of human tissues: (a) at whole observed region, (b) zoom-in view focused on human tissue layers...... 19

2.5 Near field plot (E-field) around the radiating antenna and human body layer: (a) vertical dipole antenna near the body surface, (b) horizontal dipole antenna near the body surface, (c) vertical dipole antenna in the free space, and (d) E-field magnitude along y-axis at z=0.4 λ (center oftheantenna)...... 21

2.6 Plane wave incident obliquely on a plane dielectric boundary: (a) parallel polarized wave, (b) perpendicular polarized wave...... 22

xiv 2.7 Electric far-ﬁeld pattern along the elevation angle for diﬀerent distances betweenantennaandhumanbody...... 23

2.8 The reference human torso model: (a) thorax cut image from visible human project, (b) top view, (c) front view of torso model...... 26

2.9 (a) Top view and (b) front view of the approximated concentric 4-layer model, and (c) comparison of normalized E-ﬁeld magnitude inside the humanbodyfortwomodels...... 27

2.10 Top view of (a) 4-layer model, (b) 3-layer model, (c) homogeneous model, (d) transmission coeﬃcients between two antennas for each model. 29

2.11 (a) Homogeneous human model, (b) 3-layer model (c) azimuth radiation pattern (x-y plane), (d) elevation radiation pattern (x-z plane). . 31

3.1 AMF antenna structure and dimensions for operation at 560 MHz. . . 35

3.2 Performance of the AMF antenna element when placed near the body’s torso: (a) the simulated reﬂection coeﬃcients S11 for 3 distances from the body, (b) measured and simulated radiation patterns in the horizontal (x-y) plane at 550 MHz (antenna is 12 mm from the body’s surface)...... 36

3.3 Communication channel model for the CDM...... 38

3.4 Flowchart of the diversity evaluation process using the CDM...... 40

3.5 Simple channel model for validating CDM...... 43

3.6 (a) Amplitude distribution and (b) polarization angle distribution of the incident wave in the environment depicted in Fig 3.5...... 45

3.7 Comparison of the channel capacity CCDF calculated using the Chan- nel Decomposition Method with direct simulations (no decomposition). 46

3.8 Investigated mounting positions for the AMF antenna: (a) front view, (b) back view, (c) left side view, and (d) right side view...... 49

xv 3.9 Projected upper-hemisphere radiation pattern onto the x-y plane for selected single antenna positions: (a) front torso, (b) left arm, and (c) leftshoulder...... 50

3.10 Channel capacity CCDF curves for diﬀerent antenna pairs...... 51

3.11 (a) Channel capacity per conﬁguration, and (b) list of antenna conﬁg- urationspernumberinginFig. 3.8...... 53

3.12 Schematic diagram of the implemented 4-channel diversity module. . 55

3.13 The 4-antenna diversity indoor measurement set-up...... 56

3.14 OSU-ESL hallway environment and the motion path used for the indoormeasurement...... 57

3.15 Comparison of the normalized received power between single antennas (mounted at diﬀerent location on the body) and the diversity system (measurementsweredoneinthehallway)...... 58

4.1 ConceptoftheCDM...... 61

4.2 Flow chart of the systematic antenna diversity design approach. . . . 64

4.3 (a) Investigated mounting positions for the antenna diversity conﬁgu- rations and antenna elements operating at (b) 550 MHz and (c) 350 MHz,respectively...... 67

4.4 Illustration of the communication scenario I for ground to ground communication...... 68

4.5 Illustration of the communication scenario II for all-purpose communication...... 71

4.6 Outdoor measurement setup with human mannequin and diversity module...... 75

4.7 Normalized received power from the 4-antenna diversity module as compared to each antenna’s individual response...... 75

xvi 4.8 Comparison of the eﬀective antenna gain CCDF curves between simulations and measurements for the best single antenna, 2-antenna, and 4-antennadiversitysystems...... 76

5.1 A typical vest conﬁguration [52] with selected antenna mounting locations and the proposed antenna geometries with dimensions; (a) front view, (b) rear view of the vest, (c) gorget antenna on the front, (d) rear antennas on the left back and the right back, and (e) horizontal antennaonthebackwaist...... 81

5.2 Several antenna candidates for gorget antenna mounted on the front vest; (a) antenna 1, (b) antenna 2, (c) antenna 3, and comparisons of (d)S11and(e)peakgain...... 83

5.3 Several antenna candidates for rear antennas mounted on the back vest; (a) antenna 1, (b) antenna 2, (c) antenna 3, and comparisons of (d)antennaS11and(e)peakgain...... 84

5.4 Peak gain comparison for each antenna in free space and on the body (antenna is 25 mm from the body surface); (a) gorget antenna, (b) rear antenna,(c)waistantenna...... 85

5.5 Azimuth radiation pattern comparisons between simulation and measurement; (a) simulated resuts at 250 MHz, (b) measured resuts at 250 MHz, (c) simulated resuts at 350 MHz, (d) measured resuts at 350 MHz. 86

5.6 (a) Hugo upper torso model and (b) simulation setup for SAR calculation. 87

5.7 (a) Schematic diagram and (b) operation algorithm of the diversity module...... 89

5.8 Fabricated diversity module; (a) Bottom view (RF switch), (b) Top view (micro controller and voltage regulators), (c) packaged diversity module...... 91

5.9 (a) Test setup for the diversity module and (b) test results...... 92

5.10 (a) Measurement and (b) simulation model for monopole whip antenna, and (c) peak gain comparisons between measurement and simulation. 94

5.11 Test setup for BWADS and monopole whip antenna...... 95

xvii 5.12 Test fields in OSU ESL facilities; (a) outdoor test field with several obstacles,(b)indoortestfield...... 96

5.13 Illustration of the outdoor test Scenario I (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rdrowfigures)...... 98

5.14 Illustration of the outdoor test Scenario II (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rdrowfigures)...... 99

5.15 Illustration of the outdoor test Scenario III (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rdrowfigures)...... 101

5.16 Illustration of the outdoor test Scenario IV (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rdrowfigures)...... 102

5.17 Illustration of the indoor test Scenario I (top ﬁgure), and test results forverticaltransmitter(bottomﬁgures)...... 103

5.18 Illustration of the indoor test Scenario II (top ﬁgure), and test results forverticaltransmitter(bottomﬁgures)...... 104

5.19 CCDF comparison of received power including all scenarios for (a) outdoorand(b)indoortest...... 106

6.1 Simulation model for vehicular antenna diversity system and commu- nicationscenarioconsidered...... 110

6.2 Best antenna diversity conﬁguration and antenna gain CCDF curves for single, 2-, 3-, and 4-antenna diversity system...... 111

6.3 Diversity performance comparison between the worst and the best 4- antennadiversitysystem...... 112

6.4 Simulation model of vital sign monitoring system and results; (a) side view, (b) top view of the model, and (c) path gain comparison. . . . . 114

xviii 6.5 (a) Measurement setup and (b) ambient RF power measured by 0 dBi receiver at the backyard of OSU-ESL building (Wi-Fi signal is mea- suredatthehallwayofOSU-ESLbuilding)...... 116

6.6 (a) Front view and (b) rear view of body-worn antennas, (c) antenna element structure, (d) 3-D radiation pattern and (e) peak gain of the arrayantenna...... 117

6.7 Calculated RF power received from array antenna and each single an- tennainhorizontaldirection...... 118

6.8 Schematics of the modiﬁed Greinacher rectiﬁer [61]...... 119

6.9 CCDF curves of the generated DC power from the proposed body-worn rectennasystems...... 120

xix Chapter 1: Introduction

The rapid growth in wearable sensors and wireless communications has enabled the development of a new generation of wireless sensor networks, specifically wireless body area networks (WBANs). WBANs are special sensor networks designed to autonomously connect various sensors and appliances, located inside and outside of a human body. The wireless nature of the network offers numerous new innovative applications to improve health care and quality of life. Specifically, a mobile health care system is one of the substantial applications of the WBANs [1, 2]. Various sensors are attached on the body or even implanted under the skin to measure the heartbeat, the body temperature, etc. Then, the collected information is transferred to an external medical database or a doctor through a personal communication device for remote health monitoring. Since all sensors and appliances are wirelessly connected, the patient experiences a greater physical mobility and is no longer compelled to stay in hospital. The benefit of WBANs is also applicable to communication systems for specialized occupations such as firefighters, police officers, etc. However, there is a challenge in implementing the system to provide reliable wireless links between on- body sensors, personal communication devices, and external communication nodes. In this regard, body-centric wireless communication systems (BWCS) become important in connecting all wireless devices on the body or its proximity.

1 BWCS can be classiﬁed into three communication domains as follows [3]:

1. In-body communication: This refers to communication between implanted sen-

sors and devices on the body surface.

2. On-body communication: The main goal of on-body communication is to pro-

vide a reliable wireless connectivity among wearable sensors, and mobile wireless

devices mounted on the body.

3. Oﬀ-body communication: This link provides connectivity from a personal area

network to other existing wireless communication domains (e.g. base station

for cellular communication).

Each of the above is significantly affected by the electrical properties of the human body. Therefore, device performance must be considered in presence of the human body. Chapter 2 discusses the properties of the human body and its effects on each communication devices using electromagnetic (EM) human body models. My focus throughout the dissertation is, however, on off-body communications of the wearable devices.

Electromagnetic ﬁeld interactions between the human body and antennas have been documented in the literature [4–10]. Based on these studies one may conclude the following:

1. Antenna impedance strongly depends on placement position and distance from

the body. The resonant frequency of the antenna decreases as distance de-

creases, and impedance bandwidth increases as distance decreases.

2. Antenna eﬃciency and gain are signiﬁcantly reduced at distances less than 0.1

λ from the body due to the body’s loss and negative reﬂection from the body.

2 3. Antenna pattern is also deformed by the presence of the body. One can consider

the body as a reﬂector of the antenna. Thus, it causes directional pattern instead

of a usually desired omni-directional pattern. The pattern shape and peak gain

also depend on the distance from the body.

To overcome human body effects on body-worn antenna performance addressed above, some authors have proposed metallic or electromagnetic bandgap (EBG) surfaces to reduce back radiation and increase efficiency [11–13]. However, EBG surfaces are narrowband and difficult to adapt to lower frequencies, and may also cause undesirable radiation patterns. Therefore, they are not suited for broadband UHF wearable communication.

As an alternative, diversity techniques can offer increased performance for body- worn antennas. Among previous researches in this area, Serra et al. [14] considered dual antenna diversity for the 2.45 GHz ISM band. However, they focused on the connectivity between antennas on the same body (on-body communication), and limited their study to two antennas. Other papers [15, 16] considered antenna diversity for off-body communication, but their conclusions were restricted to specific indoor environments. Moreover, all of the above cases refer to narrowband applications. For the first time, in this dissertation, the diversity techniques are adapted for wideband

UHF body-worn communications.

As body-centric communications are associated with continuously changing environments, a generalized antenna diversity evaluation method is desired when designing body-worn antennas. It is desired to include the surrounding environments as well as the eﬀect of the human body for evaluation of the body-worn antennas.

However, critical issues regarding computational resources for EM calculations arise:

3 1. The human body itself consists of various tissues with diﬀerent EM proper-

ties, and substantial computational resources are required for EM simulations

without simplifying approximations.

2. It is very diﬃcult to consider the surrounding environments with accurate nu-

merical methods like method of moments (MoM) and the ﬁnite element method

(FEM). Modeling the environment also requires signiﬁcant computational re-

sources.

3. Both the human body and the mobile environment change continuously over

time. Body movement causes rapid variations in antenna parameters (impedance,

gain, pattern, etc.) aﬀecting performance. The mobile environment can also

vary over time, however its characteristics do not change as abruptly. These

eﬀects should be considered in any evaluation method.

4. A large number of conﬁguration options should be investigated when developing

multiple body-worn antenna diversity. The total computation time required will

be multiplied by the number of diversity conﬁgurations.

With the above in mind, in this dissertation we develop an eﬃcient and versatile evaluation method for designing body-worn antennas with diversity. A channel decomposition and systematic design methods are proposed to accomplish this. The methods are subsequently employed to design a vest-mounted body-worn diversity system.

Illustrations of the research carried out in this dissertation are shown in Fig. 1.1.

Some of the key contributions of the dissertation are given below:

4 Communication channel

Human body channel Propagation channel

h1 Diversity h module 2

Rx Tx h3 SNR Re c SNR Rx SNR Tx (a)

Selected antenna Ant 1 Ant 3 Ant 2 Ant 4 Ant 1 000 gg

---1-1110000

---2-2220000

---3-3330000 Ant1- Front center (FC) Ant2- Back center (BC) ---4-4440000 Ant3- Left shoulder (LS) Ant4- Right shoulder (RS)

Normalized received [dB] power ---5-5550000 Diversity module output

000 33303000 66606000 99909000 111212220000 111515550000 111818880000 222121110000 222424440000 222727770000 3330300000 3333330000 333636660000 Azimuth angle [deg] (b)

Front of vest Back of vest Diversity module

Antennas

(c)

Figure 1.1: Brief illustration of the works in the dissertation: (a) concept of channel decomposition method (CDM), (b) best body-worn antenna diversity for omni- directional pattern, (c) vest-mounted body-worn antenna diversity system for wideband UHF operation .

5 1. Developed a channel decomposition method (CDM): The proposed CDM was

used to evaluate body-worn antenna performance in various environments. The

CDM signiﬁcantly reduced computation time (more than 10 times) without loss

of accuracy for body-worn antenna evaluations.

2. Developed a systematic body-worn antenna diversity design method: The pro-

posed method, incorporated with CDM, automatically ﬁnds the best antenna

mounting positions to achieve acceptable performance. The approach is sub-

sequently applied to design a body-worn antenna diversity system for speciﬁc

communication environments. Importantly, the proposed approach reduced the

number of diversity conﬁgurations (from 495 cases to 12 cases for 4-antenna

diversity from 12 available antenna positions) and computation time.

3. Designed and fabricated a vest-mounted UHF body-worn antenna system with

diversity: Using the above methods, a practical body-worn antenna diversity

system was developed to provide a nearly omni-directional pattern over 225-450

MHz. The light-weight and conformal antennas were hidden in a standard vest

and in a manner not to impede the wearers’ activities. A compact diversity

module (79 × 41 × 28 mm) was also constructed to automatically select the

best channel among 4-antennas in real time. The proposed system was tested

in various environments. It was demonstrated that the range of communication

was increased by 2.2 and 6.7 times as compared to a commercial whip antenna

for outdoor and indoor environments, respectively.

The organization of the dissertation is as follows:

6 Chapter 2 reviews the human body properties from VHF to X-band. Antenna

propagation on the surface of the human body is considered for in-body, on-body, and

oﬀ-body communication. The results determine which antenna type is proper for each

body-centric communication. Based on these propagation studies, a simpliﬁed human

torso model is suggested. It is shown that the simpliﬁed (homogeneous) model is

adequate for designing antenna as relates to oﬀ-body communications. This simpliﬁed

model is then employed for body-worn evaluation in the subsequent chapters.

Chapter 3 proposes and develops the channel decomposition method (CDM) for

body-worn antenna evaluation. The details of the CDM are introduced, and valida-

tions are provided by comparison to full wave simulations. The CDM is subsequently

used to ﬁnd the optimal antenna locations and design a body-worn diversity system

for isotropic pattern coverage. To realize the diversity system, a prototype 4-channel

diversity module was fabricated. Indoor measurement results are also provided to

demonstrate the eﬀectiveness of the proposed body-worn diversity system in rich

multi-path environments.

In chapter 3, the antenna locations to realize diversity are selected via an em- pirical method. Although this selection process provides some insights into eﬃcient antenna location, it requires substantial simulation time to reach an optimal solution.

With the goal of reducing computation time, chapter 4 introduces a systematic design method for body-worn diversity antennas. First, the important parameters are iden- tiﬁed. This is followed by the design method ignoring the mutual coupling between antennas. Several design examples are also given to demonstrate the eﬀectiveness of the approach. As part of the chapter, fundamental assumption of the method is also

7 justiﬁed based on full wave simulations. Measurements are also given to validate the

overall method.

Chapter 5 introduces the vest-mounted UHF body-worn antenna diversity system

(BWADS). This system is intended to provide a reliable communication system for law-enforcement officers with the goal of making the antennas unobtrusive. First, the antenna elements are designed and a compact high speed diversity module is developed to implement the diversity system. The module automatically selects the best antenna among multiple ones in real time. After integrating the antennas with the diversity module, numerous field tests are performed in outdoor and indoor environments. The benefits of the system are highlighted at the end of Chapter 5 with a comparison to a commercial whip antenna.

Chapter 6 suggests future research topics relating to the body-worn antenna system. First, a diversity system is proposed for a vehicular platform using the method proposed in chapter 4. Second, a vital sign monitoring system using body-worn antennas is suggested for monitoring the respiration and vital signs. Third, Body-worn antennas are proposed to harvest ambient RF power. These can generate dc power from ambient RF signals and provide electrical power to mobile devices.

8 Chapter 2: Human Body Model for Body-Centric Wireless Communication

2.1 Introduction

Because body-centric wireless communications always involve the interaction of electromagnetic waves with the human body, it is important to understand the body’s electromagnetic properties when we design body-worn antennas or implanted wireless sensors. As these properties also depend on signal frequency and on tissue types, it becomes necessary to consider diﬀerent human models and properties for each frequency band and application.

In this regard, the relevant properties of the human body from VHF to X-band are studied and dominant propagation modes around the body are discussed. Based on these data, approximate human torso models are built for each propagation mode using multiple concentric cylinders emulating the skin, muscle, organ, etc. The simplest model for each propagation mode is proposed for eﬃcient calculation without loss of accuracy. Furthermore, the proposed human model will be adopted for designing the body-worn antenna system.

9 2.2 Review of Electromagnetic Properties of the Human Body

The electromagnetic properties of human body tissues aﬀect the propagation,

reﬂection, attenuation, and other behaviors of electromagnetic ﬁelds around the body.

These properties depend strongly on the types of tissue and frequency. Temperature

and blood or fluid perfusion also affect these properties but these secondary effects

are normally not considered. We shall only consider relative permittivity (ǫr) and conductivity (σ). Because the body is so weakly magnetic, the relative magnetic permeability (µr) is assumed to be 1, except for special applications such as magnetic resonance imaging (MRI).

2.2.1 Numerical Expression of the Dielectric Properties for Human Tissues

The dielectric properties of body tissues have been studied by several authors

[17–22]. The most comprehensive and referenced study is [23]. Gabriel measured over 25 tissue types in the frequency range from 1 MHz to 20 GHz using an open- ended coaxial probe technique [24]. Most of these measurements were carried out with excised animal tissue from freshly killed sheep at 20◦C and 37◦C. In addition, accessible parts of the human body such as palm, sole and forearm skin were measured in vivo. To describe the frequency dependence of the tissue dielectric properties, a model based on the summation of 4-Cole-Cole expressions is used [23] and the relative dielectric constant is expressed as

4 ∆ǫm σi ǫr (ω)= ǫ∞ + + , (2.1) (1−αm) jωǫ m=1 1+(jωτm) ! 0 X

10 where ǫ∞ is the material’s relative permittivity at terahertz frequency; ǫ0 is the free

space permittivity; σi is the ionic conductivity; ǫm, τm, and αm are material parame-

ters for each dispersion region. Since the dielectric constant is generally complex, it

consists of real and imaginary components:

′ ′′ ǫr (ω)= ǫr (ω) − jǫr (ω) . (2.2)

Also, conductivity and penetration depth (or skin depth) of the tissue can be expressed as

′′ σ (ω)= ǫr (ω) · ǫ0 · ω, (2.3)

− 1 1 2 ′ ′ 2 2 1 µ ǫ0ǫr(ω) σ(ω) δ(ω)= 1+ ′ − 1 . (2.4) ω  2  ωǫ0ǫr(ω) !      The penetration depth is deﬁned as the depth at which the inte nsity of the radiation inside the material falls to 1/e of the original value at the surface.

The parameters required to determine the dielectric properties of selected human

tissues are tabulated in Table. 2.1. Using these parameters along with equation (2.1)-

(2.4), we can ﬁnd the important dielectric properties of human tissues of interest.

2.2.2 Dielectric Properties of Human Tissue

Fig. 2.1 shows the relative permittivity, conductivity, and penetration depth of

selected tissues of the human torso in the frequency range from 10 MHz to 20 GHz.

Fig. 2.2 also shows the same properties of relatively large organs inside the human

torso (heart, liver and lungs).

We can observe several important properties of the human body:

11 Table 2.1: Parameters required to ﬁnd human tissue dielectric constant and conductivity [23].

ε∞ σi ∆ε1 τ1 (ps) α1 ∆ε2 τ2 (ns) α2 ∆ε3 τ3 (µs) α3 ∆ε4 τ4 (ms) α4 Bone - cancellous 2.5 0.07 18 13.263 0.22 300 79.577 0.25 2.0e+4 159.155 0.2 2.0e+7 15.915 0 Bone - cortical 2.5 0.02 10 13.263 0.2 180 79.577 0.2 5.0e+3 159.155 0.2 1.0e+5 15.915 0 Fat 2.5 0.01 3 7.958 0.2 15 15.915 0.1 3.3e+4 159.155 0.05 1.0e+7 7.958 0.01 Heart 4 0.05 50 7.958 0.1 1200 159.155 0.05 4.5e+5 72.343 0.22 2.5e+7 4.547 0 Liver 4 0.02 39 8.842 0.1 6000 530.516 0.2 5.0e+4 22.736 0.2 3.0e+7 15.915 0.05 Lung - deflated 4 0.2 45 7.958 0.1 1000 159.155 0.1 5.0e+5 159.155 0.2 1.0e+7 15.915 0 Lung - inflated 2.5 0.03 18 7.958 0.1 500 63.662 0.1 2.5e+5 159.155 0.2 4.0e+7 7.958 0 Muscle 4 0.2 50 7.234 0.1 7000 353.678 0.1 1.2e+6 318.31 0.1 2.5e+7 2.274 0 Skin - dry 4 0 32 7.234 0 1100 32.481 0.2 0.0e+0 159.155 0.2 0.0e+0 15.915 0.2 Skin - wet 4 0 39 7.958 0.1 280 79.577 0 3.0e+4 1.592 0.16 3.0e+4 1.592 0.2

1. Human body tissues can be classiﬁed into two types: the ﬁrst type has low

water content (i.e. fat, bone, etc.), low dielectric constant, and low loss; the

second type has high water content (i.e. skin, muscle, heart, etc.), high dielectric

constant, and high loss (see Fig. 2.1).

2. Almost all internal organs with high water content tissue have very similar EM

properties. Only an inﬂated lung has lower dielectric constant and conductivity

due to increased air content (see Fig. 2.2).

3. The conductivity (or penetration depth) of all tissues increase (or decrease)

as signal frequency increases. In other words, human body becomes lossy and

incident waves are rapidly attenuated at high frequencies (see Fig. 2.1 and 2.2).

These observations about human tissues led to some insights for body-worn antenna applications. Here are some examples:

12 3 10 Skin-dry Fat ') r Muscle ε 2 Lung-deflated 10 Bone-cortical

1 10 Relative permittivity ( permittivity Relative

0 10 1 2 3 4 10 10 10 10 Frequency [MHz] (a)

2 10

1 10

0 10

-1 10 Conductivity [S/m] Conductivity

-2 10 1 2 3 4 10 10 10 10 Frequency [MHz] (b)

3 10

2 10

1 10 Penetration depth [mm] depth Penetration

0 10 1 2 3 4 10 10 10 10 Frequency [MHz] (c)

Figure 2.1: Electromagnetic properties of skin, fat, muscle, and lung (deﬂated); (a) relative permittivity (real value), (b) conductivity, and (c) penetration depth in mm.

13 3 10 Heart Liver ') r

ε Lung-deflated

2 Lung-inflated 10

1 10 Relative permittivity ( permittivity Relative

0 10 1 2 3 4 10 10 10 10 Frequency [MHz] (a)

2 10

1 10

0 10

-1 10 Conductivity [S/m] Conductivity

-2 10 1 2 3 4 10 10 10 10 Frequency [MHz] (b)

3 10

2 10

1 10 Penetration depth [mm] depth Penetration

0 10 1 2 3 4 10 10 10 10 Frequency [MHz] (c)

Figure 2.2: Electromagnetic properties of heart, liver, lung (deﬂated), and lung (in- ﬂated); (a) relative permittivity (real value), (b) conductivity, and (c) penetration depth in mm.

14 1. The human torso can be modeled by multi-layer structures using skin, fat, mus-

cle, organs, and bone in the order they appear in human body layers. Since the

organs have EM properties similar to muscle, they can be omitted in approxi-

mate human models, with the exception of inﬂated lungs.

2. The diﬀerent EM properties of lungs in diﬀerent respiratory state (exhalation

and inhalation) make it possible to sense the human’s respiratory rate and vital

sign.

3. In-body communication (between an outer antenna and implanted sensor) should

be performed at lower frequency due to loss characteristics of the human body.

Previously, in-body communication has been performed at 13.56 MHz via in-

ductive coupling and at 402-405 MHz, allocated for medical implanted commu-

nication systems.

4. Penetration depth in real human bodies will be much less than the minimum

penetration depth, because the incident waves experience a multiple reﬂections

at skin - fat - muscle interfaces (having relative permittivity 46.1 - 5.6 - 56.9 at

433 MHz).

The EM properties of the skin, muscle and fat (representing high water content and low water content) are also tabulated in Table 2.2 at selected frequencies.

2.3 Propagation Modes for Body-Centric Wireless Commu- nications

Body-centric wireless communications can be divided into three domains [3]:

1. Oﬀ-body communication (human-to-human or human-to-base station).

15 Table 2.2: Electromagnetic properties of skin (dry), muscle, and fat at selected frequencies (30 MHz, 433 MHz, 915 MHz, 2.45 GHz, and 12 GHz). Tissues EM properties f=30 MHz f=433 MHz f=915 MHz f=2.45 GHz f=12 GHz K Relative permittivity ( r’) 152.9 46.08 41.33 38.01 29.33 Skin Conductivity (σ) [S/m] 0.3415 0.7017 0.8715 1.464 10.33 - dry Penetration depth (δ) [mm] 222 54 40 23 3 K Relative permittivity ( r’) 8.11 5.57 5.46 5.28 4.46 Fat Conductivity (σ) [S/m] 0.033 0.042 0.051 0.105 0.728 Penetration depth (δ) [mm] 614 304 242 117 16 K Relative permittivity ( r’) 91.81 56.87 55.0 52.73 40.1 Muscle Conductivity (σ) [S/m] 0.658 0.805 0.948 1.739 13.540 Penetration depth (δ) [mm] 127 52 42 22 3

2. on-body communication (multiple antennas placed on the same body surface).

3. in-body communication (implanted sensor and antenna placed on the body

surface).

Since each communication domain utilizes diﬀerent propagation modes around the body, it is important to characterize each propagation mode. We can categorize the propagation modes for each body-centric wireless communication as follows:

1. Space wave that propagates away from the body (for oﬀ-body communication).

2. Creeping wave that propagates along the body surface (for on-body communi-

cation).

3. Penetrating wave that propagates into the body (for in-body communication).

To analyze each propagation mode, model of inﬁnite planar human tissue layers

(emulating human torso) with three types of radiating sources were considered as

16 shown in Fig. 2.3. Three types of tissue are used. To calculate the penetrating wave

and reﬂected wave, an ideal plane wave source is assumed as shown in Fig. 2.3(a). For

other propagating waves, vertical and horizontal dipole antennas are used as shown in

Fig. 2.3(b)-(c). Each layer is assumed to be inﬁnite in size to avoid other scattering

and diﬀraction eﬀects. The thickness of each tissue layer is t1 = 2.4 mm, t2 = 13.5 mm, and t3 = 168 mm [25]. The total thickness of the body layer is 200 mm. The

EM properties of the tissues are speciﬁed in Table 2.2.

Plane wave Q/2 dipole antenna r ) + = jk0z r E xE0e k

d d zˆ 1 2

t1 Skin xˆ Skin Skin Fat Fat Fat t2

t3 Muscle Muscle Muscle

t2 Fat Fat Fat t1 Skin Skin Skin

(a) (b) (c)

Figure 2.3: Inﬁnite human tissue layers (skin - fat - muscle - fat - skin) and radiating sources near the body surface; (a) plane wave source, (b) vertical dipole antenna, and (c) horizontal dipole antenna.

2.3.1 Penetrating and Reﬂecting Wave Analysis

To study the penetrating wave and reﬂected wave behavior of the human body,

+jk0z the plane wave represented by E~ =xE ˆ 0e (E0 = 1 V/m) is directed along the

-z axis onto an inﬁnite planar tissue layers located from z=0 mm to z=-200 mm as

17 shown in Fig. 2.3(a). The magnitude of the E ﬁeld is calculated using FEKO and

is shown in Fig. 2.4 from z=300 mm to z=-300 mm. One can readily observe large

reﬂections of incoming waves at the boundary (z=0 mm) between air and human

tissue (skin). Standing waves are formed in the air region. The magnitude of the

reﬂection coeﬃcient is calculated using the standing wave ratio (SWR) and expressed

SWR − 1 |Γ| = . (2.5) SWR + 1

Table 2.3 summarizes the calculated SWR, reﬂection coeﬃcient, and transmitted

E-field at each frequency. We note that most of the incoming waves are reflected at the boundary and incident waves cannot be efficiently transmitted into the body for in-body communication. The reflected wave also cancels the off-body radiation

Table 2.3: Calculated SWR, reflection coefficient, and transmitted E-field at each frequency. f=30 MHz f=433 MHz f=915 MHz f=2.45 GHz f=12 GHz SWR 29.29 5.84 2.45 10.89 4.94 Γ 0.934 0.707 0.421 0.832 0.663

Et [V/m] 0.0458 0.016 0.001 0.00005 0

ﬁeld when the radiation source is very close to the body (relative to its wavelength)

because the phase of the reflection coefficient is close to 180◦ (see E-field at z=0+ mm in Fig. 2.4(a)) thereby reducing the efficiency of the body mounted antenna. Overall, only a small amount of the E-field can enter the body and only small ratio of the field can transfer throught the body at higher frequency as described in Table 2.2. From

18 twoeosre ein b omi iwfcsdo ua tissu human on focused o view boundary zoom-in the (b) on region, wave observed reﬂecting whole and at Penetrating 2.4: Figure

IExI [V/m] skin (2.4mm) IExI [V/m] 2.45 GHz 2.45 915 MHz 915 MHz 433 2.45 GHz 915 MHz 433 MHz 30MHz 12GHz 30 MHz 12 fat (13.5 fat mm) GHz muscle(168 mm) z z [mm] 19 (b) (a) z [mm] fat (13.5 fat mm) ua ise:(a) tissues: human f skin (2.4 skin mm) layers. e Planewave ( E 0 =1 V/m) =1 these two observations, we can infer the design tips for in-body communication and

oﬀ-body communication antennas. The antenna for in-body communication must

be mounted directly on the body (without any gap) to reduce the reﬂection at the

boundary between air and skin. For oﬀ-body communication the gap between antenna

and body should be carefully chosen to avoid the severe gain drop caused by reﬂection

cancellation. This will be shown later.

2.3.2 Creeping Wave Analysis

Radiations from a vertical and a horizontal (relative to the body surface) dipole

antennas are compared to study the creeping wave mode at 2.45 GHz as depicted in

Fig. 2.3(b)-(c). Also, the vertical dipole antenna in free space is compared with one

on the body. All the antennas are located at z=0.4 λ and the body layers are placed between z=0 mm and z= -200 mm. Fig. 2.5(a)-(c) compares the near ﬁelds for (1) vertical dipole, (2) horizontal dipole near body, and (3) vertical dipole in free space.

Fields from both vertical antennas propagate well along the body surface, while fields from the horizontal antenna do not. Fig. 2.5(d) plots field amplitude along the body surface as a function of lateral distance from antenna. At y=5 λ, the calculated normalized E-field magnitudes are -40, -52, -42 dBV/m for the case Fig. 2.5(a), (b),

(c), respectively. It shows that the vertical antenna near the body produces stronger

field transmission than that of the horizontal antenna. It is also slightly stronger than the same antenna in free space. These observations can be explained from reflection when waves are obliquely incident as shown in Fig. 2.6 and the reflection coefficients for parallel and perpendicular polarized waves are expressed by (2.6) and (2.7) [26], respectively.

20 1 1 ENorm [dBV/m] 0

0 −10 0

− 20 − 1 − 30 −1 z in wavelength z − 40 in wavelength z

− 2 − 50 − 2 − 5.0 0 1 2 3 4 5 − 5.0 0 1 2 3 4 5 y in wavelength y in wavelength (a) (b)

1 0 ver. dipole on the body -10 hor. dipole on the body ver. dipole in free space 0 -20

-30 [dBV/m]

−1 Norm -40 E

z in wavelength z -50

− 2 -60 0 1 2 3 4 5 − 5.0 0 1 2 3 4 5 y in wavelength y in wavelength (c) (d)

Figure 2.5: Near field plot (E-field) around the radiating antenna and human body layer: (a) vertical dipole antenna near the body surface, (b) horizontal dipole antenna near the body surface, (c) vertical dipole antenna in the free space, and (d) E-field magnitude along y-axis at z=0.4 λ (center of the antenna).

η2cosθt − η1cosθi Γ// = . (2.6) η2cosθt + η1cosθi

η2cosθi − η1cosθt Γ⊥ = . (2.7) η2cosθi + η1cosθt In the case of parallel polarization (Fig. 2.5(a) and 2.6(a)), the reﬂection coeﬃcient

◦ converges to +1 when the incident angle increases to θi ≈ 90 and the incident and reflected waves add. As a result the body boundary helps to transfer EM power along the surface. In the case of perpendicular polarization, the reflection coefficient

21 r r r E// r ˆr ˆr H ⊥ k k⊥ // r r ⊗ Et ˆt E t ˆt r // k// r ⊥ k⊥ r r H // r E⊥ r θ θ t θ θ t r t H // r t H ⊥ r θ θ i i i E// ˆi r ˆi k// i k⊥ r E⊥ i H // r i H ⊥ Medium 1 Medium 2 Medium 1 Medium 2 η η η η 1 2 1 2 (a) (b)

Figure 2.6: Plane wave incident obliquely on a plane dielectric boundary: (a) parallel polarized wave, (b) perpendicular polarized wave.

◦ converges to -1 when the incident angle increases to θi ≈ 90 (Fig. 2.5(b) and 2.6(b)), causing the ﬁeld cancellation along the boundary. Thus, a vertically (relative to the body surface) polarized antenna should be selected for on-body communication.

Higher frequency (i.e. 2.45 GHz) is also preferred for this application because of the physical size limitations on the antenna.

2.3.3 Space Wave Analysis for Oﬀ-Body Communication

Space waves emitted by body-worn antennas radiating away from the body consist of a direct radiating ﬁeld from the antenna and a reﬂected wave from the body surface.

Thus, it is affected by the gap between the antenna and human body (reflecting surface). To investigate the effects of the gap, I considered a horizontal dipole antenna for off-body communication as shown in Fig. 2.3(c) placed at different distances (d2) from the body surface. Fig. 2.7 shows the electric far-field patterns at 433 MHz for

22 various antenna distances from the body surface. The free space antenna is used as a reference.

θ = 0 20zˆ in free space xˆ d =0.05λ 18 2 d =0.1λ 2 16 d2 λ

¢ d =0.2 θ = − ¡ θ = 90 2 90 Skin Fat d =0.3λ 14 2 Muscle d =0.4λ Fat 2 12 Skin d =0.5λ 2 10

Electric Electric farfield magnitude [V] 4

0 -90 -60 -30 0 30 60 90 Elevation angle, N [deg]

Figure 2.7: Electric far-ﬁeld pattern along the elevation angle for diﬀerent distances between antenna and human body.

In this case, one can observe three important phenomena:

1. There are similar radiation patterns for distances up to d2 = 0.2 λ. When

the distance is greater than 0.3 λ, the main beam is broadened or split in the

horizontal direction. This can be explained by an array model consisting of

the original antenna and an image antenna at z = -d2. Since the magnitude of

23 the reﬂection coeﬃcient is less than one and its polarity is negative, the image

antenna has a smaller excitation voltage with reversed polarity.

2. The strongest radiation ﬁeld is observed at d2 = 0.2 λ in the main beam di-

rection. It is also stronger than the reference antenna (in free space) because

the reﬂected wave is added to the direct radiation ﬁeld at this distance and the

phase of the reﬂection coeﬃcient is approximately π (negative polarity). This is

consistent with Fig. 2.4(a) in which the ﬁrst maximum of the total ﬁeld occurs

at approximately 150 mm (≈ 0.22 λ at 433 MHz) as shown in Fig. 2.4(a).

3. The radiation ﬁeld is weaker when the antenna is close to the body, because the

reﬂected wave and the direct radiation ﬁeld cancel.

From the above, one can determine the best antenna location for efficient off- body communication. The antenna should be placed at a certain electrical distance depending on the effective reflection coefficient of the human body. Since the human body can be treated as a metallic surface (large reflection with negative polarity), a distance of λ/4 is usually selected. However, this cannot be applied to wideband antennas and low-frequency antennas because the electrical distance associated with the frequency and the physical distance corresponding to λ/4 is too large to be practical in low frequency applications. In this case, it is necessary to reduce the reflection from the body using wideband electromagnetic band gap (EBG) surfaces to reduce the back radiation [13].

24 2.4 EM Human Torso Model for Body-Centric Wireless Com- munication

Many numerical human models have been developed for computational simulations of body-centric wireless communication. For accurate calculation of body effects, an anatomically realistic human model composed of a set of fine voxels is desired. Hu- man voxel models for the entire body have been developed by Dimbylow [27]. This model was based on magnetic resonance images (MRIs) of an adult male and was segmented into 37 different tissues. Subsequently, higher-resolution human models have been developed by other researchers [28–30]. Anatomical human models require large computational resources to generate accurate solutions to EM problems.

In this light, several approximate human torso models are proposed for in-body, on-body, and oﬀ-body communication. A concentric elliptical cylinder composed of multiple tissue layers is considered, and the simplest model having reasonably small error in EM analysis is selected for each application.

2.4.1 Human Torso Model for In-Body Communication

To establish a human torso model, the communication between on-body antenna and a sensor implanted on the heart is considered. The calculation is performed at

433 MHz because the antenna size is realizable and the penetration depth at this frequency is also suitable for implanted sensor applications.

First, a reference human model is constructed based on the thorax cut image from the visible human project [31] as shown in Fig. 2.8. The reference model consists of skin, fat, muscle, lung, heart, and spinal cord. The outer diameter of the model is 200

25 Muscle

Spinal cord Lung Lung

Heart Fat

Skin

(a) y

x Spinal cord Heart Lung Lung Lung Heart

z Muscle Fat Spinal cord

Skin y (b) (c)

Figure 2.8: The reference human torso model: (a) thorax cut image from visible human project, (b) top view, (c) front view of torso model.

mm (along the short axis) and 320 mm (along the long axis). To evaluate communication performance, the penetrating field is calculated inside the human model when the plane wave is incident from the +x direction. The calculated E-fields from the reference model and the approximate concentric 4-layer model (skin-fat-muscle-lung) are compared in Fig. 2.9. Overall, both results are in a good agreement except in the region between x = 0 mm to x = 48.4 mm. The difference (less than 1 dB) is caused by

EM property diﬀerences between heart and lung (deﬂated). This result shows that

E-ﬁelds inside diﬀerent organs vary slightly, but this error is small. Thus, 4-layer

26 Skin Fat Skin

Muscle Muscle Lung Lung 0 48 4. Fat 84 1. 97 6. 100 mm

(a) (b) 0

-5 y

[dBV/m] -10 x Norm E

-15

Reference model 4-layer model -20 0 10 20 30 40 50 60 70 80 90 100 x [mm]

(c)

Figure 2.9: (a) Top view and (b) front view of the approximated concentric 4-layer model, and (c) comparison of normalized E-ﬁeld magnitude inside the human body for two models.

27 model (skin-fat-muscle-lung) is suﬃcient for simulation of in-body communication antennas.

2.4.2 Human Torso Model for On-Body Communication

The main propagation mode for on-body communication is the creeping waves which are not signiﬁcantly aﬀected by the core medium (organ in this case). It is noted that the creeping wave cannot penetrate into the core medium at higher frequency as shown in Fig 2.5(a). Three human torso models (homogeneous, 3-layer,

4-layer concentric cylinder) are constructed and compared using transmission coeﬃ- cients between two antennas mounted on the same body as shown in Fig. 2.10. The homogeneous model is a phantom whose relative permittivity and conductivity are equivalent to 2/3 times that of muscle. The 4-layer model is the same as one shown in Fig. 2.9 and the 3-layer model is similar to the 4-layer model except that the lung is replaced with muscle. Fig. 2.10(d) shows the transmission coeﬃcients from

2.4 to 2.5 GHz between two antennas, it is noted that the 4 and 3-layer model show excellent agreement but the homogeneous model has about 2 dB higher values. In the 4 and 3-layer models, fat has high contrastive EM properties to skin and muscle.

Thus penetrating waves into the body layer do not propagate signiﬁcantly beyond the fat layer. This results in a loss of transmission power via creeping wave in the 4 and 3-layer models. Therefore, it is necessary to include skin, fat, and muscle layers in an approximate human model for on-body communication, but organ tissue is not important, especially at higher frequency. Finally, the 3-layer model is selected as the simplest accurate human model for on-body communication.

28 Fat Fat Skin

Lung Muscle

Muscle

0.25λ 0.48λ Skin dipole

(a) (b)

Homogeneous body

-32

-34

[dBV/m] -36 21 S

-38

-40 2.4 2.45 2.5 Frequency [GHz]

(d)

Figure 2.10: Top view of (a) 4-layer model, (b) 3-layer model, (c) homogeneous model, (d) transmission coeﬃcients between two antennas for each model.

29 2.4.3 Human Torso Model for Oﬀ-Body Communication

Space waves are the main propagation mode for off-body communication. The space wave radiating in the off-body direction is composed of a direct radiated wave from the source and a reflected wave from the body surface. Thus the reflection coefficient of the body is important in constructing a human model for off-body communication.

To evaluate the human model for oﬀ-body communication, radiation patterns for the homogeneous model and the 3 layer model at 433 MHz are calculated and compared as shown in Fig. 2.11. Both models are the same as the model presented in

Fig. 2.10(b) and (c). The λ/2 dipole antennas are located in the front of the human model at 0.1 λ distance and parallel to the body surface. The azimuth and elevation

radiation patterns are shown in Fig. 2.11(c) and (d). Overall, they are in good

agreement except for back radiation. They also have a small diﬀerence in gain (less

than 0.5 dB) in the main beam direction, but this can be compensated by modifying

the EM properties of the homogeneous body. In conclusion, the homogeneous model

with proper EM properties is suﬃcient to represent the human body model for oﬀ-

body communication.

30 z

x x

y 0.1λ

(a) (b)

φ Gain [dB] θ Gain [dB] Homogeneous model 3-layer model

Figure 2.11: (a) Homogeneous human model, (b) 3-layer model (c) azimuth radiation pattern (x-y plane), (d) elevation radiation pattern (x-z plane).

2.5 Summary

In this chapter, the human body model was investigated for body-centric wireless communication. First, electromagnetic (EM) properties of the human body were re- viewed from VHF to X-band. EM properties of the human body depend on frequency and tissue types. Human body tissues were categorized into high water content tissue (high dielectric constant and loss) and low water content tissue (low dielectric

31 constant and loss) according to their EM properties. Fat, bone and inﬂated lung are grouped with low water content tissue while skin, muscle and other main organs are grouped with high water content tissue.

Second, wave propagation modes around the body were studied for in-body, on- body, and oﬀ-body communication, respectively. Also, proper antenna type and mounting conﬁguration on the body were suggested for each body-centric communication.

Finally, approximate numerical human torso models using concentric elliptical cylinders with multi-tissue layer were suggested for each body-centric communication. A 4-layer (skin-fat-muscle-organ) model, and a 3-layer (skin-fat-muscle), homogeneous model were suggested for in-body, on-body, off-body communication, respectively. All models were simplified for calculation efficiency without loss of accuracy.

In the following, horizontally placed dipole antennas along the human body and the homogenous human model will be used in body-worn antenna design for oﬀ-body communication.

32 Chapter 3: Body-Worn Antenna Evaluation Method

3.1 Introduction

Electromagnetic field interactions between the human body and antennas have been documented in the literature [4–10]. These studies show that the antenna’s impedance depends strongly on its position and its distance from the body and conclude that the body effects cause efficiency losses. These effects are intensified for conformal wearable body-worn antenna systems, making it essential to explore viable solutions to overcome pattern deformation and gain reduction. To this end, some authors have proposed metallic or electromagnetic bandgap (EBG) surfaces to reduce back radiation and increase efficiency [11–13], reporting 63∼80 % efficiency at 2.45

GHz and 2.8 GHz. However, EBG surfaces are narrowband and diﬃcult to adapt to lower frequencies, and may also cause undesirable radiation patterns. Therefore, they are not suited for broadband UHF/VHF wearable communications.

As an alternative, diversity techniques can offer increased data throughput for body-worn antennas. Among previous works in this area, Serra et al. [14] considered dual antenna diversity for the 2.45 GHz ISM band. However, these authors focused on the connectivity between antennas on the same body, and limited their study to two antennas. Other papers [15, 16] considered antenna diversity for off-body communication, but their conclusions were restricted to specific indoor environments.

33 As body-centric communications are associated with continuously changing environ-

ments, a generalized antenna diversity evaluation method is being sought for designing

body-worn antennas.

In this respect, this chapter introduces an eﬃcient and versatile methodology for designing multiple antenna diversity systems for any environment. The multi- antenna system is integrated into a unique module that employs a diversity decision process to maximize channel capacity. The underlying design utilizes a new channel decomposition technique to account for interaction with the nearby environment.

The structure of the chapter is as follows. First, we introduce the antenna element used throughout this chapter. Next, we describe the Channel Decomposition Method

(CDM) for capacity evaluation and demonstrate its validity via simulations. Subse- quently, CDM is used to design a body-worn system for omni-directional coverage.

As such, optimum mounting positions for the body-worn antennas are determined to provide isotropic pattern coverage using the minimum number of antennas. In the last section, we provide the details of implementing a diversity module to retroﬁt existing single channel radios with diversity capability. Measurements are presented and compared with simulations to evaluate the body-worn diversity system in a multipath environment.

3.2 Body-Worn Antenna Element

Before presenting the channel decomposition method, it is important to introduce the body-worn antenna used throughout this chapter. The subject antenna is a scaled version of the Asymmetric Meandered Flare (AMF) body-worn element presented in

[32]. It is modiﬁed to operate at 560 MHz instead of 300 MHz, the center frequency

34 Figure 3.1: AMF antenna structure and dimensions for operation at 560 MHz.

of the original antenna (see Fig. 3.1 for dimensions). Fig. 3.2 depicts the antenna performance for on-body operation. Specifically, Fig. 3.2(a) presents the simulated reflection coefficients at various distances from the body and Fig. 3.2(b) provides a comparison between measurements and calculations for the pattern at 550 MHz.

These results show that the body presence aﬀects three antenna characteristics: gain, input impedance, and radiation pattern. Diversity techniques, presented in the next section, are intended to alleviate antenna performance compromises due to the body presence.

3.3 Evaluation Method for Diversity System

As mentioned in the previous section, the negative effects of the body’s presence can be reduced by employing multiple antennas. An effective way of evaluating which antenna configuration offers the highest advantages, is to study the increase in channel capacity for a given communication channel.

35 0

-5

-10

-15

-20 S11 S11 [dB] -25 2mm gap 12mm gap -30 free standing ant -35 400 450 500 550 600 650 700 Frequency [MHz]

(a) 0 zˆ -5 . yˆ xˆ -10 -15 gain ¤ -20

gain £ -25 -30

-35

Realized Realized Gain [dB] -40

-45 Solid line : Simulation Marked line : Measurement -50 0 30 60 90 120 150 180 210 240 270 300 330 360 Azimuth angle [deg]

(b)

Figure 3.2: Performance of the AMF antenna element when placed near the body’s torso: (a) the simulated reﬂection coeﬃcients S11 for 3 distances from the body, (b) measured and simulated radiation patterns in the horizontal (x-y) plane at 550 MHz (antenna is 12 mm from the body’s surface).

36 For a multi-antenna diversity system, a generalization of the channel capacity

[33, 34] is given by

C SNR H∗H bits/sec = log det 1+ Rx , (3.1) B 2 Norm2 Hz

where B denotes bandwidth, SNRRx refers to the average signal-to-noise ratio (SNR)

at the receiving antenna (see Fig. 3.3), and H is the channel transfer function matrix.

In the case of a single antenna transmitter, like the one considered in this chapter,

the channel transfer function matrix becomes a vector given by

h11 . H =  .  , (3.2) hNR1     th where hi1 represents the ratio of the voltage at the i receiver over that at the transmitter. Since this ratio includes the channel propagation eﬀects, the H matrix is typically normalized by the Frobenius norm to exclude path losses. However, as path losses don’t depend solely on the environment but also on antenna location, it is important not to exclude the antenna characteristics from capacity evaluations.

Therefore, instead of using the Frobenius norm, we considered the average SNR over all diﬀerent antenna conﬁgurations to fairly compare channel capacities. Thus, the subject norm used in (3.1) is of the form:

K N (NR)k 1 2 1 Norm = v |hi1| . (3.3) uK · N   (NR) u k=1 n=1 i=1 k u X X X t   In this, K denotes the number of diﬀerent conﬁgurations, N is the number of

samples per conﬁguration, and (NR)k refers to the number of receive antennas of the kth conﬁguration.

37 Communication channel

Human body channel Propagation channel

h1 Diversity h module 2

Rx Tx h3 SNR Re c SNR Rx SNR Tx

Figure 3.3: Communication channel model for the CDM.

To determine the channel transfer matrix, we introduce a Channel Decomposition

Method (CDM). Speciﬁcally, the communication channel is decomposed into two regions as shown in Fig. 3.3:

1. the human body, including the antennas (denoted as the human body channel

in Fig. 3.3) and

2. the surrounding environment (denoted as the propagation channel in Fig. 3.3).

Below, we introduce and validate the CDM to evaluate the channel transfer function. As such, we can eﬀectively characterize the body-worn antenna diversity system for any environment using channel statistics. As can be understood, the proposed

CDM can be readily applied to diversity systems for platforms other than the human body.

38 3.3.1 Channel Decomposition Method for Diversity Evalua- tion

Conventionally, the entire channel environment is used to evaluate the channel capacity [15, 16]. However, since the human body is a rapidly changing platform as opposed to a building or even a vehicle, this approach is not attractive for general channel evaluation. Instead, the communication channel is divided into the human body and propagation channel components, shown in Fig. 3.3. As expected, the body’s movement causes rapid variations in the antenna parameters (impedance, gain, pattern, etc.) aﬀecting performance. On the other hand, the propagation channel can vary over time, but its characteristics do not change as abruptly. Therefore, one can evaluate the propagation channel statistics separately. To do so, a ray tracing approach for outdoor channel environments can be adopted. However, for body eﬀects, a more accurate method such as the Method of Moments (MoM) or Finite Element

Method (FEM) must be used. Alternatively, measurements can be taken in place of simulated data for validation or for improved accuracy. That is, diﬀerent analysis methods can be used for evaluating the channel components, providing ﬂexibility in analyzing communication channel characteristics.

Fig. 3.4 outlines the proposed channel decomposition method. As inferred above, the human body channel consists of the body-mounted antennas in absence of other environmental obstacles or scatterers. The human body channel contribution is eval-

th uated by collecting the far-ﬁeld pattern (E~i) for the i antenna on the body with all other antennas terminated at matched loads. Concurrently, the propagation channel or environmental contributions can be estimated by means of ﬁeld measurements, large scale simulations or even tabulated statistical models assembled in the form of

39 Human body channel Propagation channel

Channel statistics Simulation / Measurement Polarization angle distribution (ϕ ) Amplitude distribution ( A)

Data building (far field pattern) Monte-Carlo method r Regenerate the A & ϕ E = θˆ()E + φˆ()E rand rand i θ i φ i from channel statistics

Data modification r = ⋅ θˆ ϕ +φˆ ϕ hi Ei ( cos rand sin rand )Arand

Selection diversity H = max{ h , ⋅⋅⋅ ,h } 1 N R

Channel capacity calculation C   HH H  bits / s = log det  I + SNR  2   Rx 2  B   Norm  Hz

Statistical analysis Find distribution of capacity Find outage capacity at 90% reliability

Figure 3.4: Flowchart of the diversity evaluation process using the CDM.

40 polarization angle (ϕ) and amplitude (A) distributions. Once the body and propa-

gation channel are characterized, they can be combined to form the channel matrix

components (hi). Generally, the environment (propagation channel) contributions

modulate the human body pattern data via a Monte-Carlo process and can be ex-

pressed as

hi = E~i · θˆcos ϕrand + φˆsin ϕrand Arand, (3.4) where ϕrand is the polarization angle and Arand is the ﬁeld amplitude random variables

generated by a Monte-Carlo process (the latter draws random values ϕrand and Arand

based on the propagation channel statistics). These random variables reﬂect the

impact of polarization mismatch and incoming attenuation due to the multipath

environment, respectively. Also, as usual,

~ ˆ ˆ Ei = θ (Eθ)i + φ (Eφ)i , (3.5)

where Eθ and Eφ are the θ and φ components of the antenna E-ﬁeld pattern on the

body, respectively. The channel capacity is subsequently calculated using (3.1) and

employed to estimate diversity performance and communication reliability.

In this section, the human body contributions are evaluated via the simulation

package FEKO (MoM) [35]. Speciﬁcally, ﬁeld patterns (500 MHz – 600 MHz at 10

MHz increments) were calculated from φ=0◦ to 360◦ at 2◦ increments in the horizon-

tal plane. The human body used in these simulations was modeled as a homogeneous

body, having a relative permittivity of ǫr=56.7 and conductivity, σ=0.94. It was placed on a ground which was emulated by ǫr=9 and a loss tangent of tan δ=0.01. To accurately apply the CDM, the propagation channel should not aﬀect the body-worn antenna impedance. This typically occurs when nearby objects are more than 0.4λ

41 away from the body at the lowest frequency of operation. This distance suﬃciently re-

duced multi-bouncing eﬀects between the human body and the adjacent objects. We

also note that use of a homogeneous human model (instead of a more detailed layered

model) should not aﬀect accuracy because our interest lies in oﬀ-body performance.

At the employed frequencies (λ ∼60 cm) the outer layer tissue losses dominate, block- ing any interaction eﬀects from the interior human organs as discussed in Chapter

Next, we proceed to validate the CDM by simulating the scattering environment and evaluating two body-worn antenna conﬁgurations.

3.3.2 CDM Validation

To validate the proposed CDM, 2-antenna and 4-antenna diversity conﬁgurations were chosen and placed in a simple channel environment (emulating a multipath channel) as illustrated in Fig. 3.5. Speciﬁcally, the channel environment consists of two lossy brick walls (ǫr=4, σ=2) and three lossless dielectric spheres (ǫr=36)

with vertically polarized transmitters, evenly distributed from 0◦ to 360◦ in 2◦ step over the horizontal plane (in far ﬁeld region). For CDM, the human body (channel 1) and propagation channel (channel 2) were simulated separately and statistics

(namely incident wave polarization and incident ﬁeld amplitude) were obtained from simulations.

To model the effects of channel 2 only, the human body and AMF antenna elements were replaced by a set of λ/2 vertical dipoles. Their far field patterns were calculated at all azimuth angles in the horizontal plane, and polarization/amplitude data were extracted. Subsequently, appropriate distribution functions were fitted to

42 CHANNEL 1 dielectric sphere

lossy brick zˆ wall yˆ × xˆ

infinite ground

(a) side view

infinite ground

Human

CHANNEL 1 yˆ ˆ zˆ . x

(b) top view

Figure 3.5: Simple channel model for validating CDM.

43 the simulated data based on a Chi-Squared test. The polarization angles (ϕrand) and

amplitudes (Arand) were then pseudo-randomly generated using a Monte-Carlo pro-

cess. The generated statistical data, shown in Fig. 3.6, allow us to conclude that the

best ﬁt for the amplitude distribution is a Rice distribution given by

r r2 + A2 Ar p (r)= exp − I , (3.6) σ2 2σ2 0 σ2

where A is the mean, σ is the standard deviation and I0 is the zeroth order modiﬁed

Bessel function of the ﬁrst kind. This result was expected since the Rice distribution

is best at describing sparsely populated scattering environments with dominant line-

of-sight contributions (as the one illustrated in Fig. 3.5). For our speciﬁc calculations

we found that A = 1.245 V/m and σ = 0.541 V/m.

The polarization angle was instead ﬁtted to an exponential distribution given by

1 r p (r)= exp − . (3.7) Γ Γ a a ◦ ◦ Here, the mean polarization angle (Γa) is 3.08 with 0 denoting vertical polarization.

From Fig. 3.6(b), it can be inferred that the given channel does not signiﬁcantly

aﬀect the incident wave’s polarization angle.

In addition to implementing the CDM, we also simulated the human model and

the environment as a single channel for validation. To adequately represent human

activity, the human model was rotated about its axis in 10◦ increments. The collected

far ﬁeld data were then used to evaluate the channel capacity to be compared to the

CDM results.

Fig. 3.7 shows the complementary cumulative distribution function (CCDF)

curves of the channel capacity obtained via the CDM and full geometry simulation.

It is evident that the CCDF curves from the two approaches are in good agreement.

44 0.05 Bar : simulation data Line: fitted Rice distribution 0.04 error χ 2 = 0.0098

0.03 Probability 0.02

0.01

0 0 1 2 3 4 E-field amplitude [V/m]

(a)

Bar : simulation data 0.25 Line: fitted exponential distribution

0.2 error χ 2 = 0.0146

0.15

Probability 0.1

0.05

0 0 10 20 30 40 50 60 70 80 90 Polarization angle [deg]

(b)

Figure 3.6: (a) Amplitude distribution and (b) polarization angle distribution of the incident wave in the environment depicted in Fig 3.5.

45 1.1 1 4-antenna 0.9 2-antenna 0.8 0.7 0.6 0.5 0.4 0.3 0.2 0.1 Solid line : no decomposition Probability (capacity abscissa) Probability> (capacity Dashed line: CDM 0 0 1 2 3 4 5 6 7 8 9 10 Channel capacity (bps/Hz)

Figure 3.7: Comparison of the channel capacity CCDF calculated using the Channel Decomposition Method with direct simulations (no decomposition).

This holds for the 2-antenna and 4-antenna diversity conﬁgurations. Of course, the

CDM was much faster by more than a factor of 10 (5.7 hours vs. 66.7 hours for a

2.21 GHz dual core CPU).

In the next section, we proceed to employ the CDM to evaluate the optimum antenna mounting conﬁguration aimed at maximizing channel capacity in a non- scattering environment. The obtained results can be easily used to evaluate the systems in a more complex environment (by adding the contribution from other statistical models).

46 3.4 Multiple Body-Worn Antenna Diversity System

Having established the CDM validity, we next proceed to evaluate specific body- worn diversity configurations. To begin with, we note that the high dielectric constant and losses due to the human body tissue [23] affect off-body radio communication in three ways: 1) the loss degrades antenna efficiency and gain, 2) the body proximity detunes the operational frequency, and 3) the body presence causes pattern deformation and polarization detuning. In spite of these issues, to ensure communication reliability, reception is needed from all incoming directions, regardless of polarization, channel environment and body posture. Ideally, the radiation pattern of the receiving antenna should fully cover the upper hemisphere for all polarizations and body orientations. Therefore, it is prudent to design around a more general propagation channel instead of a specific complex environment. These observations prompted us to adopt the following assumptions in our subsequent studies:

1. The reference channel is isotropic [36] having a uniform Angle of Arrival (AoA)

distribution over the entire upper hemisphere (0◦ to 360◦ in azimuth, 0◦ to 90◦

in elevation) and a uniform polarization distribution (between 0◦∼ 90◦) for all

incidence angles.

2. The human body and antennas are residing on an inﬁnite ground plane.

3. Three human postures (standing, kneeling, and prone) with a full 360◦ rotation

are used to evaluate capacity.

4. The SNR at the receiving antenna (SNRRx) is ﬁxed at 20 dB for all cases.

47 It is understandable that the channel assumption mentioned in 1) is not realistic.

It is nevertheless quite representative, and therefore useful for evaluating the capacity independent of the propagation environment. Furthermore, adding an alternate environment using statistical data is a straightforward process via the CDM (see previous section). In the subsequent sections optimum antenna diversity will be compared to that of isotropic reception.

3.4.1 Single Body-Worn Antenna Study

Before considering multiple antennas, it is important to determine the best body location for a single antenna. This study was done using 12 mm spacing between the antenna and the body to account for clothing. The antenna used is the AMF described in section 3.2 [32].

The considered mounting positions are illustrated in Fig. 3.8 and example performance evaluations are given in Fig. 3.9. The circular plots shown in Fig. 3.9 are two dimensional projections of the upper hemispherical gain pattern onto the x-y plane.

The center of the circle refers to zenith and the bottom edge of the circular periph- ery refers to the forward looking direction (θ=90◦, φ=0◦). As dark red represents stronger radiation, it is apparent that the best (single) antenna location is on the shoulder. This is due to the reduced obstruction by the human body and the equally good radiation of the horizontal and vertical polarizations. We also note that for this speciﬁc location, half of the antenna is on the shoulder and the other half extends along the arm.

48 (12) (5) (5) (12) (5) (12) (1) (2) (3) (8) (9) (10) (6) (13)

(4) (11)

(7) (14)

(a) (b) (c) (d)

Figure 3.8: Investigated mounting positions for the AMF antenna: (a) front view, (b) back view, (c) left side view, and (d) right side view.

49 zˆ gθ gφ [dB] . yˆ xˆ .zˆ yˆ (a)

φ xˆ

(b)

(c)

Figure 3.9: Projected upper-hemisphere radiation pattern onto the x-y plane for selected single antenna positions: (a) front torso, (b) left arm, and (c) left shoulder.

50 1.1 1 0.9 0.8 0.7 0.6 0.5 0.4 0.3 Both torsos 0.2 Both arms 0.1 Both shoulders Probability (capacity>abscissa) 0 0 1 2 3 4 5 6 7 8 9 10 Channel capacity [bps/Hz]

Figure 3.10: Channel capacity CCDF curves for diﬀerent antenna pairs.

3.4.2 Dual Body-Worn Antenna Study

The conclusions drawn from the single antenna study are further confirmed by channel capacity computations. These calculations were carried out for symmetri- cally located antennas. Specifically, in Fig. 3.10 we compare the CCDF for various antenna pairs: front/back torso, left/right shoulders, and left/right arms. As noted before, three postures (standing, kneeling, and prone) were considered in evaluating the performance. Clearly, in agreement with Fig. 3.9, the shoulder positions offered the best channel capacity (6.0 bps/Hz). By contrast, the antenna pair on the torso delivered only 4.6 bps/Hz capacity. Therefore, the shoulder antenna positions were

51 considered as the primary body-worn antenna locations in selecting the best diversity conﬁguration.

The upper hemisphere radiation patterns in Fig. 3.9 provide further insight on choosing additional antenna locations (to achieve better coverage). For instance, we observe that additional antennas should be considered to improve horizontal polarization reception (φ-polarization) and thus enhance channel capacity. Next, we proceed to add more antennas and evaluate their impact on capacity.

3.4.3 Multiple Body-Worn Antenna Diversity

Overall, four diversity conﬁgurations were investigated using 2, 4, 6, and 8 antennas. The channel capacities of all these conﬁgurations are depicted in Fig. 3.11.

Not surprisingly, the best configuration (solid circle) from each set includes antennas at both shoulders (position 5, 12 in Fig. 3.8). With the best single antenna (left shoulder positioned) used as benchmark, it is clear that all diversity configurations show channel capacity improvements of 1.4, 2.2, 2.4, and 2.7 bps/Hz for the 2, 4, 6, and 8 antenna diversity configuration, respectively. Each of these provides 4.3, 6.8,

7.4, and 8.3 dB of diversity gain. However, diminishing returns are observed as the number of antennas exceeds 4. Specifically, only 7 % increase in channel capacity was observed between 4 and 8 antennas (from 6.8 bps/Hz to 7.3 bps/Hz), concluding that the 4-antenna configuration (front/back torso and both shoulders) is the most reasonable and cost effective diversity setup.

In the next section, we describe a practical realization of a diversity module and apply the above ﬁndings to produce a wearable antenna prototype.

52 8

Channel capacity Channel at 90% reliability [bps/Hz] 2-antennas 4-antennas 6-antennas 8-ants. 0 2-2 2-4 2-5 4-1 4-2 4-3 4-4 4-5 4-6 6-1 6-2 8-2 2-1 2-3 4-7 6-3 6-4 8-1

Single Configuration

(a)

Config- uration Antenna positions Single 12 2-1 2, 9 2-2 3.10 2-3 5, 12 2-4 6, 13 2-5 7, 14 4-1 1, 3, 8, 10 4-2 2, 5, 9, 12 4-3 2, 6, 9, 13 4-4 3, 4, 10, 11 4-5 4, 5, 11, 12 4-6 4, 6, 11, 13 4-7 5, 7, 12, 14 6-1 1, 2, 3, 8, 9, 10 6-2 1, 3, 6, 8, 10, 13 6-3 3, 4, 6, 10, 11, 13 6-4 2, 5, 7, 9, 12 , 14 8-1 1, 2, 3, 6, 8, 9, 10, 13 8-2 1, 3, 5, 7, 8, 10, 12, 14

(b)

Figure 3.11: (a) Channel capacity per conﬁguration, and (b) list of antenna conﬁgu- rations per numbering in Fig. 3.8. 53 3.5 Diversity Module Realization and Measurements

To validate the ﬁndings of our previous section, a 4-channel diversity module was designed and fabricated using the selection diversity scheme operating at 100 MHz

– 1000 MHz. Using the fabricated module, the performance of the proposed body- worn antennas was measured and evaluated in an indoor setting (the Ohio State

University ElectroScience Laboratory hallway). In place of an actual human body, a full-size plastic mannequin was used, ﬁlled with body-tissue emulating liquid (water:

52 %, sugar: 47 %, salt: 1 %).

3.5.1 Diversity Module Realization

To implement a receiving diversity module, there are three choices for combining multiple signals: Maximum Ratio Combining (MRC), Equal Gain Combining (EGC), and selection diversity. In general, MRC and EGC achieve better communication performance than selection diversity, but the complexities of realizing a coherent phase circuit (having a precise and stable phase tracking) in a rapidly changing multipath fading environment are paramount. In contrast, the proposed selection diversity scheme easily achieves stable operation even in a fast multipath environment and is also simple to implement [37, 38]. Moreover, selection diversity gave better performance than other schemes in some communication environments (in which high correlated noise exists) [39]. Therefore, selection diversity was chosen for diversity module realization.

A schematic diagram of the proposed 4-channel diversity module is depicted in

Fig. 3.12. A portion of the RF signal from each antenna is routed through a directional coupler to an RF power detector. Each RF power detector produces an

54 Coupler RF_out 1

RF_in 1 RF power RSSI_1 RF_cpl 1 detector

RF_out 2 RF_in 2 RSSI_2 RF switch RF_out RF_cpl 2

RF_in 3 RF_out 3 Switch RSSI_3 control RF_cpl 3

RF_in 4 RF_out 4 RSSI_4 RF_cpl 4 Analog_in

Digital data Data Acquisition Unit Control signal Digital_out LabVIEW

Figure 3.12: Schematic diagram of the implemented 4-channel diversity module.

analog Received Signal Strength Indication (RSSI) voltage proportional to the input

RF power. This RSSI voltage is then digitized and recorded via a data acquisition

unit (DAQ) connected to a LabVIEW [40] equipped computer. LabVIEW is then

used to process the four RSSI signals and control the 4-to-1 RF switch. The RF

switch is controlled via the DAQ to route the strongest signal selected by LabVIEW.

To reduce RSSI noise, a digital smoothing ﬁlter is implemented in LabVIEW and the

signals are weighted for calibrating the power detectors. The fabricated module can

measure the RF signal and control the RF switch at a maximum rate of 106 times per second. It is, thus, suitable for real time operation. Using this module we carried out the measurements described in the next section.

55 3.5.2 Indoor Measurement

Indoor measurements were carried out to demonstrate the eﬀectiveness of the proposed diversity system in a rich multi-path fading environment. The body assembly with selected 4 antennas (12 mm thickness of styrofoam is used to emulate a gap between clothes and body) was mounted on a cart and tested in a hallway of the ElectroScience Laboratory building as shown in Fig. 3.13. The motion path

Figure 3.13: The 4-antenna diversity indoor measurement set-up.

56 33.6 m 1.2 m 2.4 m 180 °°° 180 °°°

6 m 6 m 6 m 6 m 6 m Tx 1.5 m 180 °°° 180 °°°

Figure 3.14: OSU-ESL hallway environment and the motion path used for the indoor measurement.

for data collection is illustrated in Fig. 3.14. As seen, a combination of translations and rotations were performed to deliver realistic scenarios. Speciﬁcally, the assembly

(diversity module + body-worn antenna system) was initially moved away from the transmitting antenna by 6 m for 60 seconds and then rotated 180◦ clockwise for 60 more seconds. This process was repeated four times until the assembly reached the center of the hallway. Concurrently, the individual channel data and the diversity module output were measured in real-time. The indoor measurement results are shown in Fig. 3.15. In this ﬁgure, the normalized received power from each antenna channel is plotted together with the module output. It is evident that the diversity module successfully operates to track the strongest signal throughout the entire path.

In contrast, the signals from the individual antennas suﬀer from severe drop-outs.

57 Selected antenna Ant 1 Ant 4 Ant 2 Ant 4 Ant 1 Ant4 Ant 3 Ant 2 Ant 3 Ant1 Ant1- front torso 0 Ant2- back torso Ant3- L. shoulder -10 Ant4- R. shoulder Diversity module output

-20

-30

-40

-50

Normalizedreceived power [dB] 0 60 120 180 240 300 360 420 480 540 Time [sec]

move rotate move rotate move rotate move rotate move 6 m 180 o 6 m 180 o 6 m 180 o 6 m 180 o 6 m

Figure 3.15: Comparison of the normalized received power between single antennas (mounted at diﬀerent location on the body) and the diversity system (measurements were done in the hallway).

58 Overall, the diversity module provided up to 40 dB improvement over any single

antenna conﬁguration.

3.6 Conclusion

A Channel Decomposition Method (CDM) was introduced for generalized on- platform antenna diversity evaluations and used to design a multi-antenna body- worn diversity system. The CDM separates the entire communication channel into the platform (human body) and the propagation sub-channel. It evaluates the full wave response of the platform (by means of measurements or simulations) and employs statistical data to provide the environment eﬀects. Both polarization and amplitude characteristics were considered and results showed a 10-fold computational speed increase. CDM was then used to design a body-worn antenna diversity system and found that the 4-antenna conﬁguration provided a good compromise in terms of capacity vs. number of antennas.

The designed 4-antenna diversity system (using selection diversity) was also fabricated incorporating the optimized antenna mounting conﬁguration. The indoor measurement showed the signiﬁcant improvement in communication performance.

Speciﬁcally, the proposed diversity system provided increased reliability (upto 40 dB improvement), while each single antenna suﬀered from deep nulls by the multi-path fading indoor environment.

59 Chapter 4: Systematic Design Method for Body-Worn Antenna Diversity Systems

4.1 Introduction

Antenna diversity is one of the most widely used and simplest methods for mitigating shadowing and fading eﬀects associated with rich multipath mobile radio environments. Antenna diversity becomes even more attractive for wearable applications due to human body losses and shadowing [4, 6, 15, 16]. When designing a body-worn antenna diversity system, one is likely to face the following challenges: 1) large computational burden for analyzing the antenna, body, and surrounding environment together; 2) complex optimization process that searches for the minimum number of antennas to achieve maximum performance.

Diversity performance is usually assessed by channel capacity or diversity gain, while antenna correlation data are used for diversity design [34, 36, 41]. Channel capacity provides a practical measure of communication performance (maximum data throughput), and diversity gain shows system enhancement via multiple antennas compared to a single antenna. Correlation between antenna elements presents a design guide for antenna location and orientation. To my knowledge, there is no systematic design process to select antenna mounting locations and the number of antennas needed for optimal performance in the presence of diﬀerent platform environments

60 (for example, the human body for body-worn antenna). The need to provide reliable communication viz. maximum angle and polarization coverage complicates antenna choice even further.

In this chapter, I propose a systematic design method for a body-worn antenna diversity system. The method incorporates the Channel Decomposition Method (CDM) presented in the previous chapter to reduce computation time by dividing the design domain into the human body and propagation channels (see Fig. 4.1). More significantly, the proposed design process automatically finds the antenna mounting positions for acceptable performance using the minimum number of antennas. This approach is then employed to design a body-worn antenna diversity system for specific communication environments.

The structure of this chapter is as follows. First, the systematic design method for body-worn antenna systems is introduced. Important parameters of the method are deﬁned and the fundamental assumptions of the method are described and justi-

ﬁed. Next, several body-worn diversity system design examples are presented, with the results verifying the eﬀectiveness of the method and justifying the fundamental assumptions. To validate the method measurement results are provided in a simple

True environment Human Propagation channel +

Figure 4.1: Concept of the CDM.

61 communication environment.

4.2 Systematic Design Method for Body-Worn Diversity Sys- tems

It is difficult to find optimum diversity configurations for a body-worn diversity system that accounts for both the human body and the surrounding environment (vehicle, diversity, etc.). More specifically, if there are 12 possible antenna locations for a

4-antenna diversity system, we have to consider 495 (= 12C4) diversity conﬁgurations to ﬁnd the optimal diversity system. Although we applied the CDM presented in the previous chapter, the optimization process requires a great deal of computation time and data processing. Thus, a simple and systematic design method is needed to

find an optimal diversity configuration from the multitude of possibilities. The goal is to achieve a specified antenna gain over the desired physical communication sector

(pattern coverage) using the minimum number of antennas.

4.2.1 Deﬁnition of the Design Parameters

Three parameters are used to deﬁne the optimal diversity system: 1) required antenna gain (Gmin), 2) pattern coverage (C) to gauge diversity performance, and 3)

antenna correlation coeﬃcient (ρ) to gauge design eﬃciency.

Typically, the required antenna gain is determined by the communication link

budget, and pattern coverage is set to 90 % for voice communication or 99 % for data

communication. The pattern coverage of the ith single antenna is deﬁned by

N {Gi(θ,φ) > Gmin} Ci = , (4.1) Nt

where Nt refers to the total number of pattern samples (or the total area of the

communication sector to be covered), and N represents the number of pattern samples

62 exceeding the minimum antenna gain (or the area of communication sector satisfying

the given goal). For multiple antennas, the pattern coverage becomes

N {max (Gi(θ,φ),Gj(θ,φ), ··· ) > Gmin} Cij··· = . (4.2) Nt In this expression, selection diversity is applied for the diversity module by the

max(·) operation in the numerator. The pattern coverage represents the performance

of the diversity system.

On the other hand, the antenna correlation coeﬃcient (ρ) represents the similarity

in voltage ﬂuctuations received by the two antennas. This quantity is related to the

antenna radiation patterns, their relative positions, and incoming waves’ polariza-

tions. The correlation coeﬃcient is deﬁned [41, 42] as:

∗ 2 E ViVj ρij = ∗ ∗ , (4.3) E [V iVi ] E V jVj

th where Vi is the induced voltage at the i antenna and E[·] indicates the expected value. It is also noted that the voltage Vi and Vj are phasor quantities representing amplitude and phase, thus their values are affected by not only the antenna patterns but also by the relative positions of the antennas. By using the field patterns of the antennas, this equation can be modified to 2 ~ ~ ~ ∗ ~ Ei(θ,φ) · h(θ,φ) · Ej (θ,φ) · h(θ,φ) θ,φ ρ = h i , (4.4) ij P 2 2 ~ ~ ~ ~ Ei(θ,φ) · h(θ,φ) · Ej(θ,φ) · h(θ,φ) θ,φ θ,φ P P where ~h(θ,φ) is the incoming waves’ polarization vector which can be expressed as

~h(θ,φ)= θˆcosψ + φˆsinψ. (4.5)

In the diversity system, the correlation coeﬃcient is used as a metric for diversity

design eﬃciency because it represents the similarity of the antenna patterns. For

63 example, highly correlated antennas have similar patterns, and they are not eﬃcient for the diversity system.

4.2.2 Systematic Design Process for Antenna Diversity

Fig. 4.2 describes the proposed design procedure. The design goals established at the outset are antenna gain (Gmin), pattern coverage (Cgoal) and allowable correlation coefficient (ρmax). Typically, if the correlation coefficient is less than 0.3 the diversity system consisting of antennas with low correlation is considered an efficient system

[42].

Establishment of the design goal ¦ - Required antenna gain (Gmin), pattern coverage(Cgoal), and allowed correlation ( max) CDM calculation of antenna pattern on platform Propagation channel characterization - Calculate vertical and horizontal component of gain th pattern for i antenna - Polarization angle (¥) / amplitude (A) distribution θ φ θ φ of the incident field in a given propagation channel gθ ,i ( , ), g φ ,i ( , )

Data modification using propagation channel statistics

- Regenerate the polarization angle ( ¥rand) and amplitude (Arand) via Monte-Carlo process θ φ = ()θ φ 2 ϕ + θ φ 2 ϕ ⋅ 2 Gi ( , ) gθ ,i ( , )cos rand gφ,i ( , )sin rand Arand

Correlation for pair of antennas Pattern coverage for n-antenna diversity 2 * - Define all n-antenna diversity configurations (start with n=2) 2 E[]VV ρ = ρ = i j - increase the number of antenna (n) in every repeated step ij s,ij E[][]VV * E V V * N{max()G (θ,φ),G (θ,φ),G (θ,φ)L > G } i i j j = i j k min where, V is induced voltage at ith antenna and Cijk... i Nt E[•] is expected value operator n = n +1 Finding configuration satisfying the goal > ρ ρ ρ L < ρ Cijk... Cgoal and ij , ik , jk , max No Yes Optimum diversity configuration with minimum number of antennas

Figure 4.2: Flow chart of the systematic antenna diversity design approach.

64 After initialization, CDM is employed to evaluate a single antenna in a speciﬁc

communication environment. This involves calculating the radiation patterns (gain)

for all antenna locations using a single antenna and adjusting the data to include the

th propagation channel. From this we can obtain the eﬀective gain (Gi) for the i single antenna in the given communication environment deﬁned by

2 cos sin 2 Gi(θ,φ)= gθ,i(θ,φ) ψrand + gφ,i(θ,φ) ψrand · Arand. (4.6) q q The details of this procedure were explained in the previous chapter.

Next, the pattern coverage (C) and correlation (ρ) for all 2-antenna diversity

conﬁgurations are calculated. For our case, selection diversity was used. However,

other diversity schemes can be applied by altering equation (4.2) for pattern coverage.

The decision making process involves a comparison of the calculated pattern coverage

no conﬁguration satisﬁes the goals, the antenna number is increased and the steps

are repeated until the goal is satisﬁed. Eventually, we can ﬁnd the optimum diversity

conﬁguration with the minimum number of antennas and the best pattern coverage

satisfying the given goal.

The proposed design procedure is simple and straightforward. Furthermore it

provides the best diversity conﬁguration with a minimum number of antennas. To

do so, the method assumes that mutual coupling between antennas is negligible.

This is a valid assumption in well-designed diversity conﬁgurations (e.g. having low

correlations between antennas). This will be demonstrated in the following section

with design examples.

65 4.3 Diversity Design Examples for Body-Worn Antenna

In this section, several design examples are shown for validation of the proposed

design method. Fig. 4.3 depicts the 12 available antenna positions for body-worn

antenna diversity systems and antenna elements operating at 350 and 550 MHz,

respectively. To validate the basic assumption in the proposed method, speciﬁcally

that mutual coupling between selected antennas in the well-designed diversity system

is negligible, two types of simulation are performed and compared.

1. Proposed method: The radiation pattern of each antenna at all locations is

calculated without any antennas at the other locations as suggested in the

proposed method. Thus, the pattern data does not include any mutual coupling

eﬀects with other antennas in the given diversity conﬁgurations, but only 12

simulations are required to perform the proposed design method.

2. Full evaluation method without any assumption: The radiation pattern of

each antenna at each location is calculated with other antennas terminated

by matched load (50 Ω) in the given diversity conﬁgurations. Therefore, all

mutual coupling eﬀects are included in the pattern data for the given diversity

conﬁgurations. In this case, we need to simulate 132 (= 12C2 ×2) conﬁgurations

for 2-antenna diversity, 660 (= 12C3 × 3) conﬁgurations for 3-antenna diversity,

and 1980 (= 12C4 × 4) conﬁgurations for 4-antenna diversity, respectively.

By comparing the above two cases, we can show that the assumption made in the proposed method is valid.

The proposed design method is next applied to two communication scenarios: the

ﬁrst is ground to ground communication in a rural area; the second is all purpose

66 SR SL

SL SR

FR FC FL BL BC BR

FH BH

TL TR

(a) 186 mm

23 mm

(b) 259 mm

32 mm

(c)

Figure 4.3: (a) Investigated mounting positions for the antenna diversity conﬁgura- tions and antenna elements operating at (b) 550 MHz and (c) 350 MHz, respectively.

67 communication in an urban area. The details of each communication scenario are

described in the following sub-sections.

4.3.1 Communication Scenario I (Ground to Ground Com- munication)

Fig. 4.4 illustrates the ground to ground communication scenario in a rural area.

The details of the communication scenario and design goals are described as follows:

1. Required antenna gain (Gmin), pattern coverage (C), and allowable correlation

(ρmax) are set to -8.5 dBi, 90 %, and 0.3, respectively for the antenna operating

at 550 MHz (antenna gain is set to -14 dBi for the antenna operating at 350

MHz).

2. Incoming waves are uniformly distributed from θ = 80◦ to θ = 90◦ and all

horizontal directions.

3. All incident ﬁelds are assumed to be vertically polarized.

FEKO was used to model the human body for these simulations. The human body was modeled as a lossy homogeneous body of relative permittivity ǫr = 56.7

Vertical polarization

o 80 < § < 90

Figure 4.4: Illustration of the communication scenario I for ground to ground communication.

68 and conductivity, σ = 0.94. An inﬁnite ground plane (ǫr = 9 and tanδ = 0.01) is used in all simulation cases. Antennas are located at each location shown in Fig. 4.3 with 12 mm separation from the body.

After applying the proposed design method and full evaluation method, the best diversity configurations and their pattern coverage and correlation coefficients are summarized and compared in Table 4.1(a) for 550 MHz operation. Note that the best antenna locations for each n-diversity system are same in both methods. Fur- thermore, the pattern coverage and correlation coefficients given by the two methods are within 0.2 % error for the pattern coverage and 0.0001 difference for the correlation coefficient. Finally, a 4-antenna diversity system (with antennas located on BC,

FL, TL and TR) satisfies all design goals and is the optimum. Table 4.1(b) summarizes the evaluation results for 350 MHz. The best diversity configurations found by both evaluation methods are also same for all n-antenna diversity systems, and other values are very close to each other. In all cases, the correlation coefficients are very small (the maximum correlation is only 0.153), thus all selected configurations are very efficient.

Based on the above results, we can conclude that the assumption made in the proposed method, that mutual coupling between antennas selected as the best diversity conﬁguration is negligible, is a valid assumption. From this, we can conclude that the proposed systematic design method for body-worn antenna diversity systems is very accurate. Also, the method reduces the number of conﬁgurations requiring EM simulations compared to the basic evaluation in the design process. For example,

1980 EM simulations would be required to evaluate all 4-antenna diversity conﬁgura- tions. This number is reduced to only 12 EM simulations by the proposed procedure.

69 Table 4.1: Comparison of selected antenna positions, pattern coverage, and correlation coeﬃcients for n-antenna diversity between the proposed method and full evaluation method for scenario I (Abbreviations for antenna positions; 1st letter- B: back, F: front, S: shoulder, T: Thigh, 2nd letter- C: center, H: Horizontal, L: left, R: right). (a) At 550 MHz,

Proposed Method Full evaluation method (without mutual coupling effects) (with mutual coupling)

Number (n) Pattern Pattern Best antenna Correlation Best antenna Correlation of antennas coverage coverage positions coefficient positions coefficient for diversity (%) (%) 2 TL, TR 78.6 0.0001 (TL, TR) TL, TR 78.4 0.00002 (TL, TR) 0.153 (FL, TL) 0.153 (FL, TL) 3 FL, TL, TR 84.9 0.001 (FL, TR) FL, TL, TR 84.9 0.001 (FL, TR) 0.0001 (TL, TR) 0.00002 (TL, TR) 0.0001 (BC, FL) 0.0001 (BC, FL) 0.004 (BC, TL) 0.004 (BC, TL)

BC, FL, 0.004 (BC, TR) BC, FL, 0.004 (BC, TR) 4 91.0 91.1 TL, TR 0.153 (FL, TL) TL, TR 0.153 (FL, TL) 0.001 (FL, TR) 0.001 (FL, TR) 0.0001 (TL, TR) 0.00002 (TL, TR)

(b) At 350 MHz.

Proposed Method Full evaluation method (without mutual coupling effects) (with mutual coupling)

Number (n) Pattern Pattern Best antenna Correlation Best antenna Correlation of antennas coverage coverage positions coefficient positions coefficient for diversity (%) (%) 2 BC, FC 84.1 0.022 (BC, FC) BC, FC 84.4 0.015 (BC, FC) 0.014 (BC, FL) 0.009 (BC, FL) 3 BC, FL, FR 89.6 0.013 (BC, FR) BC, FL, FR 89.6 0.008 (BC, FR) 0.039 (FL, FR) 0.039 (FL, FR) 0.128 (BL, BR) 0.127 (BL, BR) 0.004 (BL, FL) 0.005 (BL, FL)

BL, BR, 0.004 (BL, FR) BL, BR, 0.006 (BL, FR) 4 91.6 91.7 FL, FR 0.004 (BR, FL) FL, FR 0.006 (BR, FL) 0.004 (BR, FR) 0.005 (BR, FR) 0.039 (FL, FR) 0.040 (FL, FR)

70 Random linear

polarization

Figure 4.5: Illustration of the communication scenario II for all-purpose communication.

On the other hand, the best locations vary with frequency because the sum of the

line-of-sight signal and reﬂected signal from the ground plane is varies with frequency

and antenna height. Thus, we need to ﬁnd separate optimum diversity conﬁgurations

for diﬀerent operation frequencies, even when all other conditions are the same.

4.3.2 Communication Scenario II (All-Purpose Communica- tion)

Another communication scenario is described in Fig. 4.5 entailing all-purpose communication in an urban area. To apply the proposed design method, we establish the following goals and assumptions:

1. Required antenna gain (Gmin), pattern coverage (C), and allowable correlation

(ρmax) are set to -14 dBi, 90 %, and 0.3, respectively for the antenna operating

at 550 MHz (antenna gain is set to -16.5 dBi for the antenna operating at 350

MHz).

71 2. Incoming waves are uniformly distributed in all upper-hemisphere directions

(0◦ <θ< 90◦, 0◦ <φ< 360◦).

3. incident ﬁelds have random linear polarization and are of the same amplitude.

In this scenario, a 4-antenna diversity conﬁguration mounted on BH, SL, SR, and

TR is found to be the best antenna diversity satisfying the design goal. It is observed that a horizontally directed antenna element (BH) and semi-horizontal elements (SL,

SR) are included in the optimal diversity configuration, because the incoming waves are linearly polarized in random directions. All selected antennas in scenario I were vertically aligned elements because incoming waves were all vertically polarized. The best antenna locations for each n-antenna diversity are always the same in both methods, and pattern coverage and correlation coefficient are very similar as summarized in Table 4.2. It is also noted that correlation coefficients at 350 MHz operation are larger overall than those at 550 MHz operation. This result is expected since the electrical distances between the antennas are less in the 350 MHz operation increasing mutual coupling between the antennas. However, the pattern coverage and correlation coefficients are still in good agreement (maximum errors are only 0.2 % in pattern coverage and 0.017 in correlation coefficients) between the proposed method and full evaluation method for 350 MHz operation.

Evidently, the optimal diversity configurations are different for the two communication environments and operation frequencies we studied. Consequently, it is important to carefully consider propagation channel effects and operation frequencies when designing for optimal diversity. More importantly, two design examples confirm the usefulness of the proposed method in terms of accuracy and efficiency.

72 Table 4.2: Comparison of selected antenna positions, pattern coverage, and correlation coeﬃcients for n-antenna diversity between the proposed method and full evaluation method for scenario II. (a) At 550 MHz,

Proposed Method Full evaluation method (without mutual coupling effects) (with mutual coupling)

Number (n) Pattern Pattern Best antenna Correlation Best antenna Correlation of antennas coverage coverage positions coefficient positions coefficient for diversity (%) (%) 2 SL, SR 85.9 0.015 (SL, SR) SL, SR 85.9 0.016 (SL, SR) 0.015 (SL, SR) 0.016 (SL, SR) 3 SL, SR, TR 88.5 0.005 (SL, TR) SL, SR, TR 88.5 0.005 (SL, TR) 0.030 (SR, TR) 0.032 (SR, TR) 0.010 (BH, SL) 0.010 (BH, SL) 0.015 (BH, SR) 0.015 (BH, SR)

BH, SL, 0.007 (BH, TR) BH, SL, 0.007 (BH, TR) 4 90.1 90.1 SR, TR 0.015 (SL, SR) SR, TR 0.016 (SL, SR) 0.005 (SL, TR) 0.005 (SL, TR) 0.030 (SR, TR) 0.032 (SR, TR)

(b) At 350 MHz.

Proposed Method Full evaluation method (without mutual coupling effects) (with mutual coupling)

Number (n) Pattern Pattern Best antenna Correlation Best antenna Correlation of antennas coverage coverage positions coefficient positions coefficient for diversity (%) (%) 2 SL, SR 86.2 0.061 (SL, SR) SL, SR 86.2 0.056 (SL, SR) 0.097 (BL, SL) 0.080 (BL, SL) 3 BL, SL, TR 89.8 0.014 (BL, TR) BL, SL, TR 89.6 0.016 (BL, TR) 0.031 (SL, TR) 0.031 (SL, TR) 0.097 (BL, SL) 0.080 (BL, SL) 0.007 (BL, SR) 0.008 (BL, SR)

BL, SL, 0.014 (BL, TR) BL, SL, 0.016 (BL, TR) 4 91.6 91.5 SR, TR 0.061 (SL, SR) SR, TR 0.056 (SL, SR) 0.031 (SL, TR) 0.031 (SL, TR) 0.088 (SR, TR) 0.088 (SR, TR)

73 4.4 Measurement and Validation

To validate the proposed design method, outdoor range measurement is performed

at the Ohio State University ElectroScience Laboratory (OSU-ESL) facility and com-

pared with the simulation result. The proposed method is applied to the ground to

ground communication scenario from the previous section but modiﬁed to accom-

modate measurement constraints, e.g. incident waves are vertically polarized and

uniformly distributed in the horizontal plane and minimum antenna gain is set to -5

dBi. A 4-antenna diversity system with antennas located on the FC, BC, SL, and SR

is found to satisfy the given communication scenario.

For measurement, a human body mannequin ﬁlled with sugar water is used as a phantom and 4 antennas (with 12 mm thickness of styrofoam to maintain a gap between antenna and body) are attached at the optimum locations. Additionally, the diversity module, DAQ, and laptop are mounted on the rotating platform at the foot of the mannequin as shown in Fig. 4.6. The setup is similar to an antenna gain measurement setup with a vertically polarized standard horn used as a feed. Received

RF signal power from the antennas is measured both directly at their terminals and at the diversity module’s output using a network analyzer. The measurements were repeated 5 times to get individual receiving powers from 4 antennas and a combined power from diversity module. Fig. 4.7 shows the normalized received power from the diversity module and each antenna individually. As expected, each individual antenna pattern exhibits signal fade as much as 30 dB. In contrast, the module’s output provides the maximum possible signal among the 4 antennas, and experiences only a 3 dB signal drop. Note that the output of the diversity module is measured in real time. Fig. 4.8 compares the CCDF curves of eﬀective antenna gain for the

74 Antenna 4 Antenna 3

Antenna 2 Antenna 1 Tx

Diversity module

DAQ

Rotator LabVIEW

Figure 4.6: Outdoor measurement setup with human mannequin and diversity module.

Selected antenna Ant 1 Ant 3 Ant 2 Ant 4 Ant 1 000 gg

---1-1110000

---2-2220000

---3-3330000 Ant1- Front center (FC) Ant2- Back center (BC) ---4-4440000 Ant3- Left shoulder (LS) Ant4- Right shoulder (RS)

Normalized received power [dB] ---5-5550000 Diversity module output

000 33303000 66606000 99909000 111212220000 111515550000 111818880000 222121110000 222424440000 222727770000 3330300000 3333330000 333636660000 Azimuth angle [deg]

Figure 4.7: Normalized received power from the 4-antenna diversity module as compared to each antenna’s individual response.

75 1 0.9 0.8 0.7 2-ant 0.6 Diversity 0.5 4-ant Single Diversity 0.4 antenna 0.3

0.2 Solid line: simulation 0.1 Marked line: measurement Probability (GainProbability abscissa)> 0 -25 -20 -15 -10 -5 0 Antenna Gain [dBi]

Figure 4.8: Comparison of the eﬀective antenna gain CCDF curves between simulations and measurements for the best single antenna, 2-antenna, and 4-antenna diversity systems.

best single antenna and 2- and 4-antenna diversity found by the proposed method.

As observed, both measurement and simulation results are in reasonable agreement for all configurations except the single antenna configuration. The primary difference in the single antenna is caused by the inaccuracy of the pattern measurement in the null directions; this effect is negligible in the diversity design. We also note from measurement results that the 2-antenna and 4-antenna diversity configurations had diversity gains of 14.4 and 17 dB at the 90 % pattern coverage respectively, relative to the single antenna. Consequently, the measurement result also validated the proposed method for designing the body-worn antenna diversity system.

76 4.5 Conclusions

A systematic design method for body-worn antenna diversity systems was introduced. Together with the channel decomposition method, the proposed method reduced computation time and provided design ﬂexibility for diﬀerent channel environments. The fundamental assumption of the proposed method is that mutual coupling between antennas selected in a well-designed diversity system is negligible.

This assumption simplified the EM calculations and made the proposed method com- paratively simple. To validate the method, body-worn antenna diversity systems were studied for two different communication environments: ground to ground communication in a rural area and all-purpose communication in an urban area. Its results were compared with a full evaluation method including all mutual effects between antennas. We observed that the optimum diversity configurations obtained from both methods are always the same and pattern coverage and correlation coefficients of related antennas are very close. Therefore, the assumption for the proposed method is acceptable and the accuracy of the method is proved. It is also noted that the optimum diversity configuration is highly sensitive to the propagation channel environment and operation frequencies.

A 4-channel diversity system was fabricated and tested in an outdoor environment to validate the design method and simulations. Good agreement between simulations and measurements was obtained for the proposed body worn diversity system. The designed diversity system also achieved a signiﬁcant diversity gain improvement of 17 dB.

77 Chapter 5: Vest-mounted Body-Worn Antenna Diversity System for Wideband UHF Operation

5.1 Introduction

Body-worn antennas are of recent interest for a variety of commercial, medical and law-enforcement related applications. However, body-worn antennas suﬀer from lower gain than traditional monopole whip antennas. In addition, body-worn antennas must be carefully designed to deliver omni-directional patterns. The design should, of course, consider a multitude of typical human postures. Needless to mention, satisfying all these requirements is challenging, particularly due to the limited vest surface and human body eﬀects on antenna performance.

Body-worn antennas for broadband and narrowband UHF/VHF applications have been extensively studied in the literature [43–46]. The challenge of achieving omni- directional patterns using on-body antennas is evident in these papers. Eﬀorts in [47] and [48] to achieve an omni-directional pattern required an additional garment [47] or a large power combining module [48], clearly inconvenient to the wearer.

To improve body-worn antenna performance, diversity techniques were introduced in [48]. Hall et al. [14, 49, 50] performed extensive work in this area. However, their research focused more on on-body communication and narrowband applications.

78 Others [15, 16, 51] considered antenna diversity for oﬀ-body communication but their

antennas were primarily developed for narrowband applications.

In this chapter, a vest-mounted UHF body-worn antenna diversity system (BWADS)

for law-enforcement agents is proposed. The system consists of a set of light-weight,

conformal wideband UHF (225-450 MHz) antennas embedded in a standard bullet-

proof vest with mounting conﬁgurations providing the best possible performance and

a compact diversity module (79×41×28 mm) combining the signal from each antenna in a smart way. Each of the antennas is carefully designed to ﬁt in the space provided by the vest. The proposed BWADS signiﬁcantly improves communication throughput over an existing monopole whip antenna. When operated in conjunction with a diversity module, the proposed system can provide uninterrupted communication during wearers’ maneuvers.

The structure of this chapter is as follows. In section 2, antenna elements designed for the proposed system are introduced and their performances on body are provided.

In the third section, a compact and high speed diversity module designed to achieve real-time diversity operation is introduced. The details of the structure and algorithm are described. After integrating the antennas and diversity module, a variety of

ﬁeld tests are performed in realistic indoor and outdoor environments in section 4.

Speciﬁcally, the impacts of the proposed system are highlighted in comparison to an existing antenna system.

5.2 Conformal Wideband Body-Worn Antenna Design for UHF Operation

For all applications, the antennas are required to cover a wide bandwidth (225-

450 MHz), and an omni-directional radiation pattern is also desirable for reliable

79 communication irrespective of human postures and activities. Additionally, antenna performance should not be aﬀected by other equipment carried by the user. Satisfying all these requirements is a very challenging problem because of the limited space available and human body eﬀects on performance.

This section discusses a novel, very light weight antenna system operating from

225 to 450 MHz developed speciﬁcally for a standard bulletproof vest worn by law- enforcement agents. This antenna system consists of 4 antennas. Each antenna is carefully designed to be mounted inside a bulletproof vest and to provide complete pattern and polarization coverage above the horizon regardless of body posture. The technical speciﬁcations and performance of these antennas are summarized here.

5.2.1 Antenna Elements and Diversity Conﬁguration

Fig. 5.1 illustrates the novel antenna system speciﬁcally designed for installation on a standard bulletproof vest[52] consisting of 4 hard plates (one each on the front and back and one on each side), soft padding inserted into the vest (not shown in

Fig. 5.1), and the vest. Since the hard plates may or may not be used depending on the nature of the current operation, they should be avoided as antenna mounting locations.

Instead, we selected the soft padding for antenna mounting since it is always used with the vest and is easy to integrate with the antennas. The three antenna mounting locations (for the vertical elements) were carefully selected to provide an omni-directional pattern for selection diversity. Thus each antenna has a main beam at roughly 0◦, 130◦, and 230◦ in the horizontal plane. A horizontally polarized antenna is added on the back side of the vest to improve polarization diversity.

80 Gorget antenna

Rear antennas

Waist antenna

Hard plates (a) (b)

Feed point

432 450

Unit: mm 457

406 75 50

Figure 5.1: A typical vest conﬁguration [52] with selected antenna mounting locations and the proposed antenna geometries with dimensions; (a) front view, (b) rear view of the vest, (c) gorget antenna on the front, (d) rear antennas on the left back and the right back, and (e) horizontal antenna on the back waist.

The proposed antenna geometries are shown in Fig. 5.1(c)-(e). All antennas were

printed on a very thin dielectric material (ǫr = 2.2, tanδ = 0.0004, thickness=0.1 mm) via standard commercial PCB process. The antenna is quite flexible, light weight, and fits inside the vest without affecting the wearer’s mobility.

The details of antenna design and performance are described as follows.

81 All antenna elements in Fig. 5.1(c)-(e) consist of asymmetric thick dipole arms

for wideband operation. By using asymmetric dipole arms, an additional higher

order resonant mode can be excited, spectrally adjacent to the fundamental (0.48 λ) resonant mode [53].

Fig. 5.2 and 5.3 show examples of several possible antenna geometries investigated under basic shape and size constraints imposed by available space in the standard bulletproof vest. All gorget antenna structures have asymmetric dipole arms and have two adjacent resonance modes giving the antenna wide bandwidth as shown in

Fig. 5.2. However, the first antenna can be affected by hard plates (Fig. 5.2(a)) and the two resonance modes of the second antenna (Fig. 5.2(b)) are not sufficiently close for a smooth transition between the two modes. The final design in Fig. 5.2(c) has a small gap between two arms, making the antenna’s operation wideband and smoothly connecting the two resonance modes. Also, the antenna structure is not affected by the hard plates. Therefore, the third antenna in Fig. 5.2(c) is selected as the antenna for the bulletproof vest.

Fig. 5.3 shows several novel design candidates for rear antennas with vertical polarization. The step-shaped slot design feature produces two resonant modes that help achieve the desired bandwidth. As one can see, the slot arrangement of the ﬁrst design in Fig. 5.3(a) produces only a single resonant mode (green dashed line). The more asymmetry (larger t) makes the second resonant mode close to the fundamental mode and they form the smooth transition between two modes in the second antenna shown in Fig. 5.3(b). Since the physical length of the antenna is limited by the bulletproof vest, further miniaturization was achieved by adding a slot at the bottom arm to increase the current path (thus increasing the equivalent inductance of the

82 (a) (b) (c) S11 [dB] S11

Peak gain [dBi] Antenna 1 Antenna 2 Antenna 3

Frequency [MHz] Frequency [MHz]

(d) (e)

Figure 5.2: Several antenna candidates for gorget antenna mounted on the front vest; (a) antenna 1, (b) antenna 2, (c) antenna 3, and comparisons of (d) S11 and (e) peak gain.

antenna) and improve the performance in the low frequency band. Fig. 5.3(c) shows the ﬁnal design of the rear antenna. The length of the waist antenna is suﬃcient for horizontal polarization, thus we adopted the design without this additional slot.

5.2.2 Antenna Performance on Body

Fig. 5.4 shows the peak gain of each antenna in free space and on a homogeneous mannequin (ǫr = 56.7, σ = 0.94) as calculated by the FEKO simulation package.

All antennas are located in each position with 25 mm spacing to consider the vest

83 120 mm 120 mm

t=20 mm

(a) (b) (c)

S11 [dB] S11 Antenna 1

Peak gain [dBi] Antenna 2 Antenna 3

Frequency [MHz] Frequency [MHz]

(d) (e)

Figure 5.3: Several antenna candidates for rear antennas mounted on the back vest; (a) antenna 1, (b) antenna 2, (c) antenna 3, and comparisons of (d) antenna S11 and (e) peak gain.

thickness without vest model in simulation. All antennas in free space achieve 0 dBi minimum gain from 240 to 440 MHz, while the body-worn antennas have -10 dBi minimum gain in the operating frequency band (225-450 MHz). It is noted that the antenna eﬃciency/gain drop is more signiﬁcant at lower frequencies, because the spacing between the antenna and body is electrically small at lower frequency.

84 5

0 [dBi] θ -5

-10 Peakgain, G -15 200 250 300 350 400 450 500 Frequency [MHz] (a) (b)

0 [dBi] φ -5

-10 Peakgain, G -15 200 250 300 350 400 450 500 Frequency [MHz] (c)

Figure 5.4: Peak gain comparison for each antenna in free space and on the body (antenna is 25 mm from the body surface); (a) gorget antenna, (b) rear antenna, (c) waist antenna.

The azimuth radiation pattern of each antenna over an inﬁnite ground plane was also calculated and compared with measured data collected at the Ohio State Univer- sity ElectroScience Laboratory (OSU-ESL) outdoor test facility For the simulation, an inﬁnite ground plane (ǫr = 9, tanδ = 0.01) was added to emulate a realistic ground environment. For measurement, the antennas were inserted into the vest and measured with the vest on a mannequin. Fig. 5.5 compares the simulated and measured radiation patterns at 250 MHz and 350 MHz. The black line with circle markers indicates the diversity output based on selection diversity. Each main beam of the three vertical antennas (one front antenna and two rear antennas) was directed to 0◦,

130◦, and 230◦, respectively. Based on these patterns, it is clear that diversity output

85 0 0 [dBi] -10 [dBi] -10 θ θ

-20 -20

Gorget ant. -30 -30 Left back ant. Gorget ant. Right back ant. Right back ant. Left back ant. -40 -40 diversity output Realizedgain, G 0 90 180 270 360 Realizedgain, G 0 90 180 270 360 Azimuth angle, φ [deg] Azimuth angle, φ [deg] (a) (b) 0 0 [dBi]

-10 [dBi]

θ -10 θ

-20 -20

-30 -30

-40 Realizedgain, G 0 90 180 270 360 Realizedgain, G -40 φ 0 90 180 270 360 Azimuth angle, [deg] Azimuth angle, φ [deg] (c) (d)

Figure 5.5: Azimuth radiation pattern comparisons between simulation and measurement; (a) simulated resuts at 250 MHz, (b) measured resuts at 250 MHz, (c) simulated resuts at 350 MHz, (d) measured resuts at 350 MHz.

overcomes the shadowing eﬀect of the human body and achieves improved azimuth

coverage with only 4 dB gain variation at both frequencies as compared to the single

body-worn antenna. These results also show good agreement between simulation and

measurement results.

Since RF energy absorbed by body has potential for harm, it is also important to test the speciﬁc absorption rate (SAR) for body-worn antenna. SAR is a measure of rate how much energy is absorbed by the body when exposed to RF energy and is restricted by FCC [54, 55] and ICNIRP [56] for U.S. and EU, respectively. To calculate the SAR on the head and upper torso by the gorget antenna, Hugo upper

86 torso model provided by FEKO is used. This model is based on the anatomical data set of the Visible Human Project [31] and has only 4 tissues (muscle, lung, eye, and brain) with 8 mm resolution as shown in Fig. 5.6(a). When the gorget antenna transmit a 10 W (40 dBm) of RF power at 450 MHz in Fig. 5.6(b), SAR values are calculated for each tissue and are summarized in Table 5.1. Peak SAR value for 1g cube of muscle close to the limit but it still satisﬁes the guideline for USA and EU both. Therefore, one can conclude that transmitting power from the gorget antenna is allowed upto 40 dBm.

(a) (b)

Figure 5.6: (a) Hugo upper torso model and (b) simulation setup for SAR calculation.

87 Table 5.1: Summary of calculated SAR values and limitations. SAR value Whole USA limit EU limit Muscle Eyes Brain Lungs (mW/kg) body [55] [56] Average 54.7 58.4 57.2 10.524.7 80 80 Peak value for N.A. 1311.2 57.2 35.5139.5 1600 N.A. 1g cube Peak value for N.A. 896.2 57.2 31.3111.8 N.A. 2000 10g cube

5.3 Compact Diversity Module Realization

To achieve real-time autonomous diversity operation of the 4-antenna system, a

compact high-speed 4-channel diversity module was designed, fabricated, and tested

using commercial oﬀ-the-shelf (COTS) components. The purpose of the diversity

module is to transfer the optimal signal from multiple antennas to the radio system

in real time. This section discusses the operational principle, design, and test results

of the diversity module.

5.3.1 Operation of the Diversity Module

Fig. 5.7(a) shows the block diagram of the diversity module and Fig. 5.7(b)

presents the best channel selection algorithm for optimum SNR. The diversity module consists of power supply modules, a micro-controller, and a 4-channel high-speed

RF switch. For demonstration purposes, an RF power detector is used in place of a radio (it was not available to us) to provide signal quality information (feedback) such as a received signal strength indication (RSSI) voltage to the micro-controller.

88 Diversity Module

+ 5V regulator Multiple 9V antennas battery9V battery - 5V regulator

××× 2 Switch control RF switch signal (ZSWA-4-30 RF power DR ) detector Micro (MAX2016 ) controller or (Arduino pro Radio sys. RSSI mini ) analog signal

RF output

(a)

Initialization of the module and variables Searching process

Control the switch to select the input CH. n (initially n = 1)

Read and store the RSSI voltage from the n = n +1 radio system for CH. n

n = 4 ? No

Selection process Yes

Find the best input channel by comparing the stored RSSI voltages

Control the switch to select the best channel

Maintain the selection for 1 second

(b)

Figure 5.7: (a) Schematic diagram and (b) operation algorithm of the diversity module.

89 The operating algorithm of the diversity module consists of searching process and

selection process. During the searching mode, the RF switch steps through all the

signals from the antennas and routes the signals one by one to the radio system (in

this case the RF power detector) which generates feedback signals (RSSI voltages).

The generated feedback signals are sent to the micro controller to evaluate the signal

qualities of each input signal. This process takes approximately 54 µsec to evaluate the received signals from all antennas. After that, the micro controller decides which input channel is the best based on the highest RSSI voltage. It then sets the switch to select the best signal in the selection mode. Since the best signal branch changes with the communication environment and users’ activity, the processes (searching and selection process) are repeated every second to update the best signal branch.

If users are in a fast fading environment such as an urban area, the period of the process should be shortened to update the optimal channel more frequently.

5.3.2 Diversity Module Fabrication using COTS Components

Fig. 5.8 shows the fabricated module prototype using COTS components. The battery is not included in the module housing but two 9 V batteries are required for operation. The module has 4 RF input ports to connect with 4 antennas and 1 RF output port for transferring the selected signal to the radio system. As mentioned earlier, the module essentially consists of the 4-channel RF switch and micro controller. A ZSWA-4-30DR RF switch from Mini-Circuit is used for the 4-channel RF switch because it provides 45 nsec of switching time with only 0.9 dB of insertion loss, which is essential for fast searching operation. Also, an Arduino pro-mini micro controller is used for control of the module. It includes 6 analog inputs and 12 digital

90 79 (L) μμμ-controller -5V regulator

41 (W)

Control signal Switch chip +5V regulator T = 28, (unit: mm )

(a) (b) (c)

Figure 5.8: Fabricated diversity module; (a) Bottom view (RF switch), (b) Top view (micro controller and voltage regulators), (c) packaged diversity module.

I/O pins for data interface, has 32 Kbytes of ﬂash memory, and a rapid clock speed

(16 MHz) for programming and calculation [57]. Since both components operate with

5 V of DC power, ±5 V regulators are used for operation with 9 V batteries.

To check the operation of the diversity module, a simple laboratory test is performed. Fig. 5.9 shows the test setup and results for investigating the switching operation of the module. Two identical CW signals (300 MHz) are provided to channel 1 and 2 inputs of the diversity module, and the micro controller is programmed to select ﬁrst channel 1, then channel 2, and then neither. The RF switch was found to have a very short transition time (60 nsec) which is similar to the manufacturer’s speciﬁcations (45 nsec). Based on this transition time and the micro-controller’s processing time to read and store analog signals, the entire searching process takes 54

µs for 4 input channels. Other important technical speciﬁcations of the module are tabulated in Table 5.2. The module is very compact with physical dimensions only

79 mm (Length) × 41 mm (Width) × 28 mm (Thickness), thus it does not hinder the

users’ activities. It can operate for 60 hrs continuously with an 1800 mAh battery.

91 Test setup Overall view Switch on ( off CH. 1 on)

CH.2 RF power detector Signal generator CH.1 off 60 nsec RF switchSwitched signal

CH. 2 CH. 1 Original signal (300MHz CW) RF switch Micro-controller

Original switched signal signal 60 nsec 50 nsec Multi-channel Digital Oscilloscope

Switching process Ch. 1 Ch. 2 both off Transition ( CH. 1 CH. 2) Switch off ( CH. 2 off)

(a) (b)

Figure 5.9: (a) Test setup for the diversity module and (b) test results.

Table 5.2: Summary of the technical speciﬁcations of the diversity module.

Physical specifications

¨ ¨ ¨ ¨ Size (L W T, mm) 79 ¨41 28 Weight (gram) ~ 300 Operation temperature (ºC) -30 ~ 85

6 ¨ MTBF at 85 ºC for RF switch (hrs) 30 ¨10 Electrical specifications Supply voltage (V) 9 Supply current (mA) 30 Expected battery lifetime with 1800mAh capacity (hrs) 60 Insertion loss (dB) ~ 0.9 Switching time (turn-on or turn-off, nsec) 45

92 5.4 Field Test for Body-Worn Antenna Diversity System

To demonstrate the eﬀectiveness of the proposed diversity system in realistic environments with variable users activities, several outdoor/indoor measurements were performed in the Ohio State University ElectroScience Laboratory (OSU-ESL) facility. For comparison, a monopole whip antenna, which is currently used in radio systems at the same frequency band, is measured in the same environments. Mea- surements are taken in a total of 4 outdoor and 2 indoor environments with diﬀerent user movements.

First, we brieﬂy summarize the performance of the monopole whip antenna used as a reference antenna. Then several ﬁeld test results are discussed with details of the test setup and scenarios. Finally, the superiority of the proposed body-worn antenna diversity system (BWADS) is highlighted.

5.4.1 Summary of the Reference Antenna

Fig. 5.10 shows the geometry of the monopole whip antenna provided by the sponsor. The antenna is 300 mm (length) × 38 mm (diameter) in size and it is connected to a metallic box (178 mm × 76 mm × 38 mm) emulating the radio set. The measurement and simulation results for peak gain are compared and shown in Fig. 5.10 (c). Overall, the measurement and simulation results were in good agreement. It is also observed that the antenna had -3.7 to 0 dBi peak gain (from the measurement result) within the frequency band of interest (225-450 MHz). We expect that the lower gain is due to the lossy material surrounding the antenna designed for wideband operation.

93 5 Monopole whip antenna 0

-5

-10

-15 Realizedpeak gain [dBi] -20 Measurement Simulation -25 Metallic box 100 150 200 250 300 350 400 450 500 Frequency [MHz] (a) (b) (c)

Figure 5.10: (a) Measurement and (b) simulation model for monopole whip antenna, and (c) peak gain comparisons between measurement and simulation.

5.4.2 Test Setup

Fig. 5.11 illustrates the test setups for the BWADS and reference antenna. A tester wore the bulletproof vest with embedded BWADS and backpack carrying the laptop and portable network analyzer (Agilent FieldFox N9912A). The monopole whip antenna was also carried by the tester. The received power from BWADS was recorded in real time by the laptop via the portable network analyzer. In this setup, we can record the RF power data automatically. Since this setup only allows for recording one channel at a time, the measurements of BWADS and monopole were conducted separately by walking the same path and posing the same postures. The recorded data were compared after all measurements were completed. Antenna performance may be slightly aﬀected by wearing a backpack. In all tests, a log-periodic dipole antenna (LPDA) was used to transmit a single RF tone at selected frequencies (225,

94 Antennas Monopole antenna Diversity module Coupler RF out

Data RSSI logging RF power detector -12 dB PNA Laptop

Figure 5.11: Test setup for BWADS and monopole whip antenna.

300, 375 and 450 MHz). The BWADS and monopole were used as the receiving antennas.

Fig. 5.12 shows the outdoor and indoor test ﬁelds at the OSU-ESL facilities. There were several obstructions in the outdoor environment such as a car, trees, foam post, bricks and metal obstruction, which emulated the law-enforcement agents’ mission area. The ESL building corridors consisting of brick wall, metallic door, and other metallic furniture (width: 1.7 m, height: 2.3 m) were selected as the indoor test ﬁeld.

For the outdoor measurements, both horizontal and vertical polarized LPDA was used as a transmitter, while only vertical polarized antenna was used for the indoor measurements. Both the indoor and outdoor tests were performed at 4 frequencies

(225, 300, 375, and 450 MHz). In these tests, each measurement was repeated for

BWADS and monopole whip antenna, respectively, because both systems could not be measured at the same time. However, the tester (an OSU ESL student volunteer) carefully followed the prescribed scenarios with time stamps for reliable comparisons

95 Bricks Trees 2.3 m Car Foam post ESL corridor

Metal obstacle 1.7 m

(a) (b)

Figure 5.12: Test fields in OSU ESL facilities; (a) outdoor test field with several obstacles, (b) indoor test field.

between two systems. The details of the measurement scenario and results will be

described in the following two sections.

5.4.3 Outdoor Tests

In total, 4 outdoor tests were performed to evaluate the BWADS and the monopole antenna in diﬀerent environments with emulated wearers’ activities at the OSU-ESL.

Two scenarios were selected to investigate the eﬀects of wearers’ activities on antenna systems and other scenarios were chosen to study the eﬀects of obstacles on the systems. Each scenario and test results are discussed in below.

Outdoor Test I

The performances of antenna systems were investigated subject to various human

activities without obstruction in this scenario. Walking, kneeling, and lying were

repeated during the measurements as shown in the scenario illustration. The left

96 column in Fig. 5.13 represents the normalized received power recorded in real time at

300 MHz for vertical and horizontal transmitter, respectively. Overall, the BWADS showed better performance in both cases. Speciﬁcally, the monopole suﬀered from deep null for 30-40 sec due to body obstruction and cross polarization when the horizontal transmitter was used. On the other hand, the BWADS showed stable operation without deep nulls during the test for both transmitters. The right column indicates the complementary cumulative distribution function (CCDF) curves of the signal strength obtained at 4 frequencies. 5.5 dB of diversity gain was observed for the vertical transmitter and 8.0 dB of diversity gain was observed for the horizontal transmitter, when 90 % communication reliability was desired.

Outdoor Test II

This test was designed to investigate the effects of body obstruction on antenna performance. The tester walked along the specified path and sat at each designated holding position facing different directions as illustrated in Fig. 5.14. Since the monopole whip antenna was obstructed by the tester at the second sitting position

(35-45 sec), it suﬀered from severe nulls for both polarized transmitters. The BWADS showed very stable operation of the vertical transmitter because of its omni-directional pattern along the horizontal plane. However, it also suﬀered from nulls due to polarization mismatch when the horizontal transmitter was selected. In this scenario, substantial diversity gain (11.0 dB) was observed for the vertical transmitter. For the horizontal transmitter, BWADS still improved performance by 6.0 dB compared to the monopole antenna. It was also observed that overall received power for the vertical transmitter was much higher than that for the horizontal transmitter, because of polarization matching.

97 Outdoor scenario I 6. Prone (10) 4. Prone (10) 5. Walking (10) 8. Kneeling (10) 22’ 38’ 7. Walking (10) 35’ 3. Walking (10) 29’ 9. Walking (10) 38’ 1. Walking (10) Moving path 2. Kneeling (10)

Distance 60’ Duration in second Activities

Transmitter ESL building

Real-time recorded signal strength CCDF curves for signal strength 1 0 0.9 -5 0.8 -10 -15 0.7 -20 0.6 -25 0.5 -30 0.4 5.5 dB -35 0.3 diversity gain -40 Vertical Tx. Vertical 0.2

-45 Probability(power>abscissa) 0.1 -27.5 dB -22.0 dB Normalized receiving power [dB] power receiving Normalized -50 0 10 20 30 40 50 60 70 80 90 0 Measured time [sec] -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Normalized receiving power [dB] 1 0 0.9 -5 0.8 -10 0.7 -15 0.6 -20 8.0 dB 0.5 -25 diversity gain -30 0.4 -35 0.3 -40 BWADS 0.2 Probability(power>abscissa)

Horizontal Tx. Horizontal -45 Monopole antenna 0.1 -30.0 dB -22.0 dB Normalized receiving power [dB] power receiving Normalized -50 0 10 20 30 40 50 60 70 80 90 0 Measured time [sec] -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Normalized receiving power [dB]

Figure 5.13: Illustration of the outdoor test Scenario I (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rd row figures).

98 Outdoor scenario II

5. Kneeling (10)

7. Kneeling (10) 40’ 4. Walking (10) 6. Walking (10) 40’

36’ 36’ 8. Walking (10) 2. Walking (10)

15’ 9. Walking (5) 1. Walking (5)

33’

Transmitter ESL building

Real-time recorded signal strength CCDF curves for signal strength 1 0 0.9 -5 0.8 -10 0.7 -15 0.6 -20 0.5 -25 11.0 dB -30 0.4 11.0 dB diversity gain -35 0.3 diversity gain Vertical Tx. Vertical -40 0.2 Probability (power>abscissa) Probability -45 0.1 -30.5 dB -19.5 dB Normalized receiving power [dB] power receiving Normalized -50 0 0 10 20 30 40 50 60 70 80 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Measured time [sec] Normalized receiving power [dB] 1 0 0.9 -5 0.8 -10 0.7 -15 0.6 -20 0.5 6.0 dB -25 6.0 dB 0.4 -30 diversitydiversity gain gain -35 0.3

Horizontal Tx. Horizontal -40 0.2

BWADS (power>abscissa) Probability -45 0.1 -33.5 dB -27.5 dB

Normalized receiving power [dB] power receiving Normalized Monopole -50 0 0 10 20 30 40 50 60 70 80 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Measured time [sec] Normalized receiving power [dB]

Figure 5.14: Illustration of the outdoor test Scenario II (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rd row figures). 99 Outdoor Test III

In this scenario, the eﬀects of obstacles on the antenna performance were studied.

As seen in Fig. 5.15, there are three obstacles: a full size car, the ESL building, and a metallic air conditioning system at the 1st kneeling position. Several nulls were observed in the monopole antenna when it was hidden by the tester, building, the metallic obstacles (10-20 sec, 30-40 sec) and the car (50-60 sec, 70-80 sec) for both polarized transmitters. Although a similar pattern was observed in the BWADS, the received power remained strong at 6-25 dB in these durations. Overall, 9.0 and

7.5 dB of diversity gains were obtained for the vertical and horizontal transmitter, respectively.

Outdoor Test IV

This scenario emulates a combat ﬁeld environment. The tester hid behind a variety of obstacles (tree, bricks, and metallic structures) and ran between hiding points as illustrated in Fig. 5.16. For the vertical transmitter, critical nulls in the monopole were observed when the tester was hiding behind the tree and the bricks, while the BWADS received a relatively strong signal. For the horizontal transmitter, the monopole experienced severe nulls when the tester was kneeling between metal poles and the wall due to body obstruction and polarization mismatch. During the running activity (20-30 sec), both antenna systems showed unstable performance due to the fast movement. This indicates that it is necessary to select the best antenna more frequently in the BWADS. From the CCDF curves for the received signal, 10.5 dB of diversity gain was observed for the vertical transmitter and 4.0 dB of diversity gain was obtained for the horizontal transmitter.

100 Outdoor scenario III

8. Keel ing (10)

9. Walking (10) 7. Walking 44 ’ (10) 13 ’

6. Hiding (10) 48 ’

1. Running (10) 5. Walking (10) 57 ’ 50 ’ Transmitter 4. Kneel ing (10)

27 ’ 2. Standing (10) 3. Walking (10) ESL building

Real-time recorded signal strength CCDF curves for signal strength 1 0 0.9 -5 0.8 -10 0.7 -15 -20 0.6 -25 0.5 9.0 dB -30 0.4 9.0 dB diversity gain -35 0.3 diversity gain

Vertical Tx. Vertical -40 0.2 Probability(power>abscissa) -45 0.1 -34.5 dB -25.5 dB Normalized receiving power [dB] power receiving Normalized -50 0 10 20 30 40 50 60 70 80 90 0 Measured time [sec] -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Normalized receiving power [dB] 1 0 0.9 -5 -10 0.8 -15 0.7 -20 0.6 -25 0.5 -30 0.4 7.57.5 dB dB -35 0.3 diversity gain -40 diversity gain BWADS 0.2 Probability(power>abscissa)

Horizontal Tx. Horizontal -45 Monopole 0.1 -28.0 dB Normalized receiving power [dB] power receiving Normalized -35.5 dB -50 0 10 20 30 40 50 60 70 80 90 0 Measured time [sec] -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Normalized receiving power [dB]

Figure 5.15: Illustration of the outdoor test Scenario III (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rd row figures).

101 Outdoor scenario IV

2. hiding (10)

76’ 3. running (10) 1. running (10)

29’ 10.walking (10) 92’ 48’ Metal ladder Metal pole 9.walking (10) 40’

8. Kneeling (10) Transmitter 5. Walking 31’ (10) 7. Walking (10) Foam post 6. Kneeling (10) bricks 42’ 4. hiding (10) ESL building

Real-time recorded signal strength CCDF curves for signal strength 1 0 0.9 -5 0.8 -10 0.7 -15 0.6 -20 -25 0.5 0.4 10.5 dB -30 10.5 dB 0.3 -35 diversitydiversity gain gain

Vertical Tx. Vertical -40 0.2 Probability (power>abscissa) Probability -45 0.1 -30.5 dB -20.0 dB Normalized receiving power [dB] power receiving Normalized -50 0 0 10 20 30 40 50 60 70 80 90 100 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Measured time [sec] Normalized receiving power [dB] 1 0 0.9 -5 0.8 -10 0.7 -15 -20 0.6 -25 0.5 4.04.0 dB dB -30 0.4 diversitydiversity gain gain -35 0.3 -40 BWADS 0.2 Probability (power>abscissa) Probability

Horizontal Tx. Horizontal -45 Monopole 0.1 -32.0 dB -28.0 dB Normalized receiving power [dB] power receiving Normalized -50 0 10 20 30 40 50 60 70 80 90 100 0 Measured time [sec] -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Normalized receiving power [dB]

Figure 5.16: Illustration of the outdoor test Scenario IV (top figure), and test results for vertical transmitter (2nd row figures) and for horizontal transmitter (3rd row figures). 102 5.4.4 Indoor Tests

Two separate indoor tests were conducted in the ESL building corridors to evaluate the performance of both antenna systems in rich multi-path fading environments.

These tests emulate not only an indoor environment but also a dense urban environment. For these tests, a vertically polarized transmitting antenna was located at the end of the 1st ﬂoor corridor, and measurements were performed at the 1st ﬂoor and

2nd ﬂoor corridors as depicted in Fig. 5.17 and 18.

First, it was observed that the monopole antenna suffered significant loss from multi-path fading effects, represented by sharp nulls in the real-time measurement results. Second, we observed that the overall received power of the monopole on

Indoor scenario I – 1st floor

110

50 120(end) 100 90 80 Tx. start 10 sec 40 60 70

20 30

Real-time recorded signal strength CCDF curves for signal strength 0 1 BWADS Monopole -10 0.9 0.8 -20 0.7 -30 21.0 dB 0.6 21.0 dB diversity gain -40 0.5 diversity gain -50 0.4 0.3 -60 0.2

-70 Probability(power>abscissa) 0.1 -75.5 dB -54.5 dB

Normalized receiving power [dB] power receiving Normalized -80 0 0 10 20 30 40 50 60 70 80 90 100 110 120 -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0 Measured time [sec] Normalized receiving power [dB]

Figure 5.17: Illustration of the indoor test Scenario I (top ﬁgure), and test results for vertical transmitter (bottom ﬁgures).

103 Indoor scenario II – 2nd floor 10 sec

110 120(end) 100 90 80 start 20 40 50 60 70

Real-time recorded signal strength CCDF curves for signal strength 0 1 BWADS Monopole -10 0.9 0.8 -20 0.7 14.5 dB -30 0.6 14.5 dB diversity gain diversity gain -40 0.5 0.4 -50 0.3 -60 0.2 -83.0 dB

Probability(power>abscissa) -68.5 dB -70 0.1

Figure 5.18: Illustration of the indoor test Scenario II (top ﬁgure), and test results for vertical transmitter (bottom ﬁgures).

the 2nd floor was much less than on the 1st floor, because there is no line-of-sight signal to the 2nd floor. In contrast, the BWADS had smoother curves for real time received power and relatively similar power levels between the 1st and the 2nd floor measurements implying robust operation in the multi-path environment. BWADS reception was weak only at the farthest region from transmitter (between 70-80 sec duration) in the 2nd floor measurement, due to severe path loss.

104 Overall, 21 dB and 14.5 dB of diversity gains were obtained for the 1st and 2nd

ﬂoor tests, respectively. This diversity gain is much higher than in one of the outdoor environments, implying that the BWADS is quite powerful in rich multi-path environments.

5.4.5 Field Test Summary

In all the test scenarios discussed above, the proposed body-worn antenna diversity system (BWADS) repeatedly exhibited better performance than the monopole antenna. This is summarized in Fig. 5.19 which shows the CCDF curves for outdoor and indoor scenarios, respectively, compiled from all ﬁeld test data at all measured frequencies. It shows that BWADS achieved 7.0 dB of diversity gain in outdoor tests

(see Fig. 5.19(a)) and 16.5 dB of diversity gain in indoor tests (see Fig. 5.19(b)). This is primarily due to the monopole suﬀering from body obstruction and polarization mismatch in the outdoor environments. Moreover, multipath cancellation degrades the monopole’s performance. As a result the improvement with BWADS over the monopole antenna is greater in the indoor environments. These diversity gains can also be interpreted as improvements in communication range. Our results imply 2.2 times and 6.7 times range improvement in outdoor and indoor environments, respectively.

5.5 Conclusions

A wideband UHF (225-450MHz) body-worn antenna diversity system (BWADS) for typical bulletproof vests was presented. First, 4 antennas were designed to be mounted inside a bulletproof vest and to provide complete pattern and polarization coverage above the horizon regardless of the wearers’ activities. Since the antennas

105 1 0.9 BWADS Monopole 0.8 0.7 0.6 7.0 dB 0.5 diversity gain 0.4 0.3 0.2

Probability(power>abscissa) 0.1 -32.5 dB -25.5 dB 0 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Normalized receiving power [dB] (a) 1 BWADS 0.9 Monopole 0.8 0.7 16.5 dB 0.6 diversity gain 0.5 0.4 0.3 0.2 Probability (power>abscissa) Probability -80.5 dB -64.0 dB 0.1 0 -100 -95 -90 -85 -80 -75 -70 -65 -60 -55 -50 -45 -40 -35 -30 -25 -20 -15 -10 -5 0 Normalized receiving power [dB] (b)

Figure 5.19: CCDF comparison of received power including all scenarios for (a) outdoor and (b) indoor test.

106 were also very light and ﬂexible, they did not interfere with the users’ mobility.

Second, a compact diversity module (79 × 41 × 28 mm) was fabricated to achieve an omni-directional radiation pattern at UHF frequencies. It operates in high speed to ﬁnd and select the best channel among the antennas in real time. It could be also easily integrated with the previous radio system. Its low power consumption provided suﬃcient operating time (60 hrs with 1800 mAh battery).

Finally, the integrated BWADS was validated in realistic outdoor and indoor communication environments and compared to the commercial monopole whip antenna currently used in radio systems at the same frequency band. In total, 4 outdoor and

2 indoor tests were performed, and BWADS provided superior performance in all tests as summarized in Table 5.3. Measured data showed signiﬁcant improvements over the monopole whip antenna. As much as 16.5 dB diversity gain in a multi-path indoor environment and 7 dB diversity gain in outdoor environments were achieved, demonstrating that BWADS is highly eﬀective for rich multi-path environments.

Table 5.3: Summary of the test results in outdoor and indoor environments. Diversity gain (dB) Transmitter Environments polarization Scenario 1 Scenario 2 Scenario 3 Scenario 4 All scenarios

Vertical 5.5 11.0 9.0 10.5 Outdoor 7.0 Horizontal 8.0 6.0 7.5 4.0

Indoor Vertical 21 14.5 - - 16.5

107 Chapter 6: Future Works

6.1 Introduction

In this dissertation, we proposed an efficient evaluation method for body-worn antenna (Ch. 3), a systematic optimization process providing the best antenna locations for body-worn diversity system (Ch. 4), and a vest mounted body-worn antenna diversity system for off-body communication (Ch. 5). The proposed methods can be extended to other mobile platformed diversity system, and the body-worn antenna system can be used for different application such as vital sign monitoring and RF power harvest.

In this regard, this chapter suggests several future research topics as follows:

1. Vehicular antenna diversity system design.

2. Vital sign monitoring system using body-worn antennas.

3. Body-worn antenna system for RF power harvest.

Each section discusses the research issues for each application and provides simulation results showing the feasibility of each application and remaining problems to solve for improving the system.

108 6.2 Vehicular Antenna Diversity System

Antenna diversity is one of the most common and simplest methods for mitigating

shadowing and fading associated with rich multipath radio environments. However,

there are numerous challenges in evaluating and designing an optimal antenna diver-

sity system. In this regards, we proposed the channel decomposition method (CDM),

which can eﬃciently evaluate body-worn antennas in various communication environ-

ments, and the systematic diversity design method, which ﬁnds the optimal antenna

conﬁgurations with the minimum number of antennas for body-worn antenna diver-

sity system.

These two methods can be directly applied to diversity system design for other

mobile platforms such as cars, UAVs, etc. This section describes and example for

vehicular antenna diversity system for a speciﬁc communication scenario. Fig. 6.1

shows the design conﬁguration and communication scenario for a vehicular antenna

diversity system. The antenna element used in this study was developed for mini-

mizing detuning eﬀects due to diﬀerent ground plane sizes [58]. In total, 51 available

antenna positions located in front and back panels, side doors and roof of the cars

were considered for the diversity system as shown in Fig. 6.1 (top ﬁgures). For the

4-antenna diversity system, we need to evaluate 249900 conﬁgurations (=51C4) without the proposed systematic diversity design method, but we only need to calculate

51 antenna conﬁgurations using the proposed method. The diversity system goal is to provide 90 % of pattern coverage with 4 dBi minimum gain within the speciﬁed communication sector (60◦ <θ< 90◦ and 0◦ <φ< 360◦). By applying the proposed method we developed a 4-antenna diversity system located on both side doors and front and back panels which provides 90.5 % of pattern coverage as shown in

109 Humvee ’s real estate for Antenna locations (51 ea.)

antennas Roof (18, both pol.) Back Panel (9) Roof 86” 86” 86” 21” 68” Side Doors (9*2) 60” Side Door 177” Front Panel Back Panel Front Panel (6) 86” 30” 10”

Antenna element Antenna placement locations 3” 3” Recessed DBE antenna Communication scenario (3”x3” ground plane) Design goal: Vertical - Gmin = 4 dBi polarization

60 °°°<θ<90 °°°

Figure 6.1: Simulation model for vehicular antenna diversity system and communication scenario considered.

Fig. 6.2. We also compare the best and worst 4-antenna diversity system in Fig.

6.3. Each antenna is located on the same panel but in slightly diﬀerent positions.

For both the worst and best cases, however, their diversity performances are quite

diﬀerent. The worst 4-antenna diversity conﬁguration provides only 68.5 % pattern

coverage. This is mainly caused by cancellation between the direct signal and the

reﬂected signal from the ground. Note that the antenna heights from ground for the

worst conﬁguration are similar causing deep nulls at regions between θ = 74◦ and θ

= 80◦. These results emphasize that determining the proper antenna locations for a diversity system is very important, highlighting the value of the proposed systematic diversity design method.

110 There are several issues to resolve in the design process for large mobile platform diversity systems such as cars and airborne vehicles. First, the process requires substantial computational resource for accurate EM full-wave simulation. Thus, approximate numerical methods, such as physical optics (PO) and uniform theory of diﬀraction (UTD), should be considered for evaluating the diversity system. Sec- ond, optimization methods like genetic algorithm (GA) should be considered to ﬁnd optimum location and polarization of the antennas for diversity systems due to the large real estate available (hence large search space for the algorithm). To construct such an algorithm, we need to integrate the optimization method and EM calculation method to avoid the large computation burden required for evaluation.

Pattern (Top view) Single antenna 2-ant. diversity

Effective antenna gain ccdf curves

3-ant. diversity 4-ant. diversity

Figure 6.2: Best antenna diversity conﬁguration and antenna gain CCDF curves for single, 2-, 3-, and 4-antenna diversity system.

111 Optimized antenna locations Pattern comparisons

Right Door Back Panel Best 4 antenna diversity Worst 4 antenna diversity

Interested Communication Sector Front Panel Left Door

Best antenna locations for 4-antenna diversity Worst antenna locations for 4-antenna diversity

Gain [dBi] Pattern (top view)

Figure 6.3: Diversity performance comparison between the worst and the best 4- antenna diversity system.

6.3 Vital Sign Monitoring System using Body-Worn Anten- nas

One of the primary vital signs for humans is respiratory rate. Typical respiratory rates for adults are 12-20 breaths/min, and inhalation occurs over 1/3 of the respiration interval. During respiration, lungs are deflated and inflated changing their volumes (deflated: 2.2 liter, inflated: 2.7 liter on average). EM properties are another important difference between deflated lungs and inflated lungs. Relative permittivity

(ǫr) and conductivity (σ) change with air content as described in Table 6.1.

For this reason, a pair of body-worn antennas located in the front torso and the back torso can monitor the lungs’ activity as a vital sign. First, the operation

112 Table 6.1: Electromagnetic properties of deﬂated lung and inﬂated lung at selected frequencies (30 MHz, 433 MHz, 915 MHz, 2.45 GHz, and 12 GHz). Tissues EM properties f=30 MHz f=433 MHz f=915 MHz f=2.45 GHz f=12 GHz ε Relative permittivity ( r’) 98.851 54.198 51.373 48.381 35.343 Deflated Conductivity ( σ) [S/m] 0.4896 0.6945 0.8637 1.6825 12.772 lung Penetration depth ( δ) [mm] 155 58 45 22 2.6 ε Relative permittivity ( r’) 54.207 23.586 21.972 20.477 15.096 Inflated Conductivity ( σ) [S/m] 0.2628 0.3799 0.4593 0.804 5.2743 lung Penetration depth ( δ) [mm] 212 71 55 30 4

frequency is selected as 433 MHz because the penetration depth of the frequency

is acceptable and physical antenna size is also reasonable. In the simulation setup,

antennas are inserted into the skin to increase the transmitted power into the body.

The approximate human model suggested for in-body communication (4-layer model

proposed in Chapter 2) is used except lung size is altered as shown in Fig. 6.4.

The cross section area of the deflated lung is 97 × 44 mm2 and for the inflated lung is 108 × 48 mm2. Also, different EM properties for the deflated and inflated lungs (see table 6.1) are used for each simulation. Path gains are calculated between two antennas for the deflated and inflated lungs. We observed 3 dB of path gain difference between each state. Therefore, we can monitor the human respiration and health condition. This system can be applied to life and health monitoring system for firefighters, police officers, etc.

The practical issue of this system is how to insert the antennas into the human skin. One of the solutions is to use the saline gel to cover the antennas and the skin as used in breast cancer detection system [59]. Since the saline solution can have similar EM properties of human tissue, it helps increase penetrating waves on the

113 interface between the human body and the antennas. It also impedes the antennas of receiving the unwanted signals from outer sources and increases the sensitivity of the vital signal monitoring system.

Skin Fat Muscle

Ant. 1 Ant. 2 Lung

(a) (b)

-42 inflated lung deflated lung -44

-46

-48 magnitude [dB] 21 S -50

-52 429 431 433 435 437 Frequency [MHz] (c)

Figure 6.4: Simulation model of vital sign monitoring system and results; (a) side view, (b) top view of the model, and (c) path gain comparison.

114 6.4 Body-Worn Antenna System for RF Power Harvest

The most promising application of personalized sensor networks is the personal health monitoring system consisting of wearable medical sensors and body area network (BAN) wirelessly connected to a home doctor or a hospital. Dramatic changes in demography (increase of population for the aged) and lifestyle (increase of nu- clear family) exert a great pulling force for development of personal health monitoring system. Also, this is fuelled by substantial advances in microelectronics and communication technologies. To satisfy the needs and expectations of the users, all components should be integrated seamlessly with providing comfortable/easy-to-use and low maintenance of the system. To do so, the development of self powered device is essential to eliminate the necessity of wiring and batteries.

In this regards, energy harvesters generating a relatively small power are being developed for wearable applications. In general, they utilize 1) mechanical energy generated by users’ movements, 2) solar energy, 3) heat flow caused by temperature difference between user and ambient, and 4) radio frequency (RF) energy presented in ambient to generate the electrical power [60]. Availability of first three energy sources is limited by on users’ condition and environment, but RF energy presents in most time and area regardless of users environment. Thus RF energy harvester is the most promising candidate for wearable applications.

Fig. 6.5 depicts the ambient RF energy measured from 20 MHz to 3000 MHz.

Strong signals were observed at the FM radio band (87.8-108.0 MHz) and maximum receiving powers depending on the receiver positions were recorded from -17.4 to -12.5 dBm at 90 MHz using biconical receiving antenna (see Fig. 6.5(a)) with 0 dBi gain.

We also observed relatively strong signals at GSM band (869.2-894.2 MHz) and Wi-Fi

115 −10

−20

−30

−40

−50 Received power [dBm]

−60

−70 0 500 1000 1500 2000 2500 3000 Frequency [MHz] (a) (b)

Figure 6.5: (a) Measurement setup and (b) ambient RF power measured by 0 dBi receiver at the backyard of OSU-ESL building (Wi-Fi signal is measured at the hallway of OSU-ESL building).

band (2401-2495 MHz). Since the maximum ambient RF power was observed at 90

MHz and FM radio signals presents in everywhere, it is most desirable to use this frequency for RF energy harvester. Since the size of efficient antenna operating at this frequency is too big to be implemented for personal purposes, body-worn antenna will be the only practical solution for the antenna in RF energy harvester utilizing the FM radio signals. Rectenna is used as the energy harvester of RF signals and consists of receiving antenna and rectifier. For efficient rectenna, the antenna should capture the RF signals efficiently from any directions and the rectenna should have high RF to DC conversion efficiency. First, Fig. 6.6 shows the body-worn antennas designed for VHF band and calculated gain and 3-D pattern. The antennas in array

116 Inductor (100 nH)

614 mm

Feed

305 mm

(a) (b) (c)

−5

−10

−15

−20

Realized peak gain [dBi] −25

−30 60 70 80 90 100 Frequency [MHz] (d) (e)

Figure 6.6: (a) Front view and (b) rear view of body-worn antennas, (c) antenna element structure, (d) 3-D radiation pattern and (e) peak gain of the array antenna.

117 −20

−22

−24

−26 Received RF power [dBm] −28 Array antenna Front antenna Rear antenna −30 0 45 90 135 180 225 270 315 360 Azimuth angle, φ [deg]

Figure 6.7: Calculated RF power received from array antenna and each single antenna in horizontal direction.

combining has quasi omni-directional pattern and -10.1 dBi peak gain at 90 MHz.

Based on the ambient RF power measured (Fig. 6.5) and the gain of the body-worn

antenna (Fig. 6.6), the received power from array antenna and each signle antenna at

90 MHz is calculated and shown in Fig. 6.7. Since the RF signal source is grounded

radio broadcasting station, we assume that angle of arrivals of the signals are limited

from θ = 70◦ to θ = 90◦ in all horizontal directions. In this communication sector, the antennas can receive the RF power from -27.2 dBm to -22.0 dBm. Fig. 6.8 shows the schematics of the modiﬁed Greinacher rectiﬁer developed by our research group

118 Node B C1 D2

D1 C2 Iout

GND Port #1 Matching DC Load network (Antenna Port) D3 C4

C3 D4 Node C

Modified Greinacher Rectifier

Figure 6.8: Schematics of the modiﬁed Greinacher rectiﬁer [61].

[61]. The most important specification of the rectifier is the RF to DC conversion efficiency, and it depends on the RF input power, operating frequency, impedance matching, and diode properties. RF to DC conversion efficiencies of the rectifier are calculated [62] and the result shows from 43 % to 51 % of efficiency with given RF input power, operating frequency, antenna impedance, and properties of the diode

(SMS7630 schottky diode). DC power generated from body-worn antenna can be simply increased by using multiple antennas. There are two way in using the array antenna to generate DC power. The ﬁrst method is to generate the DC power after combining the RF signals, and the second method is vice versa. Finally, Fig. 6.9 shows the CCDF curves of the generated DC power from the proposed rectenna systems.

It shows that the combined DC power (DC combining in the legend) generated from each antenna is larger than DC power generated from array antenna (RF combining

119 in the legend), and it generates -25.2 dBm DC power during 50 % of time. It is enough DC power to operate a digital thermometer with backlighting LCD display

[63].

0.9

0.8

0.7

0.6

0.5

0.4

0.3 DC combining 0.2 RF combining Probability (DC power > abscissa) 0.1 Front antenna Rear antenna 0 −40 −35 −30 −25 −20 Harvested DC power (dBm)

Figure 6.9: CCDF curves of the generated DC power from the proposed body-worn rectenna systems.

120 Chapter 7: Conclusions

Body-centric wireless communication systems (BWCS) have become a key means of providing and connecting all wireless services for personal users. BWCS have a broad range of applications including health care, smart home, personal entertain- ment, communication, and unobstructed communication systems for specialized occupations such as firefighters, police officers, etc. BWCS can be divided into three communication domains: 1) in-body communication, 2) on-body communication, 3) off-body communication. Although each domain is equally important, body-worn antennas for off-body communication were the focus of this study.

Body-worn antenna systems generally suffer from antenna performance degrada- tion due to the interaction of electromagnetic waves with the human body and the surrounding environment. To address these concerns, I first proposed the approximate human body model in Chapter 2 and an efficient evaluation method in Chapter

3 for body-worn antennas. A systematic design method for antenna diversity systems was introduced in Chapter 4 to improve performance of the body-worn antenna system. Using the proposed methods, a vest-mounted body-worn antenna diversity system which showed substantial improvement compared to a commercially available antenna system for wideband UHF operation was designed in Chapter 5. Each chapter clearly identiﬁed the importance of each method in the beginning. Each method

121 was validated by simulation studies and/or experimental results with several design

examples.

The contents of each chapter are summarized as follows.

Chapter 2 presented the human body’s frequency dependent EM properties from

VHF to X-band. Human body tissues were categorized into two types (high-water content and low-water content) according to their EM properties. Dominant wave propagation modes around the body were also studied for each body-centric wireless communication domain. This study suggested proper types and mounting configurations of the antennas for each body-centric communication mode. For efficient evaluation of body-worn antennas without loss of accuracy, approximate human torso models using concentric elliptical cylinders with multi-tissue layers were proposed for each application. Dipole antennas placed horizontally on the human body and a homogenous human model were selected for off-body communication and applied to body-worn antenna design in this dissertation.

Chapter 3 introduced an eﬃcient evaluation method for the body-worn antenna system. The channel decomposition method (CDM) was proposed for eﬃciently evaluating body-worn antennas with a surrounding environment. The proposed method showed a 10-fold computational speed increase over the full channel evaluation by separating the entire communication channel into a human body channel and an environment channel. For validation, full channel evaluation results were presented and compared to the proposed method; they were found to be in good agreement. Then,

CDM was used to design a body-worn antenna diversity system and a 4-antenna diversity system was found to provide good performance. The proposed 4-antenna diversity system showed a substantial improvement (up to 40 dB) in communication

122 performance compared to the worst case of the single body-worn antenna from the

indoor measurement results.

A systematic design method for ﬁnding the best antenna conﬁgurations for body-

worn antenna diversity system was proposed in Chapter 4. The proposed method was

based on a fundamental assumption that mutual coupling between antennas selected

in an eﬃcient diversity system was negligible. This assumption simpliﬁed the EM cal-

culations and reduced the computation time signiﬁcantly. The proposed method was

validated by comparing the results with a full evaluation method including mutual

coupling effects at different operation frequencies in two different communication en-

vironments. The comparison showed that the best diversity conﬁgurations from both

evaluation methods were identical for all scenarios. Furthermore, the performance

parameters (pattern coverage and correlation coeﬃcient) were in excellent agreement

with tiny errors. The method was also veriﬁed by measurements in an outdoor envi-

ronment. In conclusion, the proposed method was very simple and straightforward,

and reduced computation time substantially without loss of accuracy for designing

body-worn antenna diversity systems.

Chapter 5 showed the performance of the body-worn antenna diversity system in

a real application. We proposed a vest-mounted body-worn antenna diversity system

(BWADS) for wideband (225-450 MHz) UHF operation. The system consisted of a set

of 4 light-weight conformal wideband antennas embedded in a standard bulletproof

vest with optimal mounting conﬁgurations providing the best possible performance

and a compact diversity module (79 × 41 × 28 mm) optimally combining the signal from each antenna in real time. The proposed BWADS had robust performance providing a nearly omni-directional pattern in the horizontal plane, even when users

123 carry backpacks with other metallic equipments. The proposed system was tested in

4 outdoor and 2 indoor realistic communication environments with real-time activity and compared to the monopole whip antenna currently used in commercial radio systems. Measured data showed signiﬁcant improvements over the monopole antenna.

As much as 16.5 dB diversity gain in rich multi-path environments (indoor) and 7 dB diversity gain in outdoor environments were achieved. These gains imply that the proposed system provided factors of 6.7 and 2.2 communication range improvements in indoor and outdoor environments, respectively.

Chapter 6 suggested several future research topics. The proposed evaluation method (CDM in Ch. 3) and systematic diversity design method (Ch. 4) could be used for other vehicular antenna diversity systems. The design example was given in the second section, highlighting the beneﬁts of the proposed diversity design method.

A pair of body-worn antennas was used for monitoring the wearer’s vital signs in section 3. Practical issues of this application were also discussed. In the last section, body-worn antennas were used for power harvesting from ambient RF signals. This system could provide suﬃcient power to operate body sensors and could increase the operation time of sensors and increase maintenance intervals.

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