ANALYSIS AND DESIGN OF AN RF FRONT END FOR A RADAR DIGITAL
RECEIVER
A Thesis
Presented to the
Faculty of
California State Polytechnic University, Pomona
In Partial Fulfillment
Of the Requirements for the Degree
Master of Science
In
Electrical Engineering
By
John O. Mortensen
2018
SIGNATURE PAGE
THESIS: ANALYSIS AND DESIGN OF AN RF FRONT END FOR A RADAR DIGITAL RECEIVER
AUTHOR: John O. Mortensen
DATE SUBMITTED: Winter 2018
Electrical and Computer Engineering Department
Dr. James Kang Thesis Committee Chair Electrical Engineering
Dr. Thomas Ketseoglou Electrical Engineering
Dr. Saloman Oldak Electrical Engineering
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ACKNOWLEDGEMENTS First, I’d like to thank my adviser Dr. James Kang for the patient help he has provided over the two years of my pursuit of my master’s degree. In addition I’d like to thank Dr. Ketseoglou and Dr. Oldak and the rest of the staff at Cal Poly for all of the great electronics courses I’ve taken both recently and back in the 1980s when I first received my BSEE. I’d should also thank MPT and its owner Dr. Rick Sturdivant for the help on this project and the funding available to complete this research and project. I’d also like to thank my wife Marie for putting up with me and the support she has given me over the years.
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ABSTRACT
This paper focuses on the use of commercial off the shelf parts in the design of an X Band receiver used in radar. The need today for a low cost flexible approach is of the utmost importance due to the trend of having a transceiver module for each element of a phased array antenna. Therefore, smaller and cheaper units are being demanded. An analysis of the given specifications is first presented. A detailed specification is developed and then the analysis and design is covered for each of the main components in the analog section. Several of the components are measured individually and the resulting test data is reported. The individual component test data provides information required for modeling of the receiver and confirms performance. The digital portion of the receiver includes an analog to digital converter (ADC) evaluation board. Since the focus of this research is on a low cost, high performance analog front end the ADC was used for capturing the data. A detailed model is developed using an Excel spreadsheet. It calculates system level performance parameters such as gain, noise figure, third order intercept point, large signal maximum before damage and dynamic range. The spreadsheet can be used to optimize the design for desired results. A circuit board of the analog section was designed, assembled and measured. It was connected to the ADC evaluation board for measurements. The measured data confirms the predicted performance from the spreadsheet analysis. The research demonstrates that a cost effective and high performance digital receiver is achievable.
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TABLE OF CONTENTS
SIGNATURE PAGE ...... ii
ACKNOWLEDGEMENTS ...... iii
ABSTRACT ...... iv
LIST OF FIGURES ...... viii
CHAPTER 1 ...... 1
Introduction ...... 1
CHAPTER 2 ...... 3
Block Diagram and Specifications ...... 3
Block Diagram ...... 3
Front End RF Section...... 4
1st IF Section ...... 4
2nd IF Section ...... 5
Separate ADC Digital Receiver Processor ...... 5
Specifications ...... 5
Frequency Input ...... 6
Power Input Max with No Damage ...... 7
RF Input Level ...... 7
LO Frequencies ...... 7
ADC Requirements ...... 7
Output Frequency...... 8
ADC Sampling Rate ...... 8
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ADC Bits ...... 8
Dynamic Range Specifications ...... 8
Noise Figure ...... 8
Linearity ...... 11
Intermodulation Distortion...... 12
3rd Order Intercept Point ...... 14
Compression ...... 18
Mixer Products ...... 19
Spurious Free Dynamic Range ...... 19
Image Rejection ...... 20
CHAPTER 3 ...... 23
Design ...... 23
Approach ...... 23
Description ...... 24
Design Simulation and Compliance...... 25
CHAPTER 4 ...... 28
Components ...... 28
Front End RF Section...... 29
Limiter...... 29
First RF Preselect and Image Filter ...... 30
First Variable Attenuator ...... 32
First Low Noise Amplifier (LNA) ...... 33
Second Variable Attenuator ...... 34
Second LNA...... 34
Second RF Preselect and Image Filter ...... 34
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RF Mixer ...... 35
Passive Attenuators ...... 35
1st IF Section ...... 36
IF Filter ...... 36
IF Amplifier ...... 38
IF Mixer ...... 39
2nd IF Section ...... 39
Voltage Amplifier ...... 39
Anti-Aliasing Filter and ADC Section ...... 40
CHAPTER 5 ...... 43
Test Results ...... 43
RF Filter ...... 43
1st IF Filter...... 44
2nd IF Filter ...... 45
Gain ...... 45
Noise Figure ...... 46
Linearity ...... 47
Mixer Products ...... 50
Image Response ...... 50
Receiver Overall Specification Compliance ...... 51
Receiver Cost ...... 52
CHAPTER 6 ...... 54
Conclusion and Future Work ...... 54
REFERENCES ...... 55
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LIST OF FIGURES
Figure 1. Block Diagram...... 3 Figure 2. Specifications ...... 6 Figure 3. Cascaded Gain and Noise Figure ...... 9 Figure 4. Equivalent Input Noise Source ...... 10 Figure 5. Intermodulation Distortion ...... 14 Figure 6. Input and Output 3rd Order Intercept Point ...... 15 Figure 7. Intermodulation Tones ...... 16 Figure 8. Graphical Representation of IMD Measurement ...... 17 Figure 9. 1dB Compression Point ...... 18 Figure 10. Image Frequency Range ...... 22 Figure 11. Simulated Compliance Matrix ...... 26 Figure 12. Overall Receiver Spreadsheet...... 27 Figure 13. Receiver Board ...... 28 Figure 14. Front End Block Diagram ...... 29 Figure 15. Front End Limiter Issue ...... 30 Figure 16. RF Preselect Filter Image Reject Specification ...... 31 Figure 17. RF Preselect Edge-Coupled Bandpass Filter ...... 32 Figure 18. 1st IF Section ...... 36 Figure 19. 2nd IF image specification ...... 37 Figure 20. Band of frequencies from RF front end that create 2nd IF images ...... 38 Figure 21. 1st IF Capacitive Coupled Quarter Wave Resonator Filter ...... 38 Figure 22. 2nd IF Section...... 39 Figure 23. Bandpass Sampling of Signal ...... 41 Figure 24. 2nd IF Filter Requirements ...... 42 Figure 25. 2nd IF Lumped Component Bandpass Filter ...... 42 Figure 26. RF Filter Response (Measured and Simulation) ...... 43 Figure 27. 1st IF Filter Response (Measured and Simulation) ...... 44 Figure 28. 2nd IF Lumped Component Filter Measurement ...... 45 viii
Figure 29. Gain vs Frequency ...... 46 Figure 30. Noise Figure vs. RF Input Frequency...... 47 Figure 31. Compression Point at Receiver Output ...... 48 Figure 32. Output and IMD Curves ...... 49 Figure 33. Image Rejection ...... 51 Figure 34. Measured Data Compliance Matrix...... 52 Figure 35 Receiver Cost...... 53
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CHAPTER 1
Introduction Humans have a need to communicate. Clay tablets or papyrus were used in the distant past then more recently there was paper. In the modern age, it was radio, TV and telephones. Today high speed networks are needed to satisfy the trend in increased data usage per user. This resulted in the need for 4G and 5G networks. 5G-PPP (5G Infrastructure Public Private Partnership) has stated that the solution to 5G requires phased array antennas [1]. Phased arrays are being designed in a wide variety of 5G systems: back haul, base stations, access points and user equipment (cell phones, lap tops). To facilitate this, transmit/receivers (transceivers) are built to accommodate elements used in the array. The first arrays built were passive where one transceiver was used to drive any number of elements. Each element had an analog phase adjustment to create the individual beams. Later, active arrays were created that allowed magnitude control being done with analog variable gain amplifiers or adjustable attenuators and the phase control was accomplished with delay lines. Now digital phased array technology is the trend due to directional requirements in both commercial uses in cell phones or a weather radar such as CHILL at Colorado State [2] and military (radar). The signal is sampled by an ADC and the gain and phase are adjusted in the digital domain using DSP algorithms. Since more elements to the phased array means tighter beam widths and since digital technology is getting cheaper the trend now is to have a transceiver module for each element of the phased array. Therefore there is a need for the transceiver cost to be as low as possible. [3] So our goal was to design a transceiver where the manufacturing and parts cost were kept to a minimum yet a flexible approach was necessary. The flexibility comes in with only needing a re-design of filters when a change in frequency band occurs for new designs. In this way repeated effort, such as new printed circuit board designs, is kept to a minimum. In light of this we needed to choose a type of module assembly method that would reflect this trend toward cheaper transceivers. Historically these transceivers were built using a “chip and wire” approach which given the manufacturing issues this can be a very expensive proposition. It turns out 1
“Commercial Off-the-Shelf” (COTS) surface mount approaches are far less expensive and easier to manufacture. For instance, in [4] a design approach was used to optimize the dynamic range of digital receivers. However, their work did not address the issues in demonstrating low cost using COTS components. Also, in [5] the authors used a system simulation approach to digital receiver development with a stage by stage description. They also included a full system simulation. Their approach did not address the cost of the digital receivers nor provide a treatment on the use of COTS parts. Other works have attempted compact digital receivers such as in [6]. Their approach was for a 3.2 GHz application and emphasized the digital functionality rather than the analog front end. Our approach is to use COTS parts and optimized design for dynamic range but due to budget constraints only the receiver section was built and tested without the ADC and FPGA. This thesis describes the architecture, design, components, test results, and analysis of a COTS surface mount digital receiver for phased array radar. Chapter 1 serves as an introduction to the receiver design. It puts the design challenge in perspective and describes the application. Chapter 2 is where the major specifications are discussed. The design approach toward meeting the specifications is examined and the block diagram is described. Chapter 3 discusses the design and analysis of the receiver including the explanation of how the specifications will be met, the spreadsheets for cascaded gain, noise figure, intermodulation distortion and spurious free dynamic range. Chapter 4 describes the components that were chosen. For example, the low noise amplifier will be explained since it dictates the system noise floor. The front-end RF filter design will be shown since they are used to attenuate out of band signals and image rejection. The RF and IF mixers will be discussed along with the potential problems they create such as mixer spurious signals. This chapter describes the critical components in the receiver. Chapter 5 describes the test results. It shows the measured data for individual components such as the filters and amplifiers. The full receiver was measured and the test results described. The measured data confirms the module performs as expected. Chapter 6 discusses possible future work.
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CHAPTER 2
Block Diagram and Specifications Block Diagram The block diagram is shown in Figure 1. A quick overview of the block diagram follows.
LO 1 Mixer PCIe Interface 3 dB LO 2
- Mixer Attenuator FPGA , LJ RX Processor Filter Separate Digital RF Image RF
r1 Transformer ADC 3 dB - Attenuator IF Filter
...-··· ···-... Amp 3 dB - Aliasing - Filter Attenuator Anti IF Section IF st 1 Amp Variable Attenuator 3 dB - Attenuator LNA Front End RF Front Section 3 dB - Amp Attenuator IF Section IF nd 2 Variable Attenuator 3 dB - IF Filter Attenuator Filter RF Image RF 3 dB - Attenuator Limiter Transformer ········ ~---·········· ·····················/
Antenna Figure 1. Block Diagram
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Front End RF Section The RF front end provides analog functions that are required for digital receivers. For our application, the RF input frequency range is in the X Band region (9300 MHz to 10000 MHz) which is too high for direct digital sampling. Therefore, a heterodyne approach is used. A similar approach was used in [7] except that their solution was for an S Band radar signal which is much lower than our X Band region. Each of the main components in the digital receiver analog section will now be briefly described. The limiter the first component a signal sees coming into the receiver. It is required to be used to meet the Power Input maximum specification and limits the amount of power that can be sent into the receiver. The RF image filter is used to preselect the band of interest which is 9300 MHz to 10000 MHz. It is also used for image suppression. There are two such filters in the RF section and both are needed to meet requirements. The variable attenuator is used to control the signal level at the input. There are two attenuator devices so as to meet the attenuator range specification which helps increase the dynamic range of the receiver. The low noise amplifier (LNA) is an amplifier which gives power gain near the input of the receiver and also has a favorable noise characteristic that allows the overall noise floor of the receiver to be low enough to detect very small input signals. An RF mixer is used to down convert any signal in the input range at X Band down to the 1st IF section at 1120 MHz. 1st IF Section The IF filter is used to shape the final information bandwidth of 30 MHz as required. It also serves to attenuate any out of band signals that could create nonlinear behavior to the section of the receiver that follows. It also is used for image suppression of the 1st IF Mixer. There are two such devices in the 1st IF section After the first IF filter there is another amplifier that boosts the power gain at this point. It helps the receiver meet the overall gain requirement. After the amplifier is the second IF filter in the chain. Two of them are used to meet the overall filter requirements of this stage. 4
Next there is a transformer and the 1st IF mixer. The mixer is a highly linear double balanced mixer which uses differential input signals. The need converting from a single ended signal to differential is the responsibility of the transformer. The double balanced mixer tactic is used to suppress many of the even harmonics and some MxN products that are typically output from a mixer. This approach eases the requirements for the anti-aliasing filter in the 2nd IF section as well. 2nd IF Section Out of the 1st IF mixer the next device is a voltage amplifier. This gives the final gain boost for the receiver and has a high enough intercept point that is well beyond the full scale input of the ADC. One of the stated goals was to make certain there would be no intermodulation distortion (IMD) products that rise above the noise floor for any two-tone signals at the output up to the full scale input of the ADC. This would help guarantee the spurious free dynamic range (SFDR) of the receiver be limited by the ADC alone and be the focal point for the non-linearity of the overall system. Given that the ADC will reside on a separate board any linearity improvements can be made by choosing a different ADC with little or any changes to the receiver itself. The last device in the chain is the anti-aliasing filter. Before an analog to digital conversion can be applied all analog signals need to be filtered to suppress any out of band signals. This prevents these signals from aliasing back into the band of interest once in the digital domain. Separate ADC Digital Receiver Processor Even though it was not physically a part of the receiver design, the ADC specifications are noted in the previous section on the 2nd IF. All of the Output Requirements from the specifications in Figure 2 such as number of bits, sampling rate, final bandwidth were all derived based on the choice of the ADC. Specifications The specifications are shown in Figure 2. A discussion of each of the specification follows.
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Requirements Specification Units RF Input Frequency Input 9300-10000 MHz Power Input Max, No Damage 37 dBm RF input level -30 dBm LO Input First LO Frequency 8180-8880 MHz 2nd LO Frequency 1030 MHz Output and ADC Requirements Output Frequency 90 MHz Bandwidth 30 MHz ADC Sample Rate 120 MSPS ADC Bits 16 Bits ADC Aliasing Rejection > 60 dB Digital Step Attenuator Attenuator Range 30 dB dB Attenuator Step Size < 1 dB Dynamic Range Specifications Noise Floor (output) < -50 dBm Noise Figure < 8 dB Image Suppression > 60 dB Spurious Free Dynamic Range > 55 dBc Figure 2. Specifications
Frequency Input There are several reasons for the choice of X Band (9300 MHz – 10000 MHz) for a radar receiver. Cost and customer needs are two of them. Specifically, an X Band range allows antenna gain to be higher for the beam area of the antenna when compared to something at a lower frequency such as at S Band (2 GHz-4 GHz). Depending on the application, the radar could need to find both the direction and speed of a potential object so a narrower beam is usually desired. Therefore even higher frequencies would be desired but this is where cost comes in. Presently at higher frequencies above X Band the costs increase significantly due to component availability and packing issues.
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Power Input Max with No Damage Most radar systems employ either a two antenna system or a one antenna system that uses a power splitter between receiver and transmitter. With either approach there is a potential for the transmitter power to leak back into the receiver. Given that most radar output power is very high, the leakage could cause damage to the input of the receiver. There is also the possibility for jamming, returns from nearby targets and clutter such as trees and landscape which could also overwhelm the receiver input and cause damage. Most receivers are designed specifically to expect very small input levels for sensitivity purposes. However, given the output power from a transmitter can be several orders of magnitude higher than what is expected at the receiver the damage will probably be irreparable. Therefore some sort of limiting functionality is needed to protect the receiver. RF Input Level The expected input level for the receiver is given as -30 dBm. This is the intended maximum input level for processing. The RF input level along with the chosen ADC’s full scale value will dictate the overall gain of the receiver. The value comes from the probable radar return from a typical target at distances that are expected for the specific situation where the radar is being used. The level is computed using the radar equation and what will be the assumed processing gain. LO Frequencies The LO frequencies are not really dictated by the customer. These are internally derived due to the designers choice of intermediate frequency band (IF). Since the 1st IF was designed for 1120 MHz the 1st LO frequency range needed to be 8180 MHz to 8880 MHz given the RF input frequency range of 9300 MHz to 10000 MHz. The 2nd LO frequency is derived so as to down convert the 1st IF of 1120 MHz to the 2nd IF of 90 MHz giving a 2nd LO frequency of 1030 MHz. ADC Requirements Due to the decision of keeping the receiver design to the analog section only it was necessary to specify the conditions on which the receiver output is to supply the
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ADC input. The specifications in this section were at the designer’s discretion or they were an internal choice based on hardware decisions. Output Frequency The output frequency of 90 MHz is a designer’s choice. It was arrived at by the decision made on the choice of ADC and its sampling rate. ADC Sampling Rate This requirement was a mixture of customer wishes and what the design team could find in terms of ADC (COTS) circuits available. ADC Bits If any of the specifications could be said to originate with the customer this is the one. The number of bits on the ADC was of primary importance and only after this was determined upon did the other specifications fall into place. The number of bits in a radar system will determine the processing gain available to detect smaller signals buried in the noise. Dynamic Range Specifications Dynamic range specifications involve parameters which include Noise Figure (NF) and Linearity and will combine to give an overall Spurious Free Dynamic Range (SFDR). In addition, the noise figure specification had to be met independently along with SFDR. Linearity is broken into different but related parameters so given the complexity of this a detailed discussion is necessary. Image rejection is not directly part of the SFDR but is separated out due to the fact the system will know where to expect it and therefore can possibly be eliminated in the digital domain. Noise Figure Noise Figure (NF) and Noise Factor (F) are defined as the degradation of Signal to Noise Ratio (SNR) from a device or system which has an input noise temperature of 290 K. NF is defined as the logarithm of the Noise Factor (F).
F = SNRin / SNRout (1)
NF = 20*log10( F ) (2)
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Most RF devices have NF listed in their data sheets which is usually plotted versus frequency. Amplifiers, mixers, filters and attenuators all have NF given in data sheets. NF can be used to calculate the input noise floor (given the bandwidth) or any other noise related issues and can be compared with other devices on the market. Also by using the Friis equation the overall NF of a system can be calculated when several amplifiers, mixers or filters are connected together. For a cascade of devices in series (see Figure 3) the Friis equation is Ft = F1 + (F 2 −1) / G1 + (F3 −1) / ( GG1* 2) ++ ... (Fn −1) / ( G 1* G2 * …* Gn −1) (3) Where Ft is the total noise factor, Fn is the noise factor of a specific device and Gn the gain of the device [8]. The parameters Ft, Fn and Gn in the Friis formula are linear and not in dB. The total Gain is Gt = G1* G2*...* Gn (4)
Total Noise Factor = Ft Total Gain = Gt
G1 G2 G3 G4 G5 F1 F2 F3 F4 F5
Figure 3. Cascaded Gain and Noise Figure
The Friis formula shows that the important setting for low overall NF is to have an amplifier that has high gain and low NF as close to the front of the chain as possible. Notice how the noise factor for the first block (F1) adds directly to the total overall noise factor Ft whereas the noise factor of the second block in the chain F2 is divided by the gain for the first block G1 before its addition to the overall noise factor. This continues down the chain where the last block’s noise factor has little to no contribution to the overall noise factor (and the NF) due to the gain that exists before it in the chain of devices.
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This makes sense if one thinks of the NF as a degradation in SNR. By definition the NF is the amount a specific device would degrade SNR but only if the input noise temperature is 290K. Noise Temperature (Ts) relates to the amount of noise that is seen by an equivalent input source. In the case of, say, a resistor it is the actual temperature of the device where its noise is defined as (5) Vs = 4*k *Ts * B * Rs Where Ts is the Noise Temperature, Rs is the resistance, k is Boltzman’s Constant
(1.38*10−23 Joules/Kelvin) and B is the bandwidth. If the device is something other than
Rnoiseless
Rs @ Ts = Vs = 4* k *Ts * B * Rs
Figure 4. Equivalent Input Noise Source
a resistor, such as an amplifier or antenna, Vs just represents an equivalent input noise source and Rs is its output impedance as shown in Figure 4. In most cases it is assumed the output impedance Rs is matched to the input impedance Rin of the next stage. Many times it is also assumed the bandwidth B is 1 Hz or, rather, Vs is in “Volts per root Hertz”. For example if an amplifier with NF of 3 dB is in a series chain such as depicted in Figure 3 and it is preceded by circuitry that has an equivalent noise temperature of 290 K the SNR after the amplifier will be 3 dB less than it was at its input per the definition of NF. However, if the amplifier is preceded by an equivalent input noise temperature that is something greater than 290 K the same amplifier would not degrade the SNR by the same 3 dB but would be somewhat less. In the case of the amplifier being far down the chain where it’s preceded by a lot of gain the amplifier may not affect the SNR at all due to the higher equivalent input noise temperature. This shows up in the Friis formula
10 for the amplifiers down the chain where their individual noise factors are diminished by the preceding gain. The opposite is true for equivalent noise temperatures less than 290 K. In those cases the amplifier could very well degrade the SNR more than its listed NF. An extreme example of this is in satellite reception. Here the input antennas are pointing to areas in space that are very quiet in terms of noise so their equivalent input noise temperatures are very low. Here an amplifier with a NF of 3 dB could degrade the SNR more than its defined NF depending on how low the noise temperature is at its input. This is the reason most satellite low noise amplifiers (LNA) have noise figures of 0.3 dB or less and are often kept in cool environments. Linearity A set of related parameters that affect dynamic range are Intermodulation Distortion (IMD), 3rd Order Intercept Point (IP3) and Compression. These interrelated parameters help define the linearity of the receiver. With larger input signals most devices exhibit some form of non-linear behavior. Usually mixers, which are highly non- linear by design, and amplifiers create the most non-linearity in a system. Filters show very little non-linear behavior but when they do it is usually from some sort of overdrive on the filter inductors. Crystal filters also have a bad habit of generating non-linear issues that are often not similar to mixers or amplifiers. Non-linear behavior will create unwanted tones at the output of the device. If you apply a 1 GHz signal to an amplifier what is wanted at the output is the same 1 GHz tone with more power. However, the non-linear performance will cause unwanted distortion that shows up in the form of tones at frequencies other than the 1 GHz input tone. For most systems 2nd and 3rd order distortion is sufficient to characterize the non- linear behavior of a device or system. The reasoning behind this is that most non-linear behavior in amplifiers and mixers are “weakly non-linear” and hence most of the distortion will occur in the lower order distortion. In addition, most distortion is inversely proportional to its order hence 3rd order distortion will be greater than 4th order and so on. Wide band systems generally use 2nd and 3rd order distortion characterization but as will be shown if the system is considered narrow band 3rd order will be enough to characterize the system. 11
Intermodulation Distortion One parameter that is often used to define linearity is Intermodulation Distortion (IMD). An ideal amplifier, for example, will have a transfer function (6) Vout ()t = AV* in where A is the gain. In reality this is really only a small signal response. Most real amplifiers will exhibit weakly non-linear behavior and will have an output equal to 2 3 (7) Vout ()t = AV* in () t + BV* in ()t + CV* in ()t + ... where A, B and C are coefficients. So if we drive the device with one sinusoid that is of an appreciable magnitude (8) Vtin ( ) = K *cos( w1 *t ) the output will consist of frequencies generated from the first, second and third order terms (9) Vout 1 ( t ) = AK* *cos( w1 * t )
2 (10) 2 2 BK* V ( t ) = B * K *cos ( wt* ) = * [1 + cos(2* w * t ) ] out 2 1 2 1 CK* 3 (11) V ( t ) = * [ 3*cos( w * t ) + cos(3* w * t ) ] out 3 4 1 1
nd rd where Vout 1 is the 1st order output, Vout 2 is the 2 order output and Vout 3 is the 3 order. As you can see the outputs are either just the fundamental output (the actual wanted output), a 2nd harmonic or a 3rd harmonic. These outputs primarily show up in wider band systems so if an input tone is applied in the X Band region any distortion products from the front end LNA and/or 1st mixer will come out well above the input band. A 2nd order harmonic will end up in the 18 GHz range and a 3rd order even higher. Most of these will be filtered by the front end X Band filter. However, if we apply two input tones (assume they are the same magnitude for simplicity) to simulate several return signals or other spurious input to the front end we could have (12) Vout 1 (t ) = A*[K *cos( w1 *t ) + K *cos( w2 *t ) ]
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K 2 *cos(2* w *t ) K 2 *cos(2* w *t ) (13) K 2 + 1 + 2 +... Vout 2 ()t = B * 2 2 2 2 ...K *cos [ ( w1 + w2 )* t]+ K *cos [ ( w1 − w2 )* t]
3* K 3 3* K 3 Vout 3 (t ) =... + *cos [ (2* w1 + w 2 )* t]+ *cos [ (2* w1 − w 2 )* t]+... 4 4
3* K 3 3* K 3 (14) *cos[ (2* w2 + w1)*t ]+ *cos[ (2* w2 − w1)*t ] 4 4
From these terms there are output frequencies similar to the situation where one
nd tone was input but there are some notable differences. In the 2 order output Vout 2 ()t you have the 2nd harmonics as before but there are also tones created at the sum and
difference frequencies w 1−w 2 and ww1+ 2 . In wide band systems these are an issue especially in direct conversion receivers.
rd st In the 3 order term Vout3 ()t there are frequencies generated such as more 1 and 3rd harmonics as in the one input tone case above. However there are 3rd order
frequencies created at 2ww1 + 2 , 2ww2 + 1 , 2w1−w 2 , 2ww2 − 1 (the sum and difference frequencies) which are known as the 3rd order intermodulation distortion or IMD. The sum tones are outside the band but the difference tones are usually within a very small percentage of the actual input tones. For example, if you use a two-tone input at, say 9300 MHz and 9400 MHz (100 MHz apart) the two difference 3rd order IMDs will be at (15) 2* f1 − f2 = 2*(9300) − 9400 = 9200MHz (16) 2* f2 −=f1 2* (9400) − 9300 = 9500 MHz
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These frequencies are in band and cannot be eliminated by filtering. These are the classic problem frequencies for typical narrow band applications. For a visual of IMD see Figure 5.
Pout Amplitude
3rd Order IMD Pout,3rd
2f1-f2 f1 f2 2f2-f1 Freq(MHz) 9400 9300 9500 9200 Figure 5. Intermodulation Distortion
The magnitude of the tones can be predicted and, more importantly, they can be shown to predictably increase at a rate depending on the increase of the input tones. 3rd Order Intercept Point For most amplifiers and mixers this predictable nature gives rise to a parameter based on IMD called the 3rd Order Intercept Point (IP3). In this case it’s the 3rd Order intercept point and can be defined at the input or output (IIP3 or OIP3). In most cases an amplifier will be specified with an output intercept and a device with loss, such as a mixer, will be specified with an input intercept point.
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40 I 7 I c.---- I IMD Slope (3:1) 30 / OIP3 I / ----- / / 20 / V I / I / / I / - 10 1..- / Actual Gain Curve ) / v / k' 0 - , i.---- dBm 1/ i....- Gain Slope (1:1) - / ,/ ,/ivf -10 ' / V I/ ,,y 3rd Order IMD Curve
// -20 I/' If ------J -30 / I Power Output ( Power V / 7 -40 V / -50 I / IIP3 I / -60 V ll I ✓ -70 -70 -60 -50 -40 -30 -20 -10 0 10 20 30 40 Power Input (dBm)
Figure 6. Input and Output 3rd Order Intercept Point
A graph which shows a typical input/output characteristic of an amplifier is shown in Figure 6. The intercept point is defined at the point where an imaginary straight line continuation of the actual gain curve and the actual IMD curve meet. The OIP3 is shown on the output y-axis and IIP3 on the input x-axis. It turns out that as you add 1 dB to the fundamental input tones the 3rd order IMD tones will rise 3 dB at the same time. Theoretically you could raise the input such that the IMD levels will gradually get closer and closer to the level of the input tones. However, as you can see the actual gain curves flatten out as the input tones are increased since the amplifier cannot supply any more output power at this point. This is the “strongly non-linear” region of an amplifier and the earlier polynomials for the weak non-linear case do not apply. Even though the intercept points are never reached by a real amplifier the intercept point parameter is used as a mathematical tool that helps describe the non-linear behavior of devices.
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The OIP3 can be measured by using a two-tone input and measuring the levels of
rd the 3 Order IMD tones. Since the IMD ( Pout3rd ) levels move up 3 dB for every 1 dB increase in the fundamental tones the difference between the IMD level and the fundamental tones is (17) ∆ OIP 3 = Pout − Pout 3 rd Then the output intercept point can be calculated using (18) ∆OIP 3 OIP 3 = P + out 2 Figure 7 and Figure 8 show intermodulation as tones and a typical measurement graph respectively.
OIP3 ··· 1·················· ··················· ∆OIP 3 OIP 3 = Pout + 2 Pout
∆OIP 3 = Pout − P out3rd Amplitude
Pout3rd 3rd Order IMD
2f1-f2 f1 f2 2f2-f1 Freq(MHz)
Figure 7. Intermodulation Tones
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There is also a similar equation to predict the overall intercept point much like the Friis formula for NF. It can be viewed as an inverse to NF in that it shows that the higher linear devices should be placed near the end of the amplifier chain rather than at the input as with NF. This is due to the large amount of gain earlier in the chain forcing larger and larger signals toward the backend of the receiver which can potentially increase the non- linear behavior. The output intercept for a chain of devices is
1 1 1 1 (19) = + ++... OIP( 3total G2 *G 3*G 4 ...* Gn −1 )* OIP 31 (G 3*G 4 ...* Gn −1)* OIP 32 OIP 3n
where Gn are the individual linear gains of each device and OIP 3n are the individual intercept points for each device. It should be noted that to do this calculation the linear (not dB) values should be used just like in the Friis formula earlier. This formula shows that the amplifier chain intercept point is affected most by the
40
30
20 Actual Gain Curve
10 ) 0 3rd Order IMD Curve dBm
-10
-20
-30 Power Output ( Power
-40 ∆OIP 3 = Pout − Pout3rd
-50
-60
-70 -70 -60 -50 -40 -30 -20 -10 0 10 20 30 40 Power Input (dBm)
Figure 8. Graphical Representation of IMD Measurement last device in the chain. If you notice the formula is very much similar to a number of 17 resistors in parallel where the smallest resistor has the most effect on the overall resistance. With the OIP3 equation the last term OIP 3n is not multiplied by any previous gain so it stands to reason that given the inversion it will have the most effect on the overall OIP3 of the chain. It should also be pointed out that since there is a lot of gain in the chain before the last amplifier it will also have a higher likelihood it will have a stronger signal at its input hence it will more likely be pressed into its own non-linear region. Compression Compression is another parameter use to describe non-linear behavior. It does not readily translate to formulas such as OIP3 does but it has its uses. If an amplifier or mixer get driven by a large signal source beyond its ability to respond at its output the device is considered to be in compression. This is where the device is driven into a region noted earlier called “strongly non-linear”. The intercept point equations no longer apply. This is shown in Figure 9. Note that the amplifier is stuck at a point which it can no longer generate any more power no matter how hard one drives the input.
10
8
6 Compressed Output
) 4 1dB Compression Point
dBm 2
0 Output down 1 dB
-2 Power Output ( Power -4
-6
-8
-10 -70 -60 -50 -40 -30 -20 -10 0 10 20 30 40 Power Input (dBm)
Figure 9. 1dB Compression Point 18
Another words the device is saturated. When the actual output is 1dB below the straight line continuation of the gain curve you are at the 1dB compression point. Although devices are not used in the saturated region it could happen in cases where a large signal is applied to the front end of the receiver such as when the transmitter output leaks back into the input. The compression point of a device can then be used as a safe guard against higher output power being applied to the next stage down the line. As long as the device’s input is not beyond the point where it could be damaged it can be used in a limiting protection capacity. This will be discussed further in the Chapter 4 section on the Limiter. Mixer Products The MxN mixer products are the harmonics created by any mixer from both the RF and LO inputs that get translated to the IF output and are considered unwanted spurious tones that are internally created. A general equation for the mixer output frequencies is M * Frf ± N * Flo (20)
Where M=1 and N=1 represent the first order output of the mixer. The magnitude of these spurious tones are often given by the mixer IC manufacturers in tables included in their data sheets. Also, testing can be done to find the actual output levels of the mixer products for a user’s specific needs. The 1st IF frequency of 1120 MHz was ultimately chosen such that a minimal number of MxN frequencies from the first mixer would not appear within the IF band or if they did they would be suppressed to a level below the dynamic range specification of 80 dBc. These need to be kept to a minimum since they cannot be eliminated by any filtering. The magnitudes of the various tones were known after extensive tests using the mixer’s own evaluation board. Spurious Free Dynamic Range The Spurious Free Dynamic Range (SFDR) is usually defined from both the intercept point and the noise figure. If you apply an input of two tones and set the levels such that the 3rd order IMD levels that occur are equal to the noise floor of the receiver
19 the value difference between the power of the two tones and the noise floor is the SFDR. The noise floor, which is depended on NF, at the output is defined as
Snf =10*log10( kTs ) + NF + 10*log10( B ) + Gt (21) where k is Boltzman’s constant, Ts is the system noise temperature, B is the bandwidth and Gt is the gain of the receiver. The SFDR (all variables in dB) is then defined as
SFDR = Pout − S nf (22) So if from (18) we have ∆ 3 P (23) OIP 3 = P +OIP 3 = P + (P − P ) = P − out3rd out 2 out out out3rd 2 out 2
Then if you let Pout 3rd = S nf OIP3 becomes
3 S (24) OIP 3 = P − nf 2 out 2 After this SFDR becomes 2 (25) SFDR = P − S = (OIP 3 − S ) out nf 3 nf So the SFDR at the output can be calculated from the output intercept point and the noise floor. As an example, if our OIP3 is designed to +32 dBm and the NF is 8 dB we will get S =10*log10( kT ) + NF +10*log10( B ) + G = (26) nf s −174 ++ 8 10*log10(30 MHz ) + 40 = −51.2dBm which gives an SFDR of 2 2 (27) SFDR = (OIP 3 − S ) = (32 −−( 51)) =55dB 3 nf 3 This just happens to meet our specification at 55 dB. Image Rejection In most receivers there are filters that are often placed at the input which eliminates any unwanted noise or spurious tones that are not part of what is needed to be processed. These front end filters also provide super-heterodyne receivers with image rejection. An image is an unwanted frequency (or band of frequencies) which end up in the IF band after a down conversion from a mixer. Mixer outputs consist of both the sum 20 and difference frequencies of the RF and LO signals. The first order (M=N=1) mixing frequency equation is
Fif = Frf ± Flo (28)
However, given the “absolute value” nature of the mixing process (specifically the negative sign) there are two bands which can be down converted to a single IF band. Depending on whether the LO frequencies are above (high-side injection) or below (low- side injection as in our case) the RF frequencies, there is at least one range of frequencies that are down converted from an unwanted RF band to the 1st IF. Since we are using a low-side injected LO the image is defined as Fim = Frf − 2* Fif (29) For example, if we had a tone at 9300 MHz we would down convert it using an LO frequency of 8180 MHz which would put it in the 1st IF range at 1120 MHz. At the same time if there had been a tone at, say, 7060 MHz the difference frequency also would appear at 1120 MHz (due to the absolute value). This would be direct interference with the wanted tone at 9300 MHz. In our case the entire frequency range from 7060 MHz to 7760 MHz will be down converted directly to the 1st IF of 1120 MHz as the 1st LO is varied across its band. It is certainly possible there may be no tones in this band but there will always be noise. So if this is not filtered adequately at the very least there will be a 3 dB rise in the noise floor without an image reject filter. So it is necessary for the front
21
1120MHz 1120MHz 1120 9650 7410 ,. ~ LO1 IF ' Image'1 RF I I Input I band I Amplitude I band
Freq(MHz) 9300 7060 7760 8880 1105 10000 1135
8180 - Figure 10. Image Frequency Range
end filters not only to filter tones outside the required input range but also eliminate any possible image frequencies. See Figure 10.
22
CHAPTER 3 Design Approach Currently several vendors have full digital transceivers that include down conversion, adjustable analog filters, ADC/DAC and some FIR filtering availability all on one IC. The Analog Devices AD9361 is such an IC. In fact, it has TWO channels of both a receiver and transmitter. It employs only one down conversion before the digital domain and has 12 bits of resolution. However, this particular solution did not meet our requirements due to the input frequency range (X band) and the ADC requirement of 16 bits. Eventually 16 bits at these frequencies will be achieved at some point in the future but not as of yet at the time of this writing. There are several possible architectures used historically in radio receiver design. In recent years direct down conversion (Zero-IF) due to its simplicity has somewhat overtaken the super-heterodyne in application deployment. This is especially true in RFIC designs of consumer products such as cell phones. Direct down conversion can be used but DC offsets due to LO leakage are a major issue with direct down conversion. This combined with analog input bandwidth constraints on our ADC made a direct down conversion receiver not achievable for our work. The super-heterodyne architecture has been used since the early 20th century. It uses a simple filter/amplifier intermediate frequency (IF) stages to do the majority of the filter processing after a mixer down conversion to a single frequency range. This approach fit to our requirements far better than the direct conversion method. Our choice then was the super-heterodyne receiver. The main reason for this choice was the fact the design required 16 bits from the ADC and the fact most ADC’s do not have an input bandwidth in the X Band range. Input capacitance limits the bandwidth and IC technology is not at a point where frequencies at 10 GHz are achievable. Therefore at least one down conversion to an intermediate frequency (IF) was needed to get the signal levels to a reasonable range that can be processed by an ADC. We then could have just one down conversion but the image rejection at X Band becomes difficult if the IF band is at too low a frequency. However, at higher IF 23 frequencies the choice of ADC with 16 bits would be very limited as well. This type of choice is a common trade-off for the IF frequency so there is a compromise. In our case since there are 16 bit ADCs readily available in the 100 MHz range for both sampling rate and input bandwidth an architecture using a second down conversion was added. This double down conversion is a common type of super heterodyne receiver and has been in use for many years. As noted earlier our approach also took into account that beamforming was to be accomplished digitally but due to specification and cost constraints it was decided that we would only design the receiver portion and it would be only the front-end analog section (including the second down conversion). The output of this receiver would be analog and drive a separate ADC board that converted the signal into the digital domain for processing. Description Our choice of a super-heterodyne receiver will include three sections: X Band RF section from 9300 MHz to 10000 MHz, the 1st IF section at 1120 MHz and the baseband section at 90 MHz as shown earlier in the block diagram. The RF front end has an input requirement from 9300 MHz to 10000 MHz. Although still considered narrow band (less than one octave percent bandwidth) the information bandwidth is only about 30 MHz so one can think of it as a series of 30 MHz bands across the input range. It should be noted there are no set bands at specific frequencies. The radar could use, for example, a chirp signal across any 30 MHz band within the 9300 MHz to 10000 MHz range. The local oscillator design (not discussed here) would have the ability to perform this over the entire RF band. So for our purposes and due to the choice of 1st IF the 1st LO frequency range was set from 8180 MHz to 8880 MHz. This accommodates the choice of the 1st IF at 1120 MHz. The RF front end consists of two LNA amplifiers to give the input signals gain and keep the Noise Figure (NF) at a reasonably low level. Two adjustable attenuators are also used to give a selection range of about 30 dB. This gain adjustment is used not only to keep the input power from overwhelming the input amplifiers it, more importantly, extends the dynamic range of the receiver.
24
It turns out that if a larger signal is input to the receiver the attenuator can be set such that it will lower the chance of potential non-linear behavior farther down the chain. Notice, however, that if the attenuation is set to a higher value the noise figure will also increase. One would think in that case there isn’t any way the dynamic range could be improved but it turns out that since the gain is also lowered the noise floor stays the same. Since the attenuators are near the input of the receiver the changes do not affect the OIP3 substantially so the receiver will have the same dynamic range even with the larger input power level. This is the main reason for the inclusion of the X Band attenuators near the front end of the receiver. There are also two preselection RF filters which attenuate all signals outside the input frequency band of 9300 MHz to 10000 MHz. These filters also serve as image rejection filters since the chosen 1st IF frequency at 1120 MHz guaranteed the image band was far enough outside the input frequency band that the RF filter can achieve both RF frequency selection and image rejection. The 1st IF section includes one amplifier and two bandpass filters. The filters are used to filter out most if not all of any mixer products which are normally the source of internal spurious tones. These filters also do a preliminary bandwidth selection of the overall final bandwidth of 30 MHz and eliminates any kind of LO or RF leakage through the mixer. In addition the filters also provide image rejection for the 2nd IF. The actual 1st IF frequency was selected after an exhaustive search to find the frequency bands that would have the least number of possible spurs. 1120 MHz was chosen as our 1st IF. The 2nd IF section at 90 MHz was primarily chosen for the Analog to Digital converter. The ADC met the 16 bit requirement and 90 MHz is at the optimized input band of the converter. At 120 MHz sampling and a bandwidth of 30 MHz, 90 MHz fits directly into the 2nd Nyquist zone for sampling. A 90 MHz bandpass filter is used as the anti-aliasing filter. Design Simulation and Compliance The design was simulated on an Excel spreadsheet originating at RF Café [9]. However it was extensively edited to include filter responses, mixer MxN product search and other frequency dependent factors that were programmed in using Visual Basic. It includes datasheet information from each of the devices and in the case of each of the 25 filters actual data was used. The cumulative G, NF, OIP3 and SFDR are all calculated at each device throughout the receiver. In Figure 12 the output values are the last ones in each column. Based on the spreadsheet the simulated compliance matrix is shown in Figure 11.
Requirements Specification Units Simulated Data RF Input Frequency Input 9300-10000 MHz 9000-10300 Power Input Max, No Damage 37 dBm -10 RF input level -30 dBm -30 LO Input First LO Frequency 8180-8880 MHz 8180-8880 2nd LO Frequency 1030 MHz 1030 Output and ADC Requirements Output Frequency 90 MHz 90 Bandwidth 30 MHz 35 ADC Sample Rate 120 MSPS 120 ADC Bits 16 Bits 16 ADC Aliasing Rejection > 60 Bits > 60 Digital Step Attenuator Attenuator Range 30 dB dB 30 Attenuator Step Size < 1 dB 1 Dynamic Range Specifications Noise Floor (output) < -50 dBm -51 Noise Figure < 8 dB 8.3 Image Suppression > 60 dB 61 Spurious Free Dynamic Range > 55 dBc 55.2 Figure 11. Simulated Compliance Matrix
The input power specification that did not meet our goals is shown in yellow. This issue will be discussed in the next chapter. The complete spreadsheet is shown in Figure 12.
26
Nom 55.19 55.19 56.60 56.60 56.60 56.60 55.19 55.19 56.60 56.60 57.09 57.09 57.09 57.09 57.09 57.09 57.10 57.10 57.45 57.45 57.10 57.46 57.46 48.36 48.36 48.37 48.37 50.00 50.00 50.00 50.00 50.01 50.01 59.55 59.55 50.01 59.57 59.57 75.91 75.91 75.54 73.98 59.63 719.90 719.90 SFDR3 (dB) Nom -50.92 -50.92 -67.92 -67.92 -67.92 -67.92 -46.92 -46.92 -67.04 -67.04 -68.07 -68.07 -67.83 -67.83 -67.83 -67.83 -65.33 -65.33 -82.36 -82.36 -58.33 -81.38 -81.38 -65.24 -65.24 -63.25 -63.25 -56.27 -56.27 -54.27 -54.27 -52.18 -52.18 -68.65 -68.65 -50.18 -66.68 -66.68 -83.86 -83.86 -83.86 -85.40 -85.40 -64.50 P[n] (dBm ) (dBm P[n] NBW (MHz) 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 30.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 700.000 999.000 999.000 700.000 700.000 999.000 999.000 700.000 9.88 9.88 2.50 2.50 Nom -7.12 -7.12 -7.12 -7.12 -6.24 -6.24 -7.24 -7.24 -7.00 -7.00 -7.00 -7.00 -4.50 -4.50 -9.00 -9.00 -7.00 -7.00 -4.90 -4.90 -2.90 -2.90 13.88 13.88 13.88 -21.50 -21.50 -20.50 -20.50 -18.00 -18.00 -16.00 -16.00 -21.30 -21.30 -19.30 -19.30 -31.20 -31.20 -35.50 -17.10 -30.00 -30.00 -33.30 P[s ig] (dBm ) (dBm ig] P[s Cumulative Cumulative 3.81 3.81 4.81 4.81 7.31 7.31 9.31 9.31 Nom 31.86 31.86 16.97 16.97 16.97 16.97 35.86 35.86 35.86 17.58 17.58 17.85 17.85 17.82 17.82 17.82 17.82 20.32 20.32 27.32 27.32 18.73 18.73 20.73 20.73 22.83 22.83 24.83 24.83 20.68 20.68 22.68 22.68 30.00 30.00 25.57 24.95 24.95 27.90 27.90 995.99 995.99 OIP3 (dBm) 8.28 8.28 8.28 8.28 8.28 8.28 8.28 8.28 8.28 8.26 8.26 8.28 8.28 8.26 8.26 8.26 8.26 8.26 8.26 8.26 8.26 8.22 8.22 8.20 8.20 8.17 8.17 8.15 8.15 8.13 8.13 8.13 8.13 8.13 8.13 8.13 8.13 8.06 8.06 8.02 8.02 1.20 1.20 5.50 8.00 8.00 3.30 3.30 0.00 0.00 Nom NF (dB) 8.50 8.50 9.50 9.50 8.70 8.70 Nom 0.00 0.00 -1.20 -1.20 -5.50 -3.30 -3.30 39.88 39.88 22.88 22.88 22.88 22.88 43.88 43.88 43.88 22.76 22.76 23.76 23.76 23.00 23.00 23.00 23.00 25.50 25.50 32.50 32.50 12.00 12.00 14.00 14.00 21.00 21.00 23.00 23.00 25.10 25.10 27.10 27.10 10.70 10.70 12.90 12.90 Gain (dB) Gain t Ou 90.00 90.00 90.00 90.00 90.00 90.00 1120.00 1120.00 1120.00 1120.00 1120.00 1120.00 1120.00 1120.00 1120.00 9300.00 9300.00 9300.00 9300.00 9300.00 9300.00 9300.00 9300.00 9300.00 9300.00 9300.00 . Frq e (MHz) 30.00 30.00 30.00 30.00 30.00 30.00 (MHz) 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 700.00 700.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 700.00 700.00 999.00 999.00 EqNBW . 999.00 Nom 40.00 40.00 25.00 25.00 37.00 37.00 20.00 20.00 25.00 25.00 30.00 30.00 43.00 43.00 25.00 25.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 999.00 . IP3 (dBm) 4.00 4.00 0.88 0.88 0.00 0.00 2.20 2.20 0.00 0.24 0.24 9.00 9.00 2.50 2.50 0.00 0.00 7.00 7.00 1.00 1.00 1.40 2.50 2.50 2.00 2.00 7.00 7.00 2.00 2.00 2.10 2.10 2.00 2.00 2.00 2.00 2.50 1.20 1.20 2.10 2.20 2.20 2.50 2.50 Nom 0.00 0.00 . NF (dB) 0.00 0.00 0.00 0.00 1.00 1.00 0.00 0.00 Nom 0.00 0.00 -4.00 -4.00 -0.88 -0.88 -0.24 -0.24 -2.50 -2.50 -7.00 -7.00 -1.00 -1.00 -2.50 -2.50 -2.00 -2.00 -7.00 -7.00 -2.00 -2.00 -2.10 -2.10 -2.00 -2.00 -2.00 -2.00 -1.20 -1.20 -2.10 -2.20 -2.20 21.00 21.00 24.00 24.00 18.40 18.40 18.40 18.40 Gain (dB) (dB) Gain Designation Designation Fixed Pad Pad Fixed ADL5531 Amplifier Amplifier ADL5531 Pad Fixed - 90BPF MHz Transformer TC4-1W+ TC4-1W+ Transformer Mixer ADL5801 ADL5801 Mixer Gain Flatness Adjust Flatness Adjust Gain Transformer TC1-1-13M+ TC1-1-13M+ Transformer BPF - 1120 MHz 1120 MHz - BPF LNA TQP3M9009 TQP3M9009 LNA Pad Fixed Fixed Pad Pad Fixed BPF - 1120 MHz 1120 MHz - BPF Fixed Pad Pad Fixed Mixer HMC558LC3B HMC558LC3B Mixer Fixed Pad Pad Fixed BPF - 9.6 GHz BPF LNA HMC564LC4 HMC564LC4 LNA Pad Fixed Fixed Pad Pad Fixed Limiter TGL2201-SM TGL2201-SM Limiter - 9.6 GHz BPF HMC1019LP4E Atten Step HMC564LC4 LNA HMC1019LP4E Atten Step Antenna Antenna Figure 12. Overall Receiver Spreadsheet
27
CHAPTER 4
Components This chapter will describe each of the major components chosen for the receiver and the reasoning behind them. Figure 13 shows a photo of the receiver.
IF Section nd 2 IF Section st 1 RF Section
Figure 13. Receiver Board
28
Front End RF Section For the front-end section refer to Figure 14 for the block diagram.
Antenna Front End RF Section .•..•···· --- ·······•..•
-3 dB
Attenuator Limiter RF Image Variable Attenuator LNA Variable Attenuator Filter RF Mixer -3 dB % -3 dB To 1st IF Attenuator Attenuator Amp RF Image ' Filter ./ •••·•············································································································································································
LO 1
Figure 14. Front End Block Diagram
Limiter The first component in the chain is the limiter. In receivers input antenna is often located near its own transmitting antenna there is a possibility for transmitted energy to be fed back to the receiving antenna. This is a common concern since most receiver inputs tend to be designed for low power inputs and the front end of a receiver usually does not use ICs that normally handle a lot of power. Receivers are used to pick out low level signals that exist in a noisy environment. These signals are very low level, often in the pico-watt range. Therefore, the amplifiers and mixers used in this way usually are designed in such a way that it is not possible to handle large signal inputs. Therefore some sort of protection needs to be provided for receiver input circuitry. This is the purpose of the limiter. The limiter that was chosen was the TGL2201-SM from Triquint (now called Qorvo). It has the ability to take up to 37 dBm (continuous 5 Watts) in the frequency range from 2 GHz to 12 GHz which easily covered our X-Band range. The output of the limiter under these conditions will typically be no more than 18 dBm. So if the electronic
29 components near the front end of the receiver, such as the LNAs and mixers, can withstand this amount there should be no damage. It should be pointed out that one must take into account any gains the LNAs give into whether the next stage could be hit with a damaging signal. For example, if the LNA has 15 dB of gain and it is hit with an 18 dBm signal the next stage will have to withstand 33 dBm coming from the LNA output. However, the LNA also has an output compression point which limits its drive capability more than this value. So if the device has a 15 dB gain LNA but its output compression is 20 dBm the next stage only has to handle 20 dBm even with the 18 dBm input to the LNA from the limiter.
Gain (dB) P[sat] (dBm) P[max] (dBm) !!! P[sat] (dBm) P[sig] (dBm) Designation Nom 1 Nom 1 Nom 1 1Nom Cumul Nom Antenna 0.00 999.00 999.00 999.00 37.00 Limiter TGL2201-SM -1.20 18.00 37.00 S,D! 18.00 18.00 BPF - 9.6 GHz -2.10 999.00 999.00 15.90 15.90 Step Atten HMC1019LP4E -2.20 24.00 27.00 13.70 13.70 LNA HMC564LC4 18.40 14.00 5.00 S,D! 14.00 14.00 Step Atten HMC1019LP4E -2.20 24.00 27.00 11.80 11.80 Fixed Pad -2.00 999.00 999.00 9.80 9.80 LNA HMC564LC4 18.40 14.00 5.00 S,D! 14.00 14.00 Fixed Pad -2.00 999.00 999.00 12.00 12.00 BPF - 9.6 GHz -2.10 999.00 999.00 9.90 9.90 Fixed Pad -2.00 999.00 999.00 7.90 7.90 Mixer HMC558LC3B -7.00 5.00 25.00 0.90 0.90 Figure 15. Front End Limiter Issue
Figure 15 shows part of a gain cascade spreadsheet. Note that in the column (!!!) the spreadsheet warns that the input power to the device in that row has more than allowed for damage. Both LNAs have this issue and will be discussed further in the LNA section below. The limiter has an insertion loss of 1.2 dB. Given the limiter is located at the very front of the receiver (before any gain from the LNA) this 1.2 dB loss will add directly to the overall receiver NF. This turns out to be a design compromise due to the necessity of the limiter. First RF Preselect and Image Filter The next component in the chain is the first of the two RF filters. These filters are used to preselect the specified input band from 9300 MHz to 10 GHz. This eliminates
30 any tones that might be picked up by the receiving antenna that is outside the required frequency range. The filter is also used for image rejection as discussed in Chapter 3. The design uses a low side injected LO (LO below RF band) so if an RF input of 9600 MHz is being tuned in using an LO of 8480 MHz (output IF at 1120 MHz) a possible image frequency at Fim = Frf − 2* Fif (30) which would be 7360 MHz. Therefore, any tone located at 7360 MHz will pass directly through the radio right along with the wanted frequency at 9600 MHz. However, since the filter is fixed so as to pass the band from 9300 MHz to 10 GHz the worst case image frequency will be at 10000 − 2*1120 = 7760 MHz (31)
This is the worst case since it is this image that is located closest to the lower passband edge of the filter at 9300 MHz. See Figure 16.
RF Filter Shape
60 dBc Image Rejection 9650 .,.. '-7410 ( RFin ' I image RFin Amplitude
Freq(MHz) 9300 7060 7760 10000 Figure 16. RF Preselect Filter Image Reject Specification
This puts a burden on the RF filter design since the filter has to roll-off enough between 9300 MHz (the lower end of the RF frequency input) and 7760 MHz (highest image frequency) so as to suppress the image to meet the specification of 60 dBc.
31
The chosen technology for the RF filter was an edge-coupled bandpass filter as shown in Figure 17.
Figure 17. RF Preselect Edge-Coupled Bandpass Filter
Here is yet another contradictory design situation. Our bandpass on this filter needed to be wider than necessary since most filters will give a higher insertion loss if the pass band is smaller relative to the center frequency. Normally this might not be a problem except we need at least one of the filters to be located before the LNA and its associated gain. Therefore any loss from the filter will directly add to NF of the whole receiver just like the limiter. On the other hand with a wider pass band the lower end for the range will come closer to the image frequency and make it more difficult to roll off the filter to attenuate the image. Hence there existed a design choice between the two specifications but we not only met the 60 dBc image suppression specification we met a goal of 72 dBc that was asked for by the customer. This was the reason that two filters were needed to make up for the fact one filter would not have meet the image rejection specification by itself. The passband attenuation ended up at 2 dB so this 1st RF filter added 2 dB of NF directly to the overall system noise. First Variable Attenuator The next device is the first of two variable attenuators. The chosen part is an Analog Device (Hittite) HMC1019. The important specifications for the variable attenuator are the attenuation range of 15 dB and step size of 0.5 dB, insertion loss of 2.2 dB in the lowest attenuation setting, input intercept point of 40 dB, maximum power input of 27 dBm and it uses Serial peripheral Interface (SPI) for control.
32
The requirements called out for the ability of the receiver to vary its gain by 30 dB in 0.5 dB steps. The HMC1019 has 16 dB total at 0.5 dB so this dictated we needed two attenuators to cover the 30 dB in range. It was also required that at least one of the attenuators be located before the first LNA. This, much like the limiter, made a compromise in the overall NF of the receiver. At the lowest attenuation state it is assumed this is where the overall system needs the most gain. The noise then would need to be low due to the need for a better signal to noise ratio. However, the insertion loss in the lowest attenuation state of the HMC1019 will be the setting that affects the NF the most. Since, as stated earlier, the NF of a device with loss is equal to its insertion loss the HMC1019 will add directly 2.2 dB to the NF to the overall system since its insertion loss in the lowest attenuation state is 2.2 dB The device can handle 27 dBm of power which is less than the maximum out from the limiter so the attenuator will not be damaged. The input intercept point of 40 dBm which far exceeds the necessary distortion levels for a device at the input of a receiver as noted earlier in Chapter 3. The second attenuator is located after the first LNA so any kind of degradation of NF due to the insertion loss setting is quelled somewhat from the gain of the LNA. The attenuators are controlled by an SPI (Serial Peripheral Interface) which uses a clock line, data input, data output and an enable line for each device. Since the data output from the attenuator is not used only four lines total will be used for the two attenuators: clock, data and an enable line for each attenuator. These lines ended up being a source of leakage in the receiver but that is beyond the scope of discussion here. First Low Noise Amplifier (LNA) The next device is the first of two low noise amplifiers (LNA). The LNA is used to give some gain at the input and keep the overall NF of the system as low as possible. There needs to be as much gain near the input to the receiver as possible before any losses. This is seen in the Friis formula as it states the best NF results when as much gain as possible is placed near the input to the receiver. In our case the limiter, the first RF filter and first variable attenuator had to be located before the LNA so their insertion loss (plus the NF of the LNA itself) added directly to the overall NF. This would lead to an overall NF of 8 dB for the receiver. However, if too much gain is used there is a strong 33 possibility there will be linearity issues farther down the chain. Therefore the design needed to be optimized to give the best overall SFDR as possible. The LNA of choice was the Analog Devices (Hittite) HMC564. This device gives about 17 dB of gain, a NF of 1.8 dB and an output intercept point of 25 dBm. Its output saturated power is about 14 dBm so this could protect any damaging power to the next stage. Unfortunately, the maximum input power before damage to the device itself was only 5 dBm. This would not work given that at this point in the receiver the limiter’s ability to keep its own output to 18 dBm was far too high to keep this LNA from damage. It turns out there was a miscommunication on the LNA’s specification data sheet which we missed during design. Later on there was a new version of the receiver proposed which not only took care of the maximum damage issue it also lowered the NF to 6 dB (instead of 8 dB) due to a replacement LNA that was found. Second Variable Attenuator The second variable attenuator is located just after the first LNA. As stated earlier the reason we used two was due to the requirement of 30 dB of attenuation adjustment but each HMC1019 only had 16 dB of range. The 2.2 dB loss here does not affect the overall NF as much since the device is located after the gain of the LNA. Second LNA For more gain in the front end stage another LNA amplifier was used (same HMC564). Since this amplifier is farther down the chain following the 1st LNA we could have used a different device with a higher NF and not really seen much of a difference in the overall system NF. Usually higher NF devices are less expensive so it was a consideration. Second RF Preselect and Image Filter The 2nd RF preselect filter is next in the chain. For us to meet the stop band attenuation for image rejection it was necessary for us to use two RF filters. Since this filter is farther down the chain with more gain behind it than the 1st filter the pass band attenuation of 2 dB is much less of a concern for this filter.
34
RF Mixer The last component for the front end RF chain is the RF mixer. The chosen device was the Hittite HMC558. The important specs are: 7 dB of loss, 15 dBm to 17 dBm LO input requirement, IIP3 of 20 dBm and a table was given for its M X N output products. Mixers cannot have compression points above the drive level of their LO ports. In fact, this device was about 12 dBm which is a bit lower than the LO drive level of 15 dBm. Since the compression level essentially dictates the intercept point we would either have to go with a lower compression level or use a larger LO drive signal. Most LO port will be driven with an amplifier at some point (not on our board) so either way power will be used. In our case we went with the low end of the LO drive at 15 dBm which is considered a moderate level as many mixers require 18 dBm or higher. We then designed the gains such that the mixer was not a critical point in the overall linearity so the 12 dBm compression (and hence the 20 dBm 3IIP) was plenty. Often the mixer turns out to be a linearity “bottle neck” due to its inherent non-linear behavior. Passive Attenuators Simple resistive attenuators were used throughout the front end. One might ask why attenuate the gain since it’s a scare quantity at X Band? The frequency range is the issue. At X Band component reflections are a great concern. The amplifiers usually have nearly 50 ohm inputs but over the wide X Band range it can vary a lot. This can create return loss issues and potential microwave modal problems. Return loss (RL) is related to the standing wave ratio Z − Z Γ= l s (32) Zl + Zs
=−Γ (33) RL ( dB ) 20*log10 (I I) Where Zi is the input impedance and Zs is the source impedance. Amplifiers can be a problem but filters and mixers are usually far worse. Mixers, for example, can have quite low input impedances both on the RF and LO ports. The RF port on the HMC558 has only about a 7 dB return loss. Since most passive mixers use
35
diodes which when driven by the LO port can have impedances in the 10 ohm range which is obviously well below a typical 50 ohm system. Manufacturers use transformers in the mixer to alleviate some of this but often it is not enough. We considered 10 dB an absolute minimum for return loss so for the mixer we added a 2 dB attenuator between the 2nd RF filter output and the mixer input. This gives directly 4 dB of return loss which when added to the 7 dB the mixer itself has gave us 11 dB of overall at that port. This same idea was used in a couple of places in the front end chain such as before the 2nd RF filter since its return loss was 9 dB at some points in the passband. Attenuation was not used at the first RF filter since the added loss at that point adds directly to the system NF and we might not have met the overall NF specification if an attenuator was used. 1st IF Section For the 1st IF section refer to Figure 18.
1st IF Section (--- _ ....--. ~ ······•...•••
From RF -3 dB % ._. -3 dB Attenuator -Attenuator IF Filter Amp
IF Mixer
-3 dB ][1--1 To 2nd IF Attenuator \ IF Filter Transformer ... I •··•························································································································· ···············
LO 2
Figure 18. 1st IF Section
IF Filter The first device in the 1st IF section is the first of the two IF filters. The primary role of this filter is to shape the information bandwidth of the receiver. The specifications call out for a 30 MHz bandwidth which this filter delivers although its pass band is a bit
36
wider. As mentioned earlier for most filters there is a compromise between bandwidth and the pass band insertion loss. The narrower that bandwidth the more loss in the pass band. So it was decided that a bit more bandwidth be given so as to limit the loss and leave the extra band shaping to the final 2nd IF filter in the next section of the receiver. Another use for the filter is to eliminate the image frequencies of the IF mixer. Figure 19 shows the 2nd IF image specification. As shown the IF filter needs to roll off about 30 dB from 1105 MHz to 955 MHz which when combined with the second IF filter in this section should give the 60 dB as called out in the specifications. It should be pointed out that tones that end up in the image band from 925-955 MHz will probably not originate in the 1st IF chain. They will originate as down converted signals from the front end. However, there is still the extra noise from the image band and that should be taken care of by the filter.
90MHz 90MHz 940 1120 .... ._ LO2 Base IF IF 'I image '• I Band I I Amplitude I I
Freq(MHz) 90 955 925 1105 1135 Figure 19. 2nd IF image specification
The frequency band of images from the front end are shown in Figure 20. This shows the band of frequencies that land in the 2nd IF image band from 925 MHz to 955 MHz from the RF front end. There is a range at the front end from a low end of 7225 MHz to a high end of 9835 MHz shown in red in Figure 20 that potentially end up in the 2nd IF image band. However, some of these frequencies are in the stop band of the RF filter between 7760 MHz and 9300 MHz so they will come under some attenuation. The worst case are the frequencies above 9300 MHz to 9835 MHz which cannot be filtered out by the RF filter (shown with pattern red). This is where the 1st IF filter image roll off comes into play. So even though the RF filter cannot reject this band the 1st IF filter will 37 create 30 dB of attenuation at 955 MHz (60 dB total for the two filters) which will guarantee the image rejection as not being an issue.
1120MHz 1120MHz 1120 9650 7410 LO1 IF Image RFin Band Amplitude
Second IF Image
Freq(MHz) 9300 7060 7760 8880 1105 10000 1135 8180 - 9835 7225 Figure 20. Band of frequencies from RF front end that create 2nd IF images
The chosen technology for the IF filter was a capacitive coupled quarter wave resonator bandpass filter shown in Figure 21.
Figure 21. 1st IF Capacitive Coupled Quarter Wave Resonator Filter
IF Amplifier The amplification in the 1st IF section is given by a Triquint TQP3M9009 amplifier. It has about 22 dB of gain at the 1st IF frequency of 1120 MHz and an OIP3 of +39 dBm. Its NF is 1.3 dB at 1120 MHz. Even though the very low NF is not really needed at this point in the receiver the device is relatively inexpensive due to its popularity.
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IF Mixer The last component in the 1st IF section is a transformer and the 1st IF mixer. The mixer is an Analog Device AD5801 which is a highly linear double balanced mixer which uses differential input signals. The need for converting from a single ended signal to differential (and vice versa) is the objective of the transformers. The double balanced mixer tactic is used to suppress many of the even harmonics and some MxN products that normally are output from a mixer. This approach eases the requirements for the anti- aliasing filter in the 2nd IF section as well. It should be noted here that there is also an output transformer that converts back from differential to single ended. This is technically part of the 2nd IF section since the output at that point is 90 MHz but it’s used as part of this mixer. 2nd IF Section For the 2nd IF section refer to Figure 22.
2nd IF Section
From 1st IF r- ][ -3 dB -3 dB % - . To ADC ==Attenuator~ ==Attenuator Transformer Amp Anti-Aliasing Filter ···...... •· ··· Figure 22. 2nd IF Section
Voltage Amplifier Out of the 1st IF mixer the next device is a voltage amplifier. The device chosen was an Analog Devices AD5531. This device a fixed gain of 20 dB and a high OIP3 of 40 dBm. The NF is 2.5 dB but that is not a real issue at this point in the receiver. This amplifier gives the final gain increase for the receiver and its high intercept point of 40
dBm is well beyond the full scale input of 2Vpp− for the ADC. One of the goals is to make certain there would be no IMD products that rise above the noise floor for any two-tone signals at the output up to the full scale input of the ADC. The SFDR specification lets the ADC to have the burden of the non-linearity
39 of the overall receiver. Any linearity issues that appear can be dealt with by possibly choosing a different ADC with little of any change to the receiver itself. Anti-Aliasing Filter and ADC Section Before any kind of analog to digital conversion can be applied an anti-aliasing filter needs to suppress any out of band signals. Once an analog signal has been converted to the digital domain there is no hope of eliminating any spurious signals that appear. The problem is that any signals that are located above half the sampling rate (Fs) will be down converted (much like what happens with a mixer) into the band that is below Fs/2. Once converted to the digital domain this cannot be filtered out by any means. Therefore the anti-aliasing filter was needed to apply this out of band attenuation. Under normal circumstances any signal greater than half the sampling rate needs to be attenuated by a filter to a specified level. In our case we are using a slightly different form of conversion. We are using a bandpass sampling scheme, often called under-sampling, so the filtering is a bandpass filter rather than low pass. Figure 23 shows how this works [10].
40
F2 = 120 MHz Sampling Rate NZ=1 NZ=2 NZ=3 NZ=4
+/- 15 MHz +/- 15 MHz +/- 15 MHz +/- 15 MHz Amplitude
Fc = 90 MHz Frequency Center Frequency IE 0.5 Fs 0.5 Fs 0.5 Fs 0.5 Fs
Figure 23. Bandpass Sampling of Signal
In choosing the ADC a suitable sampling rate was chosen that was high enough to avoid aliasing (Fs > 2 * BW). It was also chosen to be under sampled (bandpass sampling) so a Nyquist zone also needed to be chosen. For bandpass sampling the sampling rate Fs is 4F F = c (34) s 2( NZ −1) Where Fc is the center of the band and NZ is the chosen Nyquist zone. In our case we decided on Fc = 90 MHz and NZ = 2. This gives a sampling rate of Fs = 120 MHz. This also meets the requirement of Fs > 2*BW (60 MHz) and is less than the specified 125 MHz maximum sampling rate of our chosen ADC. The roll-off requirement for the filter needs to be sufficient to attenuate any tones that could alias into the processing band. The anti-aliasing requirement was given as 60 dB which is specified to be at 45 MHz. This is taken from the Nyquist rate (60 MHz = Fs/2) and drop 15 MHz from there which would be where an aliased passband would be at its highest end. See Figure 24. 41
2nd IF Filter Shape 60 dB Aliasing Rejection 90 MHz 90 MHz 30 MHz -- .L .... ( I Nyquist Nyquist I Band Amplitude Band I 1 I 2 I
Freq(MHz)
15 MHz 45 MHz 75 MHz 105 MHz 105 MHz 60 MHz Nyquist Rate
Figure 24. 2nd IF Filter Requirements
Figure 25. 2nd IF Lumped Component Bandpass Filter
The chosen technology for the 2nd IF filter was a lumped surface mount component bandpass filter using 0403 size inductors and capacitors as shown in Figure 25.
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CHAPTER 5 Test Results This chapter discusses the test results for verification. The filter responses for each of the three filters will also be shown along with tests for Noise Figure, Linearity, Mixer Products and Image Response. The Gain of the receiver will be given within each test as well as it naturally shows up with many of these tests. RF Filter The edge-coupled bandpass filter used in the front end section needed to preselect the RF band from 9300 MHz to 10000 MHz and preform the image rejection for the 1st RF mixer. The image response will be shown in a later section but the preselection is shown in Figure 26. The graph shows both the measured and simulated results of the insertion loss (filter attenuation) of one of the RF filters. Note the response at 7760 MHz. This was the worst case frequency that needed to be suppressed to meet the 60 dBc image specification. The simulation shows about 30 dB of roll off from the center frequency of the filter. The measured result is a bit better so if two of these filters are used in series and good grounding techniques are used this filter should work. As will be shown later
RF Filter Simulated and Measured 0
_I -5 9 GHz -1.122dB 9650 MHz 10.3 GHz -10 Center Frequency -1.387dB --co BW = 700 MHz l:J -15 t t --(/) (/) -20 -30 dBc @ t t 0 7760 MHz _J I -25 Simulation C 0 :p..... -30 Q) (/) / t -35 C Measurement -40 t t -45 +- +- -50 5 6 7 8 9 10 11 12 13 14 15 Frequency (GHz)
Figure 26. RF Filter Response (Measured and Simulation) 43 the image suppression was measured to 65 dBc so the image response specification was met. 1st IF Filter The capacitive coupled bandpass filter used in the 1st IF section needed to begin to shape the overall bandwidth of 30 MHz and provide image rejection for the 2nd IF mixer. Figure 27 shows the measurement and simulation for the 1st IF filter.
1st IF Filter Simulated and Measured 0 -5 Pass Band Loss -10 -2 dB -15 1120 MHz -20 Center Frequency -25 BW = 100 MHz -30 -50 dBc @ -35 955 MHz -40 -45 -50 -55 -60 -65
Insertion Loss (dB) -70 Simulation Measured -75 -80 -85 -90 900 950 1000 1050 1100 1150 1200 1250 Frequency (MHz) Figure 27. 1st IF Filter Response (Measured and Simulation)
The figure shows that the bandwidth for this filter is about 100 MHz and the roll off at 955 MHz, the more strenuous frequency specification for image rejection, was about 50 dBc. If two filters are used in series the image rejection will be more than needed. Note that the bandwidth is far more than the specified 30 MHz. This was designed in so as to minimize the insertion loss in the pass band. The loss is about 2 dB. However, if the bandwidth was made narrower the loss was have increased and another amplifier would probably have been needed to make up for the loss in gain since two
44
filters would be used in series. This larger bandwidth made it necessary that most of the information bandwidth shaping would be accomplished by the 2nd IF filter at 90 MHz. 2nd IF Filter The lumped component bandpass filter is used to shape the final information bandwidth to 30 MHz which also provides an anti-aliasing filtering for the ADC. Figure 28 shows the response. The roll-off of nearly 75 dBc at 60 MHz is more than adequate to meet the 60 dBc aliasing requirement.
90 MHz Anti-Aliasing Filter 0
-20 90 MHz Center Frequency 30 MHz BW
-40 > -60 dBc @ 45 MHz
-60
Insertion Loss (dB) Measured -80
-100 10 20 30 40 50 60 70 80 90 100 110 120 130 140 150 160 170 180 190 200 Frequency (MHz)
Figure 28. 2nd IF Lumped Component Filter Measurement
Gain The gain is measured simply by applying an input band frequency between 9300 MHz and 10000 MHz and measuring the level at the output. Figure 29 shows the gain over frequency. There was about a 3 dB flatness issue with the gain and it drops off at
45
the higher end of the range. The customer can work with this either with the variable attenuators or as an adjustment in the digital domain (equalization).
Gain vs Frequency 46.0
45.0
44.0
43.0 ~~
42.0 ...... ~ ...... - i'----. Gain(dB) 41.0 ~ 40.0
39.0
38.0
37.0 9200 9300 9400 9500 9600 9700 9800 9900 10000 10100 RF Input Frequency (MHz) Figure 29. Gain vs Frequency Noise Figure The noise figure measurement is a bit more involved due to lack of X Band equipment we had on hand. Fortunately, since the receiver itself does a down conversion the NF measurement could be made at the 90 MHz output. The graph is shown in Figure 30. The measurement was made at the receiver output but the LOs were set such that the noise input was measured at the frequencies shown. This is called a “spot” noise test and applies to a range around the frequency shown ± BW/2 ( ± 15 MHz).
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Noise Figure vs RF Input Frequency 7.00
6.80
6.60
6.40 ~ - -- 6.20 ----
6.00
Noise Figure (dB) 5.80
5.60
5.40
5.20
5.00 9200 9300 9400 9500 9600 9700 9800 9900 10000 10100 RF Input Frequency (MHz) Figure 30. Noise Figure vs. RF Input Frequency
The reason for the rise in NF over frequency has to do with the gain at the front end. As shown above the gain in the X Band section had a roll-off toward 10000 MHz. Since the NF is mostly affected by the gain near the front end of the receiver so it stands to reason the NF will rise as the gain drops. However, the noise figure change was only about 0.3 dB rise so it was not a concern. The 6.2 dB value for the NF was far better than the specified 8 dB. We had decided that our customer might want to use a receiver with less gain, perhaps to add some outside front end LNA to get an even better overall NF, so we would design in a lower NF to begin with. Linearity The linearity of the receiver is determined by three parameters: IMD, OIP3 and Compression. All three were tested and we came to the conclusion the receiver met the specifications.
47
One of the main goals was to make certain the receiver did not compress at all
with outputs up to the full scale value of the ADC. The ADC’s full scale is 2V pp− which into Rload = 50 ohm is
2 Prms = V rms / R load (35)
PdBm = 10*log10( Prms /1mW ) = 10dBm (36)
Figure 31 shows the point where an extended trend line based on the actual output power measured would be 1 dB above the actual output measured. This is the compression point and it is roughly +17 dBm at the receiver output. This meets the goal of +10 dBm.
Receiver 1dB Compression 20.0
18.0 +17 dBm 1 dB
1 dB below ideal gain
) 16.0 dBm Actual Output Power
14.0
Linear Gain Line Output Power ( 12.0
10.0
8.0 -34.0 -32.0 -30.0 -28.0 -26.0 -24.0 -22.0 -20.0 Input Power (dBm) Figure 31. Compression Point at Receiver Output
The IMD measurement is shown in Figure 32. Two tones are applied to the RF input anywhere within the input frequency range. The two tones down converted to the output is recorded as shown. The slope of this line in dB is the gain of the receiver. The 48
IMD levels at the output are also recorded and shown. Note the slope is about 3 dB/dB and shows that the IMD levels rise about 3 dB for every 1 dB the nominal inputs rise.
Output and IMD 10.0
~
0.0 Po = +5 dBm T -10.0 ~ Output -20.0 )
-30.0 dBm ∆= 73 dB -40.0
IMD ___. -50.0
~ ~ -60.0 I Output and IMD ( ______.. ~ -70.0 __:..---I
-80.0 ~
-90.0 -49.0 -47.0 -45.0 -43.0 -41.0 -39.0 -37.0 -35.0 Input Power (dBm) Figure 32. Output and IMD Curves
We then can use Equations 19 and 20 to find the OIP3. Figure 32 shows the difference between the output level of the two tones and the IMD is 73 dBc. So the OIP3 is calculated to be
OIP 3 = Pout +∆O IP 3 / 2 =−5 + 73 / 2 =+31.5 dBm (37) This shows that the OIP3 is +31.5 dBm and about 14 dB above the compression level at +17 dBm. This is typical as the OIP3 is usually somewhere from 10-12 dB above the compression level in most amplifiers or systems. More importantly we also met the goal of having the OIP3 being well above the full scale of the ADC.
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Mixer Products The mixer products proved to meet the specification of 55 dBc. The 2x2 mixer product of the 2nd IF mixer was by far the worst spurious response. It came in near its predicted value of -68 dBc at a 100 MHz output from the receiver. Other outputs measured -84 dBc for some higher order tones from the 1st IF mixer which came through the receiver at 110 MHz. All of these met specifications by a wide margin. However, with only a few samples tested it remains to be seen whether this could hold up in a production environment. Often this is a difficult issue since repeatability of the mixer data from its manufacturer is not always consistent. Image Response This test involves applying an input tone at some fixed amount at the RF input but set to a frequency in the image range from 7060 MHz to 7760 MHz. The 1st LO would then be adjusted as if the input were in the actual RF range. For example in the 7760 MHz case the 1st LO would be set to 8180 MHz since that would down convert an input at 9300 MHz of which 7760 MHz is an image. As stated earlier the worst case image rejection should come from an input frequency that is closest to the RF input range (9300 MHz to 10000 MHz) which in our case is 7760 MHz. We took data from one end of the image band to the other. Results are in Figure 33.
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Image Rejection 82.0
80.0 / -......
78.0 / ~ ) 75 dBc dBc "- 76.0 I "'-~ V ~
74.0 I 7760 MHz Worst Case Image Rejection ( 72.0 I
J I 70.0 I 68.0 7000 7100 7200 7300 7400 7500 7600 7700 7800 RF Input Frequency (MHz) Figure 33. Image Rejection
The results met the specification to 60 dBc by at least 9 dB near 7050 MHz. The worst case was supposed to be at 7760 MHz but it had a better rejection of 75 dBc. As to the 2nd IF image an RF input at the center of the band at 9600 MHz was applied and the 1st LO was adjusted such that the output from the 1st mixer was 955 MHz which is the potential worst case frequency as noted earlier. When this value is tested a signal at the output is rejected by over 90 dBc so the 2nd IF image will not be an issue in the receiver. Receiver Overall Specification Compliance As shown in the measured data compliance matrix of Figure 34 the input power requirement we did not meet is in yellow. The input power maximum was not tested for obvious reasons and as noted earlier the aliasing rejection was not going to pass the specification due to the lack of area on the board for another filter.
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Requirements Specification Units Measured Data RF Input Frequency Input 9300-10000 MHz 9000-10300 Power Input Max, No Damage 37 dBm - RF input level -30 dBm -28 LO Input First LO Frequency 8180-8880 MHz 8180-8880 2nd LO Frequency 1030 MHz 1030 Output and ADC Requirements Output Frequency 90 MHz 90 Bandwidth 30 MHz 35 ADC Sample Rate 120 MSPS 120 ADC Bits 16 Bits 16 ADC Aliasing Rejection > 60 Bits -75 Digital Step Attenuator Attenuator Range 30 dB dB 30 Attenuator Step Size < 1 dB 1 Dynamic Range Specifications Noise Floor (output) < -50 dBm -53 Noise Figure < 8 dB 6.5 Image Suppression > 60 dB 72 Spurious Free Dynamic Range > 55 dBc 56 Figure 34. Measured Data Compliance Matrix
Receiver Cost The cost for the receiver is shown in Figure 35. The figure shows that for the receiver alone a $600 cost is possible. In fact if you add an ADC and FPGA the cost will still be in the $3000 - $5000 range. If you use multiple channels with only one FPGA the “cost per channel” will be on the lower end of this range. This would depend on how many digital lines are used and the style of digital transmission (Low Voltage CMOS or other protocols). Our customer was very satisfied with these results given that some of their modules cost more than $50,000 (or more) per channel.
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QTY Component Unit Cost Total Cost Num Capacitor 0402 - $5 Num Inductor 0805CS Coilcraft - $5 Num Resistor 0201 - $5 2 BPF 9600 MHz $14 $28 2 BPF 1120 MHz $30 $60 1 BPF 90 MHz $20 $20 2 SMA-Vert Molex X-Band $3 $6 1 SMA-EndLauch J808 $7 $7 1 SMA-EndLauch J502 $3 $3 1 TC4-1W+ $2 $2 1 TC1-1-13M+ $2 $2 2 HMC564 $44 $88 2 HMC1019LP4E $120 $240 1 HMC558LC3B $50 $50 1 ADL5801 $11 $11 1 TQP3M9009 $5 $5 1 ADL5531 $5 $5 1 Housing $50 $50 1 PCB $100 $100 Total $692 Figure 35 Receiver Cost
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CHAPTER 6
Conclusion and Future Work For future work there are a number of improvements that need to be made. The maximum power input issue with the HMC564 LNAs being the most prominent. In redesign we planned on using what turned out to a better LNA for the receiver. In fact, it was an LNA that is commonly used in industry (Qorvo TGA2512) which had its own gain control. This meant we could eliminate the HMC564 and the first of the variable attenuators completely with one cheaper device. The TGA2512 also had a far higher tolerance to a large RF input at 21 dBm versus the 5 dBm of the HMC564. The down side is that we then needed to add a digital to analog converter since the TGA2512 gain control pin was analog and we only had the SPI serial input available to us. Another issue was the MxN mixer spurious outputs from the 1st RF mixer. We originally thought this was not an issue but a more thorough search found some problematic input frequencies that could in the future be an issue. So it was proposed that the 1st IF center frequency be moved from 1120 MHz to around 1300 MHz. This not only was a cleaner range for the spurious response it conveniently made the RF filter specification of image rejection more relaxed since the IF is at a higher frequency. All that would need to be done is a redesign of the 1st IF filter to a new center frequency near 1300 MHz. This is one of the advantages of a super-heterodyne receiver. It is very flexible where some components can be adjusted slightly without affecting other sections of the receiver. For example, many of the front end components are wide band the front end frequency range could be changed without too much redesign. Only the frequency dependent devices such as the filters would require redesign. One anti-aliasing 90 MHz filter was not enough to meet our goals. However, since time and money ran out we did not have time to implement another series filter. Even then the size of the module would have been prohibitive. Another solution for this needs to be addressed. Overall the design worked to our satisfaction and the customer has continued to engage with us for future work.
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REFERENCES
[1] 5G PPP Architecture Working Group, View on 5G Architecture (Version 2.0), July 18, 2017. [2] CHILL National RADAR Facility, http://www.chill.colostate.edu/w/CSU_CHILL [3] J.S.Herd, M.D. Conway, “The Evolution to Modern Phased Array Architectures,” Proceedings of the IEEE, Vol. 104, No. 3, March 2016, pp. 519-529. [4] Z. Li, et al. “Design considerations of the RF front-end for high dynamic range digital radar receivers,” International Conference on Microwaves, Radar, and Wireless Communications, Wroclaw, Poland, May 19-21, 2008. [5] S. Matah, L. Zenkouar, “A practical approach for RF system design of an S-band modern digital radar receiver” Mediterranean Microwave Symposium, Marrakech, Morocco, Dec. 12-14, 2014. [6] M. Yeary, et al. “Compact digital receiver development for radar based remote sensing,” IEEE Instrumentation and Measurement Technology Conference, Victoria BC, Canada, May 12-15, 2008. [7] K. Lauritzen, “High-dynamic range receivers for digital beam forming radar systems,” IEEE Radar Conference, Boston, MA, April 17-20, 2007. [8] Kevin McClaning and Tom Vito. Radio Receiver Design. Atlanta, GA: Noble Publishing Corporation, 2000. Print. [9] Kirt Blattenberger, RF Café, http://www.rfcafe.com [10] Walt Kester, “What the Nyquist Criterion Means to Your Sample Data System Design”, Analog Devices Tutorial MT-002 pp. 11.
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