A Low-Power Wireless Transceiver for Deeply Implanted Biomedical Devices

Home , Transceiver

A LOW-POWER WIRELESS TRANSCEIVER

FOR DEEPLY IMPLANTED BIOMEDICAL

DEVICES

STEVE MAJERUS

Submitted in partial fulfillment of the requirements

for the degree of Master of Science

Thesis Advisor: Dr. Steven L. Garverick

Department of Electrical Engineering and Computer Science

CASE WESTERN RESERVE UNIVERSITY

August, 2008 CASE WESTERN RESERVE UNIVERSITY

SCHOOL OF GRADUATE STUDIES

We hereby approve the thesis/dissertation of

______

candidate for the ______degree *.

(signed)______(chair of the committee)

______

(date) ______

*We also certify that written approval has been obtained for any proprietary material contained therein. Table of Contents

1 INTRODUCTION...... 1 1.1 Motivation...... 1 1.2 Background...... 2 1.2.1 Existing Transceivers...... 3 1.2.1.1 An RF-powered neural recording device: the Utah Array...... 3 1.2.1.2 Battery Powered Bion……………………………………..….5 1.2.1.3 A wide bandwidth, battery powered recorder: the SmartPill....7

2 BAND SELECTION AND DESIGN CONSIDERATIONS...... 9 2.1 Band Analysis...... 9 2.1.1 FCC Regulations...... 9 2.1.2 Power Considerations on Band Selection...... 11 2.1.2.1 Receiver Power Savings...... 13 2.1.2.2 Transmitter Power Savings…………………………………..18 2.2 Band Selection...... 19 2.3 Design Goals...... 19

3 ARCHITECTURE & DESIGN...... 22 3.1 Architecture...... 22 3.1.1 OOK Receiver……………………………………………………...23 3.1.1.1 Differential Amplifier………………………………………..25 3.1.1.2 Schmitt Trigger………………………………………………28 3.1.1.3 Activity Detector……………………………………………..32 3.1.2 Manchester Decoder……………………………………………….34 3.1.2.1 Method for Decoding………………………………….……..35 3.1.2.2 All-Digital Manchester Decoder……………………….…….37 3.1.3 FSK Transmitter……………………………………………………40 3.1.3.1 LC Tank Modeling…………………………………………...41 3.1.3.2 LC Oscillator…………………………………………………43 3.1.3.3 Digital Power Control………………………………………..46

4 TEST RESULTS...... 48 4.1 Test Setup...... 48 4.2 OOK Receiver...... 49 4.2.1 Fully-Differential Amplifier……………………………………….49 4.2.2 Schmitt Trigger…………………………………………………….51 4.2.3 Activity Detector…………………………………………………..54 4.2.4 Entire Receiver…………………………………………………….56 4.3 Manchester Decoder...... 58 4.4 FSK Transmitter...... 60

5 CONCLUSIONS AND FUTURE WORK...... 65

6 REFERENCES...... 68

i List of Tables

Table 1-1 Performance summary of the Utah Transceiver, as described in [2]...... 5 Table 1-2 Performance summary of the BPB in [3]...... 6 Table 1-3 Performance summary of the wireless endoscope in [6]...... 7 Table 2-1 Summary of the transceiver design specifications...... 21 Table 3-1 Transistor sizes and bias currents for the DA...... 27 Table 3-2 Transistor sizes used in the Schmitt Trigger...... 30 Table 3-3 Transistor sizes for the activity detector...... 34 Table 4-1 Comparison of designed and estimated actual bias currents and output resistance for the differential amplifier...... 50 Table 4-2 Measured Schmitt trigger thresholds...... 53 Table 5-1 Comparison of Measured Results to Design Goals...... 68

ii List of Figures

Figure 1-1 Block diagram of a multi-channel, dual-band implantable neural recorder (from [2])...... 4 Figure 1-2 Photograph of a packaged BPB (from [3])...... 6 Figure 1-3 Conceptual drawing of the SmartPill as it traverses the gastro- intestinal tract (from [5])...... 7 Figure 2-1 Table of FCC limits for maximum permissible exposure to electrical and magnetic fields at various frequencies (from [7])...... 10 Figure 2-2 Plot of the attenuation by distance versus frequency for electric fields propagating in pure water and seawater (after [3])...... 12 Figure 2-3 A two-stage differential amplifier (from [9])...... 14 Figure 2-4 The small-signal model for the output stage of the amplifier in Figure 1-5, used to determine the frequency response (after [9])...... 14 Figure 2-5 The typical frequency response of a bandpass filter, such as a parallel-resonant LC tank (from [10])...... 16 Figure 3-1 Simplified block diagram of the low-power transceiver...... 23 Figure 3-2 Block Diagram of the OOK Receiver...... 24 Figure 3-3 High-level circuit diagram of the OOK receiver...... 24 Figure 3-4 Binary approximation of an OOK-modulated signal...... 25 Figure 3-5 Circuit schematic of one of the DAs used in the OOK receiver...... 25 Figure 3-6 Half-circuit model of the DA: (a) schematic; (b) small-signal model...... 26 Figure 3-7 Transistor-level schematic of the Schmitt trigger used in the OOK receiver...... 28 Figure 3-8 Detailed simulation showing switching thresholds of ±53 mV...... 31 Figure 3-9 The activity detector used to demodulate the OOK signal to baseband...... 33 Figure 3-10 Timing diagram showing Manchester-Encoded Data...... 35 Figure 3-11 Timing diagram for a delay-based Manchester Decoder, using MED from Figure 3-10...... 36 Figure 3-12 Block diagram of a delay-based Manchester Decoder...... 37 Figure 3-13 SPICE simulation of the Manchester Decoder...... 39 Figure 3-14 Block diagram of a simple FSK modulator...... 40 Figure 3-15 Transformation of lossy LC tank with ESR to equivalent parallel RLC circuit...... 42 Figure 3-16 NMOS cross-coupled pair, with equivalent negative resistance value, from [16]...... 43 Figure 3-17 Transmitter schematic with frequency-switching network and off-chip LC tank...... 44 Figure 3-18 Digitally-controlled bias network for the FSK transmitter...... 47 Figure 4-1 Die photo of the fabricated transceiver circuits...... 48 Figure 4-2 Measured transfer function of the differential amplifiers...... 49 Figure 4-3 Schematic of the differential amplifier...... 51 Figure 4-4 Schematic of the test setup used to measure the Schmitt trigger thresholds...... 52

iii Figure 4-5 Measurement of Schmitt positive threshold...... 53 Figure 4-6 Oscilloscope plot showing the activity detector input (top trace) and output (bottom trace)...... 55 Figure 4-7 Activity detector release delay measurement showing activity detector response (bottom trace) to a step input (top trace)...... 57 Figure 4-8 OOK receiver output (top trace) when responding to a 50-mVpp input (bottom trace)...... 59 Figure 4-9 Oscilloscope trace depicting Schmitt trigger “chatter”...... 59 Figure 4-10 Plot of recovered clock (top trace) from MED (bottom trace)...... 61 Figure 4-11 Plot of recovered data (top trace) from MED (bottom trace)...... 61 Figure 4-12 Power spectral density of the FSK transmitter with modulating input held low...... 63 Figure 4-13 Power spectrum density of the FSK transmitter when modulated by a 4-kHz square wave...... 64 Figure 4-14 Power spectrum density for the FSK output when modulated with a 40-kHz square wave...... 65 Figure 4-15 700-mVpp sinusoidal oscillation sustained with 370 μW...... 66 Figure 4-16 2.75-Vpp sinusoidal oscillation sustained with 3 mW...... 66

iv Acknowledgements

I am very grateful for the opportunity to pursue this work and am forever

indebted to my thesis advisor, Dr. Steven L. Garverick, for his continual guidance and inspiration. This work would certainly not be the same without his expert advice and assistance.

I would especially like to thank Paul Fletter, Paras Samsukha, Amita Patil,

Ed Burwell, and Daniel Howe for the many enlightening conversations we have shared. Together they have greatly enriched my abilities in research and design. I would furthermore like to thank all of my friends and colleagues of the Mixed-

Signal Integrated Circuit group, for their familiar faces make long nights of work more tolerable.

Finally, I would like to thank the Dr. Margot Damaser, Brad Boggs, Jim

Buckett and the rest of the staff at the APT and FES centers of the Louis Stokes

Medical Center, Dept. of Veteran’s Affairs for their continuing help.

In closing, I would like to particularly acknowledge that this work was funded by the National Institutes of Health under grant NS047073 and the US

Dept. of Veteran’s Affairs.

v A Low-Power Wireless Transceiver for Deeply Implanted Biomedical Devices

Abstract

STEVE MAJERUS

This work presents the design of a low-power wireless transceiver optimized for use in deeply implanted biomedical devices. Potential operating frequencies were analyzed with respect to system limitations of power, size and complexity. It was reasoned that operation using low carrier frequencies provides high power efficiency while slightly increasing the size of off-chip antennas, and

125-kHz OOK and 27.12-MHz FSK were chosen for forward- and reverse- telemetry, respectively. Integrated telemetry circuits were designed to meet the power and data rate requirements for simple biomedical devices and the circuits were fabricated in the AMI 0.5-μm CMOS process. Test measurements have confirmed the functionality of the circuits, and have provided inspiration for future circuit redesign.

vi 1 Introduction

This thesis describes the design of a low-power transceiver for implantable

biomedical devices. The transceiver is suitable for monolithic integration with

other circuits and uses a minimum number of off-chip components. It achieves

full-duplex, bidirectional communication using dual frequency bands. The

operating frequencies are carefully chosen so that the transceiver may function

even when deeply implanted within human tissue.

1.1 Motivation

With the advent of microelectronics and microsensors, fully-implantable biomedical devices became feasible. By integrating “intelligent” circuits with

transducers inside biocompatible packages, devices could be fabricated that were

suitable for chronic implantation. Technological refinement has increased the

capabilities of these devices to the point where they can be used to diagnose or

heal some of the most ubiquitous of ailments. Of course, biomedical implants

must be wireless so that they can be safely implanted within the patient, and

issues related to wireless communication are key to device size, battery lifetime,

and effectiveness.

At shallow implantation depths of less than a few centimeters,

communication with an implanted device is readily accomplished. Unfortunately,

the human vital organs are located at depths that are significantly greater than a

centimeter. Muscle stimulators, for example, are designed to communicate only

over short distances and will fail to work when deeply implanted. Future

1 biomedical devices will be deeply implanted, and appropriate transceivers must be designed to communicate with them.

To increase functionality, the transceiver for a deeply-implanted device should enable full-duplex, bidirectional communication. With this feature, the biomedical implant can be externally controlled even as it is returning physiological information. As such the device could be incorporated in a number of complex systems, and the full-duplex scheme permits the use of closed-loop feedback or offline processing by an external controller.

Inductive powering of deeply implanted devices is often not feasible, so an on-board battery is necessary. To maximize the lifetime of the battery, the total power consumption of the implant must be minimized. While low-speed data acquisition is quite efficient, data transmission and reception commonly consume significant amounts of energy [1], especially when the absorptive properties of human tissue are taken into account. A low-power transceiver prolongs the battery lifetime of deeply implanted biomedical devices, and this environment presents some unusual challenges.

1.2 Background

Bi-directional, wireless transceivers have been used in a myriad of telemetric applications over the past 10 years, and are commonly customized to their specific use. Factors such as power consumption, size, operating distance, and data rate vary widely. Existing devices are typically not designed for deep implantation, but it is helpful to evaluate their potential for such an application.

2 1.2.1 Existing Transceivers

Existing biomedical transceivers can be divided into three main groups:

RF-powered, battery-powered low-bandwidth, and battery-powered high-

bandwidth. The performance characteristics of the devices within these groups

are similar, so it is instructive to analyze a single representative device from each classification. Transcutaneous neural recording arrays provide examples of RF- powered devices; the Bion stimulator is a low-bandwidth battery-powered device;

and wireless “endoscopes” such as the SmartPill represent the capabilities of the

battery-powered, high-bandwidth transceivers.

1.2.1.1 An RF-powered neural recording device: the Utah Array

Figure 1-1 is a block diagram of a 100-microelectrode neural prosthesis dubbed

the “Utah Array” [2]. A 2.64-MHz inductive link with an external controller

allows for the transmission of power and commands. The receiver in this device

uses very little power since no amplification is needed to detect the ASK-

modulated power signal used to send commands to the device.

Fig. 1-1. Block diagram of a multi-channel, dual-band implantable neural recorder

(from [2]).

From Figure 1-1 it is apparent that this device transmits reverse-telemetry

data on a high-frequency (433-MHz) channel. FSK is used to transmit the neural

signals from all 100 electrodes in the recording array. Reverse-telemetry could potentially be done using LSK over the 2.64-MHz link that is used for power and

commands, but data rate would be too low.

The performance characteristics of the Utah Transceiver are summarized

in Table 1-1.

4 Table 1-1: Performance summary of the Utah Transceiver, as described in [2].

Utah Array Scheme ASK Frequency 2.64 MHz Receiver Bit Rate < 10 kbps Power <100 μW

Scheme FSK Frequency 433 MHz Transmitter Bit Rate 330 kbps Power 6.75 mW

Total Power Consumption ~6.8 mW

1.2.1.2 Battery Powered Bion

The Bion, a therapeutic muscle stimulator, was designed with the support of the

Alfred Mann Foundation and is being commercialized by Advanced Bionics.

Since the Bion is a muscle stimulator, its original incarnation was as a shallowly implanted, inductively-powered device. Recent versions, as shown in Figure 1-2, employ an onboard rechargeable battery [3]. The use of a battery breaks the tether associated with the transcutaneous inductive link and allows for deeper implantations of up to 12 cm. Up to 1000 of these battery-powered Bions (BPBs) can be monitored by one external host controller, and a time-division multiple access (TDMA) protocol is used so the entire network of BPBs can share the same frequency band.

Fig. 1-2. Photograph of a packaged BPB (from [3]). Size is 3mm by 25mm.

The BPB transceiver operates at a center frequency of 500 MHz and communicates via quadrature phase-shift-keying (QPSK) over a 5-MHz bandwidth. Since TDMA is employed, the transceiver shares a single antenna for forward- and reverse-telemetry, but operation at 500 MHz consumes significant power. The BPB receiver and transmitter consume 9 mW each while active, and the total bidirectional bit rate for the entire network is 1.925 Mbps.

The performance characteristics of the Battery-Powered Bion are summarized in Table 1-2.

Table 1-2: Performance summary of the BPB in [3].

Battery-Powered Bion Scheme QPSK Frequency 500 MHz Receiver Bit Rate 1.155 Mbps Power 9 mW

Scheme QPSK Frequency 500 MHz Transmitter Bit Rate 770 kbps Power 9 mW

Total Power Consumption 18 mW

6 1.2.1.3 A wide bandwidth, battery powered recorder – the SmartPill

Some applications require very high data rates; notable examples include wearable physiological monitors [4] and wireless endoscopes. These devices commonly operate at UHF frequencies so that wide bandwidths are available. A commercial device called the SmartPill, shown in Figure 1-3 [5], transmits video on the UHF band. A different wireless endoscope described in [6], for example, receives command instructions via ASK signal at 2.4 GHz. Half-duplex communication is achieved through TDMA, and the transceiver uses the same antenna for command reception and to broadcast digital video.

Fig. 1-3. Conceptual drawing of the SmartPill as it traverses the gastro-intestinal

tract (from [5]).

7 Unfortunately, as in the BPB, synthesis and reception of the high- frequency signals consumes large amounts of power. These power requirements are reflected in Table 1-3.

Table 1-3: Performance summary of the wireless endoscope in [6].

Wireless Endoscope Scheme ASK Frequency 2.4 GHz Receiver Bit Rate 256 kbps Power 15 mW

Scheme ASK Frequency 2.4 GHz Transmitter Bit Rate 1 Mbps Power 14 mW

Total Power Consumption 29 mW

8 2 Frequency Selection and Design Considerations

This chapter begins with an analysis of frequency bands that are suitable for

transceiver operation. Factors such as domestic and international regulations, medium absorption, and channel noise are considered. The operating frequencies for the transceiver are analyzed based on implications of physical device size, weight, and power consumption. After the operating frequencies are selected, reasonable design goals pertaining to telemetric performance are presented.

2.1 Frequency Analysis

2.1.1 FCC Regulations

The specific absorption rate (SAR) of human tissue varies widely as a function of

frequency, and both electric and magnetic fields must be considered. For this

reason, the US Federal Communications Commission (FCC) has adopted RF

exposure limits from various ANSI and IEEE standards. No commercial device is

legally permitted to deliver RF power into human tissue in excess of these limits.

Physiologically, absorption of RF electric and magnetic fields results in

heating of human tissue. Because tissue heats rather slowly, it “integrates” the

power being absorbed. Due to the time-averaging effects of RF exposure, all

FCC power limits are defined as a maximum permissible exposure (MPE) level

averaged over six minutes. In short, this implies that a continuously transmitting

device must always remain below the MPE level, while a low duty-cycle

transmitter can briefly exceed the MPE level providing that the six-minute

average is below the limit.

9 Figure 2-1 lists the MPE levels for both electric and magnetic fields at various frequencies. The MPE for frequencies lower than 300 kHz is the highest,

since the absorption rate of the human body at these frequencies is minimal. As

the transmitted field approaches 30 MHz however, the SAR begins to increase, so

the MPE is scaled inversely with increasing frequency. Frequencies between 30 and 300 MHz cause the most heating of tissue, so very stringent MPE limits are

imposed for this band of frequencies.

Fig. 2-1. Table of FCC limits for maximum permissible exposure to electrical

and magnetic fields at various frequencies (from [7]).

10 2.1.2 Power Considerations on Band Selection

Considering the FCC limits on MPE and the frequency bands that are devoted to

short range communication for industrial, scientific, and medical (ISM) applications, a few frequencies are eligible for use by a biomedical transceiver.

Channel interference caused by commercial transmitters in the AM, FM, VHF, and UHF bands can be avoided by operating within these FCC short-range ISM bands. Narrow ISM bands exist across the frequency spectrum, but are essentially

split into low, medium and high groups (if microwave frequencies are ignored).

These ISM bands correspond to frequencies near 125 kHz, 27.12 MHz, and 433

MHz, respectively [8].

The properties of the human body make RF transmission from an

implanted device inefficient. While human tissue is mostly comprised of water,

the high concentration of mobile ions such as Na+, K+, and Cl- makes the tissue

more conductive than pure water. As a result, the electrical conductivity of the

human body is about 1/5 that of seawater. Figure 2-2 plots the attenuation per

meter in electrical field power as a function of frequency for seawater and pure

water. From this chart it is clear that electric fields propagating through saltwater at frequencies above 1 GHz experience significant attenuation due to the induction of eddy currents within the medium. Therefore, transmitters that operate at lower frequencies may transmit at a lower output power level to

achieve a given distance [3]

Fig. 2-2: Plot of the attenuation by distance versus frequency for electric fields

propagating in pure water and seawater (after [3]).

For devices that are destined for deep implantion, such as the BPB or the

SmartPill, high carrier frequencies are often used since their antennae can be made quite small. Moreover, by utilizing a wider bandwidth, these devices can transmit at very high data rates, which allows for the creation of ad-hoc networks of sensors or the transmission of real-time video. However, these benefits come at a high power cost since amplification and synthesis consume significant energy.

12 Since high data rates are not needed in single-channel biomedical sensors,

the power consumption of the transceiver can be reduced significantly by communication at lower carrier frequencies. For data reception, low-bandwidth amplifiers can be used, the low bit-rate signal causes less switching losses in digital circuits, and the Quality factor (Q) of the antenna can potentially be better at lower frequencies. For data transmission, a carrier can be synthesized without the use of a frequency synthesizer or crystal reference, less bias current can be used to transmit the signal, and the signal can penetrate environments such as saline or tissue with less losses.

2.1.2.1 Receiver Power

Reception of low-frequency carriers requires significantly less power in the electronics than reception of high-frequency carriers. This power reduction can be quantified in several ways, but small signal analysis of an amplifier is convenient.

The gain-bandwidth product of most amplifiers can generally be improved by simply increasing the transconductance of certain devices at the expense of die area and/or power consumption. Figure 2-3 depicts a model of a two-stage amplifier with a single Miller compensation capacitor, CC.

Fig. 2-3. A two-stage differential amplifier (from [9]).

The low-frequency gain of this amplifier is given by

AgrrgrrVo= m12(||) o o 4 m 55 (||) o o 6. (2.1)

The small-signal model of Figure 2-4 can be used to characterize the frequency response. In the small signal model, Resistors R1 and ROUT and capacitors C1 and CL are lumped devices comprised of several FET small-signal parameters.

Fig. 2-4. The small-signal model for the output stage of the amplifier in Figure 1-

5, used to determine the frequency response (after [9]).

14 The frequency response of this amplifier is governed by two poles, with a dominant pole leading to a gain-bandwidth of

Gm1 ω0 ≈ ' . (2.2) CC

The radial frequency of the non-dominant pole is

'' ⎛⎞GCCmLC2 ⎛⎞ p2 ≈ ⎜⎟'''''⎜⎟. (2.3) ⎝⎠CCCCCCCLLCLC⎝⎠++11

As can be seen in the above equations, the transconductance of the two stages has a significant effect on the frequency response of the amplifier. In general, to obtain wide bandwidth the input pair transconductance is increased.

To maintain stability, Gm2 and p2 must be increased in proportion. Unfortunately, the increase in transconductance is most easily accomplished by raising the bias current in the amplifier’s input and output stages. [9]

Due to the lack of large passive devices in a conventional IC process, channel selection and out-of-band noise rejection is challenging. Off-chip passives can be minimized if the transceiver’s resonant antennae are used as the main channel-selection filters. Tuned antennas have a bandpass frequency response similar to the response depicted in Figure 2-5. Such antennas have a

Quality factor that represents the ratio of the stored energy to the dissipated

15 energy per cycle. The Q of the antenna essentially represents its efficiency at selecting a single frequency and rejecting all others.

Fig. 2-5. The typical frequency response of a bandpass filter, such as a parallel-

resonant LC tank (from [10]).

It can be shown that the bandwidth of the frequency response such as shown in Figure 2-5 is given by

f ff−=0 . (2.4) 21Q

The antenna Q must be high to obtain a narrow bandpass filter. The bandwidth of a low Q antenna does not perform adequate band-limiting, so additional power mixers and high-Q IF filters must be employed to tune the transceiver.

For an inductive antenna, the Q can be calculated using

16 ωL Q = , (2.5) RS where RS is the effective series resistance of the coil. At low frequencies, this resistance is nearly equal to the DC resistance of the conductor, but the skin effect becomes prevalent at high frequencies.

The skin depth of a conductor is given by

2ρ d = (2.6) ωμ where ρ is the material resistivity, ω is a radial frequency, and μ is the material absolute magnetic permeability. At any given frequency, 70% of the current density flows at a distance less than d from the conductor outer surface. As frequency increases, the same current density is forced to flow in a narrower and narrower space. This current-crowding effect effectively increases the conductor resistance at high frequencies.

The RS of an inductor will rise as the skin depth approaches the diameter of the conductor, so at some frequency the Q of the inductor will be at a maximum. This Q peak can occur anywhere in the frequency spectrum, but usually occurs at frequencies less than 100 MHz for compact inductors wound with thin wire. Due to this effect, superior antenna performance can be attained at lower frequencies.

The dynamic power consumption of high-speed digital circuits can be significant, as evidenced by the massive heatsinks prevalent on modern computer processors. The capacitive switching power dissipation is given by the familiar equation

17 2 PCVfD = . (2.7)

From this equation it is apparent that the power dissipation of digital circuits increases linearly with frequency. Therefore, operation at low frequencies allows for data demodulation and decoding with reduced power consumption.

Digital signal processing can be used to improve the bit error rate of the receiver through the use of techniques such as Manchester Coding.

2.1.2.2 Transmitter Power Savings

Operation at low carrier frequency can also reduce transmitter power, which commonly accounts for a large percentage of the dissipation in an implanted device. Power consumption is primarily related to the generation of a stable carrier frequency. Tuned resonators such as parallel cut crystals and surface-acoustic-wave (SAW) devices used for high-frequency oscillators are bulky and difficult to modulate. Frequency synthesizers can be used to up- convert a stable reference frequency, but these circuits typically have high power consumption.

Parallel LC tank oscillators can also be used to generate a stable high- frequency reference. These circuits are more efficient than frequency synthesizers, but their power consumption is still affected by operating frequency.

These oscillators require constant bias current to sustain oscillation, and the amount of bias current needed is inversely proportional to the Q2 of the LC tank, which is generally low at high frequencies.

18 2.2 Frequency Selection

When the pros and cons of the various frequencies available for a low-power, deeply implanted transceiver are weighed, it is concluded that lower frequencies are preferable. Low frequencies only support low data rates and time-division multiplexing between data reception and date transmission would erode both the forward- and reverse-telemetry rates. Therefore, dual frequency bands are desirable to support full-duplex communication and preserve telemetry bit rates.

The proposed transceiver uses a dual-frequency design, with low bit rate commands transmitted at 125 kHz, and reverse telemetry of data at 27.12 MHz.

Both of these ISM frequencies have adequate bandwidths, acceptable attenuation in tissue and salt water, and high MPE levels. The relatively low carrier frequencies permit the use of low-power transceiver circuits. Disadvantages include the need for two antennas, each requiring relatively large tuning components.

Finally, FSK modulation has very high spectral efficiency and is chosen for 80-kbps reverse-telemetry. ASK demodulation is accomplished with simple circuitry and can be used for low bit-rate commands.

2.3 Design Goals

Given a powerful incoming signal, the implantable receiver does not require high sensitivity, which is defined as the minimum peak-to-peak voltage signal at the receiver input that yields successful signal detection. Since the receiver is designed to demodulate an OOK signal, the input sensitivity requirement specifies

19 the minimum voltage needed to receive a logic “1”. In this design, the LC tank antenna is assumed to have a Q of 10 at 125 kHz, and the receiver must detect a signal as small as 1 mV across the tank. The bandwidth of the implanted receiver must be sufficiently high to pass 125 kHz, and a –3 dB bandwidth of 2 MHz has been chosen. Finally, the receiver output must be a baseband digital signal. The incoming bit rate will be 4 kbps, but the final bit rate will be 2 kbps after

Manchester decoding is used to recover the clock and serial data.

The transmitter must drive an LC tank into resonance at 27.12 MHz. The

Q of the LC tank is assumed to be equal to 10 at this frequency. The transmitter must maintain an adequately-large signal swing across the LC tank since it is also used as the antenna. Finally, the transmitter must create a FSK-modulated waveform with a bit rate of 80 kbps.

The power consumption of the implanted transceiver is a crucial design criterion. The receiver can operate with very low power, but the transmitter must use a relatively large amount of power simply to transmit a signal from the human body. The transceiver is designed to operate from a 2.7-V supply and the total power consumption should be no higher than 1.1 mW, with 100 uW allocated to the receiver and 1 mW allocated to the transmitter.

The design specifications for this transceiver are summarized in Table 2-1.

20 Table 2-1: Summary of the transceiver design specifications.

Transceiver Specifications

Scheme OOK Frequency 125 kHz Receiver Bit Rate 4 kbps Sensitivity 1 mV Power <100 μW

Scheme FSK Frequency 27.12 MHz Transmitter Bit Rate 80 kbps Output Power Variable Power <1 mW

Power Consumption <1.1 mW

21 3 Architecture & Design

The circuit design of the implantable transceiver is made very simple to conserve power, but by simplifying the implantable circuits, a greater burden is put on the external transceiver. The size and power consumption of the external transceiver are not constrained.

3.1 Architecture

A simplified block diagram of the transceiver is shown in Figure 3-1. For full-duplex bidirectional communication, two antennas are used, and different modulation schemes are used on separate frequency bands to obtain low cross- talk and individual optimization. As can also be seen in Figure 3-1, both of the antennas are inductive and are components in resonant LC tanks. Due to the long wavelengths of low-frequency electromagnetic fields, conventional wavelength- tuned antennae would be prohibitively large for an implantable device. Inductors need not conform to a certain length, but high inductance and Quality factor is desirable. Since this transceiver uses off-chip inductors, there are a myriad of tiny devices that can meet the inductance requirement.

Fig. 3-1. Simplified block diagram of the low-power transceiver.

3.1.1 OOK Receiver

Figure 3-2 presents a block diagram of the receiver analog front-end. Signal- processing for the low-frequency OOK signal is simple. After passing the LC tank, which is used as the main channel selection filter, the received signal is amplified to a larger level and hard-limited to produce a full-scale binary signal.

At this point, the received signal is essentially comprised of bursts of binary pulses at 125 kHz (denoting logic 1) or nothing (denoting logic 0). To complete the demodulation, the 125-kHz pulses are envelope-filtered to produce a low bit- rate, baseband binary signal. Further processing may be required for decoding, packet detection, etc.

Fig. 3-2. Block Diagram of the OOK receiver.

Figure 3-3 provides a more detailed receiver block diagram. The small voltage received by the inductive antenna is amplified via cascaded Differential

Amplifiers (DAs). To avoid amplifying DC offset and to provide some band- limiting, the first two DA stages are AC-coupled. The DA gain is chosen so that the antenna signal is boosted above the Schmitt Trigger hysteresis range. When the Schmitt Trigger senses a differential voltage greater than its positive threshold

(+VST), its output goes high. When the differential input falls below its negative threshold (-VST), the output goes low.

Following the Schmitt Trigger, the received signal is a binary OOK- modulated signal with pulses occurring at the frequency of the carrier. To complete the demodulation to baseband, the short pulses are provided to an activity detector to obtain a digital baseband signal, as illustrated in Figure 3-4.

Fig. 3-3. High-level circuit diagram of the OOK receiver.

Fig. 3-4. Binary approximation of an OOK-modulated signal.

3.1.1.1 Differential Amplifier

The schematic of the DA employed by the receiver is shown in Figure 3-5. This single-stage amplifier is designed for low gain and low power. By using a fully- differential topology throughout the receiver, high CMRR and PSRR are obtained. Devices M1-M4 form the “core” of the amplifier, while all other FETs are used to set bias conditions. The gain of this amplifier is equivalent to that of a common-source amplifier with an active load, and this is best demonstrated via half-circuit analysis.

Fig. 3-5. Circuit schematic of one of the DAs used in the OOK receiver.

25 The amplifier half-circuit is shown in Figure 3-6a, along with its transformation into a simple small-signal circuit in Figure 3-6b. Half of the drain current of MS flows through each branch of the amplifier and the source of M1 is biased at potential VS, which acts as a small-signal ground.

Fig. 3-6. Half-circuit model of the DA: (a) schematic; (b) small-signal model.

For small input signals the gain of this amplifier is given by [11]

gm1 Av ≈− . (3.8) gm3

To obtain useful voltage gain, input transistor M1 needs to have gm much greater than for M3. To achieve this, Mta is used to “steal” bias current from its diode-connected counterpart M3. Thus, the gm of M3 is reduced, increasing the

gain of the amplifier. Assuming μnp= 3μ , the gain of the amplifier can be expressed as

26 WLI/ gm1 ()1 D1 Av ≈− ≈− . (3.9) g 3/WL I− I m3 ()()3 DD13

It seems counterproductive to use a PMOS input pair with diode- connected NMOS loads since the carrier mobility of an NMOS device is approximately three times that of a PMOS. However, the use of PMOS inputs allows the biasing to be simplified by referencing the input to ground, as shown in

Figure 3-3. With the PMOS input pair, this amplifier can respond linearly to inputs biased at zero volts, and can even function if the input signal is slightly negative. A center-tapped antenna could be used to provide a balanced input, but single-ended operation is envisioned.

To avoid device mismatch that would lead to offset voltage at the DA output, the amplifier FETs were carefully laid out. A unit width of 12 μm was used, and all devices were constrained to be integer multiples of the unit size.

The DA was designed for a differential gain of about 7 so that the total differential gain of the cascaded DAs is about 50.

Table 3-1: Transistor sizes and bias currents for the DA.

Transistor Name Size (W/L) μm Bias Current (μA) M1-2 72/3 1.27

M3-4 12/4 0.27

Mta-b 36/4 1.0

MS 72/4 1.0

27 3.1.1.2 Schmitt Trigger

A Schmitt trigger comparator is used to produce a binary OOK signal. Its circuit schematic is shown in Figure 3-7. M3 and M4 form a positive feedback loop that leads to hysteresis. The gain of this feedback loop is determined by the size mismatch between M3-4 and their diode-connected counterparts. For the purposes of analysis the width ratio of M3 to M5 and M4 to M6 is denoted asα .

For α < 1, the positive feedback boosts the small-signal gain of the amplifier, but for α > 1 the circuit becomes bistable. In its bistable state, the amplifier functions as a Schmitt Trigger which will hold its output state unless the differential input exceeds some threshold value. It can be readily shown that the

Schmitt Trigger thresholds are adjustable via the feedback factorα .

Fig. 3-7. Transistor-level schematic of the Schmitt trigger used in the OOK

receiver.

28 The Schmitt Trigger operation can be analyzed by studying its core consisting of FETs M1-6. Since the circuit is bistable, analysis begins by assuming that the circuit is latched into one stable state. If it is assumed that transistors M3 and M5 are turned on, then the loop is latched with M3 sinking all of the M2 bias current, keeping M4 and M6 cutoff. When a differential voltage VID is applied to

the input pair, M1 carries ()IS /2 − Δi and M2 carries(IS /2) + Δi . M3 can sink a

maximum current ofα ID51= α ID , and any residual current from M2 will flow into the M4-M6 current mirror. Once this mirror is turned on, the positive feedback loop is enabled and regeneration occurs, flipping the state of the latch. After regeneration, M3 and M5 are cutoff and the state of the latch is held until a sufficiently negative voltage reverses the process.

Assuming that small-signal models apply for the input pair, the threshold of the Schmitt Trigger is calculated by equating the current of M3 and M2, i.e. [12]

α ⎡⎤IgVIgV/2−=+ /2 /2 /2 . (3.10) ⎣⎦()S() m1,2 ST() S( m 1,2 ST )

VV− IS /2 SG1,2 Tp Solving for VST with the substitution = yields gm1,2 2

(α −1) VVST=−() SG1,2 V Tp . (3.11) ()α +1

The symmetry of the circuit assures equal but opposite thresholds, ±VST .

29 The device sizes used in the present prototype are listed in Table 3-2. The transistors in the Schmitt trigger were chosen using integer multiples of a unit size to obtain a preciseα = 4/3, and the input pair was biased using 7 μA to obtain a

predictable VVSG−= T .35 V. The Schmitt Trigger thresholds are therefore ±50 mV. Since the two-stage DA provides a gain of ~50, the minimum detectable input amplitude is ~1.0 mV.

Table 3-2: Transistor sizes used in the Schmitt trigger.

Transistor Name Size (W/L) μm MS 72/4

M1-2 9/3

M3-4 24/3

M5-8 18/3

M11-12 18/3

M9-10 36/3

M13-14 36/3

SPICE simulation was used to verify the hand-calculated Schmitt trigger thresholds. Since the positive feedback loop of the Schmitt Trigger is unstable around the switching thresholds, PSPICE did not converge during a DC sweep.

Instead, a slowly-varying differential ramp input was applied to the device and the output toggled at the switching threshold. A close-up of the simulation is shown in Figure 3-8. As can be seen in the figure, the switching thresholds for the

30 Schmitt trigger are roughly symmetrical and equal to about ±53 mV, in close agreement with the calculation.

Fig. 3-9. Detailed simulation showing switching thresholds of ±53 mV.

31 3.1.1.3 Activity Detector

It is well known that the envelope of an amplitude-modulated signal contains the baseband information, as illustrated in Figure 3-4. In this OOK system, the digital information is coded simply using the presence of the carrier as logic 1 and the absence of the carrier for logic 0. Traditionally, detection is performed using a leaky peak detector [13]. In this work, the Schmitt trigger may rest in a high or low state when the carrier is removed, so an “activity detector” is needed.

Traditional peak detectors use passive resistors and capacitors to set the decay rate. These passives can be small if the carrier is of sufficiently high frequency, but very large passives would be needed to filter out the 125-kHz carrier from the OOK signal. To avoid such large passive devices, the activity detection circuit in Figure 3-9 has been designed. This circuit can be broken into two halves: an edge-detection monostable (also called a one-shot), and a current- limited inverter.

Fig. 3-9. The activity detector used to demodulate the OOK signal to baseband.

The XOR gate and delay cell create the edge-triggered monostable that produces a digital pulse of the length of the delay for each edge in the binary input signal. This digital pulse drives the current-limited inverter on the right side of

Figure 3-9. The inverter can sink a large current using M3, but it can only source a limited amount of current through M1. When activity stops, M1 will charge the capacitor using a slow ramp whose rate is set by ID1/C1. By sizing M1 and C1 appropriately, the ramp rate can be set so if there is activity at 125 kHz, the capacitor will be drained through M3 before the logic threshold of the output inverter is reached.

The device sizes used in the envelope detector are listed in Table 2-3.

Transistor M3 carries a bias current of 100 nA, so the ramp rate is 0.2 V/μs. The logic threshold of the output inverter is approximately 1.35, so it requires ~ 6.75

μs of inactivity to produce a low output.

33 Table 3-3: Device sizes used in the activity detector.

Transistor Name Size (W/L) μm M1 6/3

M2 36/3

M3 12/3

C1 0.5 pF

3.1.2 Manchester Decoder

Wireless commands are sent to the transceiver in a serial digital format, unaccompanied by a clock. To aid clock recovery, the transmitted commands are

Manchester Coded.

In the Manchester Code, each bit is represented by an edge transition.

Figure 3-10 displays an example of serial data that has been transformed into

Manchester-Encoded-Data (MED) by XORing the “Data” and “Clock” signals. It is noted that a mid-bit transition occurs at each falling edge of the encoding clock, simplifying clock recovery. The value of the data in the second half of each encoded bit corresponds to the value of the decoded bit. For example, if a 1 is being sent, the MED will transition from low to high at mid-bit.

Fig. 3-10. Timing diagram showing Manchester-Encoded Data.

Transitions can also occur at the start of a MED bit, and the decoder must be intolerant to these spurious transitions.

3.1.2.1 Method for Decoding

A simple method to recover the clock from the MED uses a delay-based decoder [13] that locks into the guaranteed mid-bit transitions using knowledge of the bit rate. If the duration of each Manchester-coded bit is T, the mid-bit transitions are separated by T and a clock signal can be recovered from the MED using the method depicted in Figure 3-11. Each time a transition is detected, the recovered clock is reset to 0 and a ¾-T delay is triggered [13]. After ¾ T, the recovered clock is set to 1, and the current value of the MED is latched and inverted. The recovered data is delayed by ¼ T and the first bit is lost in start-up.

These discrepancies are unimportant. More importantly, if the recovered clock

35 initially locks to spurious transitions its phase will slip to the mid-bit transitions upon the first absence of a spurious transition.

Fig.3-11. Timing diagram for a delay-based Manchester Decoder, using MED

from Figure 3-10.

Figure 3-12 presents the block diagram of a delay-based Manchester

Decoder as described above. Decoding begins with an edge detector that can respond to rising or falling edges. The detection of an edge resets the clock to 0 and begins a bit-rate delay of ¾ T. After the delay expires, the clock is set to 1 and the input data is sampled. The Manchester Decoder supplies both the recovered data and the recovered clock for integration with a digital controller.

Fig. 3-12. Block diagram of a delay-based Manchester Decoder.

3.1.2.2 All-Digital Manchester Decoder

The Manchester Decoder depicted in Figure 3-12 has been designed using all digital logic. Edge detection is performed via two edge-triggered flip-flops.

When one of the two flip-flops detects an edge, the recovered clock is reset and a digital delay is triggered.

The 12-bit digital delay is implemented as a ripple counter that is driven by the 1-MHz system clock. By selecting the appropriate output bits from the counter and logically ANDing them, the required delay can be obtained. Once this delay is expired, the recovered clock is set and the input data is sampled, thereby updating the recovered data flip-flop. This cycle is repeated each time the

Manchester decoder detects an edge transition. Once the decoder is synchronized to the MED it will lock into the guaranteed mid-bit transitions and skip over the inconsistent transitions at the beginning of the MED bits.

The Manchester decoder performance was verified using a transient simulation. The decoder was simulated using a 500-kHz system clock and 8-bit

37 counter to obtain a timer delay of 384 microseconds, appropriate for MED having a data rate of 2 kbps. Figure 3-13 shows the results of the transient simulation.

The MED data rate is set by the signal labeled “Encoding Clock”. This signal is

XNORed with “NRZ Data” to create the decoder input “Manchester-Encoded

Data”.

The output signals “Recovered Clock” and “Recovered Data” are only valid after the decoder has synchronized to the MED data rate. The decoder functions properly only after it has locked onto the mid-bit transitions in the

Manchester-Encoded Data (MED), and the simulation verifies that the decoder will begin to properly function as soon as a 1-0 or 0-1 transition occurs in the

MED. In this simulation, transitions occur at the very start of the simulation, and the abnormally-long pulse in “Recovered Clock” is reset to 0 when the next bit edge is detected. This bit edge is the guaranteed mid-bit transition, and after this is detected, the decoder is synchronized.

It should be noted that the recovered clock in this implementation has a duty cycle less than 50%. When the decoder is locked the recovered clock period is determined by the bit duration T. Since the decoding delay is nominally ¾ T, a recovered clock duty cycle of ¼ T or 25% is expected. The decoder was designed using positive edge-triggered logic, so clock duty factor is not critical.

Fig. 3-13. SPICE simulation of the Manchester Decoder. The recovered clock

has a duty factor of 25% when synchronized.

39 3.1.3 FSK Transmitter

FSK-modulated signals can be created in several ways, but the simple method used here is depicted in Figure 3-14. In this circuit, an oscillator drives a parallel LC tank into resonance. The nominal frequency of oscillation is given by the familiar equation:

1 f0 = . (3.12) 2π LC

Fig. 3-14. Block diagram of a simple FSK modulator.

The inductor is implemented as a fixed passive device and the oscillator frequency is modulated by varying the tank capacitance. By switching a small capacitor in and out of the tank, i.e. C=C0 or C=C0+ΔC, the oscillation frequency is shifted. The total frequency shift can be derived from Equation 3.5 and is given by

⎛⎞ 11ΔCC/ 0 Δ=f ⎜⎟1. − ≅ f0 (3.13) 2π LC ⎜⎟1/+ΔCC 2 0 ⎝⎠()0

40 For example, a 2% change in C yields a 1% change in frequency.

In some applications, the AC voltage generated at the oscillator output ports would be amplified to drive a tuned antenna. In this case, however, the passive inductor that is required to generate oscillations can also function as the transmitting antenna to reduce component count [15].

3.1.3.1 LC Tank Modeling

Passive LC tanks contain several parasitics, but at low frequency, the main parasitic is the effective series resistance (ESR) of the components. Since inductors are formed from long spirals or coils of wire, they generally have a low to moderate DC resistance. The skin effect increases this DC resistance, however, as the frequency rises. Capacitors have some interconnect resistance, but they also have an electrostatic resistance which limits the amount of current they can supply.

As Figure 3-15 shows, these series resistances can be combined into an effective parallel tank resistance using an approximation that is valid near the resonant frequency [16].

Fig. 3-15. Transformation of lossy LC tank with ESR to equivalent parallel RLC

circuit.

ω0 L The inductor Quality factor is given by QL = while the capacitor quality RL

1 factor isQC = . The Quality factor of the capacitor is typically much ω0 RCC higher than that of the inductor, so its loss will be neglected in this analysis.

The equivalent parallel resistance of the inductor is given by [16]

2 ()ω0L 2 RQRQLPLSL===ω0 . (3.14) RS

The transmitter in this design is designed to drive an LC tank with a Q of

10 at 27.12 MHz. This is a pessimistic Quality factor since, at this low frequency, the skin effect is not severe. However, a low-Q inductor can be wound with very thin wire, thereby reducing the antenna size, and a relatively small value of L was chosen (1 μH ). Using 3.8, this corresponds to an RP of about 1.7 kΩ at 27.12

MHz.

42 3.1.3.2 LC Oscillator

Without the ESR losses, the ideal LC tank would oscillate forever once energized, since ideal capacitors and inductors do not dissipate any energy. In reality, however, the tank amplitude will eventually diminish as the stored energy is dissipated. Thus, the losses must be restored using energy supplied by the oscillator circuit. This energy restoring circuit acts as a negative resistance—as the tank amplitude diminishes it supplies extra current to maintain oscillation.

One well-known circuit that acts as a negative resistance is a cross- coupled pair of transistors biased into their active region. As shown in Figure 3-

16, the effective small-signal resistance looking into the drains of M1 and M2 is

equal to− 2 / gm , where gm is equal to the transconductance of each FET [17].

Fig. 3-16. NMOS cross-coupled pair, with equivalent negative resistance value

(from [17]).

According to the Barkhausen criteria [18] for oscillation, loop gain must be greater than unity and loop phase shift must be exactly 180 degrees at the

43 frequency of oscillation. In the case of this negative-resistance oscillator, the

criterion reduces to gm/2 >1/ RP .

In the left half of Figure 3-17, a negative-transconductance oscillator is depicted which uses two pairs of cross-coupled FETs. In this configuration, the transconductances of the PMOS and the NMOS cross-coupled pairs are added.

Since the negative conductance of each cross-coupled pair is−gm / 2, the total

loop negative conductance is −Gm = gm, assuming all FETs have equal gm .

The loop will oscillate if the loop transconductance exactly matches the inverse of the parallel tank resistance, but a safety margin is usually incorporated

into the design. When sizing and biasing the FETs, their gm is often designed to be two to three times higher than needed to guarantee oscillator function despite process variation.

Fig. 3-17. Transmitter schematic with frequency-switching network and off-chip

LC tank.

44 Using Equation 3.7 above, the effective parallel resistance of the tank near resonance can be modeled by a 1.7- kΩ resistor. Therefore, the FETs in the

oscillator must have a gm of at least 588 μS to satisfy the Barkhausen criteria.

Using a safety factor of three, each FET should contribute a gm of 1.76 mS .

As per the well-known square-law model, a FET in saturation has transconductance given by

W gCI= 2.μ (3.15) moxDSL

Assuming that process-dependent parameters are fixed, the FET transconductance can be set by the device size or bias current. There is a practical limit to the FET

size, since the Cgs will begin to load the LC tank. Therefore, to achieve the

required gm , some power must be consumed.

The oscillator core creates a negative conductance which excites the LC tank and drives it into oscillation. However, to create an FSK signal, the frequency of the oscillation must be controlled. Equations 3.5 and 3.6 govern the oscillation frequency of an LC tank as an incremental capacitance ΔC is added to the tank using the switch network of Figure 3-17. When the input labeled

“Switch” goes high, Msa and Msb are turned on, effectively adding a capacitance of ΔC to the LC tank, shifting the oscillation frequency as per Equation 3.6.

45 3.1.3.3 Digital Power Control

The oscillator FETs have been sized and biased to provide a transconductance adequate for QL=10, but it is conceivable that the antenna Q may be lower or higher. Moreover, the signal swing across the LC tank may need to be adjusted according to the desired telemetry range. Voltage swing is given by

VIZSWING= BIAS TANK , (3.16)

assuming that the oscillator core is working in a switching mode in which the inductor is driven by the full bias current [19].

To optimize the transmitter for a given application, it is therefore useful to adjust the bias current of the transmitter. Figure 3-18 depicts a bias circuit with digitally-adjustable output bias. An on-chip bandgap reference can supply a stable unit bias current. This current is copied into the transmitter bias circuit via an array of binary-weighted current mirrors. The currents from these mirrors pass through series PMOS devices and sum at the drain of MT. With all of the PMOS switches on, the drain current of MT is approximately equal to 7 times the bandgap current. By turning on/off the signals S0-S2, the amount of current flowing into the drain of MT can be varied. This current is mirrored into the tail bias source for the oscillator. MT has been sized to obtain a VGS-VT that ranges from 35 mV to 112 mV.

Fig. 3-18. Digitally-controlled bias network for the FSK transmitter.

47 4 Test Results

The transceiver circuits were fabricated in the AMI 0.5-μm process through

MOSIS and a die photo is presented in Figure 4.1. Measurements were performed on all five ICs and chip performance is closely matched among the chips, i.e. within normal chip-to-chip process variations.

4.1 Test Setup

The chips were powered from a 3.3-V DC supply, and an on-chip regulator provides a 2.7-V rail to all the transceiver circuits. A 100-nF bypass capacitor was included to ensure supply ripple less than 1 mV. An off-chip crystal oscillator provides a precise 1-MHz clock signal to the Manchester decoder. An external PIC microcontroller drives an on-chip Serial-Peripheral Interface (SPI) port that is used to serially program the FSK digital power control.

Fig 4.1. Die photo of the fabricated transceiver circuits. A SAR ADC is

highlighted for size reference.

48 4.2 OOK Receiver

4.2.1 Differential Amplifier

The receiver amplifier cascade was tested using an Agilent 4395A network analyzer. Figure 4.2 shows the measured transfer function, where bandwidth is limited by off-chip capacitive load. The low-gm amplifiers cannot easily drive the pad, package, and active probe capacitance, so the bandwidth is degraded compared to the ultimate goal of 125 kHz. The low-frequency gain of 42 dB is about 8 dB higher than the designed gain of 34 dB (~50). This was caused by a layout error in the on-chip current mirror that upset the current biasing.

Fig 4.2. Measured transfer function of the differential amplifiers. Low-frequency

gain is 42 dB and the amplifier bandwidth is limited by the test capacitance.

49 The OOK receiver uses a single reference current to set bias conditions for

PMOS and NMOS current sources. Table 4-1 compares the desired and actual bias settings of the FETs in the receiver amplifier caused by the layout error.

Figure 4.3 is included as a reference. The amplifier gain is set by the gm1,2/gm3,4 ratio. The calculated and re-simulated receiver gain with the actual bias settings closely matches the measured low-frequency gain of 42 dB. Current was estimated by measuring an off-chip current reference and scaling by on-chip current mirror ratios.

The measured 3-dB bandwidth of 3 kHz corresponds to an apparent load capacitance of 6-pF, a reasonable value given a simulated 8.6 MΩ output resistance. In the absence of this parasitic capacitance, simulation predicts a bandwidth of 830 kHz.

Table 4-1: Comparison of designed and estimated actual bias currents and output

resistance for the differential amplifier.

Bias Current (μA) ro (MΩ) Name Designed Estimated Designed Estimated

M1-2 1.27 1.24 15.8 16.1

M3-4 0.27 0.08 37.0 125

Mta-b 1.00 1.16 10.0 8.6

Ms 2.52 2.48 7.9 8.1

Fig 4.3. Schematic of the differential amplifier.

51 4.2.2 Schmitt Trigger

The Schmitt trigger was tested by applying a balanced differential input with a common-mode level approximately equal to VT of an NMOS. A schematic of the test setup is shown in Figure 4.4. Two function generators are used to overpower the output of A2, thereby driving the Schmitt trigger directly. The function generator outputs are centered at 0.7 V and synchronized such that there is a 180 degree phase difference between them. One oscilloscope channel is used to measure the differential Schmitt trigger input, while a second channel monitors the Schmitt trigger output.

Fig. 4.4. Schematic of the test setup used to measure the Schmitt trigger

thresholds.

The positive-going threshold is obtained by noting the positive differential input when the Schmitt trigger output switches from low to high. Similarly, the negative-going threshold is obtained when the output switches from high to low.

52 An oscilloscope trace depicting the measurement method is shown in Figure 4.5.

Switching thresholds were measured to the nearest 0.5 mV.

Fig 4.5. Measurement of Schmitt positive threshold.

The thresholds obtained from Figure 4.5 are +16 mV and -35 mV. The spread between thresholds is about ½ the expected 100 mV, and offset by -10 mV in this part. Simulation results closely match the hand-calculated Schmitt thresholds, and it is not presently understood why the measured thresholds are different. The MOSFET square-law model was used for hand calculation, and the

PSPICE simulator used the BSIM model. The Schmitt thresholds may differ due to moderate-inversion currents that are not easily predicted by either the square- law or BSIM models.

53 Five copies of the Schmitt Trigger were tested to investigate the threshold inaccuracy. Table 4-2 lists the measured thresholds. The absolute values of the thresholds are not well matched, but the threshold separation (difference) is fairly consistent. A possible cause of the offset may be an input-referred random offset that is effectively amplified by an attenuation factor at the Schmitt trigger inputs.

Table 4-2: Measured Schmitt trigger thresholds.

Chip V+ V- Difference Average 1 26 mV -35 mV 61 mV -4.5 mV 2 46 mV -15 mV 61 mV 15.5 mV 3 4 mV -39 mV 43 mV -17.5 mV 4 15 mV -40 mV 65 mV -12.5 mV 5 16 mV -35 mV 51 mV -9.5 mV

Testing limitations may also be skewing the measured results. The

Schmitt trigger was tested by grounding the amplifier inputs and overpowering the amplifier outputs with two function generators. Each function generator has an output resistance of 50 ohms, and the IC inputs include 100-ohm series resistors. The sink current of the differential amplifiers, however, is not constant with applied voltage. A diode-connected NMOS is connected from each amplifier output to ground, and the current drawn by this FET is related to its gate voltage. Due to the voltage drop across the 100-ohm ESD resistors, the voltage observed at the IC pins may not be the same as the input of the Schmitt trigger.

54 4.2.3 Activity Detector

The activity detector was tested by applying a modulated 100-mVpp 125-kHz sinusoid to the Schmitt trigger inputs, yielding alternating bursts of a 125-kHz square wave at the Schmitt trigger output/activity detector input. During bursts of

125-kHz activity the output is high. In between the 2-kbps bursts, the activity detector output should remain low.

Figure 4.6 presents an oscilloscope trace depicting the function of the activity detector. The top trace is the output of the Schmitt trigger, which remains constant or toggles at either 125 kHz depending on the modulating input. The bottom trace is the activity detector output. When the detector senses Schmitt trigger activity its output goes high; otherwise the activity detector output remains low.

Fig. 4.6. Oscilloscope plot showing the activity detector input (top trace) and

output (bottom trace).

55 When the Schmitt trigger output toggles, the activity detector output is set high within several tens of nanoseconds since the signal propagation path includes only a few logic gates. When the Schmitt trigger ceases its activity, however, the activity detector relies on a I / C slew rate to form an analog delay that must be greater than or equal to half a carrier period, i.e. 4 μs. The activity detector output will not reset low until this delay is over, so a useful measurement is the detector

“release delay”.

Figure 4.7 is an oscilloscope trace showing the activity detector release delay. The measurements indicate that the output is reset after 5.08 μs, which meets the requirement, including a 25% margin. The designed value was ~7 μs,

28% larger then the measured result. This discrepancy is likely related to process and lithographic variation. Fabrication test data for the run reveals that poly- poly2 capacitance was lower than the value used in design and lithographic skew may have further reduced the value of the ramp capacitor. Finally, the ramp rate is determined by the ratio of on-chip current mirrors which may not be perfectly matched.

Fig. 4.7. Activity detector release delay measurement showing activity detector

response (bottom trace) to a step input (top trace).

4.2.4 Entire Receiver

The entire OOK receiver was tested by applying a 50-mVpp sinusoidal waveform to the input amplifier cascade. The waveform was OOK-modulated using a 2- kHz square wave, corresponding to 4-kbps, the maximum bit rate for this link.

Figure 4.8 plots the receiver output (top trace) versus the receiver input (bottom trace). The desired demodulation function is observed.

Fig. 4.8. OOK receiver output (top trace) when responding to a 50-mVpp input

(bottom trace).

In Figure 4.8, the receiver was tested using a relatively large input waveform. Although the cascaded amplifiers provide sufficient gain, tests have shown that the amplifier outputs are corrupted by interaction with the Schmitt trigger when the input is low, as shown in Figure 4.9. It appears that the Schmitt trigger kicks back a small amount of charge through the CGS capacitance of the input FETs when it switches. This charge could overwhelm the outputs of the low-gm amplifiers, momentarily changing the input voltage of the Schmitt trigger.

The Schmitt trigger output toggles repeatedly following each sign change when the DA input is low.

The activity detector can filter out spurious Schmitt trigger chatter but for bit rates larger than 1 kbps the chatter duration approaches the bit duration,

58 skewing the duty cycle of the received binary data. In the present design the chatter issue is mitigated when an input waveform ≥ 50 mV is received, yielding a large signal amplitude at the Schmitt trigger inputs. In other words, the receiver input sensitivity of this prototype is ~50 mV, not the desired 1 mV.

Fig. 4.9. Oscilloscope trace depicting Schmitt trigger “chatter”. The bottom trace is the input, and the top trace is the output. The DA cascade is driven using a 10-

mVpp sinusoid.

The complete OOK receiver consumes 50 μW to receive a 4-kbps

Manchester-encoded signal. After decoding, the bit rate is halved so energy/bit is

25 nJ/bit. Although its timing is precise, the all-digital Manchester decoder accounts for approximately half of the receiver power due to the CV2f power losses inherent to digital circuits.

This receiver energy/bit is higher than present state-of-the-art receivers, which attain 1 nJ/bit [20], but this can be attributed to the use of a 2.7-V supply rail (as opposed to 1-V or lower) and a relatively low bit rate. Energy usage/bit is reduced when bit rates are increased, but total power is increased. There is no need for the larger command bit rates in a typical deeply implanted device.

4.3 Manchester Decoder

A test signal for the Manchester decoder was created by logically XORing a repeating 2-kbps bit pattern of 1100 with a 4-kHz encoding clock. This signal was applied to the input of the decoder, which was operated using a 1-MHz system clock. Figure 4.10 shows the recovered clock and the MED, while Figure

4.11 shows the recovered data and the MED. As can be seen in the figures, the decoder has successfully recovered a 25% duty cycle, 4-kHz encoding clock and the repeating 1100 data pattern.

Fig. 4.10. Plot of recovered clock (top trace) from MED (bottom trace). Clock rising edges occur 384 μs after the MED mid-bit transitions, as expected. Clock

falling edges are properly aligned to these transitions.

Fig. 4.11. Plot of recovered data (top trace) from MED (bottom trace).

61 4.4 FSK Transmitter

The FSK transmitter was tested using a 27-pF, 10% tolerance surface mount capacitor and 1.16-μH inductor, which has a Q of 25 at the resulting oscillation frequency of 27.39 MHz. The inductor is 3 mm by 5 mm and was hand-wound with 47 turns of 34 gauge wire. The inductor parameters were characterized with an Agilent 4294A impedance analyzer. The transmitted signal was received using a Fairchild Electrometrics magnetic field probe, a high-Q loop antenna, and was displayed on a Agilent 4395A spectrum analyzer. Figure 4.12 depicts a received tone corresponding to f0, i.e. with the modulating input set to 0.

Fig. 4.12. Power spectral density of the FSK transmitter with modulating input

held low.

62 To measure the frequency deviation, the FSK transmitter was modulated using a 4-kHz binary square wave. The modulated spectrum is depicted in Figure

4.13. The modulated spectrum of the FSK signal contains primary components at f0 and f1 and sideband “spurs” that are separated by the modulating frequency.

The frequency deviation of the FSK transmitter is f0-f1 and the measured Δf is equal to 169 kHz, close to the desired value of 160 kHz.

Fig. 4.13. Power spectral density of the FSK transmitter when modulated by a 4-

kHz square wave.

FSK modulation using a 4-kHz square wave is equivalent to a reverse telemetry data rate of 8 kbps. The FSK transmitter will operate in the 27.12-MHz

ISM band so the available bandwidth can support a much larger data rate. Figure

63 4.14 shows a received signal that has been FSK-modulated using a 40-kHz square wave, corresponding to an 80-kbps data rate. The two central spikes are unaffected by the modulation, while the sideband spurs are now spaced by 40 kHz.

Fig. 4.14. Power spectral density for the FSK output when modulated with a 40-

kHz square wave.

The FSK transmitter was tested at various power levels using the on-chip adjustable bias network and a 7-pF active probe across the LC tank. In Figure

4.15, the oscillator is consuming 370 μW using a bias current of 137 μA, and is sustaining a 700-mVpp signal swing at the antenna. In Figure 4.16, the power consumption has been increased to 3 mW with a bias current of 685 μA, yielding

64 a voltage swing of 2.75 Vpp. The voltage swing could be larger, but in this case is limited by the supply voltage of 2.7 V. These results closely match the

approximation that VSWING = IBIASZTANK, where ZTANK = Rp = QLω0 = 5 kΩ.

Fig. 4.15. 700-mVpp sinusoidal oscillation sustained with 137 μA.

Fig. 4.16. 2.75-Vpp sinusoidal oscillation sustained with 685 μA.

65 The FSK transmitter efficiently uses a moderate level of power to achieve a high bit rate. The corresponding energy usage is 2.5 nJ/bit, slightly better than the 3 nJ/bit consumed by state-of-the-art micro-transmitters [20]. It is very difficult to do a direct comparison of transmitter efficiency since there are so many factors that affect power efficiency, including antenna size and transmission medium.

5 Conclusions & Future Work

This document has described the design and testing of a bidirectional transceiver for deeply implanted biomedical devices. A detailed analysis of system operating frequencies was performed, and issues relating to power, size, and complexity were considered. Communication via sub-100-MHz carriers reduces medium absorption and power consumption at the expense of slightly larger off-chip components. Forward- and reverse-telemetry on separate bands allows for full-duplex bidirectional communication. Forward-telemetry commands do not require large bit rates, so OOK at 125 kHz simplifies receiver design. Reverse-telemetry data is transmitted using FSK on a 27.12-MHz ISM band to obtain moderate bit rates.

The simple OOK receiver consists of two stages of amplification and a

Schmitt Trigger. An activity detector circuit demodulates the OOK signal to baseband and a Manchester decoder improves bit-error-rate. The FSK transmitter uses a negative-transconductance circuit to stabilize an LC tank and drive it into oscillation. A 1-pF capacitance is switched in parallel with the tank to create the FSK-modulated signal.

The transceiver circuits were fabricated in the AMI 0.5-μm process and benchtop testing has confirmed that circuits function. A trivial biasing error has led to larger-than-desired receiver gain which agrees with updated simulation results and hand calculations. The activity detector, Manchester decoder, and

67 FSK transmitter all function as designed. A summary of the measured results is compared to the design goals in Table 5-1.

Table 5-1: Comparison of Measured Results to Design Goals

Design Goal Measured Frequency 125 kHz 125 kHz Bit Rate 4 kbps 4 kbps OOK Sensitivity 1 mV 50 mVpp Receiver Energy / Bit <50 nJ 25 nJ Power <100 μW 50 μW

Frequency 27.12 MHz 27.12 MHz Bit Rate 80 kbps 80 kbps FSK Output Power Variable 1.6 Vpp Transmitter Energy / Bit 2.5 nJ 2.5 nJ Power < 1 mW 1 mW

Schmitt trigger testing revealed several issues that were not predicted in simulation. The measured Schmitt trigger thresholds are lower than simulation results, and this could be due to poor modeling of MOS behavior in moderate inversion. Moreover, there is a seemingly random input-referred offset that shifts the common-mode level of the Schmitt thresholds. This offset could be due to test limitations, since the amplifier outputs must be overpowered to test the

Schmitt trigger. Finally, the switching output of the Schmitt trigger is coupling to its input, creating temporary oscillation when its input is near zero. This “chatter” is not present when the amplifier is driven using amplitudes > 50 mV.

Future work will include small adjustments to the OOK receiver circuits such that all design specifications are met. Proper layout of the differential amplifier bias network will ensure that amplifier gain matches simulation results.

68 Feedback biasing of A2 will reduce the output offset, and AC-coupling to the

Schmitt trigger inputs will eliminate low-frequency common-mode drift. The

Schmitt trigger modeling oscillations require further study. Improved SPICE models may lead to an accurate prediction of the Schmitt trigger thresholds, although the random variation requires more study. The problem may be solved via additional buffering, or by reducing the rise time of the Schmitt trigger output.

69 6 References

[1] B. W. Cook, A. Berny, A. Molnar, S. Lanzisera, K. S. J. Pister, “Low- Power 2.4-GHz Transceiver With Passive RX Front-End and 400-mV Supply.” IEEE Journal of Solid-State Circuits, Vol. 41, No. 12, Dec. 2006.

[2] R. R. Harrison, P. T. Watkins, R. J. Kier, R. O. Lovejoy, D. J. Black, B. Greger, F. Solzbacher, “A Low-Power Integrated Circuit for a Wireless 100-Electrode Neural Recording System.” IEEE Journal of Solid State Circuits, Vol. 42, No. 1, Jan. 2007.

[3] J. H. Schulman, P. Mobley, J. Wolfe, H. Stover, A. Krag, “A 1000+ Channel Bionic Communication System.” Proceedings of the 28th IEEE EMBS Annual International Conference, New York City, USA, Aug. 30 – Sept. 3, 2006.

[4] C. Park, P. H., Chou “Eco: Ultra-Wearable and Expandable Wireless Sensor Platform.” Proceedings of the International Workshop on Wearable and Implantable Body Sensor Networks, San Francisco, USA, June 26, 2006.

[5] SmartPill Corporation, “Capsule-based Technology for GI Healthcare.” Accessed at September 19, 2007.

[6] B. Chi, J. Yao, S. Han, X. Xie, G. Li, and Z. Wang, “Low-Power Transceiver Analog Front-End Circuits for Bidirectional High Data Rate Wireless Telemetry in Medical Endoscopy Applications.” IEEE Transactions on Biomedical Engineering, Vol. 54, No. 7, July 2007.

[7] Federal Communications Commission Office of Engineering and Technology Policy and Rules Division, “FCC Online Table of Frequency Allocations.” Federal Communications Commission, May 4, 2007.

[8] Federal Communications Commission Office of Engineering and Technology, “Questions and Answers about Biological Effects and Potential Hazards of Radiofrequency Electromagnetic Fields.” OET Bulletin 56, Fourth Edition, August, 1999.

[9] R. T. Howe, C. G. Sodini, “Microelectronics: An Integrated Approach.” Prentice Hall Electronics and VLSI Series, Upper Saddle River, NJ, 1997.

[10] G. A. Ybarra, C. E. Cramer, “Lab 10: Frequency Response of Filter Circuits.” Agilent Technologies, accessed online at , September 19, 2007.

[11] P. R. Gray, P. J. Hurst, S. L. Lewis, R. G. Meyer, “Analysis and Design of Analog Integrated Circuits: Fourth Edition.” John Wiley & Sons, Inc., 2001.

[12] R. Kothari, and S. L. Garverick, private conversation.

[13] B. P. Lathi, “Modern Digital and Analog Communication Systems, Third Edition.” Oxford University Press, New York, NY, 1998.

[14] K. Arabi, M. A. Sawan, “Electronic Design of a Multichannel Programmable Implant for Neuromuscular Electrical Stimulation.” IEEE Transactions on Rehabilitation Engineering, Vol. 7, No. 2, June 1999.

[15] M. Neihart, R. R. Harrison, “A Low-Power FM Transmitter for Use in Neural Recording Applications.” Proceedings of the 26th Annual International Conference of the IEEE EMBS, San Francisco, CA, USA, Sept. 1-5, 2004.

[16] T. H. Lee, A. Hajimiri, “Oscillator Phase Noise: A Tutorial.” IEEE Journal of Solid-State Circuits, Vol. 35, No. 3, Mar. 2000.

[17] H. Mostafa, M. N. El-Gamal, R. A. Rafla, “A Sub-1-V 4-GHz CMOS VCO and a 12.5-GHz Oscillator for Low-Voltage and High-Frequency Applications.” IEEE Transactions on Circuits and Systems—II: Analog and Digital Signal Processing, Vol. 48, No. 10, Oct. 2001.

[18] F. Svelto, S. Deantoni, R. Castello, “A 1.3 GHz Low-Phase Noise Fully Tunable CMOS LC VCO.” IEEE Journal of Solid State Circuits, Vol. 35, No. 3, Mar. 2000.

[19] A. Hajimiri, T. H. Lee, “Design Issues in CMOS Differential LC Oscillators.” IEEE Journal of Solid-State Circuits, Vol. 34, No. 5, May 1999.

[20] B. W. Cook, A. Berny, A. Molnar, S. Lanzisera, K. S. J. Pister, “Low- Power 2.4-GHz Transceiver With Passive RX Front-End and 400-mV Supply.” IEEE Journal of Solid-State Circuits, Vol. 41, No. 12, Dec. 2006.