Towards Long-Range Backscatter Communication with Tunnel Diode Reflection Amplifiers

UPTEC F 18039 Examensarbete 30 hp Juni 2018

Gustav Eriksson Abstract Towards Long-Range Backscatter Communication with Tunnel Diode Reflection Amplifiers Gustav Eriksson

Teknisk- naturvetenskaplig fakultet UTH-enheten Backscatter communication enables wireless communication at a power consumption orders of magnitude lower than conventional wireless Besöksadress: communication. Instead of generating new RF-signals backscatter Ångströmlaboratoriet Lägerhyddsvägen 1 communication leverages ambient signals, such as WiFi-, Bluetooth- Hus 4, Plan 0 or TV-signals, and reflects them by changing the impedance of the antenna. Backscatter communication is known as a short-range Postadress: communication technique achieving ranges in the order of meters. Box 536 751 21 Uppsala To improve the communication range, we explore the use of a tunnel diode as an amplifier of the backscattered RF-signal. We developed Telefon: the amplifier on a PCB-board together with a matching network tuned 018 – 471 30 03 to give maximum gain at 868 MHz. Our work demonstrates that the

Telefax: 1N3712 tunnel diode can achieve gains up to 35 dB compared to a tag 018 – 471 30 00 without amplification while having a peak power consumption of 48 µW. With this amplifier the communication distance can be increased by up Hemsida: to two orders of magnitude. http://www.teknat.uu.se/student

Handledare: Ambuj Varshney Ämnesgranskare: Dragos Dancila Examinator: Tomas Nyberg ISSN: 1401-5757, UPTEC F18039 Tryckt av: UPPSALA Popularvetenskaplig¨ Sammanfattning

Utbudet av tradl˚ osa¨ apparater, sa˚ som tradl˚ osa¨ horlurar,¨ mobiltelefoner, smarta klockor och sensorer okar¨ lavinartat i samhallet.¨ I hemmen skapas allt storre¨ hogar¨ av laddningskablar som vi far˚ anvanda¨ allt oare nar¨ baerierna i apparaterna blir allt samre¨ med tiden. Men alla som besokt¨ e modernt bib- liotek under senare tid, kopt¨ e ny kladesplagg,¨ anvant¨ e passerkort eller rest med e ny EU-pass kanske vet a information kan skickas kortare strackor¨ utan anvandningen¨ av baerier. Denna teknik kallas radiofrekvensidentiering (RFID). I denna studie undersoks¨ mojligheten¨ a anvanda¨ RFID for¨ a kommunicera pa˚ langre¨ distanser, med sye a minska eektforbrukningen¨ i e baeri till en tusendel av dagens forbrukning,¨ eller helt ta bort baeriet i framtida tradl˚ osa¨ apparater. Utan dagens stora baeri skulle morgondagens tradl˚ osa¨ apparater kunna bli mindre, laare¨ och smidigare a anvanda¨ da˚ de inte skulle vara beroende av a behova¨ laddas.

E baerilost,¨ passivt, RFID-system bestar˚ av en lasare¨ och en transponder (tagg). Lasaren¨ ar¨ oast en storre¨ apparat vars uppgi ar¨ a ta emot och sanda¨ barsignaler¨ till taggen, som skickar tillbaka informationen till lasaren.¨ Taggen skickar informationen genom a sprida tillbaka barsignalen¨ pa˚ e kontrollerat och eektsnalt˚ sa.¨ Dea kan liknas vid en spegel som reekterar synligt ljus, men istallet¨ for¨ en spegel anvander¨ taggen en antenn och istallet¨ for¨ synligt ljus tillbakasprider antennen RF-signaler. Skillnaden mellan hur taggen skickar information, genom tillbakaspridning, och hur exempelvis mobiltelefoner skickar information ar¨ avgorande¨ for¨ hur mobiltelefoner och andra tradl˚ osa¨ apparater ska kunna kommunicera mer energieektivt. De esta tradl˚ osa¨ apparater som anvands¨ idag maste˚ generera egna RF-signaler som skickas till moagaren. A generera dessa RF-signaler kraver¨ hog¨ eekt och ar¨ e av de primara¨ syena till varfor¨ baerier fortfarande anvandas¨ idag. I framtiden vill forskarna ta tillvara pa˚ redan bentliga RF-signaler som oa nns runtomkring oss, som exempelvis WiFi-signaler eller FM-signaler, och skicka information genom a tillbakasprida dessa istallet¨ for¨ a behova¨ generera nya RF-signaler.

En tagg bestar˚ av en antenn och e kretskort. Kretskortet kraver¨ valdigt¨ lag˚ eekt for¨ a drivas, och kan i e passivt RFID-system drivas av energi fran˚ barsignalen.¨ For¨ a mojligg¨ ora¨ langdistanskommunikation˚ har forskare utvecklat den semipassiva taggen, vilket bygger pa˚ a det interna kretskortet drivs av en ex- tern kalla¨ som till exempel e baeri eller fotodiod. Skillnaden mot de tradl˚ osa¨ apparater som nns idag ar¨ a en semipassiv tagg kan drivas pa˚ eekter i storleksordningen µW, medan till exempel mobiltelefoner kraver¨ eekter i storleksordningen mW for¨ a generera RF-signaler. Den laga˚ eektforbrukningen¨ forl¨ anger¨ baeritiden for¨ den semipassive taggen jamf¨ ort¨ med baeritiden i dagens tradl˚ osa¨ apparater.

I denna studie undersoks¨ mojligheten¨ a forl¨ anga¨ kommunikationsdistansen mellan en tagg och en lasare¨ genom a implementera en forst¨ arkare¨ i taggen. De senaste aren˚ har tunneldioden visat sig kunna ge hog¨ forst¨ arkning¨ vid lag˚ eektforbrukning,¨ och darf¨ or¨ anvands¨ tunneldioden i denna studie. Vid laga˚ spanningar¨ har tunneldioden en negativ resistans, vilket kan forst¨ arka¨ en inkommande RF-signal. Eer- som tunneldioden inte tillverkas langre¨ idag, anvandes¨ en gammal tunneldiod (1N3714) som tillverkades av General Electrics pa˚ 60- eller 70-talet. Tillbakaspridningsmatningar¨ och oscilloskopmatningar¨ visar a forst¨ arkartaggen¨ kan forst¨ arka¨ en RF-signal 35 dB jamf¨ ort¨ med en oforst¨ arkt¨ tagg, samtidigt som forst¨ arkar-taggen¨ forbrukar¨ 48 µW. Vid distansberakningar¨ visar forst¨ arkartaggen¨ en god form¨ aga˚ a forst¨ arka¨ svaga signaler, och oka¨ kommunikationsdistansen med upp till 58 ganger˚ vid barsignaler¨ pa˚ -110 dBm, och dubbla kommunikationsdistansen vid barsignaler¨ pa˚ -50 dBm. Vid -110 dBm blir kommunikationsdistansen 10 m och for¨ -50 dBm blir kommunikationsdistansen 420 m.

Forst¨ arkartaggen¨ byggs av kopparremsor som limmas pa˚ en PCB-plaa tillsammans med en DC-block- kondensator (67 pF) och en AC-blockspole (47 µH). For¨ a hia den spanning¨ dar¨ tunneldioden fungerar

1 som forst¨ arkare¨ gors¨ en strom-sp¨ anningskurva¨ med hjalp¨ av DC-kalla¨ och tva˚ multimetrar for¨ a mata¨ strom¨ och spanning.¨ For¨ a stalla¨ in forst¨ arkaren¨ pa˚ a forst¨ arka¨ signaler med frekvensen 868 MHz anvands¨ e matchande natverk.¨ Forst¨ arkartaggen¨ stalls¨ in a ge forst¨ arkning¨ pa˚ ra¨ frekvens genom anvandningen¨ av en vektornatverksanalysator¨ (VNA). Matningar¨ visar a tunneldiodens egenskaper for-¨ andras¨ vid olika RF-signalstyrkor, vilket medfor¨ a det matchande natverket¨ fungerar bast¨ i e in- tervall av barsignalstyrkor.¨ Dea ar¨ anledningen till a forst¨ arkartaggen¨ ger en forst¨ arkning¨ pa˚ pa˚ 35 dB jamf¨ ort¨ med en oforst¨ arkt¨ tagg vid sma˚ RF-signalerstyrkor.

Tunneldioden visar i denna studie en god form¨ aga˚ a framfor¨ allt forst¨ arka¨ svaga RF-signaler. Vid svaga barsignaler¨ kan en tagg med en tunneldiod av typ 1N3714 ge en maximal kommunikationsdistans pa˚ 10 m, vilket ar¨ 58 ganger˚ langre¨ an¨ for¨ en tag utan forst¨ arkare.¨ Aven¨ om 10m ej racker¨ for¨ langdistans-˚ kommunikation, racker¨ det val¨ for¨ manga˚ inomhusapplikationer. Om egenskaperna hos det matchande natverket¨ kan konstrueras pa˚ e sa¨ sa˚ a forst¨ arkningen¨ pa˚ 35 dB sker vid starkare RF-signaler skulle tunneldioden kunna mojligg¨ ora¨ langdistanskommunikation˚ vid ultralaga˚ eektforbrukningar.¨

2 Preface

is is a 30 ECTS master thesis which concludes my program in Engineering Physics, 300 ECTS, at Up- psala University. is master work was done at the department of Information Technology, at Uppsala University together with the Uppsala Networked Objects (UNO) group, where I had my supervisor Ambuj Varshney. My subject reviewer was Dragos Dancila from the division of Solid-State Electronics, and my examiner was Tomas Nyberg from the division of Solid-State Electronics. is project has given me a deeper insight in the interesting research eld of backscaer communication, which I believe will play an important part in the way to communicate between wireless devices in the future.

I would like to thank Ambuj Varshney for all the encouragement, time and belief that he has given during the whole thesis. anks to him I have had the opportunity to both improve my project and engineering skills in the most interesting and fun way. I would also like to thank the whole UNO-group and especially Professor iemo Voigt and Professor Christian Rohner for the warm welcome to the department and for all the help that I have goen during the whole thesis. I would like to thank Dragos Dancila for the help, and the opportunity for me to present my master thesis during the conference: Swedish Microwave Days, in Lund.

Finally, I would like to thank my partner, friends and family for their encouragement and support during these months.

LIST OF ABBREVIATIONS

ABT: Amplied Backscaer Tag AC: Alternating Current ADS: Advanced Design System DC: Direct Current

FET Field-Eect Transistor IoT: Internet of ings MESFET: MEtalSemiconductor Field Eect Transistor pHEMT: pseudomorphic High Electron Mobility Transistor RF: Radio Frequency RFID: Radio Frequency IDentication UBT: Unamplied Backscaer Tag

VNA: Vector Network Analyzor

KEYWORDS

Amplied backscaer tag, Reection amplier, Baery-free, Long-range communication, Tunnel diode, Ultra-low power consumption, RFID, Frequency shi.

3 Contents

1 Introduction 5

2 Background 6 2.1 Radio Frequency Identication...... 6 2.2 Monostatic and Bistatic setup...... 7 2.3 Mixing Property of the Backscaer Communication...... 8 2.4 Related work...... 9 2.5 Diodes...... 11 2.5.1 PN-Junction...... 11 2.5.2 Tunnel Diode...... 11

3 Design of Amplied Backscatter Tag 12 3.1 Motivation to Amplication at the Backscaer Tag...... 13 3.2 Gain with Amplied Backscaer Tag...... 14 3.3 State of the Art for Reection Ampliers...... 15 3.4 Tuning the Backscaer Tag to Desired Frequency...... 16 3.5 Characterization of Tunnel Diodes...... 18 3.5.1 General Electric TD104...... 18 3.5.2 Russian Ai301A...... 21 3.5.3 General Electric 1N3712...... 23 3.6 Amplied Backscaer Tag Design...... 25

4 Evaluation 26 4.1 Gain Achieved Measured with a VNA...... 28 4.2 Power Consumption of the ABT...... 29 4.3 Placing Tag between Source and Receiver...... 30 4.4 Indoor Environment...... 31

5 Discussion 33

6 Conclusion 35

7 Future Work 36

8 Appendix 40 8.1 Communication distance at Dierent Input Powers...... 40 8.2 Denition of gain...... 40 8.3 Distance Calculations...... 41 8.4 Indoor Measurement Calculations...... 43 8.5 Anechoic Chamber Calculations...... 43

4 1 Introduction

To send information is an important part of today’s society. Every day people receive and transmit information through smartphones, televisions or computers. In the last decades the ways to communicate have extended from being only between humans to nowadays also include electronic devices. Information can be sent wirelessly by encoding information into electromagnetic waves which are sent and received by antennas. Today we are standing at the edge of a new era that is called Internet of ings (IoT). e amount of connected IoT-devices worldwide is predicted to increase from 14.9 billion devices in 2016 to 82 billion devices in 2025 [1]. Researchers and entrepreneurs are talking about placing sensors in almost any places to e.g. improve security, enable smart energy management or predict ooding [2]. To power these sensors and future IoT-devices with baeries is not sustainable in terms of power consumption and maintenance cost. erefore, researchers are trying to develop a new way to communicate which is a thousands times more energy ecient than present techniques [3][4][5].

Nowadays most of the wireless communication between devices is through WiFi or Bluetooth. ese devices most likely have a baery as their source of power which can take up a signicant part of the total volume of the device. e capacity of a baery is proportional to its volume [6], and the amount of power that is needed in order to both receive and transmit WiFi-signals is in the order of W [7]. e high power consumption of generating radio signals is one reason why cellphones in the 1990s, which did not use WiFi-communication, had a longer life time compared to the smartphones used today. e high amount of power that is needed to create these RF-signals has been the boleneck in the develop- ment of IoT [8]. Eort is now paid in trying to improve present RF-technique to enable long distance communication in a more power ecient way, by using a technique called backscaer communication. Backscaer communication is based on the same concept as a mirror that reects visible light, but instead of a mirror it is an antenna that reects RF-waves. e advantage of backscaer communication is that a signal does not have to be generated by the device, instead the signal can be scaered back in a controlled way in order to send information. By using backscaer communication, present and new devices, like cellphones [9] and cameras [10], can be made simpler and lighter by removing complex and power hungry components used today. In this way communication can be made by using power in the order of µW[11] instead of in the order of mW that is used today.

Today backscaer communication is widely used in passive radio frequency identication (RFID). is technique is used in passport cards, keycards, tags on new cloths in stores and in some libraries instead of the traditional bar-code [12]. ese devices are using passive components and therefore do not need any external power to communicate. e disadvantage of using passive RFID is that the communication distance is limited to meters, and is therefore not suited for long distance communication. Recent systems improve upon existing passive RFID-systems and demonstrate a communication range of several kilometres [3]. However, these systems require a radio frequency source to be co-located with the backscaer tag in proximity which impacts practical application.

e objective in this thesis is to improve backscaer communication range by using a tunnel diode as an amplier at the tag. e goals is to characterize three tunnel diodes to nd a tunnel diode that can provide high gain at a power consumption below 100 µW. e nal measurements is done with a 1N3712 tunnel diode manufactured by General Electrics in 1960s/1970s [13], which achieves a 35 dB gain at a peak power consumption of 48 µW.

roughout this thesis, we use the notation where we call the backscaer tag with the reection amplier Amplied Backscaer Tag (ABT) and we call the backscaer tag without reection amplier Unam- plied Backscaer Tag (UBT). A collective name for both ABT and UBT will be tag.

5 2 Background

Wireless communication is information sent by devices without having them directly connected by using e.g. sound, infrared, optical or radio frequency energy [14]. Today radio frequency (RF) signals are oen used, which according to the International T elecommunication Union includes frequencies that are used in telecommunication, and covers frequencies from 300 kHz to 300 GHz [15]. e advantage of using RF-signals, compared to other type of communication signals, is that RF-signals have a wide bandwidth and can penetrate e.g. dust, fog, buildings and cars [14], which makes them suitable for communication.

is section will give the reader an introduction to RFID-systems, the monostatic- and bistatic setup, frequency shiing, general backscaer communication work and an introduction to pn-junctions and tunnel diodes.

2.1 Radio Frequency Identication Radio frequency identication (RFID) is a technique which leverage radio frequency signals to transmit data encoded into electromagnetic waves between readers and transponders (tags). A tag can be put on an object to be e.g. tracked, identied or managed by utilizing RF-signals [16]. e main components of a typical RFID system is a tag and a reader, where the reader is usually connected to a computer system [17] and the tag is the transmier of the wanted information. A schematic picture of a RFID-system is shown in Figure1. At a high level the tag communicate with the reader by either reecting an incoming carrier signal from the reader or by generating an own RF-signal, at the device, which is transmied to the reader. At the reader the information is decoded and can be sent to a computer system. e main components of a tag is an antenna and an integrated circuit. e integrated circuit contains power conversion circuits, data storage, and controls the logic in the tag [16]. Gen- erally there are two dierent types of tags, active and passive tags [17] and the classication is based on the way they transmit information. An active tag has its own source of power, e.g. a baery, to generate RF-signals at the device, which are sent to the reader. A passive tag harvest energy from the carrier signal which is used to backscaer the carrier signal in a controlled way back to the reader [16]. Backscaer communication is based on the same principle as the heliography that was developed in the 19th century, which was used to send ash signals by reecting sunlight with a set of mirrors [18]. But instead of visible light, RF-signals are used and instead of mirrors an antenna is used at the tag.

Reader Tag

Figure 1: Monostatic (RFID) system. e reader processes the message received by the tag.

Active tags can generally transmit signals over greater distances, store more data, but are baery operated and usually more expensive compared to passive tags. e advantage of passive tags is that they do not have any internal power sources and therefore passive tags need less maintenance compared to active tags [16].

6 In the last decade a third category of tags has been developed, the semi-passive tag. Semi-passive tags uses energy from baeries to power the integrated circuit, like active tags, but transmits messages by backscaer carrier signals, like passive tags [19]. By using baery power only to power the integrated circuit, the power consumption can be kept in the order of µW while the backscaer communication distance can be increased by several orders of magnitude compared to that for passive tags [3]. e reduced power consumption results in an extended life time of baeries used in a semi-passive tag compared to baeries used in an active tag.

2.2 Monostatic and Bistatic setup In passive RFID, the reader both transmits carrier signals and receives backscaered signals from the tag. is is called a monostatic setup, and is shown in Figure1. e advantage of the monostatic setup is that it is conceptually simple since one device, the reader, both transmits the carrier signal and receives the backscaered signal. e disadvantage with the monostatic setup is that it only allows communication between a reader and a tag. In a bistatic communication system, as is shown in Figure2, the reader is divided into a carrier generator (transmiing part) and a receiver (receiving part) [20]. One advantage with the bistatic setup is that the receiver does not necessarily have to be located at the same position as the carrier generator, which aects the power received at the receiver, as is shown in Figure3. In Figure 3a it is seen that in the monostatic setup the power received at the receiver from the tag is highest when the tag is in close proximity to the reader and decreases as the tag is moved away from the reader. In Figure 3b it is seen that in the bistatic setup the tag can be placed either in close proximity to the carrier generator or the receiver for the receiver to receive a high power backscaered signal from the tag. e extended eld coverage in the bistatic setup, compared to the monostatic setup, can be used in large scale, low-cost and low power sensor networks [20]. Varshney et al. demonstrated in [3] that by placing a semi-passive tag close to the carrier generator, a communication distance of 3.4 km between the tag and the receiver can be achieved.

Gt P r Gr t 1

Carrier Pr

Grt Tag r P 2 rt Receiver

Figure 2: Bistatic setup. e reader is divided into a carrier generator and a receiver.

7 Reader Location Received signal power [dBm]

Distance to tag [m]

(a) Monostatic setup

Carrier Location Receiver Location Received signal power [dBm]

Distance to tag [m]

(b) Bistatic setup

Figure 3: Dierence between monostatic and bistatic setup. In a monostatic setup the tag has to be placed in close proximity to the reader, while in a bistatic setup the tag can be placed in close proximity to either the carrier generator or the receiver in order for the receiver to receive high power.

2.3 Mixing Property of the Backscatter Communication Tags, in passive RFID, are designed to work with an extremely low power consumption. is is possible because the tag can switch the impedance of the antenna by the use of a low power consuming transistor. Commercial readers on the other hand consumes power in the order of 102 mW - 103 mW, which is primarily a reason because it needs to decode the weak backscaered signal from the stronger self- interfered carrier signal [6], and because it needs to generate the RF-signal. is means that the total power consumption in a monostatic system is high, despite the low power consumption of the tag. To solve the self-interference problem, tags can shi the backscaered signal from the frequency of the carrier signal to a nearby frequency band. e backscaered signal is frequency shied through modulation of the radar cross section of the tag antenna, which multiplies the incoming signal with the

8 modulated signal [21]. If the carrier signal, Sc, is assumed to be a perfect sinusoidal signal

Sc = sin(2πf0t) (1) and the, modulation signal, Stag, is a square wave wrien as its Fourier transformation

∞ 4 X 1 S = sin(2π∆ft) (2) tag π n n,odd the backscaered signal, B(t), is given by

∞ 2 X 1 B(t) = S × S = (cos(2π(f − n∆f)t) − cos(2π(f + n∆f)t)) (3) c tag π n 0 0 n,odd where f0 ± n∆f is the frequencies of the double sided backscaered signal. Equation3 describes the 1 1 backscaered signal as a set of harmonics where the third harmonic is reduced by 3 and the h by 5 from the rst harmonic. erefore Equation3 can be simplied to 1 B(t) ≈ (cos(2πt(f − ∆f)) − cos(2πt(f + ∆f))) (4) 2 0 0 which describes the backscaered signal and its mirror copy at the frequencies ±∆f away from the carrier signal, at the frequency f0. When the receiver receives a frequency shied signal, it sees a clean, low noise level, signal and does not have to separate the backscaered signal from the self-interfered carrier signal, which increases the performance of backscaer communication compared to when systems which have to use self- interference cancellation techniques are used [22] A disadvantage with double sided backscaered signal that is the unwanted mirror copy at the frequency −∆f can create unwanted interferences in other frequency bands which can disturb wireless transmissions in those bands. Zhang et al. presents in [23] a way to reduce the double sided backscaer signal into a single sided backscaer signal. eir solution are to rst split the incident carrier signal into two separate signals, apply a phase shi to one of the signals and then add them together, with the result that the cos(2π(f0 − n∆f)t)-term is cancelled. To get a single sided backscaered signal is out of the scope of this thesis. e choice of ∆f can be more or less general. But the choice of ∆f should be enough to make the backscaered signal not interfere with the carrier signal, but small enough because the power consumption of the tag increases with the frequency shi, ∆f [22]. A screenshot of the double sided backscaered signal together with the carrier signal, taken on a spectrum analyser, is shown in Figure 4. e impedance of the tag is changed at a rate of 250 kHz and thus the backscaered signal is moved by 250 kHz away from the carrier signal at 868 MHz.

2.4 Related work e bistatic setup can enable tag-to-tag communication, but is still in need of a carrier signals to send information. Instead of a separate carrier generator, Lui et al. demonstrates in [4] how ambient RF- signals can be used to both power the tag and to work as the carrier signal. is can enable tag-to- tag communication in an power ecient way, without the need of the carrier generator. e already existing RF-signals can for example be WiFi-signals [21], Bluetooth signals [24] or TV signals [4]. A typical ambient backscaer communication system consists of two tags, Alice (A) and Bob (B), and an already existing RF-antenna, as is shown in Figure5. At a high level an ambient communication system works as follows. In the case of a TV-antenna, it constantly sends RF-signals to a TV with a frequency f0. e two tags, Alice and Bob, also receives the RF-signal, with a frequency f0, from the antenna.

9 Figure 4: Frequency shi of backscaered signal. e screenshot is taken on a spectrum analyzer where the backscaered signal (2) is shied 250 kHz away from the carrier signal (1) at 868 MHz.

Alice can transmit a signal by backscaer the TV-signal to Bob. If the backscaered signal is frequency shied, self-interference between the stronger TV-signal and the weaker backscaered signal is avoided at Bob. erefore, a frequency shied backscaered signal allows power ecient ambient backscaer communication between tags.

RF-antenna

f 0 f0 f0

Bob Alice

f0 f

Figure 5: Tag-to-tag communication. Alice sends a frequency shied signal to Bob to prevent self- interference with the carrier signal. Alice and Bob are passive tags that communicate by reecting ambient wireless signal such as television signals.

Kotaru et al. demonstrates in [25] how ambient WiFi-signals can be used to localize low power backscaer tags with an localization error or 1 m - 1.5 m. e tag backscaer the information to a nearby smartphone where the location of the tag can be seen. An idea made by the authors is that this technique can be used by elders to nd e.g. pill boles at home. Naderiparizi et al. demonstrates in [10] a wearable camera that can steam HD-video to a smartphone by using analogue backscaer communication. e authors thinks that this could provide light weight, high quality video streaming which could be implemented in e.g. glasses.

10 Varshney et al. demonstrates in [26] how backscaer communication can be used to detect hand gestures or presence of people up to a distance of 330 m at a power consumption of 20 µW. e authors demonstrates that the ultra-low power of 20 µW can be received by the use of solar cells operating in diverse light conditions between 100 lx - 80 klx.

2.5 Diodes Diodes. or rectiers, have current rectifying characteristics for an applied bias voltage, and are common components in solar cells and transistors [27]. By changing the doping level of a diode, the current- voltage (IV) characteristics of a diode is changed. rough very high doping levels, Leo Esaki invented the tunnel diode, or Esaki diode, in 1957 which uses quantum tunnelling to conduct current [28].

2.5.1 PN-Junction A pn-junction is a n-semiconductor and a p-semiconductor merged together. When electrons from the n-region are diused into the p-region and holes from the p-region are diused into the n-region, a region of no electrons and holes are created between the n-semiconductor side and the p-semiconductor side, called the depletion region. e localize charge densities separated by the depletion region give rise to an electric potential inside the depletion region. is potential acts as a barrier to the movement of electrons and holes across the junction. Only electrons with sucient energy can overcome the potential energy and be able to move from the n-side to the p-side. An applied bias voltage across the junction changes the height of the potential barrier, which aects how easily charges carrier can move across the depletion region. When a forward bias voltage is applied the potential barrier is reduced and electrons are allowed to ow easily across the junction. When a reversed bias voltage is applied the height of the potential barrier is increased and almost no electrons can move across the junction.

2.5.2 Tunnel Diode A tunnel diode is a pn-junction with a doping level about 1000 times higher, at both the n- and p-side of the junction, compared to other diodes which results in a very thin, (≈ 100 A˚ m), depletion region [29]. According to quantum mechanics there is a nite probability that an electron can tunnel trough a potential barrier without losing energy. is is the process that governs the ow of current in a tunnel diode. At zero bias voltage across the tunnel diode, the net tunnelling between either side of the pn-junction i zero, resulting in not net ow of current [30], as is shown in Figure 6a. At small forward bias voltages the potential at the p-side decreases compared to the n-side. Now the conduction band electrons on the n-side are opposite empty states in the valance band on the p-side and valence electrons on the p-side are opposite the forbidden region on the n-side [30]. e result is a net ow of forward current through the junction, as is shown in Figure 6b. At forward voltages higher than the peak voltage, some of the electrons in the conduction band on the n-side starts to oppose the forbidden region on the p-side, which results in a decrease in the quantum tunnelling process and the net ow of current decreases [30]. A decease in current for a higher voltage, is called negative resistance. antum tunnelling stops when the forward bias voltage reaches the valley voltage, as is shown in Figure 6c. At voltages higher than the valley voltage the hight of the barrier is reduced to enable the normal current to ow [30]. Now the tunnel diode acts as a regular diode where the current increases exponentially with the voltage, as is shown in Figure 6d. A tunnel diode can be used in several applications. It has shown great performance in RF-energy harvesting because of its low power consumption [31] and that it can be used as an ecient Schoky diode [32]. A tunnel diode biased in the negative resistance region can work as an amplier to an incoming RF-signal, and thus work as a gain element in communication-type circuits [33].

11 Holes in Valence Band Free Electrons in - Free Electrons in - Conduction Band Holes in Valence Band Conduction Band - Free Electrons in Conduction Band Holes in Valence Band

n-type p-type n-type p-type

p-type n-type Current(I) Current(I) Current(I)

Voltage(v) Voltage(v) Voltage(v)

(a) No bias voltage (b) Current up to peak value (c) Current down to valley-value

- Free Electrons in Conduction Band

Holes in Valence Band

p-type n-type Current (I)

Voltage (V)

(d) Exponential increase

Figure 6: Important states for the bias voltage across a tunnel diode. e characteristic relationship between the current and voltage are due to quantum tunnelling of charge carriers across the pn-junction.

3 Design of Amplied Backscatter Tag

A general description of dierent RFID-systems and how long distance backscaer communication can be achieved with the use of the bistatic setup and the frequency shi of the backscaered signal has so far been given. Another way to increase communication distance is to implement an amplier in the tag, to increase the power of the backscaered signal. In this section a motivation to why amplied backscaer tags (ABT) should be used instead of unamplied backscaer tags (UBT) to reach long distance communication, based on the backscaer link budget equation. Dierent ampliers are compared in terms of gain and power consumption, with the conclusion that tunnel diodes will be used in this thesis. A characterization of three dierent tunnel diodes based on gain and power consumption are made with the use of a vector network analyser (VNA). A description of how to design the ABT and how to achieve gain at specic frequency by the use of a matching network is presented.

12 3.1 Motivation to Amplication at the Backscatter Tag e goal with an ABT is to increase the power of the backscaered signal to increase the communication distance between tags and receivers. e power, Pr, received at a tag from a carrier generator is given by the Friis equation [34] λ 2 Pr = PtGtGr( ) (5) 4πr1 where Pt is the power transmied from the carrier generator, Gt and Gr are the antenna gains at the carrier generator and at the tag respectively, λ is the wavelength of the RF-signal and r1 the distance between the carrier generator and the tag. e gain, G, of an antenna is given by

G = ηradD (6) where ηrad is the radiation eciency and D is the directivity. e radiation eciency models the antenna losses and the directivity models the radiation paern of an antenna. e power that is received at a receiver, Prt, from a tag is given by the backscaer link budget equation

2 2 λ 2 Prt = PtGtGr( 2 2 ) M (7) 4 π r1r2 where r2 is the distance between the tag and the receiver (r1 = r2 for monostatic setup) and M is the modulation factor which is a measurement of how ecient a tag can generate backscaered signal [21]. More about the modulation factor in Section 3.2.

Varshney et al. demonstrates in [3] how a 28 dBm carrier signal can be backscaered 3.4 km with an UBT located 1 m away from the carrier generator. For a similar experiment, but instead with an ABT, the power transmied by the tag would, in units of dBm, be

dBm dBm PABT = X + PUBT (8)

dBm dBm where PABT is the power transmied by the ABT, PUBT the power transmied by the UBT and X the gain achieved by the ABT. e received signal strength at the receiver in [3] can be calculated from the backscaer link budget equation to be -93 dBm. By assuming the same signal strength at the receiver when using the ABT an expression of the distance, r2, between the ABT and the receiver can be derived from the Friis equation, Equation5, to be

−93−X−P −G −G c − tag t r r = 10 20 (9) 2 4πf where Ptag = 2 dBm is the power received at the tag. How the distance varies with the gain X, at the frequency 868 MHz, are shown in Figure7. It can be seen that an ABT extend the backscaer communication distance by several orders of magnitude compared to an UBT. If this can be done at an ultra-low powers, e.g.below 100 µW, this would be a step towards practical use of long distance backscaer communication.

13 106

105

104

103 [km] 2 r

102

101

0 10 20 30 40 50 60 70 80 90 100 Gain [dB]

Figure 7: Communication distance as a function of gain based on results from [3]. e communication distance increases exponentially with achieved gain at the ABT. is is if the ABT can provide gain at a carrier signal strength of 2 dBm.

3.2 Gain with Amplied Backscatter Tag Mathematically an ABT achieves gain by changing the modulation factor, M, in Equation7, which is given by the dierence in the reection coecients between two operation points as 1 M = |Γ − Γ |2 (10) 4 A B

where ΓA is the reection coecient of state A and ΓB is the reection coecient of state B. In this case the reection coecient is a measurement of how well the impedance of the integrated circuit is matched to the impedance of the antenna at the tag. More about this in section 3.4. For a given backscaer system the power received at the receiver is

4 λ 2 Prt ∝ ( 2 2 )|ΓA − ΓB| (11) r1r2

where either the wavelength λ, the distances r1, r2 or the modulation factor M can be changed to aect the power received at the receiver. For a given wavelength and distances between carrier generator, tag and receiver the modulation factor is the only way to change the received power at the receiver. To maximize the dierence in reection coecients is necessary for the improvement of tag- to-tag communication [35].

14 Passive Tag Semipassive Tag

This work

Figure 8: Reection coecients for dierent tags shown in a Smith Chart. Components with negative resistance can achieve a reection coecient above unity.

When using passive loads the impedance can be switched between a short (ΓA = -1, ZA = 0) and ∗ being completely matched to the antenna (ΓB = 0, ZB = Zantenna), giving rise to a maximum modulation factor of M = 0.25 [36]. is is shown as green circles in Figure8. Semi-passive tags can switch impedance between an open (ΓA = 1, ZA = ∞) and a short (ΓB = -1, ∗ ZB = -∞ ), giving rise to a maximum modulation factor of M = 1 [36]. is is shown as red triangles in Figure8. To increase the modulation factor above unity, M > 1, and thus amplify the backscaered signal, devices with negative resistant can be used [37]. In this thesis the reection coecient are switched between, ΓA > 1 and ΓB = 0. is is shown as blue trapezoids in Figure8. Negative resistance can be found in several components, such as p-n-p-n transistors, tetrode vacuum tubes and tunnel diodes [38]. ese components uses external power in order to get a modulation factor M >1, which can be used to amplify an incoming RF signal. e dierence between devices with positive and negative resistance is that devices with positive resistance has a dependent relation between the current through and the voltage across the device, while devices with negative resistance have voltage or current regions without a dependent relation between the current and voltage [39]. A 2-terminal device, such as the tunnel diode, with negative resistance has an energy source which is dependent on either the current or the voltage, but not both [39].

3.3 State of the Art for Reection Ampliers During the last decade there has been an improvement in the state of the art of the ABTs in terms of power consumption and gain. In Table1 a summation of works with dierent ampliers is presented. Between 1979 and 2014 the power consumption was decreased from 2000 W [40] to 325 mW [41] while the gain was about the same. In 2017 Farzami et al. demonstrated in [42] that a tunnel diode can achieve a 17 dB gain at a power consumption of 200 µW. In 2018 Amato et al. demonstrated in [5] that

15 a tunnel diode can reduce the bias power consumption by another order of magnitude and at the same time perform a gain of 35 dB. Since a tunnel diode seems to be able to perform a high gain at a power consumption below 100 µW, a tunnel diode is used in this thesis as an amplier. e nal ABT, with the tunnel diode, provided a gain of 35 dB at a power consumption of 48 µW. is is well in-line with present state of the art, as is shown in Table1. e reason for the lower power consumption in [5], compared to this work, is related to the type of tunnel diode that are used in the respective works, and not the overall circuit. is is because the power consumption of a tunnel diode can be seen directly from IV-curve measurements, where no other components are included. e achieved gain is related to both the type of tunnel diode and the other internal components used in the tag.

Table 1: Related works with reection ampliers found in literature.

Ref year Amplier type Gain [dB] Power [mW] RF-input [dBm] Freq. [GHz] is work 2018 Tunnel diode 30 0.048 -110 0.868 [5] 2018 Tunnel diode 35 0.0204 -81 5.8 [42] 2017 Tunnel diode 17 0.200 -30 0.890 [41] 2014 Bipolar transistor 10.2 0.325 -50 0.920 Silicon bipolar [43] 2013 13 2 -55 5.25 transistor [44] 2013 MESFET CFY30 10.2 18 - 4.5 [45] 2008 pHEMT 14 209.3 -75 21.2 [46] 2006 pHEMT 14 330 -45 21.2 [40] 1979 FET 16 2000 - 13

3.4 Tuning the Backscatter Tag to Desired Frequency An electromagnetic wave that incident on a material with dierent permiivity or permeability can transmit into the material, be reected or a combination of both. In electrical circuits the permiivity and permeability of a material is described together as the impedance. e power that is entered into a load in a circuit as the one shown in Figure9 is given by

2 1 |V0| 2 Pavg = (1 − |Γ| ) (12) 2 ZL

where V0 is the voltage before the load, and ZL is the impedance of the load and Γ is the reection coecient. In order to maximize the power transfer from the load ZA to the load ZL a matching network can be placed between the loads. e purpose of the matching network is to match the impedance of the matching network and the load ZL to the impedance before the matching network in order to maximize the power transfer between the loads. e matching network can be constructed by either lumped components or stubs. Lumped components, such as capacitors and inductors, adds the impedances −1 Zc = jωC and ZL = jωL respectively, where ω is the angular velocity, C is the capacitance and L is the inductance. e same analogy of changing impedance is true for stubs. In this project matching networks are constructed with open stubs, which is a transmission line connected perpendicular to another transmission line, as is shown in Figure9. e open stub is placed between the load, ZA, and the load ZL. A schematic gure of a general amplier circuit can be seen in Figure9, where V0 represents the incident RF-signal, ZA represents the impedance of the antenna and ZL represents the impedance of the amplier. e complex impedance if the antenna is given by

ZA = RA + jXA (13)

16 ZA

V 0 ZL

Figure 9: Simple setup of an antenna and loads, together with a tuning stub. e tuning stub (open stub) is placed between the load ZA, and the load ZL for impedance matching.

where RA is the resistance of the antenna and XA is the reactance of the antenna. e complex impedance of the tunnel diode is given by

ZL = −RL + jXL,RL > 0 (14) where −RL is the negative resistance of the load and XL is the reactance of the load. If, at a certain bias voltage across the load, the following condition is met

XA + XL = 0 (15) the reection coecient squared, or the gain, can be wrien as Z − Z ∗ 2 R + R 2 50 + R 2 |Γ|2 = L A = A L = L (16) ZL + ZA RA − RL 50 − RL which is always greater then 1 and increases as the negative resistance converges towards 50 Ω. In Equation 16 a 50 Ω antenna has been assumed which is a typical impedance of an antenna. e meaning of Equation 16 is that a negative RF-power −IR2 can be generated from a DC power supply and added to RF-power at the end of the circuit [47]. is means that the matching network, including the circuit, should provide an impedance that matches the impedance of the 50 Ω-antennas used in this thesis. Amato et al. showed in [48] that the complex impedance of a tunnel diode changes with input power because of non-linear properties of the tunnel diode. Amato et al. showed that for their tunnel diode the negative resistance approached 50 Ω and the reactance approached 0 Ω at low input powers, and that there is a upper maximum input power threshold of where the tunnel diode works as an amplier. One advantage of using the matching network is to be able to tune the gain to a specic frequency. e frequency is tuned and the S11-parameter is studied with a VNA. e impact of using a matching network can be seen in Figure 10. It can be seen that the gain is peaking at a specic frequency when using the matching network instead of being distributed across a larger frequency interval at the cost of lower gain when no matching network is used.

17 30 No matching network 25 Matching network

| [dB] 10 11 |S 5

-5

-10 500 1000 1500 2000 f [MHz]

Figure 10: Impact of matching network on gain. By using a matching network the gain can be tuned to a specic frequency. In this case the gain is tuned to 868 MHz.

3.5 Characterization of Tunnel Diodes ree types of tunnel diodes with unknown properties are characterized in terms of gain and power consumption. e tunnel diodes that are chosen are of the models: Ai301A [49], TD104 and 1N3712 [50]. ree dierent measurements are done for each diode: IV-curve measurement, gain measurements at dierent bias voltages, and output power measurements around 868 MHz. e IV-curve measurements are done by changing the bias voltage across the tunnel diode while measuring the current with a multimeter. A VNA is used to study the gain at dierent bias voltages, in the negative resistance region of the tunnel diode, controlled from a DC-power supply. To see how the output power varied around 868 MHz a VNA is used and the tunnel diode biased from a DC-power supply. Our objective in this thesis is to use a tunnel diode that has a power consumption less than 100 µW which, together with a matching network, provides a gain at 868 MHz that is comparable to that found in the state of the art, see Table1. To see how the gain changes with the bias voltage is important to optimize the choice of operation point. In the end, the 1N3712 tunnel diode is chosen since it provides high gain (35 dB) at a very low peak power consumption (48 µW).

3.5.1 General Electric TD104 In Figure 11 the IV-curve of the TD104 tunnel diode can be seen. e peak current is about 10 mA and the valley current is about 1 mA. e negative resistance region is seen between the voltages 200 mV - 400 mV, and the power consumption there is between 0.4 mW - 1.3 mW. e negative resistance is calculated as the inverse of the derivative of the IV-curve to be 2 Ω.

18 10 1.4 Current Power 1.2 8 1

6 0.8 [m W]

I [mA] 0.6 4 Bias P 0.4 2 0.2

0 0 0 100 200 300 400 500 600 V [mV]

Figure 11: Current and DC-power for dierent bias voltages. e negative resistance region is seen between 200 mV - 400 mV, where the power consumption is between 900 mW - 1.3 mW. is is signicantly higher than the power consumption found in the state of the art where tunnel diodes have been used, see Table1.

e TD104 tunnel diode shows the expected characteristic behaviour between the current and voltage, as seen in Figure 6d. e power consumption between 0.4 µW - 1.30 µW is an order of magnitude higher than that found in the state of the art, see Table1. To see how the gain varies with bias voltage, and thus to see the power consumption at the maximum gain, a matching network is added to the TD104 tunnel diode and tuned to 868 MHz. e bias voltage is changed between 304 mV - 350 mV for VNA output powers between -80 dBm to -30 dBm, and the result can be seen in Figure 12. It can be seen that the gain increases for lower bias voltages and that maximum gain is reached at 304 mV, which corresponds to a power consumption of 1.17 mW, seen in Figure 11. In Figure 12 it can be seen that the relationship between gain and bias voltage is the same for input powers between -80 dBm and -30 dBm. At power levels below 304 mV the gain collapsed in an uncontrolled way. How the output power of the ABT changes with input power around the frequency 868 MHz, at a specic bias voltage, can be seen in Figure 13. It can be seen that the output power of the ABT is peaking around 868 MHz for all input powers. e TD104 tunnel diode provides a maximum gain of 27 dB, which is good compared to the present state of the art, seen in Table1. e collapse of the gain for bias voltages lower then 304 mV is not investigated further. e reason for this is that the TD104 tunnel diode consumed a power of 1.17 mW at the peak gain, which is signicantly higher than present state of the art, seen in Table1. is tunnel diode is therefore most likely not suitable for low-power backscaer communication, and thus no further measurements was done with the TD104 tunnel diode.

19 30 -80dBm 25 -70dBm -60dBm -50dBm 20 -40dBm -30dBm 15 Gain [dB] 10

0 300 310 320 330 340 350 V [mV] bias

Figure 12: Gain at dierent bias voltages for dierent input powers. e gain increases for lower bias voltages up to a maximum value.

-20 -30dBm -40dBm -30 -50dBm -60dBm -40 -70dBm -80dBm -50 [dBm]

out -60 P

-70

-80

-90 700 750 800 850 900 950 1000 f [MHz]

Figure 13: Output powers at dierent frequencies around 868 MHz for dierent input powers. e output power is peaking around 868 MHz for all input powers.

20 3.5.2 Russian Ai301A In Figure 14 the IV-curve of the Ai301A tunnel diode can be seen. e peak current is about 2 mA and the valley current is about 0.1 mA. e negative resistance region is seen between the voltages 120 mV - 400 mV, and the power consumption there is between 60 µW - 300 µW. e negative resistance is calculated as the inverse of the derivative of the IV-curve to be 180 Ω. e tunnel diode Ai301A shows the expected characteristic behaviour between the current and voltage, as seen in Figure 6d. e power consumption between 60 µW - 300 µW is potentially low enough to match the power consumption of less than 100 µW.

2000 400 Current 1800 Power 350 1600 300 1400 250 1200 W] µ A] [

µ 1000 200 I [ Bias

800 P 150 600 100 400 50 200

0 0 0 100 200 300 400 500 600 700 800 900 V [mV]

Figure 14: Current and power consumption of the tunnel diode for dierent bias voltages at a -50 dBm output power from the VNA. e negative resistance region is seen between 120 mV - 400 mV, where the power consumption in between 60 µW - 300 µW. is is in line with what is fond in Table1.

To see how the gain varies with bias voltage, and thus nd the power consumption at the maximum gain, a matching network is added to Ai301A and tuned to 868 MHz. e bias voltage is changed between 170 mV - 270 mV for VNA output powers between -95 dBm and -55 dBm, and the result can be seen in Figure 15. It can be seen that the gain increases for bias voltages 170 mV - 200 mV, is stable between 200 mV - 230 mV and decreases for bias voltages between 230 mV - 270 mV. e power consumption around the peak gain is therefore 250 µW - 290 µW, seen in Figure 14. From Figure 15 it can also be seen that the gain increases at lower input powers. How the output power of the ABT changes with input power around the frequency 868 MHz, at a specic bias voltage, can be seen in Figure 16. It can be seen that the output power of the ABT is peaking around 868 MHz for all input powers. e output power from the ABT is about the same regardless of the input power. e reason for this is probably because of oscillations, rather than amplication of the input signal.

21 60 -95dBm -85dBm 50 -75dBm -65dBm 40 -55dBm

30 Gain [dB] 20

0 180 200 220 240 260 280 V [mV]

Figure 15: Gain at dierent bias voltages for dierent input powers. Highest gain is achieved at the lowest input power.

-40 -55dBm -50 -65dBm -75dBm -60 -85dBm -95dBm -70 [dBm]

-80 out,VNA P -90

-100

-110 850 860 870 880 890 900 f [MHz]

Figure 16: Output powers at dierent frequencies around 868 MHz for dierent input powers. e output power is peaking around 868 MHz for all input powers.

e Ai301A tunnel diode provides a maximum gain of 50 dB which is beer than present state of the art, seen in Table1. e power consumption at the peak gain is between 250 µW - 290 µW, which is more in line with some of the present state of the art, seen in table1, but more than the 100 µW that is wanted in this thesis. is tunnel diodes could be suitable for backscaer communication with less

22 strict constraints in the power consumption. Since the tunnel diode Ai301A consumes more than the 100 µW that is wanted, no further measurements are done with the Ai301A tunnel diode.

3.5.3 General Electric 1N3712 In Figure 17 the IV-curve of the 1N3712 tunnel diode can be seen. e peak current is about 1 mA and the valley current is about 0.500 mA. e negative resistance region is seen between the voltages 70 mV - 160 mV, and the power consumption there is 48 µW - 75 µW. e negative resistance is calculated as the inverse of the derivative of the IV-curve to be 190 Ω.

1000 80 Current 900 Power 70 800 60 700 50 600 W] µ A] [

µ 500 40 I [ Bias

400 P 30 300 20 200 10 100

0 0 0 20 40 60 80 100 120 140 160 V [mV]

Figure 17: Current and DC-power for dierent bias voltages. e negative resistance region is seen between 70 mV - 160 mV, where the power consumption is between 0.48 µW - 70 µW. e power consumption is lower or similar compared to the state of the art, found in Table1.

e 1N3712 tunnel diode shows the expected characteristic behaviour between the current and voltage, as seen in Figure 6d. e power consumption between 48 µW - 75 µW fulllls the constraints to be less than 100 µW. To see how the gain varied with bias voltage, and thus nd the power consumption at the maximum gain, a matching network is added to 1N3712 tunnel diode and tuned to 868 MHz. e bias voltage is changed between 70 mV - 160 mV for VNA output powers between -60 dBm and -10 dBm, and the result can be seen in Figure 18. It can be seen that the gain increases slightly between 70 mV - 130 mV and then decreases between 130 mV - 150 mV. e power consumption around the peak gain is therefore 48 µW - 73 µW, which can be seen in Figure 17. How the output power of the ABT changes with input power around the frequency 868 MHz, at a specic bias voltage, can be seen in Figure 19. It can be seen that the output power of the ABT is peaking around 868 MHz for all input powers. For input powers around -20 dBm the ABT will absorb the incident power, and no amplication is seen. For input powers between -30 dBm - 55 dBm the output power of the ABT is about constant. e reason for this is probably because of oscillations, rather than amplication of the input signal.

23 50 -10 dbm -20 dbm -30 dbm -40 dbm -50 dbm -60 dbm

10 Gain [dB]

-10

-20

-30 70 80 90 100 110 120 130 140 150 V [mV]

Figure 18: Gain at dierent bias voltages for dierent input powers. Highest gain is achieved at the lowest input power.

-10 -20dBm -22dBm -20 -30dBm -40dBm -30 -50dBm -55dBm -40 [dBm]

out -50 P

-60

-70

-80 700 750 800 850 900 950 1000 f [MHz]

Figure 19: Output powers at dierent frequencies around 868 MHz for dierent input powers. e output power is peaking around 868 MHz for all input powers.

24 e 1N3712 tunnel diode provides a maximum gain of 35 dB which is in line with the best gain found in the state of the art, seen in Table1. e power consumption at the peak gain was between 48 µW- 75 µW which is less than the 100 µW that is wanted in this thesis. erefore the tunnel diodes has good potential to be suitable for low-power backscaer communication. e 1N3712 tunnel diode is further evaluated by backscaer measurements.

3.6 Amplied Backscatter Tag Design To use the 1N3712 tunnel diode together with a matching network an amplied backscaer tag is designed. A SMP-connector is used to to connect the ABT to an antenna. To lter out the incoming DC-part of the carrier signal an capacitor is used, and to prevent AC-signals to go into the DC-voltage source, a choke inductor is used. A tunnel diode can can used as an amplier, if the following conditions are fullled [51]:

1. e tunnel diode is biased in the negative resistant region

2. Use transmission line with a negative resistance at the centre frequency f0 3. e gain is reduced to a suitable value 4. e stability is maintained while doing a), b) and c)

From IV-measurements, see Section 3.5, the bias voltage is found to make the tunnel diode enter the negative resistance region. To calculate the width of the transmission lines to match the 50 Ω antenna, Advanced Design System (ADS) is used. e width of the transmission line is calculated to 3 mm at the frequency 868 MHz. As transmission lines, copper stripes is used. ese are glued to a FR-4 PCB board with the following properties

• ickness: 1.6mm • Dielectric constant (): 4.4 • Loss tangent (tanδ): 0.0018

• Copper thickness: 3.5 µm Several capacitors and inductors are tested to nd the capacitance and inductance that provides high gain for the ABT. In the nal ABT, a 67 pF DC-block capacitor and a 47 µH choke inductor are used. A tunnel diode of the model 1N3712, made by General Electrics in 1960s/1970s, is used as an amplier. To get high gain at the centre frequency and reduce the gain outside the centre frequency, the length and position of the stub is changed, and the S11-parameter studied by using a VNA. e stability of the ABT are checked by using a spectrum analyzer, and to see if the ABT gave an output signal without an incoming carrier signal. If that would be the case, it would be because of unwanted oscillations in the circuit. e nal setup of the ABT can be seen in Figure 20.

25 Figure 20: ABT. (1)SMP-connector, (2) DC blocking capacitor, (3)matching network,(4) Tunnel diode, (5) choke inductor, (6) Swedish 5-krona for size comparison.

4 Evaluation

e ABT is evaluated by backscaer communication measurements, and the results are compared with results obtained by a semi-passive UBT. e UBT consists of lumped components soldered to a PCB- board, with a transistor to switch the impedance of the circuit, and is shown in Figure 22b. e UBT has been used in [3], and is valid as a representative UBT. In this section the complex impedance is studied as well as the gain and the power consumption of the ABT while backscaer an incoming carrier signal. Results from two dierent setups are then presented. Firstly the result of measurements when the output power from the carrier generator is xed and the tag is placed at dierent locations between the carrier generator and the receiver is presented. Secondly the result of changing the output power of the carrier generator while keeping both the tag, carrier generator and receiver at a xed position is presented. e gain achieved from backscaer measurements are compared with the gain achieved from VNA-measurements from dierent input powers. Two omnidirectional Vert900 antennas with operation frequency between 824 MHz ∼ 960 MHz are used, as is shown in Figure 22c, at the carrier generator and the tag, with a gain of 3 dBi. e same antenna is used both for the ABT and UBT. As a receiver a cc1310 from Texas instrument, as is shown in Figure 22a, is used with a gain of 4.7 dBi [52] and a noise oor of about -120 dBm. e ABT and UBT are mounted on a tripod together with a wave generator, as is shown in Figure 21. A square wave is sent from the wave generator to the tags to frequency shi the backscaered signal, according to Equation 4, to prevent interference between the carrier signal and the backscaered signal at the receiver.

26 Figure 21: Measuring setup. (1) ABT (2) UBT (3) Wave generator.

(a) cc1310 (b) UBT

Figure 22: Antennas and UBT used in the evaluation of the ABT.

27 110 60 Real part 90 Imaginary part 40

70 20

50 0

] 30 -20 ] Ω Ω 10 -40 Im(Z) [ Re(Z) [ -10 -60

-30 -80

-50 -100

-70 -120

-90 -140 -80 -70 -60 -50 -40 -30 -20 -10 0 P [dBm] in

Figure 23: Complex impedance for dierent input powers. For low input powers -R is approaching 50 Ω and the reactance is approaching 0 Ω, which are the conditions to make ABT work as an amplier.

4.1 Gain Achieved Measured with a VNA e gain of the ABT is increased with lower input powers, as is shown in Figure 15. A potential reason for this could be because the impedance, and thus the matching, of the ABT changes with the input power, as is discussed in Section 3.4. In the derivation of Equation 16 the reactance is assumed to be 0 Ω and the resistance is said to approach -50 Ω to give high gain. To see how well these conditions are obtained, the complex impedance is measured with a VNA where the input power is changed between -87 dBm and -5 dBm in intervals of about 5 dBm. e results are shown in Figure 23, where it can be seen that the negative resistance converges to 50 Ω and the reactance converges to 0 Ω for input powers below -50 dBm. is result indicates that the ABT works best at low input powers, which is also found in [48]. is statement is supported by the VNA measurements shown in Figure 18 and by the backscaer measurements later shown in the Figure 27 and in Figure 29.

To remove self-interference between the carrier signal and the backscaered signal at the receiver, the backscaered signal is shied in frequency by the use of a square wave modulation signal at the tag, as is described in Section 2.3. To see how this inuences the gain of the ABT, a VNA is used where the output power is changed between -87 dBm to -10 dBm and the square wave is changed between 86 mV and 0 mV, instead of applying a constant bias voltage as in Section 3.5. e result is shown in Figure 24, where it can be seen is that the gain has reduced compared to when a constant DC bias voltage is applied, as shown in Figure 18. e maximum gain is 17 dB at an input power of -87 dBm. It can also be seen that a positive gain is achieve for input powers below -45 dBm. is can be related to the matching, seen in Figure 23, which shows that the matching is best for input powers below around -50 dBm. e oscillations found in Figure 19 is removed, and an amplication of the backscaered signal is seen.

28 20

Gain [dB] -5

-10

-15

-20 -87 -80 -70 -60 -50 -40 -30 -20 -10 P [dBm] in

Figure 24: VNA measurements of the output power for dierent input powers while switching bias point of the tunnel diode with 200 kHz. e ABT has a positive gain for input powers below -45 dBm.

4.2 Power Consumption of the ABT To have a low power consumption is an important part of backscaer communication in order to extend the lifetime of baeries, harvest energy from photo diodes or directly from the carrier signal. To get the power consumption of the ABT while operating, the power consumption is measured while the ABT backscaers an incoming carrier signal. e backscaered signal is measured, with a frequency shi of 200 kHz, with a spectrum analyzer. A wave-generator is used to bias the ABT between two dierent voltages. e power consumption of the ABT is calculated through P = I2R, where the current is calculated from the voltage drop across a 100 Ω resistor. e voltage drop is calculated as the dierence between the voltages measured on an oscilloscope with two probes (Aglient N2862B) on either side of the resistor. e power consumption of the ABT while backscaer a carrier signal is shown in Figure 25. It can be seen that the power consumption gets a square wave behaviour from the switching between the two voltage levels. It can also be seen that the peak power consumption of the ABT while backscaer a carrier signal is 48 µW. is is in line with the expectation from the IV-curve of the 1N3712 tunnel diode, shown in Figure 17. Compared with the state of the art, see Table1, the power consumption of 48 µW is signicantly lower than what has been achieved with dierent kind of ampliers. Compared to works where other tunnel diodes have been used, 48 µW is in line with the lowest power consumption that has been measured, and is below the 100 µW that is wanted in this thesis.

29 50

30 W] µ

P [ 20

0 0 0.5 1 1.5 2 2.5 t [µ s]

Figure 25: Power consumption of the ABT while backscaering a carrier signal with a frequency shi of 200 kHz. e peak power consumption of the ABT is 48 µW.

4.3 Placing Tag between Source and Receiver is experiment represents a typical situation for IoT-devices. e carrier generator and the receiver are at dierent distances with respect to the tag, which can represent a person moving around in a room. e setup is shown in Figure 26. e distance between the carrier generator and the receiver is 7 m, and the tag is placed in one meter intervals between the carrier generator and the receiver. e measurements are done in an anechoic chamber, which is a controlled environment that prevent reection at the walls. For each meter three measurements are done during 1 minute where the location of the tag is moved by 3 cm between each measurement. A -10 dBm carrier signal is used in order to see the typical u-shape curve of the UBT, as seen in 3b, and to try to prove that the ABT works beer at lower input powers. 7m

r r Carrier 1 2 Receiver Generator Tag

Figure 26: Setup used in the anechoic chamber. e tag is moved between the carrier generator and the receiver while having a constant signals strength of the carrier signal.

e result from the measurements is show seen in Figure 27. For the UBT the u-shape curve can be seen similar to the the theoretical curve seen in Figure 27, which is expected from the inverse relationship between the received power and the distances found in the backscaer link budget equation, see Equation7. e received power from the ABT does not show the typical u-shape behaviour. What is seen is that for distances up to 2 m away from the carrier generator, the ABT works worse than the

30 -75 ABT UBT (measured) -80 UBT (calculated)

-85 [dBm] r

P -90

-95

-100 0.2 1 2 3 4 5 6 7 r [m] 1

Figure 27: Result from measurement in an anechoic chamber. e ABT starts to perform beer than the UBT closer to the receiver.

UBT, between 2 m - 4 m the ABT and the UBT works similarly well and between 4 m - 6 m the ABT works beer than the UBT. is result is expected from the results seen in Figure 23, which shows that the complex impedance of the ABT converges to the impedance of the antenna for lower input powers, resulting in an amplication of the backscaered signal.

4.4 Indoor Environment To see how the ABT works in an indoor environment, measurements is done at the IT department in Uppsala University. e carrier generator and the receiver is separated by a distance of 19 m including 4 walls, as is shown in Figure 28. e tag is placed at a distance of 1 meter away from the receiver. An amplier is used to generate a strong carrier signal, which is measured at the tag, and varied between -115 dBm to -55 dBm. To average out contributions from reection 6 measurements are done during 1 minute for each carrier signal strength, where the tag is moved 10 cm between each measurement. e received signal strengths at the receiver from the backscaer measurements are shown in Figure 29 together with calculated values for the UBT. It can be seen that for carrier strengths below -75 dBm the UBT is not able to backscaer the carrier signal with a suciently high power above the noise oor of the receiver. From the calculated values for the UBT it can be seen that the power decreases linearly with the input power, which is expected from the backscaer link budget equation. e ABT is able to backscaer a signal above the noise oor for input powers as low as -110 dBm. In Figure 30 it can be seen that the ABT gives a gain of 35 dB at the input power -110 dBm. In Figure 29 it can be seen that the ABT performs beer for lower input powers which is expected from the non-linear behaviour of the tunnel diode, see Section 3.4. e reason the ABT performs beer at lower input powers is shown in Figure 23, which shows that the ABT is beer matched to the 50 Ω antenna at lower input powers.

31 19m

Tag

Carrier 1m generator Receiver

Figure 28: Setup used for indoor measurement. e strength of the carrier signal is changed.

-70 ABT -80 UBT (measured) UBT (calculated) -90

-100

-110 [dBm] r

P -120

-130

-140

-150

-120 -100 -80 -60 P [dBm] in

Figure 29: Power measurement from tag and ABT in indoor environment. e ABT starts to perform beer than the UBT at lower input powers.

In Figure 30 the gain seen in Figure 29 and the gain seen in Figure 24 is merged together to see how well the results from the backscaer-measurements and the VNA-measurements agrees with the other. What can be seen is that the results agrees around -87 dBm to -60 dBm. For input powers below -87 dBm it is dicult to see how well the results agrees with each other since no VNA-measurements was done for lower input power levels than -87 dBm.

32 40 VNA 30 Backscatter

10 Gain [dB] 0

-10

-20 -120 -100 -80 -60 -40 -20 0 P [dBm] in

Figure 30: Results from VNA and Backscaer measurements. A 35 dB gain is achieved at the input power -110 dBm.

5 Discussion

In this thesis an ABT has been built to increase the power of backscaered signals. ree dierent tunnel diodes have been characterized in terms of gain and power consumption to nd a tunnel diode that provides a high gain at a low power consumption. e tunnel diodes has been characterized through VNA measurements, and the ABT evaluated through backscaer measurement and compared with an UBT.

Gain to Distance: In section 3.1 it is shown in Figure7 that a gain of 35 dB can improve the backscaer communication distance in [3] from 3.4 km up to 110 km. is is true if the 35 dB gain is achieved at an input power of 2 dBm. e ABT provides a high gain at low input powers, and no gain for input powers around 2 dB, as is shown in Figure 30. To relate the gain achieved by the ABT to distance the Friis equation, Equation 5, is used where the transmied power is changed between -110 dB and -50 dB and the maximum communication distance is calculated if a receiver with a noise oor of -120 dBm is assumed. For the case of the ABT the gain, found in Figure 30, is added to the transmied power. e result is shown in Figure 31 where it is seen that the ABT achieves the longest communication distance for input powers around -55 dBm, where the ABT increases the communication distance by 320 m compared to the UBT. At the input power -110 dBm the ABT provides a maximum communication distance of 10 m while the UBT is limited to a communication distance of 0.17 m. is means that the communication distance from the ABT is increased by 58 times, at -110 dBm, and 2 times at -50 dBm compared to the UBT. Figure 31 shows the beauty of the 1N3712 tunnel diode that even at low input powers the ABT is able to provide a communication distance for 10th of meters while the UBT is limited to distances up to a few meters. To reach longer communication distances, the highest gain should have bin at a higher input power, but that would also have increased the maximum communication distance for the UBT. e increased communication distance by 58 times for the ABT compared to the UBT is the same regardless of for

33 what input power the maximum gain is achieved at, as is shown in Figure 32 in appendix. It would be best to have a high gain at a large range of input powers, which could possibly be achieved by a proper choice of the matching network, but no further investigations about that is made in this thesis.

1000 ABT 420 300 UBT 200 100 50

[m] 20 10 2,max r

0.18 -110 -100 -93 -87 -80 -75 -70 -65 -60 -55 -50 P [dBm] in

Figure 31: Maximum communication distance. e longest communication distance achieved with the ABT is for higher input powers, but relative to the UBT the ABT preforms best at lower input powers.

e use of tunnel diodes: e 1N3712 tunnel diode implemented in the ATB provided a 35 dB gain, at the input power -110 dBm, compared to an UBT, as is shown in Figure 18 and Figure 29, at a peak power consumption of only 48 µW, which as is shown in Figure 25. ese results are signicantly beer compared to the present state of the art where other type of ampliers are used and in line the with the result where tunnel diodes has been used, as is shown in Table1. High gain at a low power consumption is an important part in IoT-devices in order to reduce the use of baeries. A great advantage by using the 1N3712 tunnel diode is that the power consumption is 48 µW - 70 µW in the negative resistance region, see Figure 17, which is very low compared to the state of art. is means that the power consumption would probably not change signicantly if the matching is changed in order for the ABT to preform gain at a higher input power or another frequency. erefore, if the intention is to reach long distance communication, the 1N3712 tunnel diode could possible still be used. From Figure 23 it is argued that the ABT works beer at low input power levels. is could enable communication with very weak signal strengths, and amplify signals above the noise oor of a receiver. e fact that the ABT seems to work worse than the UBT for higher input power, as is shown in Figure 23, is a disadvantage. is would make it dicult for the ABT to work in close proximity with a carrier generator that transmit to high powers. Potential applications where the ABT could be used is in those which communicate by ambient backscaer communication. e reason for this is that people seldom come close to the antenna but are instead normally outside a specic radius of the antenna which would give a received power below the value of where the ABT starts to give gain. Indoor communication, where tenths of meters are sucient distances, are also potential places where such ABT as was designed in this thesis can be used. e evaluation of the ABT is done with one specic tunnel diode of the model 1N3712. ese tunnel diodes, or any other tunnel diode, are not fabricated in large amounts, but can in some cases be produced

34 from a special order. e fact that tunnel diodes are not made in large amounts can be a future problem of using tunnel diodes in large amounts. But since the tunnel diodes is just pn-junction with a high doping level, which has been used in larger amounts in 1960-1970s, the tunnel diode should be able to fabricate in larger amounts. e power consumption of the tunnel diodes presented in this thesis are: 48 µW (1N3712), 250 µW (Ai301A) and 1.17 mW (TD104). e reason why the tunnel diodes have dierent power consumption is related to the internal structure of the tunnel diode. Implementation: e ABT that is built in this thesis, is build with copper stripes glued to a PCB- board. e advantage of this approach is that it is simple and fast. It is easy to change the position and length of the stub in order to tune the gain to be at a specic frequency. e problem with this approach is that the properties in the glue changes with time, which in this thesis changed the matching network. e result became that a lot of time had to be spent tuning the tag to give gain at the desired frequency of 868 MHz. Instead of this approach, lumped element can been used. e advantages with this approach is that the impedance matching would not have changed so easily, but the problem is to nd the correct values of the lump components in order to match the impedance of the circuit to the antenna. e capacitor and inductors that are used in the ABT are not chosen to maximize the gain, but rather chosen to make the circuit work as an amplier. is opens up the possibility of improving the design of the ABT by using other physical sizes and values of the components.

6 Conclusion

is thesis has provided a motivation to use an amplier at a tag to increase the backscaer communication distance. Based on gain and power consumption from recent work, a tunnel diode is presented to work as an amplier. ree dierent tunnel diodes are characterized in terms of gain and power consumption by the use of a VNA. A tunnel diode of the type 1N3712 is implemented in an ABT together with a matching network to provide gain a the frequency 868 MHz. e ABT is evaluated by backscaer communication measurements, both indoors and in an anechoic chamber and compared with an UBT. e ABT is able to provide a 35 dB gain at a peak power consumption of 48 µW. is is beer both in terms of power consumption and gain compared to the state of the art where other type of ampliers have been used, and in line with the works where tunnel diodes have been used. A translation between gain and distance based on the Friis equation showed that the 35 dB gain achieved by the ABT can increase the communication distance with up to two orders of magnitude compared to an UBT.

35 7 Future Work

In Figure 27 it can be seen that the ABT preforms beer than the UBT closer to the receiver because of beer matching. It would be interesting to see is what would have happen if the carrier signal is reduced to -50 dBm, where the tunnel diodes starts to get matched to the antenna. One possible outcome could be that gain is achieved in the whole range between the carrier generator and the receiver, which potentially could result in a more at received signal strength from the ABT. In this thesis no information is send between the receiver and the carrier generator. To send information instead of just energy would support the conclusion that the backscaer communication distance increases when using the 1N3712 tunnel diode. In this thesis it has been stated that the tunnel diode works beer at low input powers because of beer matching, as is shown in Figure 23. What has not been investigated is if the reasons why it gets a beer matched at lower powers is because of the tunnel diode itself or because it was tuned with a low input power from the VNA. If the ABT is tuned with a higher input power from the VNA, a possible outcome is that the ABT can achieve gain at any input power interval. If this is true an adjustable matching network can potentially be implemented to change the input power of where the ABT gives highest gain, to make sure that the ABT always achieves peak gain.

36 References

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39 8 Appendix 8.1 Communication distance at Dierent Input Powers e communication distance between the tag and the receiver is related to the input power at the tag. How the maximum communication distance changed if the ABT achieves a gain of 35 dB compared to a UBT for dierent input powers can be seen in Figure 32. What can be seen is that if the maximum gain is at higher input powers, the communication distance increases both for the ABT and UBT, but the ratio between the maximum communication distances is still 58 times for the ABT compared to the UBT.

108 ABT UBT 106

104 [m]

2,max 2 r 10

100

10-2 -120 -100 -80 -60 -40 -20 0 20 40 P [dBm] in

Figure 32: Maximum communication distance for dierent input powers. Regardless of at for which input power the 35 dB gain is at, the ABT still gives 58 times longer communication distance compared to a UBT.

8.2 Denition of gain is section is devoted to the reader that is not very familiar to the use of dB, dBm, Gain etc. e unit dBm has the meaning decibel relative to 1 mW is is the unit of absolute power. e advantage of using the logarithmic dB scale is that both small numbers and large number can be com- pressed and wrien in a simpler way. e denition of power in terms of dBm is related to power in terms if mW in the following way P P = 10 ∗ log ( mW ) (17) dBm 10 1mW where PmW is the power in mW. In table2 a selection of some values used in this thesis can be seen.

Table 2: Conversion between wa and dBm

Power (W ) 100f 1p 10p 100p 1n 10n 100n 1 µ 10 µ 100 µ 1m Power (dBm) -100 -90 -80 -70 -60 -50 -40 -30 -20 -10 0

40 e gain is dened as a fraction between two values. On a logarithmic scale this becomes a subtrac- tion. Using the logarithmic scale, the gain has the unit dB, and is dened as

P1 RatiodB = 10 ∗ log( ) (18) P2 where P1 and P2 is e.g. output and input power. In gure 33 a selection of some gain values and how they relate to the fraction between powers in the unit of W can be seen.

102 [dB]

1 in,dBm 10 /P ut,dBm P

100 0 5 10 15 20 P /P ut,W in,W

Figure 33: Relationship between gain in dB and fraction of powers in W . A 15 times stronger output signal compared the input signal in terms of [W] is a 30dB gain.

In this thesis the gain is the fraction between the power from the ABT and the UBT

8.3 Distance Calculations

Pgen=28; %dBm GT= 3 ; %dBi Gt = 3 ; %dBi c=299792458; %m/s f=868e6; %Hz d1 = 1 ; %m Gr = 3 ; %dBi d2 = 3 4 0 0 ; %m Gcc1310=4.47; %dBi Ptag =[ −110 −100 −90 −80 −70 −60]; P r e c lorea=Ptag + Gr + Gt + 2 ∗ 1 0 ∗ l o g 1 0 ( c / ( 4 ∗ p i ∗ f ∗ d2 ) ) ; gain=0:100; g=zeros(1,length(gain )); for i=1:length(gain) syms x Pcc1310=Ptag + gain(i) + Gr + Gt + 2 ∗ 1 0 ∗ l o g 1 0 ( c / ( 4 ∗ p i ∗ f ∗ x ) ) ;

41 g(1,i)=double(solve(Pcc1310−P r e c lorea==0,x)); end f i g u r e ( 1 ) Ax=semilogy(gain ,g/1000); hold on g y=[0 10 1e2 1e3 1e4 1e5 1e6]; % user defined grid Y [start:spaces:end] g x=[0:20:100]; % user defined grid X [start:spaces:end] for i=1:length(g x ) semilogy([g x ( i ) g x ( i ) ] , [ g y ( 1 ) g y(end)],’k:’) %y grid lines hold on end for i=1:length(g y ) semilogy([g x ( 1 ) g x ( end ) ] , [ g y ( i ) g y(i)],’k:’) %x grid lines hold on end y l a b e l ( ’ r 2 [km ] ’ ) xlabel(’Gain [dB]’ ) set(gca,’YTick’,[0 10 100 1000 10000 100000 1000000] ); set(gca,’XTick’,[0 10 20 30 40 50 60 70 80 90 100] ); set(gca,’FontSize ’,13)

Ptag =[ −110 −100 −93 −87 −80 −75 −70 −65 −60 −55 −50]; p=zeros(1,length(Ptag )); for i=1:length(Ptag) syms y P r e c lorea2=Ptag(i) + Gr + Gt + 2 ∗ 1 0 ∗ l o g 1 0 ( c / ( 4 ∗ p i ∗ f ∗ y ) ) ; p(1,i)=double(solve(Prec l o r e a 2 −−120==0,y )); end f i g u r e ( 2 ) hold on gain2=[35.2 28.4 21.9 17.3 16.4 16.9 11.9 16.44 15.5 12.6 7.7]; for i=1:length(gain2) syms x Pcc13102=Ptag(i) + gain2(i) + Gr + Gt + 2 ∗ 1 0 ∗ l o g 1 0 ( c / ( 4 ∗ p i ∗ f ∗ x ) ) ; g2(1,i)=double(solve(Pcc13102 −−120==0,x )); end plot(Ptag ,g2,’ − −∗ ’) plot(Ptag ,p,’−−o ’ ) y l a b e l ( ’ r {2 , max} [m] ’ ) x l a b e l ( ’ P { in } [dBm ] ’ ) a x i s ([ −110 −50 0 4 3 0 ] ) set(gca,’YTick’,[0 20 50 100 150 200 250 300 350 400 420]); yyaxis right plot(Ptag ,g2./p,’−−v ’ ) a x i s ([ −110 −50 0 6 0 ] ) y l a b e l ( ’ r {2 ,ABT}/ r {2 ,UBT} [%]’) set(gca , ’XTick’,[ −110 −100 −93 −87 −80 −75 −70 −65 −60 −55 −50] ) ; set(gca,’YTick’,[0 5 10 15 20 25 30 35 40 45 50 55 60]);

42 set(gca,’FontSize ’,13) legend( ’ABT’ , ’UBT’ , ’ABT/UBT’) 8.4 Indoor Measurement Calculations

ABT=[R1 R2 R3 R4 R5 R6 R7 R8 R9 R10 R11 R12]; ABT std=[R1std R2std R3std R4std R5std R6std R7std R8std R9std ... R10std R11std R12std]; C a r r i e r siganl=[C1 C2 C3 C4 C5 C6 C7 C8 C9 C10 C11 C12]; UBT=[NT1 NT2 NT3 NT4 NT5 NT6 NT7]; UBT std=[NT1std NT2std NT3std NT4std NT5std NT6std NT7std];

Pr antenna=Carrier s i g a n l −1.47 −4.47+20∗ ... log10((299792458/868e6)/(4 ∗ p i ∗ 1 ) ) ; f i g u r e ( 1 ) hold on h1=errorbar(Carrier siganl ,ABT,ABT std/2,’b v’) set(h1,’linewidth ’ ,4) h2=errorbar([C1 C2 C3 C4 C5 C6 C7],UBT,UBT std/2,’k o’) set(h2,’linewidth ’ ,4) h3=plot(Carrier s i g a n l , Pr antenna ,’r’) set(h3,’linewidth ’ ,4) a x i s ([ −135 −45 −155 −65]) y l a b e l ( ’ P r [dBm ] ’ ) x l a b e l ( ’ P { in } [dBm ] ’ ) set(gca,’FontSize ’,13) legend(’ABT’,’UBT (measured)’,’UBT (calclulated ) ’) 8.5 Anechoic Chamber Calculations td1 exp0 mean=mean( td1 e x p 0 ) ; td2 exp0 mean=mean( td2 e x p 0 ) ; td3 exp0 mean=mean( td3 e x p 0 ) ; t d e x p 0 =( td1 exp0 mean + td2 exp0 mean + td3 exp0 mean ) / 3 normal1 exp0 mean=mean( normal1 exp0 ) ; normal2 exp0 mean=mean( normal2 exp0 ) ; normal3 exp0 mean=mean( normal3 exp0 ) ; normal exp0=(normal1 exp0 mean+normal2 exp0 mean + . . . normal3 exp0 mean ) / 3 t d e x p 0 e = s t d ( [ td1 exp0 mean td2 exp0 mean td3 exp0 mean ] ) normal exp0 e=std ([normal1 exp0 mean normal2 exp0 mean . . . normal3 exp0 mean ] ) td1 exp1 mean=mean( td1 e x p 1 ) ;

43 td2 exp1 mean=mean( td2 e x p 1 ) ; td3 exp1 mean=mean( td3 e x p 1 ) ; t d e x p 1 =( td1 exp1 mean + td2 exp1 mean + td3 exp1 mean ) / 3 normal1 exp1 mean=mean( normal1 exp1 ) ; normal2 exp1 mean=mean( normal2 exp1 ) ; normal3 exp1 mean=mean( normal3 exp1 ) ; normal exp1=(normal1 exp1 mean+normal2 exp1 mean + . . . normal3 exp1 mean ) / 3 t d e x p 1 e = s t d ( [ td1 exp1 mean td2 exp1 mean td3 exp1 mean ] ) normal exp1 e=std ([normal1 exp1 mean normal2 exp1 mean . . . normal3 exp1 mean ] ) td1 exp2 mean=mean( td1 e x p 2 ) ; td2 exp2 mean=mean( td2 e x p 2 ) ; td3 exp2 mean=mean( td3 e x p 2 ) ; t d e x p 2 =( td1 exp2 mean + td2 exp2 mean + td3 exp2 mean ) / 3 normal1 exp2 mean=mean( normal1 exp2 ) ; normal2 exp2 mean=mean( normal2 exp2 ) ; normal3 exp2 mean=mean( normal3 exp2 ) ; normal exp2=(normal1 exp2 mean+normal2 exp2 mean + . . . normal3 exp2 mean ) / 3 t d e x p 2 e = s t d ( [ td1 exp2 mean td2 exp2 mean td3 exp2 mean ] ) normal exp2 e=std ([normal1 exp2 mean normal2 exp2 mean . . . normal3 exp2 mean ] ) td1 exp3 mean=mean( td1 e x p 3 ) ; td2 exp3 mean=mean( td2 e x p 3 ) ; td3 exp3 mean=mean( td3 e x p 3 ) ; t d e x p 3 =( td1 exp3 mean + td2 exp3 mean + td3 exp3 mean ) / 3 normal1 exp3 mean=mean( normal1 exp3 ) ; normal2 exp3 mean=mean( normal2 exp3 ) ; normal3 exp3 mean=mean( normal3 exp3 ) ; normal exp3=(normal1 exp3 mean+normal2 exp3 mean + . . . normal3 exp3 mean ) / 3 t d e x p 3 e = s t d ( [ td1 exp3 mean td2 exp3 mean td3 exp3 mean ] ) normal exp3 e=std ([normal1 exp3 mean normal2 exp3 mean . . . normal3 exp3 mean ] ) td1 exp4 mean=mean( td1 e x p 4 ) ; td2 exp4 mean=mean( td2 e x p 4 ) ; td3 exp4 mean=mean( td3 e x p 4 ) ; t d e x p 4 =( td1 exp4 mean + td2 exp4 mean + td3 exp4 mean ) / 3

44 normal1 exp4 mean=mean( normal1 exp4 ) ; normal2 exp4 mean=mean( normal2 exp4 ) ; normal3 exp4 mean=mean( normal3 exp4 ) ; normal exp4=(normal1 exp4 mean+normal2 exp4 mean + . . . normal3 exp4 mean ) / 3 t d e x p 4 e = s t d ( [ td1 exp4 mean td2 exp4 mean td3 exp4 mean ] ) normal exp4 e=std ([normal1 exp4 mean normal2 exp4 mean . . . normal3 exp4 mean ] ) td1 exp5 mean=mean( td1 e x p 5 ) ; td2 exp5 mean=mean( td2 e x p 5 ) ; td3 exp5 mean=mean( td3 e x p 5 ) ; t d e x p 5 =( td1 exp5 mean + td2 exp5 mean + td3 exp5 mean ) / 3 normal1 exp5 mean=mean( normal1 exp5 ) ; normal2 exp5 mean=mean( normal2 exp5 ) ; normal3 exp5 mean=mean( normal3 exp5 ) ; normal exp5=(normal1 exp5 mean+normal2 exp5 mean + . . . normal3 exp5 mean ) / 3 t d e x p 5 e = s t d ( [ td1 exp5 mean td2 exp5 mean td3 exp5 mean ] ) normal exp5 e=std ([normal1 exp5 mean normal2 exp5 mean . . . normal3 exp5 mean ] ) td1 exp6 mean=mean( td1 e x p 6 ) ; td2 exp6 mean=mean( td2 e x p 6 ) ; td3 exp6 mean=mean( td3 e x p 6 ) ; t d e x p 6 =( td1 exp6 mean + td2 exp6 mean + td3 exp6 mean ) / 3 normal1 exp6 mean=mean( normal1 exp6 ) ; normal2 exp6 mean=mean( normal2 exp6 ) ; normal3 exp6 mean=mean( normal3 exp6 ) ; normal exp6=(normal1 exp6 mean+normal2 exp6 mean + . . . normal3 exp6 mean ) / 3 t d e x p 6 e = s t d ( [ td1 exp6 mean td2 exp6 mean td3 exp6 mean ] ) normal exp6 e=std ([normal1 exp6 mean normal2 exp6 mean . . . normal3 exp6 mean ] ) td =[ t d e x p 0 t d e x p 1 t d e x p 2 t d e x p 3 t d e x p 4 t d e x p 5 t d e x p 6 ] ; normal=[normal exp0 normal exp1 normal exp2 normal exp3 normal exp4 . . . normal exp5 normal exp6 ] ; t d e =[ t d e x p 0 e t d e x p 1 e t d e x p 2 e t d e x p 3 e t d e x p 4 e . . . t d e x p 5 e t d e x p 6 e ] normal e =[ normal exp0 e normal exp1 e normal exp2 e normal exp3 e . . . normal exp4 e normal exp5 e normal exp6 e ] place=[0.2 1 2 3 4 5 6 ];

45 f i g u r e ( 1 ) hold on k = 7 0 ; Gtx =3∗ ones ( 1 , k ) ; Gtx2 =3∗ ones ( 1 , k ) ; L=299792458/(868e6) ∗ ones ( 1 , k ) ; d2=linspace (0.1 ,7 ,70); Pt = ( − 4 . 3 ) ∗ ones ( 1 , k ) ; Pr normal=zeros(1,k); for n=1:1:70 d1=7−d2 ( n ) ; Pr normal(n)=(Pt(n)+Gtx(n)+Gtx2(n)ˆ2+40 ∗ ... log10((L(n)/(4 ∗ p i ∗ d1 ∗ d2 ( n ) ) ) ) ) ; end errorbar(place ,td ,td e /2 , ’ − −∗ ’) errorbar(place ,normal ,normal e /2 , ’ − −o ’ ) p l o t ( d2 , Pr normal ) y l a b e l ( ’ P r [dBm ] ’ ) x l a b e l ( ’ r 1 [m] ’ ) a x i s ( [ 0 7 −100 −75]) set(gca,’XTick’,[0.2 1 2 3 4 5 6 7] ); set(gca,’FontSize ’,13) legend(’ABT’, ’UBT (measured)’,’UBT (calcluated) ’)