Textile Antennas for Monitoring People in Danger Situations

Textile antennas for monitoring people in danger situations

CAROLINA MILLET CATALAN

Master’s Degree Project Stockholm, Sweden 2016 Textile antennas for monitoring people in danger situations

CAROLINA MILLET CATALAN

Stockholm 2016

Electromagnetism Engineering School of Electrical Engineering Kungliga Tekniska H¨ogskolan

Abstract

The fast growing ﬁeld of wearable technologies has a big impact in antenna research. Antennas integrated into clothing for body centric communications allow the user’s situation to be measured without restricting his activities. Implementing textile antennas to operate as personal transmitters for existing satellite communication systems increases its availability and uses, specially localized in places where mobile communications infrastructures are not developed but satellite communications has a full coverage.

The aim of this thesis is to design and manufacture a system of two fully textile microstrip antennas: a GPS antenna with right hand circular polarization operating at 1.575 GHz and a PLB antenna operating at 406MHz working as a distress beacon for Cospas-Sarsat search and rescue international programme. We describe the design, manufacture and performance of both antennas, as well as the material used as a key choice for textile antennas.

Antennas were measured with a near ﬁeld scanner to evaluate their performance which resulted to be working at the established operational frequency with a good matching and the expected radiation patterns from the simulations and studied literature. Further development regarding the feeding circuit of the GPS antenna is needed to ensure its circular polarization.

Keywords: Textile Antennas, Wearables, GPS, Satellites, Cospas-Sarsat.

Acknowledgements

First, I would like to express my sincerest gratitude to my supervisor Oscar Quevedo- Teruel, for giving me this opportunity, for being an inspiration and for his guidance and support during this project. He always helped when I ran into a trouble spot or had a question about my research.

I would like to thank my friends for everything we have lived together and new experiences yet to come. I am also thankful to my newest friends for this experience in Stockholm and making my stay at KTH unforgettable. A very special dedication to Laura for her patience, love and being always there for me besides the distance. Anything is possible when we are surrounded by good friends.

Finally, I would also like to thank my family for their support and love during this time. Special gratitude to my parents, Lourdes and Xavier, who have always en- couraged me to study and believed in me, to my sister Mar for her unconditional support during my whole life, to my brother Xavi for his life philosophy "Tot i res", and to my grandma Rosa for her help, fast learning with new technologies and her care despite the distance.

This accomplishment would not have been possible without them. Gràcies.

Carolina Millet Catalan,

Stockholm, February 2016

III

Contents

List of FiguresVII

List of TablesXI

1 Introduction1 1.1 Background and motivation...... 1 1.2 Aim and objectives. Boundaries of the study...... 2 1.2.1 Boundaries...... 2 1.3 Methodology. Report outline...... 3 1.3.1 Report outline...... 4

2 Background Theory5 2.1 Antenna Parameters...... 5 2.1.1 Radiation Pattern...... 5 2.1.2 Radiation Power Density...... 6 2.1.3 Radiation Intensity...... 7 2.1.4 Directivity...... 7 2.1.5 Efficiency...... 7 2.1.6 Gain...... 8 2.1.7 Bandwidth...... 8 2.1.8 Bandwidth, Quality factor and Efficiency...... 9 2.1.9 Polarization...... 9 2.1.10 Axial Ratio...... 11 2.1.11 Input impedance...... 11 2.1.12 SAR - Specific Absorption Rate...... 12 2.1.13 Scattering Matrix...... 13 2.2 Microstrip line...... 14 2.3 Microstrip Antennas...... 16 2.3.1 Advantages and disadvantages...... 17 2.3.2 Feeding Methods...... 17 2.3.3 Rectangular Patch...... 18 2.4 Communication Systems...... 20 2.4.1 Introduction to satellites...... 21 2.4.2 Global Navigation Satellite System (GNSS)...... 22 2.4.3 Cospas-Sarsat System...... 24

3 Design and Results 29

V Contents

3.1 Materials and laboratory instruments...... 29 3.1.1 Materials...... 29 3.1.2 Laboratory...... 32 3.1.3 Software...... 32 3.2 GPS Antenna...... 33 3.2.1 Circularly Polarized Microstrip Antennas...... 33 3.2.2 Quadrature 90°Hybrid...... 35 3.2.3 Design and Simulation...... 36 3.2.4 Manufacturing and matching...... 46 3.2.5 Far ﬁeld measurements...... 51 3.3 PLB - Personal Locator Beacon...... 55 3.3.1 PIFA - Planar Inverted F-Antenna...... 55 3.3.2 Design and Simulation...... 55 3.3.3 Manufacturing and matching...... 62 3.3.4 Far ﬁeld measurements...... 63 4 Conclusion and Future Work 65 4.1 Conclusion...... 65 4.2 Future work...... 66 Bibliography 67

VI List of Figures

1.1 Scheme of the relevant steps followed during the project...... 3

2.1 Directional radiation pattern [7]...... 6 2.2 Types of Polarization [10]...... 10 2.3 Circular left-hand Polarization [11]...... 10 2.4 Ellipse [7]...... 11 2.5 S11 parameter of an antenna...... 14 2.6 Geometry of microstrip transmission line [8]...... 14 2.7 Electric and magnetic fields of microstrip transmission line [8]..... 15 2.8 Equivalent geometry of Microstrip line [8]...... 15 2.9 Microstrip antenna [7]...... 16 2.10 Different patch shapes [7]...... 17 2.11 Feeds for microstrip antennas [7]...... 18 2.12 Patch electric field [7]...... 18 2.13 Patch extension [7]...... 19 2.14 Field modes for rectangular microstrip patch [7]...... 20 2.15 Orbit examples [14]...... 22 2.16 Galileo constellation [19]...... 23 2.17 Corpas-Sarsat System [22]...... 25 2.18 MEOSAR system concept [22]...... 27

3.1 Tested felts...... 30 3.2 Conductive fabric used...... 31 3.3 Conductive thread used...... 31 3.4 Circular polarization techniques...... 34 3.5 Rectangular microstrip antenna with two orthogonal feeds [29]..... 35 3.6 Geometry of a 90°hybrid [8]...... 35 3.7 GPS antenna model...... 36 3.8 S-parameters simulated of the GPS antenna...... 37 3.9 Electric field simulations at 1.57GHz...... 38 3.10 Electric field distribution evolution with the time...... 38 3.11 Far field simulations from port 1 of the GPS antenna...... 38 3.12 Far field of a patch antenna with two orthogonal ports and 90° phase shift...... 39 3.13 Complete antenna’s geometry...... 39 3.14 Top view of the 90°circuit...... 40 3.15 S-parameters of the 90° feeding circuit...... 40

VII List of Figures

3.16 Representative S-parameters of the circuit...... 41 3.17 Load patch’s geometry...... 41 3.18 |S11| of the load patch...... 42 3.19 Geometry of the final circuit...... 42 3.20 Geometry of the complete antenna...... 43 3.21 Side view of the complete antenna...... 43 3.22 |S11| of the complete design...... 44 3.23 Axial Ratio of the complete design...... 44 3.24 Electric field distribution evolution with the time...... 45 3.25 Far field simulations of the complete antenna...... 45 3.26 Far field right circular polarization simulations of the complete antenna. 46 3.27 Configuration of the dual-orthogonal feed microstrip antenna..... 46 3.28 |S11| measured with the VNA...... 47 3.29 Manufactured textile microstrip patch antenna...... 48 3.30 Configuration of the complete full textile dual-orthogonal feed microstrip...... 48 3.31 Manufactured fully textile microstrip patch...... 49 λ 3.32 Phase’s measurements to determine s ...... 49 4 3.33 Circuit’s S-parameters measurements...... 50 3.34 Load patch matching and measurement...... 50 3.35 Manufactured fully textile complete microstrip antenna with a 90° feeding circuit integrated...... 51 3.36 |S11| complete antenna...... 51 3.37 Configuration for the measurements of the dual-orthogonal feed microstrip antenna...... 52 3.38 Far field measured of the manufactured dual-orthogonal feeding microstrip antenna...... 52 3.39 Axial ratio of the manufactured microstrip antenna measured with the near field scanner...... 53 3.40 Configuration of the complete full textile dual-orthogonal feed microstrip...... 53 3.41 Far field measured of the manufactured microstrip with 90° feeding circuit integrated...... 54 3.42 Axial ratio of the measured microstrip antenna with the integrated feeding circuit...... 54 3.43 Representation of the worn belt...... 56 3.44 PIFA’s geometry...... 56 3.45 PIFA’s Top view...... 57 3.46 |S11| of the PIFA...... 57 3.47 PIFA’s electric field distribution...... 58 3.48 Far field simulation of the PIFA...... 58 3.49 Belt configuration...... 59 3.50 |S11| simulated of the PIFA integrated to the body model...... 60 3.51 Far field simulations...... 60 3.52 Far field simulations...... 61 3.53 Polar radiation pattern with different phases...... 61

VIII List of Figures

3.54 Belt sewing procedure...... 62 3.55 |S11| of the PIFA...... 62 3.56 Manufactured belt...... 63 3.57 3D radiation pattern measured of the PIFA...... 64 3.58 2D radiation pattern...... 64

IX List of Figures

X List of Tables

1.1 GPS speciﬁcations...... 2 1.2 PLB speciﬁcations...... 2

2.1 Advantages and disadvantages of microstrip antennas...... 17

3.1 Felt’s characterization...... 30 3.2 Instruments used in the lab [23], [24], [25]...... 32 3.3 GPS antenna final dimensions...... 37 3.4 Antenna’s dimensions...... 40 3.5 Dimensions of the load patch...... 41 3.6 Dimensions of the complete antenna...... 43 3.7 Dimensions of the manufactured patch...... 47 3.8 PIFA’s final dimensions...... 57 3.9 Dimensions of the final PIFA integrated to the body model...... 59

XI List of Tables

XII 1 Introduction

1.1 Background and motivation

Wearable technologies is a fast growing field in application-oriented research. The vision of wearables describes future electronic systems as an integral part of our everyday clothing serving as intelligent personal assistants without restricting the user’s activities and being aware of the user’s situation [1]. One of the dominant topics is antennas for body-centric communications. There are specialized occupa- tion segments that apply body centric communication systems such as paramedics, fire fighters, military and athletes, who already are using textile antennas integrated into their work uniform and garments [2]. Wearable antenna are required to have light weight, low cost and no installation [3].

The International Cospas-Sarsat programme is intended to provide an earth-to- satellite secure and rescue communication in case of distress alert. Distress beacons transmit emergency signals which are detected by LEOSAR, GEOSAR or MEOSAR satellites. Ground receiving stations process the satellite downlink signal and forward them to Rescue Coordination Centers available in the area were the distress signal was activated. This process allows a correct location and fast response [4].

Emergency beacons transmitters are carried aboard ships (EPIRBs), aircraft (ELTs), or used as personal locator beacons (PLBs), and activated manually when needed. EPIRBs and ELTs can be easily integrated in ships and aircraft’s structure but PLBs need to be carried by the user. PLBs are similar to portable radios both in terms of size and weight and are supposed to be carried in pockets or attached to safe vests. They are sold as self standing transmitters [5][6].

The integration of PLB transmitters as textile antennas could increase its availability to non-professionals and its uses could be diversiﬁed adding new distress situations as attacks or children and domestic abuse, specially localized in places where mobile communications infrastructures are not developed but satellite communications has a full coverage.

1 1. Introduction

1.2 Aim and objectives. Boundaries of the study

The aim of this project is to design and manufacture a system of two antennas. The ﬁrst one, is a GPS antenna responsible of deﬁning its position through any of the existing global navigation satellite system. The second one is a PLB antenna designed to be used as a distress beacon in the Cospas-Sarsat search and rescue international programme.

There are two possible ways of combining these antennas for a proper localization.

• Independent: Both antennas work independently, PLB sends the distress signal and GPS is used for localization. In this case, localization can be done with the PLB signal or using the GPS.

• Combination: GPS antenna is used to determine its location and after some processing (out of the scope of the thesis), it is codiﬁed and transmitted through the PLB distress signal allowing a simpler localization.

Each of them needs to have diﬀerent characteristics according to the requirements of the system is going to be integrated to.

Frequency 1.565-1.585GHz Axial Ratio <3dB Polarization RHCP Frequency 406MHz

S11 <-15dB S11 <-15dB

S12 <-25dB S12 <-25dB

Table 1.1: GPS speciﬁcations. Table 1.2: PLB speciﬁca- tions.

1.2.1 Boundaries

• The viability of the proposed system has not been consulted with Cospas- Sarsat professionals. • The thesis was focused on the design of the antennas, excluding the communication protocols, data processing and codiﬁcation. • Power supply and consumption of the antennas were not studied. • Bending of the antennas, material degradation and weather changes have not been taken into account. • GPS antenna has not been designed for a speciﬁc positioning on the human body.

2 1. Introduction

1.3 Methodology. Report outline

This thesis is divided in three main sections:

(a) Literature review: the ﬁrst step to start this study was reading up some literature related to wearable antennas focusing on their designs and used materials, as well as doing research about communication systems intended to rescue people in distress situations. With all these information and help from a most experienced person designing antennas as my supervisor, the purpose of the thesis was established, a project plan and schedule were deﬁned.

(b) Design and simulations: the following step was to select a suitable design model to be implemented with CST Microwave Studio simulation software. Parametrized simulations allowed us to dimension both antennas until reliable results for a correct performance were achieved. This process was simultaneous with the study of the materials that were characterized with some tests. Both processes combined concluded to the ﬁnal design for both (GPS and PLB) antennas.

(c) Manufacture, measurements and results: based on the previous step, both antennas were manufactured while checking that the characteristics followed during the design remained intact. Finally, the antennas were tested to check if the requirements for a proper performance were achieved. The use of some laboratory material as a Vector Analyser - Anritsu MS2026B and Signal Generator - HP 8665B were required, as well as the Near Field Scanner for the ﬁnal measurements.

Figure 1.1: Scheme of the relevant steps followed during the project.

3 1. Introduction

1.3.1 Report outline Chapter 1: Introduction • Background and motivation. • Aim and objectives. Boundaries. • Methodology. Report outline.

Chapter 2: Background Theory - Review of the theory related to the thesis. • Antenna parameters. • Microstrip line. • Microstrip antennas. • Communication systems.

Chapter 3: Design and Results - Detailed process of designing the antennas, including simulation, manufacturing, measuring and evaluation. • GPS Antenna. • PLB Antenna. • Materials and laboratory instruments.

Chapter 4: Conclusions and Future work - Summary of the main conclusions from the results obtained. Perspectives for future projects related to textile antennas.

4 2 Background Theory

This chapter summarizes the notions for comprehending the main concepts studied for a correct design of the antennas and their integration to the existing communication systems. Firstly, a review of the fundamental parameters of antennas and their mathematical derivations. Secondly, an overview of microstrip technology is exposed to focus later on its characteristics which determine our design. Finally, a description, functionality and requirements of the satellite communication systems used by the antennas designed and manufactured.

2.1 Antenna Parameters

Many characteristic antenna parameters are necessary to describe the performance of the designed and manufactured antennas. Some of them are interrelated and all of them are not essential for a complete study. Deﬁnitions are taken from [7], [8] and [9].

2.1.1 Radiation Pattern

An antenna radiation pattern is defined as "a mathematical function or a graphical representation of the radiation properties of the antenna as a function of space coordinates" according to [7]. Mainly, it is determined in the far-field region where the field distribution is essentially independent of the distance from the antenna. It is common to represent the radiation pattern in 2-D coordinates for an easier comprehension. Antennas can be classified according to different types of radiation patterns: • Isotropic: Equal radiation in all directions. It is ideal and not realizable but it is taken as reference. • Directional: More effective to receive or radiate in some directions where the radiation pattern has the major lobe and maximum directivity. • Omnidirectional: Isotropic antennas in one plane. It is a particular case of directional pattern.

Radiation Pattern Lobes

A radiation pattern is formed by different lobes which can be classified as main, minor, side or back lobes according to their position respect to the main lobe which has the maximum radiation, as it is shown in the next figure:

5 2. Background Theory

Figure 2.1: Directional radiation pattern [7].

2.1.2 Radiation Power Density

Electromagnetic ﬁelds transmitted or received by the antennas have a power and energy associated to them. The instantaneous Poynting vector is used to describe the power of the electromagnetic wave:

W = E × H (2.1) where W is the instantaneous Poynting vector, E the instantaneous electric field intensity and H the instantaneous magnetic field. It is more useful to find the average power density relating the instantaneous fields to the complex fields E and H:

1 1 W = E × H = Re[E × H∗] + Re[E × Hej2wt] (2.2) 2 2 The total power crossing a closed surface can be obtained integrating the normal component of the Poynting vector over the surface: ‹ P = W · ds (2.3)

Therefore, the average power radiated by an antenna can be written as ‹ ‹ 1 P = W · ds = Re(E × H∗)ds (2.4) rad 2 S S

The radiation pattern represents the average power density radiated by the antenna as a function of direction.

6 2. Background Theory

2.1.3 Radiation Intensity

Radiation intensity is the power radiated from an antenna per unit solid angle in a given direction [7]. It can be obtained using

2 U = r Wrad (2.5) where U is the radiation intensity and Wrad is the radiation density. The radiation intensity is related to the far-zone electric ﬁeld of an antenna so the total power can be derived from its mathematical expression: ‹ ˆ ˆ 2 π Prad = UdW = π Usinθdθdφ (2.6) 0 0 W where dW is the element of solid angle.

2.1.4 Directivity

Directivity of an antenna is deﬁned as "the ratio of the radiation intensity in a given direction from the antenna to the radiation intensity averaged over all directions" according to [7]. Therefore, the directivity of a nonisotropic source is equal to the ratio of its radiation intensity in a given direction over the radiation intensity of an isotropic source.

U 4πU D = = (2.7) U0 Prad where ˆ ˆ 2π π Prad = Usinθdθdφ (2.8) 0 0 and U = B0F (θ, φ) (2.9) so we can compute a general expression for the directivity

F (θ, φ) ´ ´ D(θ, φ) = 4π 2π π (2.10) 0 0 F (θ, φ)sinθdθdφ If the direction is not speciﬁed, it implies the direction of maximum directivity that can be expressed as U|max 4πUmax Dmax = D0 = = (2.11) U0 Prad

2.1.5 Eﬃciency

The eﬃciency of an antenna expresses the losses at the input terminals and within the structure of the antenna. In general can be written as et = ereced (2.12)

7 2. Background Theory

where et is the total efficiency, er is the mismatch between the transmission line and the antenna, ec is the conduction efficiency and ed in the dielectric efficiency. The mismatch efficiency can be easily computed as

2 e0 = (1 − |Γ| ) (2.13)

2.1.6 Gain

The absolute gain of an antenna in a given direction is the ratio of the intensity to the radiation intensity if the antenna radiated isotropically, in a given direction [7]. It can be expressed as

radiation intensity U(θ, φ) gain = 4π = 4π (2.14) total input power Pin Normally the relative gain is more representative which is the ratio of the power gain in a given direction to the power gain of a reference antenna in its referenced direction. The total radiated power is related to the total input power with the conduction eﬃciency and dielectric eﬃciency explained in the previous section because gain does not include mismatch losses according to IEEE Standards.

Prad = ecedPin (2.15) Therefore,

" U(θ, φ)# G(θ, φ) = eced 4π (2.16) Prad which can be related to the directivity explained before

G(θ, φ) = ecedD(θ, φ) (2.17) In a similar way, the maximum gain in the direction of maximum radiation can be developed as

G0 = G(θ, φ)|max = ecedD(θ, φ)|max = ecedD0 (2.18)

2.1.7 Bandwidth

The bandwidth of an antenna is the range of frequencies which the performance of the antenna is reliable according to some characteristics (input impedance, radiation pattern, polarization, directivity...) [7].

For narrowband antennas, the bandwidth is expressed as a percentage of the frequency diﬀerence over the center frequency of the bandwidth.

f − f BW = max min · 100 (2.19) fc

8 2. Background Theory

For broadband antennas, the bandwidth is expressed as the ratio of the upper-to- lower frequencies of operation.

f BW = max : 1 (2.20) fmin

2.1.8 Bandwidth, Quality factor and Eﬃciency

Bandwidth, quality factor and eﬃciency are interrelated and there is no complete freedom to optimize each one independently. A tradeoﬀ between them needs to be achieved deciding which one is more important to optimize.

The quality factor depends on the antenna losses created by the radiation(Qrad), conduction(Qc), dielectric(Qd) and surface wave(Qsw) losses. The total quality factor (Qt) is inﬂuenced by all this diﬀerent losses and can be written as

1 1 1 1 1 = + + + (2.21) Qt Qrad Qc Qd Qsw

For a very thin substrate (h λ0), Qrad is the dominant factor which can be calculated as 2ωrW Qrad = (2.22) hGrad for a rectangular patch operating in the dominant TM010 mode where Grad is the conductance across the gap between the patch and the ground plane. In general, the bandwidth is proportional to the volume. For a rectangular mi- 1 crostrip antenna means that the bandwidth is inversely proportional to √ . In r this case, the bandwidth increases as the substrate height increases. The radiation eﬃciency of an antenna can be also expressed in terms of quality factors

Qt ecdsw = (2.23) Qrad

2.1.9 Polarization

The polarization of an antenna is defined as "the polarization of the wave transmitted in the direction of the maximum gain" [7]. Polarization of a radiated wave is defined as the property of an electromagnetic wave describing the time varying direction and relative magnitude of the electric- field vector [7]. In other words, polarization is the curve traced by the end point of the instantaneous electric field observed along the direction of propagation. Polarization can be classified as linear, circular or elliptical depending on the figure that the electric field traces, as represented in Figure ??.

9 2. Background Theory

Figure 2.2: Types of Polarization [10].

Linear Polarization

The electric and magnetic ﬁeld vector at a point are always oriented along the same straight line at every instant of time. Linear polarization is accomplished if the electric and magnetic vector possess any of these conditions: • They have only one component. • They have two orthogonal linear components in time phase or 180°phase difference (or multiples of 180°).

Circular Polarization

The electric and magnetic field vector at a point trace a circle as a function of time as in Figure 2.3. The conditions to accomplish circular polarization are: • The field has two orthogonal linear components. • The two components have the same magnitude. • The two components have a time-phase difference of 90°or odd multiples of 90°.

Figure 2.3: Circular left-hand Polarization [11].

The electric ﬁeld can be traced according to a clockwise rotation which is designated as right-hand polarization or counterclockwise designated as left-hand polarization. For example, Figure 2.3 is left-handed polarized considering the axis origin the source of the ﬁeld.

10 2. Background Theory

Elliptical Polarization

The electric and magnetic field vector change continuously with time describing an elliptical locus in space. The conditions to accomplish elliptical polarization are: • The field has two orthogonal linear components. • The two components can have the same or different magnitude: – If the two components do not have the same magnitude, the time-phase difference between the two components must not be 0°or multiples of 180°. – If the components have the same magnitude, the time-phase difference between the two components must not be 90°or odd multiples of 90°. As mentioned for circular polarization, the same rules apply to design right-hand(clockwise) or left-hand (counterclockwise) polarization. Although linear and circular polarizations are special cases of elliptical polarization, a wave is considered elliptical polarized if it is not linearly or circularly polarized.

2.1.10 Axial Ratio

It is referred to the ratio of the major axis to the minor axis of an ellipse, represented in Figure 2.4. major axis OA AR = = , 1 ≤ AR ≤ ∞ (2.24) minor axis OB

Figure 2.4: Ellipse [7].

Axial Ratio is used to evaluate the polarization of the antenna: it tends to ∞ when linear polarization, it is 1 (0dB) when circular polarization and larger than 1 for an elliptical polarized antenna. The axial ratio tends to degrade away from the main lobe, so it is a factor that has a big inﬂuence to determine the bandwidth. It is usually indicated the angle deviation from the main beam that maintains the polarization required.

2.1.11 Input impedance

The input impedance is the impedance presented by an antenna at its terminals. The impedance of the antenna with no load attached can be calculated as

11 2. Background Theory

ZA = RA + jXA (2.25) where ZA is the impedance, RA the resistance and XA the reactance at the terminals of the antenna.

The resistive part RA is composed of two components

RA = Rr + RL (2.26) where Rr is the radiation resistance and RL is the loss resistance of the antenna.

The radiation resistance determines the radiated power of the antenna and the loss resistance the dissipated power. Therefore the power delivered to the antenna is also formed by two components:

2 2 2 Pdelivered = |I| RA = |I| Rr + |I| RL = Pradiated + Plost (2.27)

The antenna radiation eﬃciency η is deﬁned as the ratio between the radiated power and the power delivered to the antenna or in terms of impedance:

P R η = radiated = r (2.28) Pdelivered Rr + RL

2.1.12 SAR - Speciﬁc Absorption Rate

SAR is a measure of how transmitted RF energy is absorbed by human tissue. It is a function of the electrical conductivity σ, the induced E-ﬁeld from the radiated energy and the mass density of the tissue ρ. ˆ σ(r)|E(r)|2 SAR = dr (2.29) sample ρ(r)

SAR is critical to antenna design, because if the SAR is too high the antenna must be changed. Typically, if the SAR is too high the transmit power is lowered, which directly yields lower SAR. However, since there are minimum transmit power speciﬁcations for mobile devices, the SAR cannot be dropped indeﬁnitely [12]. As a result, the antenna positioning is critical. The antennas for mobile phones are typically on the bottom of the phone, to keep the radiating part of the phone as far as possible from the brain region [12]. Other methods for dropping the SAR include impedance matching changes and parasitic resonators which will disturb the antenna’s radiation pattern (hopefully lowering SAR) [12].

12 2. Background Theory

2.1.13 Scattering Matrix

Defining voltages and currents for non-TEM lines is difficult therefore some tools are used to simplify microwave circuits analysis. More intuitive ideas of circuit analysis can be used to determine voltages, currents and power flow through a specific element. As equivalent voltages and currents can be determined with this techniques, impedance and admittance matrices of circuit theory can be used in order to determine a matrix description of the network. The scattering matrix provides a complete description of the network seen as N ports. It relates the voltage waves incident on the ports to those reflected from them. + Considering a N-port network where Vn is the amplitude of the voltage wave incident − on port n and Vn is the amplitude of the voltage wave reflected from port n.

 −    + V1 S11 S12 ...S1N V1  −    + V2   S21 S22 ...S2N  V2   .  =  . .   .  (2.30)  .   . . ..   .   .   . . .   .  − + VN SN1 SN2 ...SNN VN

h i h i h i V − = S V + (2.31)

S-parameters describe the input-output relationship between ports in an electrical system. If we have 2 ports, S12 represents the power transferred from Port 2 to Port 1 and S21 represents the power transferred from Port 1 to Port 2. In general, SNM represents the power transferred from Port M to Port N in a multi-port network [12]. A port can be deﬁned as any place where voltage and current can be delivered.

The most common parameter with antennas is S11 which represents how much power is reflected from the antenna, not delivered. It is known as the reflection coefficient. A small S11 indicates a significant amount of energy has been delivered to the antenna. S11 values are measured in dB and are negative, ex: -10 dB. S11 is also sometimes referred to as return loss, which is simply S11 but made positive instead (Return Loss = - S11) If S11 = 0 dB all the delivered power is reflected from the antenna and nothing is radiated.

In general, S-parameters depend on the frequency, therefore S11 needs to be adjusted according to the operational frequency the antenna is designed for. A good response of the antenna is typically considered when S11 <-10dB. Next Figure is an example of the S11 of an antenna operating at 1.57GHz.

13 2. Background Theory

Figure 2.5: S11 parameter of an antenna.

2.2 Microstrip line

Microstrip line is one of the most popular types of planar transmission lines primarily because it is easily miniaturized and integrated [8]. The geometry of a microstrip line is a conductor of width W printed on a thin, grounded dielectric substrate of thickness d and relative permittivity r which is shown in Figure 2.6.

Figure 2.6: Geometry of microstrip transmission line [8].

The analysis of microstip line is complicated because the dielectric does not fill the region above the strip, therefore all the fields are not contained within the homogeneous dielectric region in contrast with the stripline. Some (usually most) of its field lines are in the dielectric region between the strip conductor and the ground plane and some fraction in the air region above the substrate like Figure 2.7. For this reason microstrip line cannot support pure TEM wave. The exact fields of a microstrip line constitute a hybrid TM-TE wave.

14 2. Background Theory

Figure 2.7: Electric and magnetic ﬁelds of microstrip transmission line [8].

In most practical applications, where the dielectric substrate is electrically very thin (d λ) the fields are essentially the same as the static (DC) case. So, good approx- imations for the phase velocity, propagation constant and characteristic impedance can be obtained from static or quasi-static solutions following the represented geometry in Figure 2.8. Phase velocity and propagation constant can be expressed as c vp = √ (2.32) e √ β = k0 e (2.33) where e is the effective dielectric constant of the microstrip line and as some of the field lines are the dielectric and some are in air satisfies the relation

1 < e < r (2.34) and depends on the substrate dielectic constant, the substrate thickness, the conductor width and the frequency.

Figure 2.8: Equivalent geometry of Microstrip line [8].

Approximate design formulas can be reached, eﬀective dielectric is given by [8]:

r + 1 r − 1 1 e = + s (2.35) 2 2 d 1 + 12 W

Impedance of the lines can be determined by [8]:

15 2. Background Theory

 60 8d W ! W √  ln + for ≤ 1  e W 4d d   Z0 = (2.36)   120π W  for ≥ 1 √ W W d  + 1.393 + 0.667ln + 1.444  e d d

2.3 Microstrip Antennas

Microstrip patch antennas or simply patch antennas consist of a metallic strip (patch) placed over a grounded substrate as shown in Figure 2.9.

Figure 2.9: Microstrip antenna [7].

The patch and substrate are very thin in terms of free-space wavelength t λ0 λ λ and h λ . For a rectangular patch, usually 0 < L < 0 . The substrate is 0 3 2 a dielectric sheet with a constant r usually in the range of 1 ≤ λ0 ≤ 12. Thick substrates with a low dielectric constant are desirable because the bandwidth and the eﬃciency increase. Thin substrates with higher dielectric constants are desirable for microwave circuitry because undesired radiation and coupling are minimized. A compromise has to be reached between a good performance and design. The radiating patch can have diﬀerent shapes: rectangular, square, circular, dipole, elliptical... Selecting the patch shape is important for the design as the radiation, analysis and fabrication depend on it. Figure 2.10 illustrates some of the possible shapes of the patches.

16 2. Background Theory

Figure 2.10: Diﬀerent patch shapes [7].

2.3.1 Advantages and disadvantages

Microstrip antennas technology have several advantages and disadvantages that need to taken into account according to its applications [7]. They are summarized in Ta- ble 2.1.

Advantages Disadvantages

Low proﬁle High Q Conformable to planar surfaces Dimensions-frequency dependence Simple and cheap to manufacture Narrow frequency bandwidth Versatile (frequency and bandwidth) Spurious feed radiation Easy to integrate with circuits High radiation eﬃciency

Table 2.1: Advantages and disadvantages of microstrip antennas.

2.3.2 Feeding Methods

Many conﬁgurations can be used to feed microstrip antennas. Microstrip line, coaxial probe and aperture coupling are the most popular. • Microstrip feed line is a conducting strip, with smaller width that the patch. It is easy to fabricate and match but distorts the radiation pattern. • Coaxial probe feed uses the inner conductor to connect the patch to the ground plane. It is easy to fabricate and match but it has narrow bandwidth and it is more diﬃcult to model for thick substrates. • Non-contacting aperture coupling feeds are used to avoid cross-polarized radiation produced by asymmetric feeding methods as the feed line and single coaxial probe and to increase the bandwidth.

17 2. Background Theory

In Figure 2.11 the three conﬁgurations are presented.

(a) Microstrip line (b) Probe (c) Aperture-coupled

Figure 2.11: Feeds for microstrip antennas [7].

2.3.3 Rectangular Patch

The rectangular patch has been traditionally analysed using two diﬀerent models: transmission-line which is easy to understand because of its physical insight but is less accurate, and cavity model which is more accurate but more complex.

Transmission-line Model

Basically, the transmission line model represents the mictrostrip antenna by two slots, separated by a low-impedance Zc transmission line of length L. This model and its electric ﬁeld is shown in ﬁgure 2.12.

Figure 2.12: Patch electric ﬁeld [7].

This model follows the same development described in the Section 2.2, where a microstrip line can be characterized as a new microstrip line with an effective dielectric constant that approximates the behaviour of all the electric field lines, most of them that reside in the substrate and the portion that exit through the air. The effective dielectric constant is a function of the frequency but for low frequencies, it can be approximated to the static value:

r + 1 r − 1 1 e = + s (2.37) 2 2 h 1 + 12 W

The size of the patch looks electrically greater because of the fringing eﬀects, and this extension depends on the eﬀective dielectric constant e and the width-height ratio. We can assume that

Le = L + 24L (2.38) as represented in Figure 2.13.

18 2. Background Theory

Figure 2.13: Patch extension [7].

If we do not take into account this fringing, the resonant frequency for the dominant TM010 mode can be approximated by

1 c0 (fr)010 ' √ = √ (2.39) 2L µr 2L r

For a more exact computation, the edge eﬀects can be included achieving a new formulation

1 1 c0 (frc)010 = √ = q √ = q √ (2.40) 2Le µ00e 2L µ00e 2L r therefore, we can assume that q factor is the fringe factor (length reduction factor).

(f ) q = rc 010 (2.41) (fr)010

Cavity Model

Microstrip antennas can be treated as a cavity (substrate) bounded by electric con- ductors (the patch above and the ground below) and by magnetic walls (open circuit) along the perimeter of the patch. It is an approximate model which leads to a reac- tive input impedance (zero or infinite value) and does not radiate any power but it is an accepted approach in terms of radiated pattern, input admittance and resonant frequencies. The field configuration within the cavity can be found using the vetor potential, which mus satisfy the homogeneous wave equation of

2 2 ∇ Ax + k Ax = 0 (2.42) whose solution can be written with separated variables as

Ax = [A1cos (kxx) + B1sin (kxx)] [A2cos (kyy) + B2sin (kyy)] [A3cos (kzz) + B3sin (kzz)] (2.43) where kx, ky and kz are the wavenumbers along the x, y and z directions.

Applying the boundary conditions, the ﬁnal form of the vector potential within the cavity is 0 0 0 Ax = Amnpcos (kxx ) cos (kyy ) cos (kzz ) (2.44)

19 2. Background Theory

where Amnp represents the amplitude coeﬃcients to each mnp mode.

The wavenumbers are subject to the constraint equation. Therefore, the resonant frequencies for the cavity are given by s 1 mπ 2 nπ 2 pπ 2 (f ) = √ + + (2.45) r mnp 2π µ h L W

The mode with the lowest order resonant frequency is the dominant mode. If L > W > h, the dominant mode is the TM010:

1 c0 (fr)010 = √ = √ (2.46) 2L µ 2L r

If W > L > h, the dominant mode is the TM001:

1 c0 (fr)001 = √ = √ (2.47) 2W µ 2W r Diﬀerents modes are represented in Figure 2.14.

(a) TM010 (b) TM001

2.4 Communication Systems

This section explains some communication systems operating with the use of artiﬁ- cial satellites: Global Navigation Systems as GPS, Galileo, GLONASS and COM- PASS, and the international Cospas-Sarsat programme. This summarize is needed to understand the characteristics of the antennas in order to operate and adapt to these systems correctly.

20 2. Background Theory

2.4.1 Introduction to satellites

The world’s first artificial satellite was launched by the Soviet Union in 1957, since then, thousands of satellites (approximately 6600) have been launched into orbit around the Earth [13]. A few hundred satellites are currently operational and thousands of unused satellites and satellite fragments orbit the Earth. Satellites are used for a large number of purposes: military and civilian observa- tion, communications, navigation, weather and research. Usually, they are semi- independent computer-controlled systems, therefore, satellite subsystems attend many tasks such as power generation, thermal control, telemetry, attitude control and orbit control. Satellites can be classified according to their orbit type, altitude, inclination and eccentricity.

Centric classiﬁcation

• Geocentric orbit: An orbit around the planet Earth, like the Moon. It is by far the most common type of orbit. • Heliocentric orbit: An orbit around the Sun, like all planets in the Solar System. • Areocentric orbit: An orbit around the planet Mars.

Altitude classiﬁcation

• Low Earth orbit (LEO): Geocentric orbits ranging in altitude from 0-2000km. • Medium Earth orbit (MEO): Geocentric orbits ranging in altitude from 2000km to 35,786km. • Geosynchronous Orbit (GEO): Geocentric circular orbit with an altitude of 35,786km. • High Earth Orbit (HEO): Geocentric orbits above the altitude of geosynchronous orbit 35,786km.

Inclination classiﬁcation

The orbit can be inclined in reference to the equatorial plane: • Polar orbit: An orbit that passes above or nearly above both poles of the planet on each revolution. • Polar sun synchronous orbit: A nearly polar orbit that passes the equator at the same local time on every pass.

Eccentricity classiﬁcation

• Circular orbit: An orbit that has an eccentricity of 0 and whose path traces a circle. • Elliptic orbit: An orbit with an eccentricity between 0 and 1 whose orbit traces the path of an ellipse.

21 2. Background Theory

Some of the explained orbit types are represented in Figure 2.15.

Figure 2.15: Orbit examples [14].

Doppler eﬀect

The frequency and wavelength of an electromagnetic field are affected by relative motion, this is known as Doppler effect. This effect is significant in low-earth-orbit (LEO) satellites because all LEO satellites are constantly moving relatively to each other and to points on the surface which causes variations in the frequencies and wavelengths of received signals [15]. In geostationary satellite systems, Doppler effect is not an important factor and therefore it can not be used as a tracking tool. A measurement of the Doppler Shift is used as a tracking technique to determine the distance between the satellite and the receiver at a time. A Doppler curve can be produced with the measurement of frequency against time. As a satellite approaches, the frequency appears raised relative to the actual transmission frequency. As it goes away, the frequency appears to be lowered. At the time of closest approach, the transmitted and received frequencies are usually the same [16].

2.4.2 Global Navigation Satellite System (GNSS)

GNSS is the generic term of systems of satellites that provide autonomous geo-spatial positioning with global coverage. Small electronic receivers are allowed to determine their location (longitude, latitude and altitude) to high precision (within a few me- tres) using signals transmitted from satellites [17]. The signals are used to calculate the current local time to high precision which provides time synchronisation. Curently, there are four diﬀerent functional systems: the American GPS (Global Positioning System), the Russian GLONASS (GLObal’naya NAvigasionnay Sput- nikovaya Sistema), the European Galileo and Chinese COMPASS (the evolution of Beidou Navigation Satellite System). GPS, GLONASS and Galileo’s satellites are interoperable which is beneﬁcial to all users as more satellites are available for redundancy and therefore, higher accuracy.

GPS

The Global Positioning System (GPS) is a satellite-based navigation system made up of a network of 24 satellites placed into orbit by the U.S. Department of De-

22 2. Background Theory fense. GPS was originally intended for military applications, but in the 1980s, the government made the system available for civilian use. GPS works in any weather conditions, anywhere in the world, 24 hours a day [18]. GPS satellites fly in medium Earth orbit (MEO) at an altitude of approximately 20,200 km. Each satellite circles the Earth twice a day. The 31 satellites are arranged into six orbits surrounding the Earth which ensures users can view at least four satellites from any point on the planet. The accuracy users attain depends on different factors: atmospheric effects, sky blockage and receiver quality.

• How does it work? Satellites have very stable clocks which are synchronized to each other and to ground clocks. GPS receivers have less accurate clocks. The GPS receiver compares (solving some equations) the time a satellite signal was transmitted with the time it was received, which determines the its distance from the satellite. Monitoring multiple satellites and repeating this procedure with each of them, determines the position of the receiver and its deviation from true time. Therefore, 4 satellites are needed for a reliable GPS monitoring: 3 for the position triangulation and 1 for the time synchronization.

Galileo

Galileo is Europe’s own global navigation satellite system, providing a highly accurate, guaranteed global positioning service under civilian control. It is interoperable with GPS and GLONASS, the US and Russian global satellite navigation systems to increase performance and robustness of the navigation services. By oﬀering dual frequencies as standard, Galileo is set to deliver real-time positioning accuracy down to the metre range [19]. First satellite launches were in 2011 with operational satellites to validate the Galileo concept, by the end of 2016 initial services will be available and in 2020 the constellation system will be completed and like represented in Figure 2.16. It will consist of 24 operational satellites plus six in-orbit spares positioned in three circular Medium Earth Orbit(MEO).

Figure 2.16: Galileo constellation [19].

23 2. Background Theory

Galileo MEOSAR satellites will be equipped to relay distress signals from emergency beacons according to the Cospas-Sarsat System. The incorporation of a response signal to the user informing that the situation has been detected and that help is on its way is a major upgrade to the existing system.

GLONASS

GLONASS is a satellite-based navigation system operated by the Russian Aeropspace Defence Forces that works alongside GPS (Global Positioning System) with a similar precision [20]. Since 2001, Russian Government approved a new program to modernize and restore the network that had stopped working during the crisis years (1989-1999). A big economic investment allowed the program to develop a full orbital constellation of 24 satellites in 2011.

COMPASS

The COMPASS also known as BeiDou Navigation Satellite System is a Chinese satellite navigation system. It consists of two separate satellite constellations: a limited test system that has been operating since 2000, and a full-scale global navigation system that is currently under construction [21]. The first BeiDou system consists of three satellites and offers limited coverage and applications. It has been offering navigation services, mainly for customers in China and neighbouring regions, since 2000. The second generation of the system, will be a global satellite navigation system consisting of 35 satellites that offer full coverage of the complete globe.

2.4.3 Cospas-Sarsat System

The International Cospas-Sarsat Programme is a satellite-based search and rescue distress alert detection and information distribution system. It is known as the system that detects and locates emergency beacons activated by aircraft, ships and hikers in distress [22]. It was established by Canada, France the United States and the former soviet Union in 1979. Many countries have joined the project as providers of ground segments or as user states providing a good coverage around the planet. Its mission is to provide accurate, timely and reliable distress alert and location data to help Search and rescue (SAR) authorities assist persons in distress. The objective is to reduce delays in the provision of distress alerts to SAR services and the time to locate the person in distress which increases the probability of survival. To achieve this objective, Cospas-Sarsat implement, maintain, co-ordinate and operate a satellite system capable of detecting distress alert transmissions from radio beacons. In Figure 2.17, the functionality of the system is represented.

It is composed of [22]: • Distress radio beacons like ELTs (Emergency Locator Transmitters) for avia- tion use, EPIRBs (Emergency Position Indicating Radio Beacons) for maritime

24 2. Background Theory

Figure 2.17: Corpas-Sarsat System [22].

use and PLBs (Personal Locator Beacons) for personal use, which transmit signals during a distress situation. • Instruments on board satellites in geostationary (GEOSAR) and low-altitude (LEOSAR) earth orbits, which detect the signals transmitted by distress beacons. • Ground receiving stations, referred to as Local Users Terminals (LUTs), which receive and process the satellite downlink signal to generate distress alerts. • Mission Control Centers (MCCs) which receive alerts produced by LUTs and forward them to Rescue Coordination Centers (RCCs), Search and Rescue Points Of Contacts (SPOCs) or other MCCs.

Satellites

A combination of LEOSAR, GEOSAR and MEOSAR systems are used in the Corpas-Sarsat system. Both contribute respective advantages to detection and location of activated distress beacons.

25 2. Background Theory

(a) LEOSAR: The Cospas-Sarsat LEOSAR system uses polar-orbiting satellites with a basic constrain of non-continuous coverage. Doppler positioning techniques are used to locate the distress beacon. It operates in two coverage modes: • Local Mode: When the satellite receives beacon signals, the Search and Rescue Processor (SARP) recovers the digital data, measures the Doppler frequency shift and time-tags the information. This information is stored and transferred to the any LEOLUT in view. This operation can also be performed by a repeater that reflects the beacon signal to the Earth, with the difference that the data processing will be on the ground. • Global Mode: The SARP system provides global coverage by storing data derived from onboard processing of beacon signals which is broadcasted to LEOLUTs that were not visible when the beacon was detected by the satellite. With this information, a global coverage is determined. Global mode reduces the activation response as the beacon signal does not need to be simultaneous with a LEOLUT visibility. (b) GEOSAR: Search and rescue instruments on board geostationary satellites can be used to detect the current generation of Cospas-Sarsat beacons. The GEOSAR system consists of repeaters carried on board and associated ground facilities (GEOLUTs) which process the satellite signal. As it was explained previously, GEOSAR satellites cannot use the Doppler positioning techniques to locate distress beacons, therefore, the beacon location need to come from: • An internal or external navigation receiver encoded in the beacon message. • LEOSAR system Doppler processing. GEOSAR satellites’ incapacity to locate the distress beacon needs to be com- plemented with LEOSAR system which can calculate the location, provides excellent coverage of the polar regions (where geostationary satellites have bad coverage) and is less susceptible to obstructions. With this combination, the location is guaranteed with LEOSAR system and the fast response is achived using GEOSAR satellites. (c) MEOSAR: Cospas-Sarsat is upgrading its satellite system by placing search and rescue receivers on new satellites as American’s GPS satellites, navigation satellites of Russia (GLONASS) and European navigation satellites (GALILEO). This system update will dramatically improve both speed and location-accuracy for detecting beacons. This system will complement the LEOSAR and GEOSAR systems already operating. MEOSAR system will offer the advantages of both LEOSAR and GEOSAR systems without their limitations, transmitting the the distress message with a real time global coverage and independent location of the beacon. It will also provide a confirmation to the user that the distress message has been received. Figure 2.18 is a representation of the upcoming system. MEOSAR system was planned to be completely functional by the end of 2015 but it has not been confirmed if it is operational.

26 2. Background Theory

Figure 2.18: MEOSAR system concept [22].

Distress Beacons

Since February 2009, the Cospas-Sarsat system only detects and locates distress beacons operating at 406MHz. In the past, 121.5MHz beacons were used but they were considered obsolete because the 406MHz beacons speciﬁc design for the LEOSAR system improved its performance. Some requirements are needed such as stability of the transmitted frequency (406.0-406.1 MHz) and the inclusion of a digital message which allows the transmission of encoded data such as unique beacon identiﬁcation.

Local User Terminals (LUTs)

Two types of LUTs operate in the Cospas-Sarsat system: LEOLUTs from the LEOSAR satellite constellation and GEOLUTs from the GEOSAR satellite constellation. LUTs provide reliable alert and location data without restriction on use and distribution. Strict speciﬁcations and procedures have been developed to ensure that LUTs performance is reliable and can be used by SAR community.

Mission Control Centres (MCC)

MCCs have been set up in most countries operating at least on LUT. Their main functions are: • Collecting, storing and sorting the data from LUTs and other MCCs. • Providing data exchange in the Cospas-Sarsat system • Distributing alert and location data to associated RCCs.

27 2. Background Theory

The data that MCCs manage are system information and alert data. System information keeps the Cospas-Sarsat system operating eﬀectively providing accurate and timely alert data to the users. Alert data is the data derived from distress beacons with its location and coded information.

28 3 Design and Results

This chapter explains the procedure developed in order to design, simulate, manufacture and measure the antenna system. A preliminary section is needed to explain the material choice, the laboratory instruments and software used. Even though both antennas operate together, they have been designed, manufactured and tested independently. Thus, the GPS antenna is explained in the ﬁrst place, and then the PLB antenna.

3.1 Materials and laboratory instruments

3.1.1 Materials

A very sensitive choice in every antenna design is the selection of materials, but it is even more important when the antenna is textile. Textile antenna needs to be ﬂexible, low cost, easy to manufacture and suitable to be integrated into clothing. Following the microstrip structure, it is needed an electrical conductive fabric for the ground plane and the patches; and a fabric substrate with constant thickness and stable permittivity. An accurate determination of the electrical parameters for the fabrics (dielectric substrate and conductive textile) is crucial for correct antenna simulations and agree- ment with measurements [1].

• Textile substrates:

The textile substrate defines the dielectric between the antenna patch and the ground plane, therefore we need to select it according to its permittivity and thickness. As textile materials are not specifically used for antenna manufacturing, their permittivity is not previously characterized. We used |S11| measurements of two microstrip antennas with two different dimensions of the patches for the study of 9 felts showed in Figure 3.1.

29 3. Design and Results

(a) Dark green (b) Stripped (c) Sq. green (d) Green (e) Brown

(f) Pink (g) Orange (h) Blue (i) Red

Figure 3.1: Tested felts

Once we know the operating frequency of the two antennas with each felt, we used CST simulations to estimate their permittivity. Two diﬀerent calculations were used to produce a more reliable value. The thickness of the fabric is also important as it has a direct inﬂuence on the bandwidth. In Table 3.1 the measured parameters are displayed.

Thickness(mm) r1 r2 r

Dark green 1.2 1.21 1.16 1.185 Striped 1.85 1.22 1.31 1.265 Squared green 0.5 1.25 1.22 1.235 Green 0.4 1.64 1.80 1.72 Brown 1 1.40 1.55 1.475 Pink 0.9 1.42 1.38 1.40 Orange 1 1.27 1.24 1.255 Blue 1 1.23 1.18 1.205 Red 1 1.33 1.37 1.35

Table 3.1: Felt’s characterization.

Finally, we decided that the most appropriate felt is Red from Figure 3.1i which has r=1.35 and 1mm of thickness. The reasons of this choice were the stable calculation of the permittivity as the two calculations do not diﬀer considerably, the thickness of 1mm which is not easily compressed and its robustness which is stable but easily bended. We decided to use two sewed felt layers for the GPS and three for the PLB to increase the antenna’s bandwidth. • Conductive fabric:

30 3. Design and Results

Conductive fabric is used as a ground plane and patch of both antennas. Many examples from the literature using Shieldex woven fabrics led us to contact this company for more information. They provided us with some samples to use for the antennas manufacturing. We have used two different fabrics provided by Shieldex: Norra Dell and Zell. They are woven fabric for general use (outside skin for EMI/RFI fabric over foam gaskets, shielding material for laminated flat I/O shielding panels, base material for EMI/RFI garments, EMI/RFI cable shielding). Nora Dell is a nickel copper silver plated polyamide fabric with a resistivity around 0.009W/ with 100dB shielding effectiveness from 30MHz to 10GHz. It is shown in Figure 3.2a. Zell is a tin copper silver plated polyamide ristop fabric with a resistivity around 0.02W/ with 80dB shielding effectiveness from 300MHz to 10GHz. It is shown in Figure ??. Both work in a temperature range from -30°C to 90°C. Both fabrics are suitable for any of the two operating frequencies we are working at, but to ensure its correct performance, we used Nora Dell for the PIFA antenna (lower frequency) and Zell for the GPS antenna and its feeding circuit.

(a) Shieldex Nora (b) Shieldex Zell

Figure 3.2: Conductive fabric used.

• Conductive thread:

Conductive thread is used in the manufacturing process to connect the patch of the antenna to the SMA connector pin during the measurements. It is also integrated as a deﬁnitive connector from the 90°feeding circuit to the patch of the GPS antenna. Figure 3.3 shows the conductive thread used. We have used Shieldex 235/34 a 99 % pure silver polyamide 6.6 ﬁlament yarn.

Figure 3.3: Conductive thread used.

31 3. Design and Results

3.1.2 Laboratory

Some instruments have been used during the manufacturing, material testing and measuring process in the laboratory. The most important instruments are explained in Table 3.2.

Instruments Model Image Function

Signal Generator HP 8665B Generating RF signal at a ﬁxed frequency as the input of the antenna

Vector Network Anritsu MS2026B Measurement of S pa- Analyser rameters of electrical networks

Near Field Scanner EMSCAN-RFxpert Real time performance results for antennas

Table 3.2: Instruments used in the lab [23], [24], [25].

Other elements used during the lab manufacturing and measuring were:

• SMA connectors: coaxial RF connectors used to feed the antennas. Working up to 18GHz with 50W impedance [26]. • Aluminium sheet • Coaxial cables • Power divider • Phase shifter • Textile glue • Epoxy • Solder

3.1.3 Software

Diﬀerent software tools have been used to design the model of the antenna, simulate the electrical parameters and evaluate and plot the results of its performance. These software are CST Microwave Studio 2014, Matlab R2014b and RFxpert.

• CST Microwave Studio: It is a software tool for the fast and accurate 3D simulation of high frequency devices. It enables the fast and accurate analysis of antennas, ﬁlters, couplers, planar and multi-layer structures and SI and EMC eﬀects [27].

32 3. Design and Results

3D Electromagnetic simulation integrates a wide variety of solvers into one interface enabling an easy selection of the most appropriate solver for each problem: Tran- sient, Frequency domain, Eigenmode, Resonant, Integral Equation, Asymptotic and TLM. We have used this software to implement the antennas’ models and adjust its parameters according to the simulation results. Time domain solver predicts with a single simulation the response of an antenna over a wide bandwidth enabling a clear evaluation of the results while maintaining excellent accuracy. Results data can be easily exported to an ASCII ﬁle for further evaluation and im- porting them to another software as Matlab.

• RFxpert: It is used to run the RFxpert near field scanner which instantly characterizes antennas without the need for a chamber providing far-field patterns, bisections, EIRP, S11 graph, gain and efficiency. A Circular Polarization option can be selected to calculate the right and left hand circularly polarized patterns and display axial ratio patterns. Large scan area allows designers to test antennas up to 32cm x 32cm and frequencies between 300MHz and 6GHz [24]. We have used the near field scanner to evaluate the performance of the manufactured antennas, specifically far field patterns in both 3D and bisections representations. Comparing them with the previously simulated results, we can determine if the behaviour is reliable. Results data can be easily exported to an excel file for further evaluation and im- porting them to another software as Matlab.

• Matlab: It is a high-level language and interactive environment used by millions of engineers and scientists worldwide. It lets you explore and visualize ideas and collaborate across disciplines including signal and image processing, communications, control systems, and computational ﬁnance [28]. We have used this software to evaluate the results obtained with CST simulations and RFxpert.Its high precision and versatile environment allow us to process and compare graphs to determine the reliability of each antenna.

3.2 GPS Antenna

In our study we designed and manufactured a textile microstrip antenna that to work at the GPS bandwidth (1.564 - 1.587 GHz) with right-hand circular polarization.

3.2.1 Circularly Polarized Microstrip Antennas

Using circular polarization provides some important advantages according to GPS functionality. As circularly polarized antennas transmit in all planes, it allows the transmitting and receiving antenna have a diﬀerent orientation and increases the probability of a successful link. Mobility, weather conditions, reﬂectivity, absorption and multipath have a direct impact on the polarization of the signal which are easily

33 3. Design and Results tolerated with a circular polarization. A right hand circular polarized signal becomes a left hand circular polarized after a reﬂection (and vice-versa). In an antenna, circular polarization can be achieved through a single feed in a diagonal place of an non-symmetric patch or using two feeds in the same patch with diﬀerent phases [7].

Single feed circularly polarized microstrip antennas

To avoid complexities related to dual-feed designs, a single feed can also be used to achieve circular polarization. One way to accomplish this is to feed the patch at a single point in which two orthogonal modes are generated. The 90° phase diﬀerence is induced by using this asymmetric conﬁguration. Some of the the most used techniques are trimming the ends of two opposite corners of a square patch and feed points 1 or 3 as in Figure 3.4a and cutting very thin slots as in Figure 3.4b.

(a) Trimmed square [7] (b) Square patch with thin slots [7]

Figure 3.4: Circular polarization techniques.

A single feed circularly polarized antenna is simpler than a dual-feed design and very useful when the space is a restriction of the design. On the other hand, the tolerances are very stric and the bandwidth narrow. For this reason, in this thesis we have decided to use a dual-orthogonal feed.

Dual-orthogonal feed circularly polarized microstrip antennas

The most common and direct way to generate a circular polarization is using a dual-feed technique. It is needed to generate two orthogonal modes with 90° phase diﬀerence between them like in Figure 3.5.

34 3. Design and Results

Figure 3.5: Rectangular microstrip antenna with two orthogonal feeds [29].

There are many well known power divider circuits which can be used to split the input power in two and feed the patch. Some of them, which have been successfully employed in a feed network of a patch, are: • The 180-Degree Hybrid • The Wilkinson Power Divider • The T-Junction Power Divider • The Quadrature 90°Hybrid But, as we need a 90° phase diﬀerence between the two output ports, in this thesis we have decided to use the Quadrature 90°Hybrid.

3.2.2 Quadrature 90°Hybrid

As it is described in [8], Quadrature Hybrids are 3dB directional couplers with 90° phase diﬀerence between the two outputs. Basically, the input power is divided equally to the two output ports (2 and 3) with 90°phase shift when all four ports are matched. No power is coupled to port 4 (the isolated port). It can be easily implemented with microstrip lines as it is shown in Figure 3.6.

Figure 3.6: Geometry of a 90°hybrid [8].

The high symmetry of the branch-line coupler, as any port can be used as the input port with the output ports at the opposite part of the junction. Its scattering matrix proves this symmetry: 0 j 1 0 h i −1 j 0 0 1   S = √   (3.1) 2 1 0 0 j 0 1 j 0

35 3. Design and Results

The bandwidth of a branch-line hybrid is limited to 10%–20% which won’t be a problem in our design because the GPS bandwidth is narrower.

3.2.3 Design and Simulation

The GPS antenna is designed according to the previous explanations, by using a dual-orthogonal feed circularly polarized microstrip antenna with a quadrature 90°hybrid feeding circuit. In order to reduce the dimensions of the antenna, the circuit is placed as a microstrip line below the ground plane. The design and simulations were done in separate parts: ﬁrst the patch, following with the feeding circuit 90°hybrid and ﬁnally the complete model with both parts.

Patch

The ﬁrst step to design a microstrip antenna was to determine the patch’s dimensions W and L which are inversely proportional to the frequency:

1 c0 (fr)010 = √ = √ (3.2) 2L µ 2L r

1 c0 (fr)001 = √ = √ (3.3) 2W µ 2W r As both operating frequencies needed to be the same, the patch is a square where: c λ L = √0 = √ = W (3.4) 2f r 2 r

1 Assuming f=1.575GHz and r=1.35 , L = W =81.91mm.

On the basis of these values, we built the simulation model to adjust the patch’s dimensions to the exact GPS frequency and select matched feeding points. The model follows the geometry of microstrip antennas which consists of three squared layers: ground plane, substrate and patch. Figure 3.7 is a representation of the top view and side view of the antenna model with all adjusted parameters.

(a) Top view of the antenna (b) Side view of the antenna

Figure 3.7: GPS antenna model.

1 In section 3.1, it is discussed the election of the materials and how the value of r was measured.

36 3. Design and Results

With the following dimensions:

Parameters Value(mm)

hg 0.2 hs 2 hp 0.2 W-patch 78 L-patch 78 W-substrate 100 L-substrate 100

Table 3.3: GPS antenna ﬁnal dimensions.

The feeding points of the patch are the most determinant parameters to achieve a good matching. After some attempts, satisfactory results were achieved with Port 1=(11.3, 0)mm and symmetrically Port 2=(0, 11.3)mm.

This conﬁguration was studied using the Time Domain Simulation of CST Mi- crowave Studio. The antenna was evaluated in terms of S parameters, Electric ﬁeld distribution and Far Field.

• S-parameters: S-parameters illustrate the return loss of the antenna and the coupling between port 1 and 2. As it is shown in Figure 3.8, S11 has a good performance for the whole GPS bandwidth (lower than -15dB) which indicates that the feeding points are appropriate and that the patch’s dimensions are suitable for the operating frequency. S21 determines the coupling between port 1 and port 2, since the level is lower than -35dB, the ports do not interfere each other.

Figure 3.8: S-parameters simulated of the GPS antenna.

• E-Field distribution:

37 3. Design and Results

Electric ﬁeld simulations at the operating frequency 1.57GHz are illustrated in Fig- ure 3.9. TM10 is excited using Port 1 and with port 2, TM01.

(a) Port 1 E-ﬁeld (b) Port 2 E-ﬁeld

Figure 3.9: Electric ﬁeld simulations at 1.57GHz.

Combining both modes and a 90 ° phase diﬀerence, circular polarization conditions will be achieved. Figure 3.10 shows the evolution of the electric ﬁeld distribution with the time.

Figure 3.10: Electric ﬁeld distribution evolution with the time.

• Far field: Far field simulations represent the radiation pattern of the microstrip antenna. The main lobe is formed in z-direction with 8.15dBi and 75.5 degree 3dB beam width and the back lobe is -8.4dBi. The difference between the main lobe, which will be the useful power, and the back lobe which will be dissipated power towards the body, gives us a figure of merit of the antenna. The radiation pattern is represented in 3D in Figure 3.11a and in 2D in Figure 3.11b when φ =0.

(a) 3D far ﬁeld radiation pattern (b) Polar representation when φ=0°)

Figure 3.11: Far ﬁeld simulations from port 1 of the GPS antenna.

38 3. Design and Results

Combining the two ports with a 90 ° phase diﬀerence, the obtained radiation pattern is shown in Figure 3.12.

(a) Right hand circular polarization (b) Polar representation when φ=0°

Figure 3.12: Far ﬁeld of a patch antenna with two orthogonal ports and 90° phase shift.

90° feeding circuit

Once the performance of the designed patch is reliable, we proceeded to design the 50W branch-line quadrature hybrid working at the ﬁxed GPS frequencies. We used a probe feed to connect the feeding points of the patch to the output ports 2 and 3 of the circuit, therefore we needed to adapt the theoretical geometry of the Quadrature hybrid to our design. Figure 3.13 shows the geometry of the complete antenna with the feeding circuit connected to the patch.

Figure 3.13: Complete antenna’s geometry.

We used the same materials (substrate and conductive) as for the patch so r=1.35, λ Z0 =41.09mm and if Z0=50W, √ = 35.35W. Using CST macro calculations, the 4 2 Z necessary width of the lines in order to achieve the needed impedance Z and 0 0 2 can be determined. The feeding points of the patch determined where the output ports needed to be, so we needed additional lines from the junction to these points. It is very important that both additional lines have the same length in order to respect the phase diﬀerence achieved with the quadrature hybrid. After an optimization process, the ﬁnal design is the one illustrated in Figure 3.14 with the values indicated in Table 3.4.

39 3. Design and Results

Parameters Value(mm)

hg 0.2 hs2 2 hc 0.2 w50 4.3 w35 6.3 W-substrate 100 L-substrate 100

Figure 3.14: Top view of the 90°circuit. Table 3.4: Antenna’s dimensions.

From the S-parameters, the performance of the circuit can be explained: • S-parameters: The S-parameters represent the distribution of the input power at each port. The circuit has been designed at GPS frequency band and with an adequate matching in all 4 ports, which is represented in Figure 3.15.

Figure 3.15: S-parameters of the 90° feeding circuit.

In Figure 3.20a, |S12| and |S13| show that input power from port 1 is equally dis- tributed to port 2 and 3 as both parameters are -3dB in the whole GPS bandwidth. |S11| is well matched with values below -10dB In Figure 3.16b, it demonstrated that a phase diﬀerence of 90° at the operating frequency between S12 and S13 can be obtained.

40 3. Design and Results

(a) , and |S11| |S12| |S13| (b) S12 and S13’s phase Figure 3.16: Representative S-parameters of the circuit.

The matching of port 4 at the Quadrature 90°Hybrid, presents difficulties. Different techniques can be used to adapt a port such as lumped elements and stubs with a terminated resistive load. This configurations are not appropriate for our design since one of our main goals of the project was to use 100% textile materials.

Analysing |S14| results from the Figure 3.15 we can see that the power from port 1 to port 4 of the circuit is very small. Instead of dissipating this power we decided to make it radiate. Accordingly, we designed a load patch connected with a microstrip line to port 4 which radiates all power coming from the circuit. This load patch needs to be matched and operating at the same frequency as the circuit (1.565-1.585GHz) which can be checked with the |S11| representation from Figure 3.18. We designed it separately to make sure its correct performance and adjusted its dimensions, which are represented in Figure ?? with the values displayed in Table 3.5.

Parameters Value(mm)

W-patch2 78 L-patch2 78 L-slot 32 W-slot 2 w50 4.3

Figure 3.17: Load patch’s geometry. Table 3.5: Dimensions of the load patch.

41 3. Design and Results

Figure 3.18: |S11| of the load patch.

The separation between the load patch and the circuit is crucial because the electric ﬁeld going through the circuit must not be couplet to the patch. Therefore we extended the connecting line between port 4 and the secondary patch. To increase the length of the connecting line, we also needed to increase substrate’s width from 100mm to 150mm, achieving the ﬁnal designed from Figure 3.19.

Figure 3.19: Geometry of the ﬁnal circuit.

Complete antenna

The final step of the antenna’s design was to join and unify patch and circuit’s geometry. As a consequence of the changes for the correct circuit’s design, we needed to adapt the patch design to the new substrate’s width which does not affect any other parameter preciously calculated. Therefore, the definitive model is represented in Figure 3.20 with the values from Table 3.6.

42 3. Design and Results

(a) Top view of the complete antenna (b) Back view of the complete antenna

Figure 3.20: Geometry of the complete antenna.

Parameters Value(mm)

hg 0.2 hs 2 hp 0.2 W-patch 78 L-patch 78 w50 4.5 w35 7.2 W-patch2 30 L-patch2 81.5 W-substrate 150 L-substrate 100

Figure 3.21: Side view of the complete Table 3.6: Dimensions of the antenna. complete antenna.

To corroborate the independent response of each part, we haver performed a last simulation including all the individual parts. • S-parameters:

The ﬁnal design |S11| simulation allows us to evaluate that the antenna operates at the decided frequency ban after adding the feeding circuit. In Figure 3.22, it is also noticeable that the -15dB bandwidth has become narrower compared to the initial patch’s design but still functional for the central GPS’ frequencies.

43 3. Design and Results

Figure 3.22: |S11| of the complete design.

• Axial ratio: The axial ratio of a circularly polarized antenna is lower than 3dB. The axial ratio is very frequency dependent because an exact 90° phase diﬀerence can only be achieved for a speciﬁc frequency since it depends on the length of the lines of our circuit. Therefore, and as it is shown in Figure 3.23, a reliable axial ratio covers the GPS’ bandwidth.

Figure 3.23: Axial Ratio of the complete design.

• E-ﬁeld distribution: Circular polarization can be seen with the electric ﬁeld evolution with the time.

44 3. Design and Results

Figure 3.24: Electric ﬁeld distribution evolution with the time.

• Far ﬁeld:

The radiation pattern is represented in 3D in Figure 3.25a and in 2D in Figure 3.25b when φ=0. The main lobe is formed in z-direction with 8.01dBi and 77.5 degree 3dB beam width and the back lobe is 0.25dBi. The back radiation is large because of the load patch implemented in the circuit and radiating backwards. This factor should be taken into consideration when deciding the position of the antenna respect to the human body and the SAR consequences.

(a) 3D far ﬁeld radiation pattern (b) Polar representation (φ=0)

Figure 3.25: Far ﬁeld simulations of the complete antenna.

Comparing absolute far field simulations with circular polarized simulation we can see that is practically the same, confirming the good performance in terms of polarization. It is also remarkable from the 3D representation in Figure 3.26a and the polar representation in Figure 3.26b that the main difference is the dimension of the back lobe which is not important for the design that it is circularly polarized.

45 3. Design and Results

(a) 3D far ﬁeld radiation pattern (b) Polar representation (φ=0)

Figure 3.26: Far ﬁeld right circular polarization simulations of the complete antenna.

3.2.4 Manufacturing and matching

Two GPS antennas were manufactured following the selected parameters during the simulations explained in the previous section. The ﬁrst one is a microstrip antenna with dual-orthogonal feed but without the 90° feeding circuit integrated therefore, a commercial 90° phase shifter is used to achieve the 90° phase diﬀerence between the two ports. The second one is the full textile complete microstrip with the integrated 90° hybrid circuit. Taking the simulation values as a reference, variations of some parameters were made to ensure the correct performance of the antenna. All materials and instruments used for the manufacturing process are explained in Section 3.1.

Dual-orthogonal feed microstrip antenna

The configuration in Figure fig:rff represents the elements that the dual-orthogonal feed microstrip antenna needs to operate.

Figure 3.27: Conﬁguration of the dual-orthogonal feed microstrip antenna.

A basic microstrip antenna with dual-orthogonal feed has been manufactured. In the two feeding points SMA connectors were placed to feed the patch and for the measurements. Before the manufacturing process the patch dimensions to operate at GPS frequency were chosen and the positions for 50W were determined.

46 3. Design and Results

After obtaining the required dimensions from the simulations, we proceeded to sew together two 100x100mm layers of felt which constitute the substrate. Next, we sewed the patch made of Shieldex Dell with dimensions of 80x80mm. It is 2mm larger than the established dimensions during the simulations to adjust it to the correct frequency band.

To measure the patch’s dimensions, we used the Vector Network Analyser (VNA). We placed the ongoing antenna on an aluminium sheet with a SMA connector soldered through it. We used some Shieldex conductive thread to connect the SMA pin to the patch going through the substrate. With this conﬁguration, we determined a matched feeding point (one at a time) by changing the position of the conductive thread respect to the patch. Once a good matching was achieved, the operating frequency was adjusted cutting the patch. This procedure allowed us to double check the simulation dimensions and adjusted them until an adequate performance was reached and measured with the VNA as in Figure 3.28. In Table 3.7 the values of the parameters of the microstrip antenna are displayed following the notation represented in Figure 3.7a.

Parameters Value(mm)

hg 0.2 hs 2 hp 0.2 W-patch 78.5 L-patch 78.5 W-substrate 100 L-substrate 100

Figure 3.28: |S11| measured with the Table 3.7: Dimensions of the VNA. manufactured patch.

Once the performance of the patch was demonstrated, we ﬁnished the manufacturing: the textile Shieldex ground plane was sewed and a felt layer was added to increase the robustness of the antenna, two SMA connectors were placed through holes on the ground plane and substrate and soldered to the patch.

47 3. Design and Results

(a) Top view of the patch (b) Back view of the patch

Figure 3.29: Manufactured textile microstrip patch antenna.

Dual-orthogonal feed microstrip with a 90° feeding circuit integrated

We manufactured the complete antenna: full textile dual-orthogonal feed microstrip with a 90° circuit underneath connected using conductive thread. The integration of the feeding circuits allows that the complete textile antenna operated without any additional non-textile element. Figure 3.30 represents the needed conﬁguration.

Figure 3.30: Conﬁguration of the complete full textile dual-orthogonal feed microstrip.

The patch was manufactured following the same procedure as the basic microstrip explained in the section 3.2.4, with the only diﬀerence of the substrate’s width which was 150mm instead of 100mm. The dimensions of the patch were checked once again to operate at the proper frequency and accordingly the feeding points resulting in the manufactured patch antenna in Figure 3.31a. We made a hole at the ground plane for each feeding point where the conductive thread goes through to connect the patch to the feeding circuit as it is shown in Figure 3.31b.

48 3. Design and Results

(a) Top view of the patch (b) View of the ground plane of the patch

Figure 3.31: Manufactured fully textile microstrip patch.

With the patch perfectly manufactured, we continued with the 90°feeding circuit. The main characteristics of this circuit is that the microstip lines forming the junc- λ tion need to be exactly s which also means a 90° phase difference at 1.575GHz 4 with r=1.35. The Vector Network Analyser was used to measure the phase difference between the beginning and the end of a microstrip line and adjusting it to 90° λ showed in Figure 3.32. The final value for a s line was 43mm. 4

(a) Initial phase measurement (b) Final phase measurement

λ Figure 3.32: Phase’s measurements to determine s . 4

After determining the length of the lines, we needed to decide its width. In this case, we used the VNA to measure the impedance and represent it with the Smith Chart. We adjusted each line’s width achieving the ﬁnal values of 7.5mm for 35W and 4.5mm for 50W. With the previous parameters, we implemented the circuit and analysed its performance. We measured the S-parameters of all 4 ports using the Vector Network Analyser. While measuring S-parameters of two ports, the other two ports need to be matched with 50W loads. In Figure 3.33a, port 1 and 2 are being measured while port 3 and 4 are matched. One of the measurements is shown in Figure 3.33b.

49 3. Design and Results

(a) Measurement of port 1 and 2 (b) |S22| measurement Figure 3.33: Circuit’s S-parameters measurements

To manufacture the load patch, we followed the same procedure as the main patch. Adjusting the dimensions with the |S11| results to match the feeding point and the operating frequency with the substrate placed over the aluminium sheet and connecting the patch with conductive thread using a probe feed. This process is shown in Figure 3.34a with load patch being matched and its ﬁnal measurement in Figure 3.34b. Once the feeding point was selected, two slots were cut forming the feeding microstip line which was attached to the circuit’s port 4.

(a) Load patch matching (b) |S11| measurement Figure 3.34: Load patch matching and measurement.

Finally, the final circuit was manufactured with the Shieldex fabric and attached to the load patch. The substrate was made with two layers of felt sewed together. In this case, we did not sew the circuit and the load patch to the substrate because their small dimensions made it difficult to achieve a flat layer. So we pasted them with textile glue. When both parts were ready we sewed them together and added the conductive thread from the patch feeding points to the circuit ports through the substrate and the ground plane’s holes. To finish we soldered a SMA connector to the microstrip line of port 1 to feed the circuit and therefore, the antenna. Figure 3.35a and Figure 3.35b show top and back view of the finished fully textile antenna and its feeding circuit.

50 3. Design and Results

(a) Top view (b) Back view

Figure 3.35: Manufactured fully textile complete microstrip antenna with a 90° feeding circuit integrated.

The completed antenna was measured to check that it was matched in the whole GPS band. This measurement is shown in Figure 3.36.

Figure 3.36: |S11| complete antenna

3.2.5 Far ﬁeld measurements

We analysed the radiation pattern of the two GPS antennas manufactured. We used the RFxpert Near Field Scanner to measure both antennas’ behaviour. To summarize the results, all graphs displayed are at the considered center frequency of the GPS bandwidth: 1.575GHz.

Dual-orthogonal feed microstrip antenna

To analyse the performance of this antenna we used a 90° phase shifter, according to the conﬁguration displayed in Figure 3.37.

51 3. Design and Results

Figure 3.37: Conﬁguration for the measurements of the dual-orthogonal feed microstrip antenna.

A comercial phase shifter was adjusted for a exact 90° phase at the GPS frequency with the VNA. Using RFxpert near field scanner, the radiation pattern of the antenna was measured. Comparing the two 3D plots, it is clear that its power radiation majority is right hand circularly polarized as both measurement, total radiation pattern showed in Figure 3.38a and RHCP radiation pattern showed in Figure 3.38b, are very similar. In the 2D plots, the total (blue), right hand (magenta) and left hand (red) far fields are shown in Figure 3.38c when φ=0° and in Figure 3.38d when φ=90°. There is a difference of 20dB between RHCP and LHCP for all θ directions.

(a) 3D total radiation pattern (b) 3D RHCP radiation pattern

Figure 3.38: Far ﬁeld measured of the manufactured dual-orthogonal feeding microstrip antenna.

The axial ratio is represented depending on the angle θ for two diﬀerent values of φ, 0° and 90°, in Figure 3.39. When the axial ratio is lower than 3dB, it indicates that the antenna is circularly polarized. For angles around θ= 0°, which is the more

52 3. Design and Results important direction because the maximum radiated power is in this direction, values of the axial ratio ate lower than 3dB, therefore it is circularly polarized.

Figure 3.39: Axial ratio of the manufactured microstrip antenna measured with the near ﬁeld scanner

Dual-orthogonal feed microstrip with a 90° feeding circuit integrated

In this case we only needed to connect the signal generator to the antenna port and place it on the near ﬁeld scanner, shown in Figure 3.40, as the feeding circuit is fully textile and integrated in the microstrip antenna.

Figure 3.40: Conﬁguration of the complete full textile dual-orthogonal feed microstrip.

The results form the radiation patterns measurements of the microstrip antenna with the integrated circuit, are shown in Figure 3.41. The 3D representation shows a more directive radiation than the microstrip antenna without the feeding circuit from Figure 3.38, and a similar directivity 7dBi. Also, the total radiation pattern from Figure 3.41a and the RHCP radiation pattern from Figure 3.41b, are almost identical which can easier be seen in the 2D representations from Figure 3.41c and 3.41d, as the maximum directivity from the right polarization and the total radiation pattern are 7dBi for both cases of φ=0° and φ=90°.

53 3. Design and Results

(a) 3D total radiation pattern (b) 3D RHCP

Figure 3.41: Far ﬁeld measured of the manufactured microstrip with 90° feeding circuit integrated.

The axial ratio measurement shown in Figure 3.42 indicates that the antenna is not circularly polarized as its values are not lower than 3dB. The axial ratio measurements from the microstrip antenna with the integrated circuit in Figure 3.42 have increased more than 2dB in all θ directions, compared to the the axial ratio measurements from the microstrip antenna without the circuit represented in Figure 3.39. This measurements indicate that the integration of the circuit does not have a good performance as the circular polarization is compromised.

Figure 3.42: Axial ratio of the measured microstrip antenna with the integrated feeding circuit.

54 3. Design and Results

3.3 PLB - Personal Locator Beacon

According to Cospas-Sarsat speciﬁcations for Personal Locator Beacons, ,the an- λ tenna needs to operate at 406-406.1MHz. At this frequency, s = 317.76mm which 2 is too large to be integrated with the body shape, therefore the same design strat- egy as for the GPS antenna could not be followed. Instead of a squared patch, we decided to use a Planar Inverted F-Antenna.

3.3.1 PIFA - Planar Inverted F-Antenna

A Planar Inverted F-Antenna is a short-circuited radiating patch or wire to the antenna’s ground plane with a shorting pin and can resonate at a fixed operating λ λ frequency with a size of s instead of s [30]. 4 2 Because the patch is shorted at the end, the current at the end of the patch antenna is no longer forced to be zero. As a result, this antenna actually has the same current- voltage distribution as a half-wave patch antenna. However, the fringing fields which are responsible for radiation are shorted on the far end, so only the fields separated enough from the short pin radiate. Consequently, the antenna gain is reduced, but the patch antenna maintains the same basic properties as a half-wavelength patch even with a size reduction of 50%. The Planar Inverted-F antenna (PIFA) is increasingly used in the mobile phone market, mainly because of two advantages: reduction of the space needed (λs/4 instead of λs/2) by reducing electromagnetic wave power absorption and enhancing antenna performance (good SAR)[30]. It is popular because it has a low profile. These characteristics were decisive in the selection of this model to design the PLB antenna which needs to be small enough to be integrated in the body and low backward radiation.

3.3.2 Design and Simulation

PLB antenna has designed as a PIFA operating at 406MHz. Once its performance was considered reliable, it was integrated on a belt with two identical patches, one at the front and another at the back. The belt is thought to be worn around the body like in Figure3.43. Therefore the bending produced by the shape of the body needs to be taken into consideration. The design has been done in two parts: ﬁrst it has been designed a ﬂat single PIFA and then it has been adjusted to be integrated into a belt around the body.

55 3. Design and Results

Figure 3.43: Representation of the worn belt.

PIFA

As explained before, the patch of a PIFA needs to be correctly dimensioned according to its operating frequency.

λ L = √0 (3.5) 4 r

Assuming f=406MHz and r=1.35 (we used the same materials as for the GPS antenna), then L =158.88mm. Taking this value as a reference, we proceeded to design the model presented in Figure 3.44 and Figure 3.45 with the values displayed in Table 3.8.

Figure 3.44: PIFA’s geometry.

56 3. Design and Results

Parameters Value(mm)

hg 0.2 hs 2 hp 0.2 W-patch 55 L-patch 158.2 W-substrate 90 L-substrate 300

Figure 3.45: PIFA’s Top view. Table 3.8: PIFA’s ﬁnal dimensions.

Some simulations were used to determine this parameters’ values as well as the feeding point to achieve a good matching:(-60,0)mm. We can evaluate the performance of the antenna with the following ﬁgures of merit: • S parameters:

Figure 3.46 shows that the antenna operates at 406MGz where the |S11| is bellow -10dB. It can be observed that the bandwidth is narrow because the patch’s width λ of the designed antenna is much smaller than s . 4

Figure 3.46: |S11| of the PIFA.

• E-ﬁeld distribution: Electric ﬁeld representation shows the typical distribution of a PIFA, with the shorted part at one edge of the patch

57 3. Design and Results

Figure 3.47: PIFA’s electric ﬁeld distribution.

• Far ﬁeld: The radiation pattern of the designed PIFA is represented in Figure 3.48. The main lobe is 3.85dBi and 221.2° 3dB beam width and the back lobe is -0,8dBi. The back lobe level is very low which is one of the advantages of PIFA’s design.

(a) 3D radiation pattern (b) Polar representation when φ=0

Figure 3.48: Far ﬁeld simulation of the PIFA.

PIFA integrated in the belt

To simulate the total antenna we used a model which consists of a elliptical cylinder of a 400mm width, 150mm length and 400mm height, that represents the human body. The antenna has a ground plane and substrate with elliptical shape around the body model and two shorted patch, one at the front and another at the back, as it is illustrated in Figure 3.49 with the values displayed in Table 3.9. Human body has diﬀerent relative permittivities depending on the body tissues: muscle(57), skin(20-38) and fat(5), at 400MHz . The model used is an homogeneous elliptical cylinder with r=30 [31]. Dielectric losses were added to quantify the dissipation of electromagnetic energy. The employed loss tangent was tan δ=0,001 which is the typical value at 400MHz [32].

58 3. Design and Results

(a) Belt geometry (b) Belt integrated to the lossy body

Figure 3.49: Belt conﬁguration.

Parameters Value(mm)

hg 0.2 hs 2 hp 0.2 Body’s height 400 W-patch 55 L-patch 156.92 W-substrate 90 L-body 400 W-body 150

Table 3.9: Dimensions of the ﬁnal PIFA integrated to the body model.

We can evaluate the antenna’s performance analysing the following simulated results: • S-parameters:

|S11| is represented in Figure 3.51a. Due to the lossy body the bandwidth has become narrower than at the original PIFA and the frequency was shiftet. Adjustments were made on the patch’s length to operate at the decided frequency.

59 3. Design and Results

Figure 3.50: |S11| simulated of the PIFA integrated to the body model.

• Far field: The radiation pattern of one patch of the integrated PIFA is represented in 3D in Figure 3.51a and in 2D when φ=0° in Figure 3.51b. It has a main lobe of 3.14dBi and a radiation efficiency of -0.31dB. The back lobe is -2dB, this value is lower due to the effect of the absorption of the body.

(a) 3D representation (b) Polar representation (Phi=0)

Figure 3.51: Far ﬁeld simulations

Combining both patches, we can achieve a more omnidirectional radiation pattern. It is represented in 3D in Figure 3.52a and in 2D when φ=0° in Figure 3.52b. The main lobe of 4.03dBi and the radiation eﬃciency is -0.34dB.

60 3. Design and Results

(a) 3D representation (b) Polar representation when φ=0°

Figure 3.52: Far ﬁeld simulations

In real conditions, different phases at both patches can be employed to achieve the radiation at different directions. Figure 3.53 shows the radiation pattern for 4 values of phase difference: 45°, 90°, 135° and 180°.

(a) 45° phase diﬀerence (b) 90° phase diﬀerence

Figure 3.53: Polar radiation pattern with diﬀerent phases.

61 3. Design and Results

3.3.3 Manufacturing and matching

The PIFA was manufactured and integrated in a belt as explained before. Taking the simulation values as a reference, we used the measurement of some parameters to ensure the correct behaviour of the antenna. First of all, we manufactured the substrate of the belt. To form the belt we sewed three rectangular layers of felt of 900mm x 90mm. A sewing machine was used (with my grandmother’s help) to achieve a stronger structure. A hook and loop fastener was added to attach both edges of the belt around the body. This process is represented in Figure 3.54.

Figure 3.54: Belt sewing procedure.

The next step was dimensioning the patch so the operating frequency of the PIFA was 406MHz. We used the VNA to measure |S11| and to adjust the patch’s length. As it is explained before, we used an aluminium sheet with an SMA connector soldered as a ground plane for the measurements and a conductive thread from the SMA pin to the patch. We also needed to short the patch sewing a line of conductive thread from the patch through the substrate and making sure that had a good connection with the aluminium sheet. Selecting the proper feeding point was also needed. Figure 3.55 shows the front view of the ﬁnal belt and its |S11| measurement.

(a) Front view of the PIFA (b) |S11| of the PIFA

Figure 3.55: |S11| of the PIFA.

62 3. Design and Results

Once the patch was properly dimensioned, we sewed the ground plane of Shieldex and the other shorted patch at the opposite side of the belt. We solder a SMA connector to each patch going through the ground plane and the substrate. The ﬁnal belt is presented in Figure 3.56a and its top view in Figure 3.56b.

(a) Perspective view of the belt (b) Top view of the belt

Figure 3.56: Manufactured belt.

3.3.4 Far ﬁeld measurements

We measured the manufactured PLB antenna. We used the RFxpert Near Field Scanner to evaluate its performance. Because of RFxpert’s functioning it was not possible to make on-body measurements therefore, they were performed in the next conditions: the belt was positioned on the scanner with two arms on it covering the belt total surface scanned simulating the eﬀect the body might have. All results displayed are at the operating frequency: 406MHz.

3D radiation pattern is the expected as it is very similar to the simulations. We can see that the radiated power from Figure 3.57 is larger (almost 5dBi) than the simulations represented in Figure 3.51a because it was not measured with a whole lossy body behind it.

63 3. Design and Results

(a) Total radiation pattern, combination of θ and φ polarizations.

(b) θ polarization (c) φ polarization

Figure 3.57: 3D radiation pattern measured of the PIFA.

2D representation in Figure 3.58 show that θ and φ polarizations are completely exclusive accordingly to the 3D representations.

(a) φ=0° (b) φ=90°

Figure 3.58: 2D radiation pattern.

64 4 Conclusion and Future Work

This chapter describes a general conclusion based upon the work done during this thesis project and provides suggestions for next steps in order to continue the work. The intention is to point out the most relevant ideas of the thesis and then consider how to build upon them in future researches.

4.1 Conclusion

The achieved results lead to think that the designed textile antennas could be used as GPS and PLB antennas for the Cospas-Sarsat International System. Both antennas have resulted to be working at the established and demanding operational frequency with a good matching proving correct designs and simulations. Radiation patterns from the simulations and calculated using the near ﬁeld scanner showed a resemblance which reﬂects the correct study through the literature, design, simulations and manufacture.

Even though, the achieved results regarding the axial ratio of the GPS antenna do not fulﬁl the established requirements for a proper circular polarization, they indicate that with a more precise manufacturing process of the feeding circuit they would have been accomplished. Because of the frequency dependence of the circuit, the variation of one millimetre could be the cause of this problem. As the most innovative part of the thesis and non literature available, it is accepted as an important result which needs to be more developed.

PLB antenna performance is promising because of its radiation pattern and a complete adaptability to the belt geometry around the body. Further studies need to be performed in order to study the body eﬀect more into detail, both with a more precise simulation model and with on-body measurements.

Implementing both antennas with textile materials has represented an added dif- ficulty during the manufacturing process as the laboratory instruments and the connectors normally used are not easily adapted to textiles. These difficulties were encouraging to come up with more creative ideas and configurations, extending the range of learning during the project.

Regarding the materials used, its reliability has been checked proving a good stability and durability. However, the conductive Shieldex fabric has experienced some

65 4. Conclusion and Future Work fraying after being cut. Belt sewing done with a sewing machine has resulted more reliable, stronger and faster than hand-made sewing. The solution of using textile glue for smaller pieces turned into a good alternative as it remained stable and im- mobile.

The evaluation of the antenna’s functionality was only done with the RFxpert near ﬁeld scanner and the S parameters’ measurement with the Vector network analyser, which resulted to be satisfactory. However, additional measurements as using an anechoic chamber or on-body measurements would have improved the reliability of the results.

To sum up, the proposed antennas and their operation are promising according to the results achieved. Innovation topics as textile feeding circuits proved to be decisive to accomplish a reliable circular polarization and therefore, more research is needed. The combination of Cospas-Sarsat system and the growing market of wearable antennas, is a ﬁeld to be studied as it has a wide range of applications to improve our life.

4.2 Future work

In addition to the previously stated comments, some further development is needed in order to ensure a reliable performance of the antennas so they can be integrated into Cospas-Sarsat International Programme.

• Automatized manufacturing process to reduce human imperfections. • Additional tests as far field pattern in an anechoic chamber and direct test to the Cospas-Sarsat satellites. • Better body model for the simulations. • On-body measurements. • Integration of the GPS antenna to a piece of clothes and study the effect of its positioning. • Study of the consequences of the movement and bending of the antennas. • Design of the feeding battery and the chip to achieve different phase shifts for the patches.

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