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Designing Magnetic Components for Optimum Performance Article Power Supply Design Seminar Designing Magnetic Components for Optimum Performance in Low-Cost AC/DC Converter Applications Topic Category: Magnetic Component Design Reproduced from 2010 Texas Instruments Power Supply Design Seminar SEM1900, Topic 5 TI Literature Number: SLUP265 © 2010, 2011 Texas Instruments Incorporated Power Seminar topics and online power- training modules are available at: power.ti.com/seminars Designing Magnetic Components for Optimum Performance in Low-Cost AC/DC Converter Applications Seamus O’Driscoll, Peter Meaney, John Flannery, and George Young AbstrAct Assuming that the reader is familiar with basic magnetic design theory, this topic provides design guidance to achieve high efficiency, low electromagnetic interference (EMI), and manufacturing ease for the magnetic components in typical offline power converters. Magnetic-component designs for a 90-W notebook adapter and a 300-W ATX power supply are used as examples. Magnetic applications to be considered include the input EMI filter, power inductor design, high-voltage (HV) level-shifting gate drives, and single- and multiple-output forward-mode transformers in both wound and planar formats. The techniques are also applied to flyback “transformers” (coupled inductors) and will enable lower- profile designs with lower intrinsic common-mode noise generation. I. IntroductIon Gate Drive Note: The SI (extended MKS) units system AC Filter PFC Isolated Main Drive (for some Converter Output(s) is used throughout this topic. CM Inductor, topologies), Balanced Bridge, Differential This material is intended as a high- PFC Inductor, Multi-Output Filter level overview of the primary consider- Bias Flyback ations when designing magnetic components for high-volume and cost- Isolated Feedback optimized applications such as computer Fig. 1. Generic offline topology showing the magnetic notebook adapters, gaming, consumer, applications discussed. and general AC/DC “silver box” power supplies. Two specific high-volume designs are This topic is organized as follows: discussed that conform to the generic topology • In the AC filter section, a design overview for 5 Topic shown in Fig. 1—a 90-W, high-density, low-profile the common-mode choke and the differential- notebook adapter and a 305-W multiple-output mode filter are presented. ATX power supply. • In the PFC stage, many aspects of the PFC inductor For both the adapter and “Silver Box” power are discussed, including choice of material, the supply, the topology is comprised of a power effect of the gap fringing field on AC resistance, factor correction (PFC) front end, followed by a and winding arrangements for low EMI. regulation stage that controls the power through a • An insight into the design of high-voltage (HV) 50:50 isolation transformer stage. Each achieved level-shifting gate-drive transformers is “best-in-class efficiency” at very low intrinsic presented, particularly regarding the impact the costs and also achieved very good EMI conversion of common-mode currents to performance, with conducted emissions well below differential-mode noise has in the control product standard CISPR22/EN55022 Class B. circuitry. A comparison of their performance Apart from the efficiency, EMI, and cost targets, versus a high-voltage silicon gate-driver the 90-W adapter also had to be both low in profile alternative is shown. (≤ 16 mm) and achieve a high power density (1 MW/m3). 5-1 Texas Instruments 1 SLUP265 • The design of the power transformer for low df V= N × , (1) EMI and high efficiency is presented with a dt focus on balanced structure concepts. Flyback trans form ers and achieving fractional turns ratios where N is number of turns and f = magnetic flux. through a “major-minor” transformer approach Equation (2) relates this flux (f) to flux density are also addressed. (B) via a lumped parameter Ae for an effective magnetic cross-sectional area: The design techniques employed will have gen- f eral applicability to the design of high-performance B= (2) A magnetic components across a broad range of e topol ogies. Assembly drawings or photographs are presented for many of the more novel or lower- The design processes for both applications led profile magnetic components. to maintaining relatively low switching frequen- Therefore, this topic is intended to be a holistic cies: at 100 kHz for the PFC stage and 50 kHz and and qualitative treatment of magnetic com po nent 125 kHz for the bridge in the ATX supply and the design. For a very useful treatment of basic rele vant adapter, respectively. This was a result of topology magnetic principles, the “Magnetics Design Hand- choices to “buck” the voltage early and therefore book” from Texas Instruments [1] is recommended. operate downstream stages at lower voltage. Lowering the voltage counters the increase that II. desIgnIng for effIcIency would otherwise occur as a result of the lower switching frequency on the operating volt-second The overall optimization to meet the criteria of or flux-density levels. The maintenance of lower efficiency, cost, density, and manufacturability for operating flux-density levels through this general the topologies and magnetic components discussed strategy of lower voltages, combined with lower in this topic has tended to maintain relatively low frequencies, is important when one considers that switching frequencies, in the range of 50 kHz to core loss is a strong function of AC peak flux 300 kHz. Low-loss magnetic design techniques density. The mechanisms are through core eddy and topology choices have been exploited to and hysteresis losses (see section 2 of Reference simultaneously achieve surprisingly high overall [1]). density and cost performance. Core loss for a magnetic material is generally represented by an empirical power law, usually A. Fundamental Magnetic Design named after Steinmetz [2, 3, 4] and given here in Topic 5 Topic Considerations Equation (3): Magnetic design considerations are an integral β part of the overall topology and switching fre- Pv(t)= k × fα × B , (3) quency selection. Topology selection will depend on operating voltages and their ranges, and to a where Pv(t) is the time-average power loss per high degree on whether the topology is mostly unit volume, Bˆ is the peak flux density amplitude, soft-switched quasi-resonant, resonant, or hard f is the frequency of sinusoidal excitation, and k, switched. The soft-switched or resonant genres α, and β are constants found by curve fitting. For tend to allow higher switching frequencies and will the applications and frequency ranges discussed in place greater core- and conduction-loss challenges this topic, manganese zinc (MnZn) power ferrites on the design of the magnetic component. are used, with typical values at 100 kHz of α = 1.7 Higher switching frequencies will allow lower and β = 2.7 [5]. 3C96 ferrite, which was selected volt-seconds and operating flux-density levels per for optimal efficiency in these designs, has values Faraday’s Law (Equation [1]). This benefit may α = 1.9 and β = 2.9 (extracted from curves in be capitalized on as a magnetic-power processing- Reference [6]). density increase or as a magnetic-efficiency increase. 5-2 Texas Instruments 2 SLUP265 DC flux density and core shape will also combine with a transformer action (primary with modify these curves, and manufacturers’ data equal and opposite induced secondary) at each sheets should be consulted for this impact. layer-to-layer interface to cause a successive Lower operating flux-density levels lead to increase in proximity loss effect as one traverses a beneficial reduction in current ripple levels in from layer to layer. Alternatively, consider an power-path and filter components. This derives increasing winding space field due to the increasing from a lumped magnetic component form (Equation MMF as the layers are traversed. Here the overall [4]) of Ampere’s law giving the magnetomotive field for the winding is considered to be force (MMF) where H = magnetic field intensity, predominantly tangential to each winding layer. le denotes the effective magnetic path length, N = The loss equations for a multiple layer winding number of turns, and I = current: were originally presented in Reference [13]. For an inductor, the proximity loss will depend on the H× l = N × I (4) e main AC inductor current. A reduction in inductor Equation (5) gives the direct relationship current ripple will help reduce this loss. between flux density, B, and field intensity, H, For a gapped inductor or coupled inductor where µ signifies magnetic permeability: (flyback “transformer”), the gap field will produce a component of winding space field which will be B= µH (5) normal to the main proximity-effect field. This is Equation (6) gives the correspondence between referred to here as gap-fringe proximity. It can current ripple and operating flux-density level. often be the most significant copper loss -7 µ0 = 4π × 10 H/m (permeability of free space) mechanism, particularly for foil-type windings and µr = relative permeability: where the copper paths for substantial eddy-current µ× µ × NI × flows will be present. Gap fringe proximity also B = 0 r AC (6) generates localized hot-spots. AC l e For a transformer scenario, the primary power- current MMF will be substan tially cancelled by B. Eddy Current Loss in Windings the transformed equal and opposite secondary “Skin effect” describes the tendency for AC power current in each half of the winding space. current in a conductor to concentrate towards the The winding-space MMF will therefore be outside of the conductor and is caused by its own comprised of a magnetizing field and a leakage magnetic field. Induced eddy currents cause the field, and hence the proximity effect in a Topic 5 Topic penetrating field and consequent net current transformer winding space is often associated with density to decay to 1/e times the surface current the leakage inductance. The proximity effect will density at the skin depth.
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