Development of 50-Kv 100-Kw Three-Phase Resonant Converter for 95-Ghz Gyrotron Sung-Roc Jang, Jung-Ho Seo, and Hong-Je Ryoo

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6674 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 11, NOVEMBER 2016 Development of 50-kV 100-kW Three-Phase Resonant Converter for 95-GHz Gyrotron Sung-Roc Jang, Jung-Ho Seo, and Hong-Je Ryoo

Abstract—This paper describes the development of a and high power density, the operation of high-power vacuum 50-kV 100-kW cathode power supply (CPS) for the operation devices requires low output voltage ripple with low arc energy. of a 30-kW 95-GHz gyrotron. For stable operation of the gy- This is because the output voltage ripple and the arc energy rotron, the requirements of CPS include low output voltage ripple and low arc energy less than 1% and 10 J, respec- are closely related to the stability of the output power of the tively. Depending on required specifications, a three-phase electron beam and the safety of the device, respectively. It is series-parallel resonant converter (SPRC) is proposed for clear that a higher value of the output filter capacitor allows designing CPS. In addition to high-efficiency performance a lower value of the output voltage ripple. On the other hand, of SPRC, three-phase operation provides the reduction of the energy stored in the power supply output, which may in- the output voltage ripple through a minimized output filter component that is closely related to the arc energy. For al- stantaneously be discharged to vacuum devices because of the lowing symmetrical resonant current from three-phase res- arc, is proportional to the value of the output filter capacitor. In onant inverter, the high-voltage transformers are configured order to achieve low output voltage ripple with low arc energy, as star connection with floated neutral node. This facilitates a pulse step modulator (PSM) has been proposed to operate a balanced voltage on each secondary winding. In addition, gyrotron [1], [2]. Compared with other proposed designs, based distinctive design of the high-voltage rectifier is introduced, taking into consideration the effective series stacking of largely on alternating current (ac) voltage regulation or a star- diodes by means of the parallel resonant capacitor. In partic- point controller, the PSM-based high-voltage power supply is ular, the implementation of the high-voltage part including expected to be highly reliable owing to low stored energy. Other transformer and rectifier is presented in detail. For provid- approaches that use additional solid-state switches to protect the ing high power density and high reliability, effective meth- load against the arc have been proposed [7]. A crowbar circuit ods for winding the high-voltage transformer and stacking rectifier diodes are discussed. Finally, the developed CPS connected in parallel with the load helps us to limit the energy achieves 95.5% of maximum efficiency, 0.92 of maximum from the power supply to the load, and a crowbar switch using power factor, 500 W/liter of power density, 0.6% of output a solid-state device exhibits short response time such that the voltage ripple, with 8.3-J arc energy. load can be effectively protected. Another method used to limit Index Terms—DC–DC power converters, gyrotrons, the arc energy by means of a solid-state switch is to insert a fast pulsed power supplies. opening switch between the power supply output and the load. Further research on the reduction of the ripple as well as the arc I. INTRODUCTION energy has been presented for electrostatic precipitator appli- ESEARCH on high-power vacuum devices, such as mag- cation [21]. Presented inductive adder topology which consists R netron, klystron, and gyrotron, has grown in recent times of high-voltage converter modules in series, and uses the phase owing to the development of various industrial applications in- shifting control technique between each module shows many cluding medical, military, environmental, and aerospace. The advantages including decrease of the output ripple as well as the recent proliferation of applications requiring high voltage and stored energy. power has led to a greater focus on the development of the high- Based on the basic concept of the inductive adder [21] and voltage power supplies [1]–[22]. In addition to the general re- the operating principle of the series-parallel resonant converter quirements of high-voltage power supplies, efficient operation, (SPRC) [22], the design and the implementation of a 50-kV 100-kW CPS for a 30-kW 95-GHz gyrotron are described in this Manuscript received January 12, 2016; revised March 14, 2016; ac- paper. A three-phase resonant converter based on half-bridge cepted May 23, 2016. Date of publication June 29, 2016; date of SPRC module is proposed for achieving desired specifications. current version October 7, 2016. This work was supported by the Korea Electrotechnology Research Institute Primary Research Program For balanced three-phase operation, the star configuration with of MSIP/NST (16-12-N0101-49). floated neutral node is suggested for three high-voltage trans- S. R. Jang is with the Electric Propulsion Research Center, Korea formers that are connected to each of three half-bridge resonant Electrotechnology Research Institute, Changwon 641-120, South Korea, and also with the University of Science and Technology, Daejeon 13557, inverters. In addition, a distinctive design of a high-voltage South Korea (e-mail: [email protected]). rectifier is introduced for minimizing component count. J.-H. Seo is with the Department of Energy and Power Conversion Generally, the parallel resonant capacitor is connected in Engineering, University of Science and Technology, Daejeon 13557, South Korea (e-mail: [email protected]). parallel with the transformer primary or secondary winding H.-J. Ryoo is with the School of Energy Systems Engineering, Chung- [22]. On the other hand, the proposed rectifier circuit for Ang University, Seoul 06974, South Korea (e-mail: [email protected]). CPS uses the capacitors which are connected in parallel with Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. rectifier diodes for balancing voltage of each diode as well Digital Object Identifier 10.1109/TIE.2016.2586021 as for implementing the parallel resonant capacitor. With

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TABLE I SPECIFICATIONS OF CPS FOR A 30-KW95-GHZ GYROTRON

Input Voltage 380 Vac ± 10%

Maximum output voltage, Vo,max −50 kV Maximum output current, Io,max −2A Maximum output power, Po,max 100 kW Maximum pulse width, PWmax 3s Maximum duty cycle, D max 50% Output voltage ripple at rated voltage 0.6% Arc energy, E Arc 8.3 J Maximum Efﬁciency, ηmax 95.5% Maximum Power Factor, PFmax 0.92 Protections Arc (overcurrent) Overtemperature Overvoltage Fig. 1. Proposed circuit of SPRC module for CPS.

range so that CPS can control the required output voltage from considerations such as high-voltage insulation and cooling −5to−50 kV. of components, the detailed implementation of the proposed The operating principle of SPRC have been already intro- circuit is discussed to improve power density with reliable duced and well known [13], [15], [19], [20], [22]. Therefore, operation. Especially for high-voltage part which is immersed this paper intends to omit the description about the operating in insulation oil, compact design and arrangement are presented principle of SPRC and deal with a detailed design of CPS. including the high-voltage transformer and rectifier. Section II presents the design of CPS based on three-phase SPRC with respect to the required specifications summarized A. Design of SPRC Module for CPS in Table I. Feasibility and performance of the proposed The proposed circuit of SPRC module is shown in Fig. 1. circuit is verified by PSpice simulation. In Section III, the Compared with the conventional circuit of SPRC, it is worth detailed implementation including a special winding method noting that the parallel resonant capacitor (Cp , dashed line) is ac- for high-voltage transformer and a compact design of rectifier tually not connected in parallel with the transformer primary or circuit is described. Finally, the performance of the developed secondary winding. For implementing the parallel resonant ca- CPS is experimentally proven from the point of view of its pacitor, the proposed circuit uses the capacitors (CD1 1 −CD1 N ) efficiency, power factor, output voltage ripple, and arc energy. that are originally installed for balancing the voltage between series stacked diodes. Thus, the capacitors which are connected in parallel with the diodes play the role of the parallel reso- II. DETAILED CIRCUIT DESIGN OF CPS nant capacitor. Because it should be discharged and charged SPRC which operates in continuous-conduction mode and for forward and reverse biasing of diode, respectively. From has above-resonance switching frequency range has the advan- the transformer secondary side, equivalent parallel capacitor tages of zero voltage (ZV) turn-on and relatively low conduction can be regarded as the parallel of two capacitors (CD1, CD2) loss compared with other resonant converter mode of opera- where CD1 is equivalent capacitor for series connection of tions. Moreover, the lossless snubber capacitor which can be CD1 1 −CD1 N (CD1 1 /N). Accordingly, the value of Cp is cal- 2 connected in parallel with the semiconductor switches reduces culated as n ×(CD1 +CD2) where n is the transformer turns turn off switching loss by decreasing the slope of the voltage ratio (n2/n1). By choosing suitable value of CD1 and CD2,ef- rise across the switches. However, it should be noted that there is fective balancing of voltage between the series stacked diodes additional consideration in choosing the value of snubber capac- as well as inserting the parallel resonant capacitor can be simul- itance for achieving ZVS. This is because the snubber capacitor taneously achieved with minimized component count. Another should be discharged before turning on the switch by means of point worth mentioning is that the proposed circuit does not only stored energy in resonant inductor. control the switching frequency, but also adjusts the dead time For designing high-voltage and high-power converters, the which represents the time duration between the turning off of one SPRC has many advantages compared to the other resonant con- MOSFET and the turning ON of another MOSFET. According to verter topologies that use different resonant tank structure such general control characteristic of SPRC, it is clear that the output as series resonant converter (SRC), parallel resonant converter voltage can be controlled by switching frequency modulation. (PRC), and LLC resonant converter. First of all, the SPRC has Besides controlling the switching frequency, the proposed cir- the current source characteristic of SRC and the intrinsic voltage cuit changes the dead time for allowing ZV turn on of MOSFET boost-up characteristic of the PRC. Also, the parasitic capaci- independent of the load condition [13]–[15], [19], [20]. tance of the high-voltage transformer can play a role of a parallel As shown in Fig. 1, the gate–source signal of MOSFET (VGS) resonant capacitor instead of causing adverse effect. In addition, has less pulse width compared with the frequency modulated the SPRC provides relatively wide controllable output voltage signal (Vsw) from the controller. 6676 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 11, NOVEMBER 2016

Fig. 2. Designed circuit of CPS based on the three-phase SPRC.

Fig. 3. Analysis of the proposed three-phase SPRC for CPS. (a) Operating waveforms depending on operational mode. (b) Operational mode analysis.

Thus, the gate drive (GD) circuit is designed for detecting B. Analysis and Design of CPS Based on Three-Phase ZV condition of MOSFET and providing the switching signal SPRC so that ZV turn ON of MOSFET can be always achieved Based on the proposed SPRC module, CPS is designed irrespective of the load condition. When SPRC use rela- as shown in Fig. 2. In order to minimize both arc energy tively high value of lossless snubber capacitors (C , C ) sn1 sn2 and output voltage ripple, CPS is designed based on a three- for reducing turn off loss, this additional function of the phase resonant inverter. Three switching legs consisting of proposed GD circuit provides high efficiency and reliable MOSFET (S1/S1 − S3/S3 ) operate with a 120° phase delay operation. as shown in Fig. 3(a). High-frequency ac power is transferred JANG et al.: DEVELOPMENT OF 50-KV 100-KW THREE-PHASE RESONANT CONVERTER FOR 95-GHZ GYROTRON 6677 from each switching leg to the high-voltage transformer (TR1– 2) Dashed line at inverter side: freewheeling current flowing TR3) through a series resonant inductor (Ls1−Ls3) and ca- path for allowing ZVS during switching transition. pacitor (Cs1−Cs3). The voltage on the transformer’s sec- 3) Solid line at rectifier side: current flowing path for charg- ondary winding is rectified by a voltage-doubled rectifier ing the output filter capacitor and supplying the load consisting of the diodes (Dr1 − Dr6) and the filter capaci- current. tors (Cf1-Cf6). The three rectifiers are connected in series 4) Dashed line at rectifier side: current flowing path for to generate −50 kV of output voltage, which is six times charging parallel resonant capacitor. the value of the voltage on secondary winding. The recti- When S1 is turned OFF, Mode1 starts with conducting an- fying diodes (Dr1 − Dr6) are practically implemented by a tiparallel diode (D1) and turning ON of S1 with ZV. Actually, series stacking of high-frequency diodes, and the capacitors there is an elliptical mode that charge and discharge the snubber (Cp1−Cp6) represent the equivalent capacitance of all capaci- capacitors (C1, C1) between the turning off of S1 and the turn- tors connected in parallel with each high-frequency diode [19]. ing ON of S1. During Mode1 operation, freewheeling current of In addition, Cp1−Cp6 act as parallel resonant capacitors con- switching leg1 flows through S3, TR1, and TR3 as shown in nected in parallel with the transformer’s secondary winding. Fig. 3(b) (dashed line at Mode1). Thus, the proposed converter can take advantage of an intrin- Owing to continuous current flowing through TR1, the recti- sic voltage boost-up function owing to the parallel resonant fier diode Dr1 maintains forward biased and charges the output capacitor. filter (Cf1). The solid line in Fig. 3(b) represents the current As depicted in Fig. 2, the star configuration of three high- flowing path for powering that apply input voltage (Vd )tose- voltage transformers provides neutral node (TR_N). The neutral ries connected two resonant tanks and two transformers. These node of transformer (TR_N) is typically connected to the neutral operations imply that the three-phase resonant inverter operates node of dc-link capacitor (DC_N) in order to operate three-phase not independently as half bridge. The operation is similar to legs as individual three half-bridge inverter. In other words, in- full-bridge circuit that supplies full input voltage to two reso- dependent operation of three-phase half-bridge SPRC can be nant tanks and transformers simultaneously. In other words, the achieved when two neutral nodes (TR_N, DC_N) are connected resonant current always flows through the combined resonant together. Although the aforementioned configuration is a general tank. approach for three-phase system [20], the proposed converter The resonant current, ILs3, is rectified, and it charges the differentiates between the two neutral nodes in order to over- filter capacitor (Cf6). On the other hand, additional operating come the disadvantages of the star with neutral configuration. mode of rectifier is observed for the secondary side of TR2 One disadvantage of star configuration with two neutrals con- during Mode1 operation. When the resonant current changes nected together is that the rms value of high-frequency current its polarity, it is also necessary to change the conducting diodes flowing between these two neutral nodes is relatively high. This depending on the operating principle of the voltage doubled rec- is due to the fact that SPRC does not generate pure sinusoidal tifier. As shown in Fig. 3(a), the resonant current, ILs2, changes waveform of resonant current, but trapezoidal waveform [15], its polarity from positive to negative at the starting of Mode1. [19], [20]. Therefore, the sum of three-phase resonant current is Thus, the secondary-side current flows to discharge Cp3 and not zero. This neutral current poses a challenge in choosing suit- charge Cp4, respectively. During this time duration, the res- able dc-link capacitor (Cdc1, Cdc2) due to high-frequency rms onant current has different frequency because two capacitors current. Another demerit is the unbalanced operation between (Cp3, Cp4) play the role of the parallel resonant capacitor. each phase. It was experimentally found that the output voltage Mode2 starts with S1 conducting, and the subsequent flow of each converter module could be different due to the small of positive current from the dc input to TR1. This transition in difference in resonant tank parameters. In addition to the differ- the current polarity also requires the charging and discharging ences in the three rectified voltages that are connected in series, of Cp1 and Cp2, respectively. During Mode2 operation, S3 is the unbalanced shape of three resonant currents causes signifi- turned off and the freewheeling current flows through the an- cant adverse effect on the control as well as the losses. In order tiparallel diode D3. At Mode3, the power from the switching to solve these problems, the star configuration of transformer leg1 is finally transferred to the load by means of diode Dr2. with floating neutral node is suggested. When two neutral nodes Although the explanation of the operational mode is made fo- are floated, it is not necessary to consider high-frequency neu- cusing on the switching leg1, all the operations of the proposed tral current. Moreover, the proposed three-phase SPRC is no circuit can be analyzed based on the conducting devices and longer the individual three half-bridge converters because full waveforms depicted in Fig. 3. dc-link voltage (Vd ) is applied to two resonant tanks and two In addition to the analysis and the design of SPRC, the de- transformers. The detailed operating principle of the proposed tailed design procedure and equations of three-phase SPRC have circuit can be analyzed from Fig. 3. It includes the operating been presented in [23]–[27]. Depending on the operation of waveforms of each operational mode depending on the current the proposed three-phase SPRC, the brief design guideline and flowing. consideration for CPS are as follows. The meaning of solid and dashed lines in Fig. 3(b) that rep- For designing the resonant tank parameters of CPS, the min- resent current flowing path are defined as follows: imum switching frequency should be determined depending 1) Solid line at inverter side: current flowing path from dc on the requirements of the output voltage ripple and arc en- input (Vd ) to the transformer. ergy. From the determined minimum switching frequency, the 6678 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 11, NOVEMBER 2016

TABLE II current. After charging the parallel resonant capacitor, the reso- SUMMARY OF DESIGN PARAMETERS FOR CPS nant current depends on the series resonant tank that can provide relatively low frequency compared to the resonant frequency by Resonant inductor (Ls1−Ls3)5μH parallel resonant capacitor. The value of the parallel resonant Series resonant capacitor (Cs1−Cs3)30μF Parallel resonant capacitor (Cp1−Cp6)0.38nF capacitor is closely related to not only the voltage gain, but also Snubber capacitor (C 1−C3)2nF the controllable load range. For choosing the value of the snub- Transformer turns ratio (N1:N2) 12 : 400 ber capacitor, the minimum stored energy in resonant inductor Output filter capacitor (Cf1−Cf6)40nF should be considered. When the resonant tank parameters are determined, the required transformer ratio can be calculated depending on the maximum output voltage. For designing high-voltage transformer, relatively high value of leakage inductance due to the insulation distance between primary and secondary winding is inevitable. Thus, it is preferable to implement the resonant inductor by means of transformer leakage inductor. Especially for high-power resonant converter, loss and size of resonant inductor cannot not be ignored due to the relatively high value of ac resonant current. Therefore, high power density and high efficiency can be achieved by using leakage inductance instead of additional inductor. For designing transformer, the value of desired leakage inductance can be considered [17]. Based on aforementioned design guideline and consideration of the proposed circuit, all the designed parameters for a 50-kV 100-kW CPS are summarized in Table II. The value of parallel resonant capacitor (Cp) is calculated from Cp1 which is actually implemented using 18 pieces of 6.8-nF capacitor in series. To control the output voltage and the current with desired load range, the operating switching frequency is determined from a Fig. 4. Simulation results of designed CPS at rated operation. range of 45 to 200 kHz. The value of the equivalent output filter capacitance is 6.67 nF, and it provides 8.3 J of maximum arc energy at the rated output voltage.

C. PSpice Simulation of Designed CPS The detailed design of CPS for the 30-kW 95-GHz gyrotron described in the foregone sections was verified using PSpice simulation. Based on the designed circuit and parameters in Fig. 2 and Table II, the CPS was simulated with a resistor load and the results are shown in Fig. 4. With 25-kΩ load condition, simulated waveforms including the resonant current, the MOS- FET switching voltage, the output voltage, and power confirm the fact that the rated operation of designed CPS is achieved with 500 V of input voltage, 45 kHz of switching frequency, and 300 A of resonant current peak value. From the switching voltage (VDS) and the current through the anti-parallel diode, ZV turning on of S1 is confirmed because the gate–source voltage increases after conducting antiparallel Fig. 5. Comparison of operating waveforms between star configura- diode. In addition, the output voltage ripple measurement wave- tions with floated neutral and connected neutral. form shown in Fig. 4 shows a 0.3% ripple rate with designed filter. It is clear that the frequency of output voltage ripple is required characteristic impedance as well as resonant frequency six times that of the resonant current due to the three-phase can be calculated for achieving maximum output power. It high-frequency operation. should be noted that the proposed SPRC mainly uses the res- For comparing the operational waveforms between star con- onance between the series resonant inductor and the parallel figurations with floated neutral and connected neutral, PSpice resonant capacitor. Accordingly, the characteristic impedance simulation is performed with the same operating parameters in- that is calculated from the series resonant inductor and the par- cluding the resonant tank, the switching frequency, and the load allel resonant capacitor determines the peak value of resonant condition. As depicted in Fig. 5, the proposed circuit generates JANG et al.: DEVELOPMENT OF 50-KV 100-KW THREE-PHASE RESONANT CONVERTER FOR 95-GHZ GYROTRON 6679 three-phase sinusoidal waveform which provides three-phase balanced resonant current. On the other hand, star configuration with neutral connection exhibits relatively high value of neutral current because the three-phase resonant current has trapezoidal waveform. Compared with the waveform of the proposed circuit, trapezoidal waveform of the resonant current provides an advantage related to the conduction loss because rated output power can be achieved with less rms value of resonant current. However, high-frequency neutral current causes an additional problem for designing input filter. As shown in Fig. 5, 100 A of rms current should be considered when choosing high-frequency filter capacitor (Cdc1, Cdc2).

III. IMPLEMENTATION OF CPS From the designed parameters summarized in Table II, the detailed implementation of high-voltage components is described for achieving high power density as well as high reliability.

A. Implementation of High-Voltage Transformer Depending on the design performed in Section II, the transformers with 12:400 of turns ratio are implemented by means of UU shape ferrite core (cross-sectional area: 12 cm2 ) for preventing saturation and generating 50 kV of maximum output voltage. To implement the high-voltage transformer, provid- Fig. 6. Comparison of winding method between conventional and pro- ing insulation distance between the primary and the secondary posed transformer. (a) Conventional method for secondary winding. (b) Proposed method for secondary winding. winding is essential. In addition, securing sufficient insulation between each wire of secondary winding is also important when a number of winding layers are necessary for satisfying designed turns ratio. Generally, the secondary winding of the high-voltage transformer uses multiple layer structure to wind a number of wires within restricted winding area. Fig. 6(a) shows an example of a conventional winding method for high-voltage transformer secondary winding. In order to wind 60 turns of secondary wire within restricted length of bobbin (Dbobbin), five layers are used for providing insulation distance between layers. By assuming 250 V per each turn, the maximum voltage difference between layer 1 and layer 2 is 9 kV, as shown in Fig. 6(a). Therefore, it is necessary to provide insulation distance with respect to 9 kV, and insulation material may Fig. 7. Structure of designed bobbin for a CPS transformer. need to be inserted between layers. Accordingly, it is inevitable that the height of the secondary winding will increase due to the reduction of effective winding area. On the other hand, the conventional winding method without necessarily considering proposed winding method shown in Fig. 6(b) provides more ef- the exact insulation distance with respect to the voltage. fective winding area by minimizing voltage difference between A detailed design structure of bobbin for CPS transformer adjacent wires. Compared with the conventional method, the shown in Fig. 7 includes seven partitioning flanges for pro- proposed winding method limits the maximum voltage differ- viding eight winding area on which each secondary winding ence within 2 kV. Therefore, it is not necessary to provide the is wound. The primary winding is wound on the core and the insulation distance between adjacent layers and to insert insu- bobbin envelops the primary winding in order to provide insulation material when the wire has 1 kV of intrinsic insulation lation as well as to minimize leakage inductance. Two small strength. Also, the insulation between each winding section is sections on the top and bottom of bobbin are specially designed guaranteed by partitioning flanges which is made by insulation for providing insulation distance against surface discharging be- material such as Teflon. As compared in Fig. 6, the proposed tween the secondary winding and the primary winding. Based winding method facilitates 80 turns of secondary wire within the on the aforementioned winding method, a transformer for CPS same length of bobbin (Dbobbin). It should be noted that the sim- is implemented as shown in Fig. 8 and measured magnetizing ple drawing depicted in Fig. 6 is just to explain the advantages of and leakage inductances are 1.2 mH and 5 μH, respectively. the proposed winding method by roughly comparing it with the By creating space between primary and secondary winding, the 6680 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 11, NOVEMBER 2016

Fig. 10. Picture of implemented high-voltage rectiﬁer for CPS.

Fig. 8. Picture of implemented transformer for CPS.

Fig. 9. Circuit of high-voltage rectiﬁer PCB module.

desired value of leakage inductance that is exactly matched with Fig. 11. Picture of implemented CPS. the designed value of resonant inductor is implemented. Another point worth mentioning is that we use only one side of resistor and capacitor divider is also designed in the rectifier of core for the primary and secondary winding. That is es- PCB as depicted in Fig. 9 (R1– R2, C39–C58). pecially for effective arrangement of three transformers while As shown in Fig. 2, one node (P5) of a high-voltage sensing providing sufficient insulation between them. Consequently, the circuit is connected to the high-voltage output which has same proposed winding method allows compact design as well as high potential with connector P1 on Fig. 9. That is, there are only one reliability of high-voltage transformer. connection between a high-voltage sensing circuit and a rectifier among three PCBs. Although, the other node of a high-voltage B. Implementation of High-Voltage Rectifier sensing circuit is not connected with a rectifier circuit, a sensing circuit can be implemented very closely with a rectifier circuit by A high-voltage rectifier is basically designed based on the minimizing the voltage difference between neighboring nodes voltage doubled circuit as shown in Fig. 9. From the secondary as depicted in Fig. 10. In addition, there are slits between the winding which is connected to P3 and P4, the filter capacitors series stacked diode groups for preventing surface discharging. (C37, C38) are charged through series stacked diodes (D1–D18, These implementation methods for a rectifier PCB allows com- D19–D36). For rectifying one third of the maximum output pact design and provides high reliability against high-voltage voltage, 18 pieces of diodes are stacked in series. The capaci- breakdown. tors (C1–C36) which are connected in parallel with each diode provide voltage balancing between series stacked diodes. The C. Compact Assemble of CPS other role of these capacitors is to serve as the parallel resonant capacitor that is connected in parallel with load [19]. By In order to achieve high power density and high reliability designing suitable value of these capacitors, it is possible to of CPS, the arrangement and the structural design are shown implement series-parallel resonant tank structure without using in Fig. 11. They include the resonant capacitor (Cr1−Cr3), additional capacitor on the transformer primary or secondary three-phase inverter with MOSFETs, the GD circuit, oil im- side. Therefore, the balance of voltage as well as the implemen- mersed transformers, and rectifiers. Three MOSFET modules are tation of parallel resonant capacitor can be effectively achieved attached to the heat sink located at the back side of CPS. The with reduced component count. The sensing circuit by means GD circuit which drives six MOSFETs is located on top of the JANG et al.: DEVELOPMENT OF 50-KV 100-KW THREE-PHASE RESONANT CONVERTER FOR 95-GHZ GYROTRON 6681

Fig. 12. Experimental waveforms of the unbalanced operation when two neutral nodes (TR_N and DC_N) are connected (switching voltage of S1: 50 V/div., resonant current (ILs1, ILs2, ILs3): 20 A/div., 5 μs/div.). Fig. 13. Experimental waveforms of measured voltage balancing between series stacked diodes (1kV/div., 1 μs/div., inverted polarity). MOSFETs. Three-phase resonant capacitors are arranged as “U” shape for reducing the space and allowing effective cooling. There is no additional resonant inductor because it is implemented by means of leakage inductance of transformer. Two FANs at the rear of CPS generate forced air which flows to the top front side for the cooling all components except the high- voltage part. Three high-voltage transformers and rectifiers are immersed in insulation oil. By using only one side of the core for the primary and secondary winding, three transformers can be arranged effectively while considering insulation distance between each secondary winding. In addition, the insulation distance from the secondary winding to the grounded core is also guaranteed with reduced volume. The secondary winding of TR1 is located near a Rec1 (in Figs. 2 and 11) for providing similar potential between neighboring components. Likewise, the components including Rec3 and TR3 which Fig. 14. Experimental waveforms at CPS-rated operation (output volt- have less voltage potential are situated near the grounded oil age: 10 kV/div., resonant current: 200 A/div., switching voltage (VDS): 500 V/div., 10 s/div.). tank case. For the cooling of high-voltage components, the oil μ filled tank made of duralumin material functions as a heat sink. For more effective cooling, additional heat sink is attached to the diodes, the diodes reverse biased voltage for low-voltage-side top cover of oil tank with FAN. From these compact design and rectifier (rectifier connected with TR3 in Fig. 2, Rec3 in effective cooling structure of CPS, a 50-kV 100-kW power sup- Fig. 11) was measured during −15-kV operation. As shown ply is developed within 60 L (width: 440 mm, depth: 490 mm, in Fig. 13, three voltage waveforms that has the value of 1.73, height: 270 mm). Finally, 500 W/L of power density is achieved 3.39, 5.09 kV are measured from ground to anode of diode D31, with an additional input filter for reducing low-frequency ripple. D25, D19 (see Fig. 9), respectively. This experimental waveform verifies that voltage balancing between series stacked diodes is achieved without equalizing resistors. The value of each capac- IV. EXPERIMENTAL RESULTS OF DEVELOPED CPS itor (C1–C36 in Fig. 9.) is 6800 pF with ±10% tolerance. The A prototype of CPS for a 30-kW 95-GHz gyrotron was tested rising and falling of the voltage waveform represent charging with a dummy resistor load, and the feasibility and performance of capacitors for reverse biasing and discharging of capacitors of design was verified. Fig. 12 shows the experimental wave- for forward biasing, respectively. Therefore, C1–C36 can play forms of the unbalanced operation when two neutral nodes a role of a parallel resonant capacitor. (TR_N and DC_N) are connected. Three different resonant Rated operation of CPS is shown in Fig. 14 which was tested current waveforms with same operating switching frequency with 25 kΩ of resistor load. The experimental waveforms that are represent the difference in voltage between three rectifies. Fur- exactly matched with the simulation depicted in Fig. 4 validate thermore, the triangular shape of the resonant current (ILs1) the design described in this paper. The result of the measured signifies that the rectified voltage is almost zero. On the other output voltage ripple shown in Fig. 15 verifies the superiority of hand, the proposed configurations that share the resonant tank the proposed three-phase operation. between each phase solve the problem depicted in Fig. 12.To The efficiency and the power factor were measured by power verify the effective voltage balancing between the series stacked analyzer and Fig. 16 shows 95.5% of maximum efficiency and 6682 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 63, NO. 11, NOVEMBER 2016

veriﬁed with a resistor load. Subsequently, the developed CPS will need to be tested with the gyrotron to further improve the prototype.

V. C ONCLUSION In this paper, the design and implementation of CPS to operate a 30-kW 95-GHz gyrotron were described. Depend- ing on given design considerations and speciﬁcations, a three- phase SPRC was proposed, and the detailed design and implementation were presented to lower output voltage ripple with low arc energy. Finally, the developed CPS was tested Fig. 15. Experimental waveforms of output voltage ripple at CPS-rated with a resistor load, and the proof of its superiority included operation (output voltage ripple: 200 V/div., resonant current: 200 A/div., 95.5% of maximum efﬁciency, 92% of maximum power factor, switching voltage (VDS): 500 V/div., 10 μs/div.). 500 W/L of power density, and 0.6% of output voltage ripple with 8.3 J of arc energy. For future research, we intend to improve our prototype by testing it with an actual 30-kW 95-GHz gyrotron.

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