A Photonically-Excited Leaky-Wave Antenna Array at E-Band for 1-D Beam Steering

applied sciences

Article A Photonically-Excited Leaky-Wave Antenna Array at E-Band for 1-D Beam Steering

Álvaro J. Pascual-Gracia 1, Muhsin Ali 2 , Guillermo Carpintero Del Barrio 2 , Fabien Ferrero 3, Laurent Brochier 3, Ronan Sauleau 1, Luis Enrique García-Muñoz 2,* and David González-Ovejero 1,*

1 Univ Rennes, CNRS, IETR (Institut d’Electronique et de Télécommunications de Rennes)–UMR 6164, F-35000 Rennes, France; [email protected] (Á.J.P.-G.); [email protected] (R.S.) 2 Universidad Carlos III de Madrid, 28911 Leganés, Spain; [email protected] (M.A.); [email protected] (G.C.D.B.) 3 Université Côte d’Azur, CNRS, LEAT (Laboratory of Electronics Antennas and Telecommunications)–UMR 7248, 06903 Sophia Antipolis, France; [email protected] (F.F.); [email protected] (L.B.) * Correspondence: [email protected] (L.E.G.-M.); [email protected] (D.G.-O.)

Received: 30 March 2020; Accepted: 14 May 2020; Published: 18 May 2020

Abstract: This manuscript reports the ﬁrst leaky-wave antenna (LWA) array excited by a photomixer as well as its potential application for alignment in wireless links. The designed array is manufactured in printed circuit board (PCB) technology, works at the E-band (from 75 to 85 GHz), and provides a directive beam of about 18 dBi with a frequency scanning span of 22◦. The antenna element consists of a microstrip line periodically loaded with stubs, and it has been designed employing a hybrid approach combining full-wave simulations and transmission line theory. This approach enables the optimization of the periods when the open-stopband of the LWA is mitigated or removed at the frequency of broadside emission. The proposed antenna was ﬁrst tested using a ground signal ground (GSG) probe; the measured return loss and radiation patterns of the fabricated prototype were in good agreement with full-wave simulations. Then, the LWA array was integrated with the photomixer chip using conductive epoxy threads. Measurements of the radiated power yielded a maximum of 120 µW at 80.5 GHz for a 9.8 mA photocurrent. Finally, the antenna was used in a 25 cm wireless link, obtaining a 2.15 Gbps error-free data rate.

Keywords: photomixer; photonic antenna; PCB antenna; E-band; beam scanning; beam steering; probe de-embedding; leaky-wave antenna; epoxy bonding; RF choke

1. Introduction Photonic generation of millimeter (mm) and submillimeter (sub-mm) continuous-wave radiation by photomixing is an extremely promising route to explore. In this approach, two optical wavelengths are mixed on a fast photodetector, such as a uni-traveling carrier photodiode (UTC-PD). The electrical beat note at the output of the photodetector is equal to the frequency difference of the two optical tones. Photomixing is advantageous to electronic generation in terms of coherence length and bandwidth of the generated carriers [1]. Moreover, one can use reliable and cost-effective optical components such as Erbium-doped fiber amplifiers (EDFAs), semiconductor optical amplifiers (SOAs), switches or time delay lines. Besides, due to the seamless integration with fiber optics, photonically-excited antennas are commonly used in wireless links at terahertz (THz) and sub-THz frequencies with multi-Gbps data throughput [2–5]. The main downside of photonic generation is the relatively low output power. Photonic transmitters typically use substrate lens antennas [6], horns [7], horns as feeds of quasi-optical

Appl. Sci. 2020, 10, 3474; doi:10.3390/app10103474 www.mdpi.com/journal/applsci Appl. Sci. 2020, 10, 3474 2 of 16

(QO) systems [8,9], and, less commonly, alternative antenna topologies such as tapered slot antennas [3] or monolithically integrated transverse electromagnetic horns [10]. In general, these antennas present a challenging escalation [11,12]. The difficulty of integrating several photomixer-excited antennas has obstructed the development of arrays with beam steering capabilities, and just a few demonstrations can be found in the open literature [13,14]. Beam steering would extend the range of applications of this kind of antennas. Among possible use cases, to name a few, one can find imaging systems, fine beam alignment to compensate mast misalignment or twist and sways from mounting structures [15], or reconfigurable wireless links in data centers [16]. The development of photomixer-excited antennas with beam steering could unlock the use of the E-band (71–76 GHz, 81–86 GHz), which presents several advantages. It offers a 2 5 GHz × bandwidth, adequate for tens of Gbps of data transmission; it does not present the absorption peak of oxygen compared to the 60–GHz band, and the free-space loss is lower than at higher sub-mm wave frequencies. The availability of low-loss, thin, inexpensive substrates well adapted to this frequency range makes printed circuit board (PCB) technology a promising approach for the realization of photonic transmitters/receivers (Tx/Rx). Active components with complex polarization networks can be integrated with passive components in a single multilayered board, as in [17]. Furthermore, in [3,18] we demonstrated the possibility of developing a photonic Tx with this technology. This work presents, to the best of our knowledge, the first realization of a leaky-wave antenna (LWA) excited by a photomixer, as a simple means to steer the beam with frequency in 1-D. The LWA is printed using PCB technology and works at the upper window of the E-band. It would find applications in some of the cases cited above. In particular, the case where the proposed antenna is used for automatic alignment in line-of-sight (LoS) mm-wave links will be discussed in more detail here. The realized design features a fairly broad relative bandwidth (>10%), and allows a scanning range of 22◦, including broadside. Moreover, the scalability of the presented layout is beneficial with respect to classical mm-wave antennas to combine several emitters and increase emitted power and or directivity. Section2 describes the design of the antenna. The integration of the photodiode chip with the PCB is discussed in Section3. Sections4 and5 present measurement results and a potential application, respectively. Finally, conclusions are presented in Section6.

2. Leaky-Wave Antenna This design is motivated by the necessity of increasing the emitted power by a scalable layout, with beam steering capabilities, and a 10% relative bandwidth around the center frequency (81 GHz). One of the simplest beam steering methods is frequency steering with a 1-D periodic LWA. In this configuration, a traveling wave encounters periodic perturbations in one direction, so that it leaks progressively. If the electrical path between successive perturbations is such that the traveling wave encounters them in phase, the antenna will radiate a beam at or near broadside, as is clear from the array factor. Conversely, as frequency increases, the electrical path between elements will also increase, and the array factor will produce a beam moving from the backward to the forward quadrant [19]. Figure1 shows a top view of the proposed antenna, its di fferent parts, and the unit-cell of the radiating section, all described in detail hereinafter. The geometrical features were printed on a grounded Duroid 5880 substrate (εr = 2.24, tanδ = 0.004 at 60 GHz), suitable for high-frequency applications. At mm-wave frequencies, these values are a more exact determination of the relative permittivity than the typical values used (εr = 2.20, tanδ = 0.0009 at 10 GHz) [20]. The chosen thickness is the minimum among the standards (127 µm) to minimize the radiation losses and the power lost to surface-waves [21] of the different printed structures. The antenna is excited by a 2 1 array of UTC-PD × on InP. Each UTC-PD presents a 50-Ω grounded coplanar waveguide (G-CPW) output and feeds two rows of the array. The length of each row is truncated by a matched patch, where a longer row will Appl. Sci. 2020, 10, 3474 3 of 16 feature a more directive beam. Additionally, this design is scalable in the direction perpendicular to theAppl. rows, Sci. 2020 so, that10, x FOR more PEER rows REVIEW can be added. By controlling the number of rows and their length, one3 of can 16 easily accomplish symmetrical patterns (at E- and H-planes) or increase the directivity or the emitted their length, one can easily accomplish symmetrical patterns (at E- and H-planes) or increase the power. In this respect, the proposed structure is advantageous to other classical sub-millimeter wave directivity or the emitted power. In this respect, the proposed structure is advantageous to other antennas, such as horns or on-chip antennas with silicon substrate lenses, which are more challenging classical sub-millimeter wave antennas, such as horns or on-chip antennas with silicon substrate to escalate. The antenna was designed using a full-wave simulator (ANSYS HFSS, version 19.0.0) and lenses, which are more challenging to escalate. The antenna was designed using a full-wave simulator transmission line modeling for fast and preliminary layouts. (ANSYS HFSS, version 19.0.0) and transmission line modeling for fast and preliminary layouts.

Figure 1. Layout of the designed antenna with the different sections indicated, and unit-cell of the radiatingFigure 1. section.Layout of the designed antenna with the different sections indicated, and unit-cell of the radiating section. 2.1. Feed Network 2.1. Feed network The antenna feed network comprises the section between the photodiode chip and the rows. It is composedThe antenna of a tapered feed transitionnetwork comprises to adapt the the width section diff betweenerence between the photodiode the photodiode chip and output the rows. and the It 50-is composedΩ G-CPW of on a tapered the Duroid transition 5880 substrate to adapt (asthe detailedwidth difference in Figure between 6a). After the this photodiode transition, output a section and followsthe 50-Ω in G-CPW which on the the two Duroid G-CPWs 5880 separate substrate to (as conform detailed to thein Figure row separation. 6a). After this Finally, transition, the last a section isfollows a transition in which to athe microstrip two G-CPWs transmission separate lineto conform (TL) and to a the symmetric row separation. corporate Finally, feed network the last section for the arrayis a transition rows. It to was a microstrip designed totransmission minimize the line reflection (TL) and coea symmetricfficient, cross-talk corporate between feed network photodiodes, for the andarray the rows. radiation It was losses designed that couldto minimize degrade the the reflec patterns.tion coefficient, cross-talk between photodiodes, and the radiation losses that could degrade the patterns. 2.2. RF Choke 2.2. RFAntennas choke integrated with photodiodes often require a polarization network to enhance the photodiodeAntennas responsivity. integrated with Therefore, photodiodes one must often include require an a RFpolarization choke at network the end ofto eachenhance row the to separatephotodiode the responsivity. radiating section Therefore, from theone pads must used include to polarize an RF choke the UTC-PD.at the end The of each choke row is to developed separate inthe microstrip radiating technology,section from and the it featurespads used a simulated to polarize RF isolationthe UTC-PD. better The than choke 35 dB is within developed the array in operatingmicrostrip frequency technology, band. and It it consists features of a seven simulated cascaded RF identicalisolation sections,better than as shown 35 dB inwithin Figure the2 (right). array Eachoperating section frequencyis T-shaped, band. and It theconsists horizontal of seven bar iscascaded composed identical of two sections, parallel, asλ/4 shown open-ended in Figure stubs. 2 The(right). stubs Each create section a short-circuit, is T-shaped,resulting and the horizontal in an RF choke. bar is composed Besides, the of two length parallel, of the λ T’s/4 open-ended vertical bar isstubs. chosen The to stubs translate create the a short-circuit, shorted microstrip resulting to ain large an RF inductive choke. Besides, load, so the that length it presents of the T’s high vertical input impedance,bar is chosen and to thistranslate effect isthe reinforced, shorted microstrip cascading them.to a large Figure inductive2 (left and load, center) so representsthat it presents simulated high andinput calculated impedance, isolation, and this|S 21effect|, depending is reinforced, on the ca numberscading ofthem. sections Figure as 2 well (left as and the center) magnitude represents of the reflectionsimulated coeandffi cient,calculated|S11|, forisolation, the 7-section |S21|, choke,depending which on is the better number than –0.85of sections dB in theas wholewell as band. the Inmagnitude practice, of the the stubs reflection do not coefficient, create an exact|S11|, short-circuit,for the 7-section and choke, several which sections is better need than to be –0.85 cascaded. dB in Calculationsthe whole band. are performed In practice, retrieving the stubs the do ABCD not create matrix an of exact the parallel short-circuit, stubs from and full-waveseveral sections simulations, need takingto be cascaded. the vertical Calculations bar as a section are performed of a transmission retrievi line,ng the and ABCD cascading matrix the of sections. the parallel stubs from full-wave simulations, taking the vertical bar as a section of a transmission line, and cascading the sections. Appl. Sci. 2020, 10, 3474 4 of 16 Appl. Sci. 2020, 10, x FOR PEER REVIEW 4 of 16

Figure 2. Left: Magnitude of isolation, |S |, for a choke composed of 1, 3, 5, and 7 sections. Center: Figure 2. Left: Magnitude of isolation, |S2121|, for a choke composed of 1, 3, 5, and 7 sections. Center: magnitude of the reflection coefficient, |S |, for a 7-section choke. Continuous line: calculated by a magnitude of the reflection coefficient, |S1111|, for a 7-section choke. Continuous line: calculated by a transmissiontransmission lineline (TL)(TL) equivalent.equivalent. DashedDashed line:line: simulated. Right: View View of of the the choke. choke. 2.3. Radiating Section 2.3. Radiating section The radiating section of the array comprises four identical rows. Every row consists of a microstrip The radiating section of the array comprises four identical rows. Every row consists of a line periodically loaded by groups of four stubs plus a patch that truncates it, as shown in Figure1. microstrip line periodically loaded by groups of four stubs plus a patch that truncates it, as shown in The design of the periodic structure is crucial to obtain low values of |S11|, as explained below. Figure 1. The design of the periodic structure is crucial to obtain low values of |S11|, as explained One of the difficulties encountered while designing a 1-D periodic LWA is the scan through below. broadside. At this point, all reflections from the perturbations add in phase back to the source. This can One of the difficulties encountered while designing a 1-D periodic LWA is the scan through create a bandgap in the k-β diagram; in this case the wave does not propagate into the structure broadside. At this point, all reflections from the perturbations add in phase back to the source. This and is completely reflected, degrading the input impedance matching. Following the discussion can create a bandgap in the k-β diagram; in this case the wave does not propagate into the structure in [19], the perturbations of the periodic structure are modeled in a simplified network by shunt and is completely reflected, degrading the input impedance matching. Following the discussion in impedances connected by segments of TL. At broadside emission, the wave impinges the perturbations [19], the perturbations of the periodic structure are modeled in a simplified network by shunt in phase, and the equivalent network is a shunt of infinite loads, which produces a short-circuit and the impedances connected by segments of TL. At broadside emission, the wave impinges the structure reflects the wave completely. On the other hand, if substantial losses are introduced by each perturbations in phase, and the equivalent network is a shunt of infinite loads, which produces a perturbation, the current will progressively decay, effectively breaking the infinite periodicity, and a short-circuit and the structure reflects the wave completely. On the other hand, if substantial losses short-circuit will not be achieved. Hence, the latter case will present better matching characteristics. are introduced by each perturbation, the current will progressively decay, effectively breaking the In practice, each perturbation introduces losses and reflections to some degree. They have to be infinite periodicity, and a short-circuit will not be achieved. Hence, the latter case will present better traded-off to obtain the desired radiation and impedance matching characteristics. Since the full-wave matching characteristics. simulation of electrically large structures can be a computationally intensive task, it is preferable In practice, each perturbation introduces losses and reflections to some degree. They have to be to use approximate criteria to optimize the period of the structure at the frequency of broadside traded-off to obtain the desired radiation and impedance matching characteristics. Since the full- radiation. To that end, it is assumed that coupling and surface-wave generation are negligible. In [22] wave simulation of electrically large structures can be a computationally intensive task, it is it was shown that a spacing larger than 0.1λg leads to tolerable coupling effects between stubs in a preferable to use approximate criteria to optimize the period of the structure at the frequency of loaded microscrip, where λg is the guided wavelength in the microstrip TL. The calculation consists of broadside radiation. To that end, it is assumed that coupling and surface-wave generation are computing the S-parameters of an N-periods structure (STot), from the scattering matrix of just one. negligible. In [22] it was shown that a spacing larger than 0.1λg leads to tolerable coupling effects The structure is assumed to be terminated in a matched load, and only the first bounce is considered between stubs in a loaded microscrip, where λg is the guided wavelength in the microstrip TL. The for reflections, whereas for transmission no bounces are summed up. This is true only if |S21| >> |S11|. calculation consists of computing the S-parameters of an N-periods structure (STot), from the In this manner, the following calculation yields expressions that can be readily computed and used scattering matrix of just one. The structure is assumed to be terminated in a matched load, and only for optimization: the first bounce is considered for reflections, whereas for transmission no bounces are summed up. NX1 Tot − n This is true only if |S21| >> |S11|. In this manner,S S the followingS 2 calculation yields expressions that can(1) 11 ≈ 11 21 be readily computed and used for optimization: n=0 Tot N S21 S21 (2) ≈ 𝑆 ≈𝑆 𝑆 (1) The results from Equations (1) and (2), along with the exact computations, are shown in Figure3 √ √ for a 10% and a 20% loss per pass ( S21 = 0.9, 0.8, respectively). A symmetric unit-cell with | | 𝑆 ≈𝑆 (2) S11 = S22 was assumed to compute the exact values. |S11| is taken proportional to |S21| so that the The results from Equations (1)Tot and (2), along with the exact computations, are shown in Figure infinite periodic structure presents S11 equal to –10 dB (and the bandgap can be considered effectively removed3 for a 10% in and terms a 20% of impedanceloss per pass matching). (|𝑆| = √0.9 Figure, √0.83, respectively).shows the largest A symmetric deviation unit-cell between with exact S11 = andS22 was approximate assumed to curves. compute It occurs the exact when values. thephase |S11| is of takenS11 equals proportionalπ. Second to | andS21| higher-orderso that the infinite even bouncesperiodic addstructure in phase presents with the 𝑆 traveling equal wave, to –10 yielding dB (and lower the transmissionbandgap can losses be considered than the approximate effectively removed in terms of impedance matching). Figure 3 shows the largest deviation between exact and Appl. Sci. 2020, 10, x FOR PEER REVIEW 5 of 16

Appl. Sci. 2020approximate, 10, 3474 curves. It occurs when the phase of S11 equals π. Second and higher-order 5even of 16 bounces add in phase with the traveling wave, yielding lower transmission losses than the approximate calculation. Additionally, third and higher-order odd bounces add in phase with the first bounce so calculation. Additionally, third and higher-order odd bounces add in phase with the first bounce so that S11 is underestimated in the approximate curves. As |𝑆 | decreases, the agreement is better, that S is underestimated in the approximate curves. As STot decreases, the agreement is better, while 11 while it degrades the other way around. In the latte11 r case, the effects of stationary waves should be it degradesincluded the other computing way around. multiple In the bounces. latter case, Finally, the eff ectsit is ofworth stationary mentioning waves shouldthat as bethe included losses per pass computing multiple bounces. Finally, it is worth mentioning that as the losses per pass increase, increase, |S11| can be proportionally larger that |S21| to achieve the same 𝑆 threshold (the sum in |S | can be proportionally larger that |S | to achieve the same STot threshold (the sum in Equation (1) 11 Equation (1) approaches 1). Hence,21 increasing the losse11s per period relaxes the demand on S11 for approachesadequate 1). Hence, input increasing impedance the matching, losses per in period line with relaxes the earlier the demand discussion. on S11 Notefor adequatealso that truncating input the impedancestructure matching, helpsin lineto decreasewith the earlierthe reflection discussion. coeffici Noteent. also Finally, that truncating for frequencies the structure other helpsthan that of to decreasebroadside the reflection directed coe emission,fficient. (1) Finally, and (2) for can frequencies be adapted other to introduce than that the of broadsidephase change directed per period in emission, (1) and (2) can be adapted to introduce the phase change per period in S . S21. 21

Figure 3. FigureS-parameters 3. S-parameters as a function as a offunction the number of the of number periods of in pe theriods structure. in the structure. Continuous Continuous line: exact line: exact calculation.calculation. Dashed line: Dashed approximate line: approximate calculation calculation from (1) and from (2). (1) The and percentage (2). The percentage indicates indicates the power the power loss per passloss ofper one pass period of one of period the structure. of the structure.

As an exampleAs an ofexample the usefulness of the usefulness of (1) and (2),of (1) we and simulated (2), we thesimulated unit-cell the reported unit-cell in reported [23](Figure in [23] 10) (Figure that removed10) that the removed bandgap the at bandgap broadside. at broadside. The structure The canstructure be well can described be well descri by abed traveling by a traveling wave, wave, Tot Tot following our assumptions. S , S21 were computed for N = 32 and the approximate calculations following our assumptions.11 𝑆 , 𝑆 were computed for N = 32 and the approximate calculations yielded ayielded return lossa return of 14.5 loss dB of and14.5 a dB transmission and a transmission loss of 16.3 loss dB, of 16.3 whereas dB, whereas simulation simulation of the complete of the complete structurestructure (using waveports (using waveports at both ends)at both yielded ends) yielded 15.1 dB 15.1 and dB 18.5 and dB, 18.5 respectively, dB, respectively, with a CPUwith a CPU simulationsimulation time 18 timestime 18 larger times than larger the than single the unit-cell. single un Thisit-cell. indicates This indicates that Equations that Equations (1) and (2)(1) and (2) constituteconstitute a useful toola useful for preliminarytool for preliminary optimization optimization of this kind of this of structure. kind of structure. The unit-cellThe unit-cell used in ourused design in our was design first was reported first reported in [22], and in [22], it is shownand it is in shown Figure in4a. Figure The stub 4a. The stub λ length (l)lengthis approximately (l) is approximately/2, presenting λ/2, presenting high input high impedance, input impedance, to introduce to aintroduce small perturbation a small perturbation in the TL, whereasin the TL, its widthwhereas (w) itscontrols width the(w) amount controls of the radiated amount power of radiated [22]. In turn,power the [22]. stubs In formturn, groupsthe stubs form of four: twogroups towards of four: the righttwo towards and two towardsthe right the and left, tw equispacedo towards atthe distance left, equispaceds. This configuration at distance s. This allows one to increase the return loss if ∆arg(S ) per stub plus inner separation is approximately configuration allows one to increase the21 return loss if Δarg(S21) per stub plus inner separation is π /2. In addition,approximately it introduces π/2. Inmore addition, losses it perintroduces period, more so that losses the design per period, can be so made that the more design compact. can be made On the othermore hand, compact. the stubsOn the do other not radiatehand, the in phasestubs do but not in quadrature.radiate in phase This but results in quadrature. in a pattern This that results in points offa-broadside pattern that in points the plane off-broadside perpendicular in the to plane the row; perpendicular this effect is to conveniently the row; this eliminated effect is conveniently by the arrayeliminated factor of the by complete the array four-row factor of structure. the complete The simulatedfour-row structure. and calculated The simulatedS-parameters and using calculated S- Equationsparameters (1) and (2) using are shown Equations in Figure (1) and4b for(2) broadsideare shown emission. in Figure The4b for periodic broadside structure emission. presents The periodic S < –12 dB, and the calculated curve is in agreement with simulated values. The calculated values 11 structure presents S11 < –12 dB, and the calculated curve is in agreement with simulated values. The for S are slightly overestimated due to the excluding of the higher-order bounces in the calculation, 21 calculated values for S21 are slightly overestimated due to the excluding of the higher-order bounces as also occurredin the calculation, in Figure 3as. also Once occurred the group in Figure of four 3. stubs Once isthe optimized group of four (Figure stubs4b), isa optimized segment of(Figure 4b), π transmissiona segment line is of added transmission to obtain line the is required added to length, obtainp the, for required a total phase length, change p, for ofa total 2 in phaseS21 per change of period at the central frequency of operation. To do so, the S-matrix of the four stubs is retrieved from 2π in S21 per period at the central frequency of operation. To do so, the S-matrix of the four stubs is full-waveretrieved simulations, from andfull-wave the corresponding simulations, segmentand the co ofrresponding TL is calculated. segment When of severalTL is calculated. periods When are simulated,several the periods length are of simulated, TL needs the to be length re-tuned. of TL Itsneed finals to valuebe re-tuned. changes Its by final about value 4%. changes Hence, by about effects associated with surface waves and coupling introduce only a small perturbation. To show Appl. Sci. 2020, 10, 3474 6 of 16 Appl. Sci. 2020, 10, x FOR PEER REVIEW 6 of 16

4%. theHence, suppression effects associated of the open-stopband, with surface waves Figure and4c represents coupling theintroduce dispersion only diagrama small perturbation. and attenuation To showconstant the ofsuppression the optimized of the unit-cell open-stopband, within the Figure band 4c of represents interest (solid the dispersion black line). diagram Dispersion and and attenuationattenuation constant are also of plotted the optimized for the same unit-cell unit-cell wi geometrythin the withbandw of= 50intereµm st(solid (solid grey black line), line). and for a Dispersionunit-cell and with attenuation just a 50 µm are wide also stub plotted (dashed for greythe same line). unit-cell In the latter geometry cases, p withis tuned w = to 50 obtain µm (solid∆arg( S21) greyequaling line), and 2π forapproximately a unit-cell with at 81 just GHz. a 50 For µm the wide unit-cell stub (dashed with one grey stub, line). note In the the typical latter open-stopband cases, p is tuned to obtain Δarg(S21) equaling 2π approximately at 81 GHz. For the unit-cell with one stub, note behavior around the frequency of broadside emission (β–1p/π = 0) [19]. For β–1p/π equal exactly to the typical open-stopband behavior around the frequency of broadside emission (β–1p/π = 0) [19]. For 0, the wave is not radiated (α/k0 equals 0), and it is totally reflected (S11 > –1 dB in the strop-band β–1p/frequenciesπ equal exactly when to a su0, ffitheciently wave large is not structure radiated is ( simulated).α/k0 equalsFor 0), theand four-stub it is totally unit-cell reflected with (Sw11 => 50–1 µm, dB inthe the previous strop-band behavior frequencies is mitigated when sincea sufficiently the reflections large structure from the is stubs simulated). compensate For the to four-stub some degree, unit-cellbut itwithis still w observed= 50 µm, (in the this previous case, S11 behavior= –3 dB). is Finally, mitigated for the since designed the reflections unit-cell optimizedfrom the stubs using the compensateapproach to explained some degree, before, but theit is curves still observed present a(in much this case, smoother S11 = –3 behavior, dB). Finally, and thefor stopbandthe designed effect is unit-celllargely optimized mitigated using (S11 < the–12 a dB).pproach explained before, the curves present a much smoother behavior, and the stopband effect is largely mitigated (S11 < –12 dB).

FigureFigure 4. ( 4.a) (aTwo) Two consecutive consecutive periods alongalong with with representative representative geometrical geometrical parameters. parameters. (b) Optimized (b)

Optimized|S11| and |S|11S21| |andfor broadside|S21| for broadside emissionas emission a function as ofa function the number of the of elements number inof the elements periodic in structure the periodic(lines), structure and simulated (lines), values and simulated (dots). (c) Dispersionvalues (dots). diagram (c) andDispersion attenuation diagram constant and for attenuation three structures: constantthe optimizedfor three structures: one shown the in (optimizeda) with 4 stubs one shown per unit-cell in (a) andwithw 4= stubs450 µ perm (solidunit-cell black and lines), w = 450 a 4-stub µm (solidstructure black with lines),w = a 504-stubµm (solidstructure grey with lines), w = and 50 µm a structure (solid grey with lines), a single and stuba structure per unit-cell with a with singlew stub= 50 perµm unit-cell (dashed with grey lines).w = 50 µm (dashed grey lines).

Since the antenna will be tested in a wireless link with a QO parabolic mirror of 10.16 cm focal, Since the antenna will be tested in a wireless link with a QO parabolic mirror of 10.16 cm focal, its phase center should remain constant over the operation band. Therefore, a compact design (far-field its phase center should remain constant over the operation band. Therefore, a compact design (far- at 10.5 cm), with broadside directed radiation (for simpler alignment) was targeted. For that purpose, field at 10.5 cm), with broadside directed radiation (for simpler alignment) was targeted. For that the row is truncated after four periods in a patch that radiates the remaining power, between 10% and purpose, the row is truncated after four periods in a patch that radiates the remaining power, between 25%. The resonance frequency of the patch is fixed to 85 GHz, slightly higher than the central frequency 10% and 25%. The resonance frequency of the patch is fixed to 85 GHz, slightly higher than the central (81 GHz), to compensate for the lower losses at the higher end of the operation band. It is worth frequency (81 GHz), to compensate for the lower losses at the higher end of the operation band. It is mentioning that the narrowband patch used to truncate the antenna and make it compact degrades the worth mentioning that the narrowband patch used to truncate the antenna and make it compact patterns beyond its resonance frequency, and thus, it limits the antenna bandwidth. In line with the degrades the patterns beyond its resonance frequency, and thus, it limits the antenna bandwidth. In application discussed in Section5, by including more periods in the row, the bandwidth upper limit line with the application discussed in Section 5, by including more periods in the row, the bandwidth would be increased, the side-lobes would be decreased, and also it would help to obtain a narrower upper limit would be increased, the side-lobes would be decreased, and also it would help to obtain beam for more pointing resolution. a narrower beam for more pointing resolution. The above-described row is used to obtain the final planar array comprised of four rows. First, The above-described row is used to obtain the final planar array comprised of four rows. First, the row is duplicated to get a 2 1 array. Second, this 2 1 array is mirrored and excited in phase the row is duplicated to get a 2 × 1 array.× Second, this 2 × 1 array× is mirrored and excited in phase opposition, which leads to a significant reduction of the cross-polarization level and an E-plane pattern opposition, which leads to a significant reduction of the cross-polarization level and an E-plane pointing at broadside. The final structure consists of an approximately square array with a simulated pattern pointing at broadside. The final structure consists of an approximately square array with a directivity between 18.5 and 20.0 dBi, side-lobe level (SLL) < –9 dB, a half power beam width (HPBW) simulated directivity between 18.5 and 20.0 dBi, side-lobe level (SLL) < –9 dB, a half power beam < 13 , and a radiation efficiency better than 75% in the 75–86 GHz frequency range. Figure5a shows width (HPBW)◦ < 13˚, and a radiation efficiency better than 75% in the 75‒86 GHz frequency range. the S-parameters of the whole structure, and Figure5b depicts the E- and H-plane cuts at broadside Figure 5a shows the S-parameters of the whole structure, and Figure 5b depicts the E- and H-plane (81 GHz). The UTC-PDs are simulated as 50-Ω lumped ports, which corresponds to the characteristic cuts at broadside (81 GHz). The UTC-PDs are simulated as 50-Ω lumped ports, which corresponds impedance of the output G-CPW of the photodiodes. In the future, one could envisage integration of to the characteristic impedance of the output G-CPW of the photodiodes. In the future, one could the UTC-PD with a matching network, as in [24], whose output impedance is also 50 Ω. envisage integration of the UTC-PD with a matching network, as in [24], whose output impedance is also 50 Ω. Appl. Sci. 2020, 10, 3474 7 of 16 Appl. Sci. 2020, 10, x FOR PEER REVIEW 7 of 16

FigureFigure 5. (a 5.) Simulated(a) Simulated magnitude magnitude of the of reflection the reﬂection coeffi coecientﬃcient and cross-talk and cross-talk for the for complete the complete array. array. (b) Simulated(b) Simulated E-planeE-plane (black) (black) and andH-plane H-plane (grey) (grey) cuts cutsof the of radiation the radiation pattern pattern for broadside for broadside emission emission at 81at GHz. 81 GHz.

3. Antenna3. Antenna and andphotodiode Photodiode integration Integration In thisIn thissection, section, we webriefly brieﬂy describe describe the thephotodiode photodiode parameters parameters and andstructure structure as well as well as its as its integrationintegration with with the theantenna. antenna. Additional Additional details details can canbe found be found in [3]. in [The3]. Theavailable available chip chip consists consists of of twotwo UTC-PD UTC-PD on InP. on InP.They They are separated are separated 500 µm, 500 andµm, each and photodiode each photodiode is integrated is integrated with a with50-Ω aG- 50-Ω CPWG-CPW (see Figure (see Figure 6). The6 photodiode). The photodiode was reported was reported in [25] inand [25 presents] and presents a 3 dB aelectrical 3 dB electrical bandwidth bandwidth of 110 ofGHz.110 It GHz is optically. It is optically fed by fed a 4 by µm a 4spot-sizeµm spot-size lensed lensed fiber ﬁberas shown as shown in Figure in Figure 6a,c.6 a,c.The The measured measured Appl. Sci. 2020, 10, x FOR PEER REVIEW 8 of 16 UTC-PDUTC-PD responsivity responsivity is 0.17 is 0.17 mA/mW mA/mW at 0 atV 0bias. V bias. The chip is mounted on an Aluminum Nitride (AlN) block for mechanical support, handling, and heat dissipation. During the integration process, the AlN block is positioned parallel to the PCB in such a way that the G-CPW of the chip is aligned and at the same height as the PCB (Figure 6b). Then, both are glued to a common housing, and the UTC-PD is finally bonded to the PCB using a silver-based conductive epoxy (Figure 6c). This approach results in a thick thread. Compared to wire bonding, it reduces the self-inductance of the wire, yielding a better electrical connection. This has been evaluated by bonding two sections of G-CPW on Duroid 5880 separated by 150 µm. The insertion losses (IL) obtained for ball wire bonding at 67 GHz are 1.4 dB higher than the IL using conductive epoxy.

4. Measurements and results Two different sets of measurements were carried out to evaluate the performance of the designed antenna: electronic measurements to determine return loss and radiation pattern, and photonic measurements to examine the power radiated and its performance in a wireless transmission link. FigureFigure 6. Images 6. Images of the of integration the integration process: process: (a) designed (a) designed transition, transition, (b) AlN (b) AlNblock block and PCB and PCBaligned aligned and gluedand glued to the to housing, the housing, and ( andc) detail (c) detail of thread of thread bonding bonding and fi andber ﬁberused usedto excite to excite the UTC-PD. the UTC-PD. 4.1. Electronic measurements The chip is mounted on an Aluminum Nitride (AlN) block for mechanical support, handling, ForThe theset ofreflection electronic coefficient measurements measurements, was perfor themed procedure on a two-row described array prototypein Appendix with A a wasGSG and heat dissipation. During the integration process, the AlN block is positioned parallel to the PCB employedprobe (Form to de-embedFactor ACP the 110-T), probe and and an interconnectio Agilent N5242An. Measurement vector network results analyzer are shown (VNA) in with Figure a mm- 7. in such a way that the G-CPW of the chip is aligned and at the same height as the PCB (Figure6b). Thewave antenna extension was head well (OML adapted V10VNA2). throughout The the prototype operation included band anda tapered did not interconnection present an increased from the Then, both are glued to a common housing, and the UTC-PD is ﬁnally bonded to the PCB using reflectionGSG pads coefficient to G-CPW at to the excite frequency the antenna, of broa as dsideit can emissionbe seen in (81 Appendix GHz). The Figure disagreement A2d, e. between a silver-based conductive epoxy (Figure6c). This approach results in a thick thread. Compared simulation and measurement around 78 GHz could be attributed to an inexact value of the relative to wire bonding, it reduces the self-inductance of the wire, yielding a better electrical connection. permittivity of the substrate used in the simulation, and to errors associated with the electrical This has been evaluated by bonding two sections of G-CPW on Duroid 5880 separated by 150 µm. connection between the probe and the PCB. Finally, tolerances in the fabrication of the PCB could The insertion losses (IL) obtained for ball wire bonding at 67 GHz are 1.4 dB higher than the IL using also have impacted the performance. More precisely, VIA holes were positioned with an accuracy of conductive epoxy. ± 25 µm, which is comparable to the VIA-center to edge distance (150 µm). For pattern measurements, a spherical 3D scanner and a receiving horn were additionally used. Figure 8a shows the normalized co-polar directivity of the two-row array at different frequencies. A frequency-scanning range of 22˚ was achieved. Note that the cross-polarization component, emitted largely off the scanning range of the array, vanished in the complete structure (Figure 5b). Additionally, the SLL of the two-row array was higher than in the complete structure (Figure 5b) because the patterns were optimized for the prototype excited photonically. The SLL could be improved by including more periods in the row. Figure 8b shows the simulated and the measured peak gain for the two-row prototype adapted for probe measurements. The largest deviation between simulated and measured data equaled 1.4 dB, which was acceptable considering the method used to excite the antenna. This value of measured gain was also consistent with the simulated radiation efficiency (77% at broadside). In the measurements, a standard gain horn was first characterized as a reference to retrieve absolute values of gain. Additionally, the incident power upon the antenna, using the de-embedded impedance values, was recalculated. Note the antenna did not present degraded gain at the frequency of broadside emission due to the good input matching observed in Figure 7. Therefore, the operation range was extended from the backward quadrant (θ < 0˚) to the broadside and the forward quadrant. Appl. Sci. 2020, 10, 3474 8 of 16

4. Measurements and Results Two diﬀerent sets of measurements were carried out to evaluate the performance of the designed antenna: electronic measurements to determine return loss and radiation pattern, and photonic measurements to examine the power radiated and its performance in a wireless transmission link.

4.1. Electronic Measurements The set of electronic measurements was performed on a two-row array prototype with a GSG probe (Form Factor ACP 110-T), and an Agilent N5242A vector network analyzer (VNA) with a mm-wave extension head (OML V10VNA2). The prototype included a tapered interconnection from the GSG pads to G-CPW to excite the antenna, as it can be seen in AppendixA Figure A2d,e. For the reflection coefficient measurements, the procedure described in AppendixA was employed to de-embed the probe and interconnection. Measurement results are shown in Figure7. The antenna was well adapted throughout the operation band and did not present an increased reflection coefficient at the frequency of broadside emission (81 GHz). The disagreement between simulation and measurement around 78 GHz could be attributed to an inexact value of the relative permittivity of the substrate used in the simulation, and to errors associated with the electrical connection between the probe and the PCB. Finally, tolerances in the fabrication of the PCB could also have impacted the performance. More precisely, VIA holes were positioned with an accuracy of 25 µm, which is comparable to the Appl. Sci. 2020, 10, x FOR PEER REVIEW ± 9 of 16 VIA-center to edge distance (150 µm).

FigureFigure 7. 7.Magnitude Magnitude of of the the reflection reflection coe coefficientfficient for for the the two-row two-row prototype prototype adapted adapted for for probe probe measurements.measurements. The The black black dots dots represent represent the the measurement measurement taken taken directly directly from from the the VNA, VNA, whereas whereas the the blueblue curve curve represents represents the the measurement measurement after after de-embedding de-embedding the the probe probe and and the the interconnection. interconnection. For pattern measurements, a spherical 3D scanner and a receiving horn were additionally used. Figure8a shows the normalized co-polar directivity of the two-row array at di fferent frequencies. A frequency-scanning range of 22◦ was achieved. Note that the cross-polarization component, emitted largely off the scanning range of the array, vanished in the complete structure (Figure5b). Additionally, the SLL of the two-row array was higher than in the complete structure (Figure5b) because the patterns were optimized for the prototype excited photonically. The SLL could be improved by including more periods in the row. Figure8b shows the simulated and the measured peak gain for the two-row prototype adapted for probe measurements. The largest deviation between simulated and measured data equaled 1.4 dB, which was acceptable considering the method used to excite the antenna. This value of measured gain was also consistent with the simulated radiation efficiency (77% at broadside). In the measurements, a standard gain horn was first characterized as a reference to retrieve absolute values of gain. Additionally, the incident power upon the antenna, using the Figure 8. (a) Normalized co-polar directivity at H-plane at different frequencies for the two-row prototype adapted for probe measurements. Continuous line: simulation, dashed line: measurements. (b) Simulated and measured peak gain.

4.2. Photonic measurements The antenna integrated with the UTC-PD, as described in Section 3, was used in a wireless link, as well as to measure the output power, using one active UTC-PD. The link set-up is shown in Figure 9a. A standard photonic approach [2,26] was used to generate and modulate two optical tones of two distributed feedback lasers (Toptica Photonics DFB pro BFY), with a spacing in frequency equal to that of the mm-wave carrier. After modulation using on–off keying (OOK), the signal was amplified (EDFA), monitored in an optical spectrum analyzer (OSA), and coupled to the UTC-PD input optical dielectric waveguide by a lensed fiber (OZ Optics). Subsequently, a QO system is employed to collimate the radiation. Note that the antenna radiates perpendicularly to the PCB plane (Figure 9b). Hence, it is placed at the focal point of a parabolic mirror that collimates radiation horizontally, as is most convenient for the QO link. On the receiving side, a 10 cm diameter Teflon lens focuses the radiation onto a Zero bias Schottky Barrier Diode detector (ZSBD) [27]. After the mm-wave carrier is detected, the signal, in baseband, is filtered and amplified (LNA amplifier + limiting amplifier). Finally, the detected signal is compared to the original using a BER tester (Anritsu MP2100A). The link parameters are summarized in Table 1. Figure 9c shows the corresponding eye-diagram for error- free transmission at 2.15 Gbps. At higher data-rates, variations of the antenna phase center with frequency create jitter, which in turn distorts the communication. Finally, Figure 9d shows a photograph of the assembled prototype. Appl. Sci. 2020, 10, x FOR PEER REVIEW 9 of 16

Appl. Sci. 2020, 10, 3474 9 of 16 de-embedded impedance values, was recalculated. Note the antenna did not present degraded gain at the frequencyFigure 7. of Magnitude broadside of emission the reflection due to coefficient the good inputfor the matching two-row observedprototype inadapted Figure 7for. Therefore, probe the operationmeasurements. range The was black extended dots represent from the the backward measurement quadrant taken directly (θ < 0 from◦) to the the VNA, broadside whereas and the the forwardblue quadrant. curve represents the measurement after de-embedding the probe and the interconnection.

FigureFigure 8. 8.( a(a)) Normalized Normalized co-polar co-polar directivitydirectivity atat H-plane H-plane at at di differentfferent frequencies frequencies for for the the two-row two-row prototypeprototype adapted adapted for for probe probe measurements. measurements. Continuous Continuous line: line: simulation, simulation, dashed dashed line: line: measurements. measurements. (b()b Simulated) Simulated and and measured measured peak peak gain. gain. 4.2. Photonic Measurements 4.2. Photonic measurements The antenna integrated with the UTC-PD, as described in Section3, was used in a wireless The antenna integrated with the UTC-PD, as described in Section 3, was used in a wireless link, link, as well as to measure the output power, using one active UTC-PD. The link set-up is shown as well as to measure the output power, using one active UTC-PD. The link set-up is shown in Figure in Figure9a. A standard photonic approach [ 2,26] was used to generate and modulate two optical 9a. A standard photonic approach [2,26] was used to generate and modulate two optical tones of two tones of two distributed feedback lasers (Toptica Photonics DFB pro BFY), with a spacing in frequency distributed feedback lasers (Toptica Photonics DFB pro BFY), with a spacing in frequency equal to equal to that of the mm-wave carrier. After modulation using on–off keying (OOK), the signal was that of the mm-wave carrier. After modulation using on–off keying (OOK), the signal was amplified amplified (EDFA), monitored in an optical spectrum analyzer (OSA), and coupled to the UTC-PD input (EDFA), monitored in an optical spectrum analyzer (OSA), and coupled to the UTC-PD input optical optical dielectric waveguide by a lensed fiber (OZ Optics). Subsequently, a QO system is employed to dielectric waveguide by a lensed fiber (OZ Optics). Subsequently, a QO system is employed to collimate the radiation. Note that the antenna radiates perpendicularly to the PCB plane (Figure9b). collimate the radiation. Note that the antenna radiates perpendicularly to the PCB plane (Figure 9b). Hence, it is placed at the focal point of a parabolic mirror that collimates radiation horizontally, as Hence, it is placed at the focal point of a parabolic mirror that collimates radiation horizontally, as is is most convenient for the QO link. On the receiving side, a 10 cm diameter Teflon lens focuses the most convenient for the QO link. On the receiving side, a 10 cm diameter Teflon lens focuses the radiation onto a Zero bias Schottky Barrier Diode detector (ZSBD) [27]. After the mm-wave carrier radiation onto a Zero bias Schottky Barrier Diode detector (ZSBD) [27]. After the mm-wave carrier is is detected, the signal, in baseband, is filtered and amplified (LNA amplifier + limiting amplifier). detected, the signal, in baseband, is filtered and amplified (LNA amplifier + limiting amplifier). Finally, the detected signal is compared to the original using a BER tester (Anritsu MP2100A). The link Finally, the detected signal is compared to the original using a BER tester (Anritsu MP2100A). The parameters are summarized in Table1. Figure9c shows the corresponding eye-diagram for error-free link parameters are summarized in Table 1. Figure 9c shows the corresponding eye-diagram for error- transmission at 2.15 Gbps. At higher data-rates, variations of the antenna phase center with frequency free transmission at 2.15 Gbps. At higher data-rates, variations of the antenna phase center with create jitter, which in turn distorts the communication. Finally, Figure9d shows a photograph of the frequency create jitter, which in turn distorts the communication. Finally, Figure 9d shows a assembled prototype. photograph of the assembled prototype. Table 1. Wireless link with one active PD: summary of main parameters.

Parameter Value Data rate 2.15 Gbps RF carrier 80 GHz Modulation OOK Link distance 25 cm 12 Bit error rate <10− Photocurrent 7.0 mA Appl. Sci. 2020, 10, 3474 10 of 16 Appl. Sci. 2020, 10, x FOR PEER REVIEW 10 of 16

(a) 90º off-axis Δλ parabolic mirror Intensity modulator OSA f1 f2 PC EDFA f1 f2 DFB λ1 50:50 1:99

DFB λ2 PC Quadrature A Photonic point bias V Modulator Current meter antenna Control & Driver Monitor Teflon lens Δf

Limiting BP filter LNA mm-wave radiation Ampli. Single Mode Fiber OOK Control OUT IN RF BERT ZSBD

FigureFigure 9.9. ((aa)) Schematic Schematic of of the the set-up set-up utilized utilized for for the the wireless wireless link link (PC= (PC polarization= polarization controller, controller, BP= b c BPbandpass,= bandpass LNA=, LNA low= low noise noise amplifier). amplifier). (b) ( Simplified) Simplified QO QO system, system, (c () )eye eye diagram diagram for thethe error-freeerror-free transmission at 2.15 Gbps, and (d) photograph of the prototype. transmission at 2.15 Gbps, and (d) photograph of the prototype. Power measurements of the photonic prototype with a UTC-PD excited were performed on a Table 1. Wireless link with one active PD: summary of main parameters. similar set-up as the transmission link. The Teflon lens, ZSBD, and receiving chain were replaced by a THz absolute power meter (Thomas KeatingParameter Instruments). Additionally,Value the modulator was substituted by a fiber-coupled free-space collimatorData plusrate a chopper to 2.15 enable Gbps lock-in detection. Since the broadband detector presented a large windowRF in whichcarrier pressure variations 80 GHz due to thermal heating were measured, external stimuli could alter measurements.Modulation Therefore, theOOK dependence P I2 was checked before ∝ power measurements, P being theLink detected distance power and 25I cmthe DC photocurrent. This is of utmost Bit error rate < 10-12 importance if an end-fire antenna is measured. In this case, some light, which was chopped at the same Photocurrent 7.0 mA lock-in frequency, could couple and destroy the measurements. In this case, however, the previous Power measurements of the photonic prototype with a UTC-PD excited were performed on a relationship would be linear and it could be easily noted. Figure 10 shows the frequency dependence similar set-up as the transmission link. The Teflon lens, ZSBD, and receiving chain were replaced by a THz absolute power meter (Thomas Keating Instruments). Additionally, the modulator was Appl. Sci. 2020, 10, x FOR PEER REVIEW 11 of 16

substituted by a fiber-coupled free-space collimator plus a chopper to enable lock-in detection. Since the broadband detector presented a large window in which pressure variations due to thermal heating were measured, external stimuli could alter measurements. Therefore, the dependence 𝑃 ∝ 𝐼 was checked before power measurements, P being the detected power and I the DC photocurrent. This is of utmost importance if an end-fire antenna is measured. In this case, some light, which was Appl. Sci. 2020, 10, 3474 11 of 16 chopped at the same lock-in frequency, could couple and destroy the measurements. In this case, however, the previous relationship would be linear and it could be easily noted. Figure 10 shows the offrequency the detected dependence power. It of featured the detected a 3 dB power. bandwidth It featured of 11 GHz a 3 dB (74 bandwidth GHz to 85 GHz),of 11 GHz with (74 a maximum GHz to 85 ofGHz), 120 µ withW at a 80.5 maximum GHz, for of 45120 mW µW input at 80.5 optical GHz, powerfor 45 mW at –2.5 input V bias optical voltage power (9.8 at mA –2.5 photocurrent). V bias voltage This(9.8 wasmA photocurrent). in agreement withThis was the photodiodein agreement output with the power photodiode [25] when output bonding power losses [25] when (4–5 dB) bonding and antennalosses (4–5 radiation dB) and eﬃ antennaciency radiation (77%) were efficiency accounted (77%) for. were For accounted frequencies for. out For of frequencies the operation out band, of the theoperation mirror didband, not the capture mirror all did the not emitted capture radiation, all the em emitteditted radiation, largely oﬀ emitted-boresight, largely and off-boresight, the detected powerand the decreased. detected Thispower could decreased. be solved This using could a mirrorbe solved with using lower a F-numbermirror with or lower placing F-number the power or meterplacing window the power horizontally, meter window with the appropriatehorizontally, support with the and appropriate alignment structure.support and Nevertheless, alignment thesestructure. frequencies Nevertheless, are not withinthese frequencies the range of are interest not wi andthin are the not range measured of interest in the and present are not work. measured in the present work.

FigureFigure 10. 10.Detected Detected power power after after the the QO QO system. system.

5.5. FeaturedFeatured Application application InIn a a potential potential scenario, scenario, we we discuss discuss the the application application of the of the LWA LWA for beam for beam alignment alignment in wireless in wireless data centerdata center interconnections. interconnections. Wireless Wireless interconnections interconnect inions data in centers data werecenters proposed were proposed in [28] to in provide [28] to scalabilityprovide scalability and reconfigurability, and reconfigurability, which are which difficult are todifficult achieve to byachieve fiber by interconnected fiber interconnected nodes as nodes the numberas the number of servers of increases.servers increases. Mm-wave Mm-wave links at 60links GHz at or60 sub-THzGHz or sub-THz frequencies, frequencies, better suited better for suited tens offor Gbps tens dataof Gbps rates, data could rates, provide could a provide flexible a alternative flexible alternative to an all-fiber to an scenario. all-fiber scenario. FigureFigure 11 11aa depictsdepicts aa simplifiedsimplified datadata centercenter nodenodestructure structurewith withserver serverracks racksdistributed distributed inina a hexagonalhexagonal lattice. lattice. The The serversservers formform sub-groupssub-groups thatthat areare interconnectedinterconnected by by fiber,fiber, whereas whereas the the wireless wireless linkslinks provide provide flexible flexible interconnection interconnection between between them, them, thus thus forming forming a a hybrid hybrid architecture. architecture. More More details details onon hybrid hybrid architectures architectures can can be be found found in in [ 28[28].]. ElectronicElectronic beam beam steering steering with with phased phased arrays arrays is is frequently frequently proposed proposed in in the the literature literature as as a a means means toto alignalign thethe di differentfferent line-of-sight line-of-sight (LoS) (LoS) links links [ 16[16,28].,28]. However,However, mechanicalmechanical alignmentalignment isis moremore aaffordable,ffordable, given given the the cost cost and and energy energy consumption consumption of of many many active active elements elements required required in in a a high high gain gain phasedphased array. array.As As discussed discussednext, next, these these high-frequency high-frequency data data links links feature feature high high directivity directivity and and narrow narrow beamwidth,beamwidth, which which require require dedicated dedicated hardware, hardware, as as well well as as the the corresponding corresponding protocol protocol to to control control the the stringentstringent alignment. alignment. InIn [ 18[18]] and and in in the the present present manuscript, manuscript, we we reported reported on theon LWAthe LWA that producesthat produces a beam a beam squint squint with frequency,with frequency, which iswhich one of is theone simplest of the simplest forms to form steers theto steer beam. the It beam. shows It a mediumshows a directivitymedium directivity of about 18–20of about dBi, 18–20 points dBi, within points the within 22◦ scan the range, 22˚ scan and range, works and at the works E-band at the (from E-band 75 to (from 85 GHz). 75 to Owing 85 GHz). to theOwing antenna to the scalability, antenna thesescalability, characteristics these characteristics can be easily can adapted be easily for adapted a lower for HPBW a lower and HPBW enhanced and pointingenhanced resolution. pointing resolution. Indeed, the Indeed, row composed the row co ofmposed 19 periods of 19 presents periods an presents HPBW an of 4HPBW◦ (18.5 of dBi 4˚gain (18.5 fordBi one gain row), for one while row), maintaining while maintaining good input good impedance input impedance matching. matching. Besides, Besides, the same the same concept concept can likewise be employed at other frequencies [29]. In this use case, the LWA would be co-located with the transceivers on top of the racks in a structure that would offer in-plane mechanical rotation, as shown in Figure 11b. After, the LWA could be used to establish optimum alignment between transceivers in the link. During alignment, a signal with reduced bandwidth or lock-in detection could be employed, lowering the demand for highly directive antennas and eluding a dedicated QO system. For instance, Appl. Sci. 2020, 10, x FOR PEER REVIEW 12 of 16 can likewise be employed at other frequencies [29]. In this use case, the LWA would be co-located with the transceivers on top of the racks in a structure that would offer in-plane mechanical rotation, as shown in Figure 11b. After, the LWA could be used to establish optimum alignment between transceivers in the link. During alignment, a signal with reduced bandwidth or lock-in detection could be employed, lowering the demand for highly directive antennas and eluding a dedicated QO system. For instance, for a detector NEP of 2 𝑝𝑊/√𝐻𝑧 [30], and a signal bandwidth of 1 MHz, it suffices with 21 dBi directive antennas to obtain an SNR = 10 (0 dBm noiseless radiated power, 10 m distance, f = 75 GHz, and D = G for the Rx antenna). For 10 GHz bandwidth, the required directivity would increase to 31 dBi. Appl. Sci. 2020, 10, 3474 12 of 16 This solution would offer a fast and simple option for data center reconfigurability. Additionally, the transceiver hardware, represented in blue in Figure 11b, could be partially reused. For instance, itfor would a detector suffice NEP with of 2 anpW optical/ √Hz [30switch], and for a signal a photonically bandwidth generated of 1 MHz, itsignal suffices to withselect 21 either dBi directive of the antennas.antennas toTherefore, obtain an this SNR solution= 10 (0 would dBm noiseless play little radiated role in power, the overall 10 m system distance, costf = or75 size. GHz, The and PCBD = inG whichfor the the Rx antenna antenna). has For been 10 GHzfabricated bandwidth, is 30 mm the by required 30 mm, directivity with an approximate would increase packaging to 31 dBi. footprint of around 5–10 cm3.

FigureFigure 11. 11. ((aa)) Hybrid Hybrid interconnections interconnections in in a a data data center; center; the the continuous continuous yellow yellow lines lines represent represent wired wired fiberﬁber links links whereas whereas the the dashed dashed red red lines lines illustrate illustrate reconfigurable reconﬁgurable wireless wireless LoS LoS links. links. (b (b) )LWA LWA set-up, set-up, co-locatedco-located with with the the wireless wireless transceiver transceivers,s, to to enable enable alignment alignment in in a a LoS LoS link. link.

6. DiscussionThis solution would offer a fast and simple option for data center reconfigurability. Additionally, the transceiver hardware, represented in blue in Figure 11b, could be partially reused. For instance, In this manuscript, we reported the premier work on the design, photonic integration, and it would suffice with an optical switch for a photonically generated signal to select either of the characterization of a leaky-wave antenna array in PCB technology at E-band frequencies. The antennas. Therefore, this solution would play little role in the overall system cost or size. The PCB in designed antenna exhibits beam steering with frequency and a scalable layout. Measurements of the which the antenna has been fabricated is 30 mm by 30 mm, with an approximate packaging footprint radiation pattern are performed on a two-row array, adapted for probe excitation. A 22˚ scan range, of around 5–10 cm3. including broadside, is shown across the frequency of operation of the antenna (75 GHz to 85 GHz), which6. Discussion is in agreement with full-wave simulations. Measurements of return losses are also in good agreement with simulations. In this respect, a method to de-embed the probe plus interconnection is In this manuscript, we reported the premier work on the design, photonic integration, and described in the appendix. To the best of our knowledge, this is one of the few realizations of periodic characterization of a leaky-wave antenna array in PCB technology at E-band frequencies. The designed 1-D structures in which the complete periodic structure shows adequate matching characteristics (S11 antenna exhibits beam steering with frequency and a scalable layout. Measurements of the radiation < –12 dB) at broadside. This is due to increased losses per period, which also helps to obtain a compact pattern are performed on a two-row array, adapted for probe excitation. A 22 scan range, including structure and higher radiation efficiency, and to enlarge the scanning range of◦ the antenna from the broadside, is shown across the frequency of operation of the antenna (75 GHz to 85 GHz), which is in backward quadrant to broadside and the forward quadrant. Furthermore, the antenna is tested in a agreement with full-wave simulations. Measurements of return losses are also in good agreement with wireless communications system, demonstrating error-free transmission at 2.15 Gbps and a 25 cm simulations. In this respect, a method to de-embed the probe plus interconnection is described in the link distance. Finally, the application of the LWA antenna as a simple means to align mm-wave LoS appendix. To the best of our knowledge, this is one of the few realizations of periodic 1-D structures links was outlined. in which the complete periodic structure shows adequate matching characteristics (S11 < –12 dB) at broadside. This is due to increased losses per period, which also helps to obtain a compact structure and higher radiation efficiency, and to enlarge the scanning range of the antenna from the backward quadrant to broadside and the forward quadrant. Furthermore, the antenna is tested in a wireless communications system, demonstrating error-free transmission at 2.15 Gbps and a 25 cm link distance. Finally, the application of the LWA antenna as a simple means to align mm-wave LoS links was outlined.

Author Contributions: Conceptualization, Á.J.P.-G., R.S., L.E.G.-M., and D.G.-O.; methodology, Á.J.P.-G., M.A., and G.C.D.B.; formal analysis, Á.J.P.-G.; investigation, Á.J.P.-G., M.A., G.C.D.B., L.B., and F.F.; writing—original draft preparation, Á.J.P.-G.; writing—review and editing, L.E.G.-M., R.S., and D.G.-O.; supervision, G.C.D.B., R.S., L.E.G.-M., and D.G.-O.; funding acquisition, G.C.D.B., R.S., L.E.G.-M., and D.G.-O. All authors have read and agreed to the published version of the manuscript. Appl. Sci. 2020, 10, 3474 13 of 16

Funding: Á.J.P.-G., D.G.-O., and R.S. acknowledge funding by Labex CominLabs and by the European Union through the European Regional Development Fund (ERDF), and the French region of Brittany, Ministry of Higher Education and Research, Rennes Métropole and Conseil Départemental 35, through the CPER Project SOPHIE/STIC and Ondes. The work of M.A. and G.C.D.B. has been supported by the European Union’s Horizon 2020 research and innovation programme under the Marie Sklodowska-Curie grant agreement No 642355 FiWiN5G. L.E.G.-M. and G.C.D.B. acknowledge the Spanish Ministerio de Economia y Competitividad through Programa Estatal de Investigación, Desarrollo e Inovación Orientada a los Retos de la Sociedad (grant iTWIT, TEC2016-76997-C3-3-R). Acknowledgments: The authors would like to thank X. Morvan (M2ARS platform and Université de Rennes 1), O. de Sagazan (Nano-Rennes platform and Université de Rennes 1), and T. Batté (Nano-Rennes platform and INSA de Rennes) for manufacturing the housing and for assistance in the assembly of the antenna. The authors are also grateful to F. van Dijk (III-V Labs Thales-Alcatel) for lending the photodiode chip used in this work. Conﬂicts of Interest: The authors declare no conﬂict of interest.

Appendix A. Method for Probe De-Embedding S-parameter measurements at mm-wave frequencies with coplanar probes comprise the challenge of probe and interconnection de-embedding. Indeed, the ground signal ground (GSG) probe interconnection with the device under test (DUT), as well as the probe itself, can introduce a relevant contribution to the overall measurement, and thus its effect has to be removed for an accurate determination of S-parameters. Besides, for one-port components, only one-port probe measurements might be available. In this section, we describe the procedure employed to de-embed the probe and the interconnection from S11 measurements. The difficulty of fabrication of 50-Ω loads at mm-wave frequencies in the same substrate as the DUT makes the use of the standard short-open-load-through (SOLT) calibration impractical [31]. Another of the standard methods, through-reflection-line (TRL), requires the use of two probes. Finally, one finds open-short (OS) or the short-open (SO) methods, which include the modeling of parasitic lumped elements introduced by the probes plus interconnection. These methods, however, rely on a specific probe plus interconnection, and it was shown in [31] that beyond 30 GHz substantial discrepancies existed between both for a reverse-biased Schottky diode on a silicon substrate. The method presented herein is a variant from [31] based on simulation plus measurements. Instead of imposing S11 = S22, as it is done in [31], an additional load is measured to obtain a fourth relation. It consists of a set of four measurements, three on test loads whose values are assumed, plus another one on the DUT. The three measurements plus the reciprocity condition yield four relations from which the ABCD parameters of the interconnection can be retrieved analytically. It is also assumed that its ABCD matrix is maintained through measurements. This implies that the same interconnect has to be used, in addition to landing the probes at the same positions. The measurement schematic is shown in AppendixA Figure A1. The S11 parameter at a calibrated reference plane of an arbitrary structure plus load, ZL, is given by Equation (A1), where A, B, C, and D are the ABCD-matrix elements of the structure between the load and the reference plane: ZcZLC + ZcD AZL B S11 = − − (A1) −ZcZLC + ZcD + AZL + B A short-circuit (S) and an open circuit (O) are loads that can be relatively simply manufactured in the same conditions as the DUT. Substituting in Equation (A1), the corresponding relations are found for S and O: S B 1 S11 D = − , (A2) + S Zc 1 S11 o A 1 S11 C = − . (A3) + o Zc 1 S11 Appl. Sci. 2020, 10, x FOR PEER REVIEW 14 of 16 Appl. Sci. 2020, 10, 3474 14 of 16

Figure A1. Schematic used for 1-port S11 measurements with different loads: open, short, translated shortFigure (TS), A1. translated Schematic open used (TO), for 1-port and deviceS11 measurements under test with (DUT), different as well loads: as theopen, corresponding short, translated networkshort equivalent.(TS), translated open (TO), and device under test (DUT), as well as the corresponding network equivalent. If the probe plus interconnect structure are composed of passive, isotropic materials, one can enforce reciprocityThe S11 parameter and obtain at a thecalibrated following reference relation: plane of an arbitrary structure plus load, ZL, is given by Equation (A1), where A, B, C, and D are the ABCD-matrix elements of the structure between the AD BC = 1 (A4) load and the reference plane: − Finally, a third load is measured to obtain𝐙𝐜𝐙 from𝐋𝐂+𝐙 Equation𝐜𝐃−𝐀𝐙𝐋 (A1)−𝐁 the fourth relation. It consists of a 𝐒𝟏𝟏 =− (A1) translated short (TS) or translated open (TO).𝐙 The𝐜𝐙𝐋𝐂+𝐙 surge𝐜𝐃+𝐀𝐙 impedance𝐋 +𝐁 of the transmission line, Zc, and the propagationA short-circuit constant, are (S) obtained and an open from full-wavecircuit (O) simulations. are loads that These can arebe relatively parameters simply that can manufactured be easily simulatedin the same for standard conditions low-loss as the 50- DUT.Ω transmission Substituting lines. in Equation The translated (A1), lengththe corresponding of the load is chosenrelations as are λ/8found to distribute for S and evenly O: the three loads on the Smith chart. Note that if broadband measurements are performed, different windows of frequencies will require different translated lengths. B1−𝑆 The manufactured O, S, and TS are shownD= in Appendix , A Figure A2a–c, as well as one DUT(A2) Z 1+𝑆 (two-row array) in Figure A2d. To minimize the effect of the GSG pads, and to adapt the probe pitch to the G-CPW, a tapered interconnection is optimizedA (Appendix1−𝑆 A Figure A2e). Additionally, the short C= . (A3) Z1+𝑆 and open structures are optimized using full-wave simulations to obtain nominal S11 parameters. The sameIf interconnectthe probe plus geometry interconnect is used structure among devices are comp andosed also of redundant passive, isotro samplespic arematerials, measured one to can ensureAppl.enforce Sci. that2020 reciprocity, assuming 10, x FOR PEER the and ABCDREVIEW obtainmatrix the following constant relation: is a valid supposition. 15 of 16

AD − BC = 1 (A4) Finally, a third load is measured to obtain from Equation (A1) the fourth relation. It consists of a translated short (TS) or translated open (TO). The surge impedance of the transmission line, Zc, and the propagation constant, are obtained from full-wave simulations. These are parameters that can be easily simulated for standard low-loss 50-Ω transmission lines. The translated length of the load is chosenFigure as A2. λ DifferentDi/8 ﬀtoerent distribute structures structures evenly measured measured the for forthree probe probe loads de-embedding de-embedding on the inSmith inS-parameterS-parameter chart. Notemeasurements: measurements: that if broadband (a) measurements(open-circuit,a) open-circuit (b ,()are short-circuit,b) short-circuit,performed, (c) differenttranslated (c) translated windowsshort, short, and and(ofd) DUT.frequencies (d) DUT. In this In thisexamplewill example requir it consiste itdifferent consist of a two- of translated a lengths.two-rowrow stub stubarray. array. (e) Detail (e) Detail of the of probe the probe tip and tip andtapered tapered interconnection interconnection shared shared among among structures. structures. The manufactured O, S, and TS are shown in Appendix Figure A2a–c, as well as one DUT (two- Referencesrow array) in Figure A2d. To minimize the effect of the GSG pads, and to adapt the probe pitch to the G-CPW, a tapered interconnection is optimized (Appendix Figure A2e). Additionally, the short and 1.1. PreuPreu, S.; S.; Döhler, Döhler, G.H.; G.H.; Malzer, S.;S; Stöhr, Stöhr, A; A.; Rymanov, Rymanov, V; V.; Göbel, T.; Brown, E.R.;E.R; Feiginov, M.;M.; Gonzalo,Gonzalo, R.;R; open structures are optimized using full-wave simulations to obtain nominal S11 parameters. The Beruete, M.;M; et et al al.. Principles Principles of of THz THz generation. generation. 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