Performance of Reflector Antennas by Employing

PERFORMANCE OF REFLECTOR ANTENNAS BY EMPLOYING

TRIPLE MODE FEEDHORN AND A FREQUENCY RECONFIGURABLE

SPIRAL LOADED PLANAR DIPOLE ANTENNA

______

A Thesis

Presented to the

Faculty of

San Diego State University

______

In Partial Fulfillment

of the Requirements for the Degree

Master of Science

Electrical Engineering

______

Mukund Ranga Thyagarajan

Fall 2012

iii

ABSTRACT OF THE THESIS

Performance of Reflector Antennas by Employing Triple Mode Feedhorn and a Frequency Reconfigurable Spiral Loaded Planar Dipole Antenna by Mukund Ranga Thyagarajan Master of Science in Electrical Engineering San Diego State University, 2012

This thesis focuses on two aspects: triple mode feedhorn as a feed source to reflector antennas and a compact spiral loaded planar dipole antenna with frequency reconfiguration. For the first aspect, parabolic symmetric and offset reflector antennas are adaptively illuminated using a novel triple (TE11+ TM01+ TE21) modes feedhorn with different mode combinations and impedance and radiation performances are presented. The combination of the radiating modes in a feedhorn with proper amplitude and fixed phase values helps in electronically pointing the main beam of the radiating patterns such as obtained in a beam steering antenna with limited beam scan range. This type of radiation performance virtually creates a displaced phase center location for the feedhorn, which consequently, adaptively illuminates the reflector antenna surface. Impedance matching bandwidths are preserved for both reflector antennas similar to the case of feedhorn alone. The co-polarization gain and peak cross-polarization levels are far better with the offset reflector antenna than the symmetric reflector antenna. Such reflector antennas find applications in radars. The simulation and analysis have been performed using EMSS's FEKO tool which is a Method of Moments (MOM) based Maxwell equation solver. The other design performed is a reconfigurable spiral loaded planar dipole antenna design which is frequency reconfigurable in the 0.76 GHz, 1.47 GHz and 2.2 GHz bands by employing PIN diodes as switching elements. This design is implemented to show the compactness achieved using spiraling in a planar dipole antenna. This antenna was fabricated and tested with nearly matching results. It can be used in wireless communication applications and for many other devices which operate over these frequency bands.

TABLE OF CONTENTS

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ABSTRACT ...... iv LIST OF TABLES ...... viii LIST OF FIGURES ...... x ACKNOWLEDGEMENTS ...... xiv CHAPTER 1 INTRODUCTION ...... 1 1.1 Feedhorns ...... 1 1.1.1 Pyramidal Horn ...... 4 1.1.2 Conical Horn ...... 4 1.1.3 Corrugated Horn ...... 6 1.2 Reflector Antenna and Types ...... 6 1.2.1 Plane Reflector ...... 9 1.2.2. Corner Reflector...... 10 1.2.3 Parabolic Reflector ...... 10 1.2.3.1 Front Fed Parabolic Reflector ...... 12 1.2.3.2 Offset Reflector Antenna ...... 12 1.2.3.3 Cassegrain Reflector Antenna...... 15 1.3 Reconfigurability in Spiral Loaded Planar Dipole Antenna ...... 15 1.4 Simulation Techniques and Experimental Verification Methods ...... 17 1.5 Organization of Thesis ...... 18 2 DOMINANT MODE FEEDHORN REFLECTOR SIMULATIONS USING GRASP AND FEKO ...... 19 2.1 Introduction ...... 19 2.2 TICRA Grasp Simulations ...... 19 2.3 Design and Analysis of A Cylindrical Feedhorn ...... 23 2.4 Conical Feedhorn Symmetrically Feeding A Large Reflector ...... 26 2.5 Conclusion ...... 31

3 TRIPLE MODE FEEDHORN PERFORMANCE AND PHASE CENTER DETERMINATION ...... 33 3.1 Introduction ...... 33 3.2 Design Methodology ...... 34 3.3 Simulated Results and Performance ...... 35 3.3.1 Individual Mode Analysis of the Triple Mode Feedhorn ...... 37 3.3.2 Dual Mode Analysis for Various Mode Combinations ...... 41 3.3.3 Triple Mode Analysis for Various Mode Combinations ...... 42 3.4 Phase Center Determination ...... 44 3.5 Wilkinson Unequal 1:5 Power Divider Design and Results ...... 50 3.6 Conclusion ...... 56 4 SYMMETRIC REFLECTOR ANTENNA FED BY TRIPLE MODE FEEDHORN ...... 62 4.1 Introduction to Reflector Antennas ...... 62 4.2 Design and Simulated Results ...... 66 4.2.1 Single Mode Analysis ...... 67 4.2.2 Dual Mode Analysis ...... 68 4.2.3 Triple Mode Analysis ...... 72 4.3 Analysis on Amplitude Variation on Triple Mode Combination ...... 74 4.4 Conclusion ...... 77 5 OFFSET REFLECTOR ANTENNA FED BY TRIPLE MODE FEEDHORN ...... 79 5.1 Introduction ...... 79 5.2 Design and Simulated Results ...... 80 5.2.1 Single Mode Analysis ...... 81 5.2.2 Dual Mode Analysis ...... 82 5.2.3 Triple Mode Combination Analysis...... 86 5.3 Conclusion ...... 88 6 FREQUENCY RECONFIGURABLE SPIRAL LOADED PLANAR DIPOLE ...... 95 6.1 Introduction ...... 95 6.2 Planar vs. Spiral Dipole Antenna ...... 98 6.3 Frequency Reconfigurable Spiral Loaded Planar Dipole Antenna ...... 103 6.3 Conclusion ...... 110 7 CONCLUSIONS AND FUTURE STUDY ...... 115

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REFERENCES ...... 117

viii

LIST OF TABLES

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Table 2.1: Fundamental Mode Circular Waveguide Analysis for Variable f/D ...... 20 Table 2.2: Gaussian Beam Mode Analysis for Variable f/D ...... 20 Table 2.3: The Co-Polarization Gain Values Vs Frequency for Conical Feedhorn...... 28 Table 2.4: The Co-Polarization Gain Values Vs Frequency for Conical Feedhorn- Reflector Configuration...... 31 Table 3.1. Peak Co- and Cross-Polarization Gain Values at 7.73 GHz for Triple Mode Feedhorn with Equal Amplitude ...... 41

Table 3.2: A Study of the Dual Mode Analysis for the TE11+TM01 and TE11+TE21 Mode Combinations with ±90° Phase Assigned with 50% Power Being Assigned to Each Mode ...... 44 Table 3.3. A Comparison of the Various Gain Values for the Triple Mode Combination by Power Variation with ±90° Phase Applied between Modes and at 7.73GHz ...... 48 Table 3.4. Power Divider Values Based on Calculation ...... 53 Table 3.5. Simulated Output Power Values from the Power Divider Model ...... 55 Table 3.6. Co and Cross Polarization Gain Comparison for the Power Divider Design ...... 61 Table 4.1.Individual Mode Gain Values for Symmetric Reflector-Feedhorn Design at 7.73GHz ...... 72 Table 4.2. Dual Mode Combinations of the Feedhorn Feeding a Symmetric Reflector with 50% Power Supplied to each Mode at 7.73ghz...... 74 Table 4.3. Peak Co Polarization Gain and Peak Cross-Polarization Level at 7.73GHz for +90° ...... 77 Table 5.1. Individual Mode Gain Values for Offset Reflector-Feedhorn Design at 7.73GHz ...... 86 Table 5.2. Dual Mode Combination Analysis for the Offset Fed Triple Mode Feedhorn Design at 7.73GHz with Equal Amplitude Supplied to Modes...... 88 Table 5.3. Peak Co Polarization Gain and Peak Cross-Polarization Level Values for Mode TE11 (40%) + TM01 (30%) +TE21 (30%) at +90° at Different Frequencies ...... 94 Table 6.1. Spiral Parameters ...... 99

Table 6.2. Pin Diode Implemented Spiral Loaded Frequency Reconfigurable Antenna Design Dimensions (in mm unit) ...... 105

LIST OF FIGURES

PAGE

Figure 1.1. Representation of (a) A rectangular Horn antenna, (b) Reflector antenna fed by a horn antenna...... 2 Figure 1.2. Typical Electromagnetic horn antenna configuration...... 3 Figure 1.3. Geometry of conical horn...... 5 Figure 1.4. Pyramidal horn geometry with corrugations in E-plane (a) Corrugated horn, (b) E-plane view...... 7 Figure 1.5. Design model of a corrugated horn antenna from Antenna Magus...... 8 Figure 1.6. Fabricated model of horn antenna with choke rings...... 8 Figure 1.7. Geometrical configurations of popular reflector antenna systems...... 9 Figure 1.8. Front fed parabolic dish antenna at Stanford University...... 11 Figure 1.9. General configuration of an Offset fed reflector antenna...... 13 Figure 1.10. A Cassegrain system being used in Sweden...... 16 Figure 2.1. GRASP based Result for f/D =1, Diameter = 2.4 m, f = 2.4 m...... 21 Figure 2.2. GRASP based Result for f/D = 0.8, Diameter = 2.4 m, f = 1.92m...... 21 Figure 2.3. GRASP based Result for f/D = 0.6, Diameter = 2.4 m, f = 1.44 m...... 22 Figure 2.4. GRASP based Result for f/D = 0.5, Diameter = 2.4 m, f = 1.2 m...... 22 Figure 2.5. GRASP based Results for f/D = 0.3, Diameter = 2.4 m, f = 0.72m...... 23 Figure 2.6. The conical feedhorn model generated in CADFEKO tool...... 24 Figure 2.7. (a) The meshing and excitation shown in CADFEKO model. (b) POSTFEKO model with meshing...... 25 Figure 2.8. Gain radiation patterns showing the co-polarization and cross-polarization patterns for the conical feedhorn operating at 12.5 GHz...... 26 Figure 2.9. 3-D radiation patterns of conical feedhorn at 12.5GHz...... 27 Figure 2.10. Co-polarization radiation pattern of Gain vs theta over a range of frequencies...... 28 Figure 2.11 CADFEKO model of conical feedhorn symmetrically feeding a large reflector...... 29 Figure 2.12 CADFEKO and POSTFEKO models of conical feedhorn symmetrically feeding a large reflector...... 29

Figure 2.13. Co-polarization gain radiation pattern for the feedhorn-reflector combination...... 30 Figure 2.14. 3-D radiation pattern of the feedhorn reflector configuration...... 30 Figure 3.1. Triple mode feedhorn design model in HFSS and FEKO software...... 34 Figure 3.2. CADFEKO generated model of triple mode feedhorn along with dimensions...... 36 Figure 3.3. POSTFEKO generated model of the triple mode feedhorn shown with meshing...... 36 Figure 3.4. Reflection co-efficient magnitude Vs frequency plot for designed triple mode feedhorn...... 37 Figure 3.5. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11 Mode (b) TM01 Mode and TE21 Mode at 7.73 GHz...... 38 Figure 3.6. Co-polarization and cross-polarization gain radiation patterns for the feedhorn for (a) TE11+TM01 Mode with +90, (b) TE11+TM01 Mode with -90° at 7.73 GHz...... 43 Figure 3.7. Co-polarization and cross-polarization gain radiation patterns for the feedhorn for (a) TE11+TE21 Mode with +90°, (b) TE11+TE21 Mode with -90° at 7.73 GHz...... 45 Figure 3.8. Co and cross-polarization gain radiation patterns for the feedhorn (a) TE11(20%)+ TM01(40%)+TE21(40%) mode with +90° (b) TE11(20%)+ TM01(40%)+TE21(40%) mode with -90° at 7.73 GHz...... 47

Figure 3.9. Co and cross-polarization gain radiation patterns for the feedhorn (a) TE11 (40%) + TM01 (30%) + TE21 (30%) mode with +90° (b) TE11 (40%) + TM01 (30%) + TE21 (30%) mode with -90° at 7.73 GHz...... 49 Figure 3.10. Phase center representation for triple mode feedhorn...... 50 Figure 3.11. Equal split resistive power divider...... 51 Figure 3.12. A Wilkinson power divider in microstrip form with unequal power division...... 52 Figure 3.13. A 1:4 Wilkinson equal power divider...... 52 Figure 3.14. Verification of values used for designed power divider using Wilkinson calculator...... 53 Figure 3.15. (a) Top view of HFSS models for 7-8GHz. (b) Fabricated model of designed power divider...... 54 Figure 3.16 (a) Reflection co-efficient plot and (b) Mutual Coupling plot for power divider at 7-8GHz band...... 55 Figure 3.17. Top view of HFSS model for Wilkinson unequal power divider from 1- 2GHz band...... 56

xii

Figure 3.18. (a) Reflection co-efficient plot and (b) Mutual Coupling plot for power divider at 7-8GHz band...... 57 Figure 3.19. Co and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11 (40%) + TM01 (30%) + TE21 (30%), (b) TE11 (39%) + TM01 (26%) + TE21 (26%), (c) TE11 (24%) + TM01 (11%) + TE21 (11%) at +90° phase...... 58 Figure 4.1. An illustration of a parabolic reflector antenna configuration...... 63 Figure 4.2. Reflector antenna configurations (a) Single axi-symmetric, (b) dual axi- symmetric, (c) single offset and (d) dual offset reflector...... 64 Figure 4.3. Symmetric triple mode feedhorn reflector configuration...... 67 Figure 4.4. Reflection co-efficient magnitude for the symmetric feedhorn reflector configuration...... 68 Figure 4.5. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11 Mode, (b) TM01 Mode and TE21 Mode at 7.73GHz...... 69 Figure 4.6. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11+TM01 Mode with +90° , (b) TE11+TM01 Mode with -90° at 7.73GHz...... 73 Figure 4.7. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11+TE21 Mode with +90° , (b) TE11+TE21 Mode with -90° at 7.73GHz...... 75 Figure 4.8. Co and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11+TM01+TE21 mode with +90° , (b) TE11+TM01+TE21 mode with -90° at 7.73GHz...... 76 Figure 4.9. Co and cross-polarization gain patterns for the symmetric feed-reflector combination at +90° for (a) TE11 (20%) + TM01 (40%) + TE21 (40%) mode, (b) TE11 (60%) + TM01 (20%) + TE21 (20%) mode at 7.73 GHz...... 78 Figure 5.1. Offset triple mode feedhorn reflector configuration...... 81 Figure 5.2. Reflection co-efficient magnitude for the offset feedhorn reflector configuration...... 82 Figure 5.3. Co-polarization and cross-polarization gain radiation patterns for the Offset feedhorn-reflector combination for (a) TE11 mode, (b) TM01 mode and TE21 mode at 7.73GHz...... 83 Figure 5.4. Co-polarization and cross-polarization gain radiation patterns for the offset feedhorn-reflector combination for (a) TE11+TM01 mode with +90° , (b) TE11+TM01 mode with -90° at 7.73 GHz...... 87 Figure 5.5. Co-polarization and cross-polarization gain radiation patterns for the offset feedhorn-reflector combination for (a) TE11+TE21 mode with +90° , (b) TE11+TE21 mode with -90° at 7.73 GHz...... 89

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Figure 5.6. Co and cross-polarization gain radiation patterns for the offset feedhorn- reflector combination for (a) TE11+TM01+TE21 mode with +90° , (b) TE11+TM01+TE21 mode with -90° at 7.73GHz...... 90

Figure 5.7. Gain radiation patterns for offset reflector feedhorn design for TE11 (40%) + TM01 (30%) + TE21 (30%) mode at +90° for (a) 7.45GHz, (b) 7.73GHz and (c) 7.9GHz...... 91 Figure 6.1. A general representation of (a) basic dipole antenna (b) half wave dipole antenna...... 96 Figure 6.2. A 2D and 3D radiation pattern of a standard dipole...... 97 Figure 6.3. Geometry of (a) planar microstrip dipole antenna and (b) a spiral loaded microstrip dipole antenna...... 98

Figure 6.4. Comparison of the simulated reflection coefficient magnitudes (S11, dB) of the planar conventional dipole vs. simulated and measured spiral loaded dipole antenna...... 100 Figure 6.5. Comparison of the simulated peak realized gain of planar dipole vs. spiral loaded compact planar dipole antenna...... 100 Figure 6.6. (a) Current distribution of the spiral loaded dipole at 760MHz and (b) photograph of the fabricated antenna...... 102 Figure 6.7 Simulated 3-D radiation pattern at 760MHz for spiral loaded compact dipole antenna...... 103 Figure 6.8. Normalized (a) simulated and (b) measured radiation patterns at 760 MHz for the compact spiral loaded dipole antenna...... 104 Figure 6.9. Spiral dipole antenna simulated model having required bias network...... 105 Figure 6.10. Biasing circuit shown with values for ON and OFF state...... 107 Figure 6.11. Simulated vs. measured reflection co-efficient magnitudes (S11, dB) for the spiral loaded compact dipole antenna with PIN diode switches...... 108 Figure 6.12. Current distribution plots at (a) 2.4GHz, (b) 1.8GHz and (c) 0.750GHz ...... 109 Figure 6.13. Simulated 3D omni-directional radiation patterns for (a) 0.76GHz. (b) 1.47GHz. (c) 2.2GHz for the compact spiral loaded planar dipole antenna...... 110 Figure 6.14. Measured normalized radiation patterns at (a) 0.760GHz, (b) 1.47GHz and (c) 2.2GHz for the compact spiral loaded dipole antenna...... 111

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ACKNOWLEDGEMENTS

I wish to first convey my sincere gratitude towards my advisor, Dr. Satish K. Sharma, for his advice, motivation, involvement and support throughout this research work. I am also thankful to Dr. Sharma for permitting access to the Antenna and Microwave Laboratory (AML), which allowed me to efficiently work on my research. I would also like to express my appreciation and thanks to some of the past students in Dr. Sharma’s research group especially Mehak Garg, Anup Kulkarni, Balamurugan Shanmugam, Jennifer Taylor and Rafid Damman for their help during the performance of this research work. I would also like to thank my thesis committee members Prof. Fletcher J. Miller and Prof. Long Lee for spending their time in reading this thesis and providing suggestions. I would like to thank my family for their unconditional support and the Almighty for his blessings. I would like to acknowledge funding from Dr. Satish K. Sharma’s National Science Foundation (NSF) CAREER grant # ECCS-0845822 for the period June 2010 – January 2012 for providing me financial assistance for performing the thesis research work.

CHAPTER 1

INTRODUCTION

A horn antenna comprises of a hollow waveguide structure used to convey radio waves between the transmitter and receiver. The waveguide of horn is tapered to a larger opening and the type, direction and amount of taper (flare) can have a profound effect on the overall performance as a radiator [1]. An electromagnetic horn structure behaves like a transition medium for the electromagnetic waves generated by a source and they finally propagate into free space through aperture of the horn [2]. The horn antenna can be used as a feed source to a reflector antenna and in that, this is referred as a “feedhorn”. Feedhorns have wide variety of uses from small-aperture antennas to feed sources to reflectors and they can be excited in any polarization or combinations of polarizations [3]. Feedhorn antennas are used as feed source for a reflector and is observed to have several applications due to the highly directive radiation patterns generated by this combination. Common applications of feedhorn reflectors are satellite communication, radar, remote sensing and tracking to mention a few. This combination assembly is discussed in depth in the later chapters. A basic horn antenna and combination of feedhorn-reflector are shown in Figure 1.1 (a) and (b) [4], respectively.

1.1 FEEDHORNS Feedhorn antenna comprises of a horn antenna acting as feeding element to a large reflector or other antenna geometries. Feedhorn belongs to the family of aperture antennas and hence the principle of field equivalence which states 'actual sources such as antennas are replaced by equivalent sources', can be used to compute its radiation characteristics [1]. Fields being generated at aperture of the horn can be determined by treating the horn as a radial waveguide [5], [6].The fields within the horn can be expressed in terms of cylindrical TE and TM wave functions which include Hankel functions by which the fields at the aperture of the horn and also within the horn are determined [1].

(a)

(b) Figure 1.1. Representation of (a) A rectangular Horn antenna, (b) Reflector antenna fed by a horn antenna. Source: FEKO. (2012). Homepage [Online]. Available: http://www.feko.info/.

A feedhorn is a small horn antenna used to convey radio waves between the transmitter and/or receiver and the reflector, particularly in parabolic antennas. In transmitting antennas, feedhorn is connected to the transmitter and converts the radio frequency alternating current from the transmitter to radio waves and feeds them to the rest of the antenna, which focuses them into a beam. In receiving antennas, incoming radio waves

3 are gathered and focused by the antenna's reflector on the feedhorn, which converts them to a tiny radio frequency voltage which is amplified by the receiver. Feedhorns are used mainly at microwave and higher frequencies in parabolic antennas and due to its ease of construction, durability and a high gain, it has applications in several areas [7]. The horn antenna can be manufactured in different ways depending upon applications desired and several other factors. Typical electromagnetic horn antenna configurations are shown in Figure 1.2 [1].

Figure 1.2. Typical Electromagnetic horn antenna configuration. Source: C. A. Balanis, Antenna Theory: Anal. and Design, 3rd ed. Hoboken: John Wiley & Sons, 2005.

The most widely used forms of horn antennas based on their design and applications are classified below:  Pyramidal Horn  Conical Horn  Corrugated Horn

Above mentioned are a few horn antenna classifications which have been developed over the years for a particular application and have been derived from the basic horn geometry by modifying a particular parameter which produces these different structures. A brief description about each has been stated below.

1.1.1 Pyramidal Horn A horn antenna is in the shape of a four sided pyramid, with rectangular cross-section at the aperture. They are a common type of horn used with rectangular waveguides and radiate linearly polarized radio waves and are most commonly used in industries or for research work mainly due to the fact that they provide pretty high gain characteristics. This kind of feedhorn is the most preferred one for use as a standard horn antenna in anechoic chambers for calibration and antenna measurements. A pyramidal horn antenna geometry is as shown in Figure 1.2(c). A pyramidal horn can be flared in both E-plane and H-plane and its radiation characteristics are essentially a combination of the E-plane and H-plane sectoral horns [1]. Gain of a pyramidal horn antenna is given by the Equation. 1.1. [1] (1.1) where,  A is the area of the aperture,  d is the aperture diameter of a conical horn  λ is the wavelength,

 eA is a dimensionless parameter between 0 and 1 called the aperture efficiency. The aperture efficiency ranges from 0.4 to 0.8 in practical horn antennas. For

optimum pyramidal horns, eA = 0.511 and the aperture efficiency increases with the length of the horn.

1.1.2 Conical Horn A conical antenna is horn antenna in the shape of a cone which has circular cross sections in it. Geometry of the conical horn antenna is as shown in Figure 1.3 [1]. Dominant mode present in a conical horn is TE11 mode when compared to the dominant mode of

pyramidal horn is TE10 mode and its polarization properties differ accordingly.

Figure 1.3. Geometry of conical horn. Source: C. A. Balanis, Antenna Theory: Anal. and Design, 3rd ed. Hoboken: John Wiley & Sons, 2005.

Gain of a conical feedhorn is defined by Equation. 1.2 [1] with the parameter description being same as that shown for a pyramidal horn gain. (1.2) Above defined is the standard gain for a conical horn antenna operating in any mode. For a conical horn working in the dominant mode, gain is given by Equation. 1.3 [1]. 20 (1.3)

Where, Rc is the reduction factor which accounts for loss in gain due to spherical wave phase error. Over the years, conical horn antennas have been modified in many ways and one such important modification is the addition of choke rings and having corrugations on inner walls of conical horn. Advantage of having choke rings on aperture mainly helps in reducing cross- polarization gain and also it reduces the spillover efficiency. Other advantage is since the aperture diameter increases with choke ring addition; beam width obtained here gets narrower and thus increases directivity of the design [8], [9]. This characteristic of conical feedhorn is studied in later sections.

1.1.3 Corrugated Horn Corrugated horn is a horn antenna with parallel slots or grooves covering inner surface of the horn and being transverse to horn's axis [7]. The main motive of having slots or grooves was to minimize spurious diffractions at edges of aperture [10]. The corrugated horns have advantage of wider bandwidth and smaller side lobes and cross-polarization. Hence they are widely used as feedhorns for satellite dishes and radio telescopes. The other advantage is aperture efficiency of the order 75%-80% in case of corrugated feedhorn when compared to 50-60% obtained by using a conventional feedhorn. A corrugated (grooved) pyramidal horn with corrugations in the E-plane walls is shown in Figure 1.4 (a) and (b) [1]. Figure 1.4 (a) shows the corrugated horn and its side view or the E-plane view (Figure 1.4 (b)). Since diffraction at the edges of the aperture in the H-plane is minimal corrugations are usually not placed on the walls of that plane. The surface reactance of a corrugated surface which are used on the walls of horn must be capacitive in nature in order for the surface tangential magnetic field parallel to the edge of the wall to force zero. Effect of corrugations on the walls of a horn is to modify electric field distribution in E-plane from uniform to cosine (at aperture). Through measurements it has been shown that transition from uniform to cosine distribution takes place almost at the onset of corrugations [1]. Corrugations in a horn antenna are not limited to just pyramidal horns. They can also be placed in a conical horn forming a conical corrugate horn which may also be referred to as scalar horn as stated in [10]. Several designs of conical corrugated horns have been investigated in terms of pattern symmetry, low cross polarization, low side lobe levels, circular polarization, axial ratio and phase center [11]-[13]. A designed model of corrugated horn is shown below in Figure 1.5 [4]. Another way of introducing corrugations in a horn antenna is by introducing choke rings fixed at the horn aperture and an example of this is shown in Figure 1.6 [15]. This technique is used throughout our design for triple mode feedhorn and is discussed in later sections.

1.2 REFLECTOR ANTENNA AND TYPES A reflector antenna is a passive device which reflects electromagnetic waves and has been in use since the discovery of electromagnetic wave propagation in 1888 by Hertz.

(a)

(b) Figure 1.4. Pyramidal horn geometry with corrugations in E- plane (a) Corrugated horn, (b) E-plane view. Source: C. A. Balanis, Antenna Theory: Anal. and Design, 3rd ed. Hoboken: John Wiley & Sons, 2005.

Subsequent demands of reflectors for use in radio astronomy, microwave communication and satellite tracking resulted in a huge demand. This resulted in progress in the development of sophisticated analytical and experimental techniques in shaping reflector surfaces and optimizing illumination over their apertures to maximize the gain [1]. Reflector antennas can be modeled in different configuration, the most common and popular ones are:  Plane Reflector.  Corner Reflector  Parabolic Reflector

Figure 1.5. Design model of a corrugated horn antenna from Antenna Magus. Source: FEKO. (2012). Homepage [Online]. Available: http://www.feko.info/

Figure 1.6. Fabricated model of horn antenna with choke rings. Source: T. A. Milligan. (2005). Horn Antennas (2nd ed.) [Online]. Available: http://www.microwave.gr/content/ horns.pdf

Figure 1.7 [1] shows some of the different reflector types and Figure.1.7 (c) and (d) being subtypes of parabolic reflector antennas.

Figure 1.7. Geometrical configurations of popular reflector antenna systems. Source: C. A. Balanis, Antenna Theory: Anal. and Design, 3rd ed. Hoboken: John Wiley & Sons, 2005.

1.2.1 Plane Reflector The simplest type of reflector antenna configuration is that of a plane reflector which directs energy in a desired direction. This configuration is shown in Figure 1.7 (a) where the radiating source is in a vertical or horizontal polarization.

1.2.2. Corner Reflector In corner reflectors, incoming signal is reflected back in the direction in which it came from. This type of antenna is used to better collimate the energy in the forward direction and is obtained by changing the geometry of the plane reflector so at to prevent radiation in the back and side direction. A combination of two plane reflectors can result in a corner reflector as shown in Figure 1.7 (b). If a reflector is used as a passive target for radar or communication applications then it returns the signal exactly in the same direction as it received. When its included angle is 90° and due to this unique feature military ships and vehicles is designed with minimum sharp corners to reduce detection by enemy radar. These are also used popularly as receiving elements for home television [2]. In most practical applications of corner reflectors the included angle formed by the plates is usually 90°, however other angles are also used for other specific applications. The feed element used for a corner reflector is mostly a dipole or an array of collinear dipoles placed parallel to the vertex at specific distance away and greater bandwidths achieved have been reported when feed elements are cylindrical or biconical dipoles instead of thin wires.

1.2.3 Parabolic Reflector Parabolic reflector antenna is the most well-known among the reflector antenna types and is commonly called as satellite dish antenna. Parabolic reflectors typically have very high gain values of around 35-40dBi, lower cross polarization and also have a reasonable bandwidth on commercially available antennas. Parabolic reflectors can be very wideband in the case of huge dishes like the Stanford "big dish", which can operate from 150 MHz to 1.5 GHz, as shown in Figure 1.8 [16], in which a front fed parabolic reflector is used. Since the transmitter (receiver) is placed at the focal point of the parabola, the configuration is usually known as front fed. Basic structure of the front fed parabolic reflector antenna is shown in Figure 1.7 (c) which consists of a feed antenna pointed towards a parabolic reflector. The feed antenna is often a horn antenna with a circular aperture. Figure 1.7 (d) shows a Cassegrain reflector antenna in which a sub reflector is being used to act like a beam diverter to the main parabola. This can have more gain compared to the front fed parabola systems but the higher cross-polarization factor arises in this geometry. Reflector antennas can be modeled

Figure 1.8. Front fed parabolic dish antenna at Stanford University. Source: EM Wave Inc. (2007). Choke Ring Antenna [Online]. Available: http://emwaveinc.net/ant_chokering.html

numerically in a number of ways and the various methods helps in reducing the time required in solving a particular design or in improving the performance of a design [17]. A few methods which can be considered for numerical modeling are geometric optics method; aperture method; geometric theory method of diffraction; physical optics method, integral equations method, finite elements method. Each of these methods solves the Maxwell equations in a unique way and determines the best method suitable for the antenna designed. In a number of papers different approaches are used for simplification of analytical expressions for calculation of antenna fields to deduce a mathematical model of antenna and to simplify modeling problem. In the design of reflectors performed in this thesis, we have used the FEKO tool which is a product of electromagnetic simulation software and is a Method of moments (MoM) based solver [4]. This tool has been selected due to its ability to solve large structures such as a reflector in a more efficient way and produce the necessary results in the desired method. Now we discuss front fed parabola, offset parabola and Cassegrain reflector systems in the following sections.

1.2.3.1 FRONT FED PARABOLIC REFLECTOR The reflector diameter can be on order of 100 wavelengths for very high gain reflectors (>50 dBi gain). Distance between the feed antenna and reflector is typically several wavelengths as well. This is in contrast to the corner reflector, where antenna is roughly a half-wavelength from the reflector. Relation between the focal length, f and angle subtended by feed towards the reflector θ0 and diameter of the reflector is given by relation stated in Equation. 1.4 [1]. ∗cot (1.4) For a parabolic reflector antenna, f/D ratio is an important parameter to be considered. The typical values of the f/D ratio range from 0.4-1.0. Design performed in this thesis uses a reflector of diameter =1.2m and an f/D of 0.5.The f/D ratio is chosen to be 0.5 based on investigations performed on other ratio's and this is more of an optimum value for a parabolic reflector -feedhorn geometry [2], [18]. Fields across the aperture of the parabolic reflector are responsible for this antenna's radiation. Maximum possible gain of the antenna can be expressed in terms of physical area of aperture which is shown in Equation. 1.5 [1]. (1.5) Actual gain is in terms of effective aperture, which is related to physical area by the efficiency term ( ). This efficiency term will often be on the order of 0.6-0.7 for a well- designed reflector antenna. Gain is defined by Equation. 1.6 [1]. (1.6) These are the details necessary while dealing with design of a parabolic front fed reflector antenna.

1.2.3.2 OFFSET REFLECTOR ANTENNA Offset reflector antenna is a type of parabolic reflector antenna wherein the antenna feed is offset to side of the reflector unlike in an axially fed reflector. Here the feed is located at the focal point of the reflector, but the reflector is an asymmetric segment of a paraboloid, and hence the focus is located on the side of the reflector. The offset reflector antenna designs are more popular for commercial applications mainly due to higher gains obtained

13 when compared to axially fed parabolic reflector antenna. A simple offset fed reflector antenna is as shown in Figure 1.9 [19].

Figure 1.9. General configuration of an Offset fed reflector antenna. Source: D. M. Pozar, Microwave Eng., 3rd ed. Hoboken: John Wiley & Sons, 2012.

Main motive of this design is to avoid blockage due to feed structure and to avoid support structures necessary in front of the parabola which reduces overall efficiency of the system. In an ordinary axial-fed reflector antenna, feed structure and its supports are located in the path of the beam of radio waves, partially obstructing them but in case of offset design, feed is positioned outside the area of main beam. Common application of offset reflector is in satellite communications and radar applications. Offset reflector antenna design produces a higher gain since the efficiency is also higher, and it also has an additional improvement of reduced noise temperature when compared to axial fed parabolic reflectors. Hence here the G/T ratio is higher which implies a higher signal to noise ratio which is very much desired of most antennas [18] and [20]. In an offset reflector antenna, there is a high isolation between the primary feed and the main reflector which is very desirable in many satellite communication related applications. The problem of inherent high cross-polarization of an offset configuration puts a major limitation of its use mainly in applications related to microwave radiometers where high beam efficiency is essential as well as in modern satellite communication where high cross-polarization can create cross-talk between the channels [21] A novel dual mode conical horn proposed by Potter [22] utilizes the multimode concept where in exciting the horn at throat regions produced the dominant TE11 mode and the higher

order TM11 mode. This when excited with appropriate amplitude and phase helps in the side lobe suppression and beam width equalization. This concept was then widely used for various satellites and radar application formed the start of inducing higher order modes in feedhorn designs. The idea of using this triple mode feedhorn for offset reflectors is aimed towards reducing the cross-polarization for the design which by research found to be possible by using dual mode potter horns or by inducing higher order modes with appropriate amplitude and phase distribution in conical horns. Literature from [23] describes a triple mode circular waveguide feedhorn with corrugated chokes capable of providing radiation o o pattern beam scanning in both φ=0 , 90 cut planes which has the TE11,TM01 and TE21 mode being excited with suitable amplitude and phase. Again [24] points at the study of a dual mode feedhorn having TE and TM modes being excited to generate different reflector illuminations and causing displacement of its phase center. This reflector-feed assembly, with its dual phase center capability, was developed for improving the performance of the ground moving target indicator (GMTI) radars; this concept has been used in our design to determine the displaced phase center for triple mode feedhorn and can be used for similar application. Further analysis can be seen in [25] which exhaustively deal with the determination of multiple phase center locations of a multimode horn with offset reflector antenna geometry, these feed horns with reflector assembly are suitable for radar systems, where ground moving target indicator (GMTI) mode implementation is desired. From [26] a feedhorn used to feed a reflector in offset configuration is analyzed which further provides multiple phase center which is very much similar to the work performed here to use a triple mode feedhorn to feed a reflector in offset configuration. Due to the complexity in analyzing the phase center locations, the phase center for the triple mode feedhorn is analyzed but could possibly be further extended to offset reflector configurations based on the phase center calculations. Calculations for the exact phase center location for triple mode feedhorns is not possible but a more general method, valid for small as well as large phase error (Δφ) values is described in [27] for the optimum phase center location of feed antennas used in prime- focus or dual reflector systems. In the present research work, an offset fed reflector antenna has been designed using CADFEKO for a reflector having diameter, D =1.2m and f/D = 0.5. The feed is placed at an tilt angle of 27.75° which has been determined, for achieving maximum gain value. A

detailed discussion about the design and implementation of an offset reflector has been discussed in Chapter 5.

1.2.3.3 CASSEGRAIN REFLECTOR ANTENNA Cassegrain antenna is an advancement of parabolic type of antenna where feed source is placed at or behind the surface of concave main parabolic reflector. This is aimed at a smaller convex/concave secondary sub reflector which is suspended in front of the primary reflector either in a symmetric way or an offset design based on the application needed. The collimated beam from the feed source illuminates the sub reflector, which reflects it back to the main reflector and this in turn reflects it forward again to form the desired beam. This type of reflector configuration is used to improve performance of large ground-based microwave reflector antennas for satellite tracking and communication [1]. A general geometry of the Cassegrain antenna system is shown in Figure 1.7 (d) [28]. For a Cassegrain system to achieve better collimation characteristics, the main reflector is generally larger but must be a paraboloid. The secondary reflector which is the smaller reflector is usually a hyperboloid [3]. Main advantage of Cassegrain reflectors when compared to other reflector geometries are its ability to realign the feed in any convenient location according to an application, reduction in its spillover, in the minor lobe radiation, lowering the antenna noise temperature and its capability in scanning and beam broadening by adjusting one of it reflecting surfaces. This comes in use for majority of satellite applications, radio telescopes and in defense related applications [1]-[4]. A major disadvantage of Cassegrain reflector system is that the feedhorn must have a narrower beamwidth and higher gain to focus its radiation on the smaller secondary reflector, instead of the wider primary reflector as in case of the front-fed reflectors. Hence the feedhorn must have a larger aperture for a given wavelength [5]. A Cassegrain antenna which is being used in Sweden is shown in Figure 1.10 [29].

1.3 RECONFIGURABILITY IN SPIRAL LOADED PLANAR DIPOLE ANTENNA A dipole antenna is a simple wire like antenna which is fed at the center of its geometry. A dipole antenna is widely used in many commercial applications. It mainly consists of two metal rods or wires which acts as the conducting medium and are placed

Figure 1.10. A Cassegrain system being used in Sweden. Source: Wikipedia. (2012, May 20). Feed Horn [Online]. Available: http://en.wikipedia.org/wiki/Feed_horn collinear to each other with a small spacing placed between them [29]. It finds several wireless communication applications and most commonly as television antennas and also as driven elements in Yagi-Uda antennas. Dipole antennas can be realized using finite diameter wire or using microstrip technology so it is a planar structure. Our research work in this design is to mainly study a planar dipole antenna, make it compact by adding spiral sections to it and to make the design frequency reconfigurable. A planar dipole antenna is combined with a loop having multi-band response in [30]. To understand as to how the spirals are made on a dipole and how then can be investigated we refer to [31] in which a spiral-dipole antenna with CP patterns with potential MIMO applications is proposed. This thesis work is intended to analyze the spiral loaded dipole antenna and to make it reconfigurable by proper tuning of its λ/4 sections according to the various frequency bands intended. Frequency reconfigurable half wavelength dipole antenna having lumped components to switch between the different communication bands is presented in [32]. In [33], a simple planar dipole antenna with four cantilever RF MEMS switches to reconfigure across various X-band

frequencies is presented. Similarly more research related to our design was performed in [34] where an antenna configuration that switches between dipole, slot and hybrid modes for frequency reconfiguration is presented. Presented here is a novel spiral loaded planar dipole antenna with frequency reconfiguration at760 MHz, 1.47 GHz and 2.2 GHz bands obtained by using PIN diodes as switches. First presented is a planar half wavelength dipole antenna which is designed to operate in the LTE 700MHz band. This design is then remodeled to make it more compact, by introducing spirals in the dipole ends. A frequency reconfigurable spiral loaded planar dipole antenna is presented which is designed by employing PIN diodes. Appropriate biasing circuit is modeled using the data sheet and the PIN diode values which are selected after suitable calculations on its circuit design. The full wave analysis has been performed using finite element method (FEM) based commercial tool Ansoft High Frequency Structure Simulator (HFSS) v.12 [35].

1.4 SIMULATION TECHNIQUES AND EXPERIMENTAL VERIFICATION METHODS The thesis work performed here is done using 3 tools: EMSS based FEKO, Ansoft HFSS and TICRA based GRASP tool. The method of moment (MoM) based FEKO tool is extensively used to design the triple mode feedhorn, symmetric and offset fed reflector antenna designs. FEKO tool is used because it is the right tool to solve electrically large objects with higher meshing [4]. FEKO has the ability to solve the models using different schemes like the Physical optics (PO), Uniform theory of diffraction (UTD), Geometric Theory of Diffraction (GTD), MLFMM and other methods. The advantage of this is, it makes FEKO a more efficient way of solving Maxwell's equations and is found to reduce the simulation time required for solving larger structures. Results obtained from FEKO are analyzed using the POSTFEKO option and here we can obtain the S-parameters, the radiation patterns as well as the current distributions necessary for our design. The next tool used in this design is the Ansoft HFSS v12 which is a finite element method (FEM) based 3-D full wave analysis software [35]. The unequal Wilkinson power divider design and the reconfigurable spiral loaded planar dipole antenna designs are simulated, analyzed and all the far field radiation patterns, S- parameters; current distributions have been analyzed using this tool. HFSS is more accurate for solving structure

18 which are smaller and require efficient and more accurate meshing and is thus used for these designs. The third tool used in our designs is the TICRA's student version GRASP v.9 tool which is a well renowned software tool since a couple of decades. This tool is mainly used for the analysis of various reflector antenna designs and is found to be pretty accurate. Analysis for the triple mode feedhorn has not been performed using this tool since GRASP version is trial version and it cannot be used for multimode antenna designs. Similar to FEKO this tool also provides a number of numerical methods to solve different geometrical structure based on their needs. More results obtained from the respective tools have been discussed in detail in the following chapters.

1.5 ORGANIZATION OF THESIS Thesis work performed here is broadly categorized into the following chapters. Chapter 1 presented here gives a brief introduction about the various topics being covered in this thesis. Chapter 2 deals with the design of an offset reflector antenna simulated using TICRA's GRASP software. Next section in this chapter explains the design of a single mode feedhorn using CADFEKO which is used as a feed for a symmetric fed reflector and various results obtained have been discussed. Chapter 3 presents the design of a triple mode feedhorn operating from 7-8GHz, its various mode combinations have been analyzed and the effects of phase and amplitude variation on the patterns are discussed. An unequal Wilkinson power divider has been designed (simulated and fabricated) to act as an feed network to the triple mode feedhorn. Chapter 4 presents redesign of a triple mode feedhorn, explored earlier by A. Tuteja [36] acting as feeding element for an axially placed reflector. The gain radiation patterns obtained using POSTFEKO tool is analyzed and effect of amplitude variation of the mode values on the design is discussed. Chapter 5 presents a design of an offset fed reflector antenna modeled in CADFEKO using the triple mode feedhorn. The various gain radiation patterns are analyzed again similar to the symmetric feedhorn and values are compared. Chapter 6 deals with design of a reconfigurable spiral loaded planar dipole antenna simulated using Ansoft HFSS and the fabricated model results have been presented here. Reconfigurability has been tested using PIN diodes and the fabricated model results for the various bands are compared with the simulated results. Chapter 7 concludes the research work performed here and also provides future study.

CHAPTER 2

DOMINANT MODE FEEDHORN REFLECTOR SIMULATIONS USING GRASP AND FEKO

2.1 INTRODUCTION The motivation behind designing a reflector feedhorn model in GRASP and CADFEKO tools are to understand the basic reflector-feedhorn design criteria, and to compare the results obtained. This is done so that, FEKO tool can be verified against a well known reflector antenna analysis tool GRASP, therefore, FEKO can be used for the entire analysis in this thesis.

2.2 TICRA GRASP SIMULATIONS First design presented here is an offset fed reflector antenna using student version GRASP tool v.9 .In this design, a reflector of diameter = 2.4 m which is operating at the Ku band frequency (f = 12 GHz) and having a dominant mode circular waveguide feed is shown. Aperture radius is found to be 9.75 mm and distance between main reflector and parabola axis is 1.2 m. This model is then simulated using GRASP tool and far field patterns obtained were analyzed. The peak co-polarization gain is found to be 46.68dBi and the cross- polarization gain is 29dBi making the peak cross-polarization level to be -17dBi. Next, an analysis for the reflector design based on the different f/D ratios is performed. Peak co- polarization and cross polarization gains as well as their separation for various f/D ratios are analyzed and are tabulated in Table 2.1. This analysis is done for both fundamental mode circular waveguide as well as the Gaussian beam mode as shown in Table 2.2. Based on the values observed in Table 2.1 and 2.2, the peak co-polarization gain for Gaussian beam mode analysis is much higher when compared to the fundamental mode circular wave guide analysis. The cross-polarization gain increases in Gaussian beam mode design but does not affect the design to a large extent and is hence found to be more suitable. For applications desiring lower cross-polarization values the fundamental mode circular

Table 2.1. Fundamental Mode Circular Waveguide Analysis for Variable f/D f/D Focal Peak Co- Peak Cross Peak Cross Length ,f Polarization Polarization Polarization Level (meters) Gain (dBi) (dBi) (dB) 1 2.4 43.719 16 -28 0.8 1.92 44.98 21 -24 0.6 1.44 46.18 26 -20

0.5 1.2 46.68 29 -17

0.3 0.72 46.78 37 -10

Table 2.2. Gaussian Beam Mode Analysis for Variable f/D f/D Focal Peak Co- Peak Cross Peak Cross Length ,f Polarization Polarization Polarization Level (meters) Gain (dBi) (dBi) (dB)

1 2.4 48.82 19 -29

0.8 1.92 48.603 23 -26

0.6 1.44 48.46 27 -21

0.5 1.2 48.407 31 -17

0.3 0.72 47.78 35 -13 waveguide analysis is found to be more preferred. Resultant far fields plots obtained for both the analysis are as shown in the following Figures 2.1 – 2.5. Various gains for the Gaussian beam mode as well as for fundamental mode circular waveguide are analyzed. The corresponding results for various f/D ratios are shown in Figure 2.1 - 2.5 with corresponding values as mentioned by Table 2.1 and 2.2 respectively. Next step is to use 3D simulation tool FEKO which is capable for designs involving electrically large objects. FEKO has various numerical modeling methods to solve the Maxwell's equations like Physical Optics (PO), Geometrical Theory of Diffraction (GTD), Uniform Theory of Diffraction (UTD), and Multilevel Fast Multipole Method (MLFMM) which help to obtain results in a much more efficient way. FEKO tool provides the gain patterns,

Figure 2.1. GRASP based Result for f/D =1, Diameter = 2.4 m, f = 2.4 m.

Figure 2.2. GRASP based Result for f/D = 0.8, Diameter = 2.4 m, f = 1.92m.

Figure 2.3. GRASP based Result for f/D = 0.6, Diameter = 2.4 m, f = 1.44 m.

Figure 2.4. GRASP based Result for f/D = 0.5, Diameter = 2.4 m, f = 1.2 m.

Figure 2.5. GRASP based Results for f/D = 0.3, Diameter = 2.4 m, f = 0.72m.

S-parameters, impedances and current distribution. Next section deals with designing a cylindrical feedhorn using FEKO tool. This at a later stage is used as a feed source for a reflector the results of which are discussed in Section 2.4.

2.3 DESIGN AND ANALYSIS OF A CYLINDRICAL FEEDHORN A cylindrical feedhorn operating at 12.5 GHz was initially designed. After suitable meshing criteria is given and the far field patterns have been requested, the gain patterns, current distribution and the return loss has been observed [4]. This feedhorn is used to feed an electrically large reflector antenna of 36λ diameter. The results comprising of return loss and far field gain patterns have been analyzed. The designed cylindrical feedhorn is analyzed over a frequency range of 1013GHz to observe the variation of the matching levels in this band and also to have a comparison over the co-polarization and cross-polarization gains obtained from the far field radiation patterns. The various design parameters along with their variable names used in this modeling is as shown below:  freq = 12.5e9 (The operating frequency.)

 lam = c0/freq (Free space wavelength.)  lam w = 0.0293 (The guide wavelength.)  h_a = 0.51*lam (The waveguide radius.)  h_b0 = 0.65*lam (Flare base radius.)  h_b = lam (Flare top radius.)  h_l = 3.05*lam (Flare length.)  ph centre = -2.6821e-3 (Horn phase centre.)  R = 18*lam (Reflector radius.)  F = 25*lam (Reflector focal length.)  w l = 2*lam w (The waveguide length.) After initializing CADFEKO with the above mentioned variable names and creating the model we initialize all variable values, specify frequency range and assign meshing necessary for this model. Now demonstrating the feedhorn designed using CADFEKO showing the model generated along with its POSTFEKO model and the 3D radiation patterns observed. The feedhorn model generated using CADFEKO is as shown in Figure 2.6 with its feed excitation pin placed at the bottom face of the cylinder section of the feedhorn as shown. Figure 2.7 shows the CADFEKO model involving meshing, feed pin and the POSTFEKO model of the same.

Figure 2.6. The conical feedhorn model generated in CADFEKO tool.

(a) (b) Figure 2.7. (a) The meshing and excitation shown in CADFEKO model. (b) POSTFEKO model with meshing.

CADFEKO model shown above in Figure 2.6 and Figure 2.7 are simulated. The results are observed using POSTFEKO tool. 2-D radiation patterns are shown in Figure 2.8. The radiation pattern (Figure 2.8) shows a co-polarization gain of 14.4dBi and cross- polarization gain of -54.3dBi making the peak cross-polarization separation level to be 68.7dB. This is considered a reasonably good value for a feedhorn. After observing the 2-D patterns, the 3-D patterns of the designed conical feedhorn are analyzed. Co-polarization 3-D gain radiation pattern is observed in Figure 2.9. Gain total scale shown in the corner of the Figure 2.9 clearly shows the maximum gain value of the feedhorn to be occurring at θ = 0° and φ= 0° which is the broadside gain of our design. Side lobes present in radiation pattern of the feedhorn is observed to be very narrow which shows that the designed feedhorn has optimum gain values. Feedhorn is evaluated at different frequency points to observe the variation in terms of its gain as shown in Figure 2.10. Co-polarization gain pattern is mainly depicted to illustrate the co-polarizations gain variation over the entire frequency range. Gain values obtained are shown in Table 2.3. As shown in Figure 2.10, variation of the peak co polarization gain obtained at the various frequency points is depicted. Co-polarization gain is observed to be increasing with the frequency values and gain maximum was found to be 14.95dBi which is observed to be at 14 GHz from Figure 2.10. The gain at other frequencies is mentioned in Table 2.3.

Figure 2.8. Gain radiation patterns showing the co-polarization and cross-polarization patterns for the conical feedhorn operating at 12.5 GHz.

Conical feedhorn is found to be operational at the designed frequency and functional over a wide bandwidth of frequencies. Gain values obtained here indicate a satisfactory functioning of the designed feedhorn. On observing the patterns, current distribution, and its operability over a wide band range makes this feedhorn suitable as a conventional feed source for reflector.

2.4 CONICAL FEEDHORN SYMMETRICALLY FEEDING A LARGE REFLECTOR Reflector when illuminated with a feed source can collimate the rays falling on its surface to a beam of very narrow beamwidth and of very high gain in a desired direction. The direction in which it radiates the beam depends on whether the feed source is placed symmetrically or in an offset position. Cylindrical feedhorn designed in earlier section is used as a feed source to illuminate an electrically large reflector of approximately 36λ in a

Figure 2.9. 3-D radiation patterns of conical feedhorn at 12.5GHz. symmetric fashion. For designing the feedhorn with reflector design in CADFEKO, the reflector values are used in the design equations and with meshing for the reflector as λ/6 and that of feedhorn being λ/15 so as to observe improved results from the design. Model geometry designed in FEKO is as shown in Figure 2.11 and the CADFEKO and POSTFEKO models of the geometry are shown in Figure 2.12. Total gain radiation pattern of the conical feedhorn symmetrically feeding the reflector is shown in Figure 2.13. This is the co-polarization radiation pattern showing a maximum gain of 40.12 dBi.

Figure 2.10. Co-polarization radiation pattern of Gain vs theta over a range of frequencies.

Table 2.3. The Co-Polarization Gain Values Vs Frequency for Conical Feedhorn. Frequency [GHz] Gain[dBi] 10 13.41 11 13.94 12 14.3 13.5 14.8 14 14.95

Far field 3-D radiation pattern of symmetric feedhorn reflector design is shown in Figure 2.14. Peak co-polarization gain is observed to be in a very narrow beamwidth angle at the end of the radiation pattern, the highest value of the gain occurring at gain θ=0, φ = 0 is 40.12dBi. Sidelobe level in Figure 2.14 is also observed to be pretty low which is desirable

Figure 2.11 CADFEKO model of conical feedhorn symmetrically feeding a large reflector.

Figure 2.12 CADFEKO and POSTFEKO models of conical feedhorn symmetrically feeding a large reflector.

Figure 2.13. Co-polarization gain radiation pattern for the feedhorn-reflector combination.

Figure 2.14. 3-D radiation pattern of the feedhorn reflector configuration.

for reflector feedhorn geometry and makes it more suitable for satellite communication purpose as well as other radar applications. Analysis is now needed whether this designed geometry for a symmetric feedhorn and reflector is operational over different frequency points. It is also necessary to observe how its gain varies over the frequency band of 10-14 GHz which is the designed band for the cylindrical feedhorn. This frequency range is selected as it is needed to observe the variation of the feedhorn reflector geometry values in the Ku band spectrum. CADFEKO model is loaded with these frequency points and the design is simulated. Again the co-polarization radiation patterns have been generated in POSTFEKO and their values tabulated. Co- polarization gain values obtained at different frequency points are as shown in Table 2.4.

Table 2.4. The Co-Polarization Gain Values Vs Frequency for Conical Feedhorn- Reflector Configuration. Frequency[GHz] Gain[dBi]

10 37.1

11 38.05 12 39.71 13.5 39.2

14 38.94

From Table 2.4 observed is the variation of the co-polarization gain over a frequency band 10-14GHz for the designed symmetric feedhorn-reflector configuration with pretty good peak gain which is desirable of this configuration. The maximum gain is found to be at 12.5GHz which is our design frequency.

2.5 CONCLUSION A feedhorn reflector configuration operating in the Ku band at 12 GHz is analyzed using GRASP tool. This design is done to mainly describe the modeling of an offset reflector combination, to study its gain patterns and also to perform a comparative study by varying f/D ratios and in turn to study its effect on gain values obtained. This study is initially performed using TICRA's GRASP software tool. A cylindrical feedhorn symmetrically

32 feeding a large reflector antenna is designed using EMSS's CADFEKO tool. This cylindrical feedhorn when being subjected to act as a feeding element to a large reflector antenna of 2.4m diameter, we observe that the feedhorn reflector system is having again a good radiation pattern with a high co-polarization gain value of 40dBi. The results also show that the designed model is operational over a wide bandwidth of Ku band from 10GHz-14GHz. The designed symmetric and offset configurations serves as a base for the study being done in this thesis work on the triple mode feedhorn reflector antenna analysis in symmetric and offset configurations. A triple mode feedhorn designed earlier [36] is being worked up in this thesis using a software tool EMSS's FEKO which is explained in details along with all the results obtained in Chapter 3.

CHAPTER 3

TRIPLE MODE FEEDHORN PERFORMANCE AND PHASE CENTER DETERMINATION

3.1 INTRODUCTION A microwave feedhorn antenna is an antenna that consists of a flaring metal waveguide like a horn to direct or transmit the radio waves. Feedhorns are very widely used as antennas at UHF and microwave frequencies and they can operate over a wide range of bandwidth. Useable bandwidth of horn antennas is typically of the order of 10:1, and can be up to 20:1 (for example allowing it to operate from 1 GHz to 20 GHz) [37]. Considered here is the design which mainly consists of a feedhorn which is conical in shape and dominant

mode of the feedhorn being TE11 mode. Main reason for selecting feedhorn’s as feeding elements to reflector and other antennas is due to the fact that they provide a higher co- polarization gain, good beamwidth, good efficiency, low SWR and a simple construction with respect to other antenna models. A horn antenna when excited with appropriate waveguide diameters can generate other higher order modes along with the dominant mode. This concept is used in this feedhorn design. By using appropriate diameters of conical sections with suitable diameters higher order modes could be generated. Feedhorn considered here is not just a conical feedhorn but in which the two other modes are being excited along with the dominant mode to make this a triple mode feedhorn configuration. The modes are: TE11, TM01, and TE21 respectively [23]. This triple mode horn is being redesigned from the work by Ashish Tuteja [36]. Main advantage of this type of multimode feedhorn is its use in providing displaced phase center positions. It has been shown in [24]-[27] that a multimode feedhorn offers multiple phase center positions. Pattern generated in this multimode feedhorn will be having a broadside nature due to the dominant mode and the difference mode patters will be generated due to the higher order modes. Most important aspect of this feedhorn is its ability to have beam steering possible just by controlling power to the different excited modes and

by having a fixed phase difference between the modes. This application of the designed feedhorn is explained in more detail in this thesis work.

3.2 DESIGN METHODOLOGY A triple mode feedhorn is basically a conical feedhorn which supports higher order modes and includes corrugations in the form of choke rings at the aperture of the horn. The cylindrical sections of waveguides with appropriate diameters for the modes are in accordance with the Bessel function values [38]. The designed feedhorn has various sections and each of which has a specific diameter and operates at a particular frequency range. HFSS design of the multimode feedhorn is shown in Figure 3.1.

HFSS Model FEKO Model Figure 3.1. Triple mode feedhorn design model in HFSS and FEKO software.

The triple mode feedhorn generated is shown in Figure 3.1 where in the top most section of cylinder is the place where TE21 modes are excited having two rectangular waveguide ports on either side oriented in E-plane, and similarly lower 2 sections exciting

TM01 mode and TE11 mode, respectively. At the aperture of the horn the choke rings are placed which are concentric in shape . Main reason for adding these choke rings at the

aperture is to increase co-polarization gain of dominant TE11 mode and also to reduce the -

10dB half edge beamwidth and thus increasing directivity. Modes TM01 and TE21 ports are orthogonally placed on the cylindrical waveguide. The model created in FEKO is subjected to even subtracting of the unwanted parts which are the top faces of each choke ring so that the waves can evenly propagate throughout the horn structure. The far field requests were performed along with the S-parameters request so that these results could later be viewed once model has been simulated. The meshing criterion being suitably given as λ/10 for the feedhorn and λ/25 for the waveguide p as we need finer meshing for the waveguide ports else we would be observing errors while solving the model.. The HFSS and FEKO models of the feedhorn are as shown in Figure 3.1: Various modes and their combinations from their 5 ports being analyzed are:

 TE11

 TM01

 TE21

 TE11+TM01 with ±90° Phase

 TE11+TE21 with ±90° Phase

 TE11+TM01+TE21 with ±90° Phase

3.3 SIMULATED RESULTS AND PERFORMANCE A Triple mode feedhorn designed earlier using HFSS [1] is reanalysed using

CADFEKO. Feedhorn excites the modes TE11, TM01 and TE21 modes which is then analysed to understand its various modal properties. Simulation model of the triple mode feedhorn generated in CADFEKO tool is as shown in Figure 3.2. The horn has two chokes at the aperture and three circular waveguides with the total length of horn is 24cm.The inner diameter of the outer choke is 7.2cm and outer diameter being 7.6cm. This CADFEKO model with all the necessary preconditions and meshing has been simulated. Results of designed triple mode feedhorn can be viewed in POSTFEKO tool and l is as shown in Figure 3.3.

Figure 3.2. CADFEKO generated model of triple mode feedhorn along with dimensions.

Figure 3.3. POSTFEKO generated model of the triple mode feedhorn shown with meshing.

Reflection co-efficient magnitude for triple mode feedhorn design is as shown in

Figure 3.4. For S11 on the first port exciting TE11 mode, we observe a matching bandwidth better than, S11 ≤ -10dB over the range from 7.43 GHz - 7.94 GHz which is main band of concern. The reflection co-efficient of the other ports i.e. S22, S33 is from 7.46 - 8.34 GHz &

S44, S55 is from 7.3 - 8.05 GHz and this shows that our designed model has good matching over the entire desired frequency range.

Figure 3.4. Reflection co-efficient magnitude Vs frequency plot for designed triple mode feedhorn.

Thus the designed triple mode feedhorn has relatively good matching levels with respect to all the individual modes. The S11 is found to correspond to TE11 mode, S22 and S33 corresponding to TM01 mode and S44, S55 corresponding to TE21 mode.

3.3.1 Individual Mode Analysis of the Triple Mode Feedhorn

Individual mode gain radiation patterns for the TE11, TM01 and TE21 modes at the center frequency of the common band at 7.73GHz consisting of co-polarization and cross- polarization curves are shown in Figure 3.5 (a-c), respectively. For TE11 mode radiation pattern shown in Figure 3.5 (a), the co-polarization gain lies at the broadside angle for both cuts: Eθ for φ=0° plane and Eφ for φ=90° planes and the cross polarization lies at Eθ for

φ=90° plane and Eφ for φ=0°. For TM01 mode radiation pattern shown in Figure 3.5 (b), a

Figure 3.5. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11 Mode (b) TM01 Mode and TE21 Mode at 7.73 GHz.

(a)

(b)

(c)

null in the radiation pattern is observed in the Eθ plane with the co-polarizations gains

observed in Eθ for φ=0° cut plane and Eθ for φ=90° cut plane. For the TE21 mode shown in

Figure 3.5 (c), we observe a null in the radiation pattern occurring in the Eφ plane and the co-

polarization gains in Eφ for φ=0° cut plane and Eφ for φ=90° cut planes. Various peak co- and cross-polarization values for the individual modes at 7.73 GHz is as shown in Table 3.1.

Table 3.1. Peak Co- and Cross-Polarization Gain Values at 7.73 GHz for Triple Mode Feedhorn with Equal Amplitude Peak Co- Peak Cross- Peak cross-

Mode polarization polarization polarization Gain [dBi] Gain[dBi] level [dB]

TE11 10.12 -35.42 -45.54

TM01 8.43 -15.85 -24.28

TE21 8.42 -38.82 -47.24

Gain values being shown in Table 3.1 depicts co and cross-polarization values for all the individual modes. The gain was maximum for the TE11 mode which is dominant mode for the design. The cross-polarization levels for the various modes shown can also be observed from the Figure 3.5. Cross-polarization values depicted in Table 3.1 shows that the

best obtainable low cross-polarization is that for the TE21 mode. For the TM01 mode patterns,

we observe the null of the radiation pattern to be occurring in the Eθ and it has been found to be occurring in this plane for a variety of different frequencies and is a very much desired factor. We now proceed with the dual mode analysis performed on the triple mode feedhorn.

3.3.2 Dual Mode Analysis for Various Mode Combinations

This section discusses the need for combining the TE11 mode with either TM01 mode

or the TE21 mode and shows the effects observed on varying the power and the phase given to the ports. Main intention of having dual mode combination is to show that beam can be made to scan in a fixed range from -24° to 24°. Beam scan being mentioned here is able to

scan in both Eθ φ=0° and the Eφ φ=90° cut planes but is able to scan them one at a time.

Dual mode analysis of triple feedhorn for the mode combination of TE11+TM01 with ±90° phase applied between modes is shown in Figure 3.6. All analysis shown has been performed at 7.73 GHz which is the design frequency. Beam scanning angle for the +90° case shown in Figure 3.6 (a) is found to be 11° with maximum co-polarization gain of 11.4dBi. Similarly for the -90° case shown in Figure 3.6 (b) we observe the maximum scan angle to be around -11° with the maximum co-polarization gain value being maintained around the same value. Various co-polarization and cross- polarization gain values along with the peak cross-polarization levels are shown in Table 3.2.

Coming to the next mode combination of TE11+TE21 with ±90° phase being supplied across modes, we observe radiation patterns as shown in Figure 3.7 (a) and (b), respectively. For first case in Figure 3.7 (a) we observe the beam scanning to be occurring at an angle of +9.7° with peak co-polarization gain being observed to be 11.01dBi. From Figure 3.7 (b) which is for the -90° phase being applied, we observe the beam scan angle to be -9.7° which is pretty much the same as for the +90° case and the peak co-polarization gain has been observed to be 11dBi. All the necessary co-polarization and cross-polarization values are specified in Table 3.2.

3.3.3 Triple Mode Analysis for Various Mode Combinations Triple mode feedhorn is now analyzed for its tri-mode combinations which is very

necessary condition for feed to scan a beam in both Eθ for φ=0° and Eφ for φ=90° planes. All modes specified are being excited with varying amplitude and with ±90° phase difference

being supplied to the TM01 and TE21 mode ports.

First radiation patterns demonstrated here are that of the TE11+TM01+TE21 mode

combination with 20% power supplied to TE11 mode, 40% supplied to both TM01 and TE21 modes, respectively, which has been shown for both the ±90° phase values in Figure 3.8 (a) and (b), respectively. For the pattern shown in Figure 3.8 (a), we observe the scanning is occurring at +19.2° with a peak co-polarization gain of 9.8dBi. Similarly for Figure 3.8 (b) case, we observe the scanning to be occurring at -19.7° with the co-polarization gain of 9.89dBi. The above results have been summarized at 7.73GHz in Table 3.3.

(a)

(b) Figure 3.6. Co-polarization and cross-polarization gain radiation patterns for the feedhorn for (a) TE11+TM01 Mode with +90, (b) TE11+TM01 Mode with -90° at 7.73 GHz.

Table 3.2. A Study of the Dual Mode Analysis for the TE11+TM01 and TE11+TE21 Mode Combinations with ±90° Phase Assigned with 50% Power Being Assigned to Each Mode Peak Co- Peak Cross- Peak Cross-

Mode Combination polarization polarization Gain Polarization Gain[dBi] [dBi] level [dB]

TE11(50%)+TM01(50%) with +90° Phase 11.29 2.94 -8.35

TE11(50%)+TM01(50%) with -90° Phase 11.35 3.02 -8.33

TE11(50%)+TE21(50%) with +90° Phase 11.01 -0.75 -11.76

TE11(50%)+TE21(50%) with -90° Phase 11 -0.83 -11.83

Now we proceed towards performing the same simulations by varying the power supplied to various modes to observe beam scanning angle variation as well as peak co-

polarization gain. The TE11 mode with 40% of input power and the modes TM01, TE21 are given 30% power each for a phase difference of ±90° applied between the non-dominant modes. For the pattern shown in Figure 3.9 (a), we observe that the peak gain to be occurring at -2.13° with a peak co-polarization gain of 12.95dBi and for the case shown in Figure 3.9 (b) we see the -7.3° with the peak co-polarization gain observed to be 13.04dBi. In this case we are observing that peak co-polarization gain value has increased by a factor of almost

3dBi which is occurring as in this mode we have 40% of power being supplied to TE11 mode which is the dominant mode and helps in increasing peak gain values. We can thus observe that by proper variation of power being supplied to different ports and by proper phase difference applied, we can control the beam scanning angles and range of scanning which can be used for more specific applications.

3.4 PHASE CENTER DETERMINATION Phase center is defined as the point from which it appears that an antenna radiates spherical waves. Literature study shows that the phase center is a unique point but measurements have shown that the phase center is seldom a unique point in a plane and depends on the pattern angle. Multiple phase center determination in feedhorn and feed- reflector designs has been playing a very pivotal role for these antennas as this form the basis

Figure 3.7. Co-polarization and cross-polarization gain radiation patterns for the feedhorn for (a) TE11+TE21 Mode with +90°, (b) TE11+TE21 Mode with -90° at 7.73 GHz.

(a)

(b)

(a)

(b) Figure 3.8. Co and cross-polarization gain radiation patterns for the feedhorn (a) TE11(20%)+ TM01(40%)+TE21(40%) mode with +90° (b) TE11(20%)+ TM01(40%)+TE21(40%) mode with -90° at 7.73 GHz.

Table 3.3. A Comparison of the Various Gain Values for the Triple Mode Combination by Power Variation with ±90° Phase Applied between Modes and at 7.73GHz Peak Co- Peak Cross- Peak Cross-

Mode Combination polarization polarization Polarization Gain[dBi] Gain [dBi] level [dB]

TE11(20%)+TM01(40%)+TE21(40%) 9.8 5.58 -4.22 with +90° Phase

TE11(20%)+TM01(40%)+TE21(40%) 9.89 5.10 -4.79 with -90° Phase

TE11(40%)+TM01(30%)+TE21(30%) 12.95 -2.18 -15.13 with +90° Phase

TE11(40%)+TM01(30%)+TE21(30%) 13.04 -.09 -13.13 with -90° Phase

of using these feedhorns with the reflector antennas. Determination of phase centre location is an important parameter in feed horns, which helps to locate the feed relative to the focal point of the reflector for the optimum reflector performance [31]. Phase center of a feed is mainly placed at the focus of a parabolic reflector to minimize the reflector aperture phase error loss. The location of a phase center on the reflector aperture is mainly determined by the field distribution on the surface of aperture [15]. Earlier studies were performed on determining the multiple phase center position by using a dual mode feedhorn [24]. This proposed design mainly deals with the determination of a displaced phase center position of the triple mode feedhorn design and only for a particular mode combinations and it is analyzed by studying the far fields phase vs. theta variation for various mode combinations. Displaced phase center position can similarly be found for different mode combinations too. In Figure 3.10, the Far Field E_Vertical and E_Horizontal represent the

XZ and YZ cuts Co-Polarizations parameter for the TE11 (40%) + TM01 (30%) + TE21 (30%) mode combination with the +90° Phase and the E_vertical_1 and E_Horizontal_1 represent the XZ and YZ cuts Co-Polarizations parameter for the TE11 (40%) + TM01 (30%) + TE21 (30%) mode combination with the -90° Phase. A study was performed on the mode

(a)

(b) Figure 3.9. Co and cross-polarization gain radiation patterns for the feedhorn (a) TE11 (40%) + TM01 (30%) + TE21 (30%) mode with +90° (b) TE11 (40%) + TM01 (30%) + TE21 (30%) mode with -90° at 7.73 GHz.

Figure 3.10. Phase center representation for triple mode feedhorn.

combination of TE11 (40%) + TM01 (30%) + TE21 (30%) with +90° and -90° phase being applied between the modes. At the X=0 and Y=0 position of this mode combination, we observed the peak gain for the design to be occurring at ±11° for the +90° and -90° phases, respectively. Hence to determine the phase center location of the feedhorn, where in the Phi= 0 deg Co-pol and Phi = 90 deg Co-pol are parallel to each other, we have had several parametrics to determine this position. After much analysis, we determined its position to be at X = ±7.6mm and Y = -2mm as depicted in Figure 3.10. Thus, for the phase of +90° we observe it at X = -7.6mm and for the -90° we observe it to be at X = +7.6mm. The curves are shown to be flat in this region in Figure 3.10.

3.5 WILKINSON UNEQUAL 1:5 POWER DIVIDER DESIGN AND RESULTS Power dividers are a class of passive microwave components which are used for power division or power combining [5]. They are developed often for the equal power division (3dB) type, but the unequal types of power splitter ratios are also possible and are commonly developed using Wilkinson type dividers. Simplest type of power divider is a T- junction design which is a 3-port network that can be used for power division or combining. An equal split three port resistive power divider is as shown in Figure 3.11 [39].

Figure 3.11. Equal split resistive power divider. Source: Luxorion. (n.d.). The Radio Propagation [Online]. Available: http://www.astrosurf.com/luxorion/qsl- propa6.htm

Figure3.11 shows equivalent circuit for an equal power splitter, i.e., the power input at port 1 is equally divided into ports 2 and 3. For this research work the plan is to design an unequal power divider/splitter to divide the input power into 5 output ports. Main idea behind this design is to use the outputs obtained at the end of each ports as input amplitude values to be fed to the triple mode feedhorn designed in CADFEKO.  Calculations for the various widths of the power divider were mainly performed by using Tx line software and PCAAD tool and equations to verify the values.  Power divider was designed for a power split ratio of 0.4 at Port 2 and 0.15 each for Port 3, 4, 5 and 6 respectively.  Initial stage was designing a 1:2 unequal power divider in which we obtained a 0.4 power at Port 2 and 0.6 at Port 3. This input from Port 3 is taken as input to an equal 1:4 Wilkinson power divider which will split the 0.6 power into 2 equal arms having 0.3 as output of each port. This 0.3 from each port is again split into 2 more Wilkinson equal power dividers to produce the 0.15 value at each of the four ports. This forms the basis of our power divider design. An unequal Wilkinson power divider is as shown in Figure 3.11. Impedances Z02, Z03 and resistance R, R1, R2 and K can be calculated from the formula stated in Figure 3.12 [39] The input power being supplied is split into 0.4W at port 2 and 0.6 at port 3 and the impedances here have been determined by calculations. The output of port 3 is now being fed as an input to a 1:4 Wilkinson equal power divider to deliver 0.15W to each output arm. A similar model of the Wilkinson equal power divider is as show in Figure 3.13.

Figure 3.12. A Wilkinson power divider in microstrip form with unequal power division. Source: Luxorion. (n.d.). The Radio Propagation [Online]. Available: http://www.astrosurf.com/luxorion/qsl-propa6.htm

Figure 3.13. A 1:4 Wilkinson equal power divider.

53 From the equations in Figure 3.12, we obtain the following values for parameters described in the Table 3.4.

Table 3.4. Power Divider Values Based on Calculation K=1.76dB Z02=87.491Ω Z03=58.327Ω

R2=61.235Ω R3=40.826Ω R=102.061Ω

These values are further confirmed from the online calculators and a snapshot of the obtained values is as shown in Figure 3.14 [40].

Figure 3.14. Verification of values used for designed power divider using Wilkinson calculator. Source: Microwaves101. (2012, June 3). Power Divider Calculator [Online]. Available: http://www.microwaves101.com/encyclopedia/calpowerdivider.cfm

The above values are used to create the necessary power divider design using HFSS on a Rogers 5880 Duroid 1.6mm thick substrate. Two models of power divider operating at different frequency ranges are shown here, one operating from7-8GHz which is our band of interest which is simulated and is fabricated. The other design considered here is at the lower frequency spectrum from 1-2GHz to observe its operational ability over smaller ranges. First described is the design of power divider at our band of consideration 7-8GHz along with the fabricated model of the same. The designed and fabricated models are as shown in Figure 3.15 (a) and (b). The Reflection co-efficient plots for the individual modes are plotted for both the power divider design and the aggregate plot of these has been depicted in Figure 3.16 (a). The mutual coupling obtained between these arms of the power divider is shown in 3.16 (b).

From Figure 3.16 (b) it is observed that, S12 has a value of ~-6dB which is 0.248W, similarly the quantities S13-S16 have almost matching quantities and these values are mentioned in Table 3.5.

(a)

(b) Figure 3.15. (a) Top view of HFSS models for 7-8GHz. (b) Fabricated model of designed power divider.

The results obtained after the fabrication did not match our designed values since the resistors used were not the microwave chip resistors which do not offer any isolation to microwave designs above a few MHz. Therefore, even though this design is modeled and fabricated the measured results cannot be verified. We next proceed towards our design for 1-2GHz band to see if a power divider design can be functional in this band. The designed model for 1-2GHz band is as shown in the Figure 3.17. The reflection co-efficient plot for the 1-2GHz band is as shown in Figure 3.18 (a). The result shows good matching better than -10dB for the entire band. Mutual coupling obtained from the simulated designs are shown in Figure 3.18 (b). The individual coupling

(a)

(b) Figure 3.16 (a) Reflection co-efficient plot and (b) Mutual Coupling plot for power divider at 7-8GHz band.

Table 3.5. Simulated Output Power Values from the Power Divider Model

Power divider Model S12 S13 S14 S15 S16 1-2GHz band, values at 0.3953 0.1318 0.1318 0.1318 0.1318 1.5GHz 7-8GHz band, values at 0.2483 0.11 0.08 0.08 0.11 7.73 GHz

Figure 3.17. Top view of HFSS model for Wilkinson unequal power divider from 1-2GHz band. values when converted into a unit less quantity shows the actual power being supplied to the input ports and validates the design functionality behaving as an unequal power divider.

From Figure 3.18 (b), it is observed that S12 has a value of ~-4.2dB in unit less quantity is

0.38W. Similarly the quantities S13 - S16 all have a fairly constant value and are again described in Table 3.5. Designed power ratio values to be obtained from power divider output arms to obtain the mutual coupling values are S12 = 0.4, S[13- 16] = 0.15 each. These values approximately meet the initial design consideration for the power divider which is

0.4W at TE11 port, and 0.3 W each at TM01 and TE21 ports. Markers were placed on the plot to depict the closes value corresponding to the ports. The values for mutual coupling obtained from Table 3.5 are used as an input mode values for the triple mode feedhorn model feeding a reflector in a symmetric fashion. The results obtained from here for the two bands much necessarily match that of the initial feedhorn design to validate the power divider model. The results obtained from POSTFEKO after simulating these models are shown in Figure 3.19 (a), (b) and (c) for the original model, 7-8GHz band, and 1-2GHz band respectively. The co and cross polarization values obtained in this design are shown in Table 3.6. Values indicated imply that the power divider design has coupling values which are well suitable for the triple mode feedhorn and they do not match to a large extent with the initial design consideration.

3.6 CONCLUSION A triple mode feedhorn operating from 7-8GHz is designed using CADFEKO tool and its performance over the frequency band is analyzed in terms of return loss and radiation patterns. The triple mode feedhorn has the ability to perform beam scanning and has been shown with various beam scanning angle which are obtained by suitable variation of its mode

(a)

(b) Figure 3.18. (a) Reflection co-efficient plot and (b) Mutual Coupling plot for power divider at 7-8GHz band. values and by proper phase excitation. This feedhorn is intended for use in the future sections as a feed source for reflector antenna geometries. A Wilkinson unequal 1:5 power divider is designed using Ansoft HFSS tool. The values obtained are found to be similar to what it has been designed for and the power outputs from ports S2-S6 are obtained within a very close proximity of what need to be supplied as a feed to triple mode feedhorn designed earlier. This triple mode feedhorn design is now used as a feeding source to a reflector dish surface and a detailed study of using this feedhorn and its use in the design is performed in Chapter 4.

Figure 3.19. Co and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11 (40%) + TM01 (30%) + TE21 (30%), (b) TE11 (39%) + TM01 (26%) + TE21 (26%), (c) TE11 (24%) + TM01 (11%) + TE21 (11%) at +90° phase.

(a)

(b)

(c)

Table 3.6. Co and Cross Polarization Gain Comparison for the Power Divider Design Peak Co- Peak Cross- Peak Cross-

Mode Combination polarization polarization Gain Polarization Gain[dBi] [dBi] level [dB]

TE11(40%)+TM01(30%)+TE21(30%) 36.29 25.18 -11.11

TE11(39%)+TM01(26%)+TE21(26%) 36.36 24.52 -11.84

TE11(24%)+TM01(11%)+TE21(11%) 36.19 25.57 -10.62

CHAPTER 4

SYMMETRIC REFLECTOR ANTENNA FED BY TRIPLE MODE FEEDHORN

In this chapter, the design of a reflector fed by a triple mode feedhorn in symmetric configuration is presented. Advantages of this feed design mainly is in ability of beam peak scan in a desired direction by varying the amplitude of individual modes with a fixed phase values. For this research, simulations are performed for individual mode, dual mode combinations and combined triple mode combinations with ±90° phase variation.

4.1 INTRODUCTION TO REFLECTOR ANTENNAS Reflector antennas have been of importance for decades in several areas of electrical engineering, ranging from telecommunications and radar to deep-space exploration and radio astronomy. Application of reflector antennas in these areas is mainly due to high gain of reflector antennas typically above 30dBi. The basic principle of operation of a parabolic reflector is that all rays emanating radially from a point source located at focal point are reflected as a concentrated bundle of parallel rays, which can propagate for very long distances without loss due to spreading [41]. Inversely, incident rays parallel to the axis of symmetry of the paraboloid are all reflected toward its focal point, which concentrates the received signal at a single point. In that case, if the human eye or camera is placed a little bit behind the reflector focal point, an image with enhanced luminosity and definition is formed (Figure 4.1 [18]). Figure 4.1 represent a basic symmetrically fed reflector antenna geometry with the ray scattering shown. Reflector antenna technology has gradually evolved from a basic symmetric reflector to much more complicated alignments of reflectors for various applications. Some applications include remote sensing, satellite application, radar and other defense related applications. Design schematics of reflector antenna analysis methods are shown in Figure 4.2.

Figure 4.1. An illustration of a parabolic reflector antenna configuration. Source: M. A. B. Terada. (1999, Dec. 27). Reflector Antennas [Online]. Available: http://onlinelibrary.wiley.com/doi/10.1002/0471 34608X.W1235/abstract.

Most basic form is the single axi-symmetric parabolic reflector shown in Figure 4.2 (a) [18] which is still in widespread use primarily at low frequencies and for low-cost applications [1]. Large reflectors frequently use an axi-symmetric dual reflector system with a parabolic main reflector as shown in Figure 4.2 (b). Two models shown in Figure 4.2 (b) are basically the Cassegrain and Gregorian type of axi-symmetric dual reflector systems. Axi-symmetric single and dual reflectors mainly suffer from problems such as aperture blockage due to the presence of feed/sub reflector and even supporting mechanical structures in front of the main reflector aperture [1]-[4]. This problem in the design can be solved using the offset reflector assembly system as shown in Figure 4.2 (c) and (d) respectively. Problem which arises using this offset system is that in the radiation pattern we observe an increase in cross-polarization value when compared to symmetrical reflector types, the peak gain obtained in offset reflector type is much more compared to symmetrical reflector types which again is another significant advantage of using this design. In addition, the major advantage of using the dual offset dual shaped reflector is its aperture efficiency which has been reported to be around 85% which is way higher than other reflector systems. Parabolic reflectors have been used far more widely in recent years with advent of satellite television (TV). However reflector antennas find uses in many radio and wireless applications at frequencies usually above about 1GHz where very high levels of RF antenna gain are required along with narrow beam widths. In many professional applications these

Figure 4.2. Reflector antenna configurations (a) Single axi-symmetric, (b) dual axi- symmetric, (c) single offset and (d) dual offset reflector. Source: M. A. B. Terada. (1999, Dec. 27). Reflector Antennas [Online]. Available: http://onlinelibrary.wiley.com/doi/10.1002/047134608X.W1235/abstract.

(a)

(b)

(c)

(d)

parabolic reflectors are used for satellite as well as for radio astronomy and they are used in many microwave links, often being seen on radio relay towers and mobile phone antenna masts. In all these applications very high levels of gain are required to receive the incoming signals that are often at a very low level. For transmitting this type of RF antenna design is able to concentrate the available radiated power into a narrow beam width, ensuring all the available power is radiated in the required direction. RF antenna consists of a radiating system that is used to illuminate a reflector that is curved in the form of a paraboloid. This shape enables a very accurate beam to be obtained. In this way, the feed system forms the actual radiating section of the antenna, and reflecting parabolic surface is purely passive. When looking at parabolic reflector antenna systems there are a number of parameters and terms that are of importance:  Focus: Focus or focal point of the parabolic reflector is the point at which any incoming signals are concentrated. When radiating from this point the signals will be reflected by the reflecting surface and travel in a parallel beam and to provide the required gain and beam width.  Vertex: This is the innermost point at the center of parabolic reflector.  Focal length: Focal length of a parabolic antenna is the distance from its focus to its vertex.  Aperture: Aperture of a parabolic reflector is what may be termed its "opening" or the area which it covers. For a circular reflector, this is described by its diameter. It can be likened to the aperture of an optical lens. The design of the symmetric fed reflector with the multimode feedhorn is performed here and the various analyses have been performed in the coming sections.

4.2 DESIGN AND SIMULATED RESULTS A symmetric feedhorn is designed using the CADFEKO tool and comprises of a reflector surface of diameter = 1.2 m and this is fed by a triple mode feedhorn designed in chapter 3. f/D ratio for this design has been chosen to be 0.5 with the focal length being observed to be 0.6m. Meshing for the entire reflector has been assigned as λ/8 at the highest frequency of operation. Designed model obtained in FEKO is as shown in Figure 4.3. Symmetric feedhorn reflector designed using FEKO is simulated using the method of moment (MoM) method in CADFEKO. The frequency range over which our design is simulated is 7GHz to 8GHz but frequency range under consideration is 7.47 GHz to 7.98

Figure 4.3. Symmetric triple mode feedhorn reflector configuration.

GHz as all three modes are impedance matched. This is shown in reflection coefficient

magnitude plot of Figure 4.4 where the Sii characteristics of the three modes are shown.

Dominant TE11 mode is shown to have a bandwidth of 510 MHz, the TM01 mode is shown as having a bandwidth of 710 MHz and the TE21 mode is observed to have a bandwidth of 890

MHz. In Figure 4.2, S11 corresponds to the reflection coefficient magnitude of the dominant

mode which is TE11, S22 and S33 corresponds to the reflection coefficient magnitude of the

TM01 mode, and S44 and S55 corresponds to the reflection coefficient magnitude of the TE21 mode. Reflection co-efficient plot from Figure 4.4 shows that our designed feedhorn has a good matching over the entire operating band.

4.2.1 Single Mode Analysis A single mode feedhorn analysis basically consists of the individual modes of the feedhorn being excited while keeping the other modes matched terminated. The symmetric feedhorn reflector designed has radiation pattern as shown in Figure 4.5 (a), (b) and (c), respectively. From Figure 4.5 (a), the peak co-polarization gain has been observed to be

Figure 4.4. Reflection co-efficient magnitude for the symmetric feedhorn reflector configuration.

34.12dBi and the other values relating to the co-polarization and cross-polarization are

mentioned in Table 4.1. For the patterns specified in Figure 4.5 (b) which is for the TM01 mode, it has a peak co-polarization gain of 32.86 dBi. Figure 4.5 (c) shows the TE21 mode analysis patterns which has a peak co-polarization gain of 32.89 dBi. All the remaining values are shown in Table 4.1. Individual mode gain values for co-polarization, cross-polarization along with the peak cross-polarization separation is shown in Table 4.1.

4.2.2 Dual Mode Analysis Dual mode analysis of symmetric reflector fed by triple mode feedhorn mainly consists of TE11+TM01 mode combination for ±90° cases and TE11+TE21 mode combination for ±90° phases. Simulations performed here are at the center frequency for our operating band at 7.73GHz. Mode combination shown here are with equal amplitude distributions for either mode i.e.with TE11 mode having 50% of the input power and the TM01 and TE21 mode recieving the rest 50% of the power. Radiation patterns as shown in Figure 4.6(a) shows a peak co-polarization value of 35.18dBi with the cross-polarization value as specified in

Figure 4.5. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11 Mode, (b) TM01 Mode and TE21 Mode at 7.73GHz.

(a)

(b)

(c)

Table 4.1. Individual Mode Gain Values for Symmetric Reflector-Feedhorn Design at 7.73GHz MODE Peak Co-polarization Peak Cross-polarization Peak Cross-Polarization Gain[dBi] Gain [dBi] level [dB]

TE11 34.12 -5.54 -39.66

TM01 32.86 9.43 -23.43

TE21 32.89 -3.70 -36.59

Table 4.2. Now from Figure 4.6 (b) the peak co-polarization gain value is around 35.15 dBi and the cross-polarization gain, and cross -polarization separation is as shown in Table 3.2. For +90° case, beam scanning of the main co-polarization beam is observed to be occurring at +1.2° and for the -90° case shown in Figure 4.6 (b) the beam scanning of the main beam appears to be occurring at -1.2°.

Now the dual mode combination of TE11 (50%) + TE21 (50%) for +90° and -90° phase difference is investigated. Again each mode is assigned equal amplitude values of 0.5 each along with phase values of +90° and -90°. We observe in Figure 4.7 (a), the peak gain to be occurring at +1.4° and for Figure 4.7 (b) we observe, the same peak gain to be occurring at - 1.4°. The null occurring at cross-pol Gain theta at Phi = 90° is now replaced by the Gain phi at Phi = 90° cut and the peak co-pol and cross-pol values are again shown in Table 4.2. Dual

mode analysis for the TE11+TE21 mode combination with the ±90° phases applied between the modes is as shown in Figure 4.7.

4.2.3 Triple Mode Analysis Dual mode analysis performed in earlier section shows the combinations being performed for a phase of ±90° for both cases. Performed here are the tri mode combination of symmetric reflector fed by triple mode feedhorn. The triple mode combination was mailnly performed in this design to observe the variation in the gain patterns and to observe the cross-

polarization level variation. Simulations performed mainly consists of the TE11+TM01+TE21 mode combination for the ±90° phase values. Simulations performed here are at the center frequency 7.73GHz. The mode combination shown here are with amplitude distributions as

(a)

(b) Figure 4.6. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11+TM01 Mode with +90° , (b) TE11+TM01 Mode with -90° at 7.73GHz.

74 Table 4.2. Dual Mode Combinations of the Feedhorn Feeding a Symmetric Reflector with 50% Power Supplied to each Mode at 7.73ghz. Peak Co- Peak Cross- Peak Cross-

Mode Combination polarization polarization Polarization Gain[dBi] Gain [dBi] level [dB]

TE11(50%)+TM01(50%) with +90° Phase 35.18 26.44 -8.74

TE11(50%)+TM01(50%) with -90° Phase 35.15 26.58 -8.57

TE11(50%)+TE21(50%) with +90° Phase 34.75 23 -11.75

TE11(50%)+TE21(50%) with -90° Phase 34.71 22.95 -11.76

TE11(40%)+TM01(30%)+TE21(30%) with +90° Phase 37.13 21.93 -15.2

TE11(40%)+TM01(30%)+TE21(30%) with -90° Phase 36.8 23.7 -13.1

mentioned: TE11 receives 40% of input, TM01 and TE21 receive 30% of input power each i.e.

each waveguide port of the TM01 and TE21 received 15% of input power. Results are as shown below in Figure 4.8 (a) and (b), respectively with the various gain values shown in Table 4.3.

4.3 ANALYSIS ON AMPLITUDE VARIATION ON TRIPLE MODE COMBINATION The variation of the peak co-polarization and the cross-polarization level with the

input feed to each mode is demonstrated in Figure 4.9 where TE11 + TM01 + TE21 modes combination for +90° phase has been excited with different amplitude ratios which we can compare with the case as shown in Figure 4.8 (a). The amplitude ratios, as shown in Figure

4.9 (a) is for TE11 (20%) + TM01 (40%) + TE21 (40%) and Figure 4.9 (b) is for TE11 (60%) +

TM01 (20%) + TE21 (20%). A drastic reduction is observed in cross-polarization levels when

the TE11 mode has been induced with a higher amplitude value, the various values of gain and cross-polar separation levels along with the comparison with Table 4.2 values are shown in Table 4.3. From the values mentioned, we see peak co-polarization values to be maintained at almost same level and also cross-polarization level to be reduced as and when

the input amplitude to TE11 mode is increased.

(a)

(b) Figure 4.7. Co-polarization and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11+TE21 Mode with +90° , (b) TE11+TE21 Mode with -90° at 7.73GHz.

(a)

(b) Figure 4.8. Co and cross-polarization gain radiation patterns for the symmetric feedhorn-reflector combination for (a) TE11+TM01+TE21 mode with +90° , (b) TE11+TM01+TE21 mode with -90° at 7.73GHz.

Table 4.3. Peak Co Polarization Gain and Peak Cross-Polarization Level at 7.73GHz for +90° Peak Co- Peak Cross- Peak Cross-

Mode Combination polarization polarization Gain Polarization Gain[dBi] [dBi] level [dB]

TE11(40%)+TM01(30%)+TE21(30%) 37.13 21.93 -15.2

TE11(20%)+TM01(40%)+TE21(40%) 36.34 26.09 -10.25

TE11(60%)+TM01(20%)+TE21(20%) 37.47 15.43 -22.04

4.4 CONCLUSION A symmetric reflector fed by a triple mode feedhorn is found to be satisfactory in results in accordance with the functiong of various standard reflectors. Single mode excitation, dual mode excitaion, and triple mode excitaions were studied and radiation pattern perfroamance was noted. The triple mode feedhorn is now used to analyse its use in an offset fed reflector feed combination and its results have been discussed in detail in Chapter 5.

(a)

(b) Figure 4.9. Co and cross-polarization gain patterns for the symmetric feed-reflector combination at +90° for (a) TE11 (20%) + TM01 (40%) + TE21 (40%) mode, (b) TE11 (60%) + TM01 (20%) + TE21 (20%) mode at 7.73 GHz.

CHAPTER 5

OFFSET REFLECTOR ANTENNA FED BY TRIPLE MODE FEEDHORN

This chapter introduces the idea of designing an offset fed reflector antenna fed by a triple mode feedhorn. Offset configuration considered here has a reflector with diameter = 1.2m and a f/D ratio being 0.5 which is same being used in earlier design of a symmetric reflector feedhorn system.

5.1 INTRODUCTION Offset parabolic reflector antenna is a variety of axial parabolic antenna where in the feed is located at a certain tilt angle with respect to the parabolic reflector. Of late offset parabolic reflector antennas have become the most preferred option for many practical applications in communications, remote sensing, radio astronomy and radar because of their significant advantages over prime-focal parabolic reflector antennas [42]. Main advantage of this design is its ability to reduce aperture blockage effects. Symmetric reflectors face this problem where in blocking caused by a primary feed or a sub reflector with supporting struts leads to scattered radiation which results in a loss of system gain and a degradation in the suppression of side lobe and cross-polarized radiation [38]. This design has been reported to provide a high isolation between the reflector and feed and also has higher aperture efficiency and suppressed sidelobes when compared to other design types [1]-[3]. Offset reflector design suffers from many serious drawbacks such as a high cross-polarization when illuminated by a linearly polarized primary feed and beam squinting in case of circularly polarized feed [42]. These drawbacks have been worked upon and a number of literatures show documented validation of decrease in the extent of the drawbacks by suitably making the necessary changes in the design. One such case reported is in [43] where we observe a tri- mode matched feed to illuminate a gravitationally balanced back-to-back reflector and achieve an additional 20dB improvement in cross-polarization when compared to other designs.

When compared to axi-symmetric paraboloid, the offset reflector configuration leads to the use of larger focal-length to diameter ratios (f/D) while maintaining an acceptable rigidity [38, 44]. However, larger f/D results in a bulky antenna structure and may not be suitable for spacecraft antennas in which the available space for the antenna is limited. In this design the f/D ratio are fixed as 0.5 as the need was to perform the comparison of this model with that of the symmetric parabolic reflector model presented in Chapter 4 of this book. Single mode, dual mode and triple mode analysis have been performed similar to the earlier case and the results have been reported.

5.2 DESIGN AND SIMULATED RESULTS The designed offset reflector feedhorn configuration is shown in Figure 5.1. The reflector has a diameter of 1.2m and the f/D ratio is 0.5. Triple mode feedhorn feeding the reflector is found to be having a maximum gain when the feedhorn is placed at an angle of 27° offset with respect to the reflector surface. This offset angle value can be verified here, since the f/D ratios for the design = 0.5 and on substituting it in the Equation. 5.1 [1] we get a value of 28° which is close to the designed value. ∗cot (5.1) The design performed in this case is very similar to the analysis being performed for the symmetric reflector feedhorn case. The results obtained by the two models are being analyzed. We first present the offset design as shown in Figure 5.1. A line is drawn from the rim of the reflector surface to the feed center position to show that the feed is placed at an offset to the reflector geometry and is placed at the phase center position of the feed. Offset feedhorn reflector designed using FEKO is simulated using the method of moment (MoM) method in CADFEKO. Frequency range over which our design is simulated is 7 GHz to 8 GHz but the frequency range under consideration is 7.47 GHz to 7.98 GHz as all the three modes falls in this mode. This is shown in reflection coefficient magnitude plot

shown in Figure 5.2 where the Sii characteristics of the three modes are shown. The offset design is found to be having a similar bandwidth as compared to the symmetric reflector

design’s dominant mode TE11 mode with a bandwidth of 530MHz, the TM01 mode has a

bandwidth of 720MHz and the TE21 mode is observed to have a bandwidth of 910 MHz. In

Figure 5.2, S11 corresponds to the reflection coefficient magnitude of the dominant mode

Figure 5.1. Offset triple mode feedhorn reflector configuration.

which is TE11, S22 and S33 corresponds to the reflection coefficient magnitude of the TM01 mode, and S44 and S55 corresponds to the reflection coefficient magnitude of the TE21 mode. The reflection co-efficient plot shows that our designed feedhorn has a good matching over the entire operating band. We now proceed with the designing of our triple mode feedhorn reflector design.

5.2.1 Single Mode Analysis Single mode analysis is basically analysis of individual modes for the offset reflector feedhorn design and during each analysis other two modes are matched terminated. Triple mode feedhorn designed had the TE11, TM01 and TE21 modes. Radiation patterns are as shown in Figure 5.3 (a), (b) and (c), respectively. From Figure 5.3 (a), peak co-polarization gain observed is 34.37dBi and the other values relating to co-polarization and cross- polarization are mentioned in Table 5.1. For the pattern specified in Figure 5.3 (b) (the TM01 mode) a peak co-polarization gain of 27.94 dBi is noted. Now for the patterns shown in

Figure 5.3 (c) which is for the TE21 mode has a peak co-polarization gain of 27.5 dBi.

Figure 5.2. Reflection co-efficient magnitude for the offset feedhorn reflector configuration.

The other values such as cross-polarization gain are shown in Table 5.1. The individual mode gain values showing the co-polarization, the cross-polarization gain along with the peak cross-polarization separation is as shown in Table 5.1.

5.2.2 Dual Mode Analysis Dual mode analysis of the offset fed reflector antenna has been demonstrated here.

Simulations shown here are mainly for TE11+TM01 mode with ±90° phase applied between modes and TE11+TE21 mode combination with ±90° phase applied. Combined mode combination of TE11 (50%) + TM01 (50%) with equal amplitude between modes and a phase difference of ±90° applied between then is as shown in Figure 5.4. The values for the peak gain co-pol and cross-pol are as specified in Table 5.2. Similar to the case of symmetric reflector wherein the dual mode combinations produce a beam scanning at ±1° for the phase shifts applied. The peak co-pol and cross pol separations remain close to 11.5dB in Figure 5.4 (a) and 12.9 dB for the case in Figure 5.4 (b). Figure 5.4 mainly shows the co- polarization and the cross-polarization gain, while their comparison and the peak cross polarization level have been shown in Table 5.2.

Figure 5.3. Co-polarization and cross-polarization gain radiation patterns for the Offset feedhorn-reflector combination for (a) TE11 mode, (b) TM01 mode and TE21 mode at 7.73GHz.

(a)

(b)

(c)

Table 5.1. Individual Mode Gain Values for Offset Reflector-Feedhorn Design at 7.73GHz MODE Peak Co-polarization Peak Cross-polarization Peak Cross-Polarization Gain [dBi] Gain [dBi] level [dB]

TE11 34.35 6.17 -28.18

TM01 27.9 8.82 -19.08

TE21 27.48 6.8 -20.68

The modal combination for TE11 (50%) + TE21 (50%) with equal amplitude distribution to modes and again with a phase difference of ±90° applied between then is as shown in Figure 5.5 (a) and (b). The values for the peak co-pol and cross-pol gain values are as given in Table 5.2. In this case we observe a rise in both the XZ and the YZ cross-pol levels when compared to the previous cases. The peak co-pol and cross-pol separation for case in Figure 5.5 (a) is 16.1dB and for Figure 5.5 (b) is 16.2dB which indicates an improvement when compared to that of the symmetric reflector feedhorn case with same modal combinations.

5.2.3 Triple Mode Combination Analysis

Combined mode gain patterns for TE11 (40%) + TM01 (30%) + TE21 (30%) mode with phases of +90° and -90° is as shown in Figure 5.6 (a) and (b), respectively. Again the co-pol levels are not showing that high a gain value mainly due to f/D ratio, size of reflector used as well as the elevation post most commonly used for offset reflector design. For the case in Figure 5.6 (a) observed is a co-pol gain of 37.28dBi with a peak cross-pol gain level of 16.4dBi which makes the peak cross-polarization separation to be around 21dB. The case in Figure 5.6 (b) shows a peak gain of 36.92dBi and a cross-pol peak gain value of 17.94 dBi making the peak cross-polar separation to be 19dB. These values are shown in Table 5.2. This design when compared to that of symmetric design shows a cross-polar separation improvement of 15dB for case in Figure 4.8 (a) and 13.2dB for the case shown in Figure 4.8 (b). A comparative study for the offset-reflector feedhorn configuration at different frequency points of 7.45GHz , 7.73GHz and 7.9GHz which fall in our desired band

is as shown in Figure 5.7 which is the combined mode combination of TE11 (40%) + TM01

(a)

(b) Figure 5.4. Co-polarization and cross-polarization gain radiation patterns for the offset feedhorn-reflector combination for (a) TE11+TM01 mode with +90° , (b) TE11+TM01 mode with -90° at 7.73 GHz.

Table 5.2. Dual Mode Combination Analysis for the Offset Fed Triple Mode Feedhorn Design at 7.73GHz with Equal Amplitude Supplied to Modes. Peak Co- Peak Cross- Peak Cross-

Mode polarization polarization Polarization Gain [dBi] Gain [dBi] level [dB]

TE11+TM01 with +90° Phase 34.94 23.4 -11.54

TE11+TM01 with -90° Phase 35.13 22.9 -12.84

TE11+TE21 with +90° Phase 34.63 18.33 -16.13

TE11+TE21 with -90° Phase 34.62 18.44 -16.18

TE11+TM01+TE21 with +90° Phase 37.28 16.43 -20.85

TE11+TM01+TE21 with -90° Phase 36.92 17.94 -18.98

(30%) + TE21 (30%) with a +90° phase applied. This has been done mainly to show that our design works for the entire operating band with minimal changes in peak co-polarization and cross-polarization gain. Co-polarization and cross-polarizations gain values along with peak cross- polarization levels are shown in Table 5.3.

5.3 CONCLUSION Offset reflector antenna designed using the triple mode feedhorn has been analyzed and the results obtained have been compared with that of the symmetric feedhorn reflector. This design can be used for radar applications. This concludes the research work being performed on triple mode feedhorns and its use in studying the symmetric and offset reflector performance. Next part of the research work explained in Chapter 6 deals with the study of a spiral loaded dipole antennas, making it reconfigurable by placing RF switches in appropriate locations and the results obtained from this design.

(a)

(b) Figure 5.5. Co-polarization and cross-polarization gain radiation patterns for the offset feedhorn-reflector combination for (a) TE11+TE21 mode with +90° , (b) TE11+TE21 mode with -90° at 7.73 GHz.

(a)

(b) Figure 5.6. Co and cross-polarization gain radiation patterns for the offset feedhorn-reflector combination for (a) TE11+TM01+TE21 mode with +90° , (b) TE11+TM01+TE21 mode with -90° at 7.73GHz.

Figure 5.7. Gain radiation patterns for offset reflector feedhorn design for TE11 (40%) + TM01 (30%) + TE21 (30%) mode at +90° for (a) 7.45GHz, (b) 7.73GHz and (c) 7.9GHz.

(a)

(b)

(c)

Table 5.3. Peak Co Polarization Gain and Peak Cross-Polarization Level Values for Mode TE11 (40%) + TM01 (30%) +TE21 (30%) at +90° at Different Frequencies Frequency Peak Co- Peak Cross- Peak Cross- [GHz] polarization polarization Gain Polarization Gain[dBi] [dBi] level [dB]

At 7.45 37.07 14.47 -22.6

At 7.73 37.28 16.43 -20.85

At 7.9 37.62 16.78 -20.84

CHAPTER 6

FREQUENCY RECONFIGURABLE SPIRAL LOADED PLANAR DIPOLE

6.1 INTRODUCTION The dipole antenna is one of the most important and also one of the most widely used types of antenna. It can be used on its own, or there are many other types of antenna that use the dipole as the basic element within the antenna i.e. they can be used as a driven element for an array of Yagi-Uda antenna. A dipole antenna can be made of a simple wire, with a center-fed driven element. It consists of two metal conductors of rod or wire, collinear with each other (in line with each other), with a small space between them. The radio frequency voltage is applied to the antenna at the center, between the two conductors. These antennas are the simplest practical antennas from a theoretical point of view. The basic construction of a dipole is quite straightforward - a simple dipole antenna can be constructed from a few simple pieces of wire. In this way antennas including Frequency Modulation (FM) dipole antennas or antennas for the short wave bands can easily be made. These antennas, while not having the performance of other more complicated types of antenna can nevertheless prove very effective and quite satisfactory in many applications. The name dipole means two poles and the antenna does in fact consist of two "poles" or sections. These are normally equal in length, making the antenna what is termed a center fed antenna. Sometimes a dipole may not be fed in the center, although this is not normally done in most antenna designs. The power is applied to the dipole antenna itself through a feeder. Conversely if the dipole antenna is used for receiving, the received signals are taken away to the receiver through a feeder. The feeder serves to transfer the power to or from the antenna with as little loss as possible. A basic dipole antenna configuration and a half wave dipole antenna are shown in Figure 6.1 (a) and (b) [37] respectively.

Figure 6.1. A general representation of (a) basic dipole antenna (b) half wave dipole antenna. Source: Wikipedia. (2012, Sept. 12). Horn Antenna [Online]. Available: http://en.wikipedia.org/wiki/Horn_antenna.

Dipoles whose length is half the wavelength of the signal are called half-wave dipoles, and are more efficient. This is because impedance of dipole is purely resistive at about this length. The length of the dipole antenna is approximately 95% of half a wavelength at the speed of light in free space. Dipoles have radiation pattern, shaped like a toroid (doughnut) symmetrical about the axis of the dipole. The radiation is maximum at right angles to the dipole, dropping off to zero on the antenna's axis as shown in Figure 6.2 [45]. The maximum theoretical gain of a λ/2-dipole is 10 log 1.64 or 2.15 dBi. This pattern degrades considerably when the dipole is brought closer to the ground.

Figure 6.2. A 2D and 3D radiation pattern of a standard dipole. Source: I. Poole. (n.d.). FM Dipole Antenna [Online]. Available: http://www.electronics- radio.com/articles/radio/antennas/dipole/fm-dipole-antenna.php

The half-wave length dipole antennas are one of the most extensively used antennas and finds several wireless communication applications. Dipole antennas can be realized using finite diameter wire or using microstrip technology so it is a planar structure. A novel spiral loaded planar dipole antenna is presented here with frequency reconfiguration for frequencies 760 MHz, 1.47 GHz and 2.2 GHz band by using the PIN diodes as switches. Section 6.2 presents planar dipole and spiral loaded planar antenna geometries and their performance. Section 6.3 discusses frequency reconfigurable spiral loaded planar dipole antenna by employing PIN diodes. Finally, Section 6.4 presents conclusion. The full wave analysis has been performed using finite element method (FEM) based commercial tool Ansoft High Frequency Structure Simulator (HFSS) v. 12.

6.2 PLANAR VS. SPIRAL DIPOLE ANTENNA A planar half-wavelength dipole antenna is designed to operate around 760MHz which falls in LTE 700MHz band. The dipole length, L, was found to be 92mm in each arm which together gives the resonance at 760MHz. The antenna geometry is shown in Figure 6.3

Figure 6.3. Geometry of (a) planar microstrip dipole antenna and (b) a spiral loaded microstrip dipole antenna.

(a). The upper half of the dipole arm is sitting on top part of the FR-4 substrate (r = 4.4, tan  = 0.02, thickness, h = 0.8mm) while the bottom half is on the lower part of the substrate as shown in side view of the antenna in Figure 6.3 (b). The total dimensions of the substrate is L x W =100mm X 40mm. The 50  SMA feed is being used as an excitation and is located at the center as shown in Figure 6.3 (b). Since one of the ideas is to miniaturize the antenna, we bent the end parts of the dipole as a spiral on both upper and lower parts of the dipole arms along with meandering the arms so that expected compactness could be achieved. The resulting spiral dimensions are given in Table 6.1.

Table 6.1. Spiral Parameters Spiral Parameter Value

Trace width 1.4mm Spiral Constant/ Growth Factor 0.315 mm/rad Start angle 1.5 rad End angle 23.58 rad

Number of turns 3.5

Spacing between turns 0.6mm

The spirals when uncoiled are not of the same length as that of the original dipole because spirals have been parameterized to achieve the antenna resonance. Since the spirals have the nature of creating an inductive effect, hence the antenna resonance frequency could be maintained as the original resonance of the planar dipole. The lengths of the spiral loaded dipole arms L1 and L2 are 44mm and 42mm, respectively. The lengths are not proportionate mainly to improve the matching level of the antenna which was obtained from the calculations made from a single stub tuning [6]. On comparison of the lengths of planar dipole and the spiral loaded dipole, we can see that the spiral loaded dipole is less than 50% in terms of the dipole arm hence 50% miniaturization is obtained except that spiral and meander ports require some width now. The comparison of the reflection co-efficient magnitudes (S11, dB) of the planar

dipole and the spiral loaded planar dipole are shown in Figure 6.4. The measured S11 for the Spiral loaded planar dipole (SPLD) is also shown in the Figure 6.4, it is observed that all the antennas are matched better than S11 = -10 dB criterion. The center of the resonant frequencies are at 760MHz for the simulated spiral loaded planar dipole, 770MHz for the measured SPLD and 765 MHz for the planar dipole. Further, matching bandwidth for spiral loaded dipole (from 755MHz to 765MHz) is less than the planar dipole (from 745MHz to 790MHz). The next section deals with determining the realized gains obtained from the spiral loaded planar dipole and the planar dipole designs. A comparison of the peak realized gain between the planar dipole and the spiral loaded planar dipole is shown in Figure 6.5. The

100

Figure 6.4. Comparison of the simulated reflection coefficient magnitudes (S11, dB) of the planar conventional dipole vs. simulated and measured spiral loaded dipole antenna.

Figure 6.5. Comparison of the simulated peak realized gain of planar dipole vs. spiral loaded compact planar dipole antenna.

101

peak realized gain for the planar dipole was observed to be around 2.09dBi whereas that of the spiral loaded dipole is observed to be at 0.5dBi. The reduction in the peak realized gain with the spiral loaded dipole is mainly due the compactness achieved. We have not plotted the peak realized gain measured value for the spiral loaded dipole since the range of the standard gain horns to measure it in the anechoic chambers at the AML lab does not extend till 700 MHz, we have it from 800 MHz to 12 GHz and hence the data could not be measured here and hence has not be given here. Since the realized gain has been depicted here we now proceed to show currents density on the dipole. The simulated current distribution of the spiral loaded dipole antenna is shown in Figure 6.6 (a). The Figure 6.6 (a) shows a very high current flowing through the entire length of the dipoles as well as through the spirals which makes this antenna resonate at 760MHz and this also shows that the whole antenna is radiating at the desired frequency location. The spiral loaded planar dipole was fabricated on the FR4 substrate of thickness 0.8 mm and having a ground plane of size 100mm x 40mm. Photograph of the fabricated prototype of the spiral loaded planar dipole antenna is shown in Figure 6.6 (b) which was completed by employing a LPKF milling machine available in the Antenna and Microwave Lab (AML) at San Diego State University. The 50 SMA feed is placed at the center of the dipole and the whole model is as shown. A standard dipole antenna has a typical omni-directional radiation pattern in one plane and a null in the other plane. Here the simulated 3-D radiation patterns for the spiral loaded planar dipole antenna are shown in Figure 6.7 with the gain scale on the left. The plot clearly depicts that there is null along the dipole axis (Y-axis, φ=90o and  = 90o) whereas omni-directional patterns along XZ- cut (φ=0o cut plane and =variable) and this clearly is in close agreement with that of a standard dipole antenna pattern. Illustrated above was the 3 dimensional simulated radiation pattern. The simulated and measured normalized radiation patterns obtained for the spiral loaded planar dipole are shown in Figure 6.8 (a) and (b), respectively. The simulated and measured patterns when compared show good co-polarization and cross-polarization levels which are in reasonable agreement in terms of the omni-directional patterns and the null pattern in respective planes. Now on more careful analysis of the Figure 6.7 (b) which is the normalized measured 2D radiation pattern, we observe a slight increase in the cross-polarization level which is mainly

102

Figure 6.6. (a) Current distribution of the spiral loaded dipole at 760MHz and (b) photograph of the fabricated antenna. due to the use of extended non flexible co- axial cable of 6 inches while measuring the antenna in the anechoic chamber. This cable is used to mount the antenna away from the antenna under test (AUT) post. Further, some misalignment in mounting the antenna with the feed broadband horn also contributes to the increased levels of cross-polarization. Now after observing all the above results on the spiral loaded planar dipole, its operation can be confirmed and can say that this design works pretty well for its designed functionality. Compactness in the design has also been achieved when compared to a planar dipole design. Proceeding now with the designing of a spiral loaded planar dipole by using PIN diode switches, the main intention being to achieve reconfigurabilty by proper tuning of the switches and by pre determining the bands of operation needed.

103

Figure 6.7 Simulated 3-D radiation pattern at 760MHz for spiral loaded compact dipole antenna.

6.3 FREQUENCY RECONFIGURABLE SPIRAL LOADED PLANAR DIPOLE ANTENNA A spiral loaded planar dipole designed earlier to operate in the 760MHz band is used again in this design. We now use the earlier model to implement the reconfigurabilty concept by suitably placing the switches and the biasing lines all along the model and the designed spiral dipole is as shown in Figure 6.8. We have shown the top view and the bottom view of the design, respectively. This model is designed on a FR-4 substrate having thickness of 0.8mm and the total size of the ground plane being 100mm x 40mm and the feed used here is a SMA feed similar to the one used earlier. The top view shows the whole schematic of the design and the side view shows the bottom and top spirals separated by 0.8mm thick FR-4 substrate. The green colored arms in the design represent mainly the bias lines being used here so as to have proper dc blockage and as well have proper grounding to the design too. The switches are placed at locations S1, S2, S3 and S4 between every bias line arm. In between all the arms, we again observe strips being cut which basically are again the dc blocking capacitors placed to avoid a short happening in the design. There are four such capacitors been placed in this design each on arm L1, L4 and L6 and the other is placed on

104

Figure 6.8. Normalized (a) simulated and (b) measured radiation patterns at 760 MHz for the compact spiral loaded dipole antenna.

105 the arm between the switches S3 and S4. The lengths and width of the spirals loaded antenna with the switch implementation is shown in Figure 6.9 and the designed dimensions summarized in Table 6.2, where all the units are specified in mm.

Figure 6.9. Spiral dipole antenna simulated model having required bias network.

Table 6.2. Pin Diode Implemented Spiral Loaded Frequency Reconfigurable Antenna Design Dimensions (in mm unit)

L1 L2 L3 L4 L5 L6 W1

9.7 6.75 6 13.45 17.09 7.7 14.87

The lengths L1 and L6 are not being equal again is mainly to obtain good matching levels at the desired frequencies points. The other design parameters L2-L4 and W1 and switch locations were optimized for the acceptable antenna performance. This design not only resonates at 760MHz for which this spiral dipole was originally designed, but it now works for 1.47GHz and 2.2GHz also by the switching mechanism. This is basically implementing three antennas in one which is clear benefit of the reconfigurable antennas but

106 at the cost of RF switching network and fabrication complexity associated with it. It should be pointed out that, addition of the switch and biasing components affect passive nature of the antenna performance as these components add some inductive and capacitive reactances. Bias lines are placed on arms of the dipole accordingly, since the design has four switches to tune the different frequency bands. The switching circuit mainly consists of diodes placed along the length of the dipoles in the locations where the switching of the dipole arm lengths are desired to occur. The antenna’s biasing network uses 4 blocking capacitors of 20pF each, 2 of which are placed at the center of the arm lengths L1 & L6 next to the 50 SMA to prevent the DC from flowing back into the RF input and to confine it within the diode region. The second set of blocking capacitors are added on the dipole arm lengths between the switches S1,S2, S3 and S4, respectively, so that the RF input of one switch does not enter to that of the other switch and short circuit the design in the process. Each of the bias arms mainly consist of an inductor of 86nH on the shorter arm which is then followed by a resistor of 47Ω which is placed between the two circular pads in the arms.

After the resistor, the +Vcc and –Vcc (bias voltage) are connected to the two rectangular pads on the ends of arms. Diodes which are placed are represented by the switches S1, S2, S3 and S4. The diode values used in this design are obtained from the datasheet of SKYWORKS: SMP1340-040LF. By suitably turning the switches ON and OFF states, we obtain the resonance frequencies mentioned earlier. A part of the biasing circuit with the ON and OFF states and the capacitor, inductor and resistor values in the respective state are all shown in Figure 6.10. The reflection co-efficient magnitude plot is shown in Figure 6.11. Simulated and measured plots are shown for 760 MHz band as well as the 1.47 GHz and 2.2 GHz bands which are obtained through suitably reconfiguring the switches. The simulated S11 is found to be almost matching with the measured value for all the bands except that the matching level is affected for the measured data than the simulated ones. We observe a slight drift in the resonant frequency of the 760 MHz band as it shifted to 785 MHz which is mainly because now spiral loaded dipole is resonating along with other switching networks which have parasitic values involved. This can also be inferred from the small currents on the bias padds which add up to the total current length thus making it slightly bigger pathlength for current and hence the shift in resonant frequency. We also observe that the frequency point

107

Figure 6.10. Biasing circuit shown with values for ON and OFF state. for matching for the other bands (1.47 GHz and 2.2 GHz) are not affected however, much levels and achievable bandwidths have changed attributed to the factors mentioned earlier. Percentage bandwidth obtained for the simulated switching model for three bands i.e. 0.76 GHz, 1.47 GHz and 2.18 GHz are 2.13%, 7.45% and 10.9% respectively. On comparing it with measured percentage bandwidths, we observe the measured values for three bands to be 5.19%, 3.5% and 2.1% respectively. The observed changes in the simulated and measured results are mainly due to the losses occurring in the bias lines as well as the L, C components used in the design. The current distributions for the three resonance frequencies are shown in Figure 6.12 which clearly shows that the significant portions of current are limited to dipole lengths associated with these frequencies. For 2.2 GHz case, current is primarily on the dipole arms till the switches S1 and S3 (Figure 6.8). In this case all the four switches are in the “OFF”

108

-5

-10 1.47GHz 2.18GHz -15

0.760GHz -20 1.47GHz S11 Magnitude (dB) S11 0.785GHz -25 S11 Simulated S11 Measured 2.19GHz -30 0.5 0.75 1 1.25 1.5 1.75 2 2.25 2.5 Frequency (GHz)

Figure 6.11. Simulated vs. measured reflection co-efficient magnitudes (S11, dB) for the spiral loaded compact dipole antenna with PIN diode switches.

state. Similar is the case, for the 1.47GHz band where S1 and S3 are “ON”, and S2 and S4 are in the “OFF” state so that the current flows in the spirals are turned OFF and mainly on the dipole lengths controlling 1.47GHz resonance. However, for the 760 MHz band, we observe that the current is flowing through the entire dipole lengths including the spirals. This current distribution suggests the three resonances this antenna is providing. The simulated 3D radiation patterns for the spiral loaded planar dipole antenna at these frequencies 760MHz, 1.47GHz, and 2.2GHz are shown in Figure 6.13 (a), (b) and (c), respectively. From these patterns we can observe that, all the patterns maintain null along the dipole axis while omni-directional pattern exists along the XZ-plane as expected. The realized gain scale on the left suggest that, for the 760 MHz, the realized gain is pretty low and for the 2.2 GHz the gain appears to be pretty high. The peak realized gain values are: 0.24 dBi, 1.68 dBi and 2.11 dBi which are according to the effective aperture areas attached

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Figure 6.12. Current distribution plots at (a) 2.4GHz, (b) 1.8GHz and (c) 0.750GHz . to the antenna at these frequencies which also takes into account the bias lines and switch presence. Next section explains what the measured radiation patterns would be for the reconfigurable spiral loaded planar dipole antenna. They are presented in Figure 6.14 (a), (b) and (c) at 0.76 GHz, 1.47 GHz and 2.2 GHz, respectively. Measured radiation patterns differ from conventional dipole patterns in the sense that measured patterns show high cross- polarization levels unlike the ones shown earlier in Figure 6.7 which are attributed to the following: 1. Bias source applied in each bias line is using button type battery cells which are also cause for scattering because of its finite size and can causes variations in the radiation patterns.

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Figure 6.13. Simulated 3D omni-directional radiation patterns for (a) 0.76GHz. (b) 1.47GHz. (c) 2.2GHz for the compact spiral loaded planar dipole antenna.

2. The antenna was measured using co-axial cables and 10 dB attenuators to prevent the switches from burning out and also to protect the vector network analyzer; this in turn affects the measured patterns. As mentioned above, these factors have affected the measured cross-polarization components at the three bands severely, however the co-polarization patterns at 760 MHz is also affected showing ripples in the patterns. However, it can also be concluded that these patterns are omni-directional in nature at these frequencies. Further, the cross-polarization levels are approaching the co-polarization levels which can be advantageous in case of the wireless communication application where signal receptions in any orientation are preferred.

6.3 CONCLUSION A planar dipole antenna and a spiral loaded planar dipole antenna were designed and it was found that the spiral loaded planar dipole antenna is a more compact antenna design compared to the planar straight dipole showing omni-directional radiation patterns . The realized gain and the patterns obtained for the planar dipole was found to be pretty good in

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Figure 6.14. Measured normalized radiation patterns at (a) 0.760GHz, (b) 1.47GHz and (c) 2.2GHz for the compact spiral loaded dipole antenna.

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(a)

(b)

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(c)

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terms of their matching levels and the current distribution also showed that the entire spiral dipole is resonating at the desired 760 MHz band of operation which makes it a suitable design. After the planar dipole antenna has been designed and verified, the next step was to move ahead with the concept of determining if this design is frequency reconfigurable. After calculations were performed to determine the required resonating length required for the desired frequencies be resonating, switches were places at those resonating lengths and found the spiral loaded planar dipole to be reconfigurable. This antenna was frequency reconfigured by employing PIN diodes along with the desired bias circuit to achieve three bands around 0.76 GHz, 1.47 GHz and 2.2 GHz with nearly omni-directional radiation patterns. This antenna can find applications in wireless communications where one antenna will serve for three different bands while still providing omni-directional patterns.

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CHAPTER 7

CONCLUSIONS AND FUTURE STUDY

Main purpose of this thesis work is to present two aspects of antenna work: first is to analyze reflector antenna performance by employing a triple mode feedhorn (designed earlier by our research group) as a feeding source to both the symmetric and offset reflector antennas and the second is to design a reconfigurable spiral loaded planar dipole antenna using PIN diode switches. In Chapter 2, the design of a dominant mode cylindrical waveguide based feedhorn is discussed as a feeding source for a reflector antenna. Next, reflector antenna performance results using this feedhorn and using a theoretical feed model in TICRS's GRASP is presented. A study was performed on the different f/D ratios as to what effects it has on the design. It was also found out that for the offset reflector design better gain can be obtained. Chapter 3 presented the redesign of a triple mode feedhorn from the reference [23] having the TE11, TM01 and TE21 modes excited. The beam scanning can be observed in this feed design by proper variations in the power being fed but for the fixed selected phase values between the modes. The scanning obtained here can be compared to the scanning obtainable in a phased array antenna. A Wilkinson unequal power divider was designed to excite the feedhorn. As a future enhancement, the power divider could be designed as a variable power divider and this single model could be used for feeding various mode values to the feedhorn. Chapters 4 and 5 presented the design of a symmetrical and offset reflector antennas fed by the triple mode feedhorn designed in the previous chapter. Various co-polarization and cross-polarization patterns have been analyzed and compared. The results obtained here are found to be satisfactory and can find applications in radar applications. For the future study of this design, Cassegrain and Gregorian type reflector configurations could be used to obtain a better gain and to reduce the cross-polarization effects. Chapter 6 presented a compact spiral loaded planar dipole antenna design which is a modification of a planar dipole antenna design. This design is made to achieve compactness

116 by the use of spirals at the ends of dipole arms and also to make the design reconfigurable by implementing PIN diode switching at the appropriate λ/4 sections corresponding to a specific frequency reconfiguration target. This design has been fabricated and tested and the measured radiation patterns have been found to be similar to the simulated patterns. This design could be useful for wireless communication applications where in one antenna can be used for three different frequency bands of 0.76 GHz, 1.47 GHz and 2.2 GHz. As a future study this design could also be used in array design.

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