EURASIP Journal on Applied Signal Processing 2005:3, 252–272 c 2005 Hindawi Publishing Corporation
Robert A. Scholtz Department of Electrical Engineering, University of Southern California, Los Angeles, CA 90089, USA Email: [email protected]
David M. Pozar Department of Electrical and Computer Engineering, University of Massachusetts, Amherst, MA 01003, USA Email: [email protected]
Won Namgoong Department of Electrical Engineering, University of Southern California, Los Angeles, CA 90089, USA Email: [email protected]
Received 12 May 2004
The application of ultra-wideband (UWB) technology to low-cost short-range communications presents unique challenges to the communications engineer. The impact of the US FCC’s regulations and the characteristics of the low-power UWB propagation channels are explored, and their effects on UWB hardware design are illustrated. This tutorial introduction includes references to more detailed explorations of the subject. Keywords and phrases: UWB radio, UWB propagation, UWB antennas, UWB radio architectures, selective RAKE receivers, transmitted-reference receivers.
1. ORIGINS “BA[seband]R[adars] have been ... recently demon- strated for various applications, including auto It has been said that paradigm shifts in design and opera- precollision sensing, spaceship docking, airport sur- tion of systems are necessary to achieve orders-of-magnitude face traffic control, tanker ship docking, harbor col- changes in performance. It would seem that such events lision avoidance, etc. These sensing applications cover have occurred in the world of radio communications with ranges from 5 to 5000 ft .... the advent of ultra-wideband (UWB) radio. Indeed, several remarkable innovations have taken place in the brief his- Further applications resulted in the construction of a tory of UWB radio. Initially transient analysis and time- sub-nanosecond, single coaxial cable scheme for mul- domain measurements in microwave networks (1960s) and tiplexing data between computer terminals .... More the patenting of short-pulse (often called impulse or carrierless recently baseband pulse techniques have been applied or baseband or UWB) radio systems in the early 1970s were to the problem of developing a short-range wireless major departures from the then-current engineering prac- communication link. Here, the low EM pollution and tices. (For detailed descriptions of the early work in this field, covertness of operation potentially provide the means see [1].) Marconi’s view of using modulated sinusoidal car- for wireless transmission without licensing.” (From riers and high-Q filters for channelization has so dominated the Abstract of C. L. Bennett and G. F. Ross, Time- design and regulation of RF systems since the early twenti- domain electromagnetics and its applications, Proc. eth century, that the viability of short-pulse systems often has IEEE, March 1978.) been greeted with skepticism. The early applications of UWB technology were primarily Bennett and Ross described the state of UWB engineering radar related, driven by the promise of fine-range resolution efforts near the end of the 1970s in a revealing paper. that comes with large bandwidth. In the early 1990s, con- ferences on UWB technology were initiated and proceedings This is an open-access article distributed under the Creative Commons documented in book form [2, 3, 4, 5, 6, 7]. For the most part, Attribution License, which permits unrestricted use, distribution, and the papers at these conferences are motivated by radar appli- reproduction in any medium, provided the original work is properly cited. cations. Ultra-Wideband Radio 253
Table 1: Categories of applications approved by the FCC [8].
Class/application Frequency band for operation at part 1 limit User limitations Communications and 3.1to10.6GHz(different out-of-band emission No measurement systems limits for indoor and outdoor devices) Imaging: ground penetrating < . . radar, wall, medical imaging 960 MHz or 3 1to106GHz Yes Imaging: through wall < 960 MHz or 1.99 to 10.6GHz Yes Imaging: surveillance 1.99 to 10.6GHz Yes Vehicular 24 to 29 GHz No
Beginning in the late 1980s, small companies, for exam- −40 ple, Multispectral Solutions, Inc. (http://www.multispectral. − com/history.html), Pulson Communications (later to be- 45 come Time Domain Corporation), and Aether Wire and −50 Location (http://www.aetherwire.com), specializing in UWB technology, started basic research and development on com- −55 munications and positioning systems. By the mid-1990s, −60 when the UltRa Lab at the University of Southern California was formed (http://ultra.usc.edu/New Site/), lobbying the −65 US Federal Communications Commission (FCC) to allow − UWB technology to be commercialized was beginning. At a 70 UWB EIRP emission level (dBm) US Army Research Office/UltRa Lab-Sponsored Workshop −75 in May 1998, an FCC representative indicated that a notice of inquiry (NOI) into UWB was imminent, and the compa- 100 101 nies working on UWB technology decided to band together Frequency (GHz) in an informal industry association now known as the Ultra- Indoor limit Wideband Working Group (http://www.uwb.org). The ob- Part 15 limit jective of this association was to convince the FCC to render Figure 1: FCC’s spectral mask for indoor communications appli- a ruling favorable to the commercialization of UWB radio cations [8], specifying measurements in a 1 MHz band. Different systems. masksareusedfordifferent application categories. The FCC issued the NOI in September 1998 and within a year the Time Domain Corporation, US Radar, and Zircon Corporation had received waivers from the FCC to allow lim- regulation to be short pulse in nature, but it is noteworthy ited deployment of a small number of UWB devices to sup- that testing of swept or stepped frequency systems must be port continued development of the technology, and USC’s determined with the sweep or step process turned off,mak- UltRa Lab had an experimental license to study UWB radio ing compliance unlikely (see [8, paragraph 32]). Devices sat- transmissions. A notice of proposed rule making was issued isfying the adopted UWB communication regulations will be in May 2000. In April 2002, after extensive commentary from allowed to operate on an unlicensed basis, fulfilling the po- industry, the FCC issued its first report and order on UWB tential noted by Bennett and Ross in 1978. technology, thereby providing regulations to support deploy- A further FCC memorandum opinion and order and fur- ment of UWB radio systems. This FCC action was a major ther notice of proposed rule making [9]“doesnotmakeany change in the approach to the regulation of RF emissions, al- significant changes to the now-existing UWB technical pa- lowing a significant portion of the RF spectrum, originally rameters.” allocated in many smaller bands exclusively for specific uses, to be effectively shared with low-power UWB radios. 2. OVERVIEW OF UNIQUE FEATURES AND ISSUES The FCC regulations classify UWB applications into sev- eral categories (see Table 1)withdifferent emission regula- Communication engineers know that there are several com- tions in each case. Maximum emissions in the prescribed pelling advantages to having more RF bandwidth. In all of the ff cases below, increasing RF bandwidth improves a desirable bands are at an e ective isotropic radiated power (EIRP) 1 − . − property. Many assumptions and simplifications are hidden of 41 3 dBm per MHz, and the 10 dB level of the emis- ffi sions must fall within the prescribed band (see Figure 1). In in these relations that may be di cult to justify rigorously for addition, for a radiator to be considered UWB, the 10 dB f − f bandwidth H L must be at least 500 MHz, and the frac- 1 B f − f / f f . Note that the gross bandwidth parameter RF used in the relations of tional bandwidth, 2( H L) ( H + L), must be at least 0 2, this section is defined differently in each case, and numerical values of one as determined by the −10 dB power points fH and fL (see [8, bandwidth measure cannot legitimately be substituted for another band- paragraph 30]). UWB modulations are not prescribed by this width measure when performing high-level tradeoffs[10]. 254 EURASIP Journal on Applied Signal Processing
UWB systems. For example, antennas behave like direction- in a noise-like manner. The data detector recovers all of the sensitive filters over ultra-wide bandwidths, and the signal signal power, but the total noise PSD within the data band- driving the transmitting antenna, the electric far field (even width increases from N0 to N0 +(I/BRF). Hence, the effective in free space), and the signal across the receiver load may dif- energy-per-bit-to-noise density ratio is approximately fer considerably in waveshape and spectral content. Ideally ffi E P /R matched correlation receivers are di cult to realize. b = rec data . (2) Ntot ss N0 + I/BRF 2.1. AWGN channel capacity and bandwidth efficiency The channel capacity C (in bits per second (bps)) of the A more detailed performance computation based on spread- band-limited additive white Gaussian noise (AWGN) chan- spectrum processing for interference mitigation in UWB ra- nel increases with RF bandwidth: dios is given in [14]. (ii) Interference excision works by filtering out (rejecting) P C = B rec narrowband interference. Assuming that the received wave- RF log2 1+ ,(1) BRFN0 form uniformly spans the available RF bandwidth and that the interfering signal can be eliminated by ideal notch filter- where B is the RF bandwidth of the channel, P is the re- RF rec ing that removes a fraction 1 − F of the RF bandwidth, the ceived signal power, and N is the noise power spectral den- 0 noise power density in the remaining bandwidth FB is N sity (PSD) in the RF bandwidth of the radio (see, e.g., [11, RF 0 and the remaining signal power is FP . Hence, the effec- Section 5.5]). This equation is based on an idealized rectan- rec tive energy-per-bit-to-noise density ratio for ideal interfer- gular RF filter of width B and does not account for many RF ence excision is approximately effects in real systems, including interference of all sorts, re- ceiver mismatch, and so forth. In the event that the noise E FP /R b = rec data . in the radio receiver is Gaussian but not white, Shannon’s N N (3) water filling theorem [12] indicates that capacity should in tot int ex 0 most circumstances increase with increased bandwidth un- Thus the primary effect of interference excision, in addition der a fixed received power constraint. It also suggests that the to removing the interfering signal, is to reduce the rate at distribution of power which achieves capacity in the band- which the receiver accumulates desired signal energy by a fac- limited AWGN channel corresponds to a flat PSD across the tor F. available frequency band. (iii) Selectable channelization is a form of interference ex- K 2.2. Interference in UWB receivers cision in which the RF bandwidth is divided into approx- imately nonoverlapping subchannels, and only those sub- There is no doubt that UWB radios will be sharing the channels without significant interference are processed to de- environment with other radio systems, some possibly cre- tect the transmitted data signal. This system is equivalent to ating UWB multiple access interference, and others creat- signal excision with a bank of fixed filters, each of bandwidth ing narrowband interference in the UWB radio bands. The BRF/K. When the interference bandwidth fraction 1 − F is FCC regulations have set conditions that limit the interfer- less than 1/K, then excision of the interference can be accom- ence from UWB radiators to other radio systems, by limit- − . plished by not processing one or two subchannels. In either ing UWB radios’ EIRP in any 1 MHz band to 41 3dBm. case, the aggregate signal-to-noise power ratio is the same as However, the issue of eliminating interference to UWB ra- that for signal excision, but the rate at which the receiver ac- dios from other radiating systems is left to the ingenuity of cumulates desired signal energy is reduced by a factor 1/K or the UWB radio designer. At least three standard bandwidth- 2/K when 1 − F<1/K. related approaches to the handling of interference are pos- A similar comparison of the effective energy-per-bit-to- sible, namely, spread-spectrum processing, interference exci- noise density ratios for spread-spectrum processing and in- sion, and selectable channelization. A back-of-the-envelope terference excision yields computation of the effects of these kinds of processing can be accomplished under the assumption that a reasonable model E E N I/B b > b ⇐⇒ 1 > 0 + RF . for the signal in the receiver is the sum of three terms: the (4) Ntot ss Ntot int ex F N0 desired signal with power Prec uniformly spread over the RF bandwidth BRF, an equivalent receiver noise with power den- Hence, the best processing technique is determined by com- sity N0, and a statistically independent interfering signal with paring the increase in observation time required to accumu- power I occupying a fraction 1 − F of the RF bandwidth. late a prescribed amount of signal energy when interference (i) Spread-spectrum processing works by using transmit- is excised to the increase in the receiver’s total noise floor ted waveforms that span the available RF bandwidth BRF when spread-spectrum techniques are used. Typically when as uniformly as possible, the RF bandwidth typically being the remaining band fraction F is large, signal excision will much larger than the data bandwidth Bdata. In the process be preferred, but when most of the band must be excised of ideal correlation reception, the received signal is despread to eliminate the interference and F is small, then spread- and the data recovered within a bandwidth Bdata at a rate spectrum processing will be preferred. The dilemma in ei- Rdata, while at the same time, any interference power I is ther comparison is that the designer does not usually know spread more or less uniformly across the RF bandwidth [13] the received interference power I and/or F apriori. Ultra-Wideband Radio 255
It is worth noting that as the RF bandwidth BRF in- Hence, for a specified correlation time Tco,perhapsdeter- creases in a shared spectrum environment, the likelihood mined by the requirement to collect specified amount of sig- of having more in-band interferers may increase. Signal ex- nal energy, increases in the RF bandwidth provide more se- cision of some form may be necessary for narrowbands in vere constraints on oscillator stability S. which strong interference is normally expected, and spread- spectrum processing may be desirable to handle less pre- 2.4. Adding ultra to wideband dictable and weaker sources of interference. In this case, strong narrowband interference is first excised, and the A large RF bandwidth by itself does not imply that a system remaining signal is subject to spread-spectrum processing. If is UWB. The FCC definition that a UWB signal have a frac- tional bandwidth of at least 0.2 means that of all possible sys- the interference can be completely excised, there is no added f − f benefit to the spread-spectrum processing. tems with the same bandwidth H L, those qualified as UWB have the lowest center frequencies ( fH + fL)/2. The rel- 2.3. Time resolution atively low-frequency band of UWB systems provides propa- gation advantages through many materials (see Figure 2)and ThetimeresolutionTres of a matched receiver generally is on the order of the reciprocal of the RF Gabor (RMS) bandwidth motivates the imaging applications in the FCC regulations. Large fractional bandwidths do cause implementation BRF: problems for system architectures, antennas, and circuits. 1 However, these problems can be overcome. The fundamental Tres ≈ . (5) BRF parameters that control UWB radio design are the character- istics of the channel: propagation effects, interference, and This gives a corresponding range resolution in positioning regulatory constraints on transmission. systems on the order of c/BRF,wherec is the speed of light. This time-resolution measure is well known in radar circles as a measure of the width of the peak of a matched-filter 3. MODELING THE RF CHANNEL response to a waveform of RMS bandwidth BRF. Although Woodward’s radar ambiguity function was developed for UWB channels present problems that differ somewhat from time and Doppler mismatch assessment of narrowband re- their narrowband counterparts. We will first explore a few ceiver performance, a corresponding application of this con- candidate antennas, and where analytically/computationally cept to UWB signals can be developed [15]. feasible, describe their distortion effects on the transmission ThesmallvalueofTres also can cause problems for the of a Gaussian monocycle source over a free-space channel. system designer. For example, in an ideal AWGN baseband The construction of a link budget for power or energy trans- channel, the number of measurements used in acquiring syn- mission over a free-space channel is then illustrated with chronization in a straightforward manner (e.g., a serial or both rigorous computations and Friis equation approxima- parallel search) is proportional to Tunc/Tres,whereTunc is the tions.Thenweuserealmeasurementstoillustratethecon- duration of the initial time uncertainty interval that must be siderably more complicated UWB channel structure and a searched in the acquisition process. Rapid acquisition tech- variety of indoor communication channels. niques of various types (see, e.g., [17, Section 6.8]), which usually take advantage of some property of the signal design, 3.1. Antenna design for UWB radio systems have been devised to reduce this quantity to log (T /T ). 2 unc res UWB radio systems are characterized by multioctave to mul- Whatever technique used in the acquisition process, increas- tidecade frequency bandwidths, and are expected to transmit ing the RF bandwidth of the signal generally stresses the syn- and receive baseband pulse waveforms with minimum loss chronization process. and distortion. Both transmit and receive antennas can af- Monostatic radars have the advantage of being able to ac- fect the faithful transmission of UWB signal waveforms be- cess the same clock signals on transmit and receive. Commu- cause of the effects of impedance mismatch over the oper- nication systems must have similar clocks at the transmitter ating bandwidth, pulse distortion effects, and the dispersive and receiver to provide timing structure for digital modu- effects of frequency-dependent antenna gains and spreading lation and demodulation. Differences in these clock periods factors [18, 19, 20, 21]. Some of the desirable antenna char- can be removed by voltage control in the sync-tracking mode acteristics for UWB radio systems are of the receiver, but during the sync acquisition phase in the receiver, differences in the transmit and receive clocks can (i) wide impedance bandwidth; cause problems. If two such clocks start in synchronism and (ii) fixed-phase center over frequency; if the elapsed time until the clocks are out of synchronism (iii) high radiation efficiency. by Tres is Tco, then the time over which correlations can be computed usefully is bounded by Tco. Hence the clock sta- Good impedance matching over the operating frequency bility S in parts per million (ppm) required to do acquisition band is desired to minimize reflection loss and to avoid pulse correlation computations over Tco is distortion. If the phase center (the point where spherical wave radiation effectively originates) of an antenna moves T 106 with frequency (as is the case with spiral, log periodic, S ≈ res × 106 ≈ (ppm). (6) Tco BRFTco and traveling wave antennas), pulse dispersion will occur. 256 EURASIP Journal on Applied Signal Processing
35 × Concrete block Painted 2 6board
Clay brick 30
25
3/4 plywood 20
15 3/4 pine board
Total one-way attenuation (dB) 10 Wet paper towel Glass Drywall 5 Asphalt shingle Kelvar sheet Polyethylene Paper towel (dry) 0 Fiberglass insul. 3 5 8 10 20 30 50 80 100 Frequency (GHz)
Figure 2: Total one-way attenuation through various materials (from [16] with the permission of the International Society for Optical Engineering (SPIE)).
Table 2: Some characteristics of antennas for UWB systems.
Antenna Impedance bandwidth Phase center stability Radiation efficiency Physical size Dipole Narrow Good High Small Loop Narrow Good High Small Bow tie, diamond Medium Good High Medium Vivaldi Wide Good High Large LPDA Wide Poor Medium Large Spiral Wide Poor Medium Large Loaded dipole/loop Medium Good Low Small Bicone Wide Good High Large TEM horn Wide Good High Large
The desire for high radiation efficiency is self-evident, but receive antenna is terminated with load impedance ZL(ω), several types of broadband antennas employ resistive load- with terminal voltage VL(ω). The input impedances of the ing, which reduces efficiency. Other UWB antenna concerns transmit and receive antennas are ZT(ω)andZR(ω), respec- include polarization properties (versus frequency), physical tively. The antennas are separated by a distance r,assumedto size, cost, and feeding techniques (balanced versus unbal- be large enough so that each antenna is in the far-field region anced). Table 2 summarizes several of these key features for of the other over the operating bandwidth. a number of antennas that might be considered for UWB We can define a frequency-domain transfer function that systems. relates the radiated electric field E(ω, r, θ, φ) to the transmit antenna generator voltage as 3.1.1. Transfer function for the radiated field from a UWB antenna E (ω, r, θ, φ) = FEG (ω, r, θ, φ)VG(ω), (7) Consider the canonical UWB radio configuration shown in Figure 3, where the transmit antenna is driven with a voltage where r, θ, φ are polar coordinates with origin at the trans- source VG(ω) having an internal impedance ZG(ω), and the mitting antenna. For most antennas, the transfer function, Ultra-Wideband Radio 257
This comparison shows that the use of the basic Friis formula ZG(ω) can result in link loss errors of more than 60 dB for a UWB V (ω) G E(ω, r, θ, φ) system with severely (impedance) mismatched antennas, but r may give results correct to within a few dB for well-matched ZT(ω) VL(ω) narrowband antennas, or by augmenting the formula with Z ω L( ) an impedance mismatch correction factor. We conclude that Transmit antenna the dominant limitation of the Friis formula when applied to
ZR(ω) UWB systems is not the frequency dependence of the spread- ff Receive antenna ing factor or antenna gain terms, but the broadband e ect of mismatch between the transmit/receive antennas and their Figure 3: Frequency-domain model of transmit and receive anten- source or load impedance. Pulse distortion effects also limit nas for a UWB radio system. the accuracy of the Friis approximation, but to a much lesser degree. FEG (ω, r, θ, φ), must be calculated via numerical electro- 3.2.1. Link loss based on rigorous magnetic techniques (e.g., the moment method or finite- electromagnetic analysis difference technique), as described in [21, 22, 23]. For the The calculation of UWB energy link loss in the general case simple case of an electrically short dipole located on the z- requires a complete transient electromagnetic solution for axis, however, the result can be expressed in a closed form: the transmit and receive antennas to account for the effects jωµ h θ of impedance mismatch over the operating bandwidth, pulse F ω r θ φ = θˆ · 0 · sin · e− jωr/c ff ff EG ( , , , ) πr Z ω Z ω ,(8) distortion e ects, and the e ects of frequency-dependent an- 4 G( )+ T( ) tenna gains and spreading factor [18, 21, 22, 23]. With refer- ence to Figure 3,defineHLG(ω) as the voltage transfer func- where h is the half length of the dipole, µ0 is the permeability of free space, and c is the speed of light. This result shows that tion that relates the receive antenna load voltage to the gen- the radiated electric field is related to the derivative of the erator voltage at the transmit antenna [23, 24]: dipole current, but may be a more complicated function of V ω = H ω V ω e− jωr/c the generator voltage, depending on the particular generator L( ) LG( ) G( ) ,(9) impedance and dipole input impedance functions. c Figure 4 shows the radiated electric field waveforms from where is the speed of light. Note that the exponential factor several types of antennas, for a Gaussian monocycle genera- representing the time delay between the transmit and receive tor waveform. Observe that the electrically short dipole pro- antenna has been separated from the transfer function. Al- vides good pulse fidelity, but at a relatively low amplitude. though not explicitly shown, it should be understood that The resonant dipole provides a higher amplitude, and also this transfer function is dependent on the load impedance, greater duration. The log-periodic dipole array has very good the range between the antennas, and the elevation and az- impedance and gain bandwidth, but the nonconstant phase imuth angles at each antenna. center causes considerable ringing of the radiated field. In The time-domain voltage waveform at the receive an- contrast, the constant phase center of the Vivaldi antenna tenna can be found as produces less ringing, and a very high amplitude pulse. 1 v (t) = H (ω)V (ω)e jωt dω, (10) L 2π LG G 3.2. UWB link budget analysis BW In this section we discuss the energy link loss between the where t = t − r/c is the retarded time variable. transmitter and receiver of a UWB radio system. We assume The energy delivered to the transmit antenna by the here a baseband UWB system using short pulses; the link source is given by loss of a carrier-based multichannel UWB system can gen- 2 erally be well modeled using the traditional Friis equation. 1 V (ω) R (ω) W = G T dω, (11) We first summarize the rigorous calculation of UWB en- in π 2 2 BW ZT(ω)+ZG(ω) ergy transmission based on electromagnetic analysis of tran- ff sient radiation and reception, including the e ects of antenna where RT(ω) is the real part of ZT(ω). The energy received by impedance mismatches, pulse distortion effects, and the ef- the load at the receive antenna is given by fects of frequency-dependent antenna gains and spreading factor. Next we present some closed-form approximations V ω 2 W = 1 L( ) dω. for energy link loss for the special cases of electrically small rec π Z∗ ω (12) 2 BW L ( ) dipole antennas with Gaussian or Gaussian doublet (mono- cycle) generator waveforms. We also consider the applica- The integrations in (10)–(12) are over the bandwidth of the tion of the narrowband Friis transmission formula to a UWB generator waveform. radio system, and compare these results with the rigorous To calculate energy link loss for a specific set of anten- and approximate solutions for several types of antennas. nas and a given generator waveform, the transfer function of 258 EURASIP Journal on Applied Signal Processing
4.0E +5 4.0E +5
2.0E +5 2.0E +5
0.0E +0 0.0E +0 Electric field Electric field −2.0E +5 −2.0E +5
−4.0E +5 −4.0E +5 −50 51015202530 −50 51015202530 Normalized time c(t − r/c)/L Normalized time c(t − r/c)/L
(a) (b)
4.0E +5 4.0E +5
2.0E +5 2.0E +5
0.0E +0 0.0E +0 Electric field Electric field −2.0E +5 −2.0E +5
−4.0E +5 −4.0E +5 −5 0 5 1015202530 −50 51015202530 Normalized time c(t − r/c)/L Normalized time c(t − r/c)/L
(c) (d)
Figure 4: Radiated electric field waveforms for antennas with a Gaussian monocycle generator waveform (T = 4.42 × 10−10 seconds). The field is normalized by the factor rejωr/c. (a) Short dipole (L = 1.0 cm), (b) resonant dipole (L = 15 cm), (c) log-periodic dipole array, and (d) Vivaldi antenna.
(9) is first computed over a range of frequencies that cover 3.2.2. Closed-form approximations for UWB the system bandwidth (as determined by the spectrum of the link loss for short dipole generator waveform). This can be done using a numerical Using reasonable approximations, it is possible to derive electromagnetic analysis techniques. Then the input and re- closed-form expressions for the energy link loss of a UWB ceived energies can be computed using (11)and(12). The radio system using electrically small dipoles or loops, and ei- link loss is defined as the ratio of these two quantities. Note ther a Gaussian pulse or a monocycle generator waveform that this calculation includes polarization mismatch, propa- [24]. These results appear to be the only cases that can be gation losses, antenna efficiency, impedance mismatches, and ff expressed in closed form and are therefore useful for show- waveform distortion e ects. ing the dependence of waveform shape, receiver impedance, For the results that follow, we define a Gaussian generator and gain factors in more general situations. Additionally, the waveform as accessibility of these results should be useful for systems en- 2 2 gineers working with UWB radio technology. v (t) = V e−t /2T (13) G o In the following results, we assume that both transmit and receive antennas are identical, are polarization matched, and a monocycle (Gaussian doublet) generator waveform as and are oriented so that each is in the main beam of the other. t For electrically short lossless dipoles of half length h = L/2 v t = V e−t2/2T2 . G( ) o T (14) and radius a, the energy link loss can be expressed for Gaus- sian and monocycle input waveforms, for either small or Note that the Gaussian pulse has nonzero DC content, al- large values of load resistance, RL, as shown in Table 3. In the though this does not contribute to either the input energy above results, Co =−h/120c[1 + ln(a/h)] is the capacitance W W in or received energy rec for realistic antennas. of the dipole, c is the speed of light, and ηo = 377 Ω is the Ultra-Wideband Radio 259
Table 3: Closed-form approximations for UWB link loss for short transmit and receive antenna power gains, respectively, and dipoles. λ is the wavelength at the operating frequency. Note that this result does not include propagation losses, polarization mis- R R Input waveform Link loss for small L Link loss for large L match, or impedance mismatch at either the transmit or re- W 15η R C h 2 W 3η h2 ceive antenna. Also, the Friis formula, since it applies only to Gaussian pulse rec = o L o rec = o W 16π Tr W 8πr2R the power in CW (sinusoidal) signals, does not account for in in L W η R C h 2 W η h2 pulse distortion effects at either antenna or even the type of rec = 21 o L o rec = 3 o Monocycle 2 Win 16π Tr Win 8πr RL waveform used at the generator. If the transmitted signal consists of digital data at a bit rate Rb bps, then the energies per bit on transmit and re- −55 ceive are Ebt = Pt/Rb and Ebr = Pr/Rb,respectively.Then −60 (15) can be written in terms of the transmit and receive bit energy densities as −65
−70 2 Gt(ω)Gr(ω)λ Ebr(ω) = Ebt(ω) . (16) −75 (4πr)2 −
Link loss (dB) 80 The frequency dependence of each term is explicitly shown −85 in (15)and(16). Note that the spread factor (r/λ)2 has a fre- −90 quency dependence of 6 dB per octave, but this is reduced to a maximum error of 3 dB at either end of an octave band- −95 10 100 1000 10000 100000 width for a single frequency chosen at midband. Similarly,
Receiver load resistance RL (ohms) the frequency variation of antenna gain is typically small over a wide frequency range for many practical antenna elements. Rigorous numerical solution An electrically short dipole antenna, for example, has a gain R Closed form (small L) of about 1.8 dB for all frequencies below resonance. The ef- R Closed form (large L) fect of impedance mismatch can be included (at a particular frequency ω) by multiplying (16) by the factor (1 −|Γ(ω)|2), Figure 5: Comparison of closed form versus exact (numerical) where Γ(ω) is the reflection coefficient at the receive antenna r2 energy link loss (multiplied by ) for a UWB system using two given by electrically short dipoles and a monocycle generator waveform ver- = . sus receive load resistance. Dipole length 1 0 cm, dipole radius Z − Z = . Z = Ω T = . × −10 R L 0 02 cm, G 50 ,and 4 42 10 seconds. Γ(ω) = . (17) ZR + ZL ff impedance of free space. These approximations are accurate Note that the e ect of mismatch at the generator is not W for frequencies up to where the dipole length is less than λ/20. included—this is because we have chosen to use in, the en- Over this range, the input resistance is less than 0.5 Ω, while ergy delivered to the transmit antenna, as opposed to the en- the input reactance is at least several thousand ohms. ergy available from the generator. Figure 5 shows a comparison of the closed-form energy link loss results from Table 3 compared with rigorous data 3.2.4. Examples and comments from a moment method solution for a short dipole, versus To compare specific numerical results, we consider the link load resistance, for the monocycle generator waveform. For loss for three different transmit/receive antenna pairs, with R these parameters, it is seen that the “small L”resultworks Gaussian and monocycle waveforms. Choosing T = 4.42 × R Ω R − well for L up to about 1000 , while the “large L”form 10 10 seconds for both the Gaussian pulse and the mono- works well down to about 20 000 Ω. Optimum link loss is cycle waveforms results in a 10 dB bandwidth of 550 MHz seen to occur between these values. for the Gaussian pulse, and a 10 dB bandwidth of 70 MHz to 790 MHz for the monocycle pulse. The Gaussian waveform 3.2.3. Link loss using the narrowband Friis contains power at very low frequencies (and DC), which is transmission formula not radiated by any of the antennas considered here. The pa- The Friis link equation that applies to narrowband (carrier- rameters for each of the three antennas are given below. based) radio systems is given by [25] An electrically short dipole G (ω)G (ω)λ2 = . = . Z = P (ω) = P (ω) t r , (15) Dipole length 1 0 cm, dipole radius 0 02 cm, and L r t πr 2 (4 ) ZG = 50 Ω. The 10 dB bandwidth for the magnitude of the resulting transfer function is from 10.2GHz to 18.9 GHz. where Pr(ω)andPt(ω) are the received and transmitted pow- This element is severely mismatched over the bandwidth of ers at operating frequency ω rad/s, Gt(ω)andGr(ω) are the either input signal. 260 EURASIP Journal on Applied Signal Processing
Table 4: Normalized (r = 1) energy link loss for various antennas and excitations.
Gaussian rigorous Monocycle rigorous Midband Friis and Antenna Midband frequency Midband Friis ((16)) ((9)–(12), (13)) ((9)–(12), (14)) Z-mismatch Short dipoles −85.5dB −84.0 dB 430 MHz −20.8dB −87.0dB Resonant dipoles −23.9dB −23.9 dB 500 MHz −22.1dB −22.4dB Lossy dipoles −43.1dB −41.8 dB 500 MHz −22.1dB −22.3dB
A resonant dipole (of course, it is generally undesirable to use such narrow- band antennas for a wideband system). For broadband ele- Dipole length = 30.0 cm, dipole radius = 0.02 cm, and ZL = ments, application of the Friis formula with the impedance ZG = 72 Ω. The 10 dB bandwidth for the magnitude of the resulting transfer function is from 410 MHz to 580 MHz. mismatch factor can produce results that are accurate to This is a relatively narrowband element, but is well matched about 3 dB. More complicated elements, such as arrays or ff to the source and load impedances at its resonant frequency traveling wave antennas, will likely lead to di erent conclu- of 500 MHz. sions. In a general sense, the essential problem with short-pulse ff A lossy resonant dipole radio transmission that di erentiates it from a narrowband (CW) system is the distortion introduced by practical trans- Dipole length = 30.0 cm, dipole radius = 0.02 cm, dipole mit and receive antennas. These antennas, which form the in- conductivity = 100 S/m, and Z = Z = 800 Ω.The L G terface between plane waves and circuitry at both the trans- 10 dB bandwidth for the magnitude of the resulting transfer mitter and receiver, are a direct cause of pulse distortion in function is 190 MHz to 990 MHz. This is a broadband ele- a UWB radio system. Fundamentally, this is due to non- ment, and is reasonably well matched to the source and load TEM (reactive) fields in the near zone of each antenna, which impedances over the bandwidth of the input signals. Due to lead to the impedance mismatch terms noted above as well the lossy loading, the efficiency of this element is about 10%. as the radiation mechanism itself. In principle, it is possi- The resulting energy link losses for these antennas are ble to use pure TEM mode antennas (e.g., infinite biconical shown in Table 4. The first two columns of data refer to and TEM horns) to achieve distortionless pulse transmission the rigorous calculation of link loss using a full electromag- and reception, but this is of limited practicality because of netic solution summarized by (9)–(12) for the Gaussian and the large sizes required for such antennas to avoid end reflec- monocycle input pulses. These solutions include all relevant tions. effects, including impedance mismatch, pulse distortion, and Finally, we note that the fact that the overall energy link frequency variation of gain and propagation factors. Observe loss in a UWB system depends on generator pulse shape im- that the link loss differs by a few dB for the two different in- plies that it is possible to optimize receive pulse amplitude or put pulses when broadband elements are used (short dipoles energy by proper generator waveform selection. Such opti- or lossy dipoles). In contrast, waveform shape has little ef- mal waveforms can be derived for a specific set of transmit fect on link loss when the antennas are relatively narrowband and receive antennas, and can result in an improvement of (resonant dipoles), since the relatively narrow portion of the several dB over the results obtained with Gaussian or Gaus- input spectrum that is passed by the antennas results in an sian monocycle waveforms [23]. Although it is generally not essentially sinusoidal waveform. practical to implement a UWB system with these optimal The remaining three columns present data associated waveforms, such results are useful because they set an upper with the Friis formula of (16). The midband frequency is limit on the performance of the system. the frequency at which the calculation is performed, and has been selected to be at the maximum response of the associ- 3.3. UWB propagation ated transfer function (for the resonant and lossy dipoles), or near the midband of the input waveform bandwidth (for In principle, one technique for evaluating a UWB channel the short dipoles). The gain for each antenna was assumed is to drive the transmitting antenna with a short pulse and to be constant at 1.8 dB. Note that using the basic Friis for- record the response on a digital oscilloscope (see the UltRa mula without impedance mismatch correction gives an er- Lab website). We can illustrate this in Figure 6 with a set of ror of more than 60 dB when the antennas are severely mis- measurements made in several different environments with matched (short dipoles) but gives results within a few dB the same pulse generator and antennas. Notice first that the of the correct result for narrowband matched antennas (the transmitted waveform in Figure 6a is a short pulse on the resonant dipoles). If the efficiency of the lossy dipoles is in- order of a half nanosecond in duration, so these illustrative cluded in the Friis calculation (10% efficiency, or 20 dB loss measurements are just below the FCC approved communi- for combined transmit and receive antennas), reasonable re- cations band. The measurements that we have made out- sults (−42.3 dB) are also obtained for this case. doors tend to be relatively free of dense multipath, partly be- We conclude that for narrowband antennas, the Friis for- cause the low-power nature of the signal precluded observ- mula can give results within about 1 dB for UWB systems able reflections from distant objects. One such measurement, Ultra-Wideband Radio 261 shown in Figure 6b, displays the direct path response to the ten difficult to determine if two signal components that are transmitted pulse in the first four (approximately) nanosec- close in time and angle are caused by distinct propagation onds of the response function, indicating the filtering effect effects (i.e., distinct propagation paths) or are caused by that the antenna system has on the transmitted signal. filtering effects that occur during reflections from a given Indoor communications (see Figures 6c and 6d)clearly object or transmission through a given object. Hence, the demonstrates dense diffuse multipath effects with decay time word path must be used with care in referring to signal constants of the response envelope being on the order of 50 decompositions. to 100 nanoseconds in an office/lab building. In the extreme Most of the indoor channel measurement work of which environment of the empty hold of a cargo ship (Figures 6e we are aware has been performed with identically polarized and 6f), the decay time constants exceeded one microsecond, antennas. It should be noted here that cross-polarization ef- indicating extreme RF echoing in the metallic hold of the fects are not included in these models, and more generally, ship. These shipboard measurements were made over longer there is no reliable way to substitute one antenna type or ori- distances than the other measurements and are somewhat entation for another and expect to accurately predict, in de- noisy (see http://ultra.usc.edu/new site/experiments.html). tail, the effect of the exchange from the original indoor chan- When there is an unobstructed line of sight (LOS) between nel measurement. the transmitting and receiving antenna (e.g., see Figures 6c and 6e), the initial part of the response function, correspond- ing to the direct propagation path, is usually the largest in 4. UWB CIRCUIT AND ARCHITECTURE CHALLENGES amplitude, making the direct path response readily identifi- ffi able for accurate ranging purposes. In situations in which the Although great headway has recently been made in e cient implementation of narrowband radios, the UWB signal has direct path is blocked by objects that may reduce or eliminate ff the direct path signal, there is usually an initial growth to the fundamentally di erent signal characteristics, making exist- response envelope before a decay, as illustrated in Figures 6d ing receiver circuits and architectures ill-suited for UWB use. and 6f. The signal bandwidths and fractional bandwidths of UWB Statistical models for UWB propagation [26, 27, 28, 29] radio are at least an order of magnitude greater than those have been developed for indoor channels, and UWB mea- of existing narrowband radios. Furthermore, the UWB radio surement databases currently are available over the internet must coexist with many other narrowband systems transmit- ting and receiving in the same bandwidth. Consequently, a (see http://ultra.usc.edu/new site/database.html). The IEEE ff 802.15.3a standardization group, charged with developing a UWB radio must have intrinsically di erent sensitivity, se- lectivity, and bandwidth requirements, which motivate radio UWB standard for personal area networks, has developed ff four UWB indoor channel models to support its evaluation circuit and architectural designs that are substantially di er- of proposed UWB signaling standards. The development of ent from their narrowband counterparts. these models and the details of their structure are described In a UWB receiver, the analog-to-digital converter (ADC) in [30]. These models are quite similar to the cluster models can be moved almost up to the antenna, resulting in a dra- of Saleh and Valenzuela [31]. matic reduction of the required analog circuitries, which of- Theanalysisofarraysofmeasurementsin[26]useda ten dominate the size, power, and cost of a modern receiver. version of the CLEAN algorithm to determine the angles A block diagram of the UWB receiver is shown in Figure 7. of arrival, as well as propagation delay and amplitude, for Critical to this design approach, however, is the ability for the each signal component. With the addition of the angle pa- ADC to efficiently sample and digitize at least at the signal rameter, it is possible to break up the collection of signal Nyquist rate of several GHz. The ADC must also support a components into disjoint groups (clusters) sharing similar very large dynamic range to resolve the signal from the strong angles of arrival and propagation delays. The earliest sig- narrowband interferers. Currently, such ADCs are far from nal component in a cluster (the cluster leader) serves as a being practical. reference from which to measure the relative angle of ar- The bandwidth and dynamic range requirements of the rival, amplitude, and delay of other cluster components. One UWB radio appear to have led to two alternative develop- can presume that the components in the cluster followed ment paths. In the first, the UWB system is scaled down to roughly the same path to the receiver, and that the gross operate at a greatly reduced bandwidth, compromising the properties of the path (major attenuation, e.g., caused by benefits of the UWB radio. An example of such a system is walls, and delay components) are captured by the attenu- the time-frequency interleaved (TFI) OFDM radio proposed ation and delay of the cluster leader. Deviations of other as a possible candidate for the 802.15 WPAN standard [33]. cluster components parameters from those of the cluster In the other development path, receiver functions such as leader are quite regular, with angle deviation having a trun- correlation are performed in the analog domain before dig- cated two-sided Laplacian distribution [26, 32], and rela- itizing at a much reduced sampling frequency. Such analog tive amplitude being roughly Rayleigh, the latter choice of receivers are less flexible and suffer from circuit mismatches distribution being as much for simplicity as accuracy. It is and other nonidealities. These circuit nonidealities limit the also worth noting here that while the signal with unob- number of analog correlators that can be practically realized structed propagation path from the transmitter to the re- on an integrated circuit (IC). Since many correlators are re- ceiver may be referred to as the direct-path signal, it is of- quired to exploit the diversity available in a UWB system, 262 EURASIP Journal on Applied Signal Processing
4.5 4.5 4 4 3.5 200 ns 3.5 3 3 1ns 2.5 2.5 2 2 1.5 1.5 1 1 Amplitude (V) Amplitude (V) 0.5 0.5 0 0 −0.5 −0.5 800 1200 1600 2000 2400 953 955 957 959 961 Time (ns) Time (ns)
(a) Pulse driving transmitting antenna.
1.5 1.5 1 20 ns 1 1ns 0.5 0.5 0 0
−0.5 −0.5 Amplitude (mV) Amplitude (mV) −1 −1 −1.5 −1.5 980 1020 1060 1100 994 996 998 1000 1002 Time (ns) Time (ns)
(b) Typical outdoor received signal.
150 150 100 20 ns 100 1ns 50 50 0 0
−50 −50 Amplitude (mV) Amplitude (mV) −100 −100 −150 −150 980 1020 1060 1100 981 983 985 987 989 Time (ns) Time (ns)
(c) Typical indoor received signal, clear line-of-sight.
15 15 10 50 ns 10 1ns 5 5 0 0 −5 −5 Amplitude (mV) −10 Amplitude (mV) −10 −15 −15 1000 1100 1200 984 986 988 990 992 Time (ns) Time (ns)
(d) Typical indoor received signal, blocked line-of-sight.
Figure 6: Transmitted signal and typical received signal waveforms in five different environments. The received signal in each environment is shown in two different time scales: the left plot encompasses the response decay time, and the right plot is set at one nanosecond per division to show the leading edge of the response. Ultra-Wideband Radio 263
6 6 4 200 ns 4 1ns 2 2 0 0 −2 −2 −4 Amplitude (mV) Amplitude (mV) − −6 4 −8 −6 2200 2600 3000 3400 3800 2076 2078 2080 2082 2084 Time (ns) Time (ns)
(e) Cargo-hold received signal, clear line-of-sight.
2.5 2 2 200 ns 1.5 1.5 1ns 1 1 . 0 5 0.5 0 −0.5 0 −1 −0.5 Amplitude (mV) −1.5 Amplitude (mV) −1 −2 −2.5 −1.5 2200 2600 3000 3400 3800 2070 2072 2074 2076 2078 Time (ns) Time (ns)
(f) Cargo-hold received signal, blocked line-of-sight.
Figure 6: Continued.
4.1. Digital receiver architectures Digital Since designing a single ADC to operate at the signal Nyquist Decoded signal ADC data rate is not practical, parallel ADC architectures with each processing ADC operating at a fraction of the effective sampling fre- LNA VGA quency need to be employed. To sample at a fraction of the ff Figure 7: Block diagram of a UWB receiver. e ective sampling frequency, the received UWB signal can be channelized either in the time or frequency domain. An approach that has been used in high-speed digital sampling analog receivers generally do not perform well. These circuit oscilloscopes is to employ an array of M ADCs each triggered nonidealities also preclude the use of sophisticated narrow- successively at 1/M× the effective sample rate of the parallel band interference suppression techniques, which can greatly ADC. A fundamental problem with an actual implementa- improve the receiver performance in environments with tion of such time-interleaved architecture is that each ADC large narrowband interferers such as in UWB systems. To sees the full bandwidth of the input signal. This causes great achieve high reception performance, therefore, the UWB sig- difficulty in the design of the sample/hold circuitry. Further- nal needs to be digitized at the signal Nyquist rate of several more, in the presence of strong narrowband interferers, each GHz, so that all of the receiver functions are performed digi- ADC requires an impractically large dynamic range to resolve tally. In addition, as digital circuits become faster and denser the signal from the narrowband interferers. with constant scaling of CMOS technology, simplifying the Instead of channelizing by time-interleaving, the received analog circuits as much as possible and distributing as much signal can be channelized into multiple frequency subbands of the analog operations to the digital domain should prove with an ADC in each subband channel operating at a frac- more beneficial. tion of the effective sampling frequency. The received UWB There are numerous implementation challenges in the signal is split into M subband signals using M analysis fil- UWB radio. Chief among them are the extremely high- ters. The resulting signals are sampled at feff/M,where feff sampling ADC and the wideband amplification require- is the effective sampling frequency of the receiver, and dig- ments, both of which are described in the following sections. itized using M ADCs. Based on the frequency-channelized Other design challenges include the generation of narrow signals, the receiver performs all of the receiver functions in- pulses at the transmitter and digital processing of the received cluding narrowband interference suppression and cancella- signals at high clock frequencies. tion of aliasing from sampling. 264 EURASIP Journal on Applied Signal Processing
Decoded Signal H(jΩ) ADC Σ data fADC w0[l] H(jΩ) ADC Re{·}
f w1[l] exp{− j2πfADCt} ADC ......
H(jΩ) ADC Re{·} w l fADC M−1[ ] exp{− j2π(M − 1) fADCt}
Figure 8: Block diagram of a frequency-channelized UWB receiver.
An important advantage of channelizing in the frequency Infinite b domain, instead of in the time domain, is that the dy- −10 namic range requirement of each ADC is relaxed, since the frequency-channelization process isolates the effects of large − narrowband interferers. The sample/hold circuitry, however, 15 is still very difficult to design as it sees the uppermost fre- quency in the high-frequency subband channels. In addition, −20 b = 7 sharp bandpass filters with high center frequencies, which are necessary to mitigate the effects of strong narrowband inter- ferers, are extremely difficult to realize, especially in ICs. SNR (dB) −25 b = 4 Instead of using bandpass filters with high center fre- M quencies, channelization can be achieved using a bank of −30 mixers operating at equally spaced frequencies and M low- pass filters to decompose the analog input signal into M −35 subbands. The lowpass channelization can also be achieved 12345678910 by cascading a bank of mixers and lowpass filters. A block Filter order for H(jΩ) diagram of a frequency-channelized receiver is shown in Figure 8.Atotalof2M − 1 ADCs, each operating at fADC, Time-interleaved is employed to achieve an effective sampling frequency of Frequency-channelized (2M − 1) fADC. In addition to obviating the need to design high frequency bandpass filters, channelizing the received Figure 9: SNR versus lowpass filter order. signal using this approach greatly relaxes the design require- ments of the sample/hold circuitry. The sample/hold cir- cuitry in this lowpass channelized architecture sees only the This improvement saturates when the filter order is approxi- bandwidth of the subband signal; whereas in the bandpass mately four for 7 bits and eight for 4 bits, which correspond channelization approach, the sample/hold circuitry sees the to performance improvements of roughly 7 dB and 20 dB, re- uppermost frequency in the high-frequency subbands. spectively, compared to the time-interleaved receiver. When no narrowband interferers are present, the per- This large performance difference arises for sharper fil- formance of the time-interleaved and frequency-channelized ters because the frequency-channelization process better iso- receivers are nearly identical. However, when finite resolu- lates the effects of the narrowband interferer by raising the tion ADCs are employed in the presence of large narrowband quantization noise floor mostly in the subband channels con- interferers, the frequency-channelized receiver significantly taining the interferers. Since significant interference noise is outperforms the time-interleaved receiver. This effect is il- already present in these subband channels, the additional lustrated in Figure 9, which plots the SNR after correlation quantization noise does not greatly increase the total noise of a single pulse versus the filter order of the lowpass filter power relative to the signal power. By contrast, a narrow- H(jΩ). The frequency-channelized and time-interleaved re- band interferer in the time-interleaved receiver increases the ceivers each employ nine ADCs. The narrowband interferer quantization noise floor across the entire signal spectrum. centered at the peak of the signal spectrum is assumed to Thus, even in frequencies with no interference, the quanti- have a bandwidth of 15% of the signal bandwidth and mag- zation noise floor is significantly raised relative to the signal nitude of 50 dB greater than the white noise floor. When in- power, resulting in large overall performance degradation. finite bit resolution is available, the filter order does not af- The ability to isolate the narrowband interferer significantly fect the receiver performance. However, when there are only improves the performance of the frequency-channelized re- a finite number of bits in the ADC, the performance of the ceiver compared to the time-interleaved receiver, especially frequency-channelized receiver improves steadily compared when low resolution ADCs are employed for complexity rea- to the time-interleaved receiver as the filter order increases. sons [34]. Ultra-Wideband Radio 265
The frequency-channelized receiver requires accurate The amplifier that sets the NF of the overall receiver is knowledge of the transfer functions of the analog filters. the low-noise amplifier (LNA), which is the first amplifier in Since exact knowledge is not available in practice due to the receive chain. Since the LNA interfaces with the exter- fabrication process and temperature variations, adaptive nal world, an additional constraint on the input impedance techniques are needed. An adaptive frequency-channelized is typically placed on the LNA. Designing the LNA to min- receiver, however, suffers from poor convergence speed imize the effective NF while satisfying the input impedance compared to an ideal full band receiver because of the in- requirement across the entire frequency band of interest is creased number of parameters to estimate and the slow con- in general difficult, especially when CMOS technology with vergence rates of the adaptive cross-filters that eliminate sub- on-chip passive components is employed to achieve high lev- band aliasing. This poor convergence speed can be problem- els of integration and lower cost. Most of the existing work atic in time-varying wireless systems such as in UWB. In sev- on CMOS-LNA has concentrated on optimizing at a single eral UWB systems, however, the frequency-channelized re- frequency, which is accurate and yield good performance in ceivercanbemadetoachieveconvergencespeedscompa- narrowband communication systems. For a UWB system, rable to a full band receiver. Examples of such systems are however, the UWB nature of the signal clearly invalidates described in [35, 36, 37]. the single-frequency assumption. Existing CMOS-LNA de- sign techniques therefore must be extended to efficiently sup- 4.2. Wideband amplification port a wideband source. Recently, several wideband CMOS- The performance of an amplifier is generally quantified us- LNA designs have been proposed [39, 40]. ing the noise factor (or noise figure (NF) in dB), which is defined as the ratio of the signal-to-noise ratio (SNR) at the input of the amplifier to the SNR at the output of the ampli- 5. UWB SYSTEM CHALLENGES fier. Although the use of the NF metric is straightforward in 5.1. Modulation design: efficiently staying ffi narrowband systems, its use becomes more di cult in UWB within the FCC limits systems. The main difficulty arises in defining the SNR. In a narrowband system, where both the input signal and noise One of the challenges of UWB signal design is to get the max- are assumed to be a single tone at the carrier frequency, the imum signal power to the receiver, while satisfying the FCC SNR is obtained by simply dividing the signal power by the mask on the EIRP of the UWB signal. The structure of this noise power. In a UWB system, however, the input signal is problem is illustrated here. We define the impulse response v t e t r θ φ broadband and the additive noise may be colored. The SNR from source voltage G( ) to the far electric field ( , , , ) obtained by simply dividing the signal power by the total in direction φ, θ at range r to be hEG (t) (with Fourier trans- noise power (whose bandwidth must also be defined) is less form HEG ( f )). The Fourier transform of the electric field meaningful, since a higher SNR value defined in this man- vector is ner does not necessarily translate to a higher receiver perfor- ∞ mance. This is because the performance of the receiver af- E(2πf, r, θ, φ) = e (t, r, θ, φ)e− j2πftdt ter the digital decoding process does not depend on the total −∞ (18) signal and noise power but on the PSD of the additive noise = HEG ( f )VG( f ) volts/meter-Hertz. and the impulse responses of the propagation channel and the transmit pulse. Because of the difficulty in defining the The energy density per unit wavefront area per Hertz at the SNR, existing work on broadband amplifiers defines the NF point (r, θ, φ)is|E(2πf, r, θ, φ)|2/η joules per square-meter- as the weighted average of the single-tone NF (or spot NF). Hertz, where η = 377 Ω. The effective isotropic radiated en- Although such definition of NF is an extension of a single- ergy density of this signal is equal to tone NF, minimizing such arbitrary performance metric does not necessarily improve the overall receiver performance. 2πr2 E(2πf, r, θ, φ) 2 For the NF of the amplifier to be a meaningful metric in η a UWB receiver, the SNR at the input and output of the am- (19) 2 2πr 2 2 plifier should measure the achievable performance after the = · H ( f ) · V ( f ) . eventual digital decoding process because it is ultimately the η EG G most relevant measure of performance. Hence, the SNR is v t defined as the matched-filter bound (MFB), which represents If instead the source voltage G( ) is a finite power signal with S f 2 ff an upper limit on the performance of data transmission sys- PSD G( ) volts per Hertz, then e ective isotropic radiated tems. The MFB is obtained when a noise whitened matched power density equals filter is employed to receive a single transmitted pulse. By 2πr2 defining the SNR as the MFB, the NF measures the degree · H ( f ) 2 · S ( f ). (20) of degradation in the achievable receiver performance caused η EG G by the amplifier. This NF is referred to as the effective NF. The effective NF can be shown to be the weighted harmonic mean The FCC regulation states that the EIRP measured in dBm of the single-tone NF, where the weights are proportional to in every 1 MHz bandwidth must be less than a bounding the signal spectral density [38]. mask M( f ). Hence, a reasonable model for satisfying this 266 EURASIP Journal on Applied Signal Processing requirement is that modulation switched on and with data modulation switched off,mayberequired. 6 2 fo+0.5×10 2πr 2 Determining compliance of a design with the EIRP mask 10 log HEG ( f ) · SG( f )df +30 10 η 6 fo−0.5×10 analytically requires separately substituting (24)and(25) into (21) and checking to see that the relation (21)issatisfied 2If this efficient design assumption is not valid or if the messages are not 3 selected independently, see [10] for spectral calculations. Here hEG(t) can be viewed as any component of hEG (t). Ultra-Wideband Radio 267 0.2 0 −0.2 −0.4 Amplitude (V) −0.6 −0.8 −1 100 200 300 400 500 600 700 800 900 1000 Time sample Original Digitized (a) 0 0 −20 −20 Spectra (dB) Spectra (dB) −40 −40 −60 −60 00.511.52 00.511.52 Frequency (GHz) Frequency (GHz) Radiated field Radiated field Received voltage Received voltage (b) (c) Figure 10: Quantization and spectral occupancy effects for a bow-tie antenna link. (a) Continuous and quantized source signal. (b) Nor- malized energy spectral densities of the radiated field (solid trace) and received signal (dashed trace) for the continuous source signal. (c) Normalized energy spectral densities of the radiated field (solid trace) and received signal (dashed trace) for the quantized source signal. Designing synchronous orthogonal signals are received, then The requirement for synchronous orthogonal signals is typi- ∞ cally needed for data modulation so that a receiver can sepa- vLm(t) ⊥ vLn(t) ⇐⇒ vLm(t)vLn(t)dt rate data symbol components using correlation techniques. −∞ H f ∞ Suppose that LG( ) is the system function of the linear (m) (n) ∗ 2 2 = ∆ ( f ) ∆ ( f ) PG( f ) HLG( f ) df = 0, transformation from the source voltage vG(t) to the receiver −∞ load voltage vL(t). In this situation when vGd(t)andvLd(t) (26) 268 EURASIP Journal on Applied Signal Processing and this criterion cannot be simplified for close-packed pulse 100 designs in which interpulse interference occurs in the re- ceiver. 10−1 Designing asynchronous quasiorthogonal 10−2 signals for multiple access Pulse patterns can also be used in multiple-access systems 10−3 for separation of signals from different transmitters. Since −4 the transmitters clocks typically are not synchronous, asyn- Error probability 10 chronous signal designs similar or identical to those used for direct-sequence spread-spectrum multiple access (DS- 10−5 SSMA) must be used. It is quite likely that the models from the source voltages in these transmitters to the receiver load 10−6 voltage will differ in some ways. In any event, the correla- 01020304050 tion properties of the received signals will depend on the ba- SNR at 1 meter (dB) sic pulse-shape voltages across the receiver load, as well as L = 1 L = 32 the outputs of the transmitters’ pulse pattern generators. For L = 2 L = 64 more on DS-SSMA signal designs, see [42], and for time- L = 4 L = 128 hopping variations of these designs, see [41]. L = 8 L = 1(at1m) L = 16 5.2. Receiver design: robust algorithms The indoor environments in which UWB communication Figure 11: Performance of optimal selective RAKE combiner, as a L applications are contemplated often exhibit pulse response function of the number of signal components combined. functions like those of Figure 3, that is, the delay spread of the multipath channel is much larger than the resolution ca- The quantity ηcap (ηcapSR for the SR receiver) is called the en- pability of the signal being employed. In this situation, the ergy capture efficiency of the receiver. As the energy capture resolved paths do not interfere with each other and the ef- efficiency ηcap decreases for a given available energy Ep per fects of fading are reduced (e.g., see [43, Table 1]).4 How- pulse, the probability Perr of error increases. ever, the UWB receiver must be designed to learn and track changes in the pulse response function and use that informa- 5.2.1. Selective RAKE receivers tion to efficiently demodulate the signal. This can be done A full RAKE receiver uses the signal components from all with a baseband version of a RAKE receiver which combines propagation paths, while a selective RAKE receiver performs the signals coming over resolvable propagation paths in a this process on subset of the component signals. The per- way that maximizes the energy-per-bit-to-noise density ra- formance of a selective RAKE receiver that optimally com- tio. Suppose that r(t) represents a received pulse waveform, bines L paths can be calculated from measurements using and r (t) represents the pulse template to which a correla- temp (27)and(28) (see Figure 11). The receiver parameter L af- tion detector is matched. Assume that antipodal modulation fects the performance because it affects the energy capture ef- of N pulses is used to carry one data bit and that the pulses p ficiency. The curves in Figure 11 are constructed from a set of are far enough apart in time to prevent interpulse interfer- measured pulse response functions corresponding to trans- ence in the receiver. Then it can be shown that the uncoded missions from a laboratory to an office across a hall. The path bit error probability for this stored-reference (SR) correlation strengths and delays for a measurement were determined by receiver is a basic CLEAN algorithm similar to the one mentioned in 2E N η Section 3.3, and the effective template used in the CLEAN al- P = Q p p capSR , (27) r t eSR N gorithm was the same temp( ) that was assumed to be used 0 L = by the pulse correlator ( 1) in the receiver. Note that −1/2 ∞ 2 ff where Q(x) = (2π) −∞ exp(−z /2)dz, and generally the di erence between using the strongest component path L = ( 1) and essentially the whole pulse response function ∞ 2 L = −∞ r(t)rtemp(t)dt ( 128) is roughly 10 + dB in bit-energy-to-noise power η = ∞ ∞ , cap r2 t dt r2 t dt density ratio for reasonable operating points. This indicates −∞ ( ) −∞ temp( ) (28) η < . ∞ that capSR 0 1 for the set of measurements used to generate 2 Ep = r (t)dt. Figure 11. −∞ 5.2.2. Transmitted-reference systems An alternative to a RAKE receiver’s complexity is provided 4It should be noted that while the UWB signal fades caused by antenna movement are not large in most propagation environments, narrowband by a simple transmitted-reference (TR) system [44, 45]in interference in the UWB receiver can fade or strengthen with relatively small which pulses are transmitted in pairs, one pulse p(t) (the ref- movements of the receiving antenna. erence pulse) transmitted without data modulation, followed Ultra-Wideband Radio 269 0 at a fixed time delay τdel later by a data modulated pulse 10 ±p(t − τdel). Hence, if Np pulses are transmitted per bit, then N /2 serve as reference pulses that are interleaved with N /2 p p 10−1 data-modulated (±1) pulses. The receiver cross-correlates the reference pulse with the following data-modulated pulse, and the results of this correlation are accumulated over the 10−2 N / p 2 pulse-pair transmissions that represent the same data Pe bit. The advantage of this technique is that the noisy received 10−3 reference pulse emulates the whole received pulse waveform, and the TR receiver in principle does not suffer from low − energy-capture efficiency ηcapSR. On the other hand, the gain 10 4 in the energy-capture efficiency of this TR system can be offset by the fact that the reference template is noisy, with 10−5 noise × noise products in the correlator causing performance 6101520253034 degradation. NpEp/N0 The performance of the TR system depends on two other T = N = T = N = parameters that do not appear in the idealized SR system corr 30 ns, s 1 corr 75 ns, s 100 T = N = T = N = performance model, namely, the one-sided RF bandwidth corr 75 ns, s 1 corr 140 ns, s 100 Tcorr = 140 ns, Ns = 1 SR, first path W and the duration Tcorr of the pulse-correlation operation Tcorr = 30 ns, Ns = 100 (which begins with the beginning of the signal component received over the direct propagation path). These parameters Figure 12: Comparison of TR- and SR-UWB systems. Note that limit the noise that appears on both arms of the TR system N E /N = E /N is the bit-energy-to-noise-power density ratio, T p p 0 b 0 correlator, but as corr is reduced below the received pulse and Ns = Np/2 is the number of pulse pairs used to represent a bit. length, the energy capture efficiency ηcapTR of the TR receiver also decreases. The bit error probability for this simple TR system is approximately A comparison of uncoded bit error rate performance L = / for an SR receiver using one signal component ( 1) 2 −1 2 N WT N and a TR receiver is shown in Figure 12. These curves are P ≈ Q 2 0 corr 0 . eTR + basedon(27)and(29) with identical parameters, except Np ηcapTREp Np ηcapTREp for the different energy-capture efficiencies, depending on (29) L in the SR case and Tcorr in the TR case. The perfor- mances shown here are of average over 1000 realizations of Here Tcorr appears explicitly and also implicitly because it af- fects the energy-capture efficiency by limiting the range of channel model CM4 adopted by modeling task force of the integration of the correlator. IEEE 802.15.3 Standards Committee [46]. In this compu- tation, the source pulse-signal model is of the form [1 − 4π(t/σ)2]exp(−2πt2/σ2), σ ≈ 0.29 nanosecond, and W = 5.2.3. Comparisons of TR and SR systems 2 GHz. If the pulse is modified to be narrower in time, we A rough comparison of simple transmitted- and stored- expect the curves to move to the right for the following rea- reference receivers (TR and SR) can be made using (27)and sons: (a) the TR curves will move to the right because the RF (29), with the result that bandwidth parameter W will increase, and (b) the SR curves will move to the right because more paths will be resolved, E WT and on the average less pulse energy will be collected for a p corr PeTR REFERENCES International Symposium on Antennas and Propagation, vol. 3, pp. 592–595, Columbus, OH, USA, June 2003. [1]T.W.Barrett, “Historyofultrawideband(UWB)radar& [21] L. B. Felsen, Ed., Transient Electromagnetic Fields, Springer- communications: pioneers and innovators,” in Progress in Verlag, New York, NY, USA, 1976. Electromagnetics Symposium (PIERS ’00), Cambridge, Mass, [22] J. G. Maloney, G. S. Smith, and W. R. 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Noronha, T. Bielawa, C. R. Anderson, D. G. Sweeney, Circuits and Systems I: Regular Papers, vol. 51, no. 7, pp. 1330– S. Licul, and W. A. Davis, “Designing antennas for UWB sys- 1335, 2004. tems,” Microwaves & RF Magazine, pp. 53–61, June 2003. [39] S. Andersson, C. Svensson, and O. Drugge, “Wideband LNA [20] W. Sorgel, C. Waldschmidt, and W. Wiesbeck, “Transient for a multistandard wireless receiver in 0.18 µm CMOS,” in responses of a Vivaldi antenna and a logarithmic periodic Proc. Conference on European Solid-State Circuits (ESSCIRC dipole array for ultra wideband communication,” in IEEE ’03), pp. 655–658, Estoril, Portugal, September 2003. 272 EURASIP Journal on Applied Signal Processing [40] A. Bevilacqua and A. M. Niknejad, “An ultra-wideband IEEE Signal Processing Society, the 1997 Ellersick Award from Mil- CMOS LNA for 3.1 to 10.6 GHz wireless receivers,” in Com, the 2003 S. A. Schelkunoff Prize from the IEEE Antennas IEEE International Solid-State Circuits Conference (ISSCC ’04), and Propagation Society, and Best Paper Award at the 2003 Inter- vol. 1, pp. 382–533, San Francisco, CA, USA, February 2004. national Workshop on Ultrawideband Systems, the latter two for [41] R. A. Scholtz, P. V. Kumar, and C. Corrada-Bravo, “Some works on UWB radio. In 2001, he received the Military Communi- problems and results in ultra-wideband signal design,” in Se- cations Conference Award for Technical Achievement. He has been quences and Their Applications (SETA ’01), Bergen, Norway, General Chairman of six workshops in the area of communications, May 2001. including most recently two ultra-wideband radio workshops held [42] M. Simon, J. Omura, R. Scholtz, and B. Levitt, Spread Spec- in May 1998 and October 2002 (the latter jointly with Intel). trum Communications Handbook, McGraw-Hill, New York, NY, USA, 1994. David M. Pozar received the Ph.D. degree [43] R. A. Scholtz and M. Z. Win, “Impulse radio,” in Wire- from Ohio State University, and joined the less Communications: TDMA versus CDMA,S.Glisicand faculty at the University of Massachusetts in P. Lepp anen,¨ Eds., pp. 1677–1683, Kluwer Academic Publish- 1980. In 1988, he spent a sabbatical leave at ers, Boston, Mass, USA, September 1997. Ecole´ Polytechnique Fed´ erale´ de Lausanne, [44] C. K. Rushforth, “Transmitted-reference techniques for ran- Lausanne, Switzerland. Professor Pozar is a dom or unknown channels,” IEEE Trans. Inform. Theory, vol. Fellow of the IEEE. He has served as an As- 10, no. 1, pp. 39–42, 1964. [45] R. T. Hoctor and H. W. Tomlinson, “An overview of delay- sociate Editor of the IEEE Transactions on hopped, transmitted-reference RF communications,” Tech. Antennas and Propagation (1983–1986 and Rep., General Electric Research and Development Center, 1989–1992), as a Member of the IEEE AP-S Niskayuna, NY, USA, January 2002. AdCom (1989–1991), and as an Associate Editor of the IEEE AP-S [46] J. Foerster, “Channel modeling sub-committee report fi- Newsletter (1982–1984). In 1981, he received the Outstanding Pro- nal,” IEEE P802.15-02/368r5-SG3a, December 2002, http:// fessor for 1981 Award from Eta Kappa Nu, the Student Honor So- ieee802.org/15/. ciety. In 1984, he received an NSF Presidential Young Investigator [47] E. Homier and R. A. Scholtz, “Hybrid fixed-dwell-time search Award, and the Keys to the Future Award from the IEEE Anten- techniques for rapid acquisition of ultra-wideband signals,” in nas and Propagation Society. In 1985, he received the University International Workshop on Ultrawideband Systems (IWUWBS of Massachusetts Engineering Alumni Association Outstanding Ju- ’03), Oulu, Finland, June 2003. nior Faculty Award. In 1986, he received the R.W.P. King Best Pa- [48] E. Homier and R. A. Scholtz, “A generalized signal flow graph per Award from the IEEE Antennas and Propagation Society. In approach for hybrid acquisition of ultra-wideband signals,” 1987, he received the URSI Issac Koga Gold Medal for his work on International Journal of Wireless Information Networks, vol. 10, printed antennas and phased arrays. He again received the R.W.P. no. 4, pp. 179–191, 2003. King Best Paper Award in 1988. In 1989, he received the United [49] J.-Y. Lee and R. A. Scholtz, “Ranging in a dense multipath environment using an UWB radio link,” IEEE J. Select. Areas Technologies Corporation Outstanding Teaching Award. He served Commun., vol. 20, no. 9, pp. 1677–1683, 2002. as a Distinguished Lecturer for the IEEE Antennas and Propaga- [50]J.Foerster,E.Green,S.Somayazulu,andD.Leeper,“Ultra- tion Society in 1993–1995. In 1995, he received the College of En- wideband technology for short- or medium-range wireless gineering Outstanding Senior Faculty Award. He received the Col- communications,” Intel Technology Journal, Q2, 2001. lege of Engineering College Outstanding Teacher Award in 1997. In [51]S.Roy,J.R.Foerster,V.S.Somayazulu,andD.G.Leeper,“Ul- 1998, he received the H. A. Wheeler Applications Prize Paper Award trawideband radio design: the promise of high-speed, short- from IEEE Antennas and Propagation Society. He received an IEEE range wireless connectivity,” Proc. IEEE,vol.92,no.2,pp. Third Millennium Medal in 2000. In 2003, he received the S. A. 295–311, 2004. Schelkunoff Transactions Prize Paper Award from the IEEE Anten- [52] W. Hirt, “Ultra-wideband radio technology: overview and fu- nas and Propagation Society, as well as the Chancellor’s Medal from ture research,” Computer Communications,vol.26,no.1,pp. the University of Massachusetts, and was a Distinguished Faculty 46–52, 2003. Lecturer at the University of Massachusetts. Professor Pozar has [53] USC Communication Sciences Institute, “Workshop on ul- published over 100 papers on microstrip antennas and phased ar- trawideband radio techniques,” Los Angeles, Calif, USA, May rays, and is the author of Microwave Engineering, 3rd edition (Wiley, 1998. 2004), Microstrip Antennas (IEEE Press, 1995), and Microwave and RF Design of Wireless Systems (Wiley, 2000). Robert A. Scholtz is the Fred H. Cole Professor of engineering at the Univer- Won Namgoong received the B.S. degree in sity of Southern California. In 1996, he electrical engineering and computer science founded the Ultrawideband Radio Lab- from the University of California at Berke- oratory (UltRa Lab) there. His research ley in 1993, and the M.S. and Ph.D. degrees interests include communication theory, in electrical engineering from Stanford Uni- synchronization, signal design, adaptive versity in 1995 and 1999, respectively. In processing, pseudonoise generation, and 1999, he joined the faculty of the Electri- ultra-wideband radio. He has coauthored cal Engineering Department, the University Spread Spectrum Communications (Com- of Southern California, where he is an As- puter Science Press), Spread Spectrum Communications Handbook sistant Professor. His current research ar- (McGraw-Hill), and Basic Concepts in Information Theory and Cod- eas include wireless/wireline communication systems, signal pro- ing (Plenum). Dr. Scholtz is a Fellow of the IEEE. He received the cessing systems, RF circuits, and low-power/high-speed circuits. In 1983 Leonard G. Abraham Prize and the 1984 Donald G. Fink Prize 2002, he received the National Science Foundation (NSF) CAREER for the historical article, “The origins of spread spectrum com- Award. He currently serves as the Associate Editor of IEEE Trans- munications.” He is a corecipient of the 1992 Senior Award of the actions on Circuits and Systems I—Regular Papers.