This dissertation has been microfilmed exactly as received 69-15,906
COPELAND, John Raymond, 1933- ANALYSIS OF THE TEM-LINE ANTENNA. The Ohio State University, Ph.D., 1969 Engineering, electrical
University Microfilms, Inc., Ann Arbor, Michigan ANALYSIS OF THE TEM-LINE ANTENNA
DISSERTATION
Presented in Partial Fulfillment of the Requirements for the Degree Doctor of Philosophy in the Graduate School of The Ohio State University
By
John Raymond Copeland, B.E:E., M.Sc.
The Ohio State University 1969
Approved by
Department of Electrical Engineering PLEASE NOTE:
Not original copy. Several pages have indistinct print. Filmed as received.
UNIVERSITY MICROFILMS. ACKNOWLEDGMENT
I t is a pleasure to acknowledge the assistance and encouragement toward the completion of this work by the many members and former members of the ElectroScience Laboratory. Mr. T.E. Kilcoyne, Mr.
C.H. Boehnker, and Mr. J.L. Kohli contributed especially heavily to this work, and i t is particularly gratifying to acknowledge the sug gestions and advice of Professor C.H. Walter.
The work reported in this paper was supported in part by
Contract F 33(615)-67-C -l139 between Air Force Avionics Laboratory,
Air Force Systems Command, Wright-Patterson A ir Force Base, Ohio, and The Ohio State University Research Foundation. VITA
December 4, 1933 Born - Findlay, Ohio
1956 ...... B.E.E., Ohio State University
Columbus, Ohio
1958 ...... M.Sc., Ohio State University,
Columbus, Ohio
1956-1957 ...... Research Assistant, Antenna Laboratory
Ohio State University
1957-1969 ...... Research Associate, ElectroScience
Laboratory (formerly Antenna Laboratory)
Ohio State University
PUBLICATIONS
"Radar Target Classification by Polarization Properties," Proc. IRE,
July 1960.
"Antennafiers and Antennaverters," Electronics, 6 October, 1961.
"A Proposed Lossless Electronic Phase Shifter," IRE Transactions on
Microwave Theory and Techniques, September 1962.
"Integration of Antennas and Circuits," Electronic Industries, May 1963.
"Antennafier Arrays," IEEE Transactions on Antennas and Propagation,
March 1964
i i i PUBLICATIONS (continued)
"Antennafier Arrays for Electronic Beam Control," IEEE Transactions on
Aerospace, April 1964.
"The Slotted TEM-Line Antenna," IEEE Transactions on Antennas and Prop
agation, March 1968.
U.S. PATENTS
3 041 452 - A Tunnel Diode Frequency Conversion Circuit
3 162 855 - Antenna System
3 296 536 - Combined Antenna and Tunnel Diode Converter Circuit
3 349 404 - Integrated Lobe-Switching Antenna
iv TABLE OF CONTENTS
Page
ACKNOWLEDGMENT...... i i
VITA...... i i i
LIST OF TABLES ...... v ii
LIST OF FIGURES...... v iii
Chapter
I., INTRODUCTION ...... 1
A. History 1 1. Low profile antennas 1 2. TEM-line antenna features 2 3. Low-Profile and related antennas 8 a. slot and cavity antennas 6 b. leaky waveguides 9 c. surface-wave antennas 11 d. Franklin antenna 13 e. Dallenbach antenna 14
B. Methods of Analysis 16 C. Applicability of Results 18
I I . BASIC TEM-LINE ANTENNAS ...... 20
A. Physical Characteristics 20 B. Electrical Characteristics 24 1. Radiation patterns 24 2. Brillouin diagrams 32
I I I . ANALYSIS...... 42
A. Radiation Patterns 42 B. Mutual Coupling 48 C. Impedance 51 D. Calculations 55 1. Impedance 57 2. Patterns 58
IV. MEASUREMENTS ...... 70
v Chapter Page
TABLE OF CONTENTS (cont.)
A. Finite Ground Plane 70 B. Voltage-Tuned TEM-Line Antenna 73 C. Five-Element Compact TEM-Line Antenna 77 D. Five-Element Flush-Mounted TEM-Line 84 Antenna E. Other Geometries 89
V. SUMMARY ...... 92
Appendix
A. COMPUTER. CHARACTERISTICS ...... 95
B. THE ASSEMBLER ...... 117
C. LIBRARY INDEX...... 125
D. COMPUTER PROGRAMS ...... 131
REFERENCES ...... 162
Vi LIST OF TABLES
Table Page
1 Central Processor Regi sters ...... 96
2 Beta Codes and Sense Functions ...... 98
3 Overflow Control ...... 100
4 Summary of Computer Instructions ...... 101
5 Instruction Timing ...... 106
6 Informer Control Characters ...... 109
7 Input/Output Instructions ...... I l l
8 American Standard Code for Information Interchange ...... 113
9 Plotter Functions ...... 115
v ii I
LIST OF FIGURES
Figure Page
1 Surface-mounted slotted TEM transmission line radiator...... 3
2 Elementary slot antenna ...... 7
3 Cavity-slot antenna ...... 8
4 T-bar slot antenna ...... 8
5 Leaky waveguide antenna...... 10
6 Franklin antenna, showing alternate-phase suppression ...... 13
7 Dallenbach antenna, showing alternate-phase suppression...... 14
8 Flush-mounted TEM-line antenna ...:...... 22
9 Surface-mounted TEM-line antenna with delay-line loading ...... 23
10 Standing-wave TEM-line distributions 26
11 Array of five isotropic sources along z-axis 28
12 Standing-wave power patterns vs. frequency for five elements spaced 0.2 wavelengths at design frequency ...... 30
13 Standing-wave power patterns vs. frequency for five elements spaced 0.3 wavelengths at design frequency ...... 31
14 Standing-wave power patterns vs. frequency for five elements spaced 0.4 wavelengths at design frequency ...... 33
15 Standing-wave power patterns vs. frequency for five elements spaced 0.5 wavelengths at design frequency ...... 34
v ii i LIST OF FIGURES (continued)
Figure Page
16 Frequency panorama of standing-wave power patterns for five elements spaced 0.25 wavelengths at design frequency ...... 35
17 Brillouin diagram for heavily loaded TEM-line antenna ...... 39
18 Brillouin diagram for polyethylene-loaded TEM-line antenna ...... 40
19 Rectangular HaIf-loop TEM-line radiating element ...... 43
20 Array of N point sources equally spaced along z-axis ...... 46
21 Mutual coupling geometry ...... 48
22 Segment of uniform transmission lin e ...... 51
23 Equivalent TEM-line radiating element ...... 53
24 Input impedance of capacitance-diode-tuned TEM-line antenna, 245 MHz ...... 59
25 Input impedance of capacitance-diode-tuned TEM-line antenna, 250 MHz ...... 59
26 Input impedance of capacitance-diode-tuned TEM—1ine antenna, 255 MHz •••••••*••••••••• 60
27 Input impedance of capacitance-diode-tuned TEM—11ne antenna, 260 MHz•••••••••••••••••• 50
28 Input impedance of capacitance-diode-tuned TEM-line antenna, 265 MHz ...... 61
29 Input impedance of capacitance-diode-tuned TEM-line antenna, 270 MHz ...... 61
ix LIST OF FIGURES (continued)
Fi gure Page
30 Input impedance of capacitance-diode-tuned TEM-line antenna, 275 MHz 62
31 Input impedance of capacitance-diode-tuned TEM-line antenna, 280 MHz 62
32 Input impedance of capacitance-diode-tuned TEM-line antenna, 285 MHz 63
33 Relationship between pattern modes and in put impedance of TEM-line antennas 64
34 Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C = 0 picofarad 66
35 Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C = 1 picofarad 66
36 Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C = 2 picofarad 67
37 Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C = 4 picofarad 67
38 Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C = 8 picofarad 68
39 Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C = 16 picofarad 68
40 Effect of curved ground-plane edges on E-plane power pattern of three-element TEM-line antenna 72
41 Capacitance-tuned three-element VHF TEM-line antenna 74
x LIST OF FIGURES (continued)
Figure Page
42 265 MHz E-plane power patterns of three- element, capacitance-diode-tuned TEM- line antenna...... 75
43 Best attainable VSWR and required voltage vs. frequency for three-element, capacitance- diode-tuned TEM-line antenna...... 76
44 Brillouin diagram for five-element TEM-line antenna with delay-line loading ...... 78
45 Far-field power pattern of compact five-^ element TEM-line antenna, 1.093 GHz ...... 80
46 Far-field power pattern of compact five- element TEM-line antenna, 1.192 GHz ...... 81
47 Far-field power pattern of compact fiv e - element TEM-line antenna, 1.391 GHz ...... 81
48 Input impedance with fixed short position, compact five-element TEM-line antenna (1.07 GHz—1 .332 GHz) ...... 82
49 Input impedance with fixed short position, compact five-element TEM-line antenna (1.202 GHz-1.445 GHz) ...... 83
50 Far-field power pattern of five-element flush-mounted TEM-line antenna, 1.274 GHz ...... 85
51 Far-field power pattern of five-element flush-mounted TEM-line antenna, 1.655 GHz ...... 86
52 Far-field power pattern of five-element flush-mounted TEM-line antenna, 1.762 GHz ...... 87 LIST OF FIGURES (continued)
Figure Page 53 Brillouin diagram for five-element flush-mounted TEM-line antenna...... 88
54 VSWR of five-element flush-mounted TEM-line antenna ...... 89
55 Digital computer block diagram ...... 107
x ii CHAPTER I INTRODUCTION
A. History
1. Low profile antennas
Antennas used on a irc ra ft, missiles, and other aerospace vehicles
capable of high-speed atmospheric flig h t must operate in severe en vironmental conditions. These conditions are imposed by aerodynamic
drag and heating effects on objects protruding into the airflow, by
increased mechanical stress on the antenna through acceleration
loading during maneuvering, and by mechanical vibration over a wide
range of frequencies which quickly produce metal fatigue in objects
improperly constructed with undamped mechanical resonances. All three
of these effects can be relieved greatly by use of flush-mounting or
low-profile antenna elements which minimize the protrusion into the
airflow, and which can be attached rigidly to the airframe for good
mechanical strength and vibration reduction.
Another consideration for aerospace antennas is that in certain
flig h t regimes the airflow over liftin g and control surfaces must not
be perturbed by the antenna structures. Low-profile and flush-mounted
antennas often are designed into these aerodynamic surfaces, however,
and can perform effectively where a protruding antenna could not be
used.
Weight and space limitations comprise a third important area of
1 2 design parameters for aerospace antennas. Here, too, low-profile and flush-mounted antennas can be used effectively; but i t is important
to remember that aerospace vehicles nearly always are very compact with l i t t l e unused internal space. Consequently an antenna which
presents a low profile externally at the expense of occupying a large
volume of internal space in the vehicle is considered disadvantageous.
The TEM-line antenna meets a ll of these requirements, and has the
additional potential of being easily installed on an existing vehicle
with a minimum of modification to the structure of the vehicle. Be
cause of its features, the TEM-line antenna is useful in aerospace
applications of communication, navigation, telemetry, and radar where
narrow-band but tunable antennas are required. These features and
limitations are discussed further in succeeding sections.
2. TEM-line antenna features
The original concept of the TEM-line antenna, as introduced
in 1965, was a low profile antenna composed of a coaxial (or equivalent)
transmission line several wavelengths long, operating in the transverse-
electro-magnetic (TEM) mode, with radiating gaps or non-resonant
slots cut into the outer conductor of the line at periodic intervals
approximating one electrical wavelength in the lin e .[1 ,2 ,3 ,4 ] The mod
ifie d transmission line was to be bonded securely along the length of
its outer conductor segments to an associated conducting ground surface
so that currents flowing into or out of the periodic interruptions in
the line would fringe away from the immediate gap area and, in e ffect,
excite a radiating current distribution on the ground surface itself. 3
A sketch of such a structure appears in Fig. 1.
-GROUND PLANE
SLIDING SHORT TO FEED END
IS
/ s _ /
Fig. 1 . —Surface-mounted slotted TEM transmission line radiator.
If the ground surface were planar in the vicinity of a gap, at least in the areas occupied by most of these fringing currents, and if the conductivity of the surface were high, then the radiation of the surface currents could be treated as if it arose from a mirror image of the exposed current-carrying conductors reflected in the ground plane.[5,6,7,8]
Since the geometry of the transmission line between the radiating sections was relatively unimportant to the radiation properties of the
TEM-line element, except for determining the amplitude and phase of the excitation coefficients of the several radiating sections, the 4 original concept of the TEM-line antenna was broadened subsequently to include the larger class of antennas comprised of an array of electrically small loops, or fractional-loop elements, protruding from a conducting ground surface where these elements were excited through transverse- electro-magnetic mode delay lines such as coaxial cable to obtain the desired amplitude and phase coefficients at each element of the array.
The term "electrically small" in this context is not rig id ly de fined, but may be taken to mean that the greatest dimension of the element does not exceed approximately one-quarter wavelength at the frequency of operation.
Although serial feed arrangements from element to element have received most of the study to date,antennas with parallel feed circuitry cannot be excluded from the class. Thus, the salient features of the class of TEM-line antennas are:
1) an array of electrically small radiating elements distributed
over a ground surface, and
2) an interconnecting feed network made from transverse-
electromagnetic-mode transmission line designed to provide
currents of appropriate amplitude and phase in each radiating
element.
In addition to these characteristics, it is usually necessary to provide a particular termination for the last element in serially fed
TEM-line antennas and to provide suitable impedance transformation to the input terminals, but since these features vary considerably with different designs, they cannot be listed as requisites to the class of 5
TEM-line antennas.
The low-profile nature of the TEM-line antenna, which is one of
its most important characteristics, results from the low profile height
of each of its electrically small elements. Hence, even though the
overall array of elements may be distributed over a length or area of
the ground surface which is large compared to the wavelength, the max
imum protrusion from the surface may be restricted to a distance small
compared to the wavelength. Furthermore, the interconnecting delay
lines can be placed inside or behind the ground surface so that no sur
face protrusion is required except at the locations of the radiating
elements themselves.
In some cases, to be described la te r, even the protrusion of the
elements has been eliminated by mounting them in local recesses (not
resonant cavities) in the ground surface so that no part of the ra
diating elements reached above the level of the surrounding ground
surface.
Hence, although the TEM-line antenna retains the low profile of
its electrically small elements, it is actually a full-sized array in
length, and may extend 0.5 wavelengths or several wavelengths, depending
upon its application. Accordingly, its input impedance and efficiency
are comparable to those of larger, conventional antennas, even though
the bandwidth may be small because the antenna must be operated in a
rather narrow (but tunable) resonance region in the manner of an
electrically small antenna.
This compromise between e le c tric ally small elements and full-sized 6 arrays can be a useful one, however, and many applications have been suggested fo r the TEM-line antenna in areas of airborne communications, radar, and telemetry where sufficient surface area on a vehicle can be provided for a medium-sized antenna (in terms of wavelengths at the operating frequency) but where operational requirements of the vehicle such as high-speed atmospheric flig h t, prohibit significant surface protrusions for reasons of drag and aerodynamic heating effects. Ac cordingly, only the antenna types which offer a low-profile or are
r mounted flush with the surrounding surface can be used in such ap plications, and these types w ill be reviewed in the following para graphs for a comparison with the TEM-line antenna.
3. Low-profile and related antennas
a. slot and cavity antennas
The slot antenna could be termed the ultimate in low-profile antennas, because i t consists of only an aperture in a conducting screen, and therefore has no components which could possibly protrude above the surrounding surface.[9,10] Figure 2 illustrates the simplicity of such an antenna with a coaxial cable feed which might be displaced to ward either end of the slot for impedance matching purposes.
Unfortunately, practical applications of such an elementary slot antenna are rare because radiation is usually required on only one side
of the ground surface, whereas the slot tends to radiate on both sides.
For example, i f the ground surface were the exterior skin of an aero
space vehicle, it would be undesirable for the slot to radiate a portion 7
RECTANGULAR SLOT IN LARGE METAL SHEET
Fig. 2 .—Elementary slot antenna. of its energy into the interior of the vehicle.
Radiation can be contained effectively on one side of the ground plane by use of a cavity as shown in Fig. 3. There are many variations of the cavity-slot antenna, some making use of the sharp frequency- selective resonance obtainable from cavities and others obtaining broadband performance through impedance compensation by special feed geometries. The T-bar slot is an example of the la tte r case, and is
illustrated in Fig. 4.[11]A 2:1 bandwidth (ratio of maximum frequency
to minimum frequency) can be achieved with this design.
I f narrow bandwidth is needed to aid s electivity, a conventional
cavity-slot antenna can be constructed for a half-power bandwidth less
than IX of center frequency by using a relatively small aperture in com- 8
COAX
ADJUSTABLE SHORT
Fig. 3.—Cavity-slot antenna.
65.8cm 5.2cm
5.2cm
I IX 5L2cnn
i4.2cm
Fig. 4 .—T-bar slot antenna. 9 pari son to the cavity volume so that the cavity resonance w ill be loaded only slightly by radiation, thus u tilizin g the well-known frequency-
selective properties of resonant cavities.[12,13]
The prime disadvantage of cavity-slot antennas lies in the size
of the cavity or waveguide assembly required to support the necessary
cavity modes. The characteristic dimension of simple resonant cav
itie s must be on the order of a wavelength to support these modes, thus
limiting their aerospace applications primarily to the microwave spec
trum. In the very-high-frequency (VHF) range and the lower portion of
the ultra-high-frequency (UHF) range,resonant cavities usually are too
large and too heavy to ju s tify their use on an aerospace vehicle,
especially in terms, of the competition for effective u tiliza tio n of
available on-board space and weight by other systems.
Several techniques for sfze reduction of resonant cavities are
available, such as dielectric or magnetic loading, but size reduction
is accompanied by increased weight and electrical losses, so the ob
jections to cavity-slot antennas for low-profile VHF/UHF radiators are
not altered significantly.
b. leaky-wavequides
Arrays of slot antennas can be excited for directional patterns
by placing slots in a waveguide wall and properly locating them with
respect to the electromagnetic fie ld distribution inside the waveguide.
The amplitude and phase of the fie ld coupled through each slot must
be appropriate for the position of that slot element in the array.[14,15,16]
Many variations of this principle have been used successfully at micro- 10 wave frequencies where the size of the waveguide is not a serious drawback.
In some cases the radiation can be made to occur from a continuous aperture instead of a discrete array of slots, as in the trough wave guide shown in Fig. 5. Here dielectric loading has been used to re-
WELDED ALUMINUM LAMINATEO TROUGH POLYSTYRENE SHEET
0.6'
7.3
14.6
Fig. 5 .—Leaky waveguide antenna. duce the size of the waveguide at 330 MHz, but i t can be seen that the structure remained somewhat bulky, in addition to incurring the weight penalty from the rather large quantity of dielectric material required.
Thus, although slot arrays and continuous aperture antennas can be fed from a single waveguide to form low-profile antennas at micro- n wave frequencies, their weight and bulk usually are excessive for aerospace vehicle applications in the VHF and UHF regions of the spectrum.
C. Surface-wave Antennas
Surface waves are characterized by modes obtained from discrete solutions of a vector wave equation subject to the boundary conditions of an exposed unshielded surface, in contrast to the boundary conditions of perfectly conducting walls imposed by conventional shielded wave guides which contain the fields wholly within the guide. The boundary condition required by a surface wave is that the surface impedance must be reactive, viewed at normal incidence. This condition can be obtained
in a variety of ways, but the two most common surface-wave structures are dielectric slabs layered over conducting surfaces,and corrugated
conducting surfaces having many corrugations per wavelength. A long
Yagi-uda array[17] of monopoles can be considered as a special case
of the corrugated surface, where the parasitically excited director
elements are tuned to provide the necessary reactance that supports
the surface wave.[16]
In the case of the corrugated surface, the surface impedance can
be found by treating each groove as a short-circuited parallel-plane
transmission lin e , and since there are many grooves per wavelength,
the surface impedance is approximately the input impedance of the
equivalent transmission line multiplied by the ratio of the area occupied
by the grooves to the total surface area.[18]
Depending upon the design parameters of the surface, the wave 12
can be bound more or less loosely to it. In addition, perturbations
to the surface can be made to induce radiation in a manner very similar
to the manner in which slots in waveguides induce radiation, except
that the waveguide slots are positioned with respect to a traveling wave whose velocity can be either greater or less than the free-space
velocity (depending upon the use of dielectric material and the geom
etry of the waveguide) whereas the surface wave cannot exceed the
free-space velocity without losing its required attachment to the sur
face.
Thus there is some sim ilarity between surface-wave antennas and
the leaky-wave antennas discussed previously, except that the guided
wave is carried along an exposed surface instead of being confined
to the inside of a waveguide as in the case of most leaky-wave antennas.
For this reason, surface^wave antennas can be miniaturized somewhat
through use of high dielectric-constant material to reduce the thick
ness of the dielectric layer or of the grooved surface, and therefore
aerospace applications ranging from the microwave spectrum down through
the UHF range are practical, provided the weight penalty of the die
lec tric can be overcome.
The problems in applying the surface-wave antenna in the VHF range
are primarily those of designing a suitable wave-launching feed which
must couple energy into the surface-wave mode effectively without
substantial direct radiation into free space. The usual methods of
launching the surface wave require a wavelength or more and the dimen
sions of such launchers become prohibitive at VHF. d. Franklin antenna
The Franklin antenna is a coll inear array of half wavelength di poles, end-fed in serial fashion through quarter-wavelength stubs or other reactive devices.[19] Figure 6 illustrates one of several forms
1 i
X 2
i
Fig. 6 .—Franklin antenna, showing alternate-phase suppression.
taken by this antenna. In Europe this antenna also became known as
the Marconi-Frank!in antenna, perhaps because of its early applications
in medium-wave broadcasting where the Marconi type of top-loaded ver
tical antenna has been applied widely.[20] Although the Franklin an
tenna and its relatives cannot be considered low-profile, they were 14 the first to utilize the principle of phasing adjustment in an elec trically long radiator by insertion of delay lines or lumped circuits at periodic intervals, a principle which is used in TEM-line antennas.
This phasing method of the Franklin antenna is retained in many antennas with considerable commercial significance in the fields of mobile communication.[21]
e. Da'llenbach antenna
The Dalienbach antenna shown in Fig. 7. was the f ir s t low-profile
Fig. 7 .--Da'll enbach antenna, showing alternate-phase suppression. 15 antenna to use the technique of alternate-phase suppression which originated with the Franklin antenna 122] The standing wave on the co axial transmission lin e , illustrated schematically to the same scale as the antenna, excites the side-by-side resonant cavities a ll with the same phase. Because the cavity feed points are separated by ex actly one wavelength along the transmission lin e , broadside radiation results.
The excitation amplitude of the individual resonant cavity sections can be adjusted by changing the circumferential cutout angle of the
half-wavelength gap in the outer conductor of the line. Dallenbach suggests that cutout angles smaller than 90° can be used, and that i t also may be beneficial to vary both the characteristic impedance of the transmission line and the axial length of the cutouts.
The cavity depth in the Dallenbach antenna must be approximately
an odd multiple of a quarter-wavelength so that the H01 waveguide mode can be supported with the electric-field lines parallel to the
coaxial transmission-line feed. This establishes a lim it to the low-
profile potential of the antenna, lim iting its aerospace applications
to the UHF range and above, where the size of the resonant cavities
can be accommodated either inside or on the exterior surface of an
aerospace vehicle.
From the foregoing discussions of related low-profile antennas,
i t should be concluded that the TEM-line antenna provides a better
capability for low-profile design than many previous antennas. TEM-
line antennas are more suited to aerospace vehicle uses in the VHF and 16
UHF ranges because they require very l i t t l e internal volume in the ve hicle and in many cases they protrude less from the exterior surface for better aerodynamic heating and drag force behavior at high speeds.
The penalty paid for this size reduction is a relatively narrow, but tunable, bandwidth forced by resonance in the transmission line sections which form the feed structure of the TEM-line antenna. In many applications, however, this feature may be turned to an advantage by improving the system's overall discrimination against undesired signals at nearby frequencies.
The following chapters are devoted to the analysis and evaluation of typical TEM-line antenna designs.
B. Methods of Analysis
The analysis of the radiation characteristics of TEM-line antennas is based upon fa r-fie ld relationships with the assumption that each radiating loop element in the antenna is electrically small and is isolated from its nearest neighbors by large distances in comparison to the size of the elements. All radiating elements are assumed to be identical, and small enough that the current can be considered to have nearly the same amplitude and phase at a ll points in any one element.
The radiation patterns are computed by representing the TEM-line
antenna as a linear array of electrically-small, constant-current
loops, each loop having a current flowing in i t which is obtained from
an equivalent transmission line analysis. 17
The excitation coefficients for the elements are obtained directly from the input impedance analysis, which treats the entire antenna as a collection of equivalent transmission lines with added voltage gen erators to represent mutual coupling between the various elements.
Resistive loss terms are added to the equivalent transmission line segments representing the radiating elements to account for radiation.
During the process of solving for the input impedance of the an tenna, the currents flowing in the radiating segments are found, and the values of the currents at the center of the elements are used as the constant-current excitation coefficients in, the radiation pattern expression.
In the actual computer solutions for the input impedance, i t has been convenient to le t the computer follow an iterative pro cedure in the mutual coupling problem until the results stabilize at their correct values. In this procedure the mutual coupling terms are calculated using the element currents obtained most re cently in the iterative process. Thus, the f ir s t pass through the antenna begins without regard to mutual coupling because a ll element currents are in itia liz e d to zero, but as each element current is found i t contributes an induced voltage term to subsequent element calcula tions. As the process continues, each succeeding pass improves the precision of the results obtaining better approximations of the mutual coupling terms. For the TEM-line antenna studies, i t was found that all variables converged toward their final 8-digit precision at the rate of about 1-2 decimal digits per iterative pass through the antenna. 18
Convergence of this iterative procedure toward a stable value is assured for the geometries considered here because the mutual coupling between elements always must be weaker than the direct coupling through the transmission lines. I f this condition were not met, i t would be possible for the "corrections" obtained from the mutual coupling terms in each pass through the program to be larger than the values obtained on the previous pass, thus causing the solutions to diverge. "Stability" is only relative in this context, and a discussion of computer-round off errors is deferred to Chapter III.
C. Applicability of Results
The two key assumptions in the TEM-line antenna analysis are:
i) the radiating elements are identical and electrically small,
and
i i ) the separation between elements is large compared to
their size.
The size restriction is a consequence of the need for a low-profile antenna, as discussed e a rlie r, but because of its small size the exact shape of the conductor in the radiating region is relatively unimportant and numerous geometries can be approximated by a rectangular half-loop of equivalent area. This approximation leads to the constant-current assumption which gives suitable accuracy for half-loops with total con ductor length up to about one-quarter wavelenth, but the accuracy de teriorates rapidly for conductor lengths greater than one-half wave length. 19
The separation restriction is a convenience in the mutual coupling analysis which makes the loop geometry of secondary importance and is a condition which was satisfied in all designs studied under this program where the individual radiating elements were distributed in a linear array along the surface. Good results have been obtained for elements spaced as closely as twice the element length.
Because of the separation restriction, the method used here to calculate mutual coupling does not apply to any TEM-line design in which the radiating elements are placed side-by-side or otherwise in the near fie ld of each other. The details of these other cases would depend upon the geometry of interest, but the approach could be sim ilar to the one used here for a linear array of planar half-loop elements. CHAPTER I I BASIC TEM-LINE. ANTENNAS
A. Physical Characteristics
The TEM-line antenna can have a variety of different forms, as stated in Chapter I, because it is simply an array of electrically small radiating elements distributed over a ground surface with the elements excited through an appropriate network of TEM-mode trans mission lines. All TEM-line antennas to be described here were fed seria lly , from loop element to loop element, but parallel-feed arrangements probably could be useful in some applications and could be included as members of the class of TEM-line antennas.
Figure 1 was shown previously as a sketch of a typical surface- mounted TEM-line antenna. I t showed how the low-profile antenna can be made to attach to an exterior conducting surface, requiring l i t t l e or no space on the interio r side of the surface. Several an tennas of this type, with varying numbers of radiating elements and various gap-area geometries, were constructed and tested on ground planes.
These antennas were constructed by removing the outer braid from
RG8/U coaxial cable and replacing i t by machined blocks with attach ment flanges on either side of the tubular housings as shown in Fig.
1. The flanges were bolted to the ground plane, holding the lower sur face of the cable dielectric in contact with the-ground plane along
20 21
the entire length of the antenna. A coaxial fittin g at one end and a sliding short-circuit termination at the other completed the antenna.
This method of construction produced a strong, flexible, and easily installed antenna which used a sliding short-circuit termin ation to supply the correct tuning reactance. In some applications
the movement of a short-circuit position for tuning the antenna would
be incompatible with requirements and electrical tuning would be
needed. This could be accomplished by adjustment of a bias on a
voltage-variable capacitance diode, or by electrically switching to
different length line segments. Of course, for fixed-frequency appli
cations no adjustment would be required and the terminating reactance
could be supplied by a fixed length of shorted coaxial lin e or by a
lumped reactance.
Another successful type of construction for a TEM-line antenna
is shown in Fig. 8 where the coaxial transmission line was embedded
within a thick aluminum ground plane in order to maintain a flush sur
face.[23] The center conductor of the coaxial cable extended across the
width of each transverse corrugation as shown. A sliding short-cir
cuit was used to tune this model, also. For convenience, the poly
ethylene dielectric was le f t in place around the conductor, even in
the gap region. Although the corrugations extended completely across
the ground plane, they were nonresonant and could have been much
shorter. In fact, current-probe measurements showed that the surface
current flow in the gap region decayed to a negligible value within
a distance equal to about a gap-width away from the radiating conductor. 22
FEED ADJUSTABLE END SHORT CIRCUIT 1
2cm K 40cm L10cm 122 cm
H s h -
lcm
Fig. 8 .—Flush-mounted TEM-line antenna.
An exact description of the current flowing in the ground surface
around the radiating elements of any TEM-line antenna depends strongly
on the constructional details of the element. However, for analysis,
a suitable approximation to the current distribution on the ground
plane was found to be that due to a rectangular half-loop conductor
protruding through the ground plane, with the interconnecting coaxial
lines hidden behind the ground plane, as shown in Fig. 9. z 23
TERMINATION
FEED
Fig. 9 .—Surface-mounted TEM-line antenna with delay-line loading.
I t is important to note that the short end-segments of the element which complete the half-loop approximation are required to account for radiation from the mouth area of the discontinuous coaxial line.
I t would be inadequate to base the approximation only upon the current flowing in the extended center conductor and its image in the ground plane. This subject is discussed further in Chapter IV where sup porting measurements are described.
Several 3-element and 5-element versions of TEM-line antennas were constructed with such protruding half-loops and were found to be very convenient for laboratory use because of the ease with which modifi cations could be made to antenna parameters such as element height, 24 conductor thickness, and delay-line length.
Installation of such an antenna on an existing vehicle could be more d iffic u lt than in cases discussed previously, because of the requirement for access to both sides of the mounting surface. How ever, even, with this difficulty, installation of a TEM-line antenna should compare favorably with conventional antennas for the reasons discussed in Chapter I .
B. Electrical Characteristics
The electrical characteristics of the resonant TEM-line antenna
are controlled by the distribution of a standing current wave in re
lation to the placement of the radiating elements along the trans mission lin e.
1. Radiation patterns
Some insight into the radiation properties of TEM-line antennas
can be gained from consideration of a very simple model consisting
of a uniform sinusoidal standing-wave on a transmission line exciting
a set of small radiating elements.
From a practical standpoint the existence of the purely sinusoidal
standing-wave would imply a complete absence of any radiation from the
antenna and a lack of reflections in the transmission line from any
of the radiating elements. Of course, such a set of conditions implies
poor impedance characteristics for the antenna, but a heuristic ex
planation of the TEM-line antenna using these assumptions nevertheless
w ill introduce the basic radiation characteristics and mode behavior 25 which w ill be studied further in the chapters to follow.
Figure 10 shows a schematic diagram of a TEM-line antenna represented by short radiating elements spaced at regular intervals along a trans mission line. A sliding short-circuit termination could be positioned to obtain the standing-wave patterns for various frequencies as de picted below the antenna diagram. The center frequency, f 0, is the frequency where the center-to-center distance between radiating elements, measured along the transmission lin e , is exactly one wavelength. This frequency is given by
where v is the propagation velocity in the line and t cc is the equiv
alent length of delay line between centers of adjacent radiating
elements. Here i t was assumed that the elements presented no elec
trical discontinuity to the line, so that such an equivalent length
of delay line could be defined easily.
As shown in Fig. 10 it is possible to reposition the short-circuit
for each frequency so that a current maximum is maintained in the
center element and the excitation coefficients remain symmetrical.
Furthermore, the excitation coefficients are entirely real for a pure
standing-wave because the phase switches abruptly in steps of n radians
at the current nodes, as indicated by the + and - signs in the current
loops. An odd number of elements is assumed here, but a sim ilarly
symmetrical distribution could be obtained for an array having an even STANDING-WAVE -PL _PL _PL-PL TEM-LINE ANTENNA h£
0 .0 5 f,
0 .9 0 *0
1.00 f 0
1.10 », -VWWWvVWV
Fig. TO.—Standing-wave TEM-line distributions. 27 number of elements, and the results would be sim ilar.
The standing wave of current along the line of length z may be expressed as
(2) I (li) = cos M i
where the magnitude has been normalized to unity, f is the frequency and v is the wave velocity on the line.
Since the origin was taken at the center element for symmetry, i t is convenient to denote the center element as number zero, its nearest neighbors as numbers +1 and -1 , etc. I t follows that the excitation coefficients are given by
. . . _ 2nuf (3) Ip - cos ^
n = 0, +1, +2, . . .
For the five-element array shown in Fig. 10, the excitation coef ficients for five frequencies between 0.85 f0 and 1.15 f0 can be visualized by inspection of the given standing wave patterns. At f = f Q, the distribution is uniform and in-phase, while at higher and lower frequencies the in-phase distribution becomes tapered in ampli tude toward zero at the ends. At greater frequency excursions, i t can be seen that phase reversals w ill be encountered in the outer elements.
The radiation patterns obtained from these excitation coefficients depend upon the spacing between the elements in the array and the 28 radiation pattern of the individual elements. I t w ill be shown in the
next chapter that the TEM-line element pattern is isotropic in the
principal plane containing the line of the array, so only the array
factor need be considered at this point.
The array factor for a set of five isotropic point sources may
be obtained by summing the fa r-fie ld contributions of each element as
a function of angle as shown in Fig. 11. Using the excitation coef-
TO FAR-FIELD POINT
- 2S 2S
Fig. 11.—Array of five isotropic sources along z-axis.
ficients of Eq. (3), the far-field array factor has the form:
(4) F = 1 + 2 cos (2ir £ ) cos (2ir 1. . 1 . cose) *o ^o *o
+ 2 cos (4 tt ) cos (4ir ~ - . cose) f o f o *0 29 where s/x0 is the spacing between elements of the array measured in wavelengths at the center frequency f Q.
Several plots of the square of this function were prepared to illu s tra te the variation of the power pattern with frequency for d if ferent values of spacing. Fig. 12, with s = 0.2xo, is such a two- dimensional surface with the angle e running from 0 to n from le ft to rig h t, and the frequency running from 0.5 f 0 to 1.5 f 0 from front to back. The central peak corresponds to broadside radiation at f " V Several important TEM-line antenna features are distinguishable in this plot.
1) There is a lower cutoff frequency, below which insignificant
radiation occurs. This is approximately 0.833 f Q in Fig. 12.
2) At the lower cutoff frequency, radiation is primarily in
the endfire and backfire directions (0° and 180°), and as
frequency increases these two beams scan toward broad
side until they merge into a single broadside beam.
3) The broadside beam persists over an extended frequency in
terval, but eventually splits into two beams which scan
away from each other with increasing frequency until they
disappear, one at endfire and the other at backfire (180°
from endfire). This occurs at approximately f = 1.25 f 0
in Fig. 12.
Figure 13 is a similar plot for the case where s = 0.3Xo. All
of the features mentioned for Fig. 12 are present, but it can be seen 30
1.5
1.0
0.5 180 90 B (DEGREES)
Fig. 12.—Standing-wave power patterns vs. frequency fo r five elements spaced 0.2 wavelengths at design frequency. 31
B (DEGREES)
Fig. 13.—Standing-wave power patterns vs. frequency fo r fiv e elements spaced 0.3 wavelengths at design frequency. 32
/ that the two endfire and backfire lobes at f = 1.5 f Q are too strong to be explained by the single-mode behavior of the fir s t case.
Examination of Figs. 14 and 15, which are similar plots for s = 0.4Xq and s = 0.5xo respectively, reveal that the pair of strong
lobes near f = 1.5 f 0 are really grating lobes which scan inward to -, ward broadside with increasing frequency until they have merged with the outwardly-scanning lobes of the f ir s t mode.
This introduces a fourth characteristic of TEM-line antenna pat
terns, their modal behavior. Depending upon the spacing-to-wavelength
ratio, operation in numerous higher-order modes is possible, and the overall pattern may be computed as the superposition of patterns from more than one mode of operation.
This modal character is easily seen in Fig. 16 which is a panorama of power patterns from 0.5 f 0 to 3.5 f 0 for element spacings of 0.25xo.
Modes 1, 2 and 3 are seen in their entirety, passing from backfire
through broadside to endfire, and a portion of mode 4 below its broad
side frequency is visible. Examination of Fig. 16 shows that modes
1 and 2 do not overlap in frequency, modes 2 and 3 overlap near
f = 2.5 f Q, while modes 3 and 4 show considerable overlap above
f = 3.25 f0.
The details of these modes can be presented more clearly on a
propagation-constant diagram, called a Brillouin diagram.
2. Brillouin diagrams
The behavior of traveling-wave antennas is frequently depicted on 33
9 (DEGREES)
Fig. 14.—Standing-wave power patterns vs. frequency fo r fiv e elements spaced 0.4 wavelengths at design frequency. 34
J i l f ■ -*x
9 (DEGREES)
Fig. 15.—Standing-wave power patterns vs. frequency fo r fiv e elements spaced 0.5 wavelengths at design frequency. Fig. 16.—Frequency panorama of standing-wave power patterns for five elements spaced 0.25 wavelengths at design frequency. 36 a Brillouin, or k - g, diagram in which the free-space propagation constant, proportional to the frequency, is plotted against the prop agation constant of the traveling wave.[24] I f both coordinates are multiplied by a characteristic dimension, such as the spacing be tween radiating elements, the axes may be labeled conveniently in radi ans.
I t is not d iffic u lt to show[25] that for the nth mode, sometimes called "space harmonic", the angle of maximum radiation, en, is re lated to the propagation constant by o
(5) en - k cos en
where rc\ o - « 2n tt (6 ) Bn - S0 s
and K is the free-space propagation constant
(7) K = 2-irf/c .
That is , even though the current is known to propagate along the transmission-line structure as a slow wave, one or more apparent traveling waves, or "space harmonics'^may be set up by the excitation of the ele ments, and radiation w ill occur at angles appropriate to the apparent phase constant in Eq. (6) when real solutions to Eq. (5) exist. 37
The bandwidth over which various modes operate in the "visible" or radiating region may be obtained from Eqs. (5) and (6) by observing that the onset of visible radiation for the nth mode occurs for
(8) cos en = -1 (onset)
and the extinction of that radiation occurs for
(9) cos en = +1 (extinction).
For the lowest radiating mode, n = 1, the bandwidth of this "visible region" is
(10) = . 1,+iL f min 1 - - c where c is the free-space wave velocity and v is the effective velocity of the transmission-line wave, computed by dividing the center-to-center element spacing by the tran sit time required for the wave to travel
through the delay line between those two points.
The maximum bandwidth over which a TEM-line antenna can operate in
a single mode is of interest sometimes/ and is the ratio of onset fre
quencies for mode 2 and mode 1. ' ' -
(11) BWsingle-mode = 2 :^ 38
This maximum bandwidth can be achieved only in TEM-lines with lig h t enough loading so that the fir s t mode is not extinguished be fore the second mode appears. This occurs for
(12) v/c >. j (fo r maximum single-mode bandwidth)
This condition was approximated in the two-dimensional surface plot of Fig. 13 where the element spacing was 0.3xo, and the effects of the second mode were noticeable near cutoff of the f ir s t mode.
Some of these critical points are illustrated in the Brillouin diagram of Fig. 17 which is a straight line plot representing unperturbed propagation through the delay-line sections of a TEM-line antenna de signed for an effective wave velocity of c/4. This diagram corresponds to the panorama plot of Fig. 16 where four modes were v isib le. Note that no radiation is expected from the zero order mode in the TEM-line antenna because i t is a slow-wave structure. Only the higher-order modes (n >.1) show regions of fast wave operation which permit radiation at real angles of e corresponding to
(13) —* ^ 3n <(for radiation) .
Figure 18 is a k - 0 diagram similar to that of Fig. 17 except that the loading is appropriate fo r propagation along a straight length of
coaxial line having a solid polyethylene dielectric . For this case, 39
MODE I EXTINCTION MODE 2 ONSET MODE I ONSET *
£S
Fig. 17.— Brillouin diagram for heavily loaded TEM-line antenna. 40
k = 0.659/3
^
MODE I FORWARD MODE RADIATION BACKWARD RADIATION
2 MODE ONSET ONSET/
Fig. 18.—Brillouin diagram for polyethylene-loaded TEM-line antenna. 41
(14) v = c/vie^ = 0.659 c (solid polyethylene)
and from the diagram i t can be seen that modes 1-4 could coexist over a range of frequencies between approximately 2.4 f Q and 3.0 f Q. This
condition would cause the radiation pattern to be multi-lobed and some what irregular, and probably its usefulness is lim ited. CHAPTER I I I ANALYSIS
A. Radiation Patterns
As explained in Chapter I, a far-field analysis produces satis factory results for TEM-1ine antennas because each element of the linear array is small and isolated from its neighbors. The pattern of the antenna may be calculated as the array factor multiplied by the element factor, and the excitation coefficients for the array elements then are obtained from an impedance analysis of the trans mission line structure, including mutual coupling terms between radiating elements.
The far fie ld of a single TEM-line element may be found with the aid of Fig. 19 which shows an electrically small rectangular half-loop of current with length g and height d above an infinite ground plane.
The corresponding image half-loop is shown below the ground plane.
The following five approximations are useful in fa r-fie ld cal culations because they simplify the resulting expressions somewhat.
The fir s t four approximations are used in terms involving the phase of the far fie ld :
(15) r-j & R + ^ cose
(16) ?2 & R - d sine sin«|)
42 43
x
Fig. 19.—Rectangular half-loop TEM-line radiating element. 44
(17) r3 % R - | cose
(18) r4 £ R + d sine sin
The fifth approximation is used for terms only involving amplitude of the fa r fie ld ;
(19) r-j £ r 2 £ r 3 £ r4 £ R •
Using these approximations in a vector potential expression, [26] the far electric field vector may be written in terms of the rectangular components
cose 4irR -d
where k = 2ir/x is the free-space phase constant, and y and z are the y- and z-directed unit vectors.
Because of the assumption that the element is electrically small,
(21) d « x » g/2,
and the exponential terms of the integrands may be approximated by 45 the first two terms of their power series. The result is
( 2 2 ) E £ MuI 3 * d e J' _K^ j _ z sine sin^-y cose 2ttR
The rectangular unit vectors may be replaced by their spherical coordinate equivalents to obtain a practical fa r-fie ld expression.
The necessary relationships are
/\ a (23) z = r cose - e,sine
A A (24) y = r sine sin
Substituting Eqs. (23) and (24) into Eq. (22), the radial components cancel, leaving only e- and<)>-components of electric field. Noting that the area enclosed between the current element and the ground plane is ^
(25) A = g d,
the far electric fie ld expression for the rectangular half-loop be comes
_ -up (26) E * e si n
The radiation from the complete TEM-line antenna may be computed 46 by pattern multiplication as the product of the element factor in
Eq. (26) and the array factor appropriate for point sources located at the element positions and having excitation coefficients as deter mined by separate means to be described la te r.
The array factor may be obtained from Fig. 20 which shows N iso tropic elements with uniform spacing s along the Z-axis, each element having a complex excitation coefficient of thq form
I,ejS' I2eiS*
Fig. 20.—Array of N point sources equally spaced along z-axis. 47 (27) In = I In I {" .
For the far fie ld , the array factor may be expressed as the mag- nitude of the sum fie ld of a ll sources,
(28) array factor = o
Thus, for a given set of excitation coefficients, Eqs. (26) and
(28) may be used to compute fa r-fie ld patterns of the TEM-line antenna
mounted on an in fin ite ground plane. The free-space values of the
constitutive parameter y and the velocity of light c may be inserted
as
(29) y = y0 = 4ir x 10“7 henry/meter
(30) c S 3 x 108 meter/sec
and the two principal components of the fa r fie ld patterns become
240tt2 A sin
-x, 240 ^ A cos 8 cos ei (<&-&) I |in|e^nKS cose ’ 6n^ 48
B. Mutual Coupling
Mutual coupling in a TEM-line element has the form of an open- circuit voltage induced in a rectangular half-loop element by the current flowing in another nearby rectangular half-loop element. For the purposes of this analysis, a ll elements are assumed to have the same size and shape. Figure 21 shows the geometry of the mutual-coupling
y A
..Fig, 21.—Mutual coupling geometry. problem, with the constant current flowing in element no. 1 at the origin, and the induced voltage at the terminals of element no. 2 which is displaced a distance s along the positive z-axis. Both elements terminate in the ground plane at the ends opposite their feed poi nts.
The induced voltage in element no. 2 may be found by integrating the electric fie ld set up by element no. 1 around the path defined by element no. 2. 49
The electric fie ld of the driven element was given in rectangular components in Eq. (22), and may be simplified because only the fie ld on and near the z-axis is required. Thus
(33) E- ^ -y 2irZ provided that
(34) d « Izl »
Since the electric field is entirely in the y- direction in this region, the line integral for induced voltage consists only of those portions directed along the normal to the ground plane. That is
(35) V = - J E • d a loop
2irZ s+ g/2 z = -s + g/2
j ,->* i. a.id-a3.[- !z M , ^ y=0 V 2irZ S- g/2 Z = -s - g/2
where the top values are substituted for z when element no. 2 lies in the positive-z direction from element no. 1 and the bottom values are used when it lies in the negative-z direction. In both cases, however, 50 the same expression for induced voltage is obtained, provided that g/2 is sufficiently smaller than s that it may be dropped from the de nominator of the integrands. The resulting expression is
-imu T *-2 a 2 J U t - K S ) (36) V £ . .lyjj.Li—1 ■ ------2ifS
Thus, under the same assumptions as made for the fa r-fie ld pattern analysis, ( i . e . , the radiating elements are small compared to the wave length and the spacing between the elements) the open-circuit voltage coupled into an isolated TEM-line element by a current flowing in an identical neighboring element may be written as
(37) y ^ 4 8 0 ff3 1 a 2 s X3 . where I is the current flowing into the element terminals, A is the area enclosed between the half-loop and the ground plane, and s is center-to-center distance between the driven element and the parasitic element.
Coupling between elements other than nearest neighbors is expressed by replacing s in Eq. (37) by the proper multiple of its e lf.
The same mutual-coupling result can be obtained by solving for the voltage induced in the parasitic loop by the time-changing mag netic field of the current in the driven loop.
I t is clear from the appearance of s in the denominators of
Eqs. (36) and (37) that this result should not be applied for small 51 values of s. The correct solution for that circumstance would require an accurate description'of the near-field coupling between the loops, and would be much more dependent upon the loop geometries than the cases of interest here.
C. Impedance
The input impedance of a TEM-line antenna may be obtained from repetitive applications of voltage-and-current-transformations through sections of transmission lines equivalent to corresponding segments of the antenna. Radiative losses and mutual coupling may be represented adequately by lumped values introduced at appropriate points in the transmission-line equivalent circuit.
For a TEM-mode transmission line having length % and character is tic impedance Z0, as shown in Fig. 22, with a complex wave-propagation constant given by
Fig. 22.—Segment of uniform transmission line. 52
(38) y = a + J 6 > the complex voltage and current at the. input are related to the complex voltage and current at the output by[27]
(39) V- = V° + Ip eY* + ^o ~ l o zo e“ YA 1 2 2 and
(40) I . = Jo * V Zo eyA + Jo ~ vo /zo e"YA 1 2 2
The application of Eqs. (39) and (40) to the delay lines which connect adjacent radiating elements of the TEM-line antenna is ap parent, because all voltages, currents, and impedances are readily defined. The treatment of the radiating element itself, however, re quires some interpretation.
Figure 23 shows a single rectangular half-loop radiating element protruding from a ground plane. The horizontal arm across the top is of uniform circular cross-section, but for convenience of analysis the vertical segments are right-circular cones with their apexes in the plane of the ground plane, tapering to thickness t at height d to match smoothly into the ends of the top arm.
The characteristic impedance of the top horizontal arm is that of an isolated cylindrical wire above a ground plane,[28]
(41) Z0 = 60 In ^ . 53
Fig. 23.—Equivalent TEM-line radiating element.
The characteristic impedance of the conical sections, defined in terms of the half-cone angle t|>, is [29]
(42) Z0 = 60 In ctn f
However, from Fig. 23
(43) ctn ty = ~ 54 so that for small angles
(44) Z0 = 60 In M .
Thus, for the purposes of transmission-line equivalence, the three segments of the rectangular half-loop element may be represented as three segments of uniform transmission lin e , a ll having the same characteristic impedance given by Eq. (41).
However, i t is, known that reflections occur from short-radius bends in conductors, and the transmission line equivalent c irc u it must include these effects. Ross, et. a l., have determined through studies in the time domain that an isolated right-angle bend in a thin wire can be characterized from the driving point as a nondispersive re flection coefficient[30,31]
(45) y % - 0.12 .
Thus, the equivalent circuit of Fig. 23 for the rectangular half
loop is completed by the addition of shunt resistances across the
transmission lines at the points representing the right-angle bends.
The reflection coefficient of a pure resistance shunted across a
transmission line is given by
(46) Y = -1/(1 + 2R/Z0) 55
For the thin-wire cases considered by Ross, e t.a l. this value of shunt resistance due to the right-angle bend should be
(47) Rb = 3.67 Z0 .
However, for the thick-wire cases used to represent TEM-line an tenna elements, i t was found by comparison to experiments that Ross' reflection coefficient was too great, and better agreement with measure ments could be obtained with
(48) Rb = 8 Z0 ,
corresponding to a reflection coefficient of
(49) Y = - 0.0588 .
D. Calculation
As described e a rlie r, the computer solution fo r input impedance of the TEM-line antenna proceeded from an assumed unit voltage across a terminating reactance, along with the resulting terminal current flow, through a series of transmission-line transformations appropriate for the equivalent circuits discussed in the previous section. Mutual coupling was represented by series voltage generators placed at the input (driving-point side) of each half-loop element. Inasmuch as the strength of any given induced voltage depended upon the current in the 56 element whose coupling i t represented, while simultaneously affecting the overall voltage/current distribution in the antenna, i t is clear that the complete solution could be obtained from a system of simultan eous equations.
For programming convenience, however, an iterative technique was used in which the mutual voltages were calculated from an array of element currents which were in itia lly stored as zero, but subsequently modified by the program each time a new current was calculated at the mid-point of a radiating element. This is the current denoted as
in Fig. 23.
On each repeated pass through the antenna, the values of the stored element currents approached more closely to the final, correct value.
The process was terminated a fte r the fluctuations in computed values became smaller than an a rb itra rily chosen fraction of the values being computed.
In one series of computations over a range of values in frequency and terminating reactance, i t was noted that about 90% of the cases converged to a stable value of input impedance, to the lim it of precision of the computer (approximately eight decimal digits), within four to six iterations. Most of the remaining 10% cycled between two or three values differing by 1 or 2 in the least significant d ig it, but a few cases were noted where the values differed by more than 5
in the least significant d ig it. Also, in a few cases, slower conver gence was noted with s ta b ility achieved only after as many as ten
i terati ons. 57
The reason for this anomalous behavior appeared to be that in certain ranges of parameters, one or more induced voltages became close enough to the negative of the direct-coupled voltage at that node so that the resulting cancellation caused the quantization errors of the digital computer to be significant. A consequence of this hypothesis would be that larger fluctuations in the final values would be pro duced as the cancellation of voltages became more nearly perfect, but the occurrence of perfect cancellation would be rare.
This explanation is supported by the fact that in run of about
200,000 data points the computed value for input impedance failed exactly once to stabilize within the eight least- significant binary
digits of a 27-.bit characteristic of a floating-point number. Complex numbers are actually represented by two floating-point numbers (the real part and the imaginary p art), both of which contain
27 significant b its , excluding sign and exponent; i t is not known whether the real part, the imaginary part, or both parts failed to
stabilize in the last case mentioned above.
1. Impedance
For verification of the analysis, a series of calculations was
made for a three-element half-wavelength TEM-line antenna with a var
iable capacitance termination. The parameters of the calculations were
taken to f i t those of the capacitance-diode-tuned TEM-line antenna de
scribed in the next chapter so that a direct comparison could be made
between measured and calculated results. 58
In the capacitance-diode-tuned antenna measurements, the diode
voltage was stepped between 0 and 32 volts, corresponding to maximum
capacitance and minimum capacitance, respectively. The values of the maximum and minimum capacitances could be estimated only approximately
from the manufacturer's specifications because of an unknown amount of
parasitic capacitance in the diode-retaining fixture a t ,the base of
the radiating element. The terminating capacitance was estimated to
be about 6 pf. for 0 volts and about 2 pf. for 32 volts.
For the calculated values of input impedance, the terminating
capacitance was stepped from 0 pf. to 32 pf. in steps of 1 pf.
Both the measured and the calculated sets of impedances were obtained
at 5 MHz frequency intervals from 245 MHz to 285 MHz.
Figures 24-32 show the comparison of measured and calculated in
put impedances at each of the nine frequencies. In each figure, the
open circles represent measured data taken with diode voltages of
0, 1, 2, 4, 8, 16, and 32 volts. The solid points are the calculated
impedances at each of the 1 pf. steps in terminating capacitance.
Generally good agreement between calculations and measurements
was obtained, even though this particular design was an extreme case
with relatively large elements spaced only two element-lengths apart,
thus somewhat straining two of the assumptions in the derivations.
2. Radiation patterns
A by-product of the impedance calculations is the set of complex
currents flowing at the center-point of each radiating element in the 59
32 v
0.0 pf
o •
Fig. 24.—Input impedance of capacitance-diode-tuned TEM-line antenna, 245 MHz.
o o o o MEASURED DATA • • • • CALCULATED DATA 32v
0 .0 pf
32 pf
.0.0 v
Fig. 25.—Input impedance of capacitance-diode-tuned TEM-line antenna, 250 MHz. 26.—Input impedance of capacitance-diode-tuned TEM-line antenna, 255 MHz.
o o o MEASURED DATA • • • CALCULATED DATA
32 pi 0.0 v 0.0 pf,
■32 v.
27.—Input impedance of capacitance-diode-tuned TEM-line antenna, 260 MHz. 61
Fig. 28.--Input impedance of capacitance-diode-tuned TEM-line antenna, 265 MHz.
o o o o MEASURED DATA • • • • CALCULATED DATA
p.Ov
32 pf 0 .0 pf'
32 v
Fig. 29.—Input impedance of capacitance-diode-tuned TEM-line antenna, 270 MHz. 62
32 pf 0.0 p»
32 v
Fig. 30.—Input impedance of capacitance-diode-tuned TEM-line antenna, 275 MHz.
o o o o MEASURED DATA • • • • CALCULATED DATA
32 v 0.0» > 32 pf ^ 0 0 pf
Fig. 31.—Input impedance of capacitance-diode-tuned TEM-line antenna, 280 MHz. 63
O o o o MEASURED DATA • • • • CALCULATED DATA
Fig. 32.—Input impedance of capacitance-diode-tuned TEM-line antenna, 285 MHz.
TEM-line antenna. These are exactly the currents required in Eqs.
(31) and (32) for fa r-fie ld pattern computations. The previous pat tern computations shown in Figs. 12-16 were obtained with an idealized set of excitation coefficients without regard to the input impedance they would enforce upon the antenna, but the computations described in this chapter provide the inter-relationships among excitation coefficients, input impedance, and far-field patterns.
Figure 33 is a descriptive sketch of input impedance illu stratin g the relationship between impedance and the modal behavior of the TEM- line antenna discussed previously. Typically, the point of minimum
VSWR (closest approach to the center of the Smith chart) occurs near 64
BROADSIDE REGION
VSWR = 3
FREQUENCY SCANNING INCREASING v REGION FREQUENCY
Fig. 33.—Relationship between pattern modes and input impedance of TEM-line antennas. the point where the input impedance locus crosses the real axis with series resonance (low resistive component).
This point of series resonance also marks the boundary between the frequency-scanning mode and the broadside-beam mode.
I t was seen in the impedance plots of Figs. 24-32 that an ad justment of the terminating reactance could be used to tune the in put impedance to resonance over a range of frequencies. Thus the pat tern of the TEM-line antenna would be expected to depend somewhat on the value of the terminating reactance, at least in the frequency range around the transition from the frequency-scanning mode to the broad side mode. 65
This is indeed the case, as can be seen in the two-dimensional surface plots of Figs. 34-39. Each of these figures is a plot of the power pattern (the square of the magnitude of the electric field given by Eq. (3 1 )),in the principal plane where
(50)
In each figure, the terminating capacitance is held constant,
the pattern angle 0 runs from 0 to n from le ft to rig h t, and the
frequency runs from 100 MHz to 500 MHz from front to back. Plots
are shown for capacitance values of 0, 1, 2, 4, 8 and 16 pf. The
corresponding plot for 32 pf resembled the 16 pf plot very closely,
and therefore is not shown.
The feature which most distinguishes these plots from those of
Figs. 12-16 is the lack of symmetry. Symmetry occurred in the
idealized plots only because the excitation coefficients were assumed
to be symmetrical, a condition that does not occur in a practical an
tenna which radiates well.
The modal behavior described by the idealized plots is shown
clearly in Figs. 24-39, however. In each case the first radiating
mode begins with a backfire lobe (e = i t) near 250 MHz, which quickly
scans with increasing frequency to a broadside position (e = tt/2).
The broadside mode persists for an interval of up to 100 MHz, whereupon
the main beam scans abruptly toward endfire (e = 0) and extinguishes
at a frequency near 400 MHz. 66
500
400
300
200 MHz
100 90 « 180 9 (DEGREES)
Fig. 34.—Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C=0 picofarad.
jStsyfe.m i 400
300
200 MHz
too 180 9 0 9 (DEGREES)
Fig. 35.— Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C=1 picofarad. 90 ^ 6 (DEGREES)
Fig. 36.—Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C=2 picofarads.
500
9 0 Q (DEGREES) 180
Fig. 37.—Principal-plane power patterns vs. frequency for capacitance=tuned TEM-line antenna, CM picofarads. 68
Fig. 38.—Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C=8 picofarads.
Fig. 39.—Principal-plane power patterns vs. frequency for capacitance-tuned TEM-line antenna, C=16 picofarads. 69
The change in pattern detail caused by changes in the terminating capacitance is seen readily by inspecting the plots in the area of 250
MHz-300 MHz.
Good agreement with measured patterns was obtained from these calculations, except for angles near grazing incidence where the pat tern shape was controlled by edge diffraction from the ground plane as discussed in the next chapter. CHAPTER IV MEASUREMENTS
Numerous TEM-line antenna designs were tested during the experi mental phase of this program, including antennas with various types of element design, element placement on the ground plane, and method of achieving the desired terminating impedance. Antennas with as few as one radiating element and as many as ten radiating elements were tested, covering the size range up to about six wavelengths in length.
The following four sets of measurements are presented in detail because they illu s tra te the range of topics covered and show typical results which can be obtained from practical TEM-line antenna designs.
A. Finite Ground Plane
The in fin ite ground-plane assumption, although useful in theory, cannot be realized in fact. The extent to which radiation-pattern measurements of a ground-plane-mounted antenna are perturbed by the diffraction fields from the edges of the fin ite ground plane depends
upon several factors, including the strength and polarization of the
fields illuminating the edges, and the shapes of the edges themselves.[32]
In the case of low-profile antenna elements mounted on a fin ite ground
plane with knife-edges, maximum diffraction occurs where the incident
field is polarized perpendicularly to the edge and no diffraction occurs
70 71 for parallel polarization. For loop elements, the radiation fie ld fa lls to zero in the direction which would correspond to parallel polarization. This diffraction effect was observed, in varying de grees, in the radiation pattern measurements of every antenna de sign studied under this program. In the most severe case, a single
TEM-line loop element was centered in a large, square ground plane, and the two principal-plane patterns were measured at 10,000 MHz.
The ground plane was 18 inches square and the loop element was formed simply by removing about 1/4 inch of shield braid from miniature co axial cable attached to the ground-plane surface.
The H-plane pattern showed the smooth sinusoidal variation in
nounced scallop superimposed on the predicted constant pattern
(independent of e). The greatest effect of the diffraction pattern was at angles near grazing incidence to the ground plane, where the pattern sloped sharply toward zero instead of remaining constant as
calculated. Elsewhere the scallop was about 1 dB in total fluctuation,
but the pattern shape revealed the underlying element pattern to be
nearly independent of angle.
The diffraction effects were reduced substantially by reshaping
the edges of the ground plane to a smoothly rounded, cylindrical
contour. Ordinarily, a radius of curvature of a wavelength or more
is desired for this purpose, but i t was considered unlikely that such
a large radius could be tolerated at VHF frequencies and smaller
radii of curvature should be used for TEM-line antenna measurements. 72
Figure 40 shows that a radius of curvature of only yj4 was
WITH . CURVED EDGES
/ WITH* STRAIGHT EDGES
Fig. 40.—Effect of curved ground-plane edges on E-plane power pattern of three-element TEM-line antenna. relatively effective in reducing the ground-plane-edge diffraction for a three-element TEM-line antenna. I t can be seen that edge-diffraction can obscure pattern detail to a considerable extent when the ground plane is relatively small, because the pattern which results from in terference between the fields diffracted from opposite ends of the ground plane w ill be comparable in number of lobes to that of the an tenna under study. However, the addition of cylindrical edges to this 73
ground plane reduced the diffraction effects enough to reveal the single broadside beam of this half-wavelength TEM-line antenna at
1000 MHz.
The ground plane for this antenna was only 1.8 wavelengths long,
prior to addition of the curved edges. The asymmetry of the diffraction
interference pattern indicates that the phase center of the TEM-line
antenna was not in the exact middle of the ground plane.
B. Voltage-Tuned TEM-line Antenna
A comparison between computed and measured impedance data was
made in the preceding chapter for a three-element half-wavelength
capacitance-tuned TEM-line antenna. The construction of this antenna .
is shown in Fig. 41. The ground plane was 1.5 meters long, not in
cluding the cylindrically-curved end sections with a radius of curv
ature of 37.5 cm, or approximately three-eighths wavelengths.
The dc control voltage for the capacitance diode was introduced
at the RF feed-point through a Microlab HW-02N coaxial monitor tee
fittin g designed for this purpose, containing an RF choke and a dc
blocking capacitor.
A typical set of constant-frequency power patterns for this an
tenna is shown in Fig. 42. The frequency was held at 265 MHz and
the dc control voltage was stepped between 0 and 32 volts. Reference
to the impedance data shown previously in Fig. 28 shows that the VSWR
reached its minimum of 1.0 for a control voltage of 8 volts at this
frequency. Furthermore, in accordance with Fig. 33 that point marks
the dividing line between the broadside-mode and t.'.s frequency-scanning 74
0.794 cm 10 cm 10 cm MOTOROLA MV I860 0 CAPACITANCE - DIOOE 7.75 cm 4 VOLTS)
RF
DC
50 cm R6 62 /U COAXIAL CABLE AND FITTINGS
Fig. 4 1 Capacitance-tuned three-element VHF TEM-line antenna. mode* as is evidenced by the pronounced splittin g in the beam for
16 volts and 32 volts.
A summary of the measured data is contained in Fig. 43 showing the best VSWR which could be obtained at each frequency in the range from 245 MHz to 280 MHz, along with the dc control voltage required to attain that best VSWR. The power patterns agreed closely with those calculated and presented in Figs. 34-39. Note especially that the asymmetry of the measured patterns agrees with that of the cal culations, and that the primary beam forms in the backfire direction and scans rapidly toward broadside as previously explained. RELATIVE POWER 100 100 100 40 40 40 80 60 eo 60 20 i. - H Epae oe pten o three-element, of power patterns MHz E-plane 5 6 .-2 2 4 Fig. 60 80 20 0 aaiac-id-ue TMln antenna. TEM-line capacitance-diode-tuned VOLTS 2 VOLTS 8 VOLTS 0 60 ■“ 80 g J U < J U o S > -I 100 40 01020 -90 270 180 90 -90 2 VOLTS32 6 DEGREES ( ) DEGREES( ) 90 160 16VOLTS 270 90 270
75 76
5
35 / 4 30
VSWR 25 VOLTS 3 BEST 20 VOLTAGE
BEST VSWR 2
LL 245 250 255 260 265 270 275 280 FREQUENCY (GHz)
Fig. 43.—Best attainable VSWR and required voltage vs. frequency for three-element, capacitance-diode-tuned TEM-line antenna.
An approximate check on the efficiency of the antenna was per formed by comparison to the response of a tuned half-wavelength di pole. I t was found that the TEM-line antenna gain was 0-0.5 db above the dipole. Although a numerical integration of the three-dimensional pattern necessary to find the measured d ire c tiv ity was not performed for this antenna, this value of gain is consistent with measured ef ficiencies in the range of 70% reported by Kilcoyne for other TEM-line antennas using sliding short-circuit terminations.[33] 77
C. Five-element Compact TEM-line Antenna
The profile height of TEM-line antenna elements can be reduced i f more elements are included in the design, because a reduced percentage of the total radiation is required to occur through each element. The consequences of adding more elements to the antenna are greater lengths with correspondingly narrower beamwidths and, in some cases, a reduc tion in bandwidth because of the longer total electrical length of the transmission line circu it which must be resonated.
A five-element TEM-line antenna with 5 0 - delay-line loading and rectangular half-loop elements similar to the ones discussed above was constructed so that its second mode would occur near 1.2 GHz. The element height was 0.6 cm, the length was 3.0 cm, and the center-to-
center spacing of the elements was 6.0 cm. The length of each section of delay line between adjacent elements was 29.0 cm with a velocity factor of 0.659 due to the polyethylene dielectric. A sliding short-
circuit was used for the reactive termination. Detailed measurements were made of patterns and impedance over a wide frequency range, and
gain over a half-wavelength dipole at a single mid-band frequency
of 1.192 GHz.
Figure 44 is a B rillouin, or k- 3, diagram for this antenna. The
dotted line represents unperturbed propagation of energy through the
delay-line-loaded structure, while the solid curves in the “visible"
regions of the chart show apparent propagation velocities deduced from
fa r-fie ld pattern measurements in the second-and-third space harmonic
regions. No pattern measurements were taken in the first-harmonic FREQ (GHz)
0S
-Brillouin diagram for five-element TEM-line antenna with delay-line loading.
CO
i 79 region around 0.6 GHz because the element size was chosen to be an inefficien t radiator at that frequency. Experience has shown, however, that the k- b plot in that region would resemble closely those of the second-and third-harmonic regions.
Figures 45-47 show typical power patterns obtained through the frequency of the second space harmonic, where the antenna was approx imately one wavelength long. Figure 45 shows two frequency-scanning beams near the low-frequency lim it of the visible region, with the larger beam in the backfire direction corresponding to the forward- traveling wave in the antenna. The weaker, reflected traveling wave set up the other beam. These beams scanned upward to merge at broad side in the mid-frequency range as shown in Fig. 46. At a s t ill higher frequency the broadside beam s p lit and frequency scanning be gan again with the beams moving away from each other back toward grazing angles as in Fig. 47.
The input impedance of this TEM-line antenna was adjustable at any operating frequency to within a VSWR of about 3:1 or better with
respect to 50n by proper adjustment of the sliding short-circuit
termination. This adjustment was somewhat c ritic a l, and a bandwidth
of only a few percent (of center frequency) could be obtained with a
single setting. To illustrate the impedance behavior of the antenna
over the entire second space-harmonic, the short circu it was set at
one fixed position 10.0 cm from the base of the last element, and the
input impedances were plotted as a function of frequency in Figs. 48
and 49. For c la rity , Fig. 48 covers the frequencies from broadside 80
Fig. 45.—Far-field power pattern of compact five-element TEM-line antenna, 1.093 GHz . downward, and Fig. 49 covers broadside upward, with some overlap.
The input impedance variation was large and rapid, except in the broadside mode. Note that at the broadside frequency of 1.192 GHz, for which the half-power beamwidth was shown to be 70° in Fig. 46, the input impedance was 16 + j 15a, a value that a detector could be matched to without difficulty.
Using a triple-stub tuner to match the detector at 1.192 GHz a series of gain comparisons to a tuned half-wavelength dipole was per formed, with the conclusion that at this frequency, the antenna had 81
Fig. 46.—Far-field power pattern of compact five-element TEM-line antenna* 1.192 GHz .
Fig. 47.—Far-field power pattern of compact five-element TEM-line antenna, 1.391 GHz . 82
Fig. 48.—Input impedance with fixed short position, compact five-element TEM-line antenna (1.07 GHz-1.332 GHz). 83
O.HS ,'14 Ul«i.\>T
1406 • t v
1306 iV
Fig. 49.—Input impedance with fixed short position, compact five-element TEM-line antenna (1.202 GHz-1.445 GHz). 84 a gain of 1.2 dB over the half-wavelength dipole. As with the capaci- tance-diode-tuned TEM-line antenna, this value seems consistent with
Kilcoyne’s report of TEM-line antenna efficiencies in the range of
70%.[33]
D. Five-Element Flush-Mounted TEM-line Antenna
A five-element TEM-line antenna was constructed so that the de lay-line was housed wholly within a thick skin as shown previously in
Fig. 8. The center conductor was exposed through 2 cm nonresonant gaps spaced 10 cm apart. The delay line lay in a straight line be tween the gaps for maximum spacing between radiating elements. The polyethylene dielectric gave the transmission line a velocity factor of 0.659 as in the previous case, and a sliding short-circuit term ination was also used in this design.
Power-pattern and VSWR measurements were made on this antenna throughout its fir s t space-harmonic region extending from 1.2 GHz to
2.4 GHz, and the patterns were compared to those calculated using a symmetrical set of excitation coefficients obtained from the standing- wave approximation discussed in Chapter I I .
Figures 50-52 show relatively good agreement between measured patterns and the calculated ones, even though the approximation for the element currents was somewhat crude. In Fig. 50, the effect of the fin ite ground plane on the measured pattern near grazing angles is apparent. . 50.—Far-field power pattern of five-element flush-mounted TEM-line antenna, 1.274 GHz . Fig. 51 .-Far-field power pattern of five-element flush-mounted TEM-line antenna, 1.655 GHz . 87
0 = 9 0 °
COMPUTED MEASURED
Fig. 52.—Far-field power pattern of five-element flush-mounted TEM-line antenna, 1.762 GHz .
The k - b diagram gives a concise picture of beam-angle vs. fre quency, and Fig. 53 summarizes a ll of the far-field-pattern measure ments taken on this antenna as the sequence of points covering the fir s t space harmonic region. The solid curve was computed from the pattern maxima as calculated using the approximate standing-wave dis tribution of currents. Excellent agreement was obtained in the broad side beam area represented by the vertical line at 3s = 2tt, with fa ir agreement elsewhere except near grazing incidence where the beam max imum position was influenced strongly by the ground-plane diffraction effects. The straight line from the origin with a slope of 0.659 does . KS RADIANS 27T a dtrie t b 1-% bt h VW ws dutbe o lw value low a to VSWR the 1%-2%, adjustable but was be to determinedwas t rqece i te ivsbe region. line "invisible" transmission the the in in frequencies propagation at only but radiation, represent not t n feuny n t asad y en o te sliding-short-circuit the passband means by of its in frequency any at f : o bte cud e band t ot rqece i te radiating the in frequencies most at obtained be could better or VSWR selected 3:1 at of best-attainable the of plot a is 54 Figure tuner. rqece cvrn te ein f prto ad hw ta a VSWR a shows and that operation of region the covering frequencies region. 0 h isatnos adit o ti fuhmutd E-ie antenna TEM-line flush-mounted this of bandwidth instantaneous The i. 3— iloi darm o fv-lmn flush-mounted five-element for diagram ouin rill 53.—B Fig. O O MEASURED O O O O ------COMPUTED 7T E-ie antenna. TEM-line S RADIANS jSS 27 T 3TT
88
89
VSWR
0.8 2 .0 FREQUENCY IN GHz Fig. 54.—VSWR of five-element flush-mounted TEM-line antenna.
Better impedances have been obtained from other TEM-line antennas,
but this design demonstrated the capability of a completely flu s h -_
mounted antenna with small and relatively unexposed radiating elements.
E. Other Geometries
As mentioned e a rlie r, the TEM-line-antenna characteristics described
above represent typical results of measurements performed on a series
of different antennas covering a size range from one to ten elements
spread through lengths ranging between'one-half and six wavelengths. 90
In some instances, the exposed conductor was wound into multi pie- turn loops to increase the effective radiating current in the gap area, and thereby increase the amount of radiation per gap. This technique was found to be effective for one or two extra turns in the loop, but only where the total length of conductor involved did not violate the electrically small criterion used.
I t was noted, however, that mechanical rig id ity in the gap area was reduced for such cases, and the multi pie-turn radiators should be used with discretion.
A different type of multi pi e-loop geometry was tried successfully, in which each element was replaced by a pair of orthogonal-loop elements inclined at - 45° to the longitudinal axis of the TEM-line antenna.
These orthogonal pairs were excited through separate but equal delay lines with the result that orthogonal-linear polarizations were avail able at the two feeds. With proper phase adjustments between the two feeds, e llip tic a l and nearly circular polarization could be obtained in the broadside beam.
A very interesting variation on the ground plane mounting was investigated by attaching a ten-element TEM-line antenna to the edge of ground plane, simulating an antenna along the edge of a wing or fin .
Pattern measurements showed the performance of this antenna to be generally similar to that of TEM-line antennas with conventional mountings on f la t surfaces, except that the H-plane pattern was nearly a cardioid instead of. the sinusoid obtained for other cases.
VSWR measurements of the edge-mounted antenna showed that proper 91 adjustment of the sliding short-circuit termination would match the input to 50ft to within 2:1 or better at most frequencies in the pass band. However the bandwidth was only on the order of a few percent, as is typical of TEM-line antennas.
A useful technique for extending the instantaneous bandwidth of some of the TEM-line antennas in this study was to reduce the inter connecting delay lines to approximately one-half of an electrical wave length, and to feed alternate elements from opposite directions. This f obtained the required in-phase current flow in all radiating elements with a total transmission-line length of approximately half the length required with straight serial feeds. Since the instantaneous bandwidth of a short TEM-line antenna is controlled principally by resonance in the transmission-line feed network, this technique was found to nearly double that bandwidth. CHAPTER V SUMMARY
A novel low-profile antenna called the TEM-line antenna has been introduced. Because of its simple, rugged construction, it is espec ia lly applicable to aerospace vehicles in the VHF and higher frequency ranges. Its low-profile character is a consequence of its construction as an array (usually linear) of electrically small loop or fractional- loop radiating elements. Its impedance match at its input terminals is obtained from a resonance occurring in the TEM-mode delay-line seg ments interconnecting the separate elements, and although the band width of this resonance may be as small as l%-2%, i t is tunable over ranges of 10% to 20%.
An analysis of the TEM-line antenna was performed by representing
its various parts as sections of equivalent transmission lines with
appropriate characteristic impedances. Mutual coupling was included
as a voltage generated within each radiating element by the currents
flowing in each of the other elements, subject fo the restrictions
that a ll elements were identical and small enough to consider the
current to have uniform amplitude and phase throughout each element.
The effects of radiation from the loop elements were included by placing
shunt resistances in the equivalent transmission lines at points cor
responding to the right-angle bends of the square loops.
Using an iterative technique, computer solutions were obtained
92 93 for the voltages and currents at various locations throughout the an tenna for a wide range of frequencies and reactive terminations. From these solutions it was possible to obtain the far-field pattern and input impedance of the TEM-line antenna, both of which agreed closely with measured data.
Many of the antenna designs measured in the experimental phase of the program used mechanical adjustments, such as sliding short- circuits or adjustable trimmer capacitors, to obtain the desired re active termination required for a low input voltage standing wave ratio.
A notable exception was the voltage-tuned TEM-line antenna which combined the advantages of small size and remote electrical tuning capability. Tuning was accomplished with a single capacitance diode termination. VSWR was 1.0 at 265 MHz and could be adjusted to a value below 2.0 at any frequency between 250 MHz and 275 MHz by means of the control voltage on the capacitance diode.
Other TEM-line antennas included in this study showed the fe a s ib ility of completely flush-mounted elements such as on a leading or tra ilin g edge of a wing or fin ; multi pi e-turn loop elements; and orthogonal pairs of elements for crossed-1inear or circular polarization.
To summarize,the TEM-line antenna should be useful in applications such as aerospace antenna systems because of its low-profile, its in herent mechanical strength, its flexibility, and its minimal require ments for interio r space. E lectrically, i t offers a useful compromise
between conventional resonant antennas which are awkwardly large in the 94
VHF spectrum, and electrically-sm all antennas which present poor in put impedance and/or efficiency to the remainder of the r f system.
Even though the instantaneous bandwidths of the TEM-line antennas shown here were small, they were tunable over useful frequency ranges for many types of r f systems.
A fru itfu l area of future TEM-line antenna research would be to expand the bandwidth potential, both for the basic radiating element and for TEM-line arrays. APPENDIX A COMPUTER CHARACTERISTICS
The purpose of this appendix is to describe the hardware and software associated with the IBM Minimal Informer digital computer used for a ll of the machine computations performed for the analysis of the TEM-line antenna. The description is given in sufficient detail that programs discussed elsewhere in this report are understand able.
Registers: The central processor has a total of 12 programmable registers and a core memory of 4096 37-bit (plus parity) words. The registers and their functions are tabulated in Table 1. Note that register lengths are not a ll the same. High-order bits are lost when data are transferred from memory or a long register to a shorter register. When data are transferred from a short register to a longer register, high-order bits are dummied-in.
Number system: The number system used by the computer is 36-bit
binary fractional magnitude with sign as shown below.
Bit Bit Bit 27 26 1 S I Magni tude G N
The largest number is 1-2"^®. The smallest is -(1 -2 “^ ) . Minus zero 96
TABLE 1 CENTRAL PROCESSOR REGISTERS
Name Uses Length Address zero register source of zero words, always zero, read only 37 bits 70000 * index register 1 counting, address modifica- ti on 12 bits 70001 ft index register 2 counting, address modifica- t i on 12 bits 70002ft index register 3 counting, address modifica tion 12 bits 70003ft index register 4 counting, address modifica- t i on 12 bits 700048
A register accumulator, addition sub traction multiplication division, shifting,logical operations 37 bits 700108
Q register quotient register, multi pi i cati on di vi si on ,shi f ti ng 37 bits 70011ft ..... program counter holds address of next in struction to be executed 15 bits 700138 program counter holds return address from 70014s store subroutines 37 bits also mem. loc. 14s . display register console indicator light dis play, write only 37 bits . 700168 switch register 1 console switch register, read only 37 bits 70020ft switch register 2 console switch register, read only 37 bits 70021s 97 is a valid number and in fact results when a positive number in the accumulator (A) register is reduced to zero by an addition or sub traction operation.
Addressing and instruction format: Memory and registers can be referenced by 15-bit addresses either directly or by means of in dexing. I f indexing is specified, the contents of the specified in dex register are added to the address part of the instruction word to obtain an effective address. The instruction format is shown below.
37 36 31 30 28 27 16 15 1
OP Code Y e a
Bit positions 31 through 35 contain a 6-bit operation code. Bit positions 1 through 15 (a) contain the 15-bit address of an operand in memory reference instructions. Bit positions 28 through 30 (y) specify the index register (1,2, 3 or 4)whose contents are to be added to a before the instruction is executed. A y of zero specifies no indexing.
Sense instructions: The 3 part of the instruction word (bits.
16 through 27) performs different functions depending on the instruction.
One of the functions is to specify an indicator or condition to be set, reset or tested by the sense instructions. Table 2 gives the sense functions and corresponding 3 codes. The SEN (sense), SNS (sense and set) and SNR (sense and reset) instructions have operation codes
05g, 06g and 07g, respectively. I f indexing is specified by y, the 98
TABLE 2 BETA CODES AND SENSE FUNCTIONS
Octol 3 code Function SNR SNS SEN
2-76 even numbers-I/O converter 1 in use X
100 overflow alarm X
102 interpret sign mode X X
103 continue on I/O error X X n o sense switch 1 X
111 sense switch 2 X
112 sense switch 3 X
113 sense switch 4 X
114 sense lig h t 1 XX X
115 sense lig h t 2 XXX
130 I/O converter alarm X
136 break occurred X
140 allow interrupts X X
141 allow I/O interrupts X X
142 allow CPU interrupts X X
153 write EOF X X
155 memory alarm X
156 b it error X
144 I/O converter deselect* X
174 generate b it error* X
176 complement memory parity XX ♦Transfer is forced 99 contents of the specified index register are added to a to form an effective address. For the SEN instruction, i f the condition being tested is met or the indicator being tested is set, the effective address is placed in the program counter causing a transfer to occur.
Otherwise the program counter is incremented by 1 and the program continues in sequence. For the SNS and SNR instructions the specified indicators or conditions are set and reset, respectively. For these instructions if a change in state of the specified indicator occurs the computer transfers to the effective address, otherwise i t con tinues in sequence.
Overflow: For instructions where accumulator overflow is pos sible, that is , the result of executing the instruction is greater than or equal to unity and will not fit in the A register, the action taken by the computer is controlled by the $ part of the instruction word as specified in Table 3. Two exceptions are the add 3 (ADB) and subtract s (SBB) instructions where the equivalent of 3 = 5g is forced.
Table 4 describes the operation of the central processor in structions. In order to keep the table reasonably short the following conventions have been used:
1) a means the value of the a part of the instruction word.
I f the instruction has been indexed, i t means the sum
of the a part of the instruction word and the contents
of the specified index register. TABLE 3 OVERFLOW CONTROL*
Bit = 0 Bit = 1 Bit 18 clear OA before instruction execution no action
Bit 17 set OA on overflow no action
B it 16 set OA on overflow continue on and halt overflow
* ADB and SBB force equivalent of 1012 for bits 18-16 TABLE 4 SUMMARY OF COMPUTER INSTRUCTIONS
In.st O verflow Index Address O verflow L Code Indexable Rcj)i*atjbie Possible Function A Q Regs e C onditions Com m ents Hits 16-18 do not control C (o ) * fi CiA) C(<* ) * 0 C(o) positive and ADD 24 Yes Yes Add B-j;a c - Action on overflow, O.A. 1 carry from bit Is set and computer continues [CS)‘ rt ,.,2 36 o f adder in sequence.
ADD 12 Yes Yes Yea Add C(A) . C(o ) Same sign and Bits 18-18 control action on 1 carry from o verflo w . btt 36 of adder
ADM 13 Yea Yea Yea Add C(A) * |C(o )| Sign of A posi Bits 18-18 control action on Magnitude tive and 1 carry o verflo w . from bit 36 of adder.
CAM 11 Yes Clear and Add * | C ( o ) | Magnitude
C l A 10 Yea Clear aud Add C(«)
C !£ 14 Yea Clear and Sub" 1-36: C(o)i_3G tra c t 37: Complement C(o )37
CSM IS Y i» C le a r and - |c ( « ) 1 SuU. Magni tude
C Y i. 35 Yes Cycle fxrft A, Q cycled Signs Included; If a • 0, con- * lon g le ft a mud tents of A and Q unchanged. 128 places
CYS 34 Yes Cycle Left A cycled left Sign Is not included; if a * 0 , Stiurl O mod 126 contents of A unchanged
DVD 22 Yes Yea Divide Remainder Quotient |C(A)|--|C(o )| C (A) t C(a); bits (16-18) control overflow .
22-36; 1-15: |C|A)i_l5p C(A) * Bus D V f 26 Yes Yes Divide Fast Remainder Quotient 16-16 control overflow % 1-21: 'Aero 16-36: ‘Aero !<*•>.-161
DV1. 23 Yes Yes Divide Lung Remainder Quotient |C(A)| .-|C(o)| C(A,Q) . Cfa); bits (16-18) con trol overflow. Stop computer; complete LO con H I T 00 Halt verter operations which are in process. ID X 53 Ixjud Index In 4 index register computers ul-12-I*’ * 11 > - 4, > t 1 - J R egisters fi -1 > 0.0-0 0.1*1 U1A 03 Yes Yes Logical Add Logical Sum C(a ) . C (A) 1.0*1 1.1*1 b its 1-37
1UM 02 Yes Yes li^lcal Logical Pro 0x0-0 1x0*0 M u ltip ly duct Cfa )‘C(A) 0x1*0 1 x 1 - I lutb 1-37 TABLE 4 (continued)
Inst O verflow Iiuji'X A iM ruas O verflow a tt Ct*h* InitcAjhk' Ittpeut able' Possible Function A Q H tgu Conditions Comments
H iN 04 Yes Yea Logical Nega l's comple O ita 1-37 tio n ment of C(o )
l o t ) 51 YeS loa d Cifii placed in addressable rcg
Process is performed twice for Memory Teat C(„Z) - 02 H IT 111 u \ each m em ory location at a 16- C(o2 < /II) - 02 */) (special usee rate for{a j) lim e s. I.U1>I,2 U 2 C|«2 .2/11)- a 2 . zpi sequence) tic . (1 15) High Yea M u ltip ly 3i>) f-t*w M I.K 27 order bits. C*°^I-lt>’ CiA,l- l ’J’‘S,lll‘ ol Pro£!uCl Fast • udet lilt:.. (1G-3U) Z e io (1 2 1 ) Zero in A and Q
1,11 Jt 21 Yea Multiply and High order bits low order bits C(o ) * C(A) Sign of product Kuund 1 rounded If In A and Q. «36 * 0
M I.Y 20 Yea M u ltip ly Miglt order bits low oider bits C(« ) • C(A) Sign uf product In A and Q
MOV 1)2 Yea Move 11 UPT, MOV C(<» loca tio n )1/) .. If H P T then contains » 1, m odify yp by last ** reference
ht*SK 5Si Yea Y. Mask C(A) . C(CJ) . C(o) . CM) Q Is l'u com plem ent of Q. 9
HUM 37 Yea N orm alise Itesult of shifts 1-15 Number ol. tihift C(A) left until 1 appears With 0's inaei - . h ills . 10-37: O'b in b it 3G if C(A) • 0, N - 36. lt d in vacated Sign bit unchanged. positions
HPA 1,4 Yea Yea lieplace 1-15: C (A )|. |g Address 15-37; ujichungcJ
Yea 0 - step counter. I t l l 01 Hepeat " sl* 1 unchanged A) are negative tills (16-18) do nut coutrol M ill 2t> Yea Yes Subtract ) -1* C(A) c i,id 1 c a rry from Acihm on overflow; O.A. Is set and beta e *°>l-12 - ^ 1 bl : 36 of adder computer continues lu sequence.
:a»M 17 Yea Yea Yea Subtract C(A)-|C(«)| A la negative Dlls (16-18) control over Magnitude a n d lc a rry flow action. t romblt posl- I oo 36 of adder
t»KN 05 Yea Sense
S ill. 30 Yea Yea Shllt Left C(A) ahllted I shlftedfroui’ Sign not alilflcd,Inject Q'» to le ft a mod 120 b U 36 of A rigid of A register,bits (16- r Dglster 18) c o n tro l o verflo w action. • • • Logical "A N D ” f • lacteal "OH" TABLE 4 (continued) , . = z = = < lust O verflow lnde> A ddress O verflow I L \* lr Hrpi'UtabU- P ossible Function A Q Regs. e Conditions Com m ents
S llR 32 Y es Shift Right C(A) shifted Sign not included;;nject 0's to left. rig h t o m o d 128 i»LL 31 Yes Yes Shift Left C tA.Q ) £tiiifted 1 is shifted out Signs not shifted; inject O's to U>ng le ft o m yi 128 of position 36 of right of Q register; bits (16- A re g is te r 181 control overflow action
SNR 07 Yes Sense and (£)» l,a - ‘ PC and 0 —3 Reset (0) * 0, 1 ♦ PC — PC
HUS 06 Yes Sense and WJ-O, o — PC and 1 — 0 Set ( 0 ) * I , 1 ♦ P C - PC
S R L33 Yfc£ Shift Right C(A an d Q) Signs excluded; Inject O's Ixmg Sin ft r tght o to left of A register. mod Y,IB places
STIt SO Yes Yes Store C(A)
SUB 16 Yes Yes Yes Subtract C(A D ifferent sign and Bits (16-16) control overflow 1 carry from bit action. 36 of adder If RPT, TRC C (o)>C(A), PC * 1 TRC 47 Yes Yea Compare C(A) then 1^ con C (o ) < C (A ). PC v 2 tains repeat C (0) « C(A). PC < S count remain ing
T R L 4 1 Load PCS i . - I > PC ♦ 1 -P C S r e g . o - PC R egister and T ransfer
TRN 46 Yes Transfer on (A>37 * 1 ,0 — PC Negative (A)37 - 0, (PC v 1)- PC
T K P 44 Yes T ra n s fe r on (A)37 - O .o -P C P ositive (A)37 - 1, (PC a 1J-PC
TKS 42 Transit! to C (PCS) re g —PC PCS Register
TRU 40 Yes Transfer Un u I: o - PC conditional !» • < * - i * TU X 43 Transfer on II I1 - 1) , 0 ; If 1 (> * h - 1 > 0. a - P C Index , o * 1 ) . ♦ 1)
TR Z 46 Yeb l u n s fiT on | A - O.o - PC Z ero |A v-0, PC t I- PC 104
2) C(a) means the contents of address a.
3) C(x) where x is a register name means the contents of the
register x.C(A), for example, mean the contents of the
accumulator (A) register.
4) I f a register column is le ft blank the corresponding reg
ister is unaffected by the instruction. For instructions
referencing two index registers,I(y)and I(y+1), if y is 4,
y + 1 is 1.
Several instructions are not completely specified in Table 4.
Three of these, the sense instructions, have already been described.
Another which needs further discussion is the repeat (RPT) instruc tion. The repeat instruction causes the instruction immediately fo l lowing i t to be executed a + 1 times. After each execution of the instruction its a is increased by 8 of the repeat instruction. 8 of the repeat instruction is also placed in index register 4. I f the re peated instruction calls for indexing, i t is indexed normally before the fir s t execution. Both instructions remain unchanged in memory.
For the sequence RPT, TRC the TRC (transfer on compare) instruction is repeated until the contents of the accumulator are less than or equal to the contents of address a for a maximum of a (of the RPT) + 1 times. I f the TRC is repeated the specified number of times, one in struction is skipped and the computer continues in sequence. I f the contents of address a are equal to the contents of the accumulator, two instructions are skipped. I f the contents of the accumulator are 105 less than the contents of address a the computer continues in sequence.
The remaining repeat count is placed in index register 3. For the
RPT, MOV (move) sequence the y& address is indexed by the contents of index register 2. When the sequence is completed the Q register con tains the last address where data was extracted.
Timing for instructions is given in Table 5.
Interrupt: A number of conditions can cause an interrupt. When an interrupt occurs, the contents of the program counter are placed in memory location 15g and 00200g is placed in the program counter so that the computer transfers to location 200g. For an interrupt to occur the allow-interrupts indicator (140g) must be set. When an
interrupt occurs the allow-interrupts indicator is reset to prevent
any further interrupts until the indicator is set again by the pro
gram. In addition, for interrupts to occur, at least one of two
other indicators must be set. The allow-CPU indicator must be set for
memory parity or bit errors to cause interrupts. Setting the allow-
1/0 indicator enables a number of I/O conditions to cause interrupts.
Input/Output: Input/output operations are handled by a separate
processor called the I/O converter. The relationship between the I/O
converter, the central processor and the I/O devices is shown in
Fig. 55. When an I/O instruction is recognized by the central pro
cessor it is transmitted to the I/O converter for decoding and pro
cessing. The central processor normally continues with the succeeding
instructions while the I/O converter independently processes the I/O
instruction. If the I/O converter is already in use when the central 106
TABLE 5 INSTRUCTION LIST
Average Average OP Time OP Time Code Mnamorwc Name ((/SCC ) Code Mnemonic Name (r/sec.)
00 tyj Hair 22.67 in: 40 trd Unconditional transfer 17.33 01 E£I Repeat 21.33 < 41 m Load PCS and transfer 25.33 a 02 L£M* Logical Multiply 22.67 d; « TRS Transfer ro PCS 25.33 5 03 LQA* LdflidflLAfiy 22.67 at! 43 TRX Transfer an Index 29.67 w l££l* Loaical Negation 22.67 S. « IRE Transfer on (+1 A 20 24 24 bni 05 SEN Sense 10 CIA Clear and Add 22.67 50 SIR* Stars 22.67 11 CAM Clear and Add 51 LOP Load 33.33 Magnitude 22.67 w 52 MOV* Move 33.33 12 ADD* Add 24 < 53 LB2S Load Index 18.67 a 13 ADM* Add Magnitude 24 sj 54 RPA* Rea lace Address 28 123 BYL pivids. Laos 425 «i 65 Spare 41.33 SKP Skio 21.3 §!! 24 APB* Add Beta o 66 5|25 SM* Subtract Beta 41.33 ° ' 67 6SE BflSktC MS 21.3 §j26 P.Yf Divide Fast 208 27 m i Multiply Fast 177.3 70 RAN Read Alphameric 21.3 71 RSY Regd Reygrje__ 21.3 30 stu. Shift Left 26.67 4n in, 73 RQK Rood Octal 21.3 31 i U Shift Left Loop 26.67 4n < 73 sea Search 21.3 32 SHR Shift Right 26.67 4n t i 7* WAN Write Alphameric 21.3 21.3 3 ! 33 SSL Shift Right.Lang 26.67 4n o i 75 WWA R«j»r.it? 5! 34 CYS Cycle Short 26.67 4n it 76 WOK Write Qetal 21.3 t-f ^ CYi Cycle Long 26.67 4n ' 77 m i Rewind 21.3 — ■ 36 Spare 5:37 NRM Normalize 28 4 (n-1) * Repeotable instruction*. If instruction is repeated, subtract 1.33 H sac from averoga tima for each rapetition oftar fha first. Indexing, if any, applies to first repetition only. Non index able instruction. If instruction is indexable and it is indexed, add 2.67 M sec u average time. 107 OISK MEMORY OISK ADDRESS DATA (8 BIT CHARACTERS) DEVICE CENTRAL I/O SELECTION CONVERTER (DIGITAL PROCESSOR DATA DATA MUX.) TELETYPEWRITER (3 7 BIT WORDS) ( 8 - BIT CHARACTERS) DATA PARALLEL CORE TO SERIAL MEMORY CONVERTER PLOTTER ANALOG TO DIGITAL CONVERTER PAPER TAPE READER PAPER TAPE PUNCH Fig. 55.— Digital computer block diagram. processor encounters an I/O instruction, the central processor is held up until the I/O converter is free. Data are transferred through the central processor between the core memory and the I/O converter as 37-bit words. I/O instruction format: The I/O instruction word format is shown below. 37 36 31 30 22 21 16 15 ____1 OP Code K J a 108 The a part (bits 1 through 15) specifies the starting memory address for data transfer. The K part specifies the number of words to be transferred. For write (output) instructions, bits 22 through 30 are used. For read (input) instructions, bits 22 through 29 are used to specify either a word count or block count as determined by b it 30 being a zero or a one. Blocks are made up of arbitrary numbers of words and are separated by block marks. Block marks are explained below. The J part of the instruction word is the device address used by the device-selection multiplexer. I/O registers: Two registers and a memory location are avail able for programming. The I/O instruction register contains the op code and J address of the current or last I/O instruction. The K part contains the current contents of the word or block counter and the a part contains the memory address of the next memory location from which data are to be taken or into which data are to be placed. When the I/O instruction is completed the address is one higher than the last location accessed. The address of the I/O instruction register is 70030g. The maintenance register (address 70024g) contains information on the current status of the I/O converter equipment and is not gen erally used for programming. Memory location 10g contains the current or last I/O instruction executed. Data character format: Data are transferred between the I/O con verter and the address-selection multiplexer as 8 -bit characters. The 8 bits are designated form high order to low order, as P (p arity), C (control), I 2, I ] , D3, D2, D-j, Dg. For output the P-bit is generated 109 by the I/O converter to give the character odd parity. For input the I/O converter tests for odd parity. A C-bit of 1 indicates a data character. A C-bit of 0 indicates a control character. Meaningful control characters are BLS (block s ta rt), BLE (block end), EOF (end of f ile ) and STOP. The codes for the control characters are given in Table 6 . TABLE 6 INFORMER CONTROL CHARACTERS d2 BIT P C h *i °3 D1 Do BLS 1 0 i 0 1 0 1 1 BLE 0 0 i i 0 1 0 0 EOF 1 0 i i 0 1 0 1 STOP 0 0 i 0 1 1 1 1 Each write instruction causes a BLS character to be transmitted at the beginning of output operation. Data characters are generated by breaking down the words to be output. The data characters are fol lowed by two BLE's. EOF's can be substituted for BLE's by executing a SNS 153q prior to execution of the output instruction. For input, 8 -b it data characters are assembled into 37-bit com puter words. The input data may be divided into blocks for a read- by-blocks instruction (b it 30 of the read instruction word = 1 ) by no BLS's at the beginning of each block and two BLE's or EOF's at the end of each block. I f EOF's are used and the I/O interrupt has been en abled by execution of SNS 140g and SNS 141g instructions, an interrupt w ill occur when the EOF's are read. For a read-by-words (b it 30 = 0) instruction, block marks may or may not be present. The specified num ber of words is read regardless of how the data are divided into blocks. If a STOP character is read, the input operation is terminated without regard for word or block counts. Data characters are transferred between the I/O converter and the address selection multiplexer in one of three modes; as octal data, as alphanumeric data in the interpret-sign mode, and as alphanumeric data in the not-interpret-sign mode. For octal output 37-bit com puter words are broken into thirteen data characters corresponding to the alphanumeric representations of the sign and the twelve octal digits making up the remainder of the word. The alphanumeric representation is obtained by setting the I 2 * I] and D3 bits equal to HOg and the D2, D-j and Dg bits equal to the octal number. Transmission starts at the sign b it and proceeds to the low order end of the word. For alpha numeric output in the interpret-sign mode the 37-bit word is broken into seven data characters. The sign b it is output as an alphanumeric zero or one as for octal output. The remainder of the word is output six bits at a time. Each group of six bits forms I 2 through Dg of the corresponding data character. The interpret-sign mode is set prior to the execution of an alphanumeric I/O instruction by the execution of a SNS 102g instruction. In the not-interpret-sign mode the sign ■ \ I l l b it is ignored and the 37-bit word is broken into six data characters. For input, the process is reversed and data characters are assembled into 37-bit computer words. For input in the not-interpret-sign mode the sign b it is made zero. Blocks of data to be read need not form an integral number of words. I f block mark characters are encountered, the I/O converter completes incomplete words by supplying low-order zeros. Incomplete words are sim ilarly completed i f a STOP character is read. I/O instructions: A list of input-output instructions is given in Table 7. TABLE 7 INPUT/OUTPUT INSTRUCTIONS Octal OP code Mnemonic Function 70 RAN read alphanumeric 72 ROK read octal 73 SCH search 74 WAN write alphanumeric 76 WOK write octal The search instruction performs an automatic search for data defined by up to th irty descriptors. I t hasn't been used in any of the pro grams published in this work and w ill not be described. Magnetic tape instructions have been omitted since the present computer system does not have magnetic tape. 112 Disk memory: The disk memory has a capacity of 20.5 m illion 8 -b it characters. The data are divided into ten thousand tracks of 2050 characters each. The data are written and read by a single pair of heads which are mechanically positioned to the desired track. This is accomplished by the execution of a write octal instruction with a J address of 30q and a word count of one. The a address specifies the location of a four-digit BCD track address. Access time is be tween 100 and 800 milliseconds. Data can be written by a write alphanumeric instruction with a J address of 30g. Old data are ef fectively erased. Sim ilarly, data may be read by a read alphanumeric instruction. I f more data than can be contained on one track is specified by the word or block count the heads are automatically moved to the next sequential track and the input/output operation is continued. Teletypewriter: The teletypewriter uses a modified ASCII code (American Standard Code for Information Interchange) as shown in Table 8 . The ASCII code bits actually used by the teletypewriter are given as column headings and down the le ft margin in the table. The internal code representation is given in octal along with each character. For output, bits I 2 through Dq of the output character are used directly to form bits bg through b-j, of the ASCII character. Bit by is formed by either complementing or duplicating b it bg. The mode is set to "complement" at the beginning of the output operation and does not change so long as non-zero (bits I 2 through Dq) characters are output. A zero- character is not transmitted to the printer but switches the mode to "duplicate". The f ir s t non-zero character is transmitted to the printer 113 TABLE 8 AMERICAN STANDARD CODE FOR INFORMATION INTERCHANGE ASCII Informer Control Printing characters Control Characters Characters . space 0000 0020 (00)40 (00)60 DC1 0001 0001 0021 (00)41 (00)61 DC2 0010 0002 0022 (00)62(00)42 DC3 0011 0003 0023 (00)63(00)43 DC4 BLE 0100 0004 0024 (00)44 (00)64 WRU EOF 0101 0005 0025 (00)45 (00)65 0110 0006 0026 (00)46 (00)66 BELL 0111 0007 0027 (00)67(00)47 1000 0010 0030 (00)50 (00)70 1001 0011 0031 (00)51 (00)71 LINE FEED 1010 0012 0032 (00)52 (00)72 Fsr BLS 1011 0013 0033 (00)53 (00)73 1100 0014 0034 (00)54 (00)74 CAR. RET. 1101 0015 (00)55 (00)750035 SHIFT OUT mo 0016 0036 (00)56 (00)76 SHIFT STOP RUBOUT 1111 0017 0037 (00)57 (00)77 114 in the duplicate mode and the complement made is restored. Trailing zero-characters are effectively ignored. Control characters are ignored. For input, characters from columns 0 through 5 are converted in a way analogous to the output conversion. Columns 0 and 1 produce two input characters for each keystroke, a zero-character and a character having I 2 through Dq equal to 65 through b-j. Columns 2, 3, 4 and 5 produce single characters with I 2 through D0 equal to bg through b-j. Columns 6 and 7 produce control-characters with I 2 through D0 equal to bg through b-j. The only characters from columns 6 and 7 that can be generated from the keyboard are the RUBOUT and the STOP, the la tte r of which is generated by the "CLR KYBD" key. The "CLR KYBD" key also generates an interrupt pulse and w ill cause an interrupt i f the I/O interrupt has been enabled by SNS 140g and SNS 141 g instruction. Plotter: The plotter has four basic pen motions: .01 inches in the plus-x direction, .01 inches in the minus-x direction, .01 in ches in the plus-y direction and .01 inches in the minus-y direction. In addition the pen can be raised or lowered. Each data character output to the plotter produces one or more functions as indicated by Table 9. X- and y-motion can be combined to produce 45-degree diagonal motions. Zero-characters and control-characters are ignored. Parallel/serial converter: In the parallel-to-serial converter, bits I 2 through D0 of the data characters are converted to serial NRZ data at a 500 KHz rate. A clock is provided which makes a one-to-zero transition approximately lys after a change in the data line and at 115 TABLE 9 PLOTTER FUNCTIONS Bi ts Function h h D3 °2 D1 D0 X Pen up X Pen down X 1 increment: plus-x X X 1 increment: 45 degrees X 1 increment: plus-y X X 1 increment: 135 degrees X 1 increment: minus-x XX 1 increment: 225 degrees X 1 increment: minus-y X X 1 increment: 315 degrees least lus before the next change in the data lin e. Data are not trans mitted continuously, but are limited by the maximum character rate of the I/O converter which is about 75 KHz. Control-characters are sup pressed. A/D converter: The analog-to-digital converter accepts a 0 to 1.0 volt analog signal and converts i t to 10-bit binary 0 to 1777q* The conversion is started when the first data character is read. Four zero- characters are sent followed by a character containing the two high- order octal digits and a character containing the two low-order octal digits. Data may be read at approximately a 5 kHz word rate but the input bandwidth has been limited to about 5 Hz to minimize noise. APPENDIX B ASSEMBLER Introduction The assembly-1anguage programming system used with the IBM min imal informer computer is documented in detail in an internal publica tion of the ElectroScience Laboratory, [34] and the description con tained in this appendix is only sufficient to guide an experienced assembly-language programmer in reading the computer programs con tained in this report. The instruction format in this language is less rigid than in most assemblers in that the various fields of the instructions are not required to start or end in specific columns. The fields are sep arated by one or more spaces. Instruction Fields The four fields of an instruction are: a) Label (optional) - one to six alphanumeric characters begin ning with an alphabetic character b) Operation (always required) - a two-, three-, or fiv e -le tte r alphabetic code specifying either a machine operation, an ex tended mnemonic to be interpreted in terms of machine functions, or an assembler directive causing one of the following actions: i) assignment of block storage ii) assignment of data storage 117 118 iii) linkage to external files iv) termination of assembly. c) Operand (required except for op-codes of HLT, TRS, or END) - a variable length fie ld composed of subfields separated by commas. The subfields are: i ) Alpha - address ii) Gamma - index register reference, if used i i i ) Bfeta - increment or second address, i f used. iv ) K - word or block count v) J - device number, i f used vi) textual material Zeros are supplied for any missing subfields of the operand. d) Comments (optional) - all text material following the third field on any line. Instruction Types In the following examples of various types of instruction formats, the label fie ld , i f used, must not be preceded by a space character. A central processor instruction has the form: Label OP Alpha, Gamma, Beta comments The move instruction is an exception to this form because the length of its second address requires that the Gamma and Beta portions of the address be taken together as a single subfield. I t has the form: Label MOV Alpha, Gamma-Beta 119 An input or output instruction has the form: Label OP Alpha,K,J Comments A block storage directive assigning N words of storage has the form: Label BS N Comments A tabulation directive assigning values to successive words has the form: Label TAB N1 ,N2,N3 ••• Comments A directive to assign N words of alphanumeric data with six characters per word has the form: Label Alpha N XXX... comments A directive to designate that an external f ile is required by a program where that file is indexed in the disc file directory by file name and user name, has the form: Use file,name comments The directive which indicates to the assembler that the end of the sympolic program has been reached has the form: END Special Symbols The following special symbols are recognized by the assembler: * In column 1 - entire line is comment * In column 1 - entry point for this program $ Prefixed to address - external routine * As address - current location ** As address - to be supplied by program 120 D Prefixed to number - disk address Unmodified number - decimal constant " " Enclosing characters - Alphanumeric constant 1 ' Enclosing number - octal constant [ ] Enclosing number - decimal lite ra l from lite ra l table [ " ] Enclosing number - octal literal from literal table [" "] Enclosing characters - Alphanumeric lite ra l from literal table ( ) Enclosing characters - Machine register or sense indicator An address reference may be composed of one of the above addresses plus or minus a constant to reference unlabeled instructions or storage. Hardware Operation Codes The following mnemonics are recognized as machine operation codes: Misc. Class Transfer Class HLT Halt TRU Transfer Unconditional RPT Repeat TRL Transfer and Load PCS LGM Logical Multiply TRS Transfer to PCS LGA Logical Add TRX Transfer on Index LGN Logical Negation . TRP Transfer on (+) A Reg. SEN Sense TRZ Transfer on Zero A Reg. SNS Sense and Set TRN Transfer on (-) A Reg. SNR Sense and Reset TRC Transfer on Compare 121 Add Class Shift Class (cont.) CLA Clear and Add CYL Cycle Long CAM Clear and Add Magnitude NRM Normali ze ADD Add \ Store Class ADM Add Magnitude STR Store CLS Clear and Subtract LOD Load CSM Clear and Subtract Magnitude MOV Move SUB Subtract LDX Load Index SBM Subtract Magnitude RPA Replace Address ADB Add Beta MSK Mask SBB Subtract Beta I/O Class Multiply Class RAN Read Alphanumeric MLY Multiply ROK Read Octal MLR Multiply and Round SCH Search DVD Divide WAN Write Alphanumeric DVL Divide Long WOK Write Octal DVF Divide Fast MLF Multiply Fast Shift Class SHL Shift Left SLL Shift Left Long SHR Shift Right SRL Shift Right Long CYS Cycle Short 122 A complete description of these hardware functions is presented in Appendix A. Extended Operation Codes The following mnemonics are recognized as extended operation codes in which additional information such as device number or sense indica tor is supplied by the assembler TBR - Transfer i f break occurred TIU - Transfer i f I/O in use TL1 - Transfer if sense light 1 on TL2 - Transfer if sense light 2 on TOV - Transfer if overflow indicator set RAB - Read alphanumeric by blocks RAD - Read alphanumeric from disk RAT - Read alphanumeric from typewriter RDB - Read disk by blocks RTB - Read typewriter by blocks WAT - Write alphanumeric on typewriter WAD - Write alphanumeric on disk WKD - Write octal disk (reposition R/W heads) Machine Registers The following symbols are recognized as machine registers: (Z) - Zero register (1X1) - Index register 1 (1X2) - Index register 2 (1X3) - Index register 3 (1X4) - Index register 4 (A) - A register (Q) - Q register (PC) - Program counter (PCS) - Program counter store (IPCS) - Interrupt program counter store (DISP) - Display register (SRI) - Switch register 1 (SR2) - Switch register 2 (M) - I/O maintenance register (10) - I/O instruction register (D) - Disk address register Sense Codes The following symbols are recognized as sense indicators (IU) - I/O in use (OVA) - overflow alarm (ISN) - Interpret-sign mode (CIO) - Continue on I/O error (SW1) - Sense switch 1 (SW2) - Sense switch 2 (SW3) - Sense switch 3 (SW4) - Sense switch 4 124 (SL1) - Sense lig h t 1 (SL2) - Sense lig h t 2 (IOA) - I/O alarm (BRK) - Break occurred (AI) - Allow interrupt (AIO) - Allow I/O interrupt (ACPU) - Allow CPU interrupt (VJEF) - Write end of f ile Data Formats Floating point numbers are represented as a signed binary fraction with a nine-bit exponent. The least-significant nine bits of the word represent the power of two multiplier plus 400g. Thus the octal floating point representation of -1.0 would be written as -'400000000401' where the primes denote octal notation. ASCII control characters are represented internally as 12-bit characters with zeros for the high-order six bits. This is done to distinguish them from the ASCII printing characters which are represented as six-bit characters. APPENDIX C LIBRARY INDEX The system library contains a large number of useful subroutines which may be called from user programs to perform a variety of commonly needed functions. These subroutines are grouped in several file s under the user name "LIB", and the subroutines in a given f ile share some common features or applications. For example, the f ile "PLOT, LIB" contains a ll of the library subroutines which pertain to the on-line plotter. Most subroutines are called through a TRL instruction (transfer and load PCS reg ister), which utilizes the PCS register (program counter storage) as explained in Appendix A, to indicate where the re turn from the subroutine should be directed. In addition, if an index register is specified in the TRL in struction, that index register will be loaded with the address specified in the Beta part of the instruction. This feature is used in many subroutine calls to indicate to the subroutine where i t must find or place additional data beyond that which i t finds or places in the A- and Q-registers. Index register 1 has been chosen for this purpose in a ll library subroutines which require such additional data. The re maining three index registers are undisturbed by library subroutines, but the contents of index register 1 may be lost i f used in the calling sequence. 125 126 The following lis t of available library subroutines is arranged according to the grouping within the several file s of the lib rary. This lis t shows the appropriate calling sequence for each library subroutine, along with a brief description of the action taken by the subroutine. The following calling sequences may be used with UTIL, LIB: TRL $FADD,1, ADDEND Floating add ADDEND to accumulator. TRL $FSUB,1,SUBTR Floating subtract SUBTR from accumulator. TRL $FMLY,L,MPLR Floating multiply accumulator by MPLR,return result in accumulator. TRL $FDVD,1,DVSR Floating divide accumulator by DVSR, return result in accumulator. TRL $AFTR,1,TRADD Arm floating trap. Spill will force transfer to TRADD, cell 16 will contain address of instruction causing spill. TRL $DFTR Disarm floating trap. TRL $TFTR Test floating trap. Return skips one instruction i f trap is armed. 127 TRL $PRFL Print floating point number in accumulator, using format -1 .OOOOOOOOE-OO TRL $PRI Print integer in accumulator, using format -1234 TRL $CNVTF,1,BUFFER Convert 5 cells of teletype code beginning with BUFFER into a floating point number in accumulator. Return skips one instruction i f conversion was successful, continues in sequence i f not. TRL $CNVTI,1,BUFFER Same as $CNVTF, except an integer is returned in the accumulator. TRL $INTFL Convert integer in accumulator to a floating point number. TRL $FLINT Convert floating point number in accumulator to an integer. For debugging purposes, i f sense switch 1 is on, floating point arithmetic routines will halt just prior to the return to the calling program. Note that other library routines may use floating arithmetic and have several halts i f sense switch 1 is on. The following calling sequences may be used with CMPLX,LIB: 128 TRL $CL0D,1,Z Load A&Q registers with the complex number in cell Z and the next following c e ll. TRL $CSTR,1 ,Z Store the complex number in A&Q registers in cell Z and the next following c e ll. TRL $CADD,1,ADDEND Complex floating add ADDEND pair to A&Q registers. TRL $CSUB,1,SUBTR Complex floating subtract SUBTR pair from A&Q registers. TRL $CMLY,1,MPLR Complex floating multiply A&Q registers by MPLR pair. TRL $CDVD,1,DVSR Complex floating divide A&Q registers by DVSR pair. TRL $CMAG Return the magnitude of the complex number in the A&Q registers in the accumulator. TRL $CEXP Return the complex exponential of the complex floating point number in the A&Q registers. TRL $PRCX Print the complex number in A&Q registers, using format -1 .OOOOOOOOE-OO -J 1 .OOOOOOOOE-OO 129 The following calling sequences may be used with MATH!,LIB: TRL $SQRT Return the square root of the magnitude of the floating point number in the accumulator. TRL $SIN Return the sine of the floating point number (in radians) in the accumulator. TRL $C0S Return the cosine of the floating point number (in radians) in the accumulator. TRL $SINC0S Return the sine of the floating point number (in radians) in the accumulator and the cosine of the same number in the Q-register. TRL $EXP Return the exponential of the floating point number in the accumulator. TRL $LN Return the natural logarithm of the floating point number in the accumulator. The following calling sequences may be used with PLOT,LIB: TRL $UP Lift pen of plotter if it was down. 130 TRL $PL0T,1,X Move pen in straight line from present position to the point (X,Y) represented by the integers in cell X and the next following c e ll, provided fewer than 4096 increments are required. The pen is lowered at the new position i f i t was up previously. TRL $M0VE,1,X Same as $PL0T, except pen is not lowered. TRL $0RI6,1,X Reset the plotter origin so that the present position of the pen is the point (X,Y) specified by the integers in cell X and the next following cell. TRL $WHERE,1,X Return present position of pen as the point (X,Y) represented by the integers in cell X and the next c e ll. The following calling sequence may be used with EXRET.LIB: TRU $EXEC Return to executive system. APPENDIX D COMPUTER PROGRAMS *A PTEMS *P 1 * THIS PROGRAM PLOTS THE E-PLANE POWER PATTERN 2 * AS A FUNCTION OF FREQUENCY FOR A PARTICULAR 3 * THREE-ELEMENT* CAPACITANCE-TUNED TEM-LINE ANTENNA 4 * WITH A FIXED VALUE OF CAPACITANCE* OVER THE 5 * FREQUENCY RANGE FROM 100 MHZ TO 500 MHZ. 6 **************************************************** 7 *** INTERRUPT PROCESSOR 8 KLT $START 9 #START SNS GO**CAIO> 10 TRL CLNUP 11 SNR GO** 12 *** GET TERMINATING CAPACITANCE AND PLOT-INCREMENT SIZES* 13 *** AND INITIALIZE PLOT ROUTINE 14 GO WAT C * 1500120012*3*1 15 WAT MSG3*2 16 TRL READF*1 * CAP 17 WAT C *150012*3*1 18 WAT MSG1*2 19 TRL READF*i*XINC 131 132 20 WAT C*150012'3*1 21 WAT MSG2*2 22 TRL READF*1*XINC+1 23 WAT i'150012'3*1 24 SNS 5^+ 1 * * CAI ) 25 TRL SPL31*1*LIST 26 *** 3-D PLOT 27 TRL SPL3 * 1 * FXY 28 TRU START+1 29 *** TAKE NEW PAGE AND SET PEN TO NEW ORIGIN 30 EXEC TRL $WHERE*1*X 31 CLA X+l 32 TRZ *+2 33 TRL CLNUP 34 TRU SEXEC 35 CLNUP MOV CPCS)*PCS 36 MOV CZ)*X+1 37 MOV C8503*X 38 TRL $UP 39 TRL SMOVE* 1*X 40 MOV 41 TRL SORIG*1*X 42 MOV PCS*CPC) 43 * * * GET X AND Y 133 44 FXY MOV CPCS)*PCS 45 CLA 0*1 46 STR XP 47 CLA 1*1 48 *** GO TO PATTERN CALC IF OLD FREQUENCY 49 SUB XP+1**7 50 TRZ SCAN 51 *** GET NEW FREQ AND PERFORM INITIALIZATIONS 52 CLA 1*1 53 STR XP+1 54 TRL SFADD*1*C'600000000401'3 55 TRL SFDVD*1*C'500000000403'3 56 STR FREQ 57 STR MDATA+1 58 TRL SFMLY*1 *C '655165200376'3 2*PI/29.97925 59 TRL SFDVD*1*C'656050754400'3 .84 60 STR DATA+5 61 TRL $TMLPI*1*SPECS 62 CLA CZ) 63 RPT 5**1 64 STR IA 65 *** SET TERMINATION V AND I 66 ITRAT MOV CZ)*DATA+2 67 MOV C Z > * DATA+1 134 68 MOV C *400000000401 * 3 * DATA 69 CLA C '633614557371 '3 2*PI/1000 70 TRL $ FMLY*1 * DATA-1 FREQ 71 TRL $FMLY*1*CAP 72 STR DATA+3 73 74 LDX 6*1 3 RADIATING ELEMENTS 75 LOOP TRL STMLP*1 * DATA-1 TRANS THRU ELEMENT 76 CLA 0*1 i 3> * H to 77 STR to 78 CLA 1*1 79 STR IA-1*2 80 STR (Q) 81 CLA 0* 1 82 CLA (1X2) 83 SHR 1 84 STR MDATA 85 SEN *+4** 86 TRL $VM*1*MDATA CALC MUTUAL COUPLING 87 TRL SCADD* 1*DATA 88 TRL SCSTR* 1 * DATA 89 CLA CIX2) 90 SUB C23 91 TRN ZIN I 135 92 TRL $TMN*1 * DATA TRANS THRU DELAY LINE 93 SBB (1X2)** 1 94 TRX LOOP*1 95 *** CALCULATE INPUT IMPEDANCE* 96 *** COMPARE WITH SAVED VALUE* AND ITERATE 97 *** IF DIFFERENT IN HIGH-ORDER 19 BITS. 98 ZIN SEN UNLOOP**CSW3) 99 TRL SCLOD*1* DATA 100 TRL SCDVD*1 * DATA+2 101 LGM C-*777777400777'2 102 STR TEMP 103 CLA CQ) 104 LGM C-*777777400777*3 105 STR TEMP+1 106 SUB Z+1 * * 7 107 TRZ *+4 108 MOV TEMP*Z 109 MOV TEMP+1*Z+1 110 TRU ITRAT 111 CLA TEMP 112 SUB Z**7L 113 TRZ *+2 114 TRU *-6 115 *** SUM MAGNITUDES OF ELEMENT CURRENTS 116 *** FOR PATTERN NORMALIZATION 117 UNLOOP CLA FREQ IfB TRL SFMLY*1#C '655165200376'3 2*PI/29.97925 119 TRL SFMLY* 1# MDATA+ 5 120 STR KD 121 TRL $CLOD>1» IA+4 122 TRL SCADD*1>IA 123 TRL SCSTR*1*TEMPI 124 TRL SCLOD*1 * IA+4 125 TRL SCSUB*1 *IA 126 TRL SCSTR,1,TEMP2 127 TRL SCLOD*1#IA 128 TRL SCMAG 129 STR SUM 130 TRL SCLODs1» IA+2 131 TRL SCMAG 132 TRL $FADD.»1»SUM 133 STR SUM 134 TRL SCLODs1• IA+4 135 TRL SCMAG 136 TRL SFADD*1•SUM 137 STR SUM 138 *** GET ANGLE, CALC POINT ON POWER PATTERN 139 SCAN CLA XP 140 TRL SFADD#1#C'400000000401'3 141 TRL SFMLY#1#C*622077326401*3 PI/2 142 TRL $C0S 143 TRL SFMLY#1#KD 144 TRL SSINCOS 145 MOV 146 ADB 147 TRL SCMLY# 1 #TEMP2 148 TRL SCSTR#1»TEMP 149 CLA COS 150 MOV 151 TRL $CMLY*\*TEMP1 152 TRL $CADD#1#TEMP 153 TRL SCADD*1#IA+2 154 TRL SCMAG 155 TRL SFDVD#1»SUM NORMALIZE 156 STR TEMP 157 TRL $FML Y#1# TEMP 158 SUB Cl] 159 MOV PCS# 160 *** READ FLOATING POINT NUMBER FROM TTY# 161 *** ELSE KEEP OLD NUMBER IF NO KYBD ENTRY 162 READF MOV CPCS)#PCS 163 MOV (Q)#AQ+1 138 164 STR AQ 165 CLA <1X1) 166 RPA STORE 167 RPA KEEP 168 TRL $READ*1*BUF 169 TRU SEXEC 170 CLA BUF 171 TRZ KEEP 172 TRL $CNVTF*1*BUF 173 TRL ERR 174 STORE STR ** 175 MOV AQ+1 *(Q) 176 CLA AQ 177 MOV PCS* CPC) 178 KEEP CLA ** 179 TRL SPRFL 180 TRU STORE+1 181 ERR WAT C *405277524040*3*1 182 SBB (PCS)**6 183 TRS 184 ********** 185 AQ BS 2 186 FREQ BS 1 187 DATA BS 4 139 188 TAB 0 189 BS 1 190 TAB '564000000407'*'740000000406* 191 MDATA BS 2 192 TAB 3 193 HLT IA 194 TAB '466000000407***500000000405' 195 IA BS 6 196 SPECS TAB ’500000000404'*'626416254400'*'760000000403' 197 CAP BS 1 198 Z BS 2 199 COS BS 1 200 SUM BS 1 201 KD BS 1 202 TEMP BS 2 203 TEMPI BS 2 204 TEMP2 BS 2 205 MS63 ALPHA 2 CCPF> = 206 MS61 ALPHA 2 XINC * 207 MS62 ALPHA 2 YINC * 208 BUF BS 5 209 LIST TAB 350*200 210 XINC BS 2 211 PCS BS 1 212 X BS 2 213 XP BS 2 214 USE UTIL,LIB 215 USEPLOT,LIB 216 USE CMPLX,LIB 217 USE MATH1,LIB 218 USEREAD,RAY 219 USE PL3,RAY 220 USE EXRET,LIB 221 USE TMN,RAY 222 USETMLP,RAY 223 USE VM,RAY 224 END * 141 *A ZTEMS *P 1 * THIS PROGRAM PLOTS THE INPUT IMPEDANCE OF A 2 * PARTICULAR THREE-ELEMENT* CAPACITANCE-TUNED 3 * TEM-LINE ANTENNA AT A FIXED FREQUENCY FOR VALUES 4 * OF TERMINATING CAPACITANCE FROM ZERO TO 32 PF 5 * IN STEPS OF 1 PF. 5 **************** if:********:*:************* ************ 7 *** DRAW SMITH CHART TO DESIRED SIZE 8 HLT SSTART 9 #START WAT C •1500120012 *3#1 10 WAT SIZE#6 11 TRL READF#1#RADIUS 12 CLA RADIUS 13 SUB Cl] 14 STR RADIUS 15 CLA C'400000000377*3 16 TRL SFADD#1#RADIUS 17 TRL SFMLY#1#C'620000000407'3 18 TRL $FLINT 19 STR XORG 20 CLA RADIUS 21 TRL $CHART#1#XORG 22 *** GET FREQUENCY AND PERFORM INITIALIZATIONS 142 23 WAT C •1500120012*3*1 24 WAT Q1 * 2 25 TRL READF*1*DATA-1 26 CLA DATA-1 27 STR MDATA+1 28 TRL $FMLY*1*C*655165200376*3 2+PI/29.97925 29 TRL $FDVD*1 *C'656050754400 * 3 .84 30 STR DATA+5 31 TRL STMLPI*1,SPECS 32 *** SET CAPACITANCE TO 0 AND LOOP 33 TIMES 33 MOV CZ)*CAP 34 MOV ARK*MARK 35 LDX 33*2 36 CLOOP CLA 37 RPT 5**1 38 STR IA 39 MOV C1X3)* CDISP) 40 *** SET TERMINATION V AND I 41 ITRAT MOV CZ)*DATA+2 42 MOV < Z)* DATA+1 43 MOV C*400000000401*3*DATA 44 CLA C *633614557371 * 3 2*PI/1000 45 TRL SFMLY*1 *DATA-1 FREQ 46 TRL $FMLY* 1 * CAP 47 STE DATA+3 48 ********** 49 LDX 6/1 3 RADIATING ELEMENTS 50 LOOP TRL STMLPz1/DATA-1 TRANS THRU ELEMENT 51 CLA 0/1 52 STR IA-2/2 53 CLA 1/1 54 STR IA-1/2 55 STR 56 CLA 0/1 57 CLA ( 1X2 ) 58 SHR 1 59 STR MDATA 60 SEN *+4//CSU4> SKIP MUTUAL COUPLING 61 TRL SVM/l/MDATA CALC MUTUAL COUPLING 62 TRL SCADD/1/DATA 63 TRL SCSTR/I/DATA 64 CLA (1X2) 65 SUB Z22 66 TRN ZIN 67 TRL STMN/l/DATA TRANS THRU DELAY LINE 68 SBB (1X2)//1 69 TRX LOOP/1 70 *** CALC POINT ON SMITH CHART/ 71 *** COMPARE WITH SAVED POINT* 72 *** ITERATE IF DIFFERENT 73 ZIN TRL SCLOD* 1*DATA 74 TRL SCDVD*1 * DATA+2 75 MOV 76 TRL SFADD*1 * C *620000000406 77 STR Z 78 TRL SFSUB*1 * C'620000000407 79 TRL SCDVD*1 *Z 80 TRL SFMLY*1 * C*620000000407 81 TRL SFMLY* 1* RADI US 82 TRL SFLINT 83 ADD XORG 84 STR XX 85 SUB x 86 TRZ *+2 87 MOV XX* X 88 CLA 89 TRL SFMLY*1 * C * 620000000407 90 TRL SFMLY*1*RADIUS 91 TRL $ FLINT 92 ADD XORG+l 93 STR XX 94 SUB X+l 145 95 TRZ *+3 96 MOV XX*X+1 97 TRU ITRAT 98 ********** 99 MARK TRL SMARK2*1*X PLOT POINT ON SMITH CHART 100 MOV ARK+1 * MARK 101 *** INCREMENT CAPACITANCE BY 1 PF AND LOOP UNLESS DONE 102 CLA CAP 103 TRL SFADD*1*C'400000000401'3 1.00 104 STR CAP 105 TRX CL00P*2 106 ********** 107 TRL SUP 108 SEN START+13**CSW4) NO NEW CHART 109 *** TAKE NEW PAGE AND BEGIN AGAIN 110 MOV CZ>*X+1 111 MOV C85Q 3*X 112 TRL SMOVE*1*X 113 MOV 114 TRL SORIG*1/X 115 TRU START 116 *** READ FLOATING POINT NUMBER FROM TTY* 117 *** ELSE KEEP OLD NUMBER IF NO KYBD ENTRY 118 READF MOV CPCS)*PCS 146 119 MOV 120 STR AQ 121 CLA <1X1 ) 122 RPA STORE 123 RPAKEEP 124 TRL SREAD*1*BUF 125 TRU SEXEC 126 CLA BUF 127 TRZ KEEP 128 TRL SCNVTF*1 *BUF 129 TRL ERR 130 STORE STR ** 131 MOV AQ+1* 132 CLA AQ 133 MOV PCS* CPC) 134 KEEP CLA ** 135 TRL SPRFL 136 TRU STORE+1 137 ERR WAT C '40 5277524040 * 3*1 138 SBB (PCS)* *6 139 TRS 140 •jc ifc jjg 141 PCS BS 1 142 AQ BS :2 147 143 BUF BS 5 144 01 ALPHA 2 FREQ(GHZ) = 145 SIZE ALPHA 6 CHART DIAMETER IN INCHES? 146 FREQ BS 1 147 DATA BS 4 148 TAB 0 149 BS 1 150 TAB *564000000407','740000000406' 151 MDATA BS 2 152 TAB 3 153 HLT IA 154 TAB '466000000407 ', '500000000405' 155 IA BS 6 156 SPECS TAB * 500000000404', '626416254400' •760000000403* 157 CAP BS 1 158 RADIUS BS 1 159 XX BS 1 160 X BS 2 161 X0R6 TAB , 500 162 Z BS 2 163 ARK TRL $MARK2,1 ,X 164 TRL SMARK,1/X 165 USE PLOT,LIB 166 USE UTIL,LIB 748 167 USE CMPLX* LIB 168 USEEXRET*LIB 169 USE CHART* RAY 170 USE READ*RAY 171 USE TMN*RAY 172 USE TMLP*RAY 173 USE MARK* DEAN 174 USE VM*RAY 175 END * 149 *A TMLPS *P 1 * THIS SUBROUTINE TRANSFORMS COMPLEX V AND I THROUGH 2 * A RECTANGULAR TEM-LINE LOOP ELEMENT* INCLUDING THE 3 * RADIATION TERMS. ITS INITIAL CALLING SEQUENCE IS: A * TRL STMLPI*1*SPECS WHERE SPECS LIST IS: 5 * LENGTH OF LOOP ELEMENT CCM) 6 * DIAMETER OF WIRE CONDUCTOR CCM) 7 * HEIGHT OF LOOP ELEMENT CCM) 8 *' ITS REGULAR CALLING SEQUENCE IS: 9 * TRL STMLP*1 * DATA WHERE DATA LIST IS: 10 * FREQUENCY CGHZ) 11 * RECV3 12 * IMCV3 13 * RECI] 1A * IMCID 15 #TMLPI MOV CPCS)*PCS 16 STR AQ 17 C L A %0*1 18 TRN ERR 19 STR LL 20 CLA 1*1 21 TRN ERR 22 STR DIAM 150 23 CLA 2* 1 24 TRN ERR 25 STR LV 26 ADD C23 27 TRL SFDVD*1*DIAM 28 TRL SLN 29 TRL SFMLY*l/[*740000000406 * 3 30 STR ZL 31 TRL $FMLY*1*C'400000000404*3 32 STR RB 33 CLA LL 34 TRL SFMLY*1*LV 35 STR AREA 36 CLA AQ 37 MOV PCS*(PC) 38 ERR WAT C *1500120012'3*1 39 WAT NEGVAL*7 40 WAT C * 1500120012'3*1 41 TRU ERR-2 42 NEGVAL ALPHA 7 NEGATIVE LOOP DIMENSION FOUND. 43 DIAM BS 1 44 HLT IA 45 #TMLP MOV (PCS)*PCS 46 MOV 47 MOV C Q} * AQ+1 48 STR AQ 49 CLA 1*1 RECV] 50 STR LIST 51 CLA 2* 1 IMCV] 52 STR LIST+1 53 CLA 3* 1 REC I] 54 STR LIST+2 55 CLA 4* 1 IMCI] 56 STR LIST+3 57 CLA 0*1 58 STR FREQ 59 TRL SFMLY*1*C•655165200376’] 2*PI/C 60 STR LIST+5 IMCGAMMA] 61 MOV CZ)*LIST+4 RECGAMMA] 62 MOV ZL*LIST+6 63 MOV LV*LIST+7 64 TRL STMN*1 *LIST 65 TRL BEND 66 CLA LL 67 SUB CU 68 STR LIST+7 69 TRL STMN*1*LIST 70 MOV HST+2* IA 71 MOV LIST+3/IA+1 72 TRL STMN/ 1/LIST 73 TRL BEND 74 MOV LV/LIST+7 75 TRL STMN/1/LIST 76 LOD IX//<1X1> 77 CLA LIST 78 STR 1/ 1 79 CLA LIST+1 80 STR 2/1 81 CLA LIST+2 82 STR 3/1 83 CLA LIST+3 84 STR 4/ 1 85 LOD TMLP-1//CIX1) 86 CLA AQ 87 MOV AQ+1/ 88 MOV PCS/ 89 BEND MOV CPCS)/PCS+1 90 CLA LIST 91 TRL SFDVD/1/RB 92 TRL SFADD/1/LIST+2 93 STR LIST+2 94 CLA LIST+1 95 TRL SFDVD,1,RB 96 TRL SFADD, LLIST+3 97 MOV PCS+1,CPC> 98 IA BS 2 99 PCS BS 2 100 IX BS 1 101 AQ BS 2 102 AREA BS 1 103 FREQ BS 1 104 LIST BS 8 105 ZL BS 1 106 LL BS 1 107 LV BS 1 108 RB BS 1 109 USE UTIL,LIB 110 USE MATH1,LIB 111 USE TMN,RAY 112 END * 154 *A TMNS *P 1 * THIS SUBROUTINE TRANSFORMS COMPLEX V AND I THROUGH 2 * A LENGTH OF TRANSMISSION LINE HAVING ZNOT = REAL 3 * CHARACTERISTIC IMPEDANCE* GAMMA = COMPLEX PROPAGATION 4 * CONSTANT. ITS CALLING SEQUENCE IS: 5 * TRL STMN*1*V WHERE V LIST IS: 6 * RECV3 7 * IMCV3 8 * REC I 3 9 * IMCI3 10 * RECGAMMA3 (PER CM) 11 * IMC GAMMA3 (PER CM) 12 * ZNOT 13 * LENGTH (CM) 14 #TMN MOV (PCS)*PCS 15 MOV (1X2)*1X2 16 MOV (1X3)*1X3 17 MOV (Q)*Q 18 STR (A) 19 LDX 4*2 20 MOV (1X1)*(1X2) 21 GET CLA 3*2 22 TRZ *+4 155 23 SUB C 13 24 TRP *+2 25 ADD C23 26 STR LI ST# 3 27 TRX GET# 2# -1 28 CLA 10#2 L 29 STR LIST 30 CLA 8#2 BETA 31 TRL SFMLY#1#LI ST 32 STR 33 CLA 7#2 ALPHA 34 TRL SFMLY#1#LI ST 35 TRL SC EXP 36 STR CEXPL 37 MOV CQ>#CEXPL+1 38 CLA 9# 2 ZNOT 39 STR LIST 40 TRL SFMLY#1#LIST+4 41 STR 42 STR F2+1 43 CLA LIST 44 TRL SFMLY#1#LIST+3 45 STR F2 46 TRL SCADD#1#LIST+1 47 TRL SCMLY,1*CEXPL 48 STR F1 49 MOV (Q)/Fl + 1 50 CLS F2+1 51 STR (Q) 52 CLS F2 53 TRL SCADDj IjLIST+1 54 TRL SCDVDj 1* CEXPL 55 TRL $CADD« 1 * FI 56 STR 3*2 VCOUT) 57 CLA CQ) 58 STR 4* 2 59 CLA LIST+2 60 TRL SFDVD* 1*LIST 61 STR CQ) 62 STR F2+1 63 CLA LIST+1 64 TRL SFDVD*i*l i s t 65 STR F2 66 TRL SCADD*l*LIST+3 67 TRL SCMLY*1*CEXPL 68 STR FI 69 MOV 70 CLS F2+1 71 STR CQ > 72 CLS F2 73 TRL SCADD*l*LIST+3 74 TRL SCDVD*1*CEXPL 75 TRL $CADD*1*F1 76 STR 5* 2 ICOU' 77 CLA (0) 78 STR 6#2 79 MOV 1X3*C1X3) 80 MOV 1X2* (1X2) 81 MOV Q*CQ) 82 CLA A 83 MOV PCS*(PC) 84 PCS BS 1 85 A BS 1 86 0 BS I 87 1X2 BS 1 88 1X3 BS 1 89 LIST BS 5 90 CEXPL BS 2 91 FI BS 2 92 F2 BS 2 93 USE MATH1*LIB 94 USE CMPLX*LIB 95 USE UTIL*LIB 96 END 158 *A VMS *P 1 * THIS SUBROUTINE RETURNS THE TEM-LINE MUTUAL VOLTAGE 2 * AS A COMPLEX NUMBER IN CA) AND CQ). ITS CALLING SEQUENCE • 3 * IS m 4 * TRL $VM#1#MDATA WHERE MDATA LIST IS* 5 * ELEMENT NUMBER 6 * FREQUENCY (GHZ) 7 * MAX ELEMENT NUMBER 8 * IA ADDRESS 9 * LOOP AREA CSQ CM) 10 * LOOP SPACING CCM# CENTER-TO-CENTER) 11 * THE IA LIST IS THE SET OF COMPLEX ELEMENT CURRENTS* 12 * RECIACD3 13 * IMCIACD3 14 * * 15 * * 16 * RECIAC N)3 17 * IMCIACN)] 18 #VM MOV CPCS)#PCS 19 MOV C1X3)#1X2+1 20 MOV C1X2)#1X2 21 MOV C1X1)# C1X2) 22 CLA 3#2 159 23 SUB C13 24 RPA II 25 SUB Cl] 26 RPA 11+2 27 MOV 28 MOV C Z > *VMC+1 29 LOD 2*2* (1X3) 30 LOOP CLA CIX3) 31 SUB 0* 2 32 TRZ END 33 CAM (A) 34 TRL SINTFL 35 STR TEMP 36 CLA 5* 2 37 TRL SFMLY*1*TEMP 38 STR DENOM 39 CLA 1*2 40 TRL SFMLY*1*C'655165200375'3 PI/C 41 STR TEMP+1 42 — ADD C 1 3 43 TRL SFMLY*1*DENOM 44 CLS CA) 45 ADB CZ) 46 TRL SCEXP 160 47 TRL SCSTR,1, CEXP 48 CLA TEMP+1 49 TRL SFMLY,1,TEMP+1 50 TRL SFMLY,1,TEMP+1 51 TRL SFMLY,1,C-’740000000411'D -480 52 STR TEMP 53 CLA 4,2 54 STR TEMP+1 55 TRL SFMLY,1,TEMP+1 56 TRL SFMLY,1,TEMP 57 TRL SFDVD,1,DENOM 58 ADB CZ) 59 TRL SCMLY,1,CEXP 60 TRL SCSTR,1,TEMP 61 CLA C1X3 ) 62 SHL 1 63 STR CIX1) 64 11 CLA **,1 65 STR 66 CLA **,1 67 TRL SCMLY, 1, TEMP 68 TRL SCADD, 1,VMC 69 TRL SCSTR,1,VMC 70 END TRX LOOP,2 71 TRL SCLOD,1*VMC 72 MOV 1X2*C1X2) 73 MOV 1X2+1,(IX3> 74 MOV PCS* 75 PCS BS 1 76 1X2 BS 2 77 VMC BS 2 78 TEMP BS 2 79 DENOM1 BS 1 80 CEXP BS 2 81 USE UTIL,LIB 82 USE CMPLX,LIB 83 END REFERENCES 1. Interim Engineering Report, 28 February 1965, Report 1566-17, ElectroScience Laboratory, Department of Electrical Engineering, Ohio State University (AD 461 378). 2. Final Engineering Report, 20 December 1965, Report 1566-24, ElectroScience Laboratory, Department of Electrical Engineering, Ohio State University (AD 476 943). 3. Copeland, J.R. "A Surface-Mounted Slotted TEM-Line Antenna," Presented at the 15th Annual Symposium on USAF Antenna Research and Development, Monti c e llo ,Illin o is , October 1965. 4. Copeland, J.R ., "The Slotted TEM-Line Antenna," IEEE Transactions on Antennas and Propagation, Vol. AP-16, No. 2, March 1968, pp. 260-262. 5. Stratton, J.A ., Electromagnetic Theory, McGraw H ill Book Co., New York, 1941, pp. 193-4. 6. Jeans, Mathematical Theory of Electricity and Magnetism, 5th Edition, Chapter V I I I , Cambridge University Press. 7. Mason and Weaver, The Electromagnetic Field, pp. 109ff, University of Chicago Press, 1929. 8. Terman, F.E ., Radio Engineer's Handbook. McGraw H ill Book Co., Inc. New York, 1943, pp. 774-6. 9. Kraus, J .D ., Antennas, McGraw H ill Book Co., In c ., New York, 1950, Chapter 13. 162 163 10. Booker, H.6., "Slot Aerials and Their Relation to Complementary Wire Aerials," IIEE (London) Vol. 93, Part III-A , No. 4, 1946. 11. Radio Research Lab S taff, "Very High Frequency Techniques," McGraw H ill Book Co., In c., New York, 1947, Chapter 7, by Dome and Lazarus. 12. Jasik(ed), Antenna Engineering Handbook, McGraw H ill Book Co., Inc., New York 1961, Chapter 8 by Judd Blass. 13. Stratton, J.A., op. c it., Section 9.24, pp. 560-563. 14. Watson, W.H., "The Physical Principles of Waveguide Transmission and Antenna Systems," Oxford University Press, London, 1947. 15. Jasik,(ed), op. c i t . , Chapter 9 by M.J. Ehrlich. 16. Walter, C.H., Traveling Wave Antennas, McGraw H ill Book Co., In c ., New York, 1965, Chapter 5. 17. Uda, S. and Mushiake, Y., Yagi-Uda Antenna, Maruzen Co., Ltd., Tokyo, 1954 (in English). 18. Jasik (ed), o p .c it., Chapter 16, by F.J. Zucker. 19. Franklin, Charles Samuel, and Green,Ernest, of London,England, U.S. Patent 1 957 949 issued May 8, 1934, Assigned to Radio Cor poration of America. Prior British Specification Nos. 242324, 281762 and 285106. 20. Br’uckmann, Hellmut, Antennen Ihre Theorie und Technik, Verlag von S. Hirzel in Leipzig, 1939, p. Ill (in German). 21. Cohn, Stanley I . , Hoehn, Alfred J ., and Cohn, George I . , "Coaxial Omnidirectional Slot Antenna Arrays," Proceedings of the National Electronics Conference, Vol. 10, 1954. 22. Dallenbach, Walter, of Bern Germany, Bundesrepublik Deutschland Patent No. 902510 issued December 10, 1953. 23. Final Technical Report, "Investigation of a Low Profile Flush Mounted Antenna, 14 March 1967, Report 2147-4, ElectroScience Laboratory, Department of Electrical Engineering, Ohio State University, (AD 810 255). 24. Brillouin, L.N.,Wave Propagation in Periodic Structures, McGraw H ill Book Co., In c ., New York 1946. 25. Walter, C.H., op. cit.,Chapter 8. 26. Kraus, J.D., op. c i t . , Chapter 5. 27. Terman, F.E., op. c it ., p. 179. 28. Ib id ., p. 174. 29. Kraus, J.D ., op. cit.,Chapter 8. 30. Ross, Gerald, F ., "A Time Domain Determination of the Driving Point Characteristics of Certain Bent Wire Radiating Elements," presented at the 1967 IEEE G-AP Symposium, Ann Arbor, Michigan, October 1967. 31. Ross, G., Bates, R.H.T., Susman, L ., e t .a l., "Transient Behavior of Radiating Elements, " U.S.A.F. RADC-TR-66-790, AF30(602)-4050, February 1967. 32. Rudduck, R.C., "Application of Wedge Diffraction to Antenna Theory," Report NASA CR 372 prepared under Contract NsG-448, For sale by the Clearinghouse for Scientific and Technical Information, Springfield, Virginia, 22151. 165 33. Kilcoyne, T.E ., "The Slotte.d TEM-Line Antenna - A Low P rofile, Beam Scanning, Traveling Wave Antenna," 3 April 1967, Report 2341-1, ElectroScience Laboratory, Department of Electrical En gineering, Ohio State University (AD 811 090). 34. "Informer Assembler Manual," Internal publication of the Ohio State University, ElectroScience Laboratory, October 1968.