Rake Reception for Uwb Communication Systems with Intersymbol Interference

RAKE RECEPTION FOR UWB COMMUNICATION SYSTEMS WITH INTERSYMBOL INTERFERENCE A.G. Klein†, D.R. Brown III‡, D.L. Goeckel§, and C.R. Johnson, Jr.† † Cornell University, School of Electrical and Computer Engineering, Ithaca, NY 14853 ‡ Worcester Polytechnic Inst., Dept. of Electrical and Computer Engineering, Worcester, MA 01609 § University of Mass., Dept. of Electrical and Computer Engineering, Amherst, MA 01003 ABSTRACT have demonstrated delay spreads far beyond 30 ns [2][3], which indicate that significant intersymbol interference (ISI) Recently, ultra wideband (UWB) technology has been pro- is unavoidable. A RAKE receiver, the preferred structure posed for use in wireless personal area networks (WPANs). for collecting multipath energy in UWB systems, does not Under the conditions where such transceivers are expected combat ISI. The vast majority of published results on UWB to operate, intersymbol interference (ISI) will become a sig- have overlooked this fact as most performance analyses em- nificant performance limitation, and improvements to con- ploy a RAKE receiver under the assumption that channel ventional RAKE reception will be necessary. We propose a delay spreads are much less than the pulse repetition period modified RAKE receiver that finds an optimal balance be- [4]. In this paper, the RAKE structure is modified to one tween the goal of gathering multipath signal energy, avoid- that considers ISI and narrowband interference. By mini- ing ISI, and suppressing narrowband interference. For fixed mizing the mean squared error (MSE), we arrive at a choice RAKE finger delays, we develop a closed-form expression of RAKE delays and combining weights that yield supe- for the minimum mean squared error (MMSE) combining rior performance when compared to maximal ratio combin- weights that account for ISI. We then examine the optimal ing (MRC). A generalized RAKE for combatting interfer- choice of RAKE finger delays, and show that significant ence was previously proposed in the context of DS-CDMA performance gains can be achieved, particularly in an un- [5], where the authors require the noise to be white Gaus- dermodeled situation when there are more channel paths sian, and they employ a different optimization criterion (i.e. than RAKE fingers. Several numerical examples are pre- maximum-likelihood) but arrive at a similar result. A mod- sented which compare our proposed scheme to a conven- ified RAKE for suppressing narrowband interference was tional RAKE with maximal ratio combining (MRC). proposed in [6], but no consideration is given to ISI. 1. INTRODUCTION Ultra wideband (UWB) communication has attracted recent 2. SYSTEM MODEL interest in the research and standardization communities, due to its promising ability to provide high data rate at low Two popular modulation types being considered for UWB cost and power consumption. Another frequently cited ben- are pulse-position modulation (PPM) and pulse amplitude efit of UWB transmission is its ability to resolve individual modulation (PAM). Here, we consider systems of the PAM multipath components (see, for instance, [1]). This feature type, and a model for single-user M-PAM UWB transmis- motivates the use of RAKE multipath combining techniques sion over channels with ISI is shown in Figure 1. We have to provide diversity and capture as much energy as possible assumed a baseband transmission system, and hence all sig- at the UWB receiver. nals are assumed to be real-valued. The zero-mean i.i.d. UWB is being considered for use in wireless personal data symbols {sn} are passed through a unit energy pulse- area networks (WPANs), where required data rates are in shaping filter p(t) which includes the effects of the trans- excess of 110 Mbps per user. To obtain such rates with bi- mit antenna. Note that we require the pulse shape to be nary signaling, the required symbol period will need to be unchanging from symbol period to symbol period, but our less than 10 ns even without coding or spreading. Mean- model is general enough to include either time hopping (TH) while, several UWB indoor channel measurement campaigns or direct sequence (DS) block spreading if, for example, † p t Supported in part by Texas Instruments and a Lockheed Martin Fellowship. ( ) is the sum of several delayed Gaussian monocycles. § Supported in part by a MURI Project under Contract DAAD10-01-1-0477. After pulse shaping, the signal undergoes the effects of a w t ( ) finger are orthogonal (as is the case when there is no ISI), MRC attains the matched filter bound [7]. However, when sn pulse shapechannel r(t) RAKE yn p(t) h(t) f(t) nT the ISI becomes significant the orthogonality of the paths is violated unless care is taken in the design of the pulse shape (as in long-coded DS-CDMA systems where succes- Fig. 1. System model. sive symbols are nearly orthogonal). For UWB-based high rate WPANs it is anticipated that spreading codes will be channel with L paths whose response given by: short, and therefore MRC is suboptimal even when the noise is AWGN. This motivates a smarter choice of combining L−1 weights βm and finger delays θm to combine the signal en- h(t)= αδ(t − τ) ergy while compensating for the effects of ISI and narrow- =0 band interference. where α and τ are the gain and delay introduced by the th path of the channel. The received signal can then be 3. ISI-AWARE RAKE expressed as: In the presence of ISI, the optimal receiver is the maximum- ∞ L−1 likelihood sequence estimator (MLSE). Since the computa- r(t)= si αp(t − τ − iT )+w(t) tional complexity grows exponentially with channel length, i=−∞ =0 most channels of practical interest require too much com- putation for MLSE to be feasible. Consequently, system where T is the symbol rate and w(t) is additive noise. The designers typically resort to suboptimal schemes like lin- noise is assumed to be a zero-mean wide sense stationary ear equalization to compensate for ISI. A linear equalizer process that is uncorrelated with the data, and it may be and a RAKE are structurally quite similar, as both are lin- colored due to narrowband interferers. ear combiners. However, an equalizer operates on sam- In the case of no ISI and when the noise is AWGN, the pled data, whereas a RAKE does not need to be confined optimal receiver is a filter matched to the received wave- to a “grid” of delays imposed by sampling. Furthermore, form (i.e. the combined response of the channel and trans- an equalizer attempts to suppress ISI by minimizing some mit pulse shapes). Typically, this is implemented in a RAKE metric like MSE, while a MRC-RAKE ignores ISI and at- receiver structure with M fingers, which can be represented tempts to gather all the signal energy to maximize the SNR. as a filter with response: In this section, we propose a RAKE where the finger delays M−1 and combining weights are chosen to minimize the MSE, f(t)= βmp(−t − θm) thereby arriving at a RAKE that finds an optimal balance be- m=0 tween the goal of gathering multipath signal energy, avoid- ing ISI, and combatting narrowband interference. In the where our model places no restrictions on the spacing of analysis that follows, we assume complete knowledge of the the RAKE delays θm. The sampled output of the RAKE channel path delays and gains, the transmit pulse shape, and receiver is then: the autocorrelation of the additive noise. yn =[r(t) f(t)] t=nT 3.1. Choosing the Combining Weights ∞ L−1 M−1 = si αβmRp (nT − iT + θm − τ) Define b =[β0 β1 ... βM−1] to be the vector of RAKE i=−∞ =0 m=0 combining weights, and define Φ[n] ∈ RM as: +˜w(nT ) (1) R nT − τ θ p( + 0) ∞ L−1 R t p τ p τ t dτ Rp nT − τ θ1 where p( ) −∞ ( ) ( + ) is the time-autocorrelation ( + ) Φ[n] α of the pulse shape and . =0 . R nT − τ θ M−1 ∞ p( + M−1) w˜(t)= βm w(τ)p(−t + τ − θm)dτ b n m=0 −∞ so that Φ[ ] represents the combined response of the pulse shaping, channel, and RAKE, and (1) can be repre- is filtered noise. sented as: It is well known that the optimal combiner for the AWGN ∞ M L β multipath channel is MRC, where = fingers, m = yn = sib Φ[n − i]+w ˜(nT ) αm, and θm = τm. When the received signals on each i=−∞ R ∈ RM×M R ∞ R τ R τ Define the matrix as [ ]i,j −∞ w( ) p( + finger delays for the generalized RAKE receiver. Specif- β θi − θj)dτ, where Rw(τ) E[w(t)w(t + τ)] is the statis- ically, in the case when m is selected optimally (3), the tical autocorrelation of the noise. Due to the assumption of MSE cost can be expressed strictly as a function of the fin- a time-invariant channel, the MSE is given by: ger delays: 2 ∞ −1 JΘ,b = E |sn − yn| J σ2 − 1 R i i Θ = s 1 Φ[0] σ2 + Φ[ ]Φ [ ] Φ[0] 2 s i=−∞ = σs 1 − 2b Φ[0] and the optimal RAKE finger delays can be found via: ∞ b 1 R i i b + 2 + Φ[ ]Φ [ ] (2) σs i=−∞ 1 Θopt = arg max Φ[0] R Θ σ2 where we have used the fact that the data symbols and sam- s σ2 E s2 pled noise process are uncorrelated, and s = [ n] is the ∞ −1 power of the source symbols. + Φ[i]Φ[i] Φ[0] (4) b We see from (2) that the MSE is quadratic in , and i=−∞ hence the globally optimal solution to minimize the MSE is given by: where Θ=[θ0 θ1 ..

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