A Coherent Carrier Technique for Self-Timed

Home , Carrier system

A COHERENT CARRIER TECHNIQUE

FOR

SELF-TIMED DIGITAL REGENERATIVE REPEATERS

a thesis

submitted for the degree

Doctor of Philosophy

The University of New South Wales

William Carroll, B.E., M.I.R.E.E.

October 1970 UNIVERSITY OF N.S.W.

02056 29.JUN.71 LIBRARY PREFACE

This thesis is primarily an experimental investigation concerned

with the fundamental problem in long distance digital communication systems

of maintaining synchronisation of the digital data after its passage through

many hundreds of regenerative repeaters and dispersive transmission lines.

Existing self-timed regenerative repeaters derive the timing wave from the

base band digital data and are subject to systematic timing jitter ac

cumulation because of their data pattern dependency; hence communication

systems employing these techniques are limited to contain a maximum of

100-300 repeaters.

By utilizing a simple technique of phase-locking the data pulse

repetition frequency to the carrier frequency a new type of self-timed digital

regenerative repeater has been designed, implemented and tested which derives

a pattern insensitive timing wave from the carrier frequency thus preventing

the occurrence of the systematic timing jitter accumulation. Also the spectral

purity of the timing wave frequency impulse function can be controlled by

appropriate filtering in each regenerator. This reduces the noise power in

the vicinity of the carrier frequency by an amount which produces an overall

convergent signal-to-noise ratio for the system of such a value that the

average number of zero crossings of the carrier frequency remains constant.

This allows synchronisation to be maintained in extremely long systems con

taining large numbers of repeaters..

Wherever possible suitable analytic material is presented so that the experimental results may be appropriately evaluated, with emphasis being placed upon prediction and performance analysis from measurements rather than from pure mathematical models. To serve this purpose a simulation network was constructed and the manner in which its performance differs from a real system is detailed. Envelope Bi-stable

Detector

Filter - Self - Adapt Limiter SampleTime Network

Modulator

Regenerated AM Output

FIGURE 1. Block diagram of the Coherent Carrier Self-timed Regenerative Repeater Front View, all test points shown connected.

Rear View, cover panels removed.

FIGURE 2. Front and Rear Views of the racks containing the Test/Simulator Control Equipment and the two Regenerators with the Delay Line Network ACKNOWLEDGEMENTS

I am indebted to Professor A.E. Karbowiak for his introduction to the problem area and criticisms during the investigation. Discussion with other members of staff, particularly Mr. G.T. Poulton, and fellow research students is also recognised as being essential to the progress and success of the project.

Professors A.E. Karbowiak and M.W. Allen, Associate Professor

R.M. Huey together with Dr. E.H. Fooks are thanked for their recommendations which enabled me, through my employer, the Postmaster-General's Department, to obtain sufficient study leave from the Commonwealth Public Service Board to conduct the investigation. LIST OF PRINCIPAL SYMBOLS

a = Any odd number greater than 2.

A = Arbitrary constant.

A = Amplifier voltage gain.

A.G.C. = Automatic Gain Control.

A.P.C. = Automatic Phase Control.

B = Communication channel half bandwidth. c = Timing filter half bandwidth.

= Effective bandwidth of cascaded filter networks.

= Jitter cumulation factor.

= Synchronisation code group.

= Carrier frequency.

= Digital data transmission speed.

= Combined noise factor of limiter.

= Noise factor of first limiter stage.

= Noise factor of second limiter stage.

= Power gain of first limiter stage.

G = Filter conductance.

H(s) = Timing filter transfer function.

K = Reciprocal of filter inductance (I/L). m = Modified noise reduction ratio.

M = Positive integer equal to number of carrier cycles per data pulse.

N r = Number of repeaters.

= r.m.s. random jitter accumulation rate.

= r.m.s. systematic jitter accumulation rate. r

N o = Average number of zero crossings of sine wave plus noise per second.

NRZ = Non-return to zero digital data format. ON/OFF - Signal peak-to-carrier leak ratio.

P = Average number of time slots containing pulses

P = Positive integer.

PCM = Pulse Code Modulation.

PST = Paired Selected Ternary code. q = Effective noise increase due to mistuning.

Q = Timing filter Q-factor.

r = Noise reduction ratio.

= Maximum number of early/late section data bits SM

SASTN = Self Adaptive Sample Time Network.

T = Total loop delay time (measured). D T Transducer Gain. g = v . = r.m.s. noise input voltage. ni

V r.m.s. noise output voltage. no =

Vi(s) = Regenerator input signal voltage.

V (s) Regenerator input noise voltage. n = V (s) Regenerator output signal plus noise voltage. o =

= Input noise spectral power density.

a = Standard deviation for noise spectrum.

Factor by which r.m.s. timing deviation at nth repeater Yn = than that at first repeater.

T = Total loop delay time (theoretical).

03 = Filter center frequency. O

0) = Carrier frequency. c l = Signal loss due to mistuning. CONTENTS

Chapter Page

1 INTRODUCTION

1.1 Area of Research...... 1

1.2 Arrangement of Thesis...... 2

1.3 Originality and Significance of Thesis ...... 3

2 LITERATURE SURVEY

2.1 Introduction ...... 6

2.2 Self-Timed Digital Regeneration ...... 6

2.3 Simulation Networks ...... 22

2.4 Conclusions ...... 39

3 SYSTEM DESCRIPTION

3.1 Introduction...... 40

3.2 The Coherent Carrier Self-Timed Digital Regenerative Repeater 40

3.2.1 Design Criterion ...... 42

3.2.2 Functional Description ...... 43

3.2.3 System Synchronisation ...... 56

3.3 The Simulation Network ...... 60

3.3.1 Design Criterion ...... 60

3.3.2 Control and Test Functions...... 68

3.4 Conclusions ......

4 THEORETICAL CONSIDERATIONS

4.1 Introduction...... 72

4.2 Timing Filter Characteristics ...... 73

4.2.1 Filter Transfer Function ...... 74

4.2.2 Noise Reduction Ratio r...... 76 Chapter Page

4.3 Signal-to-Noise Analysis ...... 79

4.3.1 Cascaded Repeater System ...... 79

4.3.2 Noise Convergency Criterion ...... 80

4.4 Number of Zero Crossings of a Sine Wave plus Narrow Band

Gaussian Noise ...... 82

4.5 Signal-to-Noise Ratios in Band Pass Limiters...... 86

4.6 Mistuning ...... 87

4.6.1 Modified Noise Reduction Ratio m ...... 87

4.6.2 Frequency Spectra ...... 90

4.7 Simulation Loop Transfer Function ...... 92

4.8 Conclusions ...... 99

5 EXPERIMENTAL RESULTS

5.1 Introduction...... 102

5.2 Simulation Network Characteristics ...... 102

5.2.1 Natural Response ...... 102

5.2.2 Forced Response ...... 108

(a) Alignment 108

(b) Synchronisation ...... 112

5.2.3 Loop-Induced Spectral Characteristics ...... 118

5.3 Timing Filter Characteristics ...... 121

5.3.1 Noise Reduction Ratio r ...... 121

5.3.2 Synchronisation Level...... 123

5.3.3 Limiter Characteristics ...... 124

5.3.4 Noise Convergency ...... 129

5.4 Timing Jitter Characteristics ...... 132

5.4.1 Pulse Pattern Dependency ...... 134 Chapter Page

5.4.2 Number of Regenerations...... 137

5.4.3 Signal-to-Noise Ratio ...... 138

5.5 Error Rate...... 139

5.5.1 Pulse Pattern Dependency ...... 142

5.5.2 Number of Regenerations...... 145

5.5.3 Signal-to-Noise Ratio ...... 146

5.6 Mistuning ...... 150

5.6.1 Timing Jitter ...... 150

5.6.2 Error Rate ...... 151

5.7 Phase Discontinuities ...... 154

5.8 Conclusions...... 156

6 CONCLUSIONS AND RECOMMENDATIONS ...... 158

BIBLIOGRAPHY ...... 165

APPENDIX 1 170

2 172

3 175

PUBLICATIONS 181 1

CHAPTER 1

INTRODUCTION

1.1 Area of Research

The most important advantage of transmitting information in digital-

pulse form results from the fact that these pulses can be regenerated each

time the information passes through a repeater. The function of each re

generative repeater is to remove undesired amplitude effects by reshaping

the pulses and to restore the pulses to their assigned positions in equally

spaced time slots. Any random or systematic departure of the pulses about

their equally spaced time slots is described as TIMING JITTER. If the mag

nitude of the jitter becomes too great, because of noise or system imperfections,

the repeaters will lose synchronisation and hence cannot transmit the correct

information. Synchronisation is thus a basic problem in digital data trans mission systems.

Many types of synchronisation techniques exist, divisible into two main groups. One group uses a .separate timing signal and suffers in varying degrees the drawback that extra power, time and bandwidth has to be expended to achieve the degree of synchronisation required. Also the communication channel may affect differently the information signal and the timing signal owing to frequency and time selective fading or because of the frequency-phase dispersion characteristics of the medium.

The other group attempts to synchronise the repeaters by extracting the timing signal from the information signal itself. It is these self-timed synchronisation techniques that define the main problem area for this thesis.

The deficiencies of existing systems will be discussed and proposals, design, 2

implementation and testing of a simple but powerful technique of self-timing a digital regenerative repeater will be given.

A network simulation analysis applicable to the above class of re peater systems is also developed and presented as a supporting area of research.

1.2 Arrangement of Thesis

The thesis is primarily an experimental investigation into the design and analysis of the synchronisation of self-timed digital repeaters suitable for communication systems employing large numbers of repeaters. Because there exist two main areas of research the various chapters will be found to be split into two main parts, the first main part dealing specifically with those aspects relating directly to the digital repeaters whilst the second main part deals with the simulation considerations. Most chapters have their own intro duction and conclusion.

Chapter 1 serves as a general introduction by discussing the problem area and arrangement of thesis as well as pointing out the originality and significance of the thesis.

Chapter 2, the literature survey, is given in a form which allows an overview of the research conducted by previous workers to be developed. Using the comments and general conclusions made in the available literature a dis cussion of the limitations and deficiencies of existing systems will be presented.

In Chapter 3 action is taken to implement the conclusions and re commendations made in the previous chapter. A particular system description is developed and all the essential characteristics that will have to be in corporated into the design and testing discussed. 3

The purpose of Chapter 4 is to allow presentation of the theoretical considerations, and further results available in the literature but not given

in Chapter 2, that serve as the background against which the experimental re sults may be compared.

Chapter 5 details the results of the experiments from which the necessary conclusions are drawn to assess the success or failure of the re generation technique and method of simulation.

Chapter 6 concentrates specifically on the advancement in knowledge obtained and makes recommendations for future investigations and applications.

Because of the dynamic behaviour of the system a large number of photographs has been included to best illustrate the points under consideration.

In general any detailed circuit diagrams or analysis will be found in the appendix, the main body of the thesis contains only the relatively broad out line which is sufficient to provide continuity of the discussion.

1.3 Originality and Significance of Thesis.

The major portion of the investigations for this thesis is original work carried out by the author. Any assistance given by other persons has been acknowledged and those parts of the thesis which utilise the results of investigations conducted by other authors, particularly Chapter 2 and part of

Chapter 4, are marked by references to the Bibliography.

The significance of the thesis lies in the development of a new type of self-timing technique for digital regenerative repeaters which enable a much larger number of repeaters to be connected in cascade thereby increasing the previously attainable circuit length for a digital communication system. 4

The thesis thus contains the necessary information on the design, implementat ion and testing of a repeater employing this new technique of self-timing. It will be shown that this repeater does not produce any systematic timing jitter components in its output waveform because the timing wave, unlike existing self-timed digital regenerative repeaters, is not derived from the digital pulse pattern.

The technique developed possesses the major advantage that a spectral line impulse at the desired timing frequency is derivable from the input wave form in every regenerator in the system. The spectral purity of this frequency impulse is directly controllable by simple LCR networks which produce an over all convergent signal-to-noise ratio of such a value that the average number of zero crossings of the timing wave remains constant and keeps the whole digital communication system in synchronism, provided the signal-to-noise is not degraded below a threshold value.

The motivating reason for conducting this particular investigation arose as a consequence of a research project conducted mainly by this author into the feasibility of employing incoherent GaAs light emitting diodes as relatively high speed digital data transmitters for an optical communication 42 system. A major disadvantage of the optical communication system was that its single-hop length was range-limited to such an extent that regeneration of the digital data was required every half-a-mile. Thus any long range optical communication system employing these light emitting diodes would re quire a large number of repeaters. With the large distances between cities in

Australia the envisaged number of repeaters would readily exceed 500 to 1000.

This number of repeaters exceeds the normally recommended number of repeaters that can be connected in cascade if only simple self-timing techniques are to 5

be employed. Therefore the interest in being able to produce a self-timed digital regenerator which would maintain synchronism in a cascaded network of many hundreds of repeaters was established. 6

CHAPTER 2

LITERATURE SURVEY

2.1 Introduction

The first papers dealing with the problems associated with timing in

digital regenerative repeaters were published in 1956. However, it is not

the purpose of this chapter to present a historical survey of all the avail

able literature. What has been done is to select only those papers which

contain any fundamental analysis and/or detailed experimental results. By

using the results of the calculations, comments and conclusions referred to

in those selected papers present an overview of the problems associated with

self-timing techniques and make recommendations as to the nature of an im

proved digital regenerator (an extended list of references is available in

the Bibliography).

Wherever appropriate, critical comments have been made to emphasise

any deficiencies or commend any strong points contained in-the selected papers.

2.2 Self-timed Digital Regeneration

The November, 1958 issue of the Bell System Technical Journal con

tained 4 papers dealing with various aspects of self-timed digital regeneration.

In particular the companion papers presented in that issue by H.E. Rowe and

W.R. Bennett form the analytical background for the work that is presented by most other research workers who have examined this problem area and are thus discussed first.

In W.R. Bennett's paper the following specific problems were 7

/"\A AM-' OIL AJJDl O II I O I o O I I IOl o OlIIOIO O I II Ol o

timinq wave V

FIGURE 2.1 Functional block diagram of baseband timing wave recovery network

utilising the transitions in the binary pulse train. 8

considered:

1. Properties of a digital message pulse train as a, random noise

source

2. Derivation of the pulse repetition frequency from a pulse train

by shock excitation of a tuned circuit. Under this topic, the

following effects were studied:

(a) message statistics

(b) message pulse shape

(d) mistuning

(e) noise

3. Effect of time jitter in received pulse train on recovered analog

signal.

Bennett shows for a single repeater, using statistical Fourier transform techniques, that the frequency spectrum of the actual pulse en semble contains line spectral components at harmonics of the pulse rate and a continuous density function, both with intensity proportional to the square of the absolute value of the Fourier transform of the standard pulse at the frequency considered. The continuous component has many properties similar to thermal noise but differs in that it can exhibit regular spaced axis crossings (phase structure), can be exactly predicted over finite intervals and is capable of producing discrete components when non-linear operations are performed on it, even though no line spectral terms are originally present in the pulse spectrum at those frequencies. Figure 2.1 depicts the non linear operations required to produce the spectral line component at the de sired pulse rate when a non-return to zero (NRZ) data format is employed. The absence of this spectral line is shown in Figure 2.2 where both NRZ and Uni polar spectral curves are given together with the probability of occurrence FIGURE mean square voltage per unit bandwidth or discrete line

multiply multiply probability ordinate 2.2

Continuous on-off (from

ordinates ordinates scale

Bennett that signalling

and

any

for

4 by by

p.1511) line

pulse sin 4p 4p(l-p) p

with

2 = frequency

frequency spectral ttf

h for present

NRZ

\ discrete for 2

and

continuous density =

Unipolar p

terms -=

components

units spectrum units pulses.

for

independent

for both the continuous and discrete components as determined by Bennett.

Although Bennett provides the necessary information which is required to compute the probability of occurrence of a particular spectral line for the technique of shock-exciting a narrow band tuned filter, he does not draw attention to the fact that since the probability of occurrence for the discrete line component is always less than unity they cannot be single frequency im pulses. Impulses are only obtained if a continuous periodic time function 20-25 exists. Thus the discrete line components produced by the random patterns actually consist of a narrow band distribution of spectral energy instead of the indicated single frequency. Because of this continuous spectral energy distribution the output from a tuned filter will contain both amplitude and phase variations and thus the desired constant amplitude single frequency timing wave component is not obtained. The probability factor can be con trolled by the pulse data content and it is for this reason that coding re dundancy techniques can be employed to guarantee a minimum level of spectral energy at the pulse repetition frequency. The higher the number of trans itions in the pulse stream, for the NRZ and Unipolar data format, the greater the probability that a pulse exists and hence the larger the amount of

"continuous" spectral information at the timing frequency. This is an al ternate way of indicating that the timing information resides only in the leading and trailing edges of the pulse patterns.

Bennett's analysis of the effect of the Q indicates that a trade off is necessary and that an optimum Q should exist. He does not determine the optimum value but does show that the effect of noise on the recovered timing varies inversely with Q whereas the mistuning error becomes worse as

Q is increased. The final section of Bennett's paper deals with the effects of timing errors. The main issue he discusses is the seriousness of this effect when related to different types of messages. At one extreme the recovery 11

of printed text from a telegraph message is considered briefly whilst at the other extreme is situation where an analog signal is to be transmitted, such as a group of frequency-multiplexed voice channels transmitted by PCM. Timing errors in the latter example would lead to interchannel cross-talk and must be minimised. The other effect of timing error mentioned briefly by Bennett is the difficulty of reading messages correctly because, due to the finite rise and fall times of the bandwidth limited pulses, the decisions are made at a less favourable point on the pulse waveform thus giving rise to an increase in the error rate. This is called alignment error and is particularly important in a long chain of regenerative repeaters. This effect is analysed in the com panion paper given by H.E. Rowe.

Rowe studies some of the statistical properties of the random timing deviations, or position modulation of the signal pulses, in a long chain of regenerative binary repeaters. The random timing deviations of the output signal pulses result from input noise, tuning error, random timing deviations of the input signal pulses (introduced by preceding repeaters) and other sources at each repeater. He determines the power spectra and total power of the timing noise, spacing noise (random deviations in spacing of two consec utive pulses from an integral number of pulse periods) and alignment noise

(random deviations in alignment between an input signal pulse and its correspond ing timing pulse). The particular unipolar pulse patterns analysed by Rowe were:

(1) all pulses present

(2) every pulse present

(3) random pulse pattern.

Because of the approximations that had to be made to yield tractable mathematics the analysis for the long chain of repeaters turned out to be 12

W- Limiting power spectrum \ for large N (a) Timing noise

_ 1.265

(b) Spacing noise sM(f)

AJf)

FIPURE 2.3 Timing (a), spacing (b) and alignment (c) noise power spectra for a chain

of N repeaters with zero tuning error and complete retiming, with a white

timing noise of power No added at the input of each repeater; 0>>1, N>>1.

(Rowe^) 13

exactly that required if an idealised locked oscillator, synchronised by the signal pulses, replaced the tuned filter network. It was found that the low frequency timing noise does grow along the repeater chain whilst both the spacing and alignment noise approached limiting values rather quickly. Figure

2.3 depicts the results obtained by Rowe. The low frequency growth occurs because each repeater acts as a discrete low-pass filter to the timing de viations, transmitting the low frequencies with little loss whilst alternating the high frequencies. This total timing noise power is shown to grow as the square root of the number of repeaters, , whilst the power of the timing 2 wave is proportional to p , where p is the average number of time slots con taining pulses. Again it has been shown that the timing wave characteristics is controlled by the statistics of the pulse pattern distribution and hence any attempt to use the pulse pattern as the timing wave source must necessarily result in undesired variations in timing wave amplitude and phase which lead to the accumulation of timing jitter along a system. Rowe describes a few specific cases, his description follows below, and for each case he considers the power N of the equivalent timing noise at the input of a repeater and o the total output timing noise in a single repeater containing a tuned cir cuit timing filter of a particular Q.

"1. Baseband repeater, signal pulses drive tuned circuit directly. The real output noise power remains constant for different pulse patterns; since t proportional to the output noise to signal ratio, or — » and from (124), 1 P N cp - . o p 2. Baseband repeater, signal pulses passed through square-law or similar nonlinear device before driving tuned circuit. The real noise in the vacant time slots is now substantially suppressed; the real noise power at the output 1 might be expected to vary as p. Consequently oc — and is approximately independent of p. 3. Carrier frequency repeater, linear envelope detector. This case is somewhat different from case 1 above. During a time slot containing a pulse, only the in-phase component of the real input noise contributes to the random component of the envelope of the input wave. In the absence of a signal pulse, both the in-phase and quadrature components of the real input noise contribute equally to the output random phase modulation of the tuned circuit. Consequently, as p decreases, will increase somewhat faster than 1 1 —— , and N will increase somewhat faster than — . 2 o p P 4. Carrier frequency repeater, square law or similar non-linear de tector. This type of repeater behaves essentially the same as a baseband repeater with a square-law device, case 2 above; is approximately independ ent of p. "

Rowe thus concludes that improved performance is attained ly using a non-linear detector in both baseband and carrier frequency repeaters, using either tuned circuit or locked oscillator timing circuits. It should be pointed out at this stage that the conclusions for the carrier repeaters are for systems which use the detected data envelope as the source for the self-timing in formation. No attempt has been made to utilise the spectral information of the carrier frequency to avoid the jitter growth.

The first publication (July 1957) appearing in the literature which provided a detailed analytic examination of some of the specific problems g associated with self-timed regenerative repeaters was written by E.D. Sunde.

He discussed the timing principles for a particular type of self-timed re generative repeater invented by L.R. Wrathalf-^September 1956) in which a timing wave, derived from either the received (forward-acting) or the regener ated (reverse-acting) pulse train, was combined in a particular way with the received pulse train to provide either complete or partial control of the re timing process. Sunde resolved the pulse train passing through the regenerator 15

into three components:

1. steady state or systematic component at the pulse repetition

frequency (line spectrum)

2. random component which when combined with 1 yields the particular

pulse train (continuous spectrum) but gives rise to a variation

with time in the amplitude and phase response of any mistuned

resonant circuit.

3. dipulse component which is a second order effect of 2 above which

accounts for the random pulse displacements occuring in the re

generator output.

Component 1 represents the source for the ideal unperturbed timing frequency wave. Component 2 always will give rise to both amplitude and phase variations in the timing wave, even when a correctly tuned resonant circuit is employed. This is a direct consequence of the random nature of Sunde's pro posed Component 2. Thus it is not necessary to mistune the filter to produce the undesired timing deviations. Sunde utilised these separate components to initially analyse the effect of a single tuned LCR filter in establishing the r.m.s. deviation in the timing wave as a function of the Q factor of that filter network. This information was then applied to a repeater chain so that the accumulation of random timing deviations and the resultant reduction in tol erance to noise of the repeaters towards the end of the chain may be evaluated.

To provide a satisfactory upper and lower limit to the results the calculations were made with direct and root-mean-square additions. From these results a cumulation factor C was defined, 16

C = r.m.s.

direct

where N = number of repeaters in chain r Y = the factor by which the rms timing deviation at the n n repeater is smaller than at the first repeater, with the

timing deviation originating at the first repeater only.

With a Q = 100 and limiting the maximum phase deviation to approx imately 1 radian, C factors of 1.25 (r.m.s.) and 2.5 (direct) were computed.

Based upon these results the timing deviations are virtually completed in a chain of 3 to 6 repeaters and thus Sunde concluded that the accumulation of random timing deviations could for practical purposes be disregarded, and also any experimental observations for a long system analysis could be con ducted on a simulated system operating with that small number connected in tandem. The minimum number of repeaters required to simulate a real system will be discussed further in Section 2.3 where the results obtained from simulation networks and actual systems are presented. Of further interest to this thesis is Sunde's comments relating to the accumulation C of random timing deviations increasing indefinitely with N if complete retiming employed. 4 \ C = (— N ) r.m.s. accumulation, which he states is the same as would be ob- tt r tained if a timing wave was transmitted on a separate path, with a resonant circuit at each repeater to limit noise and with amplification to compensate for losses. This slow rate of accumulation , must necessarily limit the circuit length of any system employing these techniques. Extending Sunde's 7T Q ^ analysis where he obtained that N = — (—) if C = 1, it is possible to show r 4 tt 17

that with a separate timing path, narrower bandwidth filters than is possible for the data path can be employed. This would enable a larger number of repeaters to be connected in cascade before the same magnitude of r.m.s. jitter occurred.

By having different bandwidths (Q-factors) in each channel would not be desirable because it would increase the difficulty of keeping the timing wave in phase with the data envelope because of the different phase delay characteristics produced by that particular configuration. Thus the separate timing path does not provide the solution for the jitter accumulation. As a final performance analysis Sunde computes the number of repeaters required to produce a timing deviation at the output of the chain equal to that at the input to the chain. For perfect tuning in all repeaters, = 800 when Q = 100. With mistuning and maximum allowed phase shift of 30° the N = 300 for the same Q. In his summary Sunde indicated that if his suggested timing principles were implemented by appropriate repeater in strumentation, a performance could be realised which approached that of ideal regenerative repeaters. To this end it would be necessary to meet certain require ments with regard to the Q-factor of the resonant circuit, its frequency precision, the shape of the received pulses and the amplitude of the timing wave in relation to that of received pulses. The results of other researchers in this field will be used to show that this optimistic "ideal" performance cannot be realised in practice because of the systematic jitter produced by the signal dependent con version processes within the regenerators.

3 O.E. De Lange (November 1958) recognised the importance of these con version processes, "Our experience up to the present time leads us to believe that, unless some specific remedy is applied, this problem may be more serious than that posed by random noise. The frequency spectrum of the phase deviations resulting from changes of pulse pattern is determined by the rates at which the pattern changes and the bandwidth of the timing circuit in which the deviations are produced. The spectrum at the output of any repeater will be further modified 18

by all succeeding repeaters. The solution to th

bandwidth of 320 MHz, pulse repetition frequency 160 Mbits/s, timing Q = 200 and N = 200 repeaters. Use of a much higher Q network, such as that produced by r a phase-locked oscillator, distributed throughout the network was also suggested as means of extending the circuit length or improving a given performance al ready obtained with simple LCR filter networks. On present day standards, utilising integrated circuit techniques, every regenerator could contain the phase- locked oscillators in lieu of the LCR networks without any major increase in overall system costs.

Although the effect of changing digital patterns had been recognised as a source of timing jitter its degree of seriousness had not been exposed until the 7 work of C.J. Byrne, B.J. Karafin and D.B. Robinson, Jr. (November 1963) was pre sented. From their analysis the amplitude-to-phase conversion processes introduced the same pattern dependent jitter in each regenerator. Therefore, the jitter accumulates systematically and from the results of the spectral distribution of the timing jitter energy, the low frequency components increase without bound in a long chain of repeaters. The r.m.s. value of this jitter was shown to increase as the square root of the number of repeaters in the chain, N^ for the systematic 2 3 jitter is larger than for random jitter already determined by Rowe and De Lange % as . Therefore for long regenerator chains the systematic jitter must dominate. 15 It is pertinent at this point to refer to research of Kinariwala , who produced the observations that although the total r.m.s. jitter is unbounded, it is bounded for the jitter components located near any one frequency. The explanation for the paradox lies in the fact that the bound for the very low frequency jitter com ponents becomes very large as the chain grows longer thus validating Byrne et al’s comments about the unbounded growth of the low frequency systematic jitter.

To maximise the length of any long distance chain of digital repeaters every step must be taken to avoid the above systematic jitter build-up otherwise the system will rapidly lose synchronism. One particular method has been 20

0 developed by L.E. Zegers (August 1967) wherein the application of pattern trans formations to the output signals of successive repeaters is applied to break up the coherent jitter accumulation. In fact Zeger demonstrates using both theoret ical data and measurements on an experimental network that his coding technique does convert the r.m.s. systematic jitter N accumulation rate to the random

u- N 4 rate. According to the measurements conducted by Zegers, see Figure 2.9 in r Section 2.3, the systematic jitter accumulation for only 7 repeaters equalled that obtained for 100 repeaters with random jitter accumulation.

9 The next paper to be considered is that of J.M. Manley (March 1969).

He studied, both theoretically and experimentally, three sources of timing noise in a self-timed regenerative PCM repeater. These were tank circuit mistuning, amplitude-to-phase conversion and pulse shape. He discusses how these noises accumulate and combine along a chain of repeaters. The theoretical work is de veloped from a frequency analysis viewpoint which leads to the timing noise spectrum.

The spectrum for mistuning is shown to have no energy at zero frequency and hence cannot build up indefinitely along a chain of repeaters but soon reaches a limit of approximately four times the peak of the first repeater. This important result is utilised later in Section 4.6.2 where use is made of the fact that in an Amp litude Modulation system the spectral characteristics of 1he baseband modulating signal is preserved in the sideband spectra of the modulated signal. The zero energy at zero frequency for the baseband system must produce zero interfering energy at the carrier frequency. Mistuning can then only have limited effect upon the carrier frequency component. The spectrum of timing noise caused by amplitude- to-phase conversion is shown to have energy at zero frequency; thus timing noise from this source increases indefinitely along a repeater chain. Some of the timing noise is attributed to pulse shape alone and in some cases, aliasing of harmonics near the pulse rate down to very low frequencies by the sampling process 21

occurs. No analysis was conducted into the effect of random noise. Figures 2.12 and 2.13, in next section, represent the results obtained by Manley for the spectra, both theoretical and experimental, of mistuning and amplitude-to-phase conversion

for different numbers of repeaters. No investigation into error rate character

istics was performed.

19 A.E. Karbowiak (July 1962) considered the proposal that instead of

attempting to utilise perfect slicers in a repeater, the desired regeneration

process could still be produced provided the amplitude transfer characteristic

for the non-linear amplifier could be made satisfy a particular law over its entire

operating range:

v (2.1) out

where A = arbitrary constant

a = any odd number greater than 2.

If the above law is satisfied the signal-to-noise ratio will be improved

by at least 3 db per repeater and will remain above a controllable minimum; there

is no limit on permissible circuit length provided that, "if at every repeater the

timing of the pulses is improved so that the r.m.s. jitter with respect to the mean pulse repetition frequency is decreased by 3 db or more, then timing errors will not be cumulative and the system can be satisfactorily operated". Karbowiak

suggests a suitable partial amplitude regeneration transfer function but leaves

the choice of the partial timing improvement network as a problem for the reader.

Based upon the previously presented material in this section it is not possible

to employ a simple LCR filter network to perform this task because all components

of the spectrum passing through the filter are not uniformly attenuated, in par

ticular the low frequency systematic jitter components are coherently added almost without bound. If the systematic jitter sources can be removed from the 22

regeneration process then there is every possibility that the system length can be increased by several orders of magnitude.

The basic aim of this thesis is to develop a self-timing regeneration technique which produces a pattern insensitive timing wave. This will avoid the systematic jitter accumulation thereby enabling larger numbers of digital regenerative repeaters to be connected in cascade. The conclusion to be drawn from the information presented in this part of the literature survey is that any attempt to use the pure baseband binary digital data must necessarily re sult in the undesired systematic jitter being produced. Hence an entirely new approach will be required to overcome the problem. Without the systematic jitter being produced the advantages of using simple LCR bandpass filters could then be applied.

2.3 Simulation Networks

The purpose of this section is to provide details on the various methods employed by different research workers to test the performance character istics of self-timed digital regenerators. In general the mathematical analysis which is required to produce a general model for the a communication system employing these self-timed digital regenerators is complex. To yield tractable mathematics approximations are required which limit the extent of applicability for the general model initially proposed. In a survey paper written by B.K. 16 Kinariwala and S.L. Freeny of the Bell Telephone Laboratories (1964), the authors' comment that a general solution to the timing problem would involve first characterising each source in such a way that, given the pulse pattern and given a complete description of the repeater, it would he possible to cal culate the jitter contributed by each source. In addition it would be necessary to know quantitatively how the various jitter components propagate through successive repeaters and how they combine with the new jitter at each repeater. 23

So far such a complete solution as this has steadfastly resisted the many determined and elegant approaches made toward it. This is true for a variety of reasons. First of all, the transformation performed by the tuned circuit on the jitter of the incoming pulse wave is non-linear with memory and is time-varying, i.e., about the most general type of transformation one can have. Moreover, the memory is large, as it must be if the tuned circuit is to perform its function properly. The transformation depends on the pulse pattern, hence the time-varying property. Although a general solution has not been achieved because of the difficulty associated with the characterisation of the noise sources, particularly those arising from the non-linear operations within the regenerators, important and sufficiently general information has been obtained from the ’’linearised" models utilised in the Section 2.2 to pre dict the performance of repeater chains. These analytic performance predictions have been checked by different investigators by conducting measurements on both laboratory simulation models and actual field installations. Returning to the survey paper of Kinariwala and Freeny the authors conclude that, "there are many areas of the timing problem which are yet unsolved. At present there are various experimental investigations being carried out and, at this point, these seem to hold the most promise for a better understanding of these unsolved aspects". For these reasons an experimental simulation network was selected for this investigation and hence a survey of the available literature on the experimental methods employed by other research workers is provided in this section.

The first research paper which specifically dealt with the simulation 5 of a digital regenerator was written by O.E. De Lange in January 1956. His tests were conducted to evaluate a simple device which had been produced for regenerating digital data at microwave frequencies. To determine the 24

capabilities of the devices one of them was connected into a circulating test loop in which groups of pulses were passed through the device a large number of times and thus avoided the necessity of building a large number of devices. The operating carrier frequency was 4 GHz and either a 20 or 40 MHz pulse repetition frequency was employed. The loop time was 400 nanoseconds and the pulse word was limited to contain a maximum of 5 bits. The normal number of regenerations produced by this system was 24 but 100 was tried and found successful. The particular performance characteristic being investigated was the partial amplitude regeneration, at the microwave frequency, employing a differential bias on 2 diodes being controlled by the timing wave. De Lange’s results show that it is not necessary to perform the amplitude regeneration at the baseband frequency. Satisfactory performance can be obtained by partial amplitude regeneration at the carrier frequency. However, this technique does not solve the difficulty associated with obtaining a suitable timing wave. This basic problem still exists and until it has been solved, neither baseband nor direct carrier amplitude regeneration will produce the desired increase in the number of repeaters that can be connected in cascade. De Lange's timing wave was not derived from the input waveform but was obtained from external gener ators which were phase locked to enable the stringent requirements placed upon the loop synchronisation and gating to be satisfied.

De Lange utilised a series of photographs to present some of his results and these clearly establish the capability of loop testing to reveal any noise build and timing variations. No analysis of the effect of the sim ulation network on the spectral characteristics of the regenerated pulse train was performed. The recommended peak signal-to-r.m.s. noise ratio for satis factory operation was 23 db and also De Lange made the following comment, "It should be pointed out that although half amplitude is the preferred slicing for the baseband pulses this is not the case for carrier pulses. W.R. Bennett of 25

Bell Telephone Laboratories has shown that for carrier pulses the probability that noise of a given power will reduce signal pulses below half amplitude is less than the probability that this same noise will exceed half amplitude.

This comes about from the fact that for effective cancellation there must be a

180° phase relationship between noise and pulse carrier. For this reason the slicing level should be set slightly above half amplitude for a carrier pulse system."

4 In November 1958 O.E. De Lange combined with M. Pustelnyk to produce further experimental evidence on the behaviour of the digital regenerative re peater. Because the recovered timing information is at baseband frequency in a carrier system as well as in a baseband system the authors, even though their main interest was in microwave systems, built a chain of baseband repeaters to simulate, as nearly as possible, the microwave repeaters of interest. The pulse repetition frequency was 10 MHz and the experiment had 2 main objectives;

1) to determine the effects of noise in producing pulse errors through its action on the timing circuit

2) to determine the effects of noise in producing time or phase de viations of signal pulses.

They found that the number of errors produced by the effects of noise upon timing was negligibly small in comparison to the number produced by other effects of noise. Their results also indicated that as far as the random noise was concerned the amount of time deviation of pulses at the output of a long chain of repeaters can be kept within tolerable bounds. To avoid the recog nised problem which was common to all forms of self-timing repeaters, of con version of changes in pulse pattern into changes in the time of occurrence of the pulses out of the repeater, the experiments were conducted with fixed pulse patterns. 26

average number of errors in 10 pulses Figures 2.4 & 2.5. ______FIGURE FIGURE

2.5 2.4

Error both (De respectively Average noise (b) (a)

Lange, externally self-timed,

timing

build-up in

pulse Signal-to-Noise timing

Pustelnyk and

except

error

timed, slicing path noise

chain 4

non-linear rate

) early; ______noise in

paths;

with both of

1 present.

into (d) self-timed out

linear slicing db

timing (b) and

slicing of

9 noise

(e),

timing pulses amplifier. and

repeaters:

path same in timing

amplifier;

timing

as only;

paths; (a)

path

(a) (c) and

noise

self-timed only. (c)

in 27

The experimental analysis was initially performed on a single repeater which possessed the capability of allowing the noise to be selectively fed into various sections of the repeater. Their results are plotted in Figure

2.4 where the reduction in errors due to a non-linear amplifier in the timing path is observable. The next phase of their experiment was to increase number of repeaters in cascade to 5. The peak signal-to-average noise level was set at 22 db and the errors obtained with different number of repeaters were measured, see Figure 2.5. It should be observed that errors due to noise in the timing path is negligible when compared to the effects in the amplitude slicing path. Following these results the phase or time deviations were con sidered and, based upon the well-established principle that as the number of repeaters is increased the effective bandwidth of the system is reduced, i.e.

1 77 "^3^0son3.nc0 B = —1------, a comparison between calculated and measured phase eff Q S--- r deviation was established. The number of repeaters was finally increased to

22 and employing the same analytic approach it was estimated that, if the jitter I* growth is proportional to , then with Q = 80 and a 20 db input peak signal- to-average noise ratio in each repeater, the r.m.s. phase deviation in the timing wave as measured to be 4.5° at the 22n<^ repeater would be only 8° for 200 re peaters operating under the same signal-to-noise conditions. An important point which should not be overlooked with these measurements and predicted re sults is that, for the reasons indicated earlier, a fixed test pattern was employed with one pulse out of nine being present. Some simple tests were con ducted with changes in digital pattern. The phase delay through the particular repeater employed amounted to 10° phase change in the timing wave when the pattern was switched from one pulse out of nine to seven pulses out of nine. No results of tests were given on the cumulative effect of this pattern dependent phase shift or how these shifts could upset the synchronisation of a loop simulation network due to the phase delay change changing the overall loop time 28

delay. De Lange provides no explanation for changing over from one form of simulation testing to another nor does he utilise the opportunity to draw com parative conclusions for the different techniques. He originally favoured the use of only one repeater in a loop because it avoided the construction of many repeaters and then in this latter investigation avoided the loop test and con

structed 22 repeaters. The paper is mainly descriptive and no reason has been

given why 22 repeaters is an adequate set to use for the analysis. It will be

shown later that this number of repeaters is more than the minimum number re

quired for prediction purposes. However, within the indicated bounds of the

experimental investigation De Lange and Pustelynk were able to show that the 6 ^Li random jitter, as predicted initially by Sunde , does accumulate at the rate.

The effect of this slow rate of accumulation is such that this jitter can be

held within small enough limits to enable at least 200 repeaters to be employed

in a system. The selection of the minimum operating signal-to-noise ratio

could be based primarily on the action of the noise in the amplitude-slicer path because this path yielded a higher error rate than the equivalent amount of noise in the timing path.

7 The analytic work of Byrne et al has been previously referred to

in Section 2.2. They also were able to conduct measurements in both the

laboratory and in an actual field installation constructed between Passaic and

Newark, New Jersey, U.S.A. In the laboratory, trains of one to ten repeaters were employed whilst trains of 14 to 84 repeaters, in steps of 14 because of the line construction and switching centers, in the actual field installation were utilised. The test equipment was located at one end of the system and therefore the line had to be looped back to the sending end for phase com parison and error measurements. The objective of this study was to examine the systematic jitter accumulation in the system as a result of the pattern dependent phase shifting mechanisms which existed in the self-timed repeaters 29 360

0db=1 degree2 per Hz

100 Frequency H z •~J FIGURE 2.7 Measured spectral distribution for a random pattern (Byrne7) 30

employing simple tuned timing filters. Eight bit word patterns were selected yielding a total of 35 non-duplicated formats. Figure 2.6 shows the results obtained for steady-state phase shift versus pattern changes for different numbers of repeaters. A quasi-random pulse pattern was also applied and the output from a phase detector was recorded on tape and a spectral analysis was performed on this information to yield Figure 2.7. Also of interest is the result plotted in Figure 2.8 which shows the r.m.s. jitter amplitude in degrees versus number of repeaters for the laboratory system, the field installation and the proposed model. This is the only investigation which brings together the results from analytic, laboratory simulation and actual field installation observations. Because of the close agreement between these three results the important conclusion can be drawn that laboratory simulation networks can en able adequate predictions to be made about the behaviour of a real system. If a sufficient number of the repeaters is utilised in the laboratory tests then the performance of long repeater chains can also be predicted by a simple ex tension of the results obtained for the particular phenomenon under investigation.

With reference to Figure 2.8 the minimum number of repeaters in the simulation network would have to be between 5 and 10. This minimum number of repeaters will be shown later to exceed that number required to produce the maximum jitter due to mistuning. Two other later investigations, employing this same type of 13 8 cascaded simulation network, used ten and twelve repeaters for their per formance analysis. Returning to the investigation of Byrne et al they also demonstrated that the low frequency jitter power was mostly located well below the 3 db half bandwidth of the timing filters (below 400 Hz for half bandwidth

10,000 Hz with 84 repeaters of Q = 80). The r.m.s. jitter increased in proportion to the square root of the number of repeaters, . The amplitude dis tribution of the jitter was Gaussian. No measurements were conducted by the authors for random jitter accumulation. 31

corr --asymptote

FIGURE 2.8 Root-mean-sauare jitter due to a random pattern versus number 7 of repeaters in the chain, measured and calculated (Byrne )

FIGURE 2.9 Theoretical data compared with the measured r.m.s. values of Q jitter as a function of the number of repeaters (Zepers ) 32

8 7 Zegers (August 1967) utilised the model proposed by Byrne et al for his pattern transformation method of breaking-up the coherent jitter trans port mechanism. To verify his results he constructed a transmission chain with 12 repeaters and used a binary pseudo-random word generator of variable period length. By simple switching arrangements within each regenerator Zegers was able to study the systematic and random r.m.s. jitter accumulation and % ^ Figure 2.9 shows the results confirming the N and N respective rate of pro portionality over the number of repeaters employed.

In actual Pulse Code Modulation systems the code of all zeroes is sometimes avoided to guarantee that there always exists at least one digit transmitted, say every 7 or 15 digit positions, to keep the timing circuits active. Another technique is to employ multilevel or n-nary codes. In par- 13 ticular Dorros et al (September 1966) utilised a ternarny code which provided sufficient redundancy (0.59 bits/symbol) to allow transmission of unrestricted binary sequences while still providing timing information to the repeaters.

Also the ternary code eliminates D.C. components from the transmitted spectrum, thus permitting A.C. coupling in the repeaters and powering with D.C. down the same line. The particular objective of the study was to derive the performance characteristics for a single repeater from the overall objectives for a 4000 mile coast-to-coast digital repeatered line operating at 224 Mb/s with an overall —6 “d-O error rate below 10 , or below 2.5 x 10 per link. The repeater used forward acting complete retiming with non-linear extraction of the timing information from the paired selected ternary (P.S.T.) code. Ten repeaters were operated in tandem, at one mile intervals, to form ten miles of repeatered line.

To meet the design objectives with an economical repeater design, appropriately spaced high Q jitter reducers were required along the line. A jitter reducer includes an automatic phase control (A.P.C.) loop to smooth the 33

jittered timing wave, and buffer storage for the information pulses. The information pulses are read into the store by the jittered timing wave and read out by the smoothed wave. The recommended repeater Q was 80 whilst the jitter 6 reducer required, if only one reducer per 1000 miles was employed, a Q of 10 .

If the jitter reducers are spaced closer together, say at all main switching centers, then their Q may be lowered. The authors do not give any estimates of the buffer storage required to compensate for the jitter accumulation. Based on previous observations it would seem likely that as the overall circuit length was increased, the number of repeaters would have to be increased which in turn increases the pattern dependent systematic jitter. To compensate for the large amount of low frequency timing deviations the buffer storage capacity could become uneconomically large.

41 The buffer storage capacity was examined by F.J. Witt (November

1965). The phase-locked loop was adjusted so that the "elastic store would be half full on the average, and the elastic store is made large enough so that overflow does not occur often enough to be a problem". For the system described by Witt for a total number of 3,600 repeaters with a "dejitterizer" every 360 repeaters, one overflow per minute, week and century would be 13, 15 and 17, respectively. The analysis was performed for an oscillator instability of

4 p.p.m.,, r.m.s. jitter = 11.2° per repeater, repeater Q = 80 and dejitterizer

Q = 10^. No digital errors occurred during the experiments conducted on an actual dejitterizer "because the input phase deviation was limited (by clipping) to a level which prevented store overflow". The very act of limiting the phase deviation must necessarily invalidate the results obtained ly Witt for his system containing one 8 bit store dejitterizer and a noise spectrum which had been shaped to represent the output from a chain of repeaters. Also the shaped spectrum would not contain the systematic components, produced by an actual system. It is the low-frequency portion of the jitter spectrum that propagates 34

calculated

approximation

100 Frequency

41 FIGURE 2.10 Spectrum of dejitterizer: input jitter (Witt )

--Spectral density of input jitter. __ •• •• output •' , calculated for Q=1Q6 • Measured output jitter.

100 Frequency

FIGURE 2.11 Spectrum of dejitterizer: output jitter (Witt ) 35

through the network and causes loss of synchronism. Filtering cannot remove this effect. Some photographs were included in the paper to illustrate the smoothing effect of the dejitterizer with a lKHz sinusoidal jitter component which was producing a 7 time slots peak-to-peak jitter. Figures 2.10 and 2.11 depict the shaped jitter spectrums employed for the tests and it should be ob served that the lKHz tone is 20 db down on the maximum calculated input jitter and 40 db down on the maximum measured and calculated output jitter. No in formation was provided to enable correlation between the lKHz sine wave and the r.m.s. input jitter spectrum. Insufficient storage must produce situations where the store could be empty or full and thus the desired smoothing flywheel action of the jitter reducer would be lost. It seems feasible that since during the course of this investigation an advantage in reduced error rates was obtained by employing a non-linear device to limit the introduced phase deviations, a similar advantage could be obtained in the dejitterizer itself to combat the overflow problem. The transfer characteristics of the phase locked control loop could be modified to include a particular non-linear transfer device which would increase or reduce the speed of the elastic store at a non-linear rate proportion al to the store occupancy, to ensure that overflow effects were minimised without having to uneconomically increase the size of storage required per jitter re ducer. Thus by placing greater emphasis on circuit stability, precision and overall'complexity, longer circuit lengths can be employed but this does not solve the basic systematic jitter problem.

9 As previously discussed, Manley , the coherent jitter of the pattern dependent processes cannot be eliminated by filtering because of the non-zero energy distribution around the timing frequency produced by these systematic accumulation processes. The measurements conducted by Manley to verify his theoretical results were not performed on chains of varying numbers of real re peaters. Instead a simulation network containing one real repeater, of special 36 FIGURE

Degrees r.m.s. in 0.625 Hz band. S Degrees r.m.s. in 0.625 Hz band. 2.13 2.12

Timing and tank Timing

trigger Q

noise noise =

100

circuit spectra spectra (Manley

offset

caused caused q p-. —

544)

•

from

measured calculated by by

timing 0.1% zero

mistuninp

(Manley^ wave

62.5

amplitude

p.546) of Hz

timing

variations 37

calculated measured r.m.s. me from distribution

FIGURE 2.14 Total timing noise caused by 0.1% mistuning of timing tanks in 9 chains of like regenerative repeaters (Manley p.602) 38

design, and a 5-track tape recorder. Two tracks contain previously recorded pulse trains and timing information. These two tracks serve as the input to the repeater whilst two other tracks are employed to record the pulse train and timing wave information from the output of the repeater. These latter two tracks then serve as the input for the next regeneration process; the cycle is repeated until the required number of regenerations are performed. The fifth track was employed for speed servo control. Because no recorder had steady enough speed the timing wave could not be recorded directly. Instead the phase deviations were recorded and during playback the timing wave was reconstructed, inclusive of any phase jitter produced by previous regenerations, with a phase modulator.

In general the 240 bit pulse train consisted of random unipolar pulses at a IKHz rate. The record length was approximately 15 minutes which provided a sufficient ly long statistical sample set from which the spectral results were obtained.

The results for mistuning are plotted in Figure 2.14. Manley's experimental arrangement is equivalent to a loop simulator and no account has been taken of the periodic sampling procedure which is produced by this technique. Because he operated at such a low frequency he was able to use test equipment with a much higher degree of resolution than would have been obtainable if the experiments were conducted at microwave frequency.

This completes the survey of the experimental investigations. Two main techniques have been shown to exist. The loop test has been successfully applied at both microwave and audio frequencies and offers the investigator the opportunity to minimise the number of repeaters required to perform his analysis. The effect of the periodic circulation of data through the one re generator and subsequent modification to the spectral information has not been considered by the investigators who used this technique. The other method of long repeater chain simulation analysis utilized a cascaded connection of re peaters (non-looped). The highest number used in the laboratory was 22 whilst the highest number in a field installation was 84. The minimum number required 39

to enable adequate predictions to be made has been shown to be between 5 and

10. These numbers are consistent with the theoretical predictions of the minimum number of repeaters required to enable fhe systematic jitter sources to be made clearly distinguishable from the non-systematic or random sources.

2.4 Conclusions

The major defect in existing self-timed digital regenerative repeaters is that they produce pattern dependent timing jitter components in their output spectrum. This jitter accumulates systematically as the number of repeaters

in a system is increased, r.m.s. systematic jitter proportional to , until

its magnitude becomes too large and synchronisation is lost. Without employing sophisticated jitter reducers and high coding redundancy the number of repeaters connected in cascade is normally limited to 100-300 repeaters. To overcome the 3g length limitation being produced by the factor, the causes of the low fre quency jitter accumulation must be eliminated. This would require that the new type of self-timing regenerator must not contain any pattern dependent jitter production sources and be capable of deriving its timing frequency component, which ideally is a spectral line impulse, from the regenerator input (forward acting) or output (reverse acting) spectrum. The reduction of the overall jitter from the systematic component to the random would, for the same tolerable timing phase margin, enable a system increase from 1,000 to 1,000,000 repeaters.

The specific aim of this thesis is to design, implement and test a regenerator which will enable a simple direct method of self-timing, not em ploying coding restraints, to be produced which does not suffer from pattern dependent systematic jitter accumulation. 40

CHAPTER 3

SYSTEM DESCRIPTION

3.1 Introduction

In this chapter the design criterion which has to be satisfied for the digital regenerative repeater and simulation test network is presented.

A general description, void of detailed mathematical analysis, of how each function has been achieved is also given together with discussions on al ternative configurations and implementations. Where applicable references are given to the section containing the specific analysis. Full details of the actual circuitry employed in the regenerators for the investigation are presented in Appendix 2.

3.2 The Coherent Carrier Self-Timed Digital Regenerative Repeater

The main emphasis of the investigation has been placed on being able to produce a self-timing technique suitable for digital communication systems containing many hundreds of regenerative repeaters. Existing tech niques have already been shown to be adequate for the systems containing less than 100 regenerative repeaters. Because of the large number of repeaters envisaged, each individual repeater should not be too complex otherwise the total system cost may become prohibitive and service reliabilitv rendered too difficult to achieve because of the large number of components, circuit stability and precision required.

Whatever the method chosen it must provide a simple, inexpensive and reliable regenerative digital repeater. Also because of the state of the technological development with electronic integrated circuits the implementation of the circuit design should utilize these devices as a guide for a network 41

constructed entirely of integrated circuits.

Before proceeding to the design criterion which has to be satisfied for this particular digital regenerative repeater, two simplified design philosophies are presented for consideration. The significance attached to these philosophies should not be underated otherwise a totally wrong concept will be applied to the design, implementation and testing procedures.

1. Because present systems are limited to contain a maximum number of repeaters proportional to the degree of perfection that each individual re peater can obtain, then if the individual repeater circuit design can be improved the number of repeaters connected in cascade can be increased. With an ideal or perfect regenerator an infinite number of regenerations can be performed because the timing wave would be jitter free.

2. If the individual repeater is designed to provide only a specific improvement in the timing jitter between its input and output, then provided the improvement is such that the equivalent of a convergent geometric series progression is obtained for a whole chain of repeaters, the timing jitter will remain bounded, never exceeding a predetermined amount which can be set below that level which would normally cause loss of synchronisation due to the de gree of spectral impurity of the timing wave. With the controlled level of timing jitter an infinite number of repeaters could be cascaded even though the timing wave would not be jitter free.

The latter philosophy has been chosen as the guide for the overall design objective of this investigation. It will be shown that the baseband retiming wave is derived directly from a spectrally impure carrier frequency but there still exists a coherent phase relationship between the carrier 42

frequency and the retimed envelope. Provided no pattern dependent process exists within the regenerators this coherence will be produced in the output

of all the regenerators. For this reason the title "A Coherent Carrier Tech nique for Self-Timed Digital Regenerative Repeaters" has been applied to the

thesis.

3.2.1 Design Criterion

In order to synchronise a chain of self-timed digital regenerative

repeaters each individual repeater must be capable of extracting from the in

formation signal a timing signal which possesses sufficient spectral purity

to perform the necessary retiming process. If the repeater chain is to con

sist of many hundreds of repeaters then it is absolutely essential that the

timing signal contain no spectral components that will lead to a systematic

accumulation of the timing jitter. If the information signal contains no

large amounts of timing code redundancy then any attempt to utilize the pure

baseband signal must result in a pattern dependent timing wave due to repeater

imperfections. If the digital information is modulated on to a carrier fre

quency, as can quite often be the case, then whatever comments and conclusions

have been made about the baseband signal must be applicable to the sidebands of

the modulated signal.

The need to produce an impulse type response (discrete line) in the

frequency domain has already been discussed in Chapter 2 and the technique of

coding redundancy to achieve this result, with consequent reduction in avail

able channel capacity commented upon. However, the frequency spectrum of a

modulated pulse stream does contain a discrete line that can be made independent

of the data pattern being transmitted. For an angular modulation process

certain modulation indexes would have to be avoided to guarantee the existence 43

of the line. For amplitude modulation processes no restriction on data patterns is necessary to guarantee the presence of the line and its independence of the data content. This spectral line is, of course, the carrier frequency.

The proposal is then to use an amplitude modulated carrier system and to utilize the carrier frequency as the means of obtaining the timing signal. It will be necessary to guarantee coherence between the carrier fre quency f and the digital data transmission speed f^. This concept of relating the fc and f^ by an integer ratio is certainly not new, see Appendix 1, however what is original and significant is the fact that self-timed digital regener ative repeaters can be implemented using this integer ratio between f and f^ as the fundamental timing design criterion.

3.2.2 Functional Description

With the type of modulation process chosen, the problem of obtaining the timing wave will now be discussed. The signal received by each regenerator will be a pulsed AM wave. To reduce system bandwidth requirements a non-return to zero (NRZ) baseband binary data modulating signal is employed. With 100 percent modulation of the carrier the difficulty of obtaining a constant amplitude continuous timing wave during periods of long zeroes still exists with the associated pattern dependent jitter accumulation. It was initially contemplated that each regenerator would contain its own carrier frequency oscillator. A second order phase-locked loop with a sample-hold network was also to be included which would provide the essential corrections to each local carrier frequency oscillator by deriving an error signal each time that an AM pulse arrived and holding that correction until the next pulse arrived.

This would cause all the regenerators to track (ideally) the original trans mitter frequency. This technique was never implemented mainly because of its 44

complexity but more importantly because a simpler solution to the problem was found.

The basis of the simpler solution is directly concerned with the 20—25 interpretation of the Fourier Transform theory . Analysis of the initially proposed system revealed that without including any external noise or system imperfections, the random interruption of the carrier frequency due to the data variations must lead to a dispersion of the carrier frequency line width and its associated modulation sidebands, plus the effect of producing an additional continuous Mnoise-like'T spectrum derivable from the slightly different

(random) local oscillator frequencies. The amount of line width dispersion and "noise" is a function of the number of carrier cycles per pulse, the pulse pattern variation and the corrections applied at each regenerator. The Fourier

Transform theory is explicit in detailing that if a line (impulse) spectrum is required in the frequency domain then a continuous time function must exist.

In other words the transmitted carrier frequency should never be completely switched off. This condition can be satisfied by allowing a low level of carrier to "leak-out" through the system during the OFF periods. The pulse

ON level to pulse OFF carrier leak level will be referred to as the ON/OFF ratio and the actual ratio required for optimum performance is the subject of part of the experimental investigations to be presented in Chapter 5.

Since the carrier signal is now available at all times the re generators do not have to have local oscillators with phase-locked control loops. This results in a marked simplification of design and implementation but more importantly an improved performance will be shown to have been achieved. However, the regenerators still have to extract the carrier frequency from the incoming AM spectrum, provide a constant amplitude signal for re timing and amplify the carrier wave to a level which is suitable for the 45

Channel To Envelope Amplifier Detector

Hi- Q B.P.F.

Amplitude Limiter

Low-0 To Frequency B.PF. Divider M

To Modulator

♦

FIGURE 3.1. The carrier frequency filter-limiter network 46

remodulation process. To provide these functions the regenerator contains a high Q (narrow bandwidth) filter followed by an amplitude limiter which in turn is followed by a low Q filter, see Figure 3.1. Both filters are tuned to the carrier frequency.

Because communication systems operate in the presence of noise the carrier leak signal level will be such that the signal-to-noise ratio during the OFF periods will be low. The tolerable level of carrier leak buried in the noise is the subject of several parts of the investigation, Sections 4.4-

4.5, 5.3-5.6. The extraction of the leak signal from the noise is performed by the high Q filter. In performing this latter task the filter will also be found to satisfy another important aspect of the operation. This is the con trol of the noise build-up along the repeater chain. An analysis of this noise convergency criterion is presented in Sections 4.2-4.3.

Thus the first filter in the timing section has a triple task to perform.

1. Extract the carrier frequency from the AM spectrum and reject the sidebands

2. Extract the carrier frequency from any noise

3. Control the noise build-up along the repeater chain.

Because the first filter has a finite bandwidth, even if it is perfectly tuned, the spectral purity of its output will be such that it contains both amplitude and phase variations. The tolerable amount of phase variation is a function of the signal-to-noise ratio at the filter output, Sections 4.2-4.5.

The amplitude variation can be minimised by passing the filter out put through an amplitude limiter. Apart from the desirable effect of producing 47

a constant amplitude output over a relatively wide frequency range, the highly non-linear operation of the limiter modifies the signal-to-noise ratio improve ment, Section 4.5, and produces a multitude of harmonics of the carrier fre quency together with any adjacent frequency components. The desired part of the spectrum is filtered out by using a second filter. The bandwidth of this second filter could also be made very narrow so as to further improve the signal-to-noise ratio of the timing section but in actual fact this is not necessary provided the first filter has been selected properly. A low Q filter will suffice. From qualitative observations made on the particular implemen tation used for the investigation, this second filter could actually be removed.

The resultant carrier frequency output waveform would be more rectangular than sinusoidal. The output channel amplifier bandpass filters would then have to be relied upon to remove the unwanted harmonics from the regenerator output.

If filtering is not performed before the limiting, the full AM and noise spectrums will be passed through to the non-linear limiter and produce a complex output waveform consisting of all the original frequency components plus their harmonics and intermodulation products. Although filtering will readily remove the harmonics, some intermodulation products would be passed through the finite bandwidth of a filter producing a waveform that had pattern dependent components. The propagation of these additional frequency compon ents situated very close to the carrier frequency is a potential source of systematic timing jitter accumulation and can be avoided by prefiltering the input to the limiter and ensuring linearity of operation for all stages pre ceding the filter.

With the carrier frequency present at all times, all the regenerators are locked to the transmitter carrier frequency with the data being an integer sub-multiple of that frequency. All the tuned networks are being forced to 48

respond to the transmitted frequency rather than operate in some isolated and random natural response mode such as that determined by local oscillator net works or shock-excited filters "ringing" at slightly different frequencies.

A direct and highly desirable effect of the forced response mode of operation is that the alignment (tuning) of the timing filters is not ex tremely critical. What is important is;

1. that the improvement between S/N input and S/N output of the filter is maintained above that value which produces an overall convergent S/N ratio for the whole system, Section 4.3,

2. that in any particular regenerator the S/N ratio is such that the average number of zero crossings of the carrier frequency is maintained con stant, Section 4.4.

The improvement in S/N ratio is a direct consequence of the spectral area contained within the timing filter response curve. Any normal mistuning will not greatly change the spectral area, thus the noise reduction, r, is changed only marginally. More importantly, the carrier frequency component will be reduced by an amount proportional to the degree of mistuning leading to a reduction in S/N ratio at the output of the mistuned filter. This could lead to loss of synchronisation if the S/N ratio becomes too low.

The action of the limiter following the filter will be that, pro vided the limiting level is still exceeded, the carrier frequency output power will remain constant over a wide frequency range. However, the same decreased S/N ratio will be transmitted through the limiter, see Section 4.5, to the next regenerator. The effect of the mistuning is to add "noise" to the system because it modifies the reduction ratio r and hence the S/N ratio output, Section 4.2. This additional "noise" is reduced at the same rate as the noise which normally enters the next timing filter input. 49

Provided the mistuning is not severe the overall S/N ratio will still converge and remain above that value required 1to guarantee synchronisation.

The next important point to consider is the choice of the value for the integer ratio M relationship between the carrier frequency f and the digital data frequency f^.

f= f (3.1) d c M

where M = positive integer and is equal to the number of carrier cycles

per data pulse.

Two factors influence the minimum value of M.

1. M must be large enough to prevent the overlapping of the frequency components of the baseband signal and the lower sidebands of the modulated signal. This condition is satisfied if the carrier frequency is greater than twice the highest frequency transmitted in the baseband signal.

2. The smaller the value of M, say 3 carrier cycles per pulse as com pared to 9 carrier cycles per pulse of the same carrier frequency, the wider the communication channel bandwidth must be in order to allow transmission of a recognisable waveform.

It is always desirable to minimise the value of M. This leads to maximum utilization of the available channel capacity and also simplifies the task of the filter network in being able to extract the carrier frequency from the input AM spectrum.

With the value of M selected, the output of the filter-limiter net work is passed through a M times frequency divider and thereby provides a timing wave at the correct data frequency but not necessarily of the correct 50

A m p litu d e FIGURE Timing N.B.

3.2. Impulse Baseband

Wave Carrier Timing frequency Sketch

Wave illustrating Wave

dividing

Impulse filtered

(tM) at the

before

the the technique

data

Carrier frequency

frequency Carrier

Impulse

Wave providing division.

Impulse Wave f^

the

f^ 51

phase relationship, Figure 3.2. Before presenting the technique employed to correct the timing wave phase relationship the amplitude detection process will be discussed.

Detection processes are in their own right a major problem in communication systems. . Because the main emphasis of this investigation has been placed on timing characteristics, no detailed analysis of the detection processes has been conducted. A simple envelope detector employing a non linear element and low pass filter has been employed.

The non-return to zero (NRZ) data format, although it minimises bandwidth consumption, suffers from the defect that a pattern dependent d.c. level component is transmitted. Rather than compensate for this effect, by using a d.c. level restoration derived from the input carrier level (equivalent to an A.G.C. system except that the signal gain is not being adjusted), it was decided to investigate what problems would be associated with the simple en velope detector and d.c. level shift.

In order to minimise the error rate the timing wave should sample for the presence of a pulse at the center of the input pulse. Even though the timing wave frequency may be correct, if its phase relationship with the data waveform is not correct additional errors will be made. To correct for this phase error a Self-Adaptive Sample Time Network, SASTN, has been included in the regenerator. This network measures the time interval between the leading edge of the detected pulse and shifts the timing wave until (ideally) a half- a-pulse period separation exists. Because the carrier frequency is M times the data frequency a time quantisation is derivable from the carrier frequency which will be much finer than that available from the relatively coarse data envelope. In actual fact the carrier frequency has segmented each data pulse into M parts. The SASTN utilizes these time segments to adjust the centering 52 NORMAL OPERATION

Divider input, f

t 2 output

t 4 output

t 8 output, TIMING PULSES

TIMING PULSE EARLY

Divider input, inhibit 1 cycle f

•f 2 output, delayed by 1 segment

t 4 output, it it

t 8 output, " " TIMING PULSE DELAYED

TIMING PULSE LATE ■■ Delayed carrier segment BBBBBHHI t 2 output plus delayed segment

t 4 output, decreased by 1 segment

t 8 output, " " " TIMING PULSE SPED-UP

FIGUPE 3.3 Photographs illustrating the 3 separate SASTN operations H = 0.2 ySec/div, V = 2 Volts/div. 53

of the timing wave. Essentially the SASTN modifies the divider frequency count rate by adding or subtracting carrier frequency segments to the timing wave un til the desired centering is achieved. Figure 3.3 shows the effect of adding and subtracting the time segments on the frequency divider. Full details of actual circuit and explanation of operation given in Appendix 2 and Subsection

3.2.3.

Regeneration of pulses will occur every time that both the timing wave

is applied (samples) and the output of the envelope detector exceeds some pre determined threshold level. With a band-limited data pulse waveform the noise level residing on both the leading and trailing edges will cause the threshold level to be exceeded or depressed in a random manner in the vicinity of the threshold level of the noise-free waveform. If the timing samples are applied

in these critical noise prone regions errors will be made. The susceptibility of the amplitude regeneration threshold network to make errors can be modified by the non-linearities in the regenerator. During the OFF periods the noise can be compressed by the non-linear envelope detector whilst during the ON periods the noise can also be compressed by any overloading in stages preceding the thres hold network or even in the threshold network itself. The ON noise compression can be gainfully utilized, provided the saturation is not excessive and lead to pulse widening associated with charge storage effects in the active elements

(transistors). With the ON level compressing the noise during the higher level parts of the pulse, the centering of the timing sample is not extremely critical.

All that is required is that the timing samples be applied somewhere near the equivalent center-of-gravity of the pulse but nowhere near the leading or trail ing edges. However, the narrower the bandwidth the slower the pulse rise and fall times and hence the closer the timing samples must be made to the pulse center. The relaxation on the position of the timing wave with respect to the pulse center allows the requirements on the spectral purity of the timing wave 54

AM Input

Data Envelope Channel Envelope Bi-s table Amplifier Detector

B.R F. Trigger Pulses

Amplitude SASTN Limiter

Low-0 fi. P F.

Filtered Retimed Modulator Carrier Envelope

Channel Filter

Regenerated AM Output

FIGURE 3.4 Functional Block Diagram of Complete Regenerator 55

to also be relaxed. Provided the time dispersion associated with the impure timing wave is bounded no excessive build-up in timing jitter will occur.

The detected envelope and the "centered" timing wave must now be com bined to produce the retimed baseband signal. This is readily achieved by using a triggered bistable multivibrator (JK Flip-Flop). The detected envelope is applied to the set input with the timing wave applied to the clock input.

Some logic control is necessary on the reset or clear input to ensure that the correct transfer of data pattern results.

The retimed envelope and the filtered carrier frequency are then com bined via a modulator and thus the desired regenerated AM wave is obtained.

It is most important that during the OFF periods a minimum level of the carrier frequency be allowed to leak-out through the output. This may be achieved by using some intrinsic property of the modulating device which normally prevents 100 percent modulation being obtained or an appropriate net work, with adjustable signal attenuation and phase control, can be connected so as to by-pass the modulator and insert the correct amplitude and phase for the carrier leak in the output. The former method would allow use of devices for modulators which are sometimes rejected, particularly in the microwave region, because of a poor ON/OFF ratio. The result of other non-ideal modulator performance characteristics, such as a modulating level dependent phase shift will be discussed in Section 5.7.

The final network of the regenerator is the channel filter, suitably designed to match the bandwidth requirements and impedance levels of the system.

Dependent upon whether the modulation has occurred in a low or high level sig nal stage, a final power amplifier may be combined with the channel filter.

The complete block diagram of the regenerator is presented in Figure 3.4. A careful study of the delay paths through the regenerator will reveal that even 56

though the data envelope has been stripped from the input carrier wave, re timed and then recombined with a section of the cleaned-up carrier wave different from what it had at the input, the desired coherent phase separation between various discrete sections of the various patterns has been maintained by the phase lock existing between the cleaned-up carrier frequency, retiming wave and modulated output.

3.2.3 System Synchronisation

In the previous section a description of the functions necessary to incorporate in the individual regenerators were discussed. An essential function of being able to position the timing wave samples near the center of the input pulse envelope was also presented. The particular network which pro duced the centering action for each individual regenerator is also used to enable synchronisation of the entire cascaded network of regenerators.

When the total system is initially switched on and the transmitter frequency is allowed to leak out through the network, after the appropriate propagation delays, all the timing filter networks will be locked to the carrier frequency and generating internal trigger pulses at the proposed data trans mission speed. Because of the random start-up action in each regenerator, and the various propagation delays between regenerators, there is no guarantee that when a pulse is transmitted out through the system that all the trigger pulses will be located near the center of that pulse when it passes through any particular regenerator. To ensure that the correct sampling occurs, a special synchronisation code group can be transmitted. This code group should be employed every time that the system, or any section of it, is switched on.

Unless there is some storage routine available at the transmitter to ensure that any new message sequence, whether it be of several seconds or several hours duration, has a commensurable starting time with the previous message Timing wave pulses

Early and Late gate HI pulses

FIGURE 3.5 Multiple exposure indicating the action of the SASTN keeping the timing wave pulse "centrally located" between the Early and Late gate pulses which have been derived from the Data Pulse Envelope H = 0.2 ySec/div V = 2 Volts/div for Early, Late and Timing Wave pulse = 0.5 Volts/div for Data Pulse Envelope 58

sequence, the special code group would have to be used as a header, or lead-in, for the new message sequence to ensure synchronisation of all regenerators in the system. Storage routines may be needed at the sender's end to enable the header to be synchronised with the information code before transmission occurs.

It should be recalled that the higher frequency carrier has segmented the basic pulse interval into M parts. Those segments located immediately after the beginning of the pulse or just before the end of the pulse form regions where the sampling should not occur. As an example these regions each may con sume 30 percerft of the total pulse interval with the 40 percent central region remaining being used as the accepted portion where the sampling should be applied. If an M value of 8 is being employed each segment is 12.5 percent of the total pulse interval. Normally the Self-Adaptive Sample Time Network

(SASTN) is capable of only up-dating the divider counter routine by one segment at a time. For the example given a maximum of 3 segments would be used to pro vide the worst case condition required to shift the trigger pulse to an accepted part of the pulse central region. The central region contains 3 allowed seg mental positions and allows time dispersion about these positions without pro ducing any segmental shifts. If the SASTN up-dates the divider counter by more than one segment at a time it is essential that enough segments are allocated to the central region to prevent the trigger pulse from overstepping the total central region in one hop and then start oscillating backwards and forwards across it. Too many segments could be allocated and the trigger pulse might never increment to the center. As mentioned above, 3 segments for the center, out of a possible 8 was selected as a suitable compromise for the investigation, see Figure 3.5. For a correction to be applied a transition from OFF to ON must occur. If a 2 segment shift is required, 2 separate OFF to ON transitions must occur in the code group consuming, for the NRZ pattern employed in this K investigation, 4 pulse bit intervals in the form 0101 or 1010. 59

To determine the length of the code group which will synchronise the entire system, the worst case condition is assumed i.e. the trigger pulse is coincident with either the leading or trailing edge of the data pulses in each regenerator. Initially the maximum number of segments in either the early or late section is noted, S^. Because 2 bits of data, 0 and a 1, are required for the NRZ data format to produce a segment shift of the timing wave,

2 Sw bits of data are necessary to guarantee synchronisation for each regenerator. Multiplying the above result by the number of regenerators, , gives the maximum length of the synchronisation code group, C^.

Hence C^, = 2 SM N , bits (3.2) G Mr

The pattern transmitted is 101010 ......

The regenerators are synchronised sequentially beginning at the transmitter end. The first 2 bits synchronise the first regenerator which will then remain synchronised for all the remaining bits. The first 2 Sw bits will now be distorted due to the addition or subtraction of segments to the timing wave and resultant output pattern of the first regenerator. Thus this first 2 subgroup cannot be relied upon to give accurate synchronisation for M any further regenerators. The 2 subgroup will pass through the pre viously. synchronised N-l regenerators and guarantee synchronisation of the N**1 regenerator.

If the synchronisation code group is omitted the system will self- synchronise after a maximum of C /2 transitions of data but a high initial O error rate will occur.

In the event of noise causing an unsynchronised pulse to be inserted within the normal data stream, it can cause a perturbation of the timing wave in all the subsequent regenerators. However, the very next following data 60

transition will step-back all those regenerators which may have had their timing wave segmented out into the undesired early-late regions of the pulse interval. The synchronisation is thus self-healing. As is the case with any digital data transmission systems, a high density of noise spikes will lead to errors; there is no claim to be made that this system is insensitive to noise.

Since it is not necessary to continually add code into the normal data stream to maintain synchronism, the coherent carrier technique offers maximum utilization of the available channel capacity for the transmission of information.

3.3 The Simulation Network

Since the technique being investigated could involve a large number of regenerators, the economics of time and cost are such that a simulation of a "real" communication network employing this self-timed technique has been implemented. The capability of the simulated system to test thoroughly the proposed principle will be discussed and where applicable references are given to the section of the thesis containing the specific analysis which is necessary to validate any comments made.

3.3.1 Design Criterion

The real system has to be simulated in such a manner that whatever performance statistics are derived from the simulation system, they must con tain sufficient information to enable the performance of any real system, using the same type of regenerators, to be effectively predicted.

For successful operation of the simulated network, it may be found necessary to introduce operational restraints which will not exist in the real X) u) 73 ro XI i « i -H X

3" 1 r-*- O "I ii “ fO 73 fP 3 a rt> 3 za 3 IQ

;

it —

II r-+- O Q_ o Q X C

II n> o Z in 11' CL Q- Z

FIGURE 3.6 Simplified block diagram for cascaded network of regenerators illustrating

the continuous analog carrier frequency filter and modulator leak path. 61 62

system. The type of restraint will depend upon which type of simulation scheme is found desirable to implement.

The loop method of simulation was chosen because it does offer the opportunity of examining some aspects of the characteristics of the commun ication channel as well as the performance of the regenerative repeaters themselves. In order to couple the transmission line characteristics with the regenerator characteristics, but still be able to separate the deficiencies which may exist in either, two repeaters and two delay lines were connected in series to form the basic loop. The delay lines were made with different de lay times (length) as a step towards randomising the system operation and periodic result sampling procedure which exists with a simulation network con taining only one regenerator and one delay line.

The major difference between the simulation loop and the real network is that with the former a feedback path has been established which would not normally exist in a real system. However, the possibility of establishing feedback loops in an interconnected communication system mesh would exist and thus any results obtained from the simulation could be readily applied to such configurations.

It should be observed that the coherent carrier self-timed regener ative system offers a combination of both analog and digital techniques functioning simultaneously.over the one channel. The digital aspect is in volved with the amplitude response and is effective only when the input amplitude exceeds a predetermined threshold. The analog section is concerned with the extraction of the carrier frequency at a controlled level of spectral purity. This network is effective at all times and provides a signal path from the input to the output of every regenerator. Figure 3.6 is a simplified illustration of the carrier leak path through a cascaded network of regenerators. 63

It is important to note that the transmitter is the carrier frequency source and is applied only to the first regenerator; all other regenerators obtain the carrier frequency from the previous regenerator and operate in a forced response mode to that frequency.

For satisfactory operation of the simulation loop a restraint must be imposed upon the choice of total loop delay T^, which includes both the re generator and the transmission line delay, the carrier frequency f and the data speed f . This will be referred to as the COMMENSURABILITY CRITERION and states simply,

The total loop delay T^ must be an EXACT integer multiple of the reciprocal of the data speed f^.

seconds (3.3) td = f

where P = positive integer.

The choice of the P value is detailed in Section 3.3.2.

The implication of the above criterion is that when the total delay around the loop is being adjusted, the phase of carrier frequency being in serted into any point on the loop must be the same as the phase of that fre quency after it has travelled around the loop. If this condition is not satisfied the phase discontinuity will produce a build-up of timing errors which will prevent stable operation from being obtained. With only one regener ator and one delay line, and remembering that the real system operates in a forced response mode, with the transmitter frequency being inserted at the in put to the regenerator the situation could exist that during the passage of the carrier frequency around the loop some pattern dependent phenomena had perturbed the carrier pulse. The signal at the regenerator input would then 64

Perturbed Envelope Perturbed Bi-stable

AM Input Detector Envelope

Filt e r - Limiter SA ST N

Timing Wave Jitter Perturbed" f,

ModuLator Link

Retimed Envelope Modulator with Jitter Carrier inserted with same S/N ratio as in unmodified regenerator Regenerated AM Output with Jitter

FIGURE 3.7 Modified Regenerator indicating technique of carrier frequency

insertion for the simulation network forced response mode of operation.

Propagation of jitter not prevented by the pseudo-carrier frequency

insertion. 65

contain two components, see Figure 3.8(a), one being the original "clean"

carrier frequency being used to drive the system whilst the other would be the

circulated perturbed wave. To resolve this undesirable situation, but still

maintain the forced mode of operation, the circuit of the regenerator was

modified to allow the "clean" carrier frequency to be inserted at the input to

the modulator. The normal connection between the output of the filter-limiter

and the modulator was open-circuited. This technique open-circuited the loop

to the carrier frequency but still allowed any perturbation of the carrier

frequency, after it had passed around the loop, to be passed through the timing

network and envelope detection section and re-appear on the inserted carrier

during the modulation process. Thus if any jitter is being generated by any

part of the network it will be allowed to circulate and accumulate without the

inserted carrier masking the effect. Figure 3.7 is provided to illustrate the

necessary modifications which still allow transfer of any perturbations.

Although the preceding comments were introduced and explained using a single regenerator with a single delay line, the same technique of carrier

frequency insertion, and thus open-circuiting the analog feedback path, can be applied to achieve the same result if only one regenerator is modified in a simulation network containing many regenerators and many delay lines. As mentioned previously, it is desirable to be able to separate the effects of the regenerator from the effects of transmission medium and also to break-up the periodic structure which exists with only one regenerator and line. The minimum configuration which will satisfy this criterion is that consisting of two "identical" regenerators and two unequal length delay lines. The particular configuration implemented for the investigation had one normal regenerator and one regenerator modified for the carrier frequency insertion together with the unequal length delay lines. To enable a clearer interpretation for the be haviour of the signal as it circulates around the simulation loop, both simple inserted at modulator 68

loop sketches and ’’unfolded" signal flow illustrations are given for each of

the four examples chosen, Figures 3.8(a) - (d) inclusive. The "unfolded" sig nal flow illustrations enable a direct comparison to be made with the real

system as illustrated in Figure 3.6 and show the open-circuiting of the feed back path formed by the carrier frequency insertion into the simulation loop.

Experiments and analysis will be presented in Sections 5.2 and 4.7 respectively

to explain the behaviour of the simulated systems with the modulator-bandpass

filter link open and closed, with the carrier frequency either present or ab

sent to give both the forced and natural response modes of operation.

The commensurability criterion prevents the simulator containing net works which would simulate such things as time variable transmission delays and doppler frequency shifts. However, it is possible to test some aspects of the transmission medium. At each normal regenerator output the carrier frequency

and data envelope are phase-locked. A mechanism MAY be present in the trans mission medium which could destroy this coherence; this is the frequency de pendent phase characteristic dispersion which can cause significant differences between phase and group velocities. If this effect is present it can be con- | trolled by decreasing the spacing between regenerators. However, it is im portant to note that any minor velocity difference existing in the transmission medium between regenerators would not accumulate because each regenerator guarantees an updated coherent output even though the input envelope may have

"slipped" with respect to the phase established at the previous regenerator.

The "slippage" is taken up by the retiming process involved in triggering the bistable multivibrator that follows the envelope detector.

3.3.2 Control and Test Functions

A feature of the coherent carrier technique is that it is applicable to an extremely wide frequency range, the upper frequency limit being governed 69

by the technological development of suitable components to perform the necessary functions. In particular the commercially available digital devices, at present, are frequency limited to about 300 Million bits per second (MECL-3).

The features which governed the choice of operating frequency for the sim ulation study was the availability within the research department of certain items of test equipment and circuit components which were held as standard store's items.

A carrier frequency of 8 Megahertz together with a NRZ binary data speed of 1 Million bits per second was nominally selected (M = 8). A test word length of 8 bits gave a minimum loop delay of 8 microseconds before the end of a word would be overlapped by its beginning. To avoid "end effects" of the circulating word the loop delay was increased to 13 microseconds which, with approximately half a microsecond delay per regenerator, yields an equiv alent two repeater spacing of about one and a half miles in coaxial cable.

This choice of loop delay versus word length controls the choice of the P in teger value mentioned in Section 3.3.1, Equation (3.3).

The Table of Contents lists for Chapter 5 all the tests considered essential to assess the performance of the coherent carrier self-timed digital regenerative technique. In order to conduct these tests special test and con trol equipment had to be designed and constructed to provide the necessary variation and measurement of the parameters under investigation. The single most important test involves determination of the presence of any pattern de pendent systematic timing jitter accumulation. The technique employed involves regenerating specific data patterns (words) for various number of times and measuring any phase variation of the timing wave that may occur. This is 7 similar to the method used by Byrne, et al , and exactly the same patterns were chosen so that a direct comparison could be made between their 70

regeneration method and the coherent carrier technique.

Error detection and counting networks were also constructed. Of particular importance is that the bit-by-bit "exclusive or" network did not use the original word first transmitted into the loop as the basis for error detection. This word is stored in the word generator and provided no errors are made in the regenerators it is the same as the word circulating around the loop. If, however, the regenerator does make, say, one mistake and adds an additional 1 to the circulating word, any comparisons between it and the un changed word generator pattern would indicate that every time the word circulated the loop an error has been made, where in actual fact only one mis take had been made. Therefore the error comparison must be made within the loop. In this investigation the retimed baseband signals from both regenerators are compared, after one has been suitably delayed to match the transmission delay between the regenerators. If only one regenerator and delay line is em ployed then the input, with a matching delay, and the output could be compared so as to prevent multiple counting of the one error. If dispersion on the delay line is causing a smearing of the pulses then one of the sample points would have to be moved away from the regenerator and varied along the line to test the effect.

One important advantage of performing the study in the frequency region selected is that a real time display of the words circulating the loop can be made. This is achieved by phase-locking all the test equipment to the trigger pulses being generated by the regenerators. If the oscilloscope has a variable delayed time base network it is then possible with the periodic repetition of the same static word pattern being fed to the loop and circulating it, to vary the time base delay and scan across the oscilloscope display to select any particular regeneration value and examine in detail any 71

perturbations.

Details of the actual test and control circuits are provided in

Appendix 3.

3.4 Conclusions

A deliberate attempt was made throughout this chapter to avoid any- detailed theoretical analysis or to present too many specific details about the particular implementation employed for the investigation. This was done to provide a relatively free-flowing development of the general concepts in volved with the coherent carrier technique. This should allow the various sections of the work to be viewed in proper perspective when both the analytic and experimental results are presented in the next two chapters.

The main considerations have been concentrated upon the design, implementation and testing of a technique for retiming digital regenerative repeaters which does not use the baseband information as the source for the self-timing signal. 72

CHAPTER 4

THEORETICAL CONSIDERATIONS

4.1 Introduction

The purpose of this chapter is to allow the development and presentation of suitable theoretical material which can be used as the com parison basis for the system timing and synchronisation performance. The theoretical analysis of an optimum amplitude characteristic for the system

is beyond the intended scope of the investigation and is thus not presented.

Formulae will be derived, discussions about their particular sig nificance given, together with the results computed from them. Reference has also been made to information published by other research workers, see Sections

4.4, 4.5 and 4.6, where their results have been found to be directly applicable to specific sections of the analysis.

An exhaustive theoretical treatment of the many concepts involved is not given. A selection has been made of those concepts which have been found to be most important. Their analysis is considered necessary in order to pro vide sufficient information for accurate conclusions and predictions to be made about the feasibility of employing the coherent carrier technique as a suitable means of timing and synchronising a regenerative digital communication system.

Of main interest is the spectral purity of the extracted carrier frequency wave which is used for the retiming in each regenerator. The re lationship between the timing filter bandwidth and input signal-to-noise ratio in being able to produce a specific signal-to-noise ratio at the output of each regenerator is examined closely, together with any noise build-up which may occur because of the cascaded connection cof the regenerators. In order to 73

maintain synchronism of the entire system the signal-to-noise ratio at each timing filter output must be such that the average number of zero crossings of the extracted carrier frequency wave remains constant. The lack of spectral purity, because of the noise passing through the finite bandwidth of the filter leads to a dispersion of the desired frequency impulse shape and results in perturbations of the zero crossing intervals. In actual fact occasional break through of the zero crossing interval can take place but provided the variance of the average number of zero crossings remains small, with no build-ups in close proximity, then no break-down in synchronisation will occur leading to high error rates.

For the purpose of the analysis, the timing filter-limiter network will be subdivided; major emphasis being placed upon the characteristics of the first bandpass filter (high Q). Subsequent work will then reveal what allowances will have to be made for the separate effects of the amplitude limiter as well as any mistuning that may be present. To conclude this chapter the re sult of the feedback path in the simulation loop will be presented and compared with how a real system should perform.

It was felt that no advantage would be gained by starting the analysis with the time variable random processes involved, then performing the con volution and auto-correlation integrations together with Fourier transforms to obtain the power spectrum. To simplify the analysis, the power spectral densities will be assumed to be known and the effect of the regeneration technique upon the power spectrum is then developed.

4.2 Timing Filter Characteristics

For the regeneration technique implemented in this investigation, the transfer characteristic H(juj) of the first filter (High Q) in each 74

regenerator’s timing filter network is the critical function that has to be analysed in order to determine the S/N ratio and consequent spectral purity of the carrier frequency wave being transmitted through the network.

Initially all the filters are assumed to be tuned to the carrier

frequency. Any sidebands of the input AM spectrum that pass through the

finite bandwidth of the first filter will be symmetrical in respect to both

amplitude and phase lead-lag such that no angular modulation will result;

amplitude variations can exist and will be removed by the limiter immediately

following the first filter. However, the random amplitude and phase character

istic of any noise added to the system will perturb the carrier frequency zero

crossing intervals and it is this source of disturbance that is of major

interest.

As far as the carrier frequency is concerned each regenerator may be replaced by:

1. a high Q bandpass filter,

2. a noise source at the input,

3. a limiter to provide a constant amplitude output,

4. an amplifier to compensate for attenuation between regenerators

4.2.1 Filter Transfer Function

The bandpass filter consists of a parallel LCR network and its

impedance transfer relationship is,

H(s) V0UT<-S> Vs)

(4.1) 1 ♦ sCR + R sL 75

By substituting s = ju) and then Q = oo^ RC with u)q LC = 1,

Equation (4.1) becomes

H( joi) : (4.2) 1 + jQ (W__ Wo_) W U) o

For frequencies near this expression may be modified to,

H(joo) = (4.3) 1 + i 2Q (oo - to ) —- o 0) o

(4.4) 1 + j (w - CO ) o

where BT is the timing filter half bandwidth. To simplify further calculations the transfer function is normalised with respect to the center frequency w

A H( jco) (4.5) 1 +

The square of the magnitude then becomes,

I H( j a)) | 2 (4.6) 1 +

Since the analysis is concerned with power spectrums, the phase characteristic of the filter is not of great interest, but it is presented because it will be used for the analysis of the simulation loop system.

From Equation (4.1), with substitution of s = jto , G = 1_ and K = 1_. R L

H( jo)) JO) (4.7) (K - u)2C) + jo>G 76

When this equation is rationalised the phase versus frequency behaviour may be obtained as,

Tan = K - oo2C (4.8) o)G

4.2.2 Noise Reduction Ratio r

Because the timing filter bandwidth is less than the communication channel bandwidth, a reduction in the noise power passing through the timing network will occur.

r = total output noise power from filter total input noise power into filter

H( joj) 3>((d) dm N -B r (4.9) B re $>(u)) du) N -B

where B Half communication channel bandwidth

#M(w) Input noise spectral power density N 2

(4.10)

Substitution of Equations (4.6) and (4.10) into (4.9) yields 77

2 a 2B

FIGURE 4.1(a) Noise Spectral Power distribution of input and output of Timing Filte:

FIGURE 4.1(b) Noise reduction ratio r versus ratio of communication channel

bandwidth Bc and timing filter bandwidth BT 78

R1 R2 R3 Rn s

N N

n=$nM

S =82(

Rn=n^ Repeater

FIGURE 4.2 Block diagram of cascaded connection for the repeater network

indicating the signal S and additive noise N ports 79

B r c 2 1 a dto 2B c

-B c r

dto

Bm -1 B^ T tan C (4.11)

Figure 4.1 depicts the spectral distributions and provides a graph of the noise reduction r achieved for various ratios of communication and timing filter bandwidths.

4.3 Signal-to-Noise Analysis

Now that the amount by which the noise is reduced in each generator is known, the next step is to determine what the combined effect of the cas caded connection of all the regenerators, each with additive noise being fed into their input. Figure 4.2 illustrates the particular connection to be analysed. The essential feature is that the noise must not be allowed to pro gressively build-up along the repeater chain until the extracted carrier signal- to-noise ratio becomes too small and synchronisation is lost because the number of zero crossings of the extracted carrier frequency becomes random.

4.3.1 Cascaded Repeater System

The output power of the first repeater is 0^, 80

o = s |H(j«)r + rN (4.12)

With the filters aligned correctly u) = u) , thus the output carrier o c 2 frequency is not modified by the filter because |H(j)| = 1

s|H(ju> >r =s (4.13)

The output power of the second repeater is 0^,

02 = S + r2N + rN (4.14)

Similarly the output power of the n repeater is 0 ,

= S + N l r} (4.15) k=l

The second term on the RHS of equation (4.15) is a geometric progression and may be summed as follows;

Noise Sum to n terms = N r (1-r ) (4.16) 1-r

Noise Sum to infinity = N r (4.17) 1-r

4.3.2 Noise Convergency Criterion

Provided the noise reduction ratio r is less than unity expression

(4.16) and (4.17) converge. If r = ^, that is, a 3db reduction in noise, the limiting sum (4.17) produces a N output level which is the same as the first

N input level. If r < ^, then the limit noise level will be accordingly smaller.

Figure 4.3 depicts various r factors together with their limit sum value. For all v < h the filter output noise reaches at least 97 percent of its final value ^fter 5 regenerations and 99.9 percent after 10 regenerations. The smaller the r value the less the noise build-up. However, it must be guaranteed that S/N level at any regenerator filter output must be greater than that value Convergent Limit Sum Coefficient of Noise Term, Sum = 81 g FIGURE

4.3 ratio Noise

r term

Noise 0.05 per

limit regenerator

Reduction

0.1 10db sum

coefficient 0.2

versus

noise

reduction 82

necessary to provide synchronisation. The preceding calculations assumed that all the N levels were the same, and at some predetermined maximum value.- Thus any N level may be reduced without destroying the desired operation. However, no particular N level can be increased without causing serious local disturb ances. Although the latter condition would have little effect on the overall

N limit sum, if only one N source was increased, it must be realised that the total system synchronisation would be lost if any one regenerator loses its synchronisation. To use a reduction factor r = \ (3db improvement) the input maximum N level must satisfy the minimum S/N requirement, This could be readily achieved but the timing filter bandwidth would be so wide that a large percent age of the AM spectrum would be accepted by the timing network. Normally the timing filter bandwidth would be chosen so as to minimise the AM spectrum trans fer, and this choice would provide a r factor of sufficiently small value to guarantee convergence of the N noise power level.

4.4 Number of Zero Crossings of a Sine Wave plus Narrow Band Gaussian Noise

The spectral purity of the extracted carrier frequency is determined by the S/N ratio at the output of the timing filter network. To guarantee that the system remains in synchronism a minimum S/N output ratio must always be exceeded. In this section the results of other research workers who have ex amined the statistical properties of a sine wave plus narrow band gaussian noise are presented in order that may be determined. Because of the narrow band filtering action the output waveform will appear to be quasi-sinusoidal; its amplitude and phase can only change slightly during any cycle or adjacent cycles because of the inertia associated with the high Q resonant circuit.

(26—34) From the available sources of informationv which deal with (31) the zero crossing problem the work of J.S. Bendat was found to be most suitable. For the special problem of a single fixed sine wave coupled with 83

1.0009

1.0006

b = -rr^- = bandwidth ratio

b=1-25 21.0003

b=2-5

b =12 5 1.0000

Signal - to-Noise Ratio

FIGURE 4.4 Expected number of zero crossings N of a sine wave plus narrow band

gaussian noise versus (adapted from Bendat pp.388-389) bt 84

a strong assumption of ergodicity, which permits replacement of ensemble averages with time averages, the problem reduces to the examination of,

y(t) = Q sin (

where the noise n(t) is described by its normal probability density function,

2 2 -n /2c f (n) e n (4.19) a /2tT n

The mean value of the noise is assumed to be zero, whilst its variance is derivable from the power spectral density function $^(a>), where J yU) da) (4.20)

The associated auto correlation function of 1he noise, R(x) is,

R(t ) = COS 0) T (co) do) (4.21) N

The initial problem is to find a suitable expression for the ex pected number of zeroes of y(t) per second, denoted by N^. Using Bendat's

10.3 results suitably modified to allow the parameters of this system to be displayed, Figure 4.4 is presented. Unlike the situation which can exist for low pass filtering of a sine wave plus noise, the bandpass condition always results in an increase in the expected number of zero crossings Nq above that for the pure sine wave. The actual increase is a function of both the signal- to-noise ratio and the bandwidth of the filter network, as indicated in Figure

4.4. 85

FIGURE 4.5 The relationship between (S/N)^ and (S/'NK for both the

ideal limiter and linear amplifier for different (S/NK ratios

(Davenport35) 86

4.5 Signal-to-Noise Ratios in Band Pass Limiters

Because the filter network contains a limiter, which helps minimise

the amplitude variations in the output waveform, the results of an investigation

on the modification to the S/N ratio passing through a band pass limiter is

presented. These results are a summary of the paper MSignal-to-Noise Ratios

(35) in Band Pass Limiters” by W.B. Davenport Jr. It is necessary to know the

change in S/N ratio due to the limiting action so that any degradation can be

allowed for when determining the (S/N) ^ to be produced at the timing network

output by the first filter.

Essentially the paper deals with the correlation functions produced

by the signal S and noise N interacting individually and together to form SxS,

NxN and SxN products. Their Fourier transforms are performed with specific

transfer characteristics for the limiter. Davenport's results are plotted in

Figure 4.5. His general conclusion is that the output S/N ratio is essentially

directly proportional to the input S/N ratio for all values of the latter. This type of behaviour is due primarily to the band pass filtering of the system out put in the immediate vicinity of the input frequency. This result differs from

the square law S/N relationship which exists for low pass filtering of non

linear detectors which gives rise to the small signal suppression effect

(page 267 reference (28)).

From Figure 4.5 it can be seen that even though the output S/N ratio

is essentially directly proportional to the input S/N ratio it is level de pendent upon the input S/N ratio. For (S/NK > -4db the (S/N)q is improved, up to a maximum of 3db with the ideal limiter, otherwise it is degraded by approximately ldb if (S/NK < -4db. Thus, dependent upon the actual S/N level out of the first filter an increase or decrease of the noise reduction ratio will have to be made so as to incorporate the limiter effect. 87

4.6 Mistuning

The presence of the limiter situated between the first (high Q) filter and the second (low Q) filter complicates the effect of mistuning.

Because the first filter has the highest Q any mistuning of it will produce the greatest effect on the overall timing filter characteristics. As dis cussed in Section 3.2.2 the mistuning will produce a reduced S/N ratio which can be accounted for by considering that the signal S level remains constant, due to the limiter, but additional "noise’1 has been added. This "noise” in creases the numerical value of the reduction ratio r, thus the final limit sum for the noise term will be modified by the allowed amount of mistuning per regenerator.

As a specific example, the limiting level could be set at 3db below the tuned u) = u)^ level. For this setting the input carrier frequency could be altered from oo - Bm to w + B_ and even though the first filter output o I o I would follow the rise and fall due to the deliberate mistuning, the amplitude limiter would present to the second filter a constant amplitude S signal. The final output S would then vary according to the response shape of the lower Q filter.

The noise level N would remain largely unaffected because, as dis cussed previously in Section 3.2.2,the area under the filter response curve would not be altered appreciably due to the mistuning. The change in input

S/N ratio could also produce a small change in the output S/N ratio of the limiter, the amount being calculated from Figure 4.5.

4.6.1 Modified noise reduction ratio m

The modified noise reduction ratio m will be shown to incorporate both the effect of mistuning as well as the normal noise reduction. To simplify 88 -fa cto r q-factor FIGURE FIGURE

4.6 4.7

q-factor ^ q-factor

increase increase Mistuninq

of versus

noise mistuning

reduction % for

due different

to Q=100

mistuning,

Q-factors

w co

the calculations, the worst case condition with all regenerators mistuned by the same amount is assumed. It is not necessary that the mistuning be always to the same side of resonance with respect to the carrier frequency.

The signal S loss 1 due to mistuning is readily calculated from

IH(jw)|2 .

(4.22)

1 + / 0) - 00 v o c

Instead of the first filter output signal and noise being represented as,

0 = lSt rN (4.23)

it can be represented, by allowing for the limiter action ss,

0 = S + mN (4.24)

where m = r_ T

- qr (4.25)

"th The output of the n regenerator,

n , 0 = S + N £ (qr) (4.26) n k=l

As before qr ^ ^ to guarantee that the final output noise is not greater than the initial input noise level. Figures 4.6 and 4.7 are presented to show the variation in q as a function to - w as well as the percentage mistune co B T combined with various Q factors for the timing network's first filter.

For the normal stability and accuracy of tuning of the filter network the modified noise reduction factor m will differ little from the initial r value. Thus mistuning is not envisaged as a major problem for the noise 90

convergency control.

4.6.2 Frequency Spectra

A principal advantage of the coherent carrier technique is that the

originating transmitter's carrier frequency forces all the timing filter net

works to respond to that particular frequency. The regenerators do not con

tain individual oscillators, which require frequency correction and/or tracking

feedback networks, nor do they contain filters which are allowed to "ring" at

their own, but all slightly different, natural frequencies. However, it is

also unlikely that all the timing filters for this new technique will be tuned

precisely to the carrier frequency. The passage of an AM spectrum through a

mistuned filter produces both amplitude and phase distortion of all the side

band frequencies with respect to the carrier frequency. These sideband fre

quencies can be either the unwanted modulation terms which have not been

rejected or the noise components which pass through the finite bandwidth of

the filters.

Several research workers have published results upon the effect of

mistuned timing filters employed in self timed digital regenerative repeaters, 9 Sections 2.2-2.3.In particular a paper published in the Bell System Technical

Journal by J.M. Manley is of direct interest to this investigation because this

analysis can be readily extended to cover the effect ofnistuning for this

technique. In his Sections 2.2 - 2.5 the response of a mistuned parallel LCR

circuit is analysed by considering one representative modulation term which

contains only two sidebands from a quantised noise spectrum. He develops a

series of equations which express the phase variation produced by the asym

metric sidebands at the mistuned filter output. His next step was to consider

how this phase modulation accumulated in a chain of repeaters when the same

amount and kind of phase modulation was generated at each repeater in the 91

FIGURE 4.8 Calculated total timing noise power caused by tank circuit

mistuning as a function of the number of repeaters in a chain

(Manley9 p.565) 92

chain. Manley's Figure 15, which is reproduced as Figure 4.8 shows the calculated timing noise caused by tank circuit mistuning as a function of the number of like repeaters in a chain. The mean square jitter power reaches its peak value after 4 regenerations and then levels out to become essential constant and independent of the number of regenerations. This effect is due to the spectral distribution of the timing jitter (phase modulation) having zero energy at zero frequency. Manley shows that this absence of low fre quency energy is totally different to the energy distribution associated with the pattern dependent amplitude-to-phase process of an offset trigger in the shock excited filter networks.

The significance of Manley's work in relation to this investigation is that whatever results he obtained for the mistuned baseband spectra centered around the pulse repetition frequency will be applicable to the sideband spectra of the amplitude modulated wave used in the coherent carrier technique of self timing. Thus the mistuning equivalence will be zero energy at the carrier frequency. Even though the mistuning does produce a finite amount of phase modulation it remains bounded. For the worst case, with all networks mistuned the same way, the jitter power will never exceed the amount produced after 4 regenerations. Thus mistuning is not envisaged as a major source of timing jitter because there is no build-up of energy in that region of the spectrum which will cause a significant change in the spectral purity of the carrier frequency.

4.7 Simulation Loop Transfer Function

From the results previously obtained in Section 4.2 the transfer function of each regenerator, as far as the carrier frequency is concerned, may be approximately represented by Equation (4.1). The analysis in this section is concerned directly with the effect of the closed loop employed in 93

*4 V„(s)

FIGURE 4. Functional sketch of regenerator without feedback loop 94

the simulation network upon the characteristics of the carrier frequency which is passing through each regenerator. Although the actual simulation network loop contained two regenerators and two lines, this analysis will be performed for one regenerator with one line providing the feedback path due to the closed loop configuration, see Figure 3.8. These results can be readily

extended to any closed loop configuration with any number of regenerators

connected in cascade. To include them at this stage of the analysis would only

serve to confuse the interpretation of the basic mechanism of operation and

behaviour of this particular type of simulation network.

Figure 4.9 shows the simplified sketch of a regenerator without the

feedback path formed by the simulation loop. The output voltage V^(s) from the regenerator is given by

Output Voltage = Input Voltage x Transducer Gain x Filter Transfer

Function

V (s) V. (s) x T x H(s) (4.27) o i g

V.(s) T s J- g (4.28) Cs^ + Gs + K

Without the loop feedback, the normal input voltage V^(s) consists of the modulated carrier spectrum plus noise.

With the loop feedback, the input voltage becomes the normal input voltage plus the output voltage delayed in time by the total loop delay time x.

V.(s) = V (s) + V (s) e"TS (4.29) l no

Substitution of Equation (4.29) into (4.28) yields,

V (s) T s V (s)e XST s V (s) n g + _o______g__ (4.30) o 2 2 Cs + Gs + K Cs + Gs + K 95

By rearranging (4.30)

V (s) T s o g V (s) Cs2 + (G - T e“TS)s + K (4.31) n g

-i03T Substitute s = ju) and e = cos cox - j sin got 9

V (joj) jo)T o J (4.32)

V (ju>) -Cco + (G - T (cos got - j sin GOT))jco + K n 6

u)T ______i______2 (goG - ojT cos got) - j(K - Cgo - T go sin got) (4.33) g g

The phase-frequency function of the simulation network is then obtained by

rationalising Equation (4.33) to obtain

2 Tan ip = K - Coo - T a) sin got (4.34) ______g______o)G - T gocos cox g

When this last expression is compared with the phase-frequency function of the 2 regenerator alone, i.e. Equation (4.8), Tan = K - Cco it can be observed ooG that the delay line feedback has produced a periodic phase function which will

give rise to a multimode oscillatory behaviour for the simulation network. The

frequency separation of the discrete mode lines will depend upon the loop delay

x. Because of the amplitude-frequency response of the filter network only

certain modes can satisfy the gain requirements for oscillation. These are the spectral lines located near to the center frequency of the filter. In

Section 5.2.1 the results of the experimental investigation into this multimode behaviour are presented.

The next point to consider about the behaviour of the simulation network is to what extent, if any, does the periodic circulation of any given 96

continuum

_n_ *• t aperiodic

(a) Single pulse applied to cascaded network of repeaters

CL discrete lines, £ spacing = o

0 1 A 1 a a a _nH K_a n_ n_t l«-T periodic

(b) Single pulse applied to simulation network

FIGURE 4.10 Simulation network loop-induced spectral line modification

to the frequency spectra presented to a repeater 97

test word pattern around the loop modify the frequency spectra of that pattern.

Figure 4.10(a) depicts the condition for a real communication system, operating at the baseband frequency, where an aperiodic pulse is applied to the network.

The frequency spectra of an aperiodic pulse is continuous. The timing filter of each regenerator for the baseband system must extract sufficient energy from the continuous spectra so as to provide the necessary timing wave. By looping this same pulse back through the same regenerator, the original aperiodic pulse is converted into a periodic pulse stream, the pulse separation being equal to the loop delay t. .This immediately modifies the original con tinuous spectrum and creates a line spectrum, with spectral line separation equal to the reciprocal of the loop delay t. In general, any pulse stream x(t) which is being coherently repeated at time intervals t, may be represented by

y(t) = £ x(t - n ) (4.35) n integer

Now the Fourier Transform of x(t) is X(f),

where X(f) x(t)e ^27Tftdt (4.36)

From reference (39), pages 122-129, the Fourier Transform of x(t-m) is

X(f)e j^fm. This result may be extended to provide the Fourier Transform of y(t). i.e. Y(f) = X(f) limit 1 £ e"^27TfnT (4.37) n -* 00 (2n + 1 )t

The geometric series contained in this last expression can be summed to infinity by,

Sum = e-j2"fnx 1 _ ej2irf(2n + 1 )t t4.38) 00 . i 2Trf t 1 - eJ

- sin 2irf(n + %)t (4.39) sin 2frf t_ 2 98

Thus Y(f) = X(f) 1 . Sun (4.40) limit n -► 00 (2n + 1)t 00

= 0 , except at f = n_ (4.41) T

Using L’Hopital’s Rule, the limiting value yields,

±n Y(f) = X(f) E 6(f - n) (4.42) T T

Figure 4.10(b) depicts the loop-induced spectral lines. Physically

it is impossible to have a perfect line spectrum because infinite precision would be required. This lack of repetition at precise intervals prevents the

interference pattern from producing perfect impulses, a smearing of the lines actually results. For all practical purposes the resultant lines which are

formed after 30 - 100 repetitions can be considered as impulses.

To satisfy the commensurability criterion, Section 3.3.1, the loop

time delay and the reciprocal of the timing frequency bear an integer ratio relationship. This centers the timing filter response curve exactly over one of the loop-induced spectral lines and produces a mode of operation for the regenerator with conditions that do not exist in a real system.

The magnitude of the loop-induced spectral line at the timing fre quency, for the baseband system, will depend upon,

i. the basic digital data pulse format (e.g. NRZ, Unipolar)

ii. the particular digital data word pattern (e.g. position and

number of 1 and 01s)

iii. any non-linear operations performed upon the envelope which

will change the frequency spectra (e.g. full wave rectification).

The technique of operating the simulation loop developed for this in vestigation, see Sections 5.2 and 3.3.1, enable the oscillatory behaviour near 99

the carrier frequency to be stopped. Further consideration of the operation of the regenerators reveal that although the carrier frequency path is ter minated, the baseband signal is still being periodically produced and sampled.

This cyclic regeneration at the baseband frequency of the same test word pattern produces the same discrete frequency spectrum as that already discussed for the pure baseband system, if the same loop delay exists. This modified spectrum then modulates the carrier and appears in the regenerated AM output as the upper and lower sideband spectrums.

Thus the loop method of simulation, by the very nature of its operation, will always produce a modification to the spectral energy distribution which would normally exist in any real communication system employing the same type of regenerators. This shaping of the spectral information which is presented to the timing filters is more serious for the simulation performance of a pure baseband system because it produces a frequency impulse, which normally does not exist, at the desired timing frequency. With the simulated coherent carrier system, the carrier frequency impulse function is normally present and so the loop-induced modification to the energy distribution does not introduce a re finement which normally does not exist at that frequency which is being used to produce the timing wave. The final spectral purity of the carrier wave will actually be determined by the amount of noise passing through the finite band width of the timing filters.

4.8 Conclusions

A set of analytic results has been obtained by extending a few simple concepts combined with the results of investigations conducted by other re searchers. A brief outline of their work has been given and this provides the reader with the opportunity of either accepting the simplified presentation or going to the reference cited so as to obtain the detailed and lengthy analysis 100

generally provided. This method of presentation was chosen to indicate that specific analysis had already been conducted in these areas which had been con sidered necessary to understand the operation of this system. Any attempt to produce an analysis on these separate points would have resulted in a large amount of duplication of previous efforts.

The analysis has revealed that the S/N ratio of the timing filter outputs can be readily controlled by relatively narrow bandwidth filters located in each regenerator. Also the total system noise level can be main tained below that level which could cause a loss in synchronisation because the number of zero crossings of the carrier frequency becomes random. This noise level convergency characteristic is essential so as to maintain the spectral purity of the carrier frequency above a level which has yet to be quantitatively defined. Although this critical level presents a difficult mathematical exercise to define, it can be readily provided by a simple ex perimental technique. The results are presented in the next chapter together with the effect of mistuning and the multi-mode oscillatory behaviour of the simulation network.

With the carrier frequency being present at all times it is a relatively simple task to provide a 'fconstanf amplitude carrier frequency sig nal at the filter-limiter output. This signal is pattern independent and the major problem associated with the amplitude-to-phase conversion mechanism associated with an offset trigger does not exist.

The main issue arising from these theoretical considerations, as far as the simulation network is concerned, is the discussion centered around the modification of the spectral energy distribution of the transmitted information from which the timing wave has to be extracted. In all the available literature

102

CHAPTER 5

EXPERIMENTAL RESULTS

5.1 Introduction

A feature of the research conducted in the digital regenerative re

peater field is that detailed analytic work is assisted where possible by

utilizing the results of experimental observations to establish the bounds of

operation for the particular system; the same technique has been applied in

this investigation.

To assist with the logical development of the prediction and per

formance analysis, the characteristics of the simulation network is presented

first instead of last, as in the other chapters. This is then followed by

those sections dealing with the repeater performance and they contain the

specific experimental results, together with an outline of the measurement

technique employed and a discussion of the significance of the results obtained.

Because of the dynamic behaviour of the system,photographs have been included

to best illustrate the points under discussion.

5.2 Simulation Network Characteristics

The reason for presenting the performance characteristics of the

simulation network first is to enable an assessment to be made on the capability

of the simulation network to test adequately those parameters which are necessary to predict the performance of a real communication system containing

the same type of regenerative repeaters.

5.2.1 Natural Response

The multimode oscillatory behaviour of the simulator was examined 103

Mode Number Natural Resonant Mode Frequency MHz and Loop Delay Loop Delay Loop Delay Frequency 12.60 ySecs 12.61 ySecs 12.62 ySecs Difference

1 8.1501 8.1430 8.1359

1-2 = 78.7 KHz 78.6 KHz 78.6 KHz

2 8.0714 8.0644 8.0573

2-3 = 79.2 KHz 79.1 KHz 78.9 KHz

3 7.9922 7.9853 7.9784

3-4 = 79.2 KHz 78.7 KHz J-8,4 KHz

4 7.9130 7.9066 7.9000

4-5 = 77.8 KHz 77.7 KHz 77.4 KHz

5 7.8352 7.8289 7.8226

TABLE 5.1 Variation in Natural Resonant Mode

Frequency and Separation for Different

Loop Delay Times. FIGURE 104 relative relative relative

5.1

(a)linear (c) (b) Effect either in

delay of producing combined non-linear

tuned side of

non-linear line

phase-frequency

of uneven

resonance filter phase-frequency

phase-frequency

spacing phase-frequency

network f o

. of

relative

characteristics

zero characteristics

phase

(mod.

of filter 2 tt )

separation

network 105

by removing the external transmitter oscillator drive to the regenerators and closing the circuit link between the timing filter and modulator. The link closure allowed a complete feedback path to be formed through the analog section of the network. With no noise or data patterns being deliberately fed into the loop it was observed that discrete frequency changes of the oscillatory frequency occurred. This mode line jumping was aggravated if any switching of adjacent power supplies or test equipment took place which would induce transient energy into the simulator. The stability of each oscillatory mode 4 was 3 parts in 10 , i.e. ± 150 Hz in 1 MHz, with each mode being separated by a frequency difference which was less than that calculated by using the loop delay time measured at the center frequency. This difference was due to the non-linear phase characteristics of the band pass filters around the center frequency. Figure 5.1 depicts the phase versus frequency characteristics of a simulation loop with linear phase characteristics for the delay line and the typical non-linear phase for a bandpass filter in a regenerator. These separate phase characteristics are combined to produce the crowding together of the spectral mode lines on either side of resonance.

To examine the behaviour of the modes the timing filter bandwidth was increased from 160 KHz (Q = 50, f = 8 MHz) to 800 KHz (Q = 10). The loop delay for this configuration was 12.6 microseconds which yielded a mode line g separation, for a linear phase system, of 10 = 79.4 KHz. Because of the 12.6 amplitude response characteristics of the filters, even though a bandwidth of

800 KHz could accommodate 11 mode lines above the 3 db response points, the spectral lines nearest to the resonant frequency would be preferentially selected. Table 5.1 lists the modes, their frequency separation, and the effect of increasing the delay line lengths by a known amount for the wider bandwidth configuration. It should be observed that the line length increase lowered the frequency of a particular mode and brought the modes closer 106

.Regenerators'

Combined

Frequency MHz t model mode2

FIGURE 5.2 Combination of the separately measured phase-frequency characteristics

for the 2 regenerators and 2 delay lines versus the measured natural

resonant mode frequencies of the simulation loop. 107

together. The amount of variation is a function of the operating point of the oscillatory frequency on the frequency-phase curve for the overall net work .

This latter effect was examined in greater detail by returning the filters to the normal operating narrow bandwidth (Q = 50). The number of modes dropped from 5 to 2. The frequency-phase characteristic of each delay line and regenerator was measured separately, with a Hewlett-Packard Vector

Voltmeter, and their combined characteristics over the frequency spectrum of interest is plotted in Figure 5.2. The actual decrease in mode frequency separation is readily traceable to the non-linear phase relationship in the regenerators and not in the delay lines for these particular frequencies.

With the natural response mode of operation an opportunity to run the network without the transmitter carrier frequency being undesirably forced into the loop on every second regeneration existed, see Figure 4.8. Tests were made to determine if data patterns could be supported by the network.

Although all the test patterns could be introduced into the loop for this par ticular configuration, the results obtained were not reproducable, mainly because of the random time during the test interval when a "line jump" or frequency change would occur which would change the data pattern. Also the low stability of the natural oscillatory frequencies would cause undesirable switching of the SASTN network with consequent timing jitter being recorded on the timing wave output and transmitted pattern. It was also noted that with sufficient noise energy being deliberately fed into each regenerator's in put, the filtering action of the timing networks produced a quasi-sinusoidal output which would force the regenerators to function in a mixed mode of operation depending upon the noise level. With low noise levels the character istic natural response multimode behaviour existed. With high noise levels 108

the line jumping could not be detected because of the masking effect of the noise in distributing the available energy over a band of frequencies controlled by the filter characteristics.

It is thus observed that the natural response configuration for the simulation network is not suitable for obtaining the reliable performance data required to perform the necessary analysis. What is important about the tests conducted for this configuration, with respect to the frequency-phase character istics, is that the network performed in a totally deterministic manner, an essential feature for any research tool.

5.2.2 Forced Response

(a) Alignment

With the transmitter oscillator frequency being fed into the loop, via the modulator of that particular regenerator which had the modulator link open-circuit, the simulation network is forced to respond to that particular frequency. The open-circuit link eliminated the self-oscillation capability of the simulator. Initially the network had to be aligned so as to satisfy the commensurability criterion, see Section 3.3.1. Because of the interaction which exists between the group delay around the loop and the frequency of operation due to,

(i) Tuning (L - C ratio) of the tuning filters,

(ii) Phase-frequency characteristics of the delay line,

(iii) Phase-frequency characteristics of the regenerator, other than the tuning filter circuit,

(iv) Loading effects of ancillary test equipment, signal sources and their point of connection,

a "cut and try" method had to be resorted to with the aim of pro ducing an overall convergent network tuning. 109

The 50£2 delay line constructed for the simulator consisted of 1000 discrete constant-K L (0.66 pH) C (270 pF) sections to form a 24 MHz low pass filter. Each section yielded a delay of 13.1 nanoseconds to provide a total delay of 13.1 microseconds and attenuation of 34 db at 8 MHz. Although the delay line phase-frequency characteristic is sufficiently linear at this fre- _4 quency (L.P.F. phase = 2 sine f ), the available delay time quantisation ^cut off of 13.1 nanoseconds compared to the 125 nanosecond period of the carrier fre

quency was too coarse to enable precision adjustments to be made. This problem was overcome by the following adjustment procedure;

1. Select the approximate carrier frequency.

2. Align all filters to that frequency.

3. Activate loop by feeding in a test word pattern.

4. Adjust tap on delay line until closest approximation to the commensurable loop delay time achieved as indicated by the maximum number of regenerations possible to produce in the network.

5. Reset carrier frequency so as to further increase the number of regenerations which can occur in the network.

6. Retune all filters to the new frequency.

7. Repeat 5 and 6 until the interaction between tuning and forcing frequency minimised.

Because of the capability of the SASTN circuit in each regenerator to break synchronisation if the timing sample is not located in the central region of the data pulse, a situation which occurs readily whenever the loop delay time and forcing frequency do not produce an exact alignment of cir culating pulses with previous pulses, a false mode of operation can occur.

Provided the overlap misalignment is not severe the SASTN will add or subtract an appropriate number of pulses to the frequency divider which will compensate 110

Retimed Envelope of

Circulating Pattern

10010000

Digital Phase Comparator

Output

a) Correct Alignment, = 8.056 MHz

Retimed Envelope of

Circulating Pattern

10010000

Digital Phase Comparator

Output

b) Incorrect Alignment, f = 7.994 MHz c

FIGURE 5.3 Four Photographs illustrating one correct and three false

modes of stable operation for the simulation network due to

incorrect choice of carrier frequency for a given loop delay time. Ill

Retimed Envelope of

Circulating Pattern

10100100

Digital Phase Comparator

Output

c) Incorrect Alignment, f = 7.983 MHz

Retimed Envelope of

Circulating Pattern

10100100

Digital Phase Comparator

Output

d) Incorrect Alignment, = 7.938 MHz

All photographs, H = 20 ySec/div

V = 2 Volts/div 112

for the error that would, without the SASTN characteristic, cause a progressive creep of the data envelope with respect to the loop time delay and produce eventual loss of synchronisation. For a given amount of frequency error a specific number of corrections to the frequency divider has to occur in order that synchronisation can be maintained. Because a correction can only be applied whenever a data 0-1 transition occurs, the false mode of operation will be pattern dependent. The above mechanism also enables the same pattern to be supported by more than one incorrect frequency. It is therefore essential to test simultaneously;

1. the amplitude response, to establish that no pattern changes occur,

2. the timing wave frequency, to establish if any additional segments are being added or subtracted,

to guarantee that the correct set-up procedure has occurred. Figure

5.3 consists of four photographs illustrating one correct and three false modes of operation for the simulation network. In particular the measurement of the timing wave jitter, performed by a bistable multivibrator which has its set input derived directly from the transmitter frequency and its reset input driven by the regenerator timing frequency, provides the necessary sensitive indication that a correct alignment has been achieved because, as indicated in Figure 5.3, any frequency error will produce a cyclic pulse width modulated output waveform dependent upon the amount of error and the data test pattern circulating in the loop.

(b) Synchronisation

With the simulation network aligned, a check on the synchronisation characteristics of the regenerators can be performed. The SASTN circuit was switched out so as to provide a regenerator without the capability of internal synchronisation correction. Any attempt to circulate data test patterns re sulted in high error rates and loss of pattern before the required number of 113

Regenerator Input 200 mV/div

Retimed Envelope 2V/div

■

a) Pattern changes due to lack of synchronisation H = 5 ySec/div

Regenerator Input 200 mV/div

Retimed Envelope 2V/div

b) Correct synchronisation H = 5 ySec/div

FIGURE 5.4 Pattern Synchronisation and Simulator Quench Control. 114

Regenerator 1 Retimed Envelope

Initial Pattern Feed-in after Quench Period

c) Simulator Quench and Number of Regenerations Control H = 100 ySec/div, V = 1 V/div

Regenerator 1

Regenerator 2

d) Regenerator AM input waveforms, 11001000 pattern H = 50 ySec/div, V = 200 mV/div. 115

regenerations were performed.

With the SASTN circuit switched in, any test pattern would syn

chronise the regenerators. Because of the available error counter character

istics, when combined with the data speed of transmission, it was not possible

to check accurately the initial error rate versus test pattern and time taken

to correctly synchronise.

After initial synchronisation, and with no high noise level being

deliberately fed into the system, the SASTN circuit could be switched off and no difference in performance could be detected; the initial synchronisation held and many different test patterns could be switched in and out. Figure

5.4 indicates the pattern changes due to lack of synchronisation, the correct

synchronisation with the retimed envelope being established near the center of the input pulse, the initial feed-in ofallOOlOOO test pattern together with both of the AM input waveforms that have been regenerated 61 times. The restriction on the high noise level previously mentioned arises because the amplitude detection network enables large spikes to pass through and be occasion ally recognised by the retiming process. This then introduces random pulses into the circulating data pulse stream which produces an error count due to an effect not being considered at this stage of the analysis.

Thus the synchronisation characteristics are such that the SASTN is effective and necessary.

The alignment finally obtained for the investigation required a carrier frequency of 8.056 MHz, a data speed, with M = 8, of 1.007 M bits/ second and a loop delay time of 12.9 microseconds which permits 13 bits of data to fit precisely into each loop time delay interval. No equalisation 116

Regenerator

FIGURE 5.5 Regenerator RF input waveforms after one hour of circulation of the pattern 10101100

H = 5 ySec/div V = 200 mV/div 117

networks were connected into the loop to compensate the line characteristics.

Every time that a word circulates around the loop it is regenerated twice, thus for every second the 8 bit test word is regenerated approximately

155,000 times or in other words it takes 6.45 seconds to enable 1 million regenerations to be performed.

The long term frequency stability of the transmitter oscillator, 0 on load, was measured to be 4 parts in 10 and this was sufficient, when coupled with the up-dating capability of the SASTN to keep the synchronisation within proper bounds, to allow test patterns to circulate for several hours provided a short initial warm-up period of five minutes was allowed. Figure

5.5 is a photograph of the two regenerator inputs after 1 hour of circulation of the test pattern 101001100 , upon which over 500 Million regen erations had taken place. This number of regenerations is equivalent to

4 complete trips around the world in a digital regenerative communication system containing one regenerator every one foot length.

Although the simulation network is restricted to the conditions determined by the commensurability criterion, the periodic sampling system which is produced demands operating characteristics that are more stringent than those occurring in a real communication system containing the same re generators. During the alignment and synchronisation procedures some of the important aspects of the regenerator’s timing characteristics were re vealed together with the capability of the simulation network to enable detection of any departures from correct operation.

The important question to answer at this stage of the thesis is, "Does the simulation network developed provide a suitable means by which

an accurate assessment of the timing characteristics for this particular

type of self-timed digital regenerative repeater can be performed.

This is best answered by considering the following facts,

1. All aspects of the system behaviour examined thus far have

been totally deterministic.

2. No outstanding departure of anticipated behaviour exists.

3. The test configuration developed guarantees that the re

generators derive their timing wave from their own input AM waveform and

from nowhere else, as would be the situation existing in the real communi

cation system.

4. The non-real system oscillatory behaviour of the simulation

loop has been eliminated.

5. A method of forcing the carrier frequency into the loop without preventing the free propagation around the loop of any digital

pattern disturbance, either random or systematic, exists.

6. Sufficiently sensitive test equipment is being utilized which enables measurement of all critical parameters.

For these reasons the answer to the question must be yes.

Further discussion upon the validity of the results obtained from the

simulation network will be presented in the following sections.

5.2.3 Loop-induced Spectral Characteristics

To examine the effect of the loop-induced spectral line at the 119

carrier frequency upon the composition of the timing wave, the carrier fre quency signal at the timing filter output was measured with a Tektronix 1L10

Spectrum Analyser tuned to that frequency. Over the range of resolution avail able from the 1L10, 10 Hz to 2 KHz/cm, no difference could be recorded in the spectral characteristics of the carrier frequency with and without data being circulated around the loop. This effect was also independent of whether or not the actual loop circuit was open or closed.

Although shaping of the spectrum must exist due to the periodic re generation of the data, see Section 4.7, no improvement beyond those conditions which would be found in a normal regenerator operating in a real system was produced for this self-timing technique. This is a direct consequence of having a frequency impulse function to start with in the first instance. The same situation does not exist in a pure baseband system where considerable re finement of the spectral energy distribution near the timing frequency is produced by the circulating test word pattern.

It should be observed that the initiating digital data pulse word generator also repetitively produces the same test word at a periodic rate.

This repetition rate is normally synchronised to the same rate at which the data circulates the loop. Thus any test sequence which utilises the technique of circulating the same test pattern on a periodic basis, whether it occurs in the external test equipment or internally in the loop, must have the basic test word energy distribution modified to include the line spectra. The 120

FIGURE S/N Improvement db 2 Noise Reduction Ratio

5.6 5.7

Comparison factor to Signal-to-noise timing experimental

versus

filter of

theoretical

timing

ratio bandwidths Q-factor

improvement filter

theoretical B and t

experimental Q-factor

versus

ratio noise

of reduction

channel

121

importance of this observation is not generally recognised and no allowance has been made for this effect by other research workers who, in utilising baseband retiming techniques, must be operating in a frequency region where a loop-induced spectral line exists.

5.3 Timing Filter Characteristics

5.3.1 Noise Reduction Ratio r

In Section 4.2.2 an analysis was performed which predicted the noise reduction ratio for various ratios of timing filter bandwidth and channel bandwidths. To simplify the analysis a "flat" noise spectrum over the input bandwidth was assumed. However, in the experimental investigation the noise spectrum was shaped by a 2 MHz bandpass filter centered on the 8 MHz carrier frequency in order to represent the typical band limited conditions which could exist in a real system. This has led to a difference between the ex perimental and theoretically predicted value for r, the wider the bandwidth of the timing filter the greater the error. It will be observed from Figure

5.6 that the error is negligible for the Q = 50 filter employed in this in vestigation and thus either the theoretical or experimental reduction r may be correctly applied for any further analysis.

The noise reduction was obtained by measuring the voltage gain of the timing filter at the center frequency, the r.m.s. noise voltages at the input v . and output v ni no

Reduction ratio r (5.1) A v . v ni 122

b = 2.5

b=5.0

Mistuning %

FIGURE 5.8 Minimum acceptable signal-to-noise ratio at timing filter output,

which still maintains synchronism, versus percentage mistuning for

various ratios of channel to timing filter bandwidth. 123

This value was then checked by feeding in the carrier signal and measuring the input (S/N^) and output (S/N^) signal-to-noise ratios. By com paring the results of equations (5.2) and (5.3) an excellent check on the linearity of the timing filter stage could be performed.

Noise Reduction R = 20 log 1_ , db (5.2) r

Signal-to-Noise Improvement = S/N^ - S/N^ db (5.3)

The noise reduction, or improvement in S/N ratio for various ratios of channel and timing filter bandwidths are plotted in Figure 5.7.

5.3.2 Synchronisation Level

The next important aspect tested, which involved a variation of the timing filter characteristics, was that of determining the maximum tolerable noise level at the output of the timing filter stage which would not cause a measurable change in the value of the number of zero crossings of the carrier frequency. This particular S/N ratio output versus bandwidth of the timing filter determines the spectral purity of the timing wave frequency impulse function required to keep the whole communication system in synchronisation.

Under all conditions this particular S/N^ must always be exceeded at every regenerator's timing filter output.

Figure 5.8 presents the minimum acceptable S/N ratios at the output of the timing filter stage for various bandwidth ratios together with the effect of mistuning the carrier frequency. The necessary increase in S/N ratio required to prevent the loss of synchronisation due to mistuning may be read directly off this same graph and indicates that provided the mistuning is kept to less than 0.1% no difficulties should be encountered with loss of 124

synchronisation due to changes in S/N ratio.

5.3.3 Limiter Characteristics

Dependent upon the S/N ratio entering the limiter stage and the

shape of the limiter amplitude input-output transfer curve, a variation in the total S/N ratio improvement of the filter-limiter-filter will occur. The minimum S/N ratio applied will be determined by the selected bandwidth and

signal levels from the first narrow band filter. Since the second filter bandwidth is greater than that of the first filter, the results of Section 4.5 may be validly applied.

To determine if the limiter was modifying the S/N improvement pro duced by the first filter, the S/N input to the first filter was measured in

order to determine what minimum S/N ratio would be required to maintain the

same average mumber of zero crossings as the carrier frequency at the output

of both the first and second filters. A direct measurement of the S/N out put of the second filter is difficult to perform because of the action of the limiter and second filter, (the main reasons for this can be best discussed in the next section after the results of this section have been presented), thus an indirect technique of measuring the spectral purity in terms of frequency counting had to be applied.

The actual implementation being examined, see Appendix 2, utilized a 2N3693 NPN silicon transistor as the active element in the first filter stage with a bandwidth of 160 KHz (Q = 50). For this combination a minimum

S/N output of 11.4 db was required to guarantee that no change in the average number of zero crossings occurred. The corresponding S/N input was -7.6 db thus indicating that the first filter stage network provided a 19 db improve ment (r = 0.11) in S/N ratio. From Section 4.5 the 11.4 db input to the 125

a) 0 db S/N^ ratio, output zero crossing rate not constant

b) 4 db S/N^ ratio, output zero crossing rate constant

FIGURE 5.9. Regenerator input and output waveforms, Q=50 H = 200 nanosecs/div V = 20 mV/div lower trace, 50 mV/div upper trace. 126

limiter-second filter combination should produce a S/N output of 13.4 to

14.4 db due to its own improvement characteristic, Figure 4.5. Thus the in put S/N ratio to the first filter stage could be lowered by the 2 to 3 db so as to yield a -9.6 to -10.6 db minimum operating level. When the actual test was applied, the minimum S/N was measured to be +2 db, a maximum discrepancy of 12.6 db. Figure 5.9 shows the regenerator input and output waveforms for

S/N levels above and below the 2 db level. Separate tests were then conducted upon each of the Fairchild 4L914 Integrated Circuits being employed in the 2 stage limiter sections. Their noise figures were measured to be 8 db at 8 MHz over a bandwidth of 800 KHz, the same bandwidth as the second filter stage

(Q = 10). The effect of the noise figure of the first filter stage has been already included in the 19 db improvement figure. The individual noise figures were then combined, according to Equation (5.4), so that a check on the S/N discrepancy could be made.^

11 1 F = F - 1 + F (5.4)

where Noise Factor = antilog (Noise Figure) 10 F combined 2 Stage limiter noise factor

Noise factor of first limiter stage

Noise factor of second limiter stage

Power gain of first limiter stage

The unknown variable parameter is G^.G^ of a limiter is non-linear and thus a variation in F must occur if is signal level dependent. This variation of G^, and thus F, should produce a change in the minimum input S/N ratio required to maintain synchronisation and the effect does exist, the re sults being plotted in Figure 5.10. The modulation system developed in the regenerator was peak power limited, and since various levels of ON/OFF ratios 127

29db 1/db input ON/OFF range

mV r.m.s.

FIGURE 5.10(a) Filter-Limiter amplitude voltage characteristic

Input ON/OFF Carrierleak-to-noise ratio db

FIGURE 5.10(b) Change in minimum S/N ratio due to change in

saturation of limiter altering effective gain G^ 128

were to be employed, the carrier leak OFF level had to be the variable. This meant that the range of operation upon the limiter characteristics would change according to changes in carrier leak level, thus producing a variation of G^. Intuitively the larger the operating range swing, the greater the saturation and hence the smaller the effective G^. This would then require an increase in the input S/N ratio (the carrier leak level has to be increased to provide the smaller ON/OFF ratios) so as to compensate for the increased effect of the 2nd Stage of the limiter noise contribution. This change in minimum S/N input level versus various ON/OFF ratios is plotted on Figure 5.10 together with the limiter amplitude transfer characteristics. It should be noted that although a specific measurement of the effective G^ has not been obtained, the results shown on Figure 5.10 clearly indicate that an allowance must be made for a variation in the S/N performance due to variations in the operating region of the non-ideal limiter characteristics brought about by different carrier leak levels. These corrections will be applied in Sections

5.5 - 5.6. Based upon the measurements taken for these experiments, the max imum discrepancy of 12.6 db can be used to give a measure of the effective G^ for the particular input ON/OFF ratio.

F = antilog 12.6 =18.2 10

F^ = F^ = antilog 8_ = 6.3 10

From (5.4) ,

G 1 1 1 (5.5) F

= 5.3 11.9

0.45 129

This value of G^ is consistent with the square of the slope of the saturation section of the voltage transfer curve at the minimum ON/OFF level, or maximum carrier leak, of 17 db.

The variation in is inherent in the limiter action and provided the variation is noted in the design procedure, a worst case example would adequately enable the designer to allow for this effect. In actual fact a trade off between G^ and limiter output amplitude variation could be made.

However, that procedure is not recommended particularly when it is considered that a better gain would be obtained if active devices with a lot smaller noise figures were used in any given implementation. The problem of high noise figures in linear integrated circuits is a serious drawback for any communication system which is finally envisaged as having every repeater com pletely fabricated into a single integrated circuit chip. The poor noise figures must lead to a conflict between such things as transmitted power out of each repeater, the spacing between repeaters, the necessity for narrower bandwidths together with the stability problems of the timing filter networks.

For the implementation utilized in the simulation network, the poor noise figures (8 db) produced a degradation of 12.4 db in achievable S/N improvement for the bandwidth in use. If these noise figures could be reduced to 4 db, the resultant noise factor F, using the same value for G^, would decrease from

18.2 to 5.8, a saving of 5 db. This is equivalent to almost a quartering of the output power from the regenerators.

5.3.4 Noise Convergency

Before presenting the noise convergency results, the following dis cussion is presented upon the direct measurement of the S/N levels at the output of the regenerator or timing filter section. S/N Ratio Normalised Meter Deflection

Signal Alone 1.000

20 db 1.005

17 db 1.010

15 db 1.015

13 db 1.025

Meter Order of Accuracy: ± 5% at 10 MHz ± 3% at 5 MHz

TABLE 5.2 Comparison of required meter deflection

for different S/N ratios with relevant

order of accuracy figures for the R.M.S.

meter employed for the measurements.

(Hewlett Packard True Reading R.M.S. Meter). 131

The gain and noise reduction mechanism of the first filter network operating in the linear portion of its characteristics is not signal level de pendent. Thus an accurate measurement of the noise reduction can be made without the carrier signal being present. If the carrier has to be present, the order of accuracy of the results is degraded because the difference between the r.m.s. meter deflection of different signal to signal plus noise levels becomes progressively smaller as the S/N ratio increases. Table 5.2 lists some S/N ratios with the corresponding changes in meter deflection. Although a change in meter deflection will occur if the noise level changes, an accurate measurement of whether the new S/N ratio is, say, 18 or 19 db, cannot be made with this method. No meter offset facility was available to enable a lower or expanded scale range to be used so that the difference could be more accurately measured.

It was observed in the last section, 5.3.3, that the limiter per formance is highly signal level dependent. If the carrier signal is removed, the gain increases because the same degree of saturation does not exist.

Although the increase minimises the second limiter stage noise contribution, the change in gain amplifies the input noise to a greater extent. Thus to avoid these problems the indirect method of measuring the spectral purity in terms of the average frequency count was employed.

When the simulation network was being tested for the noise build-up or convergency characteristics, the just mentioned constraints upon measure ment and operation of signal levels had to be observed. Also it should be recalled that the analog section of one of the regenerators in the simulation network has the link between the timing filter and modulator open circuit.

The reasors for this have been previously discussed in Section 5.2. This open circuit prevents the noise from passing through that particular regenerator 132

and circulating freely around the test loop. When conducting the forced response noise measurements the re-inserted carrier was combined with an equivalent amount of noise so as to simulate the normal spectral purity of the Mcleaned-upM carrier feeding the modulator, see Figure 4.8(c). The validity of that procedure depends a great deal upon the noise build-up and the presence of any pattern dependent frequency components which appear at the input to the modulator from out of the timing filter.

In order to test the noise build-up the open circuit link was closed, the transmitter carrier frequency removed and the system allowed to oscillate.

The signal levels, without noise, were adjusted to give the same levels as those appearing when the system was operating in the forced mode. The loop was then opened at both regenerator outputs and independent precise (band- limited) noise levels were fed into each regenerator input. The output S/N of each independent regenerator was checked in order to determine the r.m.s. meter deflection for each separate S/N input. After each input S/N ratio had been established and checked, the loop was closed and the change in S/N level at each regenerator input noted and compared with the meter deflection differ ences recorded for the independent S/N output measurements. These tests were confined to the narrow band filter configuration, where any additional noise input to the regenerator is reduced by 19 db. For all the S/N ratios, over the operating range of 17 to 26 db input ON/OFF ratios, no departure in de flection except for the expected and previously measured feed through occurred.

These results were cross checked, by usins the indirect method of measuring the spectral purity, and confirmed that the noise converged to a stable and controllable level as predicted by the analysis in Section 4.3.

5.4 Timing Jitter Characteristics

The most important test that has to be carried out in order to 133

a) Pattern 10110100

Retimed Envelope

Digital Phase Comparator Output Waveform

b) Pattern 00000000

FIGURE 5.11 Digital Phase Comparator Waveform versus Pulse Pattern. H = 2 ySec/div, V = 2 V/div 134

evaluate the capability of the coherent carrier technique to self-time a

chain of digital regenerative repeaters is a measurement which will detect departures, both random and systematic, of the timing wave phase in any re

generator from the phase characteristics established by the originating trans mitter's timing wave. These phase departures, referred to as timing jitter,

are readily observed by comparing a timing wave derived directly from the transmitter carrier frequency oscillator with any regenerator's timing wave through a phase comparator. The method of operation of the digital phase

comparator, utilizing a R - S multivibrator, has already been referred to in

Section 5.2.2(a) where this measurement provided the sensitive indication

that correct alignment had been achieved.

The timing jitter is tested by allowing static digital test word patterns to circulate around the simulation loop a controlled number of times.

During the test sequence a direct measurement of the r.m.s. value of the out put of the digital phase comparator is made.

5.4.1 Pulse Pattern Dependency

Systematic jitter is carried through a system by any word pattern, or portion thereof, which allows the system deficiencies to create accumulative phase disturbances. The presence of this pattern dependent process is in itially tested without any additional noise being deliberately fed into the input of the regenerators.

With the exception of a few test words, the phase difference between the timing wave of the regenerator under test and the transmitter timing wave remained constant for a fixed number of regenerations. The pulse pattern could be changed without any change in the phase difference being recorded thus indicating that the regeneration retiming process was insensitive to 135

FIGURE 5.12 Line reflections of pattern 11010000 at pulse position 5 (Line 1). H = lOpSec/div V = lOOmY/div 136

pattern changes. Figure 5.11 shows the retimed envelope of a regenerator together with the output of the digital phase comparator for different test word settings. The initial test word patterns which produced phase variations were those with al---01-- sequence (the - being either a 1 or a 0).

Of further importance was the observation that when the SASTN circuit was switched off for Regenerator 1 the phase disturbance disappeared. All test word patterns could then be switched in and out with no variation in the phase measurement being recorded. This effect was independent of the SASTN circuit setting for Regenerator 2.

Examination of the input waveform into Regenerator 1 revealed that a reflection from the first pulse on Line 1 was appearing at the crossover interval between pulse position 5 and 6 of the test word and adding to the leading edge of the sixth pulse. The resultant effect was that this modified leading edge no longer possessed the initial time coherence between the other leading edges of the pulses and itself. The SASTN circuit was then adding and subtracting appropriate time segments to the timing wave in order to keep the sample point located at half a pulse interval from every leading edge. Figure 5.12 is a photograph showing reflections on Line 1. When the delay line was constructed, with component tolerances of ±5%, no attempt was made to grade the values so that a gradual transition in component value difference existed along the line length. The random placement of the large number (1000) of discrete and different valued components proved to be a serious drawback for the delay line pulse performance. By crossing over the connection of the delay lines to the regenerators, the pattern dependent effect was then controlled by Regenerator 2 and independent of the SASTN circuit setting in Regenerator 1. This confirmed the diagnosis that the fault existed in Line 1. 137

To avoid the problems associated with the discrete section line, and still achieve the required delay, one and a half miles of coaxial cable

is required. This cable length was not available for the project and so the reflections on the discrete section line had to be tolerated. Further adjust ments were made to the line in order to minimise the reflections. Provided they could be made greater than 20 db below the pulse peak their effect was not troublesome. During these adjustments it was observed that a large amount of interaction existed; the curing of one reflection allowed another to occur which in turn changed the particular test word pattern which produced the phase variations.

When the reflection problem had been overcome all 36 different test word patterns could be switched in and out at random with no variation in the phase difference being measured between the timing wave derived from the trans mitter carrier frequency oscillator and the regenerator under test. Thus for a fixed number of regenerations the retiming process is pattern insensitive.

5.4.2 Number of Regenerations

The next test conducted, which involved a measurement of the timing jitter, consisted of varying the number of regenerations for each particular test word pattern. For every pattern the number of regenerations could be selectively switched and repeated up to two million times. It was also possible to select any particular pattern, insert it into the loop, and then switch out the loop quenching control so that the pattern would circulate for any desired length of time.

In every case no variation in the timing wave could be measured clearly indicating that this regeneration technique does not produce any systematic timing jitter accumulation. 138

5.4.3 Signal to Noise Ratio

In the previous two subsections the timing jitter performance for the regeneration technique in a minimum noise level environment was pre sented. The next question to answer is whether or not the timing wave syn chronism can be maintained in the presence of additive noise together with the various test word patterns. Because no pattern dependency could be

found, the synchronisation is most severely tested during long periods of zeroes when the S/N ratio is at its minimum value. For periods of successive ones the S/N ratio is at its maximum value. The effect of the non-ideal limiter characteristics upon the S/N performance has already been discussed and it should be observed, from Figure 5.10, that the variation in is re stricted to the knee area of the limiter curve where the low OFF levels exist.

Thus the higher ON levels do not produce any further marked degradation in

S/N performance such that the synchronisation would not be maintained.

The tests conducted for this section required that after each ON/OFF level had been adjusted from 17 to 26 db, in 3 db steps, that particular S/N level which just maintained synchronisation for the OFF period was obtained.

Each test word was then fed into the loop and the timing jitter phase com parator output was measured. The noise level was increased until the syn chronisation was broken. Below this noise level the phase comparator output remained constant and independent of test word pattern, number of regenerations and S/N ratio. Most importantly, the network exhibited a threshold effect for any given ON/OFF ratio, and provided the effect of the non-ideal limiter was taken into account this threshold was independent of the OFF level for ON/OFF ratios greater than 19 db. The lower ON/OFF produced amplitude errors before the synchronisation breakthrough occurred. These random amplitude components caused undesired switching of the SASTN circuits which masked the true synchronisation break through. 139

Thus for the test sequences employed no relationship between S/N ratio and digital word pattern could be found. The measurement of the timing jitter on the R.M.S. voltmeter remained constant, independent of the

S/N ratio until a synchronisation break-through occurred or non coherent amplitude components were generated.

No accumulation of timing jitter, either systematic or random, occurred. The spectral purity of the timing wave, as determined by the timing

filter bandwidth, remained within established and controllable bounds thus

satisfying the major design objective, as outlined in Sections 3.2 and 3.2.1.

5.5 Error Rate

In this section the results of the investigation into the number of errors that the simulated network made as a function of pulse pattern, number of regenerations and S/N ratio are presented. These error rates include the effects due to:

(a) the break down in synchronisation due to a change in the average number of zero crossings of the carrier frequency

(b) the recognition of random pulses produced by the high noise level exceeding the threshold level whenever a timing sample pulse was applied in the baseband amplitude regeneration section of a regenerator.

Because the specific objective for this thesis was to investigate the timing characteristics of a particular regeneration technique, it was de cided to design the timing filter characteristics so that a breakdown in synchronisation occurred, over the major portion of the range of measurements, before the high noise level produced recognizable pulses in the amplitude de tection network output. This was readily achieved by making the bandwidth of the timing filter wide enough to permit sufficient noise energy to pass through the filter and cause the required degradation in spectral purity of the 140

0 Test Pattern Errors in 10 bits

00000000 0 10000000 0-10 11000000 0-50 10100000 0-30 10010000 0-30 10001000 0-10 11100000 70-350 11010000 0-40 11001000 0-40 11000100 0-40 11000010 0-40 10101000 0-30 10100100 0-30 11110000 1,100-3,000 11101000 30-600 11100100 70-350 11100010 70-350 11011000 80-180 11010100 0-60 11010010 0-60 11001100 0-60 11001010 0-50 10101010 0-70 11111000 3,500-6,000 11110100 1,000-3,500 11110010 1,000-3,000 11101100 400-1,000 11101010 170-420 11100110 270-700 11011010 20-200 11111100 5,200-7,000 11111010 3,100-5,500 11110110 1,500-3,500 11101110 1,000-4,000 11111110 7 ,000-11,000 11111111 10,000-15,000

TABLE 5.3. Pattern Dependent Error Rate; 23 db ON/OFF. ratio, 2 db S/N ratio, N = 100 r FIGURE 10000000

5.13 of Pattern l's

contained Test dependent

Pattern

in error pattern.

rate

associated

with

density

11111111 142

timing wave.

In a real communication system an error in a regenerator due tb either (a) or (b) above is equally incorrect. However, as far as the total real network behaviour is concerned the filter bandwidths should be made sufficiently narrow in order to prevent condition (a) occurring before con dition (b). Whenever synchronisation is lost in any particular regenerator, the timing of all succeeding regenerators will be perturbed until the dis turbance has propagated through the rest of the system. This perturbation would yield a higher error count than if condition (b) alone caused the initial error.

5.5.1 Pulse Pattern Dependency

Two important pattern dependent error processes were carefully checked during this phase of the experimental analysis. The 36 different test word patterns were employed with selected ON/OFF ratios (17 to 26 db) and S/N ratios (+12 to -2 db) during the OFF period. One process of interest was that which might produce a particular class of patterns to which other patterns would be selectively changed due to errors. These new patterns would indicate that a class of patterns exist which is more immune to noise than the remaining patterns. This selective mechanism could not be found indicating still further that both the regeneration technique and the simulation network are pattern insensitive. The other process involved the existence of a class of patterns which was more susceptible to noise than the remaining patterns.

Table 5.3 lists a particular test sequence for an ON/OFF ratio of 23 db and

S/N ratio of 2 db, just on the verge of synchronisation break-through.

Analysis of this table reveals that the error rate is definitely pattern de pendent. The higher the number of l's in a pattern, the higher the error count provided that no 0*s were inserted between the lls. Wherever a 0 occurs 143

Threshold Circuit Input, Regenerator 1 V = 0.5V/div

RF input pulses, Regenerator 2 V = 0.2V/div

a) Stable operation H = 20 ySec/div

Threshold Circuit Input, Regenerator 1 V = lV/div

RF input pulses Regenerator 2 V = 0.2V/div

b) Unstable operation, pulse pattern change H = 50 ySec/div

FIGURE 5.14 Effect of D.C. level shift at input to Amplitude threshold circuit. FIGURE actual ideal actual ideal

5.15

Change circuit (b) (a)

in wide due narrow

effective to

D.C. pulses

pulses pulse level shift height

associated applied

with threshold pulse

pattern decision threshold level 144 145

between the l’s the error count is reduced. Figure 5.13 is a plot of the patterns with continuous l’s versus error rate. This pattern dependent error count is caused by the D.C. shift associated with the N.R.Z. data format and method of detection, without D.C. restoration, employed in the investigation.

Figure 5.14 contains two photographs of the input to the amplitude threshold

(Schmitt trigger) circuit showing the change in effective height of the input because of the D.C. level shift. The time constant of the capacitive coupling and by-passing controlled the rate of change of the D.C. level and this effect is observable in the two photographs. Figure 5.15 details the mechanism by which the error rate is linked to the D.C. shift and effective pulse height by allowing the noise residing on top of the jpulses to produce the pattern de pendent error count.

To allow the higher bit density pulses to circulate the required number of times, the D.C. shift could be offset by adjusting the A.C. gain of the detection network. This procedure was checked before each test sequence to establish that the "noise free" patterns wouls circulate correctly (these problems associated with the D.C. level shift would be readily overcome in any production model regenerator employing this self timing technique). The trigger level was always adjusted to occur between 0.5 and 0.6 of the maximum pulse height.

No other pattern dependent error mechanism could be found to exist.

5.5.2 Number of Regenerations.

A difficulty encountered in the interpretation of Ihe error results is the fact that when testing a particular word pattern with a specific noise level that is causing changes in the timing wave, the particular word pattern changes progressively to some other and continuously changing pattern. This 146

means that the latter part of an error count sequence for a large number of regenerations is not testing the initial pattern any longer. In general it was observed that, with the synchronisation breakdown occurring first, the error count reduced as the number of regenerations increased. This process was caused by the progressive loss of the 1 bits from a word because a slip

in the synchronisation leads to a bit loss with a greater probability than the addition of a bit to a word. This effect was readily observed by scanning across the regenerator outputs with the delayed time base facility of a suit able oscilloscope (Tektronix 585A). If the regenerator outputs had not been observed, a series of misleading results would have been obtained. In some cases, with the number of regenerations exceeding 100, the error count reduced considerably because the initial pattern had been completely lost due to the lack of synchronisation. It is therefore essential to observe the patterns circulating around the loop when recording the amplitude error rate or cross check the error rate with the word generator test patterns.

The most important feature revealed from these particular tests is the existence of the synchronisation threshold. Provided the S/N ratio is kept above this level, the regeneration technique does not produce any pattern dependent error count which is also dependent upon the number of regenerations.

Once the S/N ratio falls below this threshold level then either the synchro nisation is lost or amplitude recognition of noise spikes are made, or both occur. No reliable data could be obtained upon pattern dependent error rate versus number of regenerations for these low S/N ratios.

5.5.3 Signal-to-Noise Ratio

As mentioned in Section 5.5.1 the error measurements were conducted for specific ON/OFF ratios, varying from 17 to 26 db, for the S/N ratios of

+12 to -2 db for the 36 different test word patterns. These combined results are plotted in Figures 5.16 and 5.17 where the existence of the threshold (•2000

0 FIGURE 5.16 Errors in 10 pulses versus carrier-leak S/N and ON/OFF ratio

Note lower rate of errors with change in S/N ratio for

lower ON/OFF ratio because of amplitude errors occurring

before synchronisation break-through. 148

a=17 db ON/OFF b = 20 •• " c =23 •• d = 26 " •*

6 =29 n "

S/N ratio db ( i) Measured error characteristics

31000

S/N ratio db (ii)Corrected error characteristics FIGURE 5.17 Correction of measured error characteristics due to chanpe in (S/N) . by variation of limiter G (see FIGURE 5.10(b)). mm l 149

is observable.

The restraint imposed upon the simulation network operation, see

Section 3.3.1, which requires a commensurable time relationship for pulses travelling around the loop proved to be a serious drawback for the validity of the results obtained in the error analysis. Previous discussions, Section

5.2.2(a), indicated the susceptibility of the simulation loop to produce false operation, with pattern dependent switching of the timing wave, if the correct number of zero crossings of the carrier wave did not exist. Thus when the high noise level produces a break in synchronisation, the resultant change in zero crossing count leads immediately to the situation where the simulation network cannot support the pattern which is being circulated around the critically timed loop.

In a real system this loss of synchronisation, due to a change in the average number of zero crossings, is not as critical because no feedback path normally exists. The real system has an inbuilt high inertia to changes in number of zero crossings because of the timing filter narrow bandwidth and all neighbouring cycles of the carrier frequency can only differ by a small amount. Although the regenerators in the simulation network have the same in ertia characteristics, their performance suffers because the exact fit of the correct number of zero crossings must occur. Any change in the zero crossing count leads to a phase perturbation which will circulate around the loop and eventually cause loss of loop alignment because of the progressive build-up of timing error. This change in zero crossing count in a real system would still cause a perturbation in the timing wave, but because the same regenerators are not being forced to successively track this error, no build up in timing jitter will occur in any one regenerator leading to the high error rates measured on the simulation network for any given S/N ratio. 150

The slower rate of error count is observable for the 17 db ON/OFF ratio where amplitude errors are made before the synchronisation is lost due to the low S/N ratio.

The best that can be said for the error analysis obtained from this particular simulation network implementation is that it validly enables a measurement of the threshold level to be determined. Beyond this threshold the error results are suspect because the successive regenerations are not sufficiently independent to represent the real situation.

5.6 Mistuning

The results in this section are given as a guide to the variation which will occur to those results already presented in Sections 5.4 and 5.5 whenever a limited amount of mistuning exists in the timing filter of the re generators. As predicted in Section 4.6.1 no variation in the noise reduction ratio was observable if the mistuning was limited to less than ±0.1% (Q = 50).

5.6.1 Timing Jitter

Provided that the synchronisation break-through level was not ex ceeded, no variation in the digital phase comparator output could be observed as a function of mistuning over the range of ±0.1% of the nominal carrier fre quency. This effect was independent of the test word pattern being circulated around the loop and the ON/OFF level.

When the mistuning was increased to 1% (Q = 50), the output level from the timing filter was no longer constant. Dependent upon which test word pattern was being circulated, and the 0N/0FF level, the timing wave amplitude fell below the triggering level for the frequency divider (trigger level set at 50% up the positive half cycle of the carrier). This immediately produced 151

loss of synchronisation and large variations in the digital phase comparator occurred. The severity of these deflections increased with an increase in the number of l’s in a word and the increase of the ON/OFF ratio (lower OFF level).

Another effect which was readily observed under the condition of

gross mistuning was the presence of a compression and rarefication of the

timing wave. This effect was pattern dependent and extended, under the con

ditions of 0.1 to 1% mistune, to approximately one bit interval after the

digital pulse which produced it. As the mistuning was reduced so did the ex

tent of the compression and rarefication until it was no longer visually ob

servable on the oscilloscope display at 0.1% mistune. Even under the gross

mistune setting this effect was bounded. The disturbance was localised to the

region of the digital pulses, wherever they were present, and did not produce

any perturbations of the timing wave outside of those regions. The compression

was exactly compensated for by the equal and opposite amount of rarefication

thus maintaining the average number of zero crossings of the carrier frequency

at the correct value. This frequency modulation of the resultant timing wave

is the direct result of the amplitude and phase distortion of the sideband

frequencies produced by the mistuned filter. See Section 4.6.2.

When the system was operated within the ±0.1% limit these two effects

were not observed. The main difference which was recorded, see Figure 5.8,

was the change in synchronisation break-through level as a function of the

timing filter bandwidth and mistuning. Thus by setting a maximum allowable

limit to the amount of mistuning, a simple allowance for degradation in S/N

ratio can be made to prevent loss of synchronisation.

5.6.2 Error Rate

Provided an allowance was made for the change in synchronisation

break-through level due to the mistuning, no change in measured error rate •2000

region ------of errors less than 1 in

FIGURE 5.18 Effect of gross mistiming (1%) upon error rate

versus carrier-leak S/N and ON/OFF ratio. FIGURE

5.19

No Number for

£A//0/r

allowance various

errors

ON/OFF for

limiter in

ratios 10

pulses

with 1 variation

versus gross

mistuning carrier-leak included.

(1%).

S/N

ratio 153 154

between correctly aligned and ±0.1% mistiming was observed. N.B. See comments in Section 5.5 upon the validity of error rate measurements when utilizing the loop method of simulation.

Figures 5.18 and 5.19 depict the error measurements obtained when the regenerators are mistuned by 1%. The presence of the optimum ON/OFF ratio is directly traceable to the amplitude variation in the timing filter producing loss of trigger level to the frequency dividers. These two Figures should be compared with Figures 5.16 and 5.17 to reveal the difference in recorded error rate between incorrectly and correctly aligned filters. Not only have the threshold levels changed but the error rate has increased for any chosen S/N ratio.

5.7 Phase Discontinuities

Essentially the various circuits within the regenerator operate at either of two signal levels, ON or OFF. Associated with this two level switch ing is a change in the phase shift characteristics of the active circuit elements, particularly those transistors in the limiter circuit and the F.E.T. modulator, because of the inherent change in capacitance and resistance of their depletion regions with change of ON to OFF levels. Thus when the circuits switch from ON to OFF, or OFF to ON, a stepped-phase change is produced which advances or retards the phase of carrier frequency in a particular direction, dependent upon circuit configuration, for the given transition and then re verses that direction by an exact equal amount when that transition is removed.

These phase perturbations of the carrier frequency produce angular modulation frequency components. It is conceivable that the situation could exist, with poor circuit design, where the amplitude of the carrier frequency can be made to change according to what effective modulation index is produced by the phase perturbation. Figure 5.20 contains the phase variation versus 155

enerator

o-105

Filter- limil

mV r.m.s.

FIGURE 5.20 Voltage dependent phase shift characteristics of

limiter-filter network and full regenerator 156

carrier frequency signal level for the limiter section of the regenerator

together with the total regenerator input-output phase transfer characteristic.

The maximum phase change through the limiter was 4 degrees whilst the same

change in signal level produces an overall phase change of 11 degrees through

the regenerator. With the timing filter narrow bandwidth (Q = 50) no

variation in the spectral purity of the extract carrier frequency could be

measured on the Tektronix 1L10 Spectrum Analyser. Provided the timing filter

can reject these undesired angular modulation frequency components, no de

terioration in network performance will be produced.

5.8 Conclusions

Where relevant throughout this chapter, conclusions have been made

at the end of each specific subsection. The loop simulation network has been

shown to have deficiencies, namely multimode oscillatory capability, spectrum

shaping of the circulating digital data, and the existence of a threshold

operating S/N ratio. If too low a S/N ratio is employed the simulator is

not capable of producing reliable data about the characteristics of the de

vice being tested within the loop structure because the random number of zero

crossings of the circulating wave destroys the required timing coherence.

However, sufficient information has been obtained to enable meaning

ful conclusions to be drawn from the experimental results. Under the operating

conditions, without and with additive random noise at the regenerator input, no mechanism was found to exist which produced coupling between changes in digital data and the timing wave which had been derived from the carrier frequency.

This particular type of self-timing regeneration technique does not produce

systematic jitter accumulation.

With a rearrangement of circuit connections to provide continuous Number of Effective Effective Repeaters Bandwidth KHz Q-factor

1 160 50

10 89 90

100 28 280

1,000 8.9 900

10,000 2.8 2,800

100,000 0.9 9,000

TABLE 5.4 Dependency of system effective

bandwidth and Q-factor upon the

different number of repeaters connected

in cascade. 157

circulation of the noise, Section 5.3.4, no build-up of noise could be measured thus indicating that the noise convergency predictions of 4.3.2 had been satis fied. This then led to the implementation of the circuit depicted in Figure

3.8 (c) with the filter-modulator link open circuit and band limited noise being added to the carrier to produce the same signal-to-noise ratio as obtained in an unmodified regenerator at the input to the modulator. Now the real system shapes the final noise spectrum due to the cascaded filter effective bandwidth reduction. Table 5.4 lists the reduction in effective bandwidth and the in crease in effective Q for different numbers'of repeaters. For very long systems the effective bandwidth is so narrow that the passage of random noise through the system would produce a pseudo-carrier frequency wave whose amplitude and phase would vary so slowly, because of the total system inertia, that extreme difficulty would be experienced in being able to measure the difference between a pure sinusoidal carrier wave and the noise wave. Hence the single timing channel bandwidth noise being added to the carrier in the modified regenerator does not accurately represent the conditions which would exist in a real system because no spectrum shaping versus number of regenerations being simulated has been included. The spectral characteristics of the carrier frequency could be represented as a narrow band FM signal; the longer the circuit length, the larger the number of repeaters, the narrower the effective bandwidth, the more intense concentration of transmitted noise energy around the carrier frequency.

Further conclusions are left until the next chapter where a discussion upon the overall characteristics of the technique are presented. 158

CHAPTER 6

CONCLUSIONS AND RECOMMENDATIONS

The outstanding advantage possessed by the coherent carrier tech nique of self-timing is that it produces a continuously available frequency impulse at the desired timing frequency. The spectral purity of this impulse can be controlled so that it maintains a constant average number of zero crossings of the timing wave per unit time interval and contains no accumulative amplitude-to-phase, pulse width or adjacent pulse pattern dependent frequency components. This will enable digital communication systems to be constructed with a greater number of self-timed regenerative repeaters because no sys tematic jitter is produced by this technique. Thus the major synchronisation problem which has been encountered by other investigators has been overcome by this relatively simple technique of integrally relating the digital data speed to the carrier frequency. The carrier frequency is then utilized as the self timing frequency source instead of, as in all other systems, the baseband data.

The selection of the carrier frequency as the timing frequency source for the AM spectrum centers the carrier frequency in the middle of the fre quency spectrum being utilised for the communication channel. Any normal dispersion mechanism existing in the channel has the least effect for this symmetrical configuration with the timing frequency source centered between the upper and lower sidebands. Also any time-variable mechanisms which produce equivalent frequency shifts (e.g. continuous path length change due to temp erature variations or even between an orbiting satellite and fixed ground station) forces all the timing filters to track the frequency change and thus the data envelope remains locked to effective carrier frequency. The technique is thus dynamic enough to track these forms of frequency instabilitv whether 159

they were derived by frequency drift in the originating transmitter or by such things as path length changes in the communication channel.

The synchronisation during periods of binary zeroes is achieved by allowing a low level of carrier frequency signal to leak out amongst the noise.

Suitable band pass filters of only moderate Q factors, (Q= 50), tuned to the carrier frequency in each regenerator, extract the carrier wave embedded in the noise from which the desired timing impulse is derived. These regenerators require no coding restraints, no storage, no critically tuned filter or os cillator networks, are not pattern sensitive and employ a technique which, in principle, is applicable to most of the communication frequency spectrum. The only limitation being the present date state of the art for certain technological developments of sufficiently high speed devices.

Because of the ability of the Self Adaptive Sample Time Network

(SASTN) to continuously align the timing wave samples approximately near the center of the input pulses, by modifying the M times frequency divider count rate in every regenerator, the occasional addition of an extra zero crossing of the carrier wave will not produce a break down in synchronisation because the perturbation can be absorbed by the timing network, by shortening a pulse in terval by a one segment shift in the timing wave. This perturbation is propa gated through to the end of the system but the very next pulse transition after the additional zero crossing will enable the timing networks to segment back to the correct timing phase relationship; the synchronisation is self-healing.

The relatively simple band pass timing filter network has been shown to be adequate for extracting the required timing information from the input spectrum to each repeater and controlling the noise buildup along a long chain of repeaters. With a channel bandwidth to timing filter bandwidth ratio of

10:1, an ON/OFF ratio of 23 db, a carrier leak signal-to-noise ratio of 0 db, 160

the equivalence of a 20 db baseband peak signal-to-r.m.s. noise ratio is -7 attained. This yields a theoretical error rate of 5 x 10 . To reduce this error rate,

a) the ON/OFF ratio could be increased for the same noise level and bandwidth ratio

b) the timing filter bandwidth could be reduced for the same noise level and ON/OFF ratio, which would enable a lower carrier leak signal-to-noise ratio to be employed and still maintain the same average number of zero cross ings of the extracted timing wave.

Comparing the coherent carrier system to a conventional pulsed carrier system, to produce the same theoretical error rate with a 0 db carrier leak, signal-to-noise ratio and all other system parameters equivalent, an additional

3 db of peak power would be required. The ability to narrow the timing filter bandwidth, as suggested in previous paragraph (b), would enable a lower carrier leak signal-to-noise ratio to be used, say -6 db. This in turn would reduce the additional peak power to less than 0.3 db whilst still maintaining the same error rate as that obtained for a conventional pulsed carrier system operating in the same noise level environment. Thus the necessity to provide a con tinuous carrier during the OFF periods does not lead to any major increase in the RF power requirements of the system.

Excluding the D.C. power supply, the cost of the repeaters employed in this investigation was $^35. The total power consumption was 1 watt for a

R.F. peak output power of 100 milliwatts.

The loop method of simulation has been shown to provide a limited range of results from which valid performance analysis and prediction can be made. Specifically whenever any variation in the precise number of zero cross ings of the carrier frequency occur, the simulation loop network is no longer 161

operating in a stable manner and any results obtained during this mode of

operation do not represent the conditions which would exist in a real system

if the same perturbation had been introduced. The simulation network thus

exhibits a threshold phenomenon. Another feature of the loop method of sim

ulation is that it can produce multi-modal oscillatory behaviour because of

the delay line loop feedback. This behaviour could be produced in a real

communication system if interconnected loops were formed through various

switching centers. The forced mode of operation for the filter networks in

the simulator prevented the natural resonant modes from altering the desired

operation. Extension of the results obtained from this loop simulator indicate

that if a real system is to contain interconnected loops then the problem of

random self oscillations can be avoided by allowing all sections of the net

work to respond to a forcing signal derived from a reference oscillator. The

problem of phasing the forcing signal through various unstable path lengths

has not been considered. A further feature of the loop feedback is the mod

ification which this produces in the distribution of the spectral energy

presented to the timing filters. These loop-induced spectral lines lead to an

improved spectral resolution being available to the timing filters and thus an

accurate representation of the real system performance does not exist if the real system doesn't intrinsically possess these frequency impulses in its input

spectra.

Many research departments are now able to produce their own special

ised integrated circuits. If the frequency of operation for the simulation network is kept low then no advanced or expensive techniques have to be em ployed to produce the integrated circuits. This would overcome the major advantage of cost saving exhibited by the loop method of testing. The difference in production cost between 1, 10 or 100 integrated circuit repeaters would be negligible. Also the balance between increased loop delay time and 162

decreased operating frequency would not be as difficult to maintain with the non-looped simulator containing the cascaded connection of repeaters. The advantages of this last mentioned method over the loop method would be;

1) no multimodal oscillatory behaviour

2) no loop induced spectral line modification to the frequency spectra

3) no critical frequency relationship between operating speed and loop time delay (COMMENSURABILITY CRITERION)

4) no break-down in simulator operation due to high noise levels thereby enabling an increased range of error rate versus signal-to-noise ratio to be performed

5) no modification to any particular regenerator is required thereby allowing the same type of noise or timing jitter spectrum to build up, without requiring further alteration of any connections, as would occur in a real system.

The main difference which would then exist between the non-looped laboratory simulation model and the real system would be the transmission channel network. Also each repeater in the simulator would have to provide a sufficient number of control points on each circuit chip to mable the required range of parameter variations to be introduced during the tests. Another difference would be the data entering the systems. In general, periodic test word patterns are employed for the experimental measurements. The output spectrum from the equipment generating the periodic test word pattern must con tain spectral line components at harmonics of the periodic repetition frequency.

This modifies the distribution of the spectral energy presented to the repeaters and is different from the real situation where the pulse patterns would, if no coding redundancy used, be random thereby producing a dispersive spectral energy distribution. 163

The recommendation is then, if the economics of time and money allow a choice between the loop and the non-looped cascaded simulation networks, the non-looped should be chosen together with a combination of both periodic and random test word pattern measurements. Although direct evidence has not been obtained in this thesis, nor is there evidence published elsewhere, to enable comparison between a loop simulator and a real system employing the same re- 7 generators, the comparitive results of Byrne et al for a non-looped and a real system indicate that sufficient information can be obtained from a short, non- looped chain of 5-10 repeaters to adequately predict the performance of a long chain of repeaters in a real system. Instead of conducting the investigation at a relatively high frequency where implementation, instrumentation, measure ment analysis and recording difficulties are experienced, the simulator should be a low frequency version of the desired repeater configuration. This would allow a more detailed experimental analysis to be performed because of the re latively higher resolution test equipment being readily available.

The particular regenerator implemented for this study took the form of a forward acting, complete retiming regenerator containing a narrow bandwidth analog channel for the timing filter and a digital section for frequency di vision, sampling, adaptive sample time control and synchronisation. An alternate configuration could exist where part of the digital section could be replaced by an analog network, such as the frequency divider or the amplitude regeneration network. In particular further research is being conducted, in the same de partment, by Mr. G.N. French into the possibility of combining this same co herent carrier self-timing technique with an optimum non-linear partial amplitude 19 regeneration method, as previously suggested by A.E. Karbowiak.

Other problems of interconnected systems, how best to optimise the consumption and feed of the D.C. power into a large number of remote repeaters, 164

the construction of suitable integrable cable and repeater configurations, the design of fail-safe fault self-compensating circuitry to yield a high enough overall system service reliability figure all exist and are to be considered further by a research group of which this author is a member. A factor in the formation of this group was the ability to synchronise a digital communication system containing a large number of repeaters. This synchronisation capability

is made possible by the utilisation of "A Coherent Carrier Technique for Self-

Timed Digital Regenerative Repeaters". 165

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Data Transmissions", Radio and Electronic Engr., Vol.25, pp.155-165,

February 1963.

(24) MC EVOY, W.J. and CRIST, P.W., "Spectrums of Modulated Pulse Trains",

Proc. IEEE, Vol.53, p.1244, September 1965.

(25) LENEMAN, O.A., "Some Remarks on 'Spectrums of Modulated Pulse Trains'",

Proc.IEEE, Vol.54, p.792, May 1966.

(26) WAX, N., "Selected Papers on Noise and Stochastics Processes",

Dover (N.Y.), 1954.

(27) LANING, J.H. and BATTIN, R.H., "Random Processes in Automatic

Control", McGraw Hill, 1956.

(28) DAVENPORT, W.B. and ROST, W.L., 'An Introduction to the Theory of

Random Signals and Noise", McGraw Hill, 1958. (29) LAWSON, J.L. and UHLENBECK, G.E., "Threshold Signals", Radiation

Laboratory Series M.I.T., McGraw Hill, 1950.

(30) FREEMAN, J.J., "Principles of Noise", Wiley, 1958.

(31) BENDAT, J.S., "Principles and Applications of Random Noise Theory"

Wiley (N.Y.), 1958.

(32) RICE, S.O., "Mathematical Analysis of Random Noise", B.S.T.J.,

Vol.23, pp.282-332, July 1944; Vol.24, pp.44-156, January 1945.

(33) RICE, S.O., "Properties of a Sine Wave plus Random Noise", B.S.T.J

Vol.27, pp.109-157, January 1948.

(34) RAEMER, H.R. and BLYTH, R., "The Probability Density of the Phase

Difference of a Narrow Band Gaussian Noise with Sinusoidal Signal",

IRE Trans. Information Theory, IT-7, pp.265-267, October 1961.

(35) DAVENPORT, W.B., "Signal-to-Noise Ratios in Band Pass Limiters",

Journal Appl. Phys., Vol.24, No.6, pp.720-727, June 1953.

(36) BLACHMAN, N.M., "The Output Signal-to-Noise Ratio of a Power-Law

Device", Journal Appl.Phys., Vol.24, No.6, pp.783-785, June 1953.

(37) SCHWARTZ, M., BENNETT, W.R. and STEIN, S., "Communication Systems

and Techniques", McGraw Hill, 1966.

(38) BENNETT, W.R. and DAVEY, J.R., "Data Transmission", McGraw Hill,

1965.

(39) JAVID, M. and BRENNER, E., "Analysis, Transmission and Filtering

of Signals", McGraw Hill, 1963. 169

(40) MAYO, J.S., ’’Experimental 224 Mb/s PCM Terminals”, B.S.T.J., Vol.44,

pp.813-1841, November 1965.

(41) WITT, F.J., "An Experimental 224 Mb/s Digital Multiplexer using Pulse

Stuffing Synchronisation”, Ibid, pp.1843-1886.

(42) CARROLL, W. and P0R0NNIK, K., ”An Infrared, Solid-State Optical

Communication System”, Proc. I.R.E.E., Vol.31, No.7, July 1970, pp.212-219. 170

APPENDIX 1

The technique of relating the carrier frequency f to the data

speed f^ by an integer ratio is referred to on page 262 in the book "Data

Transmission", Bennett, W.R. and Davey, J.R., McGraw Hill, 1965. No refer

ence to any available literature on the subject was made by the authors. A

letter requesting the references was sent to them and the reply is included

in this appendix.

It is interesting to note the comment on the use of the carrier

for purposes other than for the desired timing signal and also the significance of the last paragraph when it is considered that the writer is the head of a department of that organisation which is a leader in the research area under

investigation for this thesis. 171

Bell Telephone Laboratories INCORPORATED

HOLMDEL. NEW JERSEY 07733

Telephone

Area Code 201 949-3000

February 2, 1968

MR. W. CARROLL, Post Graduate Research University of N.S.W. N.S.W., Australia

Dear Sir:

Your letter of January 12 to W. R. Bennett was referred to me since Mr. Bennett has retired from Bell Telephone Laboratories. I am sorry to say that I know of no published reports on the types of synchronous data systems briefly mentioned in the center paragraph of page 262 of "Data Transmission." Some years ago we learned of an experimental system using FM mod ulation in which a marking symbol consisted of one cycle of carrier and a spacing symbol two cycles of carrier. It had been constructed and tested in a laboratory and made use of the zero crossings of the signal to establish symbol timing. It came to our attention after they found that it would not operate over actual telephone circuits when there was fre quency off-set because they lost count of the actual symbols transmitted. I also believe similar systems have been pro posed, and possibly used, over metallic pairs where the wave form is preserved. A form of synchronous demodulation can also be employed by multiplying the signal by the proper phase of the derived timing signal. Where signal-to-noise performance is not important, such signals can also be detected by simple logic type circuitry with essentially no filtering. In such case the use of carrier would be mainly to permit insertion of isolating transformers to provide balance to ground and freedom from ground potentials. To the best of my knowledge it is highly unlikely that any of your work in this area would be a duplication of previous work.

Yours truly, s ^ ^

t JAMES R. DAVEY Head Data Terminals and JRD/je Transmission Dept. 172

APPENDIX 2

This appendix contains the circuit diagram and block schematic diagrams for the regenerator employed in this investigation.

All digital integrated circuits were RTL devices.

Fairchild pL 900 Buffer Amplifier

Fairchild pL 914 Dual-input Gate (Monostable Multivibrators, Trigger Circuits,

Gates)

Fairchild pL 923 JK Multivibrator (Frequency Divider, Envelope Retiming)

See Figures 1, 3.1, 3.4 and 3.7 for further details on Regenerator circuitry. I 173 Ll. CL i_ a) CD -4-' i E O — L. O > O 5 O => ? > ■*-* “O -4—' c ^

D CL A2

FIGURE

(see

network

SASTN

for

except

regenerator

diagram

Circuit

A2.1

FIGURE 174

IvW^oo UJ CD P. w i—i p o •H •H TO rC £ o -p ■w| w c p •H p o T3 J o , i—1 o o 0) o p o a) P P p •H X o •H JC

— 3 o , o >* i—i Q) a) — Q O p 4-> C/D X

j w pJ bo P p bC X •H o p (U fr-> CP FIGURE A2.2 Self Adaptive Sample Time Network (SASTN) w PC. 1 r X X o •H 4h 175

APPENDIX 3

This appendix contains schematic block diagrams of the simulation control and test networks which were used for this investigation. Each separate test configuration is shown together with the connections required to enable that specific test to be performed.

All digital integrated circuits were RTL devices.

Fairchild yL 900 Buffer Amplifier

Fairchild yL 914 Dual-input Gate (Monostable Multivibrators, Trigger-Circuits,

Gates)

Fairchild yL 923 JK Multivibrator (Shift Register Delay Line, Frequency Divider) Regenerator 1

Data Input

Line 2 Simulator Test Control Line 1

Quench

Clock Initial RFInput Regenerator 2 Test Word , Patterr Ext.Synch. GR Word Generator Tx. Osc.

FIGURE A3.1 Basic simulation loop network interconnections Envelope Detector

Schmitt Trigger

Quench Input if Inhibit Regenerator 2 Data Input if Regenerator 1

Inverter

to SASTN

to Modulator

Trigger Pulse

FIGURE A3.2 Modification to regenerators to enable initial Data Word

Input entry and Quench control functions to be performed. 178

CO o • 1 CP 03 P CO rC P P p. O 03 o G X cn CP G p bo O G O' C P CP 1 •H 0) p CP o P G cp G CP 03 CP 03 CO i—1 G ip p P CP X O o e> CP o G •H Ci in •H X i—1 CP o G CX rQ CP o CX e p: X 03 p E-< CP CP c O E G P O X • X) G P

. rC > i—1 C g •H CO PC • a) P P X P r—1 0 CO *P P •H (P G < e > > O (P o P CP g E 03 • I-1 cx CP G CO rQ G CP CP 03 CO G co P CP G CP 1—1 CO > 0 bL p G o P CP a. O E 03 CX 6 CP G G rQ 03 O rC •H •H o O x: > > i—i G o •H o CP •H p cx P rG H o CN o 3 P o E 1—1 G O CM o P CP G -H G i—1 CP 03 X O rQ P G CP P 03 G 0) •H 03 P CP G •—1 G CO a) cx CP O o tx. cx C C p 03 03 CP O cx hi E X CP CP G co CX X O rQ 1—1 G 3 p 0) O G cx •H 3= 3 CP > CP > X P G •H CP CX O g P O CP CP X 03 o 1—1 P •H 1—1 CP CP G P CO i—1 o G G CX p CP O CP E 03 G •H rC o G CP P p a CP cp 03 G Mh rH r> CP (U •H P CO G O X) E p o •H CP X CO CO (0 GG CP CP o 3s o G i—i p rQ CO rC 03 G X o G • • O CP G CP CP •H •H CP o P i—1 P X c 03 CX o G CP G cx 03 p CP 03 G cr1 G CP P (P CP CO rC P C/3 Wj •H 3 CX

FIGURE A3.3 Simulator Test Control Unit; Quench, initial word selection

and number o^ regenerations. o

o o £ _c o c

Q_ 9-

Regenerator u o LlI O

FIGURE A3.4 Gated "Exclusive-ORM error count network. Gating required

to prevent false counting due to misaligned word patterns. 180

Regenerator Low under test. Foss Clock Filter CR0.& r.m.s. meter, Set-Reset Bistable M.V.

Tx. Osc.

FIGURE A3.5 Timing Jitter Analyser (digital phase comparator) FENNEY and WILHELM : Preview/Edit Controller for Electronic Editing of Videotapes

A pnp/npn pair was adopted as such a combination allows electronic edits on videotape to be previewed and san be shown to have a lower output impedance than provides a means of fast sequence editing. >he more common Darlington connected emitter pair, Although the unit has been specifically designed to n this circuit (fig. 7), SC25 conducts the majority of the operate with Ampex VR1000C and VR1100 videotape mrrent and the circuit configuration adopted provides recorders equipped with Electronic Editors and Switch or the collector of this transistor to be at earth potential. Kits, it should not be difficult to modify the equipment The output impedance is approximately l/gTO /?, where to operate with VR2000 machines also equipped with rm is the mutual conductance of SC26 and ft is the current Electronic Editors. rain of SC25. The Controller provides many of the facilities of Editec Short circuit current is limited by R80, which in the case at very low cost but it is not intended that it replace of a long-term short circuit will burn out. Editec as it does not provide for frame by frame ani mation. 3.6.2 Mechanical Construction The unit has been designed to replace the vision monitor panel mounted on the overhead bridge assembly of Acknowledgment Ampex VR1000C and VR1100 videotape recorders. It The author would like to thank Mr. R. Carden of the will be seen from fig. 1 that the left hand side of the videotape staff at ABN for his valuable suggestions and controller unit has two rows of monitor switches which assistance in satisfactorily integrating the unit to our provide the same switching functions as the unit it videotape recorders. Also, thanks are due to Mr. K. replaces. Parkyn of the A.B.C.’s Engineering Laboratory for the mechanical layout and construction of the prototype. 4. Conclusion The authors also express their appreciation to the A Preview/Edit Controller has been designed, based on Controller of Technical Services, A.B.C., for giving appro original work undertaken in the Hobart studios, which val for this paper to be published.

July, 1970 Proceedings I.R.E.E. Australia 211 An Infrared, Solid-State Optical Communication System

W. CARROLL, Member I.R.E.E.* and K. PORONNIK, Member I.R.E.E.f

Summary The effects of atmospheric disturbances upon the trans A feasibility study of a completely solid-state, room temperature mission channel characteristics are such as to produce operated, optical communication system suitable for moderately intensity (amplitude) variations or scintillation, which high speed digital data transmission has recently been completed. would distort conventional amplitude modulation of the A discussion of the important design parameters and system compromises is presented together with results of meteorological optical beam, thus pulse-time modulation techniques were effects upon the service reliability of the link systems. chosen as being the most suitable for this investigation. 1. Introduction The pulse modulation system offers other advantages, especially, in this case, with internal modulation of the The particular objectives of the study were : light source being applicable. Because power is only 1. Investigation of commercially available light- needed to generate pulses during pulse time, a modulation emitting diodes and photodiode detectors. efficiency greater than that obtainable with analog tech 2. Setting up of a simple optical link to study trans niques is achieved. Also non-critical modulator and mission characteristics and reliability over a path receiver circuitry may be employed because of the easing of the order of one half mile. of the linearity requirements. For the same available optical power, pulse systems offer a direct trade-off in 3. Study of modulation methods with particular em terms of peak-power and duty cycle versus longer path phasis on pulse and digital signals. lengths. In this project it was the intention to concentrate on solid-state devices suitable for continuous operation at 2. Solid-state Device Characteristics room temperatures and for this reason the gallium arsenide electroluminescent infrared light-emitting diode was 2.1 Light Emitting Diodes chosen as the source.1-3 Although solid-state emitters A typical GaAs e.d., operating continuously at room are available which produce coherent light at room tem temperature will radiate incoherent energy at a wavelength peratures, they cannot be continuously operated ; a duty of 9000 ± 200 A with a possible spread, at the 3 dB cycle of 0.02 percent is typical together with a maximum points, of 200 to 600 A. The radiation, without the use “ ON ” time of 0.2 microseconds. The spectrally- of optics, spreads out with a conical apex angle varying matched solid-state detector for the GaAs e.d.J radiation source is the silicon photo-voltaic diode.4-7 Although 1. Minden, H. T., “ Gallium Arsenide Electroluminescent and the emission source produces incoherent radiation, the Laser Diodes ”, Semiconductor Products, Vol. 6, No. 8, August 1963, p. 34. information presented may be applied to coherent (lasing) 2. Lamorte, M. F. and Liebert, R. B., “ P-N Junctions as Radia sources without loss of generality ; the only major differ tion Sources ”, Electronics, Vol. 37, No. 20, July 13, 1964, p. 61. ence is that, with incoherent sources, very low f-number 3. Lorenz, M. R. and Pilkuhn, M. H., “ Semiconductor-diode optics are required to make up for their inherent beam Light Sources ”, I.E.E.E. Spectrum, Vol. 4, No. 4, April 1967, divergence. p. 87. 4. Di Domenico, M. and Svelto, O., “ Solid-State Photodetection : *Postmaster-General's Department, Sydney, on study leave at A Comparison Between Photodiodes and Photoconductors ”, School of Electrical Engineering, University of New South Wales. Proc. I.E.E.E., Vol. 52, No. 2, February 1964, p. 136. fSchool of Electrical Engineering, University of New South 5. Van Vliet, K. M., “ Noise Limitations in Solid-State Photo- Wales. detectors ”, Applied Optics, Vol. 6, No. 7, July 1967, p. 1145. Manuscript received by The Institution September 29, 1969. 6. Lucovsky, G. and Emmons, R. B., “ High Frequency Photo Revised manuscript received by The Institution February 6, diodes ”, Applied Optics, Vol. 4, No. 6, June 1965, p. 697. 1970. 7. Johnson, K. M., “ High-Speed Photodiode Signal Enhance U.D.C. number 621.396.49 : 535. ment at Avalanche Breakdown Voltage ”, Trans. I.E.E.E., JGallium arsenide electroluminescent diode. Vol. ED-12, No. 2, February 1965, p. 55.

212 Proceedings I.R.E.E. Australia July, 1970 CARROLL and PORONNIK : An Infrared, Solid-State Optical Communication System

•om 100 tf 10 million bits per second. The d.c. power Except for the very large area diodes (area greater than nput to optical power output conversion efficiency is 0.1 cm2) the pulse response characteristics of the detectors Ipproximately 1%. can be made to match the modulation speed of the GaAs The outstanding advantage of these light-emitting emitters by appropriate choice of detector load resistance liodes is their ease of modulation by simple current control. and reverse bias. To obtain microwave demodulation The optical power output bears an almost linear relation capabilities some detecting diodes have light sensitive ship with the forward current flowing through the emitter areas less than 0.5 mm2. This presents a severe optical liode. When designing a modulator circuit it should be matching problem and, since the GaAs emitters are not observed tihat the emitter diode presents a non-linear low capable at room temperature, of being modulated at micro- mpedance to that circuit. Also, the d.c. resistance of wave frequencies, a compromise between detector active bhe emitter diode decreases as the junction temperature area, frequency response and optical matching can be rises and tlhis can cause thermal runaway unless a current readily made. mpply bias configuration is utilised.8-10 This is easily Manufacturers of light detecting diodes generally achieved by connecting a low value resistor, whose resist employ the dark current shot noise as the criteria for ance is 3 to 10 times the emitter diode resistance, in series calculating the minimum optical signal power that can with the diode, supply voltage and modulating (switching) be detected but, as is the case for the background radiation transistor. shot noise, this noise level is less than the thermal noise Any temperature variations will produce a positive generated by the particular diode load resistor. The temperature co-efficient shift in the peak wavelength of calculations should be based upon the equation which approximately 3 A per degree Centigrade. This wave gives that input optical power irradiating the detector length shift is important whenever an optical bandpass which produces a unity s.n.r. including both thermal filter is fitted to the receiving system. and shot noise components ; this is the “ noise equivalent : Selection of the optical transmitter must take into power ” (n.e.p.) : account tlhe relationships which exist between optical ^ ^ p (Thermal noise current2 + Shot noise current2)* power output, beam spread and emitting area. This Responsivity relationship is termed radiance and is probably the least understood and appreciated fact which governs the An alternate figure of merit commonly employed is the design of optical link systems (see appendix). detectivity D*. (Diode light sensitive area)* Optical power output Radiance N.E.P. tt [“Beam spread angle-]2 Emitter area X —------4 L 57.3 J (W cm-2 steradian-1) 3. Link System 3.1 System Description 2.2 Light Detecting Diodes The optical mounting frames used in the investigation There exists a wide range of silicon photovoltaic diodes were designed to enable the capsule containing the in pn, pin and avalanche type junction structures which light emitting or detecting diodes with their associated provide a correct spectral match for the GaAs emitters. electronic circuitry to be placed at, or near, the focus of The silicon diodes have a spectral response region covering the 12 inch diameter, 4| inch focal length, parabolic the wavelengths from 4000 to 11 000 A, thus the receiving mirrors (fig. 1). These removable capsules permit a quick system has a relatively wide response to background radia change from one type of measurement to another. The tion. The responsivity at 9000 A is approximately 0.4 amps yoke assembly has been dimensioned to allow the mounting per watt, the peak occurring at 8000 A with 0.5 amps of telescopes used for alignment purposes, as well as an per watt. image converter tube for visual studies on the infrared Optical bandpass filters can be employed to reduce the focussed spot or a photomultiplier tube with associated undesired pick-up of background energy and associated infrared bandpass filters and aperture stops. shot noise but, in all the experiments conducted on the To achieve the compactness necessary to mount the link systems, it was observed that the noise level at the electronic circuitry near the focus, wide-band linear receiver output was primarily due to thermal noise being integrated circuits were employed. All available inte 8. Bonin, E. L., “ Drivers for Optical Diodes ”, Electronics, grated circuits had poor noise figures, typically between Vol. 37, No. 22, August 10, 1964, p. 77. 8 to 12 dB. It was also necessary to maintain specific 9. Everest et al., “ The Generation of High Current Pulses of Short Duration for Use with GaAs Lamps and Lasers ”, impedance matches to avoid further degradation of “ Lasers and their Applications ”, I.E.E., 1964 (London). s.n.r. and prevent spurious oscillations. To overcome 10. “ Pulse; Power Supplies for Gallium Arsenide Injection these problems, interface stages were designed using low Lasers ”, R.C.A. Application Note ODL-100, September 15, 1966. noise, wide-band transistors (RCA 2N3478) and discrete

July, 1970 Proceedings I.R.E.E. Australia 213 CARROLL and PORONNIK : An Infrared, Solid-State Optical Communication System

wide angles enable simplified alignment but produce unwanted losses and background radiation pick-up. 3.3 System Optics The range of available large aperture, low f-number op tics is very restricted and generally is in the form of reflect ing rather than refracting elements. High quality surface specifications introduce cost factors that would outweigh other advantages of the system and so a compromise choice was made. It was found that the optical imperfec tions in the low cost, readily available, back-surface silvered, parabolic mirrors employed in carbon-arc pro jectors could be tolerated for distances up to one mile.’ The major difficulty encountered with the mirrors used in the experiments is that more than one spatial focal position occurs, resulting in a variation of at least 20 dB in the received signal level depending upon which focus is encountered. The multi-focussing can be attributed to sections of the relatively large surface area of the non- ideally parabolic shaped mirrors, each section with its own and separate focus point, collecting sufficient energy

Figure 1.—Mirror frame, yoke assembly with capsule and tripod mount used for either transmitting or receiving the optical signal.

components which then provided the desired impedance levels and reduced the overall noise figure to approxi mately 3 dB for the 1 MHz bandwidth employed in the link system. The total cost of materials and components for the two terminal (single-hop) one way link system was $A340. The physical size of each terminal, ignoring the tripod support stand, is less than 1 cubic foot. The total power consumption less than 1 watt. Figure 2.—Tripod assembly with mirror and capsule removed, arranged for initial visual alignment. 3.2 System Alignment The initial alignment of the system is performed visually which lifts the signal level above the output noise. The and is done by removing the mirror and capsule from the final focus position employed is actually coincident with mounting frame, inserting a telescope into the yoke, the circle of least confusion and, if the detector employed in this case an 8 times magnification cross-hair telescope has a light sensitive area which is equal to or greater than, was used (see fig. 2), and moving the appropriate horizon the confusion circle, no power should be lost clue to this tal and vertical adjustments until the image of the mirror effect. No facilities were available to measure the optical assembly at the opposite end of the proposed link path imperfections (surface tolerance and shape). is located centrally. The telescopes at each end are then The non-ideal parabolic shape of the mirror leads also removed, the parabolic mirrors mounted and the emitter to increased spreading of the transmitted beam above the and detector capsules inserted at their approximate focus theoretical size-limited beam divergence, a (radians), position. The system is then energised and final adjust originating from the finite area size, AE , of the emitter ments are made on the focus, horizontal and vertical direc and the focal length, fT , of the transmitter optics; where tions to obtain maximum signal at the receiver output. Due to the distance between the link terminals and the A E one-way nature of transmission, focussing was accomp lished with the help of instructions by portable radio Experimental evidence has been obtained which shows transmitters. The transmitter and receiver beam diver that the signal power loss due to the increased beam spread gence and acceptance angles were measured and found to is less for the relatively larger area emitters and thus a vary from 12 to 50 milliradians depending upon which compromise between optical quality, emitter area and emitter or detector was being used. These relatively radiated power can be made. To illustrate this, the

214 Proceedings I.R.E.E. Australia July, 1970 CARROLL and PORONNIK : An Infrared, Solid-State Optical Communication System

“ollowing example is given. Fig. 3 shows the comparison infrared communication systems. The test site, the ►etween a small area (0.035 mm2) TIXL06 emitter and a campus of the University, is in a residential area of Sydney arger area (0.5 mm2) GAL2 emitter. The main point to which is not subjected to any extreme climatic effects ; onsider is that when these emitters were both used in the rain and fog being the principal factors studied. Although ame optics, the 22 dB theoretical advantage of the small meteorological investigations have been conducted on rea TIXL06 is reduced to an experimental difference optical communication links by other research groups,11-15 f 12 dB. However, fig. 3 does not indicate that the little data on meteorological effects has been published 1AL2 is capable of radiating 10 mW of optical power as for the 9000 A system, the majority of workers concen ompared to the smaller TIXL06 radiation of 1.2 mW. trating on the coherent wavelengths of 6 300 and 106 000 A. Another factor of interest is scintillation which is produced by temperature gradients along the optical path. These gradients are modified by wind velocity [ and produce unwanted amplitude modulation of the optical beam, the frequency of variation extending up to /x'i2dB 2 kHz, with depths of modulation exceeding 80%.16-19 To analyse these latter effects, micro-meteorological equipment of fast response times would be necessary. In general meteorological test equipment which was available for the project was of an integrating type and was only useful for evaluating relatively gross, slow changes. It RANGE (FEET) must be noted that during the period of operation of the figure 3.—Comparison of theoretical and experimental signal link, exceptionally dry weather persisted with only a attenuation for TIXL06 and GAL2 emitters. little rain and fog in the first weeks of the experiment. This optical power difference amounts to a 18 dB signal The results which are presented below were obtained jower difference at the output of the silicon photodiode from continuous recordings made from July to November, letector, thus the GAL2 has a net 6 dB gain for our par- 1968 (see fig. 4 for typical recordings). iicular optical system. Calculations for radiance, N, (a) Rain jased upon ideal optics gives the TIXL06 an N of For path lengths of approximately 0.5 mile, heavy [100 mW cm-2 steradian-1, the GAL2 an N of 640 mW rain will produce signal attenuation up to 20 dB. im-2 steradian-1, indicating that the TIXL06 should A heav}7- drizzle or light rain up to 6 dB, with a light lave been the superior device by approximately 4 dB. drizzle producing up to 3 dB. On the average 1.4 Optical Link. Paths during a wet day, the signal attenuation would The choice of suit able link sites and paths was governed vary between 3 and 6 dB. ly factors such as ease of access, availability of power (b) Fog Ind height of surrounding buildings with reference to the The one factor which most seriously limits the fuildings used. For topological reasons (many high-rise service reliability at 9000 A is fog. A severe fog, Holdings in a confined area) it was generally necessary to one in which visibility is reduced to less than 1000 srect the transmitter and receiver at roof-top level, vhich suited the nat ure of the investigation. Path lengths 11. Chu, T. S. and Hogg, D. C., “Effects of Precipitation on )f 100, 200, 600, 900 and 1320 feet (0.25 mile) and 0.8 mile Propagation at 0.63, 3.5 and 10.6 Microns ”, Bell Syst. Tech. J., vere employed, with the main body of investigation con Vol. 47, No. 5, May-June 1968, p. 723. 12. Goodwin, F. E. and Nussmeier, T. E., “ Optical Heterodyne sentrated on the 0.25 and 0.8 mile paths. Owing to Communication Experiments at 10.6 Micron ”, Hughes .ransmission losses caused by dew condensing on the out- Research Labs., Malibu, California, U.S.A. ikle unmounted mirrors, which could be as high as 12 dB, 13. Knestrick, G. L. and Curcio, J. A., “ Atmospheric Propagation of Laser and Non-Laser Light ”, Applied Optics, Vol. 6, No. 8, t was found necessary to install the mirrors in special August 1967, p. 1420. veatherproof huts or behind windows in a laboratory. 14. Reisman, et al., “ Comparison of Fog Scattered Laser and ‘ Monochromatic ’ Incoherent Light ”, Applied Optics, Vol. 6, £ach window glass introduced an optical power loss of No. 11, November 1967, p. 1969. |0% which represents an approximate signal power loss 15. Arendt, J. R. M., “ A Summary of Papers Presented at the if 3 dB. Conference on Atmospheric Limitations to Optical Propaga tion ”, Radio Science, Vol. 1, No. 3, March 1966, p. 405. It is important for any east/west paths that the angle 16. Ryznar, E., “ Dependency of Optical Scintillation Frequency >f declination of the sun does not come within the field of on Wind Speed”, Applied Optics, Vol. 4, No. 11, November 1965, p. 1416. riew of the optics, otherwise thermal destruction of the 17. Subramanian, M. and Collinson, J. A., “ Modulation of Laser smitter or detector will result. Beams by Atmospheric Turbulence ”, Bell Syst. Tech. J., Vol. 44, No. 3. March 1965, p. 543. !.5 Meteorological Effects 18. Hogg, D. C., “ On the Spectrum of Optical Waves Propagated Through the Atmosphere ”, Bell Syst. Tech. J., Vol. 42, Optical signals suffer attenuation in propagation on No. 6, November 1963, p. 2967. .ccount of the same factors which affect visibility 19. Hohn, D. H., “ Effects of Atmospheric Turbulence on the tamely, fog, rain, snow and industrial smoke. These Transmission of a Laser Beam at 6328A : I—Distribution of Intensity, II—Frequency Spectra ”, Applied Optics, Vol. 5, actors must be considered when evaluating open-air No. 9, September 1966, p. 1427.

July, 1970 Proceedings I.R.E.E. Australia 215 CARROLL and PORONNIK : An Infrared, Solid-State Optical Communication System

yards, will attenuate transmission on a half-mile Dependent upon which noise source predomirates, the link by more than 40 dB, causing signal “ black optical link is said to be operating in that particular noise- out Fig. 4 shows the effect of heavy fog on the limited environment. The additional noise proluced by experimental link ; the record is, however, comjdi- the avalanche mechanism and diode defects is accounted cated by the effect of condensation as the trans for by allowing the exponent, n, of the mult.plication mitting mirror was not at that time protected. factor, M, in the shot noise terms to vary in fhe range 2 < n < 5, dependent upon the particular diode in use.i Ideally n should equal 2 otherwise further degradation of the s.n.r. will result. For the wide frequency bandwidth (B) operation employed for the digital transmission system (B = 1 MHz), resistive loading of the detector diode is employed* to avoid the detector diode RC time constant limitations. The signal current, Is, developed by the detector diode from the incident optical power, Ps, is

Is = p PSM hv where r\ is the quantum efficiency, e is the electronic charge, h is Planck’s constant, v is the optical frequency, M is the avalanche multiplication factor, I,, is the peak signal current, P, is the peak optical power. The peak signal power, S, developed across the load! resistor, RL , becomes Figure 4.—Typical recordings obtained during period of observation, showing the effects of weather on the 0.8 mile path. S = Is2Hl (a) fine day. (b) transmission attenuated by smolce. = (£)‘r,‘M‘RL (c) transmission attenuated by sudden rain showers on a stormy spring afternoon. (d) complete loss of transmission due to early morning fog. Consideration of the noise sources reveals that the! thermal noise power, with RL not a conjugate match foil Statistics on meteorological conditions for the the diode, is Sydney area, given in the 1967 Commonwealth of Nthermal = 4kTB Australia Year Book indicate that the average where k is Boltzmann’s constant, number of the above type of foggy days per year, taken over a 44 year period, is 17.8 days. This T is the absolute temperature, gives rise to a service reliability figure of 99.5% B is the system bandwidth. due to fog interference. Before providing the s.n.r. equation which includes all the noise contributions, the separate issues surrounding! (c) Smolce the interpretation of the different shot noise sources will Although the area is relatively free of industrial be further developed. smoke an occasion did arise when, due to bushfire smoke, the visibility was reduced to 1 mile, pro (a) Background Radiation (P^) ducing signal attenuation of the order of 4 dB. Dependent upon the spectral response of the detector] the problem of background radiation collected by the 4. Signal-to-Noise Analysis detector diode and receiver optics field of view will vary, In this section a signal-to-noise analysis for direct For wavelengths greater than 15 000 A, severe absorption photon-detection of a pulse modulated optical carrier occurs at discrete wavelength bands giving rise to regions including avalanche multiplication effects in the detector termed “ atmospheric windows ” where this absorption photodiode is given. The received signal must be detected does not occur. Below 20 000 A the wavelengths are short in the presence of noise originating from the following enough to enable both Rayleigh and Mie scattering to main sources : occur, thus sending radiation into the receiver’s field ol (1) Thermal noise in the detector load, R, , (amplifier view. This difference between absorption and scattering noise is treated separately as noise figure). is significant because, for wavelengths greater thar 15 000 A the difference between day-time and night-time (2) Shot noise from (a) background radiation, background radiation is relatively small when compared (b) dark current, to the scattering process differences between day and (c) signal radiation. night evident belowT 20 000 A.

216 Proceedings I.R.E.E. Australia July, 1970 CARROLL and PORONNIK : An Infrared, Solid-State Optical Communication System

The use of 9000 A as the emission line source locates the If this occurs, R7 should be made as large a value as stem in a spectral region subject to scattering and, to possible, within the restrictions of system bandwidth rovide a worst case value for background radiation, the requirements, otherwise unnecessary degradation of the ay light sky should be used as the radiance source20 s.n.r. will result. — 10~4 W steradian-1 ft-2 A-1). The background radiation power, PB , collected by the eceiver is calculated then as

Pb = Nb . Qp . Ar . A\ rhere is the aperture area of the receiver, AX is the spectral response region, including optical filtering,

Qf is the receiver field of view, n (Detector’s light sensitive area) ~ 4 (Focal length of optics)2 The average current, IB , and its associated shot noise urrent component, in6 , produced by this background adiation is OPTICAL SIGNAL POWER Ps Figure 5.—SB/N plotted against optical power, Ps, as a function ye of the combined noise-limited condition, for the L4503 detector diode. Is = —— Pb M h v Figs. 6 and 7 are presented to show an actual signal X72 = 2e Pb] M« B which was transmitted and received over the 0.8 mile path. A TIXL06 emitter was driven by a 0.7 amp peak b) Dark Current (1^) pulse, with a 6 jus pulse width, the peak optical j>ower The shot noise associated with the detector diode’s emitted being 1.5 mW. A Philco L4503 detector oper lark current is ating into 5 kf2 load amplifier (input impedance) produced = 2e ID M« B a 13 mV peak signal above the 0.8 mV r.m.s. thermal c) Signal Radiation (Ps) Because of the quantum nature of the optical signal adiation, P9 , collected by the detector, associated with he desired signal current, Is , is a shot noise component, r rte Is = ~ Fs M hr

L? = 2e Q? Ps) M» B

The combined s.n.r. equation then becomes

M2 -M2P52Rb S.N.R. o Figure 6.—Emitter current of the TIXL06 used as transmitter on the 0.8 mile path. Pb + Ib B Rz, + 4kTB 230 mA/div. vertical [(S) (£)Ps]M' 20 p s/div. horizontal For all the high speed silicon photodiode detectors ested, the dark current Ic component was insignificant. )epending upon the various values of P?, R7 and M he system will be found to operate in different noise- mited environments. These conditions are depicted in g. 5 where the characteristics of the Philco L4503 diode re utilised to present the s.n.r. per unity bandwidth 3B/N) as a function of P5 and R7 (M = 1). To convert he SB/N ratio to include the actual system bandwidth, B, 0 log B has to be subtracted from the vertical axis read- ig. It should be observed that although the shot noise n.r’s are independent of R7 , the value of P5 and Rz kely to be experienced with wide-band networks, the ystem will operate in the thermal noise-limited region. Figure 7.—-Signal received by L4503 detector over the 0.8 mile path. 5 mV/div. vertical ). Monte R., “ Laser Receivers ”, Wiley (p. 277). 20 p s/div. horizontal

July, 1970 Proceedings I.R.E.E. Australia 217 CARROLL and PORONNIK : An Infrared, Solid-State Optical Communication System

noise. This s.n.r. of 24 dB was subsequently improved computer paper tape and card readers, decoupling net to 34 dB by increasing the current drive into the emitter works, level indicators, photographic film annotation diode before any overheating effects became noticeable in and as high-speed light sources for testing detectors. the transmitter modulator circuits ; the emitter diode It is hoped that the above discussion has outlined suffi was not affected provided the duty cycle was controlled ciently the engineering problems associated wilh optical to maintain its average power below the rated maximum. link systems and indicated the compromises permissible in arriving at a suitable design configuration.

5. Conclusion Acknowledgments The feasibility of utilising commercially available solid- The authors wish to thank the University of New South state components for an optical communication link Wales for permission to publish the paper and Professor system has been established. As is the case for all open A. E. Karbowiak for his advice and criticism air transmission systems, the service reliability is affected The project was financially assisted by a grant from the by the weather, fog being the most severe attenuating Australian Research Grant Committee and a contract condition. Quantitative measurements, coupled with pro from the Postmaster-General’s Department. jected meteorological conditions, indicate that the signal at Thanks are also due to Dr. E. H. Fooks for helpful tenuation for a 0.5 mile optical link system on 9000 A discussion and to Mr. D. Irving for his technical assistance. wavelength, located in climatic conditions similar to that of Sydney, is 6 dB or less for 98% of the year and 20 dB Appendix or less for 99% of the year. The system was successfully A simplified derivation, assuming ideal optics and loss-! tested over a 0.8 mile path using one single channel of the less transmission medium, is presented so that the relation Philips 24 Channel Delta Modulation Telephone system. ship between source radiance (brightness) and received Subjective tests over the one-way optical transmission optical power can be examined (fig. 8). and automatic telephone network connection revealed no impairment of audio signal quality due to optical link. From the results there is every indication that all 24 Transmitter channels could have been transmitted. These results Optics are especially encouraging when comparison26 is made Emitter with tests conducted on a 12 channel (60 to 108 kHz) Detector analog system over a path length of 500 metres utilising similar optical components. Because of the principal advantages of this optical system for short-haul paths, such as low cost, portability, low power consumption and ease of installation there must be many applications21-26 where the short-coming Figure 8.— Basic diagram of an optical link sysbm. of weather dependency need not be a major obstacle. The future of free-space optical communication links List of Symbols depends to a large extent on the developments in solid- N Source radiance (W cm-2 steradian-1). state emitters and detectors. Devices of much higher Pr Transmitted optical signal power (watts). efficiency than those used in this research are certainty possible ; present devices operate nowhere near the P/; Received optical signal power (watts). theoretical limit and it is estimated that more than 20 dB Ae Emitter source area (cm2). could be gained with reasonable ease following improve 0 Cone half-angle subtended by transmitter optics at ments to manufacturing techniques. This is understand each point on source (radians). able since the bulk of devices produced commercially so far are not specifically designed for communication pur Solid angle subtended by transmitter optics at each poses but for such purposes as lamps for optical encoders, point on source (steradians). Dr Diameter (aperture) of transmitter optics (an). 21. Shah, B. R., “ Experimental Optical Links ”, I.E.E.E. Int. Communs. Conf., 1966, Philadelphia, Optical Society of fr Focal length of transmitter optics (cm). America, March 1966. D^ Diameter (aperture) of receiver optics (cm) 22. Boerschig, B. A., “ A Light-Modulated Data Link ”, Trans. I.E.E.E., Vol. BC-10, No. 1, February 1964, p. 4. ty Focal length of receiver optics (cm). 23. Davies, A. T., “ A Technique for the Transmission of Digital Information over Short Distances using Infrared Radiation ”, R Distance between optics (cm). The Radio and Electronic Eng., Vol. 29, No. 6, June 1965, p. 369. Flux density at range, R (W cm-2), 24. Gross, A. J. and Sarson, A. E., “ The Light Emitting Diode, its a Size-limited beam divergence angle (radians). Application to a Short Path TV Link ”, Wireless World, Vol. 70, No. 8, August 1964, p. 377. AR Cross-sectional area of beam at R (cm2). 25. Karlson, et al, “ Room Temperature GaAs Laser Voice Com munication System ”, Proc. I.E.E.E., Vol. 52, No 11, Novem Power transmitted = Source radiance X Soure area X ber 1964, p. 1345. Solid angle subtended ly trans 26. Williamson, W. J. and Tai, K. C., “ Optical Communications Using Semi-Conductor Lamps as the Transmitter Source ”, mitter optics at each point or Aust. Telecom. Research, Vol. 3, No. 1, May 1969, p. 10. source.

218 Proceedings I.R.E.E. Australia July 1970 CARROLL and PORONNIK : An Infrared, Solid-State Optical Communication System

PT — N X AE X Qe (1) 7r2 NDr2D*2 (10) 16 R2 here Qe = 277(1 — cos 9) (2) Thus for any range, R, with fixed optics, Dr and D^ , = 477 sin2 - the received signal power is directly proportional to the 2 emitter source radiance, N. The fact that Pw is indepen

~ 77 02 dent of the focal length of the transmitter optics, fr , may tend to be confusing when it is considered that the longer fT , the narrower the transmitted beam (equation 5). This should result in a higher flux density, , at any given range R. This latter condition is only satisfied 77 rDrl2 Thus Pr = NAe - — (4) if the transmitter optics collect all the emitted energy. 4 L fr J Whenever the aperture, Dr , is too small for the given fr , Due to the size-limited beam divergence, a, this power a “ spillover ” of the emitted radiation occurs. This rill spread out over an area, , at distance, R. mismatch results in a loss of radiation which exactly (A/Q» compensates for the anticipated increase in HR of the (5) fr narrower beam, that is, remains constant. When DT is large enough to collect all the emitted radiation no Then Ar = (aR)2 further increase in P^ due to increases in Dr will result. A£R2 li\\ Upper bounds also exist for the aperture of the receiver optics, Dfl . The flux density, , at distance, R, equals the trans- (a) When D^ equals the beamwidth at range R, no nitted power divided by the cross-sectional area at R. further increase in P^ can occur with D^ being made larger. P T Hfl = — (7) (b) It should be recognised that the detector’s field of A R view should be larger than the solid angle subtended at 77 N the detector by the receiver optics. Increase of =------Dt2 (8) 4 R2 results in an increase of the solid angle until it equals the detector’s field of view. Further increase of will The received optical signal power is, collect more optical power but no more will enter the Pfl = Hji X Area of receiver optics (9) detecting diode. For a fixed D/j , if a choice is available, the focal length, fw , of the receiver optics should be 77 N 77 =------Dt2 X - D/;2 made as long as possible so as to reduce the optical 4 R2 4 matching problems found with small photodiode detectors.

July, 1970 Proceedings I.R.E.E. Australia 219 Measurement of First-Order Probability Density Functions Using a General Purpose Analogue Computer

J. K. BARGH*

Summary 1. Kretzmer, E. R., “ Statistics of Television Signals ”, Bell Syst\ Tech. J., Vol. 31, No. 3, July 1952, p. 751. The growing application of statistical design techniques calls for 2. Davies, H. E. and Cooper, G. R., “ Direct Reading Probability adequate description of system signals and one valuable property Distribution Meter ”, Proc. I.R.E. Nat. Elec. Conf., Vol. 10 is the probability density function. While a number of analysers 1954, p. 358. for measuring probability density functions have been described in 3. Zoll, D. J., “ Simple Plotter Analyser for Radar Noise ” the literature, almost all require the construction of special purpose Electronics, Vol. 31, No. 11, March 14, 1958, p. 162. hardware. This paper describes a method of measuring first-order 4. Sullivan, A. W. and Well, J. D., “ Probability Computer foi probability density functions of stationary signals using standard Noise Measurements ”, Electronics, Vol. 30, No. 10, October 1, computing elements incorporated in modern analogue computers. 1957, p. 208. 5. White, H. E., “ An Analog Probability Density Analyser ” Tech. Rept. 326, Res. Lab. of Elec., M.I.T., April 1957. 6. Dray-son, M., “ An Amplitude Distribution Meter ”, Electronic 1. Introduction Engg., Vol. 31, No. 380, October 1959, p. 578. 7. Bell, D. A., “ Distribution Functions of Semiconductoi Numerous authors1-20 have discussed methods of Noise ”, Proc. Phys. Soc., Vol. 68B, No. 9, 1955, p. 690. determining the probability density functions of random 8. Lien, H., “ Probability Density Measurements with an Elec trode Mounted in the Face of a Cathode Ray Tube ”, Rev. Sci signals. Most of the analysers described have been single Instrum., Vol. 30, No. 12, December 1959, p. 1100. channel devices in which an amplitude window is scanned 9. Easter, B., “ An Electronic Amplitude Distribution Analy ser ”, S M Thesis, Dept, of Elec. Eng., M.I.T., 1953. through the range of interest. The proportion of the time 10. Knudtzon, N. H., “ Experimental Study of the Characteristics that the signal lies within the amplitude window for each of Filtered Random Noise”, Tech. Rept. 115, Res. Lab. ol setting is then determined using analogue or digital tech Elec., M.I.T., July 1949. 11. Davenport, W. B., “A Study of Speech Probability Distribu niques. Some multi-channel analysers have been de tions ”, Tech. Rept. 148, Res. Lab. of Elec., M.I.T., Augusi scribed17-20 but these have been expensive units to con 1950. struct. 12. Hoffman, D. and Schutzman, E., “ Statistical Analysis oi Noise Signal Amplitudes ”, Electronics, Vol. 32, No. 30 Although two of the analysers15* 20 have incorporated July 24, 1959, p. 48. 13. Jordan, K. L., “ A Digital Probability Density Analyser ”, standard analogue computer elements, they have required S M Thesis, Dept, of Elec. Eng., M.I.T., 1956. specially constructed hardware elements in addition. 14. Singleton, H. E., “ A Digital Electronic Correlator ”, Tech, The method described in this paper requires standard Rept. 152, Res. Lab. of Elec., M.I.T., February 1950. 15. Lampard, D. G. and Harvey, I. K., “ A Probability Distribu analogue computer elements only and the analyser can tion Analyser ”, J. Electronics and Control, Vol. 9, No. 3, be set up on any analogue computer having integrators, September 1960, p. 233. potentiometers, comparators, diode AND gates, track-store 16. Brubaker, T. A. and Korn, G. A., “ Accurate Amplitude Distribution Analyser Combining Analog and Digital Logic ”, units and logic mode control. Such elements are incor Rev. Sci. Instrum., Vol. 32, No. 3, March 1961, p. 317. porated in most modern analogue computers, including 17. Stoddard, J. C., “ Measurement of Second Order Probability Distributions of Pictures by Digital Means ”, Tech. Rept. 302, the smaller units. These machines are often available Res. Lab. of Elec., M.I.T., July 1955. in laboratories which do not possess a hybrid computer. 18. Schreiber, W. F., “ Probability Distribution Measurements ol Determination of probability density functions on hybrid Television Signals ”, I.R.E. Nat. Conv. Rec., Pt.4, 1953, p. 35. 19. Schreiber, W. F., “ The Measurement of Third Order Proba machines has been treated elsewhere.21 bility Distributions of Television Signals ”, Trans. I.R.E., Vol. IT-2, No. 3, September 1956, p. 94. 20. Bargh, J. K., “ The Analysis of Random Signals ”, Ph.D. ♦Senior Lecturer in Electrical Engineering, University of Thesis, Cambridge University, 1962. Canterbury, Christchurch, New Zealand. 21. Cameron, W. D., “ Hybrid Computer Techniques for Deter Manuscript received by The Institution, February 2, 1970. mining Probability Distributions ”, Electronic Associates Inc., U.D.C. number 681.33 : 519.2. Application Note 2.4.2h, 1965.

220 Proceedings I.R.E.E. Australia July, 1970