DOCSLIB.ORG

United States Patent (19) 11 Patent Number: 4,503,404 Racy 45) Date of Patent: Mar

Total Page：16

File Type：pdf, Size：1020Kb

Download full-text PDF Read full-text
United States Patent (19) 11 Patent Number: 4,503,404 Racy 45) Date of Patent: Mar United States Patent (19) 11 Patent Number: 4,503,404 Racy 45) Date of Patent: Mar. 5, 1985 54 PRIMED MICROWAVE OSCILLATOR cludes a single tank conductor (16) coupled to a cou 75 Inventor: Joseph E. Racy, Hudson, N.H. pling conductor (17) by an interdigitated coupler (26). The coupling conductor (17) is connected to the cath 73 Assignee: Sanders Associates, Inc., Nashua, ode of an IMPATT diode (22) which is triggered by the N.H. application of a back-biasing trigger pulse that biases it 21 Appl. No.: 473,173 into its negative-resistance region. When a keying pulse is applied to the IMPATT diode (22), the diode couples 22 Filed: Mar. 7, 1983 power through the interdigitated coupler (26) to the 51) Int. Cl. .......................... H03B5/00; H03B 7/00 tank circuit (16) to cause oscillations that are initially in 52 U.S. Cl. ............................... 331/96; 331/107 SL; phase with any incoming signals, but the frequency of 331/107 G; 330/287 the oscillations is determined by the configuration of 58 Field of Search ................. 331/55, 56, 96, 107 G, the tank circuit (16), not by the frequency of the incom 331/107 DP, 107 SL, 173; 330/286, 287 ing signal. If the incoming signal is near enough to the 56) References Cited resonant frequency, and if the duration of the keying pulses is short enough, the output of the primed oscilla U.S. PATENT DOCUMENTS tor appears to a band-limited receiver to be an amplified 4,056,784 11/1977 Cohn ................................... 330/287 version of the input signal. In another version, the am 4,259,647 3/1981 Chang et al. ............. 331/107 SL X plifying element is a GaAs FET (66) with a tank con Primary Examiner-Eugene R. LaRoche ductor (70) connected to its gate. The tank conductor is Assistant Examiner-Robert J. Pascal in turn coupled to a coupling conductor (62) for posi Attorney, Agent, or Firm-Louis Etlinger; Richard I. tive feedback at the resonant frequency of the tank (70). Seligman 57 ABSTRACT A primed oscillator for microwave amplification in 11 Claims, 4 Drawing Figures U.S. Patent Mar. 5, 1985 Sheet 1 of 2 4,503,404 2 - 777-77 77 . O - lyr a Nt I-V4)4-4444 4. FG. TO HD ANTENNA FG, 4. U.S. Patent Mar. 5, 1985 Sheet 2 of 2 4,503.404 4,503,404 1. 2 By virtue of the coupling from the matching circuit PRIMED MICROWAVE OSCILLATOR to the tank circuit, the amplified signal from the drain terminal is fed back to the tank circuit to maintain oscil BACKGROUND OF THE INVENTION lation. In this version, there is only one resonant circuit, The present invention is directed to microwave oscil so tuning is necessary for only one. The transistor lators. It has particular application to primed oscillators power voltage appears on the matching circuit, but it is employing low-impedance active elements. low enough that it can be blocked by an ordinary micro Primed oscillators are often used to provide high wave capacitor. levels of amplification for microwave signals. When the The invention is defined more specifically in the primed oscillator is activated, it will begin oscillation in 10 claims. phase with an incoming signal if the amplitude of the BRIEF DESCRIPTION OF THE DRAWINGS signal is sufficient and the coupling to the amplifying These and further features and advantages of the device is great enough. One of the difficulties encoun present invention are described by reference to the tered in the use of primed oscillators is that the impe 15 accompanying drawings, in which: dance of the amplifying element typically differs greatly FIG. 1 is a side elevational view, partly in section, of from the impedance of the antenna, and steps must be a stripline oscillator that follows the teachings of the taken to achieve the requisite coupling. present invention; The matching of impedances has been accomplished FIG. 2 is a plan view of the oscillator of FIG. 1 with by employing separate resonant circuits with different 20 the upper dielectric and ground-plane layers removed impedances. However, this has presented the problem to expose the intermediate conductor layer; that the two circuits have to be tuned in tandem during FIG. 3 is a plan view of another embodiment of the manufacturing. This tuning operation complicates the invention with the upper dielectric and ground-plane manufacturing process. layers removed to expose the intermediate conductor SUMMARY OF THE INVENTION 25 layer; and FIG. 4 is a schematic diagram of the oscillator of According to one version of my invention, a planar FIG. 3. transmission-line conductor is connected at one end to an amplifying element such as an IMPATT diode, DESCRIPTION OF THE PREFERRED which transfers power onto the conductor to cause 30 EMBODEMENTS oscillations at the resonant frequency of the conductor. FIG. 1 is a side view of the oscillator showing its The resonant conductor, which thus acts as the tank stripline construction. The oscillator 10 includes upper circuit for the oscillator, has a characteristic impedance and lower ground-plane conductors 12 and 14, respec that is relatively close to that of the amplifying device tively, between which generally planar conductors 16 for efficient coupling of power into the tank circuit. 35 and 17 are disposed, being separated from the ground To transfer power to a transmission line leading to an planes 12 and 14 by a dielectric layer 18. Signals travel antenna, a second planar transmission-line conductor is ing to and from an antenna (not shown) propagate by coupled to the first conductor. It is a resonant line so way of a coaxial cable connected by a coaxial connector that it can be used as an impedance transformer, and its 20 to ground plane 14 and intermediate conductor 16. characteristic impedance is different from that of the An IMPATT diode 22 is connected between ground oscillator tank conductor, having such a value that the plane 14 and intermediate conductor 17, and triggering second conductor can readily achieve an approximate pulses for the diode 22 are applied to it by way of a match of the characteristic impedance of the antenna conductor 24 and intermediate conductor 17. transmission line to the impedance of the diode. The layout of the intermediate conductors 16 and 17 The coupling of the two resonant conductors is pref. 45 is depicted in FIG. 2. Conductor 16 is elongated and erably produced by both edge coupling and capacitive relatively narrow. Its width ordinarily is chosen to coupling for transfer of power between the antenna and minimize its dissipation, i.e., maximize its Q. This nar the oscillator. The coupling between the two resonant row width, together with its thickness and spacing from circuits is at least critical. As a result, the oscillator can the ground planes 12 and 14, results in a relatively high be tuned during manufacturing by adjusting only one of 50 characteristic impedance. Because of the relatively high the resonant circuits, e.g., by adjusting the length of one impedance, conductor 16 can be tapped reliably to of the conductors. match its input impedance to the characteristic impe A different version of my invention also uses a planar dance of the coaxial feed line from the antenna. On the transmission-line conductor as a tank circuit. The tank other hand, conductor 17 is relatively wide, so it and circuit is coupled to a transistor control terminal, such 55 ground planes 12 and 14 form a transmission line having as the gate terminal of a field-effect transistor in a com a lower characteristic impedance. In the middle of its mon-source configuration. The tank circuit is edge cou negative-resistance region, IMPATT diode 22 has a pled to an impedance-matching circuit, which is also a negative resistance that is lower in magnitude than the planar transmission-line conductor, and which is con characteristic impedance of conductor 17 but closer to nected at one end to the transistor's amplifying ter 60 that impedance than to the characteristic impedance of minal-that is, to the drain terminal in the case of a conductor 16. common-source configuration. The characteristic impe At one end of intermediate conductor 16, which acts dance of the tank matches the input impedance of the also as an input tank circuit, is an interdigitated capaci transistor, while the matching circuit is a quarter-wave tor 26 comprising alternating fingers 28 and 30 on con line whose characteristic impedance matches the output 65 ductors 17 and 16, respectively. The capacitor 26 acts as impedance of the transistor to the characteristic impe a low-impedance coupler to the low impedance of the dance of a transmission line that leads from the match diode 22. A high impedance is presented by the other, ing circuit to the antenna. open end of conductor 16. The length of conductor 16 4,503,404 3 4. is approximately one quarter wavelength at a frequency imity of the arms of conductor 17 to the longitudinal within a range of frequencies to which the primed oscil edges 39 of conductor 16. This coupling widens the lator is intended to respond. Therefore, it acts as a reso bandwidth of the oscillator. Thus, the high degree of nant circuit. coupling, in addition to presenting a significant-ampli The center conductor of a coaxial feed line from the tude signal on conductor 17, increases the sensitivity of antenna is electrically connected to conductor 16 at a the oscillator to frequencies on the outer edge of the connection point 32 at which the impedance presented intended frequency range.
Load more

Recommended publications
  • Design and Analysis of Band Pass Filter for Wireless Communication Viswavardhan Reddy
    Volume III, Issue VI, June 2014 IJLTEMAS ISSN 2278 - 2540 Design and Analysis of Band Pass Filter for Wireless Communication Viswavardhan Reddy. K1, Mausumi Dutta2, Maumita Dutta3 1,2Department of Telecommunication Engineering, RV College of Engineering, Bangalore - 560059 [email protected], [email protected],[email protected] Abstract— There are applications in wireless communications, where a particular band of frequencies are needed to be filtered full sections can be used to increase the filter roll-off factor. from a wider range of mixed signals. Filter circuits can be Method [4] includes equal-ripple and maximally flat passband designed to accomplish this task by combining the properties of filters with general stopbands, as well as equal-ripple low-pass filter and high-pass filter into a single filter which is stopband filters with general passbands. To solve the referred as band-pass filter. A band-pass filter works to screen out frequencies that are too low or too high, giving easy passage approximation problem and to improve numerical only to certain frequencies within a specific range. At high conditioning, the design is carried out exclusively in terms of frequencies, the behavior of the discrete components changes. one or two transformed frequency variables. Hence discrete components are replaced by microstrip transmission lines. Microstrip transmission line is the most used II. LITERATURE SURVEY planar transmission line in Radio Frequency (RF) applications. A microstrip transmission line consists of a thin conductor strip The concept of coupling coefficients has been a very useful over a dielectric substrate along with a ground plate at the one in the design of small-to-moderate bandwidth microwave bottom of the dielectric.
    [Show full text]
  • Development of Planar Microstrip Resonators for Electron Spin Resonance Spectroscopy 2 Per Cladding Is Present on Either Side of the Substrate
    Development of planar microstrip resonators for electron spin resonance spectroscopy Preprint, compiled April 2, 2020 Subhadip Roy1, a, Sagnik Saha1, a, Jit Sarkar1, and Chiranjib Mitra1 1Department of Physical Sciences, Indian Institute of Science Education And Research Kolkata,India. Abstract This work focuses on the development of planar microwave resonators which are to be used in electron spin resonance spectroscopic studies. Two half wavelength microstrip resonators of different geometrical shapes, namely straight ribbon and omega, are fabricated on commercially available copper clad microwave laminates. Both resonators have a characteristic impedance of 50 Ω.We have performed electromagnetic field simulations for the two microstrip resonators and have extracted practical design parameters which were used for fabrica- tion. The effect of the geometry of the resonators on the quasi-transverse electromagnetic (quasi-TEM) modes of the resonators is noted from simulation results. The fabrication is done using optical lithography technique in which laser printed photomasks are used. This rapid prototyping technique allows us to fabricate resonators in a few hours with accuracy up to 6 mils. The resonators are characterized using a Vector Network Analyzer. The fabricated resonators are used to standardize a home built low-temperature continuous wave electron spin resonance (CW-ESR) spectrometer which operates in S-band, by capturing the absorption spectrum of the free radical DPPH, at both room temperature and 77 K. The measured value of g-factor using our resonators is consistent with the values reported in literature. The designed half wavelength planar resonators will be even- tually used in setting up a pulsed electron spin resonance spectrometer by suitably modifying the CW-ESR spectrometer.
    [Show full text]
  • Equivalent-Circuit Models for Split-Ring Resonators And
    IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 53, NO. 4, APRIL 2005 1451 Equivalent-Circuit Models for Split-Ring Resonators and Complementary Split-Ring Resonators Coupled to Planar Transmission Lines Juan Domingo Baena, Jordi Bonache, Ferran Martín, Ricardo Marqués Sillero, Member, IEEE, Francisco Falcone, Txema Lopetegi, Member, IEEE, Miguel A. G. Laso, Member, IEEE, Joan García–García, Ignacio Gil, Maria Flores Portillo, and Mario Sorolla, Senior Member, IEEE Abstract—In this paper, a new approach for the development expected for left-handed metamaterials (LHMs); namely, inver- of planar metamaterial structures is developed. For this pur- sion of the Snell law, inversion of the Doppler effect, and back- pose, split-ring resonators (SRRs) and complementary split-ring ward Cherenkov radiation. It is also worth mentioning the con- resonators (CSRRs) coupled to planar transmission lines are investigated. The electromagnetic behavior of these elements, as troversy originated four years ago from the paper published by well as their coupling to the host transmission line, are studied, Pendry [2], where amplification of evanescent waves in LHMs and analytical equivalent-circuit models are proposed for the is pointed out [3]–[6]. isolated and coupled SRRs/CSRRs. From these models, the In spite of these interesting properties, it was not until 2000 stopband/passband characteristics of the analyzed SRR/CSRR that the first experimental evidence of left-handedness was loaded transmission lines are derived. It is shown that, in the long wavelength limit, these stopbands/passbands can be interpreted as demonstrated [7]. Following this seminal paper, other artifi- due to the presence of negative/positive values for the effective cially fabricated structures exhibiting a left-handed behavior and of the line.
    [Show full text]
  • Static and Dispersion Analysis of Strip-Like Structures
    EE 694 Seminar Report on Static and Dispersion Analysis of Strip-like Structures Submitted by Ravindra.S.Kashyap 06307923 [email protected] Under the guidance of Prof. Girish Kumar Dept. of Electrical Engineering Indian Institute of Technology Bombay 1 CONTENTS I Introduction 2 II Static analysis 2 II-A The Conformal Mapping method . 3 II-B Variational approach . 4 II-C Finite Difference Method (FDM) . 6 II-D Losses . 8 III Dispersion analysis 9 III-A Spectral Domain Analysis . 10 III-B Method of Lines . 10 IV New Techniques for Microwave CAD 10 V Static Analysis for Various Microstrip line Configurations 11 V-A Tri-plateR . 11 V-B Microstrip Lines . 12 V-C Coupled Microstrip line . 14 V-D Suspended Microstrip Lines . 15 V-E Inverted Microstrip lines . 16 V-F Slot Lines . 16 V-G Coplanar Waveguide . 17 VI Dispersion Analysis for Various Microstrip line Configurations 17 VI-A Microstrip lines . 17 VI-B Coupled Microstrip Lines . 18 VI-C Suspended Microstrip lines . 19 VI-D Inverted Microstrip Line . 19 VII Conclusions 19 Appendix 19 References 20 2 Static and Dispersion Analysis of Strip-like Structures Ravindra.S.Kashyap (06307923) Abstract— Different models for the static and dispersion velocity (vp). These three parameters are related to the analysis of strip-like structures will be studied. We will be Capacitance (C) of the structure [2]. We know that studying, in particular, Tri-plate stripline, Micro stripline, the wave propagates through a medium with velocity p Coupled micro stripline, Inverted and suspended micro v = 1= ¹². If the medium is not free space but a striplines, Slot lines, and Coplanar waveguide.
    [Show full text]
  • Planar Transmission Line Method for Char- Acterization of Printed Circuit Board Di- Electrics
    Progress In Electromagnetics Research, PIER 102, 267{286, 2010 PLANAR TRANSMISSION LINE METHOD FOR CHAR- ACTERIZATION OF PRINTED CIRCUIT BOARD DI- ELECTRICS J. Zhang CISCO Systems, Inc. CA, USA M. Y. Koledintseva Missouri University of Science & Technology Rolla, MO, USA G. Antonini Department of Electrical Engineering University of L'Aquila Poggio di Roio, 67040 AQ, Italy J. L. Drewniak Missouri University of Science & Technology Rolla, MO 65401, USA A. Orlandi Department of Electrical Engineering University of L'Aquila Poggio di Roio, 67040 AQ, Italy K. N. Rozanov Institute for Theoretical and Applied Electromagnetics Russian Academy of Sciences Moscow 125412, Russia Corresponding author: M. Y. Koledintseva ([email protected]). 268 Zhang et al. Abstract|An e®ective approach to characterize frequency-dispersive sheet materials over a wide RF and microwave frequency range based on planar transmission line geometries and a genetic algorithm is proposed. S-parameters of a planar transmission line structure with a sheet material under test as a substrate of this line are measured using a vector network analyzer (VNA). The measured S-parameters are then converted to ABCD matrix parameters. With the assumption of TEM/quasi-TEM wave propagation on the measured line, as well as reciprocity and symmetry of the network, the complex propagation constant can be found, and the corresponding phase constant and attenuation constant can be retrieved. Attenuation constant includes both dielectric loss and conductor loss terms. At the same time, phase term, dielectric loss and conductor loss can be calculated for a known transmission line geometry using corresponding closed- form analytical or empirical formulas.
    [Show full text]
  • Copyrighted Material
    k 1 Introduction to Design Using Microstrip and Planar Lines 1.1 Introduction The microstrip line is the most widely used interconnect at radio frequency (RF) and microwave frequencies. The microstrip line, shown in Figure 1.1, is the main member of a broad class of transmission lines that are built using printed circuit board technology. Here the microstrip line is typically created starting with a low loss dielectric slab or substrate that has a metal sheet bonded to both sides. The strip of the line is formed by patterning the top conductor and etching away the unwanted metal. Sometimes the metal pattern forming k the strips is patterned by silk screening or by growing metal in the appropriate place. At RF, k microwave and high-speed digital frequencies it is necessary to provide a return current path as well as the signal current path defined by the strip. At low frequencies, below a few tensof megahertz, it is also necessary to provide a signal return path but then it is less critical that the cross-sectional geometries be precisely established. The cross-sectional geometry, the width, , of the strip and the thickness, h, of the substrate define the ratio of the voltage and current signals traveling along the microstrip line. This ratio is called the characteristic impedance of the line and it is critical for reliable signal transmission, that is, good signal integrity, that the cross-sectional geometry be the same along the line as then the characteristic impedance of the line is constant. When microwave engineers refer to microstrip design they are referring to the design of RF and microwave circuits using the major types of planar transmission line technologies.
    [Show full text]
  • AN-1014 Microstrip Vs Stripline: Crosstalk and RMS Phase Jitter
    Microstrip vs Stripline: Crosstalk AN-1014 and RMS Phase Jitter Application Note Introduction This application note discusses the effects of using different transmission line structures to minimize crosstalk. As higher data rate and lower phase noise are required, electromagnetic interference (EMI) on PCBs becomes more significant. On PCBs that use IDT Timing Devices, there can be multiple adjacent clock traces that have different frequencies. Without proper layout guidelines, current flowing down one trace (aggressor), can induce current on the other (victim), to cause unwanted coupling (crosstalk). Microstrip and Stripline Figure 1. Microstrip Microstrip is a planar transmission line, consisting of a conducting layer and a ground plane separated by dielectric. As there is no dielectric above the conducting layer, this structure is usually used at the top or bottom of the PCB. Figure 2. Stripline Stripline is another planar transmission line, consisting of a conducting layer and a pair of ground planes separated by dielectric. Comparison Due to its structure, microstrip is usually easier and less expensive to manufacture than stripline. On the other hand, compared to microstrip, stripline is better isolated due to shielding ground planes on both sides. Of the same characteristic impedance, stripline has a shorter trace width than microstrip, allowing for more densely packed traces. © 2018 Integrated Device Technology, Inc. 1 November 1, 2018 Microstrip vs Stripline: Crosstalk and RMS Phase Jitter S-Parameters Figure 3. S-Parameters: Two-Port Network a1 S11 S12 a2 b1 S21 S22 b2 S11 and S22 are the reflection coefficient, meaning the response at port 1 due to a signal at port 1 and the response at port 2 due to a signal at port 2.
    [Show full text]
  • Analysis, Design, Optimization and Realization of Compact High Performance Printed RF Filters
    Analysis, Design, Optimization and Realization of Compact High Performance Printed RF Filters Dissertation zur Erlangung des akademischen Grades Doktoringenieur (Dr.-Ing.) von M.Sc. Atallah Balalem geb. am 15. January 1978 in Nablus, Palestine genehmigt durch die FakultÄatfÄurElektrotechnik und Informationstechnik der Otto-von-Guericke-UniversitÄatMagdeburg Gutachter: Prof. Dr.-Ing Abbas S. Omar Prof. Dr.-Ing Marco Leone Prof. Ing Jan Machac, DrSc. Promotionskolloquium am: 15. MÄarz2010 Dedications This work is dedicated to my Father and to the memory of my Mother Atallah i Acknowledgement First of all, I would like to thank God, who gives us every thing and without him nothing can be done. First and foremost, I would like to thank my advisor, Professor Abbas S. Omar for his constant help and advising me. His encouragement and insight were extremely valuable throughout my graduate career. I would also like to thank Professor Jan Machac from Czech Technical University for his cooperation and discussions, and for allowing me to join his research group in Prague for six month. My thanks also go to Professor Wolfgang Menzel from the University of Ulm for his cooperation, discussions and for the fabrication of my structures at his department. I also thank Professor Samin Amari from Royal Military Collage in Canada for his cooperation and his discussions. I would like to thank all of my colleagues and friends who have helped me through the di±cult task of creating this Thesis, namely: Ali R. Ali, Ihab Hamad, Abdallah Fared, Julia Braun, Hasan Salem, Michael Maiwald, Mohammed Salah, Adel Abdel- Rahman, Fawzy Abujarad and his wife Raesa.
    [Show full text]
  • On-Wafer Characterization of Electromagnetic Properties of Thin-Film RF Materials
    On-Wafer Characterization of Electromagnetic Properties of Thin-Film RF Materials Dissertation Presented in Partial Fulfillment of the Requirements for the Degree Doctor of Philosophy in the Graduate School of The Ohio State University By Jun Seok Lee, B. S., M. S. Graduate Program in Electrical and Computer Engineering The Ohio State University 2011 Dissertation Committee Professor Roberto G. Rojas, Adviser Professor Patrick Roblin Professor Fernando L. Teixeira Copyright by Jun Seok Lee 2011 ABSTRACT At the present time, newly developed, engineered thin-film materials, which have unique properties, are used in RF applications. Thus, it is important to analyze these materials and to characterize their properties, such as permittivity and permeability. Unfortunately, conventional methods used to characterize materials are not capable of characterizing thin-film materials. Therefore, on-wafer characterization methods using planar structures must be used for thin-film materials. Furthermore, most new, engineered materials are usually wafers consisting of thin films on a thick substrate. Therefore, it is important to develop measurement techniques for on-wafer films that involve the use of a probe station. The first step of this study was the development of a novel, on-wafer characterization method for isotropic dielectric materials using the T-resonator method. Material characterization using a T-resonator provides more accurate extraction results than the non-resonant method. Although the T-resonator method provides highly accurate measurement results, there is still a problem in determining the effective T-stub length, which is due to the parasitic effects, such as the open-end effect and the T-junction effect. Our newly developed method uses both the resonant effects and the feed-line length of the T-resonator.
    [Show full text]
  • FEM Analysis of Quasi-Planar Transmission Lines
    FEM Analysis of Quasi-Planar Transmission Lines A Thesis Submitted to the College of Graduate Studies and Research in Partial Fulfillment of the Requirements For the Degree of Master of Science In the department of Electrical Engineering University of Saskatchewan Saskatoon 3/1 By Yue Yan November, 99 © Copyright Yue Yan, 1999, All right reserved ABSTRACT The high frequency wireless communication and high-speed electronics industries are demanding increasingly more sophisticated microwave integrated circuits. Today, conventional microwave integrated circuits have used single substrate planar microstrip line or suspended and inverted microstrip lines. Present day microwave integrated circuits, however, cannot rely solely on conventional microstrip line or its derivatives. During the past five years new quasi-planar microstrip lines, known as microshields, have been proposed. Electrical properties of such lines have not been adequately studied either experimentally or theoretically. There is a strong need for such studies. This thesis aims to fill this gap for some newly emerged quasi-planar lines, using the finite element method. By implementing the incremental inductance rule, the finite element method is used for transmission line loss calculation. It is then used in estimating the effects of metal penetration in microshield transmission lines used in modern day microwave circuits. This research will help advance the art of microwav6 integrated circuit technology. ii ACKNOWLEDGMENTS The author would like to express his gratitude and appreciation to Dr. Protap Pramanick for his guidance throughout the course of this work and also for hisl assistance in the co-authoring, preparation and submission of the various papers and l publications referenced in this work.
    [Show full text]
  • Chapter 21 Using Transmission Lines
    hspice.book : hspice.ch22 1 Thu Jul 23 19:10:43 1998 Chapter 21 Using Transmission Lines A transmission line delivers an output signal at a distance from the point of signal input. Any two conductors can make up a transmission line. The signal which is transmitted from one end of the pair to the other end is the voltage between the conductors. Power transmission lines, telephone lines, and waveguides are examples of transmission lines. Other electrical elements which should be thought of as transmission lines include traces on printed circuit boards and multichip modules (MCMs) and within integrated circuits. With current technologies that use high-speed active devices on both ends of most circuit traces, all of the following transmission line effects must be considered during circuit analysis: ■ Time delay ■ Phase shift ■ Power, voltage, and current loss ■ Distortion ■ Reduction of frequency bandwidth ■ Coupled line crosstalk Star-Hspice provides accurate modeling for all kinds of circuit connections, including both lossless (ideal) and lossy transmission line elements. This chapter covers these topics: ■ Selecting Wire Models ■ Performing HSPICE Interconnect Simulation ■ Understanding the Transmission Line Theory ■ References Star-Hspice Manual, Release 1998.2 21-1 hspice.book : hspice.ch22 2 Thu Jul 23 19:10:43 1998 Selecting Wire Models Using Transmission Lines Selecting Wire Models Various terms are used for electrical interconnections between nodes in a circuit. Common terms are ■ Wire ■ Trace ■ Conductor ■ Line The term “transmission line” or “interconnect” generally can be used to mean any of the above terms. Many applications model electrical properties of interconnections between nodes by their equivalent circuits and integrate them into the system simulation to make accurate predictions of system performance.
    [Show full text]
  • Efficient Integration of Planar Antennas Considering Electromagnetic Interactions at Board Level
    Efficient Integration of Planar Antennas Considering Electromagnetic Interactions at Board Level vorgelegt von Diplom-Ingenieur Florian Ohnimus aus Lüneburg von der Fakultät IV - Elektrotechnik und Informatik der Technischen Universität Berlin zur Erlangung des akademischen Grades Doktor der Ingenieurwissenschaften - Dr.-Ing. - genehmigte Dissertation Promotionsausschuss Vorsitzender: Prof. Dr. Bernd Tillack 1. Berichter: Prof. Dr. Dr. E.h. Herbert Reichl 2. Berichter: Prof. Dr. Ege Engin Tag der wissenschaftlichen Aussprache: 1.4.2011 Berlin 2011 D 83 Danksagung Die vorliegende Arbeit entstand während meiner Tätigkeit als wissenschaftlicher Mitarbeiter in der Arbeitsgruppe RF & High-Speed System Design (RSD) der Abteilung System Design and Integration (SDI) am Fraunhofer-Institut für Zuverlässigkeit und Mikrointegration (IZM) und dem Forschungsschwerpunkt Technologien der Mikroperipherik der Technischen Universität Berlin. Ich möchte mich besonders bei meinem Doktorvater und Erstgutachter, Prof. Dr. Dr. E.h. Herbert Reichl, für die Betreuung meiner Arbeit, das Feedback und die wertvollen Ratschläge bedanken. Meinem Zweitgutachter, Prof. Dr. Ege Engin, möchte ich insbesondere für die Aufnahme und Betreuung während meines zweimonatigen Aufenthalts an der San Diego State University und die Fachdiskussionen danken. Für die Übernahme des Vorsitzes des Promotionsausschusses danke ich außerdem Prof. Dr. Bernd Tillack. Des Weiteren danke ich meinem Abteilungsleiter, Dr. Stephan Guttowski, für die Unterstützung und Schaffung eines produktiven
    [Show full text]