Compact Slotted Waveguide Antenna Array Using Staircase Model of Tapered Dielectric-Inset Guide for Shipboard Marine Radar

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Article Compact Slotted Waveguide Antenna Array Using Staircase Model of Tapered Dielectric-Inset Guide for Shipboard Marine Radar

Kyei Anim , Henry Abu Diawuo and Young-Bae Jung *

Electronics Engineering Department, Hanbat National University, Daejeon 34158, Korea; [email protected] (K.A.); [email protected] (H.A.D.) * Correspondence: [email protected]; Tel.: +82-42-821-1136

Abstract: This paper presents a new configuration of a slotted waveguide antenna (SWA) array aimed at the X-band within the desired band of 9.38~9.44 GHz for shipboard marine radars. The SWA array, which typically consists of a slotted waveguide, a polarizing filter, and a metal reflector, is widely employed in marine radar applications. Nonetheless, conventional slot array designs are weighty, mechanically complex, and geometrically large to obtain high performances, such as gain. These features of the conventional SWA are undesirable for the shipboard marine radar, where the antenna rotates at high angular speed for the beam scanning mechanism. The proposed SWA array herein reduces the conventional design’s size by 62% using a tapered dielectric-inset guide structure. It shows high gain performance (up to 30 dB) and obtains improvements in radiation efficiency (up to 80% in the numerical simulations) and weight due to the use of loss and low-density dielectric material.

Citation: Anim, K.; Diawuo, H.A.; Keywords: beam scanning; dielectric-inset guide; high gain; low weight; marine radar; radiation Jung, Y.-B. Compact Slotted efficiency; slotted waveguide array; X-band; size reduction; tapering Waveguide Antenna Array Using Staircase Model of Tapered Dielectric-Inset Guide for Shipboard Marine Radar. Sensors 2021, 21, 4745. https://doi.org/10.3390/s21144745 1. Introduction Marine radars have been proven useful in commercial and military settings to provide Academic Editors: Youchung Chung information about other seacrafts and land targets [1] for diverse applications, such as and Cynthia M. Furse coastal surveillance and surface target detection, collision avoidance, weather, and bird migration monitoring [2–5]. In this regard, X-band marine radars have become more Received: 24 May 2021 attractive as they offer a high spatial and temporal resolution, low cost, flexibility, and Accepted: 8 July 2021 installation ease [6]. The marine environment imposes severe constraints on antennas’ Published: 12 July 2021 performance and mechanical design. Thus, the marine radar antenna needs to obtain fan-beam radiation patterns to operate successfully under these harsh conditions. Publisher’s Note: MDPI stays neutral Several microstrip array antennas have been extensively employed in radar and other with regard to jurisdictional claims in wireless communications systems [7–17]. Although they have promising features, such as published maps and institutional affil- low profile, low cost, and fabrication ease, their low-power handling capacity and high iations. losses often make them less suitable for marine radar [18–20]. Consequently, the slotted waveguide array (SWA) antennas [21–25], based on the pioneering work published by Elliot [26–28], are widely employed in marine radar applications. This is because of their high-power handling capacity, low losses, and good phase stability. The conventional Copyright: © 2021 by the authors. designs of SWA arrays typically consist of a slotted waveguide structure, a polarizing filter, Licensee MDPI, Basel, Switzerland. and a large metal reflector to enhance directivity, especially in the vertical plane. By using This article is an open access article this reflector device, especially in the lower frequency bands, the conventional SWA arrays distributed under the terms and become weighty, geometrically large, and mechanically complex [29]. They are, therefore, conditions of the Creative Commons undesirable for marine radar systems of which the antenna rotates at high-angular speed. Attribution (CC BY) license (https:// A miniaturization technique is thus required to make the traditional SWA more suitable creativecommons.org/licenses/by/ for the marine environment without resorting to gain degradation. Nonetheless, reducing 4.0/).

Sensors 2021, 21, 4745. https://doi.org/10.3390/s21144745 https://www.mdpi.com/journal/sensors Sensors 2021, 21, x FOR PEER REVIEW 2 of 15

Sensors 2021, 21, 4745 2 of 14 SWA more suitable for the marine environment without resorting to gain degradation. Nonetheless, reducing the reflector size as a simple miniaturization process results in considerable antenna gain reduction. the reflectorIn this paper, size asa 62% a simple size reduction miniaturization of the SWA process array results was obtained in considerable by incorporating antenna again tapered reduction. dielectric-inset guide structure in the conventional slot array design. At the same time, Indirectivity this paper, in athe62% verticalsize reduction plane was of improved the SWA array significantly was obtained without by the incorporating inclusion of a thetapered reflector dielectric-inset device. The guide function structure of the indielec the conventionaltric structure slotwith array the staircase design. Atmodel the same is to smoothlytime, directivity transform in the the vertical incident plane and wasstrongly improved bound significantly surface waves without from thethe inclusionwaveguide of intothe reflectorfree space device. characterized The function by minimum of the dielectric reflection structure and phase with disparity. the staircase Thus, model the results is to smoothly transform the incident and strongly bound surface waves from the waveguide indicated that the proposed SWA array produces antenna gain up to 30 dB comparable to into free space characterized by minimum reflection and phase disparity. Thus, the results the conventional SWA designs. This new configuration of the SWA array was compared indicated that the proposed SWA array produces antenna gain up to 30 dB comparable with its conventional counterpart to prove the practicality of the approach used in this to the conventional SWA designs. This new configuration of the SWA array was com- paper. pared with its conventional counterpart to prove the practicality of the approach used in this paper. 2. Antenna Configuration 2. AntennaFigure 1 Configuration compares the geometry of the conventional SWA array, set as a benchmark here, Figureto that1 of compares the proposed the geometry antenna based of the on conventional the design specifications SWA array, set in as Table a benchmark 1. In both cases,here, tothe that antenna of the consists proposed of a antennaslotted wave basedguide on the arrayed design with specifications ninety-four inslot Table radiators1. In toboth produce cases, a the fan-beam antenna radiation consists pattern of a slotted at the waveguide center frequency arrayed f with= 9.41 ninety-four GHz. The fan- slot beam,radiators due to to produce the long a waveguide fan-beam radiation axial length pattern (L) along at the the center y-axis, frequency has a narrowf = 9.41 beam- GHz. widthThe fan-beam, of 1.2° in due the tohorizontal the long waveguide(𝑥) plane and axial a broader length (L)beamwidth along the ofy -axis,20° in has the a vertical narrow ◦ ◦ (beamwidth𝑥) plane to compensate of 1.2 in the for horizontal the roll of the (x) ship plane [30]. and The a broader design, performances, beamwidth of and 20 analy-in the sisvertical of the ( xantennas) plane to presented compensate in this for articl the rolle have of the been ship performed [30]. The using design, CST performances, Microwave Studioand analysis software. of the antennas presented in this article have been performed using CST Microwave Studio software.

(a) (b)

Figure 1. Full 3D views of slotted waveguide array antenna. ( a)) Conventional case; ( b) Proposed case.

Table 1. Antenna design specifications for X-band marine radar. Table 1. Antenna design speciﬁcations for X-band marine radar. Parameters Value Parameters Value Parameters Value Parameters Value Frequency 9.4 GHz ± 30 MHz Polarization Horizontal Frequency 9.4 GHz ± 30 MHz Polarization Horizontal Horizontal: 1.2° or less Gain ≥28 dB Beamwidth Horizontal: 1.2◦ or less Gain ≥28 dB BeamwidthVertical: 20° or higher Vertical: 20◦ or higher Voltage standing wave Horizontal: −10 dB or less Voltage≤1.5 standing Sidelobelevel (SLL) Horizontal: −10 dB or less ratio (VSWR) ≤1.5 SidelobelevelVertical: (SLL) −10 dB or higher wave ratio (VSWR) Vertical: −10 dB or higher

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2.1. Slotted Waveguide Design 2.1. Slotted Waveguide Design The slotted waveguide structure shown in Figure2 consists of a WR-90 standard The slotted waveguide structure shown in Figure 2 consists of a WR-90 standard waveguide (a = 22.86 mm, b = 10.16 mm) with aluminum material and a wall thickness, waveguide (𝑎 = 22.86 mm, 𝑏 = 10.16 mm) with aluminum material and a wall thick- t = 1.27 mm. An array of tilted slots is milled into the narrow wall of a rectangular ness, 𝑡 = 1.27 mm. An array of tilted slots is milled into the narrow wall of a rectangular waveguide in order to generate the desired horizontally polarized electric fields [30], waveguide in order to generate the desired horizontally polarized electric fields [30], as as specified in Table1. The slots introduce discontinuities in the waveguide’s conduct- specified in Table 1. The slots introduce discontinuities in the waveguide’s conducting ing walls to interrupt the flow of electric currents along the waveguide axial. Hence, walls to interrupteach the slot flow acts of electric as a dual currents electric along dipole, the accordingwaveguide to axial. Babinet’s Hence, principle, each slot to elicit radia- acts as a dual electric dipole, according to Babinet’s principle, to elicit radiations from the tions from the traveling waves propagating in the fundamental TE10 (transverse electric) traveling waves propagating in the fundamental TE (transverse electric) mode in the mode in the waveguide [31]. The cutoff frequency for the TE10 mode is computed as waveguide [31]. The cutoff frequency for the TE mode is computed as 𝑓 = 𝑐/(2𝑎) = fc = c/(2a) = 6.56 GHz (c is the speed of light in a vacuum). The guided wavelength in q 6.56 GHz (c is the speed of light in a vacuum). The 2guided wavelength in this case is 𝜆 = this case is λg = λ0/ 1 − ( fc/ f ) = 44.43 mm (λ0 is the free space wavelength). 𝜆/1−(𝑓⁄)𝑓 = 44.43 mm (𝜆 is the free space wavelength).

Figure 2. ConfigurationsFigure 2. Configurations of the slotted of waveguide the slotted antenna waveguide and theantenna polarization and the filter polarization used in the filter design used of in the the SWA arrays. design of the SWA arrays. The tilted slots wrap around the edges of the narrow wall and into the broad wall (see The tilted Figureslots wrap2). Thus, around each the slot edges has the of slot the tiltnarrow angle wallθ and and slot into depth theδ broadparameters. wall By adjusting (see Figure 2). θThus,, the excitation each slot strengthhas the slot from tilt each angle slot 𝜃 can and be slot controlled depth independently.𝛿parameters. By The excitation is adjusting 𝜃, thesmall excitation for a large strengthθ and from zeroes each for slotθ = can0◦ bewhich controlled may be independently. used in a non-uniformly The excited excitation is smallarray for design a large with 𝜃 differentand zeroesθ values, for 𝜃 as = depicted°0 which may in Figure be used2, to in decrease a non-uni- the sidelobe levels formly excited(SLLs). array design To reduce with the different number θ of values, design as parameters, depicted in all Figure slots have 2, to the decrease same width w = 3 mm. the sidelobe levelsThe (SLLs). slot length To reduce for each the slot number in this of case design is L rparameters,= (b + t)/ all cos slotsθ + 2have(δ − thet/2 ) = 0.4625λg . same width w The= 3 mm. set of The values slot forlengthδ is centeredfor each slot around in this 1.73 case mm is at 𝐿 resonance = (𝑏 after+ 𝑡)/𝑐𝑜𝑠𝜃 optimization. + 2(𝛿 − 𝑡/2) = In 0.4625𝜆 the . designThe set of of a non-uniformlyvalues for 𝛿 is SWAcentered array around in Figure 1.732, eachmm at slot’s reso- dimensions are nance after optimization.computed and optimized independently to attain the desired amplitude coefficient an at In the designresonance. of a non-uniformly Once an for each SWA slot array is set, in theFigure array 2, factoreach slot’s (AF) baseddimensions on Taylor are distribution is computed and estimatedoptimized using independently (1) and (2) to [32 attain]. the desired amplitude coefficient 𝑎 at resonance. Once 𝑎 for each slot is set, the array factor (AF) based on Taylor distribution M is estimated using (1) and (2) [32]. (AF)2M = an cos[(2n − 1)u] (1) M ∑ ()AFanu=−cos ( 2n= 11 ) 2M  n=1 n  (1) u = πd/λ cos θ (2) ud= πλcos θ M is an integer, n is the number of slots, and d is the slot spacing (see(2) Figure2). In M is an integer,this case, n isµ the= 0.5 numberπ/ cos ofθ isslots, used and for thed is chosenthe slot slot spacing spacing (seed =Figure0.5λ g2).i.e., In λ = λg . The this case, µ = 0.5𝜋/𝑐𝑜𝑠𝜃 is used for the chosen slot spacing 𝑑 = 0.5𝜆 (i.e.,𝜆 = 𝜆 ). The AF is thus dependent on amplitude coefficient 𝑎 and the slot tilt angle 𝜃 of each

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slot to achieve the desired distribution to result in the radiation pattern computed as follows: AF is thus dependent on amplitude coefﬁcient an and the slot tilt angle θ of each slot to achieve the desired distribution to result in the radiation pattern computed as follows: Etotal=⋅EAFsin gle () (3)

Since the slotted waveguide Estructuretotal = Esingle in Figu· (AFre) 2 is fed at the excitation port and(3) terminated in a matched load (absorber) to achieve a traveling wave antenna, its equivalent circuitSincethe becomes slotted a waveguideparallel conductance structure G in at Figure resonance,2 is fed whereas at the excitation the reactive port suscep- and terminated in a matched load (absorber) to achieve a traveling wave antenna, its equivalent tance B cancels out. Hence, the resonant conductance 𝑔 of N slot is computed from the circuit becomes a parallel conductance G at resonance, whereas the reactive susceptance B amplitude coefficient 𝑎 using (4), while the slot tilt angle 𝜃 is related to conductance cancels out. Hence, the resonant conductance g of N slot is computed from the amplitude 𝑔 by (5). Once the initial 𝜃 values are set, the nslot tilt angle is further optimized to obtain coefficient a using (4), while the slot tilt angle θ is related to conductance g by (5). Once the final rangen of values from −10° to 15°. n n the initial θ values are set, the slot tilt angle is further optimized to obtain the final range of ◦ ◦ a2 values from−10 to 15 . g ==n ;nN 1,2,...... , n N 2 (4) 2 a an i=1 i gn = ; n = 1, 2, . . . , N (4) ∑N a2 θ =−i=1 i 2 + + nnn900.7gg 237.6 2 2 (5) θn = −900.7gn + 237.6gn + 2 (5) The main beam, which is perpendicular to the waveguide axis, lies in the forward The main beam, which is perpendicular to the waveguide axis, lies in the forward direction along the x-axis (see Figure 1) because of the slot spacing 𝑑 = 0.5𝜆 and oppo- direction along the x-axis (see Figure1) because of the slot spacing d = 0.5λ and opposite site tilt angles among adjacent slots. Thus, the co-polarization due to the desiredg horizon- tilt angles among adjacent slots. Thus, the co-polarization due to the desired horizontally tally polarized electric fields 𝐸 (see Figure 2) is superimposed in the phase, whereas the polarized electric fields EH (see Figure2) is superimposed in the phase, whereas the verti- vertically polarized electric fields 𝐸, which contribute to undesired cross-polarization, cally polarized electric fields EV, which contribute to undesired cross-polarization, among among the adjacent slots, cancel out because they assume an equal amplitude but opposite the adjacent slots, cancel out because they assume an equal amplitude but opposite direction.direction. This phenomenon,This phenomenon, in theory, in theory, inherently inhere reducesntly reduces the sidelobe the sidelobe levels levels of a waveguide of a wave- slotguide array slot due array to due the lowto the levels low oflevels cross-polarized of cross-polarized signals. signals.

2.2.2.2. Polarization FilterFilter TheThe instantaneousinstantaneous electric fieldsfields E fromfrom thethe inclinedinclined slotsslots decomposedecompose intointo thethe hori-hori- zontalzontal E𝐸H andand vertical verticalE V𝐸fields, fields, as as depicted depicted in in Figure Figure2. As2. As mentioned mentioned earlier, earlier, the the vertical verti- electriccal electric fields fieldsEV among𝐸 among the adjacentthe adjacent slots slots cancel cancel out asout the as adjacentthe adjacent slots slots have have equal equal and oppositeand opposite tilt angles. tilt angles. However, However, in practice, in practice, the slotthe tiltslot angle tilt angleθ among 𝜃 among the adjacent the adjacent slots variesslots varies for optimization for optimization and amplitude and amplitude distribution distribution purposes. purposes. In addition, In addition, due to fabricationdue to fab- errors,ricationθ mayerrors, differ 𝜃 may from differ the numerical from the value numerical in the simulation.value in the Hence, simulation. the EV amongHence, thethe adjacent𝐸 among slots the doadjacent not cancel slots outdo not entirely cancel due out to entirely the varying due to phases the varying and magnitudes. phases and Therefore,magnitudes. high Therefore, sidelobes high are sidelobe observeds are in the observed horizontal in the (xy horizontal) plane due (𝑥𝑦 to) plane the measurable due to the levelsmeasurable of unwanted levels of cross-polarized unwanted cross-polarized signals (see Figuresignals3a). (see Figure 3a).

(a) (b)

Figure 3. Radiation patterns of conventional SWA array.array. ((aa)) HorizontalHorizontal ((xy𝑥𝑦)) plane;plane; ((bb)) VerticalVertical ( xz(𝑥𝑧)) plane. plane.

InIn orderorder toto significantlysignificantly reducereduce thethe sidelobes,sidelobes, aa polarizationpolarization filterfilter isis positionedpositioned inin frontfront ofof thethe slottedslotted waveguide,waveguide, asas shownshown inin FigureFigure2 .2. The The filter filter constitutes constitutes equispaced equispaced orthogonal walls with a spacing parameter ψ ≤ 0.5λg to ensure adequate suppression

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of cross-polarized signals. The filter is positioned ρ distance away from the waveguide wall (see Figure2). In principle, the effectiveness of the filter is inversely proportional to wall spacing ψ and proportional to the distance ρ between the filter and the waveguide wall. However, the attenuation of the EV denoted by γ is determined primarily from the wall spacing ψ as expressed in (6) [33]. Referring to Figure2, the filter’s orthogonal walls are positioned in such a way that they stand in between the adjacent slots to limit the efficient propagation of undesired EV to guarantee the sidelobes’ suppression, as observed in Figure3a. s 5.46 2 · ψ 2 γ = · 1 − [dB/m] (6) 2 · ψ λ It should be noted that both the ψ and ρ parameters have an influence on the radiating slot and the radiation. Therefore, they are optimized and fixed as ψ = 6 mm and ρ = 20 mm.

2.3. Reﬂector Referring to Figure1, the conventional slot array design has a reflector device with a flare-out angle to increase the antenna’s aperture to increase the gain from 24 dB to 30 dB, as shown in Figure3b, to satisfy the requirement in Table1. Thus, the conventional SWA array has a dimension of 1920 mm × 165 mm × 84.3 mm (L × W × H). However, the SWA arrays with the reflector device are large in the antenna’s cross- section (i.e., yz-plane), mechanically complex, and weighty to rotate at a very high angular speed. The proposed antenna in Figure1b is the result of the effort to design a reduced size SWA array by using a dielectric-inset structure, over which the bound surface waves are excited. Thus, its vertical size is about half that of the conventional case (see Figure1a), giving the same vertical ( xz) plane beamwidth, and shows im- provement in weight and mechanical simplicity. After the full analysis of the SWA arrays in Figure1 was performed, the following are the optimized design parameters: a = 22.86, b = 10.16 mm for WR − 90, L = 1920 mm, t = 1.27 mm, θ = −10◦ ∼ 15◦, δ = 1.73 mm, w = 3 mm, ψ = 6 mm, and ρ = 20 mm.

3. Design and Analysis of the Proposed Slotted Waveguide Array Antenna 3.1. Evolution of the Proposed Case To facilitate the development of the proposed antenna while still providing a physical understanding of its geometrical structure, the three-dimensional SWA arrays in Figure1 are mirrored in their two-dimensional analogues in the xz-plane, as shown in Figure4. Thus, the evolution of the proposed case is demonstrated in Figure4. The conventional antenna, Type A, with a wide flare-angle reflector gives an ideal gain of more than 30 dB in Figure5 at the expense of large antenna size. The simplest approach to reduce the antenna’s physical size is by reducing the reflector size. Unfortunately, this approach results in a significant decrease in antenna gain. The solution to this problem was accomplished by incorporating a tapered dielectric- inset guide structure in the SWA array to produce antenna Type B in Figure4. This approach immediately reduces the height of the conventional antenna by about 50%. However, in the Type B antenna conﬁguration, an optimum taper proﬁle can only be achieved when the length parameter Sx in the forward direction is very long, up to 2.5λ, which undermines the antenna’s compactness. The relationship between Sx and the gain of the antenna can be expressed mathematically in (7) [34].

Gain = 8Sx/λ (7) Sensors 2021, 21, x FOR PEER REVIEW 6 of 15

Thus, the incident, strongly bound surface wave field from the slotted waveguide, which is butted to the base of the dielectric structure, is transformed smoothly into a radi- Sensors 2021, 21, 4745 6 of 14 ation field that is characterized by maximum antenna gain (see Figure 5, Type B) with an in-phase field.

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Figure 4. Two-dimensional analogue of four different SWA array models in the xz (vertical) plane. Figure 4. Two-dimensional analogue of four different SWA array models in the xz (vertical) plane.

(a) (b)

FigureFigure 5. 5.Simulated Simulated results results for for different different SWA SWA array array models. models. (a )(a Gain;) Gain; (b )(b Patterns) Patterns in in vertical vertical (xz ()xz plane.) plane.

Thus,To solve the incident,the beam strongly splitting bound problem surface of antenna wave fieldType from C without the slotted resorting waveguide, to an in- whichcrease is in butted the antenna’s to the base size, of the the dielectric-ins dielectric structure,et guide structure is transformed was modified smoothly by tapering into a radiationthe termination field that with is characterized a staircase approximation by maximum antennato form the gain end-gradient (see Figure5 ,in Type antenna B) with Type anD. in-phase The staircase field. model of the tapered dielectric is synthesized as a series of short slab segments, wherein the uniform cross-sectional areas in the forward direction are diminished progressively, as shown in Figure 4. Thus, the width and height parameters of each dielectric slab segment of the staircase model of the taper profile were optimized individually to allow for the redistribution of propagating signals in the dielectric medium so that at the dielectric–air interface, all the signals exiting the medium assume the same approximated phase velocity to maximize radiation efficiency. In effect, a conformal beam pattern is restored for the proposed SWA array Type D as the gain improves to about 29 dB in Figure 5. By incorporating the dielectric-inset guide structure into the SWA design, the proposed antenna, Type D, achieves a 62% size reduction while maintaining the antenna’s high gain result, which is comparable to the conventional case-Type A. It should be noted that the long axial length (L) of the antenna is preserved in all the designs to ensure that the beam has a narrow width in the horizontal plane to satisfy the required half power beamwidth (HPBW) of 1.2°.

3.2. End Gradient Optimization The optimization of the dielectric-inset guide structure significantly improves the radiation pattern of the antenna. The taper profile of the optimized dielectric structure is depicted in Figure 6, wherein the end gradient of the dielectric is synthesized as a series of dielectric slab segments of equal width ∆w and uniform cross-sectional areas of gradually smaller heights 𝐻,𝑖 = 1,2,…,6. In other words, the tapered dielectric region is modeled as a sequence of sufficiently short (in terms of wavelength), uniform slab segments of diminishing cross-section in the forward direction, i.e., along the x-axis, as shown in Figure 6. The taper is segmented into five regions (𝑖 = 2,3,…,6) with successively smaller heights. In this step-synthesis technique, the step discontinuity on the dielectric structure is regarded as a radiating aperture of a serial uniform slabs. At each step discontinuity, the fundamental transverse electric (TE) surface wave, which is assumed to be incident in the +𝑥 direction from 𝑥 = − ,∞ is perturbed to cause redistribution of the instantaneous phases of the surface wave field on the uniform cross-section for phase matching. Consequently, the optimum taper profile of the dielectric is taken as one which smoothly transforms the strongly bound surface wave field into a radiation field with minimal phase disparity and reflection. This suggests that surface wave fields constructively interfere as they assume the same approximated phase velocity to maximize radiation efficiency.

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By truncating the long tapered section of the dielectric structure to produce antenna Type C (see Figure4), the width of the antenna is reduced to 108 mm. Thus, a 29% size reduction is achieved in the forward direction as Sx decreases to about 1.7λ. However, the gain of antenna Type C drops considerably to 26 dB in Figure5a, since the phase disparity of the incident surface wave on the uniform cross-section of the truncated dielectric structure increases significantly. This causes the destructive interference of the surface wave field, transforming into a radiation field at the termination to result in a beam split (see Figure5b, Type C). To solve the beam splitting problem of antenna Type C without resorting to an increase in the antenna’s size, the dielectric-inset guide structure was modified by tapering the termination with a staircase approximation to form the end-gradient in antenna Type D. The staircase model of the tapered dielectric is synthesized as a series of short slab segments, wherein the uniform cross-sectional areas in the forward direction are diminished progressively, as shown in Figure4. Thus, the width and height parameters of each dielectric slab segment of the staircase model of the taper profile were optimized individually to allow for the redistribution of propagating signals in the dielectric medium so that at the dielectric–air interface, all the signals exiting the medium assume the same approximated phase velocity to maximize radiation efficiency. In effect, a conformal beam pattern is restored for the proposed SWA array Type D as the gain improves to about 29 dB in Figure5. By incorporating the dielectric-inset guide structure into the SWA design, the proposed antenna, Type D, achieves a 62% size reduction while maintaining the antenna’s high gain result, which is comparable to the conventional case-Type A. It should be noted that the long axial length (L) of the antenna is preserved in all the designs to ensure that the beam has a narrow width in the horizontal plane to satisfy the required half power beamwidth (HPBW) of 1.2◦.

3.2. End Gradient Optimization The optimization of the dielectric-inset guide structure significantly improves the radiation pattern of the antenna. The taper profile of the optimized dielectric structure is depicted in Figure6, wherein the end gradient of the dielectric is synthesized as a series of dielectric slab segments of equal width ∆w and uniform cross-sectional areas of gradually smaller heights Hi, i = 1, 2, ... , 6. In other words, the tapered dielectric region is modeled as a sequence of sufficiently short (in terms of wavelength), uniform slab segments of diminishing cross-section in the forward direction, i.e., along the x-axis, as shown in Figure6. The taper is segmented into five regions (i = 2, 3, . . . , 6) with successively smaller heights. In this step-synthesis technique, the step discontinuity on the dielectric structure is regarded as a radiating aperture of a serial uniform slabs. At each step discontinuity, the fundamental transverse electric (TE) surface wave, which is assumed to be incident in the +x direction from x = −∞, is perturbed to cause redistribution of the instantaneous phases of the surface wave field on the uniform cross-section for phase matching. Consequently, the optimum taper profile of the dielectric is taken as one which smoothly transforms the strongly bound surface wave field into a radiation field with minimal phase disparity and reflection. This suggests that surface wave fields constructively interfere as they assume the same approximated phase velocity to maximize radiation efficiency. Sensors 2021, 21, 4745 8 of 14 Sensors 2021, 21, x FOR PEER REVIEW 8 of 15 Sensors 2021, 21, x FOR PEER REVIEW 8 of 15

Figure 6. Geometry of thethe staircasestaircase modelmodel ofof thethe taperedtapered dielectric-insetdielectric-inset guideguide structurestructure depictingdepicting theFigure end gradient6. Geometry andand of feedfeed the gradient.gradient. staircase model of the tapered dielectric-inset guide structure depicting the end gradient and feed gradient. The instantaneous phase phase of of the the radiation radiation fiel fieldd from from the the dielectric dielectric surface surface with with no nota- taperper profileThe profile instantaneous received received at points at phase points A, of B, A,the and B, radiation and C, assumed C, assumedfield tofrom be to withinthe be dielectric within the far-field the surface far-field region with region nofrom ta- 2 fromtheper antenna, profile the antenna, received i.e., 𝑅=2𝐷 i.e., at Rpoints/𝜆= 2 (maximumD A,/ B,λ and(maximum C,linear assumed dimension linear to dimensionbe ofwithin an antenna the of anfar-field is antenna D), isregion shown is D from), in is shownFigurethe antenna, 7a. in FigureIt can i.e., be7 a.𝑅=2𝐷 noted It can that/𝜆 be (maximumnotedthere is that a si therelineargnificant is dimension a significantphase difference of an phase antenna between difference is D ),the is between shownreceived in thefieldsFigure received at 7a. these It fieldscan points, be at noted these resulting that points, there in resultinga isscatteri a significantng in awave scattering phase to cause difference wave reflection to causebetween and reflection thedestructive received and destructiveinterference.fields at these interference. However, points, resultingthe However, introduction in thea scatteri introduction of thnge wavetaper of toregion the cause taper of reflection the region optimized of and the optimizeddestructive dielectric dielectricstructureinterference. modeled structure However, with modeled staircasethe introduction with approximation, staircase of approximation,the taperin Figure region 6, leads in of Figure the to aoptimized 6substantial, leads to dielectric areduc- sub- stantialtionstructure in the reduction modeled phase disparity in with the phasestaircase at these disparity approximation, received at these points received in within Figure points the 6, leads desired within to aband, thesubstantial desired as shown reduc- band, in asFiguretion shown in 7b. the inThe phase Figure radiation disparity7b. The field radiationat atthese these received fieldpoints at , points thesetherefore, points,within assumes therefore,the desired the same assumes band, approximated as the shown same in approximated phase velocity to mimic the behavior of a plane wave that maximizes phaseFigure velocity 7b. The to radiation mimic the field behavior at these of points a plane, therefore, wave that assumes maximizes the radiationsame approximated efficiency radiation efficiency due to the constructive interference. The optimal taper profile of the duephase to thevelocity constructive to mimic interference. the behavior The of aoptima plane lwave taper thatprofile maximizes of the dielectric radiation in efficiency Figure 6 dielectricdue to the in constructive Figure6 with interference. the staircase The model optima has geometricall taper profile parameters of the dielectric fixed as in follows: Figure 6 with the staircase model has geometrical parameters fixed as follows: ∆𝑤 = 4 mm, 𝐻 = ∆withw = the4 staircasemm, H2 model= 39.48 hasmm geometrical, H3 = 36.33 parametersmm, H4 fixed= 31.07 as follows:mm, H 5∆𝑤= 26.64 = 4mm mm, and 𝐻 = 39.48 mm, 𝐻 = 36.33 mm, 𝐻 = 31.07 mm, 𝐻 = 26.64 mm and 𝐻 = 18.45 mm. H 39.486 = 18.45mm, 𝐻 mm. = 36.33 mm, 𝐻 = 31.07 mm, 𝐻 = 26.64 mm and 𝐻 = 18.45 mm.

(a) (b) (a) (b) Figure 7. Simulated phase responses at three different points from the antenna’s dielectric rod structure with (a) No taper Figureprofile;Figure 7. (7.bSimulated )Simulated Taper region phase phase modeled responses responses with at at the three three staircase different different approximation. points points from from the the antenna’s antenna’s dielectric dielectric rod rod structure structure with with ( a(a)) No No taper taper proﬁle;profile; ( b(b)) Taper Taper region region modeled modeled with with the the staircase staircase approximation. approximation. It should also be noted that the base of the dielectric-inset guide structure, where the slottedItIt shouldwaveguideshould alsoalso isbe butted, noted noted thathas that alsothe the basebeen base of ta of thpered thee dielectric-inset dielectric-inset using the staircase guide guide structure,approximation structure, where where (see the theFigureslotted slotted 6) waveguide to waveguideresult in is a butted, minimum is butted, has reflectionalso has been also andta beenpered gradually tapered using usingtransitionthe staircase the the staircase approximation incident, approxima- strongly (see tionboundFigure (see field 6) Figureto toresult the6) dielectric toin a result minimum inmedium. a minimum reflection This feed and reflection taper,gradually andtherefore, transition gradually significantly the transition incident, reduces the strongly inci- the dent,Voltagebound strongly fieldStanding to boundthe Wave dielectric field Ratio to medium. (VSWR) the dielectric onThis the feed medium. uniform taper, section Thistherefore, feed of the taper,significantly guide therefore, to improve reduces signifi- the the radiationVoltage Standing characteristics Wave Ratioof the (VSWR) antenna. on The the matchinguniform section slab segments of the guide at the to improvebase of the the radiation characteristics of the antenna. The matching slab segments at the base of the

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dielectric structure in Figure 6 have been optimized and fixed as: ∆𝑤 = 2 mm, ∆𝑤 = 3 mm, ∆𝑤 = 5 mm, 𝐻 = 10 mm, 𝐻 = 20 mm, and 𝐻 = 35 mm. Sensors 2021, 21, 4745 9 of 14 3.3. Dielectric Material Selection The selection of a proper dielectric material for the design of the antenna’s dielectric structurecantly reduces is primarily the Voltage based Standing on the Wave material’s Ratio (VSWR)relative onpermittivity the uniform (𝜀 section) , loss of tan- the gentguide (tan𝛿 to improve), and most the importantly, radiation characteristics weight or density. of the It antenna.is evident Thethat as matching 𝜀 increases, slab seg-the bandwidthments at the reduces, base of and the dielectricthe radiation structure efficiency in Figure decreases6 have due been to optimizedthe increase and in the fixed qual- as: w = w = w = H = H = H = ity∆ factora 2 (Q-factor).mm, ∆ b It should3 mm, ∆be cnoted5 mmthat, as athe 10𝜀mm increases,, b the20 mm dielectric, and componentc 35 mm . no longer works under the TE10 mode required for a low sidelobe and high gain. This 3.3. Dielectric Material Selection suggests that the lower the 𝜀, the higher the radiation power for a fixed tan𝛿, as shown in FigureThe selection8. Furthermore, of a proper extremely dielectric lightw materialeight formaterials the design are ofhighly the antenna’s recommended dielectric for thestructure design is of primarily a dielectric based component, on the material’s since heavy relative materials permittivity undermine(εr), loss the tangent compactness(tan δ), achievedand most by importantly, the proposed weight SWA or array. density. Thus, It isto evident choose thatthe right as εr materialincreases, for the the bandwidth dielectric component,reduces, and several the radiation property efﬁciency data among decreases commercial due to dielectric the increase materials in the are quality compared factor in(Q-factor). Table 2. It It should should be be noted noted that that model as the Dεr hasincreases, the lightest the dielectric weight in component terms of density no longer and works under the TE10 mode required for a low sidelobe and high gain. This suggests that the lowest ɛ among the materials. Therefore, it is chosen as the material of the dielectric componentthe lower the of theεr, theproposed higher antenna. the radiation power for a ﬁxed tan δ, as shown in Figure8. Furthermore, extremely lightweight materials are highly recommended for the design of Tablea dielectric 2. Comparison component, of the sinceproperties heavy of different materials dielectric undermine materials. the compactness achieved by the proposed SWA array. Thus, to choose the right material for the dielectric component, several propertyMaterial data among commercialA dielectricB materials areC compared in Table D 2. It shouldRelative be permittivity noted that model (𝜀) D has2.1 the lightest 2.53 weight in terms 1.7 of density and the 1.53 lowest εr amongLoss the tangent materials. (tan𝛿 Therefore,) 0.00015 it is chosen 0.0001 as the material 0.0001 of the dielectric component 0.00035 of the proposedDensity (kg/cm antenna.) 2.2 1 × 10 1 × 10 8 × 10

FigureFigure 8. 8. NormalizedNormalized radiation radiation power power from from the the dielectric dielectric rod rod structure structure with with va variedried relative relative permit- permit- tivity,tivity, εɛr..

4.Table Results 2. Comparison and Discussions of the properties of different dielectric materials. Figure 9a,b show a photograph of the fabricated SWA array and outdoor antenna Material A B C D installation with a radome, respectively. Relative permittivity (εr) 2.1 2.53 1.7 1.53 Loss tangent (tan δ) 0.00015 0.0001 0.0001 0.00035 Density (kg/cm3) 2.2 1 × 10−3 1 × 10−3 8 × 10−5

4. Results and Discussion Figure9a,b show a photograph of the fabricated SWA array and outdoor antenna installation with a radome, respectively.

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Sensors 2021, 21, x FOR PEER REVIEW 10 of 15 Sensors 2021, 21, 4745 10 of 14

(a) (a)

Radome Radome

(b) (b) Figure 9. Photographs of the fabricated SWA array with a compact dielectric structure. (a) Assem- FigurebledFigure antenna 9. 9.Photographs Photographs prototype of ofwithout the the fabricated fabricated a radome; SWA SWA (b array) Outdoorarray with with a proposed compact a compact dielectric antenna dielectric structure.installation structure. (a )in Assembled a(a radome.) Assem- antennabled antenna prototype prototype without without a radome; a radome; (b) Outdoor (b) Outdoor proposed proposed antenna antenna installation installation in a radome. in a radome. The experimental results of the reflection coefficients shown in Figure 10a exhibit a broadTheThe bandwidth experimental experimental for 𝑆 resultsresults < −20 ofof the the dB reflection reflectionto cover coefficientsthe coefficients entire frequency shown shown in in band Figure Figure of 10 interest.10aa exhibit exhibit The a a broadexperimentalbroad bandwidthbandwidth results forfor ofS 𝑆11 the< < gain− −2020 dBand dB to radiationto cover cover the theefficiency entire entire frequency frequencyare shown band band in ofFigure of interest. interest. 10b. The TheThe experimentalmeasuredexperimental gain resultsresults curve, of whichof thethe gain ironicallygain and and radiation slightlyradiation outperforms efficiency efficiency are arethe shown shownsimulated in in Figure Figureone, has 10 10b.b. a peak The The measuredgainmeasured of 30.06 gain gain dB curve, curve, at 9.41 which which GHz. ironically ironicallyThe radiation slightly slightly efficiency outperforms outperforms is above the the simulated 85%simulated in the one, one, desired has has a a peakband peak gainofgain 9.38 of of− 30.069.44 30.06 GHz. dB dB at at 9.41 9.41 GHz. GHz. The The radiation radiation efficiency efficiency is is above above85% 85%in the in the desired desired band band of 9.38of 9.38−9.44−9.44 GHz. GHz.

(a) (b) (a) (b) FigureFigure 10.10. MeasuredMeasured andand simulated simulated results results of of proposed proposed SWA SWA array. array. (a )(a Reflection) Reflection coefficients; coefficients; (b) ( Gainb) Gain curves curves and and radiation radia- efficiency.tionFigure efficiency. 10. Measured and simulated results of proposed SWA array. (a) Reflection coefficients; (b) Gain curves and radiation efficiency. FigureFigure 1111a–fa–f showshow thethe measuredmeasured andand simulatedsimulated radiationradiation patternspatterns ofof thethe proposedproposed SWASWA arrayFigurearray inin 11a–f the xy 𝑥𝑦show-plane-plane the (horizontal) (horizontal) measured and andand xz simulated𝑥𝑧-plane-plane (vertical)(vertical) radiation atat 9.38,patterns9.38, 9.41,9.41, of andand the 9.449.44 proposed GHz. TheySWA show array that in the measured 𝑥𝑦-plane beam (horizontal) patterns, and to a𝑥𝑧 large-plane extent, (vertical) agree at well 9.38, with 9.41, the and simulated 9.44 GHz. results. However, the measured xz-plane beam patterns have SLL lower than −12 dB,

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Sensors 2021, 21, 4745They show that measured beam patterns, to a large extent, agree well with the simulated 11 of 14 results. However, the measured 𝑥𝑧-plane beam patterns have SLL lower than −12 dB, which is substantially higher than that of the simulated results at the frequency points. This is attributablewhich to is the substantially complexity higher of the than antenna’s that of theconfiguration. simulated results Thus, atthe the combined frequency points. This effect of the differentis attributable structural to the layers complexity of the antenna of the antenna’s has a higher conﬁguration. tendency Thus,for errors the combinedto effect occur during ofmanufacturing the different structuraland assembly. layers This ofthe could antenna easily has be acorrected higher tendency when the for an- errors to occur tenna is assembledduring properly. manufacturing The measured and assembly. 𝑥𝑦-plane This HPBWs could easilyare all be 1.2 corrected° at these when three the antenna is ◦ frequency points.assembled The corresponding properly. The measured measured 𝑥𝑧xy-plane-plane HPBWs HPBWs are are 28 all°, 1.227.8°atand these 27.7 three° frequency points. The corresponding measured xz-plane HPBWs are 28◦, 27.8◦and 27.7◦.

(a) (b)

(e) (f)

Figure 11. MeasuredFigure 11. and Measured simulated and radiation simulated patterns radiation of thepatte proposedrns of the SWA proposed array. (SWAa) Horizontal array. (a) ( xyHorizontal)-plane, 9.38 GHz; 𝑥𝑦 𝑥𝑦 𝑥𝑦 (b) Horizontal( ()-plane,xy)-plane, 9.38 9.41 GHz; GHz; (b) (Horizontalc) Horizontal ( ()-plane,xy)-plane, 9.41 9.44 GHz; GHz; (c) Horizontal (d) Vertical ( (xz)-plane,)-plane, 9.44 9.38 GHz; GHz; (d (e) ) Vertical Vertical (𝑥𝑧)-plane, 9.38 GHz; (e) Vertical (𝑥𝑧)-plane, 9.41 GHz; (f) Vertical (𝑥𝑧)-plane, 9.44 GHz. (xz)-plane, 9.41 GHz; (f) Vertical (xz)-plane, 9.44 GHz.

As the proposed SWA array operates in X-band, the conventional SWA array, set as a benchmark, should also operate in X-band for comparative physical size to ensure a fair comparison. Under this condition, an SWA array with a high-proﬁle reﬂector for a high gain result was simulated. The two-dimensional equivalents in the xz or vertical plane Sensors 2021, 21, 4745 12 of 14

of the proposed and conventional models are shown in Figure 12a. The simulated gain curves are plotted in Figure 12b. • Ant-A: Proposed SWA array comprising of a slotted waveguide, compact dielectric Sensors 2021, 21, x FOR PEER REVIEW structure, and miniaturized reﬂector. 13 of 15 • Ant-B: Commercial SWA array composed of a slotted waveguide and a relatively larger reﬂector.

(a) (b)

FigureFigure 12. 12.(a )(a Two) Two dimensional dimensional analogues analogues in xz in(vertical) 𝑥𝑧 (vertical) plane; plane; (b) Gain (b) Gain results results of the of simulated the simulated conventional conventional and proposed and pro- SWAposed arrays. SWA arrays.

5. ConclusionsThe proposed antenna possesses an average gain of 29 dB, which has been found to be comparableIn summary, to that of a thenew conventional configuration type. of slo However,tted waveguide the proposed array antenna (SWA) antennas, configuration com- achievesprising aa 50%slottedsize waveguide, reduction in polarizing lateral size filter, that liesand along tapered the dielectric-insetz-axis and 29% guidein the forwardstructure, orwas radiation proposed direction for use along in a theshipboardx-axis. Thus,marine the radar antenna’s in X-band. total The size concept is reduced of using by 62% the, whiledielectric-inset keeping its guide high gainstructure within to the reduce band th ofe interest.size of an Moreover, SWA array a low was loss, presented. low-density Com- materialparison wasbetween chosen the forproposed the design and the of the conventional dielectric structureSWA arrays to maintainwas also presented, this physical con- advantagefirming the over practicality the conventional of this approach. counterpart The without experimental resulting results in a indicated heavy weight that a and size lowreduction efficiency. of 62% could easily be achieved for the SWA array, while preserving the antenna’sTable superior3 compares performance, the requirements i.e., a higher of the gain marine and radiation radar to efficiency the measurement at the same data time. ofA the staircase proposed model SWA of array.the tapered It shows section that of the the proposed implemented SWA dielectric array can structure fulfil all ofwas the in- requirementstroduced. The except incident, for the guided SLL insurface the vertical wave plane.field on This the isuniform primarily cross-section due to fabrication was per- andturbed assembly for phase errors, matching which may purposes result fromto ensure misalignment that it transformed between the into different a radiation structural field componentsthat is characterized of the proposed by the SWA maximum array. Thus, intens properity with assembly little reflection. of the various Thus, components the antenna willexhibited easily alleviate a peak gain the of high 30.06 SLL dB in and the radiation measured efficiency results. above 80% in the desired band of 9.38~9.44 GHz. The measured horizontal plane HPBWs are about 1.2° at the various frequency points. The corresponding measured 𝑥𝑧-plane HPBWs are about 28°. The selected dielectric has low-density properties to maintain low weight. This proposed SWA array has promising physical and electrical features, making it more suitable for modern marine radar applications than the conventional case.

Author Contributions: K.A. designed, simulated, and optimized the proposed slotted waveguide antenna array. H.A.D contributed to the preparation of this article. K.A. prepared this article, including its revision. Y.-B.J. provide the suggestions regarding the design, measurement, and preparation of this article and is the supervisor of the research group. All authors have read and agreed to the published version of the manuscript. Funding: This research received no external funding. Institutional Review Board Statement: Not applicable. Informed Consent Statement: Not applicable. Data Availability Statement: Not applicable.

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Table 3. Comparison between the requirements and the measured data.

Parameters Requirement Numerical Data Measured Data Frequency 9.4 GHz 9.4 GHz 9.4 GHz o Gain ≥ 28 dB ≥ 29.62 dB ≥ 29.4 dB o VSWR ≤ 1.5 ≤ 1.5 ≤ 1.5 o Polarization Horizontal Horizontal Horizontal o ◦ ◦ ◦ Horizontal (Az) ≤ 1.2 ≤ 1.2 ≤ 1.2 o Beamwidth ◦ ◦ ◦ Vertical (El ) ≥ 20 ≥ 27.7 ≥ 27.8 o

Horizontal (Az) ≤ −20 dB ≤ −20 dB ≤ −20.8 dB o SLL Vertical (El) ≤ −30 dB ≤ −31 dB ≤ −12 dB X Az and El denote azimuth and elevation planes, respectively.

5. Conclusions In summary, a new configuration of slotted waveguide array (SWA) antennas, comprising a slotted waveguide, polarizing filter, and tapered dielectric-inset guide structure, was proposed for use in a shipboard marine radar in X-band. The concept of using the dielectric-inset guide structure to reduce the size of an SWA array was presented. Com- parison between the proposed and the conventional SWA arrays was also presented, confirming the practicality of this approach. The experimental results indicated that a size reduction of 62% could easily be achieved for the SWA array, while preserving the antenna’s superior performance, i.e., a higher gain and radiation efficiency at the same time. A staircase model of the tapered section of the implemented dielectric structure was introduced. The incident, guided surface wave field on the uniform cross-section was perturbed for phase matching purposes to ensure that it transformed into a radiation field that is characterized by the maximum intensity with little reflection. Thus, the antenna exhibited a peak gain of 30.06 dB and radiation efficiency above 80% in the desired band of 9.38~9.44 GHz. The measured horizontal plane HPBWs are about 1.2◦ at the various frequency points. The corresponding measured xz-plane HPBWs are about 28◦. The selected dielectric has low-density properties to maintain low weight. This proposed SWA array has promising physical and electrical features, making it more suitable for modern marine radar applications than the conventional case.

Author Contributions: K.A. designed, simulated, and optimized the proposed slotted waveguide antenna array. H.A.D. contributed to the preparation of this article. K.A. prepared this article, including its revision. Y.-B.J. provide the suggestions regarding the design, measurement, and preparation of this article and is the supervisor of the research group. All authors have read and agreed to the published version of the manuscript. Funding: This research received no external funding. Institutional Review Board Statement: Not applicable. Informed Consent Statement: Not applicable. Data Availability Statement: Not applicable. Conﬂicts of Interest: The authors declare no conﬂict of interest.

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