DOCSLIB.ORG

Switched-Capacitor Circuits

Total Page：16

File Type：pdf, Size：1020Kb

Download full-text PDF Read full-text
Switched-Capacitor Circuits Switched-Capacitor Circuits David Johns and Ken Martin University of Toronto ([email protected]) ([email protected]) University of Toronto 1 of 60 © D. Johns, K. Martin, 1997 Basic Building Blocks Opamps • Ideal opamps usually assumed. • Important non-idealities — dc gain: sets the accuracy of charge transfer, hence, transfer-function accuracy. — unity-gain freq, phase margin & slew-rate: sets the max clocking frequency. A general rule is that unity-gain freq should be 5 times (or more) higher than the clock-freq. — dc offset: Can create dc offset at output. Circuit techniques to combat this which also reduce 1/f noise. University of Toronto 2 of 60 © D. Johns, K. Martin, 1997 Basic Building Blocks Double-Poly Capacitors metal C1 metal poly1 Cp1 thin oxide bottom plate C1 poly2 Cp2 thick oxide C p1 Cp2 (substrate - ac ground) cross-section view equivalent circuit • Substantial parasitics with large bottom plate capacitance (20 percent of C1) • Also, metal-metal capacitors are used but have even larger parasitic capacitances. University of Toronto 3 of 60 © D. Johns, K. Martin, 1997 Basic Building Blocks Switches I I Symbol n-channel v1 v2 v1 v2 I transmission I I gate v1 v p-channel v 2 1 v2 I • Mosfet switches are good switches. — off-resistance near G: range — on-resistance in 100: to 5k: range (depends on transistor sizing) • However, have non-linear parasitic capacitances. University of Toronto 4 of 60 © D. Johns, K. Martin, 1997 Basic Building Blocks Non-Overlapping Clocks I1 T Von I I1 Voff n – 2 n – 1 n n + 1 tTe delay 1 I fs { --- delay V 2 T on I Voff 2 n – 32e n – 12e n + 12e tTe • Non-overlapping clocks — both clocks are never on at same time • Needed to ensure charge is not inadvertently lost. • Integer values occur at end of I1. • End of I2 is 1/2 off integer value. University of Toronto 5 of 60 © D. Johns, K. Martin, 1997 Switched-Capacitor Resistor Equivalent I I 1 2 Req V V V1 V2 1 2 C1 T 'QC= V – V every clock period Req = ------ 1 1 2 C1 Qx = CxVx (1) • C1 charged to V1 and then V2 during each clk period. 'Q1 = C1 V1 – V2 (2) • Find equivalent average current C V – V I = ------------------------------1 1 2 (3) avg T where T is the clk period. University of Toronto 6 of 60 © D. Johns, K. Martin, 1997 Switched-Capacitor Resistor Equivalent • For equivalent resistor circuit V1 – V2 Ieq = ------------------ (4) Req • Equating two, we have T 1 Req ==------ ---------- (5) C1 C1fs • This equivalence is useful when looking at low-freq portion of a SC-circuit. • For higher frequencies, discrete-time analysis is used. University of Toronto 7 of 60 © D. Johns, K. Martin, 1997 Resistor Equivalence Example • What is the equivalent resistance of a 5pF capacitance sampled at a clock frequency of 100kHz. • Using (5), we have 1 R ==---------------------------------------------------------2M: eq –12 3 510u 100u 10 • Note that a very large equivalent resistance of 2M: can be realized. • Requires only 2 transistors, a clock and a relatively small capacitance. • In a typical CMOS process, such a large resistor would normally require a huge amount of silicon area. University of Toronto 8 of 60 © D. Johns, K. Martin, 1997 Parasitic-Sensitive Integrator vc2()nT I1 I2 C vcx()t 2 I1 vci()t vco()t vc1()t C1 v ()n = v ()nT vi()n = vci()nT o co • Start by looking at an integrator which IS affected by parasitic capacitances • Want to find output voltage at end of I1 in relation to input sampled at end of I1. University of Toronto 9 of 60 © D. Johns, K. Martin, 1997 Parasitic-Sensitive Integrator C2 C2 v ()nT– T ci vci()nT– T e 2 C 1 v ()nT– T C1 co vco()nT– T e 2 I1 on I2 on • At end of I2 C2vco()nT– T e 2 = C2vco()nT– T – C1vci()nT– T (6) • But would like to know the output at end of I1 C2vco()nT = C2vco()nT– T e 2 (7) • leading to C2vco()nT = C2vco()nT– T – C1vci()nT– T (8) University of Toronto 10 of 60 © D. Johns, K. Martin, 1997 Parasitic-Sensitive Integrator • Modify above to write C1 vo()n = vo()n – 1 – ------ vi()n – 1 (9) C2 and taking z-transform and re-arranging, leads to Vo()z §·C1 1 Hz(){ ------------ = –¨¸------ ----------- (10) Vi()z ©¹C2 z – 1 • Note that gain-coefficient is determined by a ratio of two capacitance values. • Ratios of capacitors can be set VERY accurately on an integrated circuit (within 0.1 percent) • Leads to very accurate transfer-functions. University of Toronto 11 of 60 © D. Johns, K. Martin, 1997 Typical Waveforms I1 t I2 t vci()t t vcx()t t vco()t t University of Toronto 12 of 60 © D. Johns, K. Martin, 1997 Low Frequency Behavior • Equation (10) can be re-written as §·C z–12/ Hz()= – ------1 ---------------------------- (11) ¨¸12/ –12/ ©¹C2 z – z • To find freq response, recall ze==jZT cos ZT + jsin ZT (12) ZT ZT z12/ = cos§·------- + jsin§·------- (13) ©¹2 ©¹2 ZT ZT z–12/ = cos§·------- – jsin§·------- (14) ©¹2 ©¹2 ZT ZT cos§·------- – jsin§·------- jZT §·C1 ©¹2 ©¹2 He()= –¨¸------ --------------------------------------------------- (15) C ZT ©¹2 j2sin§·------- ©¹2 University of Toronto 13 of 60 © D. Johns, K. Martin, 1997 Low Frequency Behavior • Above is exact but when ZT « 1 (i.e., at low freq) jZT §·C1 1 He()# –¨¸------ --------- (16) ©¹C2 jZT • Thus, the transfer function is same as a continuous- time integrator having a gain constant of C1 1 KI # ------ --- (17) C2T which is a function of the integrator capacitor ratio and clock frequency only. University of Toronto 14 of 60 © D. Johns, K. Martin, 1997 Parasitic Capacitance Effects Cp3 Cp4 C2 I1 I2 I1 vi()n vo()n C1 Cp1 Cp2 • Accounting for parasitic capacitances, we have §·C1 + Cp1 1 Hz()= –¨¸---------------------- ----------- (18) ©¹C2 z – 1 • Thus, gain coefficient is not well controlled and partially non-linear (due to Cp1 being non-linear). University of Toronto 15 of 60 © D. Johns, K. Martin, 1997 Parasitic-Insensitive Integrators C2 I1 I C 2 1 I v ()t 1 ci vco()t I2 I1 vi()n = vci()nT vo()n = vco()nT • By using 2 extra switches, integrator can be made insensitive to parasitic capacitances — more accurate transfer-functions — better linearity (since non-linear capacitances unimportant) University of Toronto 16 of 60 © D. Johns, K. Martin, 1997 Parasitic-Insensitive Integrators C 2 C2 vci()nT– T vci()nT– T e 2 + - C - 1 C1 vco()nT– T + vco()nT– T e 2 I1 on I2 on • Same analysis as before except that C1 is switched in polarity before discharging into C2. Vo()z §·C1 1 Hz(){ ------------ = ¨¸------ ----------- (19) Vi()z ©¹C2 z – 1 • A positive integrator (rather than negative as before) University of Toronto 17 of 60 © D. Johns, K. Martin, 1997 Parasitic-Insensitive Integrators Cp3 Cp4 C2 I1 I C 2 1 I1 vi()n vo()n I2 I1 Cp1 Cp2 • Cp3 has little effect since it is connected to virtual gnd • Cp4 has little effect since it is driven by output • Cp2 has little effect since it is either connected to virtual gnd or physical gnd. University of Toronto 18 of 60 © D. Johns, K. Martin, 1997 Parasitic-Insensitive Integrators • Cp1 is continuously being charged to vi()n and discharged to ground. • I1 on — the fact that Cp1 is also charged to vi()n – 1 does not affect C1 charge. • I2 on — Cp1 is discharged through the I2 switch attached to its node and does not affect the charge accumulating on C2. • While the parasitic capacitances may slow down settling time behavior, they do not affect the discrete- time difference equation University of Toronto 19 of 60 © D. Johns, K. Martin, 1997 Parasitic-Insensitive Inverting Integrator C2 I1 I1 C1 I1 vi()n vo()n Vi()z I2 I2 Vo()z C2vco()nT– T e 2 = C2vco()nT– T (20) C2vco()nT = C2vco()nT– T e 2 – C1vci()nT (21) • Present output depends on present input(delay-free) Vo()z §·C1 z Hz(){ ------------ = –¨¸------ ----------- (22) Vi()z ©¹C2 z – 1 • Delay-free integrator has negative gain while delaying integrator has positive gain. University of Toronto 20 of 60 © D. Johns, K. Martin, 1997 Signal-Flow-Graph Analysis C1 V1()z I I 1 C 2 2 CA V2()z I I 2 1 I1 V ()z I1 I1 o C3 V3()z I2 I2 –1 –C1 1 – z V1()z –1 C2z 1 1 §·------- ---------------- V ()z –1 V ()z 2 ©¹CA 1 – z o –C3 V3()z University of Toronto 21 of 60 © D. Johns, K. Martin, 1997 First-Order Filter Vin()s Vout()s • Start with an active-RC structure and replace resistors with SC equivalents. • Analyze using discrete-time analysis. University of Toronto 22 of 60 © D. Johns, K. Martin, 1997 First-Order Filter I I 1 1 I1 I1 C2 C3 I2 I2 I2 I2 CA I C1 1 Vi()z Vo()z –C3 –C2 1 1 §·------- ---------------- V ()z Vi()z –1 o ©¹CA 1 – z –1 –C1 1 – z University of Toronto 23 of 60 © D. Johns, K. Martin, 1997 First-Order Filter –1 –1 CA 1 – z Vo()z = – C3Vo()z – C2Vi()z – C1 1 – z Vi()z (23) §·C1 §·C2 ¨¸------- 1 – z–1 + ¨¸------- ©¹CA ©¹CA –------------------------------------------------------ Vo()z C Hz(){ ------------ = 1 – z–1 + ------3- (24) V ()z i CA §·C1 + C2 C1 ¨¸------------------- z – ------- ©¹CA CA = –----------------------------------------- §·C3 ¨¸1 + ------- z – 1 ©¹CA University of Toronto 24 of 60 © D. Johns, K. Martin, 1997 First-Order Filter • The pole of (24) is found by equating the denominator to zero CA zp = -------------------- (25) CA + C3 • For positive capacitance values, this pole is restricted to the real axis between 0 and 1 — circuit is always stable.
Load more

Recommended publications
  • Run Capacitor Info-98582
    Run Cap Quiz-98582_Layout 1 4/16/15 10:26 AM Page 1 ® Run Capacitor Info WHAT IS A CAPACITOR? Very simply, a capacitor is a device that stores and discharges electrons. While you may hear capacitors referred to by a variety of names (condenser, run, start, oil, etc.) all capacitors are comprised of two or more metallic plates separated by an insulating material called a dielectric. PLATE 1 PLATE 2 A very simple capacitor can be made with two plates separated by a dielectric, in this PLATE 1 PLATE 2 case air, and connected to a source of DC current, a battery. Electrons will flow away from plate 1 and collect on plate 2, leaving it with an abundance of electrons, or a “charge”. Since current from a battery only flows one way, the capacitor plate will stay BATTERY + – charged this way unless something causes current flow. If we were to short across the plates with a screwdriver, the resulting spark would indicate the electrons “jumping” from plate 2 to plate 1 in an attempt to equalize. As soon as the screwdriver is removed, plate 2 will again collect a PLATE 1 PLATE 2 charge. + BATTERY – Now let’s connect our simple capacitor to a source of AC current and in series with the windings of an electric motor. Since AC current alternates, first one PLATE 1 PLATE 2 MOTOR plate, then the other would be charged and discharged in turn. First plate 1 is charged, then as the current reverses, a rush of electrons flow from plate 1 to plate 2 through the motor windings.
    [Show full text]
  • Switched Capacitor Concepts & Circuits
    Switched Capacitor Concepts & Circuits Outline • Why Switched Capacitor circuits? – Historical Perspective – Basic Building Blocks • Switched Capacitors as Resistors • Switched Capacitor Integrators – Discrete time & charge transfer concepts – Parasitic insensitive circuits • Signal Flow Graphs • Switched Capacitor Filters – Comparison to Active RC filters – Advantages of Fully Differential filters • Switched Capacitor Gain Circuits • Reducing the Effects of Charge Injection • Tradeoff between Speed and Charge Injection Why Switched Capacitor Circuits? • Historical Perspective – As MOS processes came to the forefront in the late 1970s and early 1980s, the advantages of integrating analog blocks such as active filters on the same chip with digital logic became a driving force for inovation. – Integrating active filters using resistors and capacitors to acturately set time constants has always been difficult, because of large process variations (> +/- 30%) and the fact that resistors and capacitors don’t naturally match each other. – So, analog engineers turned to the building blocks native to MOS processes to build their circuits, switches & capacitors. Since time constants can be set by the ratio of capacitors, very accurate filter responses became possible using switched capacitor techniques Æ Mixed-Signal Design was born! Switched Capacitor Building Blocks • Capacitors: poly-poly, MiM, metal sandwich & finger caps • Switches: NMOS, PMOS, T-gate • Op Amps: at first all NMOS designs, now CMOS Non-Overlapping Clocks • Non-overlapping clocks are used to insure that one set of switches turns off before the next set turns on, so that charge only flows where intended. (“break before make”) • Note the notation used to indicate time based on clock periods: ... (n-1)T, (n-½)T, nT, (n+½)T, (n+1)T ..
    [Show full text]
  • Printed Circuit Board Mount Switches Shock Proof • Waterproof • Explosion
    shock proof • waterproof • explosion proof Printed Circuit Board Mount Switches These versatile switches are a great choice for many applications due to their small size and variety of connection styles. Although these switches are built to connect to a printed circuit board, they can also be retrotted for nearly any application by connecting wire leads. These switches can sense a pressure ranging from 6 inches of water all the way up to 65 PSI. For a frame of reference, a trumpet player blows 55 inches of water (or 2 PSI) on average, so these switches have a broad range. They can also be used to sense a vacuum ranging between 6 inches of water to 65 inches of water. Therefore, these miniature single pole and double pole switches are used as pressure, vacuum, or dierential pressure switches where moderate accuracy is sucient. Presair switches deliver critical benefits: CSPSSGA - PC Mount Switch SAFE: Presair switches deliver complete electrical isolation with zero voltage at the actuator to shock the user or spark an explosion. 1”x1”x1.5” ECONOMICAL:Presair’s switching system costs are comparable or lower than digital controls meeting the needs of original equipment manufacturers. ACCURATE: All switches are 100% tested to meet 100,000+ cycle life. General Specification: RATING: 1 amp resistive 250 VAC Ratings are dependent on actuation pressure and must be derated at lower pressures. APPROVAL: UL Recognized, CUL Recognized. File #E80254 MATERIAL: Lower body: Rynite w/ pin terminals molded Upper body: Acetal Diaphram material is dependent on application. PRESSURE RANGE: When used as a pressure or vacuum switch the actuation point must be factory set.
    [Show full text]
  • Kirchhoff's Laws in Dynamic Circuits
    Kirchhoff’s Laws in Dynamic Circuits Dynamic circuits are circuits that contain capacitors and inductors. Later we will learn to analyze some dynamic circuits by writing and solving differential equations. In these notes, we consider some simpler examples that can be solved using only Kirchhoff’s laws and the element equations of the capacitor and the inductor. Example 1: Consider this circuit Additionally, we are given the following representations of the voltage source voltage and one of the resistor voltages: ⎧⎧10 V fortt<< 0 2 V for 0 vvs ==⎨⎨and 1 −5t ⎩⎩20 V forte>+ 0 8 4 V fort> 0 We wish to express the capacitor current, i 2 , as a function of time, t. Plan: First, apply Kirchhoff’s voltage law (KVL) to the loop consisting of the source, resistor R1 and the capacitor to determine the capacitor voltage, v 2 , as a function of time, t. Next, use the element equation of the capacitor to determine the capacitor current as a function of time, t. Solution: Apply Kirchhoff’s voltage law (KVL) to the loop consisting of the source, resistor R1 and the capacitor to write ⎧ 8 V fort < 0 vvv12+−=ss0 ⇒ v2 =−= vv 1⎨ −5t ⎩16− 8et V for> 0 Use the element equation of the capacitor to write ⎧ 0 A fort < 0 dv22 dv ⎪ ⎧ 0 A fort < 0 iC2 ==0.025 =⎨⎨d −5t =−5t dt dt ⎪0.025() 16−> 8et for 0 ⎩1et A for> 0 ⎩ dt 1 Example 2: Consider this circuit where the resistor currents are given by ⎧⎧0.8 A fortt<< 0 0 A for 0 ii13==⎨⎨−−22ttand ⎩⎩0.8et−> 0.8 A for 0 −0.8 e A fort> 0 Express the inductor voltage, v 2 , as a function of time, t.
    [Show full text]
  • Switched Capacitor Instrumentation Amplifier
    LTC1043 Dual Precision Instrumentation Switched Capacitor Building Block FEATURES DESCRIPTIO U ■ Instrumentation Front End with 120dB CMRR The LTC®1043 is a monolithic, charge-balanced, dual ■ Precise, Charge-Balanced Switching switched capacitor instrumentation building block. A pair ■ Operates from 3V to 18V of switches alternately connects an external capacitor to ■ Internal or External Clock an input voltage and then connects the charged capacitor ■ Operates up to 5MHz Clock Rate across an output port. The internal switches have a ■ Low Power break-before-make action. An internal clock is provided ■ Two Independent Sections with One Clock and its frequency can be adjusted with an external capacitor. The LTC1043 can also be driven with an external APPLICATIOU S CMOS clock. The LTC1043, when used with low clock frequencies, ■ Precision Instrumentation Amplifiers provides ultra precision DC functions without requiring ■ Ultra Precision Voltage Inverters, Multipliers precise external components. Such functions are and Dividers differential voltage to single-ended conversion, voltage ■ V–F and F–V Converters inversion, voltage multiplication and division by 2, 3, 4, 5, ■ Sample-and-Hold etc. The LTC1043 can also be used for precise V–F and ■ Switched Capacitor Filters F–V circuits without trimming, and it is also a building block for switched capacitor filters, oscillators and modulators. The LTC1043 is manufactured using Linear Technology’s enhanced LTCMOSTM silicon gate process. , LTC and LT are registered trademarks of Linear Technology
    [Show full text]
  • Capacitive Voltage Transformers: Transient Overreach Concerns and Solutions for Distance Relaying
    Capacitive Voltage Transformers: Transient Overreach Concerns and Solutions for Distance Relaying Daqing Hou and Jeff Roberts Schweitzer Engineering Laboratories, Inc. Revised edition released October 2010 Previously presented at the 1996 Canadian Conference on Electrical and Computer Engineering, May 1996, 50th Annual Georgia Tech Protective Relaying Conference, May 1996, and 49th Annual Conference for Protective Relay Engineers, April 1996 Previous revised edition released July 2000 Originally presented at the 22nd Annual Western Protective Relay Conference, October 1995 CAPACITIVE VOLTAGE TRANSFORMERS: TRANSIENT OVERREACH CONCERNS AND SOLUTIONS FOR DISTANCE RELAYING Daqing Hou and Jeff Roberts Schweitzer Engineering Laboratories, Inc. Pullman, W A USA ABSTRACT Capacitive Voltage Transformers (CVTs) are common in high-voltage transmission line applications. These same applications require fast, yet secure protection. However, as the requirement for faster protective relays grows, so does the concern over the poor transient response of some CVTs for certain system conditions. Solid-state and microprocessor relays can respond to a CVT transient due to their high operating speed and iflCreased sensitivity .This paper discusses CVT models whose purpose is to identify which major CVT components contribute to the CVT transient. Some surprises include a recom- mendation for CVT burden and the type offerroresonant-suppression circuit that gives the least CVT transient. This paper also reviews how the System Impedance Ratio (SIR) affects the CVT transient response. The higher the SIR, the worse the CVT transient for a given CVT . Finally, this paper discusses improvements in relaying logic. The new method of detecting CVT transients is more precise than past detection methods and does not penalize distance protection speed for close-in faults.
    [Show full text]
  • Switched-Capacitor Integrator
    EE247 Lecture 10 • Switched-capacitor filters (continued) – Switched-capacitor integrators • DDI & LDI integrators – Effect of parasitic capacitance – Bottom-plate integrator topology – Switched-capacitor resonators – Bandpass filters – Lowpass filters – Switched-capacitor filter design considerations • Termination implementation • Transmission zero implementation • Limitations imposed by non-idealities EECS 247 Lecture 10 Switched-Capacitor Filters © 2008 H. K. Page 1 Switched-Capacitor Integrator C φ φ I φ 1 2 1 Vin - φ 2 Cs Vo + T=1/fs C C φ I φ I 1 2 Vin Vin - - C C s s Vo Vo + + φ High φ 1 2 High Æ C Charged to Vin s ÆCharge transferred from Cs to CI EECS 247 Lecture 10 Switched-Capacitor Filters © 2008 H. K. Page 2 Switched-Capacitor Integrator Output Sampled on φ1 φ φ 1 2 Vin CI φ - 1 Cs Vo Vo1 + φ φ φ φ φ Clock 1 2 1 2 1 Vin VCs Vo Vo1 EECS 247 Lecture 10 Switched-Capacitor Filters © 2008 H. K. Page 3 Switched-Capacitor Integrator ( (n-1)T n-3/2)Ts s (n-1/2)Ts nTs (n+1/2)Ts (n+1)Ts φ φ φ φ φ Clock 1 2 1 2 1 Vin Vs Vo Vo1 Φ 1 Æ Qs [(n-1)Ts]= Cs Vi [(n-1)Ts] , QI [(n-1)Ts] = QI [(n-3/2)Ts] Φ 2 Æ Qs [(n-1/2) Ts] = 0 , QI [(n-1/2) Ts] = QI [(n-1) Ts] + Qs [(n-1) Ts] Φ 1 _Æ Qs [nTs ] = Cs Vi [nTs ] , QI [nTs ] = QI[(n-1) Ts ] + Qs [(n-1) Ts] Since Vo1= - QI /CI & Vi = Qs / Cs Æ CI Vo1(nTs) = CI Vo1 [(n-1) Ts ] -Cs Vi [(n-1) Ts ] EECS 247 Lecture 10 Switched-Capacitor Filters © 2008 H.
    [Show full text]
  • Practical Issues Designing Switched-Capacitor Circuit
    Practical Issues Designing Switched-Capacitor Circuit ECEN 622 (ESS) Fall 2011 Practical Issues Designing Switched-Capacitor Circuit Material partially prepared by Sang Wook Park and Shouli Yan ELEN 622 Fall 2011 1 / 27 Switched-Capacitor practical issues Practical Issues Designing Switched-Capacitor Circuit MOS switch G G S Cov Cox Cov D S D Ron o Excellent Roff o Non-idea Effect Charge injection, Clock feed-through Finite and nonlinear Ron ELEN 622 Fall 2011 2 / 27 Switched-Capacitor practical issues Practical Issues Designing Switched-Capacitor Circuit Charge Injection G Qch1 Qch2 C VS o During TR. is turned on, Qch is formed at channel surface Qch = WLC OX (VGS −Vth ) When TR. is off, Qch1 is absorbed by Vs, but Qch2 is injected to C o Charge injected through overlap capacitor o Appeared as an offset voltage error on C ELEN 622 Fall 2011 3 / 27 Switched-Capacitor practical issues Practical Issues Designing Switched-Capacitor Circuit Charge Injection Effect CLK Ideal sw. Vout MOS sw. 0.1pF 1V CLK o When clock changes from high to low, Qch2 is injected to C o Compared to ideal sw., MOS sw. creates voltage error on Vout ELEN 622 Fall 2011 4 / 27 Switched-Capacitor practical issues Practical Issues Designing Switched-Capacitor Circuit Decrease Charge Injection Effect (1) CLK Vout W/L = 1/0.4 0.1pF 1V W/L = 10/0.4 o Decrease the effect of Qch o Use either bigger C or small TR. (small ratio of Cox/C) o Increased Ron ELEN 622 Fall 2011 5 / 27 Switched-Capacitor practical issues Practical Issues Designing Switched-Capacitor Circuit Decrease Charge Injection Effect (2) CLK CLKb 10/0.4 3.1/0.4 Vout With dummy sw.
    [Show full text]
  • Introduction How Circuit Protection Devices Work?
    Introduction Adding extra strain to any person or object can be a recipe for disaster. This is especially the case for electrical circuits. When they’re tasked with carrying more current than they were designed to handle, the added burden can lead to detrimental and dangerous circumstances. Not only could overloading your circuits damage or destroy your sensitive electronic equipment, but it could also generate extra heat in wires that weren’t meant to carry the load. If this happens, it can cause a fire in a matter of seconds. This is where an overcurrent protection device, such as a fused disconnect switch or a circuit breaker, comes in. Though they serve a similar purpose, these two components each have a unique design. Today, we’re sharing how they work and how to choose the right one for your project. Ready to learn more? Let’s dive in. How Circuit Protection Devices Work? The field of circuit protection technology is vast, designed to prevent circuits from risks associated with overvoltage, overcurrent, reverse-bias, electrostatic-discharge (ESD) and overtemperature events. Such risks include: • High-voltage transients • Capacitive coupling • Inductive kickback • Ground faults • High inrush currents From your smartphone battery to your steering wheel, these components are necessary to ensure safe and reliable electronics. While they all serve a valuable purpose, we’re delving deeper today into the specific field of overcurrent protection. Overcurrent protection devices are designed to disconnect or open a circuit quickly in the event that an overload or short-circuit occurs. This helps to mitigate any damage to the connected equipment and can also reduce the risk of electrical fires.
    [Show full text]
  • Aluminum Electrolytic Vs. Polymer – Two Technologies – Various Opportunities
    Aluminum Electrolytic vs. Polymer – Two Technologies – Various Opportunities By Pierre Lohrber BU Manager Capacitors Wurth Electronics @APEC 2017 2017 WE eiCap @ APEC PSMA 1 Agenda Electrical Parameter Technology Comparison Application 2017 WE eiCap @ APEC PSMA 2 ESR – How to Calculate? ESR – Equivalent Series Resistance ESR causes heat generation within the capacitor when AC ripple is applied to the capacitor Maximum ESR is normally specified @ 120Hz or 100kHz, @20°C ESR can be calculated like below: ͕ͨ͢ 1 1 ͍̿͌ Ɣ Ɣ ͕ͨ͢ ∗ ͒ ͒ Ɣ Ɣ 2 ∗ ∗ ͚ ∗ ̽ 2 ∗ ∗ ͚ ∗ ̽ ! ∗ ̽ 2017 WE eiCap @ APEC PSMA 3 ESR – Temperature Characteristics Electrolytic Polymer Ta Polymer Al Ceramics 2017 WE eiCap @ APEC PSMA 4 Electrolytic Conductivity Aluminum Electrolytic – Caused by the liquid electrolyte the conductance response is deeply affected – Rated up to 0.04 S/cm Aluminum Polymer – Solid Polymer pushes the conductance response to much higher limits – Rated up to 4 S/cm 2017 WE eiCap @ APEC PSMA 5 Electrical Values – Who’s Best in Class? Aluminum Electrolytic ESR approx. 85m Ω Tantalum Polymer Ripple Current rating approx. ESR approx. 200m Ω 630mA Ripple Current rating approx. 1,900mA Aluminum Polymer ESR approx. 11m Ω Ripple Current rating approx. 5,500mA 2017 WE eiCap @ APEC PSMA 6 Ripple Current >> Temperature Rise Ripple current is the AC component of an applied source (SMPS) Ripple current causes heat inside the capacitor due to the dielectric losses Caused by the changing field strength and the current flow through the capacitor 2017 WE eiCap @ APEC PSMA 7 Impedance Z ͦ 1 ͔ Ɣ ͍̿͌ ͦ + (͒ −͒ )ͦ Ɣ ͍̿͌ ͦ + 2 ∗ ∗ ͚ ∗ ͍̿͆ − 2 ∗ ∗ ͚ ∗ ̽ 2017 WE eiCap @ APEC PSMA 8 Impedance Z Impedance over frequency added with ESR ratio 2017 WE eiCap @ APEC PSMA 9 Impedance @ High Frequencies Aluminum Polymer Capacitors have excellent high frequency characteristics ESR value is ultra low compared to Electrolytic’s and Tantalum’s within 100KHz~1MHz E.g.
    [Show full text]
  • Measurement Error Estimation for Capacitive Voltage Transformer by Insulation Parameters
    Article Measurement Error Estimation for Capacitive Voltage Transformer by Insulation Parameters Bin Chen 1, Lin Du 1,*, Kun Liu 2, Xianshun Chen 2, Fuzhou Zhang 2 and Feng Yang 1 1 State Key Laboratory of Power Transmission Equipment & System Security and New Technology, Chongqing University, Chongqing 400044, China; [email protected] (B.C.); [email protected] (F.Y.) 2 Sichuan Electric Power Corporation Metering Center of State Grid, Chengdu 610045, China; [email protected] (K.L.); [email protected] (X.C.); [email protected] (F.Z.) * Correspondence: [email protected]; Tel.: +86-138-9606-1868 Academic Editor: K.T. Chau Received: 01 February 2017; Accepted: 08 March 2017; Published: 13 March 2017 Abstract: Measurement errors of a capacitive voltage transformer (CVT) are relevant to its equivalent parameters for which its capacitive divider contributes the most. In daily operation, dielectric aging, moisture, dielectric breakdown, etc., it will exert mixing effects on a capacitive divider’s insulation characteristics, leading to fluctuation in equivalent parameters which result in the measurement error. This paper proposes an equivalent circuit model to represent a CVT which incorporates insulation characteristics of a capacitive divider. After software simulation and laboratory experiments, the relationship between measurement errors and insulation parameters is obtained. It indicates that variation of insulation parameters in a CVT will cause a reasonable measurement error. From field tests and calculation, equivalent capacitance mainly affects magnitude error, while dielectric loss mainly affects phase error. As capacitance changes 0.2%, magnitude error can reach −0.2%. As dielectric loss factor changes 0.2%, phase error can reach 5′.
    [Show full text]
  • Power Sense Switch & Fan Control Thermostat
    Speed Controllers & Switches_8_DR_ref.qxd 6/11/2015 10:34 AM Page 17 POWER SENSE SWITCH & FAN CONTROL THERMOSTAT POWER SENSE SWITCH TYPE - VZ-IISNSE TECHNICAL DATA Switched Output Model Number Load Sense Range Max. Amps Enclosure Size, mm On = 2.5 - 15 amp VZ-ISNSE 5 153W x 110H x 60D Off = 0 - 1.5 amps SUGGESTED WIRING ARRANGEMENT The VZ-ISNSE is a 240V AC power sense switch that provides automatic on/off control of a fan or exhaust fan system. The unit detects the current of an appliance such as a clothes dryer, closing and opening the switched output 240V relay when the load sense range Switched Output parameters are met. Can be used with the VZ2-10TS, VZM0-28TS, VZ6-4PL or as a stand alone unit. In LED’s indicate when load is activated. 3-pin Socket Warning Not to be used with fans with EC motors Appliance or inverters. FAN CONTROL THERMOSTAT TECHNICAL DATA Model Temperature Maximum Number range Mounting Amps Dimensions, mm TFC6 5ºC to 30ºC Wall mountable 6.0 86W x 86H x 33D SELECTION TABLE The TFC6 is suitable for most 240-volt fans. The TFC6 can be set in two modes: Cool mode - will start the fan when the room temperature is higher than the set point. The Fantech Fan Control Thermostat has Heat mode - will start the fan when the room temperature is lower than the set point. been developed to control the operation of a 240-volt fan based on the setting made on the thermostat dial. It can be set to turn on a fan when the WIRING DIAGRAM room temperature is either higher or lower than the set point.
    [Show full text]