AN66-1 Application Note 66

Application Note 66 August 1996

Linear Technology Magazine Circuit Collection, Volume II Power Products Richard Markell, Editor

INTRODUCTION Application Note 66 is a compendium of “power circuits” included here are circuits that provide 300W or more of from the first five years of Linear Technology. The objective power factor corrected DC from a universal input. Battery is to collect the useful circuits from the magazine into chargers are included, some that charge several battery several applications notes (another, AN67, will collect types, some that are optimized to charge a single type. signal processing circuits into one Application Note) so MOSFET drivers, high side switches and H-bridge driver that valuable “gems” will not be lost. This Application Note circuits are also included, as is an article on simple thermal contains circuits that can power most any system you can analysis. With these introductory remarks, I’ll stand aside imagine, from desktop computer systems to micropower and let the authors describe their circuits. systems for portable and handheld equipment. Also

ARTICLE INDEX REGULATORS—SWITCHING (BUCK) High Power (>4A) Big Power for Big Processors: The LTC®1430 Synchronous Regulator ...... 4 Applications for the LTC1266 Switching Regulator ...... 5 A High Efficiency 5V to 3.3V/5A Converter ...... 7 High Current, Synchronous Step-Down Switching Regulator ...... 8 Medium Power (1A to 4A) 1MHz Step-Down Converter Ends 455kHz IF Woes ...... 10 High Output Voltage Buck Regulator ...... 11 The LTC1267 Dual Switching Regulator Controller Operates from High Input Voltages...... 12 High Efficiency 5V to 3.3V/1.25A Converter in 0.6 Square Inches...... 13 LT ®1074/LT1076 Adjustable 0V to 5V Power Supply...... 14 Triple Output 3.3V, 5V and 12V High Efficiency Notebook Power Supply ...... 15 The New SO-8 LTC1147 Switching Regulator Controller Offers High Efficiency in a Small Footprint...... 17 The LT1432: 5V Regulator Achieves 90% Efficiency ...... 20 Low Power (<1A) Applications for the LTC1265 High Efficiency Monolithic Buck Converter...... 22 REGULATORS—SWITCHING (BOOST) Medium Power (1A to 4A) High Output Current Boost Regulator...... 24 Low Power (<1A) Applications for the LT1372 500kHz Switching Regulator ...... 25

, LTC and LT are registered trademarks of Linear Technology Corporation.

AN66-1 Application Note 66

REGULATORS—SWITCHING (BUCK/BOOST) ±5V Converter Uses Off-the-Shelf Surface Mount Coil...... 27 Switching Regulator Provides Constant 5V Output from 3.5V to 40V Input Without a Transformer ...... 28 Switching Regulator Provides ±15V Output from an 8V to 40V Input Without a Transformer ...... 29 REGULATORS—SWITCHING (INVERTING) High Efficiency 12V to –12V Converter ...... 32 Regulated Charge Pump Power Supply...... 34 Applications for the LTC1265 High Efficiency Monolithic Buck Converter...... 22 LTC1174: A High Efficiency Buck Converter ...... 35 REGULATORS—SWITCHING (FLYBACK) Applications for the LT1372 500kHz Switching Regulator ...... 25 REGULATORS—SWITCHING (POWER FACTOR CORRECTED) The New LT1508/LT1509 Combines Power Factor Correction and a PWM in a Single Package ...... 37 REGULATORS—SWITCHING (DISCUSSION) Adding Features to the Boost Topology...... 39 Sensing Negative Outputs ...... 40 REGULATORS—SWITCHING (MICROPOWER) 3-Cell to 3.3V Buck/Boost Converter ...... 41 LT1111 Isolated 5V Switching Power Supply...... 41 Low Noise Portable Communications DC/DC Converter...... 43 Applications for the LT1302 Micropower DC/DC Converter ...... 44 Clock-Synchronized Switching Regulator Has Coherent Noise ...... 49 Battery-Powered Circuits Using the LT1300 and LT1301 ...... 51 LTC1174: A High Efficiency Buck Converter ...... 35 Battery-Powered Circuits Using the LT1304 Micropower DC/DC Converter with Low-Battery Detector ...... 54 Automatic Load Sensing Saves Power in High Voltage Converter...... 57 REGULATORS—SWITCHING (MICROPOWER) Backlight High Efficiency EL Driver Circuit...... 58 A Low Power, Low Voltage CCFL Power Supply ...... 60 All Surface Mount EL Panel Driver Operates from 1.8V to 8V Input ...... 61 A Dual Output LCD Bias Voltage Generator ...... 62 LCD Bias Supply...... 63 REGULATORS—SWITCHING (MICROPOWER) Switched Capacitor Regulated Charge Pump Power Supply...... 34 REGULATORS—SWITCHING (MICROPOWER) VPP Generator LTC1262 Generates 12V for Programming Flash Memories Without Inductors ...... 64 Flash Memory VPP Generator Shuts Down with 0V Output ...... 64

AN66-2 Application Note 66

REGULATORS—LINEAR Low Noise Wireless Communications Power Supply ...... 65 An LT1123 Ultralow Dropout 5V Regulator ...... 66 REGULATORS—LINEAR Microprocessor Power LT1580 Low Dropout Regulator Uses New Approach to Achieve High Performance ...... 67 LT1585: New Linear Regulator Solves Load Transients ...... 68 BATTERY CHARGERS Charging NiMH/NiCd or Li-Ion with the LT1510 ...... 70 Lithium-Ion Battery Charger ...... 71 Simple Battery Charger Runs at 1MHz ...... 73 A Perfectly Temperature Compensated Battery Charger...... 74 A Simple 300mA NiCd Battery Charger ...... 75 High Efficiency (>90%) NiCd Battery Charger Circuit Programmable for 1.3A Fast Charge or 100mA Trickle Charge...... 76 POWER MANAGEMENT LT1366 Rail-to-Rail Amplifier Controls Topside Current ...... 78 An Isolated High Side Driver ...... 79 LTC1163: 2-Cell Power Management ...... 80 LTC1157 Switch for 3.3V PC Card Power ...... 81 The LTC1157 Dual 3.3V Micropower MOSFET Driver ...... 82 The LTC1155 Does Laptop Computer Power Bus Switching, SCSI Termination Power or 5V/3A Extremely Low Dropout Regulator ...... 82 A Circuit That Smoothly Switches Between 3.3V and 5V...... 84 A Fully Isolated Quad 4A High Side Switch ...... 85 The LTC1153 Electronic Circuit Breaker ...... 86 LTC1477: 0.07Ω Protected High Side Switch Eliminates “Hot Swap” Glitching ...... 87 MISCELLANEOUS Protected Bias for GaAs Power Amplifiers ...... 88 LT1158 H-Bridge Uses Ground Referenced Current Sensing for System Protection...... 89 LT1158 Allows Easy 10A Locked Antiphase Motor Control ...... 91 All Surface Mount Programmable 0V, 3.3V, 5V and 12V VPP Generator for PCMCIA ...... 92 A Tachless Motor Speed Controller ...... 93 LT1161...And Back and Stop and Forward and Rest—All with No Worries at All ...... 95 Simple Thermal Analysis—A Real Cool Subject for LTC Regulators ...... 98 ALPHABETIC INDEX By Major Categories ...... 101

AN66-3 Application Note 66

Regulators—Switching (Buck) similar class processor and the input is taken from the system 5V ±5% supply. The LTC1430 provides the pre- High Power (>4A) cisely regulated output voltage required by the processor BIG POWER FOR BIG PROCESSORS: without the need for an external precision reference or THE LTC1430 SYNCHRONOUS REGULATOR trimming. Figure 1 shows a typical application with a by Dave Dwelley 3.30V ±1% output voltage and a 12A output current limit. The power MOSFETs are sized so as not to require a heat The LTC1430 is a new switching regulator controller sink under ambient temperature conditions up to 50°C. designed to be configured as a synchronous buck con- Typical efficiency is above 91% from 1A to 10A output verter with a minimum of external components. It runs at current and peaks at 95% at 5A (Figure 2). a fixed switching frequency (nominally 200kHz) and provides all timing and control functions, adjustable current Pentium is a registered trademark of Intel Corporation. limit and soft start, and level shifted output drivers de- 100 signed to drive an all N-channel synchronous buck con- VCC = 5V ° verter architecture. The switch driver outputs are capable 90 TA = 25 C VOUT = 3.3V of driving multiple paralleled power MOSFETs with 80 submicrosecond slew rates, providing high efficiency at very high current levels while eliminating the need for a 70 heat sink in most designs. The LTC1430 is usable in EFFICIENCY (%) 60 converter designs providing from a few amps to over 50A of output current, allowing it to supply 3.3V power to the 50 most current-hungry arrays of microprocessors. 40 0.1 1 10

A Typical 5V to 3.3V Application LOAD CURRENT (A) AN66 F02

The typical application for the LTC1430 is a 5V to 3.xV Figure 2. Efficiency Plot for Figure 1’s Circuit. Note That converter on a PC motherboard. The output is used to Efficiency Peaks at a Respectable 95% power a Pentium® processor, Pentium® Pro processor or

VIN 4.5V TO 5.5V

D1 C1 R1 R2 1N4148 0.1µF 16k 100Ω

SVCC PVCC2 IMAX PVCC1 M1B C3 MTD20N03HL C2+ 0.1µF M1A 10µF G1 L1 R3 MTD20N03HL µ SGND LTC1430 2.5 H/15A 1k V I OUT FB 3.3V NC FREQ G2 M2 MTD20N03HL SHUTDOWN SHDN +SENSE

COMP VTRIM NC + CIN + COUT CC* 220µF 330µF 3300pF SS –SENSE 10V 6.3V 100pF* × SGND PGND 4 × 6 RC* CSS 33k 0.01µF AN66 F01

PGND L1 =6 TURNS #16 WIRE ON MICROMETALS T50-52B CORE AND SGND CIN =4 EACH AVX TPSE 227M010R0100 SGND CONNECTED AT PGND COUT = 6 EACH AVX TPSE 337M006R0100 A SINGLE POINT *TRIM TO OPTIMIZE TRANSIENT REPONSE

Figure 1. Typical 5V to 3.3V, 10A LTC1430 Application

AN66-4 Application Note 66

The 12A current limit is set by the 16k resistor R1 from PVCC to IMAX and the 0.035Ω ON resistance of the MTD20N03HL MOSFETs (M1A, M1B).

The 0.1µF capacitor in parallel with R1 improves power 20mV/DIV supply rejection at IMAX, providing consistent current limit performance when voltage spikes are present at PVCC. 5A/DIV Soft start time is set by CSS; the 0.01µF value shown reacts with an internal 10µA pull-up to provide a 3ms start-up time. The 2.5µH, 15A inductor is sized to allow the peak AN66 F03 Figure 3. Transient Response: 0A to 5A Load Step current to rise to the full current limit value without Imposed on Figure 1’s Output saturating. This allows the circuit to withstand extended output short circuits without saturating the inductor core. largest value, lowest ESR capacitors that will fit the The inductor value is chosen as a compromise between design budget and space requirements. Several smaller peak ripple current and output current slew rate, which capacitors wired in parallel can help reduce total output affects large-signal transient response. If the output load capacitor ESR to acceptable levels. Input bypass capaci- is expected to generate large output current transients (as tor ESR is also important to keep input supply variations large microprocessors tend to do), the inductor value will to a minimum with 10AP-P square wave current pulses need to be quite low, in the 1µH to 10µH range. flowing into M1. AVX TPS series surface mount tantalum Loop compensation is critical for obtaining optimum capacitors and Sanyo OS-CON organic electrolytic ca- transient response with a voltage feedback system like pacitors are recommended for both input and output the LTC1430; the compensation components shown bypass duty. Low cost “computer grade” aluminum here give good response when used with the output electrolytics typically have much higher series resistance capacitor values and brands shown (Figure 3). The ESR and will significantly degrade performance. Don’t count of the output capacitor has a significant effect on the on that parallel 0.1µF ceramic cap to lower the ESR of a transient response of the system. For best results use the cheap electrolytic cap to acceptable levels.

APPLICATIONS FOR Figure 5 shows an LTC1266 in the charge pump configu- THE LTC1266 SWITCHING REGULATOR ration designed to provide a 3.3V/10A output from a single by Greg Dittmer supply. The Si4410s are new logic level, surface mount, N-channel MOSFETs from Siliconix that provide a mere Figures 4, 5 and 6 show the three basic circuit configura- 0.02Ω of on-resistance at VGS = 4.5V and thus provide a tions for the LTC1266. The all N-channel circuit shown in 10A solution with minimal components. The efficiency Figure 4 is a 3.3V/5A surface mount converter with the plot shows that the converter is still close to 90% efficient internal MOSFET drivers powered from a separate supply, at 10A. Because the charge pump configuration is used, PWR VIN. The VGS(ON) of the Si9410 N-channel MOSFETs the maximum allowable VIN is 18V/2 = 9V. Due to the high is 4.5V; thus the minimum allowable voltage for PWR VIN AC currents in this circuit we recommend low ESR is VIN(MAX) + 4.5V. At the other end, PWR VIN should be OS-CON or AVX input/output capacitors to maintain effi- kept under the maximum safe level of 18V, limiting VIN to ciency and stability. 18V – 4.5V = 13.5V. The current sense resistor value is chosen to set the maximum current to 5A according to the Figure 6 shows the conventional P-channel topside switch formula IOUT = 100mV/RSENSE. With VIN = 5V, the 5µH circuit configuration for implementing a 3.3V/3A regula- inductor and 130pF timing capacitor provide an operating tor. The P-channel configuration allows the widest pos- frequency of 175kHz and a ripple current of 1.25A. sible supply range of the three basic circuit configurations,

AN66-5 Application Note 66

3.5V to 18V, and provides extremely low dropout, exceed- The three application circuits demonstrate the fixed 3.3V ing that of most linear regulators. The low dropout results version of the LTC1266. The LTC1266 is also available in from the LTC1266’s ability to achieve a 100% duty cycle fixed 5V and adjustable versions. All three versions are when in P-channel mode. In N-channel mode the duty available in 16-pin SO packages. cycle is limited to less than 100% to ensure proper start- up and thus the dropout voltage for the all N-channel converters is slightly higher.

VIN 3.5V TO 14V + CIN D1 100µF 100 Si9410DY MBRS140T3 20V OSCON VIN = 5V × 2

95 1 16 0.1µF TDRIVE BDRIVE Si9410DY PWR V 2 15 IN PWR V PGND 90 (SEE TEXT) IN 3 LTC1266-3.3 14 PINV LB OUT L EFFICIENCY (%) 4 13 5µH 85 BINH BINH LBIN 5 12 VIN SGND 6 11 80 CT SHDN SHDN 0.01 0.1 1 5 7 10 LOAD CURRENT (A)

ITH NC COUT

CT CC µ AN66 F04b + 330 F 130pF 3300pF 8 9 SENSE – SENSE + 10V × 2 Figure 4b. Efficiency for Figure 4a’s Circuit R R C 1000pF SENSE 470Ω 0.02Ω VOUT 3.3V/5A AN66 F04a Figure 4a. All N-Channel 3.3V/5A Regulator with Drivers Powered from Seperate Power VIN (PWR VIN) Supply

VIN 4V TO 9V D1 CIN MBR0530T1 + µ Si4410DY MBRS340T3 100 F 10V 0.1µF OS-CON 100 × 3 VIN = 5V

1 16 TDRIVE BDRIVE Si4410DY 95 2 15 PWR VIN PGND 3 LTC1266-3.3 14 PINV LB 90 OUT L 4 13 5µH BINH BINH LB IN EFFICIENCY (%) 5 12 85 VIN SGND 6 11 CT SHDN SHDN 7 10 COUT 80 ITH NC 330µF 0.01 0.1 1 10

CT CC 220pF 3300pF 8 9 + 10V LOAD CURRENT (A) SENSE – SENSE + × 3 AN66 F05b RC R Ω SENSE 470 1000pF 0.01Ω VOUT 3.3V Figure 5b. Efficiency for Figure 5a’s Circuit 10A AN66 F05a

Figure 5a. All N-Channel Single Supply 5V to 3.3V/10A Regulator

AN66-6 Application Note 66

VIN 3.5V TO 18V + CIN D1 100µF Si9430DY MBRS140T3 25V 100 VIN = 5V

1 16 95 0.1µF TDRIVE BDRIVE Si9410DY 2 15 PWR VIN PGND 3 LTC1266-3.3 14 90 PINV LB OUT L 4 13 10µH BINH BINH LBIN EFFICIENCY (%) 5 12 85 VIN SGND 6 11 CT SHDN SHDN 80 7 10 COUT 0.01 0.1 1 3

ITH NC 220µF

CT CC LOAD CURRENT (A) + 10V 220pF 3300pF 8 – + 9 SENSE SENSE × 2 AN66 F06b R R C 1000pF SENSE 1k 0.033Ω Figure 6b. Efficiency for Figure 6a’s Circuit VOUT 3.3V

3A AN66 F06a

Figure 6a. Low Dropout 3.3V/3A Complementary MOSFET Regulator

A HIGH EFFICIENCY 5V TO 3.3V/5A CONVERTER High efficiency is mandatory in these applications, since by Randy G. Flatness converting 5V to 3.3V at 5A using a linear regulator would require dissipating over 8W. This wastes power and board The next generation of notebook and desktop computers space for heat sinking. is incorporating more 3.3V ICs alongside 5V devices. As the number of devices increases, the current require- The LTC1148 synchronous switching regulator controller ments also increase. Typically, a high current 5V supply is accomplishes the 5V to 3.3V conversion with high effi- already available. Thus, the problem is reduced to deriving ciencies over a wide load current range. The circuit shown 3.3V from 5V efficiently in a small amount of board space. in Figure 7 provides 3.3V at efficiencies greater than 90%

VIN 5V C1 = TANTALUM + C1 C2 + C3 µ C3 = SANYO (OS-CON) 20SA100M ESR = 0.037Ω I = 2.25A 1µF 0.1µF Q2 100 F RMS 3 C6 = AVX (TA) TPSE227K01R0080 ESR = 0.080Ω I = 1.285A Si9430DY 20V RMS Ω × 2 Q1, Q2 = SILICONIX PMOS BVDSS = 20V DCRON = 0.100 Qg = 50nC VIN 0V = NORMAL 10 1 Q1 L1 Q3 = SILICONIX NMOS BV = 30V DCR = 0.050Ω Q = 30nC SHDN PDRIVE R2 DSS ON g >2V = SHUTDOWN Si9430DY 27µH 0.02Ω VOUT D1 = MOTOROLA SCHOTTKY VBR = 30V LTC1148-3.3 8 3.3V R2 = KRL NP-2A-C1-0R020J Pd = 3W SENSE + 5A L1 = KOOL Mµ® CORE, 16 GAUGE C7 6 7 0.01µF COILTRONICS (408)241-7876 – ITH SENSE KRL BANTRY (603) 668-3210 C6 R1 4 14 + SILICONIX (800) 554-5565 Ω Q3 220µF 470 CT NDRIVE KOOL Mµ IS A REGISTERED TRADEMARK OF MAGNETICS, INC. C5 Si9410 D1 10V SGND PGND C4 680pF MBRS140T3 × 2 3300pF NPO 11 12

AN66 F07

Figure 7. LTC1148-3.3 High Efficiency 5V to 3.3V/5A Step-Down Converter

AN66-7 Application Note 66

100 maximize the operating efficiency at low output currents, Burst ModeTM operation is used to reduce switching losses. Synchronous switching, combined with Burst Mode op-

90 eration, yields very efficient energy conversion over a wide range of load currents. The top P-channel MOSFETs in Figure 7 will be on 2/3 of EFFICIENCY (%) 80 the time with an input of 5V. Hence, these devices should be carefully examined to obtain the best performance. Two MOSFETs are needed to handle the peak currents safely 70 and enhance high current efficiency. The LTC1148 can 1 10100 1000 10000 OUTPUT CURRENT (mA) drive both MOSFETs adequately without a problem. A AN66 F08 single N-channel MOSFET is used as the bottom synchro- Figure 8. Efficiency for 5V to 3.3V Synchronous Switcher nous switch, which shunts the Schottky diode. Finally, adaptive anti-shoot-though circuitry automatically pre- from 5mA to 5A (over three decades of load current). The vents cross conduction between the complementary efficiency of the circuit in Figure 7 is plotted in Figure 8. MOSFETs which can kill efficiency. At an output current of 5A the efficiency is 90%; this The circuit in Figure 7 has a no-load current of only 160µA. means only 1.8W are lost. This lost power is distributed In shutdown mode, with Pin 10 held high (above 2V), the among R , L1 and the power MOSFETs; thus heat SENSE quiescent current decreases to less than 20µA with all sinking is not required. MOSFETs held off DC. Although the circuit in Figure 7 is The LTC1148 series of controllers use constant off-time specified at a 5V input voltage, the circuit will function from current mode architecture to provide clean start-up, accu- 4V to 15V without requiring any component substitutions. rate current limit and excellent line and load regulation. To Burst Mode is a trademark of Linear Technology Corporation.

HIGH CURRENT, SYNCHRONOUS The circuit’s operation is as follows: the LTC1149 provides STEP-DOWN SWITCHING REGULATOR a P-drive output (Pin 4) that swings between ground and by Brian Huffman 10V, turning Q3 on and off. While Q3 is on, the N-channel MOSFET (Q4) is off because its gate is pulled low by Q3 The LTC1149 is a half-bridge driver designed for syn- through D2. During this interval, the Ngate output (Pin 13) chronous buck regulator applications. Normally a P- and turns the synchronous switch (Q5) on creating a low N-channel output stage is employed, but the P-channel resistance path for the inductor current. device ON resistance becomes a limiting factor at output currents above 2A. N-channel MOSFETs are better suited Q4 turns on when its gate is driven above the input voltage. for use in high current applications, since they have a This is accomplished by bootstrapping capacitor C2 off substantially lower ON resistance than comparably priced the drain of Q4. The LTC1149 VCC output (Pin 3) supplies P-channels. The circuit shown in Figure 9 adapts the a regulated 10V output that is used to charge C2 through LTC1149 to drive a half-bridge consisting of two D1 while Q4 is off. With Q4 off, C2 charges to 5V during the N-channel MOSFETs, providing efficiency in excess of first cycle in Burst Mode operation and to 10V thereafter. 90% at an output current of 5A.

AN66-8 Application Note 66

V IN + CIN 12V TO 36V µ D1 R4 1000 F 1N4148 220Ω 63V Q1 2N3906 C2 + R2 R3 0.1µF C1 Ω µ 10k 470 0.1 F Q2 2 D2 2N2222 VIN 1N4148 3 1 Q4 V PGATE CC MTP30N06EL + 5 4 C3 Q3 VCC P-DRIVE L1 R 3.3µF VN2222LL SENSE 16 50µH 0.02Ω CAP LTC1149-5 R5 5V 100Ω 10 + 9 + 5A SHDN1 SENSE C C4 OUT 0V = NORMAL 15 220µF SHDN2 – 8 0.001µF >2V = SHUTDOWN SENSE 10V 7 I × 2 TH R6 Q5 Ω 6 13 100 IRFZ34 R1 CT NGATE D3 1k C4 CT SGND PGND RGND MBR160 3300pF 820pF X7R NPO 11 12 14

C3(TA) LOW ESR Q4, Q5NMOS, BVDSS = 60V, RDSON = 0.05Ω CINNICHICON (AL) UPL1J102MRH, ESR = 0.027Ω, IRMS = 2.370A D1, D2SILICON, VBR = 75V COUTSANYO (OS-CON) 10SA220M, ESR = 0.035Ω, IRMS = 2.360A D3MOTOROLA SCHOTTKY, VBR = 60V Q1PNP, BVCEO = 30V RSENSE =KRL NP-2A-C1-0R020J, PD = 3W Q2NPN, BVCEO = 40V L1 = COILTRONICS CTX50-5-52, DCR = 0.21Ω, IRON POWDER CORE Q3 SILICONIX NMOS, BVDSS = 60V, RDSON = 5Ω ALL OTHER CAPACITORS ARE CERAMIC AN66 F09

Figure 9. LTC1149-5 (12V-36V to 5V/5A) Using N-Channel MOSFETs

When Q3 turns off, the N-channel MOSFET is turned on by 100 the SCR-connected NPN/PNP network (Q1 and Q2). Re- sistor R2 supplies Q2 with enough base drive to trigger the 90 SCR. Q2 then forces Q1 to turn on, supplying more base 12V drive to Q2. This regenerative process continues until both 80 transistors are fully saturated. During this period, the 24V 70 source of Q4 is pulled to the input voltage. While Q4 is on, EFFICIENCY (%) its gate source voltage is approximately 10V, fully enhanc- 36V 60 ing the N-channel MOSFET.

Efficiency performance for this circuit is quite impressive. 50 0.1 15 Figure 10 shows that for a 12V input the efficiency never OUTPUT CURRENT (A) drops below 90% over the 0.6A to 5A range. At higher AN66 F10 input voltages efficiency is reduced due to transition Figure 10. LTC1149-5 (12V-36V to 5V/5A) High Current Buck losses in the power MOSFETs. For low output currents efficiency rolls off because of quiescent current losses.

AN66-9 Application Note 66

Regulators—Switching (Buck) ciency buck topology switching regulator. The switch is internally grounded, calling for the floating supply ar- Medium Power (1A to 4A) rangement shown (D1 and C1). The circuit converts inputs 1MHz STEP-DOWN CONVERTER of 8V through 30V to a 5V/1A output. ENDS 455kHz IF WOES The chip’s internal oscillator operates at 1MHz for load by Mitchell Lee currents of greater than 50mA with a guaranteed tolerance There can be no doubt that switching power supplies and of 12% over temperature. Even wideband 455kHz IFs are radio IFs don’t mix. One-chip converters typically operate unaffected, as the converter’s operating frequency is well in the range of 20kHz to 100kHz, placing troublesome over one octave distant. harmonics right in the middle of the 455kHz band. This Figure 12 shows the efficiency of Figure 11’s circuit. You contributes to adverse effects such as “desensing” and can expect 80% to 90% efficiency over an 8V to 16V input outright blocking of the intended signals. A new class of range with loads of 200mA or more. This makes the circuit switching converter makes it possible to mix high effi- suitable for 12V battery inputs (that’s how I’m using it), but ciency power supply techniques and 455kHz radio IFs no special considerations are necessary with adapter without fear of interference. inputs of up to 30V. The circuit shown in Figure 11 uses an LT1377 boost converter operating at 1MHz to implement a high effi-

8V TO 30V INPUT D1 + 1N5818 100µF V = 5V 100 O 58 V+ V SW 90 VIN = 8V 4 SHDN 3.57k 1N4148 10Ω LT1377 2 3 PFB 80 NC NFB VIN = 12V SG V PG 1.24k 100nF + C 70 V = 16V C1 EFFICIENCY (%) IN 2.2µF 617 60 2k 4.7nF 50 47nF 0200 400600 800 1000 CTX20-2P* 5V IOUT (mA) 1A AN66 F12 + 150µF 6.3V Figure 12. Efficiency Graph of the MBRS130 OSCON** Circuit Shown in Figure 3

AN66 F11 *CTX20-2P, COILTRONICS 20µH **OS-CON, SANYO VIDEO COMPONENTS

Figure 11. Schematic Diagram: 1MHz LT1377-Based Boost Converter

AN66-10 Application Note 66

HIGH OUTPUT VOLTAGE BUCK REGULATOR common mode problems. The circuit in Figure 13 can be by Dimitry Goder used in applications that do not lend themselves to this approach. High efficiency step-down conversion is easy to implement using the LTC1149 as a buck switching regulator Figure 13 shows a special level shifting circuit (Q1 and U2) controller. The LTC1149 features constant off-time, cur- added to a typical LTC1149 application. The LT1211, a rent mode architecture and fully synchronous rectifica- high speed, precision amplifier, forces the voltage across tion. Current mode operation was selected for its R5 to equal the voltage across current sense resistor R8. well-known advantages of clean start-up, accurate current Q1’s drain current flows to the source, creating a voltage limit and excellent transient response. across R6 proportional to the inductor current, which is now referenced to ground. This voltage can be directly Inductor current sensing is usually implemented by plac- applied to the current sense inputs of U1, the LTC1149. ing a resistor in series with the coil, but the common mode C12 and C4 are added to improve high frequency noise voltage at the LTC1149’s Sense pins is limited to 13V. If a immunity. Maximum input voltage is now limited by the higher output voltage is required, the current sense resis- LT1211; it can be increased if a Zener diode is placed in tor can be placed in the circuit’s ground return to avoid parallel with C12.

VIN 26V TO 35V + C13 C9 0.068µF R9 100Ω C12 0.1µF 1 16 Q2 P-GATE CAP L1 R8 RFD15P05 150µH Ω 15 0.05 2 U1 24V C8 VIN SHDN LTC1149 R5 2A 0.047µF 3 14 D1 V RGND 100Ω + CC MBRS140 4 13 Q3 1% C1 P-DRIVE N-GATE RFD14N05 C7 D3 8 5 12 3 1µF VCC PGND 1N4148 + 6 11 Q1 1 U2A CT SGND C5 LT1211 7 10 VN2222LL 220pF R9 2 ITH VFB Ω C10 – C6 R13 100 µ 8 9 C11 R12 R6 0.1 F 4 3300pF – + 12k SENSE SENSE 100pF 220k 100Ω 1% R4 1% 1% 510Ω C2 1000pF R10 100Ω

AN66 F13

Figure 13. High Output Voltage Buck Regulator Schematic Using LTC1149

AN66-11 Application Note 66

THE LTC1267 DUAL SWITCHING REGULATOR Adjustable Output 3.6V and 5V Converter CONTROLLER OPERATES FROM The adjustable output LTC1267-ADJ shown in Figure 16 is HIGH INPUT VOLTAGES configured as a 3.6V/2.5A and 5V/2A converter. The resis- by Randy G. Flatness tor divider composed of R1 and R2 sets the output voltage Fixed Output 3.3V and 5V Converter according to the formula VOUT = 1.25V (1 + R2/R1). The input voltage range for this application is 5.5V to 28V. A fixed LTC1267 application circuit creating 3.3V/2A and 5V/2A is shown in Figure 15. The operating efficiency 100 LTC1267 shown in Figure 14 exceeds 90% for both the 3.3V and 5V VIN = 12V 5V SECTION sections. The 3.3V section of the circuit in Figure 15 90 comprises the main switch Q1, synchronous switch Q2, inductor L1 and current shunt RSENSE3. 80 The 5V section is similar and comprises Q3, Q4, L2 and EFFICIENCY (%) RSENSE5. Each current sense resistor (RSENSE) monitors LTC1267 70 VIN = 12V the inductor current and is used to set the output current 3.3V SECTION according to the formula IOUT = 100mV/RSENSE. Advan- 60 tages of current control include excellent line and load 0.001 0.01 0.1 1A 2A transient rejection, inherent short-circuit protection and OUTPUT CURRENT controlled start-up currents. Peak inductor currents for L1 AN66 F14 and L2 are limited to 150mV/RSENSE or 3.0A. The EXT VCC Figure 14. LTC1267 Efficiency vs Output Current pin is connected to the 5V output increasing efficiency at of Figure 15 Circuit high input voltages. The maximum input voltage is limited by the MOSFETs and should not exceed 28V.

5.5V < VIN < 28V

+ + CIN5 C 3.3µF 100µF + IN3 + 0.15µF 0.15µF 100µF 33µF 50V 1N4148 50V Q1 1N4148 Q3 P-CH 83 12 27 26 28 21 P-CH Si9435DY Si9435DY VCC3 CAP3VCC VIN MASTER CAP5 EXT VCC VCC5 L1 4 25 L2 RSENSE5 µ PGATE3 SHDN PGATE5 µ VOUT3 20 H 33 H 0.05Ω VOUT5 3.3V 5 5V µ PDRIVE3 24 µ 2A RSENSE3 0.1 F PDRIVE5 0.1 F 2A 0.05Ω 14 18 SENSE+3 SENSE+5 D1 1000pF LTC1267 1000pF MBRS140T3 13 17 SENSE–3 SENSE–5 12 C SHDN3 19 OUT5+ SHDN5 220µF 6 23 NGATE3 NGATE5 D2 10V MBRS140T3 × 2 + COUT3 PGND3 SGND3 CT3 ITH3 ITH5 CT5 SGND5 PGND5 220µF Q2 Q4 7 11 9 10 15 16 20 22 10V N-CH N-CH × 2 Si9410DY RC3 RC5 Si9410DY 1k 1k

0V = RUN 0V = RUN C C C C >2V = SHUTDOWN T3 C3 C5 T5 >2V = SHUTDOWN 270pF 3300pF 3300pF 270pF

RSENSE,:KRL SL-C1-1/2-R050J KRL (603) 668-3210 L1:COILTRONICS CTX20-4 COILTRONICS (407) 241-7876 L2:COILTRONICS CTX33-4 AN66 F15

Figure 15. LTC1267 Dual Output 3.3V and 5V High Efficiency Regulator

AN66-12 Application Note 66

5.5V < VIN < 28V

+ + CIN2 C 3.3µF 100µF + IN1 + 0.15µF 0.15µF 100µF 33µF 50V 1N4148 50V 1N4148 P-CH 73 12 27 26 28 21 P-CH Si9435DY Si9435DY VCC1 CAP1VCC VIN MASTER CAP2 EXT VCC VCC2 L1 4 25 L2 RSENSE2 µ PGATE1 SHDN PGATE2 µ VOUT1 20 H 33 H 0.05Ω VOUT2 3.6V 5 5V µ PDRIVE1 24 µ 2.5A RSENSE1 0.1 F PDRIVE2 0.1 F 2A 0.04Ω 13 18 SENSE+1 SENSE+2 D1 1000pF LTC1267-ADJ 1000pF MBRS140T3 12 17 SENSE–1 SENSE–2 11 C SHDN1 23 OUT2+ NGATE2 220µF 6 22 NGATE1 PGND2 D2 10V N-CH MBRS140T3 × 2 C V C V + OUT1 R2 FB1 SGND1 CT1 ITH1 ITH2 T2 SGND2 FB2 Si9410DY R2 220µF N-CH 100k 14 10 8 9 15 16 20 19 150k 10V Si9410DY 1% 1% × 2 RC1 RC1 1k 1k R1 R1 100pF 100pF 52.3k 0V = RUN 49.9k C C C C 1% >2V = SHUTDOWN T1 C1 C2 T2 1% 270pF 3300pF 3300pF 270pF

RSENSE1,: KRL SL-C1-1/2-R040J KRL (603) 668-3210 RSENSE2,: KRL SL-C1-1/2-R050J COILTRONICS (407) 241-7876 L1: COILTRONICS CTX20-4

L2: COILTRONICS CTX33-4 AN66 F16

Figure 16. LTC1267 Dual Adjustable High Efficiency Regulator Circuit. Output Voltages Set at 3.6V and 5V

HIGH EFFICIENCY 5V TO 3.3V/1.25A CONVERTER 5V supply is already available. Thus, the problem is IN 0.6 SQUARE INCHES reduced to deriving 3.3V from 5V at high efficiency in a by Randy G. Flatness small amount of board space. The next generation of notebook and desktop computers High efficiency is mandatory in these applications since will incorporate a growing number of 3.3V ICs along with converting 5V to 3.3V at 1.25A using a linear regulator 5V devices. As the number of 3.3V devices increases, the would require dissipating over 2W. This is an unnecessary current requirements increase. Typically, a high current waste of power and board space for heat sinking.

+ + C V IN IN 0.1µF 47µF 4V TO 10V 1 16V V IN 8 P-CH PDRIVE Si9433DY L1 0V = NORMAL 6 RSENSE SHDN 10µH >1.5V = SHUTDOWN 0.068Ω VOUT LTC1147-3.3 5 3.3V + SENSE 1.5A 0.01µF 3 4 + C I SENSE– OUT TH 100µF RC 2 10V 1K CT D1 GND CC CT MBRS130LT3 3300pF 120pF 7 RS: KRL SP-1/2-A1-0R068J L: SUMIDA KRL/BANTRY (603) 668-3210 CDR74 SUMIDA (708) 956-0666 (ALT: CD54) AN66 F17 Figure 17. High Efficiency Controller Converts 5V to 3.3V in Minimum Board Area

AN66-13 Application Note 66

The LTC1147 SO-8 switching regulator controller accom- efficiency; for lower cost an Si9340DY can be used at a plishes the 5V to 3.3V conversion with high efficiencies slight reduction in performance. over a wide load current range. The circuit shown in Figure The circuit in Figure 17 has a no load current of only 17 provides 3.3V at efficiencies greater than 90% from 160µA. In shutdown, with Pin 6 held high (above 2V), the 50mA to 1.25A. Using all surface mount components and quiescent current is reduced to less than 20µA with the a low value of inductance (10µH) for L1, the circuit of MOSFET held off. Although the circuit in Figure 17 is Figure 17 occupies only 0.6 square inches of PC board specified at a 5V input voltage the circuit will function area. The efficiency of the circuit in Figure 17 is plotted in from 4V to 10V. Figure 18.

At an output current of 1.25A the efficiency is 90.4%; this 95 means only 0.4W are lost. This lost power is distributed 90 among RSENSE, L1 and the power MOSFETs; thus heat 85 LTC1147-3.3 sinking is not required. SUMIDA CDR74 V = 5V The LTC1147 series of controllers use constant off-time 80 IN LTC1147-3.3 current mode architecture to provide clean start-up, accu- 75 SUMIDA CD54 VIN = 5V rate current limit and excellent line and load regulation. To EFFICIENCY (%) 70 maximize the operating efficiency at low output currents, Burst Mode operation is used to reduce switching losses. 65

60 The P-channel MOSFET in the circuit of Figure 17 will be 1mA 10mA100mA 1A 2A on 2/3 of the time with an input voltage of 5V. Hence, this OUTPUT CURRENT (A) device should be carefully selected to obtain the best AN66 F18 performance. This design uses an Si9433DY for optimum Figure 18. 5V to 3.3V Conversion Efficiency

LT1074/LT1076 ADJUSTABLE 0V TO 5V output voltages power losses in these regulators can be a POWER SUPPLY problem. For example, if an output current of 1.5A is by Kevin Vasconcelos required at 1.25V from an input of 8V, the regulator dissipates more than 10W. Figure 19 shows a DC/DC Linear regulator ICs are commonly used in variable power converter that functionally replaces a linear regulator in supplies. Common types such as the 317 can be adjusted this application. The converter not only eliminates power as low as 1.25V in single-supply applications. At low

VIN = 10V TO 20V 5 C4 L1 V 0.1µF R6 VIN OUT CTX100-5A-52 2.2k 4 V 5%

SW R2 R4 7 – 2 LT1076 3.65k 3.01k + C1 1% µ 1 1% 6 U1

330 F FB

LT1006 R5 35V + 3 C3 5k LT1029 GND VC + 470µF 25T 3 2 4 50V R5 R1 R3 D1 220 2.7k 10.65k MBR340P 1/4W 1% C2 5% 0.01µF

AN66 F19 L1 = COILTRONICS (407) 241-7876 Figure 19. Adjustable LT1074/LT1076 0V to 5V Power Supply

AN66-14 Application Note 66 loss as a concern, but can be adjusted for output voltages substituted for the LT1076. This change accommodates as low as 25mV while still delivering an output current of outputs up to 5A but at the expense of a heftier diode and 1.5A. coil (D1, L1). An MBR735 and Coiltronics CTX50-2-52 are recommended for 5A service. The circuit of Figure 19 employs a basic positive buck topology with one exception: a control voltage is applied through R4 to the feedback summing node at Pin 1 of the 10 LT1076 switching regulator IC, allowing the output to be LT317 adjusted from 0V to approximately 6V. This encompasses 8 the 3.3V and 5V logic supply ranges as well as battery pack 6 combinations of one to four D cells. 4

As R4 is driven from 0V to 5V by the buffer (U1) more or POWER LOSS (W) less current is required from R2 to satisfy the loop’s desire 2 LT1076 to hold the feedback summing point at 2.21V. This forces the converter’s output to swing over the range of 0V to 6V. 0 0 1 2 3 4 5 Figure 20 shows a comparison of power losses for a linear OUTPUT VOLTAGE (V) regulator and the circuit of Figure 19. The load current is AN66 F20 1.5A in both cases although the LT1076 is capable of Figure 20. Power Loss Comparison: Linear Regulator 1.75A guaranteed output current in this application and 2A vs Figure 19’s Power Supply typical. If more current is required the LT1074 can be

TRIPLE OUTPUT 3.3V, 5V AND 12V When the output current for either regulator section drops HIGH EFFICIENCY NOTEBOOK POWER SUPPLY below approximately 15mV/RSENSE, that section auto- by Randy G. Flatness matically enters Burst Modeoperation to reduce switching losses. In this mode the LTC1142 holds both MOSFETs off LTC1142 Circuit Operation and “sleeps” at 160µA supply current while the output capacitor supports the load. When the output capacitor The application circuit in Figure 22 is configured to provide falls 50mV below its specified voltage (3.3V or 5V) the output voltages of 3.3V, 5V and 12V. The current capability LTC1142 briefly turns this section back on, or “bursts,” to of both the 3.3V and 5V outputs is 2A (2.5A peak). The recharge the output capacitor. The timing capacitor pins, logic-controlled 12V output can provide 150mA (200mA peak), which is ideal for flash memory applications. The 100 operating efficiency shown in Figure 21 exceeds 90% for 95 both the 3.3V and 5V sections. 90 LTC1142 The 3.3V section of the circuit in Figure 22 comprises the 85 VIN = 8V main switch Q4, synchronous switch Q5, inductor L1 and 5V SECTION 80 current shunt RSENSE3. The current sense resistor RSENSE 75 LTC1142 EFFICIENCY (%) monitors the inductor current and is used to set the output VIN = 8V 70 3.3V SECTION current according to the formula IOUT = 100mV/RSENSE. Advantages of current control include excellent line and 65 load transient rejection, inherent short-circuit protection 60 0.001 0.010.1 1 2.5 and controlled start-up currents. Peak inductor currents OUTPUT CURRENT (A) for L1 and T1 of the circuit in Figure 22 are limited to AN66 F21 150mV/RSENSE or 3.0A and 3.75A respectively. Figure 21. LTC1142 Efficiency

AN66-15 Application Note 66

+ 22µF VIN + 22µF + + 6.5V TO 14V 1µF 0V = NORMAL 1µF 25V 25V >1.5V = SHUTDOWN × 2 VOUT5 VOUT3 × 2 5V 3.3V Q4 242 16 10 Q2 2A Si9430DY Si9430DY T1 2A VIN3 SHDN3 SHDN5 VIN5 L1 23 9 RSENSE 3 P-DRIVE 3 P-DRIVE 5 RSENSE 5 0.05Ω 33µH 30µH 0.04Ω 1 15 SENSE+ 3 SENSE+ 5 100Ω µ LTC1142 R1 0.01 F Ω 1000pF 100 SENSE– 3 SENSE– 5 28 14 D1 D2 N-DRIVE 3 N-DRIVE 5 MBRS140 20 MBRS140 6 R5 Q5 PGND3 SGND3 CT3 ITH3 ITH5 CT5 SGND5 PGND5 Q3 18k + 100µF Si9410DY Si9410DY 4 3 25 27 13 11 17 18 10V µ × 2 220 F+ 510Ω 510Ω 10V Q1 × 2 VN2222LL CT3 3300pF 3300pF CT5 390pF 200pF

12V ENABLE 0V = 12V OFF >3V = 12V ON (6V MAX) 12V 150mA 1 + C9

µ R3 VOUT Ω L1: COILTRONICS CTX33-4 22 F 20pF 22 22µF + 25V 649k 5 D3 T1: DALE LPE-6562-A026 SHDN MBRS140 35V 1% 2 PRIMARY: SECONDARY = 1:1.8 ADJ 1000pF RSENSE 3: KRL SL-1R050J LT1121 8 V RSENSE 5: KRL SL-1R040J R4 IN 294k GND COILTRONICS (407) 241-7876 1% 3 DALE (605) 665-9301

KRL/BANTRY (603) 668-3210 AN66 F22

Figure 22. LTC1142 High Efficiency Power Supply Schematic Diagram which go to 0V during the sleep interval, can be monitored inductor. The output from this additional winding is recti- with an oscilloscope to observe burst action. As the load fied by diode D3 and applied to the input of an LT1121 current is decreased the circuit will burst less and less regulator. The output voltage is set by resistors R3 and R4. frequently. A turns ratio of 1:1.8 is used for T1 to ensure that the input voltage to the LT1121 is high enough to keep the regulator The timing capacitors CT3 and CT5 set the off-time ac- 4 out of dropout mode while maximizing efficiency. cording to the formula tOFF = 1.3 (10 )(CT). The constant off-time architecture maintains a constant ripple current The LTC1142 synchronous switch removes the normal while the operating frequency varies only with input limitation that power must be drawn from the primary 5V voltage. The 3.3V section has an off-time of approxi- inductor winding in order to extract power from the mately 5µs, resulting in a operating frequency of 120kHz auxiliary winding. With synchronous switching, the auxil- with an 8V input. The 5V section has an off-time of 2.6µs iary 12V output may be loaded without regard to the 5V and a switching frequency of 140kHz with an 8V input. primary output load, provided that the loop remains in continuous mode operation. Auxiliary 12V Output When the 12V output is activated by a TTL high (6V The operation of the 5V section is identical to the 3.3V maximum) on the 12V enable line, the 5V section of the section with inductor L1 replaced by transformer T1. The LTC1142 is forced into continuous mode. A resistor 12V output is derived from an auxiliary winding on the 5V

AN66-16 Application Note 66 divider composed of R1, R5 and switch Q1 forces an 5V output. The 100% duty cycle inherent in the LTC1142 offset, subtracting from the internal offset at Pin 14. When provides low dropout operation limited only by the load this external offset cancels the built-in 25mV offset, Burst current multiplied by the sum of the resistances of the 5V Mode operation is inhibited. inductor, Q2 RDS(ON) and current sense resistor RSENSE5. Auxiliary 12V Output Options Extending the Maximum Input Voltage The circuit of Figure 22 can be modified for operation in The circuit in Figure 22 is designed for a 14V maximum low-battery count (6-cell) applications. For applications input voltage. The operation of the circuit can be extended where heavy 12V load currents exist in conjunction with to over 18V if a few key components are changed. The low input voltages (<6.5V), the auxiliary winding should parts that determine the maximum input voltage of the be derived from the 3.3V instead of the 5V section. As the circuit are the power MOSFETs, the LTC1142 and the input input voltage falls, the 5V duty cycle increases to the point capacitors. With the LTC1142 replaced by an LTC1142HV, when there is simply not enough time to transfer energy an 18V typical (20V maximum) input voltage is allowable. from the 5V primary winding to the 12V secondary wind- Since the gate drive voltages supplied by the LTC1142 and ing. For operation from the 3.3V section, a transformer LTC1142HV are from ground to VIN, the input voltage with a turns ratio of 1:3.25 should be used in place of the must not exceed the maximum VGS of the MOSFETs. The 33µH inductor L1. Likewise, a 30µH inductor would re- MOSFETs specified in Figure 22 have an absolute maxi- place T1 in the 5V section. With these component changes, mum of 20V, matching that of the LTC1142HV.1 Finally, the duty cycle of the 3.3V section is more than adequate for the input capacitor’s voltage rating will also have to be full 12V load currents. The minimum input voltage in this increased above 12V. case will be determined only by the dropout voltage of the 1For improved efficiency, CT5 should be charged to 270pF.

THE NEW SO-8 LTC1147 SWITCHING REGULATOR 500mW. The efficiency plotted as a function of output CONTROLLER OFFERS HIGH EFFICIENCY current is shown in Figure 24. IN A SMALL FOOTPRINT by Randy Flatness V + CIN + IN µ (4V TO 0.1µF 15 F 12V) 25V 1 × 2 Introduction V 0V = NORMAL 6 IN 8 P-CH SHDN PDRIVE The LTC1147 switching regulator controller is a high >1.5V = SHUTDOWN Si943ODY RSENSE 0.1Ω VOUT efficiency step-down DC/DC converter. It uses the same 3.3V LTC1147-3.3 L 1A 100µH current mode architecture and Burst Mode operation as 3 5 SENSE + the LTC1148/LTC1149 but without the synchronous ITH 1000pF switch. Ideal for applications requiring up to 1A, the RC 2 4 – 1k CT SENSE LTC1147 shows 90% efficiencies over two decades of COUT+ GND CC D1 220µF output current. 3300pF C 7 MBRD330 6.3V T 560pF High Efficiency 5V to 3.3V in a Small Area AN66 F23 RS = KRL SP-1/2-A1-0R100 L = COILTRONICS CTX100-4 The LTC1147 5V to 3.3V converter shown in Figure 23 COILTRONICS (407) 241-7876 has 85% efficiency at 1A output with efficiencies greater KRL/BANTRY (603) 668-3210 than 90% for load currents up to 500mA. Using the Figure 23. This LTC1147 5V to 3.3V Converter Achieves LTC1147 reduces the power dissipation to less than 92% Efficiency at 300mA Load Current

AN66-17 Application Note 66

100 is the ideal application for the LTC1147. As the output LTC1147-3.3 current increases the diode loss increases. At high input- VIN = 5V 90 to-output voltage ratios, the Schottky diode conducts most of the time. In this situation, any loss in the diode will have a more significant effect on efficiency and an LTC1148 80 might therefore be chosen.

EFFICIENCY (%) Figure 26 compares the efficiencies of LTC1147-5 and 70 LTC1148-5 circuits with the same inductor, timing capacitor and P-channel MOSFET. At low input voltages and 1A 60 0.001 0.01 0.1 1 output current the efficiency of the LTC1147 differs from LOAD CURRENT (A) that of the LTC1148 by less than two percent. At lower

AN66 F24 Figure 24. The LTC1147 5V to 3.3V Converter Provides Better 100

Than 90% Efficiency from 20mA to 500mA of Output Current 2 GATE CHARGE I R

) 95 Giving Up the Synchronous Switch? % LTC1147 IQ

SCHOTTKY DIODE The decision whether to use a nonsynchronous LTC1147 90 design or a fully synchronous LTC1148 design requires a careful analysis of where losses occur. The LTC1147 switching regulator controller uses the same loss reduc- EFFICIENCY/LOSS ( 85 ing techniques as the other members of the LTC1148/

LTC1149 family. The nonsynchronous design saves the 80 0.01 0.03 0.1 0.3 1 3 N-channel MOSFET gate drive current at the expense of OUTPUT CURRENT (A) increased loss due to the Schottky diode. AN66 F25 Figure 25 shows how the losses in a typical LTC1147 Figure 25. Low Current Efficiency is Enhanced by Burst Mode Operation. Schottky Diode Loss Dominates at High Output application are apportioned. The gate-charge loss Currents (P-channel MOSFET) is responsible for the majority of the efficiency lost in the midcurrent region. If Burst Mode 100 LTC1147-5 operation was not employed, the gate charge loss alone LTC1148-5 would cause the efficiency to drop to unacceptable levels 90 at low output currents. With Burst Mode operation, the DC ILOAD = 1A supply current represents the only loss component that increases almost linearly as output current is reduced. As 80 2

expected, the I R loss and Schottky diode loss dominate EFFICIENCY (%) at high load currents. 70 ILOAD = 100mA In addition to board space, output current and input voltage are the two primary variables to consider when 60 4 61214810 deciding whether to use the LTC1147. At low input-to- INPUT VOLTAGE (V) output voltage ratios, the top P-channel switch is on most AN66 F26 of the time, leaving the Schottky diode conducting only a Figure 26. At High Input Voltages Combined with Low Output small percentage of the total period. Hence, the power lost Currents, the Efficiency of the LTC1147 Exceeds That of the in the Schottky diode is small at low output currents. This LTC1148

AN66-18 Application Note 66 output currents and high input voltages the LTC1147’s P-channel switch (off-time is constant) thereby keeping efficiency can actually exceed that of the LTC1148. the inductor ripple current constant. Eventually the on- time extends so far that the P-channel MOSFET is on at DC Low Dropout 5V Output Applications or at a 100% duty cycle. Because the LTC1147 is so well-suited for low input-to- With the switch turned on at a 100% duty cycle, the output voltage ratio applications it is an ideal choice for dropout is limited by the load current multiplied by the low dropout designs. All members of the LTC1148/LTC1149 sum of the resistances of the MOSFET, the current shunt family (including the LTC1147) have outstandingly low and the inductor. For example, the low dropout 5V regu- dropout performance. As the input voltage on the LTC1147 lator shown in Figure 27 has a total resistance of less than drops, the feedback loop extends the on-time for the 0.2Ω. This gives it a dropout voltage of 200mV at 1A output current. At input voltages below dropout the output C + VIN + IN voltage follows the input. This is the circuit whose effi- 15µF (5.5V 0.1µF 25V ciency is plotted in Figure 28. TO 12V) 1 × 3 V 0V = NORMAL 6 IN 8 P-CH SHDN PDRIVE 100 >1.5V = SHUTDOWN Si943ODY RSENSE Ω 0.05 VOUT LTC1147-5 5V 95 LTC1147-5 L 2A VIN = 6V 50µH 3 5 + 90 ITH SENSE VIN = 10V 1000pF RC 2 4 – 85 1k CT SENSE C CC OUT+ GND µ EFFICIENCY (%) 3300pF D1 220 F 80 CT 7 MBRD330 10V 470pF × 2 75

AN66 F27 R = KRL SL-1-C1-0R050J S 70 L = COILTRONICS CTX50-4 1 10 100 1000 COILTRONICS (407) 241-7876 KRL/BANTRY (603) 668-3210 LOAD CURRENT (mA) AN66 F28 Figure 27. The LTC1147 Architecture Provides Inherent Low Dropout Operation. This LTC1147-5 Circuit Supports a 1A Load Figure 28. Greater Than 90% Efficiency is Obtained for Load with the Input Voltage Only 200mV Above the Output Currents of 20mA to 2A (VIN = 10V)

AN66-19 Application Note 66

THE LT1432: 5V REGULATOR critical. Ordinary 5V switchers draw quiescent currents of ACHIEVES 90% EFFICIENCY 5mA to 15mA for these light loads. The efficiency of a 12V by Carl Nelson to 5V converter with 10mA supply current and 1mA load is only 4%. Clearly, some method must be provided to Power supply efficiency has become a highly visible issue eliminate the quiescent current of the switching regulator in many portable battery-powered applications. Higher control section. efficiency translates directly to longer useful operating time—a potent selling point for products such as note- An additional requirement for some systems is full shut- book computers, cellular phones, data acquisition units, down of the regulator. It would be ideal if a simple logic sales terminals and word processors. The “holy grail” of signal could cause the converter to turn off and draw only efficiency for 5V outputs is 90%. a few microamperes of current. For a number of reasons, older designs were limited to The combination of battery form factors, their discrete efficiencies of 80 to 85%. High quiescent current in the voltage steps and the use of higher voltage wall adapters control circuitry limited efficiency at lower output cur- requires a switching regulator that operates with inputs rents. Losses in the power switch, inductor and catch from 6V to 30V. Both of these voltages present problems diode all added up to limit efficiency at moderate-to-high for a MOS design because of minimum and maximum gate output currents. Each of these areas must be addressed in voltage requirements of power MOS switches. a design that is to have high efficiency over a wide output The LT1432 was designed to address all the requirements current range. described above. It is a bipolar control chip that interfaces Some portable equipment has the additional requirement directly to the LT1070 family of switching regulators and of high efficiency at extremely light loads (1mA to 5mA). is capable of operating with 6V to 30V inputs. These ICs These applications have a sleep mode in which RAM is have a very efficient, quasisaturating NPN switch that kept alive to retain information. The instrument may spend mimics the resistive nature of MOS transistors with much days or even weeks in this mode, so battery drain is smaller die areas. The NPN is a high frequency device with

VIN

VSW VIN + C1 330µF LT1271 35V FB VC GND D2 1N4148 C6 µ 0.02 F R1 V C5 OUT 680Ω C3+ 5V 0.03µF µ 4.7 F L1 3A C4 TANT R2* µ 50µH 0.1 F 0.013Ω + C2 D1 390µF MBR330P 16V + VC DIODE V VIN VLIM < 0.3V = NORMAL MODE > 2.5V = SHUTDOWN LT1432 OPEN = BURST MODE MODE INPUT MODE V OUT AN66 F29 200pF GND

*R2 IS MADE FROM PC BOARD COPPER TRACES L1 = COILTRONICS CTX 50-3-MP (3A) (407) 241-7876

Figure 29. High Efficiency 5V Buck Converter

AN66-20 Application Note 66 an equivalent voltage and current overlap time of only The LT1271 normally draws about 50µA to100µA in its 10ns. Drive to the switch is automatically scaled with shutdown state. A shutdown command to the LT1432 switch current, so drive losses are also low. Switch and opens all connections to the LT1271 VIN pin so its current driver losses using an LT1271 with a 12V input and a 5V, drain is eliminated. This leaves only the shutdown current 500mA load are only about 2%. of the LT1432 and the switch leakage of the LT1271, which µ To reduce quiescent current losses, the LT1271 is pow- typically add up to less than 20 A—less than the self- ered from the 5V output rather than from the input voltage. discharge rate of NiCd batteries. For many applications the This is done by pumping the supply capacitor C3 from the on/off function is under keystroke control. Digital chips output via D2. Quick minded designers will observe that which draw only a few microamps are available for key- this arrangement does not self-start; accordingly, a paral- stroke recognition and power control. lel path was included inside the LT1432 to provide power There is no way to design around inductor losses. These to the IC switcher directly from the input during start-up. losses are minimized by using low loss cores such as Equivalent quiescent supply current is reduced to about molypermalloy or ferrite, and by sizing the core to use wire 3.5mA with this technique. with sufficient diameter to keep resistive losses low. The µ Catch diode losses cannot be reduced with IC “tricks” 50 H inductor shown has a core loss of 200mW with type- unless the diode is replaced with a synchronously driven 52 powdered iron material and 28mW with molypermalloy. MOS switch. This is more expensive and still requires the For a 1A load this represents efficiency losses of 4% and diode to avoid voltage spikes during switch nonoverlap 0.56% respectively—a major difference. Ferrite cores times. The question is, is it worth it? would have even lower losses than molypermalloy, but the “moly” has such low losses that ferrites should be chosen The following formula was developed to calculate the for other reasons, such as height, cost, mounting and the improvement in efficiency when adding a synchronous like. DC resistance of the inductor shown is 0.02Ω. This switch. represents an efficiency loss of 0.4% at 1A load and 0.8% 2 at 2A. Significant reduction in these resistance losses (VIN – VOUT)(Vf – RFET • IOUT)(E) Efficiency change = would require a somewhat larger inductor. The choice is (VIN)(VOUT) yours. With VIN = 10V, VOUT = 5V, Vf (diode forward voltage) = The LT1432 has a high efficiency current limit with a sense Ω 0.45V, RFET = 0.1 and IOUT = 1A the improvement in voltage of only 60mV. This has a side benefit in that printed efficiency is only 2.8%. This does not take into account circuit board trace material can be used for the sense the losses associated with MOS gate drive, so real resistor. A 3A limit requires a 0.02Ω sense resistor and improvement would probably be closer to 2%. The this is easily made from a small section of serpentine trace. availability of low forward voltage Schottky diodes such The 60mV sense voltage has a positive temperature coef- as the MBR330P makes synchronous switches less ficient that tracks that of copper so that the current limit is attractive than they used to be. flat with temperature. Foldback current limiting can be To achieve higher efficiency during sleep, the LT1432 has easily implemented. Burst Mode operation. In this mode the LT1271 is either The LT1432 represents a significant improvement in high driven full on, or completely shut down to its micropower efficiency 5V supplies that must operate over a wide range state. The LT1432 acts as a comparator with hysteresis of load currents and input voltages. Its efficiency has a instead of a linear amplifier. This mode reduces equivalent very broad peak that exceeds 90%, requiring a new input supply current to 1.3mA with a 12V battery. Battery definition of the “holy grail.” Logic controlled shutdown, life with NiCd AA cells is over 300 hours with a 1mA 5V millipower Burst Mode operation and efficient, accurate, load. Burst Mode operation increases output ripple, espe- current limiting make this regulator extremely attractive cially with higher output currents, so maximum load in this for battery-powered applications. mode is 100mA.

AN66-21 Application Note 66

Regulators—Switching (Buck) a frequency of 100kHz. Figure 33 is the efficiency plot of the circuit. At a load current of 100mA the efficiency is at Low Power (<1A) 92%; the efficiency falls to 82% at a 1A output. APPLICATIONS FOR THE LTC1265 HIGH EFFICIENCY MONOLITHIC BUCK CONVERTER 2.5mm Typical-Height 5V to 3.3V Regulator by San-Hwa Chee Figure 34 shows the schematic for a very thin 5V to 3.3V converter. For the LTC1265 to be able to source 500mA Efficiency output current and yet meet the height requirement, a Figure 30 shows a typical LTC1265-5 application circuit. small value inductor must be used. The circuit operates at The efficiency curves for two different input voltages are a high frequency (500kHz typically) increasing the gate shown in Figure 31. Note that the efficiency for a 6V input charge losses. Figure 35 is the efficiency curve for this exceeds 90% over a load range from less than 10mA to application. 850mA. This makes the LTC1265 attractive for all battery operated products and efficiency sensitive applications. Positive-to-Negative Converter Besides converting from a positive input to positive out- 5V to 3.3V Converter put, the LTC1265 can be configured to perform a positive- Figure 32 shows the LTC1265 configured for 3.3V output to-negative conversion. Figure 36 shows the schematic with 1A output current capability. This circuit operates at for this application.

VIN 5.4V TO 12V VOUT 5V 1 13 1A * 100 L1 R ** PWR VIN PWR VIN µ SENSE VOUT = 5V 33 H 0.1Ω VIN = 6V †† 2 14 + CIN VIN SW 95 68µF 0.1µF D1 C † LTC1265-5 + OUT 20V 10 12 MBRS130LT3 220µF VIN = 9V SHDN PGND 10V 90 11 SGND 130pF 85 5 CT EFFICIENCY (%) 80 6 9 ITH NC L = 33µH 1k VOUT = 5V 3900pF 7 8 75 Ω SENSE – SENSE+ RSENSE = 0.1 C = 130pF *COILTRONICS CTX33-4 1000pF T **KRL SL-C1-OR100J 70 †AVX TPSE227K010 AN66 F30 0.01 0.10 1.00 ††AVX TPSE686k020 COILTRONICS 407-241-7876 LOAD CURRENT (A) KRL/BANTRY 603-668-3210 AN66 F31

Figure 30. High Efficiency Step-Down Converter Figure 31. Efficiency vs Load Current

AN66-22 Application Note 66

VIN 5V

VOUT 100 113 + CIN† 3.3V 0.1µF 100µF L1* 1A PWR VIN PWR VIN 10V µ Ω 95 2 14 47 H 0.1 ** VIN SW D1 C †† LTC1265-3.3 + OUT 90 4 12 MBRS130LT1 220µF LBIN PGND 10V 3 11 85 LB SGND 270pF OUT

5 10 EFFICIENCY (%) 80 CT SHDN SHUTDOWN L1 = 47µH 6 9 NC VOUT = 3.3V ITHR 75 Ω 1k RSENSE = 0.1 3900pF 7 8 C = 270pF SENSE – SENSE+ T 70 1000pF 1 10 100 1000 AN66 F32 LOAD CURRENT (mA)

*COILCRAFT D03316-473 COILCRAFT 708-639-6400 AN66 F33 **KRL SL-C1-OR100J KRL/BANTRY 603-668-3210 †AVX TAJD100K010 ††AVX TAJD226K010

Figure 32. High Efficiency 5V to 3.3V Converter Figure 33. Efficiency vs Load Current

VIN 5V

VOUT 95 113 + CIN† 3.3V 0.1µF 15µF L1* 500mA PWR V IN PWR VIN 10V × 2 18µH Ω 2 14 0.20 ** 90 VIN SW C †† D1 OUT LTC1265-3.3 + 22µF 4 12 MBRS0520LT1 LB PGND 6.3V 85 IN × 2 3 11 LBOUT SGND 51pF 80 5 10 EFFICIENCY (%) CT SHDN SHUTDOWN L1 = 18µH 1k 6 9 75 V = 3.3V ITHR NC OUT RSENSE = 0.20Ω 3300pF 7 8 C = 50pF SENSE – SENSE+ T 70 1000pF 1 10 100 500 AN66 F34 LOAD CURRENT (mA) *SUMIDA CLS62-180 SUMIDA 708-956-0666 AN66 F35 **KRL SL-C1-OR200J KRL/BANTRY 603-668-3210 †AVX TAJB155K010 ††AVX TAJB225K06

Figure 34. 2.5mm High 5V to 3.3V Converter (500mA Output Current) Figure 35. Efficiency vs Load Current

AN66-23 Application Note 66

VIN 3.5V TO 7.5V SHUTDOWN

CIN† 113 + 22µF 0.1µF COUT†† VIN (V)OUT(MAX) I (mA) RSENSE** µ 25V PWR VIN PWR VIN 100 F/10V 3.5 360 TP0610L 0.1Ω × 2 2 14 VOUT 4.0 430 VIN SW D1 L1* + –5V 5.0 540 LTC1265-5 µ 6.0 630 4 12 MBRS130LT3 47 H 7.0 720 LBIN PGND 7.5 740 3 11 LB SGND 220pF OUT †AVX TPSD226K025 5 10 100k C SHDN ††AVX TPSD106K010 2200pF T 1k *L1 SELECTION 6 9 MANUFACTURER PART NO. I NC THR COILTRONICS CTX50-4 7 – + 8 COILCRAFT D03316-473 SENSE SENSE DALE LPT4545-500LA 1000pF SUMIDA CD75-470 **KRL SL-C1-OR100J

AN66 F36 Figure 36. Positive (3.5 to 7.5V) to Negative (–5V) Converter

Regulators—Switching (Boost) switch currents of up to 10A are available, providing a convenient means for power conversion over wide input Medium Power (1A to 4A) and output voltage ranges. If higher switch currents are HIGH OUTPUT CURRENT BOOST REGULATOR required, a controller with an external power MOSFET is a by Dimitry Goder better choice. Low voltage switching regulators are often implemented Figure 37 shows an LTC1147-based 5V to 12V converter with self-contained power integrated circuits featuring a with 3.5A peak output current capability. The LTC1147 is PWM controller and an onboard power switch. Maximum a micropower controller that uses a constant off-time

VIN 5V L1 15µH

C6 D2 + R5 220µF BAT54 100Ω 10V VOUT × 2 D1 12V/3A + C7 MBR735 3.5A PEAK 3.3µF Q3 TP0610L R6 C5 56k Q2 R8 + 150µF IRL2203 100k 16V × 1 8 Q1 R2 1% 2 VIN PDRIVE VN2222LL 11.5k 2 7 1% CT GND U1 C4 3 LTC1147 6 100pF ITH VFB C1 4 5 3300pF SENSE – SENSE + C2 R1 R4 R7 Ω Ω 180pF 510Ω C3 100 0.01 0.01µF 2%

R3 AN66 F37 C5, C6 SANYO 0S-CON 100Ω EFFICIENCY AT 3A ≥ 90%

Figure 37. LTC1147-Based 5V to 12V Converter

AN66-24 Application Note 66 architecture, eliminating the need for external slope com- transferred to the output capacitor C5. Timing capacitor pensation. Current mode control allows fast transient C2 sets the operating frequency. The controller is pow- response and cycle-by-cycle current limiting. A maximum ered from the output through R5 providing 10V of gate voltage of only 150mV across the current-sense resistor drive for Q2. This reduces the MOSFET’s ON resistance R7 optimizes performance for low input voltages. and allows efficiency to exceed 90% even at full load. The feedback network comprising R2 and R8 sets the output When Q2 turns on, current starts building up in inductor voltage. Current sense resistor R7 sets the maximum L1. This provides a ramping voltage across R7. When output current; it can be changed to meet different circuit this voltage reaches a threshold value set internally in the requirements. LTC1147, Q2 turns off and the energy stored in L1 is

Regulators—Switching (Boost) space. Figure 39 shows the circuit’s efficiency, which can reach 89% on a 5V input. Low Power (<1A) The reference voltage on the FB pin is trimmed to 1.25V APPLICATIONS FOR THE LT1372 500kHz and the output voltage is set by the R1/R2 resistor divider SWITCHING REGULATOR ratio (VOUT = VREF • (R1/R2 + 1). R3 and C2 frequency by Bob Essaff compensate the circuit. Boost Converter Positive-to-Negative Flyback with Direct Feedback The boost converter in Figure 38 shows a typical LT1372 A unique feature of the LT1372 is its ability to directly application. This circuit converts an input voltage, which regulate negative output voltages. As shown in the posi- can vary from 2.7V to 11V, into a regulated 12V output. tive-to-negative flyback converter in Figure 40, only two Using all surface mount components, the entire boost resistors are required to set the output voltage. The converter consumes only 0.5 square inches of board reference voltage on the NFB pin is –2VREF, making VOUT = – 2VREF • (R2/R3 + 1). Efficiency for this circuit D1 5V L1* µ MBRS120T3 reaches 72% on a 5V input. 4.7 H † VOUT 5 12V VIN R1 4 8 100 ON V 53.6k OFF S/S SW 1% VIN = 5V LT1372/LT1377 2 90 + C1** FB + C4** 22µF 22µF GND VC 80 6, 7 1 R2 6.19k C2 1% 0.047µF 70 R3 C3 2k 0.0047µF EFFICIENCY (%) AN66 F38 60

† *COILCRAFT DO1608-472 (4.7µH) OR MAX IOUT µ COILCRAFT DT3316-103 (10 H) OR L1 I 50 SUMIDA CD43-4R7 (4.7µH) OR OUT 0.01 0.1 1 4.7µH0.25A SUMIDA CD73-100KC (10µH) OR OUTPUT CURRENT (A) 10µH 0.35A **AVX TPSD226M025R0200 AN66 F39

Figure 38. 5V to 12V Boost Converter Figure 39. 12V Output Efficiency

AN66-25 Application Note 66

Dual Output Flyback with Overvoltage Protection regulation under varying load conditions. For evenly loaded outputs, as shown in Figure 42, cross regulation can be Multiple-output flyback converters offer an economical quite good, but when the loads differ greatly, as in the case means of producing multiple output voltages, but the of a load disconnect, there may be trouble. Figure 43 power supply designer must be aware of cross regulation shows that when only the 15V output is voltage sensed, issues, which can cause electrical overstress on the sup- the –15V quasi-regulated output exceeds –25V when ply and loads. Figure 41 is a dual-output flyback converter unloaded. This can cause electrical overstress on the with overvoltage protection. Typically, in multiple-output output capacitor, output diode and the load when recon- flyback designs only one output is voltage sensed and nected. Adding output voltage clamps is one way to fix the regulated. The remaining outputs are “quasi-regulated” by problem but the circuit in Figure 41 eliminates this require- the turns ratios of the transformer secondary. Cross ment. This circuit senses both the 15V and –15V outputs regulation is a function of the transformer used and is a and prevents either from going beyond its regulating measure of how well the quasi-regulated outputs maintain value. Figure 44 shows the unloaded –15V output being held constant. The circuit’s efficiency, which can reach VIN 2.7V TO 16V 79% on a 5V input, is shown in Figure 45.

+ 2 T1 4 C1 D2 30 22µF + P6KE-15A C4 25 VIN = 5V 5 47µF D3 20 1N4148 VOUT 4 VIN 13 ON 8 –VOUT 15 S/S VSW OFF R2–5V 10 LT1372 D1 2.49k 2 3 MBRS130LT3 5

NC FB NFB 1% (V) 0 I V R3 VC GND OUT IN OUT 2.49k V –5 1 6, 7 0.3A 3V 0.5A 5V 1% –10 –VOUT C2 0.75A 9V AN66 F40 –15 0.047µF R1 T1 = COILTRONICS CTX10-2 –20 C3 2k COILTRONICS (407) 241-7876 0.0047µF –25 –30 110100 Figure 40. LT1372’s Positive-to-Negative Converter OUTPUT CURRENT (mA) with Direct Feedback AN66 F42 Figure 42. Cross Regulation of Figure 41’s Circuit. VOUT and –VOUT Evenly Loaded R2 R1 1.21k 13k 1% 1%

VIN 30 V = 5V 2.7V TO 13V MBRS140T3 25 IN V 20 + T1 OUT VOUT C1 2, 3 5 + 15V 15 22µF P6KE-20A C4 47µF 10 2 5 1N4148 4 5 6, 7 FB V 8 (V) ON 4 IN 8 + 0 S/S V C5 SW OUT

OFF 47µF V –5 LT1372 1 –V 3 OUT –10 NFB –15V R4 –15 VC GND MBRS140T3 12.1k –V –20 OUT 1 6, 7 1% C2 –25 R5 0.047µF –30 R3 T1 = DALE LPE-4841-100MB 2.49k 110100 C3 1% 0.0047µF 2k DALE (605) 665-9301 OUTPUT CURRENT (mA) AN66 F41 AN66 F43 Figure 41. LT1372 Dual Output Flyback Converter Figure 43. Cross Regulation of Figure 41’s Circuit. with Overvoltage Protection –VOUT Unloaded; Only VOUT Voltage Sensed

AN66-26 Application Note 66

30 85 V = ±15V V = 9V 25 VIN = 5V OUT IN 20 80 V 15 OUT V = 5V 10 IN 5 75 (V) 0 VIN = 3V OUT

V –5 70

–10 EFFICIENCY (%) –VOUT –15 65 –20 –25 –30 60 110100 5 10 100 200 OUTPUT CURRENT (mA) OUTPUT CURRENT (mA)

AN66 F44 AN66 F45 Figure 44. Cross Regulation of Figure 41’s Circuit. Figure 45. Efficiency of Dual Output Flyback Converter –VOUT Unloaded; Both –VOUT and VOUT Sensed in Figure 41

VIN = 9V Regulators—Switching FB LT1176CS-5 (Buck/Boost) CTX100-5P 5V V V IN SW 800mA ±5V CONVERTER USES OFF-THE-SHELF 1 8 2 7 SURFACE MOUNT COIL VC GND + 3 6 By Mitchell Lee and Kevin Vasconcelos + 470µF 100µF 2k 4 5 Single-output switching regulator circuits can often be adapted to multiple output configurations with a minimum 10nF 1N5818 of changes, but these transformations usually call for + custom wound inductors. A new series of standard induc- 1N5818 470µF 1 –5V tors, featuring quadrifilar windings, allows power supply 100mA designers to take advantage of these modified circuits but AN66 F46 without the risks of a custom magnetics development Figure 46. 5V Buck Converter with –5V Overwinding program. 1:1:1:1 sections. In the application of Figure 46, three The circuit shown in Figure 46 fulfills a recent customer sections are paralleled for the main 5V winding and the requirement for a 9V to 12V input, 5V/800mA and remaining section is used for the –5V output. This concen- –5V/100mA output converter. It employs a 1:1 overwind- trates the copper where it is needed most—on the high ing on what is ostensibly a buck converter to provide a current output. –5V output. The optimum solution would be a bifilar Efficiency with the outputs loaded at 500mA and –50mA wound coil with heavy gauge wire for the main 5V output is over 80%. Minimum recommended load on the –5V and smaller wire for the overwinding. To avoid a custom TM output is 1mA to 2mA, and the –5V load current must coil design, an off-the-shelf JUMBO-PAC quadrifilar always be less than the 5V load current. wound coil is used. This family of coils is wound with 1 JUMBO-PAC is a trademark of Coiltronics Inc. (407) 241-7876.

AN66-27 Application Note 66

SWITCHING REGULATOR PROVIDES Figure 48 shows the operating waveforms for the circuit. CONSTANT 5V OUTPUT FROM 3.5V TO 40V In this architecture the capacitor C2 serves as the single INPUT WITHOUT A TRANSFORMER energy transfer device between the input voltage and by Brian Huffman output voltage of the circuit. While the LT1171 power switch is off, diode D1 is forward biased, providing a path A common switching regulator requirement is to produce for the currents from inductors L1 and L2. Trace A shows a constant output voltage from an input voltage that varies inductor L1’s current waveform and trace B is L2’s current above or below the output voltage. This is particularly waveform. Observe that the inductor current waveforms important for extending battery life in battery-powered occur on top of a DC level. The waveforms are virtually applications. Figure 47 shows how an LT1171 switching identical because the inductors have identical inductance regulator IC, two inductors and a “flying” capacitor can values and the same voltages are applied across them. The generate a constant output voltage that is independent of current flowing through inductor L1 is not only delivered input voltage variations. This is accomplished without the to the load but is also used to charge C2. C2 is charged to use of a transformer. Inductors are preferred over trans- a potential equal to the input voltage. formers because they are readily available and more economical. When the LT1171 power switch turns on, the VSW pin is pulled to ground and the input voltage is applied across the The circuit in Figure 47 uses the LT1171 to control the inductor L1. At the same time, capacitor C2 is connected output voltage. A fully self-contained switching regulator across inductor L2. Current flows from the input voltage IC, the LT1171 contains a power switch as well as the source through inductor L1 and into the LT1171. Trace C control circuitry (pulse-width modulator, oscillator, refer- shows the voltage at the V pin and Trace D is the current ence voltage, error amplifier and protection circuitry). The SW flowing through the power switch. The catch diode (D1) is power switch is an NPN transistor in a common-emitter reverse biased and capacitor C2’s current also flows configuration; when the switch turns on, the LT1171’s through the switch, through ground and into inductor L2. V SW pin is connected to ground. This power switch can During this interval C2 transfers its stored energy into handle peak switch currents of up to 2.5A. inductor L2. After the switch turns off the cycle is repeated. C2 150µF D1 Another advantage of this circuit is that it draws its input L1 50V MBR350 50µH VOUT current in a triangular waveshape (see Trace A in Figure 5V 5 + 0.5A 48). The current waveshape of the input capacitor is L2 identical to the current waveshape of inductor L1 except VIN 50µH R2 V 4 that the capacitor’s current has no DC component. This SW 3.01k LT1171 1% type of ripple injects only a modest amount of noise into VIN 2 + C3 (3.5V FB 470µF the input lines because the ripple does not contain any TO 40V) 16V GND VC sharp edges. 3 1 + C1 R3 R1 56µF 1.00k 1k 50V 1% A = 1A/DIV C4 IL1, IC1 1µF B = 1A/DIV AN66 F47 IL2

C1 = NICHICON (AL) UPL1H560MEH, ESR = 0.250Ω, IRMS = 360mA C = 10V/DIV Ω C2 = NICHICON (AL) UPL1H151MPH, ESR = 0.100 , IRMS = 820mA VSW C3 = NICHICON (AL) UPL1C471MPH, ESR = 0.090Ω, IRMS = 770mA L1, L2 = COILTRONICS CTX50-4, DCR = 0.090Ω, D = 1A/DIV I COILTRONICS (407) 241-7876 SW

EQUATION 1: VOUT = 1.25V (1 + R2/R3) AN66 F48 µ Figure 47. LT1171 Provides Constant 5V Output from 5 s/DIV 3.5V to 40V Input. No Transformer Is Required Figure 48. LT1171 Switching Waveforms

AN66-28 Application Note 66

Figure 49 shows the efficiency of this circuit for a 0.5A load equal. (The duty cycle is determined by multiplying the and maximum output current for various input voltages. switch ON time by the switching frequency.) The RC The two main loss elements are the output diode (D1) and network (R1 and C4 in Figure 47) connected to the VC pin the LT1171 power switch. A Schottky diode is chosen for provides sufficient compensation to stabilize this control its low forward voltage drop; it introduces a 10% loss, loop. Equation 1 (see Figure 47) can be used to determine which is relatively constant with input voltage variations. the output voltage. At low input voltages the efficiency drops because the LT1171 power switch’s saturation voltage becomes a 1.2 80 higher percentage of the available input supply. 1.0 75 EFFICIENCY

This circuit can deliver an output current of 0.5A at a 3.5V EFFICIENCY (%) 0.8 I 70 input voltage. This rises to 1A as input voltage is in- OUT(MAX) (A) creased. Above 20V, higher output currents can be achieved 0.6 65 OUT(MAX)

by increasing the values of inductors L1 and L2. Larger I inductances store more energy, providing additional cur- 0.4 60 rent to the load. If 0.5A of output current is insufficient, use 0.2 55 a higher current part, such as the LT1170. 0.0 50 The output voltage is controlled by the LT1171 internal 0 5101520 253035 40 INPUT VOLTAGE (V) error amplifier. This error amplifier compares a fraction AN66 F49 of the output voltage, via the R1 to R2 divider network shown in Figure 47, with an internal 1.25V reference Figure 49. Efficiency and Load Characteristics voltage, and varies the duty cycle until the two values are for Various Input Voltages

SWITCHING REGULATOR PROVIDES pin swings between the input voltage (VIN) and the nega- ±15V OUTPUT FROM AN 8V TO 40V INPUT tive output voltage (–VOUT). (The ability of the LT1074’s WITHOUT A TRANSFORMER VSW pin to swing below ground is unusual—most other by Brian Huffman 5-pin buck switching regulator ICs cannot do this.) Trace ± A shows the waveform of the VSW pin voltage and Trace B Many systems derive 15V supplies for analog circuitry is the current flowing through the power switch. from an input voltage that may be above or below the 15V output. The split supply requirement is usually fulfilled by While the LT1074 power switch is on, current flows from a switcher with a multiple-secondary transformer or by the input voltage source through the switch, through multiple switchers. An alternative approach, shown in capacitor C2 and inductor L1 (Trace C), and into the load. Figure 50, uses an LT1074 switching regulator IC, two A portion of the switch current also flows into inductor L2 inductors and a “flying” capacitor to generate a dual- (Trace D). This current is used to recharge C2 and C4 output supply that accepts a wide range of input voltages. during the switch OFF time to a potential equal to the This solution is particularly noteworthy because it uses positive output voltage (VOUT). The current waveforms for only one switching regulator IC and does not require a both inductors occur on top of a DC level. transformer. Inductors are preferred over transformers The waveforms are virtually identical because the induc- because they are readily available and more economical. tors have identical values and because the same voltage The operating waveforms for the circuit are shown in potentials are applied across them during the switching Figure 51. During the switching cycle, the LT1074’s VSW cycles.

AN66-29 Application Note 66

VOUT C2 15V µ 470 F 0.5A 25V L1 50µH

5 4 VIN VSW + C6 VR1 L2 D1 0.01µF µ MUR410 LT1074 50 H R4 1 FB 20k GND V VIN C 8V 32 C7 R2 µ C3 + TO 40V 0.01 F R5 7.50k R1 470µF 20k 1% 3.3k 25V C1 + C5 R3 1000µF 0.01µF 1.30k 50V 1%

C4 + D2 EQUATION 1: VOUT= 2.21V* (1 + R2/R3) 470µF MUR410 VOUT = –V OUT 25V C1 = NICHICON UPL1H102MRH C2, C3, C4 = NICHICON UPL1E471MPH D1, D2 = MOTOROLA MUR410 –VOUT L1, L2 = COILTRONICS CTX50-2-52 (407) 241-7876 –15V 0.5A AN66 F50 Figure 50. Schematic Diagram for ±15V Version

interval the voltage on the VSW pin is equal to a diode drop A = 20V/DIV below the negative output voltage (–VOUT). L2’s current VSW then circulates between both D1 and D2, charging C2 and C4. The energy stored in L1 is used to replace the energy B = 2A/DIV ISW, IC1 lost by C2 and C4 during the switch ON time. Trace G is capacitor C2’s current waveform. Capacitor C4’s current C = 1A/DIV IL1, IC3 waveform (Trace F) is the same as diode D2’s current less the DC component. Assuming that the forward voltage D = 1A/DIV IL2 drops of diodes D1 and D2 are equal, the negative output voltage (–VOUT) will be equal to the positive output voltage (VOUT). After the switch turns on again the cycle E = 1A/DIV is repeated. ID1, IC3 Figure 52 shows the excellent regulation of the negative output voltage for various output currents. The negative F = 1A/DIV ID2, IC4 15.3

15.2

G = 1A/DIV 15.1 IC2 IOUT = 0.5A 15.0 (V) AN66 F51

µ OUT 5 s/DIV 14.9 –V I = –I Figure 51. LT1074 Switching Waveforms 14.8 OUT OUT

14.7 When the switch turns off, the current in L1 and L2 begins 14.6 to ramp downward, causing the voltages across them to 0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.5 reverse polarity and forcing the voltage at the VSW pin –IOUT (A) below ground. The VSW pin voltage falls until diodes D1 AN66 F52 (Trace E) and D2 (Trace F) are forward biased. During this Figure 52. –15V Output Regulation Characteristics

AN66-30 Application Note 66 output voltage tracks the positive supply (VOUT) within in Figure 50, with an internal 2.21V reference voltage and 200mV for load variations from 50mA to 500mA. Negative then varies the duty cycle until the two values are equal. output load current should not exceed the positive output The RC network (R1 and C5 in Figure 50) connected to the load by more than a factor of 4; the imbalance causes loop VC pin along with the R4/R5 and C6/C7 network provides instabilities. For common load conditions the two output sufficient compensation to stabilize the control loop. Equa- voltages track each other perfectly. tion 1 can be used to determine the output voltage. Another advantage of this circuit is that inductor L1 acts as Figure 54 shows the circuit’s –5V load regulation charac- both an energy storage element and as a smoothing filter teristics and Figure 55 shows its efficiency. for the positive output (V OUT). The output ripple voltage Refer to the schematic diagram in Figure 56 for modified has a triangular waveshape whose amplitude is deter- component values to provide ±5V at 1A. mined by the inductor ripple current (see trace C of Figure 51) and the ESR (effective series resistance) of the output 5.7 capacitor (C3). This type of ripple is usually small so a post 5.6 filter is not necessary. 5.5 Figure 53 shows the efficiency for a 0.5A common load at 5.4 various input voltages. The two main loss elements are the (V) 5.3 OUT 5.2 IOUT = 1A output diodes (D1 and D2) and the LT1074 power switch. –V At low input voltages, the efficiency drops because the 5.1 switch’s saturation voltage becomes a higher percentage 5.0 IOUT = –IOUT of the available input supply. 4.9 4.8 The output voltage is controlled by the LT1074 internal 0 0.10.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 error amplifier. This error amplifier compares a fraction of –IOUT (A) the output voltage, via the R2 to R3 divider network shown AN66 F54 Figure 54. –5V Output Regulation Characteristics

75 75

70 70

65 65

EFFICIENCY (%) 60 EFFICIENCY (%) 55 55

50 0 510 15 20 25303540 50 0 51015 2025 30 35 40 INPUT VOLTAGE (V) AN66 F53 INPUT VOLTAGE (V) AN66 F55 Figure 53. ±15V Efficiency Characteristics with 0.5A Common Load Figure 55. ±5V Efficiency Characteristics with 1A Common Load

AN66-31 Application Note 66

VOUT C2 5V 680µF 1A 16V L1 50µH

5 4 VIN VSW + C6 VR1 L2 D1 0.01µF LT1074 50µH MBR360 R4 1 20k FB V GND V IN C C7 R2 8V 32 2.80k 0.01µF R5 C3 + TO 40V 1% R1 20k 680µF + 2k 16V C1 C5 R3 µ 1000 F 0.033µF 2.21k 50V 1%

EQUATION 1: V = 2.21V* (1 + R2/R3) C4 + OUT D2 µ V = –V 680 F OUT OUT MBR360 16V C1 = NICHICON UPL1H102MRH C2, C3, C4 = NICHICON UPL1C681MPH AN66 F56 D1, D2 = MOTOROLA MBR360 L1, L2 = COILTRONICS CTX50-2-52 (407) 241-7876 –VOUT –5V 1A Figure 56. Schematic Diagram for ±5V Version

Regulators—Switching A unique EXT VCC pin on the LTC1159 allows the MOSFET (Inverting) drivers and control circuitry to be powered from the output of the regulator. In Figure 57 this is accomplished by HIGH EFFICIENCY 12V TO –12V CONVERTER grounding EXT VCC, placing the entire 12V output voltage by Milton Wilcox and Christophe Franklin across the driver and control circuits (remember the ground pins are at –12V). This is permissible with the It is difficult to obtain high efficiencies from inverting LTC1159, which allows a maximum of 13V between the switching regulators because the peak switch and induc- Sense and Ground pins. During start-up or short-circuit tor currents must be roughly twice the output current. conditions, operating power is supplied by an internal Furthermore, the switch node must swing twice the input 4.5V low dropout linear regulator. This start-up regulator voltage (24V for a 12V inverting converter). The adjustable automatically turns off when the output falls below –4.5V. version of the LTC1159 synchronous stepdown controller is ideally suited for this application, producing a combina- A cycle of operation begins when Q1 turns on, placing the tion of better than 80% efficiency, low quiescent current 12V input across the inductor. This causes the inductor and 20µA shutdown current. current to ramp to a level set by the error amplifier in the LTC1159. Q1 then turns off and Q2 turns on, causing the The 1A circuit shown in Figure 57 exploits the high input- current stored in the inductor to flow to the –12V output. voltage capability of the LTC1159 by connecting the con- At the end of the 5µs off-time (set by capacitor C troller ground pins to the –12V output. This allows the T), Q2 simple feedback divider between ground and the output turns off and Q1 resumes conduction. With a 12V input the (comprising R1 and R2) to set the regulated voltage, since duty cycle is 50%, resulting in a 100kHz operating fre- the internal 1.25V reference rides on the negative output. quency. The inductor connects to ground via the 0.05Ω current- sense resistor.

AN66-32 Application Note 66

INPUT +30% 12V –10%

330µF + 35V 1N4148 0.1µF Q1 NICHICON Si9435 UPL1V331M 0.15µF

1 16 PGATE CAP 2 15 VIN SHDN2

µ 3 14 0.1 F VCC EXT VCC LTC1159 MBRS140 L1 4 13 µ PDRIVE NGATE Q2 100 H Si9410 DALE 1N5818 3.3µF 5 12 TJ4-100-1µ VCC PWR GND

6 + 11 OUTPUT CT SGND –12V CT 1A 390pF 7 10 R1 I V TH FB 200pF 10.5k 6800pF 150µF 8 9

SENSE– SENSE+ 16V + OS-CON 5V OR 3.3V 1000pF R2 × 2 Q4 90.9k Q3 2N7002 SHUTDOWN TP0610L 1k 100Ω 100Ω 20k 0.05Ω

5.1V 510k 1N5993

AN66 F57

Figure 57. LTC1159 Converts 12V to –12V at 1A

The LTC1159, like other members of the LTC1148 family, 100 automatically switches to Burst Mode operation at low output currents. Figure 57’s circuit enters Burst Mode 90 operation below approximately 200mA of load current. 80 This maintains operating efficiencies exceeding 65% over two decades of load current range, as shown in Figure 58. 70 Quiescent current (measured with no load) is 1.8mA. EFFICIENCY (%) Complete shutdown is achieved by pulling the gate of Q3 60 low. Q3, which can be interfaced to either 3.3V or 5V logic, 50 creates a 5V shutdown signal referenced to the negative 10 100 1000 output voltage to activate the LTC1159 Shutdown 2 pin. OUTPUT CURRENT (mA) AN66 F58 Additionally, Q4 offsets the VFB pin to ensure that Q1 and Q2 remain off during the entire shutdown sequence. In Figure 58. Efficiency Plot of Figure 57’s Circuit shutdown conditions, 40µA flows in Q3 and only 20µA is taken from the 12V input.

AN66-33 Application Note 66

REGULATED CHARGE PUMP POWER SUPPLY In this application less than 5mA output current is re- by Tommy Wu quired. As a result, charge pump capacitor C1 is reduced to 1µF from the usual 10µF. Curves of output voltage with The circuit shown in Figure 59 uses an LTC1044A charge and without feedback are shown in Figure 60. The equiva- pump inverter to convert a 5V input to a –1.7V potential as lent output impedance of the charge pump is reduced from required for a certain LCD panel. Output regulation is approximately 100Ω to 5Ω. provided by a novel feedback scheme, which uses components Q1, R1 and R2. Without feedback the charge pump A variety of output voltages within the limits of the curve would simply develop approximately –5V at its output. in Figure 60 can be set by simply adjusting the VBE With feedback applied, VOUT charges in the negative multiplier action of Q1, R1 and R2. Tighter regulation or direction until the emitter of Q1 is biased by the divider a higher tolerance could be obtained by adding a refer- comprising R1 and R2. Current flowing in the collector ence or additional gain, at the expense of increased tends to slow the LTC1044A’s internal oscillator, reducing complexity and cost. the available output current. The output is thereby main- –5 tained at a constant voltage. VIN = 5V –4 NO FEEDBACK 1 8 5V INPUT 2 7 + 1µF –3 + LTC1044A C1 3 6 Q1* 1µF ZTX384 4 5 –2 R1 R2 OUTPUT (V) 47k 100k

–1.7V –1 WITH FEEDBACK OUTPUT

*ZETEX (516) 543-7100

µ + 10 F –0

AN66 F59 0510 15 20 LOAD (mA) Figure 59. Regulated Charge Pump AN66 F60

Figure 60. Effect of Feedback on Output Voltage

AN66-34 Application Note 66

LTC1174: A HIGH EFFICIENCY BUCK CONVERTER If higher output currents are desired Pin 7 (IPGM) can be by San-Hwa Chee and Randy Flatness connected to VIN. Under this condition the maximum load current is increased to 450mA. The resulting circuit The LTC1174 is an 8-pin SO “user-friendly” step-down and efficiency curves are shown in Figures 63 and 64 converter. (A PDIP package is also available.) Only four respectively. external components are needed to construct a complete high efficiency converter. With no load it requires only 100 130µA of quiescent current; this decreases to a mere 1µA L = 100µH VOUT = 5V 95 upon shutdown. The LTC1174 is protected against output IPGM = 0V shorts by an internal current limit, which is pin selectable 90 V = 6V V = 9V to either 340mA or 600mA. This current limit also sets the IN IN inductor’s peak current. This allows the user to optimize 85 the converter’s efficiency depending upon the output EFFICIENCY (%) 80 current requirement. In dropout conditions, the internal 0.9Ω (at a supply 75 voltage of 9V) power P-channel MOSFET switch is turned 70 on continuously (DC), thereby maximizing the life of the 1 10 100 200 LOAD CURRENT (mA) battery source. (Who says a switcher has to switch?) In AN66 F62 addition to the features already mentioned, the LTC1174 boasts a low-battery detector. All versions function down Figure 62. Efficiency vs Load Current to an input voltage of 4V and work up to an absolute VIN maximum of 13.5V. For extended input voltage, high 9V 6 + 100µF* voltage parts are also available that can operate up to an VIN 0.1µF 3 8 20V absolute maximum of 18.5V. LBIN SHDN 2 LTC1174-5 1 5V Output Applications LBOUT VOUT 100µH† 7 5 5V Figure 61 shows a practical LTC1174-5 circuit with a IPGM SW 425mA GND 1N5818 + 220µF* minimum of components. Efficiency curves for this circuit 4 10V * SANYO OS-CON at two different input voltages are shown in Figure 62. Note †COILTRONICS CTX100-4 that the efficiency is 94% at a supply voltage of 6V and load COILTRONICS (407) 241-7876 AN66 F63 current of 175mA. This makes the LTC1174 attractive to all Figure 63. Typical Application for Higher Output Currents power sensitive applications and shows clearly why switching regulators are gaining dominance over linear regulators in battery-powered devices. 100

95 VIN 9V 6 + 100µF* V 0.1µF 90 V = 6V IN 20V IN V = 9V 3 8 IN LBIN SHDN 85 LTC1174-5 2 1 LBOUT VOUT 100µH† EFFICIENCY (%) 80 7 5 5V I SW L = 100µH PGM 175mA + 75 VOUT = 5V GND 220µF* I = V 1N5818 PGM IN 4 10V COILTRONICS = CTX100-4

AN66 F61 70 * SANYO OS-CON 1 10 100 500 †COILTRONICS CTX100-4 COILTRONICS (407) 241-7876 LOAD CURRENT (mA) AN66 F64 Figure 61. Typical Application for Low Output Currents Figure 64. Efficiency vs Load Current

AN66-35 Application Note 66

More Applications A 5V to 3.3V Converter

Positive-to-Negative Converter The LTC1174-3.3 is ideal for applications that require 3.3V at less than 450mA. A minimum board area surface mount The LTC1174 can easily be set up for a negative output 3.3V regulator is shown in Figure 66. Figure 67 shows that voltage. The LTC1174-5 is ideal for –5V outputs as this this circuit can achieve efficiency greater than 85% for configuration requires the fewest components. Figure 65 load currents between 5mA and 450mA. shows the schematic for this application with low-battery detection capability. The LED will turn on at input voltages INPUT VOLTAGE 4V TO 12.5V + 15µF* below 4.9V. The efficiency of this circuit is 81% at an input 6 25V 0.1µF × voltage of 5V and output current of 150mA. VIN 3 7 8 IPGM SHDN INPUT VOLTAGE LTC1174-3.3 4V TO 7.5V 3 1 LBIN VOUT 50µH† V 33µF* 2 5 OUT + + LBOUT SW 3.3V 4.7k 6 0.1µF 16V 450mA × 2 VIN GND + 33µF** 1N5818 LOW- 7 8 4 16V 270k I SHDN × BATTERY PGM 2 LTC1174-5 INDICATOR 2 1 * (3) AVX TPSD156K025 LBOUT VOUT ** (2) AVX TPSD336K016 † 50µH† COILTRONICS CTX50-4 3 5 LBIN SW COILTRONICS (407) 241-7876 AN66 F66 + 33µF* GND 39k 1N5818 16V Figure 66. 5V to 3.3V Output Application 4 × 2 VOUT –5V * AVX TPSD336K016 150mA † COILTRONICS CTX50-4 100 COILTRONICS (407) 241-7876 AN66 F65

Figure 65. Positive to – 5V Converter 90 with Low-Battery Detection VIN = 5V

70 EFFICIENCY (%) L = 50µH 60 VOUT = 3.3V IPGM = VIN COILTRONICS = CTX50-4 50 1 10 100 500

LOAD CURRENT (mA) AN66 F67

Figure 67. Efficiency vs Load Current

AN66-36 Application Note 66

90 Regulators—Switching 300W (Power Factor Corrected) 85 THE NEW LT1508/LT1509 COMBINES POWER FACTOR 150W CORRECTION AND A PWM IN A SINGLE PACKAGE by Kurk Mathews 80 EFFICIENCY (%)

Typical Application 75 30W Figure 68 shows a 24VDC, 300W, power-factor corrected, universal input supply. The continuous, current mode 70 100 150 200250 300 boost PFC preregulator minimizes the differential mode VIN (AC) input filter size required to meet European low frequency AN66 F71 conducted emission standards while providing a high Figure 71. Efficiency Curves for Figure 68’s Circuit power factor. The 2-transistor forward converter offers many benefits, including low peak currents, a current step. Regulation is maintained to within 0.5V. nondissipative snubber, 500VDC switches and automatic Efficiency curves for output powers of 30W, 150W and core reset guaranteed by the LT1509’s 50% maximum 300W are shown in Figure 71. The PFC preregulator alone duty-cycle limitation. An LT1431 and inexpensive has efficiency numbers of between about 87% and 97% optoisolation are used to close the loop conservatively at over line and load. 3kHz with excess phase margin (see Figure 69). Figure 70 Start-up of the circuit begins with the LT1509’s VCC shows the output voltage’s response to a 2A to almost 10A bypass capacitors trickle charging through 91kΩ to 16VDC, overcoming the chip’s 0.25mA typical start-up 80 90 current (VCC ≤ lockout voltage). PFC soft start is then 60 75 released, bringing up the 382VDC bus with minimal overshoot. As the bus voltage reaches its final value, the 40 60 forward converter comes up powering the LT1431 and

20 45 closing the feedback loop. A 3-turn secondary added to the 70-turn primary of T1 bootstraps VCC to about LOOP GAIN (dB) 0 30 15VDC, supplying the chip’s 13mA requirement as well PHASE MARGIN (DEGREES) –20 15 as about 39mA to cover the gate current of the three FETs and high side transformer losses. A 0.15Ω sense resis- –40 0 10 100 1K10K 100K tor senses input current and compares it to a reference FREQUENCY (Hz) current (IM) created by the outer voltage loop and multi- AN66 F69 plier. Thus, the input current follows the input line voltage Figure 69. Bode Plot ot the Circuit Shown in Figure 68 and changes, as necessary, in order to maintain a constant bank voltage. The forward converter sees a voltage input of 382VDC unless the line voltage drops out, in µ 5A/DIV which case the 470 F main capacitor discharges to 250VDC before the PWM stage is shut down. Compared to a typical off-line converter, the effective input voltage

0.5A/DIV range of the forward converter is smaller, simplifying the design. Additionally, the higher bus voltage provides greater hold-up times for a given capacitor size. The high side transformer effectively delays the turn-on spike to AN66 F85 the end of the built-in blanking time, necessitating the Figure 70. Figure 68’s Response to a 2A to ≈10A Load Step external blanking transistor.

AN66-37 Application Note 66 T2 0.005" Cu Ω OUT AN66 F68 × ETD44-P 1W 10 1000pF T2 7 TURNS G1 FEP 30DP (DUAL) 26AWG 26AWG LPRI = 3.1mH 12.5A 24V 0.9" TRI-FILAR TRI-FILAR 17 TURNS 17 TURNS , Ω , 2W 2000pF 2) Ω 10 2W × MUR150 ( RPS2 IRF840 RG ALLEN 0.51 H µ 67 T150-52 39T 12AWG MUR150 IRF840 + 4700pF "Y" 3) F, 50V × µ + ( NICHICON 20k PL12,5X25 470 + F 2220pF "Y" µ 1 63V FILM 20k 2) × 15V 1N965 ( 30.1k 1% 3.4k 1% OUTPUT COM 24.9k 1% 10k 1% Ω 8 7 F REF 10 µ V MID REF Ω Ω R 0.0022 20 T3 10:15 100 TURNS Ω TOP R F FILM 2N2907 10 µ F 1 – + µ F µ 0.1 + V GND-S 470 450V 20k + F µ 382VBUS C1 1 400V Ω Ω 100pF 2N2222A 220 220 LT1431 1k COMP 2.2k 2.2k Ω (DUAL) Ω 470 IRFP450 MURH860CT COLL GND-F 134 62 5 330 F µ 1N5819 330 35V F 2N2222A FUJI ERA82-004 0.6A/40VR µ 100pF 20k CNY17-3 15V 0.01 + F F 20 19 µ µ Ω 1N5819 20 2.2 50V 2.2 50V C 2k V RAMP GTDR1 REF GTDR2 + + V OUT 1.2V F T1 CA µ – + 1 FILM 17 18 ERA82-004 ERA82-004 CC 20k F µ F µ F 300pF µ 2W 91k T1 200 0.047 + – 4.02k 1% SENSE 0.001 I SS1 V + Ω 15V R1 OUT 5W 4.02k 1% 0.15 M 15k SET LIM PK F µ 1.8k 1 FILM 10k LT1509 BR1 12 F REF REF µ RT1 =KETEMA S65T SURGE GARD T1 =COILTRONICS CTX02-12378-2, (407) 241-7876 T3 =BI TECHNOLOGY HM41-11510, (714) 447-2345 C1 =ELECTRONIC CONCEPTS 5MP12J105K R1 =JW MILLER/FUKUSHIMA MPC71 BR1=GENERAL INSTRUMENTS KBU6J SS2 R V V F µ F µ 0.001 0.1 IN 10 5 8 7 6 1 V RT1 0.001 OUT VA SET 0.1 "X" 330k F Danger!! Lethal Voltages Present µ + – 4700pF "Y" 0.1 "X" 4700pF "Y" 0.047 23 4 131516 SENSE AC V OVP I GND2 GND1 C F 9 F µ 14 11 µ 20k 1% 0.47 0.0047 0.1 "X" 499k 1% 499k 1% 20k 1% 1M 1/2W NOTE: UNLESS OTHERWISE SPECIFIED 1. ALL RESISTORS 1/4W, 5% 2. ALL CAPACITANCE VALUES IN MICROFARADS 6A 382VBUS FAST 499k 1% 499k 1% IN V 90VAC TO 264VAC

Figure 68. Schematic Diagram of 300W 24VDC Output Power Factor Corrected Universal Input Supply

AN66-38 Application Note 66

Regulators—Switching If the input voltage increases above 4V, the internal error (Discussion) amplifier, acting to keep the output at 5V, boosts the voltage on C1 to a level greater than 1V above the input. ADDING FEATURES TO THE BOOST TOPOLOGY This voltage controls Q1 to provide the desired output with by Dimitry Goder the transistor operating as a linear pass element. The output does not change abruptly during the switch-over A boost-topology switching regulator is the simplest so- between step-up and step-down modes because it is lution for converting a 2- or 3-cell input to a 5V output. monitored in both modes by the same error amplifier. Unfortunately, boost regulators have some inherent dis- advantages, including no short-circuit protection and no Figure 73 shows efficiency versus input voltage for shutdown capability. In some battery-operated products, 5V/100mA output. The break point at 4.25V is evidence of external chargers or adapters can raise the battery voltage Q1 beginning to operate in a linear mode with an attendant to a potential higher than the 5V output. Under this roll-off of efficiency. Below 4.25V the circuit operates as a condition a boost converter cannot maintain regulation— boost regulator and maintains high efficiency across a the high input voltage feeds through the diode to the broad range of input voltages. output. The circuit can be shut down by pulling the LT1301’s The circuit shown in Figure 72 overcomes these problems. Shutdown pin high. The LT1301 ceases switching and Q1 An LT1301 is used as a conventional boost converter, automatically turns off, fully disconnecting the output. preserving simplicity and high efficiency in the boost This stays true over the entire input voltage range. mode. Transistor Q1 adds short-circuit limiting, true shut- Q1 also provides overload protection. When the output is down and regulation when there is a high input voltage. shorted the LT1301 operates in a cycle-by-cycle current When the input voltage is lower than 4V and the regulator limit. The short-circuit current depends on the maximum is enabled, Q1’s emitter is driven above its base, saturating switch current of the LT1301 and on the Q1’s gain, the transistor. As a result, the voltages on C1 and C2 are typically reaching 200mA. The transistor can withstand roughly the same and the circuit operates as a conven- overload for several seconds before heating up. For sus- tional boost regulator. tained faults the thermal effects on Q1 should be carefully considered.

R1 1.5k 100 BOOST LINEAR STEP- L1 MBR0520L RANGE DOWN RANGE 22µH V VOUT 90 IN 5V 2V TO 9V Q1 100mA ZTX788B + C3 80 33µF 6 7 VIN SW 2 4 SELECT SENSE 70

LT1301 EFFICIENCY (%) 3 5 SHUTDOWN SHDN I + + LIM C1 C2 8 1 47µF 100µF 60 PGND GND

R2 3.3k 50 23456789 INPUT VOLTAGE (V)

AN66 F73 AN66 F72 Figure 73. Efficiency vs Input Voltage Figure 72. Q1 Adds Short-Circuit Limiting, True Shutdown and Regulation for 5V/100mA Output When There Is a High Input Voltage to the LT1301 in Boost Mode

AN66-39 Application Note 66

SENSING NEGATIVE OUTPUTS Q2 is connected as a diode and is used to compensate for by Dimitry Goder Q1’s base-emitter voltage change with temperature and collector current. Both transistors see the same collector Various switching regulator circuits exist to provide posi- current and their base-emitter voltages track quite well. tive-to-negative conversion. Unfortunately, most control- Because the base-emitter voltages cancel, the voltage lers cannot sense the negative output directly; they require across R2 also appears on the LT1172’s Feedback pin. a positive feedback signal derived from the negative output. This creates a problem. The circuit presented in Figure The resulting output voltage is given by the following 74 provides an easy solution. formula: The LT1172 is a versatile switching regulator that contains R3 V = V – V an onboard 100kHz PWM controller and a power switch- OUT FB R2 BE ing transistor. Figure 74 shows the LTC1172 configured to provide a negative output using a popular charge pump where VFB is the LT1172 internal 1.244V reference and VBE ≈ technique. When the switch turns on, current builds up in is Q1’s base/emitter voltage ( 0.6V). The VBE term in the the inductor. At the same time the charge on C3 is equation denotes a minor output voltage dependency on transferred to output capacitor C4. During the switch off- input voltage and temperature. However, the variation due time, energy stored in the inductor charges capacitor C3. to this factor is usually well below 1%. A special DC level-shifting feedback circuit consisting of Essentially, Q1 holds its collector voltage constant by Q1, Q2, and R1 to R4 senses the negative output. changing its collector current and will function properly as Under normal conditions Q1’s base is biased at a level long as some collector current exists. This puts the about 0.6V above ground and the current through resistor following limitation on R1: at mi