Proposed Improvements to the Neutral Beam Injector Power Supply System

ABSTRACT

PROPOSED IMPROVEMENTS TO THE NEUTRAL BEAM INJECTOR POWER SUPPLY SYSTEM

by Zhen Jiang

The tokamak fusion reactor is one of the most promising and well-developed designs for fusion energy production. Scientists around the world use tokamaks to research methods of generating electrical energy from the fusion reaction. Furthermore, efforts are now underway to design a new large sized tokamak, Chinese Fusion Engineering Test Reactor (CFETR), with the aim of demonstrating fusion energy as a viable source of power. As this project is just beginning, it is necessary to evaluate new and emerging technologies that can be used in this endeavor. Power electronics play a crucial role in fusion energy research and are the focus of the thesis. The High Power, Power Supplies (HPPS) transforms electrical energy from the grid into AC and DC signals with extremely high voltage and current magnitudes. For example, the Neutral Beam Injectors (NBI) require a power source that is capable of generating 110 kVDC at nearly 10 MW. This thesis evaluates two new types of technology for use in the NBI power supply; the utilization of new semiconductor switching devices and the application of new circuit topologies. The switching devices used in the existing HPPS all utilize silicon based semiconductors. Within the last ten years, new devices created from Wide Bandgap (WBG) semiconductors have become commercially available. This thesis demonstrates that gains in efficiency are possible by utilizing WBG based power devices in the NBI power supply. It also explores the application of a Modular Multilevel Converters (MMC) as a replacement to the existing topology used in the NBI power supply. It shows that the MMC can be used as both a rectifier for the NBI and an active filter for the electrical grid. The case study for this work is the NBI power supply of Experimental Advanced Superconducting Tokamak in Hefei, China. However, the findings are applicable to CFETR as well as other future designs of tokamak reactors.

PROPOSED IMPROVEMENTS FOR THE NEUTRAL BEAM INJECTOR POWER SUPPLY SYSTEM

Thesis

Submitted to the

Faculty of Miami University

in partial fulfillment of

the requirements for the degree of

Master of science

Zhen Jiang

Miami University

Oxford, Ohio

2017

Adviser: Dr. Mark J. Scott

Committee member: Dr. Dmitriy Garmatyuk

Committee member: Dr. Donald R.Ucci

This thesis titled

THE PROPOSED IMPROVEMENT FOR NEUTRAL BEAM INJECTOR POWER SUPPLY SYSTEM

Zhen Jiang

has been approved for publication by

College of Engineering and Computer

and

Department of Electrical and Computer engineering

______Dr. Mark J. Scott

______Dr. Dmitriy Garmatyuk

______Dr. Donald R.Ucci

Table of Contents Acknowledgements ...... 1

Chapter 1: Introduction ...... 1

1.1 Motivation ...... 1

1.2 Overview of Power Electronics ...... 3

1.3 Thesis Statement ...... 5

1.4 Chapter Outlines ...... 6

Chapter 2 Current and Emerging Power Electronic Devices ...... 8

2.1 Power Loss Analysis of Different Switching Devices ...... 8

2.1.1 Diode Power Loss ...... 8

2.1.2 IGBT Power Loss ...... 9

2.1.3 MOSFET Power Loss ...... 11

2.2 Wide Bandgap Semiconductor ...... 12

2.2.1 Comparison of GaN, Si, SiC ...... 13

2.2.2 SiC Power Devices ...... 14

2.3 Summary of the Chapter ...... 15

Chapter 3: Overview of Tokamak Reactors ...... 16

3.1 Nuclear Fusion Reactions ...... 16

3.2 The Tokamaks ...... 17

3.2.1 The EAST Tokamak ...... 19

3.2.2 The CFETR Tokamak ...... 21

Chapter 4: NBI Power Supply ...... 23

4.1 Background ...... 23

4.2 Power Supply System...... 24

4.3 Summary of the Chapter ...... 26

Chapter 5: NBI Power Loss Analysis ...... 27

5.1 Active Rectification ...... 27

5.2 PSM Physical Calculation ...... 29

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5.2.1 Rectifier Power Loss ...... 31

5.2.2 Power Module Power Loss ...... 39

5.3 PSIM Simulation Results...... 42

5.3.1 Existing NBI PSM ...... 42

5.3.2 Si IGBT PSM ...... 44

5.3.3 SiC MOSFET PSM ...... 46

5.4 Summary of the Chapter ...... 47

Chapter 6: Improved NBI Topology based on MMC ...... 49

6.1 Voltage Source Converter Technology...... 49

6.2 Modular Multilevel Converter ...... 50

6.3 MMC Benefits to NBI Power Supply System ...... 53

6.3.1 Simplified Distribution System ...... 54

6.3.2 Simplified PSM Structure ...... 54

6.3.3 RPC and HF ...... 55

6.3.4 WBG Power Devices ...... 60

Chapter 7 ...... 68

7.1 Conclusion ...... 68

7.2 Future Work ...... 69

Reference ...... 72

List of Tables

TABLE 1.1 PARAMETER OF CHEMICAL, FISSION AND FUSION ENERGY [7]

TABLE 2.1 COMPARISON OF SI , SIC, AND GAN SEMICONDUCTOR PROPERTIES [16], [18]

TABLE 3.1 PARAMETERS OF DIFFERENT COILS IN EAST [5], [52]

TABLE 3.2 PARAMETERS OF ITER AND CFETR POWER SUPPLY SYSTEM [4], [55]

TABLE 4.1 THE PARAMETERS OF THE NBI POWER SUPPLY [5]

TABLE 5.1 RATED PARAMETERS FOR DIODE (DDB6U215N18) TABLE 5.2 RESULTS OF POWER LOSS CALCULATIONS FOR THE SI DIODE RECTIFIER TABLE 5.3 THEORETICAL CURRENTS IN THE SI BASED ACTIVE RECTIFIER TABLE 5.4 PARAMETERS OF SI DEVICE (FF300R17KE4)

TABLE 5.5 RESULTS OF POWER LOSS CALCULATIONS FOR THE SI BASED ACTIVE RECTIFIER

TABLE 5.6 PARAMETERS OF SIC DEVICE (CAS300M17BM2)

TABLE 5.7 RESULTS OF POWER LOSS CALCULATIONS FOR THE SIC BASED ACTIVE RECTIFIER

TABLE 5.8 COMPARISON OF RECTIFIER CALCULATION RESULTS TABLE 5.9 DEVICE PARAMETERS OF SI POWER MODULE (FF300R17KE4) TABLE 5.10 RESULTS OF POWER LOSS CALCULATIONS FOR THE SI BASED POWER MODULE

TABLE 5.11 DEVICE PARAMETER OF SIC POWER MODULE (CAS300M17BM2)

TABLE 5.12 RESULTS OF POWER LOSS CALCULATIONS FOR THE SIC BASED POWER MODULE

TABLE 5.13 COMPARISON OF THE POWER MODULE’S POWER LOSSES FOR SI AND SIC VERSION

TABLE 5.14 NBI PARAMETERS USED FOR SIMULATION

TABLE 5.15 SIMULATION RESULT OF NBI PSM

TABLE 5.16 SIMULATION RESULT OF SI IGBT PSM

TABLE 5.17 SIMULATION RESULT OF SIC MOSFET PSM TABLE 5.18 POWER LOSS COMPARISON FOR ONE NBI PSM TABLE 6.1 ONE SUBMODULE COMPARISON BETWEEN NBI AND MMC

TABLE 6.2 MMC HARMONIC FILTER PARAMETERS IN SIMULATION

TABLE 6.3 THEORETICAL CURRENTS IN THE SI BASED DEVICES

TABLE 6.4 PARAMETERS OF SI DEVICE (FF300R17KE4)

TABLE. 6.5 POWER LOSS ESTIMATES FOR ONE SI PSM IN MMC

TABLE 6.6 THEORETICAL CURRENTS IN THE SIC BASED DEVICES

TABLE 6.7 PARAMETERS OF SIC DEVICE (CAS300M17BM2)

TABLE 6.8 POWER LOSS ESTIMATES FOR ONE SIC PSM IN MMC

TABLE 6.9 MMC PARAMETERS USED FOR SIMULATION v

TABLE 6.10 SIMULATION RESULT OF SI MMC

TABLE 6.11 SIMULATION RESULT OF SIC MMC

List of Figures

FIGURE 2.1 THE EQUIVALENT CIRCUIT OF A DIODE WHILE CONDUCTING. FIGURE 2.2 THE EQUIVALENT CIRCUIT OF AN IGBT WHILE CONDUCTING. FIGURE 2.3 THE EQUIVALENT CIRCUIT OF A MOSFET WHILE CONDUCTING. FIGURE 3.1 DIAGRAM OF THE FUSION REACTION PROCESS. FIGURE 3.2 DIAGRAM OF TOKAMAK MAGNETIC FIELDS [49].

FIGURE 3.3 DIAGRAM OF THE DESIGN FOR THE EAST POWER SUPPLY SYSTEM [5].

FIGURE 4.1 NBI POWER SUPPLY SYSTEM AND ONE OF THE POWER SUPPLY MODULES [58].

FIGURE 5.1 COMPARISON OF INPUT CURRENT OF PASSIVE AND ACTIVE RECTIFIER IN TIME DOMAIN.

FIGURE 5.2 COMPARISON OF INPUT CURRENT OF PASSIVE AND ACTIVE RECTIFIER IN FREQUENCY DOMAIN.

FIGURE 5.3 THE CIRCUIT OF SINGLE PHASE. FIGURE 5.4 SCHEMATIC OF ONE POWER SUPPLY MODULE USED IN NBI. FIGURE 5.5 THE CURRENT WAVEFORM THROUGH DEVICE IN DIODE RECTIFIER.

FIGURE 5.6 THE PHASE ANGLE OF INPUT CURRENT AND DUTY CYCLE.

FIGURE 5.7 THE PWM CONTROL TECHNIQUE.

FIGURE 5.8 THE CURRENT WAVEFORM THROUGH EACH SWITCHING DEVICE IN IGBT RECTIFIER.

FIGURE 5.9 THE CURRENT WAVEFORM THROUGH EACH SWITCHING DEVICE IN MOSFET RECTIFIER.

FIGURE 5.10 SIMULATION SCHEMATIC OF NBI PSM.

FIGURE 5.11 SIMULATION RESULTS FOR THE EXISTING NBI PSM’S INPUTS AND OUTPUTS.

FIGURE 5.12 SIMULATION SCHEMATIC OF IGBT PSM.

FIGURE 5.13 SETTING PARAMETERS IN IGBT FILE. FIGURE 5.14 SIMULATION RESULT OF SI PSM’S OUTPUTS AND INPUTS. FIGURE 5.15 SIMULATION SCHEMATIC OF MOSFET PSM.

FIGURE 5.16 SIMULATION RESULT OF SIC PSM’S OUTPUTS AND INPUTS. FIGURE 6.1 THE SCHEMATIC OF DOUBLE-STAR MMC AND ONE HALF BRIDGE SUBMODULE. FIGURE 6.2 THE SCHEMATIC OF STAR MMC.

FIGURE 6.3 THE SCHEMATIC OF DELTA MMC. FIGURE 6.4 (A), (B) NEGATIVE AND (C), (D) POSITIVE CURRENT FLOW INSIDE A SUBMODULE. FIGURE 6.5 CIRCUIT AND CONTROL DIAGRAM OF PROPOSED HARMONIC FILTER.

FIGURE 6.6. THE SIMULATION SCHEMATIC OF HARMONIC FILTER SYSTEM.

FIGURE 6.7. THE SIMULATION SCHEMATIC OF CONTROL SYSTEM FOR THE ACTIVE POWER FILTER

FIGURE 6.8. THE SIMULATION RESULTS OF CURRENT IN TIME DOMAIN.

FIGURE 6.9. THE SIMULATION RESULTS OF CURRENT IN FREQUENCY DOMAIN.

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FIGURE 6.10 THE CURRENT WAVEFORM THROUGH EACH SI BASED SWITCHING DEVICE.

FIGURE 6.11 THE SIMULATION SCHEMATIC OF MMC STRUCTURE.

FIGURE 6.12 THE SIMULATION SCHEMATIC OF (A) SI IGBT SM AND (B) SIC MOSFET SM FIGURE 6.13 THE SIMULATION RESULTS OF SI IGBT MMC. FIGURE 6.14 THE SIMULATION RESULTS OF SIC MOSFET MMC.

viii

Acknowledgements

The researcher would like to express the deepest gratitude to the following for their help. First deepest thanks to my thesis advisor, Dr. Mark J. Scott of the Department of Electrical and Computer engineering at Miami University, who always leading and helping me to finish this thesis. I am extremely grateful for his expert sincere, valuable guidance and encouragement extended to me until this thesis came to existence. It is a great honor to work under his supervision. I would like to acknowledge Dr. Donald R. Ucci and Dr. Dmitriy Garmatyuk at Miami University as the committee members of this thesis, and I am gratefully indebted to them for their very comprehensive comments and supports on this thesis. I would also like to express sincere appreciation to the experts who were involved in the validation survey for this work: Dr. Lei Yang and Dr. Peng Fu of Institute of Plasma Physics, Chinese Academy of Science. I must express my very profound gratitude to my friends, especially to Yujie Bai, Yixing Liao, Austin Lowder and Hassan Hassan providing me with kind endless help, generous advice and support during the study. Finally, I would express a deep sense of gratitude to my dearest darling mom, who has always stood by me like a pillar in times of need and to whom I owe my life for her constant love, encouragement, moral support and blessings. Special thanks for the love and encouragement are due to my one and only loving boyfriend, Yu Sun, without whom I would never have enjoyed so many opportunities. Thank you very much

Chapter 1 Introduction

Nuclear fusion is one of the most promising options for generating large amounts of energy in the future. It is the process where atomic nuclei collide together and release energy. Fusion scientists and engineers are developing the technology to harness this reaction in future power plants. To produce plasma, power electronics provide high voltage, high power signals to a tokamak reactor 1. Thus, any improvements to the power electronics used in a tokamak reactor, will subsequently benefit fusion energy production. This thesis explores new developments in power electronics and evaluates their potential for integration into the design of future fusion reactors.

1.1 Motivation

The energy of the sun powers the earth by a process known as 'fusion', wherein light atoms, such as hydrogen, fuse together and release energy in the form of neutrons. A fusion reaction requires considerably high pressures and temperatures, primarily between the hydrogen isotopes in the core of the sun (15 million ºC) [1]. At such high temperatures, any gas in the sun exists in a plasma state, which can be described as an ‘electrically-charged’ gas [1]. The negative electrons are separately charged from the positive atomic ions. Plasma is rarely found on earth: however, 99 % of the elements in the universe exist in a plasma state.

Fusion may be a promising method for producing energy on earth, but the reaction must be performed in a controlled manner. It is the objective for nuclear fusion scientists and engineers to harness this mechanism in future power plants. Considerable effort has already been put into achieving this goal, as research began in the 1950s [2]. While many issues with plasma production have been solved, further research is required to turn this into a viable energy source.

As the world continues to industrialize and economies become more interconnected, society needs to find an alternative option to satiate its increased energy consumption. Currently, 80 %

1 **Tokamak is toroidalnya kamera ee magnetnaya katushka – a Russian phrase meaning a ring-shaped magnetic chamber.

of the world’s energy production comes from fossil fuels [1]. This energy source is finite in nature and may become scarce at some point in the future. Furthermore, energy created from fossil fuels produces greenhouse gases that threaten to change the climate. Environmental problems are a serious concern and every effort should be made to combat them. Zero or low emission sources are promising alternatives. To that end, nuclear fusion is being explored as a panacea for the world’s energy problems [3].

When compared to traditional energy sources generated by fossil materials, fusion offers a secure supply of power with several key advantages. First, the fuel supplies will last for millions of years [1]. Relying on fossil fuels in the short term is fine, but nuclear energy is much more abundant and will be available over a relatively longer period of time. Second, fusion power is a sustainable source of power. Deuterium is one of the basic elements in a fusion reaction, and it can be refined from water. Third, fusion is clean with nearly no carbon emission. The only by-product is a small amount of helium, which is an inert gas that does not contribute to the emission of greenhouse gases [4], [5]. Fourth, a major advantage of the fusion based reaction is that there is no possibility of a core meltdown [6]. In this way, it can provide a highly efficient supply of energy. It is noted, however, that the plant would be a nuclear facility; hence, great care needs to be taken during construction and operation. After the end of its operation, only the reactor’s radioactive components would need to be stored in order to safeguard society from radiation effects. This effect will be ameliorated in about 100 years. Finally, the energy of a fusion reaction is estimated to be much larger than that obtained from a fission reaction or chemical fossil fuel production. As shown in Table 1.1, the difference is as high as five orders of magnitude.

TABLE 1.1 PARAMETER OF CHEMICAL, FISSION AND FUSION ENERGY [7] Chemical Fission Fusion 235 2 3 4 Reaction C + O= CO2 N + U = H + H = He + n Ba143 + Kr31 + 2n

Fuel Coal UO2 Deuterium + Tritium Temperature 700 K 1000K 100,000,000 K Energy (J/Kg) 3.3 X 107 2.1 X 1012 3.4 X 1014

1.2 Overview of Power Electronics Power electronics is the application of semiconductor switching devices in circuits that control the conversion of electrical power [8]. The energy conversion process is performed using devices such as diodes, thyristors and transistors. For a diode, the electrical characteristics of the system control whether it is on or off. It conducts forward current and blocks negative current. As for the thyristor, it blocks forward and reverse voltage, and conducts forward current. A control signal turns it on, but the system turns it off as the current commutates in the reverse direction. The family of controllable switches includes several types of devices: Bipolar Junction Transistor (BJT), Metal–Oxide–Semiconductor Field-Effect Transistor (MOSFET), Gate Turn-off Thyristor (GTO), and Insulated Gate Bipolar Transistor (IGBT). Diodes and IGBTs are the major devices applied in the fusion reactor. When this technology is quite mature now, IGBTs were state of the art components when many of the current generations of tokamaks were being design. In this thesis, MOSFETs based on wide bandgap (WBG) are studying as a substitution of existing compoents. Detailed analysis is provided in later chapters of some of these devices.

Pulse-width modulation (PWM) is a technique used to control the switching devices so that the output can be regulated. A PWM signal consists of two main components that define its behavior: a duty cycle and a frequency. The duty cycle is the ratio of switch’s conduction time over the period [9]. The frequency determines how fast the PWM completes a cycle, and

therefore how often it switches between high and low states and back again. PWM is crucial to operate power electronics, but it is also used in other fields, such as communication [10].

Power electronics can play an important role in power transmission. High Voltage Direct Current (HVDC) systems are one example of how this hardware can be used in the grid. The circuits take an AC voltage source and transformer it to a DC source. As DC voltage only transfers active power, a long distance HVDC transmission generally has lower losses and cost than an equivalent AC transmission scheme [11]. HVDC also requires less conductor volume per unit distance than an AC line, because there is no skin effect [12]. Furthermore, HVDC transmission is selected for other technical benefits and performance characteristics that are not achievable with AC transmission systems. It provides asynchronous interconnections between separate AC grid networks [13]. HVDC power flow can be automatically controlled to support either network during transient conditions [11]. The combination of economic and technical advantages offered by HVDC can make it a suitable choice for connecting electricity sources.

In HVDC systems, the equipment that performs the conversion between AC and DC is referred to as the converter. An AC/DC converter is a type of power electronics that is commonly found in many consumer goods, such as personal laptop adapters and cellphone battery chargers. Regardless of the topology, the station that operates with power flowing from AC to DC is referred to as the rectifier and the station that operates with power flowing from DC to AC is referred to as the inverter. Besides the rectifier and inverter, there are also DC to DC converters and AC to AC cycloconverters. The power supply systems applied in the tokamak rely on rectification.

When it comes to the voltage conversion, there are two common types of power supplies: the linear regulator (LR) and the switching mode power supply (SMPS). While both of them are available in power supplies, the main difference between them is in the ways they operate. The SMPS regulates the output by utilizing a transistor in a switching fashion. When the transistor is conducting current, the voltage drop can be minimized. Otherwise, almost no current flows through its power path [14]. Since the semiconductor transistor works ideally, the power loss can

be minimized. This results in the more efficient performance. By contrast, the regulating device of LR is made to act like a variable resistor in accordance with the load resulting in a constant output voltage [15]. Therefore, the traditional LR power supplies have a simplified circuit structure and generate little-to-no Electromagnetic Interference (EMI). However, a major drawback is excessive power dissipation of its series transistor [14].

1.3 Thesis Statement Fusion energy relies on power electronics to operate the tokamak. Since the construction of previous generations of tokamak reactors, new technology has developed that can enhance the performance of the High Power, Power Supplies (HPPS) used in fusion energy systems. This thesis presents analysis on potential improvements for Neutral Beam Injection (NBI) power supply used in a tokamak reactor.

For example, Wide Bandgap (WBG) based switching devices have emerged with ratings suitable for application within some of the power supplies in a tokamak reactor [16]. Furthermore, Modular Multilevel Converters (MMC) are a topology that have gained in popularity and are maturing quickly [17]. This research explores the utilization of both of these technologies in an HPPS for the NBI auxiliary heating systems of a tokamak reactor. The Experimental Advanced Superconducting Tokamak (EAST) is used as the case study for this work. However, the results are applicable to other tokamak based fusion energy systems, such as the Chinese Fusion Engineering Test Reactor (CFETR).

WBG devices have a unique molecular structure that enables power devices to be created with lower on-resistance [16]. Additionally, WBG components have faster switching speeds and can achieve lower switching losses [18]. WBG devices also have the potential to realize higher breakdown voltages. These benefits make WBG an attractive candidate for implementation into the HPPS. The thesis extends a previous study [16] to include the entire Power Supply Module (PSM) in the NBI power system, and considers the benefits of using WBG components in all of the auxiliary heating systems. The existing PSM structure of NBI power system uses a passive rectifier to convert the AC to DC. This can generate a significant amount of harmonics. Another

contribution of this work is to apply active rectification in the PSM to get a better harmonic performance.

The current NBI in EAST requires a 100 kV/100A HPPS. By utilizing an MMC topology over the existing method, several advantages can be realized for the NBI system. A single MMC can eliminate all but one of the transformers that are used in the existing NBI supply. It also eliminates the rectifiers in the existing PSM. Additionally, it is possible for the MMC to act as a hybrid power supply and harmonic filter [19]. This scenario is studied in this thesis. The inclusion of WBG power devices into an MMC also provides opportunities to modify and improve the submodule performance [20].

For a fusion power plant to be integrated into the existing grid structure, it will have to comply with relevant grids codes. This is an issue of concern as the harmonics generated by the poloidal field (PF) and toroidal field (TF), are huge. Therefore, EAST utilizes power electronics to perform Reactive Power Compensation (RPC) and harmonic filtering (HF) system. It enjoys a high priority and big cost in the EAST tokamak. In the AC power distribution systems, harmonics occur when the normal current waveforms are distorted by non-linear loads [21]. Therefore, in this thesis, the harmonics produced by non-linear loads are used to represent the harmonics occurring in the EAST system. This is necessary because creating the whole tokamak system inside of a simulator is extremely difficult.

This thesis analyzes the benefits of several modifications to the NBI system. First, alterations to the PSM are explored. This includes changing the Si based components for comparably rated WBG components. Furthermore, active rectification is examines as a second form of improvement. Next, a complete topological shift is explored. An MMC based rectifier is examined as a replacement for the existing power supply topology. The thesis presents methods for using the MMC as a rectifier and an active filter simultaneously. Finally, it presents the benefits achieved by combining all three of these technologies together.

1.4 Chapter Outlines In Chapter 2, the power loss analysis of different switching devices is reviewed. Current and

emerging power devices are also introduced and compared in detail. Silicon Carbide (SiC) MOSFETs are studied as a replacement for existing silicon (Si) IGBTs. The atomic structure of WBG semiconductors, in particular, SiC and Gallium Nitride (GaN), enables power devices to be developed with a number of advantages [18].

Chapter 3 presents an overview of the basics of fusion reactions. It discusses the structure of tokamak power supply systems, and provides some details on two of the Chinese tokamaks. The first (EAST) has been in operation for some time while the second is entering its initial design phase (CFETR).

Chapter 4 reviews the operating principles of the NBI power supply system. In the chapters that follow, the benefits of applying active rectification and using WBG switching devices in the PSM of existing NBI power supply are explored. This is followed by an examination of using alternative circuit topologies for the NBI power supply.

Chapter 5 focuses on the analysis of modifying the PSM of NBI power supply system. The harmonic performance is compared between the passive rectifier and the active rectifier. The analysis is supported with simulation results. The chapter concludes by showing how changes in PSM’s efficiency occur as a result of using the new type of switching devices.

In Chapter 6, a new structure is proposed for NBI power supply system. It is based on using an MMC as an AC/DC converter. Power loss estimates and simulation results are analyzed step-by-step. The method of accomplishing HF in MMC is another major point emphasized in this chapter.

Chapter 7 provides the conclusion of this thesis. A summary of the research objectives is provided along with their outcomes. Finally, the future directions generated by this research topic are discussed in detail.

Chapter 2 Current and Emerging Power Electronic Devices

Compared to circuits built with Si based semiconductors, WBG based devices can operate at much higher voltages, switch as faster speeds, and tolerate higher temperatures. These properties enable them to perform better than Si in many power electronic applications. This chapter begins with a review of power loss for different types of switching devices. This is provided to add context for the following section where the benefits and challenges of WBG semiconductors are compared with Si. A case is made for using SiC devices in fusion energy applications so the chapter concludes with a discussion on the different types of SiC based switching devices.

2.1 Power Loss Analysis of Different Switching Devices Diodes, IGBTs, and thyristors are mainly used in tokamak fusion reactors. Furthermore, MOSFETs are considered in this thesis as replaces for some of the existing switching devices. To provide context for the rest of the thesis, a brief discussion on these components is provided. Thyristors are not explored further in this work, but details about their operating principles can be found in tradition textbooks on power electronics.

One factor that causes power loss in a power supply system is the loss that occurs within the semiconductor device. This occurs either during the switching events, or during the conduction time [22]. Switching losses are due to the charging and discharging of semiconductor junctions during each switching cycle. The conduction loss occur because of the on-state voltage drop of the device. This value depends on the current and the voltage rating of the device.

2.1.1 Diode Power Loss A diode is a unidirectional device created from the junction of two dissimilar materials. When the diode is forward biased, it will allow the current to flow from the anode to the cathode. When the diode is reverse biased, it will block the current. When a diode is on, it can be approximated as a voltage source (푉퐹) and a resistor (푟푐푒) (see Fig. 2.1) in series. The conduction loss is due to the on-state voltage drop across the device [8]. Therefore, the conduction loss

(푃푐표푛_퐷푖표푑푒) can be represented as: 8

1 푇푠 푃푐표푛_퐷푖표푑푒 = ∗ ∫ 푉퐷(푡) ∗ 푖퐷(푡)푑푡 푇푠 0 2 = 푉퐹 ∗ 퐼푎푣푔_D + 푟푐푒 ∗ 퐼푟푚푠_퐷, (2.1)

Here, 퐼푎푣푔_D is the average value of the current through the diode, and 퐼푟푚푠_D is its RMS value.

The diode’s reverse recovery mechanism is responsible for generating switching losses during the turn-off transition. Therefore, the switching loss for a diode (푃푠푤_퐷푖표푑푒) is calculated by the reverse recovery charge (푄푟푟) and the voltage that the diode blocks (푉퐷).

푃푠푤_퐷푖표푑푒 = 푄푟푟 ∗ 푉퐷 ∗ 푓푠. (2.2) The total power loss is calculated by adding the conduction loss and the switching loss.

FIGURE 2.1 THE EQUIVALENT CIRCUIT OF A DIODE WHILE CONDUCTING. 2.1.2 IGBT Power Loss There are two types of conduction losses in an IGBT. The first is the on-state losses that occur when the IGBT is conducting first quadrant current and the second happens when it conducts third quadrant current. In the case of the former, the total power dissipation during conduction time (푃푐표푛_퐼퐺퐵푇) is computed by multiplying the on-state voltage and the on-state current [23]

1 푇푠 푃푐표푛_퐼퐺퐵푇 = ∗ ∫ 푉푐푒(푡) ∗ 푖푐푒(푡)푑푡. (2.3) 푇푠 0

When an IGBT is on, it can be approximated as a voltage source (푉푐푒) and a resistance (푟푐푒) in series (see Fig. 2.2). Therefore, the power dissipated by the IGBT is given as:

2 푃푐표푛_퐼퐺퐵푇 = 푉푐푒 ∗ 퐼푎푣푔_푐푒 + 푟푐푒 ∗ 퐼푟푚푠_푐푒. (2.4)

The time it takes an IGBT to turn-on is given as both the delay time (푡푑(표푛)) and the rise time (푡푟). The delay time is the interval between the time when the gate signal is applied and just before the drain-to-source voltage transition states. The rise time is a measurement of the time. It takes the drain-to-source voltage to go from 90 % of its block voltage to 10 %. Using this information, the energy lost during the turn-on (퐸표푛) phase is given as:

푡표푛 = 푡푑(표푛) + 푡푟 (2.5)

푡 퐸 = 표푛 푉 (푡) ∗ 푖 (푡) 푑푡. (2.6) 표푛 ∫0 푐푒 푐푒

The turn-off delay time (푡푑(표푓푓)) is the amount of time that elapse between the gate-to-source voltage changing and the drain-to-source voltage changing. The fall time (푡푓) is the interval of time from wen the drain-to-source voltage is 10 % and when it is 90 %. The energy dissipated during turn-off (퐸표푓푓) is given as:

푡표푓푓 = 푡푑(표푓푓) + 푡푓 (2.7)

푡 퐸 = 표푓푓 푉 (푡) ∗ 푖 (푡) 푑푡. (2.8) 표푓푓 ∫0 푐푒 푐푒

Since energy is lost at each interval, the switching losses on the IGBT (푃푠푤_퐼퐺퐵푇 ) are proportional to the switching frequency (푓푠):

푃푠푤_퐼퐺퐵푇 = (퐸표푛 + 퐸표푓푓) ∗ 푓푠. (2.9) The total power loss of IGBT is calculated from the sum of both the conduction losses and the switching losses. An antiparallel diode is always used along with an IGBT to conduct the freewheeling current. Its power loss can be determined using the equations given in Section 2.1.1 for the conduction loss and switching loss of the diode.

FIGURE 2.2 THE EQUIVALENT CIRCUIT OF AN IGBT WHILE CONDUCTING. 2.1.3 MOSFET Power Loss A MOSFET is a bidirectional switch that is capable of blocking current and voltage in the forward direction. When it is on, it can be approximated as a resistive element (see Fig.2.3). Thus, it dissipates power as current is conducted through the device [24]. The resistive parameter is described as on-resistance (푟퐷S_on). 퐼퐷S_on is the drain current thought the MOSFET when it is on. The conduction loss is shown by:

1 푇푠 푃푐표푛_푀푂푆퐹퐸푇 = ∗ ∫ 푉퐷푆(푡) ∗ 푖퐷푆(푡)푑푡 푇푠 0 2 = 푟퐷푆_푂푁 ∗ 퐼푟푚푠_퐷푆 . (2.10) When its freewheeling diode conducts the third quadrant current, the third quadrant losses

(푃푐표푛_푀푂푆퐹퐸푇_3푄) are calculated using the reverse RMS current (퐼푟푚푠_퐷푆_푟푒푣) and on-resistance of the third quadrant (푟퐷푆_푂푁_3푄).

2 푃푐표푛_푀푂푆퐹퐸푇_3푄 = 푟퐷푆_푂푁_3푄 ∗ 퐼푟푚푠_퐷푆_푟푒푣. (2.11)

The other source of power loss is through switching losses. Switching losses come from the dynamic voltages and currents the MOSFETs must handle during the time it takes to turn on or off. The methods to calculate the switching loss of MOSFETs stay the same as those equations shown in Section 2.1.2. Besides the turn-on and turn-off switching loss, its intrinsic parasitic capacitance stores and then dissipates energy during each switching transition [25]. Switching loss ( 푃푠푤_푀푂푆퐹퐸푇_퐶푂푆푆 ) occurs due to the charging and discharging of the output capacitance. 퐸퐶푂푆푆 is the loss associated capacitance. 11

푃푠푤_푀푂푆퐹퐸푇_퐶푂푆푆 = 퐸퐶푂푆푆 ∗ 푓푠. (2.12)

FIGURE 2.3 THE EQUIVALENT CIRCUIT OF A MOSFET WHILE CONDUCTING. 2.2 Wide Bandgap Semiconductor A goal of this thesis is to improve the performance of the existing NBI power supply system. One way to achieve better performance is to reduce the energy loss in the system by using switching devices based on new semiconductor materials. The atomic structure of these new materials has several advantages, including the ability to operate at higher temperatures, realize lower on-resistance, and switch at faster speeds [16]. These factors are widely being applied to existing systems including converters for server applications [26], electric vehicles [27], grid-tied inverters [28], and other areas [29]. However, only recently have they started to receive attention as candidate for the power supply systems of a fusion reactor [16], [30], [31]. Two emerging WBG materials that show great promise for future SMPS designs are GaN and SiC. Like Si, they can be applied in many power electronic areas. Homo-epitaxial SiC devices are fabricated in a way that is analogous to Si in that an SiC epi-layer is formed on an SiC substrate [32]. This enables complex switching structures to be created using SiC, such MOSFETs, JFETs, and IGBTs. The strengths of SiC provide new opportunities for the application of SMPS.

One of the objectives of this thesis is to examine the power loss of the NBI power supply system. WBG based power devices are analyzed in Chapter 5 to verify that they can perform as well or better than Si in the NBI power supply system. The optimal choice of a semiconductor switching device will achieve higher efficiency during power generation.

2.2.1 Comparison of GaN, Si, SiC Table 2.1 shows the material properties of Si, SiC, and GaN. It can be seen that the bandgap of a WBG device is much higher than Si. This makes them have a higher critical electric field strength, which enables the channel length of the WBG transistors to be decreased. Since resistance is proportional to length [33], WBG technology results in a smaller on-resistance over comparably rated Si devices. Thus, the device exhibits a reduction in the conduction loss. Furthermore, since WBG devices have a higher saturation velocity and smaller parasitic capacitances, when compared to Si based devices, they have the capability to switch at a faster speed [34]. This results in a lower switching loss [18]. This benefit is suitable for NBI which requires the power supply to provide fast turn-on and turn-off voltage for sustaining the beam current [5]. Because the power loss is the sum of the conduction loss and switching loss, lower total loss can be achieved. Theoretically, applying WBG devices enables the power supply system to realize better efficiency under existing operating conditions.

TABLE 2.1 COMPARISON OF SI , SIC, AND GAN SEMICONDUCTOR PROPERTIES [16], [18] Materials Property Unit Si SiC GaN

Band Gap (Eg) eV 1.1 3.2 3.4 6 Critical Electrical Field (Ec) 10 V/cm 0.3 3 3.5 6 Electron Saturation Velocity (Vsat) 10 cm/sec 10 22 25 -3 10 -9 -10 Intrinsic Carrier Concentration (ni) cm 1.5x10 8.2x10 1.0x10 Thermal Conductivity (λ) W/cm2K 1.5 3.3 1.3

Homo-epitaxial WBG vertical devices will dominate in switching applications above 600 V, especially for higher power applications, such as the NBI system. Among WBG devices, SiC and GaN materials have become readily available through vendors such as Mouser and Digikey. Comparing the states-of-the-art of SiC and GaN technologies, SiC power devices are closer to a mature stage and have better commercialization when compared with GaN devices [18]. The breakdown voltage of SiC is much higher than GaN, which leads to more SiC based applications

in high power motor drives, power supplies, and high voltage transmissions [18]. Generally, SiC devices are designed for a medium to high voltage (600 V – 1700 V) range with a higher device power rating, while GaN devices are more suitable for low to medium voltage (15 V – 600 V) applications [18], [35]. This is due in part to low breakdown voltage of GaN devices [16], [36]. Higher thermal conductivity enables SiC devices to dissipate the larger amounts of heat when operating at extremely high power levels. Additionally, the smaller intrinsic carrier concentration of SiC results in lower leakage current at higher junction temperatures, which enables SiC power devices to operate at higher junction temperatures [16], [37]. The extended operating temperature range makes it possible for SiC based power devices to operate with smaller heatsinks over similarly rated Si components. Therefore, SiC devices can be used as a substitution for the Si IGBTs that are currently employed in HPPS of NBI, and other auxiliary heating power supply systems [32].

2.2.2 SiC Power Devices With the rapid advancement of the semiconductor industry, SiC power devices have evolved into a viable alternative for Si devices in many power electronic applications. The circuit designer has many device types to select from. Each one has advantages and disadvantages.

The SiC Junction Field Effect transistor (JFET) was commercialized earlier than the other SiC switches because it is relatively easy to implement [38]. It can operate in either depletion or in enhancement mode [39]. In an "enhancement mode", voltage applied to the gate terminal increases the conductivity of the device. In a "depletion mode", voltage applied at the gate reduces the conductivity [24]. Therefore, voltage-controlled converters typically use normally off devices, while current source converters prefer normally on devices. In the contrast, the SiC BJT functions as current controlled switch [40]. It has no oxide layer and can operate at higher temperatures than the SiC MOSFET [38].

Even though SiC MOSFETs have large threshold voltage instability in reverse operation, the reverse recovery charge is lower than the corresponding Si devices [41]. Another main advantage of a MOSFET is that it does not require continuous driving current to maintain the conduction

state. IGBTs achieve lower on-resistance than MOSFETs by injecting minority carriers into the drift region, a phenomenon called conductivity modulation [42]. However, SiC MOSFETs generate no tail current, and the results in lower switching loss and faster operation than IGBTs [42].

Economic viability of SiC-based devices is one of the challenges because their price is about 3 to 5 times higher than Si devices [43]. Prices are so high because the size of SiC wafers is rather small. SiC manufacturers have steadily reduced the defects in the material while increasing the wafer size [44]. Even though it has limitations, the materials are finding more applications where the benefits of SiC technology can offer system advantages of lower on- resistance, less reverse recovery charge, higher switching frequency and higher temperature operation., which are significant enough to offset the increased device cost.

2.3 Summary of the Chapter SiC power devices are an attractive alternative for Si MOSFETs and Si IGBTs. Their benefits include higher efficiency, higher power density, and the ability to operate in more rugged environments. These strengths are suitable for the NBI because it requires a power supply system that provides high efficiency and reliability.

Chapter 3 Overview of Tokamak Reactors

A tokamak fusion reactor is a device that uses a powerful magnetic field to confine plasma in a toroidal shape. The tokamak is a magnetic confinement device that was developed to contain the hot plasma needed for producing controlled fusion power. This method of generating plasma is the leading candidate for a reactor for a fusion power plant. This chapter introduces the fusion reactor and discusses two implementations of a tokamak.

3.1 Nuclear Fusion Reactions

Fusion produces enormous amounts of energy. It is the source of power for the sun and other stars. To harvest energy from fusion reactions, two light atoms combine to form a heavier atom and release a neutron in the process. To achieve this, two types of hydrogen isotopes are used in the fusion reaction: Deuterium (D), which is extracted from water, and Tritium (T). To accomplish a useful fusion reaction rate, the hydrogen needs to be heated to an extremely high temperature that can be as high as 100 million degrees Celsius [45]. At these high temperatures, the atomic nuclei separate from electrons, and appear in a plasma state. During the fusion process, a helium molecule, a neutron and a significant amount of heat energy (17.6 MeV) are released per reaction (see Fig. 3.1). This is estimated to be 10 million times more energy than compared to a chemical reaction.

2 3 4 1 1퐷 + 1푇 → 2퐻1 + 0푛 + 17.6 푀푒푉. (2.1)

FIGURE 3.1 DIAGRAM OF THE FUSION REACTION PROCESS.

To achieve these extreme temperatures, there must be a powerful heating mechanism. The incredibly hot plasma produced by this heating process is also extremely fragile. The plasma may become cool through contact with other surface materials of the container. Therefore, minimizing the thermal losses is also an objective.

Today’s fusion researchers emphasize two main experimental approaches to confine the plasma. One way to prevent the particles from escaping is by using a magnetic confinement system. This is achieved by creating a magnetic cage with strong magnets to contain the hot plasma [46]. A plasma controlled by strong magnetic fields is heated and confined in a circular bottle known as a tokamak. The other approach is inertial confinement. In this approach, laser phase or particle beams are used to compress a small amount of fuel to extremely high densities [47]. It is felt that the tokamak is the most promising method for creating a fusion based power plant.

The difficulty with these methods is in the process used to heat the fuel to a high enough temperature and confine it in a sufficiently long period for fusion to occur. In a commercial fusion power station, the heat is generated by the neutrons and slowed down by a blanket of denser material so as to provide electricity for residential consumption.

3.2 The Tokamaks The tokamak was designed in 1951 by Soviet physicists Andrei Sakharov [48] and soon adopted by researchers around the world. Tokamaks operate within limited parameters, outside of which a sudden losses of energy confinement can occur, causing major thermal and mechanical stresses to the structure and surfaces. Nevertheless, the tokamak is a mechanical device based on powerful magnetic parameters. It is one of the most promising and well- developed designs for fusion energy [30]. Research on several tokamaks around the world is ongoing including the Joint European Torus (JET), the Mega Amp Spherical Tokamak (MAST) in the UK and the Tokamak Fusion Test Reactor (TFTR) in the USA. When it is complete, the International Thermonuclear Experimental Reactor (ITER) project is currently the largest physical model in the world. The case study for this work is EAST. However, the intent is to

explore alternative designs for future reactors, such as CFETR. This is another tokamak based fusion energy system that is projected to be larger than ITER, and will be completed in 2030 [4], [5].

The main tokamak components and functions are shown in the Fig. 3.2.

FIGURE 3.2 DIAGRAM OF TOKAMAK MAGNETIC FIELDS [49]2. The plasma is contained in a vacuum vessel at some typical orders of magnitude density. The vacuum is maintained by external pumps. The plasma is heated by driving a current through it. Since plasma consists of free electrons and ions, the hot plasma is contained by a magnetic field that produces an external force to prevent it from escaping from the machine walls [46]. The combination of two sets of magnetic coils, known as toroidal and poloidal field coils, creates a field in both vertical and horizontal directions, acting as a magnetic ‘cage' to hold and shape the plasma [50]. The toroidal field is created by coils outside the plasma and the poloidal field is created by running a toroidal current through the plasma. Therefore, it is a kind of a low-β device where β is the ratio of the plasma pressure and the magnetic energy density [51].

2 **Figure from M. Kikuchi, "A Review of Fusion and Tokamak Research Towards Steady-State Operation: A JAEA Contribution", Energies, vol. 3, no. 11, pp. 1741-1789, 2010.

Large power supplies are used to generate the magnetic fields and plasma currents. Plasma current is induced by a transformer with the central magnetic coil acting as the primary winding and the plasma as the secondary winding. The heating provided by the plasma current, known as ohmic heating, supplies up to a third of the temperature required to make fusion occur. Therefore, additional plasma heating is necessary to reach the necessary temperatures. This heating is provided by the neutral beam injectors and three other systems which will be detailed described in next section. In this process, neutral hydrogen atoms are injected at high speed into the plasma and then, ionized and trapped by the magnetic field. As they are slowed down, they transfer their energy to the plasma and heat it.

For the tokamak to operate, the HPPS converts the grid’s high voltage AC to levels necessary to support the magnetic coils and the heating system of the reactor [5]. Since the tokamak needs to work in extreme conditions with high voltage and high current, it is a challenging task to develop the power electronic systems for this application. Extending the operational reliability would accelerate the research continuing on the next generation tokamak. An improved controller design and physical understanding has attracted researchers to this field to improve the performance of the existing power supply system by implementing new materials or proposing new structures for it [50].

3.2.1 The EAST Tokamak

EAST is a national research and engineering project that aimed to test the proposed technologies of fusion power. Construct started on EAST in 1998 and concluded around 2005 [5]. It is the first tokamak in the world to utilize superconductive coils. Because of this, EAST serves as a testbed for technologies proposed for the ITER project. This reactor, which is locate at China’s Institute of Physical Science in Hefei, is reported to have produced hydrogen plasma at 50 million degrees Centigrade and held it for 102 seconds.

The main design for EAST’s power supply system is shown in Fig. 3.3. It can be considered as five separate electrical loads. First, EAST has 16 sets of superconducting TF coils, which are connected in series to balance the toroidal magnetic field [5]. Second, it also has another 14 sets

of superconducting PF coils, which produce the poloidal magnetic field to initiate plasma current and control its position and shape [5]. Third, the Lower Hybrid Current Drive (LHCD), Ion Cyclotron Range of Frequencies (ICRF), Electron Cyclotron Resonance Heating (ECRH) and NBI are used to heat and drive the 1 MA plasma current. Each part of this system has its own power supply, but the structures are almost identical. Fourth, the Central Solenoid (CS) coils are required to decrease the voltage for breakdown and maintain, as well as ohmic heat the plasma. Finally, the Inner Vessel (IV) coil is supplied by AC/DC-DC/AC power supply. The rated parameter and different requirement of each of the coils are presented in Table 3.1. The ratings for the associated power supplies can be inferred from this table.

3 FIGURE 3.3 DIAGRAM OF THE DESIGN FOR THE EAST POWER SUPPLY SYSTEM [5] .

3 **Figure Citation with authorization from P. Fu, "Power supply system of EAST superconducting tokamak," 2010 5th IEEE Conference on Industrial Electronics and Applications, Taichung, 2010, pp. 457-462.

TABLE 3.1

PARAMETERS OF DIFFERENT COILS IN EAST [5], [52] Coil Design Solution Rated Others Parameter TF 12-pulse converter 30V/16kA Magnetic field: 3.5T PF 12-pair of coils, 4-quadrant 15kA/1kV Total power: 200MVA CS 12-pair of coils, 4-quadrant 15kA/1kV Total power: 200MVA IV 24 set of H bridge IGBT 800V/±5kA Heating system AC/DC converter, IGBT chopper 100kV Total power: 60MVA

Thyristors are used in the PF and TF because they are the only device available to operating under these extreme conditions [5]. Nonlinear currents created by thyristors in the grid are applied in the HPPS. Recently, WBG based switching devices have become available with ratings suitable for some of the HPPS required for EAST’s operation [16]. In particular, the voltage and current capabilities of SiC MOSFETs are suitable for Pulsed Power Electrical Network (PPEN). However, existing WBG devices cannot accommodate the power levels of the PF and TF supplies [16]. During operation, engineers at EAST have to ensure that the tokamak operates safely and reliably in order to interface with the existing AC grid and satisfy relevant high voltage grid codes [53]. One of the requirements is to reduce the presence of reactive power, harmonics, and unbalanced components. Therefore, RPC and HF are indispensable [16]. Active rectification [16] and power factor correction [16] may also be used to improve the HPPS and minimize the size of the RPC and HF system [30], [54]. WBG components provide extra advantages in this application as well.

3.2.2 The CFETR Tokamak ITER is an international nuclear fusion research and engineering megaproject, which will be the world's largest magnetic confinement plasma physics experiment. It will be the largest in terms of power, grid voltage and energy capacity [4]. CFETR is the first reactor to be based on ITER-like superconducting coils. It is under design by the China National Integration Design 21

Group, which the parameters can be seen from Table. 3.2. The main goal of CFETR is to generate fusion power at 50-200 MW with a duty cycle around 0.3-0.5 [52], [55]. The concept of engineering design includes a magnet system, vacuum vessel system, and maintenance system as well as other components [55]. The rated parameters of CFETR are similar to ITER. Most of the operation parameters of EAST provide a good reference of the CFETR design work [55].

TABLE 3.2 PARAMETER OF ITER AND CFETR POWER SUPPLY SYSTEM [4], [55] Parameter CFETR ITER Plasma Current 8.5/10 MA 15 MA Central magnetic field 4.5/5.0 T 5.3 T TF coils 16 18 TF stored energy 35.8 GJ 41 GJ

The AC power system consists of independent subsystems: PPEN that supplies the pulsed loads and a Steady State Electrical Network (SSEN) that feeds the steady loads [56]. Like other tokamak fusion reactors, TF coils, the PF coils, the CS coils and ohmic heating system are the main power supply needed in system [55]. Several AC/DC converters are required to meet the different voltage and current ratings. Another important system in ITER is the RPC and HF system, which is used to improve the stability of AC Power System. Load compensation and voltage support are two functions of RPC and HF system [54]. Voltage support is required to reduce voltage fluctuation at the grid terminal. Load compensation involves balancing the real power, minimizing the reactive power, and eliminating harmonic current components [54].

Chapter 4 NBI Power Supply

NBI power supply is one of the auxiliary heating systems used in the tokamak. The chapter provides an overview of the background and operation principles of the NBI power supply system. This knowledge will be utilized in Chapter 5 when the power loss analysis is conducted on the PSM of the NBI power supply system.

4.1 Background

The ability to heat the plasma to temperatures high enough for fusion reaction is critical in magnetic confinement fusion research. In a future fusion reactor, practice requires that most of the heating should come from the slowing down of the charged fusion products.

A tokamak cannot just rely on ohmic heating to reach the ignition temperature [57]. It needs effective auxiliary heating methods to achieve safe and efficient steady-state operation. Therefore, the ability to heat the plasma to a high enough temperature is important in magnetic confinement fusion research. The auxiliary heating system is comprised of four different systems, namely: LHCD, ECRH, ICRH and NBI. Their circuit topologies are nearly identical. Each part of this system has its own power supply. The PSMs in each of the auxiliary heat power supply systems all have the same configuration (see Fig. 4.1). The NBI system works by injecting energetic ions from an external source, as an accelerator [58]. The neutron emission spectrum from neutral beam-heated plasmas of EAST is investigated with the first principle of the fast deuteron energy distribution.

To characterize the beam plasma interaction within EAST, it is important to diagnose the injected neutral beams as well as possible. The diagnostic technique applied to the neutral beam injector is limited due to the high power density, and because some common techniques, such as calorimetry, electricity, and mass spectrometry, require stopping most or all of the beam and cannot be utilized during actual injection operation.

4.2 Power Supply System As an important part of the power supply systems, NBI requires that the power supply can provide fast turn-on and turn-off voltages for sustaining the beam current. A l00 kV / 100 A HVPS is needed to the power the NBI [5]. Table 4.1 presents the principal parameters of NBI power supply and the main configuration is shown in the Fig. 4.1. There are two isolated 10 kV AC oil-transformers with three sets of windings. Each secondary set of windings is connected to one of the four dry-type transformers. Each dry-type transformer has 26 secondary windings that are used to supply AC voltage to the PSM. In total there are 104 PSM [58]. The insulation voltage of the oil transformers is 145 kVDC. A dry-type transformer cannot insulate against this high level of voltage without suffering a penalty in physical size. Therefore, realizing the insulation of the voltage over 35 kV is the main objective of oil transformer [5]. The input to the PSM is 900 VAC, which is obtained from one secondary windings of dry-type transformer. The output voltage of the PSM is 1,150 VDC. The outputs of all the PSMs are connected in series to obtain a high voltage. Therefore, the output current is limited to 100 A, and the highest voltage is around 110 kVDC on load. For the unloaded case, the maximum on should be around 145 kVDC. Each PSM shares the same configuration and has an individual module control system [5], [58].

TABLE 4.1

THE PARAMETERS OF THE NBI POWER SUPPLY [5]

Rated voltage 110 kV Rated current 100 A Rising time 50 μs Switching off time 5 μs Adjusts precision 1 % Stability precision 1 % Efficiency 96 % Ripple peak to peak 2 %

FIGURE 4.1 NBI POWER SUPPLY SYSTEM AND ONE OF THE POWER SUPPLY MODULES [58].

Fig. 4.1 also shows a detailed configuration of one submodule. The PSM module is the main part in the NBI power supply system. Each module has the same topological configuration which consists of a soft start section, silicon diode rectifier, DC filter, IGBT chopper and control system. 25

[5], [58]. The individual control circuit is responsible for measuring all necessary signals, controlling the soft start switch and IGBT, and monitoring of the communication. The soft start connection protects the capacitance from an over-voltage and prevents the DC circuit from experiencing an overvoltage when the AC supply switches on. The diode based rectifier is the main circuit of the PSM. When the IGBT power modular switches on, this PSM will be in series with the whole system. Otherwise, it will be disconnected. The IGBT chopper is also a protection device when a power supply or load output is in a fault condition [5]. The snubber, consisting of an inductor and free-wheeling diode, is used to limit the over current in the output fault or load fault. The SCR paralleled to filter capacitor performs as a protection when the submodule at fault. Each submodule has an individual module control system by DSP. The controller can measure all possible signals containing soft start switch performance, digital signals and thyristor connection [5].

4.3 Summary of the Chapter The chapter provides an introduction of NBI power supply system. Operation parameters of HPPS of NBI and the detailed configuration of one PSM structure have been included. This knowledge will be important principles that can aid with understanding the research presented in the following chapters.

Chapter 5 NBI Power Loss Analysis

The system’s power loss is calculated by using the switching losses and conduction losses of power devices. The models presented in Chapter 2 are employed for this analysis. Next, the passive rectifiers are replaced with active rectifiers to examine any benefits that may be obtained. Prior to both of these tasks, the principles of power flow in AC systems are reviewed and background information for active rectification is given. 5.1 Active Rectification

The original design of NBI is based on a passive rectifier, which is not an ideal process. It results in distortion of the input current drawn from the power line. The character of these input currents creates many problems. It produces a large spectrum of harmonic signals that may disturb the whole system [59]. As a result, there is a need to apply active rectification in NBI PSM to meet the demand of better harmonic performance.

Replacing a diode with actively controlled switches, such as Si IGBTs or SiC MOSFETs, is the key for achieving active rectification [60]. This chapter explores replacing the diodes with controllable devices, such as 1700 V/300 A Si IGBT with 1700 V/225 A SiC MOSFETs. The disadvantage of the diode is the power loss that occurs during conduction from the forward voltage drop. This drop also reduces the available supply voltage, which is sometimes important when the AC voltage at the input is in the lower range of its limits [61]. Pass transistors, which are actively driven by the PWM control, are evaluated as replacements for the existing diodes. Especially for SiC MOSFETs, they maintain a low on-resistance when conducting, thus resulting in the lower forward voltage drop and power loss across the transistor. The fast turn-off switching behavior also prevents reverse current from reaching a damaging level. In this way, a better performance can be achieved unlike a discrete diode with a recovery time measured in hundreds of nanoseconds [61]. Furthermore, the passive rectifier produces a large spectrum of harmonic signals that may disturb the whole system. Then, the active rectification properly enables efficiency level to be improved although at the expense of additional complexity [62]. A fast Fourier transform (FFT) algorithm computes the discrete Fourier transform (DFT) of 27

a sequence. In the PSIM simulation, Fourier analysis has been applied to convert a signal from its time domain to a representation in the frequency domain. The time domain results are shown in Fig. 5.1, and the frequency domain results are shown in Fig. 5.2. As what can be seen from this comparison, input phase current from a diode based the passive rectifier has higher discrete amplitude and more harmonics components compared with the results obtained by an active rectifier. Furthermore, the amount of Total Harmonic Distortion (THD) is 36 % higher for the passive rectifier, which also implements it has much more harmonic distortion.

FIGURE 5.1 COMPARISON OF INPUT CURRENT OF PASSIVE AND ACTIVE RECTIFIER IN TIME DOMAIN.

FIGURE 5.2 COMPARISON OF INPUT CURRENT OF PASSIVE AND ACTIVE RECTIFIER IN FREQUENCY DOMAIN. 5.2 PSM Physical Calculation Prior to the power loss analysis, principles of power flow in AC systems need to be introduced. Active power is basically a function of dissipative circuit elements. Usually resistors are dissipative in an electronic circuit. The reactive power is a function of reactive elements in the circuit, usually inductors and capacitors. Reactive power is assumed to be zero in this case study. Since apparent power (푆푝푢 ) is basically function of total circuit impedance, it is the combination of real power (푃3휑) and reactive power (푄3휑). Then, the relationship of apparent power, active power and reactive power can be derived as the following equations shown.

푆3휑 = 푃3휑 + 푄3휑푗. (5.1)

In a balanced, three-phase system, it is possible to perform the analysis using a simplified single-phase circuit like the one shown in Fig. 5.3. Here, the input phase current (퐼퐼푁) flows from the grid side to rectifier circuit through an inductance (L). ∝ is the phase angle of the grid AC voltage (푉퐴퐶) relative to the single phase voltage (푉퐴). Apparent power of three phase is three times the product of the current and voltage of the single phase circuit. In this way, 푃3휑 and

푄3휑can be derived as shown:

FIGURE 5.3 THE CIRCUIT OF SINGLE PHASE.

∗ |푉퐴퐶|∠−∝−|푉퐴|∠0 푆3휑 = 3푉퐴(퐼퐼푁) = 3|푉퐴| ∗ . (5.2) −푗푋퐿

3|푉퐴|(|푉퐴퐶|푐표푠∝−|푉퐴|) 푄3휑 = . (5.3) 푋퐿

3|푉퐴퐶||푉퐴|푠푖푛∝ 푃3휑 = . (5.4) 푋퐿

Based on the equations, the grid voltage and its phase angle can be used to control the real and reactive power in the system. That is the way this thesis achieve zero reactive power. It uses the grid parameters to realize conditions that minimize the reactive power in the system. Detailed power loss calculations of implementing different switching devices are provided in the following sections. The chapter will discuss the completed power loss estimates for existing devices as well as the Si IGBT and SiC MOSFETs. The rectifier and power module in Fig. 5.4 will be implemented with different switching devices and an active rectifier.

FIGURE 5.4 SCHEMATIC OF ONE POWER SUPPLY MODULE USED IN NBI.

5.2.1 Rectifier Power Loss A. Existing NBI system power loss

The existed NBI system uses a diode based passive rectifier and an IGBT power module. For the rectifier shown in the Fig. 5.4, the conduction losses (푃푐표푛_푆푖_퐷푖) are calculated for each of the six diodes (where k=1, 2, … 6 for each diode) where 푟퐷0 is on-resistance, and 푉퐷0 is the forward voltage the diode:

2 푃푐표푛_푆푖_퐷푘 = 푉퐷0 ∗ 퐼푎푣푔_Dk + 푟퐷0 ∗ 퐼푟푚푠_퐷푘. (5.5)

In this analysis, it is assumed that the load current 퐼표푢푡 is constant in time. As we can see from Fig. 5.5, the current through each diode conducts one-third of the whole period. Therefore, the diode current waveforms have the same average value [63] and its RMS value can be shown as:

1 퐼 = 퐼 . (5.6) 푎푣푔_퐷푘 3 표푢푡

1 퐼 = √ 퐼2 . (5.7) 푟푚푠_퐷푘 3 표푢푡

FIGURE 5.5 THE CURRENT WAVEFORM THROUGH DEVICE IN DIODE RECTIFIER.

The switching loss (푃푠푤_푆푖_퐷푘) for one diode is given as:

푃푠푤_푆푖_퐷푘 = 퐸푅푅 ∗ 푓푠, (5.8)

where 퐸푅푅 is the reverse recovery loss in the diode. For a diode based rectifier, the switching frequency (푓푠) is the same as the line frequency (푓푐). The detailed properties of diode in the existing NBI power supply are given in the Table 5.1.

The total power loss of the diode rectifier (PD_REC) is the summation of the conduction losses and the switching times the number of devices (k):

푃퐷_푅퐸퐶 = k ∗ (푃푐표푛_푆푖_퐷푘 + 푃푠푤_푆푖_퐷푘). (5.9) TABLE 5.1

RATED PARAMETERS FOR DIODE (DDB6U215N18)

푉퐷0 푟퐷0 퐸푅푅 푓푠 1.61 V 1.6 mΩ 50 Hz

For each PSM in NBI, the total power loss can be obtained by combining both the power loss of active rectifier and power module. Table 5.2 presents the existing NBI system power loss.

Since the reverse recovery energy cannot be estimated by the datasheet, 퐸푅푅 = 36 푚퐽 is taken as the worst case. This value was obtained from another diode’s datasheet. It was used because the two diodes have similar ratings. It can be seen that the switching loss is small with such low frequency. Therefore, the exact value of 푃푠푤_푆푖_퐷푘 does not significantly influence the power loss estimate.

TABLE 5.2

RESULTS OF POWER LOSS CALCULATIONS FOR THE SI DIODE RECTIFIER

푃푐표푛_푆푖_퐷푘 푃푠푤_푆푖_퐷푘 푃퐷_푅퐸퐶 354 W 1.8 W 355.8 W

B. Si IGBT based rectifier

To create an active rectifier, the diodes are replaced with active switches. Si IGBTs are examined in this section. IGBTs cannot be implement alone, they need a diode connected in antiparallel to deal with the freewheeling current. Therefore, calculating the conduction losses for an active rectifier necessitates calculating the conduction losses for both the IGBTs and the

diodes. For each of the IGBTs ( 푃푐표푛_푆푖_푄푘 ), and their antiparallel diode ( 푃푐표푛_푆푖_퐷푘 ), the conduction losses for the IGBT are given as:

2 푃푐표푛_푆푖_푄푘 = 푉퐶퐸0 ∗ 퐼푎푣푔_푄푘 + 푟퐶퐸0 ∗ 퐼푟푚푠_푄푘, (5.10)

where rCE0 is on-resistance of the IGBT, 푉퐶퐸0 is the saturation voltage of IGBT, 퐼푎푣푔_푄푘 is the average value of the current through the IGBT and 퐼푟푚푠_푄푘 is its RMS value, and where k =1, 2, … 6 for each IGBT. The conduction losses for the diode will use the equation form the previous Section. 5.2.1-A.

The instantaneous current through the diode and IGBT is the product of the input current and the duty cycle. Different legs with upper and lower switching device have various duty cycles (푑휃) and input voltages (see Fig. 5.7). The phase angle (휑) of the input current relative to the duty cycle can be seen in Fig. 5.6. The input current is the reference current, and the peak value (퐼푝푘) is based on its RMS value (퐼푅푀푆). The modulation index (푚푎) of the PWM control is the ratio of the peak of the reference waveform to the peak of the carrier waveform.

휑

FIGURE 5.6 THE PHASE ANGLE OF INPUT CURRENT AND DUTY CYCLE.

FIGURE 5.7 THE PWM CONTROL TECHNIQUE.

퐼푝푘 = √2 ∗ 퐼푅푀푆 (5.11)

푖(휃) = −퐼푝푘 ∗ 푠푖푛(휃 − 휑) (5.12) 1 푑휃 = ∗ (1 + 푚 ∗ 푠푖푛휃) (5.13) 2 푎 Therefore, the current through the device can be represented by multiplying 푖(휃) and d휃.

For example, the current through different switching devices (Q1, Q2, D1, D1) can be seen in the Fig. 5.8.

FIGURE 5.8 THE CURRENT WAVEFORM THROUGH EACH SWITCHING DEVICE IN IGBT RECTIFIER. 34

The equation of the average current (퐼푎푣푔_퐷1) and the RMS current (퐼푟푚푠_퐷1) are as follows [64]: 1 휋+휑 1 퐼 = ∫ ∗ (1 + 푚 ∗ 푠푖푛휃) ∗ (퐼 ∗ 푠푖푛(휃 − 휑)) 푑휃 . 푎푣푔_퐷1 2휋 휑 2 푎 푝푘 (5.14) 1 1 퐼푝푘 + 퐼푝푘푚푎cos (휑)휋 = 4 2 휋

1 휋+휑 1 퐼 = √ ∫ ( ∗ (1 + 푚 ∗ 푠푖푛휃) (퐼 ∗ 푠푖푛(휃 − 휑)))2푑휃 . 푟푚푠_퐷1 2휋 휑 2 푎 푝푘 (5.15)

2 2 1 2 √2 √ 퐼푝푘 푚푎cos (휑) + 퐼푝푘 휋 = 3 4 2 휋 The method to calculate the current through the IGBT should be similar to the diode current. Since both the IGBT and its antiparallel diode conduct for one half of the period, the conduction period should be different. Take Q1 for instance, the equation of the average current (퐼푎푣푔_푄1) and the RMS current (퐼푟푚푠_푄1) are. 1 2휋+휑 1 퐼 = ∫ ∗ (1 + 푚 ∗ 푠푖푛휃) ∗ (−퐼 ∗ 푠푖푛(휃 − 휑)) 푑휃. 푎푣푔_푄1 2휋 휋+휑 2 푎 푝푘 (5.16) 1 1 퐼푝푘 − 퐼푝푘푚푎cos (휑)휋 = 4 2 휋

1 2휋+휑 1 퐼 = √ ∫ ( ∗ (1 + 푚 ∗ 푠푖푛휃) (−퐼 ∗ 푠푖푛(휃 − 휑)))2푑휃. 푟푚푠_푄1 2휋 휋+휑 2 푎 푝푘 (5.17)

2 2 1 2 √2 √ 퐼푝푘 푚푎 cos(휑) − 퐼푝푘 휋 = 3 4 2 휋

퐸푇푂푇 is the energy loss of IGBT to the on-and-off switching transition. Based on 퐸푇푂푇, the switching loss of IGBT (푃푠푤_푆푖_푄푘) is given as:

푃푠푤_푆푖_푄푘 = 퐸푇푂푇 ∗ 푓푠. (5.18)

The total power loss (푃푆푖_푅퐸퐶) for the rectifier is the summation of the individual conduction losses and switching losses for all of the switches:

푃푆푖_푅퐸퐶 = k ∗ (푃푐표푛_푆푖_푄푘 + 푃푐표푛_푆푖_퐷푘 + 푃푠푤_푆푖_푄푘 + 푃푠푤_푆푖_퐷푘). (5.19)

The operating conditions and a long with the switching device currents are for the analysis of the IGBT devices (see Table 5.3). The device’s parameters are given in Table 5.4. It can be seen that both the average and RMS current through the freewheeling diode is greater than the current through the IGBT.

TABLE 5.3 THEORETICAL CURRENTS IN THE SI BASED ACTIVE RECTIFIER

퐼푟푚푠_퐷푘 퐼푎푣푔_퐷푘 퐼푟푚푠_푄푘 퐼푎푣푔_푄푘 푚푎 휑 166.31 A 83.17 A 125.42 A 49.44 A 1 71.1°

TABLE 5.4

PARAMETERS OF SI DEVICE (FF300R17KE4)

푉퐷0 푉퐶퐸0 푟퐷0 푟퐶퐸0 퐸푅푅 퐸푇푂푇 푓푠 1.50 V 1.45 V 2 mΩ 2.5 mΩ 42.5 mJ 80.1 mJ 1000 Hz

Table 5.5 presents the results of the power loss calculations for the Si based active rectifier. It can be seen that the IGBT rectifier approach results in higher power loss when compared to the results of passive rectification, which are shown in Table 5.2. However, the diode based passive rectifier does generate lower harmonics components as can be seen from the comparison in Fig. 5.1, which is a big concern needed to be dealt with.

TABLE 5.5

RESULTS OF POWER LOSS CALCULATIONS FOR THE SI BASED ACTIVE RECTIFIER

푃푐표푛_푆푖_푄푘 푃푐표푛_푆푖_퐷푘 푃푠푤_푆푖_푄푘 푃푠푤_푆푖_퐷푘 푃푆푖_푅퐸퐶 713.22 W 1246.4 W 480.6 W 255 W 2695.22 W

C. SiC MOSFET based rectifier

This case extends the work of the previous section by applying WBG devices in active rectification. The conduction losses (푃푐표푛_푆푖퐶_푄푘) of the SiC MOSFET are estimated by using the on-resistance (푟퐷푆_푂푁 ) of the switching devices and calculating the RMS current (퐼푟푚푠_푄푘 ) through it (Where k=1,2,…..6 for each MOSFET). The third quadrant losses (푃푐표푛_푆푖퐶_3푄) are calculated using the reverse RMS current (퐼푟푚푠_푄푘_푟푒푣) and on-resistance of the third quadrant

(푟퐷푆_푂푁_3푄). In this instance, SiC MOSFETs are functioning as synchronous rectifiers. As shown in Fig. 5.9, each switching device has its conduction interval. The currents through the reverse direction are shown as negative values. However, the calculation used the absolute value of magnitude.

FIGURE 5.9 THE CURRENT WAVEFORM THROUGH EACH SWITCHING DEVICE IN MOSFET RECTIFIER.

2 푃푐표푛_푆푖퐶_푄푘 = 푟퐷푆_푂푁 ∗ 퐼푟푚푠_푄푘. (5.20)

2 푃푐표푛_푆푖퐶_3푄 = 푟퐷푆_푂푁_3푄 ∗ 퐼푟푚푠_푄푘_푟푒푣. (5.21)

The method to calculate 퐼푟푚푠_푄푘 and 퐼푟푚푠_푄푘_푟푒푣 are the same as (5.14) to (5.17). Since it is assumed that the currents are the same in both cases, the values in Table. 5.3 are used for the analysis below. The total switching loss for SiC MOSFET is the sum of one on-and-off switching transition loss (퐸푇푂푇) and the energy loss associated output capacitance (퐸퐶푂푆푆): 37

푃푠푤_푆푖퐶_푄푘 = 퐸푇푂푇 ∗ 푓푠 (5.22)

푃푠푤_푆푖퐶_퐶푂푆푆 = 퐸퐶푂푆푆 ∗ 푓푠, (5.23) where 푓푠 is the switching frequency.

The total power loss for SiC MOSFET is calculated as the summation of the individual and three-quadrant conduction losses and switching losses for all of the switches:

푃푆푖퐶_푆퐸퐶 = k ∗ (푃푐표푛_푆푖퐶_푄푘 + 푃푐표푛_푆푖퐶_3푄 + 푃푠푤_푆푖퐶_푄푘 + 푃푠푤_푆푖퐶_퐶푂푆푆). (5.24)

The operating conditions of MOSFET and its parameters are shown in Table 5.6. Table 5.7 presents the calculation result.

TABLE 5.6

PARAMETERS OF SIC DEVICE (CAS300M17BM2)

푟퐷푆_푂푁 푟퐷푆_푂푁_3푄 퐸퐶푂푆푆 퐸푇푂푇 푓푠 퐼푟푚푠_푄푖 푚푎 휑 10.5 mΩ 6.5 mΩ 1.9 mJ 16 mJ 1000 Hz 124.05 A 1 71.1°

TABLE 5.7

RESULTS OF POWER LOSS CALCULATIONS FOR THE SIC BASED ACTIVE RECTIFIER

푃푐표푛_푆푖퐶_푄푖 푃푐표푛_푆푖퐶_3푄 푃푠푤_푆푖퐶_푄 푃푠푤_푆푖퐶_퐶푂푆푆 푃푆푖퐶_푆퐸퐶 969.48 W 1087.7 W 96 W 11.4 W 2164.58 W

D. Summary of the rectifier power loss analysis.

Based on the calculation results, the active rectification creates the higher power loss compared with the passive rectification. However, diode rectifier does generate harmonics which must be dealt with in order to comply with grid standards. Replacing a diode with actively controlled switches, as Si IGBT, enables rectification to be performed with minimal harmonic content. Thus, a tradeoff exists between the power loss and harmonic content. Moreover, if one compares the Si based active rectifier to the SiC active rectifier, the power loss drops significantly showing an improvement of over 19 % lower for SiC implementation (see Table

5.8).

TABLE 5.8

COMPARISON OF RECTIFIER CALCULATION RESULTS

푃퐷_푅퐸퐶 푃푆푖_푅퐸퐶 푃푆푖퐶_푆퐸퐶 354 W 2695.22 W 2164.58 W

One needs to consider the entire tokomak system and determine whether or not the tradeoff between lower harmonic content is worth the additional power loss. Future research should explore the power loss associated with the RPC and HF system to quantify the merit of this approach.

5.2.2 Power Module Power Loss A. Si IGBT based power module

In order to get the total power loss of one submodule, the losses in the power module also needed to be considered. The power loss analysis is mainly calculated using the switching devices in the power module shown in Fig. 5.1. The conduction loss of Q1 is estimated by the diode’s forward voltage (푉퐹푊퐷_푆푖), the load current (퐼표푢푡), and the duty cycle.

푉 푑 = 표푢푡 (5.25) 푁푂푚표푑∗푉퐷퐶

푃푐표푛_푆푖_푄1 = 푉퐹푊퐷_푆푖 ∗ 퐼표푢푡 ∗ (1 − 푑). (5.26)

Here, 푉표푢푡 is the output voltage of each PSM and 푁푂푚표푑 is the total number of PSM. For the switching device Q2, the conduction loss is calculated by the saturation voltage (푉퐶퐸_푆퐴푇),

퐼표푢푡, and its duty cycle.

푃푐표푛_푆푖_푄2 = 푉퐶퐸_푆퐴푇 ∗ 퐼표푢푡 ∗ 푑. (5.27)

The switching loss of Q1 and Q2 are given as:

푃푠푤_푆푖_푄1 = 퐸푅푅 ∗ 푓푠 (5.28)

푃푠푤_푆푖_푄2 = 퐸푇푂푇 ∗ 푓푠. (5.29)

Then, the total power loss can be calculated as:

푃푆푖_푃푀 = 푃푐표푛_푆푖_푄1 + 푃푐표푛_푆푖_푄2 + 푃푠푤_푆푖_푄1 + 푃푠푤_푆푖_푄2. (5.30)

The operating conditions and the device’s parameters for the analysis of the IGBT devices are shown in Table 5.9.

TABLE 5.9 DEVICE PARAMETERS OF SI POWER MODULE (FF300R17KE4)

푉퐶퐸 푉퐹푊퐷_푆푖 퐸푅푅 퐸푇푂푇 푁푂푚표푑 푉퐷퐶 푉표푢푡 퐼표푢푡 d 푓푠 1.35 V 1.28 V 32 mJ 70 mJ 104 1150 V 110 kV 100 A 0.92 1000 Hz

Table 5.10 provides the results of the calculations. It can be seen that the Si power module experiences large conduction losses due to the high forward voltage of the device. Since Si has a slower switching speed compared to SiC devices, it also suffers from high switching loss.

TABLE 5.10

RESULTS OF POWER LOSS CALCULATIONS FOR THE SI BASED POWER MODULE

푃푐표푛_푆푖_푄1 푃푐표푛_푆푖_푄2 푃푠푤_푆푖_푄1 푃푠푤_푆푖_푄2 푃푆푖_푃푀 10.27 W 124.16 W 32 W 70 W 236.43 W

B. SiC MOSFET based power module

For the SiC MOSFETs in the power module, the conduction loss of Q1 is estimated by the

load current (퐼표푢푡), the on-resistance (푅퐷푆_푂푁) and the duty cycle (d).

2 푃푐표푛_푆푖퐶_푄 = 퐼표푢푡 ∗ 푅퐷푆_푂푁 ∗ 푑. (5.31)

Since the SiC MOSFET is operating as a synchronous rectifier, the power loss is calculated using the third-quadrant on resistance.

2 푃푐표푛_푆푖퐶_3푄 = 푅퐷푆_3푄 ∗ 퐼표푢푡 ∗ (1 − 푑). (5.32)

The switching loss for SiC MOSFET are obtained using the following relationships:

푃푠푤_푆푖퐶_푄 = 퐸푇푂푇 ∗ 푓푠 (5.33)

푃푠푤_푆푖퐶_퐶푂푆푆 = 퐸퐶푂푆푆 ∗ 푓푠. (5.34)

The total power loss for SiC MOSFET is given as:

푃푆푖퐶_푃푀 = 푃푐표푛_푆푖퐶_푄 + 푃푐표푛_푆푖퐶_3푄 + 푃푠푤_푆푖퐶_푄 + 푃푠푤_푆푖퐶_퐶푂푆푆. (5.35)

The operating parameters for the analysis of the MOSFET devices are shown in Table 5.11. The calculated results are presented in Table 5.12.

TABLE 5.11 DEVICE PARAMETER OF SIC POWER MODULE (CAS300M17BM2)

푉퐷푆_3푄 푉퐹푊퐷_푆푖퐶 퐸퐶푂푆푆 퐸푇푂푇 푅퐷푆_푂푁 푅퐷푆_3푄 0.9 V 1.2 V 1.9 mJ 13 mJ 10 mΩ 6.5 mΩ

TABLE 5.12

RESULTS OF POWER LOSS CALCULATIONS FOR THE SIC BASED POWER MODULE

푃푐표푛_푆푖퐶_푄 푃푐표푛_푆푖퐶_3푄 푃푠푤_푆푖퐶_푄 푃푠푤_푆푖퐶_퐶푂푆푆 푃푆푖퐶_푃푀

91.9 W 5.22 W 13 W 1.9 W 112.09 W

C. Results of the power module power loss analysis

With regards to the power module, the SiC MOSFETs can offer a significant advantage over the Si IGBTs that are currently being used. Comparing the total power loss for the power module, it enjoys a 55 % reduction in power loss by applying SiC devices (see Table 5.13).

TABLE 5.13

COMPARISON OF THE POWER MODULE’S POWER LOSSES FOR SI AND SIC VERSION

푃푆푖_푃푀 푃푆푖퐶_푃푀 236.43 W 112.09 W

5.3 PSIM Simulation Results Since calculations are the theoretical method used to prove the proposed improvement, they need to be verified. Simulation results can serve to validate these results in cases where building a prototype is prohibitive. The experimental system is simulated using PSIM software, and the related simulation parameters for NBI simulation are shown in Table 5.14. A higher capacitor is applied to drive the IGBT and MOSFET, which makes the lower output ripple. Each specific simulation schematic can be found separately in the corresponding sections.

TABLE 5.14

NBI PARAMETERS USED FOR SIMULATION System parameter Value

Rated DC voltage 1150 V Rated DC current 100 A Test system Rated AC line voltage 900 V Fundamental frequency 60 Hz AC system inductance 1.06 mH Submodule DC capacitance 0.3 mF AC system inductance (SiC) 1.25 mH DC capacitance (Si/SiC) 5 mF

5.3.1 Existing NBI PSM The existed NBI system (see Fig. 5.1) consists of diode rectifier and IGBT power module. The same structure was built in simulation (see Fig. 5.10). The input side is a three-phase voltage source, while the output side is a DC load resistance and an energy storage capacitor. PSIM is

used to measure the input and output currents and the input and output voltages. The results are shown together in Fig. 5.11. The data from these plots was exported to calculate the power loss for this case. Then, the active power was calculated for both sides.

FIGURE 5.10 SIMULATION SCHEMATIC OF NBI PSM.

FIGURE 5.11 SIMULATION RESULTS FOR THE EXISTING NBI PSM’S INPUTS AND OUTPUTS.

The total power loss of one PSM can be calculated by the differences between the input

power and the output power. The results are shown in Table 5.15. It can be seen that the PSM achieves a high efficiency.

TABLE 5.15

SIMULATION RESULT OF NBI PSM Input power Output power PSM power loss Efficiency

114,293.10 W 113,774.34 W 518.8 W 99.5%

5.3.2 Si IGBT PSM The simulation schematic of one IGBT PSM is shown in Fig. 5.12. It can be seen that the passive rectifier has been replaced by the IGBT active rectifier, while the power module remains the same structure. The detailed parameter settings in the text file can be found in Fig. 5.13. In the simulation, another diode was place in antiparallel with one IGBT so that the current through the free-wheeling diode can be measured in this method. Fig. 5.14 shows the output and input results of one Si IGBT based PSM by PSIM simulation. The data from these plots was exported to calculate the power loss for the case.

FIGURE 5.12 SIMULATION SCHEMATIC OF IGBT PSM. 44

FIGURE 5.13 SETTING PARAMETERS IN IGBT FILE.

FIGURE 5.14 SIMULATION RESULT OF SI PSM’S OUTPUTS AND INPUTS.

The total power loss of this PSM can also be calculated by the difference between the powers on the input side and the output side. The results are shown in Table 5.16. One can see that the results are not as good as the results of the passive rectifier given in Section 5.3.1.

TABLE 5.16

SIMULATION RESULT OF SI IGBT PSM Input power Output power PSM power loss Efficiency

117,509.60 W 114,745.32 W 2764.3 W 97.6%

5.3.3 SiC MOSFET PSM The image of the simulation schematic for the SiC MOSFET PSM is shown in Fig. 5.15. It can be seen that MOSFETs have been applied in both the rectifier and power module. Fig. 5.16 shows simulations results for the output and input results of one SiC MOSFET based PSM. The data from these plots was exported to calculate the power loss for the case.

FIGURE 5.15 SIMULATION SCHEMATIC OF MOSFET PSM.

FIGURE 5.16 SIMULATION RESULT OF SIC PSM’S OUTPUTS AND INPUTS.

The total power loss of one PSM can be calculated by the difference of input power and the output power. The results are shown in Table 5.17. Regarding the power dissipation, the results of active rectification do not show an improvement compared to the results of passive rectification. However, as for the whole PSM of NBI, SiC devices can be substituted for Si IGBTs to achieve the higher efficiency. The simulation results verify the theoretical analysis in Section 5.2.

TABLE 5.17

SIMULATION RESULT OF SIC MOSFET PSM Input power Output power PSM power loss Efficiency

116,351.0 W 114,251.72 W 2099.3 W 98.1%

5.4 Summary of the Chapter This chapter calculated the total power loss of the PSM in the NBI power supply. It looked at the losses generated by rectifier and power module power individually. It proposed changing the passive rectifier to an active rectifier. Then it examined the benefits accrued by using SiC MOSFETs in the rectifier and the power module. Simulation results for the whole PSM were used to validate this analysis.

Table 5.18 shows the results for the existing NBI PSM structure and the first proposed method, where just the power module is replaced with an SiC MOSFET. In this case, the power losses improved by over 20 % through utilizing a WBG device. Even though the passive rectifier approach offers the best performance in power dissipation, lower THD is achieved and minimal harmonic is generated by using active rectification. Especially, if SiC MOSFETs are used. In the case of an active rectifier, the power losses 21 % lower for the SiC MOSFETs over the Si IGBTs based active rectifier. Theoretically, the SiC approach does increase the efficiency of each PSM to some extent. If the whole NBI system is considered, then the total power loss of circuit is greatly reduced by using the SiC based approach. The analysis presented in Section 5.2 was validated by the results of simulation given in Section 5.3. For the simulation results, SiC MOSFETs also achieved 24 % lower power loss over the Si based implementation of NBI’s PSM. The differences between the calculated results and the simulated results can be attributed to the fact that the switching losses are not modelled in PSIM. To summarize the results of this chapter, active rectification with SiC MOSFETs can provide lower harmonic content compared to the existing passive rectification, and lower power loss compared to Si IGBT devices, which makes them a possible for substitution in PSM of the NBI power supply.

TABLE 5.18

POWER LOSS COMPARISON FOR ONE NBI PSM

푷푹푬푪 (W) 푷푷푴 (W) 푷푻푶푻 (W) Existing NBI 355.8 (Si Diode) 236.43 (Si IGBT) 592.23 Proposed system 1 355.8 (Si Diode) 112.09 (SiC MOSFET) 467.89 Proposed system 2 2695.22 (Si IGBT) 236.43 (Si IGBT) 2931.65 Proposed system 3 2164.58 (SiC MOSFET) 112.09 (SiC MOSFET) 2296.67

Chapter 6: Improved NBI Topology based on MMC

The MMC topology offers several advantages over the existing NBI power supply system methods. For example, distribution system within the tokamak may be simplified and PSM structure of the MMC topology has fewer components. Additionally, it can act as both an active rectifier and an active power filter. The inclusion of WBG in MMC can also provide benefits regarding the improvement in the efficiency. Prior to analyzing those benefits, the background for Voltage Source Converters (VSC) and MMC topology is given.

6.1 Voltage Source Converter Technology

The VSC transmission technology had captured a significant proportion of the HVDC market [11]. Embedding VSC into the power grid has the potential to improve the power grid’s reliability and efficiency [19]. First, flexible control of the power flow meets the demand of effective management and security requirements. Furthermore, rapid and independent response of active and reactive power modulation can improve the dynamic response under disturbances. Though all power electronic converters generate some harmonics, HVDC systems based on VSC normally apply high-frequency PWM to improve the harmonic distortion of the converter. Some HVDC systems have been built with three level converters, but current installations of VSC technology are based on variants of a multi-level converter, most commonly the MMC. Multilevel converters have the advantage that they allow harmonic filtering equipment to be reduced or eliminated. Lower THD, higher operating voltage levels, higher effective switching frequency and better distribution of power losses are the main benefits of the multilevel configurations [65], [66]. MMC converters permit the switching frequency to be reduced by an order of magnitude, while simultaneously achieving better harmonic performance than is possible with two-level and three-level converters [67]. For the MMC based transmission, the AC/DC converter arms act as a controllable voltage source with a high number of possible voltage steps [19]. As the next section explores, the MMC shows its further benefits based on the VSC topology.

6.2 Modular Multilevel Converter The MMC was invented by A. Lesnicar and R. Marquardt as a proposed development from the Cascaded Half-Bridge Converter [63]. The MMC is one of the next generation of multilevel converters that are being applied in HVDC transmissions. The advantages include both a modular structure and independent control of the voltage levels. Worldwide, there are many large HVDC systems using MMC technology [68] due to high voltage capabilities. Based on the concept of this topology, it can generate variable sources of voltage and power by applying a different number of identical, but individually controllable submodules. Increasing the number of submodules can expand the level of voltages, which result in reduced harmonic distortion within the system [69]. Additionally, flexible results of MMC can be achieved by utilizing different parameters in the submodule, such as capacitor size and voltage level. The MMC is based on a cascading connection of bidirectional submodules in a phase leg. The phase legs can be arranged in three different types of configurations. A double-star configuration, which is shown in Fig. 6.1, a star-configured MMC which is shown in Fig. 6.2, and the delta-configured MMC shown in Fig. 6.3. To operate as an active rectifier, the double- star configuration is used. The star and delta configurations have no common DC output, so they cannot achieve power conversion. However, by connecting them in parallel with the system grid, these two configurations are suitable for energy storage systems [70] The submodules in each arm can be a Half-Bridge (HB) or full bridge structure; the HB topology is shown in Fig. 6.1. Each leg also contains a pair of current sharing inductors along with the cascaded connection of the bidirectional submodules (see in Fig. 6.1) [70]. The input current of each phase flows through each multivalve. The capacitor voltage of each submodule is balanced around the designed voltage [71].

FIGURE 6.1 THE SCHEMATIC OF DOUBLE-STAR MMC AND ONE HALF BRIDGE SUBMODULE.

FIGURE 6.2 THE SCHEMATIC OF STAR MMC. FIGURE 6.3 THE SCHEMATIC OF DELTA MMC.

The submodule is the core of MMC system. The number of submodules required for a given system is calculated using the desired output voltage for both the system and each individual submodule. Each HB submodule is a two-terminal component that can be switched between states; one with a full module voltage (푉푐) and a second state with zero module voltage. Both

states are capable of bidirectional current flow [19]. As shown in Fig. 6.1, the HB submodule consists of a DC storage capacitor and two IGBTs (Q1 and Q2) and their corresponding freewheeling diodes (D1 and D2). By using the two switches in a submodule, the capacitor can either be connected in series with the output or be bypassed. This allows the valve to synthesize a stepped voltage with very low levels of harmonic distortion. When only one IBGT is switched on, either the IGBT or its freewheeling diode will conduct. Whether it is the IGBT or the diode that conducts depends on the direction of the current. For this reason, it makes sense to define the submodule as on to indicate that either the IGBT Q1 or the diode D1 is conducting [72]. The four possible switching states of each submodule circuit can be defined [19], [72]-[74]. 1) State 1: Q1 is on and Q2, D1, D2, are off. The submodule output voltage equals the capacitor voltage

(Vc), and the capacitor discharges when the multivalve current (see Fig. 6.4A) flows in the negative direction 2) State 2: D2 is on, Q1, Q2, D1, are off. The arm has negative polarity (see Fig. 6.4 B). The capacitor will charge constantly and this submodule is bypassed. 3) State 3: D1 is on, Q2, Q1, D2, are off. As the multivalve side current flows in the positive direction

(see Fig. 6.4 C), the capacitor is charging and the voltage of capacitor is designed voltage (Vc) 4) State 4: Q2 is on, Q1, D1, D2, are off and the arm has positive polarity (see Fig. 6.4 D). The submodule is bypassed, and the capacitor voltage is constant.

퐼푣푎푙푣푒 퐼푣푎푙푣푒

(A) (B)

퐼푣푎푙푣푒 퐼푣푎푙푣푒

FIGURE. 6.4 (A), (B) NEGATIVE AND (C), (D) POSITIVE CURRENT FLOW INSIDE A SUBMODULE. In this research, 1700 V Si IGBT and SiC MOSFET are applied in the HB submodule. The normal voltage behavior can reach as high as around 1100 V per submodule. Since the voltage for each submodule is the same, the number of inserted sub-modules can be derived by the output voltage.

푉 110푘푉 N = 퐷퐶 = ≈ 100. (6.1) 푉퐷퐶−푠푚 1100푉 In this case, the system needs 100 submodules for each arm. There will need to be around 600 submodules in the three-phase system. The number of submodules can be decreased by increasing the voltage on each submodule, but to do this depends on the maximum voltage that each switching device can withstand.

6.3 MMC Benefits to NBI Power Supply System

The following sections show the improvements of MMC topology, which can be utilized in

the NBI power supply system. Each benefit has been analyzed in detail in a separate section. Regarding to the whole tokamak, further discussions of improvement are also mentioned.

6.3.1 Simplified Distribution System For MMC configuration, it can directly rectify AC voltage to the DC voltage. This enables the distribution structure of the tokamak to be simplified. First, all but one of the transformers used in the current NBI structure can be eliminated. In terms of the current tokamak, it applies 100 kV grid voltage to supply the whole system, while the level of AC grid can be reduced to reach the desired output power level. Based on the following equation, 66 kV could be an option for MMC input voltage.

66푘푉∗ 2 푉 = 푉 ∗ 2.34 = √ ∗ 2.34 ≈ 126푘푉. (6.2) 퐷퐶 퐴퐶 √3 Thus, the advantages of the MMC’s performance, flexible structure and high efficiency on the tokamak infrastructure are highlighted [19]. 6.3.2 Simplified PSM Structure

Based on the previous analysis, the number of submodules increased significantly. However, in regards to the structure of each submodule (see Table 6.1), the traditional circuit has two switching devices, one capacitance and one rectifier with six switching devices. The MMC has just two transistors and one capacitor, which eliminates the rectifier. By applying the WBG devices, lower energy loss can also be accomplished. The analysis for this point is presented in Section 6.3.4.

TABLE 6.1

ONE SUBMODULE COMPARISON BETWEEN NBI AND MMC

SM Structure Switching device Capacitance Existed NBI 8 1 MMC 2 1

6.3.3 RPC and HF Based on the transmission requirements and objectives, MMC has many advantages that it can offer to a DC system. First, the DC voltage generated by the MMC is constant, which allows converters to operate continuously and reliably. Furthermore, this new structure does not require active power filters because when the MMC topology is used in HVDC converters, it minimizes harmonic content generated by the conversion process. The normal thyristor converters with high voltage change require extensive filtering methods at the AC grid. Therefore, accomplishing harmonic filtering in the MMC is a considerable benefit.

Due to the large amount of harmonic current generated by the other power supply systems during the tokamak operation and other harmonic components produced by the module’s capacitor voltage ripple, the whole distribution system may become polluted with noise. The power supply system can also fluctuate with multiple harmonic components [17]. Since the RPC and HF system for the tokamak is expensive and requires a lot space, accomplishing HF by using an NBI power supply based on the MMC topology would be a considerable benefit for a fusion reactor [75].

A load is considered non-linear if its impedance changes with the applied voltage, resulting in a series of positive and negative pulses [21]. These pulses can create harmonic currents in addition to the original current. Those non-sinusoidal currents contain harmonic currents that interact with the impedance of the power distribution system to create distortion [76]. Therefore, in the test simulation systems, a non-linear load is used to represent the harmonics that naturally occur in the tokamak system [21].

Fig. 6.5 is the proposed system. It consists of the series connection of an active filter and a LC passive damping filter that is connected in parallel with a non-linear load and a voltage source. The non-linear load is implemented using a passive rectifier. The purpose is to model the harmonics created by the thyristor based converters used for the TF and PF power supplies. The passive filter, which can be replaced with LCL filter [77], provides a low impedance for specific harmonic frequencies and power-factor correction of inductive load. The LC filter provides better harmonic reduction than the L filter, and it reduces the output ripple. A simple resistor is connected in series with LC filter to perform passive damping and solve the resonance problem [78]. The active power filter (APF) is connected with a passive filter at the point of common coupling (PCC) [79]. The MMC inverter is utilized to regulate the dc-link voltage and generate the active filter’s reference current [80].

FIGURE 6.5 CIRCUIT AND CONTROL DIAGRAM OF PROPOSED HARMONIC FILTER. The source harmonic current is calculated by applying three phase load currents and terminal

voltages into instantaneous p-q theory [81]. The Clarke transformation matrix transforms the currents and voltages from three to two-phase quantities:

1 1 1 − − 푋푎 푋훼 2 2 2 [ ] = √ [ ] ∗ [푋푏]. (6.3) 푋 3 √3 √3 훽 0 − 2 2 푋푐

The instantaneous real power and the instantaneous imaginary power are calculated as:

푝 푉α 푉β 푖 [ ] = [ ] ∗ [ α]. (6.4) 푞 −푉β 푉α 푖β

The reference filter current could be determined by applying reference power and compensatory current into equation (6.3) and inverse Clarke transformation:

푖cα 1 푉α 푉β −푝푐 [ ] = 2 2 [ ] [ ]. (6.5) 푖cβ 푉α +푉β −푉β 푉α −푞푐

1 0 퐼퐶푎 1 √3 2 − 푖cα [퐼퐶푏] = √ 2 2 [ ]. (6.6) 3 푖cβ 1 3 퐼퐶푐 − − √ [ 2 2 ]

There are various current controlled techniques for the active power filter, such as hysteresis current control (HCC) and phase disposition pulse width modulation (PD-PWM). The proposed system uses space vector PWM (SVPWM) generated by reference filter current, which can provide higher voltage with lower THD. The MMC based APF was prepared for the test simulation. The major experimental parameters are listed as Table 6.2. The simulation structure is shown in Fig. 6.6. As we mentioned above, the MMC is acts as APF and its output provides energy to a DC load. Fig. 6.7 shows the control system in the simulation.

TABLE 6.2

MMC HARMONIC FILTER PARAMETERS IN SIMULATION

System parameter Value Non-linear load Capacitance 2 mF Inductance 1.5 mH LC passive Inductance 0.1 mH damping filter Capacitance 10 mF Resistance 10 Ω

FIGURE 6.6. THE SIMULATION SCHEMATIC OF HARMONIC FILTER SYSTEM.

FIGURE 6.7. THE SIMULATION SCHEMATIC OF CONTROL SYSTEM FOR THE ACTIVE POWER FILTER.

Two sets of plots showing the experimental results are included in Fig. 6.8 and Fig. 6.9. 퐼퐼푁 is the input grid current, 퐼푛표푛_푙푖푛푒푎푟 is the current waveform containing harmonic contents, which is distorted by the non-linear load. 퐼푑푐 is the output current of DC load. 퐼푛표푛_푙푖푛푒푎푟 shows lots of harmonics, while the leading current of the passive filter flow through the MMC-APF presents the accurate compensation characteristics. Furthermore, MMC APF demonstrates its capability to prevent the grid current from becoming distorted. The FFT analysis confirms the reduction of harmonic content as a result of using the MMC-APF. Therefore, it demonstrates that the MMC can be used as a harmonic filter.

FIGURE 6.8. THE SIMULATION RESULTS OF CURRENT IN TIME DOMAIN.

FIGURE 6.9. THE SIMULATION RESULTS OF CURRENT IN FREQUENCY DOMAIN. 6.3.4 WBG Power Devices The inclusion of WBG power devices into an MMC also provides opportunities to modify the existing submodule construction [30], [82]. Using the same procedure outlined in Chapter 5, the SiC MOSFETs can be substituted for the Si IGBT that are used in each submodule. Then the power loss of MMC can be calculated by the summation of the conduction loss and the switching loss.

The advances of the MMC innovated the market for VSC-HVDC by greatly eliminating harmonics and dramatically decreasing the power losses, which can be accomplished by faster switching frequencies. The switching loss currently enjoys secondary significance in the MMC, 60

and the power loss is dominated by the conduction loss in the IGBT and diodes. The behavior of MMC is much more complicated than that of the two-level converter. Calculating the conduction losses and switching losses in one submodule is difficult, which makes it hard to obtain an exact analysis [67].

A reduction in switching frequency results in lower switching losses in the semiconductor device, so a modulation method which produces a low switching frequency is preferable. Generally, for power semiconductors there is a trade-off between switching losses and conduction losses [83], so if low switching frequency can be ensured, then a semiconductor manufacturer can design a semiconductor device with higher switching losses to bring down the conduction losses.

A. PSM Physical Calculation Since the current through every switching device is unique, the operating conditions of the Si IGBT need to be recalculated. They are shown in Table 6.3 and Table 6.4. In this case study, one submodule per leg is analyzed. Fig. 6.10 presents the upper multivalve current waveform through each switching device. The multivalve current contains the DC component; therefore, the positive and negative half cycle is not a symmetric sine wave [17]. The method to calculate

퐼푟푚푠_푄푘 and 퐼푟푚푠_푄푘_푟푒푣 are similar to (5.14) to (5.17), while the value of the reference current, modulation index, and duty cycle are different. This shown in Table 6.3. A PWM control technique can ensure stable performance of the back-to-back operation of switching devices in a submodule. Each submodule has a dedicated triangular carrier waveform with the same magnitude, but with a different phase shift [73].

TABLE 6.3

THEORETICAL CURRENTS IN THE SI BASED DEVICES

퐼푟푚푠_퐷푖 퐼푎푣푔_퐷푖 퐼푟푚푠_푄푖 퐼푎푣푔_푄푖 푚푎 휑 235.52 A 106.96 A 230.65 A 102.9 A 0.98 64.8°

TABLE 6.4

PARAMETERS OF SI DEVICE (FF300R17KE4)

푉퐷0 푉퐶퐸0 푟퐷0 푟퐶퐸0 퐸푅푅 퐸푇푂푇

Q1/Q2 1.7 V 2.5 mΩ 110 mJ

D1/D2 1.65 V 2 mΩ 25 mJ

FIGURE 6.10 THE CURRENT WAVEFORM THROUGH EACH SI BASED SWITCHING DEVICE.

Based on the methods and equations for the power loss calculations of the NBI supply given in Chapter 5, the results can be obtained for one power submodule in the MMC (see Table 6.5).

TABLE 6.5

POWER LOSS ESTIMATES FOR ONE SI PSM IN MMC

Device Pcon_Si_Ii Psw_Si_Ii Pcon_Si_Di Psw_Si_Di PSi_SM_TOT Si 615.86W 220W 574.84W 50W 1460.7W

The SiC devices properties are given in Table 6.6 and Table 6.7. The input current for WBG system can be smaller compared with the existing system.

TABLE 6.6

THEORETICAL CURRENTS IN THE SIC BASED DEVICES

퐼푟푚푠_퐷푖 퐼푎푣푔_퐷푖 퐼푟푚푠_푄푖 퐼푎푣푔_푄푖 104.65 A 53.25 A 74.65 A 28.59 A

TABLE 6.7

PARAMETERS OF SIC DEVICE (CAS300M17BM2)

푟퐷푆_푂푁 푟퐷푆_푂푁_3푄 퐸퐶푂푆푆 퐸푇푂푇

Q1/Q2 10 mΩ 7 mΩ 1.9 mJ 13 mJ

Based on the approaches and equations for the power loss calculations of the NBI supply given in Chapter 5, the result of SiC based PSM can be calculated and are shown in Table 6.8.

TABLE 6.8

POWER LOSS ESTIMATES FOR ONE SIC PSM IN MMC

Device Pcon_SiC Pcon_SiC_3Q Psw_SiC_COSS Psw_SiC_Q PSiC_SM_TOT SiC 111.4W 153.8W 3.8W 26W 295.19 W

It can be concluded that the SM power loss is reduced by over 78 % by applying SiC switching devices. If total 600 sub modules are applied as the substitution for the existing NBI system, the total power loss is around 177.12 kW. In regards to the whole system efficiency, 24 % improvement is possible by using WBG devices in the MMC, compared with the results obtained in the conclusion of Chapter 5.

B. PSIM Simulation Result Fig. 6.11 shows the schematic of MMC structure built for this analysis. In this case study, only one SM per arm is analyzed in the simulation. This is done because of the long time required to perform the simulation so a smaller system was used to reduced the computational burden on the hardware during simulation. The result is a total of 6 SMs in the simulation system. The structures of each SM are presented in Fig. 6.12. As can be seen in the submodule structure,

there is a diode connecting in series to IGBT or MOSFFET in the same direction of the current flow. It is used to conduct the first quadrant current. The same methods are constructed in the reverse direction to conduct the third quadrant current. The experimental system is simulated using PSIM software, and the related simulation parameters shown in Table 6.9.

FIGURE 6.11 THE SIMULATION SCHEMATIC OF MMC STRUCTURE.

(A) (B) FIGURE 6.12 THE SIMULATION SCHEMATIC OF (A) SI IGBT SM AND (B) SIC MOSFET SM.

TABLE 6.9

MMC PARAMETERS USED FOR SIMULATION

System parameter Value Rated DC voltage 1100 V Test system Rated DC current 100 A Rated AC line voltage 660 V Fundamental frequency 60 Hz AC system inductance (Si) 0.77 mH AC system inductance (SiC) 8 mH Submodule Arm inductance 3 mH SM capacitance 6.5 mF DC capacitance 1.8 mF

The simulation results are shown in Fig. 6.13 for the model displayed in Fig. 6.11. The top graph shows the output current and output voltage. It can be seen that the waveforms are quite stable. The bottom graphs show the input current and input voltage. There is no distortion in the

phase current of the grid.

FIGURE 6.13 THE SIMULATION RESULTS OF SI IGBT MMC. The total power loss of one PSM is calculated by the difference between the input power and the output power. The results are shown in Table 6.10. One can notice that this loss, which is for six modules, is close to six times the loss calculated for an individual module in Section 6.3.4(A).

TABLE 6.10

SIMULATION RESULT OF SI MMC

Input power Output power PSM power loss Efficiency

121,324.2 W 112,279.8 W 9044.3 W 92.5%

The simulation results of WBG structure are shown in Fig. 6.14 and Table 6.11. The top graph shows the waveform of output current and output voltage, and the bottom graphs show the waveform of input current and input voltage. There is no ripple shown in the DC output and no distortion in the phase current of the AC grid.

FIGURE 6.14 THE SIMULATION RESULTS OF SIC MOSFET MMC.

TABLE 6.11

SIMULATION RESULT OF SIC MMC

Input power Output power PSM power loss Efficiency

109,267.64 W 107,290.24 W 1977.4 W 98.1%

The simulation results of the MMC confirm that the power loss is significantly reduced by applying SiC switching devices. A comparison of MMC approach to the existing NBI approach shows the improvement with regards to one submodule. When we consider the whole system, MMC structure with SiC approach also offers a better performance with efficiency. However, MMC structure does have some limitations. Over 600 submodules needed to be built and this makes the system huge and complex. Economic viability of MMC circuit is another challenge because the price of SiC-based devices is high.

Chapter 7 7.1 Conclusion This thesis performed a theoretical study to evaluate several improvements for the NBI power supply that is currently used in the EAST reactor. The purpose of these changes was to improve the operation of the NBI power supply system and to provide additional benefits to the tokamak system as a whole. The latter came in the form of reduced harmonic content via an active filter. The improvements to the entire PSM in the exiting NBI power supply system came from applying active rectification and using WBG components. Next, the MMC topology was studied as a replacement to the existing multilevel structure. Finally, the inclusion of WBG power devices into this MMC NBI based power supply was evaluated.

First, the thesis applied controlled device, Si IGBTs and SiC MOSFETs, as substitutes for the passive rectifier currently used in the of PSM structure in NBI. A diode based rectifier is currently used, which produces significant harmonic contents. With the active rectifier, lower harmonic content means it is easier for the system to comply with grid codes. The value of THD showed over a 36 % reduction by this approach.

SiC devices provide a major benefit in system performance because lower conduction and switching power losses are realized over the current Si devices being used in the NBI system. Both the simulated analysis and the calculated analysis have indicated that the efficiency can be improved by 55 % using SiC MOSFETs instead of Si IGBTs in the power module of existing NBI PSM. In the analysis, the amount of efficiency was also calculated for applying SiC based devices in the whole PSM. This included both as an active rectifier and in the power module. This approach showed a 20 % improvement over the same approach that used Si. However, the losses were still higher than the case were the passive rectifier was used in conjunction with the SiC based power module. Therefore, a tradeoff exists between lower harmonic contents and lower power loss. Furthermore, it was shown that the SiC MOSFETs are a substitute for existing NBI power supply system over comparably rated Si devices.

Replacing the existing NBI structure with one based on an MMC topology showed 68

additional benefits for the tokamak distribution system. Despite the relatively complex structure with a total of 600 submodules, which may cause higher construction cost, the proposed MMC is still worth researching. The submodules have a simplified structure over the existing PSM as it includes only a power module and capacitor without any other rectifier in the system. A single MMC can eliminate all but one of the transformers that are used in the existing NBI supply system. Furthermore, applying SiC devices in the MMC structure makes the power loss 24 % lower compared to the whole NBI system, which means higher efficiency can be achieved with the inclusion of WBG materials in MMC structure. Additionally, this thesis provides a method of using the MMC to act as an active harmonic filter hybrid and has proved its possibility. Since the RPC & HF system enjoys big cost and high priority in tokamak, this approach provides a big benefit.

Though the NBI power supply system is the case study of this thesis, the analysis of the improvements can be further explored in other auxiliary heating systems of tokamak. Currently, as power electronics designers are developing future HPPS for the next generation of the fusion energy systems, especially for those larger fusion rectors like CFETR and DEMO systems, the benefits WBG semiconductor switching devices bring to these systems could be significant. Furthermore, the MMC structure is being deployed widely in HVDC transmission systems. This is another application area where WBG devices may be able to improve the performance of the hardware of HVDC.

7.2 Future Work In recent years, WBG semiconductors have attracted increasing attention in the area of power electronics. Mitigation of the high economic cost and limitations of WBG devices is one challenge that the future work could focus on. Furthermore, more analysis is needed on the long- term reliability of these devices. Before they can be applied in mission critical systems, like a tokamak or HVDC system, relevant failure mechanisms need to be fully qualified. As was mentioned in the conclusion, there is a tradeoff between better harmonic performance and higher power efficiency when applying the active rectifier. Since the future tokamak reactors have to

realize both stability and efficiency in the power supply system, this is another direction requiring further analysis.

MMC are widely used in high voltage applications due to its modularity and the capability of high voltage output. Based on this work, the utilization of a lower distribution voltage (i.e. moving from 110 kV to 66 kV) needs to be better understood. Despite many essential advantages of using MMCs in this system, some points still need clarification. For one, the complexity of control and its impact on the tokamak’s operation should be evaluated. Furthermore, advantages may be gained from optimization of the capacitors, the number of submodules, and their structure remains an important research task.

Submodules with full-bridge structure and an improved control scheme are another proposed area of research. This would enable fast dynamic and tight tolerance bands of the arm currents of the converter [84]. The full-bridge submodule allows the capacitor to be inserted in either polarity, which provides additional flexibility in controlling the converter and allows it to block the over current.

Optimizing the voltage levels of the submodules is another area to be investigated. More than 100 submodules were used for each arm and a large number of capacitors were also utilized. The voltage of each capacitor in the converter has to be controlled in order to keep the capacitor and the switches from experiencing an overvoltage. Thus, the capacitor voltage balance control and minimization of capacitor size will become a very hot issue in the studies of MMC [84]. Both hardware and software methods can be applied to balance the capacitor voltage [85].

Further investigations concerning optimization of the control scheme, waveforms and modulation schemes should also follow. The voltage magnitude can be adjusted by the modulation index of the PWM. Thus, it is possible to control both the delay angle and the voltage magnitude influencing the active power and reactive power [72]. Mainly, controller gains and limitations of set values have to be modified.

SVPWM techniques for MMC APF topologies had been mentioned in literature, while other modulation and harmonic stepped-waveform technique could be optimized. A new pulse position 70

technique can be explored for harmonic elimination. Then, harmonic filtering for a high voltage power converter in high frequency applications can be analyzed.

Finally, fault analysis also needs to be performed on the proposed MMC based NBI power supply. Mechanical switches or fuses can be considered. Electronic current limitation on the AC and DC side including overvoltage clamping and DC breaker functionality needs to be considered to stabilize the system [86].

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