A HYBRID FLYBACK LED DRIVER WITH UTILITY GRID AND SOLAR PV

INTERFACE

A Thesis

Presented to

The Graduate Faculty of The University of Akron

In Partial Fulfillment

of the Requirements for the Degree

Master of Science

Awab Ali

December, 2017

A HYBRID FLYBACK LED DRIVER WITH UTILITY GRID AND SOLAR PV

INTERFACE

Awab Ali

Thesis

Approved: Accepted:

______Advisor Interim Department Chair

Dr. Yilmaz Sozer Dr. Joan Carletta

______

Co-Advisor Dean of the College Dr. Jose A. De Abreu-Garcia Dr. Donald P. Visco Jr.

______Committee Member Dean of the Graduate School

Dr. Malik E. Elbuluk Dr. Chand Midha

______Date

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ABSTRACT

In renewable energy systems, maximum utilization of the available power is a desirable objective. In this thesis, a hybrid flyback converter with a Photovoltaic Panel

(PV) port, an AC grid port and a DC Load port is proposed. The converter has the capability to achieve two major objectives: to maintain sustainable operation for a load such as Light

Emitting Diodes (LED) lighting system, and to achieve maximum utilization of the solar

PV panel output. Conventionally, PV panel power is injected into the grid using a converter, and then imported back to support the LED lighting system using another separate converter. A single converter capable of handling bi-directional power flow could be used to reduce the power processing compared to a system that uses multiple power converters. The LED lighting system can have its power supplied primarily by the solar

PV. The balance of the power can be processed through the utility interactive port in both directions.

There are systems already available to achieve the proposed modes of operation for a higher power range. However, these systems are not cost effective for low power renewable energy based lighting systems, such as LED lighting. This thesis proposes a single stage power converter that can host multiple energy interface ports through a single flyback transformer.

The converter design procedure specifies the required conditions to achieve full functionality. The converter topology, operating principle, modes of operation and control

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structure are presented in this research. The operation of the proposed converter is verified through Matlab Simulink® simulations. An experimental prototype was designed and developed for a 120 W system using a 35 VDC solar PV, a 120 Vrms 60 Hz grid, and 24

VDC LED lights.

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DEDICATION

To my Family To my Friends To you

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ACKNOWLEDGMENT

I would like to show my profound gratitude to my advisor Dr. Yilmaz Sozer, for his courage in concurring any problem; in research and any other life aspects. His ability to find pearls in between sand is remarkable. Because of his insights all this came true.

I would like to acknowledge my committee members Dr. Jose A. De Abreu-Garcia and Dr.

Malik E. Elbuluk for their support and follow up.

Special thanks go to the family of the Electrical and Computer Engineering Department at

The University of Akron, their cooperation and patience are invaluable.

Finally, I am thankful for my family that supported me in all situations for better, for worse and for whatever is yet to come.

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TABLE OF CONTENTS

I...... 1

1.1 Energy Sources and Challenges ...... 1

1.2 Renewable Energy Sources ...... 2

1.3 Thesis Organization ...... 4

...... 5

2.1 Introduction ...... 5

2.2 Converter Topologies for the AC Grid ...... 5

2.3 LED Lights Supply ...... 11

2.4 AC Grid Interface to LEDs ...... 12

2.5 Research Motivation ...... 16

2.6 Conclusion ...... 23

...... 24

3.1 Introduction ...... 24

3.2 Topology and Principle of Operation ...... 24

3.3 Grid Only Mode ...... 25

3.4 LED Only Mode ...... 31

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3.5 Grid and LED Mode ...... 34

3.6 Grid Support Mode ...... 37

3.7 HFC Controller ...... 39

3.8 Filter Type Selection and Design ...... 40

3.9 Conclusion ...... 44

...... 45

4.1 Introduction ...... 45

4.2 Transformer and Inductor Design ...... 45

4.3 Power Device Selection ...... 52

4.4 Conditioning Circuit for Sensors ...... 55

4.5 Gate Driver Development ...... 58

4.6 Printed Circuit Board Layout ...... 59

4.7 Conclusion ...... 60

...... 61

5.1 Introduction ...... 61

5.2 Simulation Results ...... 61

5.3 Experimental Results ...... 73

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5.4 Conclusion ...... 80

...... 81

6.1 Conclusion ...... 81

6.2 Future Work ...... 81

REFERENCES ...... 83

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LIST OF FIGURES

Figure 2.1: Single phase H-bridge inverter circuit...... 6

Figure 2.2: (a) The bipolar PWM reference signal, carrier and output voltage waveforms. (b)The unipolar PWM reference signal, carrier and output voltage waveforms...... 7

Figure 2.3: Frequency spectrum of the output voltage of a bipolar PWM H-bridge at full modulation index...... 8

Figure 2.5: Phase shifted full bridge and full bridge inverter topology...... 9

Figure 2.4: Boost converter and full bridge topology...... 9

Figure 2.6: Three-level inverter with neutral point clamped (NPC) topology...... 10

Figure 2.7: Interleaved flyback converter with an unfolding H-bridge topology...... 11

Figure 2.8: Load requirements matching system...... 12

Figure 2.9: AC supply characteristics and LED load characteristics. (a), (b) and (c) show the per unit AC grid supply voltage, current and power respectively at unity power factor. (d), (e) and (f) show the per unit voltage, current and power of the LED load respectively...... 13

Figure 2.10: General block diagram of a grid-connected LED driver...... 14

Figure 2.11: General block diagram of a grid-connected LED driver with a DC bus filter capacitor...... 14

Figure 2.12: AC and DC power waveforms...... 14

Figure 2.13: Solar panel with a DC/DC converter to supply a LED Load...... 17

Figure 2.14: Daily power profile of a PV panel and a LED lighting system...... 17

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Figure 2.15: PV, grid, battery and LED structure proposed in [16]...... 18

Figure 2.16: Conceptual realization of the proposed converter...... 18

Figure 2.17: Initial realization of the proposed converter...... 19

Figure 2.18: Inductor or flyback based converters modes of operation. (a) Continuous Conduction Mode CCM. (b) Boundary Conduction Mode (BCM) (c) Discontinuous Conduction Mode DCM...... 20

Figure 2.19: Conventional grid tied flyback inverter...... 21

Figure 2.20: Grid tied flyback inverter proposed in[19] with reactive power capability. 21

Figure 2.21: Proposed topology for power decoupling in [23]...... 22

Figure 2.22: Proposed power factor correction topology in [17]...... 23

Figure 2.23: Proposed HFC topology...... 23

Figure 3.1: Grid only mode involved ports...... 25

Figure 3.2: Grid only mode steps of operation, (a) Step1, (b) Step2, Positive half cycle,

(c) Step2, Negative half cycle...... 27

Figure 3.3: Simplified magnetization inductance current in grid only mode operation. .. 28

Figure 3.4: PWM generation scheme for grid only mode...... 28

Figure 3.5: Generic waveforms of the current in the HFC for a 60 Hz cycle in grid only mode...... 29

Figure 3.6: LED only mode operation...... 31

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Figure 3.7: LED only mode steps of operation, (a) Step1, (b) Step2...... 32

Figure 3.8: Generic waveforms of the current in the HFC for a 60 Hz cycle in LED only mode...... 33

Figure 3.9: Grid and LED mode steps of operation, (a) Step1, (b) Step2 and (c) Step3. . 35

Figure 3.10: Simplified magnetization inductance current in grid and LED mode steps of operation...... 36

Figure 3.11: Generic waveforms of the current in the HFC for a 60 Hz cycle in grid and LED mode...... 36

Figure 3.12: Simplified magnetization inductance current in grid support mode steps of operation...... 37

Figure 3.13: Generic waveforms of the current in the HFC for a 60 Hz cycle in grid support mode...... 38

Figure 3.14: The proposed controller of the HFC...... 39

Figure 3.15. The HFC and its controller...... 40

Figure 3.16: The objective of the filter to filter (a) HFC current 푖푆퐴퐶 to (b) the fundamental current 푖퐺푟푖푑...... 41

Figure 3.17: Block diagram of the filter interface between the grid and the HFC...... 42

Figure 3.18: CL filter transfer function block diagram as a function of inverter current 푖푆퐴퐶 and grid voltage 푉푎푐...... 42

Figure 3.19: Block diagram of the filter interface between the grid and the HFC with the grid as 푍푔...... 42

Figure 3.20: CL filter transfer function block diagram as a function of inverter current 푖푆퐴퐶 and grid equivalent impedance 푍푔 ...... 43

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Figure 3.21: Filter pole locus as 푍푔 the changes...... 43

Figure 4.1: Typical BH curve...... 46

Figure 4.2: Flux accumulation and discharge into and from the transformer core during switching states ...... 46

Figure 4.3: Different core shapes for transformers and inductors...... 48

Figure 4.4: Ferrite E-shaped core and its bobbin...... 49

Figure 4.5: Sample of a Litz wire ...... 51

Figure 4.6: Operation regions for MOSFETs and IGBTs in terms of voltage and power.53

Figure 4.7: Schematic of grid voltage sensing chain...... 55

Figure 4.8: Schematic of PV panel voltage sensing chain...... 56

Figure 4.9: Schematic of LED voltage sensing chain...... 57

Figure 4.10: Schematic of PV panel current sensing chain ...... 57

Figure 4.11: Printed circuit board final layout...... 59

Figure 5.1: Block diagram of the simulated system in Matlab Simulink®...... 61

Figure 5.2: Flowchart of P&O MPPT algorithm...... 63

Figure 5.3: I-V characteristic (top), P-V characteristic (bottom) of the modeled PV panel...... 63

Figure 5.4: MPPT algorithm command of the main switch S1 duty cycle...... 64

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Figure 5.5: PV panel voltage development during MPPT...... 65

Figure 5.6: PV Panel current development during MPPT...... 65

Figure 5.7: Power sharing and distribution between the different ports in grid only mode...... 65

Figure 5.8: LED port output voltage during grid only mode...... 66

Figure 5.9: Grid voltage and current during grid only mode...... 66

Figure 5.10: Power sharing between the different ports in LED only mode...... 67

Figure 5.11: LED voltage during LED only mode...... 68

Figure 5.12: Power sharing and distribution between the different ports in grid and LED mode...... 69

Figure 5.13: LED voltage during grid and LED mode...... 69

Figure 5.14: Grid voltage and current during grid and LED mode...... 69

Figure 5.15: Duty cycle control for AC side switches in grid and LED mode...... 70

Figure 5.16: MPPT duty cycle the main switch at grid support mode...... 71

Figure 5.17: Duty cycle control to the AC side switches at grid support mode...... 71

Figure 5.18: PV panel voltage development during MPPT in weak irradiance and grid support mode...... 71

Figure 5.19: PV panel current development during MPPT in weak irradiance and grid support mode...... 71

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Figure 5.20: Power sharing and and distribution between the different ports in grid support mode...... 72

Figure 5.21: LED voltage during grid support mode...... 72

Figure 5.22: Grid voltage and current during grid and LED mode...... 73

Figure 5.23: Experimental setup of the proposed HFC...... 74

Figure 5.24: TerraSAS photovoltaic simulator model ETS80...... 74

Figure 5.25: Variable resistor as LED...... 74

Figure 5.26: (a) Transformer currents in a half 60 Hz cycle in grid only mode. (b) Zoomed at the peak of (a)...... 75

Figure 5.27: Grid voltage and current of the experimental setup in grid only mode...... 76

Figure 5.28: Transformer currents in a half 60 Hz cycle in LED only mode...... 77

Figure 5.29: Zoom into the peaks of transformer currents of Fig. 5.28...... 77

Figure 5.30: Transformer currents in a half 60 Hz cycle in grid and LED mode...... 77

Figure 5.31: Zoom into the peaks of transformer currents of Fig. 5.30...... 78

Figure 5.32: Grid voltage and current of experimental setup in grid and LED mode...... 78

Figure 5.33: (a) Transformer currents in a half 60 Hz cycle in the grid support mode. (b) Zoomed at the peak of (a)...... 79

Figure 5.34: Grid voltage and current in grid support mode...... 80

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LIST OF TABLES

Table 2.1: Bipolar PWM switching operation...... 6

Table 2.2: Unipolar PWM switching criteria using high switching frequency...... 7

Table 2.3: Unipolar PWM switching criteria using low switching frequency in two switches...... 7

Table 3.1: PV panel and grid parameters...... 25

Table 4.1: Preferred operating conditions of some switches...... 52

Table 4.2: Final selected parameters for the HFC switches...... 54

Table 4.3: Gate driver chip capabilities and specification...... 58

Table 5.1: The HFC parameters...... 62

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INTRODUCTION

1.1 Energy Sources and Challenges

Throughout our daily usage of any tool or device, energy is what keeps them working. Energy availability assures the continuity of development, prosperity and comfort of human lifestyle [1]. With every discovery of a new energy source, new horizons are opened for exploration to enhance the human life. The impact of the discovery of a new source of energy can be realized through history, from extracting energy from wood and dry leaves to create fire, to utilizing animals in agriculture, to the use of fossil fuels during the industrial revolution.

Initially, the main concern was to secure energy from sources that were available in abundance, had high energy density and were easy to extract the energy from. Fossil fuels have been the dominant source of energy since the 1950s. However, the realization of the scarcity and the drainage of the world’s reserve of fossil fuels has sparked the search for alternate energy sources that are clean and environment friendly. These new sources of energy have to sustain the growing demand and be cost effective. Additionally, research efforts have focused on efficient energy production and utilization, as well as reducing, mitigating and controlling the effects of fossil fuels pollution via the usage of electric vehicles.

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1.2 Renewable Energy Sources

Renewable energy sources are clean and available in abundance. Different renewable energy sources like wind, solar and geothermal are currently being utilized.

However, renewable energy sources are intermittent in nature. The intermittence period may be yearly, seasonal, daily and/or hourly. For example, hydro dams are considered a yearly intermittent renewable energy source; that is, dam reservoirs are used to regulate that intermittence across the year to sustain a continuous water head and flow through the hydro generation units. Solar energy fluctuates on a yearly basis, from the summer’s long sunlight hours and high sun irradiance to the winter’s shorter sunlight hours and lower sun irradiance. On a daily basis, the availability of solar energy changes from zero at night to a maximum in the afternoon, with temporary fluctuations due to shading.

Solar energy is one of the major sources of energy utilized in places where the extension of the regular AC grid is not feasible. This infeasibility is not only based on the capital cost of the power system infrastructure but also on the availability of raw material of energy generation such as water in dams and fossil fuel in thermal generation. Solar panels can provide power to people living off the grid, and it can also help the grid to have extra surplus power that can be supplied to other customers. If grid connected customers reduced their consumption from the grid, and generated part of their energy demand locally, or even exported power to the grid, the coverage and resiliency of the grid would increase.

The usage of solar panels for electricity generation is being promoted by federal governments and local authorities. Legislations and bills are passed through law makers to pave the way for solar panels by adding incentives, like removing the net metering cap on

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solar power injected into the grid from homeowners. The development of solar energy will keep flourishing through such initiatives. The “California one million solar roofs initiative” is one of the projects promoting and facilitating solar renewable clean energy [2], with the objectives to achieve:

 Emission control and pollution reduction

 Job creation

 Grid resiliency

Solar panels will provide maximum power, that is, extract energy most efficiently from sunlight, only under specefic condition. The point where this happens is referred to as the Maximum Power Point (MPP). Loads require a constant supply, hence, if the solar panels are used to support a load, they have to work at exactly the same power level required by the load. If this condition is not met, it will lead to an inefficient use of the solar PV capabilities. Therefore, a converter is needed to utilize the unused energy from the PV. Since the solar PV will be unable to sustain the load under very low irradiance levels, another energy source is needed to support the load during these periods. To decrease costs and increase efficiency, the power processing through the electrical components has to be reduced.

The objective of this research is to develop a single unit that can interface to the grid, the solar PV and the load to reduce complexity and increase efficiency. The proposed system will utilize the PV panel efficiently and work at the MPP at all times while continuously supporting the load. To achieve this objective, a converter topology is proposed with a specific control strategy for optimum solar energy utilization. The

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proposed topology specifically targets indoor lighting applications, with LED lamps as the load.

1.3 Thesis Organization

The thesis is organized as follows:

Chapter 2 provides a general discussion of topics related to the realization of the topology and application of the converter. These topics include DC/AC conversion and the different circuits and topologies used to convert DC voltage to AC voltage or to inject AC current into AC grids, LED development and the industrial boost of LED lamps over the regular tungsten and fluorescent lamps, and the flyback converter topologies and their applications. In Chapter 3, the proposed hybrid flyback converter is detailed in terms of its operation modes, and the required operating conditions to achieve all of the projected features of the proposed topology. While Chapter 4 shows insight about the design of the converter, especially the hardware implementation, focusing on some of the critical parts of the proposed topology components like the transformer design, printed circuit board design and component selection. Chapter 5 contains the simulation and experimental results. Finally the conclusions and future work are detailed in Chapter 6.

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LITERATURE REVIEW

2.1 Introduction

The concern for the continuity of life for generations yet to come is one of the driving forces behind the use of renewable energy sources. Among renewable energy sources, solar PV is one of the promising technologies that has the potential for widespread usage. Breakthroughs in the solar panel market, in terms of manufacturing costs and ease of installation, have led to a high number of rooftop installations of solar panels.

The intermittent nature of solar power is a concern for customers and, as a result, interfacing with the AC grid or even establishing an AC supply is required in off grid installations. This Chapter presents Different circuits and topologies that can be used to interface incompatible systems of different characteristics. This will result in the proposed topology that can interface the PV, the LED and the grid together.

2.2 Converter Topologies for the AC Grid

The H-bridge topology is commonly used in many applications to generate an AC output [3], as shown in Fig. 2.1. There are two main fundamental switching methods used for H-bridge control. These are the bipolar Pulse Width Modulation (PWM) and the unipolar PWM [4].

5

푆 1 푆3

푉퐷퐶 푉푎 푣표푢푡 푉푏

푆4 푆2

Figure 2.1: Single phase H-bridge inverter circuit.

 The bipolar PWM

A low frequency reference signal (푉푟푒푓) is compared to a high frequency triangular carrier signal (푉푐푎푟) to generate the PWM commands. In this PWM method, all the power devices switch at the carrier frequency. The switching commands are assigned as per the criteria set in Table 2.1. Accordingly, the voltages applied to the output are either (+VDC) or (–VDC) at every instant of time. Fig. 2.2(a) shows the output voltage of the converter using a bipolar PWM.

Table 2.1: Bipolar PWM switching operation.

Case 푆1 푆2 푆3 푆4 푉푎 푉푏 푉푟푒푓 > 푉푐푎푟 ON ON OFF OFF 푉퐷퐶 0

푉푟푒푓 < 푉푐푎푟 OFF OFF ON ON 0 푉퐷퐶

 The Unipolar PWM

This PWM method utilizes an additional voltage level compared to the bipolar

PWM. In the unipolar method three voltage levels are applied to the output terminals. An additional voltage level of zero voltage is applied to the output terminal beside ±VDC. Fig.

2.2(b) presents the output voltage of the unipolar PWM. This output can be obtain by applying both the switching criteria in Table 2.2 and Table 2.3. The switching criteria in

Table 2.3 operates two switches at low frequency and the other two at low frequency.

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(a) (b) Figure 2.2: (a) The bipolar PWM reference signal, carrier and output voltage waveforms. (b)The unipolar PWM reference signal, carrier and output voltage waveforms. Table 2.2: Unipolar PWM switching criteria using high switching frequency.

Case 푆1 푆4 Case 푆2 푆3 푉푟푒푓 > 푉푐푎푟 ON OFF −푉푟푒푓 > 푉푐푎푟 OFF ON 푉푟푒푓 < 푉푐푎푟 OFF ON −푉푟푒푓 < 푉푐푎푟 ON OFF

Table 2.3: Unipolar PWM switching criteria using low switching frequency in two switches.

High Frequency Low Frequency Case 푆1 푆4 Case 푆2 푆3 푉푟푒푓 > 푉푐푎푟 ON OFF 푉푟푒푓 > 0 ON OFF 푉푟푒푓 < 푉푐푎푟 OFF ON 푉푟푒푓 < 0 OFF ON The output of the H-bridge inverter is a sequence of varying width square pulses of voltage, it can be expressed, using Fourier transformation,

∞

푣표푢푡(푡) = ∑ 푉푛 sin(푛휔표푡) (2.1) 푛=1 where 푛 is the harmonic number, 휔표 is the angular frequency of the fundamental component and 푉푛 is the Peak of the nth harmonic. Fig. 2.3 shows the frequency spectrum of the bipolar PWM output voltage, where the frequency modulation ratio (mf ),

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퐹푐푎푟 푚푓 = (2.2) 퐹푟푒푓

where (퐹푐푎푟) is the carrier frequency at1800 Hz and (퐹푟푒푓) is the fundamental component of the signal at 60 Hz or the reference signal frequency. The fundamental component at 퐹푟푒푓 is the main intended output of the system, so by utilizing the right filter in the output of the bridge, these other harmonics can be attenuated significantly leaving the system with a dominant fundamental signal at 퐹푟푒푓. In several applications, the nature of the connected load, like motors, can provide the required filtering effect on the signal because motors inductances act as filtering elements.

Grid interfacing circuits can be categorized as isolated and non-isolated topologies.

Most of the topologies are based on the H-bridge converter with or without additional converters or components through which the converter is able to push power to the grid or provide reactive power compensation. The exact condition that allows the H-Bridge to push power into the grid is that the DC bus has to be higher than the grid peak voltage.

2 3

1 푚 2푚 3푚 푓 푓 푓

Figure 2.3: Frequency spectrum of the output voltage of a bipolar PWM H-bridge at full modulation index.

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Dbypass

PV

AC

Figure 2.5: Boost converter and full bridge topology.

PV VDC AC

Figure 2.4: Phase shifted full bridge and full bridge inverter topology. In [5] several H-bridge based topologies are provided, starting with the boost converter and full bridge topology as shown in Fig. 2.4. Here, a boost converter acts as the initial stage of voltage amplification so that the DC bus is higher than the grid voltage.

Another advantage of this topology is the bypass diode that will support the DC bus directly when the PV panels’ voltage is higher than the DC bus voltage, significantly reducing the losses in the boost converter.

Phase shifted full bridge and full bridge inverter topologies are also used for grid interfacing as shown in Fig. 2.5. The transformer, in this topology, provides isolation between the grid and the PV panels, as well as a voltage gain so that the DC bus voltage

(VDC) in the DC Link capacitor is regulated at the right level. The efficiency of this topology is less than that of the boost converter and full bridge topology as per [5], but an optimized design can significantly improve the efficiency.

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PV

PV

AC

Figure 2.6: Three-level inverter with neutral point clamped (NPC) topology.

Multilevel inverters are one of the topologies used in supplying power to the grid with fault tolerant capabilities [6]. Multi-level inverters can be considered as cascaded H- bridges or as a more sophisticated converter with additional features. Fig. 2.6 shows a three-level inverter with the neutral point clamped (NPC)[7]. Multilevel inverters also need to have a total voltage greater than that of the grid voltage to be able to inject power into the grid.

The interleaving principle along with isolation capabilities can be combined as shown in the topology of Fig. 2.7, using an interleaved flyback converter and an unfolding

H-bridge. This topology provides low current ripple on the DC bus capacitor [5]. Various different grid connected topologies were introduced based on the topologies provided above, some modifications in the topologies or the control strategy were applied for specific reasons, like zero voltage switching.

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AC

PV string PV

Figure 2.7: Interleaved flyback converter with an unfolding H-bridge topology. 2.3 LED Lights Supply

One of the evolving appliances in industry is that of indoor and outdoor lighting.

This is especially true of LED based lighting which is becoming more prominent in the lighting market. Starting from the reports of light emission from the crystal detector by

Henry Round, in 1907, to the first practical LED that provides emission in the visible spectrum range by Nick Holonyak, in 1962, at GE’s advanced semiconductor laboratory

[8]; there are different milestones in the LED development timeline. The development expanded the capabilities of LED to be used in different applications; such as, computers and mainframes, indoor and outdoor lighting and auto headlights. This almost exponential development has essentially rendered incandescent light bulbs obsolete as they were completely phased out in 2014 [9]. LED development has also improved their efficiency and lifetime compared to conventional lamps.

While LEDs are DC loads that work and provide light at certain DC levels, the available electrical supplies come mainly as AC and DC supplies, albeit different specifications. The main AC supply comes from the utility grid. DC supply is obtained mainly from different sources such as solar, fuel cells, batteries or a rectified filtered AC input. 11

Matching System Supply LED

Figure 2.8: Load requirements matching system.

The electrical supply source needs to be compatible with the LED requirements.

For example, an 18 V LED would require that the DC voltage of the battery, fuel cell or the solar panel be in that vicinity, moreover be able to supply the required current. The output of an AC supply source would need to be rectified. Furthermore, if the rectified voltage is incompatible with the LED voltage, then a matching system is needed. A matching system is needed to interface the different electrical supply sources to the specifications and requirements of the system being supported. These matching systems are referred to as converters, as shown in Fig. 2.8.

2.4 AC Grid Interface to LEDs

The main difference between the AC supply characteristics and the LED load requirements is shown in Fig. 2.9. The power coming from the AC supply has a ripple at double the grid frequency, while the power required by the LED is steady. A converter is needed to bridge the difference. Even after rectification; the rectified power, changes the current and voltage from bipolar to positive only but still both of them vary and exhibit rippling power at double the grid frequency.

Capacitors are used to provide load support when the grid voltage is low, and current filter inductors are used to smooth out the current, this solves the ripple problem.

To address the mismatch between the rectified grid voltage level and the LED voltage level a DC/DC converter is used to regulate and reduce the voltage level.

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Figure 2.9: AC supply characteristics and LED load characteristics. (a), (b) and (c) show the per unit AC grid supply voltage, current and power respectively at unity power factor. (d), (e) and (f) show the per unit voltage, current and power of the LED load respectively. Fig. 2.10 shows a general block diagram of a general LED driver. The intermediate stage denoted as a filter in Fig. 2.10 works mainly as a buffer to interface the pulsating waveform of the rectified signal to the DC/DC converter side input. Usually, bulk capacitors are used to filter out the ripple in voltage and power, as shown in Fig. 2.11. The changing role of the DC bus capacitor is shown in Fig. 2.12, where PAC is the instantaneous

AC power, 푃푎푐푝 is the peak value of the instantaneous AC power, 휔 is the angular frequency of the AC supply and 푃퐷퐶 is the instantaneous equivalent DC power. In region

(a), the capacitor stores the instantaneous extra power and then utilizes that stored energy in supporting the load in region (b).

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AC

LED Utility Rectifier Filter DC/DC Supply Convertor

Figure 2.10: General block diagram of a grid-connected LED driver.

AC 퐶퐵

LED Utility Rectifier DC BUS DC/DC Supply Convertor Figure 2.11: General block diagram of a grid-connected LED driver with a DC bus filter capacitor.

푃푎푐

푃푎푐푝

(푎) 푃퐷퐶 (푏)

o 휋 휋 3휋 휋 휔푡

4 2 4 Figure 2.12: AC and DC power waveforms. The capacitor is sized such that it stores the excess energy shown in region (a) of

Fig. 2.12, and then uses it to support the load in region (b). The total energy that needs to be stored (Ea) can be derived through the following equations:

2 푃퐴퐶(푡) = 푃푎푐푝(푠푖푛 (휔푡))

3휋 4휔 퐸 = ∫ [푃 (sin2(휔푡)) − 푃 ] 푑푡 푎 휋 푎푐푝 퐷퐶 4휔

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3휋 1 4 퐸 = ∫ [푃 (sin2(휔푡)) − 푃 ] 푑휔푡 푎 휔 휋 푎푐푝 퐷퐶 4

휋 + 2 2휋 퐸 = 푃 − 푃 푎 4휔 푎푐푝 4휔 퐷퐶

Since

푃푎푐푝 = 2 푃퐷퐶

푃 퐸 = 퐷퐶 (2.3) 푎 휔

The voltage ripple on the capacitor is affected by the capacitor size 퐶퐵 and the energy stored in the capacitor. If ∆푉 is the peak to peak voltage ripple, 푉푎푣푔 is the average voltage

∆푉 ∆푉 across the capacitor, 푉 = + 푉 and 푉 = 푉 − then the energy consumed 푚푎푥 2 푎푣푔 푚푖푛 푎푣푔 2 or gained 퐸푐 from 퐶퐵 between two voltage level is

1 퐸 = 퐶 (푉2 − 푉2 ) 푐 2 퐵 푚푎푥 푚푖푛

1 퐸 = 퐶 (푉 + 푉 ) (푉 − 푉 ) 푐 2 퐵 푚푎푥 푚푖푛 푚푎푥 푚푖푛

퐸푐 = 퐶퐵푉푎푣푔 ∗ ∆푉 (2.4)

Equating Eqn. 2.3 and Eqn. 2.4, the required capacitor size for the specific load power 푃퐷퐶, with certain voltage ripple on 퐶퐵 is given by [10][11].

푃퐷퐶 퐶퐵 = (2.5) 휔 푉푎푣푔 ∆푉

The voltage ripple in the DC bus must be minimized so that the regular DC/DC converter stage in Fig. 2.11 is designed for low ripple input source, but that comes at the

15

expense of using bigger capacitors. The rectified voltage in the DC bus is at the level of the grid peak voltage, then for the relatively low voltage loads like LED, a DC/DC converters is need to significantly reduce the voltage level, thus the need for isolated

DC/DC converters. Most LED drivers proposed in the literature, [12][13], are supplied from the grid and follow, for the most part, the topology introduced in Fig. 2.10.

Recognizing the difference between the nature of AC and DC, the impact of the topology and many other relevant parameters; different modifications, topological variations and control strategies have been proposed to mitigate some of the drawbacks of the main topology. Chief among these drawbacks are the size of the main DC bus capacitor, the power factor, the total harmonic distortion (THD) and the voltage rating of the DC bus capacitor.

2.5 Research Motivation

The main objective in this thesis is to come up with a converter that utilizes a PV solar power energy to supply an LED load lighting system while maintaining full solar panel utilization. Based on Fig. 2.13, different converter topologies are available for either

LED lighting, or for interfacing PV panels to the grid.

The power characteristics of the solar PV panel and the LED lighting system cannot be matched at all times as shown in Fig. 2.14. For example, before 10 A.M., the solar panel is unable to support the LED, while after that there is a power surplus that cannot be used as it exceeds the LED lighting system requirements. In order to be able to get the most benefit from the solar PV and to extract all of the available energy; the converter has to

16

LED PV Panel DC/DC Convertor Regulator

Figure 2.13: Solar panel with a Figure 2.14: Daily power profile DC/DC converter to supply a LED Load. of a PV panel and a LED lighting system.

have a n energy buffer between the varying supply (The solar PV) and the semi-constant

load of the LED. This buffer can be realized using different structures. Battery-based [14],

semi-based or full supercapacitor-based structures [15] can be used for this purpose.

Battery-based or supercapacitor-based structures store the excess energy from the solar

panel, to use it at moments when the solar panel does not generate enough power. These

structures have some critical points:

 The size of the energy storage elements dictate the maximum energy that

can be stored.

 If there is not enough energy stored and the PV produces less than the LED

needs, then the system would fail.

To solve these problems, and to overcome these limitation, the buffer system has

to have the capability to store enough energy while being able to support the LED when

even when there is no adequate power supply from the solar panel. The grid can be an

option to provide an energy buffer between the solar PV and the LEDs. Fig. 2.15 shows

17

LED

y a

r Grid

r A

AC

V P

Battery

Figure 2.16: PV, grid, battery and LED structure proposed in [16].

LED PV Panel

Grid

Figure 2.15: Conceptual realization of the proposed converter. the structure proposed in [16], which has the combined feature of battery storage and grid interface. The battery provides the option of energy storage instead of selling it to the grid, which gives the option of targeting high price periods. The main point of energy exchange here is the DC Bus capacitor. The DC link here is on the high voltage side, where several

PVs are connected to achieve such high voltage.

For low power low voltage systems, the PV panel cannot provide a DC bus that is high enough to interface to the grid. Another issue with the topology in Fig. 2.15 is the high number of required converters. The presence of the battery is redundant and it gives no additional benefits for low power applications, in the range of 100 W. To overcome these challenges, an alternative topology needs to be introduced. The block diagram of the

18

concept of a multiport converter topology proposed in this thesis is shown in Figure 2.15ig.

2.16.

Due to the relatively low voltage of the PV panel compared to the grid voltage; a voltage boost shall be achieved, and due to the large gap between the low power PV panel voltage and the grid peak voltage; the isolated converter comes as the main component in the system. The proposed converter, given in Fig. 2.16, can be realized with the three converter system given in Fig. 2.17. The roles of the three converters are, respectively:

 Converter#1 or the PV converter: Connects the PV and the DC link and

tracks the MPPT.

 Converter#2 or the Grid converter: Interfaces the grid and makes up the

balance of the power between the PV and the LED.

 Converter#3 or the LED converter: Supports power to the LED.

Different options to configure the individual converters can be suggested. To combine these converters and reduce the number of components, different topologies need to be investigated. The different voltage levels within the system suggest a multiport inverter.

Converter#3 LED

PV Panel

Grid

Converter#1 DC Bus Converter#2

Figure 2.17: Initial realization of the proposed converter.

19

n

n n

o

o o

i

i i

t

t

t t

t t

a

n

a a

n n

z

e

z z

i

e e

i i

r

t

r r

t t

r

r r

e

e e

u

u u

n

n n

C

g

C C

g g

a a

DT a DT DT

M M t M t t T T T (a) (b) (c) Figure 2.18: Inductor or flyback based converters modes of operation. (a) Continuous Conduction Mode CCM. (b) Boundary Conduction Mode (BCM) (c) Discontinuous Conduction Mode DCM.

The flyback converter is one of the topologies that can be used for different

applications and multipurpose usage. Different topologies and inverter configurations are

discussed and proposed in the literature [10], [17]–[21]. The different operating modes and

the flexibility provided by the presence of the transformer in the topology is a main

advantage for the flyback converter.

The flyback converter can be operated mainly in three different modes. Continuous

Conduction Mode (CCM), where the current in the magnetization inductance does not go

to zero when the next switching cycle starts. Boundary Conduction Mode (BCM), where

the second cycle starts immediately when the current in the inductor reaches zero.

Discontinuous Conduction Mode (DCM), where the current reaches zero and stays at zero

for a while before the next switching cycle starts. Fig. 2.18 shows the three modes of

operation.

Through these different modes of operation, researchers have been able to apply

different control strategies to achieve specific objectives, especially through the current

shaping feature of the flyback converter [20]. Current shaping and the transformer have

20

made the flyback converter a favorable candidate for grid-connected inverters [22], especially for low power levels and for house scale renewable energy systems [22].

In [19], a flyback inverter with power factor correction capability was proposed.

The authors added another winding, as in Fig. 2.20, to the conventional grid-tied flyback converter topology shown in Fig. 2.19. The inverter was operated in the BCM mode, thereby making the inverter more energy efficient because of the low switching losses due to zero current switching (ZCS).

푉푔푟푖푑 AC

푉푝푣

Figure 2.19: Conventional grid tied flyback inverter.

푉푔푟푖푑 AC

푉푝푣

Figure 2.20: Grid tied flyback inverter proposed in[19] with reactive power capability.

21

PV AC Port

AC DC Port

Ripple Port

Figure 2.21: Proposed topology for power decoupling in [23]. In [20], researchers proposed a solution to the impact of the change in the operating conditions. Utilizing a different switching reference; the current distortion was reduced along with the harmonics components. The bulky DC bus capacitor problem discussed in

Fig. 2.12 was addressed in [23]. The third port topology in Fig. 2.21 was used to decouple the grid frequency power ripple from the PV bulky capacitor, allowing the electrolytic capacitor to be replaced with a film capacitor, leading to a reduced system volume and increased lifetime. This work can be compared to that proposed in [17], shown in Fig. 2.22, where the grid input power ripple is smoothed out by using an extra decoupling capacitor.

Harmonic injection in the control signal was used in [18] to reduce the third harmonic, generated by the power factor controller in BCM mode.

In an attempt to combine the three converters in Fig. 2.17 and overcome the problems stated earlier, the topology shown in Fig. 2.23 is proposed to address all the objective stated in the motivation of supporting the LED load, and sustain it from the grid or the

PV panel while maintaining MPPT. From here on, this topology is referred to as Hybrid

Flyback Converter (HFC).

22

푉푔푟푖푑 AC

Figure 2.22: Proposed power factor correction topology in [17].

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷퐿퐸퐷 푆퐷퐶 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

Figure 2.23: Proposed HFC topology.

2.6 Conclusion

Different topologies used for grid interfacing were reviewed in this chapter. The flexibility and capability of the flyback converter concept were illustrated. The flyback converter is used to achieve different design goals by utilizing its physical features or applying different control strategies. A multiport HFC is introduced to develop a cost effective, compact and reliable converter for MPPT of solar PV panel and supporting a

LED based lighting system.

23

PROPOSED HYBRID FLYBACK CONVERTER

3.1 Introduction

The proposed HFC has four modes of operation; the first mode is grid only mode, the second mode is LED only mode, the third mode is grid and LED mode and the forth mode is grid support mode. In this chapter, the four modes of operation will be illustrated and analyzed separately to ease and simplify the analysis of the HFC. The outcomes of the simplified analysis of each mode will be taken into consideration in any further mode analysis. Eventually, the results of each analysis will be considered in the final parameters selection and design. In this design, the power required by the supported LED lighting system will vary based on the user usage up to 120 W.

3.2 Topology and Principle of Operation

The base topology of the proposed multiport converter architecture is the flyback concept. The flyback topology and concept enhance the capabilities of converters and provide more flexibility and degree of freedom in the designing and the interfacing converters. The selected concept of operation for the analysis of the HFC is the DCM. The main reason for selecting this mode is the ease of implementation and control [20]. The targeted operating conditions of the HFC are listed in Table 3.1. As shown in Table 3.1, the designed system will be able to handle power exchange up to 121 W. The provided parameters are based on a typical industrial PV panel.

24

Table 3.1: PV panel and grid parameters. PV Panel Utility or Grid

Parameter Value Parameter Value

Power 푃푃푉_푀푃푃 121 W RMS voltage 푉푎푐,푅푀푆 120 V

Open circuit Peak voltage 푉푃푉_푂퐶 42 V 푉푎푐,푝 170 V voltage

Voltage at MPP 푉푃푉_푀푃푃 35 V Frequency 푓 60 Hz

Current at MPP Switching 퐼푃푉_푀푃푃 3.45 A 퐹푠 20 kHz Frequency

3.3 Grid Only Mode

This mode focuses only on the case where all the power from the PV panel is being injected into or exported to the grid. The configuration is shown in Fig. 3.1 where the LED port is dimmed while the remaining, where n is the secondary’s turns ratio and m is the tertiary turns ratio. The system will be treated as a grid tied flyback converter[22].

Tertiary Secondary 퐷푝 푖푆퐴퐶 퐿푓 푖푆퐷퐶 퐷퐿퐸퐷 푉퐿퐸퐷 푆퐴퐶푝 퐶푓 푚 AC 푉푔푟푖푑

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

Figure 3.1: Grid only mode involved ports.

25

This mode has two steps of operation:

 Step 1: S1 is ON, the current in the primary side (푖푆1)starts to increase into

the primary winding of the transformer untill 푖푆1 reaches 푖푝1; that is, when

S1 is turned OFF, this step is shown in Fig 3.2 (a) as and the corresponding

current waveform in the top part of Fig. 3.3. This charges the magnetization

푉푝푣 푑 푇 inductance from the PV panel as shown. 푖푝1 can be found by 푖푝1 = 퐿1

where 푉푝푣 is the PV panel average voltage, 퐿1 is the magnitization

inductance of the transfomer referred to the primary side, 푑 is the duty ratio

1 and 푇 = . 퐹푠

 Step 2: S1 is OFF, and either 푆퐴퐶_푝 or 푆퐴퐶_푛 is on, the stored energy in the

magnetization inductance of the transformer is discharged into the grid.

푆퐴퐶_푝 is turned ON during the positive half cycle of the grid voltage and

푆퐴퐶_푛 is ON during the negative half cycle as Fig. 3.2 (b) and (c) shows. The

푖푝1 secondary current will be discharged in to the grid in period T2, governed 푛

2 푛 퐿1 푖푝1 by the equality 푇2= , where 푉푎푐 is the grid voltage shown in the 푛 푉푎푐

bottom of Fig. 3.3.

Fig. 3.3 summarizes the charging and discharging currents of the magnetization inductance. In order for a sinusoidal current to be supplied to the grid, a sinusoidal reference signal is used to generate the duty cycle for the main switch S1 as shown in Fig.

3.4.

26

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷퐿퐸퐷 푆퐷퐶 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(a)

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷 푆퐷퐶 퐿퐸퐷 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(b)

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 푆퐷퐶 퐷퐿퐸퐷 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(c) Figure 3.2: Grid only mode steps of operation, (a) Step1, (b) Step2, Positive half cycle, (c) Step2, Negative half cycle.

27

Current 푖푝1 푖푠1 t

퐷1 푇 푖 푝1 푛 푖푆퐴퐶 t Step1 Step 2

푇2 푇

Figure 3.3: Simplified magnetization inductance current in grid only mode operation.

푉푔푟푖푑 Scaling abs > 푆1

Yes 푆퐴퐶푝 >0 No 푆퐴퐶푛

Figure 3.4: PWM generation scheme for grid only mode. The result of such control PWM signal is a set of triangular current pulses following a sinusoidal envelope. Fig. 3.5 shows the generic current waveforms of the HFC in grid

only mode for one complete 60 Hz cycle where 푖푆퐴퐶 is the AC side current as indicated in

Fig.3.2 , 푖푆퐴퐶_푝 is the AC side current in 푆퐴퐶푝 and 푖푆퐴퐶_푛 is the AC side current in 푆퐴퐶푛.

28

(a) 푖푆1 (b) 푖푆퐴퐶

(c) 푖푆퐴퐶_푝 (d) 푖푆퐴퐶_푛 Figure 3.5: Generic waveforms of the current in the HFC for a 60 Hz cycle in grid only mode. DCM is chosen as the operating mode, so the volt second rule must be sustained all times. The volt second when S1 is ON must be less than or, at maximum, equal to the

volt second when the AC side switches 푆퐴퐶푝or 푆퐴퐶푛 are ON, especially at the peak of the grid voltage,

푉푎푐,푝 푉 푑 푇 ≤ (1 − 푑 )푇 푝푣 푝1 푛 푝1

So

1 푑 < 푝1 푛 푉 (3.1) 1 + 푝푣 푉푎푐,푝

The maximum duty ratio (푑푝1) assures that the average voltage across the transformer inductance is zero over each switching cycle. The max duty ratio depends on the turn ratio

(n), the average PV voltage ( 푉푝푣) and the peak voltage of the grid 푉푎푐,푝. The maximum

29

duty cycle can be used to find the maximum power the inverter can process in this mode.

The HFC shall not be operated beyond the max duty cycle.

The maximum average current injected to the grid ( 푖푎푐), in Step 2, can be found by

1 푇 푖푝1 푖 =< 푖 > = 2 푎푐 푆퐴퐶 2 푇 푛

푉 푖 The power injected to the grid is given by 푃 = 푎푐푝 푎푐푝, therefore the maximum power 푎푐 2 that can be injected in DCM mode is given by

2 2 1 푑푝 푉푝푣 푃 = (3.2) 4 퐹푠 퐿1

The maximum power that the inverter can process depends on 푑푝, the switching frequency 퐹푠, the PV panel voltage 푉푝푣 and the primary magnetization inductance 퐿1 as per

Eqn. 3.2. This equation can be used to find the max duty cycle for a certain power level.

The current and voltage stresses on the components are some of the most important aspects of the design, the two main stresses in the HFC – in this mode – are the voltage stress across the AC side switches ( 푆퐴퐶_푝 푎푛푑 푆퐴퐶_푛 ) and the current stress on the main

푉푝푣 푑푝1 switch S1. Substituting Eqn. 3.1 and Eqn. 3.2 into 푖푝1 = gives the maximum current 퐹푠 퐿1 stress on the primary side switch

퐼푑푐,푝 푉푝푣 = 4 [1 + 푛 ] (3.3) 퐼푝푣 푉푎푐,푝 while the voltage stress on the secondary side switches is

푉푆퐴퐶 = 2 푉푎푐,푝 (3.4)

30

The secondary side switches shall be able to withstand a voltage stress of double the grid peak voltage 푉푎푐,푝 [24][25]. The required capacitor size for the bulk PV capacitor depends on the PV voltage ripple, the grid frequency and the amount of power,

푃 퐶푝푣 = (3.5) 휔 ∆푉푝푣 푉푝푣 where ∆푉푝푣 is the peak to peak voltage ripple on the PV panel.

Different approaches or starting conditions can be used for designing the HFC. The approach taken in this work starts by selecting a peak current stress of 28 A on S1, so the turns ratio is n=5, from Eqn. 3.3. Then, using Eqn. 3.4 the voltage stress on the AC side switches can be found as 푉푆퐴퐶=340 V. For the inverter to handle 130W, at the selected duty ratio of 0.485, that is less than the ultimate duty ratio 푑푝1 of 0.493, the required primary inductance is found to be 퐿1=27.7 µH. 퐶푝푣 is chosen to be 9.17 mF to give a voltage ripple of 0.5 V when the PV panel power is 121 W.

3.4 LED Only Mode

In this mode, the grid port will be completely ignored, as shown in Fig. 3.6

Tertiary Secondary 퐷푝 푖푆퐴퐶 퐿푓 푖푆퐷퐶 퐷퐿퐸퐷 푉퐿퐸퐷 푆퐴퐶푝 퐶푓 푚 AC 푉푔푟푖푑

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

Figure 3.6: LED only mode operation.

31

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷퐿퐸퐷 푆퐷퐶 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(a) 푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷 푆퐷퐶 퐿퐸퐷 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(b) Figure 3.7: LED only mode steps of operation, (a) Step1, (b) Step2. This mode can be analyzed or designed as a normal DC/DC flyback converter to operate at all operation modes; that is, CCM, BCM and DCM. To make it simple and compatible with the control structure provided in Fig. 3.4; it has to be designed based on the AC cycle PWM. So there are two operating steps in this mode as shown in Fig. 3.7.

Step1 is the same as that of the grid only mode, but Step2 is different, where instead of discharging the stored energy in the magnetization inductance to the grid it will be discharged to the LED port based on the flyback concept through 퐷퐿퐸퐷. Fig. 3.8 shows the generic waveforms for the currents in this mode where 푖푆퐷퐶 is the waveform of the current following through 퐷퐿퐸퐷.

32

(a) 푖푆1 (b) 푖푆퐷퐶 Figure 3.8: Generic waveforms of the current in the HFC for a 60 Hz cycle in LED only mode. Similarly to the Grid only mode, and using the volt second rule, the maximum allowable duty cycle for this mode 푑푝2 is dictated by

1 푑 < 푝2 푚 푉 (3.6) 1 + 푝푣 푉퐿퐸퐷

The current stress on 퐷퐿퐸퐷 and the voltage stress on the AC side switches

(푆퐴퐶_푝 푎푛푑 푆퐴퐶_푛) are calculated using

퐼푑푐,푝 푖 = (3.7) 푆퐷퐶 푚

푉 푛 푉 = 푉 + 퐿퐸퐷 (3.8) 푆퐴퐶 푎푐,푝 푚

To continue the design of the HFC, the parameters that have been selected in the grid only mode will be used as the base, then any necessary condition or modification will be satisfied or modified to make sure that all modes of operation can be implemented in the same hardware.

The LED port voltage was chosen to be 24 V as one of the regular voltage levels used for LEDs. To keep 푑푝2 less than 푑푝1, m had to be chosen such that 푚 < 0.75; so m was chosen to be 0.5. The current stress on the LED side diode is 56 A peak, and the voltage

33

stress on the AC side switches is 520 V. The required capacitor size for the LED port is calculated using

푃 퐶퐿퐸퐷 = (3.9) 휔 ∆푉퐿퐸퐷 푉퐿퐸퐷 where ∆푉퐿퐸퐷 is the peak to peak voltage ripple on the LED. This gives a 6.68 mF for a voltage ripple of 1 V.

3.5 Grid and LED Mode

In this mode, all ports of the HFC are active and involved in the energy exchange.

The PV panel power will be distributed between the grid port and the LED port. Actually this mode is the result of combining the two previous modes. So the HFC will have three steps of operation.

 Step1: S1 is ON, storing the energy from the PV in the transformer similarly

to Step 1 in other modes as in Fig. 3.9 (a).

 Step2: 푆퐴퐶_푝 표푟 푆퐴퐶_푛 is ON, discharging some of the stored energy to the

푖 grid side. The secondary current 푝1 will be discharged in to the grid till it 푛

reach 푖푝2 then the AC side switch is turned OFF as shown in Fig. 3.9 (b).

 Step3: All switches are OFF, 퐷퐿퐸퐷 discharges the remaining energy in the

n transformer to the LED port. The current starts at 푖 until the current is 푝2 푚

fully distinguished as in Fig. 3.9 (c).

The corresponding current of the magnetization inductance for the three steps is shown in Fig. 3.10. A generic waveform for the currents in the drive in this mode is shown in Fig. 3.11.

34

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷퐿퐸퐷 푆퐷퐶 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(a)

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷 푆퐷퐶 퐿퐸퐷 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(b)

푖 Tertiary Secondary 퐷푝 푆퐴퐶 퐿푓 푖 퐷 푆퐷퐶 퐿퐸퐷 푆 퐶 푚 퐴퐶푝 푓 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

(c) Figure 3.9: Grid and LED mode steps of operation, (a) Step1, (b) Step2 and (c) Step3.

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푖푝1 푖푠1 Current t

퐷1 푇 푖 푝1 푖 푛 푆퐴퐶 푖푝2

t 퐷 푇 2 푛 푖푝2 푚 푖푆퐷퐶 t Step1 Step 2 Step 3 푇 Figure 3.10: Simplified magnetization inductance current in grid and LED mode steps of operation.

(a) 푖푆1 (b) 푖푆퐴퐶

(c) 푖푆퐴퐶_푝 (d) 푖푆퐴퐶_푛

(e) 푖푆퐷퐶 Figure 3.11: Generic waveforms of the current in the HFC for a 60 Hz cycle in grid and LED mode.

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Only one condition must hold to make sure that the HFC is able to work in this mode which is exporting some of the power to the grid and some to the LED. The condition is that the diode of the LED 퐷퐿퐸퐷 must remain reverse biased when the grid is taking the power. This is shown as

푚 푉 > 푉 (3.10) 퐿퐸퐷 푛 푎푐,푝 for m=0.5, n=5, 푉푎푐,푝=170 V and 푉퐿퐸퐷=24 V, the condition in Eqn. 3.10 holds.

3.6 Grid Support Mode

This mode covers the case when either the PV power is not enough to supply the

LED or the PV panel is not working at all, so the deficit in power will be supplied from the grid. This mode has also three steps like the grid and LED modes. The main difference is in Step2 where, instead of discharging into the grid, the grid will keep charging the transformer, then discharge to the LED port, as shown in Fig. 3.12.

푖 Current 푖푠1 푝1 t

퐷1 푇 풊 푖푆퐴퐶 풑ퟏ 풏 푖푝2 t 퐷2 푇 푛 푖 푝2 푚 푖푆퐷퐶 t Step1 Step 2 Step 3 푇

Figure 3.12: Simplified magnetization inductance current in grid support mode steps of operation.

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(a) 푖푆1 (b) 푖푆퐴퐶

(c) 푖푆퐴퐶_푝 (d) 푖푆퐴퐶_푛

(e) 푖퐷_퐿퐸퐷 Figure 3.13: Generic waveforms of the current in the HFC for a 60 Hz cycle in grid support mode.

Unlike the grid and LED mode, the activation of the AC side switches 푆퐴퐶_푝 or

푆퐴퐶_푛 is reversed; that is, 푆퐴퐶_푝 is activated during the negative half cycle and 푆퐴퐶_푛 is activated during the positive half cycle. The generic current waveforms of a 60 Hz cycle of the HFC in this mode is shown in Fig. 3.13.

Some additional conditions need to be sustained at all times, in addition to the previous conditions stated in other modes. The condition of

1 푉 > 푉 (3.11) 푝푣 푛 푎푐,푝 needs to hold to reverse bias the AC side diodes at Step1, and to reverse bias the body diode of S1 at Step2. To reverse bias the LED diode in Step1 and Step2 the conditions

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푉퐿퐸퐷 > 푚푉푝푣 { 푚 } (3.12) 푉 > 푉 퐿퐸퐷 푛 푎푐,푝 shall be satisfied. All these conditions are satisfied using the parameters selected previously for the design.

3.7 HFC Controller

The proposed controller of the HFC is shown in Fig. 3.14 where the MPPT block gives the command of D1 to Track the MPP, 퐺 is a gain that depends on 푑푝1 . The gain 퐺 has to be less than 1/푑푝1, because the maximum time required for the AC side switches to discharge the magnetization inductance equals (1 − 푑푝1 )푇. The Control on the AC side switches is based on D2. When D2 is positive, power will be injected to the grid. When D2 is negative, power will be imported from the grid. Fig. 3.15 shows the complete HFC and its controllers.

푆1 퐼푝푣 퐷 MPPT 1 |sin(wt)| 푉푝푣

푉푔푟푖푑 > 0 → 푆퐴퐶푝 푉푔푟푖푑 < 0 → 푆퐴퐶푛 G 퐷 2 퐷2 > 0 푉푟푒푓 PI 푉푔푟푖푑 < 0 → 푆퐴퐶푝 푉푔푟푖푑 > 0 → 푆퐴퐶푛

푉퐿퐸퐷

Figure 3.14: The proposed controller of the HFC.

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Tertiary Secondary퐷푝 푖푆퐴퐶 푉퐿퐸퐷 퐿푓 푖 푆퐷퐶 푆 푚 퐴퐶푝 푉푔푟푖푑 AC LED 퐶푓

푖 푆1 푛 1 푆퐴퐶푛 Primary 퐷푛 푉푝푣 푆1

푉푝푣 퐼푝푣 퐷 푆1 MPPT 1 > x 푆 abs x 퐴퐶푝 푉푔푟푖푑 > PLL G 퐷 푆퐴퐶푛 2 x > 푉푟푒푓 PI abs

Figure 3.15. The HFC and its controller. 3.8 Filter Type Selection and Design

Many types of filters are used in connecting inverters to grids, such as LCL, CLC,

CL, LC. The Converter topology and the principle of operation allows a fast selection between filters. Since our inverter works as a current source inverter for the grid side, a current filter rather than a voltage filter is used. One of the common suitable filters for this

CSI tied to the grid is the CL filter [10][19][23].

The grid is interfaced to the AC side switches of the HFC through a CL filter. The

CL filter extracts the fundamental 60 Hz current from the triangular pulsating current wave shown in Fig. 3.16(a) 푖푆퐴퐶, in other words the CL filter filters out the high order harmonics and extracts the fundamental component, as shown in Fig. 3.16(b) 푖퐺푟푖푑 .

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For the analysis of the CL filter, the HFC in grid only mode will be represented as a current source, as in Fig. 3.17. From Fig. 3.17, the transfer function of the filter as a current filter is

푖퐺푟푖푑(s) 1 푠푐 푉푔(푠) 퐹(푠) = = 2 − 2 (3.13) 푖푆퐴퐶(s) 푠 퐶퐿 + 1 푠 퐶퐿 + 1 푖푆퐴퐶(s)

Eqn. 3.13 shows that the filter transfer function does not depend on the inverter current only but also on the grid voltage and the inverter current. Eqn. 3.13 can be represented in a block diagram as shown in Fig. 3.18.

(a) 푖푆퐴퐶

(b) 푖퐺푟푖푑 Figure 3.16: The objective of the filter to filter (a) HFC current 푖푆퐴퐶 to (b) the fundamental current 푖퐺푟푖푑.

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푖퐺푟푖푑 푖푆퐴퐶

퐿푓

푖퐶푓 푉푔푟푖푑 AC 퐶푓

Figure 3.17: Block diagram of the filter interface between the grid and the HFC.

푖푆퐴퐶 1

푠2퐶퐿 + 1 + 푖퐺푟푖푑 푠푐 - 푉푔 푠2퐶퐿 + 1

Figure 3.18: CL filter transfer function block diagram as a function of inverter current 푖푆퐴퐶 and grid voltage 푉푎푐. Another way to analyze the filter is to look at the grid as a specific impedance value

푉푔푟푖푑 for specific grid current; that is 푍푔 = ⁄ . So, the circuit model is modified as 푖푔푟푖푑 shown in Fig. 3.19 and the transfer function will be

푖퐺푟푖푑(s) 1 퐹(푠) = = 2 (3.14) 푖푆퐴퐶(s) 푠 퐶퐿 + 푍푔푠퐶 + 1 while the block diagram model will be reduced to Fig. 3.20.

푖퐺푟푖푑 푖푆퐴퐶

퐿푓

푖퐶푓 푍푔 퐶푓

Figure 3.19: Block diagram of the filter interface between the grid and the HFC with the grid as 푍푔.

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푖푆퐴퐶 1 2 푖퐺푟푖푑 푠 퐶퐿 + 푍푔푠퐶 + 1

Figure 3.20: CL filter transfer function block diagram as a function of inverter current 푖푆퐴퐶 and grid equivalent impedance 푍푔 .

Figure 3.21: Filter pole locus as 푍푔 the changes.

Assuming that a unity power factor is supplied to the grid; 푍푔 will be a pure resistor.

Fig. 3.21 shows the locus of the poles of the transfer function of Fig. 3.20 as 푍푔 changes.

This represents the change in the grid current effect on the system poles locations. As the injected power to the grid increases the system poles locations approach the natural resonance poles of the CL filter more and more. The worse damping performance is when the 푍푔 term is neglected. So the transfer function will be considered as[26]

푖퐺푟푖푑(s) 1 퐹(푠) = = 2 (3.15) 푖푆퐴퐶(s) 푠 퐶퐿 + 1

To select the filter parameters, first the attenuation at the switching frequency 푘푎 is

푖푔(푗휔푠푤) 1 = 2 = 푘푎 (3.16) 푖푆퐴퐶 (푗휔푠푤) 1 − 퐶퐿휔푠푤

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1 If 휔 = then the ratio of the switching frequency to the resonance frequency 푟 can 푟푒푠 √퐿퐶 be calculated as

1 푟 = √1 + (3.17) 푘푎

푉2 As mentioned earlier the grid can be represented as a load with value of 푍 = 푔푟푖푑, 푔 푃 which is also the base impedance of the filter [27]. The base capacitance can be found as 퐶푏 = 1/(휔푔푟푖푑 푍푔).

Despite the grid synchronized injection of current to the grid, the filter adds a power factor effect on the injected power, so a reactive component appears. Based on the selection of the filter capacitance the reactive power contribution of the filter is determined, and so the capacitor filter is determined as a ratio (α) of the base capacitance depending on the maximum allowed power factor so

퐶푓 = α 퐶푏 (3.18)

1 Accordingly the filter inductor may be found using 휔 = . The selection may need 푟푒푠 √퐿퐶 more than one try based on the initial selection of 푘푎 and the final selection of α. The selection process resulted in the filter parameters to be 퐶 푓 = 1.6푢퐹 푎푛푑 퐿푓 = 11 푚퐻.

3.9 Conclusion

This chapter presented the different operating modes of the proposed converter topology. In each mode, the required physical and operational conditions to allow for the operation of the converter in the specific mode were illustrated and explained. The grid side filter design was detailed due to the high impact on the THD and the power factor.

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HYBRID FLYBACK CONVERTER HARDWARE DESIGN

4.1 Introduction

In Chapter 3, the design parameters, condition and requirements of the proposed

HFC were outlined. In order for these parameters to be realistic for a practical implementation, specific components shall be designed before proceeding into further analysis or implementations. This chapter covers the design consideration for some of the critical and essential components in the HFC such as the selection of the transformer and switches. After ensuring that those components are practically viable, the printed circuit board (PCB) design is implemented.

4.2 Transformer and Inductor Design

A 4-winding transformer is the heart of the converter design. The energy transfer and exchange between all windings is achieved by storing the energy in the magnetic field of the transformer, then guiding this energy to a port that is selected based on the operating conditions. Transformers have two operational modes; symmetrical and asymmetrical operation [28]. Fig. 4.1 shows the difference between the two operation modes. The asymmetrical operation utilizes the first quarter of the BH curve; that is, AC line in Fig.

4.1, while the symmetrical operation utilizes the whole BH curve line, denoted as CAB points in Fig. 4.1.

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B

A

C

Figure 4.1: Typical BH curve.

S1 ON SAC or DLED ON t

Flux Φ

t

Figure 4.2: Flux accumulation and discharge into and from the transformer core during switching states

For the HFC, the asymmetrical operation will be utilized while operating in one of the four modes of operation. For example, when S1 is turned ON the flux (Φ) or the flux density (B) will accumulate in the transformer until the switch is turned OFF. The accumulated flux density (B) can be calculated by

1 퐵 = ∫ 푉푝푣 푑푡 푁 퐴푐 (4.1) 1 푑푝1 퐵 = 푉푝푣 푁 퐴푐 퐹푠

46

2 where N is the number of turns in the winding, Ac is the core cross sectional area in m .

When S1 is turned OFF, the flux will be discharged to any other selected port as shown, generically, in Fig. 4.2. .

The inductance of the transformer and the turn ratios (n and m) between its windings is critical in defining the capability of the HFC and the operating range of the proposed concepts as described in the design equations of Chapter 3.

4.2.1 Core Materials

Transformers are composed of a magnetic core and windings around this core. In transformer design, the material of the core is one of the most important criteria in the initial selection. Every material is recommended to be used in specific applications, based on its power level and frequency range [29]. There are many types of materials used in building transformers such as iron powder, ferrite, Nano-crystalline and silicon steel. In each type of material there are different alloys with different characteristics.

Mainly ferrite cores are used for power electronics applications [30][29], which is preferred for its low losses, especially in the frequency operation region of switch mode power supplies (SMPS). Ferrite material has a wide range of permeability and relatively low maximum flux density when compared to iron steel, iron powder and Nano-crystalline

[29]. The iron powder core has relatively low permeability but higher flux densities. Core losses are higher for iron powder compared to ferrite at high frequencies.

The main parameters that differentiate materials from each other are the permeability of the material (µ), the maximum flux density (Bsat) and the core losses at different frequencies. Another parameter that comes into play in applications is the

47

available winding area. Some of these parameters are lumped together to form another parameter, for example, the inductance per square turn (AL) as in

휇표휇푟 퐴푐 휇표 퐴푐 퐴퐿 = + (4.2) 푙 푙푔

AL represents the permeability (휇), the core cross-sectional area (퐴푐), the length of the magnetic circuit (푙) and the air gap length in air gapped cores (푙푔) and the air permeability (휇푟). 퐴퐿 is one of the main parameters used in the design to get the required inductance value and used in the following form.

2 퐿 = 푁 퐴퐿 (4.3)

4.2.2 Core Shape

Transformers’ cores are manufactured in different shapes such as E, EI, Toroidal,

C, Q, PM and planar. Fig. 4.3 shows several core shapes. Each core shape has its own characteristics and capabilities. The main used ones are toroid and E shaped cores. The E shaped core comes in two parts and uses a bobbin for the windings formation. Fig. 4.4 shows an E shaped core and its bobbin. The shapes that come with bobbins are easy to

Figure 4.3: Different core shapes for transformers and inductors.

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Figure 4.4: Ferrite E-shaped core and its bobbin. wind by machines and even by human hands, unlike the toroidal cores that are difficult to handle and require a more complicated machinery to wind. An E core shape was used for the output filter inductor while a toroid core was used for the main transformer due to winding area availability.

4.2.3 Flux Density and BH Curve Operating Region

The operation of the HFC relies on two main assumptions; the first is that the

푑푖 inductance is fixed, thus 푣 = 퐿 is satisfied and the system is operating within the linear 푑푡 operating region and below the saturation limit. The selected toroid core for the 4 winding

2 transformer has a maximum flux density (Bsat) of =1.3T, Ac =1.68 cm . This is the maximum applied Bsat calculated using Eqn. 4.1. The primary winding has 18 turns in the primary; so the maximum operating flux density Bopt =0.287 T.

The output filter inductor uses an E core of the same material used for the transformer. The maximum operating flux density of the output filter inductor can be calculated using

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휇 휇 푁퐼 퐵 = √2 푟 ° (4.4) 푙

where 휇푟 is the relative permeability of the core material, 휇° is the air permeability, and

(I) is the RMS output current. The output filter E core has 200 turns, 휇푟 of 75 and l =13.2 cm and is operated at maximum flux density of 0.202 Tesla. Both cores are operating at relatively low flux density compared to the saturation flux density. This ensures operation in the linear region.

4.2.4 Winding wires cross-section

The selection of wire size for each winding initially depends on the RMS current following through that winding, the allowable current density ( 퐽) and the maximum temperature rise. Generally, the required cross sectional area (푊퐴) for certain amount of current is

퐼 (푊 ) = 푟푚푠 (4.5) 퐴 퐽

As a rule of thumb, the current density of copper wires is considered to be 4.5 A/mm2.

This can be modified in the design process [28].

The operating frequency is another factor that has to be considered for the skin effect. The skin effect is the tendency of the current to flow through the outer layer of the conductor. The skin depth (훿) can be calculated using

2휌 (훿) = √ (4.6) 휔휇

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Figure 4.5: Sample of a Litz wire where 휌, 휇 are the conductor resistivity and permeability, and 휔 is the angular velocity of

−8 the current. For copper of 휌=1.68×10 Ωm and 휇 = 휇표 ; 훿=0.46 mm at 20 kHz signal.

That means that the largest wire strand to be used should be 20 AWG of cross sectional area 0.51 mm2. The advantage of considering the skin effect is the efficient utilization of the available winding area of the core, because the unused area of the wire, without considering the skin effect, will reduce the effective available winding area. The AC resistance is higher than the DC resistance of the wire, so after considering the skin effect, the wire size and type can be selected accordingly to get better area utilization.

4.2.5 Litz wire

Litz wire is a wire made of several fine insulated strands, as shown in Fig. 4.5. The usage of Litz wire in high-frequency applications guarantees that the required wire, as calculated in Eqn. 4.5, will be effectively utilized.

Applying Eqn. 4.6 will specify the maximum wire radius that can be used effectively. The winding factor (K) is another parameter that indicates the effective utilization of the available winding area. The winding factor considers the gaps created between the wires and each layer of wires wounded in the coil. Considering all of these effects the required winding area can be calculated for a 4 winding transformer using

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푊 푁 + 푊 푁 + 푊 푁 + 푊 푁 퐴 = 퐴1 1 퐴2 2 퐴4 4 퐴4 4 (4.7) 푤 퐾

where N1 and N4 are the primary and tertiary number of turns respectively, while N2 and

N3 are the number of turns of the secondary windings.

The transformer design and core selection process is not straightforward. Several passes of design and reselection need to be made before settling on a final design, especially for off the shelf components. The T300-26 iron powder toroidal core from Micrometals

2 was chosen to function as the main transformer core, with 퐴푙=80 nH/N , N1=18, N4=9,

N2=N3=90, thus the primary inductance is L=25.9 µH. The E220-26 iron powder E core

2 from Micrometals was chosen to function as the output filter core, with 퐴푙=275 nH/N with

N=200 turns, therefore the filter inductance L=11mH.

4.3 Power Device Selection

MOSFETs and IGBTs are the main types of switches used in power electronics circuits and applications. Each type has the upper hand in certain operating conditions than others, as shown in Table 4.1 [31].

Table 4.1: Preferred operating conditions of some switches.

Characteristics MOSFET IGBT Frequency More than 200 kHz Less than 20 kHz Voltage 250 V > 1000 V Duty Cycle Long Short

MOSFET switching periods; that is, turn ON and turn OFF times, are short compared to those of IGBTs, which reduces the switching losses significantly and makes

MOSFETs more efficient, especially in low voltage applications. IGBTs are used mainly

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OVERLAP

Figure 4.6: Operation regions for MOSFETs and IGBTs in terms of voltage and power. in high power applications. Fig. 4.6 [31] shows the operating regions of these types of power devices. MOSFETs have been chosen as switches in this application. To specify the switch parameters three factors are considered:

 Maximum voltage stress

o The MOSFET must be able to withstand the maximum reverse voltage

normally without sustaining any damage. The voltage stress is practically

higher than the theoretically calculated stress due to circuit parasitics such

as stray inductances, stray capacitance and mainly leakage inductance,

especially during switching and transitional states. Thus the maximum

reverse voltage that the MOSFET has to withstand is chosen to be multiples

of the one calculated in the ideal case.

 RMS current and Peak current stresses.

o MOSFETs have a continuous ON current rating and pulse current rating.

The nature of the current in the HFC is a pulsating triangular current. The

current pulses are indigenous to the system, so with the parasitics included

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the pulses would be higher. As in the voltage rating, the current rating of

the MOSFET should be multiples of the theoretically calculated rating.

 Losses and Drain source ON resistance RDS_ON.

o The conduction losses are mainly controlled by RDS_ON, so the chosen

switch should have a small RDS_ON.

The final selection of the switches is shown in Table 4.2. The table also shows the theoretically calculated voltage and current stresses.

Table 4.2: Final selected parameters for the HFC switches.

S1 SAC

Voltage stress 83 V 580 V

Peak current stress 28 A 5.6 A

RMS current stress 7.5 A 1.1 A

IPP110N20NA of Selected Switch IXFK24N100 of IXYS INFINEON

Max voltage stress 200 1000 V

Max. Continuous 88 24 A current

Pulse current 352 A for 1usec 96 A for 300usec

RDS_ON 10.7mΩ 390 mΩ

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Figure 4.7: Schematic of grid voltage sensing chain.

4.4 Conditioning Circuit for Sensors

Four signals shall be sensed and fed into the controller to achieve the objectives of the HFC.

4.4.1 Grid Voltage Sensor

The grid voltage has to be sensed for the Phase Locked Loop (PLL) to track the grid voltage. Fig. 4.7 shows the grid voltage processing chain. The grid voltage is scaled down using the voltage divider resistors Rx and Ry. The isolation amplifier (AD202JN) is used to sense voltage across Ry. A Sallen Key filter is used to filter any unwanted noise or distortions. Since the grid voltage is bipolar, the measured signal is pivoted around 1.75 volts by adding an offset voltage using a summing amplifier.

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Figure 4.8: Schematic of PV panel voltage sensing chain. 4.4.2 PV Panel Voltage Sensor

The solar panel voltage will be measured and fed into the MPPT algorithm in the controller.

The PV panel voltage measurement processing chain is the same as the grid voltage except for the offset part. The PV voltage is a DC voltage so there is no need for the offset. A low pass Sallen Key filter is used to remove any high-frequency signals that can appear due to the switching nature of the converter and due to the sizing of the PV panel input capacitor.

Fig. 4.8 shows the schematic of the PV panel voltage processing chain.

4.4.3 LED voltage Sensor

The main purpose for the HFC is to supply the LED load with a constant voltage and the required power. The LED voltage will also be measured and fed into the controller.

Fig. 4.9 shows the LED voltage measurement processing chain which contains voltage scaling, isolation sensing and a low pass Sallen Key filter. The offset voltage can be used in this scenario to avoid a false zero measurement.

4.4.4 PV Panel current Sensor

The MPPT algorithm uses the PV panel current. The sensor used (LTS-6NP of

LEM) provides the measured current signal pivoted around 2.5 V, and thus 2 V is deducted

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in the summing amplifier as shown in Fig. 4.10. This will center the current measurement on 0.5 V to utilize the voltage span of the controller ADC bin much better. For better utilization of the ADC voltage span the signal can be amplified further.

Figure 4.9: Schematic of LED voltage sensing chain.

Figure 4.10: Schematic of PV panel current sensing chain

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4.5 Gate Driver Development

The gate driver is one of the important parts in any converter. The design of the gate driver circuitry around a certain gate driver chip defines the capability of the gate driver and hence the overall circuit performance. The role of the gate driver is to drive the gate or the base of the MOSFET or the IGBT respectively using the appropriate voltage levels and supply the required current. Thereby controlling the turn-on time and the turn- off time while also having an impact on the ringing phenomenon. Through the accumulated experience of the researchers in the Alternative Energy Lab at The University of Akron, different gate driver circuits have been designed. With continuous enhancement in every version designed, the latest design of the gate driver circuitry was used to drive the switches. Table 4.3 shows the main parameters and capabilities of the gate driver circuitry developed in the Lab.

Table 4.3: Gate driver chip capabilities and specification.

Input voltage 5 VDC

Signal isolation Opto-coupler, Isolated DC/DC converters

Driver voltage +15VDC,-15VDC

Maximum current 3 Amps

Main gate driver FOD8318 of Fairchild chip

Other features Fault detection and indication that can be used in inverter protection scheme. Compact design: vertical connection that saves space on PCB, Plug-in capability, Easy to debug.

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4.6 Printed Circuit Board Layout

The final designed board is a PCB with 2 layers and 35 µm of copper thickness. By using the French norm NF C 93-713 ANNEX C, of January 1989, the board would be considered as class 6. Moreover, for the design of the board additional guidelines were followed:

 The minimal width is 0.12 mm.  The minimal distance between two traces is 0.12 mm.  The minimal width of each via is 0.2 mm.  The board traces were routed to be as short as possible. The DC supply voltage for sensors and chips were distributed in a balanced way to assure a consistent voltage is supplied to all chips. Special jumper wires were added to provide access to the current waveform to allow monitoring using probes. Test points in different locations were added to enhance board debugging and maintenance capabilities.

Fig. 4.11 shows the final layout for the inverter. The dimensions of the board are given as

(34.5 X 16.3) cm.

Figure 4.11: Printed circuit board final layout.

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4.7 Conclusion

A practical design and the hardware implementation of the converter were presented in this chapter. The transformer design parameters were detailed as the transformer is the heart of operation in the proposed topology. The effects of selected switch parameters were emphasized to ensure safe operation and efficiency. The sensors used in the printed circuit board were outlined with the signal processing details.

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SIMULATION AND EXPERIMENTAL RESULTS

5.1 Introduction

To verify the capability of the proposed HFC, the simulation results and the control strategy are presented initially, followed by the hardware set up test results, for the different modes of operation. The experimental results shall verify and clarify the practicality of the operation principle.

5.2 Simulation Results

Following the guidelines and constraints provided in Chapter 3 and using the system parameters in Table 3.1, the parameters of the converter were selected to ensure the capability of the HFC to work in the different modes or scenarios.

The selected parameters are shown in Table 5.1. The HFC is simulated using

Matlab Simulink® as shown in Fig. 5.1.

Figure 5.1: Block diagram of the simulated system in Matlab Simulink®.

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The converter maintains and follows the MPPT by controlling S1. Different MPPT algorithms are mentioned in the literature [32],[33] such as Hill Climbing or perturb and observe (P&O), incremental conductance and ripple correlation control (RCC). The P&O algorithm is a well-known algorithm, especially for its simplicity and its PV array independence [33]. The P&O algorithm can be implemeted with a reasonable response time to allow for system dynamics. Filters and time delays can be used to control the response of the algorithm. The flow chart of the P&O algorithm is shown in Fig. 5.2.

Table 5.1: The HFC parameters.

Parameter Value 9.17 mF 퐏퐕 퐏퐚퐧퐞퐥 퐂퐚퐩퐚퐜퐢퐭퐨퐫 푪풑풗 퐓퐫퐚퐧퐬퐟퐨퐫퐦퐞퐫 퐩퐫퐢퐦퐚퐫퐲 퐢퐧퐝퐮퐜퐭퐚퐧퐜퐞 28 µH 푳풑풓풊

5 풏 0.5 풎 퐆퐫퐢퐝 퐬퐢퐝퐞 퐟퐢퐥퐭퐞퐫 퐢퐧퐝퐮퐜퐭퐨퐫 11 mH

푳풇풊풍풕풆풓 퐆퐫퐢퐝 퐬퐢퐝퐞 퐟퐢퐥퐭퐞퐫 퐜퐚퐩퐜퐢퐭퐨퐫 1.1 µF

푪풇풊풍풕풆풓 6.68 mF 퐋퐄퐃 퐂퐚퐩퐚퐜퐢퐭퐨퐫 푪푳푬푫

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START

Input Vn,In

Pnew=Vn.In ∆ P =P_new-P_old

NO Yes Dir =-Dir Dir=Dir P > 0 D_new=D_old+ ∆ d D_new=D_old+ ∆ d

Figure 5.2: Flowchart of P&O MPPT algorithm.

Figure 5.3: I-V characteristic (top), P-V characteristic (bottom) of the modeled PV panel. The I-V and P-V characteristics of the modeled PV panel are shown in Fig. 5.3 for radiation levels of 1000 W/m2 and 500 W/m2. The LED Lighting system is simulated using different resistance values. This emulates different numbers of LED lamps simultaneously turned ON. The number of activated LED lamps depends on the user preferences and the architectural design of the targeted facility.

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Figure 5.4: MPPT algorithm command of the main switch S1 duty Cycle.

5.2.1 Grid Only Mode

As aforementioned in Chapter 3, all of the power will be exported to the grid from the PV in grid only mode. This mode or scenario shows the capability of the HFC to utilize the PV panel even when there is no load demand. The economic point of connecting to the grid can be achieved by selling the power to the utility, or by utilizing the utility grid as storage, using the net metering feature, where the consumer is charged or compensated for the net power consumed or produced.

The LED turned off condition is simulated using a very high resistance. The simulation results of this mode are shown in Figures 5.4 to 4.9. Fig. 5.4 shows the MPPT algorithm command for the duty cycle on S1. The duty cycle starts increasing until t=3 sec, then the P&O action around the MPP is observed; the duty cycle fluctuates around a fixed value, 0.46 precisely to maintain the MPP.

At the beginning, the system is left idle to charge the PV capacitor (Cpv). This period can be seen in Fig. 5.5, Fig. 5.6 and Fig. 5.7 as a step voltage on the PV panel, a high current drawn from the PV and a pulsed power to the peak respectively.

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Figure 5.5: PV panel voltage development Figure 5.6: PV Panel current development during MPPT. during MPPT.

Figure 5.7: Power sharing and distribution between the different ports in grid only mode.

The PV panel voltage and current are shown in Fig. 5.4 and Fig. 5.5 respectively, also here the effect of the P&O algorithm can be noticed as the voltage fluctuates around

35 V, while the current fluctuates around 3.5 A. Fig. 5.7 and Fig. 5.8 provide a complete picture of the simulation steps. At the start, the PV panel is left to charge 퐶푝푣 only. While the MPPT algorithm is increasing the duty cycle, the control is charging the LED voltage

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around the required voltage of 24 V. At around t=1s, the MPPT algorithm starts to increase the duty cycle, thus increasing the amount of power drawn from the PV panel and increasing the amount of power exported to the grid while maintaining the LED voltage.

Finally, Fig. 5.9 presents both the grid voltage and current at the MPP.

Figure 5.8: LED port output voltage during grid only mode.

Figure 5.9: Grid voltage and current during grid only mode.

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5.2.2 LED Only Mode

In this mode, all of the power will be directed to the LED port. The controller will prevent any power to be delivered to the grid side by commanding zero duty cycle to the

AC side switches. The LED only mode can be controlled in two different ways; the sinusoidal PWM or the regular constant PWM. The sinusoidal PWM method is chosen so that, the controller complexity is reduced and an easy transition between the modes can be achieved. For the full power delivered to the LED, the LED was simulated as a 4.8 Ω resistor at 24 V so that the total power is 120 W. This case assumes that the supplied LED lighting system is operating at full power.

For the LED only mode, the responses of the MPPT duty cycle, PV voltage and current are the same as those of the previous mode in Fig. 5.4, Fig. 5.5 and Fig. 5.6 respectively. This confirms that the MPPT algorithm works independently regardless of the modes of operation. Fig. 5.10 shows that the PV power is being delivered to the LED port only. The LED voltage profile in Fig. 5.11 shows that the LED voltage reaches the required voltage level at t=2.5 seconds. As predicted by Eqn. 3.9; the voltage ripple in the

LED port in this mode is around 1V.

Figure 5.10: Power sharing between the different ports in LED only mode.

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Figure 5.11: LED voltage during LED only mode. 5.2.3 Grid and LED Mode

In this scenario, the power of the PV panel is used to support the LED as the main load while any excess power shall be directed to the grid. In this simulation the LED was simulated as a 9.6 Ω resistance to resemble less usage of the lighting system supplied. The objective is to maintain 60 W of power on the 24 V LED.

The MPPT duty cycle control, PV panel voltage and current responses remain the same as in Fig. 5.4, Fig. 5.5 and Fig. 5.6 respectively as expected.

The main difference in this mode is the power split between the LED and the grid.

LED support is the main objective in this mode as shown in Fig. 5.12. While the MPP is being tracked, all of the power is directed to the LED port until t=1.75s when the extracted power from the PV exceeds the LED requirement. Fig. 5.13 presents the LED voltage kept at 24 V. Fig. 5.14 shows the grid voltage and current injected into the grid, so the excess power is directed towards the grid.

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Figure 5.12: Power sharing and distribution between the different ports in grid and LED mode.

Figure 5.13: LED voltage during grid and LED mode.

Figure 5.14: Grid voltage and current during grid and LED mode.

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Figure 5.15: Duty cycle control for AC side switches in grid and LED mode.

The duty cycle control command for the AC side switches (D2) is shown in Fig.

5.15. It is worth mentioning that the relation between D2 and the LED voltage is inversely proportional. When D2 is higher the LED voltage will get lower unless more power is injected from the PV side and vice versa.

5.2.4 Grid Support Mode

The PV power may not be enough to sustain the LED voltage and load due to many reasons, such as weak irradiance, shading or temperature changes. As a result, the PV panel characteristics can change as shown in Fig. 5.3; which shows the impact of the solar radiation change from 1000 to 500 W/m2. In this case the grid will be used as a supplemental source for the power deficit.

This scenario starts normally allowing the MPPT algorithm to track the MPP. At t=1.7s, the controller starts working to sustain a 24 V on the LED load by taking power from the AC side. Fig. 5.16 and Fig. 5.17 show the MPPT duty cycle and AC side switches duty cycle respectively.

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Figure 5.16: MPPT duty cycle for S1 at Figure 5.17: Duty cycle control to the AC grid support mode. side switches at grid support mode.

Figure 5.18: PV panel voltage Figure 5.19: PV panel current development during MPPT in weak development during MPPT in weak irradiance and grid support mode. irradiance and grid support mode.

The PV voltage in this mode is shown in Fig. 5.18. The effect of weak irradiance is mostly visible in Fig. 5.19 where the maximum current is around 1.8 A, unlike the regular

3.5 A of the rated conditions. The power sharing is shown in Fig. 5.20, as stated, at t=1.7s the controller starts to maintain the 24 V on the LED side, so the deficit in power is taken from the grid. Around t=2.4s, the controller reaches the targeted voltage of the LED as seen in Fig. 5.21. The current drawn from the grid is 180 degrees phase shifted as power is being consumed from the grid as shown in Fig. 5.22.

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Figure 5.20: Power sharing and distribution between the different ports in grid support mode.

Figure 5.21: LED voltage during grid support mode.

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Figure 5.22: Grid voltage and current during grid and LED mode.

5.3 Experimental Results

The designed PCB was populated and installed as shown in Fig. 5.23. The four modes of operation were tested in the setup. The transformer primary inductance value L was designed to be 28 µH. After the final assembly and testing of the transformer the inductance was found to be around 48 µH. That limits the amount of power the system can process to 54 W.

An AMETEK TerraSAS photovoltaic simulator Model ETS80 is used to simulate the PV panel as shown in Fig. 5.24. This model has the capabilities to simulate PV panel and arrays up to 80 V and 10 A. Different irradiance and temperature profiles can be programmed as well as shading effects can also be simulated [34]. The LED was simulated using a variable resistor as shown in Fig. 5.25.

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Figure 5.23: Experimental setup of the proposed HFC.

Figure 5.24: TerraSAS photovoltaic simulator model Figure 5.25: Variable ETS80. resistor as LED.

5.3.1 Grid Only Mode

In this mode, almost all of the power coming from the PV panel was directed to the grid side except for a small amount to sustain a 24 V on the LED port on 250 Ω resistor.

The PV panel generated 53 W at 1.23 A and 43.25 V. The power delivered to the LED port was 2.3 W to sustain the 24 V for the LED port. While the rest of the power was pushed into the grid.

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The currents through the transformer windings in the three main stages of operation are shown in Fig. 5.26. Firstly, The PV is charging the transformer’s primary inductance.

Secondly, some of the power is discharged or delivered to the grid side and finally the remaining energy is discharged into the LED port. Fig. 5.26 (a) shows the transformer current in a half 60 Hz cycle, while (b) shows a zoomed version for three switching cycles.

The grid voltage and the current injected into the grid are shown in Fig. 5.27. The current was 0.3A RMS so the total power injected to the grid was 36 W. That gives the HFC a total efficiency of around 72.3% overall.

(a)

(b) Figure 5.26: (a) Transformer currents in a half 60 Hz cycle in grid only mode. (b) Zoomed at the peak of (a).

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Figure 5.27: Grid voltage and current of the experimental setup in grid only mode. 5.3.2 LED Only Mode

All of the power is directed to the LED port. A 26 V was maintained across 17 Ω resistance, that is a 39.8 W, while the PV was producing 52.8 W at 43.25 V, 1.23 A. This resulted in a total efficiency of 75%. Fig. 5.28 shows the current waveforms of the primary and the tertiary sides. Fig. 5.29 shows a zoomed view of Fig. 5.28, Fig. 5.29 is similar to the theoretical waveforms in Fig. 3.3.

5.3.3 Grid and LED mode

In this mode the LED resistance was reduced to 30 Ω. The power extracted from the PV is 53.5 W at 43.22 V and 1.24 A. The LED is maintained at 24.1 V using 19.3 W.

The rest of the power was pushed into the grid. Fig. 5.30 shows the transformer windings current. A zoom on the peak of Fig. 5.30 is shown in Fig.5.31. Fig. 5.32 shows the grid voltage and current. The grid RMS current is 0.17A so the total power pushed into the grid is 20.4 W, leading to a 74% efficiency.

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Figure 5.28: Transformer currents in a half 60 Hz cycle in LED only mode.

Figure 5.29: Zoom into the peaks of transformer currents of Fig. 5.28.

Figure 5.30: Transformer currents in a half 60 Hz cycle in grid and LED mode.

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Figure 5.31: Zoom into the peaks of transformer currents of Fig. 5.30.

Figure 5.32: Grid voltage and current of experimental setup in grid and LED mode.

The similarity between Fig. 5.31 and Fig 3.10 prove the capability of splitting the power between the Grid and the LED.

5.3.4 Grid Support mode

In this scenario, a 25 V is maintained across the LED resistor of 17 Ω; the PV panel was providing 25.4 W while the rest was coming from the grid side current. Fig. 5.33 shows the transformer windings current. A zoom on the peak of Fig. 5.33(a) is shown in Fig.

5.33(b).

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(a)

(b) Figure 5.33: (a) Transformer currents in a half 60 Hz cycle in the grid support mode. (b) Zoomed at the peak of (a). Fig. 5.33 is alike to Fig. 3.12 in terms of the general current profile. That shows the good agreement between the theoretical concept and the experimental result in the grid support mode. The grid voltage and current are shown in Fig. 5.34. From Fig. 5.34 the grid

RMS current is 0.226 A; that is, the power from the grid is about 26.4 W. So the efficiency in this mode is about 71%.

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Figure 5.34: Grid voltage and current in grid support mode.

5.4 Conclusion

The simulation results of the proposed HFC were provided in this chapter. The simulation results were verified experimentally using the designed prototype. Both the simulation and the experimental results, prove the capability of the converter to work on the different modes of operation. That ensures the main objective of the converter – continuous maximum utilization of the PV panel power regardless of the amount the PV panel can support while sustaining the load requirements.

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CONCLUSION AND FUTURE WORK

6.1 Conclusion

A hybrid flyback converter (HFC) topology is proposed in this thesis. The HFC interfaces three ports; solar PV port, LED lights port and the utility grid port. The proposed

HFC does not only assure MPPT of the solar panel at all times but also sustains the load requirements by compensating for any power deficit from the grid or by exporting the excess power to the AC grid. The analysis provides detailed information about the required conditions for such converter topology to work in each operating mode and to achieve the aimed objective.

The proposed HFC relies on the transformer magnetization inductance as the common point for energy exchange so special care has to be taken for the design and selection of the transformer. An important fact to be considered is that all of the energy transfer is achieved by allowing a single port to make use of the transformer magnetic circuit at each instant of time; this imposes high stress on the transformer so the core losses must be accounted for.

The ability of this topology to handle higher power levels pushes the design to hard limits. These limits include designing a transformer of a very low magnetization inductance while keeping the leakage inductance relatively low. Achieving low magnetization inductance is usually linked to the transformer working closely to the saturation region, also the current stress on the primary switch will increase significantly. Additionally, a high current ripple will be drawn from the bulk capacitor on the PV side, and this will lead

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to higher losses on the capacitor. Furthermore, higher current and voltage stresses are expected in some of the components used in the HFC.

The operation of this HFC requires a special starting condition or procedure to achieve the required voltage on the LED side, which ensures that all of the conditions are satisfied and all operating modes are possible. The proposed topology has been verified through simulations. An experimental setup and PCB has been built for the converter. The

HFC was tested for specified operating conditions and achieved an overall efficiency in the range of 71 to 75 %.

6.2 Future Work

The possible future investigation that can be extrapolated from this work include:  The inverter was designed to work on 20 kHz, higher frequency operation

can be investigated to see the impact on the overall system.

 The investigation of BCM operation can come at a benefit due to the usage

of lower rating switches.

 The HFC can be utilized as a common inverter in a hybrid microgrid by

utilizing the LED port as the DC microgrid side.

 In the hybrid microgrid, an additional switch can be added to facilitate the

capability of taking power from the DC side to another part of the converter.

 The stability of the converter has to be studied for a better selection of the

control parameters.

 The output current of the inverter to the AC side suffered from a noticeable

ripple at the resonance frequency of the output filter, which can be

investigated to have lower THD.

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