Reconfigurable Electromagnetics Through Metamaterials

International Journal of Antennas and Propagation

Reconfigurable Electromagnetics through Metamaterials

Guest Editors: Giacomo Oliveri, Douglas Werner, Filiberto Bilotti, and Christophe Craeye Reconfigurable Electromagnetics through Metamaterials International Journal of Antennas and Propagation

Reconfigurable Electromagnetics through Metamaterials

This is a special issue published in “International Journal of Antennas and Propagation.” All articles are open access articles distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Editorial Board

Mohammod Ali, USA Se-Yun Kim, Republic of Korea Matteo Pastorino, Italy Charles Bunting, USA Ahmed A. Kishk, Canada Massimiliano Pieraccini, Italy Felipe Catedra,´ Spain Selvan T. Krishnasamy, India Sembiam R. Rengarajan, USA Dau-Chyrh Chang, Taiwan Ju-Hong Lee, Taiwan Ahmad Safaai-Jazi, USA Deb Chatterjee, USA Byungje Lee, Republic of Korea Safieddin Safavi-Naeini, Canada Z. N. Chen, Singapore Joshua Le-Wei Li, China Magdalena Salazar-Palma, Spain Michael Yan Wah Chia, Singapore J.S. Mandeep, Malaysia Stefano Selleri, Italy Shyh-Jong Chung, Taiwan Atsushi Mase, Japan Zhongxiang Shen, Singapore Lorenzo Crocco, Italy Giuseppe Mazzarella, Italy John J. Shynk, USA TayebA.Denidni,Canada C. F. Mecklenbrauker,¨ Austria Seong-Youp Suh, USA Francisco Falcone, Spain Mark Mirotznik, USA Parveen Wahid, USA Miguel Ferrando Bataller, Spain A. S. Mohan, Australia Yuanxun Ethan Wang, USA Vincenzo Galdi, Italy P. Mo h a n a n , In d i a Tat Soon Yeo, Singapore Wei Hong, China Pavel Nikitin, USA Young Jo ong Yo on, Korea Tamer S. Ibrahim, USA Symeon Nikolaou, Cyprus Jong-Won Yu, Republic of Korea Nemai Karmakar, Australia A. D. Panagopoulos, Greece Anping Zhao, China Contents

Reconfigurable Electromagnetics through Metamaterials, Giacomo Oliveri, Douglas Werner, Filiberto Bilotti, and Christophe Craeye Volume2014,ArticleID215394,2pages

Bandwidth Reconfigurable Metamaterial Arrays, Nathanael J. Smith, Dimitris Papantonis, and John L. Volakis Volume2014,ArticleID397576,17pages

Reconfigurable and Tunable Metamaterials: A Review of the Theory and Applications, Jeremiah P. Turpin, Jeremy A. Bossard, Kenneth L. Morgan, Douglas H. Werner, and Pingjuan L. Werner Volume 2014, Article ID 429837, 18 pages

MEMS-Reconfigurable Metamaterials and Antenna Applications,TomislavDebogovicand Julien Perruisseau-Carrier Volume 2014, Article ID 138138, 8 pages

Switchable Electromagnetic Bandgap Surface Wave Antenna,QiangBai,KennethL.Ford, and Richard J. Langley Volume2014,ArticleID693852,7pages

Tunable Plasmonic and Hyperbolic Metamaterials Based on Enhanced Nonlinear Response, Christos Argyropoulos, Francesco Monticone, Nasim Mohammadi Estakhri, and Andrea Alu` Volume 2014, Article ID 532634, 11 pages

Voltage Controlled Intertwined Spiral Arrays for Reconfigurable Metasurfaces, A. Vallecchi, R.J.Langley,andA.G.Schuchinsky Volume 2014, Article ID 171637, 10 pages

Mechanically Reconfigurable Microstrip Lines Loaded with Stepped Impedance Resonators and Potential Applications,J.NaquiandF.Mart´ın Volume 2014, Article ID 346838, 8 pages Hindawi Publishing Corporation International Journal of Antennas and Propagation Volume 2014, Article ID 215394, 2 pages http://dx.doi.org/10.1155/2014/215394

Editorial Reconfigurable Electromagnetics through Metamaterials

Giacomo Oliveri,1 Douglas Werner,2 Filiberto Bilotti,3 and Christophe Craeye4

1 ELEDIA Research Center@DISI, University of Trento, 38123 Trento, Italy 2 Department of Electrical Engineering, The Pennsylvania State University, University Park, PA 16802, USA 3 Department of Engineering, Roma Tre University, 00146 Rome, Italy 4 Ecole Polytechnique de Louvain, Universite´ catholique de Louvain, 1348 Louvain-la-Neuve, Belgium

Correspondence should be addressed to Giacomo Oliveri; [email protected]

Received 17 April 2014; Accepted 17 April 2014; Published 15 June 2014

Copyright © 2014 Giacomo Oliveri et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

In recent years there has been a growing interest in the technological, methodological, and experimental viewpoint, development of artificial materials, in particular in meta- and it highlights some of the potentialities and future trends materials, engineered in order to exhibit electromagnetic in this research area. properties that do not occur in nature. The high degree More in detail, the paper by J. P. Turpin et al. presents a of enthusiasm over electromagnetic metamaterials stems comprehensive review of the developments of reconfigurable from their ability to manipulate and channel electromag- andtunablemetamaterialsaswellasoftheapplicationsof netic waves in unprecedented ways, which is derived from such technology. A survey of several different tuning meth- their subwavelength structure/granularity, rather than being ods, including circuit-enabled tuning, geometrical tuning, based solely on their constitutive materials. These features and material tuning, is presented, and their applicability over of electromagnetic metamaterials promise a path toward different frequency bands (from RF to optical frequencies) completely new devices with properties and functionality is discussed. A set of applications is also reported that com- not possible with currently available technologies. This is prises tunable filters and antennas, devices with adjustable specifically true whenever reconfigurability is of interest scattering parameters, tunable GRIN lenses, and antennas (e.g., for nonconventional and transformative applications with reconfigurable propagation features. Additionally, some in telecommunications, medicine, and security). Indeed, due of the open challenges towards their applicability in practical to their intrinsic structure, electromagnetic metamaterials devices are discussed. represent a key tool to implement the “reconfigurable electro- Areviewofsomeoftherecentcontributionstomaterials magnetics” paradigm that is to design and fabricate devices where the dynamic control is enabled by micro-electro- which can be controlled through a change of the physical mechanical systems (MEMS) technology is presented in the properties of their constitutive materials at a subwavelength paperbyT.DebogovicandJ.Perruisseau-Carrier.More scale. specifically, efficient reconfigurable phase shifters and leaky- In this framework, the special issue is aimed at reviewing wave antennas (LWA) based on reconfigurable composite the most recent advances of metamaterials as an enabling right/left-handed transmission lines are illustrated. More- technology for the design and realization of reconfigurable over, very low loss metasurface designs with reconfigurable devices at RF and THz/optical frequencies, and it includes reflection properties are discussed along with their applica- 7 papers representing the state-of-the-art work being car- tion in reflectarrays and partially reflective surfaces. Fabrica- ried out in this topic area by some of the top university tion and experimental validation of the presented devices in research labs around the world. This collection provides a theX-andKu-bandsarealsoreported. comprehensive overview of some of the most interesting N. J. Smith et al. present a conformal wideband meta- advancements in reconfigurable metamaterials from the material array achieving a 10 : 1 continuous bandwidth. To 2 International Journal of Antennas and Propagation accomplish this goal, a wideband Marchand-type balun the authors for their patience with us and with the review spanning the bandwidth from 280 MHz to 2800 MHz was process. We hope that you will find this special issue on designed and measured; its reconfiguration capabilities are the subject of reconfigurable electromagnetics interesting. obtained by means of circuit changes in the balanced The work reported in these manuscripts demonstrates that feed integrated with the wideband metamaterial array. The such a topic represents an extremely active interdisciplinary potentialities of the metamaterial array’s reconfiguration are research field with the promise of significant scientific and demonstrated through five example bandpass and band- industrial opportunities. rejection responses. ThepaperbyJ.NaquiandF.Martinisconcernedwith Giacomo Oliveri the exploitation of microstrip transmission lines loaded with Douglas Werner stepped impedance resonators (SIRs) etched on top of a signal Filiberto Bilotti strip to achieve mechanically reconfigurable metamaterials. Christophe Craeye More in detail, it is shown that the notch frequency and depth of the transmission line can be mechanically controlled by acting on the symmetry of the transmission line itself. Such a property is then exploited for the implementation of sensors and electronic barcodes. An innovative surface wave antenna is presented in the paper by Q. Bai et al. More in detail, the novel low-profile switchable antenna is based on band gap materials that can support both surface waves and normal modes of communication. The techniques for generating a dual mode switchable antenna are reported, and the performance of the antenna is investigated. The surface wave communication mode is also assessed by using an EBG to couple antennas/sensors together around the body. The paper by A. Vallecchi et al. presents single and dual polarisation operation arrays based on reconfigurable bistate metasurfaces composed of interwoven spiral arrays with embedded PIN diodes. More specifically, the array response between transmission and reflection modes at the specified frequencies is changed by means of the PIN diodes, which are controlled through the DC bias supplied by the spiral conductors forming the metasurface. A set of simulation results is reported to illustrate the good isolation between transmission and reflection states, as well as the excellent angular and polarisation stability of the proposed active metasurfaces over a large fractional bandwidth. Finally, the paper by C. Argyropoulos et al. discusses several tunable and reconfigurable designs of linear and nonlinear plasmonic and hyperbolic metamaterials. More in detail, rich scattering features of multilayered composite nanoparticles are demonstrated; they include exotic scattering signatures that combine multiple dipolar Fano resonances and electromagnetic induced transparency (EIT) features. Moreover, the exploitation of nonlinear hyperbolic metamaterial designs is addressed to realize tunable positive-to- negative refraction at the same frequency. Several devices areenvisioned,basedontheproposedtunablemetama- terials, including ultrafast reconfigurable imaging systems, tunable sensors, novel nanotag designs, and efficient all- optical switches and memories.

Acknowledgments WewouldliketothanktheInternationalJournalofAntennas and Propagation for the opportunity to serve as Guest Editors of this special issue. Moreover, we would like to thank all of Hindawi Publishing Corporation International Journal of Antennas and Propagation Volume 2014, Article ID 397576, 17 pages http://dx.doi.org/10.1155/2014/397576

Research Article Bandwidth Reconfigurable Metamaterial Arrays

Nathanael J. Smith, Dimitris Papantonis, and John L. Volakis

ElectroScience Laboratory, Department of Electrical and Computer Engineering, The Ohio State University, 1330 Kinnear Road, Columbus, OH 43210, USA Correspondence should be addressed to Nathanael J. Smith; [email protected]

Received 2 December 2013; Accepted 1 March 2014; Published 12 June 2014

Academic Editor: Douglas H. Werner

Copyright © 2014 Nathanael J. Smith et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Metamaterial structures provide innovative ways to manipulate electromagnetic wave responses to realize new applications. This paper presents a conformal wideband metamaterial array that achieves as much as 10 : 1 continuous bandwidth. This was done by using interelement coupling to concurrently achieve significant wave slow-down and cancel the inductance stemming from the ground plane. The corresponding equivalent circuit of the resulting array is the same as that of classic metamaterial structures. In this paper, we present a wideband Marchand-type balun with validation measurements demonstrating the metamaterial (MTM) array’s bandwidth from 280 MHz to 2800 MHz. Bandwidth reconfiguration of this class of array is then demonstrated achieving a variety of band-pass or band-rejection responses within its original bandwidth. In contrast with previous bandwidth and frequency response reconfigurations, our approach does not change the aperture’s or ground plane’s geometry, nor does it introduce external filtering structures. Instead, the new responses are realized by making simple circuit changes into the balanced feed integrated with the wideband MTM array. A variety of circuit changes can be employed using MEMS switches or variable lumped loads within the feed and 5 example band-pass and band-rejection responses are presented. These demonstrate the potential of the MTM array’s reconfiguration to address a variety of responses.

1. Introduction of dielectrics, magnetic materials, conductors, or lumped circuit elements also suffer from narrow bandwidth [11–18]. Engineered materials, also referred to as metamaterials Bandwidth reconfiguration is addressed using tunable leaky (MTMs) [1], have been of strong interest to the electromag- wave antennas in [19–23]. However, these approaches employ netics and optics communities. The last decade has provided varactors or external field biasing with ferrite substrates a large body of papers on realizing metamaterials to exploit resulting in narrow instantaneous bandwidths. their novel phenomena for all sorts of applications from A popular approach to achieve bandwidth or frequency antennas to radio frequency (RF) filters, frequency selective reconfiguration is that of using switches to alter the geome- surfaces, novel optical devices, and terahertz. The special try/current flow of the antenna itself [24, 25]. This approach October 2011 IEEE Proceedings issue [1–6] provides a large relies on many different technologies including RF-MEMS cross section of the theory and applications of metamaterials [26–32], pin diodes [33, 34], varactors [35–37], and optical as well as their use in achieving miniaturization, wave slow- switches [38, 39].Theuseoffrequencyselectivesurfacesfor down, ultrathin ground planes, bandwidth control, cloaking, reconfiguration and/or enhancing antenna bandwidth has and other special phenomena using optical nanocircuits and also been pursued [40]. Recently, microfluidic liquid metals other MTM techniques. andplasmahavebeenusedtoreconfigureMTMsfortunable Typically, metamaterial and other miniaturization tech- sensors and resonators [41–44] implying a strong potential niques [7] result in narrow bandwidths. This is also true when for their application to antennas. more traditional techniques such as meandering/shaping and While there is a large body of reconfigurable antennas lumped loading [8–10] are used. Miniaturizations based on (including MTM antennas), their instantaneous bandwidth metamaterials using combinations of periodic arrangements continues to be small. Specifically, during the past decade 2 International Journal of Antennas and Propagation most efforts on reconfiguration focused on extending the TCDA-IB ∙ 7.35:1 bandwidth antenna bandwidth or in shifting the aperture’s frequency 4.5 ∘ ∙ >2x size reduction 70 ∙> response using switches or piezoelectric materials. By con- 4 5x weight reduction trast, herewith we propose a reconfiguration approach that 60∘ ∙>10x cost reduction 3.5 begins with a wideband metamaterial aperture [45, 46]. TCDA-IB ∘ Because of this, the bandwidth, defined by1 ( )(where𝜔max 3 45 𝜔 and min arethemaxandminfrequencyofoperationresp.) 2.5 A (𝜔 −𝜔 ) P 2 𝐵= max min , (1) 1.5 √𝜔max𝜔min 1 for each reconfiguration is limited only by that of the original 0.5 aperture. An essential aspect of this aperture [46]isits 0 conformal nature, simplicity (planar array of dipoles or 0123456 bowtie elements), relatively small thickness, and wideband k h Marchand balun. It is referred to as the tightly coupled dipole 0 array (TCDA) [47–49] and when integrated with a wideband Fund. limit (constant pol.) BAVA balun, we will note it as the TCDA-IB array [49–51]. Fund. limit (arbitrary pol.) Multilayer/lossy It is important to note that the TCDA array and its TCDA-IB Vivaldi feed variations, even though conformal in nature, have been Patch ISPA shown to deliver instantaneous bandwidth that is comparable TCDA and even greater than the nonconformal Vivaldi arrays [52, Figure 1: Performance factors of selected dual-pol. arrays, alongside 53]. A version of the Vivaldi arrays such as the BAVA [54] fundamental limit from [56]. Here, 𝑃𝐴 is a factor proportional to does have much improved performance and achieved an the antenna’s bandwidth, 𝐵, with penalizations due to mismatches 𝜆 /2 𝜃 Γ impressive10:1bandwidthwitha hi thick profile at when measured or computed at the scan angle max. max refers to broadside. However, this decade bandwidth comes at the the maximum reflection coefficient at the feed across the bandwidth. expense of matching as the array is scanned (VSWR = 2.75 Note that the symbol “O” refers to broadside performance and ∘ at broadside, and VSWR < 4at𝜃 =45scan). Another symbol “X” refers to the bandwidth performance at the scanning recent BAVA design [52] delivers 1.8–8 GHz at broadside angle of the array as indicated in the figure (see58 [ ] for details). ∘ (1.875–7.5GHz at 45 scan) with 4.44 : 1 and 4 : 1 bandwidths, Qualitatively, when the plot symbol is closer to the theoretical optimal line, the antenna’s performance is also closer to having respectively, when VSWR < 2. optimal performance. When considering conformal and/or planar wideband arrays,amajorchallengeisthesuppressionofthecommon mode that may appear in the feed structure of these arrays. This requires a balanced feed that retains its performance We note that the optimal values for 𝑃𝐴 at broadside are across the entire bandwidth. Various feeding arrangements [57, 58] were employed in [47, 48, 55, 56] to suppress the mode and ensure a balanced feed. However, in all these cases, retaining 𝑃𝐴 ≤𝜋𝜇0𝑘0ℎ, constant polarization, a low VSWR implied narrower bandwidths that ranged from (4) 2:1uptoatmost5:1.Thatis,theactualdeliveredbandwidth 𝑃𝐴 ≤2𝜋𝜇0𝑘0ℎ, arbitrary polarization. of wideband arrays depends on several parameters: isolated array bandwidth, thickness, feed impedance, common mode Figure 1 plots 𝑃𝐴 as a function of thickness, ℎ,forvarious suppression, maximum acceptable VSWR, losses, and achiev- wideband arrays, including the MTM TCDA-IB array. It is able beam scanning range (𝜃max). With this in mind, Doane seen that variations of the TCDA-IB have larger 𝑃𝐴 values et al. [57, 58]usedthefollowingmetrictocomparewideband for the same thickness (the multilayer/lossy array in Figure 2 arrays: refers to the arrays in [49, 59] as they include losses in 󵄨 󵄨 the substrate). Specifically, the TCDA-IB delivers 6.3 : 1at 𝐵 (1/ 󵄨Γ 󵄨) ∘ ∘ log 󵄨 max󵄨 broadside, 7.2 : 1 at 30 ,and6.1:1at60 with VSWRs 2.25, 2.9, 𝑃𝐴 = , lossless arrays, (2) cos (𝜃max) and 3.9, respectively. For these TCDA arrays, the superstrate makes them just about 𝜆 /2 thick, where 𝜆 refers to the 𝐵 (1 − 𝜂 ) hi hi log min wavelength at the highest operational frequency. However, 𝑃𝐴 = , lossy arrays. (3) 2 cos (𝜃max) if reflections from the ground plane are partially absorbed, much larger bandwidths can be achieved. As an example, In this expression, 𝐵 is the bandwidth noted above, Γmax theTCDAarraysin[49–51] achieved 21 : 1 bandwidth with refers to the max reflection coefficient at the feed within 70% efficiency. Indeed, this is a small loss in gain of only theapertureband,𝜂min is the efficiency which accounts 1.5 dB, but the bandwidth is nearly tripled as compared to 2 for various losses and includes the term (1 − |Γmax| ),and the lossless case. The ISPA consists of interweaved spirals “log” implies the natural logarithm. As such, the array’s producing several different polarizations across its UWB and performance is penalized for larger VSWRs or higher losses. thus is measured against (3)metricfor𝑃𝐴. International Journal of Antennas and Propagation 3

Balanced port introduce constructive addition of the fields directly radiated at antenna from the array with those generated by the ground plane. terminal 8×8array with tightly coupled elements Alternatively, loss within the substrate [63]canbeintroduced Dipoles/bowties to suppress ground plane reflections. Electromagnetic band Balun gap (EBG) ground planes [64, 65]canalsobeintroducedto h reduce array thickness, typically at the expense of bandwidth. In contrast to modifying or suppressing ground plane Balun detail reflections, Munk et al. [66](seealso[67]) proposed cance- Microstrip lation of the ground plane’s inductance by introducing strong Ground feed line capacitance between the dipoles. This concept is depicted Ground in Figure 3, showing an array of closely placed dipoles, Possible frequency plane Unbalanced selective surface capacitivelycoupledtoeachother.Theequivalentcircuit feeding port (FSS) for additional of the unit cell for this array configuration is represented bandwidth and in Figure 3(b) and consists of shunt inductance (𝐿2)and polarization control capacitance (𝐶2) along with a series capacitance (𝐶1)and Figure 2: Geometrical configuration of the tightly coupled array inductance (𝐿1). A resistor (𝑅) is included to account for discussed in [48, 49]. the array’s radiation resistance. We note that the shunt inductance (𝐿2) representing the effect of the ground plane and the series capacitance (𝐶1) representing the interdipole coupling are the most critical components of the circuit. Given that recent MTM arrays can achieve large band- To achieve large bandwidths, it is necessary for the capac- width, these apertures can serve as common front-end itance 𝐶1 to cancel the effect of the inductance 𝐿2 across hardware to accommodate a variety of back-end receivers a large bandwidth. Specifically, 𝐶1 must vary as a function operating from VHF up to several GHz. Concurrently, there of frequency to cancel the ground plane inductance of the 𝑍 =𝑗2𝑅 (2𝜋ℎ/𝜆) 𝑅 is strong interest to control the array bandwidth for software ground plane 1+ 𝐴0 tan ,where 𝐴0 refers radio applications, specifically, ensuring that the aperture to the characteristic impedance of the medium below the 𝑍 only operates within the band of interest. As such, it is ground plane. That is, referring to Figure 3(c), 1+ must 𝑗𝑋 advantageous to reconfigure its operation or even reduce be approximately equal to the conjugate of 𝐴.Assuch, its large bandwidth, if not needed, to minimize background the input impedance seen by the array ports remains nearly noise into the receiver. In this paper, we provide a tech- real across the bandwidth. We remark that the equivalent niqueforreducingthebandwidthofwidebandarraysby circuit shown in Figure 3 isthesameasthatofmetamaterial reconfiguring its balanced feed placed between the array and structures and is the reason for referring to these arrays as groundplane.ThisisdoneusingswitchesandLCloadsto MTM structures. control the operation of the Marchand [60, 61]balunas Munk [45, 64] chose to realize the capacitive coupling displayed Figure 2. It is important to note that our approach between the dipoles by placing them close to each other departs from traditional methods that shift the bandwidth or by interweaving the adjacent dipoles. This is depicted in by integrating switches on the aperture. In the latter case, Figure 4 showingthataVSWRbandwidthof3:1isachieved. the switches change the aperture geometry or reconnect the More recently, Munk’s array was improved in bandwidth by as aperture elements to shift the operational bandwidth. Instead, much as 10 : 1 bandwidth or more using overlapping dipoles ourapproachonlyreconfiguresthebalunbehindthearray as depicted in Figure 5. This array and feed configuration is with a goal to limit its bandwidth or possibly forbid reception discussed in [49, 50]. over a section of the aperture’s original bandwidth. Summarizing, the TCDA array achieves wide bandwidth Below, we begin with a presentation of the MTMs by employing the following features. operational concept (Section 2). This is followed by the integrated balun-aperture design of the tightly coupled dipole (i) Interelement Coupling.Typically,arrayelementsareplaced array (TCDA-IB) and measured performance (Section 3). apart to minimize mutual coupling. However, in MTM The reconfiguration concepts and performance are presented arrays, coupling is not only desired but also enhanced and adjusted to increase array bandwidth. in Section 4. (ii) Much Smaller Array Elements.Thecoupledarrayconcept 2. Wideband Metamaterial Arrays is concurrently used for miniaturization. The highest frequency operation occurs when the feed-to-feed element sep- Wideband low profile UWB arrays are of interest for many aration becomes 𝜆o/2 sincemultiplelobeswillappearbeyond communication and radar functions. As already noted, a this frequency. As an example, to achieve 10 : 1 bandwidth, challenge in designing UWB arrays is that of maintaining low the feed-to-feed distance between dipole elements must be 𝜆 /20 profiles (small thickness) without reducing bandwidth. It is o or less at the lowest operating frequency. That is, the well known that the bandwidth of low profile apertures can be interleaving or overlapping of dipoles is not only essential to increased by using magnetic loading [62]tochangethephase cancelling the ground plane’s inductance but is also critical to of the ground plane’s reflection coefficient and therefore achieving large bandwidths. 4 International Journal of Antennas and Propagation

2RA0

Z1−

R C1 L ··· 1 ··· Coupling jXA capacitors d Z1+ C2 L2 h 2R h A0

Unit cell GP GP

(a) (b) (c) Figure 3: (a) General representation of the metamaterial (MTM) array with definition of unit cell, (b) equivalent circuit of MTM array unit cell, and (c) transmission line circuit representation of the array on a ground plane from [46].

VSWR deteriorates at low frequencies due to 4 ground plane shorting Feed

mm Ground plan improves

8.2 3 array VSWR at higher frequencies Unit cell VSWR 8.2 mm 2

0 0 5 10 15 20 Frequency (GHz) Free space 7 mm above GP (a) (b)

Figure 4: VSWR for a tightly coupled dipole array (TCDA) when radiating in free space or in presence of a metallic ground plane from [46].

(iii) Ground Plane (GP) Impedance. In standard arrays, the particularly when the array is scanning towards lower angles. GP is used as a reflecting surface with often undesirable The printed balun is therefore essential for good scanning results. However, in MTM arrays the GP impedance (which performance and for impedance matching, namely, for tran- is inductive) becomes part of the design and is tuned to sitioning the coax feed’s 50 Ω impedance to the array’s port increase array bandwidth. As depicted in Figures 4 and 5, impedance across the entire bandwidth. Typically, TCDA a simplistic interconnected dipole array exhibits much better array’s impedance is around 180 Ω. For the feed approach performance at higher frequencies (and over wide band- used in Figure 5, a single Wilkinson divider is used to feed width) when placed above a ground plane. The ground plane apairofdipoleswiththeirMarchandbaluns.Thus,theunit height typically becomes ℎ=𝜆o/7 at the highest operational cell becomes a pair of dipoles and not the typical single dipole. frequency and as small as ℎ=𝜆o/70 at the lowest operational Thisisdonetodoubletheimpedanceofthebalunitselffor frequency to achieve a 10 : 1 bandwidth. improved matching at the dipole terminals. The details of this concept are depicted in Figure 6 [50]. (iv) Balanced Feeds across the Entire Bandwidth.Theprinted Marchand-type balun shown in Figure 5 is essential to (v) Superstrate.ThesuperstrateoftheTCDAarray(see suppressing the common mode whose presence would Figure 5) plays an important role. It improves the MTM array generate traveling waves within the substrate. Such waves bandwidth by reducing the effective size of the array elements would destructively interfere with the MTM’s array radiation, andreducesthearrayimpedancetofacilitatematchingwith International Journal of Antennas and Propagation 5

dunitcell

Superstrate hsuperstrate Dipoles

height Array unit cell R-card showing integrated zRCard feed with Ground Wilkinson plane power driver Power hPD divider 50 Ω unbalanced port dipolefeedgap

dipoleedgegap squarelength dipolelength squarewidth

dipolewidth Overlapping dipole details

Figure 5: Illustration of the tightly coupled dipole array (TCDA) integrated with a balun to create the TCDA-IB array. Enhanced coupling between dipole is achieved using overlapping dipoles at the array’s surface. The Marchand-type balun is depicted to the left and a single feed is used to feed a pair of dipoles (creating the unit cell) to double the impedance from the Wilkinson divider for improved matching at the dipole terminals. the lower feed impedance. It further improves scanning, a Table 1: Values of the all parameters used for the construction of the concept that has been well exploited in arrays and creates a TCDA-IB array depicted in Figures 5–9. lensing effect to reduce substrate fields that may cause unde- Model parameter Value Model parameter Value sirable surface effects. As is usually the case, its permittivity cannot be too large so as not to negatively affect the array’s dunitcell 56.6 mm dipolelength 7.8 mm VSWR and bandwidth. height 103.1 mm diplewidth 26.7 mm hsuperstrate 33.3 mm squarelength 16.7 mm In the next section, we proceed to discuss a specific zRCard 51.6 mm squarewidth 26.7 mm TCDA-IB array, providing information on its geometry and hPD 60 mm dipoleedgegap 1.2 mm measured performance. This array is designed to operate extralength 3.3 mm subthickness 24 mil from 280 MHz to 2800 MHz using a ground plane that is only dgnd 4.1 mm dipolesubthickness 12 mil 𝜆o/10 in thickness at 2800 GHz and 𝜆o/100 thick at the lowest dshort 8.3 mm dipolegap 1.2 mm operational frequency. Subsequently, the balun of this class of arrays is reconfigured using switches and LC loads to realize zshort 45.9 mm subedge 10 mil smaller operational bandwidths or even band rejections. zopen 21.3 mm ddipolecon 2.5 mm dopen 2.0 mm zdipolecon 2.2 mm dtrace1 7 mil dtransition 3.1 mm 3. Design and Performance of dtrace2 11 mil subthickness 24 mil a 10 : 1 MTM Array dtrace3 13.2 mil dipolesubthickness 12 mil rvias 8.3 mil dipolegap 1.2 mm The successful TCDA-IB arrays in [50, 51]provideda dvias 13.3 mil methodology for designing conformal ultra wideband MTM arrays. In this section, we follow this procedure to design a lower frequency TCDA-IB array with 10 : 1 bandwidth operating across 280 MHz to 2800 MHz. This class of arrays parameters, we obtained the dimensions listed in Figure 8 will be used in the next section for reconfiguration. To and Table 1. improve the bandwidth and scanning, parameters of the The overall TCDA-IB structure was constructed using a TCDA-IB array, we pursued a reoptimization of its various total of sixteen printed circuit boards (PCBs) placed vertically parameters. The size of this redesigned array was 10.36 cm (as below the aperture as depicted in Figure 5.Outofthese compared to 6.22 cm in [51]) to achieve greater than 0 dB gain PCBs, 8 constituted the feed baluns (see Figure 7). The other from 280 MHz to 2800 MHz. Also, the array aperture size was 8wereassociatedwiththeprinteddipoles.Asillustrated increased to 54.33 cm × 54.33 cm. The details are depicted in Figure 10, the whole structure is then enclosed in a in Figures 7 and 8. After reoptimization of all geometrical frame with a Styrofoam molding (if needed) to support the 6 International Journal of Antennas and Propagation

377 Ω 13.69 square unit cell 10.36

Dipole arm Dipole arm dH Figure 7: Detail of the 8 PCB boards balun feeds with the 200 Ω Winkinson divider below the ground plane. All dimensions are in cm. Balun and transformer

50 Ω

(a)

dE/2dE/2

188 Ω 188 Ω rectangular rectangular 54.33 unit cell unit cell

dH 100 Ω 100 Ω

Balun Balun

100 Ω 100 Ω 54.33 50 Ω Figure 8: Dipole arrangement of the TCDA-IB array in Figure 6 (b) forming the top of the 8 × 8 unit cell array (16 × 16 bowtie dipoles). All dimensions are in cm. Figure 6: Impedance of array unit cell is proportional to the aspect ratio of 𝑑𝐸/𝑑𝐻. Splitting the cell vertically into two halves, the impedance is reduced by a factor of two. Combining the halves in parallel reduces the impedance by another factor of two (power was placed on top of the array PCB [68] for improving divider from 50 Ω to 100 Ω). (a) Square 377 Ω unit cell and (b) two efficiency and scanning. If needed, a resistive sheet canbe 188 Ω “half” unit cells from [48]. placed between the dipole array and the ground plane to improve bandwidth at the expense of aperture efficiency. For the case discussed in [49], the aperture efficiency is greater than 70%. It should be noted that the efficiency with the R- aperture. Photos at different stages of the assembly process card included is no worse than 70% and reaches its lowest for constructing the TCDA-IB are provided in Figure 10.As value when the ground plane thickness is around 𝜆/2. shown in these photos, the PCBs for the feeds are placed Measurements were conducted by exciting each array vertically with respect to the array plane. These feeding port separately while the other 63 ports were terminated in boards are then held in place by a set of metal rods placed just 50 Ω. The UEAEP method69 [ ] was then used to synthesize under the ground plane. The gaps between the feeding boards the gain and patterns at different scan angles. The cross pol arethenfilledwithStyrofoamtomakethestructurestable was also calculated using Ludwig’s 3rd definition70 [ ]. We and easy to handle. It is important to note that the Styrofoam remark that 2 identical versions of the TCDA-IB array were does not impact the array’s performance as its permittivity constructed and measured. These are referred to as Antenna is comparable to air. The dipole PCBs are placed on top and 1andAntenna2inthegaindatagiveninFigure 11.As parallel to the ground plane. It is important to note that the demonstrated, the measured 8 × 8 array measurements are dipole PCB boards include strategically placed slots within in complete agreement with calculations, delivering >0dB the bowtie dipoles for anchoring the balun feed ends (see realized gain across a 10 : 1 bandwidth. It is important to note parameter ddipolecon in Figure 9). A low loss polyethylene that the VSWR bandwidth can be defined to show greater superstrate 3.33 cm thick and having a relative dielectric bandwidths. For our design, the simulated VSWR is depicted constant of 𝜀𝑟 = 2 : 25, with a loss tangent of tan 𝛿 = 0.0007 in Figure 12 showing a VSWR < 2.5 from 200 MHz to about International Journal of Antennas and Propagation 7

ddipolecon dvias dopen zdipolecon zopen zshort

dipolesubthickness dshort dtransition rvias

dtrace 1

dipolegap dgnd

dtrace 2 extralenght

dtrace 3

Figure 9: Parameter definitions for the TCDA-IB array are shown in Figure 5. This figure provides a listing of the various geometrical parameters that define the integrated balun. Values for the shown parameters are given in Table 1.

Figure 10: Fabrication stages of the TCDA-IB array depicted in Figures 5–9. The lower right figure refers to the measurement setup in the Ohio State’s compact range. The feeding antenna during measurements is a linearly polarized horn as the array itself is also linearly polarized. Note that the array was measured in the presence of a metallic ground plane of 360 cm × 240cminsize. 8 International Journal of Antennas and Propagation

Measured versus simulated broadside gain 30 to the H-plane patterns. It is indeed impressive to observe the nearly perfect agreement between the measured and 20 simulated scan patterns. Moreover, the cross-polarization 10 levels are 20–30 dB below the copolarization curves. It is well known that the worst cross-polarization occurs 0 while scanning in the diagonal plane (D-plane). Therefore, Bandwidth 10.0 : 1 −10 our measurements in this plane were carefully completed. −20 Specifically, time gating was used during measurements to

Broadside gain (dB) gain Broadside remove diffractions from the ground plane edges. The mea- −30 sured patterns are depicted in Figure 15.Asinthecaseofthe 280 2 345 2 800 −40 , , co-pol measurements, we again observe distinct lobes while 0 500 1,000 1,500 2,000 2,500 3,000 scanning. Moreover, the cross-polarization levels remain Frequency (MHz) 20dBbelowthecopolarizedgainlevel. 1st antenna Having validated this TCDA-IB array design, we next 2nd antenna proceed to introduce methods for reconfiguring the band- 8×8simulation width of this class of arrays. Specifically, we introduce methods to reduce its bandwidth or to simply reject a defined Figure 11: Measured and simulated realized gain at broadside (co- set of frequencies across its wide operating bandwidth. pol/E-pol data) for the TCDA-IB array whose geometry is presented in Figures 5–10. The agreement among all 3 gain curves provides substantial confidence in the validity of the measured performance. Small fluctuations in the realized gain are attributed to the horn used 4. Bandwidth Reconfigurable for calibration during measurements. Metamaterial Array Bandwidth reconfiguration of the MTM array in Figures Infinite versus 8×8simulated VSWR 2 and 5 can be accomplished in one of three locations 5 outlined in Figure 16. Specifically, we can choose to (1) reconfigure the geometry of the dipoles across the aperture 4 using switches to control the interelement capacitance, (2) integrate filters or geometrically reconfigure the Marchand (3) 3 balun feed depicted more simplistically in Figure 2,and

VSWR reconfigure the ground plane’s location, impedance or by introducing a frequency selective surface (FSS) in place of the 2 resistive sheet depicted in Figure 5. Indeed, the possibilities for array bandwidth/impedance reconfiguration are rather 1 0 500 1000 1500 2000 2500 2800 extensive and have yet to be exploited for this class of arrays.WemayactuallyviewtheEBGgroundplanes[64] Frequency (MHz) and FSS insertions [63] as reconfigurations that belong to 8×8simulated VSWR the 3rd category noted above and in Figure 16.However,in Infinite VSWR the authors’ opinion, reconfiguration of the aperture or the × ground planes can be challenging to accomplish in practice Figure 12: VSWR comparison of the infinite versus 8 8 finite andmaynotofferasmuchflexibility.Specifically,switches TCDA-IB array defined in Figures 5–9. It is noted that the truncation effectsoftheedgeelementsofthe8× 8 array are responsible for the within the aperture are difficult integrate with the superstrate. higherVSWRassociatedwiththe8× 8array. In the case of ground plane reconfigurations, the number of parameters to control is limited and cumbersome to implement. By contrast, reconfiguration of the Marchand balun offers several control parameters. Among them are 2400 MHz. This implied a 12 : 1 bandwidth for the infinite the balun’s stripline impedance, short circuit, and open array. Because of the array’s optimization, this bandwidth is a circuit stublengths and impedances. Below, we consider these bit greater than that given in [51] which reaches a bandwidth options in more detail and provide reconfiguration examples. of 9.2 : 1 with a VSWR < 3. Of course, the finite 8 × 8arraywill To better assess the reconfiguration options, it is impor- have a larger VSWR due to truncation effects of the periphery tant to first cast the array aperture and its feeding configu- elements. Indeed, simulations of the 8 × 8arrayshownin ration into an equivalent circuit. The steps to doing so are Figure 12 demonstrate the larger VSWR. depicted in Figure 17.Indeed,thetransmissionlinerepresen- The beam scanning performance of the fabricated TCDA- tation at the top of Figure 17 is precise as it provides circuit IB array in Figure 10 was measured at 600 MHz, 1200 MHz, parameters to account for the aperture (dipole elements), 1800 MHz, and 2400 MHz. Figures 13 and 14 show the superstrate, substrate, ground plane, and balun. We note that scanned patterns at 0, ±15, ±30, and ±45 degrees for these thebalancedfeedcanbecharacterizedbythemicrostrip 𝑍 𝑍 frequencies. Specifically, Figure 13 shows the E-plane (polar- line impedance, 1,theopenstub’simpedance, OC,andthe 𝑍 ization parallel to the dipoles) patterns and Figure 14 refers short stub’s impedance, SC. The corresponding geometrical International Journal of Antennas and Propagation 9

0 0

−10 −10 −20 −20 −30 −30 −40 −40 −50

−50 −60 Normalized scanning patterns (dB) patterns scanning Normalized Normalized scanning patterns (dB) patterns scanning Normalized −70 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 Scanning angle (deg) Scanning angle (deg) (a) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg (b) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg from broadside at 2.4 GHz from broadside at 1.8 GHz

0 0

−10 −10

−20 −20

−30 −30

−40 −40

−50 −50 Normalized scanning patterns (dB) patterns scanning Normalized Normalized scanning patterns (dB) patterns scanning Normalized −60 −60

−70 −70 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 Scanning angle (deg) Scanning angle (deg)

Co polarization measured Co polarization measured Co polarization simulated Co polarization simulated Cross-polarization measured Cross-polarization measured (c) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg (d) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg from from broadside at 1.2 GHz broadside at 0.6 GHz Figure 13: E-plane scanning for the TCDA-IB array defined in Figures 5–9. Comparison of measured versus simulated patterns at various scan angles. The measured cross-polarization levels are also included.

𝑍 componentsarenotedbythecirclednumbers2,3,and4at As a first step in modifying OC we consider the sim- the top of Figure 17. We assert that modification and control plified equivalent circuit in Figure 19 implemented on a of these parameters allow for substantial and full control substrate using microstrip line traces as depicted in Figures 2 of the MTM array’s band-pass and band-rejection features. and 5. The goal in designing this wideband balun is to ensure 𝑍 𝑍 𝑍 However, in modifying the parameters OC and SC,itis that in is maintained close to the feedline’s characteristic important to concurrently keep in mind the fundamental impedance 𝑍1. For the Marchand balun, it is necessary that 𝑍 𝑍 𝑍 ≪𝑍 ≪𝑍 operation of the balun as depicted in Figure 18. Specifically, OC and SC satisfy the condition OC bal SC, 𝑍 the short circuit stub (closed loop in the geometry) ensures where bal is the impedance at the antenna terminals and 𝑍 that the common mode is suppressed using a symmetric must be ideally kept equal to the complex conjugate of TCDA 𝑍 geometry. Therefore, modification of SC canonlyberealized across the band of interest. In reconfiguring the MTM array’s 𝑍 𝑍 by making the loop or the stub shorter. However, both sides performance our goal is to ensure that bal matches TCDA of the loop need to be kept identical to ensure cancellation of only for the band of interest or is highly mismatched across 𝑍 the common mode. On the other hand, OC can be modified thebandrejectionofinterest. in a more arbitrary fashion. The possibilities are rather broad, To demonstrate MTM bandwidth reconfiguration, we 𝑍 and below we demonstrate that modification/control of OC proceed to use the equivalent circuit in Figure 19 and opti- allows for substantial control. mize its parameters within Agilent’s Advanced Design System 10 International Journal of Antennas and Propagation

0 0 −10 −10 −20 −20 −30 −30 −40 −40 −50

−50 −60 Normalized scanning patterns (dB) patterns scanning Normalized Normalized scanning patterns (dB) patterns scanning Normalized −60 −70 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 Scanning angle (deg) Scanning angle (deg) (a) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and (b) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg 0 deg from broadside at 2.4 GHz from broadside at 1.8 GHz

0 0

−10 −10

−20 −20

−30 −30

−40 −40

−50 −50

Figure 14: H-plane scanning for the TCDA-IB array defined in Figures 5–9. Comparison of the measured versus simulated patterns at various scan angles. The measured cross-polarization levels are also included.

(ADS). As a first step, we select the parameters as follows: State 2: 𝐶2 = 1.75 pF is the only load turned on using 𝑍 = 100 Ω 𝑍 =𝑍 =𝑍 = 188.5 Ω 𝐿 = Feed , 0 Sub Sup , dipole SW1 and a switch that activates the load. 2.2 𝐶 = 2.3 ,ℎ =60∘ ℎ =90∘ nH, coupling pF sup , sub ,character- State 3: 𝐿2 =2.5nHand𝐶2 =1.75pF,bothturnedon ∘ istic impedance of 𝑍 = 312 Ω, 𝑙 =58all defined (but not 𝐿1 and 𝐶1). SC SC ∘ at 2.5GHz. Also, we chose 𝑍1 =78Ωand 𝑙1 =66. State 4: only 𝐿1 =1.3nHand𝐶1 = 3 pF are activated. Using these parameters, we next proceeded to only modify 𝑍 State5:completelyabsentloadattheterminationof OC and observe the bandwidth of the MTM array for various choices. Specifically, as depicted in Figure 20, a load theopenstub(unconfigured). is inserted at the end of the open stub consisting of a series Thelaststate(State5)issimplythebroadbandMTMarray inductor (𝐿1)andcapacitor(𝐶1)andashunt(fromthestub as originally designed and shown in Figure 20.TheVSWR to the ground) inserted inductor (𝐿2)andcapacitor(𝐶2). responses for the other 4 states are depicted in Figure 21. For greater flexibility, a switch SW1 can be closed or open to We observe that States 1 and 2 correspond to narrow band- effectively short out the 𝐿1 and 𝐶1 loads. pass bandwidths at the edges of the original unconfigured Table 2 provides 5 different reconfiguration states of the VSWR response of the MTM array. State 3 refers to another 𝐿1, 𝐶1, 𝐿2,and𝐶2 and SW1 parameter choices. The chosen band-passresponseinthemiddleoftheMTM’sarrayoriginal states include the cases as follows. unconfigured bandwidth. Finally, State 4 shows a band- State 1: 𝐿2 =2.5nHistheonlyloadturnedonusing rejection response realized with 𝐿1 =1.3nHand𝐶1 =3pF SW1 and a switch that activates the load. attheterminalsoftheopenstubinthebalun. International Journal of Antennas and Propagation 11

0 0

−10 −10

−20 −20

−30 −30

−40 −40

−50 −50

−60 −60 Normalized scanning patterns (dB) patterns scanning Normalized Normalized scanning patterns (dB) patterns scanning Normalized −70 −70 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 Scanning angle (deg) Scanning angle (deg) (a) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg (b) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg from broadside at 2.4 GHz from broadside at 1.8 GHz

0 0

−10 −10

−20 −20

−30 −30

−40 −40

−50 −50

−60

Normalized scanning patterns (dB) patterns scanning Normalized −60 Normalized scanning patterns (dB) patterns scanning Normalized

−70 −70 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 −90 −75 −60 −45 −30 −15 0 15 30 45 60 75 90 Scanning angle (deg) Scanning angle (deg) Co polarization measured Co polarization measured Cross-polarization measured Cross-polarization measured (c) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg (d) Radiation pattern of the TCDA-IB array at ±45, ±30, ±15, and 0 deg from broadside at 2.4 GHz from broadside at 1.8 GHz Figure 15: Measured diagonal plane scanning for the TCDA-IB array defined in Figures 5–9. No simulation data is available due to demanding computational complexity as symmetry cannot be applied to ease the computational burden. The cross-polarization levels are also included.

Wideband element (spiral, bowtie, TCDA) High passBand-pass Low pass

(1) Balun

(2) Tunable filter

Microstrip (3) VSWR

Ground plane Coax unbalanced feed Un-configured Frequency (a) (b)

Figure 16: Bandwidth reconfiguration concepts: (a) tunable filtering integrated in the aperture (1), feed (balun) (2), or the ground plane (3); (b) representative bandwidth response due to reconfiguration. 12 International Journal of Antennas and Propagation

1-1 correspondence to the actual balanced feed (balun) to its equivalent circuit representation Z0 5 5 Z superstrate

Z0 Ground plane Z0 2 3 4 4 Ground plane 3 2

Ground plane 3 Open stub 11 4 Short stub

Ground plane

(a)

Z0 Z =Z open OC

Inter-element capacitance Z Wilkinson divider Dipole 𝜀 superstrate inductance r

C Zfeed Z1 par Z0

Z reconfigurable Microstrip ground plane Z feed 0

Z =Z short SC

(b)

Figure 17: Equivalent circuit representation of the TCDA-IB array in Figure 5. The top graphic provides a one to one correspondence between the Marchand balun in Figure 2 and its equivalent transmission line circuit representation. The bottom circuit is a recasting of the equivalent 𝑍 𝑍 transmission line circuit into a more typical RF circuit showing the open ( OC) and short circuit ( SC) impedances of the balun itself.

Table 2: MTM array performance for various choices of the load termination of the open stub.

Chosen state SW1 𝐶1 𝐿1 𝐶2 𝐿2 Configuration Bandwidth (%) Bandwidth (MHz) 1 X X high Pass 9.3 400 2X X lowPass58515 3X XXbandpass531,300 ∗ ∗ 4XXband-stop22565 5 unconfigured UWB 161 3,960 X = activated (on)/closed switch; no marker = deactivated (off)/open switch. ∗ Bandwidth is listed for VSWR < 3; indicates band rejection bandwidth. 𝐶1 = 3 pF, 𝐿1 =1.3nH,𝐶2 =1.75pF,and𝐿2 =2.5nH. AllotherMTMarraycircuitparametersaregiveninFigure 20. International Journal of Antennas and Propagation 13

Bottom Top Common mode +− rejection Z Balanced bal Current vectors

Z Z OC SC

Ground plan Z Currents cancel Unbalanced feed (a) (b) (c)

Figure 18: MTM array microstrip balun functionality depicting balanced currents and common mode rejection.

. . . . Z Z l OC 0 OC L C Z dipole coupling Z h feed sup sup Unbalanced Z 1 l Z feed SC 0 h

l1 Z Z Z SC Z Z L in bal TCDA

Figure 19: Simplified Marchand balun equivalent circuit (see48 [ ]).

+−

Z Balanced bal

8 480 MHz to 4.44 GHz 9.25: 1 BW 6 Switch C1

VSWR L 4 SW1 1 Switch

C2 L2 2

Unbalanced Z 12345 feed Frequency (GHz) Switch (a) (b)

Figure 20: MTM array bandwidth reconfiguration by varying the loads at the termination of the open stub. For the VSWR response to the 𝐶 /𝐿 𝑍 = 100 Ω 𝑍 =𝑍 = left stub it is simply left open, implying SW1 is open and 2 2 are off (unconfigured case). The other parameters: Feed , 0 Sub 𝑍 = 188.5 Ω 𝐿 = 2.2 𝐶 = 2.3 ℎ =60∘,ℎ =90∘ 𝑍 =11Ω 𝑙 =87∘ Sup , dipole nH, coupling pF, sup sub , characteristic impedance of OC , oc , characteristic 𝑍 = 312 Ω 𝑙 =58∘ impedance of SC , SC with all electrical lengths given at 2.5 GHz. 14 International Journal of Antennas and Propagation

10 10

8 8

6 6 VSWR VSWR

4 4

2 2

12345 12345 Frequency (GHz) Frequency (GHz)

State 5 (unconfigured) State 5 (unconfigured) State 1 State 3 State 2 State 4 (a) (b)

Figure 21: VSWR versus frequency for Table 2 reconfiguration states.

The above reconfiguration choices indicate that we Equivalent circuit representations of the array were pre- have significant control in generating band-pass and band- sentedtoaidinthedesignandreconfigurationofthearray. rejection responses starting with the broadband MTM array These circuit representations were used to develop simple in Figures 2 and 5. Moreover, bandwidth reconfiguration is bandwidth reconfiguration schemes. Instead of reconfiguring achieved using simple loads at the terminal of the open stub thegeometryoftheantenna’sapertureorthegroundplane, of the balun feed behind the MTM array. We can achieve we instead resorted to simple changes in the microstrip any response across the original MTM array bandwidth by lines present in the balanced feed. It was shown that the simply using varactors to change 𝐶1 and 𝐶2 and/or MEMS addition of simple lumped capacitors or inductors at the stub’s switches as reported in [71] or pin diodes to change the termination can provide substantial bandwidth control by length of the open or shorted stubs in the balun. Also, the simply switching on or off the load inductors or capacitors. load’s reactive impedance can be realized in practice using variable length stubs. Both, the MEMS switches and capacitor banks are known to have low losses. Therefore, their impact Conflict of Interests on the overall antenna performance should be quite small. The authors declare that there is no conflict of interests If varactors are necessary for continuous precise tuning, regarding the publication of this paper. care in choosing the optimum low-loss component would be imperative. Maximum allowable loss would dictate the highest tuning ratio attainable by the varactor diode. References [1] G. V.Eleftheriades and N. Engheta, “Metamaterials: fundamen- 5. Concluding Remarks tals and applications in the microwave and optical regimes,” Proceedings of the IEEE,vol.99,no.10,pp.1618–1621,2011. Inthispaperwereviewedtheconceptofconformalwideband [2]N.EnghetaandR.W.Ziolkowski,Electromagnetic Metamate- metamaterial (MTM) arrays with as much as 10 : 1 continuous rials: Physics and Engineering Explorations,JohnWiley&Sons, bandwidth in the presence of realistic feeds. MTM array Hoboken, NJ, USA, 2006. design guidelines and performance validation were provided via measurements. Array scanning was demonstrated and we [3] G. V. Eleftheriades and K. G. Balmain, Negative Refraction notedthatakeyaspectofthebroadbandresponsewasdueto Metamaterials: Fundamental Principles and Applications,John (a) the array’s interelement coupling to cancel the inductance Wiley & Sons-IEEE Press, Hoboken, NJ, USA, 2005. stemming from the ground plane and (b) the balun feed [4] D. H. Werner and D. H. Kwon, Transformation Electromagnetics that suppressed common modes and achieved impedance and Metamaterials: Fundamental Principles and Applications, matching across the entire bandwidth. Springer, London, UK, 2014. International Journal of Antennas and Propagation 15

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Review Article Reconfigurable and Tunable Metamaterials: A Review of the Theory and Applications

Jeremiah P. Turpin, Jeremy A. Bossard, Kenneth L. Morgan, Douglas H. Werner, and Pingjuan L. Werner Department of Electrical Engineering, The Pennsylvania State University, University Park, PA 16802, USA

Correspondence should be addressed to Douglas H. Werner; [email protected]

Received 15 November 2013; Accepted 7 January 2014; Published 22 May 2014

Academic Editor: Giacomo Oliveri

Copyright © 2014 Jeremiah P. Turpin et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Metamaterials are being applied to the development and construction of many new devices throughout the electromagnetic spectrum. Limitations posed by the metamaterial operational bandwidth and losses can be effectively mitigated through the incorporation of tunable elements into the metamaterial devices. There are a wide range of approaches that have been advanced in the literature for adding reconfiguration to metamaterial devices all the way from the RF through the optical regimes, but some techniques are useful only for certain wavelength bands. A range of tuning techniques span from active circuit elements introduced into the resonant conductive metamaterial geometries to constituent materials that change electromagnetic properties under specific environmental stimuli. This paper presents a survey of the development of reconfigurable and tunable metamaterial technology as well as of the applications where such capabilities are valuable.

1. Introduction electromagnetics for the construction of devices with complex spatial- or frequency-domain behavior [2, 3]. Recent Modern trends in technological development have increased developments in reconfigurable and tunable metamaterials demands for multifunctional components across the spec- have extended the possibility for fabricating metadevices [4] trum. In the radio frequency (RF) regime, wireless communi- and unique, subwavelength devices with practical function- cations necessitate efficient, reconfigurable, tunable, inexpen- ality. In addition to exhibiting electromagnetic responses not sive, and electrically small antennas that can be implemented readily available in nature, these metadevices offer the pos- in increasingly space-limited devices. In the terahertz band, sibility for improved performance characteristics in smaller, many materials do not respond to in-band radiation and multifunction applications. the components required to construct complex systems of Strictly speaking, metamaterials are collections of terahertz devices, such as lenses, switches, and modulators, far-subwavelength (<𝜆/10)resonatingstructures,typically do not exist. Significant efforts are going into filling this “gap” aligned in a regular crystal lattice, and may be characterized in the spectrum [1]. Additionally, the emerging use of trans- as possessing either effective material parameters for bulk, formation electromagnetics/optics, particularly with regard 3D structures, or effective surface impedances for planar, to cloaking, requires spatial gradients that natural materials 2D structures [5].Theseresonatorsaredesignedtocouple do not possess. With each of these challenges, designers and interact with the free-space propagating electromagnetic must compromise among size, functionality, complexity, and waves, rather than be excited directly by a waveguide fabrication cost. or transmission line. Metamaterials may be designed to Artificially constructed materials, metamaterials, have yield a desired refractive index and intrinsic impedance or emerged as an attractive option for addressing many of permittivity and permeability profile or to match a desired these issues and have become a useful tool in optics and frequency-dependent scattering response, which may be 2 International Journal of Antennas and Propagation viewed as a dispersive constraint on the effective material approaches and a reconfigurable metamaterial-inspired parameters. In practice, the term “metamaterial” is applied transmission line or antenna may be viewed as the first step to any subwavelength resonator, whether in a collection or as in the development of bulk tunable metamaterial-enabled an individual structure. devices. Many challenges have impeded the implementation of metamaterial-based devices, including the bandwidth lim- 2.2. Tuning Method versus Frequency. Avarietyoftuning itations of strongly resonant devices, as well as fabrication methods have been examined in the literature to generate limitations and tolerances. Many solutions to these prob- dynamic changes in a metamaterial’s performance. These lems have been presented which involve changes to meta- include direct changes to the unit cell’s circuit model by material structure, composition, and constituent material varying capacitance or conductance, using electrically, chem- properties. Tunable or reconfigurable metamaterials have ically, thermally, or optically sensitive materials to change the great potential to alleviate many of the complications with constituent material properties of a structure and therefore passive metamaterials at the cost of increasing fabrication change its electrical response and altering the geometry of complexity and expense. Within this review, the definition of the unit cell through stretching, shifting, or deforming all a tunable metamaterial is taken to mean a structure whose or part of the structure. Some of these techniques (such as electromagnetic behavior is intentionally modified as part of varactor diodes) have been applied for operation at particular the ordinary operation of the device through the influence wavelengths, while others (such as phase-change materials) of a change in, for example, the unit cell effective circuit, have been applied across the electromagnetic spectrum. Each constituent material properties, or geometry. In addition to of these tuning methods is described in more detail in the conventionally defined free-space metamaterials, we also Section 3. Figure 1 shows a diagram of tuning methods versus consider related structures including electromagnetic band- operational frequency for all reconfigurable metamaterial gap (EBG) materials or high-impedance surfaces (HIS), since and metamaterial-inspired devices reviewed in this paper. they may be designed to mimic the properties of an effective magnetic conductor [6, 7]. This paper is intended to provide a comprehensive survey 2.3. Common Unit Cell Designs. The base unit cell on which of published metamaterial tuning methods over all frequency a reconfigurable metamaterial device is designed determines bands and to give comparison information between compet- the fundamental behavior of the structure. In most cases, ing techniques and their applications. the starting point for the reconfigurable device is a static design which is then augmented with the tunable component, material,orstructure.Thephysicalgeometryoftheunitcell 2. Overview of Efforts and Techniques determines the electromagnetic coupling into the metamaterial, provides (potentially desirable or undesirable) frequency 2.1. Metamaterials and Metamaterial-Inspired Devices. selectivity, and can allow for cancellation of electric (in the Many antenna [8, 9], microstrip/transmission line [10], case of the SRR) or magnetic (for the electric LC resonator) and frequency-selective surface (FSS) applications use excitations as well as for strongly coupling the electric and the term “metamaterial-inspired” to describe their use of magnetic responses. Especially at optical wavelengths, many resonant structures as loading or filter elements within unit cells do not explicitly couple to the electric or magnetic the design, especially when the resonator in question has fields but are tuned to resonate simultaneously in the same been repurposed from a metamaterial structure. The design geometry [11]. Unit cells with well-defined resonant and andmeasurementprocedureandgoalsmaybeviewedas EM-coupling properties are commonly used to implement a loose discriminator between a “metamaterial-enabled” metamaterials with desired material properties, while designs and a “metamaterial-inspired” device. Metamaterial-enabled for scattering control or antenna enhancements may be less devicesmakeuseoftheeffectivebulkorscatteringproperties sensitive to cross-polarization or E-H field (magneto-electric) of a metamaterial and will typically employ these desired coupling. Most tunable metamaterials are based on common effective properties in simulations during the design process. elemental building blocks or particles, such as the widely used A metamaterial-inspired device, however, will rely on the SRR [12, 13], the complementary split-ring resonator (CSRR) exact behavior of individual resonators and will generally [14–16], and their electric-field coupled cousin the electric LC not utilize the metamaterial unit cells so as to obtain an resonator (ELC) [17–21]. Often, the structure is modified to effective bulk behavior. Although these structures are not a greater or lesser degree to accommodate the constraints of true metamaterials, the fabrication approaches and design the tuning mechanism [22, 23]. Tunable impedance surfaces decisions share many similarities with metamaterial design, are generally constructed from grounded square patches especially in the case of reconfigurable devices. Many of the (mushroom-type artificial magnetic conductor (AMC) struc- examples in the literature of tunable metamaterials are more tures) [24–33]. Although basing a design on an existing properly metamaterial-inspired devices which use individual resonator can be useful, many studies have developed new resonators (most commonly a split-ring resonator (SRR)), structures subject to limitations or to specifically leverage the often strongly coupled to transmission lines or antennas, capabilities of a given tuning mechanism; this includes the use as the proof of concept or even the final application for of reorientable or variable distributions of colloidal nanopar- the design. We include these papers in this review since ticles [34–38], interconnected grid structures [39–41], and the challenges towards implementation are shared for both optimized binary FSS patterns [42, 43]. Transmission-line International Journal of Antennas and Propagation 3

Tuning mechanisms [94, 102, 109] [93] [91, 92, 96, 98] [101] [97] Shift [100, 108] Deformation [38]

Physical Microﬂuidics [111] 117 118 136 34 42 120 Liquid crystal [ , , ] [35] [ , , ][119] Ferromagnetic/ 13 115 122 19 [125] ferroelectric [ ][ , ] [ ] 123 [17, 18, 124, 128] Semiconductor [ ] [126]

Material Phase change [129, 130] [131] [43, 107] Open/short [73–76, 78] [22, 77] Semiconductor 69 70 21 devices [ , ][] MEMS [80, 95][79, 81][103–106, 110]

Non-Foster [89] [45, 87, 88] 12

Circuit [ ] [66, 71, 133] (∗) [28, 31, 41, 47] Varactor 137 [15, 48, 55, 57][60, 63, 67, 132] [30] [ ]

Frequency (Hz) RF THz IR Optical 107 108 109 1010 1011 1012 1013 1014 1015

Wavelength (m) 101 100 10−1 10−2 10−3 10−4 10−5 10−6

∗[14, 16, 20, 23–27, 29, 32, 33, 39, 40, 46, 49, 51–54, 56, 58, 59, 64, 65, 68, 134, 135, 138, 143, 144]

Figure 1: Examples of metamaterial tuning mechanisms in the literature plotted against their operational frequency. Groups of references are centered on their shared operational frequency. metamaterials may not use such a recognizable geometric inductive, capacitive, and resistive elements. This decompo- unit cell pattern as an SRR or spiral but achieve periodic sition of complex geometric structures into a well-defined behavior instead based on their equivalent circuit model circuit model is highly useful for predicting the response [44–46]. of modified or perturbed designs. SRRs, for example, may be represented as a parallel LC or RLC tank circuit (pos- 3. Tuning Mechanisms sibly with additional parasitic elements depending on the frequency and exact geometry) where the inductive element The various tuning mechanisms that have been demonstrated is coupled to the incident field. We classify circuit tuning of in the literature may be divided into three general categories. metamaterials as those methods which insert, modify, and Circuit tuning involves the targeted insertion or modification control individual elements in the metamaterial’s equivalent of individual impedances into the unit cell circuit model; this circuit. includes the use of variable capacitors and switches within and between unit cells. We classify geometric tuning as those 3.1.1. Varactor Tuning. Varactor diodes represent the single methods that physically perturb the structure of the unit cell most popular tuning technique due to their simple inte- in such a way that many or multiple changes occur in the gration into many kinds of metamaterials. By acting as equivalent circuit model. This would include moving subsets a voltage-controlled capacitance, varactors are convenient of the unit cell relative to a fixed point or using MEMS for controlling the resonant frequency of SRRs or other devices to change orientation or location of a significant typesofresonantunitcells.Limitationstotheiruniversal fraction of the unit cell. Finally, material tuning leverages application include losses due to nonzero series RF resistance, changes in material parameters of a substrate, patterned layer, reduced performance at higher frequencies, and challenges or small region of the unit cell to modify the response and with distributing the required DC bias signals throughout properties. Phase change materials and liquid crystal devices a metamaterial. Varactors are very good approximations to are examples of this technique. The remainder of this section ideal tunable capacitors at MHz frequencies, but significant parasitic impedances accumulate and limit many metamate- is devoted to describing the use of these methods as reported rial applications above 4–10 GHz. Bias distribution has been in the literature. implemented for several specific metamaterial design styles. Using varactors to reconfigure planar metamaterials (such as 3.1. Circuit Tuning. The electromagnetic behavior of real, HIS or AMC) alleviates the bias distribution issue by allowing passive transmission lines, antennas, and metamaterials may bias signals to be delivered through the vias and traces behind be represented as an equivalent circuit composed of lumped the ground plane, isolated from the incident RF signals. 4 International Journal of Antennas and Propagation

Applying varactors to transmission-line metamaterials also actuator exists that can satisfy the design requirements for a simplifies the bias distribution [14–16, 23, 47–51], since given structure. additional traces to provide bias levels can be easily sited Micro-electro-mechanical systems (MEMS) can be used to avoid distortion of the desired frequency response. Bulk to fabricate RF switches with very high efficiency. Here, 3D metamaterials or transmissive planar metamaterials are we consider the use of MEMS as localized (relatively) self- more challenging, but several implementations have showed contained switches between two points in the metamaterial success by basing the geometry on an interconnected grid, equivalent circuit, rather than tuning by changing the geom- potentially with lumped resistance or inductance to prevent etry of the unit cell (which is discussed in Section 3.2). Apply- RF coupling [52]. Figure 2 shows an example of a tunable ing multiple switches per unit cell or group of unit cells allows AMC structure with integrated bias lines. multiple responses to be generated, as illustrated in Figure 3. Varactors are used to augment capacitive coupling within MEMS switches that physically make and break contact metamaterial structures, for example, across an existing between two terminals can be manufactured, but most capacitive gap [20, 31, 54–60] or between adjacent unit cells RF MEMS switches are better analyzed as high-magnitude [24, 26, 27, 29, 32, 33, 39–41, 61], as well as to completely capacitance modulators. In electromagnetics terms, tuning change the equivalent circuit by introducing a tunable capac- between a large and a small capacitance will yield a good itance in place of an inductor, open, or short, as done approximation to an ideal RF switch. Unlike varactor diodes, in combined right-left handed (CRLH) transmission line the capacitance ranges achieved by MEMS actuation can metamaterials [62, 63]. The former, by tuning the values in an often be sufficient to yield nearly ideal switching behavior essentially unchanged equivalent circuit, adjusts the resonant and are used as more realistic alternatives to open/short frequency of the device, without the dramatic change in actuation [22]. behavior seen in the latter case. MEMS switching performance is exceptional [79–81]. As a semiconductor device, use of a varactor in a metama- Limitations on the use of MEMS switches include high terial design is occasionally referred to as an “active” metama- actuation voltages (on the order of 70–150 V) that are difficult terial [33, 53, 54, 64, 65] although the device remains passive to interface with CMOS control circuits, high manufacturing for RF frequencies and is only active in its requirement for variability in the “off”-state capacitance between individual a DC bias. Most examples have, with reasonable accuracy, devices [80], and either complex fabrication requirements treated the varactors in their designs as ideal or nearly ideal for an integrated design or expensive per-unit costs for linear capacitors, but several studies have examined in more the purchase of commercially packaged devices. For these detail the nonlinear effects of the varactor at low and high reasons, the use of MEMS as RF switches in metamaterials has power levels [55, 56]. been limited, with published examples demonstrating either Some initial descriptions of new tunable metamaterials asingleorasmallnumberofunitcells. have used explicit lumped capacitors with different discrete values to approximate the varactor capacitance tuning 3.1.3. Active, Non-Foster Metamaterials. Non-Foster refers behavior and demonstrate the desired tuning effects [66– to active circuitry that is not subject to Foster’s reactance 68]. This strategy represents a first step to a full, tunable theorem [82–84] which states that the reactance slope versus implementation of the metamaterial. frequency for any real, passive circuit must be nonnegative. The use of PIN diodes [69–71] and digital potentiometers Non-Fostercircuits,then,throughtheuseofactivecircuitry [72] in a metamaterial shares similarities with the use of in the form of a negative impedance converter [85, 86], can varactors in terms of their application and ease of integration, achieve a negative impedance slope. This unique characteris- but these actuators affect resistance rather than capacitance. tic allows true broadband impedance manipulation through The advantages of simple control and liabilities of control complete or partial cancellation of reactance dispersion over signal distribution and frequency limitations are also compa- some bandwidth, rather than only at a point as is possible with rable to varactors. By tuning resistance, these methods could combinations of inductive and capacitive loads. Partial canbe applied to control or augment losses in a metamaterial cellation of reactance dispersion in a metamaterial through absorber design. the use of a negative inductance or capacitance may be used to maintain a “resonant” effect over a wide operation band 3.1.2. (Nearly) Ideal Switches. An ideal RF switch would be [62]. capable of instantly changing states from a 0 Ohm perfect Applying a non-Foster load to a metamaterial or AMC zero-electrical-length connection to a perfect open circuit. structure [83, 87] to broaden a resonance band can be viewed Since this device does not exist, several approximations to as a type of frequency-selective tuning [12], making such this ideal have been applied for the design of reconfigurable devices [45, 62, 88, 89] a close cousin to other active [21] metamaterials. and tunable metamaterials. Both share many of the same The simplest approximation of an ideal switch is to replace design considerations as to the inclusion of active devices, the prototype containing a switch by a pair of prototypes or control, and bias circuitry into the limited physical space of simulations, one with a metallic short and the other an open a metamaterial unit cell. Future work that includes tunable circuit at the desired location of the switch [22, 73–78]. This or reconfigurable non-Foster element may yield many advan- techniqueissimpleandcanbeusefultoprovideaproof- tages in terms of bandwidth and shared complexity. Figure 4 of-concept or intermediate design demonstration when no demonstrates the dramatically increased AMC bandwidth International Journal of Antennas and Propagation 5

−5 0 V 13 V 5 V 20 V −10 7 V Dipole

(dB) 10 V −15 S11

−20

−25 1 1.5 2 2.5 3 Frequency (GHz) (a) (b)

Figure 2: (a) Bow-tie AMC structure with bias lines to varactor diodes running throughout the structure. (b) When placed beneath a dipole antenna, tuning the varactors in the AMC controls the resonant frequency of the dipole. Used by permission [53].

ABCD magnetic resonance or other effective behavior. Because the metamaterial properties typically depend on the shape, size, orientation, and proximity of the conducting elements, techniques that alter the geometry of the conducting elements can provide an excellent means for tuning or switching the metamaterial response. At optical wavelengths, one nascent technique for geo- RF-MEMS Polarization metrical tuning is to use self-assembly to align resonant Resistive pads ABCDswitches lines nanorods in solution [36, 37]. In this method the entire conducting elements are aligned by applying a local electric (a) field gradient via electrodes. The gold (Au) nanorods have 0 a different resonant wavelength depending on whether the incident electromagnetic wave is aligned with the electric field oriented along the length of the rod or orthogonal to −10 the length. Hence, the metamaterial response can be changed

(dB) by aligning the nanorods in different orientations or by f D f A allowing them to disperse in the solution. Another related S21 −20 technique disperses Au nanorods in liquid crystal or other anisotropic fluids, so that the Au nanorods can be aligned f f C B with the liquid crystal molecules [90]. This technique helps −30 to align the conducting nanorods within a larger volume and 91011points toward a capability of realizing bulk reconfigurable Frequency (GHz) metamaterials. (b) Another related geometry tuning technique that has been demonstrated in the THz regime is tilting the resonant con- Figure 3: (a) The SRR-loaded microstrip line with RF-MEMS ducting elements in order to alter the metamaterial response switches controlling the resonances acts as a bandstop filter. (b) The [91, 92]. In both of these demonstrations, SRR elements stopband frequency can be altered by changing the combination of were tilted using MEMS [91]orthermal[92]control.The switch states. Used with permission [79]. SRR element is anisotropic in both effective permittivity and effective permeability, so tilting the element causes it thatcanbeachievedwithnon-Fosterloadsoverasimilar to couple differently with an incident wave. Hence, tilting AMC implemented with varactor loading. techniques are useful for reconfiguring the absorption or As with all active circuits, individual non-Foster devices filtering spectrum of the metamaterial as well as causing must be frequency-limited to preserve stability. The examples electric or magnetic resonances to appear and disappear in of non-Foster metamaterials have been implemented primar- regions of the electromagnetic spectrum. ily in the MHz range for size reasons as well as amplifier In addition to reorienting entire elements, metamaterials stability limitations. can be geometrically tuned by moving conducting elements in relation to each other [38, 81, 93–108]. MEMS are fre- 3.2. Geometrical Tuning. Many metamaterials rely on con- quently employed to perform the mechanical movement of ducting elements that can couple with impinging electro- conducting elements for THz metamaterials [38, 80, 93, 95, magnetic waves in order to achieve a desired electric or 101]. When conducting elements are moved closer or further 6 International Journal of Antennas and Propagation

Outer coaxial cone

40 cm Inner coaxial cone

AMC under test

27 cm

(a) (b) 150

125

100

75 bandwidth (%) bandwidth

∘ Unstable 50 NFC ±90 operation Stable NFC operation 25

0 100 150 200 250 300 350 400 450 500 AMC resonant frequency (MHz) NFC Varactor Simulation (c)

Figure 4: (a)-(b) Coaxial TEM reflection test cell for cylindrical non-Foster circuit (NFC) negative-impedance-loaded AMC structure. (c) The non-Foster-loaded AMC shows greatly increased bandwidth compared to a similar varactor-loaded AMC design. Used by permission [87]. apart, the coupling between them changes, which can result antenna system [102, 109]. Several tunable HIS have also been in shifts in resonance frequency or changes in resonance demonstrated that employ MEMS to raise or lower capacitive strength. Moving conducting elements can also reconfigure conducting plates in the unit cell [103, 104, 106, 110]. This the shape of the element. For example, in [101]onearmof technique effectively changes the capacitance in a part of the a cross can be disconnected and moved to reconnect with element and operates similar to the circuit tuning techniques the neighboring cross element. This kind of reconfiguration described in Section 3.1. changes the response of the metamaterial from being polar- Perhaps the most impressive form of geometric tuning is ization independent to being dichroic. One novel technique described in [111], where mercury (Hg) SRRs were formed for moving conducting elements in a metamaterial involves by using microfluidics to inject Hg into SRR molds to form stretching the substrate, so that the elements on the substrate the resonators. This system can place the Hg resonators become further separated [97], producing large shifts in and remove them in real time in order to reconfigure the the resonant frequency of the elements. Several tunable metamaterial response. HIS have been demonstrated that operate by mechanically Geometrical tuning can provide drastic changes in the shifting an upper plate of elements along the surface102 [ , metamaterial properties because the shape of the conducting 109]orvertically[105] in order to adjust the phase of the elements has such a large influence on the corresponding reflected wave, illustrated in Figure 5.Tuningthephaseof resonance. However, implementing geometrical tuning is a HIS is effective for steering a radiated beam from an challenging because a physical control mechanism is needed International Journal of Antennas and Propagation 7

Cell pitch: 350 𝜇 m ≈𝜆/10 Conductive patches Elements tuned for phase gradient

V Incoming microwave beam

Ground plane V Substrate Suspended membrane V Dielectric layer V Substrate Steering the reﬂected beam Metal V Dielectric (a) (b)

Figure 5: (a) Vertical motion of a suspended conductive membrane changes the resonant frequency of the high-impedance surface. (b) Varying the MEMs actuation voltage across the surface allows the metamaterial to act as a reflectarray. Used by permission [105].

to manipulate the conducting elements. Typically, the control wavelengths. Liquid crystals are anisotropic in permittiv- mechanisms, whether they are electrodes for self-assembly or ity due to their composition of long, aligned molecules MEMS actuators, are complicated to design and fabricate and and exhibit changes in permittivity due to applied voltage, they need to be accounted for in simulation. induced magnetic field, optical excitation, and even thermal change, which can be exploited to tune metamaterials. There are a number of theoretical and practical demonstrations 3.3. Material Tuning. Whilechangingtheshapeofresonant showing the tuning of negative index materials and magnetic elements provides a range of opportunities for tuning, the resonances at optical wavelengths [34, 35, 42, 117–120], constituent materials that make up the unit cell ultimately including the example shown in Figure 5.Thesameshift control the properties of the metamaterial. In the literature, in magnetic resonance has been explored for designing a a variety of constituent materials have been evaluated and metamaterial that switches between being highly transmis- exploited for tuning metamaterials by controlling the permit- sive and reflective by reorienting liquid crystal molecules tivity, permeability, and conductivity of parts of the unit cell. included in the unit cell [119]. GST phase-change materials The electrical size of the conducting resonant elements is have discrete crystalline and amorphous phases that possess affected by the permittivity of the surrounding medium. For distinct electrical properties at infrared wavelengths and that instance, a dipole element in free space is resonant when its can be switched between by subsequently melting and cooling length is approximately half of the wavelength. However, if the material. Several papers have investigated the use of this dipole is embedded in a dielectric, it becomes resonant GST as a constituent material for tuning metamaterials [43, athalfthewavelengthsizeinthedielectric.Thismeansthat 107, 121]. GST was used as the resonant element in [43]to if tunable dielectric materials can be incorporated in the change scattering behavior among being highly reflective, unit cell, the resonance wavelength of the metamaterial can transmissive, or absorptive. GST was also explored as a be tuned as the constituent material permittivity changes. substrate in [107, 121] that can switch a metamaterial mirror Several papers have theoretically investigated the effects between reflection polarization states. of changing the substrate permittivity on a metamaterial Permeability tuning can also be achieved at RF using response [112–114]. There are a few candidate materials that ferrite materials. In [122], yttrium iron garnet (YIG) rods have been used for permittivity tuning of metamaterials, having a negative permeability are incorporated into a meta- including Ba0.5Sr0.5TiO3 (BST) ferroelectric films, liquid material unit cell. Applying a magnetic field bias allows the crystal, and Ga-Sb-Te (GST) phase-change materials. BST permeability of the YIG rods to be tuned from negative to films provide a permittivity change under voltage bias and positive, causing a change in response of the metamaterial. have been used as a substrate for tuning SRR elements along While negative permeability constituent materials are not a transmission line at microwave frequencies [115, 116]. BST available at THz or higher frequencies, ferrite materials can can also be tuned via change in temperature as demonstrated be a useful tuning method at RF. in [13] for tuning the resonance frequency of SRRs. Liquid The third general area for constituent material tuning crystals offer dielectric tuning over a broad range of the is conductivity. Conductivity change is frequently accom- electromagnetic spectra from RF all the way through optical plished using semiconductor materials with applied voltage 8 International Journal of Antennas and Propagation

[17] or optical pumping [124–128]. The semiconductor can that manipulate wave propagation characteristics including be incorporated as a substrate under all elements [17, 128] directivity, radiation pattern, polarization, and propagation or as an inductive load over part of the resonant elements mode. [125–127] to tune their resonant frequency. Graphene has also been used to inductively load resonant elements in a 4.1. Tunable Filters and Antennas. Amajorfocusoftunable HIS when under a voltage bias [129]. A variety of techniques and reconfigurable metamaterials lies with designing tunable have also been employed to change the conductivity of the filters and frequency-reconfigurable antennas. One desirable resonant elements themselves. Semiconductor SRRs have feature of such devices is tunable frequency band selectivity. been used with a change in conductivity achieved thermally In wireless communications, using tunable metamaterials [18] or by applying a magnetostatic field [19]. Conduct- enables wider ranges of operating bands within a minimal ingpolymershavealsobeenusedasresonantelements geometry. Bossard et. al. have applied liquid crystal super- in metamaterial absorbers [75]. The conducting polymers strates to shift the frequency of the transmission band in an exhibit large changes in conductivity (i.e., from resistive to infrared FSS device, as illustrated in Figure 6. Reynet and conductive) when stimulated by certain chemical analytes. Acher demonstrated tunable varactor-loaded metamaterials This enables the metamaterial to reconfigure from reflecting based on conducting coils [57]. Later works introduced to absorbing or to change resonant frequencies. VO2 phase- this concept into transmission lines and antennas, where change materials also exhibit large conductivity changes with tuning was achieved by varying capacitance, typically of avoltagebiasandhavebeenusedtoformtheresonant SRRs and CSRRs [16, 23, 46–50, 62, 123, 124, 133]. Zhu et elements in several metamaterial experiments [130–132]. al. demonstrated an electrically small, tunable SRR antenna Bulk material-based tuning is ultimately limited by the operating in the UHF frequency band [123]. As shown in range of electromagnetic responses available in the con- Figure 7, this varactor-loaded dipole-like structure provides stituent substances, where each material system poses unique a tunable narrowband (notch filter) alternative to wideband implementation challenges. For instance, BST and VO2 offer antennas in RF communication systems. The antenna has useful permittivity and conductivity changes, respectfully, the advantage of compact size, low cost, and easy fabrication buttheyarebothsensitivetotemperatureandthusmustbe as well as implementation. Researchers have attained similar used in temperature controlled environments. GST phase- RF frequency tunability through a variety of other methods, change material on the other hand does not suffer from such as through MEMS [22, 73, 79, 81, 93, 94], tunable temperature variation, but incorporating the heating and EBG surfaces [59, 78, 95, 112, 134, 135], ferroelectric rods cooling mechanisms into the metamaterial to control the and films [13, 115, 116, 122], vanadium dioxide switches [130, phase transitions between crystalline and amorphous states is 131],andliquidcrystals[42, 117, 118, 136]. These proposed challenging. Research needs to continue in the development designs exhibited varied tunable RF filter types, such as band-stop, band-pass, and notch filters. Whereas the filters of tunable material systems that can be harnessed to meet discussed above are actively voltage-controlled, designers application specific reconfigurable metamaterial needs. have also demonstrated nonlinear varactor-loaded (or p-i-n- loaded) metamaterial filters that vary with incident power, 4. Applications and Tuning Goals making them ideal for nonlinear devices such as RF limiters [55, 56, 69]. Frequency tunability is especially desirable for The inherent nature of tunable and reconfigurable metama- fabricating flexible devices in the terahertz and optical bands. terialsaffordsthemtheabilitytobeusedinawidevariety Such devices could include sensitive platforms for measuring of applications. The distinctions between these end goals chemical or biological agents, temperature sensors, and are largely a matter of perspective. Shifting the frequency intelligent detectors among other applications [18, 19, 96, of a passband is functionally identical to changing from a 97, 125, 126, 137]. Furthermore, in contrast to scanning the transmissive to reflective surface at a fixed frequency. Inter- center frequency of a desired response feature, another essen- pretation of an identical change in response can be made in tial functionality is tunable bandwidth, or reconfigurable terms of scattering parameters, effective material parameters, filtering type, that is, dynamically switching between band- or the metamaterial’s loading effect on an adjacent antenna pass, band-stop, notch, and so forth. Several authors have or transmission line. We acknowledge the discretionary demonstrated metamaterial structures that exhibit recon- nature of categorizing tunable metamaterials into intended figurable filter types, as well as tunable center frequencies applications; however, in this literature survey, we define [14, 39, 63]. Tunable filters, FSSs, and antennas have wide four broad common application areas. First, we discuss the design of tunable metamaterials as an enabling technology ranges of applications across the electromagnetic spectrum. for tunable filters and antennas; this focuses on metamaterials Tunable metamaterials offer a cost-effective, compact, and that are primarily designed to shift the resonant frequency flexible option for implementing tunable band selectivity or alter the device’s bandwidth. We then note metamaterials and reconfigurable filter functionality in applications ranging that are optimized to modulate the scattering (transmis- from RF communication system design to optical sensors and sion, reflection, and absorption) of a material at a given detectors. resonant frequency. We classify designs meant to spatially vary the index of refraction with a focus on transformation 4.2. Scattering Parameter Tuning. Tuning the transmission, electromagnetics. Finally, we recognize tunable structures reflection, and absorption characteristics of a material at International Journal of Antennas and Propagation 9

Wavelength (𝜇m) 3.75 3.33 3 2.73 2.5 0

−5 Liquid crystal superstrate

FSS screen elements −10

Polyimide substrate

Transmission (dB) Transmission −15

−20

Quartz or glass slide 80 90 100 110 120 Frequency (THz) Rubbing layer O-ring seal 𝜀 = 2.0 LC Liquid crystal 𝜀 =3.0 LC 𝜀 =4.0 Polyimide FSS screen LC (a) (b)

Figure 6: (a) An illustration of a liquid-crystal-based tunable FSS unit cell. (b) Changing the applied potential difference between the FSS and the quartz slide changes the liquid crystal orientation and the scattering response of the FSS. Used by permission [42].

−5

−10

−15

magnitude (dB) magnitude −20 1 V 3 V S11 −25 1.5 V 4 V 2 V 9 V −30 2.5 V 350 400 450 500 550 Frequency (MHz) (a) (b)

Figure 7: (a) This varactor-loaded dipole-like antenna has a (b) narrow-band but tunable notch-filter response, allowing operation overa wide range of frequencies while avoiding spurious signals. Used by permission [123]. a particular frequency affords the designer the creation were used to drive parallel strings at megahertz frequencies to of devices such as switches, modulators, FSSs, absorbers, modify the transmission and reflection spectra of the device. sensors, and more. Tunable metamaterials offer avenues for Thecharacteristicsofthisdevicemakeitidealforprotec- realizing these devices in low-profile mediums with ultrafast tive optical circuitry and reconfigurable optical networks. tuning capabilities. A fundamental goal for dynamically Tunablescatteringcharacteristicsarealsovitaltodeveloping controlling the scattering from a material is amplitude mod- tunable cloaks and electromagnetically transparent materials. ulation. Tunable metamaterials enable designers to create In particular, by controlling the reflection and transmission, high-speed modulators that alter the transmission and/or designers can create perfect absorbers, reduce radar cross reflection amplitude of an incident electromagnetic wave17 [ , section, or achieve complete transmission and reflection 21, 67, 111, 127, 138]. Practical devices emerge from this funda- from a surface [29, 68, 70, 75, 91, 92, 119, 139]. Tao et al. mental functionality, such as high-speed switches operating designed a MEMS-based structurally tunable metamaterial in the terahertz and near-infrared bands [38, 98, 128]. Ou et al. that can dynamically modify its transmission amplitude in demonstrated a switchable metamaterial that operates in the the terahertz regime [91]. Rotating the unit cell plane with optical band at three orders of magnitude faster than previ- respect to the incident electric and magnetic fields alters the ously achieved [38]. As shown in Figure 8, electrostatic forces coupling efficiency. Structural reconfiguration time can be 10 International Journal of Antennas and Propagation

10 𝜇m 6

3 (%) 0 OFF ΔT/T −3 on individual strings individual on

Electrostatic forces acting acting forces Electrostatic −6 1.0 1.2 1.4 1.6 1.8 2.0 U + − Wavelength (𝜇m) (a) (b) 9 2.4 V 1.8 V E 2.3 1.5 1 V V 2.2 1.0 6 V V 2.1 V 0.5 V 2.0 0.0 nm V V (%)

1000 3 OFF Normalized Normalized

0 electric field optical 800 OFF ON ΔR/R nm e 30 0 ) −1

−3 MV m ( 1.0 1.2 1.4 1.6 1.8 2.0

0 electric field Static OFF ON Wavelength (𝜇m) (c) (d)

Figure 8: (a) Scanning electron microscope (SEM) image of photonic metamaterial device and schematic of driving circuit. (c) SEM image of single metamolecule and plasmonic field distribution for OFF and ON states. (b) Transmission and (d) reflection spectra of photonic metamaterial at varied induced static voltages. Used by permission [38].

on the order of milliseconds, allowing this design to be used enables designers to create a wide variety of common devices in applications ranging from reconfigurable filters, thermal across the electromagnetic spectrum. cantilever-based detection, or tunable absorbers, cloaks, or concentrators. Controlling scattering also has several applications in the optical band; varying transmission and reflection 4.3. Spatial Tuning for GRIN Lenses. The recent explosion enables the development of optical temperature sensors, of research in transformation electromagnetics techniques switches, modulators, and other planar optical metamaterial- [140, 141] has increasingly put emphasis on spatially modi- enabled devices [90, 99]. One particular application is in fying the parameters of materials. In particular, these spa- compressive sensing. Werner et al. demonstrated a phase tial gradients are primarily focused on altering index of change material that could spatially modify its complex refraction. Sheng and Varadan investigated the effect of index of refraction to dynamically change the effective size substrate dielectric (relative permittivity) on a metamaterial and shape of an aperture [43]. This beam-forming aperture, structure [113]. By varying the relative permittivity from 1 to with its ultrafast response in the infrared band, could then 14, the structure’s resonant frequency dropped from 16 GHz be used for compressive sensing, a technique where few to 6 GHz [114]. This discovery provided a foundation for information rich measures are sampled to construct high- designing voltage tunable dielectric substrates, which led resolution images. This technique is a step towards achieving to the development of microfabricated artificial dielectrics thedesiredresolutionwhilemaintainingtheareacoverage for microwave circuits [100]. While this research addressed and minimizing the cost, size, weight, and power of the sensor only modifying the overall index of refraction of a substrate, system. Controlling the scattering characteristics of a material several authors developed methods for creating materials International Journal of Antennas and Propagation 11

R2 R 1 H

E K z

y Repeat x

(a)

Hz Hz 0.05 0.05

0 0 (m) y

−0.05 −0.05 −0.05 0 0.05 −0.05 0 0.05 x (m) x (m)

1 0.5 0−1−0.5

(b)

Figure 9: (a) Configuration of TM mode reconfigurable cloak; independent control over the effective material properties of each unit cell within the cylindrical region is achieved through independent biasing of each column of diodes. (b) Magnetic field distributions surrounding and within the dielectric cylinder with and without the cloak shell at 9.5 GHz. The material properties throughout the structure may be tuned to act as a cloak around the inner cylinder. Used by permission [41]. with spatially varying indices of refraction [34, 36, 37, 72, 4.4. Antenna Propagation Tuning. Tunable metamaterials 120, 132]. Spatial gradients are the fundamental mechanisms also provide the potential for increasing existing compo- in electromagnetic cloaking. Researchers have demonstrated nent functionality and enhancing propagation properties, spatially varying material parameters that are intended to such as scanning range, radiation pattern, and directivity, guide electromagnetic waves around a desired region [41, 45, while simultaneously reducing costs. In particular, tun- 54]. Wang et al. designed a reconfigurable cloak that could able metamaterials have found applications in fabricating be tuned for multiple operating frequencies. Furthermore, reconfigurable directive antennas and beam steering devices, this cloak could be switched from a visible to invisible status especially for radar and communication systems: radomes, [41]. The mechanism for attaining this cloak’s spatial gradient radar absorbent materials, reflectors, electromagnetic inter- index of refraction relaxes fabrication precision constraints ference shielding, and terrestrial and satellite communica- because each unit cell can be tuned independently. These tions. Many of the efforts towards achieving beam steering features make it ideal for cloaking in RF communication focus on developing electrically and mechanically tunable systems [142]. The structure, material parameters, and field high-impedance or artificial magnetic surfaces [24–28, 30, 31, distribution of this cloak are depicted in Figure 9. In addition 33, 53, 64–66, 80, 95, 102–106, 109, 110, 129, 143, 144]. Arti- to varying index value, researchers have also demonstrated ficial magnetic surfaces typically consist of planar periodic metamaterials that vary in anisotropy [101]. Such features can metallic elements backed by a PEC ground plane. Coupled be used in creating displays, beamsplitters, isolators, cloaks, with tuning elements, they are commonly utilized as tun- and hyperlenses and in controlling luminescence. able high impedance ground planes to achieve low-profile 12 International Journal of Antennas and Propagation

Diode Diode

Wires Vias (a) (b) 150 Cj =1pF C Cj = 1.1 pF 4 j constant 100 Cj = 1.2 pF Cj negative gradient C = 1.3 j pF C C = 1.4 j positive gradient j pF 2 50 Cj = 1.5 pF Cj = 1.6 pF

Cj =1.7pF

0 Cj = 1.8 pF 0

Cj = 1.9 pF C =2

j pF (dB) Gain −50 −2 Reﬂection phase (deg) phase Reﬂection −100 −4 −150 2.22.42.62.83 −150 −100 −50 0 50 100 150 Frequency (GHz) 𝜃 (deg) (c) (d)

Figure 10: (a)-(b) Active AMC bow-tie surface showing varactor diode tuning between the bias lines and the vias to the ground plane. (c) Tuning the varactor state changes the frequency response of the AMC. (d) Applying a constant gradient to the AMC capacitances allows beamforming by a bow-tie antenna placed above the surface. Used by permission [64]. antenna configurations. Costa et al. employed this concept This increases the functionality of microwave waveguide to create a steerable bow-tie antenna [64]. Figure 10 displays systems by allowing designers to broaden the scanning range the artificial magnetic surface, the reflection phase at var- or control the radiation pattern of leaky wave antennas, ied capacitance values, and the resulting beam deflections. which increases the link capacity for multi-input/output Ratajczak et al. demonstrated a directive antenna using a communication systems. planar electromagnetic band-gap reflector based on the same design principles [28], which is shown in Figure 11.This 5. Remaining Challenges proposed design offers advantages over traditional directive array antennas, which suffer from complex, expensive, and Based on the existing capabilities described in the literature, bulky feeding networks with high losses. This design is easier several broad goals are seen for future developments. Extend- ing tuning concepts that have been demonstrated at an indi- tofabricateandsimpletouseinlargefrequencybands,has vidual or small number of unit cells to tile or fill large regions low losses due to a waveguide feed structure, and has a tunable is necessary for the implementation of many metamaterial radiation pattern. With average tuning ranges of approxi- or transformation electromagnetics/optics devices, such as ± mately 45 degrees, these tunable impedance surfaces offer lenses. Very few studies have demonstrated spatial tuning low-cost, low-profile, and light-weight alternatives to tradi- of metamaterials for the creation of reconfigurable gradient- tional scanning antennas and reflectors. Similar functionality index lenses, for example. Expanding the reconfigurable unit can be achieved with digitally addressable, anisotropic, and cells to large regions of independently tunable elements SRR-based metamaterials [20, 76, 77, 107, 121]. Several papers requires the associated development of tuning and control also document work towards creating partially reflective sur- signal distribution throughout the metamaterial. Finally, faces that alter the reflection and transmission phases for use extending the tunable range and application flexibility of in Fabry-Perot cavity systems to obtain directive emissions reconfigurable metamaterial devices will enhance the useful- [32, 40, 60, 61]. Tunable metamaterials also offer increased ness and applicability of this design strategy to solving real- functionality when incorporated into existing devices. Meng world problems. et al. demonstrated a reconfigurable magnetic metamaterial- Another general area for future development is the loaded waveguide in which the propagation mode could expansion of available tunable materials. In particular, phase- be switched between left-handed and right-handed108 [ ]. change materials are under continued development to expand International Journal of Antennas and Propagation 13

(a)

Wp C Wc =14.00mm pi Z = i/C 𝜔 Wp = 12.80 mm i pi Wp2 Wp1 = 6.00 mm Ch W = 8.00 Cp 𝜙 p2 mm via Wc Cpi h = 3.175 mm Wp Wp1 𝜀r = 2.2 𝜙 = 1.5 via mm Ch = 3.00 mm Lp C vi Rs

h Zi =Ri +iXi Wc 1/Cpi𝜔 ∼Xi

(b)

Figure 11: (a) Reconfigurable reflectarray that changes the reflection coefficient from the surface by varying the capacitance betweenthe central via and the ground plane. (b) Reflectarray unit cell geometry and equivalent circuit model. Used by permission [28].

therangeofmaterialformulationsandtorefinetheswitching development, we have illustrated that the study of tunable and tuning circuitry. While phase-change materials have metamaterials is a vibrant and active subfield, based on the already been commercialized for high speed memory appli- breadth and depth of the applications and methods that have cationsincomputerhardware,theirsuitabilityforincorpo- been reported on in the literature. By classifying the tuning ration into practical, high-speed switchable metamaterials mechanisms and applications into groups based on function- is a topic of ongoing research. The development of smart ality and capability, rather than operational frequency, our materials that respond to environmental stimuli is also an main goal is to provide a comprehensive overview of the state area of future development. Multiple works were highlighted oftheartintunableandreconfigurablemetamaterials.New in the literature looking at conducting polymers that can tuning methods and analysis techniques may be applied to change properties under stimulus by chemical analytes. existing static metamaterial designs to dramatically increase Furtherworkisrequiredinboththematerialscienceand electromagnetic fields to identify candidate sensing materials their capability and effectiveness. Although there is more and develop metamaterial platforms to bring other smart progress that must be made before many of the techniques devices into practice. discussed here may be practically applied, rapid developments in tunable metamaterials hold great promise for future Finally, extending the tunable range and application flexibility of reconfigurable metamaterial devices will enhance the implementations. usefulness and applicability of this design strategy to solving real-world problems. There is room for advances in material and switch development as well as identifying metamaterial Conflict of Interests geometries that are sensitive to the switching property in The authors declare that there is no conflict of interests order to increase the device tuning range and functionality. regarding the publication of this paper. 6. Conclusion Acknowledgment In this review, we have extensively documented the state of the metamaterials field as it applies to the generation and This work was supported in part by the EMERALD Project, usage of tunable or reconfigurable electromagnetic responses. which is funded by the Provincia Autonoma di Trento under Although metamaterials are themselves in the early stages of the“BandoUnita` Ricerca 2011.” 14 International Journal of Antennas and Propagation

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Review Article MEMS-Reconfigurable Metamaterials and Antenna Applications

Tomislav Debogovic1 and Julien Perruisseau-Carrier2

1 Laboratory of Electromagnetics and Acoustics, Ecole Polytechnique Fed´ erale´ de Lausanne (EPFL), EPFL-STI-IEL-LEMA, ELB 030 (Batimentˆ ELB), Station 11, 1015 Lausanne, Switzerland 2 Group for Adaptive MicroNanoWave Systems, Ecole Polytechnique Fed´ erale´ de Lausanne (EPFL), EPFL-STI-IEL-GR-JPC, ELB 030 (Batimentˆ ELB), Station 11, 1015 Lausanne, Switzerland

Correspondence should be addressed to Tomislav Debogovic; [email protected]

Received 27 February 2014; Accepted 22 March 2014; Published 30 April 2014

Academic Editor: Giacomo Oliveri

Copyright © 2014 T. Debogovic and J. Perruisseau-Carrier. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

This paper reviews some of our contributions to reconfigurable metamaterials, where dynamic control is enabled by microelectromechanical systems (MEMS) technology. First, we show reconfigurable composite right-/left-handed transmission lines (CRLH- TLs) having state of the art phase velocity variation and loss, thereby enabling efficient reconfigurable phase shifters and leaky-wave antennas (LWA). Second, we present very low loss metasurface designs with reconfigurable reflection properties, applicable in reflectarrays and partially reflective surface (PRS) antennas. All the presented devices have been fabricated and experimentally validated. They operate in X- and Ku-bands.

1. Introduction consumption (electrostatic control), high linearity, and possibility of monolithic integration. Metamaterials [1–5] allow efficient manipulation of guided In this paper, we show reconfigurable metamaterial and free-space electromagnetic waves, thereby potentially devices with direct applications in antenna systems. In improving existing and enabling novel microwave com- Section 2, we present a Ku-band MEMS-reconfigurable component and antenna designs [6–11]. On the other hand, posite right-/left-handed transmission lines (CRLH-TLs) dynamic reconfiguration [12–17] has become a prime need in with applications in leaky-wave antennas (LWA) and feed modern communication and sensing systems, for instance, to networks for antenna arrays. In Section 3,wepresentX-band scan space and polarization, to dynamically compensate for MEMS-reconfigurable metasurfaces whose reflection prop- varying system conditions thereby guaranteeing optimal per- erties can be reconfigured. These devices can be applied for formanceinrealtime,ortosupportahighernumberoffunc- dynamic beam-scanning/-forming in reflectarrays and dyna- tionalities through a single and compact device. In this con- mic beamwidth control in partially reflective surface (PRS) text, the implementation of reconfigurable metamaterials [18, antennas. The underlying reconfiguration technology is 19] for antenna applications has become a topic of intense MEMS due to the need for low loss, convenient control, and practical relevance. For example, reconfigurable metamate- ability to monolithically integrate a large number of controll- rials can enable operating frequency reconfiguration13 [ , 20], ing elements into an electrically large structure. Finally, con- beam steering without the need for a beam-forming network clusions are drawn in Section 4. [16, 21], and dynamic beamwidth control [22, 23]. The technology used to enable reconfiguration has apro- found impact on the performance of reconfigurable metama- 2. MEMS-Reconfigurable CRLH-TLs terials and consequently on the quality of microwave components and antennas utilizing them. Microelectromechanical AcompositeCRLH-TLstructure[4, 5, 31–33]isaclassof1-D systems (MEMS) technology [24–30]canprovideexcellent metamaterial, which can be implemented by lumped “dual” properties thanks to very low loss, virtually zero power elements (series capacitors 𝐶𝑠 and shunt inductors 𝐿𝑝) 2 International Journal of Antennas and Propagation

Cs Cs

Lp Lp

(a)

Interdig. C on CPW signal

TRANSPARENT

(b)

Figure 1: MEMS-based CRLH-TL unit cell from [34] (a) circuit model and (b) layout. loading a usual TL, as shown in Figure 1(a). It is interesting the cell input/output. Several unit cells can be cascaded by because it can provide both positive and negative phase shifts, using dc-block capacitors and high resistivity bias lines. The corresponding to the left-handed and right-handed region unit cell was fabricated using MEMS process developed at of the Bloch-Floquet propagation constant, respectively, with Middle East Technical University on 500 𝜇mthickglass seamless transition between them. In addition, it is inherently wafers,detailedin[27]. wideband structure, thereby being well suited for low loss Simulated propagation constant 𝛾𝐵 (without MEMS actu- phase shifting and antenna applications. MEMS technology ation) is shown in Figure 2(a). A typical CRLH-TL disper- ∘ enables relatively straightforward phase shift reconfiguration sion is observed, with the frequency of the 0 phase shift by dynamically controlling the series capacitance 𝐶𝑠 [34, 35] (corresponding to the transition between the left- and right- or both series capacitance 𝐶𝑠 and shunt inductance 𝐿𝑝 [36]. handed bands) at 𝑓0 =14GHz. Simulated and measured One possible unit cell design, implementing the circuit results of the transmission phase upon reconfiguration are model of Figure 1(a),isshowninFigure 1(b) [34]. A simi- shown in Figure 2(b). Reconfigurable CRLH-TLs have at least larly operating design can be found in [35]aswell.Series twoapplicationsinantennas,bothbasedonthetransmission MEMS capacitors and folded shunt stub inductors load a phase manipulation shown in Figure 2(b).Thefirstone coplanar waveguide (CPW) line. The profile of the MEMS concerns leaky-wave antennas scanning around broadside areaisshownintheinsetofFigure 1(b).ThebottomMEMS [4, 5, 7] and it relies on the ability to generate a reconfig- electrodeisapartoftheCPWsignallineinthemiddleofthe urable negative/zero/positive phase shift at a given operating unit cell, and it is dc grounded through inductive stubs. The frequency, as symbolized by the “A” arrow in the figure. The movable MEMS membrane, essentially operating as the series second main application concerns series feed networks or ∘ capacitor 𝐶𝑠, is connected to the CPW signal line at the unit dividers [4, 5]. These devices operate at the frequency of 0 ∘ cell input/output by the main anchor. Electrostatic MEMS phase shift. Thus, their frequency of 0 phase shift 𝑓0,infact, actuation, enabling analog capacitance control, is achieved by their operating frequency, could be reconfigured as shown by applying a voltage between CPW signal line and grounds at the “B” arrow in Figure 2(b).OwingtotheMEMStechnology, International Journal of Antennas and Propagation 3

3 40 Stop 2 band Left-handed Right-handed 20 1 Real [] Light line ) ∘ (

(—) A 0 ) d B

21 0 B 𝛾 S −1 Light line arg( −2 −20 Image [] −3 −40 4 6 8 10 12 14 16 18 20 12 13 14 15 16 17 18 Frequency (GHz) Frequency (GHz)

Up, measurement (0 V) Down, measurement (25 V) Up, simulation (a) (b)

Figure 2: MEMS CRLH-TL unit cell results from [34] (a) the simulated Bloch-Floquet propagation constant without MEMS actuation and (b) simulated and measured transmission phase upon reconfiguration. Arrows “A” and “B” illustrate possible modes of utilization. very low insertion loss is achieved, being less than 0.7 dB when the transmission phase is reconfigured and less than ∘ 0.8 dB when the frequency of 0 phaseshiftisreconfigured. ∘ A differential phase shift over losses is 38 /dB at 14 GHz. Apart from the analog MEMS control presented above, digital MEMS control (where MEMS elements take two discrete states, namely, “up” and “down”) can also be implemented. In fact, this type of control is less sensitive to MEMS fabrication tolerances and vibrations, and nowadays it is widely accepted. A CRLH-TL unit cell with digital MEMS control [36]isshowninFigure 3.AdigitalMEMScapacitor located in the horizontal CPW line acts as the series capacitor, Figure 3: Digitally controlled MEMS-based CRLH-TL unit cell while the remaining two (identical) MEMS elements, located from [36]. Series capacitance and shunt inductances are controlled in the vertical shorted CPW stubs, serve to effectively change in two discrete states. their length and hence the shunt inductance value. As a result, ∘ ∘ a CRLH-TL line with the phase shift of −50 /+50 is obtained. beamwidth of such an antenna can be dynamically controlled by the MEMS elements. This functionality is useful for instance in satellite communications from elliptical orbits, 3. MEMS-Reconfigurable Metasurfaces where the antenna coverage should remain constant despite variable platform altitude. This section presents unit cells of two MEMS-reconfigurable metasurface types. The first type is designed to provide 3.1. Metasurface Unit Cell with Reflection Phase Control. reflection phase reconfiguration and it has direct application Figure 4 shows the reconfigurable reflection phase metasur- in reconfigurable reflectarray antennas17 [ ], where shaping of face unit cell [38]. In essence, it is a tunable resonator that reflection phase profile of a reflector allows dynamic beam- consists of two pseudorings backed by a ground plane and scanning. Such functionality and essentially flat reflectarray loaded by digital series MEMS capacitors. As the structure is antenna geometry are very useful in space communications backed by the ground plane, its reflection loss is only a subject and radars. to the dissipation in the materials and MEMS elements, being The second metasurface type presented is a partially generally very low. The MEMS elements are organized in five reflective (or a partially transparent) design whose reflection pairs in order to retain symmetry and thereby lower the cross- magnitude can be reconfigured owing to the embedded polarization level. The resonance is tuned by altering the MEMS elements. Placing such a metasurface approximately capacitance values of the MEMS pairs. This implies that the 5 halfawavelengthaboveasourceantennabackedbyaground metasurfacereflectionphasecanbereconfiguredin2 =32 plane results in a reconfigurable partially reflective surface discrete states at a given operating frequency. (PRS) antenna [37], whose directivity (beamwidth) depends The metasurface reflection phase values for all 32 combi- on the metasurface reflectivity. Therefore, directivity and nations of the MEMS states are shown in Figure 5.Itcanbe 4 International Journal of Antennas and Propagation

The unit cell topology is based on a ring-like shape, loaded by digital series MEMS capacitors 𝑆𝑥 and 𝑆𝑦.TheMEMS elements come by pairs to preserve symmetry. Owing to this property, the MEMS pairs 𝑆𝑥 and 𝑆𝑦 affect reflection properties independently in 𝑥-and𝑦-polarizations, respectively, thereby enabling a 1-bit dual-polarization operation. This unit cell is fabricated by RF Microtech using the process described in [44]. It also includes the possibility to pattern high- resistivity polysilicon bias lines which do not impact on the Figure 4: MEMS-reconfigurable metasurface unit cell with reflec- microwave performance (visible in Figure 6(a) as narrow tion phase control [38]. Ten digitally controlled MEMS elements are lines). organized in five pairs. The unit cell reflection coefficient is shownin Figure 7(a), for incident 𝑥-polarization at the design frequency 𝑓0 = 0 11.2 GHz. The reflection magnitude in this polarization is 180 controlled (in two discrete states) by the MEMS pair 𝑆𝑥 and 135 −0.2 independent of the states of the MEMS pair 𝑆𝑦.Asthemeta- 90 −0.4 surface is practically lossless and single-layered, the reflec- )

∘ 45

( tion magnitude reconfiguration implies a reflection phase ) 𝜌 0 −0.6 dB) variation as well. However, in a properly assembled recon- |𝜌| ( Arg( −45 figurable PRS antenna, the amplitude reconfiguration has a −0.8 −90 stronger impact [23]. Predicted radiation patterns of the assembled PRS −135 −1 antenna, formed by placing such a metasurface above a source −180 −1.2 antenna (and thereby creating a Fabry-Perot´ resonator) are 5 10 15 20 25 30 shown in Figure 7(b). It can be noticed that the antenna MEMS state beamwidth is reconfigured in two discrete states by MEMS 𝑥 fC (12 GHz) elements. Since the result is shown for the -polarization, 𝑆 fL (11.7 GHz) only the MEMS elements 𝑥 affect the beamwidth. Thanks to fH (12.3 GHz) the MEMS technology and the unit cell design, the predicted radiation efficiency is above 75%, which is considered to be Figure 5: Simulation results from [38] metasurface reflection phase very high for a reconfigurable antenna operating in X-band. and loss as a function of the MEMS state, at the limit of a 5% Results for the 𝑦-polarization are not shown here for space bandwidth at 12 GHz. consideration, but they are essentially the same owing to the unit cell symmetry. Reflection measurement results [43]areshownin noticed that the reflection phase can be reconfigured within ∘ Figure 8. An orthomode transducer (OMT) was employed for full 360 range. In addition, the phase curves at the limits of this measurement. Being a square waveguide with couplers 5% bandwidth are quasi-parallel to the one observed at the designed to separate its orthogonally polarized modes, the operating frequency, as required to avoid beam squint in a reconfigurable reflectarray39 [ ]. The observed reflection loss OMT allows validation of the unit cell reflectivity in two is very low, mainly below 0.3 dB. orthogonal linear polarizations. A very good agreement The main design effort in such metasurfaces is to obtain between measurements and simulations is observed, both in the uniform phase step between MEMS states and to retain magnitude and phase, thereby validating the unit cell design itinagivenbandwidth,whilealsokeepingthephaserange and confirming the predicted reconfigurable PRS antenna ∘ close to 360 and loss as small as possible. This is a matter performance. of a compromise, but it can be nearly achieved by employing The main design effort concerning such metasurfaces is an optimization routine developed in [40], where full-wave to obtain a considerable reflection magnitude range while simulation results are combined with postprocessing data [41] keeping the loss as low as possible. This can be achieved at the in order to fulfill the desired goal. unit cell level. Then, owing to the modelling procedure devel- The fabrication process of this unit cell is based on the oped by authors [45], the whole PRS antenna simulation can depositionofa500nmthickgoldlayeranda1500nmthick be performed accurately and rapidly despite potentially large aluminum one on a quartz substrate [42]. It includes the electrical size of the metasurface. possibility to pattern high-resistivity polysilicon bias lines which do not impact on the microwave performance, which is a very important feature when designing MEMS-based 4. Conclusion metamaterials. Monolithic MEMS integration enables structures to have a 3.2. Metasurface Unit Cell with Reflection Magnitude Control. potentially large number of embedded MEMS elements, at Figure 6 shows the layout and photography of the recon- a price that does not depend directly on their number. figurable reflection magnitude metasurface unit cell [43]. This property, combined with other advantages of MEMS International Journal of Antennas and Propagation 5

w y

Sx S y x D=10 D mm S y S x s=3.3mm s/2 p w = 0.35 mm MEMS g g = 0.23 mm p = 0.345 mm

𝜀r = 3.78; tan𝛿≈0 Thickness = 0.5 mm

Metallization: Au Substrate: quartz Bias: polysilicon (a) (b)

Figure 6: MEMS-reconfigurable metasurface unit cell with reflection magnitude control43 from[ ] (a) layout and (b) photography of the fabricated device.

100 90 80 0 110 1.00 70 120 60 −30 −3.00 30 130 50 −6.00 140 40 −9.00 150 30 −60 −12.00 60 160 20 −15.00 0.20 3.00 −18.00 10 170 −21.00 0.00 0.20 0.50 1.00 2.00 5.00 −90 180 0.00 0 90

−170 −10 m3, 4 −5.00 −160 −0.20 −20 m1, 2 −150 −30 −120 120 −140 −40 −0.50 −2.00 −130 −50 −120 −150 −1.00 −60 150 −110 −70 −100 −90 −80 −180 ∘ S = S = m1. . . Sx = on; Sy = off; Γ = 0.49∠−121.4 x off; y off ∘ Sx = off; Sy = on m2. . . Sx = on; Sy = on; Γ = 0.49∠−121.5 Sx = on; Sy = off ∘ m3... Sx = off; Sy = off; Γ = 0.88∠−157.1 Sx = on; Sy = on ∘ m4... Sx =Soff; y = on; Γ = 0.88∠−155.7 (b) (a)

Figure 7: Simulation results from [43] (a) metasurface reflection coefficient as a function of the MEMS state at the operating frequency 𝑓0 = 11.2 GHz and (b) PRS antenna beamwidth as a function of the MEMS state. 6 International Journal of Antennas and Propagation

0.00 200.00 −2.00 150.00 100.00 −4.00 50.00 −6.00 (dB) 0.00

|Γ| −8.00 −50.00 Phase (deg) −100.00 −10.00 −150.00 −12.00 −200.00 10.70 10.95 11.20 11.45 11.70 10.70 10.95 11.20 11.45 11.70 Frequency (GHz) Frequency (GHz) x -pol; measurement SxSy = “01”; sim SxSy = “00”; meas x -pol; simulation SxSy = “01”; meas SxSy = “10”; sim y -pol; measurement SxSy = “00”; sim SxSy = “10”; meas y-pol; simulation (a) (b)

Figure 8: Simulation and measurement results in the OMT environment, from [43] (a) metasurface reflection magnitude and (b) metasurface reflection phase. technology, such as extremely low power consumption, References linearity, and availability of highly resistive bias lines, makes large reconfigurable 1-D and 2-D metamaterial designs with [1] N. Engheta, “An idea for thin subwavelength cavity resonators hundreds or even thousands of MEMS elements feasible. The using metamaterials with negative permittivity and permeability,” IEEE Antennas and Wireless Propagation Letters,vol.1,pp. main limitation of this technology lies in the actuation speed 10–13, 2002. which is typically in the range of microseconds. [2] G. V.Eleftheriades, A. K. Iyer, and P.C. Kremer, “Planar negative MEMS-reconfigurable metamaterials, if properly desig- refractive index media using periodically L-C loaded transmis- ned, can bring valuable improvements to antenna systems, sion lines,” IEEE Transactions on Microwave Theory and Tech- which was demonstrated in this paper. In Section 2,effi- niques,vol.50,no.12,pp.2702–2712,2002. cient MEMS-based CRLH-TLs were presented. It was shown [3] C. Caloz and T. Itoh, “Transmission line approach of left- that they can enable leaky-wave antennas with dynamic handed (LH) materials and microstrip implementation of an beam-scanning and improved array feed networks. In artificial LH transmission line,” IEEE Transactions on Antennas Section 3, MEMS-based metasurface designs with reconfig- and Propagation,vol.52,no.5,pp.1159–1166,2004. urable reflection properties were presented. Their utilization [4] G. V. Eleftheriades and K. G. Balmain, Negative-Refraction in reflectarrays and PRS antennas, resulting in dynamic radi- Metamaterials: Fundamental Principles and Applications,Wiley- ation pattern control (beam-scanning and beamwidth recon- IEEE Press, Hoboken, NJ, USA, 2005. figuration), was discussed and demonstrated. These function- [5]C.CalozandT.Itoh,Electromagnetic Metamaterials, Transmis- alities are welcome in radar systems and space communica- sion Line Theory and Microwave Applications, Wiley-IEEE Press, tions. Hoboken, NJ, USA, 2006. Current trends within the area of reconfigurable metama- [6] S. Hrabar, J. Bartolic, and Z. Sipus, “Waveguide miniaturiza- terials for antenna applications are mainly oriented toward tion using uniaxial negative permeability metamaterial,” IEEE higher operating frequencies. MEMS technology can already Transactions on Antennas and Propagation,vol.53,pp.110–119, be considered as a mature technology, with prominent 2005. properties up to mm wave frequencies. Emerging new tech- [7] M.A.Y.Abdalla,K.Phang,andG.V.Eleftheriades,“Printedand nologies such as electrically actuated elastomers [46, 47], integrated CMOS positive/negative refractive-index phase shif- liquid crystals [48, 49], and graphene [50, 51] promise good ters using tunable active inductors,” IEEE Transactions on Mic- reconfigurable antenna solutions at higher frequencies, up to rowave Theory and Techniques,vol.55,no.8,pp.1611–1623,2007. terahertz bands. [8] F. Bilotti, A. Alu, and L. Vegni, “Design of miniaturized metamaterial patch antennas with 𝜇-negative loading,” IEEE Trans- actions on Antennas and Propagation,vol.56,no.6,pp.1640– Conflict of Interests 1647, 2008. [9] P.-S. Kildal, E. Alfonso, A. Valero-Nogueira, and E. Rajo- The authors declare that there is no conflict of interests Iglesias, “Local metamaterial-based waveguides in gaps bet- regarding the publication of this paper. ween parallel metal plates,” IEEE Antennas and Wireless Prop- agation Letters, vol. 8, pp. 84–87, 2009. Acknowledgment [10] M. Bosiljevac, M. Casaletti, F. Caminita, Z. Sipus, and S. Maci, “Non-uniform metasurface Luneburg lens antenna design,” This work was supported by the Swiss National Science IEEE Transactions on Antennas and Propagation,vol.60,pp. Foundation (SNSF) under Grant no. 133583. 4065–4073, 2012. International Journal of Antennas and Propagation 7

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Research Article Switchable Electromagnetic Bandgap Surface Wave Antenna

Qiang Bai, Kenneth L. Ford, and Richard J. Langley

DepartmentofElectricalandElectronicEngineering,UniversityofSheffield,MappinStreet,SheffieldS13JD,UK

Correspondence should be addressed to Richard J. Langley; [email protected]

Received 3 December 2013; Accepted 20 February 2014; Published 10 April 2014

Academic Editor: Giacomo Oliveri

Copyright © 2014 Qiang Bai et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

This paper presents a novel switchable electromagnetic bandgap surface wave antenna that can support both a surface wave and normal mode radiation for communications at 2.45 GHz. In the surface wave mode, the antenna has a monopole-like radiation ∘ pattern with a measured gain of 4.4 dBi at ±49 and a null on boresight. In the normal mode, the antenna operates like a back-fed microstrip patch antenna.

1. Introduction viasareremovedfromthemushroom-likeEBGstructure. Hence instead of being suppressed, a strong surface wave Manytypesofwearableantennashavebeenproposedin can be exited and radiated on the via-less EBG material. recent years designed for body area networks (BANs). Anten- Basedonthisfeature,severalsurfacewaveantennas(SWA) nas may operate in either of two modes, on-body and off- have been designed to achieve a monopole-like radiation body. On-body communication refers to the transmission pattern on a thin, planar structure [14, 15]. However, in of signals across the human body as a surface wave, which previous designs, the antenna could only support surface requires the wearable antenna to have maximum directivity wave communication, and it was difficult to achieve the along the body surface while, in the off-body (normal) mode, normal mode communication feature as the antenna was the communication occurs away from body to a node such fully covered by the EBG cells. as a base station, which demands that the antenna have the This paper presents a novel switchable surface wave maximum radiation in the boresight direction. Therefore antenna based on bandgap materials which keeps the planar ideally such antennas have to be designed to support each structure and low thickness but can support both surface mode of operation. For many antennas, it is useful if they are wave (on-body) and normal (off-body) modes of com- able to operate without being affected by their environment, munication at 2.45 GHz. Although not designed on textile for example, radiating into the human body or placing them materials, this paper shows the techniques for generating on metal surfaces such as vehicles. Hence, a large ground adualmodeswitchableantenna.Theperformanceofthe plane or electromagnetic bandgap (EBG) structure may be antennas is investigated based on numerical and experi- also necessary to reduce the detuning effect and the backward mental methods. CST Microwave Studio was used for the radiation. Some antenna designs using EBGs as artificial antenna simulations. The surface wave communication mode magnetic conductors (AMCs) have been reported in [1– is further investigated by studying the use of an EBG to couple 5] for off-body communication and in [6–8] for on-body antennas/sensors together around the body. communication. Only a small number of papers have studied a switchable system which can support both on- and off-body 2. Antenna Design communications [9–11]. The mushroom-like EBG structure has been well studied The proposed switchable surface wave antenna comprises and widely used in various designs to improve antenna three layers. The radiating antenna is a 27 mm × 26.5 mm performance [1, 12, 13]. Recent research also observed that microstrip patch printed on a grounded slab and back-fed by the surface wave bandgap may disappear when the vertical aSMAconnector(Figure 1). The feeding point is 6 mm offset 2 International Journal of Antennas and Propagation

190 mm

mm PIN diode Parasitic patch 4 25 mm EBG 100 4 EBG FR z FR4 Feed y

(a) (b)

Figure 1: Geometries of the switchable SWA: (a) top view, EBG based antenna layer with PIN diode switches, (b) side view.

180

135

−45

−90 Reﬂection phase (deg) −135

−180 012345 Frequency (GHz) (a) (b)

Figure 2: (a) Photograph of the fabricated model (without PIN diodes), (b) EBG reflection phase, (c) simulated radiation patterns for antenna with EBG, (d) simulated radiation patterns for antenna without EBG. from the patch centre. The top layer consists of a parasitic 100 × 190 mm. The bias network can be printed on the back identical patch aligned above the lower patch antenna and an side of the top layer to actuate the PIN diodes. And also the optimized EBG surface (Figure 1(a)). The EBG, with 25 mm × PIN diodes can be replaced by varactor diodes to tune the 25 mm unit cell, was designed, so that a surface wave was surface response. excited and propagated when the reflection phase of the The simulated radiation patterns for the antenna are ∘ EBG cell was approximately −90 (Figures 2(a) and 2(b)) shown in Figures 2(a) and 2(b),withandwithouttheEBG [15].Therearesixstripsattheedgesofthesurfaceandsix surface. These show that the radiation pattern with the EBG PIN diodes switching the EBG surface to the strips to give produces a mode that directs the peak radiation away from the radiation characteristics. Both layers are made of FR4 boresight. Based on this, the next section describes the material with a total thickness of 3.2 mm and an overall size of performance. International Journal of Antennas and Propagation 3

0 0 330 30 −5

−15 −10 300 60 −15 −25 −20

−35 −25 −15 (dB) -parameters 270 90 S −25

−30 1.8 2 2.2 2.4 2.6 2.8 3

240 120 Frequency (GHz)

S11 Without EBG With EBG, t=3 With EBG, t=2 With EBG, t=4 210 150

180 Figure 4: S11 of switchable surface wave antenna (when t =2,3, 4mmandwithoutEBG). Without EBG With EBG, t=4 With EBG, t=3 With EBG, t=5

Figure 3: Simulated radiation patterns of switchable surface wave caused by the inaccuracy of the manufacture as a thin air antenna (when t = 3, 4, 5 mm and without EBG). gap is noticed when the antenna top layer and lower layer are assembled together. Unlike in previous EBG surface wave antennas reported in [14, 15], where the resonant frequency 3. Simulation and Measurement highly depends on the EBG, the matching performance of proposed antenna in this paper is only slightly affected when 3.1. Simulation. Initially, the EBG layer, with parasitic patch the top EBG structure is switched, giving an opportunity to inclusion, was spaced a distance, t,fromthefedpatch. tailor the antenna radiation pattern without changing the Figure 3 plots the simulated y-z plane radiation patterns of resonant frequency. the patch antenna alone and the patch antenna with the The measured y-z plane radiation patterns for the switch- EBG surface for varying distances, t. For easy comparison, able surface wave antenna are plotted in Figure 7.Inthe all radiation patterns in this paper are normalized. From surfacewavemode,theantennahasthemaximumdirectivity ∘ Figure 3, the peak power from the radiation pattern varies at ±49 withameasuredgainof4.4dBiandanullis ∘ from boresight for the patch antenna to approximately 50 also obtained in the boresight direction. When switched to with the EBG when the separation, t,issetto4mmand the normal mode, the antenna mainly radiates towards the a null appears at boresight. There is an appreciable amount boresight direction (red dotted line in the figure) with a gain of radiation directed sideways from the antenna, along the of 3.7 dBi. It is also noted, due to the large ground plane, EBG. It is also found in Figure 4 that the antenna matching that the antenna has low backward radiation in both modes. performance was not significantly affected, maintaining the Overall,thesimulatedandmeasuredperformanceofthe resonance at about 2.4 GHz. antenna are in good agreement. The operation of the EBG antenna is clearly shown by the current distributions on the surface in the two modes (Figure 5). For the PIN diodes shorted in Figure 5(a),the 4. EBG Waveguide currents are confined to the parasitic patch and the six The switchable antenna presented in the previous sections elements of the EBG closest to it which form the antenna allows reconfiguration of the radiation pattern. For applica- radiating away from the patch. With the PIN diodes open tions such as body worn antennas, there is a need to acquire circuit, the whole of the EBG is excited directing the radiation data from a number of sensors and antennas around the along the antenna surface. body to a central node for transmission to a base station far away from the body. This is difficult to achieve by line of 3.2. Measurement. Figure 6 presents the measured reflection sight due to blockage by the body and losses tend to be very coefficient of the switchable surface wave antenna to be high at >40 dB. Transmission lines could be used to connect compared with the simulation in Figure 4. In both modes (pin the sensors but in this section, the potential use of an EBG open and short circuit), the antenna has −10 dB bandwidth propagatingthesurfacewavemodewillbepresented.This covering 2.4 GHz to 2.47 GHz. The antenna working band follows directly from the study above. The concept is shown is shifted upwards by 20 MHz, covering the band from in Figure 8 where two switchable antennas are connected 2.42 GHz to 2.49 GHz in the measurements. This may be together using the EBG designed in Section 2 and operating 4 International Journal of Antennas and Propagation

8 8 7.27 7.27 6.55 6.55 5.82 5.82 5.09 5.09 4.36 4.36 (m) 1 1 (m) A 3.64 3.64 A 2.91 2.91 2.18 2.18 1.45 1.45 0.727 0.727 0 0

PIN short PIN open

(a) (b)

Figure 5: Surface currents on EBG antenna. (a) PIN diodes shorted. (b) PIN diodes open.

0 0 330 30 −5 −5 −10 −15 −15 300 60 −20 −25 −25 -parameters (dB) -parameters

S −35 −25 −15 −5 −30 270 90 −35

−40 1.8 2 2.2 2.4 2.6 2.8 3

Frequency (GHz) 240 120 PIN open PIN short 210 150 Figure 6: Measured S11 of switchable surface wave antenna. 180

PIN open in the surface wave mode. The antennas remain switchable PIN short andareseparatedfromcentretocentreby315mm.The Figure 7: Measured radiation patterns of switchable surface wave bandgap surface is shown as 3 elements wide by 6 elements antenna in y-z plane for pin diodes open and short circuit. long but this is for demonstration only and can be extended lengthwise as required. Three unit cell elements in width are sufficient for the EBG to operate satisfactorily rather than 2 elements. Five-element width was also tested, but there was no significant improvement obtained on transmission performance. Figure 9 shows the surface currents for the cases with the PIN diodes both open and short circuit and it is clear that the EBG propagates a wave from one antenna to the other when the diodes are open circuit. The transmission coefficients, 315 mm S21,forbothcasesareplottedinFigure 10 where simulations and measurements are compared. When the PIN diodes are Figure 8: Antennas coupled using an EBG. International Journal of Antennas and Propagation 5 0 109 182 218 400 327 145 255 291 364 72.7 36.4 V (m) (a) Upper plot PIN open circuit 0 109 182 218 400 327 145 255 291 364 72.7 36.4 V (m) Figure 11: Radiation pattern for complete antenna/EBG with PIN (b) Lower plot PIN short circuit diodes short circuit, normal mode.

Figure 9: Surface currents on EBG coupled antenna system. switch between the normal mode (radiation away from 0 antenna)andthesurfacewavemodewiththeEBGactingas −10 a transmission medium as in Figure 9. To further demonstrate propagation across the antennas −20 with and without the connecting EBG, the antenna system is −30 bent around a tissue equivalent phantom [6] with a diameter of 100 mm as shown in Figures 12(a) and 12(b) respectively. −40 In Figure 12(c) the surface currents excited on the antenna −50 structure are shown in both cases. From Figure 12(c),itis clear that the EBG propagates the wave around the bend to −60 the second antenna.

Transmission coefficient (dB) coefficient Transmission −70 The simulated transmission coefficients21 (S )areplotted −80 in Figure 13 for the bent geometry and compared with the 1.8 2 2.2 2.4 2.6 2.8 3 value when the surface was flat. The coupling for the curved Frequency (GHz) surface with the EBG increases from −45 dB without the EBG to −23 dB, although it does not reach the value of −18 dB Measured PIN open cct Simulated PIN open cct fortheflatsurfacewithEBG.TypicalS21 between antennas Measured PIN short cct Simulated PIN short cct located at top and bottom of this tissue equivalent phantom − Figure 10: Measured and simulated transmission coefficientsS ( 21) is below 40 dB [6]. between switchable mode antennas. 5. Conclusion open circuit, the measured transmission was −21 dB between This paper presents a novel EBG surface wave antenna which the antennas while the calculated value was about −23 dB. can be switched by PIN diodes to support surface wave and When the PIN diodes were short circuited, the transmission normal mode communications at 2.45 GHz. In the surface between the antenna and the EBG is considerably reduced wave mode, a monopole-like radiation pattern is obtained ∘ (see Figure 5(a)) and hence the EBG carries little signals to the with a measured gain of 4.4 dBi at ±49 .Inthenormal other antenna and the measured transmission falls to −33 dB mode, the antenna mainly radiates towards the boresight for the measurement and −41 dB in the simulation. Manu- direction. Coupling two antennas together using the EBG facturing difficulties probably account for the discrepancy design in surface propagation mode allows the wave to travel between the measured and calculated results as there was a round a bent surface while maintaining the radiation pattern small uneven air gap between the layers of the structure. In switching ability. Due to the thin and planar structure, the addition, for the simulation, the PIN diodes are assumed to EBG surface wave antenna could be easily incorporated into be perfect switches. clothing for body worn applications. Coupling losses using Figure 11 shows the simulated 3D radiation pattern from theEBGarebetterthandirectradiationcouplingbetweenthe the whole structure when operating in the normal mode antennas (typically −50 dB) when the signal travels round the showing that the full structure with the EBG was able to torso. 6 International Journal of Antennas and Propagation

D = 100 mm

(a) Antennas without EBG connection (b) Antennas with EBG 400 400 364 364 327 327 291 291 255 255 218 218 182 (m) 182 (m) V V 145 145 109 109 72.7 72.7 36.4 36.4 0 0

Figure 12: Antennas bent on a radius of 100 mm.

0 Acknowledgment −10 This work is funded by the UK Engineering and Physical −20 Sciences Research Council (EPSRC) Grant Reference no. −30 EP/G056633.

−40 References −50 [1] S. Zhu and R. Langley, “Dual-band wearable textile antenna −60 on an EBG substrate,” IEEE Transactions on Antennas and

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[6]G.A.ConwayandW.G.Scanlon,“Antennasforover-body- surface communication at 2.45 GHz,” IEEE Transactions on Antennas and Propagation,vol.57,no.4,pp.844–855,2009. [7] M. R. Kamarudin, Y. I. Nechayev, and P. S. Hall, “Antennas for on-body communication systems,” in Proceedings of the IEEE International Workshop on Antenna Technology (IWAT ’05),pp. 17–20, March 2005. [8] L. Akhoondzadeh-Asl, P. S. Hall, and Y. Nechayev, “Novel conformal surface wave Yagi antenna for on-body communication channel,”in Proceedings of the IEEE International Symposium on Antennas and Propagation (AP-S/URSI ’10),pp.1–4,July2010. [9] A. R. Chandran, G. A. Conway, and W. G. Scanlon, “Pattern switching compact patch antenna for on-body and off-body communications at 2.45 Ghz,”in Proceedings of the 3rd European Conference on Antennas and Propagation (EuCAP ’09),pp. 2055–2057, March 2009. [10] J. Kelly, L. Ford, and R. Langley, “Slotline structure for on/off- body communications at 2.45 GHz,” in Proceedings of the 5th European Conference on Antennas and Propagation (EUCAP ’11), pp. 525–529, April 2011. [11] R. J. Langley, K. L. Ford, and Hyung-Joo Lee, “Switchable on/off- body communication at 2.45 GHz using textile microstrip patch antenna on stripline,” in European Conference on Antennas and Propagation (EuCAP ’12),pp.728–731,2012. [12] P. Salonen and Y. Rahmat-Samii, “Textile antennas: effects of antenna bending on input matching and impedance bandwidth,” IEEE Aerospace and Electronic Systems Magazine,vol. 22,no.12,pp.18–22,2007. [13]D.Sievenpiper,L.Zhang,R.F.JimenezBroas,N.G. Alexopolous,¨ and E. Yablonovitch, “High-impedance electromagnetic surfaces with a forbidden frequency band,” IEEE Transactions on Microwave Theory and Techniques,vol. 47, no. 11, pp. 2059–2074, 1999. [14] F. Yang, A. Aminian, and Y. Rahmat-Samii, “A novel surface- wave antenna design using a thin periodically loaded ground plane,” Microwave and Optical Technology Letters,vol.47,no.3, pp.240–245,2005. [15] R. Khouri, P.Ratajczak, P.Brachat, and R. Staraj, “Athin surface- wave antenna using a via-less EBG structure for 2.45 GHz on-body communication systems,” in Proceedings of the 4th European Conference on Antennas and Propagation (EuCAP ’10), pp. 1–4, April 2010. Hindawi Publishing Corporation International Journal of Antennas and Propagation Volume 2014, Article ID 532634, 11 pages http://dx.doi.org/10.1155/2014/532634

Research Article Tunable Plasmonic and Hyperbolic Metamaterials Based on Enhanced Nonlinear Response

Christos Argyropoulos, Francesco Monticone, Nasim Mohammadi Estakhri, and Andrea Alù DepartmentofElectrical&ComputerEngineering,TheUniversityofTexasatAustin,Austin,TX78712,USA

Correspondence should be addressed to Andrea Alu;` [email protected]

Received 2 December 2013; Accepted 18 February 2014; Published 6 April 2014

Academic Editor: Giacomo Oliveri

Copyright © 2014 Christos Argyropoulos et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

We present here tunable and reconfigurable designs of linear and nonlinear plasmonic and hyperbolic metamaterials. Rich scattering features of multilayered composite nanoparticles are demonstrated, which include complex and exotic scattering signatures combining multiple dipolar Fano resonances and electromagnetic induced transparency (EIT) features. These dipole- dipole multi-Fano scattering responses can be further tuned through altering the plasmonic properties of the concentric layers or the permittivity of the core, for instance, by the presence of nonlinearities. Strong third-order nonlinear effects, such as optical bistability, may also be induced in the scattering response of nonlinear nanoparticles due to the highly enhanced and confined fields inside their core. Nonlinear hyperbolic metamaterial designs are also explored, which can realize tunable positive-to-negative refraction at the same frequency, as a function of the input intensity. Negative Goos-Hanchen¨ shift is demonstrated based only on the hyperbolic dispersion properties of these layered metamaterials without the usual need of negative index metamaterials. The Goos-Hanchen¨ shift may be tuned from positive-to-negative values, when the structure is illuminated with different frequencies. A plethora of applications are envisioned based on the proposed tunable metamaterials, such as ultrafast reconfigurable imaging devices, tunable sensors, novel nanotag designs, and efficient all-optical switches and memories.

1. Introduction fields [16], temperature [17, 18], liquid crystals [19, 20], graphene [21, 22], phase-change media [23, 24], and elec- Metamaterials are artificially constructed materials that can trooptical effects25 [ –27]. exhibit novel functionalities not available in nature. Unusual Nonlinear optical effects can also be strongly enhanced electromagnetic properties, such as negative refraction [1] with plasmonic metamaterial structures, mainly due to the and invisibility [2, 3], have been achieved with differ- highly localized fields obtained in these structures. For exam- ent metamaterial structures. These interesting properties ple, we recently demonstrated that strong optical bistability have been obtained with various metamaterial designs at can arise when nonlinear plasmonic waveguides operate at microwaves, terahertz, optical, and ultraviolet frequencies their cut-off frequency [28]. Furthermore, inspired by the [4–10]. Recently, increased attention has been dedicated to recent interest in the concept of Fano scattering resonances the study of reconfigurable and tunable metamaterials and [29], we reported giant all-optical scattering switches based nonlinear plasmonic devices, where even more exotic and on nonlinear core-shell plasmonic nanoparticles [30]. The breakthrough functionalities may be achieved [11, 12]. Sev- proposed nanoparticles exhibited purely dipolar Fano reso- eral reconfigurable and tunable metamaterial and plasmonic nances, in contrast to more conventional Fano resonances devices have been presented in the recent literature. Their based on the interaction and coupling of dipolar and higher- operation may be controlled with an applied voltage (varac- order scattering modes supported by complex subwavelength tors) [13, 14], electromagnetic forces [15], external magnetic plasmonic systems [31–35]. The concept of purely dipolar 2 International Journal of Antennas and Propagation

ac2 ac4 ac3 ac2 ac1 ac1

𝜀 a 𝜀 core core a

Plasmonic shells

(a) (b)

Figure 1: Geometry of composite nanoparticles consisting of (a) two and (b) four plasmonic layers, respectively. The core is made of dielectric material.

Fano resonances has also been extended to multilayered permittivity 𝜀𝑐2, which follows a similar Drude dispersion plasmonic nanoparticles, inducing Fano-comb scattering with a plasma frequency increased by Δ𝜔𝑝.Onewayto responses [36, 37]. Cloaking and resonant scattering states achieve this effect could be to modify the doping level of excited within the proposed plasmonic shells may be used, plasmonic semiconductors, similar to our previous work [37] respectively,asthedarkandbrightmodestoinducemultiple in which nanoparticles with four plasmonic shells based dipolar Fano-like scattering features. on aluminum-doped zinc oxide (AZO) semiconductors [41] In this paper, we further study the interesting scattering were utilized to produce Fano scattering combs. properties of multilayered plasmonic nanoparticles in order Following the analysis presented in [36], we calculate the to demonstrate tunable Fano-comb operation for sensing quasi-static conditions for resonant scattering and cloaking and nanotagging applications. Third-order optical nonlinear based on Mie theory [42]. The contour plots of the total materialsloadedinthecoreoftheseplasmoniccomposite SCS of this composite nanoparticle with two plasmonic layers nanoparticlesareshowntoinducelargebistabilityforeach are shown in Figures 2(a)–2(d) as a function of the aspect dipolar Fano resonance, due to the enhanced and strongly ratio 𝜂𝑐 and the wavelength of operation, where the plasma localized electric fields inside the core of the device. More- frequency of the second plasmonic layer is increased by (a) over, reconfigurable nonlinear hyperbolic layered metama- Δ𝜔𝑝 = 750 THz, (b) Δ𝜔𝑝 = 500 THz, (c) Δ𝜔𝑝 = 250 THz, terial structures will be studied, which may achieve tunable and (d) Δ𝜔𝑝 =50THz in each case, respectively. The red positive-to-negative refraction as a function of the input radi- and blue thick curves in these plots indicate the dispersion of ation intensity [38]. We will also demonstrate that hyperbolic quasi-static conditions for resonant scattering and cloaking, metamaterials can realize strong negative Goos-Hanchen¨ respectively. We fix the geometry picking a large aspect ratio shifts [39] without the need of negative refractive index meta- 𝜂𝑐 = 0.91 and calculate the dynamic normalized scattering 2 materials. Reconfigurable operation for Goos-Hanchen¨ shift response (SCS/𝜆 ),whichisshowninFigures2(e)–2(h) for will be presented based on layered metamaterial structures as each increased plasma frequency of the second plasmonic a function of the frequency of the impinging radiation. layer. The dimensions of the plasmonic layers are equal to 𝑎𝑐1 = 𝑎/𝜂𝑐 =55nm and 𝑎𝑐2 =60nm. In Figures 2(e)–2(h) it canbeseenthattwopurelydipolarsharpFanoresonancesare 2. Plasmonic Composite Nanoparticles formed in the scattering response of the composite plasmonic nanoparticle for different plasma frequencies of the second Two composite plasmonic nanoparticle geometries are con- layer. It is interesting that the second Fano resonance is sideredinthefollowing,asshowninFigure 1.First,we getting closer to the first as we reduce Δ𝜔𝑝,whereasthe calculate the scattering response of the geometry depicted first one stays almost unperturbed at the same wavelength in Figure 1(a), consisting of a dielectric core with radius position. This demonstrates the high tunability potential of 𝑎=50 𝜀 =2 nmandrelativepermittivity core and two the proposed double-layer nanoparticle design, especially for 𝜂 = 𝑎/𝑎 plasmonic coating layers. The aspect ratio 𝑐 𝑐1 is the second Fano-like resonance. defined as the ratio between the radius of the dielectric core Next, the same nanoparticle of Figure 1(a) is used, fixing andtheradiusofthefirstplasmoniclayer.Bothplasmonic the plasmonic properties of the second layer to be similar layers are considered to have the same thicknesses. The to the first layer, but with an increased plasma frequency first layer is assumed to follow a lossless Drude permittivity Δ𝜔𝑝 = 325 THz. Moreover, the dimensions of the core are 𝜀 =𝜀 −𝜔2 /[𝜔(𝜔 + 𝑖𝛾)] 𝜔 = 2175 dispersion with 𝑐1 ∞ 𝑝 , 𝑝 THz, changed to 𝑎=10nm. The contour plots of the quasi-static 𝛾=0THz, and 𝜀∞ =5,fittedtotherealpartofthe SCS are plotted in Figures 3(a)–3(d) as a function of the −𝑖𝜔𝑡 experimentally retrieved silver dispersion [40], under an 𝑒 aspect ratio 𝜂𝑐 and of the wavelength of operation, similar time convention. The second plasmonic layer has a relative to Figure 2, but here the permittivity of the dielectric core International Journal of Antennas and Propagation 3

1.0 0 0

0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 Δ𝜔p = 750 THz −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (a) (e) 1.0 0 0

0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 Δ𝜔p =500THz −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (b) (f) 1.0 0 0

0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 Δ𝜔p =250THz −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (c) (g) 1.0 0 0

0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40

Δ𝜔p =50 THz −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (d) (h)

Figure 2: (a)–(d) Contour plots of the total SCS (normalized to the maximum value) as a function of the wavelength and the aspect ratio 𝜂𝑐 2 and (e)–(h) normalized SCS/𝜆 versus wavelength, for different plasma frequencies in the two plasmonic layers of the structure in Figure 1(a): (a), (e) Δ𝜔𝑝 = 750 THz, (b), (f) Δ𝜔𝑝 = 500 THz, (c), (g) Δ𝜔𝑝 = 250 THz, and (d), (h) Δ𝜔𝑝 =50THz.Theaspectratioisfixedto𝜂𝑐 = 0.91 (indicated by the dashed arrow in panels (a)–(d)). Cloaking (blue lines) and resonant scattering (red lines) quasi-static conditions are also shown in the contour plots (a)–(d). 4 International Journal of Antennas and Propagation

𝜀 = 1.5 is different in each case, taking the values (a) core , the degeneracy between cloaking and resonant states, in both 𝜀 = 4.5 𝜀 = 7.5 𝜀 =10 (b) core ,(c) core , and (d) core .Now, cases the fields and power level are enhanced and strongly a lower aspect ratio is picked, 𝜂𝑐 = 0.33,andthedynamic confined inside and around the dielectric core. Conversely, 2 normalized scattering response (SCS/𝜆 ) is computed and outside the nanoparticle the field distributions are dras- plotted in Figures 3(e)–3(h) for each permittivity value of tically different: the resonant state causes large scattering the core. In these cases, the dimensions of the plasmonic around the nanoparticle (point I), whereas at the cloaking layers are equal to 𝑎𝑐1 = 𝑎/𝜂𝑐 =30nm and 𝑎𝑐2 =50nm. wavelength the power flow is almost unperturbed (point Figures 3(e)–3(h) demonstrate that, by changing the core II), even right outside of the outer plasmonic layer of the permittivity, we are able to modify and control the frequency shell. positionsofbothcloakingandresonantscatteringstates, The strong field enhancement inside the core represents but selectively tuning only the lower-frequency narrowband an ideal condition to boost weak optical nonlinear effects. Fano resonance. Similar to the previous results, the Fano Based on this idea, we propose a nonlinear plasmonic design resonance found at higher frequencies stays fixed, with its in which the core is composed of third-order Kerr nonlinear shape being unaffected. Hence, tunable operation can also 𝜀 =2.2+𝜒(3)|𝐸|2 𝜒(3) = material with permittivity core ,where −20 2 2 be obtained by changing the core permittivity, leading to 4.4 × 10 m /V is typical value of nonlinear dielectric reconfigurable Fano scattering responses. Finally, we note materials [43]and|𝐸| is the magnitude of the mean value of that the complex scattering signature shown in Figure 3(h) the local complex electric field in the core of the nanoparticle, 𝜆 ≅ 450 𝜆≅ includes both EIT ( nm) and Fano responses ( calculated using full-wave Mie theory. It is expected that a 350 nm) in the same scattering spectrum. Excitingly, these small change in permittivity of the nonlinear core, induced features can be observed from any angle of observation, by an increase in light intensity, will strongly modify the due to the purely dipolar nature of the modes involved in scattering spectrum of the plasmonic nanostructure. The these effects. The EIT scattering signature is characterized nonlinear scattering response of the proposed nanoparticle by a symmetric scattering response, featuring a sharp dip is calculated and plotted in Figure 4 (red line), when it in between two resonant peaks, and it typically arises when is illuminated with a moderate input intensity radiation two strong resonant states closely interact in frequency. The 𝐼 = 173 / 2 in MW cm . Bistable scattering performance is Fano response, on the contrary, arises when a broad dipolar obtained, with broader bistability induced at the second scattering state, or continuum, interacts with a dark state [29], Fano resonance. Both narrowband resonances experience and it is typically characterized by an asymmetric scattering an abrupt switching effect spanning almost 50 dB of total signature. It is interesting that almost 50 dB scattering con- scattering reduction. Note that in this plot we also present the trast, across a very narrow frequency range, may be obtained calculated unstable branch indicated by the dashed red line. in the dipolar Fano resonance examples discussed here. Note The nonlinear nanoparticle has a reconfigurable response and that the broad cloaking dip shown in Figures 3(e)–3(h) at its scattering behavior is dependent upon the previous values approximately 300 nm coincides with the “dark” scattering of input radiation intensities, similar to an optical memory. state of the plasmonic cloak [3, 30]. It is based on a single The scattering signature can switch between different values, dipolarmodeand,quiteinterestingly,itisweaklyaffectedby which correspond to different branches of the bistability the core permittivity values. excursion for both dipolar Fano resonances. Hence, the scat- The previous examples clearly demonstrate that the de- tering response is characterized by a nonlinear Fano comb generate states of cloaking and resonant scattering in com- with reconfigurable and tunable exotic scattering features. A posite nanoparticles are tunable and strongly modifiable plethora of potential applications are envisioned based on by just changing the core permittivity. The degeneracy these nonlinear nanoparticles, such as reconfigurable optical between the two scattering states guarantees strong field nanotags and tunable sensors. enhancement inside the nanoparticle’s core at both the Next, a four-layer plasmonic composite nanoparticle is cloaking and resonant frequency, which is the ideal con- analyzed, with geometry shown in Figure 1(b). The relative 𝜀 =10 dition for enhanced optical nonlinear operation. Now, a permittivity of the core is chosen to be core and its 𝑎 = 150 double-layer plasmonic composite nanoparticle is consid- radius is nm. In this case, the plasmonic layers are ered, with geometry shown in Figure 1(a) and dimensions made of AZO semiconductors with different doping levels. 𝑎=3nm, 𝑎𝑐1 =15nm, and 𝑎𝑐2 =45nm. The scat- The first plasmonic layer follows a lossless Drude dispersion 2 2 tering response of this nanoparticle with core permittivity with relative permittivity 𝜀𝑐1 =𝜀∞ −𝜔𝑝1/𝜔 ,where𝜀∞ =3.3 𝜀 = 2.2 core is computed and plotted in Figure 4 (blue line) in and 𝜔𝑝 = 2213 THz [37]. The other plasmonic layers follow a narrow wavelength range. This geometry was optimized in the same Drude dispersion, but assuming different doping order to sustain two closely spaced dipolar Fano resonances levels, leading to an increase in plasma frequency by Δ𝜔𝑝 with similar shape. The dimensions of the core were chosen foreachlayer,thatis,theplasmafrequencyofeachlayer to be small in order to further increase the field intensity. will be 𝜔𝑝𝑛 =𝜔𝑝1 +(𝑛−1)Δ𝜔𝑝,where𝑛 is the index of The electric field (left insets in Figure 4)andpowerflow plasmonic layers of the composite nanoparticle. In this case, (right insets in Figure 4) distributions are plotted in the the aspect ratio 𝜂𝑐 is again equal to the ratio between radius 𝐸 plane at the Fano-like resonant peak (point I, upper of the dielectric core and radius of the first plasmonic layer insets) and cloaking dip (point II, lower insets), for the first 𝜂𝑐 = 𝑎/𝑎𝑐1. All four plasmonic layers are assumed to have the dipolar Fano resonance. As we discussed above, thanks to same thickness. International Journal of Antennas and Propagation 5

1.0 0 0 𝜀 = 1.5 core 0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 𝜀 = 1.5 core −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (a) (e) 0 0 1.0 𝜀 =4.5 core 0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 𝜀 =4.5 core −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (b) (f) 0 0 1.0 𝜀 =7.5 core 0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ −40 0.2 𝜀 =7.5 core −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (c) (g) 1.0 0 0 𝜀 =10 core 0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 𝜀 =10 core −45 −50 250 300 350 400 450 500 200 250 300 350 400 450 500 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (d) (h)

Figure 3: (a)–(d) Contour plots of the total SCS (normalized to the maximum value) as a function of the wavelength and the aspect ratio 𝜂𝑐 /𝜆2 𝜀 = 1.5 and (e)–(h) normalized SCS versus wavelength, for different core permittivities of the structure in Figure 1(a): (a), (e) core ,(b),(f) 𝜀 = 4.5 𝜀 = 7.5 𝜀 =10 𝜂 = 0.33 core ,(c),(g) core , and (d), (h) core . The aspect ratio is fixed to 𝑐 (indicated by the dashed arrow in panels (a)–(d)). Cloaking (blue lines) and resonant scattering (red lines) quasi-static conditions are also shown in the contour plots (a)–(d). 6 International Journal of Antennas and Propagation

3 +2 3 +5

(I) (I) 2 2

1 1

c2 0 c2 0 z/a z/a

−1 −1

−2 −2

−3 −2 −3 0 −3 −2 −1 0 1 2 3 −3 −2 −1 0 1 2 3

x/ac2 x/ac2

2 I =173MW/cm 0 in

(I) −10

−20 (dB)

2 −30 𝜆

SCS/ −40

−50 (II) −60 336 340 344 348 352 𝜆 (nm)

Linear Non linear 3 +2 3 +5

(II) (II) 2 2

1 1

c2 0 c2 0 z/a z/a

−1 −1

−2 −2

−3 −2 −3 0 −3 −2 −1 0 1 2 3 −3 −2 −1 0 1 2 3

x/ac2 x/ac2

Figure 4: Normalized scattering response versus wavelength, for linear (blue line) and nonlinear (red line) dielectric core for the geometry in Figure 1(a). The electric field (left insets) and power flow (right insets) distributions are plotted onthe 𝐸 plane at the higher-frequency Fano-like resonant peak (I) and cloaking dip (II) when linear operation is considered. The nonlinear plasmonic nanoparticle is illuminated 𝐼 = 173 / 2 with input intensity in MW cm . The unstable branch of the nonlinear bistable response is plotted with a dashed red line. International Journal of Antennas and Propagation 7

1.0 0 0

0.8 −10

−20

0.6 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 Δ𝜔p =500THz Δ𝜔p =500THz −45 −50 1000 1200 1400 1600 1800 2000 1000 1200 1400 1600 1800 2000 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (a) (d) 1.0 0 0

0.8 −10

0.6 −20 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ 0.2 −40 Δ𝜔p =275THz Δ𝜔p =275THz −45 −50 1000 1200 1400 1600 1800 2000 1000 1200 1400 1600 1800 2000 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (b) (e) 1.0 0 0

0.8 −10

−20

0.6 (dB) c 2 𝜂 𝜆 0.4 (dB) −30 SCS/ −40 0.2 Δ𝜔p =50THz Δ𝜔p =50THz −45 −50 1000 1200 1400 1600 1800 2000 1000 1200 1400 1600 1800 2000 Wavelength 𝜆 (nm) Wavelength 𝜆 (nm) (c) (f)

Figure 5: (a)–(c) Contour plots of the total SCS (normalized to the maximum value) as a function of wavelength and aspect ratio 𝜂𝑐.(d)–(f) 2 normalized SCS/𝜆 versus wavelength, for different plasma frequencies in the four plasmonic layers of the geometry of Figure 1(b): (a), (d) Δ𝜔𝑝 = 500 THz, (b), (e) Δ𝜔𝑝 = 275 THz, and (c), (f) Δ𝜔𝑝 =50THz. The aspect ratio is fixed in these cases to 𝜂𝑐 = 0.952 (indicated by the dashed arrow in panels (a)–(c)). Cloaking (blue lines) and resonant scattering (red lines) quasi-static conditions are also shown in the contour plots (a)–(c).

2 Applying Mie theory consistent to the analysis presented scattering response (SCS/𝜆 ) is calculated for a large aspect in [37], we calculate the quasi-static conditions for resonant ratio 𝜂𝑐 = 0.952 and plotted in Figures 5(d)–5(f) for different scattering and cloaking states of each dipolar Fano resonance. plasma frequencies of each plasmonic layer. The dimensions The contour plots of the total SCS of this four-layer plasmonic of the four plasmonic layers are equal to 𝑎𝑐1 = 𝑎/𝜂𝑐 = composite nanoparticle are shown in Figures 5(a)–5(c) as 157.5 nm, 𝑎𝑐2 = 165 nm, 𝑎𝑐3 = 172.5 nm, and 𝑎𝑐4 = 180 nm. a function of the aspect ratio 𝜂𝑐 and the wavelength of Multiple purely dipolar sharp Fano resonances can be seen in operation, and the plasma frequency of each plasmonic Figures 5(d)–5(f), creating a Fano-comb scattering response. layer is increased by (a) Δ𝜔𝑝 = 500 THz, (b) Δ𝜔𝑝 = Note that the narrowband Fano resonances, produced at near 275 THz, and (c) Δ𝜔𝑝 =50THz, respectively. Rich and exotic infrared wavelengths, get closer together as we reduce Δ𝜔𝑝, scattering spectra are obtained for this multilayered plas- whereas the broader conventional dipolar resonance stays monic nanoparticle, with multiple dipolar Fano resonances almost unperturbed at the same wavelength position 𝜆= (Fano-comb scattering signature). The dynamic normalized 1650 nm. Hence, tunable Fano comb scattering signatures 8 International Journal of Antennas and Propagation

Positive Negative refraction refraction

0.8

0.6

𝜆=311nm 0.4 Transmission

0.2

0 0 0.5 1 1.5 2 2.5 3 I 2 in (GW/cm ) x (b) k0 20 E 𝜃 z 𝜃 15

y 10 H d2 d1 5

(3) 𝜒 -NOM 0 Ag Angle of transmission Angle of (a) −5

−10 0 0.5 1 1.5 2 2.5 3 I 2 in (GW/cm ) (c)

Figure 6: (a) Geometry of a nonlinear hyperbolic metamaterial with tunable response. (b) Transmittance and (c) angle of transmission (in degrees) as a function of the input intensity for the structure shown in panel (a), with third-order Kerr nonlinearity introduced in the dielectric layers. The results are computed for incident radiation wavelength of 𝜆 = 311 nm.

maybeobtainedaswemodifythedopinglevelsofthe 𝜔𝑝 = 2175 THz and 𝜀∞ =5, based on the experimental semiconductor material used for each plasmonic layer of data retrieved from [40]. The third-order nonlinear dielectric (3) 2 the composite nanoparticle. Reconfigurable optical tagging material has a relative permittivity 𝜀𝑑 =𝜀𝐿 +𝜒 |𝐸| ,where (3) −18 2 2 applications may be achieved with these multilayer nanopar- 𝜀𝐿 = 0.1 and 𝜒 = 4.4 × 10 m /V , a common value ticle designs, exhibiting tunable scattering response based on for semiconductor nonlinear materials [43]. The nonlinear the doping level of each plasmonic semiconductor shell. permittivity directly depends on the local electric field inside the nonlinear layers induced by the incident radiation intensity. This device is characterized by hyperbolic dispersion and, 3. Hyperbolic Metamaterials as a result, negative refraction without requiring a negative refractive index may be obtained over a relatively broad Inthefollowing,westudytunableandreconfigurableeffects frequency range, as shown in [38]. in hyperbolic metamaterial structures. The geometry of The structure is illuminated by a monochromatic plane the layered nonlinear metamaterial structure is shown in wave operating at 𝜆 = 311 nm and at an incidence angle 𝜃𝑖 = ∘ Figure 6(a). The structure may be characterized as an inho- 45 . The transmittance and the angle of transmission (relative mogeneous anisotropic slab composed of 20 alternating to normal direction), varying the input radiation intensity, layers of Kerr-nonlinear dielectric material and lossless are plotted in Figures 6(b) and 6(c), respectively. Two peaks plasmonic silver (Ag). The thicknesses of both layers are in transmission (Figure 6(b)) are obtained at two different 𝑑 =50 𝑑 =25 𝐼 = 0.78 / 2 𝐼 = 2.1 / 2 subwavelength with values 1 nm and 2 nm, input intensities: in GW cm and in GW cm . respectively. The dispersion of the plasmonic Ag material is These peaks correspond to positive and negative angles of 𝜀 =𝜀 −𝜔2 /𝜔2 modeled by a Drude permittivity Ag ∞ 𝑝 ,where the transmitted wave, as seen in Figure 6(c). The response of International Journal of Antennas and Propagation 9

𝜆=323nm 𝜆 = 352 nm PEC 3 3

PEC 2 2

x k 0 d E 3 𝜃 𝜃 z 1 1 H y d d1 2

𝜀d Ag 0 0 (a) (b) (c)

Figure 7: (a) Geometry of a hyperbolic metamaterial terminated by a PEC (mirror) slab with thickness 𝑑3 = 300 nm. Tunable (b) positive and (c) negative Goos-Hanchen¨ shifts obtained at two different illumination wavelengths: 𝜆 = 323 nm and 𝜆 = 352 nm, respectively.

the layered nonlinear hyperbolic metamaterial may drasti- on double negative metamaterials [44]. To sum up, tunable cally change, as we increase the input intensity, from positive Goos-Hanchen¨ shift, positive to negative, may be obtained to negative refraction at the same frequency. Hence, tunable with the same hyperbolic metamaterial as a function of and reconfigurable refraction may be achieved as a function the frequency of operation. With the addition of proper of the input radiation intensity. nonlinearity or tunable materials, it may be also possible to Next, a different layered hyperbolic metamaterial is stud- induce these effects at the same frequency of operation. ied, terminated this time by a perfect electric conductor (PEC), operating as a mirror for electromagnetic waves. The geometry of this structure is shown in Figure 7(a).Inthis case, the transmission is equal to zero and only reflected 4. Conclusions waves exist. The layers are now composed of dielectric material with relative permittivity 𝜀𝑑 =2and thickness To conclude, we have presented here tunable and reconfig- 𝑑1 =50nmandsilverwiththesameDrudedispersion urable plasmonic metamaterial designs using linear and non- described before and thickness 𝑑1 =25nm. The PEC slab is linear materials. Multilayered plasmonic composite nanopar- used to terminate the layered metamaterial and has thickness ticles have been demonstrated to achieve exotic and complex 𝑑3 = 300 nm. The back mirror may be realized with different scattering responses, which may be tuned at will, as we change metals, which approximate the PEC properties at optical the doping level of the semiconductor layers or the permittiv- frequencies when their thickness is much larger than their ity of the dielectric core. Dipolar Fano resonances, EIT-like skin depth (𝛿 ≈ 30 nm). effects, and Fano scattering combs have been obtained based The Goos-Hanchen¨ shift39 [ ]ofthereflectedwavefrom on these composite nanoparticles. Their scattering response this structure is analyzed for two different wavelengths of may be further reconfigured when Kerr nonlinear materials operation. Figure 7(b) demonstrates the operation of this areincludedintheircore,leadingtostrongopticalbistability, device at 𝜆 = 323 nm, when negative refraction is not all-optical switching, and memory operation. Furthermore, realized inside the hyperlens. It can be seen that the Goos- layered hyperbolic metamaterials have been studied con- Hanchen¨ shift is positive and rather small at this frequency sidering third-order nonlinear materials introduced in their point, as it is expected for the usual operation of conventional layers. Tunable operation with positive-to-negative refraction dielectrics, which refract light in a positive way. However, has been demonstrated as the input radiation intensity is when the device is illuminated at 𝜆 = 352 nm, strong negative increased. Finally, reconfigurable Goos-Hanchen¨ shift has Goos-Hanchen¨ shift is obtained, which is clearly shown in been presented in a linear layered hyperbolic metamaterial. Figure 7(c). This effect causes a distinctively strong focal spot Theshiftmaychangefrompositivetonegativewhenthesame of radiation in front of the hyperbolic metamaterial. The metamaterial structure is illuminated at different frequencies. negative shift is a direct outcome of the negative refraction The proposed sharp Fano scattering signatures and the the waves experience inside the hyperlens, which is dominant negative refraction properties of the proposed hyperbolic and strong for this particular frequency point. Moreover, metamaterial are expected to be affected when realistic losses this effect can arise without the usual requirements of areincludedinthemetallicparts[36–38]. Nevertheless, gain negative refractive index metamaterials. It is interesting that and active media may be in principle introduced in the hyperbolic metamaterials may also lead to “trapped rainbow” dielectric layers [38], expectedly compensating the inherent configurations with lower losses compared to devices based metallic losses. The incorporation of gain and active materials 10 International Journal of Antennas and Propagation

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Research Article Voltage Controlled Intertwined Spiral Arrays for Reconfigurable Metasurfaces

A. Vallecchi,1 R. J. Langley,1 and A. G. Schuchinsky2

1 Department of Electronics and Electrical Engineering, University of Sheffield, Sheffield S1 3JD, UK 2 Institute of Electronics, Communications and Information Technology, Queen’s University of Belfast, Belfast BT3 9DT, UK

Correspondence should be addressed to A. Vallecchi; [email protected]

Received 2 December 2013; Accepted 9 February 2014; Published 20 March 2014

Academic Editor: Douglas H. Werner

Copyright © 2014 A. Vallecchi et al. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

Reconfigurable bistate metasurfaces composed of interwoven spiral arrays with embedded pin diodes are proposed for single and dual polarisation operation. The switching capability is enabled by pin diodes that change the array response between transmission and reflection modes at the specified frequencies. The spiral conductors forming the metasurface also supply the dc bias for controlling pin diodes, thus avoiding the need of additional bias circuitry that can cause parasitic interference and affect the metasurface response. The simulation results show that proposed active metasurfaces exhibit good isolation between transmission and reflection states, while retaining excellent angular and polarisation stability with the large fractional bandwidth (FBW) inherent to the original passive arrays.

1. Introduction that a large number of unit cells are usually required for the array to effectively interact with an incident field7 [ ]. Planar metamaterials formed by a single layer of electrically Moreover, the small electrical size of the unit cells plays the small scatterers arranged into two-dimensional periodic pivotalroleintheangularandpolarisationstabilityofthe arrays have recently attracted increasing interest owing to the array response to the incident fields. Both of these aspects are ease of fabrication and the intricate properties appealing for essential for FSS applications to wireless communication sys- potential applications. Metasurfaces, as planar metamaterials tems operated in the controlled electromagnetic architecture are usually called, allow, for example, for effective control of of buildings with enhanced spectral efficiency and security. polarization, frequency selectivity, and multiband operation Infact,ontheonehand,thewavelengthsinthefrequency while being deployed on conformal surfaces; see for exam- bands used in indoor communications can be comparable ple, recent review [1]. They can also be used in ultrathin with the size of building interiors and ordinary office rooms. absorbers of substantially sub-wavelength thickness [2–4], to On the other hand, the propagation characteristics of the implement slow-light effects in compact configurations5 [ ], built environment can be complex due to diffraction, reflec- and designed to create mantle cloaking devices capable of tion, scattering, and multipath propagation, as demonstrated drastically suppressing the scattering from both planar and in [8] by the finite-difference time-domain simulations of 3D objects [6]. The distinctive feature of metasurfaces is the electromagnetic wave propagation in buildings. Thus it can sub-wavelength size of their unit cells, which provides the be argued that only FSSs with a very stable frequency advantages of angular invariance of the array response and response over a wide range of incidence angles are suitable suppression of the higher order diffraction effects. for walls controlling interference and shielding in the built Miniaturization of periodic structures is an important environment. prerequisite for their integration in small mobile terminals, Among the many different types of layouts that have been RF front-ends, conformal frequency selective surfaces (FSSs), proposed and used in the design of FSSs, periodic arrays andotherapplicationswhereoneofthemainchallengesis of convoluted and interwoven elements have demonstrated 2 International Journal of Antennas and Propagation many unique and attractive properties [9–14]. The main by switching on and off a control signal, usually a dc bias. distinctive feature of such arrays is that their fundamental The major challenge in the implementation of such AFSSs resonance occurs at substantially sub-wavelength size of the is related to delivery of the dc control signal throughout the unit cell, thus qualifying them as metasurfaces. In addition, AFSS without distorting its RF performance. the interleaved conductors extended into adjacent unit cells In a bistate dipole FSS, the ends of adjacent dipoles are broaden fractional bandwidth (FBW) of the fundamental res- connected by pin diodes to form columns that are linked onance. For example, the free-standing intertwined quadri- together with biasing lines. The surface behaves as a conven- filar spiral arrays were shown to achieve the −10 dB FBW of tional reflective array when the diodes are off,whereaswhen ∼55% at the unit cell size ∼1/40th of the resonance wavelength thediodesareintheon state, the array columns behave [13]. The latter property of the interwoven patterns is in as continuous conducting strips presenting inductive impe- stark contrast to the conventional periodic structures with the dance and acting as a high-pass filter. If the high-pass band constituent elements confined to a single unit cell of the size is low enough in frequency, the surface can be substantially commensurate with half a resonance wavelength, where FBW transparent to the incident waves [19, 20]. rapidly narrows as the resonance frequency lowers [15, 16]. Adifferentapproach,similartothatappliedtoannular The use of FSS and metasurfaces for efficient control of ring FSS in [24], is explored in this work for the implemen- the electromagnetic environment of buildings faces addi- tation of bistate voltage-controlled interwoven spiral arrays. tional challenges. Namely, passive FSSs, although suitable for In particular, we initially focus on FSSs formed by interwo- many fixed frequency applications, become restrictive in the ven bifilar spiral elements. These elements are conceptually buildings with diverse spectral requirements and propagation similar to the original interwoven quadrifilar spiral layout conditions changeable due to the physical movement of presented in [12, 13]. But since only two spiral conductor arms dividing walls or partitions. This implies that a FSS needs extend from the adjacent unit cells into a reference unit cell the capability of adjusting its response in accordance with and are counter-wound in the gaps between the turns of the the variable architectural requirements. Active FSSs (AFSSs) primary reference spiral, they form either column- or row- with embedded pin diodes serve to this purpose by switching wise patterns depending on whether the number of spiral between the transmission and reflection modes [17]. folds is even or odd. As a result, these surfaces are nearly In this paper, we demonstrate that the interwoven spi- transparent to one of the incident field polarisations, whereas ral arrays with periodic and tessellated conductor patterns they exhibit a strong resonant response to the orthogonal proposed in [12, 13, 18] are particularly apt for realization polarisation. Similarly to the interwoven quadrifilar spirals, of bi-state switchable AFSSs with integrated voltage control the fundamental resonance of interwoven bifilar spiral FSS circuitry. Initially, the concept of the switch-mode AFSS is has broad FBW and occurs at wavelengths large as compared elucidated using an example of entwined bifilar spirals with to the unit cell size. embedded pin diodes. In this arrangement, the AFSS exhibits To enable the desired reconfigurability of this type of a voltage-controlled band-stop response to one polarisation AFSS, gaps are introduced in the arms of the bifilar spirals of the incident field while remaining always transparent to first with the aim of suppressing the fundamental resonance the orthogonal polarisation. This approach is then extended by such discontinuities. The strip breaks can be positioned, to the AFSSs responsive to both TE and TM polarisations of forexample,attheperipheryofeachunitcell,asshownin the incident waves. Figure 1, but alternative locations and a different number of The rest of the paper is organised as follows. The concept breaks can be also used. Then pin diodes are inserted across of the proposed AFSS is described and elucidated in Section 2 the strip breaks. When the diodes are in the off state, the AFSS using an example of intertwined bifilar spiral AFSS. The is transparent because it is no longer resonant at its original effects of the pin diode parameters in the on and off states and design frequency. On the contrary, switching the diodes in theunitcelllayoutontheAFSSperformancearediscussed the on state reconnects the spiral arm segments and the AFSS here for single polarisation of the incident field. The dual provides its original stop-band response. In the transparent polarised bistate AFSSs composed of intertwined quadrifilar state, the AFSS still exhibits resonance, which however can spirals are presented in Section 3 where their performance is be shifted at other frequency depending on the number of discussed in detail for the normal and oblique incidence of breaks and switches introduced in the spiral arms, in order to TEandTMwaves.Themainfindingsandobtainedresults achieve the required level of transparency in the operational aresummarisedinSection4. band. In this arrangement, spiral arms themselves act as current carrying conductors supplying the dc bias for switching the 2. Anisotropic Bistate Interwoven pin diodes. This way there is no need in external bias lines that Spiral Metasurface can cause multiple resonances and interference which may drastically affect the FSS response at oblique incidence. The 2.1. Topology and Operational Principle of Bifilar Spiral AFSS. continuity of dc path is realized by means of suitably chosen Semiconductor switches, such as pin diodes, are commonly inductors which are connected between the open ends of used to realize bistate switching of AFSSs comprised of interwoven spiral arms while being isolated from the pair of dipoles and other element types; see for example, [17, 19– spiralarmsconnectedatthecentreofthereferenceunitcell. 24]. The term bistate means that the FSS can be made Inductors present high impedance at the design frequency either reflective or transparent at the frequency of interest andworkasRFchokes,toensuretheisolationbetween International Journal of Antennas and Propagation 3

+V on the transmission curve is ∼63%. For an x-polarised (TM) incident wave instead the FSS is practically transparent. The response of the FSS in the off state, when the L pin diodes are considered as open circuits, is shown in L L Figure 2(b). It can be observed that for the TE wave with the electric field polarization along the 𝑦-axis, which is affected bythepresenceoftheFSS,theresonanceoccursatapprox- imately 1.6 GHz and has narrowband (FBW ∼4%). As a result the transmission loss in the off state is less than 0.5 dB over the stop-band of the FSS in the on state. The presence of the breaks has practically negligible effect on an incident field polarized along the 𝑥-axis. The transparency of the FSS in the LL off state can be further improved by increasing the number of y breaks and pin diodes in the spiral arms, which can shift the unwanted FSS resonance upward in frequency. This is shown in Figure 3 forthemodifiedunitcellinthefigureinsetwitha x total of four breaks introduced in the spiral arms. It must be notedthatwhilealargernumberofbreaksandpin diodes can −V be beneficial for FSS transparency in the off state, they will Figure 1: Layout of 2×2array portion of sample free-standing also cause higher dissipative losses and increase fabrication bistate interwoven 9-fold bifilar spiral elements. The reference basic costs. spirals in adjacent unit cells are alternatively marked dark and light grey to highlight the intertwining scheme. Two pin diodes are connected across breaks in the outer folds of the spiral conductors 2.2. Effect of Nonideal Switches on the Performance of Bifi- and the choke inductance is placed at the centre of each unit lar Spiral AFSS. Whilefeasibilityofthebistateswitching cell between the ends of disconnected spiral arms. Geometrical response of the interwoven bifilar spiral elements has been 𝑝 = 10.2 parameters of the unit cell: lattice period mm, spiral pitch proven, assuming an ideal behaviour of the biasing and 𝑠 = 0.2 𝜇 2.4 mm, and strip width mm and strip thickness 17.5 m. switching components, it is important to estimate the impact of imperfect components on the AFSS performance. The main effects of switch parasitics are the dissipative loss the resonant surface and bias voltages that will however and the resonance frequency shift caused by the off state impact upon FSS operation. As observed in [24], it could capacitance of switches. To model these effects, we use the be advantageous for improved isolation to exploit the larger simplified equivalent circuits of the forward and reverse impedance presented by inductors as the minimum self- biased diodes shown in Figure 4.Intheforwardbiascase(on resonant frequency due to the presence of stray reactances state), the diode presents a resistance 𝑅𝑠 in series with the canbeapproached. packaging inductance 𝐿𝑠.Atthereversebias(off state), the circuit becomes a parallel combination of 𝑅𝑝 with 𝐶𝑡 in series The free-standing FSS with the square unit cell shown 𝐿 𝐶 in Figure 1 has been investigated as a proof of concept. with 𝑠,where 𝑡 is actually a combination of the device The periodicity of the array is 𝑝 = 10.2 mm and the 9- junction capacitance and its package parasitic. fold bifilar spiral is formed by conductor strips with 0.2 mm The main criteria for specification of pin diodes are the size,thecost,andtheleastforwardresistanceandoff state width separated by gaps of 0.4 mm. Initial assessment of capacitance. In principle, different technologies are available the operation of this bistate interwoven spiral AFSS has for the realization of pin diodes with low on state resistance been performed by CST MWS assuming ideal electrical and low off state capacitance. These characteristics however components. Specifically, the pin diodes across the breaks in are among the main factors driving the pin diodes cost, the spiral conductor arms are represented as short or open and pin diodes with larger off capacitance are generally less circuits, depending whether they are in the on or off state, expensive. Since in the design of an AFSS a balance has to respectively, and the inductances as open circuits, without be struck between the cost of individual devices and the additional stray reactances or losses associated with these numbers required, for our simulations, we have assumed lump components. The spiral array has been modeled using to use reasonably low cost pin diodes, for example, the a single unit cell with doubly periodic boundary conditions. device HMSP3862 from Avago [25], which have been recently The transmittance and reflectance at normal incidence are employed also in [24]. The values used for forward bias are plottedinFigure2(a) for the two orthogonal polarizations of 𝑅𝑠 =3Ωand 𝐿𝑠 =0.3nH, while for reverse bias 𝐶𝑡 = the incident field when the pin diodes are in the on state; that 0.15 pF is added to the 𝑅𝑠-𝐿𝑠 series circuit. The 𝑅𝑝 resistance is, the breaks in the spiral arms are short circuited, as shown in the off state, usually larger than 10 kΩ,canbeneglected in the inset. For a y-polarised (TE) wave, the FSS exhibits in the simulations. The inductors that provide a dc path pronounced stop-band resonance at about 0.8 GHz, which for the pin diode control signal must be suitably chosen for corresponds to the packaging density 𝜆𝑟/𝑝 ∼ 37 (𝜆𝑟 is the RF choke to provide the sufficiently high impedance at the resonance wavelength); FBW measured at the −10 dB points design frequency. In the simulations, we have used 100 𝜇H 4 International Journal of Antennas and Propagation

0 0

−10 −10

−20 −20

−30 −30

Magnitude (dB) Magnitude

Magnitude (dB) Magnitude −40 −40

−50 −50 0 0.5 1 1.5 2 0 0.5 1 1.5 2 Frequency (GHz) Frequency (GHz)

Ryy Tyy Ryy Tyy Rxx Txx Rxx Txx

(a) (b)

Figure 2: Transmittance and reflectance of the AFSS with the unit cell of Figure 1 at normal incidence of x-andy-polarized waves when ideal pin diode switches are in the (a) on and (b) off states.

0 0

−10 −10

−20 −20 −30 −30

Magnitude (dB) Magnitude

Magnitude (dB) Magnitude −40 −40 −50 0 0.5 1 1.5 2 −50 0 0.5 1 1.5 2 Frequency (GHz) Frequency (GHz) Ryy -on Ryy -off Tyy -on Tyy -off Ryy Tyy Rxx Txx Figure 5: Transmittance and reflectance of the AFSS with the unit Figure 3: Off state transmittance and reflectance of the AFSS of cell of Figure 1 at normal incidence of a y-polarized wave when pin Figure 1 at normal incidence of x-andy-polarized waves when two diode switches are in the on (solid lines) and off (dashed lines) states. additional breaks and ideal switches are contained in the spiral arms.

inductors available, for example, within the SIMID 1210-100 inductor series from EPCOS [25]. RF choke inductors used in the simulation present a high impedance of approximately Ls 50 kΩ at the AFSS resonance frequency of 0.8 GHz. The RF choke inductance has been selected by trial and error to minimise distortion of the ideal FSS response (current Rs magnitude across inductors is about 40 dBA smaller than Ls maximum current flowing across conductor strips). Smaller inductance values could be used in practical implementations with just slight degradation of the FSS performance. Ct Rp The equivalent circuits of the pin diodes and RF choke Rs inductors shown in Figure 4 were incorporated in the CST MWS model of the AFSS shown in Figure 1.Figure5 displays the simulated transmittance and reflectance of the AFSS illu- (a) (b) minated by a normally incident plane wave at both on and off Figure 4: Pin diode equivalent circuits in the (a) forward and (b) states of the pin diodes. Only the response to the incident field reversed bias states. polarized along the 𝑦-axis is presented because the realistic International Journal of Antennas and Propagation 5

0 the off state attenuation. For example, GaAs pin diodes typically have an off capacitance of only 0.036 pF along with −10 a2.5Ω forward resistance [26]. Alternatively, RF MEMS switches can provide the key capability of low control power −20 and low insertion loss required in active and reconfigurable FSSs [27, 28] and periodic arrays [29, 30]. The representative −30 values of MEMS off capacitance are in the range 0.05–0.08 pF. Forexample,theAFSSwiththelayoutofFigure1,usingtwo Magnitude (dB) Magnitude 𝐶 = 0.056 −40 switches per unit cell with an off capacitance 𝑡 pF (either low off capacitance pin diodes or RF MEMS) exhibited −50 the performance similar to that of the AFSS with four low cost 0 0.5 1 1.5 2 pin diodes with 𝐶𝑡 = 0.15 pF. Frequency (GHz) The transmission and reflection characteristics of the bistate interwoven bifilar spiral AFSS with four pin diodes R on R off yy - yy - have been simulated in CST MWS Studio at oblique incidence T on Tyy -off yy - of TE and TM waves. Similar to the normal incidence, Figure 6: Transmittance and reflectance of the AFSS with the unit the transmittance and reflectance of TM waves at oblique cell of Figure 1 at normal incidence of a y-polarized wave when the incidence remain practically indistinguishable in the on and four pin diode switches in the spiral arms are in the on (solid lines) off states of the switches—the FSS is practically transparent and off (dashed lines) states. The on and off states of pin diodes are and the corresponding plots are not shown. In the case of TE modelled with their equivalent circuits of Figure 4. waves, the AFSS in the on state exhibits the high stability of the resonance frequency in a broad range of incidence angles, as illustrated in Figure 7.Intheoff state, the response is also model of pin diodes does not affect the AFSS response at the reasonably stable, with the transmission slowly decreasing incident field polarized along the 𝑥-axis. The transmittance with angles of incidence, yet the attenuation remaining below ∘ andreflectanceintheon state of pin diodes, plotted as solid 1dB at45 obliqueincidence.Thetransparencyintheoff red and blue lines, respectively, look very similar to the case state slightly decreases at larger incidence angles for the TE of ideal switches and absence of the RF choke inductors. polarized waves due to broadening the rejection resonance at Namely, the resonance frequency remains unchanged with about 2 GHz. Such resonance broadening is typical for the TE thebroadstop-bandaroundit,andresonancecurveisjust waves incident at slant angles on the interwoven spiral arrays, slightly shallower than in Figure 2 due to dissipative losses, cf. [13, 31]. which also marginally increased FBW (64%). However, when pin diodes are reverse biased, it is apparent that the off state capacitance of the pin diodes causes shift of the resonance 3. Dual Polarised Bistate Interwoven for approximately 0.2 GHz to lower frequency than that in Spiral Metasurface Figure 2(b) of ideal switches. At the same time, the FBW of thesecondarystop-bandresonanceincreasesfromabout4% 3.1. Concept and Topology of Dual Polarised AFSS. The to 12%. Both of these factors contribute to reducing the AFSS concept of interweaving spiral arrays can be further extended transparency in the off stateascomparedtothatforideal to realising switchable isotropic AFSS sensitive to both switches, with the transmission attenuation in the off state polarisations of the incident field. Intertwined quadrifilar increasing to about 2.6 dB at the higher frequency edge of the spiralscanbeusedforthispurposeasillustratedbythelayout on state reflection band. of a 2×2cell array depicted in Figure 8.Similarlytothe To improve the AFSS transparency in the off state, intertwined bifilar spiral AFSS, a number of strip breaks are one approach is to increase the number of breaks in the cutinthespiralarmsattheperipheryoftheunitcells,and spiral arms that shift the undesired FSS resonance to higher switches are inserted across these breaks. The switch state frequencies. This can be achieved by introducing cuts in both is controlled by a suitable bias that is supplied by the same the reference spiral and the intertwined spiral arms extended conductors forming the spiral arms. The control signal is from the surrounding cells, as shown in insert of Figure 3. appliedinagridformat,alongbothrowsandcolumnsof Figure 6 illustrates the transmittance and reflectance of the the array. It is distributed over the whole surface through the AFSS with the unit cell comprising four pin diodes connected continuous dc current path provided by the pairs of lump across a corresponding number of breaks in the spiral arms, inductors at the unit cell centre which act as RF chokes at as depicted in the inset. It can be observed that against a small the design frequency. The inductors are connected between increase of dissipative losses in the on state, in the off state, the diagonally opposite open ends of the extended spiral arms this surface exhibits good transparency with a transmission and are isolated from each other and the crossing arms of the loss smaller than 0.5 dB over the whole stop-band of the reference spiral. surface in the on state. TheAFSSshowninFigure8 is isotropic if all the switches Instead of increasing their numbers, pin diodes with are in either on or off states. When the bias is symmetrically smaller off capacitance can effectively shift upward the applied along both rows and columns of the array to keep frequency of the parasitic resonance and consequently reduce all the switches in the on state, the AFSS is reflective in the 6 International Journal of Antennas and Propagation

0 0

−10 −10

−20 −20 (dB) (dB) TE −30 TE −30 T T

−40 −40

−50 −50 0 0.5 1 1.5 2 0 0.5 1 1.5 2 Frequency (GHz) Frequency (GHz)

∘ ∘ 𝜃=15-on/off 𝜃=15-on/off ∘ ∘ 𝜃=30-on/off 𝜃=30-on/off ∘ ∘ 𝜃=45-on/off 𝜃=45-on/off (a) (b) Figure 7: (a) Transmittance and (b) reflectance of the AFSS with the unit cell of Figure 1, modified with the introduction of two additional pin diode switches, at oblique incidence of TE waves when the pin diodes switches are in the on (solid lines) and off (dashed lines) states. The on and off states of pin diodes are modelled with their equivalent circuits of Figure 4. specified frequency band and becomes transparent at these 3.2. Effect of Nonideal Switches on the Performance of Quadri- frequencies when switches are set in the off state. In the filar Spiral AFSS. The effect of switch parasitics and finite latter mode, the AFSS still exhibits resonance at a higher impedance of lumped inductances on the predicted bistate frequency, which can be placed far enough from the operating response of the interwoven quadrifilar spiral AFSS needs to band by altering the number of switches inserted in the spiral be assessed. For this purpose, the equivalent circuits of the conductor arms. Dual polarized operation and independent pin diodes shown in Figure 4 and the RF choke inductors control over orthogonally polarized incident waves can be areincorporatedintheanalysisofthearraystructurein realised by applying different biases at rows or columns of the Figure 8 in both the forward and reverse bias states. The diode array, depending on the targeted polarization sensitivity. equivalent circuit parameters are the same as in Section 2;that The bistate characteristics of the AFSS depicted in Fig- is, for forward bias 𝑅𝑠 =3Ohm, 𝐿𝑠 =0.3nH, and for reverse ure 8 have been simulated in CST MWS assuming first bias, a series capacitance 𝐶𝑡 =0.15pF.Andapairof100𝜇H the ideal lossless lumped components, that is, pin diodes choke inductors is also included in the unit cell simulation modelledasshortoropencircuitsintheon or off state, model [25]. respectively, and inductances as open circuits. The simulated ThesimulatedtransmittanceandreflectanceoftheAFSS transmittance and reflectance at normal incidence shown with nonideal lumped components illuminated at normal in Figure 9 demonstrate that when the pin diodes are in incidence are shown in Figure 11 for both on and off states the on state, the AFSS exhibits a wide rejection resonance of the pin diodes. In the on state, the dissipative losses of at ∼0.7 GHz, which corresponds to the packaging density switches have negligible effect on the AFSS response which 𝜆𝑟/𝑝 ∼ 42,FBW∼53% at the transmission level −10 dB. hasonlyslightlyshorterstop-banddipandFBWincreasedto Conversely, when pin diodes are switched off and act as ideal 56%. The effect of the off state capacitance of pin diodes at the open circuits, the resonance shifts upward above 2 GHz, and reverse bias is more pronounced and causes downward shift the AFSS becomes practically transparent—the transmission of the secondary resonance for ∼0.7 GHz and broadening its loss is less than 0.24 dB at the high frequency edge of the bandwidth as compared with the ideal case in Figure 9.Asa AFSS stop-band in the on state. The AFSS responses shown result, the AFSS transparency in the off state is reduced and in Figure 9 are invariant to the incident field polarization transmission loss increases to ∼1.3 dB at the upper frequency provided that symmetric bias is applied to both rows and edge of the reflection band in the on state. columns of the array. The capability of this AFSS for dual Similarly to the case of the intertwined bifilar spirals, polarised operation is illustrated in Figure 10. It is assumed the transparency of the isotropic AFSS in the off state can that a nonsymmetric bias voltage scheme is applied to the be improved by using switches with lower off capacitance AFSS to switch on only a pair of pin diodes, whereas the or/and increasing the number of breaks in the spiral arms of otherpairremainintheoff state. As a result, the AFSS both the reference spiral and the spirals extended from the provides an anisotropic response to the incident field; namely, surrounding cells. This effect is illustrated in Figure 12 which for a y-polarised incident wave, the AFSS exhibits a wide shows the transmittance and reflectance at normal incidence rejection resonance at ∼0.7 GHz also seen in Figure 9, while it on the modified AFSS with four additional pin diode switches is practically transparent to the orthogonal polarization along connected across a corresponding number of breaks in the the 𝑥-axis. spiral arms, as depicted in the inset of Figure 12,atbothon and International Journal of Antennas and Propagation 7

L 0

−10 L +V −20 +V

−30 L L L (dB) Magnitude −40 L L L −50 0 0.5 1 1.5 2 Frequency (GHz)

R-on R-off T on T off L L - -

L L y Figure 9: Transmittance and reflectance at normal incidence of the AFSS with the unit cell of Figure 8 when ideal pin diode switches are x in the on and off states. −V −V 2×2 Figure 8: Layout of array portion of sample free-standing 0 bistate interwoven 7-fold quadrifilar spiral elements. The reference basic spirals in adjacent unit cells are alternatively marked dark and −10 light grey to highlight the intertwining scheme. Four pin diodes are connected across breaks in the outer folds of the spiral conductors and the RF choke inductances are placed at the centre of each unit −20 cell between the open ends of the spiral arms extended from adjacent unit cells. Geometrical parameters of the unit cell: lattice period −30 𝑝 = 10.2 mm, spiral pitch 1.6 mm, and strip width 𝑠 = 0.2 mm and L 𝜇 (dB) Magnitude thickness 17.5 m. −40 L

−50 off states of the pin diodes. It is noteworthy that transmission 0 0.5 1 1.5 2 loss of this surface in the off state is less than 0.5 dB over Frequency (GHz) the entire operating band, though improved transparency is R obtained at the cost of a small increase of dissipative losses in yy Tyy R T the on state. xx xx For comparison, Figure 13 shows the improved trans- Figure 10: Transmittance and reflectance at normal incidence of the parency of the AFSS in the off state that can be obtained AFSS with the unit cell of Figure 8 illuminated by two orthogonal with only four switches with the smaller off capacitance 𝐶𝑡 = 0.056 polarizations when a pair of ideal pin diode switches (depicted in pF. This performance is very similar to that achievable red) are in the on state and the other pair (depicted in blue) is in the by using eight low cost pin diodes switches with the larger off state. off capacitance 𝐶𝑡 = 0.15 pF. It is necessary to note that the performance of these AFSSs in the off state can also be adjusted by altering the positions of the spiral arm breaks. surfaceintheoff state slightly decreases at larger angles of Identification of the optimal locations of switches is therefore incidence of TE polarised waves, whereas it even improves a part of the design process. for obliquely incident TM polarised waves. Similarly to the The response of the bistate dual polarised AFSSs at oblique incidence of TE and TM waves has been simulated response of bifilar spiral AFSSs at oblique incidence of TE for the unit cell configuration with eight pin diodes shown waves, the transparency of quadrifilar spiral AFSSs in the off in insert of Figure 12. The transmittance and reflectance state is affected by broadening (TE waves) or narrowing (TM characteristics are displayed in Figure 14 at variable incidence waves) of the bandwidth of the rejection resonance at about angles of TE and TM waves for both the on and off states of 2 GHz. Such resonance broadening for TE waves and narrow- the switchable diodes. These plots demonstrate that, in the ing for TM waves incident at slant angles on the interwoven on state, the AFSS exhibits high stability of the resonance spiral arrays are the inherent property of this type of FSS, cf. frequency in a broad range of incidence angles at both TE [13, 31]. At any rate, for both polarizations, transmission losses ∘ and TM wave polarisations. The resonance response remains are found to be less than 1 dB at 45 oblique incidence at the fairly stable also in the off state. The transparency of the upper frequency edge of the reflection band in the on state. 8 International Journal of Antennas and Propagation

0 0

−10 −10

−20 −20

−30 −30 L

L (dB) Magnitude

Magnitude (dB) Magnitude −40 L −40 L

−50 −50 0 0.5 1 1.5 2 0 0.5 1 1.5 2 Frequency (GHz) Frequency (GHz) R on R off R-on R-off - - T on T off T-on T-off - -

Figure 11: Transmittance and reflectance at normal incidence of the Figure 13: Transmittance and reflectance of the AFSS with the unit AFSS with the unit cell of Figure 8 when pin diode switches are in cell of Figure 8 at normal incidence when switches, with low off 𝐶 = 0.056 the on (solid lines) and off (dashed lines) states. capacitance 𝑡 pF, are in the on (solid lines) and off (dashed lines) states.

−10 small parasitic reactances have minor effect on the array RF response in both on and off states as demonstrated by the −20 presented simulation results. The detailed analysis has shown that parasitic reactances −30 of the switches strongly influence the overall response of the L AFSS, thus making it progressively more difficult to achieve

Magnitude (dB) Magnitude −40 L the desired off state transparency at the upper edge of the operational band. Nonetheless, the AFSS transparency in −50 the off state can be improved by increasing the number of 0 0.5 1 1.5 2 switches in the spiral arms, but this inflicts small increase Frequency (GHz) of dissipative losses in the on state. Alternatively, pin diodes with smaller off capacitance can be used for this purpose for R-on R-off T-on T-off expense of higher cost. It has been demonstrated that the presented AFSS con- Figure 12: Transmittance and reflectance at normal incidence of the figurations can achieve good isolation between transmission AFSS with the unit cell of Figure 8, with four additional pin diode and reflection states over a broad FBW, while retaining the switches introducing the spiral arms, in the on (solid lines) and off substantial subwavelength response of the original passive (dashed lines) states. metasurfaces. The latter property enables high polarization and angular stability of AFSS. Also owing to the extremely small subwavelength size of their unit cell, the interwoven 4. Conclusion multifilar spirals are particularly suitable for the design of The bistate voltage-controlled metasurfaces composed of reconfigurable homogeneous metasurfaces. the interwoven spiral arrays with integrated pin diodes are proposed for both single and dual polarisation operation. Conflict of Interests The primary feature of the presented architecture is the dual use of the spiral conductors as the constituent elements The authors declare that there is no conflict of interests of the AFSS and as the path for supplying dc bias to pin regarding the publication of this paper. diodes. It has been demonstrated that switching pin diodes between their on and off states enables toggling the AFSS response between the transmission and reflection modes at Acknowledgment the specified frequency. Integration of the biasing circuitry with the metasurface conductor pattern has enabled us to This work was supported by the European Commission eliminate the need of separate wires supplying bias voltage 7th Framework Program Marie Curie IAPP Project Grant to each switchable diode. The proposed metasurface archi- no. 286333, “Wireless Friendly Energy Efficient Buildings tecture has dramatically simplified the AFSS topology where (WiFEEB). ” International Journal of Antennas and Propagation 9

0 0

−10 −10

−20 −20 (dB) (dB) TE TE

−30 T −30 R L L −40 L −40 L

−50 −50 0 0.5 1 1.5 2 0 0.5 1 1.5 2 Frequency (GHz) Frequency (GHz) (a) (b) 0 0

−10 −10

−20 −20 (dB) (dB) TM −30 TM −30 R L R L −40 L −40 L

−50 −50 0 0.5 1 1.5 2 0 0.5 1 1.5 2 Frequency (GHz) Frequency (GHz)

∘ ∘ 𝜃=15-on/off 𝜃=15-on/off ∘ ∘ 𝜃=30-on/off 𝜃=30-on/off ∘ ∘ 𝜃=45-on/off 𝜃=45-on/off (c) (d)

Figure 14: Transmittance and reflectance of the AFSS with the unit cell layout of Figure 8, modified with the introduction of four additional switches, at oblique incidence of TE (a)-(b) and TM (c)-(d) waves when the pin diode switches are in the on (solid lines) and off (dashed lines) states.

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[12] A. Vallecchi and A. G. Schuchinsky, “Entwined spirals for ultra [28] G. M. Coutts, R. R. Mansour, and S. K. Chaudhuri, “Microelec- compact wideband frequency selective surfaces,” in Proceedings tromechanical systems tunable frequency-selective surfaces and of the 4th European Conference on Antennas and Propagation electromagnetic-bandgap structures on rigid-flex substrates,” (EuCAP ’10), pp. 12–16, Barcelona, Spain, April 2010. IEEE Transactions on Microwave Theory and Techniques,vol.56, [13] A. Vallecchi and A. G. Schuchinsky, “Entwined planar spirals no.7,pp.1737–1746,2008. for artificial surfaces,” IEEE Antennas and Wireless Propagation [29] Y. Raedi, S. Nikmehr, and A. Poorziad, “A novel band- Letters, vol. 9, pp. 994–997, 2010. width enhancement technique for X-band RF mems actu- [14] Z. Bayraktar, J. P. Turpin, and D. H. Werner, “Nature-inspired ated reconfig-urable reflectarray,” Progress in Electromagnetics optimization of high-impedance metasurfaces with ultrasmall Research, vol. 111, pp. 179–196, 2011. interwoven unit cells,” IEEE Antennas and Wireless Propagation [30] F. A. Tahir, H. Aubert, and E. Girard, “Equivalent electrical cir- Letters, vol. 10, pp. 1563–1566, 2011. cuit for designing mems-controlled reflectarray phase shifters,” Progress in Electromagnetics Research,vol.100,pp.1–12,2010. [15] E. A. Parker and A. N. A. El Sheikh, “Convoluted array elements and reduced size unit cells for frequency-selective surfaces,” IEE [31] A. Vallecchi and A. G. Schuchinsky, “Artificial surfaces of inter- Proceedings H,vol.138,no.1,pp.19–22,1991. twined square spirals: a CPW model,”in Proceedings of the IEEE International Microwave Symposium Digest,pp.1–3,Montreal, [16] E.A.Parker,A.N.A.ElSheikh,andA.C.C.Lima,“Convoluted Canada, June 2012. frequency-selective array elements derived from linear and crossed dipoles,” IEE Proceedings H,vol.140,no.5,pp.378–380, 1993. [17] T. K. Chang, R. J. Langley, and E. A. Parker, “Active frequency selective surfaces,” IEE Proceedings H,vol.143,no.1,pp.62–66, 1996. [18] A. Vallecchi and A. G. Schuchinsky, “Artificial surfaces formed by tessellations of intertwined spirals,” in Proceedings of the 5th European Conference on Antennas and Propagation (EUCAP ’11),pp.1846–1848,Rome,Italy,April2011. [19] P. Edenhofer and A. Alpaslan, “Active frequency selective surfaces for antenna applications electronically to control phase distribution and reflective/transmissive amplification,” in Pro- ceedings of the IEEE/ACES International Conference on Wireless Communications and Applied Computational Electromagnetics, pp. 237–240, April 2005. [20] B. M. Cahill and E. A. Parker, “Field switching in an enclosure with active FSS screen,” Electronics Letters,vol.37,no.4,pp.244– 245, 2001. [21] A. Tennant and B. Chambers, “A single-layer tuneable microwave absorber using an active FSS,” IEEE Microwave and Wireless Components Letters,vol.14,no.1,pp.46–47,2004. [22] G.I.Kiani,K.L.Ford,K.P.Esselle,A.R.Weily,C.Panagamuwa, and J. C. Batchelor, “Single-layer bandpass actlve frequency selective surface,” Microwave and Optical Technology Letters, vol. 50, no. 8, pp. 2149–2151, 2008. [23]G.I.Kiani,K.L.Ford,L.G.Olsson,K.P.Esselle,andC.J.Pan- agamuwa, “Switchable frequency selective surface for reconfigurable electromagnetic architecture of buildings,” IEEE Transactions on Antennas and Propagation,vol.58,no.2,pp. 581–584, 2010. [24]P.S.Taylor,E.A.Parker,andJ.C.Batchelor,“Anactiveannu- lar ring frequency selective surface,” IEEE Transactions on An- tennas and Propagation,vol.59,no.9,pp.3265–3271,2011. [25] http://www.avagotech.com/pages/en/rf microwave/diodes/pin/ hsmp-3862. [26] K. W. Kobayashi, A. K. Oki, D. K. Umemoto, S. Claxton, and D. C. Streit, “GaAs HBT PIN diode attenuators and switches,” in ProceedingsoftheIEEEMicrowaveMillimeter-WaveMonolithic Circuits Symposium Digest, pp. 349–352, Atlanta, Ga, USA, June 1993. [27] B. Schoenlinner, A. Abbaspour-Tamijani, L. C. Kempel, and G. M. Rebeiz, “Switchable low-loss RF MEMS Ka-band frequency- selective surface,” IEEE Transactions on Microwave Theory and Techniques,vol.52,no.11,pp.2474–2481,2004. Hindawi Publishing Corporation International Journal of Antennas and Propagation Volume 2014, Article ID 346838, 8 pages http://dx.doi.org/10.1155/2014/346838

Research Article Mechanically Reconfigurable Microstrip Lines Loaded with Stepped Impedance Resonators and Potential Applications

J. Naqui and F. Martín

GEMMA/CIMITEC, Departament d’Enginyeria Electronica,` Universitat AutonomadeBarcelona,08193Bellaterra,Spain`

Correspondence should be addressed to F. Mart´ın; [email protected]

Received 15 January 2014; Accepted 16 January 2014; Published 20 February 2014

Academic Editor: Giacomo Oliveri

Copyright © 2014 J. Naqui and F. Mart´ın. This is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited.

This paper is focused on exploring the possibilities and potential applications of microstrip transmission lines loaded with stepped impedance resonators (SIRs) etched on top of the signal strip, in a separated substrate. It is shown that if the symmetry plane of the line (a magnetic wall) is perfectly aligned with the electric wall of the SIR at the fundamental resonance, the line is transparent. However, if symmetry is somehow ruptured, a notch in the transmission coefficient appears. The notch frequency and depth can thus be mechanically controlled, and this property can be of interest for the implementation of sensors and barcodes, as it is discussed.

1. Introduction appears, and such notch widens if the line is loaded with several resonators. The stopband characteristic exhibited The topic of metamaterials has experienced an exponential in a transmission line loaded with an array of electrically growth in the field of science and technology since 2000, small resonators has been interpreted as due to the negative when the first metamaterial structure (a one-dimensional effective permeability (SRRs) or permittivity (CSRRs) of the left-handed medium) was reported [1]. In particular, meta- lines [9–11]. material transmission lines (first reported in [2–4]) have attracted the attention of RF/microwave engineers, since such However, it is not necessary to invoke metamaterials the- artificial lines exhibit further controllability than ordinary ory to interpret and understand the stopband functionality lines, and they can be used as building blocks for the ofSRRorCSRRloadedlines.Inmostapplications,thelines implementation of microwave devices with small size and/or areloadedwithasingleresonatorstagebecausethisreduces high performance and with novel functionalities as well [5– size or suffices to achieve the required functionality. Indeed 7]. the modeling through equivalent circuits of transmission There are nowadays many different approaches and types lines loaded with electrically small resonators, such as those of metamaterial transmission lines. Among them, the so- aforementioned, has been a subject of an intensive study in called resonant type approach has been revealed to be very the last years [11–16]. Certainly, transmission lines loaded useful for the implementation of microwave components with a single or with few resonators cannot be considered (including filters, dividers, and leaky wave antennas) [7]. to be effective media metamaterials. However, in certain Resonant type metamaterial transmission lines are imple- applications, the controllability of the dispersion and char- mented by loading a host line with electrically small res- acteristic impedance of the loaded lines are fundamental to onant elements, typically, although not exclusively, split achieve the required functionality, and for this reason such ring resonators (SRRs) [8, 9] and complementary split ring lines are typically referred to as metamaterial transmission resonators (CSRRs) [10]. By loading a host line with a coupled lines [17]. In other applications, the key point is to achieve (electrically, magnetically, or both) resonator, a notch in stopband functionality. In these cases, as long as the consid- the transmission coefficient (at the fundamental frequency) ered resonators are electrically small and potentially useful 2 International Journal of Antennas and Propagation

w2 line-to-resonator capacitances and thin substrates), the dominant line-to-resonator coupling mechanism is electric. Regardless of the coupling mechanism, if both the resonator and the line are symmetric, there is a necessary E ts H + + + + condition for particle excitation: either the symmetry planes Electric wall l2 ofthelineandresonatorarenotaligned,or,iftheyare −−−− w1 aligned, they must be of the same electromagnetic nature (either an electric wall or a magnetic wall). In other words, if the symmetry planes of the line and resonator are aligned and they are of distinct electromagnetic nature, then elec- l1 tromagnetic coupling capable of exciting the resonator does not arise. The result is a transmission coefficient for the Figure 1: Typical topology of a folded SIR and relevant dimensions. loaded line close to 1 even at the fundamental resonance of The resonator can be excited by a time varying electric or magnetic the considered particle. This operating principle is universal, field oriented in the indicated directions. The distribution of current being independent of the line type, the resonator, and the and charges is also sketched. resonator-to-line coupling mechanism [19, 20]. To gain more insight into this feature, let us consider a symmetric structure consisting of a microstrip line loaded with a folded SIR on top of the signal strip, with aligned for the implementation of effective media metamaterials, symmetry planes. As mentioned above, the particle cannot the lines are usually designated as transmission lines with be excited at the fundamental resonance, by neither the metamaterial loading. The structures considered in this magnetic field nor the electric field generated by the line. Due work belong to this category: microstrip lines loaded with to symmetry, the magnetic field components at both sides of stepped impedance resonators (SIRs). We have considered thesymmetryplaneoftheparticleareperfectlycancelled,and both folded SIRs and conventional SIRs coupled to the lines the magnetic field is not able to induce circulating currents and separated from the signal strip of the line by means of in the SIR. Analogously, for electric field excitation, a net a dielectric slab. It will be seen that the response of the SIR component of the electric field orthogonal to the symmetry loaded lines can be tailored through mechanical actuation planeisrequired,andthisisimpossibleduetothesymmetry and potential applications of this effect are discussed. ofthelineandtothefactthatthesymmetryplaneof the line is a magnetic wall. However, if the symmetry is broken,theperfectcancellationoffieldsvanishes,theparticle 2. Microstrip Lines Loaded with Stepped is excited, and the incident power is partially reflected at Impedance Resonators (SIRs) the fundamental resonance, thus providing a notch in the transmission coefficient at that frequency. Stepped impedance resonators are common planar resonators in microwave engineering [18]. Essentially SIRs are metallic open-ended strip resonators where the width of 3. Mechanical Reconfigurability of the the strip is varied abruptly. In a trisection SIR, it is well SIR Loaded Microstrip Lines and known that by narrowing the central section and widening Possible Applications the external ones, the resonator can be made electrically small as compared to a uniform half wavelength resonator [18]. In order to symmetrically load a microstrip line with an Additional miniaturization can be achieved by folding the SIR,threemetallevelsarerequired.Apossiblemultilayer SIR, as Figure 1 illustrates. It is important to mention, for structure is the one depicted in Figure 2, with top and bottom thepurposesofthispaper,thattheSIRexhibitsanelectric substrates. To mechanically rupture the symmetry of the wall at the indicated (symmetry) plane (see Figure 1)atthe structure and thus achieve a notch in the transmission coeffi- fundamental resonance. Hence, the resonator exhibits an cient, there are several possibilities. For instance, the SIR and electric dipole perpendicular to the symmetry plane, and thelinecanbeetchedondifferent(movable)substrateswitha itcanbeexcitedthroughanelectricfieldorientedinthat relative motion. Thus, by laterally displacing the top substrate, direction. Moreover, in a folded SIR, as the one depicted in symmetry is broken. Another interesting approach to rupture Figure 1, the current loop induces a magnetic dipole moment symmetry is by implementing the SIRs with deflectable and therefore the resonator can also be magnetically driven cantilever type (movable) arms through microelectrome- (this is not the case in unfolded SIRs). chanical systems (MEMS) (similar to the reconfigurable SRR According to the previous statements, in a transmission loaded lines reported in [21]). Any physical variable able line loaded with a folded SIR, the line-to-resonator coupling to deflect the suspended parts of the SIR can potentially may be electric, magnetic, or mixed, whereas in a transmis- be detected/sensed through its effect on the transmission sion line loaded with an unfolded SIR the magnetic coupling coefficient of the loaded line. Of course the MEMS-based is prevented. For instance, it was demonstrated in [19]that SIRs can be actuated electronically, with the potential to in coplanar waveguide (CPW) transmission lines loaded with implement reconfigurable notch filters. However, it is also foldedSIRsonthebacksideofthesubstrate(withmeaningful possible to break symmetry through the effects of an external International Journal of Antennas and Propagation 3

SIR

Top substrate Signal strip

Bottom substrate Ground plane

Figure 2: Cross-sectional view of a microstrip line loaded with an SIR, etched on top of it. force, pressure, torque, and so forth. Alternatively, the effects to these conventional approaches is to align the symmetry of these variables can also be potentially sensed/detected by planesofthelineandresonator.Thus,byusingamicrostrip using malleable/flexible substrates, rather than deflectable line loaded with folded SIRs, if the resonator symmetry plane SIRs. This eases the fabrication process and reduces costs. is aligned with that of the line, the resonator is not coupled Finally, dielectric loading may be another strategy to rupture to the line and the notch at the fundamental resonance symmetry. The idea behind this approach is to add a material does not arise. Conversely, the logic “1” can be achieved, for or substance on top of the SIR or in specific test regions of the instance, by laterally displacing the resonator. Under these structure (for instance, by creating symmetric holes inside the conditions, the magnetic wall of the line and the electric top substrate). As long as the material or substance exhibits wall of the resonator are not aligned, and the resonator is a symmetric distribution of the dielectric permittivity, the excited. Interestingly, as long as the SIRs can be laterally line is transparent to signal propagation. However, if the shifted independently, a reconfigurable barcode is potentially dielectric loading is not symmetric, this can be detected by possible. Figure 3(a) depicts the top view of a microstrip line the appearance of a notch in the transmission coefficient. One loaded with three folded SIRs, each designed to resonate at possibility that this approach opens concerns the detection of a different frequency. Notice that in this configuration the defects or imperfections in certain substances by comparison central SIR is aligned with the line, whereas the SIRs of the to a reference or standard. This approach may be useful extremes are nonaligned. Therefore the line is encoded with for analysis of substances, including organic tissues and the code “101.” Figure 3(b) depicts the simulated frequency microfluidics. Sensors for analysis of organic tissues and response (i.e., the spectral signature) of this structure and also dielectric monitoring of microfluidic channels based on the the one corresponding to the complementary code, that is, frequency or quality-factor variation of resonant elements “010,” where the aligned SIRs are those of the extremes. have recently been proposed (see, e.g., [22–24]). Intheproposedbarcode,thegroundplaneiskept unaltered, this being an advantage as compared to other proposals [26]. An important aspect to highlight is that the 4. Illustrative Preliminary Examples design of the line and SIRs does not require special attention, since the main purpose is to achieve narrow notches. This In this section, two application examples of SIR loaded micr- allows accommodating the maximum number of resonators ostrip lines are reported. Firstly, the symmetry properties of (resonance frequencies) in a predefined frequency spectrum, such loaded lines are exploited as an encoding mechanism thus embedding as many bits as possible. Narrow notches to implement an RF (radiofrequency) barcode. Secondly, the typically arise if the coupling between the line and the symmetry rupture is used as a sensing mechanism to design resonatorisweak,asshowninthestructureofFigure3 an alignment/displacement sensor. (the reason is that the dominant electric coupling is poor because of the extremely narrow line). Conversely, in the next 4.1. RF Barcode. A multiresonator-based chipless radiofre- example, the aim is to obtain a high sensitivity of the notch quency identification (RFID) tag is an RF barcode where magnitude (depth) with the lateral displacement, and this the information is stored in the so-called spectral signature requires a specific design. of the tag. Such spectral signature is obtained by loading a transmission line with several resonators, each resonating at a different frequency [25]. Each resonator corresponds to a bit, 4.2. Alignment/Position Sensor. Let us now consider the and the logic “1” or “0” is chosen by the presence or absence application of SIR loaded microstrip lines to detect lack of of the notch at the resonance frequency (i.e., the encoding lateral alignment or to measure lateral shift between two is in amplitude). By coupling the resonator to the line, the surfaces with relative motion to each other (other proposals notch appears, giving the logic level “1.” On the contrary, can be found in [27, 28]). In this case, a figure of merit is the notch can be prevented (logic level “0”) by removing the sensitivity of the notch magnitude and/or frequency with or short-circuiting the resonator (this shifts the resonance the lateral displacement. For the implementation of these frequency outside the region of interest) [25]. An alternative sensors, it is critical to adequately choose the topology of 4 International Journal of Antennas and Propagation

−3 (dB) 1| 2

W |S Port 1 Port 2 −6

−9 2.0 2.5 3.0 3.5 Frequency (GHz)

“010” “101” (a) (b)

Figure 3: Microstrip line loaded with folded SIRs for the code “101” (a) and lossy electromagnetic simulation of the transmission coefficient corresponding to the indicated codes (b). The top and bottom substrates are Rogers RO3010 with thickness ℎ = 0.254 mm, dielectric constant 𝜀𝑟 = 11.2, and loss tangent tan 𝛿 = 0.0023. The dimensions are line width 𝑊 = 0.2 mm, corresponding to a characteristic impedance close to 50 Ω; for the folded SIRs, 𝑤1 (2.5 GHz) = 2 mm, 𝑤1 (2.6 GHz) = 1.8 mm, 𝑤1 (2.7 GHz) = 1.64 mm, 𝑤2 = 0.2 mm, 𝑠 = 1.2 mm, 𝑙1 (2.5 GHz) = 3.26 mm, 𝑙1 (2.6 GHz) = 3.06 mm, 𝑙1 (2.7 GHz) = 2.9 mm, and 𝑙2 = 6 mm.

Ws L/2 L/2

Port 1 Port 2 C Ls

l s C sl Csl ·Csg }C = s C +C Csg sl sg

(a) (b)

Figure 4: Proof of concept demonstrator of an SIR loaded microstrip line useful as alignment/position sensor for a displacement corresponding to the strongest notch (a) and lumped element equivalent circuit model (b). The microstrip line dimensions are 𝑊=2mm, 𝑊𝑠 = 5.8 mm, and 𝑙𝑠 = 7.8 mm; the lengths of the SIR sections are 2 mm, and the widths are 𝑤1 = 7.6 mm and 𝑤2 = 0.2 mm. The top and bottom substrates are Rogers RO3010 with ℎ = 0.254 mm, 𝜀𝑟 = 11.2,andtan𝛿 = 0.0023. The inductance and capacitance of the line section are 𝐿 and 𝐶, the inductance of the SIR is 𝐿𝑠, the SIR-to-line capacitance is 𝐶𝑠𝑙, and the SIR-to-ground capacitance is 𝐶𝑠𝑔. both the SIR and the line. In order to boost the electric in Figure 4(a), corresponding to the lateral shift providing the coupling between the line and the SIR, it is important to widest and deepest notch. To enhance the electric coupling enhancethepatchcapacitanceoftheSIRtotheline(as for nonsymmetric loadings, the width of the line has been well as to the ground), and for this reason the line width chosen to be identical to the length of the wide strip sections must be widened. Since this reduces the line characteristic oftheSIR.Wehavethusetchedaslotinthegroundplane(see impedance, in order to match the ports to 50 Ω,awindow thedimensionsinFigure4) to achieve good matching to 50 Ω in the ground plane below the position of the signal strip can in the symmetrically loaded line. be etched. This enhances the line inductance and drops the A lumped element equivalent circuit model of the struc- line capacitance simultaneously, compensating the reduction ture of Figure 4(a) is proposed and depicted in Figure 4(b). of the characteristic impedance. Moreover, since the electric According to this simple model, it is clear that to strengthen coupling is generally stronger than magnetic coupling in the notch it is necessary to increase the ratio between the SIRs, we have opted to use an unfolded configuration. Specifi- resulting patch capacitance, 𝐶𝑠 (given by the capacitance cally, we have designed the proof of concept prototype shown between the SIR and line, 𝐶𝑠𝑙,andbythatbetweenthe International Journal of Antennas and Propagation 5

0 S21 S11 (dB) |

21 −20 |S , | 11 |S −40 180

Port 1 Port 2 0 S phase (deg) 21 S 21 11 S , 11 −180 S 0.0 0.5 1.0 1.5 2.0 2.5 Frequency (GHz)

EM sim. Circuit sim. (a) (b)

Figure 5: Topology of an electrically small SIR loaded microstrip line (a) and transmission/reflection coefficients from the lossless electromagnetic and circuit simulations (b). The structure is the same as that of Figure 4(a) with the exception of the SIR length, with the lengths of the wide and narrow sections being, respectively, 4 mm and 6 mm. The lumped element values are 𝐿 = 4.43 nH, 𝐶 = 1.58 pF, 𝐿𝑠 = 6.90,and𝐶𝑠 = 2.66 pF.

0 3.0 0 mm 0

2.8 −10 0.2 mm −10

2.6 (dB) 0 (dB) 0.4 mm f | −20 −20 (GHz) 21 @ 0 | |S 2.4 0.6 f

mm 21

0.8 |S −30 mm −30 1 mm 2.2 2 mm −40 −40 2.0 2.0 2.2 2.4 2.6 2.8 3.0 3.2 0.0 0.4 0.8 1.2 1.6 2.0 Frequency (GHz) Displacement (mm)

|S21| f0

(a) (b)

Figure 6: Transmission coefficient (a) and dependence of the notch magnitude and frequency (b) with lateral displacement corresponding to the structure of Figure 4, inferred from the lossy electromagnetic simulation. The displacement for maximum notch magnitude, that is, the dynamic range, is 2 mm.

SIR and ground, 𝐶𝑠𝑔), and the SIR inductance, 𝐿𝑠.Hence, sensitivity due to the increase in the transversal dimension). the equivalent circuit confirms that the electric coupling The circuit parameters have been extracted in an analogous should be enhanced as much as possible, and for this reason manner as in the procedure reported in [29]. In particular, the step in width in the considered SIR is substantially such systematic method is based on a mapping from the extreme. With a view to validate the circuit model, we electromagnetic simulation to the circuit response as follows: have increased the inductance 𝐿𝑠 and the capacitance 𝐶𝑠 (i) the impedance of the series resonator 𝐿𝑠-𝐶𝑠 nulls at the by lengthening, respectively, the narrow and the wide SIR notch frequency, (ii) when the series and shunt impedances ∘ sections, as shown in Figure 5(a).Bythismeanstheunitcell are conjugate the transmission coefficient phase is −90 ,and length is shortened at the resonance frequency and the circuit (iii) the intersection between the reflection coefficient and model is expected to predict the frequency response of the the unit normalized resistance circle provides the resonance structure more accurately (note that this is at the expense of frequency of the parallel resonator 𝐶-𝐿𝑠-𝐶𝑠 and also (iv) the 6 International Journal of Antennas and Propagation

in Figure 6(b).Ascanbeobserved,bothelectricalvari- ables are quite sensitive to the displacement, indicative of a significant increase in the capacitive line-to-SIR-to-ground coupling with the displacement. Note that the sensitivity is particularly very high in magnitude for small displacements, and hence the sensor is very suitable for alignment (a) purposes. Specifically, the average sensitivity in frequency is 42.95 MHz/100 𝜇minthefulldynamicrange,whilein magnitude is −5.42 dB/100 𝜇mintherangeupto400𝜇m. The previous proof of concept demonstrator was fabricated and measured in an experimental setup that enables 3D calibrations through manual positioners (see Figure 7). Thus, the measurement of the transmission coefficient was carried out for different displacements, manually performed. Note that it was not possible to perfectly align the SIR with thelineduetothefactthatthesensorfeaturesaveryhigh sensitivity for small shifts (for this reason a measured sample close to the symmetric structure was labeled as the reference one). Furthermore, the measured notch frequencies were rel- ativelymuchhigherthanthoseobtainedbyelectromagnetic simulations, and this is primarily attributed to a drop of the capacitance 𝐶𝑠 caused by a thin air layer between the top and (b) bottom substrates (indeed further simulations have suggested that an in-between air layer of 75 𝜇mshiftsthenotch 0 Reference frequency upwards up to the measured results). Nevertheless, the proof of concept is experimentally validated. In the near future, we plan to characterize the structure by considering a −10 specific distance between the line and the SIR (using a setup similar to that reported in [30]). It is also worth to bear in

(dB) mind the acquisition of a more sophisticated experimental

| −20 0.635 mm

21 setup to perform 3D calibrations with higher accuracy. |S Toendthissection,wewouldliketomentionthatthe −30 notch magnitude and spectral position are also sensitive to 1.27 mm the force applied on top of the SIR substrate. This opens the possibility to implement force or pressure sensors based on 1.905 −40 mm the reported approach.

3.03.23.43.63.84.0 Frequency (GHz) 5. Conclusions (c) In conclusion, it has been shown that the SIRs are useful Figure 7: Photograph of the fabricated SIR loaded microstrip line of resonant particles as loading elements in microstrip lines Figure 4 on separate substrates, where matched input/output lines 𝑊 = 0.2 aimed to act as microwave sensors or RF barcodes based on of mm have been added to solder the connectors (a). symmetry rupture. Specifically, for sensing purposes, SIRs Photograph of the experimental setup with the substrates attached provide high sensitivity due to the strong electric coupling to Teflon slabs (a material with a low dielectric constant, 𝜀𝑟 = 2.08) and with manual positioners labeled in inches (b). Measured between the line and the resonator. This is an advantage transmission coefficient for some lateral displacements of the SIR over other proposed sensors based on symmetry properties (c). (e.g., displacement sensors that use split ring resonators). Moreover, these SIR loaded lines can also be potentially useful for other types of sensors, such as pressure sensors, or sensors for dielectric characterization, where the high series inductance 𝐿.AscanbeappreciatedinFigure5(b),the sensitivity of SIR loaded lines can represent an advantage over circuit simulation fits the lossless electromagnetic simulation other exiting approaches. Work is in progress in this regard. in a very good approximation. Concerning barcodes, a proof of concept has been developed. Figure 6(a) plots the transmission coefficient inferred Although the number of bits achievable with the reported from the lossy electromagnetic simulation of the proof approach is limited (unless the spectral bandwidth is very of concept structure shown in Figure 6(a) for different high), tags based on the reported approach can be of interest lateral displacements. The notch magnitude and frequency in systems that do not require to store too much information versus the displacement were recorded and are indicated and do not need a high level of security. Moreover, these International Journal of Antennas and Propagation 7 tagscanbeprogrammable(symmetrycanbeachievedby [10] F. Falcone, T. Lopetegi, J. D. Baena, R. Marques,F.Mart´ ´ın, adding metallic strips to the SIRs) and can be implemented in and M. Sorolla, “Effective negative-𝜀 stopband microstrip lines flexible substrates by means of printed techniques (including based on complementary split ring resonators,” IEEE Microwave low cost paper substrates). Thus, the potential can be very and Wireless Components Letters,vol.14,no.6,pp.280–282, high (they can be useful, e.g., for document identification, 2004. product identification, electoral processes with small voting [11] J. D. Baena, J. Bonache, F. Mart´ın et al., “Equivalent-circuit members, etc.). In summary, several applications of SIR models for split-ring resonators and complementary split- loaded lines have been discussed, and a proof of concept ring resonators coupled to planar transmission lines,” IEEE (experimentally validated) of an alignment/position sensor Transactions on Microwave Theory and Techniques,vol.53,no. has been provided. 4, pp. 1451–1460, 2005. [12] L. J. Rogla,J.Carbonell,andV.E.Boria,“Studyofequivalent´ circuits for open-ring and split-ring resonators in coplanar Conflict of Interests waveguide technology,” IET Microwaves, Antennas and Propa- gation,vol.1,no.1,pp.170–176,2007. The authors declare that there is no conflict of interests regarding the publication of this paper. [13] F. Aznar, J. Bonache, and F. Mart´ın, “Improved circuit model for left-handed lines loaded with split ring resonators,” Applied Physics Letters,vol.92,no.4,ArticleID043512,2008. Acknowledgments [14] J. Bonache, M. Gil, O. Garc´ıa-Abad, and F. Mart´ın, “Parametric analysis of microstrip lines loaded with complementary split This work has been supported by MINECO (Spain) (Projects ring resonators,” Microwave and Optical Technology Letters,vol. TEC2010-17512, CSD2008-00066, and TEC2011-13615-E) and 50,no.8,pp.2093–2096,2008. by AGAUR (Generalitat de Catalunya), through Project 2009SGR-421. Jordi Naqui is also in debt to MECD (Spain) for [15]J.Naqui,M.Duran-Sindreu,´ A. Fernandez-Prieto,´ F. Mesa, F. Medina, and F. Mart´ın, “Multimode propagation and complex supporting his work through the FPU Grant AP2010-0431. waves in CSRR-based transmission line metamaterials,” IEEE Antennas and Wireless Propagation Letters,vol.11,pp.1024–1027, References 2012. [16] J. Naqui, M. Duran-Sindreu,´ and F. Mart´ın, “Modeling split [1] D. R. Smith, W. J. Padilla, D. C. Vier, S. C. Nemat-Nasser, and ring resonator (SRR) and complementary split ring resonator S. 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