Generalized Poisson Summation Formula for Tempered Distributions

Total Page:16

File Type:pdf, Size:1020Kb

Generalized Poisson Summation Formula for Tempered Distributions 2015 International Conference on Sampling Theory and Applications (SampTA) Generalized Poisson Summation Formula for Tempered Distributions Ha Q. Nguyen and Michael Unser Biomedical Imaging Group, Ecole´ Polytechnique Fed´ erale´ de Lausanne (EPFL) Station 17, CH–1015, Lausanne, Switzerland fha.nguyen, michael.unserg@epfl.ch Abstract—The Poisson summation formula (PSF), which re- In the mathematics literature, the PSF is often stated in a lates the sampling of an analog signal with the periodization of dual form with the sampling occurring in the Fourier domain, its Fourier transform, plays a key role in the classical sampling and under strict conditions on the function f. Various versions theory. In its current forms, the formula is only applicable to a of (PSF) have been proven when both f and f^ are in appro- limited class of signals in L1. However, this assumption on the signals is too strict for many applications in signal processing priate subspaces of L1(R) \C(R). In its most general form, ^ that require sampling of non-decaying signals. In this paper when f; f 2 L1(R)\C(R), the RHS of (PSF) is a well-defined we generalize the PSF for functions living in weighted Sobolev periodic function in L1([0; 1]) whose (possibly divergent) spaces that do not impose any decay on the functions. The Fourier series is the LHS. If f and f^ additionally satisfy only requirement is that the signal to be sampled and its weak ^ −1−" derivatives up to order 1=2 + " for arbitrarily small " > 0, grow jf(x)j+jf(x)j ≤ C(1+jxj) ; 8x 2 R, for some C; " > 0, slower than a polynomial in the L2 sense. The generalized PSF then it can be shown [6], [7] that (PSF) holds pointwise with will be interpreted in the language of distributions. both sides converging absolutely. It is also known [8], [9] ^ that (PSF) holds pointwise when f; f 2 L1(R) \C(R) and I. INTRODUCTION f^ has bounded total variation. Other versions of the PSF on L1(R) can also be found in [9]–[11]. The main limitation of A widely used result in engineering is that sampling in the above mathematical forms of the PSF is that they do not time corresponds to periodization in frequency and vice versa. apply to functions that are non-decreasing, such as our above This beautiful relation is captured in the celebrated Poisson example of a ramp signal or any realization of a stationary summation formula stochastic process, not to mention the non-stationary ones like X X the Brownian motion which may even grow at infinity [12]. f(k)e−2πikξ = f^(ξ + k): (PSF) Intuitively, however, there does not seem to be any compelling k2 k2 Z Z reason why a continuous signal cannot be sampled even if it In other words, the discrete-time Fourier transform of the sam- is not decaying. pled sequence is equal to the periodization of the continuous- In this paper we attempt to bridge the gap between the time Fourier transform of the analog signal. This identity is engineering and mathematics statements of the PSF. We will the key behind many results in sampling theory including show that (PSF) holds in the sense of distribution theory [13] the classical Shannon’s sampling theorem [1] for bandlimited for a broad class of continuous signals included in a weighted signals. What appears to be missing, however, is a proof of s Sobolev space L2;1=w(R), whose weak derivatives up to order f (PSF) for a general function beyond the classical hypotheses, s are in the weighted-L2 space L2;1=w(R). Here, the decaying which will be detailed later. In engineering texts, see for ex- weight 1=w allows growing signals. In particular, when the ample [2]–[5], the common “proof” of (PSF) for an arbitrary weight 1=w decays polynomially and the order of smoothness function f is based on the following observations: (1) the s is above the threshold 1=2, both sides of (PSF) are well- sampled sequence is obtained by multiplying the analog signal defined periodic distributions and they are equal when acting with a Dirac comb; (2) multiplication maps to convolution on periodic test functions. In light of our new result, it is in Fourier-domain; and (3) the Fourier transform of a Dirac easy to see that (PSF) holds for the ramp function, and more comb is also a Dirac comb. The problem with this argument generally, for any piecewise polynomials that are continuous. is that the multiplication of a Dirac comb with a general For brevity, most of the proofs in this paper are omitted or function is not necessarily a tempered distribution, so that the just sketched. Detailed proofs can be found in our extensive convolution theorem may not be applicable. Take, for example, papers [14], [15] that are currently in preparation.1 x+jxj the popular ramp signal f(x) = 2 . Since f(x) is not The rest of the paper is organized as follows. Section II differentiable at 0, the multiplication between f and the Dirac introduces some notations and definitions that will be used comb P δ(· − k) is prohibited. It means that the above k2Z throughout the paper. Section III discusses the sampling in argument for (PSF) does not even work for this seemingly weighted Sobolev spaces. The generalized PSF is stated in well-behaved signal. 978-1-4673-7353-1/15/$31.00 c 2015 IEEE 1The manuscripts are available from the authors upon request. 978-1-4673-7353-1/15/$31.00 ©2015 IEEE 1 2015 International Conference on Sampling Theory and Applications (SampTA) p Section IV with a sketched proof. Finally, some concluding w(0)=Cw ≥ 1=Cw, for all x. This means that every submul- remarks are made in Section V. tiplicative weighting function is lower-bounded. A prototypical example of submultiplicative weights is w(x) = (1+jxj2)α/2, II. NOTATION AND DEFINITION for some α ≥ 0. For this weight, we can choose Cw to be α/2 Throughout the paper, we deal with complex-valued func- Cα = 2 . tions and sequences. Following the convention in signal pro- Definition 2 (Weighted L and ` spaces). For p ≥ 1 and cessing, we use parentheses for functions and square brackets p p a weighting function w, a function f is in L ( ) if fw is for sequences to specify their values at a particular point. Espe- p;w R in Lp(R); a sequence c is in `p;w(Z) if fc[k]w(k)gk2Z is in cially, for a function f, f[·] denotes the sequence ff(k)gk2 . Z `p(Z). The corresponding weighted norms are defined as For p ≥ 1, the spaces Lp(R) and `p(Z) include functions and sequences, respectively, whose p-norms are finite. The scalar kfk := kfwk ; kck := kc w[·]k : Lp;w(R) Lp(R) `p;w(Z) `p(Z) product between two functions in L2(R) is defined as Z Note that if w is submultiplicative then Lp;w(R) ⊂ Lp(R), hf; gi := f(x)g(x)dx; for f; g 2 L ( ) 2 R and `p;w(Z) ⊂ `p(Z) because w is lower-bounded. R As usual, the notation h·; ·i is also used for the action of a Definition 3 (Weighted hybrid norm spaces). For p; q ≥ 1, the distribution on a test function. Sometimes, it is useful to write hybrid norm space Wp;q(R) includes all functions f : R ! C the complex sinusoid e−2πikξ as e−2πih·;ki to imply that it is a whose following norm is finite function in variable ξ. We also adopt standard notations used 0 !q=p 11=q in Schwartz’s distribution theory [13]. We use the symbols ^ Z 1 X kfk := jf(x + k)jp dx ; and _ to denote the forward and inverse Fourier transforms, Wp;q (R) @ A 0 respectively, of a tempered distribution f 2 S0(R), i.e. k2Z D E ^ with usual adjustments when p or q is infinity. The weighted f;^ ' = h(f) ;'i := hf; '^i ; 8' 2 S(R); hybrid norm space Wp;q;w(R) w.r.t. a weighting function w is _ f;ˇ ' = h(f) ;'i := hf; 'ˇi ; 8' 2 S(R) defined according to the following weighted norm where kfkW ( ) := kfwkW ( ): Z p;q;w R p;q R '^(ξ) = (')^(ξ) := '(x)e−2πiξxdx; These hybrid (mixed) norm spaces were mentioned in [9]. ZR _ 2πiξx They are similar to Wiener amalgam spaces Wfp;q(R) [16]– 'ˇ(x) = (') (x) := '(ξ)e dξ: [18], where the discrete and continuous norms are mixed in R reverse order, i.e. The set of continuous functions is denoted by C(R), whereas 1 1=p Cc (R) = D(R) denotes the space of bump functions that 1 p=q! X Z are infinitely differentiable and compactly supported. For an kfk := jf(x + k)jq dx : Wfp;q ( ) 1 R 0 interval T of R, C (T) denotes the space of T-periodic test k2Z functions that are infinitely differentiable. Here, a function These amalgam spaces play a central role in many important ' is T-periodic if ' = '(· + jTjk); 8k 2 Z. The constants throughout the paper are denoted by C with some subscripts results of the non-uniform sampling theory in shift-invariant denoting the dependence of the constants on those parameters. (spline-like) spaces [19]–[23]. Deeper results regarding Wiener amalgam spaces and their generalizations can be found For example, Cx;y;z is a constant that depends only on x; y and z.
Recommended publications
  • Convolution! (CDT-14) Luciano Da Fontoura Costa
    Convolution! (CDT-14) Luciano da Fontoura Costa To cite this version: Luciano da Fontoura Costa. Convolution! (CDT-14). 2019. hal-02334910 HAL Id: hal-02334910 https://hal.archives-ouvertes.fr/hal-02334910 Preprint submitted on 27 Oct 2019 HAL is a multi-disciplinary open access L’archive ouverte pluridisciplinaire HAL, est archive for the deposit and dissemination of sci- destinée au dépôt et à la diffusion de documents entific research documents, whether they are pub- scientifiques de niveau recherche, publiés ou non, lished or not. The documents may come from émanant des établissements d’enseignement et de teaching and research institutions in France or recherche français ou étrangers, des laboratoires abroad, or from public or private research centers. publics ou privés. Convolution! (CDT-14) Luciano da Fontoura Costa [email protected] S~aoCarlos Institute of Physics { DFCM/USP October 22, 2019 Abstract The convolution between two functions, yielding a third function, is a particularly important concept in several areas including physics, engineering, statistics, and mathematics, to name but a few. Yet, it is not often so easy to be conceptually understood, as a consequence of its seemingly intricate definition. In this text, we develop a conceptual framework aimed at hopefully providing a more complete and integrated conceptual understanding of this important operation. In particular, we adopt an alternative graphical interpretation in the time domain, allowing the shift implied in the convolution to proceed over free variable instead of the additive inverse of this variable. In addition, we discuss two possible conceptual interpretations of the convolution of two functions as: (i) the `blending' of these functions, and (ii) as a quantification of `matching' between those functions.
    [Show full text]
  • The Convolution Theorem
    Module 2 : Signals in Frequency Domain Lecture 18 : The Convolution Theorem Objectives In this lecture you will learn the following We shall prove the most important theorem regarding the Fourier Transform- the Convolution Theorem We are going to learn about filters. Proof of 'the Convolution theorem for the Fourier Transform'. The Dual version of the Convolution Theorem Parseval's theorem The Convolution Theorem We shall in this lecture prove the most important theorem regarding the Fourier Transform- the Convolution Theorem. It is this theorem that links the Fourier Transform to LSI systems, and opens up a wide range of applications for the Fourier Transform. We shall inspire its importance with an application example. Modulation Modulation refers to the process of embedding an information-bearing signal into a second carrier signal. Extracting the information- bearing signal is called demodulation. Modulation allows us to transmit information signals efficiently. It also makes possible the simultaneous transmission of more than one signal with overlapping spectra over the same channel. That is why we can have so many channels being broadcast on radio at the same time which would have been impossible without modulation There are several ways in which modulation is done. One technique is amplitude modulation or AM in which the information signal is used to modulate the amplitude of the carrier signal. Another important technique is frequency modulation or FM, in which the information signal is used to vary the frequency of the carrier signal. Let us consider a very simple example of AM. Consider the signal x(t) which has the spectrum X(f) as shown : Why such a spectrum ? Because it's the simplest possible multi-valued function.
    [Show full text]
  • The Discrete-Time Fourier Transform and Convolution Theorems: a Brief Tutorial
    The Discrete-Time Fourier Transform and Convolution Theorems: A Brief Tutorial Yi-Wen Liu 25 Feb 2013 (Revised 21 Sep 2015) 1 Definitions and interpretation 1.1 Units Throughout this semester, we will use the integer-valued variable n as the time variable for discrete-time signal processing; that is, n = −∞, ..., −1, 0, 1, ..., ∞. In this convention, the unit of n is dimensionless, but be aware that in reality the sampling period is 1/fs, where fs denotes the sampling rate. In some text books, 1/fs is denoted as T , and its unit is second. In this course we do not do this, thereby favoring conciseness of notations. 1.2 Definition of DTFT The discrete-time Fourier transform (DTFT) of a discrete-time signal x[n] is a function of frequency ω defined as follows: ∞ ∆ X(ω) = x[n]e−jωn. (1) n=−∞ X Conceptually, the DTFT allows us to check how much of a tonal component at fre- quency ω is in x[n]. The DTFT of a signal is often also called a spectrum. Note that X(ω) is complex-valued. So, the absolute value |X(ω)| represents the tonal component’s magnitude, and the angle ∠X(ω) tells us the phase of the tonal component. Also note that Eq. (1) defines the DTFT as the inner product between the signal and the complex exponential tone e−jωn. Because complex exponentials {e−jωn,ω ∈ [−π, π]} form a complete family of orthogonal bases for (practically) all signals of interest, what Eq. (1) essentially does is to project x[n] onto the space spanned by all complex exponen- tials.
    [Show full text]
  • Observational Astrophysics II January 18, 2007
    Observational Astrophysics II JOHAN BLEEKER SRON Netherlands Institute for Space Research Astronomical Institute Utrecht FRANK VERBUNT Astronomical Institute Utrecht January 18, 2007 Contents 1RadiationFields 4 1.1 Stochastic processes: distribution functions, mean and variance . 4 1.2Autocorrelationandautocovariance........................ 5 1.3Wide-sensestationaryandergodicsignals..................... 6 1.4Powerspectraldensity................................ 7 1.5 Intrinsic stochastic nature of a radiation beam: Application of Bose-Einstein statistics ....................................... 10 1.5.1 Intermezzo: Bose-Einstein statistics . ................... 10 1.5.2 Intermezzo: Fermi Dirac statistics . ................... 13 1.6 Stochastic description of a radiation field in the thermal limit . 15 1.7 Stochastic description of a radiation field in the quantum limit . 20 2 Astronomical measuring process: information transfer 23 2.1Integralresponsefunctionforastronomicalmeasurements............ 23 2.2Timefiltering..................................... 24 2.2.1 Finiteexposureandtimeresolution.................... 24 2.2.2 Error assessment in sample average μT ................... 25 2.3 Data sampling in a bandlimited measuring system: Critical or Nyquist frequency 28 3 Indirect Imaging and Spectroscopy 31 3.1Coherence....................................... 31 3.1.1 The Visibility function . ................... 31 3.1.2 Young’sdualbeaminterferenceexperiment................ 31 3.1.3 Themutualcoherencefunction....................... 32 3.1.4 Interference
    [Show full text]
  • Fourier Transforms & the Convolution Theorem
    Convolution, Correlation, & Fourier Transforms James R. Graham 11/25/2009 Introduction • A large class of signal processing techniques fall under the category of Fourier transform methods – These methods fall into two broad categories • Efficient method for accomplishing common data manipulations • Problems related to the Fourier transform or the power spectrum Time & Frequency Domains • A physical process can be described in two ways – In the time domain, by h as a function of time t, that is h(t), -∞ < t < ∞ – In the frequency domain, by H that gives its amplitude and phase as a function of frequency f, that is H(f), with -∞ < f < ∞ • In general h and H are complex numbers • It is useful to think of h(t) and H(f) as two different representations of the same function – One goes back and forth between these two representations by Fourier transforms Fourier Transforms ∞ H( f )= ∫ h(t)e−2πift dt −∞ ∞ h(t)= ∫ H ( f )e2πift df −∞ • If t is measured in seconds, then f is in cycles per second or Hz • Other units – E.g, if h=h(x) and x is in meters, then H is a function of spatial frequency measured in cycles per meter Fourier Transforms • The Fourier transform is a linear operator – The transform of the sum of two functions is the sum of the transforms h12 = h1 + h2 ∞ H ( f ) h e−2πift dt 12 = ∫ 12 −∞ ∞ ∞ ∞ h h e−2πift dt h e−2πift dt h e−2πift dt = ∫ ( 1 + 2 ) = ∫ 1 + ∫ 2 −∞ −∞ −∞ = H1 + H 2 Fourier Transforms • h(t) may have some special properties – Real, imaginary – Even: h(t) = h(-t) – Odd: h(t) = -h(-t) • In the frequency domain these
    [Show full text]
  • Fourier Analysis
    Chapter 1 Fourier analysis In this chapter we review some basic results from signal analysis and processing. We shall not go into detail and assume the reader has some basic background in signal analysis and processing. As basis for signal analysis, we use the Fourier transform. We start with the continuous Fourier transformation. But in applications on the computer we deal with a discrete Fourier transformation, which introduces the special effect known as aliasing. We use the Fourier transformation for processes such as convolution, correlation and filtering. Some special attention is given to deconvolution, the inverse process of convolution, since it is needed in later chapters of these lecture notes. 1.1 Continuous Fourier Transform. The Fourier transformation is a special case of an integral transformation: the transforma- tion decomposes the signal in weigthed basis functions. In our case these basis functions are the cosine and sine (remember exp(iφ) = cos(φ) + i sin(φ)). The result will be the weight functions of each basis function. When we have a function which is a function of the independent variable t, then we can transform this independent variable to the independent variable frequency f via: +1 A(f) = a(t) exp( 2πift)dt (1.1) −∞ − Z In order to go back to the independent variable t, we define the inverse transform as: +1 a(t) = A(f) exp(2πift)df (1.2) Z−∞ Notice that for the function in the time domain, we use lower-case letters, while for the frequency-domain expression the corresponding uppercase letters are used. A(f) is called the spectrum of a(t).
    [Show full text]
  • Fourier Transform, Convolution Theorem, and Linear Dynamical Systems April 28, 2016
    Mathematical Tools for Neuroscience (NEU 314) Princeton University, Spring 2016 Jonathan Pillow Lecture 23: Fourier Transform, Convolution Theorem, and Linear Dynamical Systems April 28, 2016. Discrete Fourier Transform (DFT) We will focus on the discrete Fourier transform, which applies to discretely sampled signals (i.e., vectors). Linear algebra provides a simple way to think about the Fourier transform: it is simply a change of basis, specifically a mapping from the time domain to a representation in terms of a weighted combination of sinusoids of different frequencies. The discrete Fourier transform is therefore equiv- alent to multiplying by an orthogonal (or \unitary", which is the same concept when the entries are complex-valued) matrix1. For a vector of length N, the matrix that performs the DFT (i.e., that maps it to a basis of sinusoids) is an N × N matrix. The k'th row of this matrix is given by exp(−2πikt), for k 2 [0; :::; N − 1] (where we assume indexing starts at 0 instead of 1), and t is a row vector t=0:N-1;. Recall that exp(iθ) = cos(θ) + i sin(θ), so this gives us a compact way to represent the signal with a linear superposition of sines and cosines. The first row of the DFT matrix is all ones (since exp(0) = 1), and so the first element of the DFT corresponds to the sum of the elements of the signal. It is often known as the \DC component". The next row is a complex sinusoid that completes one cycle over the length of the signal, and each subsequent row has a frequency that is an integer multiple of this \fundamental" frequency.
    [Show full text]
  • On the Duality of Regular and Local Functions
    Preprints (www.preprints.org) | NOT PEER-REVIEWED | Posted: 24 May 2017 doi:10.20944/preprints201705.0175.v1 mathematics ISSN 2227-7390 www.mdpi.com/journal/mathematics Article On the Duality of Regular and Local Functions Jens V. Fischer Institute of Mathematics, Ludwig Maximilians University Munich, 80333 Munich, Germany; E-Mail: jens.fi[email protected]; Tel.: +49-8196-934918 1 Abstract: In this paper, we relate Poisson’s summation formula to Heisenberg’s uncertainty 2 principle. They both express Fourier dualities within the space of tempered distributions and 3 these dualities are furthermore the inverses of one another. While Poisson’s summation 4 formula expresses a duality between discretization and periodization, Heisenberg’s 5 uncertainty principle expresses a duality between regularization and localization. We define 6 regularization and localization on generalized functions and show that the Fourier transform 7 of regular functions are local functions and, vice versa, the Fourier transform of local 8 functions are regular functions. 9 Keywords: generalized functions; tempered distributions; regular functions; local functions; 10 regularization-localization duality; regularity; Heisenberg’s uncertainty principle 11 Classification: MSC 42B05, 46F10, 46F12 1 1.2 Introduction 13 Regularization is a popular trick in applied mathematics, see [1] for example. It is the technique 14 ”to approximate functions by more differentiable ones” [2]. Its terminology coincides moreover with 15 the terminology used in generalized function spaces. They contain two kinds of functions, ”regular 16 functions” and ”generalized functions”. While regular functions are being functions in the ordinary 17 functions sense which are infinitely differentiable in the ordinary functions sense, all other functions 18 become ”infinitely differentiable” in the ”generalized functions sense” [3].
    [Show full text]
  • Algorithmic Framework for X-Ray Nanocrystallographic Reconstruction in the Presence of the Indexing Ambiguity
    Algorithmic framework for X-ray nanocrystallographic reconstruction in the presence of the indexing ambiguity Jeffrey J. Donatellia,b and James A. Sethiana,b,1 aDepartment of Mathematics and bLawrence Berkeley National Laboratory, University of California, Berkeley, CA 94720 Contributed by James A. Sethian, November 21, 2013 (sent for review September 23, 2013) X-ray nanocrystallography allows the structure of a macromolecule over multiple orientations. Although there has been some success in to be determined from a large ensemble of nanocrystals. How- determining structure from perfectly twinned data, reconstruction is ever, several parameters, including crystal sizes, orientations, and often infeasible without a good initial atomic model of the structure. incident photon flux densities, are initially unknown and images We present an algorithmic framework for X-ray nanocrystal- are highly corrupted with noise. Autoindexing techniques, com- lographic reconstruction which is based on directly reducing data monly used in conventional crystallography, can determine ori- variance and resolving the indexing ambiguity. First, we design entations using Bragg peak patterns, but only up to crystal lattice an autoindexing technique that uses both Bragg and non-Bragg symmetry. This limitation results in an ambiguity in the orienta- data to compute precise orientations, up to lattice symmetry. tions, known as the indexing ambiguity, when the diffraction Crystal sizes are then determined by performing a Fourier analysis pattern displays less symmetry than the lattice and leads to data around Bragg peak neighborhoods from a finely sampled low- that appear twinned if left unresolved. Furthermore, missing angle image, such as from a rear detector (Fig. 1). Next, we model phase information must be recovered to determine the imaged structure factor magnitudes for each reciprocal lattice point with object’s structure.
    [Show full text]
  • Understanding the Lomb-Scargle Periodogram
    Understanding the Lomb-Scargle Periodogram Jacob T. VanderPlas1 ABSTRACT The Lomb-Scargle periodogram is a well-known algorithm for detecting and char- acterizing periodic signals in unevenly-sampled data. This paper presents a conceptual introduction to the Lomb-Scargle periodogram and important practical considerations for its use. Rather than a rigorous mathematical treatment, the goal of this paper is to build intuition about what assumptions are implicit in the use of the Lomb-Scargle periodogram and related estimators of periodicity, so as to motivate important practical considerations required in its proper application and interpretation. Subject headings: methods: data analysis | methods: statistical 1. Introduction The Lomb-Scargle periodogram (Lomb 1976; Scargle 1982) is a well-known algorithm for de- tecting and characterizing periodicity in unevenly-sampled time-series, and has seen particularly wide use within the astronomy community. As an example of a typical application of this method, consider the data shown in Figure 1: this is an irregularly-sampled timeseries showing a single ob- ject from the LINEAR survey (Sesar et al. 2011; Palaversa et al. 2013), with un-filtered magnitude measured 280 times over the course of five and a half years. By eye, it is clear that the brightness of the object varies in time with a range spanning approximately 0.8 magnitudes, but what is not immediately clear is that this variation is periodic in time. The Lomb-Scargle periodogram is a method that allows efficient computation of a Fourier-like power spectrum estimator from such unevenly-sampled data, resulting in an intuitive means of determining the period of oscillation.
    [Show full text]
  • Numerical Analysis of Elastic Contact Between Coated Bodies
    Hindawi Advances in Tribology Volume 2018, Article ID 6498503, 13 pages https://doi.org/10.1155/2018/6498503 Research Article Numerical Analysis of Elastic Contact between Coated Bodies Sergiu Spinu 1,2 1 Department of Mechanics and Technologies, Stefan cel Mare University of Suceava, 13th University Street, 720229, Romania 2Integrated Center for Research, Development and Innovation in Advanced Materials, Nanotechnologies, and Distributed Systems for Fabrication and Control (MANSiD), Stefan cel Mare University of Suceava, Romania Correspondence should be addressed to Sergiu Spinu; [email protected] Received 31 May 2018; Revised 28 September 2018; Accepted 11 October 2018; Published 1 November 2018 Academic Editor: Patrick De Baets Copyright © 2018 Sergiu Spinu. Tis is an open access article distributed under the Creative Commons Attribution License, which permits unrestricted use, distribution, and reproduction in any medium, provided the original work is properly cited. Substrate protection by means of a hard coating is an efcient way of extending the service life of various mechanical, electrical, or biomedical elements. Te assessment of stresses induced in a layered body under contact load may advance the understanding of the mechanisms underlying coating performance and improve the design of coated systems. Te iterative derivation of contact area and contact tractions requires repeated displacement evaluation; therefore the robustness of a contact solver relies on the efciency of the algorithm for displacement calculation. Te fast Fourier transform coupled with the discrete convolution theorem has been widely used in the contact modelling of homogenous bodies, as an efcient computational tool for the rapid evaluation of convolution products that appear in displacements and stresses calculation.
    [Show full text]
  • The Poisson Summation Formula, the Sampling Theorem, and Dirac Combs
    The Poisson summation formula, the sampling theorem, and Dirac combs Jordan Bell [email protected] Department of Mathematics, University of Toronto April 3, 2014 1 Poisson summation formula Let S(R) be the set of all infinitely differentiable functions f on R such that for all nonnegative integers m and n, m (n) νm;n(f) = sup jxj jf (x)j x2R is finite. For each m and n, νm;n is a seminorm. This is a countable collection of seminorms, and S(R) is a Fréchet space. (One has to prove for vk 2 S(R) that if for each fixed m; n the sequence νm;n(fk) is a Cauchy sequence (of numbers), then there exists some v 2 S(R) such that for each m; n we have νm;n(f −fk) ! 0.) I mention that S(R) is a Fréchet space merely because it is satisfying to give a structure to a set with which we are working: if I call this set Schwartz space I would like to know what type of space it is, in the same sense that an Lp space is a Banach space. For f 2 S(R), define Z f^(ξ) = e−iξxf(x)dx; R and then 1 Z f(x) = eixξf^(ξ)dξ: 2π R Let T = R=2πZ. For φ 2 C1(T), define 1 Z 2π φ^(n) = e−intφ(t)dt; 2π 0 and then X φ(t) = φ^(n)eint: n2Z 1 For f 2 S(R) and λ 6= 0, define φλ : T ! C by X φλ(t) = 2π fλ(t + 2πj); j2Z 1 where fλ(x) = λf(λx).
    [Show full text]