Broadband Optical Beam Forming for Airborne Phased Array Antenna

H. Schippers, J. Verpoorte, P. Jorna, A. Hulzinga National Aerospace Laboratory NLR Anthony Fokkerweg 2, 1006 BM Amsterdam, the Netherlands [email protected] L. Zhuang, A. Meijerink, C. G. H. Roeloffzen, D. A. I. Marpaung, W. van Etten Telecommunication Engineering group, Faculty of Electrical Engineering University of Twente, P.O.Box 217, 7500 AE, Enschede, the Netherlands [email protected] R. G. Heideman, A. Leinse LioniX bv P.O. Box 456, 7500 AH Enschede, the Netherlands [email protected] M. Wintels Cyner Substrates Savannahweg 60, 3542 AW Utrecht, the Netherlands [email protected]

Abstract—For enhanced communication on board aircraft, novel antenna systems with broadband satellite-based 1. INTRODUCTION capabilities are required. The technology will enhance airline operations by providing in-flight connectivity for For enhanced communication on board of aircraft novel flight crew information and will bring live TV and high- antenna systems with broadband satellite-based capabilities speed Internet connectivity to passengers. The installation are required. The technology must bring live weather of such systems on board aircraft requires for aerodynamic reports to pilots, as well as live TV and high-speed Internet reasons the development a very low-profile aircraft antenna, connectivity to passengers. Satellite communication which can point to satellites anywhere in the upper services can be provided by Low Earth Orbiting (LEO) hemisphere. Major keystones for the success of steerable systems and Geostationary (GEO) systems. Well-known low-profile antennas are multi-layer printed circuit boards LEO systems are Iridium and Globalstar. The satellite orbits (PCBs) with an array of broadband antenna elements, and are optimised to provide highest link availability in the area compact micro-wave systems with appropriate beam between ±70 degrees latitude on earth. For geostationary steering capabilities. The present paper describes the communication systems much fewer satellites are required development of a prototype 8x1 optical beam forming to provide coverage on earth. For instance, Inmarsat uses network using cascades of optical ring resonators as part of only two I4 satellites to provide coverage to around 85 per a breadboard Ku-band phased array antenna.12 cent of the world's landmass and 98 per cent of the world's population. Today there are more than 300 operational TABLE OF CONTENTS geostationary satellites. These satellites are fixed at an 1. INTRODUCTION...... 1 altitude of approximately 36.000 km at the equator. These 2. SYSTEM ASPECTS ...... 2 satellites are being used for television broadcasting, 3. DEVELOPMENT OF OPTICAL BEAMFORMER ...... 3 communication and weather forecasting. In general, 4. DEVELOPMENT OF KU-BAND ANTENNA ...... 9 receiving and transmitting antennas on the earth do not need 5. DEVELOPMENT OF DEMONSTRATOR...... 13 to track such a satellite. These antennas can be fixed in 7. CONCLUSIONS ...... 13 place and are much less expensive than tracking antennas. ACKNOWLEDGMENT ...... 13 However, when the terminal for geostationary satellite REFERENCES ...... 14 communication is moving (for instance when applied on a BIOGRAPHIES ...... 16 flying aircraft) a tracking antenna is required in all circumstances. Many studies are going on worldwide to employ these Ku-band geostationary satellites for communication with mobile terminals on cars, trains, ships and aircraft. For a short period broadband internet was available on aircraft via Connexion by Boeing (CBB) 1 services. Lufthansa installed the CBB system on some of 1 978-1-4244-2622-5/09/$25.00 ©2009 IEEE their long-haul aircraft. In 2006 the CBB services ended, 2 IEEEAC paper #1507 Version 1, Updated January 26, 2009

Authorized licensed use limited to: UNIVERSITEIT TWENTE. Downloaded on October 7, 2009 at 07:26 from IEEE Xplore. Restrictions apply. because the consumer market for this service had not - AES receive band 1: 10.70 – 11.70 GHz (primary materialized as was expected. In August 2007 the US allocation to fixed satellite service) government awarded a contract to Boeing for providing Boeing Broadband Satcom Network (BBSN) services to the - AES receive band 2: 12.50 – 12.75 GHz (primary U.S. Air Force Air Mobility Command. BBSN has to allocation to fixed satellite service) provide high-speed Internet communications and direct broadcast satellite TV service as a cost effective solution for - AES transmit band: 14.00 – 14.50 GHz (secondary the U.S. government's aircraft and airborne customers. allocation to mobile satellite service)

Ku-band terminals on board moving platforms require The Aeronautical Earth Stations (AES) have to comply with broadband antennas with high gain in which the main beam ITU-R RECOMMENDATION M.1643 [3] and with ETSI can be steered continuously to geostationary satellites. EN 302 186 [4], a harmonised European Norm for satellite Major keystones for the success of steerable low-profile mobile Aircraft Earth Stations (AESs) operating in the antennas are multi-layer printed circuit boards (PCBs) with 11/12/14 GHz frequency band. an array of stacked patch antenna elements, and compact micro-wave systems with appropriate beam steering In addition, reception of commercial satellite broadcasts is capabilities. This requires the optimisation and of interest: manufacturing of a broadband multi-layered Ku-band antenna array, and development of a broadband beam - Satellite TV: 11.70 – 12.50 GHz (primary allocation to forming network. Electronic beam steering can be realized broadcast satellite service) by adding RF-phase shifters and Low Noise Amplifiers (LNA’s) to the antenna elements of the array. However, the In the Dutch FlySmart project, only the receive antenna traditional phase shifters have in general narrow band system has been developed. The objective was to develop a characteristics, and hence do not yield the required conformal phased array antenna having an instantaneous broadband capability. Alternative technologies for bandwidth of 2 GHz, covering the whole frequency range of broadband beam forming are switched beam networks 10.7 to 12.75 GHz. (using Butler matrices), the use of innovative designs for RF-components such as phase shifter & LNA components Satellites operating in this band are geostationary satellites in (M)MIC technology, or beam forming by using optical spaced 2o apart in the United States and 3o in Europe. In ring resonators. In the Dutch FlySmart project a national order to be able to receive these satellites also at high consortium (consisting of University of Twente, Lionix BV, latitudes (e.g. during inter-continental flights) the antenna National Aerospace Laboratory NLR and Cyner Substrates) system should have sufficient performance at low elevation is developing a broadband optical beam forming network angles. for a broadband phased array antenna to be mounted on the fuselage of an aircraft. For the steering of the beam of the Therefore the antenna system is required to have a small phased array a squint-free, continuously tunable mechanism beamwidth (to discriminate between the satellite signals) has been developed that is based on a fully integrated and a high gain (>30 dB) also at low elevation angles. Since optical beam forming network (OBFN) using cascades of the gain of the antenna is related to the effective aperture of optical ring resonators (ORRs) as tunable delay elements. the antenna in the direction of the satellites, a conformal Such an OBFN can be realized on a single-chip. The proof- antenna also covering side parts of the fuselage could be an of-concept has been shown by manufacturing a chip for an advantage. 8x1 OBFN. Essential components of the OBFN are the optical modulators, which are used for the RF signal to The phased array antenna shall maintain the proper (linear) optical signal conversion. The present paper describes the polarization during all attitudes and at all positions of the development of the prototype 8x1 OBFN and a breadboard aircraft (also at high latitudes). Ku-band antenna (consisting of 8x8 antenna elements on a multilayer planar structure with stacked microstrip patches, An antenna to be used on aircraft has to be able to operate feeds and slots on substrates). The broadband capabilities of in severe environmental conditions concerning temperature, the prototype 8x1 OBFN and the breadboard Ku-band pressure, vibration and humidity. The environmental antenna are presented in this paper. requirements for civil airborne equipment are given in RTCA DO-160 or EUROCAE ED-14 [5].

2. SYSTEM ASPECTS In general, the antenna system consists of a phased array antenna, electrical-to-optical conversion, optical beam

In the ITU Radio Regulations [2] portions of the Ku-band forming (and beam steering) and optical-to-electrical are allocated to aeronautical services: conversion (Figure 1).

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(TTD), and therefore result in a frequency-dependent beam angle and shape (beam squint). Switchable delay matrices are not continuously tunable: they show a trade-off between beam angle resolution and complexity. An alternative that offers both continuous tunability and TTD is based on chirped fibre gratings (CFGs) [14]–[16], but this technique

has the disadvantage of requiring bulky optical components and an (expensive) tunable laser. Figure 1 System design of Ku-band receive antenna with Optical Beam Forming Network (OBFN) In References [27], [28] a squint-free, continuously tunable The phased array antenna contains a two-dimensional array OBF mechanism for a phased array receiver system was of dual linear polarised broadband antenna elements. Each proposed that does not require a tunable laser. It is based on antenna is followed by a Low Noise Amplifier (LNA) and a fully integrated optical beam forming network (OBFN) down-converter (together a Low Noise Block converter, using cascades of optical ring resonators (ORRs) as tunable LNB). The Local Oscillator (LO) signals of the LNBs are delay elements. A dedicated system architecture has been synchronised to maintain an appropriate phase relation proposed that relaxes the requirements on optical between the OBFN channels. The Intermediate Frequency modulators and detectors, and on the OBFN itself. It has a (IF) signal from the LNB is subsequently fed to optical potential for full beam forming integration. modulators which perform the electrical-to-optical conversion. In the Optical Beamforming Network (OBFN) This section is organized as follows. In the next subsection, each individual signal is attenuated and delayed in order to the theoretical principles of ORR-based delay elements will shape and direct the antenna beam. The sum of all signals is be summarized. After that it will be explained how these converted back from the optical to the electrical domain. delay elements should be grouped into an OBFN. In the third subsection the complete system architecture around The tracking algorithm will use the aircrafts position and this OBFN will be described, with particular focus on how attitude to determine the appropriate polarization and the electro-optical and opto-electrical conversions should be azimuth and elevation angle for the antenna beam. performed. The section will end with a description of the fabrication technology for the optical chips. To reach the objective of a 2 GHz bandwidth, both the antenna front-end and the beamforming network should Ring Resonator-Based Delays have broadband characteristics. Therefore, the antenna A narrowband continuously tunable optical TTD device can front-end consists of an array of stacked patch antennas. be realized as a circular waveguide (also called optical ring The beamforming network consists of an optical network resonator, or ORR) coupled parallel to a straight waveguide with True Time Delays (TTD) which have inherently large (see references [17]-[21]). When propagation losses are bandwidth. To have a 2o beamwidth and high gain antenna neglected, such configuration can be considered as an all- (approx. 36 dB), a large array antenna is needed. The pass filter, with a periodic, bell-shaped group delay current design is based on an array of 40 by 40 antenna response, as illustrated by the dotted lines in Figure 2. The elements (1600 in total). In order to limit pointing loss, the period or free spectral range (FSR) is equal to the inverse of 2o beamwidth requires tracking accuracy of about 0.2o. the ORR's round-trip time T. Achieving this accuracy with open loop track techniques is challenging. A closed loop technique could provide the φ φ φ required accuracy and can be implemented into the array 1 2 3 design. in TTTout κ κ κ 1 2 3 group delay

3. DEVELOPMENT OF OPTICAL BEAMFORMER → 0 Implementation of the beam forming network in the optical f1 f2 f3 → f domain shares many common advantages with other RF photonic signal processing techniques [10],[11], such as Figure 2 – Theoretical group delay response of three compactness and light weight (particularly when integrated cascaded ORR sections. The dashed lines represent the on a chip), low loss, frequency independence, large group delay responses of the individual sections. (Inset: instantaneous bandwidth, and inherent immunity to cascade of three ORRs with round-trip delay T, electromagnetic interference. Most previously proposed additional round-trip phase-shifts φ and power coupling optical beamformer systems are either based on optical i coefficients κ .) phase shifters [12] or switchable delay matrices [13]. i However, phase shifters do not provide true time delay

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The maximum group delay occurs at the resonance stage 1 stage 2 stage 3 φ frequency, which can be varied by tuning the round-trip 7 out 1 κ 15 7 out 2 φ φ φ phase shift of the ORR, using thermo-optical tuning 5 6 κ κ 7 14 φ elements. Similarly, the maximum delay can be varied by 56 8 out 3 κ κ κ 8 tuning the coupling coefficient κ between waveguide and in 5 6 16 out 4 κ κ 8 ORR. The width of the bell shape is more or less inversely 13 φ φ φ φ φ 11 out 5 proportional to its height, resulting in a trade-off between 1 2 3 4 κ 18 11 out 6 12 34 φ φ peak delay and optical bandwidth. 9 10 κ κ κ κ κ κ 11 1 2 3 4 17 φ 9 10 12 out 7 κ κ κ 12 The bandwidth can be extended by cascading multiple ORR 9 10 19 out 8 κ sections, as shown in the inset of Figure 2. The resulting 12 × group delay response (the solid line in Figure 2) is equal to Figure 3 – Binary tree-based 1 8 optical beam the sum of the individual group delay responses (dashed forming network for a phased array transmitter lines), so the group delay curve can be flattened by tuning system, consisting of twelve ORRs and seven tunable splitters. the ORRs to different resonance frequencies. Such a multi- stage delay element has a trade-off between peak delay, optical bandwidth, relative delay ripple and number of ORR The first single-chip realization of an ORR-based OBFN, sections (see references [17],[20],[21]). based on a 1×4 binary tree topology, was presented in [25], and later extended to a 1×8 OBFN [26],[27]. Optical Beam Forming Network When the optical delay elements are combined with tunable Optical Beamformer System Architecture signal processing circuitry (power splitters or combiners), When an OBFN is applied in a phased array receiver an OBFN is formed. Integrating such an OBFN into one system, the individual antenna element signals first have to single optical chip has many advantages compared to be converted from the electrical to the optical domain. The connecting separate optical devices, such as compact size, optical signals are then re-aligned and combined by the light weight, low loss, and reduced costs. Moreover, OBFN, resulting in one output signal that has to be integration on chip facilitates coherent optical combining, converted back to the electrical domain. so that only one laser and one detector are required in a complete phased array receiver system, as we will see in the In order to minimize the loss, the combining of the optical next subsection. signals in the OBFN should preferably be done coherently, which requires the use of a common laser. The output light × Figure 3 shows an ORR-based 1 8 OBFN for a transmitter of the laser should first be split, and then be modulated by phased array, based on a binary tree topology. It consists of the antenna element signals, using external modulators. The three stages and has eight outputs. In this case twelve ORRs most straightforward way of doing so is to apply optical and seven tunable power splitters are involved. The OBFN double-sideband (DSB) modulation, for example using is arranged in such a way that an increasing number of Mach-Zehnder modulators (MZMs). The output signal of ORRs is cascaded for Outputs 1 to 8, to satisfy the delay the OBFN can then be converted to the electrical domain by requirement for beam forming in a linear phased array. direct optical detection, using a photodiode. This is Compared to the parallel topology, which has independent illustrated in Figure 4. cascades of ORRs for each output, the binary tree-based OBFN is more efficient with respect to the required number AE of rings, and therefore has a reduced tuning complexity. Moreover, the binary tree-based OBFN is easy to extend: LNA more outputs can be obtained by simply adding more stages. MZM

It() AE OBFN out LNA

MZM Figure 4 – Optical beamformer architecture using DSB modulation and direct detection (AE=antenna element, LNA=low-noise amplifier, MZM=Mach- Zehnder modulator, OBFN=optical beam forming network)

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A drawback of this approach is the large bandwidth of the modulated optical signal [28]. Its spectrum consists of the optical carrier and —depending on the modulation depth— at least two sidebands. With satellite TV operating in the 10.7–12.75 GHz band, it follows that the optical bandwidth is at least 25.5 GHz.

The most straightforward way of reducing the bandwidth of the DSB-modulated optical signals is to apply frequency down conversion (FDC) to an intermediate frequency (IF) range prior to electro-optical conversion. This can be done Figure 5 – Optical beamformer architecture using filter- by mixing the element signals with local oscillator (LO) based SSB-SC modulation and balanced coherent signals and low-pass-filtering. This has the additional detection (AE=antenna element, LNA=low-noise advantage that slower (and hence less expensive) optical amplifier, FDC=Frequency-Down-Conversion, modulators can be used. Notice that the local oscillators MZM=Mach-Zehnder modulator, OBFN=optical beam (LOs) in the down-converters need to have tuneable phase forming network, OSBF=optical sideband filter) differences (corresponding to the delays in the OBFN) in order to prevent IF phase offsets. When the antenna element Advantages of this optical modulation/detection scheme are: signals are down-converted to an IF range with a relative bandwidth of more than one octave, the spectrum will be • The optical bandwidth is significantly reduced, thereby distorted by second order intermodulation. This can be reducing the complexity of the OBFN (number of avoided by keeping the modulation index sufficiently small rings), and bringing the bending loss and tuning (see [28]). efficiency in each ring to an acceptable level; The second approach to reduce the bandwidth of the • The balanced detector cancels most of the laser’s modulated optical signal is to remove one of the sidebands, relative intensity noise (RIN), thereby significantly by applying optical single-sideband (SSB) modulation. increasing the dynamic range of the system [31]; Optical SSB modulation has been previously proposed as a means to overcome the bandwidth-limiting effect of • In case down conversion is performed prior to optical chromatic dispersion in single-mode fiber-based Radio- modulation, IF phase offsets can now be corrected by over-Fiber (RoF) transmission systems [14] and CFG-based simple optical phase shifts in the OBFN (which are optical beam forming systems [15]. The advantage of required anyway for coherent combining), and IMD-2 optical SSB modulation compared to optical DSB is canceled by the balanced detector [28]. modulation is namely that optical detection of SSB- modulated signals results in only one beating product at the desired RF frequency, whereas DSB-modulated signals give Optical SSB-SC modulation can be implemented in various two beating products at the desired RF frequency, which are ways [28], but the simplest way for this particular generally not in phase in case of chromatic dispersion, application is to use DSB modulators followed by optical resulting in RF power fading. sideband filters (OSBFs). DSB modulation can be performed by MZMs, which can be biased in such a way that the carrier is inherently suppressed (DSB-SC The main reason for using optical SSB modulation in our modulation) [32]. The OSBF is then only required to system, however, is to reduce the bandwidth of the suppress one of the sidebands. Since the OBFN and OSBF modulated optical signal: SSB modulation exactly halves are both linear devices, their order can be reversed, so that the bandwidth compared to optical DSB modulation. The only one common OSBF is required, as illustrated in Figure bandwidth can be even further reduced by also removing 5. the optical carrier, resulting in single sideband suppressed carrier (SSB-SC) modulation. The optical bandwidth then equals the RF bandwidth, which is the smallest that can be A logical design choice for the OSBF is to use a filter achieved without splitting the RF signals in sub-bands prior structure based on the same building blocks as the OBFN to electro-optical conversion. This is illustrated in Figure 5. (couplers and ORRs). The OSBF can then be realized in the same technology as the OBFN (see the next subsection) and, hence, be integrated on the same chip. We chose to use an unbalanced Mach-Zehnder interferometer with an ORR in the shortest branch, as shown in Figure 6. The circumference of this ORR is twice the path length difference of the MZI. The advantage of such a filter is that it has flattened pass bands and stop bands that can be made relatively wide, with steep transitions in between [29],[33].

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φ 1 T φ 2 κ 2T κ 1 2 OBFN OSBF κ 3 Figure 6 – Optical sideband filter (OSBF) consisting of a Mach-Zehnder interferometer (MZI) and an optical ring resonator (ORR) Figure 7 –Waveguide structure of the 1×8 OBFN chip and OSBF chip. (Top image: drawing of maskfile; Design and Realization of the Optical Chips Bottom image: photo of realized device with the The Optical Side-band Filter (OSBF) can be realized in the bondpads and electrodes clearly visible, the dimensions of same technology as the OBFN, which reduces the which is 6.85 cm long and 0.95 cm wide) fabrication complexity. Using the same technology for different components is a large advantage in VLSI The chips fabricated so far have waveguides of 2 µm wide, integration. An important parameter in the realization of the with a bending radius of 700 µm and directional coupler optical chips is the loss induced by the waveguide lengths of 150 µm. The heaters are 2.8 mm long, with a structures. A very promising technology that can meet the width of 20 µm and a thickness of 150 nm. They allow high requirements for the optical chip fabrication is the tuning of the resonance frequency and peak delay of each TriPleX technology of LioniX [22], [24], [34], [35]. This ORR, and tuning of the splitting ratios of the splitters, technology consists of a multi-layer stack of LPCVD silicon within 1 ms. For instance a 1×8 OBFN (as shown in Figure oxide and nitride. Since all composing materials are end 7) uses 31 heaters (two tuning elements for each ORR, and products during the deposition, the material properties are one for each splitter). Each heater consumes approximately very stable and reproducible and the device properties are 0.25 W, which brings the total power consumption of the therefore only determined by the geometry of the device. entire chip to approximately 8 W. Further research into The basic steps of the fabrication of the TriPleX lowering the amount of optical power is still in progress. waveguides have been presented in Ref. [30].

The actual chip design (for instance a 1×8 OBFN based on Measurements on Optical Beam Forming Network Chip couplers and ORRs) has been divided into a set of basic building blocks (BBB): The optical group delay at the outputs of the 1×8 OBFN chip were measured while varying the optical wavelength, - bent waveguides; over one FSR of 14 GHz. (This corresponds to waveguide group index of 1.8, and a ring circumference of 1.2 cm.) - Mach Zehnder Interferometers (MZIs) for tunable This was done by means of a network analyzer, using the coupling and splitting functions; phase shift method; details are given in [25]–[27]. The results for Output 2 to 8 are shown in Figure 8, and - tapered waveguides for coupling in and coupling demonstrate the delay generation of one single ring up to out. seven cascaded rings, respectively. The rings are tuned such that the group delay responses are flat in a common The BBBs were designed and their fabrication process was frequency band with a width of roughly 2.5 GHz, which is simulated with software of PhoeniX bv [36] after which more than enough to support the satellite TV band (10.7– they were realized in order to verify their behaviour before 12.75 GHz). The delay values of the respective output ports combining them into a complete optical chip. After are linearly increasing, corresponding to an eight-element selecting the proper BBBs from measurements on the linear phased array. The largest delay value is realized test-structures, the complete functional chip is approximately 1.2 ns (corresponding to 36 cm in air), and designed with these BBBs. Heaters are applied on the chip has a maximum ripple of approximately 0.1 ns (about one in order to thermally tune the optical properties of the ORR wavelength in Ku-band). (the resonance wavelength), the optical delay in the OBFN and the splitting ratio in the directional couplers. Figure 7 shows an image of the designed three stage cascaded 1×8 OBFN , followed by the OSBF (including a realized chip).

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1.4 From the graph we can conclude that the measured response Out 2 2.5 GHz Out 3 fits in the theoretical curve rather well. We believe that the Max. ripple 0.1 ns 1.2 Out 4 difference (especially the asymmetry) should mainly be Out 5 Out 6 attributed to the fact the circumference of the ring might not 1.0 Out 7 Out 8 be exactly twice the path length difference of the MZI. The 0.8 pass band ripple is below 1 dB and the stop band suppression is more than 25 dB, with pass band and stop 0.6 band both having a width of 15 GHz. Based on this OSBF a measurement to demonstrate the idea of sideband filtering Group Delay (ns) Delay Group 0.4 and carrier suppression for RF frequencies from 1 to 2 GHz are shown in Figure 10. For this measurement the optical 0.2 heterodyning technique is used before optical detection, to shift the spectrum of the modulated optical signal down into 0.0 the frequency range of the RF spectrum analyzer, by mixing 1549.97 1549.98 1549.99 1550.00 1550.01 1550.02 1550.03 the modulated light with CW light. The peak between two Wavelength (nm) sidebands in Figure 10 indicates the frequency difference Figure 8 – Measured group delay responses at between the two heterodyning optical carriers. It is shown different outputs of the 1×8 OBFN chip. that the magnitude of one sideband of the signal is 25 dB suppressed by the OSBF. When the OSBF is working properly, the ORRs of each signal channel of the OBFN can Note that 1.2 ns is more than enough for an eight-element be tuned such that a flat group delay response covers the linear phased array in the Ku band. These measurements frequency range of the remaining sideband of the modulated were actually done within the framework of a project in optical signals. which a different (lower) frequency range was considered, but clearly demonstrate that sufficient delay tuning range can be generated for satellite TV communication. Scaling to larger arrays is simply a matter of adding ports and -20 DSB signal 0 SSB-SC signal cascading more rings to achieve higher delays. OSBF response -30 -5 Measurements on Optical Sideband Filter Chip -40 -10 Figure 9 shows the measured cross port transmission of the OSBF chip (solid line). It was normalized to the maximum -50 -15 power transmission in order to enable comparison to theory (dotted line), without taking into account the relatively high -60 -20 OSBF Response (dB) fiber-chip coupling losses (10 dB). These should be RF signal Magnitude (dB) -70 -25 attributed to the fact that this chip has no tapered end faces, so we expect to significantly improve this in the future. -80 -30

45678910 Frequency (GHz) ~15 GHZ Figure 10–Measured spectrum of modulated optical signal, with and without side-band filtering.

Measurements on Optical Beamformer System > 25 dB To demonstrate the idea and feasibility of the broadband ~15 GHZ optical beamformer system. A setup with four RF input channels are built for simplicity. It uses a section of the complete OBFN chip and single-ended optical detection. A schematic of the setup is shown in Figure 11 .

Figure 9 –Normalized cross port power transmission of the OSBF chip. The solid line is the measured result, and the dotted line corresponds to the theoretical value.

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For the receive antennas the delay-synchronized antenna signals on the OBFN channels should be coherently combined in the OBFN to maximize the output signal. A setup consists of MZM-based intensity modulation, the OBFN, and direct detection is used for the measurement of coherent optical signal combination, as shown in Figure 13. One RF source is equally split into four RF Figure 11 – Measurement specification of RF phase channels for the OBFN. A delay setting of ORRs in the response of one beamformer channel. OBFN is made to compensate the signal path length differences between the four channels. The demonstration of signal recovery after optical SSB-SC modulation and OBFN is shown in Figure 12. A RF signal RF input 0-1 GHz over frequency range from 1 to 2 GHz is applied to one OBFN channel. Three group delay responses of the channel CH 1 are shown in the inset of Figure 12, with the maximum CH 2 value of 1.5 ns (45 cm delay distance in air). By means of Output coherent optical detection, signal recovery is performed after the combination between the delayed sideband and the CH 3 unmodulated optical carrier as shown in Figure 11. In CH 4 Figure 12 the recovered RF signals are shown in terms of RF-to-RF phase responses, after the processing of optical Figure 13 – Measurement specification of RF phase SSB-SC modulation, channel group delay and coherent response of four channels. optical detection. The phase response for 0 ns group delay is regarded as zero phase response, and the other two phase Figure 14 demonstrates the signal combination in the OBFN responses show good match to the corresponding delay through RF-to-RF measurement over 1 GHz signal values. Though not shown in the figure, the corresponding bandwidth. The RF magnitude differences between magnitude responses of the RF signal are flat over the signal individual channel outputs are due to the optical path loss band, but with larger loss for higher delay, because the differences, which can be removed by means of an optical loss increases with delay value [17], [21]. Besides, equalized setting of the optical couplers in the OBFN. The the ripples in the results are mainly due to the optical phase magnitude levels illustrate that the four channels are fluctuation at the optical carrier reinsertion, which comes coherently combined in the OBFN. The fluctuation in the from the slight fluctuation in the position and temperature signal band comes from the imperfection of the applied RF of the optical fibers before the chip. In the future connections and low-frequency response of the modulators. implementation this will not be a problem because the full beamformer will be integrated to a single chip including laser splitter and modulators. -20 ch 1 ch 2 0.0 ch 3 for 0 ns -30 ch 4 ch 1&2

) -0.5 ch 3&4 o ch All -40 2 for 0.75 ns -1.0 Signal sideband 1.5 ns -1.5 -50 for 1.5 ns 1

-2.0 0.75 ns (dB) magtitude signal RF Measured response -60 Desired response Signal Phase Shift ( x 360 ( x Shift Phase Signal -2.5 0 ns 0 -70 0.5 1.0 1.5 2.0 2.5 0.0 0.2 0.4 0.6 0.8 1.0 -3.0 Channel Group Delay Response (ns) Relative Frequency ( GHz ) Frequency (GHz) 1.0 1.2 1.4 1.6 1.8 2.0 Signal Frequency (GHz) Figure 14 –Measured output of RF power of beamformer with intensity modulation and direct Figure 12 – Measured RF phase response of one detection, for 1 channel, 2 combined channels, and 4 beamformer channel, for different delay values. combined channels.

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4. DEVELOPMENT OF KU-BAND ANTENNA The Ku-band antenna design has been optimised by using ANSOFT HFSS simulation software. The ANSOFT HFSS The objective is to develop a very low-profile aircraft model for the design is shown in Figure 16. The dimensions antenna, which can point to geostationary satellites of the patches, dog bone aperture and thicknesses of foam anywhere in the upper hemisphere. Traditionally, reflector- layers have been optimised with the aim to get an antenna based solutions have been proposed which are unattractive which could span the frequency-band from 10.7 to to airlines since they create significant drag and push up 12.75 GHz. fuel costs. Instead conformal phased array antennas of which the main beam can electronically be steered are recommended. In general an array antenna consists of a multiple of active antenna elements coupled to a common

source to produce a directive radiation pattern. The antenna Substrate element could be any type, but should have an omni- Foam replacement Patch directional radiation pattern. Since the aircraft antenna Patch should have a low profile most suitable antenna elements Substrate

are microstrip patches, which are fed by apertures in a Ground

ground plane. The main disadvantage of these microstrip Dogbone Substrate patch antennas is their limited bandwidth which is in the

Feed order of a few percent for a typical patch radiator. This Substrate property makes the classical patch antennas less attractive for broadband satellite communication. In order to increase Shielded vertical feedline the bandwidth, stacked patches have been advocated in Feedline routed on special feedline layer(s) literature. In the stacked patch configuration a parasitic element is placed above a lower patch, separated by foam or other space filler. In this manner, bandwidths on the order Figure 15 Design of stacked Ku-band antenna element of 30–35% can be achieved [6]. In this section we present the design of stacked patch Ku-band antenna element and we discuss the development of a planar phased array of Some results of the design and optimization process are stacked Ku-band antenna elements. shown in Figure 17, Figure 18 and Figure 19. The design has a ground plane at a distance of 5 mm from the planar Design of single Ku-band antenna element layer with feed traces. The function of this ground plane is to shield the antenna element from the layers below the A common approach for increasing the bandwidth is to add ground plane in order to minimise the influence on the parasitic elements to the antenna structure (e.g. a stacked element. From a manufacturing point of view, a maximum patch). This reduces the impedance variation of the antenna distance of 5 mm was recommended. Figure 17 and Figure with the frequency, thus enhancing bandwidth performance. 18 show the computed radiation patterns for co- and cross Various arrangements of stacked structures have been polarization (according to Ludwig’s third definition) for two investigated in [7] and [8]. In practice, it is difficult to sections in the hemisphere (at φ = 0 and φ = 90 degrees). optimise the bandwidth of these structures due to their sensitivity with respect to many physical parameters (patch The computed gain of this stacked patch antenna element is sizes, substrate thicknesses, and feed-point position). about 9dBi. Furthermore, it can be observed that the Research has focused on the choice of the materials for the radiation pattern of this stacked antenna element shows fair dielectric layers in the stacked configuration. Thick omni-directional behavior, which is required for good laminates of low-dielectric constant provide the largest performance in a large antenna array. Figure 19 displays the bandwidth and surface wave efficiency (see [8]). reflection coefficients of the antenna element. Notice that the return loss is below -10 dB in the frequency range between 10.7 to 12.75 GHz, as aimed, which indicates that Figure 15 shows the design of the present Ku-band antenna element consisting of a multilayer structure where the this element has sufficiently large bandwidth for broadband parasitic and radiating patches are mounted on data transmission. This Ku-band antenna element satisfies commercially available Duroid substrates. The space the preset requirements (see the inset in Figure 19). between the patches is filled with typical space filler that is being developed for this purpose. The lowest patch is being fed by an aperture in a lower ground plane, again mounted on a substrate. On the lower side of this substrate are horizontal feed lines, which are connected to shielded vertical feed lines to provide connections with the beam forming network on a lower layer.

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Figure 16 ANSOFT HFSS model for stacked Ku-band Figure 19 Reflection coefficients of stacked Ku-band patch antenna patch antenna with ground plane at 5 mm distance.

Design of Ku-band phased array

The Ku-band elements are part of the of the Ku-band phased array antenna. The preliminary design contains 24 square building blocks (see Figure 20). Each building block contains 8x8 Ku-band elements. In this 64 element array the distance between the centers of the patches is a half wavelength for the maximum frequency of 12.7 GHz; that is, the element centers are 11.8 mm apart from each other. Both height and width of the tile are 8 x 11.8 mm = 94.4 mm and hence the overall size of the tile is 94.4 mm by 94.4 mm.. The array shown in Figure 20 crudely

approximates a circularly shaped array with radius Figure 17 Radiation patterns for dual-polarized Ku- R = 410λ . The array is built with 24 square tiles band patch antenna (ϕ = 0o and ϕ = 90o ), with ground a containing a total of 1536 elements. plane, excited at port 1.

Figure 20 Conformal antenna array composed of 24 square 8x8 Ku-band building blocks crudely Figure 18 Radiation patterns for dual-polarized Ku - approximating an antenna with circular boundary band patch antenna (ϕ = 0o and ϕ = 90o ), with ground

plane, excited at port 2. The ANSOFT HFSS model for the building block with 8x8 antenna Ku-band antenna elements is shown in Figure 21. This model contains vertical transmission lines for each antenna element. Once again a lower ground plane was defined on 5 mm below the layer with feed traces. The radiation pattern of the 8x8 Ku-band antenna array has been calculated by means of a simple summation (without tapering) of the excited fields of single Ku band elements.

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The coupling between the elements was neglected. The represent the outer conductor in a transmission line. The computed radiation pattern of the 8x8 Ku-band array is dimensions of the vias were optimized using ANSOFT displayed in Figure 22 for co- and cross polarization HFSS and CST software. Several samples of this vertical (according to Ludwig’s third definition) for two sections in transmission line were manufactured. So far, most samples the hemisphere (at φ = 0 and φ = 90 degrees). All antenna showed some misalignment between vias and feeds. elements are excited at port 1. The computed gain of the Research is going on to improve the process for main lobe is about 27 dBi. manufacturing the vertical transmission lines. The focus is on the use of stable substrates which are less sensitive to temperature variations in the manufacturing process.

Figure 21 ANSOFT HFSS design of 8x8 Ku-band antenna array

National Aerospace Laboratory (NLR) Radiation Pattern 9 KuDual

0 Curve Inf o dB(RealizedGainL3X) -30 30 Setup6 : LastAdaptive Figure 23 ANSOFT HFSS model for vertical 20.00 Freq='11.7GHz' Phi='0deg' dB(RealizedGainL3X) Setup6 : LastAdaptive transmission line 10.00 Freq='11.7GHz' Phi='90deg' dB(RealizedGainL3Y) -60 0.00 60 Setup6 : LastAdaptive Freq='11.7GHz' Phi='0deg' dB(RealizedGainL3Y) -10.00 Setup6 : LastAdaptive Freq='11.7GHz' Phi='90deg' Development of breadboard Ku-band array antenna -20.00

-90 90 To start with, a single prototype Ku-band antenna element with dual polarization has been manufactured. The building components of the prototype antenna as well as the -120 120 assembled antenna are shown in Figure 24. Notice that two trace feeds are on the lowest substrate, which terminate on -150 150 -180 the edge. Two connectors were attached to verify the dual linear polarization properties of the antenna element. This Figure 22 Radiation pattern of 8x8 Ku-band antenna antenna element does not have a ground plane at 5 mm array distance.

Vertical transmission lines To achieve sufficient gain the final antenna must have about 1600 elements. Each element has two feed lines, one for each polarization. Every feed line has to be connected to the beam forming network. This means that the connections cannot be routed to one of the four sides of the antenna. Beneath the feed lines there must be a separation of at least 5mm to have an acceptable decrease in return loss. The larger this distance gets, the lesser the influence on the return loss. This 5 mm in air is much larger as the rule of thumb of lambda/100 for not needing a transmission line connection. There are several solutions possible to bridge this gap with a transmission line. Figure 24 Building components of dual-polarized Ku- band antenna element, and assembled prototype element The leading choice was the capabilities of the PCB with connectors. manufacturer. A solution with vertical vias was chosen. Figure 23 shows the HFSS model. The green cylinder The performances of the dual-polarized Ku-band antenna represents the centre conductor. The four outer cylinders element are displayed in Figure 25 and Figure 26. Figure 25

Authorized licensed use limited to: UNIVERSITEIT TWENTE. Downloaded on October 7, 2009 at 07:26 from IEEE Xplore. Restrictions apply. shows the measured and computed reflection coefficient. problems with the manufacturing of vertical transmission The computed reflection coefficient of this antenna element lines in suitable substrates, it was decided to replace these still satisfies the requirements. The measured data of the lines by a combiner feed network on the layer just below the prototype antenna shows a noticeable difference with the feeding slots, and to delete the lower ground plane at 5 mm computed design, which cannot be explained so far. distance. This feed network consists of 8 combiners, where Apparently the bandwidth of the assembled antenna is each combiner coherently sums 8 antenna elements (see smaller than the bandwidth of the computed design. At the Figure 27). The eight feed lines terminate at the edge of upper side of the band, between 12 and 12.7 GHz, the substrate, so that 8 connectors can be attached. In measured reflection coefficient is a bit too high. The combination with the prototype 8x1 OBFN, a Ku-band measured and computed isolation (S21 parameters) between phased array antenna is obtained of which the beam can be the two ports of the Ku-band antenna elements are steered in one direction. displayed in Figure 26. For the frequency band of interest a good correlations is observed between measurements and computation.

Return Loss Dual Polarised

-5

-10 Meas S11 Meas S22 Sim S11 Sim S22

Return loss (dB) -15

-20

-25 10 11 12 13 14 Figure 27 Feed network: 8 times 8x1 combiners Frequency (GHz) The 8x8 feed network is being considered as an Figure 25 Measured and computed reflection intermediate solution. In the future, it will be replaced by a coefficients of dual-polarized Ku-band prototype system of vertical transmission lines. Due to limited antenna element (no ground plane) available space for the 8x8 feed network on just one single layer, it was also decided to replace the dual-polarized

Isolation Dual Polarised elements by single polarized ones. The dimensions of the

0 dual-polarized stacked Ku-band antenna element were accepted also for the single polarised element. Also the -5 absence of the vertical transmission lines and the lower

-10 ground plane were accepted. No re-design was performed.

S21 Simulated -15 S21 Measured

Isolation (dB) Isolation

-20

-25

-30 10 11 12 13 14 Frequency (GHz) Figure 26 Measured and computed isolation of dual- polarized Ku-band prototype antenna element (no ground plane)

In spite of the increased reflection coefficient of the assembled single prototype, this Ku-band element has been taken as basic element for the manufacturing of the 8x8 Ku- band breadboard antenna. The antenna with eight Figure 28 Breadboard Ku-band antenna array mounted connectors is shown in Figure 28. on demonstrator box

The measured and computed radiation pattern of the 8x8 The breadboard Ku-band antenna array consists of 8x8 Ku-band breadboard antenna are displayed in Figure 29. A stacked Ku-band antenna elements (see Figure 16). Due to

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good comparison is observed for the beam width of this 7. CONCLUSIONS antenna. Radiation Pattern 8*8 Array @ 11.7 GHz (SinglePolElm) For broadband satellite communication on board an aircraft 0 the development of an advanced conformal antenna array -10 has been discussed. Such antennas require high gain with

-20 large bandwidth of which the beam can be steered continuously to the communication satellite. To reach the Measured -30 Simulated objective of a 2 GHz bandwidth at Ku-band, both the Normalized Gain (dB) -40 antenna front-end and the beam forming network should have broadband characteristics. Key subsystems which are -50 needed to achieve this objective are: Ku-band antenna

-60 elements with sufficiently large band with and an optical -40-30-20-100 10203040 Angle (deg) beamforming network with True Time Delays which have inherently a large bandwidth. The feasibility of these Figure 29 Measured and computed radiation pattern of subsystems has been presented. Ku-band breadboard antenna (normalized gain) Next, the gain of the main lobe has been measured in The Ku-band antenna is a stacked patch configuration relation to the frequency between 10.7 GHz and 12.7 GHz. where a parasitic element is placed above a lower patch, The measured gain has been normalized with respect to the separated by dedicated space filler. The first manufactured measured gain at 11.7 GHz. From Figure 30 it can be prototype antennas indicate that the bandwidth is observed that the variations in gain are less than 2 dB. sufficiently large. Furthermore, the required gain can be Hence, it can be concluded that this breadboard antenna has achieved by putting a sufficiently large number of these good broadband properties. antenna elements in an array. It has been shown that the Normalized Gain @ 11.7 GHz mutual coupling between these antenna elements is low.

10 8 For the steering of the beam of the conformal phased array 6 an advanced, squint-free, continuously tunable, bandwidth- 4

2 conserving, optical beamformer system has been proposed

0 which consists of a fully integrated broadband optical beam

Delta Gain (dB) -2 forming network using cascades of optical ring resonators -4 as tunable delay elements, a filter-based optical SSB-SC -6 modulation, and balanced coherent optical detection. A chip -8

-10 containing both OBFN and OSBF has been fabricated in 10.7 11.2 11.7 12.2 12.7 Frequency (GHz) TriPleX technology by LioniX BV. The delay measurements on the chip shows optical group delay Figure 30 Deviations in gain of Ku-band breadboard responses with maximal delay of 1.2 ns over a bandwidth antenna in frequency band 10.7-12.7 GHz of 2.5 GHz, Together with other measurements it is demonstrated that this system satisfies the required properties, so that it can be used for tracking satellites in a 5. DEVELOPMENT OF DEMONSTRATOR broadband Ku-terminal. The demonstrator antenna that will be built in the FlySmart project will have limited capabilities compared with a future CKNOWLEDGMENT airborne antenna. The main objective of the demonstrator is A to show the broadband characteristics of the antenna front- end and the optical beam former. In addition the beam This work was part of the Broadband Photonic Beamformer forming and beam steering algorithms will be verified. For project, the FlySmart project, the IO-BFN project, and the this purpose an 8 by 8 breadboard array antenna has been IO-BFNSYS project, all supported by the Dutch Ministry of developed. In one dimension the output of all antenna Economic Affairs, SenterNovem project numbers IS052081 elements has been coherently summed. In the other and ISO53030, NIVR project numbers PEP61424 and th dimension the 8 combiner outputs will be fed to an 8 PEP61629, respectively, and EU 6 Framework project channel OBFN. For demonstration purposes, this antenna ANASTASIA. The FlySmart project is part of the Eureka + will be installed on a vehicle. Since the beamwidth and PIDEA project SMART. antenna gain are not appropriate for reception of real satellite signals, a local satellite repeater will be used for the Robert Wijn, Rineke Groothengel, and Melis Jan Gilde of demonstration. LioniX BV are acknowledged for technical assistance during the fabrication of the optical devices.

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Hans van Gemeren (Cyner Substrates) is acknowledged for REFERENCES technical assistance during the fabrication of the prototype antennas. [1] P. Jorna, H. Schippers, J. Verpoorte, “Beam Synthesis for Conformal Array Antennas with Efficient Tapering”, Eduard Bos of the Telecommunication Engineering Group Proceedings of 5th European Workshop on Conformal is acknowledged for technical assistance during the Antennas, Bristol, September 11-12, 2007. characterization of the optical devices. [2] The Radio Regulations, edition of 2004, contain the complete texts of the Radio Regulations as adopted by the World Radio-communication Conference (Geneva, 1995) (WRC-95) and subsequently revised and adopted by the World Radio-communication Conference (Geneva, 1997) (WRC-97), the World Radio- communication Conference (Istanbul, 2000) (WRC- 2000), and the World Radio-communication Conference (Geneva, 2003) (WRC-03) including all Appendices, Resolutions, Recommendations and ITU-R Recommendations incorporated by reference.

[3] RECOMMENDATION ITU-R M.1643, Technical and operational requirements for aircraft earth stations of aeronautical mobile-satellite service including those using fixed satellite service network transponders in the band 14-14.5 GHz (Earth-to-space), 2003

[4] ETSI EN 302 186 v1.1.1 (2004-01); Satellite Earth Stations and Systems (SES); Harmonised European Norms for satellite mobile Aircraft Earth Stations (AESs) operating in the 11/12/14 GHz frequency bands covering essential requirements under article 3.2 of the R&TTE directive

[5] EUROCAE ED-14E; Environmental Conditions and Test procedures for Airborne Equipment, March 2005.

[6] F. Croq and D. M. Pozar, “Millimeter wave design of wide-band aperture-coupled stacked microstrip antennas,” IEEE Trans. Antennas Propagat., vol. 39, pp. 1770–1776, Dec. 1991.

[7] S. D. Targonski, R. B. Waterhouse, D. M. Pozar, "Design of wide-band aperture-stacked patch microstrip antennas ", IEEE Transactions on Antennas and Propagation, vol. 46, no. 9, Sep. 1998, pp. 1245-1251.

[8] R. B. Waterhouse, "Design of probe-fed stacked patches", IEEE Transactions on Antennas and Propagation, vol. 47, no. 12, Dec. 1999, pp. 1780-1784.

[9] http://www.ansoft.com

[10] R. A. Minasian, “Photonical signal processing of microwave signals,'' IEEE Trans. Microwave Theory Tech., vol. 54, no. 2, pp. 832–846, Feb. 2006.

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[11] A. J. Seeds, K. J. Williams, “Microwave photonics,'' J. [22] R. G. Heideman, A. Melloni, M. Hoekman, A. Lightwave Technol., vol. 24, no. 12, pp. 4628–4641, Borreman, A. Leinse, F. Morichetti, “Low loss, high Dec. 2006. contrast optical waveguides based on CMOS compatible LPCVD processing: technology and experimental [12] G. Grosskopf et al., “Photonic 60-GHz maximum results,” Proc. 9th IEEE/LEOS Symp. Benelux, Mons, directivity beam former for smart antennas in mobile Belgium, 1–2 Dec. 2005, pp. 71–74. broad-band communications,'' IEEE Photon. Technol. Lett., vol. 14, no. 8, pp. 1169–1171, Aug. 2002. [23] R. G. Heideman, D. H. Geuzebroek, J. A. Walker, “High contrast surface waveguide technology for [13] M. A. Piqueras et al., “Optically beamformed beam- biochemical sensing and telecom applications,” Proc. switched adaptive antennas for fixed and mobile broad- IEEE/LEOS Annual Meeting 2006, Montreal, Canada, band wireless access networks,” IEEE Trans. 29 Oct.–2 Nov. 2006, paper ThDD1. Microwave Theory Tech., vol. 54, no. 2, pp. 887–899, Feb. 2006. [24] R. G. Heideman, D. H. Geuzebroek, A. Leinse, A. Melloni, F. Morichetti, C. G. H. Roeloffzen, A. [14] J. L. Corral, J. Marti, J. M. Fuster, R. I. Laming, Meijerink, L. Zhuang, W. van Etten, E. Klein, A. “Dispersion-induced bandwidth limitation of variable Driessen, “Low loss, high contrast optical waveguides true time delay lines based on linearly chirped fibre based on CMOS compatible LPCVD processing,” Proc. gratings,” Electron. Lett., vol. 34, no. 2, pp. 209–211, 13th Europ. Conf. Integr. Opt. (ECIO'2007), Jan. 1998. Copenhagen, Denmark, 25–27 April 2007, paper WB0.

[15] B. Ortega, J. L. Cruz, J. Capmany, M. V. Andrès, D. [25] L. Zhuang, C. G. H. Roeloffzen, R. G. Heideman, A. Pastor, “Variable delay line for phased-array antenna Borreman, A. Meijerink, W. van Etten, “Single-chip based on a chirped fiber grating,” IEEE Trans. optical beam forming network in LPCVD waveguide Microwave Theory Tech., vol. 48, no. 8, pp. 1352– technology based on optical ring resonators,” Proc. of 1360, Aug. 2000. the International Topical Meeting on Microwave Photonics (MWP'2006), Grenoble, France, 3–6 Oct. [16] D. B. Hunter, M. E. Parker, J. L. Dexter, 2006, paper F1.4. “Demonstration of a continuously variable true-time delay beamformer using a multichannel chirped fiber [26] L. Zhuang, C. G. H. Roeloffzen, R. G. Heideman, A. grating,” IEEE Trans. Microwave Theory Tech., vol. 54, Borreman, A. Meijerink, W. van Etten, “Ring no. 2, pp. 861–867, Feb. 2006. resonator-based single-chip 1 × 8 optical beam forming network in LPCVD waveguide technology,” Proc. 11th [17] G. Lenz, B. J. Eggleton, C. K. Madsen, R. E. Slusher, IEEE/LEOS Symp. Benelux, Eindhoven, The “Optical delay lines based on optical filters,” IEEE J. Netherlands, 30 Nov.–1 Dec. 2006, pp. 45–48. Quantum Electron., vol. 37, no. 4, pp. 525–532, Apr. 2001. [27] L. Zhuang, C. G. H. Roeloffzen, R. G. Heideman, A. Borreman, A. Meijerink, W. van Etten, “Single-chip [18] J. E. Heebner, V. Wong, A. Schweinsberg, R. W. Boyd, ring resonator-based 1 × 8 optical beam forming D. J. Jackson, “Optical transmission characteristics of network in CMOS-compatible waveguide technology,” fiber ring resonators,” IEEE J. Quantum Electron., vol. IEEE Photon. Technol. Lett., vol. 15, no. 15, pp. 1130– 40, no. 6, pp. 726–730, June 2004. 1132, Aug. 2007.

[19] M. S. Rasras et al., “Integrated resonance-enhanced [28] A. Meijerink, C. G. H. Roeloffzen, L. Zhuang, D. A. I. variable optical delay lines,” IEEE Photon. Technol. Marpaung, R. G. Heideman, A. Borreman, W. van Lett., vol. 17, no. 4, pp. 834–836, Apr. 2005. Etten, “Phased array antenna steering using a ring resonator-based optical beam forming network,” Proc. [20] L. Zhuang, C. G. H. Roeloffzen, W. C. van Etten, 13th IEEE/CVT Symp. Benelux, Liège, Belgium, 23 “Continuously tunable optical delay line,” Proc. 12th Nov. 2006, pp. 7–12. IEEE/CVT Symp. Benelux, Enschede, The Netherlands, 3 Nov. 2005, paper P23. [29] L. Zhuang, A. Meijerink, C. G. H. Roeloffzen, D. A. I. Marpaung, J. Peña Hevilla, W. van Etten, R. G. [21] C. G. H. Roeloffzen, L. Zhuang, R. G. Heideman, A. Heideman, A. Leinse, M. Hoekman, “Phased array Borreman, W. van Etten, “Ring resonator-based tunable receive antenna steering using a ring resonator-based optical delay line in LPCVD waveguide technology,” optical beam forming network and filter-based optical Proc. 9th IEEE/LEOS Symp. Benelux, Mons, Belgium, SSB-SC modulation,” Proc. of the International Topical 1–2 Dec. 2005, pp. 79–82. Meeting on Microwave Photonics (MWP'2007), Victoria, BC Canada, 3–5 Oct. 2007, to be published.

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[30] H. Schippers, J. Verpoorte, P. Jorna, A. Hulzinga, A. Meijerink, C. G. H. Roeloffzen, L. Zhuang, D. A. I. Marpaung, W. van Etten, R. G. Heideman, A. Leinse, A. BIOGRAPHIES Borreman, M. Hoekman M. Wintels, “Broadband Conformal Phased array with Optical Beamforming for Harmen Schippers is senior scientist Airborne Satellite Communication”, Proc. of the IEEE at the National Aerospace Laboratory Aerospace Conference, March 2008, Big Sky, Montana, NLR. He received his Ph. D. degree in US. applied mathematics from the University of Technology Delft in [31] G. L. Abbas, V. W. S. Chan, T. K. Yee, “A dual- 1982. Since 1981 he has been detector optical heterodyne receiver for local oscillator employed at the National Aerospace noise suppression,” J. Lightwave Technol., vol. 3, no. 5, laboratory NLR. He has research pp. 1110–1122, Oct. 1985. experience in computational methods for aero-eleastics, aeroacoustic and electromagnetic [32] R. Montgomery, R. DeSalvo, “A novel technique for problems. His current research activities are development double sideband suprressed carrier modulation of optical of technology for integration of smart antennas in aircraft fields,” IEEE Photon. Technol. Lett., vol. 7, no. 4, pp. structures, and development of computational tools for 434–436, Apr. 1995. installed antenna analysis on aircraft and spacecraft.

[33] K. Oda, N. Takato, H. Toba, K. Nosu, “A Wide-Band Guided-Wave Periodic Multi/Demultiplexer with Ring Jaco Verpoorte has more than 10 Resonator for Optical FDM Transmission Systems,” J. years research experience on Lightwave Technol. vol. 6, no. 6. pp. 1016–1023, June antennas and propagation, 1988. Electromagnetic compatibility (EMC) and radar and satellite navigation. [34] R. G. Heideman, M. Hoekman, “Low modal He is head of the EMC-laboratory of birefringent waveguides and method of fabrication,” NLR. He is project manager on United States Patent US 7,146,087 B2, Dec. 5, 2006. several projects concerning EMC- analysis and development of [35] R. G. Heideman, M. Hoekman, “Surface waveguide advanced airborne antennas. and method of manufacture,” United States Patent US 7,142,759 B2, Nov. 28, 2006. Adriaan Hulzinga received his BEng [36] Phoenix BV, FlowDesigner, tool for 2D process degree in electronics from the visualization, http://www.phoenixbv.com/. hogeschool Windesheim in Zwolle. Since 1996 he has been employed at the [37] D. H. Geuzebroek, A. Driessen, “Ring-resonator-based National Aerospace laboratory (NLR) wavelength filters,” in Wavelength filters in fibre optics, as a senior application engineer. He is H. Venghaus, Ed. Berlin: Springer, 2006, pp. 341–379. involved in projects concerning antennas and Electromagnetic compatibility (EMC).

Pieter Jorna received the M.Sc. degree in applied mathematics from the University of Twente in 1999. From 1999 to 2005 he was with the Laboratory of Electromagnetic Research at the University of Technology Delft. In 2005 he received the Ph.D. degree for his research on numerical computation of electromagnetic fields in strongly inhomogeneous media. Since 2005 he is with the National Aerospace Laboratory (NLR) in the Netherlands as R&D engineer.

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working towards the PhD degree in the Telecommunication Arjan Meijerink was born in Almelo, Engineering Group at the University of Twente. His The Netherlands, in 1976. He research is related to the development of ring resonator- received the MSc and PhD degrees based optical beam forming networks for phased array (both with honours) in electrical antenna systems engineering from the University of Twente, Enschede, The Netherlands, in 2001 and 2005, respectively. From 2001 to 2005 he carried out David A. I. Marpaung was born in research on Coherence Multiplexing Balikpapan, Indonesia in 1979. He for Optical Communication Systems, received the BSc degree in physics in the Telecommunication Engineering Group at the (with honours) from Institut Teknologi University of Twente. He worked as a Postdoctoral Bandung, Indonesia, in February Researcher in that same group from 2005 to 2007, carrying 2002, and the MSc degree in applied out research on RF Photonic signal processing techniques, physics from the University of Twente, especially on the design and performance analysis of ring Enschede, The Netherlands, in resonator-based optical beamformers. Currently he is an December 2003. He is now working Assistant Professor in the Telecommunication Engineering towards the PhD degree in the Group. He teaches an undergraduate course on random Telecommunication Engineering Group at the University of signals and noise, and is involved in research on short- Twente. His research is directed towards the development range radio transmission techniques for wireless ad hoc of efficient modulation methods to increase the dynamic networks. range of analog optical links.

Chris G. H. Roeloffzen was born in Almelo, The Netherlands, in 1973. He received the MSc degree in applied Wim van Etten was born in physics and PhD degree in electrical Zevenbergen, The Netherlands, in engineering from the University of 1942. He received the MSc and PhD Twente, Enschede, The Netherlands, degrees in electrical engineering in 1998 and 2002, respectively. From from Eindhoven University of 1998 to 2002 he was engaged with Technology, Eindhoven, The research on integrated optical add- Netherlands, in 1969 and 1976, drop demultiplexers in Silicon respectively. From 1969 to 1970, he Oxinitride waveguide technology, in the Integrated Optical was with Philips Electronics, MicroSystems Group at the University of Twente. In 2002 developing circuits for he became an Assistant Professor in the Telecommunication oscilloscopes. In 1970, he became an Assistant Professor at Engineering Group at the University of Twente. He is now Eindhoven University of Technology, Faculty of Electrical involved with research and education on optical fiber Engineering. From 1970 to 1976, he was engaged in communications systems. His current research interests research on the transmission of digital signals via coaxial include optical communications and RF photonic signal and multiwire cables. Since 1976, he has been involved with processing techniques. research and education on optical fiber communications. In 1985, he was appointed Associate Professor at the Eindhoven University of Technology. In 1994, he became a Full Professor of Telecommunications at the University of Leimeng Zhuang was born in Twente, Enschede, The Netherlands. His current interests Beijing, China, in 1980. He received comprise optical communications, mobile communications, the BSc degree in detection, and simulation of communication systems. Telecommunication Engineering from the University of Electronic Science and Technology of China, Chengdu, China, in June 2003, and the MSc degree in electrical engineering (with honours) from the University of Twente, Enschede, The Netherlands, in June 2005. His master thesis was about the time delay properties of optical ring resonators. He is now

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René G. Heideman was born in Goor, The Netherlands, in 1965. He received the MSc and PhD degrees in applied physics from the University of Twente, Enschede, The Netherlands, in August 1988 and January 1993, respectively. After his PostDoc positions he applied his extensive know-how in the industry. Since 2001, he is co-founder and CTO of LioniX BV, Enschede, The Netherlands. He is an expert in the field of MST, in which he has more than 20 years of experience. He specializes in Integrated Optics, covering both (bio-)chemical sensing and telecom applications.

Arne Leinse was born in Enschede, the Netherlands, in 1977 and studied applied physics at the University of Twente where he received a M.Sc. degree at the integrated Optical Microsystems group in 2001. In this same group he started his PhD work on the topic of active microring resonators for various optical applications. His PhD work was carried out in the framework of a European project (IST 2000-28018 “Next generation Active Integrated optic Sub-systems”) and his thesis was titled: “Polymeric microring resonator based electro-optic modulator". In 2005 he joined LioniX BV where he is now involved as a project engineer in several integrated optical projects.

Marc Wintels was graduated in business administration. Then he fulfilled several commercial and financial jobs. With this background he became an entrepreneurial partner in a PCB manufacturing company, of which he became full owner several years later. From the beginning Cyner substrates had its focus on the production of prototyping and non-conventional Printed Circuit boards. Working mainly for design and research centers Cyner got involved in many high tech projects and from this developed a great expertise in the use of different (RF) materials. In the FlySmart project Marc and his colleagues are able to do what they like most: In close cooperation with designers, creatively working on substrate solutions.

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