Software-Defined Radio Beamforming System for 5G/Radar Applications

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Article Software-Deﬁned Radio Beamforming System for 5G/Radar Applications

Diogo Marinho 1,2,*, Raul Arruela 1,2, Tiago Varum 1 and João N. Matos 1,2

1 Instituto de Telecomunicações, Campus Universitário de Santiago, 3810-193 Aveiro, Portugal; [email protected] (R.A.); [email protected] (T.V.); [email protected] (J.N.M.) 2 Departamento de Eletrónica, Telecomunicações e Informática, Universidade de Aveiro, 3810-193 Aveiro, Portugal * Correspondence: [email protected]

Received: 2 September 2020; Accepted: 10 October 2020; Published: 15 October 2020

Abstract: Based on the ﬂexibility of software-deﬁned radio (SDR) techniques applied to an array of antennas, this article presents a beamforming architecture designed to operate in millimeter-wave bands (28 GHz), with possible applications in radar and 5G systems. The system structure, including its main constituents such as the radio frequency (RF) frontend modules, the radiating elements as well as the baseband processing on the host computer are widely described. Beamforming is achieved by digitally controlling the signals that feed the antennas. The experimental measurements performed in an anechoic chamber validate the proposed approach.

Keywords: digital beamforming; SDR; 5G; radar; beam shaping; phased array antenna

1. Introduction The world is gradually undergoing the age of communications, facing an enormous density of traffic and connections, where everyone is online, people and objects, interacting with each other, through a multitude of applications [1]. The modern generation of mobile communications (5G) is launched, which brings new challenges and opportunities, and will allow the creation and integration of new networks such as the internet of things (IoT) and vehicular networks, in an increasing global coverage. These systems impose a set of new requirements such as high speed data transfer, low latency, high network capacity and high connections density [2], which involve the operation in a frequency range little used so far, in the zone of millimetre waves. Radio detection and ranging (RADAR) technology, that was mainly used for military purposes, recently has suffered increasing attention, and its application is nowadays widespread in sports, biomedical, military or in transportation systems. Radars are used for surveillance, navigation, localization, traffic control or for weapons guidance. Currently, with the evolution of vehicular networks, smart cities and other intelligent systems, more advanced radar technologies are being developed. In the near future, the extension of the radar concept is expected for two purposes, reflectometry and communications. The scenario of Figure1 is foreseeable, a panoply of interconnected systems sharing information, and experiencing a huge communications density, with high risk of interferences.

Appl. Sci. 2020, 10, 7187; doi:10.3390/app10207187 www.mdpi.com/journal/applsci Appl. Sci. 2020, 10, 7187 2 of 14

Appl. Sci. 2020, 10, x FOR PEER REVIEW 2 of 14

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ThereThere havehave been been technological technological advances advances in this in field, this wherefield, low-costwhere mm-wavelow-cost mm-wave radars composed radars ofcomposed several antenna of several elements antenna integrated elements inintegrated single board in single and withboard high and resolution, with high resolution, have been presented.have been Inpresented. [5] is presented In [5] is an presented example ofan aexample low-cost of digital a low-cost beam digital steering beam receiver steerin (Rx)g phasedreceiver array (Rx) forphased IoT devicearray for connectivity, IoT device aconnectivity, cheap application a cheap to application get over the to communication get over the communication problems. In problems. [6], a frequency In [6], modulateda frequency continuousmodulated continuous wave (FMCW) wave radar (FMCW) is described. radar is described. This system This generates system generates multiple multiple digital beamsdigital withbeams high with gain, high low gain, side low lobe side level, lobe narrow level, beamnarrow width beam and width high and angular high angular resolution resolution control. Thecontrol. principles The principles of the digital of the beamforming digital beamforming applied to applied synthetic to aperturesynthetic radar aperture (SAR) radar systems (SAR) are systems shown inare [7 ],shown as well in as [7], the as improvement well as the inimprovement the resolution in achievedthe resolution regarding achieved conventional regarding radars. conventional radars.In [8 ,9], a wide variety of future applications are shown, as well as the progress that technology is facingIn in [8,9], reducing a wide the variety cost of of phased future array applications antennas. are Massive shown, antenna as well arraysas the usingprogress up tothat 128 technology elements, increasingis facing in performance reducing the and cost gain of phased of radar array communication antennas. Massive systems antenna are presented arrays using in [10 up], takingto 128 advantageelements, increasing of digital beamforming performance basedand gain on codedof radar aperture communication radar (CAR) systems technique are presented at 77 GHz. in [10], takingIn advantage [11–13], the of importancedigital beamforming of the beamforming based on coded spectral aperture efficiency radar (CAR) is demonstrated, technique at with 77 GHz. the cancelationIn [11–13], of undesired the importance interferences of the by beamforming placing nulls spectral in their directionsefficiency thatis demonstrated, can lead to a lowerwith the bit errorcancelation rate (BER). of undesired interferences by placing nulls in their directions that can lead to a lower bit errorIn rate [14 (BER).], there is a software-defined phased array radio operating at 28 GHz that uses software to controlIn [14], multiple there is a beam software-defined characteristics. phased However, array radio additional operating hardware at 28 GHz was that required, uses software a field to programmablecontrol multiple gate beam array characteristics. (FPGA) board forHoweve digitalr, control additional and an hardware Ettus B200 was mini required, software-defined a field radioprogrammable (SDR) for gate data array waveform (FPGA) board control. for digital In [15 control], a highly and an compact Ettus B200 28 GHzmini software-defined complementary metal–oxide–semiconductorradio (SDR) for data waveform (CMOS) control. integrated In [15], a circuit highly (IC) compact with multiple28 GHz complementary input multiple outputmetal– (MIMO)oxide–semiconductor and beamforming (CMOS) capabilities integrated with circuit a single (IC) with wire multiple for baseband input multiplexingmultiple output is reported. (MIMO) and beamforming capabilities with a single wire for baseband multiplexing is reported. However, it requires heavy digital signal processing (DSP) MIMO operations to retrieve the information. A Appl. Sci. 2020, 10, 7187 3 of 14

However, it requires heavy digital signal processing (DSP) MIMO operations to retrieve the information. A similar concept was recently shown in [16], proposing a digital array receiver for satellite applications, also integrating DSP algorithms, operating in a single-chip RF system on chip (SoC) solution installed on the Xilinx ZCU1275 prototyping platform; however, operating in a fixed frequency range between 27.5 and 28.35 GHz. A 64-channel massive MIMO transceiver with a fully digital beamforming (DBF) architecture is presented in [17]. The MIMO transceiver operates in time division duplex (TDD) mode, at 28 GHz with a 500 MHz signal bandwidth. A high data rate in the communication was achieved using the beam-tracking technique and two streams of 64-QAM (Quadrature Amplitude Modulation) signals. However, one of the great difficulties of this architecture is the high cost of the hardware structure that supports the 64 channels. Due to the high power consumption of the RF frontends, additional space in the circuit is required for heat dissipation. A Ka band DBF array transmitter using a direct digital synthesizer (DDS) for millimeter-wave applications is proposed in [18], operating at 24 GHz. This system reveals a complex calibration and has fluctuation over different scanning angles that unbalance the gain of different beam peaks in the 1 15 antenna array. In [19], the implementation × of an X band 4 4 digital phased array module is shown, which includes one custom CMOS RF SoC. × This system has high side lobes due to the transmitter (Tx) calibration process that involves random errors both in the phase and magnitude. The concept of beamforming, in the sense of the ability to control the radiation shape of an antenna array, can be performed either digitally, like the examples presented previously [14–19], analogically (by using the frontend analog phase shifters [20]) or in a hybrid way [21]. A millimeter-wave hybrid beamforming system architecture based on analog phased subarray is reported in [22]. This work uses a 32-element antenna array with a hybrid beamforming transceiver to operate at the 28 GHz band in TDD mode, with 500 MHz bandwidth. An important aspect is the reduction in the number of intermediate frequency (IF) channels compared to [14–19], reducing the cost comparing to fully digital architectures. However, one issue with this architecture is the analog phase shifting, which is critical to increasing the insertion loss up to 19.5 dB, requiring the use of additional amplifiers to compensate such losses. Additionally, this architecture has a higher complex design of the control signal because of the non-linearity between the control signals (amplitude weighting control) and output phase of this vector-sum phase. Authors in [23] explored the use of a Universal Software Radio Peripheral (USRP) X310 connected to a computer capable of calculating the direction of arrival (DoA) and generating beamforming in the sub-6 GHz frequency band for a satellite transponder structure. Nevertheless, this test bed presents considerable DoA estimation errors in a lower frequency of the operation scenario. Millimeter-wave RF systems are evolving rapidly, to some extent supported by software-defined radios. These systems provide a straightforward way to rapidly interact with hardware, and to test their use. Therefore, the combination of digital technology and versatile architectures, with adaptive antenna processing techniques, allows for powerful systems that can be used in communications and radar approaches. In this work, a versatile and reconfigurable 4 4 SDR digital beamforming phased array system × operating in the Ka band at 28 GHz is proposed. This system has the capability of electronically and digitally manipulating and steering the radiation pattern of an antenna array, and adjusting it to the environment, maximizing its performance and efficiency. Besides that, this system also has the versatility to, with low complexity, change the frequency of the local oscillator (LO) and the IF to implement communication systems and radar in the millimeter waves (Ka band), enabling the rapid testing of new solutions based on software-defined radio. It has also the advantage of carrying out full-duplex communication. This architecture has the potential to be used in a great variety of future scenarios such as in 5G communications, radar applications, IoT or surveillance. It is an extension work regarding the previous papers [24,25] with improvement in the digital processing unit and practical measurements validation. Appl. Sci. 2020, 10, 7187 4 of 14

This paper is organized into ﬁve sections, starting with the Introduction in Section1, stating the objectives and providing a brief state-of-the-art on SDR beamforming systems. Section2 presents both digital and analog components that constitute the proposed architecture of the software-deﬁned radio (SDR) phased array system. Section3 describes the beamforming processing system while in Section4 Appl. Sci. 2020, 10, x FOR PEER REVIEW 4 of 14 the experimental setup and the measurement results are presented. Finally, the last section reports the mainSection conclusions IV the experimental of the work. setup and the measurement results are presented. Finally, the last section reports the main conclusions of the work. 2. System Architecture 2. System Architecture Figure2 shows the proposed architecture of the 28 GHz SDR system, using a digital beamforming phased arrayFigure structure. 2 shows This the architectureproposed architecture has two complementary of the 28 GHz domains, SDR system, the digital using anda digital the analog, whichbeamforming are described phased below. array structure. This architecture has two complementary domains, the digital and the analog, which are described below.

FigureFigure 2. 2.Block Block diagram diagram of the propos proposeded SDR SDR beamforming beamforming system. system. 2.1. Digital Component 2.1. Digital Component TheThe digital digital component component of of the the proposed proposed system includes includes one one Ettus Ettus SDR SDR Universal Universal Software Software Radio Radio PeripheralPeripheral (USRP) (USRP) N310 N310 from from National National Instruments Instruments and and one one Host-PC Host-PC (Personal (Personal Computer). Computer).The TheUSRP is a networkUSRP is SDRa network equipment SDR equipment (uses gigabit (uses ethernet gigabit interface)ethernet interface) which has which four has independent four independent transmitting and receivingtransmitting channels, and receiving operating channels, at a frequencyoperating at that a frequency can be chosen that can from be chosen 10 MHz from up 10 to MHz 6 GHz, up withto up to 1006 GHz MHz, with of bandwidth. up to 100 MHz This of bandwidth. equipment This is responsible equipment foris responsible digitally generatingfor digitally anygenerating required any signal for therequired desired signal application for the desired and modulating application itand with modulating a RF microwave it with a carrier.RF microwave The baseband carrier. The signal is digitallybaseband decomposed signal is intodigitally two decomposed orthogonal componentsinto two orthogonal that feed components a phaseand that quadraturefeed a phase modulator. and quadrature modulator. The block diagram of the signal modulation performed inside the USRP is The block diagram of the signal modulation performed inside the USRP is shown in Figure3. At the shown in Figure 3. At the USRP output, the resulting modulated intermediate frequency (IF) signal USRPis output,chosen to the have resulting a 3.84 GHz modulated carrier frequency intermediate (fc). An frequency equivalent (IF) reverse signal process is chosen is applied to have to athe 3.84 IF GHz carrierreception frequency signals. (fc). The An equivalentreceived IF reversesignal is processdemodulated is applied in phase to the and IF quadrature, reception signals. generating The I/Q received IF signalsamples is demodulated that are later insent phase out andover quadrature,a 10 GbE connection generating to Ithe/Q samplesHOST-PC that forare processing. later sent This out over a 10equipment GbE connection is controlled to the by HOST-PC the HOST-PC, forprocessing. where a Python This script equipment based on isthe controlled USRP hardware by the drivers HOST-PC, where(UHD) a Python library script was developed. based onthe USRP hardware drivers (UHD) library was developed. The N310 architecture is divided into two segments, as shown in Figure3: a pair of daughterboards, each based on the Analog Devices AD9371 transceiver, and one motherboard composed by one Arm Cortex A9 and one FPGA (Kintex-7). In this work, all the core digital signal processing is performed in the USRP from the selection of frequency of the local oscillator, the sample rates, decimation/interpolation and the selection of appropriate ﬁlter bandwidth. The remaining processing of the In-phase and Quadrature (IQ) samples is performed on the PC. Regarding the communication between the HOST-PC and the USRP, the UHD library used in the PC to control USRP parameters is illustrated in Figure4, which provides a set of important functions for the communication. Figure4 focuses with high detail on the block diagram of the software structure developed on the PC to calibrate the system and manage the beamforming. A Python program adapted to the USRP N310 was created, to control the generated and received waveform signals. In this script, eight threads were created, each one Figure 3. Block diagram of software interface between Host-PC and Universal Software Radio Peripheral (USRP) N310. Appl. Sci. 2020, 10, x FOR PEER REVIEW 4 of 14

Section IV the experimental setup and the measurement results are presented. Finally, the last section reports the main conclusions of the work.

2. System Architecture Figure 2 shows the proposed architecture of the 28 GHz SDR system, using a digital beamforming phased array structure. This architecture has two complementary domains, the digital and the analog, which are described below.

Figure 2. Block diagram of the proposed SDR beamforming system.

2.1. Digital Component The digital component of the proposed system includes one Ettus SDR Universal Software Radio Peripheral (USRP) N310 from National Instruments and one Host-PC (Personal Computer). The USRP is a network SDR equipment (uses gigabit ethernet interface) which has four independent transmitting and receiving channels, operating at a frequency that can be chosen from 10 MHz up to 6 GHz, with up to 100 MHz of bandwidth. This equipment is responsible for digitally generating any required signal for the desired application and modulating it with a RF microwave carrier. The Appl.baseband Sci. 2020 signal, 10, 7187 is digitally decomposed into two orthogonal components that feed a phase5 ofand 14 quadrature modulator. The block diagram of the signal modulation performed inside the USRP is independentlyshown in Figure controlling 3. At the USRP each channel output, ofthe the resulting USRP. The modulated process startsintermediate by requesting frequency samples (IF) signal from theis chosen USRP, to encoding have a 3.84 them GHz with carrier the channel frequency number (fc). An and equivalent sending them reverse over process the Transmission is applied to Control the IF reception signals. The received IF signal is demodulated in phase and quadrature, generating I/Q ProtocolAppl./Internet Sci. 2020, 10 Protocol, x FOR PEER (TCP REVIEW/IP)socket. In this case, the Matlab program is the client, which5 of for 14 each channelsamplesdecodes that are the later samples sent inout an over IQ vector. a 10 GbE A calibration connection routine to the was HOST-PC also developed, for processing. calculating This the phaseequipment andThe amplitude is controlledN310 architecture diﬀ byerences the HOST-PC, is existing divided amongwhere into twoa the Python RFsegments, frontend script as based channels.shown on thein FigureUSRP hardware3: a pair driversof (UHD)daughterboards, library was developed. each based on the Analog Devices AD9371 transceiver, and one motherboard composed by one Arm Cortex A9 and one FPGA (Kintex-7). In this work, all the core digital signal processing is performed in them USRP fro the selection of frequency of the local oscillator, the sample rates, decimation/interpolation and the selection of appropriate filter bandwidth. The remaining processing of the In-phase and Quadrature (IQ) samples is performed on the PC. Regarding the communication between the HOST-PC and the USRP, the UHD library used in the PC to control USRP parameters is illustrated in Figure 4, which provides a set of important functions for the communication. Figure 4 focuses with high detail on the block diagram of the software structure developed on the PC to calibrate the system and manage the beamforming. A Python program adapted to the USRP N310 was created, to control the generated and received waveform signals. In this script, eight threads were created, each one independently controlling each channel of the USRP. The process starts by requesting samples from the USRP, encoding them with the channel number and sending them over the Transmission Control Protocol/Internet Protocol (TCP/IP) socket. In this case, the Matlab program is the client, which for each channel decodes the samples in an IQ vector. FigureA calibration 3.3. BlockBlock routine diagram diagram was ofalso software developed, interface calculating between the phase Host-PC and and amplitude Universal differences SoftwareSoftware existing RadioRadio Peripheralamong the (USRP)(USRP) RF frontend N310.N310 .channels.

FigureFigure 4. Flow 4. Flow control control between between the the receiver receiver and and transmittertransmitter during during Python Python and and Matlab Matlab interfaces. interfaces.

2.2. Analog2.2. Analog Component Component The analogThe analog component component of theof the proposed proposed system system is a transceiver transceiver composed composed of frequency of frequency converter converter modules,modules, between between IF andIF and RF, RF, and and also also by by the the antenna antenna arrays that that transmit transmit and and collect collect the theinvolved involved signals.signals. The The operating operating frequency frequency waswas selected in in the the Ka Ka band, band, at 28 at GHz, 28 GHz,and the and chosen the chosenIF was 3.84 IF was GHz. 3.84 GHz. Therefore, the RF upconverter module translates the modulated 3.84 GHz IF signal to the output Therefore, the RF upconverter module translates the modulated 3.84 GHz IF signal to the output operating frequency of 28 GHz. The block diagram of the upconverter is presented in Figure 5a, and operatingits measurements frequency of are 28 GHz.shown The in Figure block diagram6. The RF of downconverter the upconverter transforms is presented the received in Figure 285 a,GHz and its measurementssignals down are to shown 3.84 GHz in Figure IF, to be6. processed The RF downconverter in the USRP. The transforms block diagram the receivedof the downconverter 28 GHz signals downcan to be 3.84 seen GHz in Figure IF, to be7a, processedand its measured in the results USRP. are The provided block diagramin Figure of8. the downconverter can be seen in Figure7a, and its measured results are provided in Figure8. 2.2.1. RF Frontend Modules 2.2.1. RF FrontendUSRP channels Modules can operate with frequencies up to 6 GHz (max), so to allow the system to operate USRPat 28 GHz, channels frequency can operate conversion with units frequencies are requir uped toto 6be GHz attached (max), to sotheto USRP allow channels, the system connected to operate at 28 GHz,to the antennas. frequency Therefore, conversion twounits RF frontend are required units we tore be developed, attached allowing to the USRP this conversion channels, from connected 3.84 to to 28 GHz and vice versa. Appl. Sci. 2020, 10, 7187 6 of 14 the antennas. Therefore, two RF frontend units were developed, allowing this conversion from 3.84 to 28 GHz and vice versa. The chosen topology uses the heterodyne configuration. The upconverter is composed, in each channel, by an active mixer, a bandpass filter and an amplifier. To distribute the signal from the local oscillator (LO) to each channel mixer, discrete power splitters were also used. The key element for the up/down conversion is the mixer. The chosen mixer was the HMC264LC3B from Analog Devices. It is a subharmonic mixer that operates from 21 to 31 GHz at the RF port, using a signal from Direct Current (DC) to 6 GHz at the IF port, and from 10.5 to 15.5 GHz at the LO port. The used amplifier is the MAAL-011129 from MACOM, which is a low noise broadband amplifier (18 to 31.5 GHz) with a gain of 18 dB @ 28 GHz [19]. The local oscillator signal is generated by the phased locked loop (PLL) LMX2595 evaluation board which operates over a wideband (from 10 MHz upAppl. to 19 Sci. GHz) 2020, 10 with, x FOR low PEER noise REVIEW characteristics. The power splitter that divides the LO into two channels6 of 14 is the Mini-Circuits EP2k1+, which works from 1.8 to 28 GHz. The selected LO was 12.08 GHz. Finally, The chosen topology uses the heterodyne configuration. The upconverter is composed, in each a fifth order parallel coupled microstrip band pass filter was designed with 5 GHz of bandwidth, channel, by an active mixer, a bandpass filter and an amplifier. To distribute the signal from the local an attenuation of approximately 40 dB at the image frequency (20 GHz) and 0.9 dB of insertion loss at oscillator (LO) to each channel mixer, discrete power splitters were also used. 28 GHz. The key element for the up/down conversion is the mixer. The chosen mixer was the The upconverter module comprises four IF inputs, four RF outputs and two LO inputs. It was HMC264LC3B from Analog Devices. It is a subharmonic mixer that operates from 21 to 31 GHz at designed for a maximum power of about 10 dBm at the RF outputs, and for a maximum IF input power the RF port, using a signal from Direct Current (DC) to 6 GHz at the IF port, and from 10.5 to 15.5 of 7 dBm, resulting in an expected conversion gain of 3 dB. GHz at the LO port.

(a) (b)

FigureFigure 5. 5.Upconverter Upconverter designed designed module: module: (a ()a block) block diagram, diagram, (b ()b fabricated) fabricated prototype prototype board. board.

TheThe purpose used amplifier of the designed is the filterMAAL-011129 is to mitigate from the MACOM, image frequency which signalis a low and noise other broadband spurious signalsamplifier generated (18 to by31.5 the GHz) mixer. with It isa gain assumed of 18 that dB @ the 28 IF GHz signals [19]. at The the local inputs osci arellator bandpass signal at is 3.84 generated GHz withby the phased proper filteringlocked loop done (PLL) in the LMX2595 USRP. A evaluati photographon board of the which fabricated operates prototype over a wideband can be seen (from in Figure10 MHz5b). up to 19 GHz) with low noise characteristics. The power splitter that divides the LO into two channelsThe upconverter is the Mini-Circuits was tested EP2k1+, using which a 12 GHz works signal from in 1.8 the to LO 28 portGHz. and The withselected an IFLO of was 4 GHz. 12.08 FigureGHz.6 aFinally, shows thea fifth output order spectrum parallel atcoupled one of themicrostrip four RF band outputs pass of filter the upconverter. was designed It iswith possible 5 GHz to of find out the image signal (24 4 GHz = 20 GHz), the LO signal (2 12 GHz) and the desired RF signal bandwidth, an attenuation− of approximately 40 dB at the image× frequency (20 GHz) and 0.9 dB of (24insertion+ 4 GHz loss= 28 at GHz). 28 GHz. The behaviour of the filter can also be observed, since both the image and LO signalsThe are highlyupconverter attenuated module compared comprises to thefour RF IF signal, inputs 28.8, four and RF 26.4 outputs dBm, and respectively. two LO inputs. It was designedFigure 6forb presents a maximum the measured power of conversion about 10 dBm gain at of the the RF upconverter outputs, and at di forfferent a maximum RF frequencies. IF input It should be noted that the RF variation was made through the LO. The measured gain was 1 dB at power of 7 dBm, resulting in an expected conversion gain of 3 dB. − 28 GHz,The slightly purpose below of the the designed expected filter 3 dB. is Regarding to mitigate the the 1 dB image compression frequency point signal (P1dB) and atother the output,spurious 8.6signals dBm wasgenerated obtained, by the a value mixer. relatively It is assumed close tothat the the expected IF signals (10 at dBm, the asinputs can beare seen bandpass on the at link 3.84 powerGHz budgetwith the in proper Figure 5filteringa). done in the USRP. A photograph of the fabricated prototype can be seen in Figure 5b). The upconverter was tested using a 12 GHz signal in the LO port and with an IF of 4 GHz. Figure 6a shows the output spectrum at one of the four RF outputs of the upconverter. It is possible to find out the image signal (24 − 4 GHz = 20 GHz), the LO signal (2 × 12 GHz) and the desired RF signal (24 + 4 GHz = 28 GHz). The behaviour of the filter can also be observed, since both the image and LO signals are highly attenuated compared to the RF signal, 28.8 and 26.4 dBm, respectively. Figure 6b presents the measured conversion gain of the upconverter at different RF frequencies. It should be noted that the RF variation was made through the LO. The measured gain was −1 dB at 28 GHz, slightly below the expected 3 dB. Regarding the 1 dB compression point (P1dB) at the output, 8.6 dBm was obtained, a value relatively close to the expected (10 dBm, as can be seen on the link power budget in Figure 5a). Appl. Sci. 2020, 10, 7187 7 of 14 Appl.Appl. Sci.Sci. 20202020,, 1010,, xx FORFOR PEERPEER REVIEWREVIEW 77 ofof 1414

2828 GHzGHz

2020 GHzGHz 2424 GHzGHz

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FigureFigureFigure 6. 6.Measurements6. MeasurementsMeasurements of ofof upconverter upconverterupconverter module: module:module: (a )((aa RF)) RFRF output outputoutput spectrum spectrumspectrum (M1@20 (M1@20(M1@20 GHz, GHz,GHz, M2@24 M2@24M2@24 GHz, GHz,GHz, M3@28M3@28M3@28 GHz; GHz;GHz; (b ()(bb upconverter)) upconverterupconverter conversion conversionconversion gain gaingain at at diat ﬀdifferentdifferenterent RF RFRF frequencies. frequencies.frequencies. The downconverter module was designed similarly as the upconverter. The purpose of the ﬁlter TheThe downconverterdownconverter modulemodule waswas designeddesigned similarlysimilarly asas thethe upconverter.upconverter. TheThe purposepurpose ofof thethe filterfilter is to reject any image signal that can be captured at the RF inputs. In this structure, the mixer is the isis toto rejectreject anyany imageimage signalsignal thatthat cancan bebe capturedcaptured atat thethe RFRF inputs.inputs. InIn thisthis structure,structure, thethe mixermixer isis thethe element that imposes the maximum power on the RF inputs since it has a P1dB compression point elementelement thatthat imposesimposes thethe maximummaximum powerpower onon thethe RFRF inputsinputs sincesince itit hashas aa P1dBP1dB compressioncompression pointpoint ofof of 3 dBm. Thus, the maximum RF input power will be 13 dBm and the maximum power at the IF 33 dBm.dBm. Thus,Thus, thethe maximummaximum RFRF inputinput powerpower willwill bebe− −−1313 dBmdBm andand thethe maximummaximum powerpower atat thethe IFIF output will be 10 dBm, resulting in an expected conversion gain of 3 dB. The photograph of the outputoutput willwill bebe− −−1010 dBm,dBm, resultingresulting inin anan expectedexpected conversionconversion gaingain ofof 33 dB.dB. TheThe photographphotograph ofof thethe fabricated prototype is shown in Figure7b. fabricatedfabricated prototypeprototype isis shownshown inin FigureFigure 7b.7b.

((aa)) ((bb))

FigureFigureFigure 7. 7.Downconverter7. DownconverterDownconverter designed designeddesigned module: module:module: (a ()(aa block)) blockblock diagram; diagram;diagram; (b ()(bb fabricated)) fabricatedfabricated prototype prototypeprototype board. board.board.

InInIn Figure FigureFigure8a 8a8a is isis shown shownshown the thethe output outputoutput spectrum spectrumspectrum atatat oneone IF IF portport ofof of thethe the downconverterdownconverter downconverter module.module. module. TheThe Themeasurementmeasurement measurement testtest test waswas was performedperformed performed usingusing using anan an LOLO LO ofof 12 of12 12GHzGHz GHz andand and RFRF RF signalsignal signal ofof 28 of28 28GHz.GHz. GHz. MarkerMarker Marker M1M1 M1 isis atat isthe atthe the desireddesired desired IFIF frequency, IFfrequency, frequency, 44 GHzGHz 4 GHz (28(28 (28 −− 2424 24GHz)GHz) GHz) andand and M3M3 M3 isis atat is the atthe the LOLO LO frequencyfrequency frequency signalsignal signal (12(12(12 GHz).GHz). GHz). TheThe − Thedownconverterdownconverter downconverter accomplishedaccomplished accomplished aa aconversionconversion conversion gaingain gain ofof of 44 4 dB,dB, too too closeclose tototo thethethe expected expectedexpected 3 33 dB. dB.dB. As AsAs for forfor the thethe P1dBP1dBP1dB at atat the thethe RF RFRF input, input,input, 7 −− dBm77 dBmdBm was waswas measured, measured,measured, a valueaa valuevalue well wellwell above aboveabove the thethe expected expectedexpected13 −−1313 dBm. dBm.dBm. The TheThe third thirdthird − − orderorderorder intersection intersectionintersection point pointpoint is is 4.4is 4.44.4 dBm, dBm,dBm, and andand the thethe intermodulation intermodulationintermodulation ratio ratioratio is is39.2is 39.239.2 dB. dB.dB. AsAsAs in inin the thethe upconverter, upconverter,upconverter, the thethe results resultsresults suggest suggestsuggest a shiftaa shiftshift in inin the thethe central centralcentral frequency frequencyfrequency of ofof the thethe mixer, mixer,mixer, since sincesince the thethe resultsresultsresults at atat 25 2525 GHz GHzGHz are areare better betterbetter than thanthan at atat 29 2929 GHz. GHz.GHz. In InIn any anyany case, case,case, at atat the thethe frequency frequencyfrequency of ofof interest, interest,interest, the thethe results resultsresults are areare quitequitequite satisfactory satisfactorysatisfactory and andand close closeclose to toto what whatwhat was waswas intended. intended.intended. Appl. Sci. 2020, 10, x FOR PEER REVIEW 8 of 14 Appl.Appl. Sci. 20202020,, 1010,, 7187x FOR PEER REVIEW 88 of 1414

4 GHz 4 GHz 12 GHz 8 GHz 12 GHz 8 GHz

(a) (b) (a) (b) Figure 8. Measurements of downconverter module: (a) RF output spectrum (M1@4 GHz, M3@12 GHz; Figure(Figureb) down 8. converterMeasurements conversion of downconverter gain at differe module:module:nt RF frequencies. (a)) RF output spectrum (M1@4 GHz, M3@12 GHz; ((bb)) downconverter downconverter conversionconversion gaingain atat didiffereﬀerentnt RFRF frequencies.frequencies. 2.2.2. Planar Antenna Array 4 × 4 2.2.2. Planar Antenna Array 4 4 2.2.2. Planar Antenna Array 4 × 4 The antenna used at the interface between the free space and the RF frontend modules operates The antenna used at the interface between the free space and the RF frontend modules operates at 28 TheGHz antenna and is a used 4 × 4 atseries the interfacefeed planar between antenna the array free spacedescribed and inthe [26] RF andfrontend shown modules in Figure operates 9. The at 28 GHz and is a 4 4 series feed planar antenna array described in [26] and shown in Figure9. antennaat 28 GHz consists and is ofa 4 N × × =4 4series individual feed planar series antenna feed linea arrayr arrays, described each using in [26] four and microstrip shown in Figurepatches. 9. TheThe The antenna consists of N = 4 individual series feed linear arrays, each using four microstrip patches. arrayantenna can consists provide of controlN = 4 individual of the radiation series feed pattern linea r inarrays, the horizontal each using plane.four microstrip In [26] is patches. shown Thethe The array can provide control of the radiation pattern in the horizontal plane. In [26] is shown the simulatedarray can radiationprovide controlpattern ofon thethe horizontalradiation pattern axis. It isin possible the horizontal to observe plane. that Inthe [26] radiation is shown pattern the simulated radiation pattern on the horizontal axis. It is possible to observe that the radiation pattern cansimulated be steered radiation in different pattern directions on the horizontal by applying axis. Ita isprogressive possible to phase observe shift that in the the radiation feeding ofpattern each can be steered in diﬀerent directions by applying a progressive phase shift in the feeding of each subarray.can be steered The gainin different decreases directions progressively by applying when a theprogressive direction phase of radiation shift in themoves feeding out ofof eachthe subarray. The gain decreases progressively when the direction of radiation moves out of the boresight. boresight.subarray. The gain decreases progressively when the direction of radiation moves out of the boresight.

Figure 9. Planar antenna array 4 4. Figure 9. Planar antenna array 4 × 4. Figure 9. Planar antenna array 4 × 4. 3. Beamforming Processing 3. Beamforming Processing 3. BeamformingIn the previous Processing section, the architecture of the SDR system was presented, as well as the description In the previous section, the architecture of the SDR system was presented, as well as the of each analog and digital component used. In this section, taking advantage of the absolute control descriptionIn the ofprevio eachus analog section, and the digital architecture component of the used. SDR In system this section, was presented, taking advantage as well ofas thethe over each of the four Tx/Rx channels and the processing capacity that this system has, an example absolutedescription control of each over analog each of and the digitalfour Tx/Rx component channe lsused. and Inthe this processing section, capacitytaking advantage that this system of the demonstrating some of its potentialities is presented. In this work, some examples will be presented has,absolute an example control demonstratingover each of the some four of Tx/Rx its potentia channelitiesls and is presented.the processing In this capacity work, thatsome this examples system using two possibilities, beam shaping and beam steering, to highlight the functionality of the proposed willhas, anbe examplepresented demonstrating using two possibilities,some of its potentia beam shapinglities is presented. and beam In steering, this work, to some highlight examples the system. Nevertheless, a multitude of techniques may still be explored in the future. Figure 10 shows functionalitywill be presented of the proposedusing two system. possibilities, Nevertheless, beam ashaping multitude and of beamtechniques steering, may tostill highlight be explored the a block diagram of a beamforming system, applied to a linear array with N number of isotropic infunctionality the future. ofFigure the proposed 10 shows system. a block Nevertheless, diagram of a abeamforming multitude of system,techniques applied may tostill a linearbe explored array elements, in reception. In this structure, each antenna element has an adjustment, by multiplying within the N future. number Figure of isotropic 10 shows elements, a block indiagram reception. of a Inbeamforming this structure, system, each appliedantenna toelement a linear has array an each received signal by a weight factor, W = a ejα, where a is the amplitude excitation, and α is with N number of isotropic elements, in reception.n n In this structure,n each antenna element has an adjustment, by multiplying each received signal by a weight factor, 𝑊 =𝑎𝑒 , where 𝑎 is the the relative phase excitation to the previous element. By varying the weight factor of each channel, amplitudeadjustment, excitation, by multiplying and α iseach the rerelativeceived phasesignal excitation by a weight to the factor, previous 𝑊 =𝑎 element.𝑒 , where By varying 𝑎 is thethe andamplitude properly excitation, estimating and its α values, is the relative it is possible phase to excitation control the to radiation the previous pattern element. of the By array. varying the Appl. Sci. 2020, 10, x FOR PEER REVIEW 9 of 14

Appl. Sci. 2020, 10, 7187 9 of 14 weight factor of each channel, and properly estimating its values, it is possible to control the radiation pattern of the array.

Figure 10. Block diagram of Rx antenna array. Figure 10. Block diagram of Rx antenna array. The far ﬁeld of the array factor can be written according to the Equation (1), which relates the contributionThe far field of each of elementthe array for factor the totalcan be radiation written of according a linear array. to the Equation (1), which relates the contribution of each element for the total radiation of a linear array. XN = [(j[(n)1)kdcosθ+α ] 𝐴𝐹=AF 𝑎 𝑒ane − (1)(1) n=1 In Equation (1), the summation of independent phasors (weights), the progressive phase is In Equation (1), the summation of independent phasors (weights), the progressive phase is represented by ψ = kd cosθ+α, where k = (2π⁄λ) and d the distance between elements of the array. represented by ψ = kd cosθ+α, where k = (2π⁄λ) and d the distance between elements of the array.

3.1. Beam Steering Steerin Considering an array of N == 4 elements, with equal and unitary excitation amplitude for all elements, a =1, the respective phase delay to apply to each channel in order to have the array steer elements, an= 1, the respective phase delay to apply to each channel in order to have the array steer its radiationits radiation to the to positionthe positionθ is givenθ is given by α =by 𝛼=−𝑘𝑑kd cosθ. Table 𝑐𝑜𝑠𝜃1. Tableshows 1 the shows estimated the estimated phase delays phase for delays each − channel,for each channel, considering considering the set of the chosen set of locations chosen locations (θ), using (θ an), arrayusing ofan half-wavelength array of half-wavelength spaced elements. spaced elements. Table 1. Beamforming phase weights. Table 1. Beamforming phase weights. Steering Angle θ (◦) α1 (◦) α2 (◦) α3 (◦) α4 (◦) Steering angle 𝜽 (°) 𝛂𝟏 (°) 𝛂𝟐 (°) 𝛂𝟑 (°) 𝛂𝟒 (°) 40 0 173 346 519 ◦40° ◦ 0° 17◦3° 346◦° 519°◦ 30◦ 0◦ 135◦ 270◦ 405◦ 20◦30° 0◦ 0° 9213◦5° 184270◦° 405276°◦ 10◦20° 0◦ 0° 4792◦ ° 1894◦4° 276141°◦ 0◦ 10° 0◦ 0° 047◦ ° 940◦ ° 1401◦° 10 0 47 94 141 − ◦0° ◦ 0° − 0°◦ −0°◦ −0° ◦ 20 0 92 184 276 − ◦ ◦ − ◦ − ◦ − ◦ 30−10° 0 0° 135−47° −27094° −141405° − ◦ ◦ − ◦ − ◦ − ◦ 40−20° 0 0° 173−92° −347184° −276519° − ◦ ◦ − ◦ − ◦ − ◦ −30° 0° −135° −270° −405° 3.2. Beam Shaping −40° 0° −173° −347° −519°

3.2. BeamThe conceptShaping of beam shaping is vast and brings together all the techniques that, through feeding of the array, allows the modification of its radiation shape. Typically, this concept is used in communications to mitigateThe the concept influence of beam of interfering shaping signalsis vast throughand brings side together lobes (reducing all the techniques or eliminating that, them) through or confining feeding theof the width array, of the allows main lobe.the modification of its radiation shape. Typically, this concept is used in communicationsThere are diff toerent mitigate reported the methods influence in theof literature,interfering such signals as the through Schelkuno sideff polynomiallobes (reducing method, or minimumeliminating mean-square them) or confining error (MMSE)the width weight of the main and manylobe. others, and in this work, two methods, Dolph–TschebyscheThere are differentff and reported binomial methods [27], were in appliedthe literature, in the proposedsuch as the digital Schelkunoff system and polynomial tested to demonstratemethod, minimum their impact mean-square on the radiation error (MMSE) of the antenna weight arrayand many when comparedothers, and to in the this uniform work, array. two methods, Dolph–Tschebyscheff and binomial [27], were applied in the proposed digital system and Appl. Sci. 2020, 10, x FOR PEER REVIEW 10 of 14 testedAppl. Sci. to2020 demonstrate, 10, 7187 their impact on the radiation of the antenna array when compared to10 ofthe 14 uniform array. UsingUsing th thee theoretical theoretical expressions of [27 [27], for the array of N == 44 elements, elements, the the feeding feeding distributions distributions werewere estimated estimated for for the the binomial binomial method method and and for for the the Dolph Dolph–Tschebysche–Tschebyscheffff techniquetechnique using using different different valuesvalues of of s sideide l lobeobe l levelevel (SLL), such as −1515 dB, −2020 dB dB and and −2525 dB. dB. The The SLL SLL represents represents the the signals signals − − − beingbeing radiated radiated in in the the unwanted direction. direction. Applying Applying such such method methodss in in practical practical application application could could be be veryvery helpful helpful to to reduce reduce the interferences. TheseThese distributions areare providedprovidedin in Table Table2 and2 and were were applied applied to theto the theoretical theoretical Equation Equation (1). (1) In. the In theMatlab, Matlab, the the diff erentdifferent radiation radiation patterns, patterns considering, considering allthe all techniques,the techniques were, were calculated calculated using using (1) and (1) andmultiplying multiplying bythe by the theoretical theoretical beam beam pattern pattern of aof microstrip a microstrip patch patch antenna antenna (element (element factor), factor), which which is isthe the radiating radiating element element chosen. chosen The. The results results are are shown shown in in Figure Figure 11 11..

TableTable 2. EstimatedEstimated amplitude amplitude distribution distribution.. Weight method a1 a2 a3 a4 Weight Method a1 a2 a3 a4 Uniform 1 1 1 1 Uniform 1 1 1 1 BinomialBinomial 11 3 33 11 DolphDolph–Tschebysche–Tschebyscheffff ( (−1515 dB) dB) 11 1.33 1.331.33 11 − DolphDolph–Tschebysche–Tschebyscheffff ( (−2020 dB) dB) 11 1.74 1.741.74 11 − Dolph–Tschebyscheff ( 25 dB) 1 2 2 1 Dolph–Tschebyscheff −(−25 dB) 1 2 2 1

FigureFigure 11. 11. TheoreticalTheoretical r radiationadiation pattern patternss considering considering the the distribution distribution weights weights..

AccordingAccording to to Figure Figure 11,11, it it is is possible possible to to observe observe several several aspects aspects resulting resulting from from the the use use of of each each method.method. First, First, using using the the uniform uniform distribution, distribution, with with all all elements elements fed fed with with equal equal amplitude amplitude and and phase, phase, aa higher higher side side lobe lobe is is observable, observable, and and the the amplitude amplitude of of this this lobe lobe reduces reduces with with the the application application of of the the differentdifferent methods,methods, namely namely using using the Dolph–Tschebyschethe Dolph–Tschebyscheffff technique technique (for the di(forfferent the SLL different estimations), SLL estimations),and in particular and usingin particular the binomial using the method, binomial forwhich method, the for side which lobes the even side disappear. lobes even It candisappear. also be Itproved can also that be theproved amplitude that the distribution amplitude distribution from the application from the application of the Dolph–Tschebysche of the Dolph–Tschebyscheffff method for methodvery small for SLLvery values small SLL will tendvalues towards will tend the towards binomial the distribution. binomial distribution. AnotherAnother important aspectaspectthat that can can be be taken taken from from Figure Figure 11 is11 the is variationthe variation of the of half the powerhalf power beam beamwidth width (HPBW), (HPBW), using using the di thefferent different methods, methods, observing observing that that this valuethis value increases increases as the as the value value of the of theSLL SLL decreases. decreases. TheThe standard standard definition definition of of beam beam width width is is the the angle angle aperture aperture from from which which most most of ofthe the pow powerer is radiatedis radiated from from antennas antennas [27]. [ 27It ].is a It very is a important very important characteristic characteristic of an antenna of an antenna array, and array, a crucial and a Appl. Sci. 2020, 10, 7187 11 of 14

Appl. Sci. 2020, 10, x FOR PEER REVIEW 11 of 14 crucial characteristic in phased array antennas and beamforming, because of the useful advantage of characteristic in phased array antennas and beamforming, because of the useful advantage of minimizing unwanted interference signals by controlling the main lobe characteristics, beam width minimizing unwanted interference signals by controlling the main lobe characteristics, beam width and the side lobes. and the side lobes. The concept of half power beam width (HPBW) is commonly used and reﬂects the angular The concept of half power beam width (HPBW) is commonly used and reflects the angular aperture in which the gain of the antenna falls 3 dB. There are theoretical expressions reported in the aperture in which the gain of the antenna falls 3 dB. There are theoretical expressions reported in the literature [27] that estimate this value. literature [27] that estimate this value. 4. Measurement Setup and Results 4. Measurement Setup and Results The measurements scenario of the proposed system is illustrated in Figure 12. The setup was placedThe in anmeasurements anechoic chamber, scenario and of includedthe proposed a power system supply, is illustrated providing in theFigure DC 12. voltage The setup to the was RF upplaced/downconverter’s in an anechoic boards chamber, and theand PLLincluded LMX2595 a power (LO supply, frequency providing of 12.08 the GHz). DC Involtage addition to the to the RF frontendup/downconverter’s boards and theboards PLL, and the the setup PLL also LMX2595 included (LO the frequency Host-PC, of the 12.08 Tx GHz). and Rx In antenna addition arrays, to the andfrontend the USRP boards N310, and as the can PLL, be identiﬁedthe setup also in the inclu Figureded 12the. Host-PC, the Tx and Rx antenna arrays, and the USRP N310, as can be identified in the Figure 12.

TX RX

HOST-PC

USRP

PLL

Figure 12. Experimental setup to measure antenna radiation pattern. Figure 12. Experimental setup to measure antenna radiation pattern. Before starting the measurement process, and after carefully placing all the elements of the setup, Before starting the measurement process, and after carefully placing all the elements of the the first task was to calibrate the system, in order to ensure that all signals reached the antennas setup, the first task was to calibrate the system, in order to ensure that all signals reached the antennas in phase, and all signals received maintained their properties until they were read by the USRP. in phase, and all signals received maintained their properties until they were read by the USRP. The The principle of calibration is the same as beamforming, and consists of adding certain weight (phase principle of calibration is the same as beamforming, and consists of adding certain weight (phase and and amplitude) to each channel with the purpose of compensating for the hardware impairments amplitude) to each channel with the purpose of compensating for the hardware impairments in the in the upconverter/downconverter modules, the phase differences in the USRP daughterboard and upconverter/downconverter modules, the phase differences in the USRP daughterboard and the the temperature drifts. Using the Python/Matlab interface, a calibration routine was developed to temperature drifts. Using the Python/Matlab interface, a calibration routine was developed to compensate for the phase and amplitude differences of the channels. compensate for the phase and amplitude differences of the channels. The four elements of the transmitter phased array antenna were placed in the rotor arm of the The four elements of the transmitter phased array antenna were placed in the rotor arm of the anechoic chamber, 65 cm away from the receiver (ensuring the far field distance). In Matlab/Python, anechoic chamber, 65 cm away from the receiver (ensuring the far field distance). In Matlab/Python, a function that calculates the correct phase to apply to each channel based on the desired direction of a function that calculates the correct phase to apply to each channel based on the desired direction of beamforming θ( ) was developed. The USRP modulated the baseband signal into two orthogonal beamforming 𝜃(°)◦ was developed. The USRP modulated the baseband signal into two orthogonal carrier waves, thus generating a 3.84 GHz IF signal. This signal was then applied to an upconverter carrier waves, thus generating a 3.84 GHz IF signal. This signal was then applied to an upconverter module that converted it to 28 GHz. The resultant signal was fed into one element of the antenna array. module that converted it to 28 GHz. The resultant signal was fed into one element of the antenna Since there is a channel digitally controlled (in amplitude and phase) for each element of the array, it is array. Since there is a channel digitally controlled (in amplitude and phase) for each element of the possible to use various beamforming algorithms. In the tests carried out, the radiation beam varied array, it is possible to use various beamforming algorithms. In the tests carried out, the radiation (in relation to the perpendicular plane of the array) between 40 and +40 with one degree increments. beam varied (in relation to the perpendicular plane of the− array)◦ between◦ −40° and +40° with one This procedure was repeated for nine different positions of the transmitter. For each step, the received degree increments. This procedure was repeated for nine different positions of the transmitter. For signal in the antennas were captured, downconverted to 3.84 GHz, and then the IQ demodulation each step, the received signal in the antennas were captured, downconverted to 3.84 GHz, and then the IQ demodulation was performed in the USRP to obtain the IQ samples. From these received samples, it was possible to measure the amplitude and phase of the signal coming from each channel. Appl. Sci. 2020, 10, 7187 12 of 14 was performed in the USRP to obtain the IQ samples. From these received samples, it was possible to Appl.measure Sci. 2020 the, 10 amplitude, x FOR PEER and REVIEW phase of the signal coming from each channel. 12 of 14 The first measurement using the system was related to beam steering, in which the phases estimatedThe first in Table measurement1 were applied using to the the foursystem channels, was related to steer to the beam beam steering, to di fferent in which angles. the Figure phases 13a estimatedshows the in normalized Table 1 were measured applied radiation to the four patterns channels, of the to received steer the signal beam for to nine different different angles. positions Figure of 13thea transmitter,shows the normalized using uniform measured amplitude radiation in thefeeding patterns of of the the array. received signal for nine different positions of the transmitter, using uniform amplitude in the feeding of the array.

(a) (b)

FigureFigure 13. 13. MeasurementMeasurement results: results: (a ()a )p polarolar plot plot of of the the measured measured radiation radiation patterns patterns;; ( (bb)) m measuredeasured aantennantenna radiation radiation pattern pattern for for many many amplitude amplitude variations variations..

TheThe second second measurement was devoteddevoted toto thethebeam beam width width of of the the radiation radiation pattern. pattern It. It is is possible possible to toobserve observe good good agreement agreement between between theestimated the estimated curves curve of thes arrayof the factor array (Figure factor 11(Figure) and the11) measured and the measuredvalues of thevalues radiation of the patternradiation of pattern Figure 13 ofb. Figure It is noticeable 13b. It is noticeable that the radiation that the patternradiation with pattern the widest with thebeam widest is the beam one thatis the uses one the that binomial uses the distribution binomial distribution (27◦) and the (27° narrowest) and the narrowest beam is related beam with is related those withthat usethose uniform that use distribution uniform distribution (17◦). (17°). TableTable 33 comparescompares thethe informationinformation ofof thethe measuredmeasured beambeam widthswidths ofof radiationradiation patternspatterns fromfrom FigureFigure 13b13b and and those those taken taken from from the the theoretical theoretical results results of F Figureigure 11.11. It It is is possible possible to to verify verify good good correspondencecorrespondence between between the the evolution evolution of of these these values values..

TableTable 3. 3. HHalfalf power power beam beam width width ( (HPBW),HPBW), m measuredeasured and and s simulated.imulated.

Weight Method Beam Width MeasuredBeam width ( ) BeamBeam Width width Theoretical ( ) Weight Method ◦ ◦ Measured (°) Theoretical (°) Uniform 17◦ 24◦ BinomialUniform 27◦ 17º 24°32 ◦ Dolph–Tschebyscheff ( 15 dB) 19◦ 27º 32°26 ◦ Binomial− Dolph–Tschebyscheff ( 20 dB) 21 28 Dolph–Tschebyscheff− (−15 dB) ◦ 19º 26° ◦ Dolph–Tschebyscheff ( 25 dB) 22.5 30 Dolph–Tschebyscheff− (−20 dB) ◦ 21º 28° ◦ Dolph–Tschebyscheff (−25 dB) 22.5º 30° 5. Conclusions 5. ConclusionIn this work,s a 28 GHz SDR beamforming antenna system was proposed for 5G, radar or IoT applications.In this work All, thea 28 analog GHz SDR and digitalbeamforming modules antenna were properly system was developed proposed and for described, 5G, radar allowing or IoT applications.us to obtain aAll system the analog operating and digital in the modules Ka band. were The properly use of USRP developed enabled and a flexible described, and allowing versatile usarchitecture to obtain a since system all itsoperating characteristics in the Ka could band be. The adjusted use of digitally. USRP enabled The experimental a flexible and results versatile have architectureshown good since agreement all its betweencharacteristics the simulated could be and adjusted measured digitally. radiation The patterns.experimental It was results possible have to shownverify techniquesgood agreement to shape between the beam the simulated of the array, and as measured well as steering radiation it, in patterns. a simple It way, was throughpossible theto verifycontrol techniques of the signal to amplitudeshape the beam and phase of the of array, each channelas well as in thesteering USRP. it, This in a system simple demonstrates way, through great the controlflexibility of the and signal scalability amplitude to be usedand phase in a wide of each range channel of equipment, in the USRP. especially This system in radar, demonstrates mobile and greatsatellite flexibility communication and scalability systems. to be used in a wide range of equipment, especially in radar, mobile and satellite communication systems.

Author Contributions: Conceptualization, D.M., R.A., T.V.and J.N.M.; methodology, D.M., R.A. and T.V.; software, D.M., and R.A.; validation, D.M., R.A., T.V. and J.N.M.; formal analysis, D.M., and R.A.; investigation, D.M., R.A. and T.V.; resources, T.V. and J.N.M.; data curation, D.M.; writing—original draft preparation, D.M., R.A. and T.V.; writing—review and editing, T.V. and J.N.M.; visualization, T.V.; supervision, T.V. and J.N.M.; project administration, J.N.M.; funding acquisition, J.N.M. All authors have read and agreed to the published version of the manuscript. Funding: This work was partially supported by FCT/MCTES through national funds and when applicable cofunded EU funds under the project UIDB/50008/2020-UIDP/50008/2020 and the European Regional Development Fund through the Competitiveness and Internationalization Operational Program, Regional Operational Program of Lisbon, Regional Operational Program of the Algarve, in component FEDER, and the Foundation for Science and Technology, Project RETIOT, POCI-01-0145-FEDER-016432. Conﬂicts of Interest: The authors declare no conﬂict of interest.

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