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Tecknical Journal of the lnternaticraal Telephone and Telegraph Corporation and Asso£iate Companies
•
MANUFACTtIRE. (}F COAXIAL CABLES
FAULT LOCATIDN BY PULSE... TIME MODULATION
LIFE. OF VALVES WITH OXIDE..-COATED, CATHODES
SECONDARY EMISSION IN THE PRESENCE. OXIDE...COATED CATH1JllMlS OF .DIRECT-WIRE. REMOTE CONTROL AT LOCH SL0¥ POWER STATION
TELEVISION-LINK SOUND mPLEXER
TRANSMISSlfiN-:UNE BALANCE INDICATOR FOR TRANSMIT'l�E-RS
SUPPRESSIDN OF HABMONU::S IN RADIO TRANSMITTERS
trANTIZATION PULSE-COUNT MODULATION WITH NONUN"IFORM LEVELS Q IN TIME-DIVISION-MELTJ.PLEX TELEGRAPH SYSTEM FOR. RADIO CIBCUITS
INTERFERENCE IN MULTI-CHANNEL. CIRCUITS
Volume 28 JUNE, 1951 NumheT 2
www.americanradiohistory.com ELECTRICAL COMMUNICATION Technical Journal of the INTERNATIONAL TELEPHONE AND TELEGRAPH CORPORATION and Associate Companies
H. P. WESTMAN, Editor F. J. MANN, Managing Editor J. E. ScmAIKJER, Assistant Editor
EDITORIAL BOARD P. Bourget H. Busignies H. H. Buttner E. M. Deloraine W. Hatton F. B. C. Holding J. S. Jammer E. Labin A. W. Montgomery E. D. Phinney E.G. Ports Haraden Pratt G. Rabuteau T. R. Scott C. E. Strong A. E. Thompson E. N. Wendell H.B. Wood
Published Quarterly by the
INTERNATIONAL TELEPHONE AND TELEGRAPH CORPORATION BROAD STREET, NEW YORK 4, NEW YORK, U.S.A. 67 Sosthenes Behn, Chairman William H. Harrison, President Charles D. Hilles, Jr., Vice President and Secretary
50 ElectricalSubscription, Communication $2.oo per is year;indexed single in Industrial copies, Artscents Index Copyrighted 1951 by International Telephone andTelegraph Corporation
Volume JUNE, Number 28 1951 2
CONTENTS MANUFACTURE OF COAXIAL CABLES 83 FAULT LOCATION BY PULSE-TIME MODULATION . . . . • • ...... • . . . . • . . . . . • ...... • . By W. Hughes and Nelson Weintraub . 90
LIFE OF VALVESR. WITH OXIDE-COATED CATHODES . . . . • • ...... • . . . . . • • . • • ...... • By Eaglesfield 95 C. C. SECONDARY-EMITTING SURFACES IN THE PRESENCE OF OXIDE-COATED CATHODES ...... 103 By S. Nevin and H. Salinger
MINIATURE DIRECT-WIRE REMOTE CONTROL AT LOCH SLOY POWER STATION ...... 106 TELEVISION-LINK SOUND DIPLEXER •...... •..•...... •...... •.... By L. Staschover and H. G. Miller 108
TRANSMISSION-LINE BALANCE INDICATOR FOR TRANSMITTERS ...... •..•••...... 110 By George Royden
SUPPRESSION OF HARMONICS IN RADIO TRANSMITTERS • . T...... 112 By George Royden
QUANTIZATION DISTORTIONT. IN PULSE-COUNT MODULATION WITH NONUNIFORM SPACING
OF LEVELS . . . . • • • • • • ...... • ...... • • . . . . . • • . . . 121 By P. Panter and W. Dite
FouR/Two-CHANNELF. TIME-DIVISION-MULTIPLEX TELEGRAPH SYSTEM FOR LONG DISTANCE RADIO CIRCUITS ...... • • • . • ...... • ...... • By W. Peterman and Anatole Mine 127 C. INTERFERENCE IN MULTI-CHANNEL CIRCUITS .•••....•••...•.....•..••••..•.•.•..... 142 By L. Lewin
IN MEMORIAM-HERMAN T. KoHLHAAS •...... •...... •.•...... •.....•...... 157 RECENT TELECOMMUNICATION DEVELOPMENTS
MAXIMUM-AMPLITUDE MEASURING SET ...... ••...... ••.•..•.••....•.. 111 MINIATURE TOROIDAL INDUCTOR •..•...... •...... •...... · · · · · . • . · · · 155 TELEVISION SCANNER FOR SLIDES .....•...•....•.....•.•.•.....•... · • • · · · • · • · · · 156 ...... CONTRIBUTORS TO THlS ISSUE ··············· ...... 158
www.americanradiohistory.com www.americanradiohistory.com Manufacture of
Coaxial Cables
EDERAL Telephone and Radio Corporation, at Clifton, New ersey, is one of the largest manu J facturers of coaxial-type cables in the world. These cables are extensively used for the trans balanced line with three separate braided shieJ Fmission of radio-frequency power, particularly at the A higher frequencies where low losses and good shielding from external interference are important. The following few pages show pictorially the various stages in the manufacture of one of the simpler types of this cable. In the illustration above, the end of a reel of cable has been stripped to show the central conductor, a High-impedance delay line using a spiraled conduct solid copper wire; next is a layer of polyethylene plastic insulation, over which is a braided-wire shield, and finally, a noncontaminating vinylite plastic sheath for over-all protection. If the cable is intended for rough use in an exposed location, a braided layer of aluminum or steel armor wires may be applied over the sheath-the bottom reel in the above picture contains such armored Balanced air-spaced line for high-impedance circui cable. Some of the other types of cable that are fabricated on the versatile machinery at the plant are shown at the right in approximately actual size. The extrusion machine shown in numbers and of 3 4 the series may be considered the heart of the entire process. This machine must be kept running continu ously by splicing the end of one reel of wire to the start Balanced twinax line; two conductors with one shit of the next. The extruder is stopped only when a dif ferent type of cable is to be made. At one time, a con tinuous run of miles of type cable was 400 RG-8/U made on one of the four extruders at the plant. Wire passes through an extruder at roughly the speed of a fast walk, and very close coordination of speeds and temperatures must be maintained constantly. Double-shielded coaxial line with stranded conduc
June 1951 ELECTRICAL COMMUNICATION 83
www.americanradiohistory.com Both polyethylene plastic for insulation and vi coloring matter in the steam-heated mixer, and the nylite1. for the sheaths are compounded in the Ban doughlike mass is rolled out in thin strips. These are bury mixer in the background and then milled on the cut to convenient sizes, and after being ground to a heated rolls. Plastic in the form of a powder is mixed granular form in a chopper the plastic is ready to with a liquid plasticising agent and any necessary be placed in the hopper of the extrusion machine. www.americanradiohistory.com MOTOR DRIVE FOR FEED SCREW
:-';7----HOPPER FOR FEEDING PLASTIC FEED SCREW----._
HOT OIL HOT OIL
HINGE TO OPEN HEAD FOR CLEANING
PLASTIC-COVERED COPPER WIRE WIRE
- -
An over-all view of the front end of an extruder is in the extruding head, the wire goes through a 300-foot shown2. below. The wire passes from a reel at the left, trough of water. The temperature of the water in the through a tension-maintaining device, and then various parts of the trough is adjusted so that the through a series of gas flames that warm the wire and extruded plastic is cooled very slowly. Details of the ex drive off any moisture. After the hot plastic is applied truding head are shown in the diagram and view above.
www.americanradiohistory.com The view above shows the 3.back end. of the extruder, the large wheel in the foreground being the capstan that pulls the wire from the reel through the extrusion head and the long trough of water. On the table in the center of the picture are an air-blast drier, footage-count ing machine, and a a high-voltage breakdown tester, in which 10 to 20 kilovolts are applied to check the homogeneity of the polyethylene insulation.
The next step is the applica 4.tion of the braided-wire shield to the cable. The shield consists of a number of ribbons of wire interwoven to form a complete covering over the polyethylene insulation. Each of these rib bons is formed of anywhere from 2 to or separate wires of 7 10 very small diameter that must he laid exactly parallel to each other to form the ribbon. Per haps the most critical operation in braiding is the filling of the bobbins with these thin wires. The operation is illustrated at the left, where wires are being 7 wound on a bobbin. www.americanradiohistory.com A close-up of a braider is sh5. own at the right. The insu lated wire is pulled upward through the center of the cone. Two circles of 12 bobbins each rotate in opposite directions around the wire. The strands from the lower circle of bobbins are alternately passed above and below the strands of the upper circle. Thus, 24 separate ribbons are tightly interwoven on the cable. This same machine may he used to apply a braid of steel wire over the outside of a small armored cable.
A single operator is capable of6. replacing depleted bobbins and performing other necessary duties on a series of 10 hraiders. Such a row of machines is shown below. The insulated wire is pulled from the reel below the circles of bobbins hy the large wheel at the top of each ma chine. The braided cable is re wound on the second reel under the braider. The machines oper ate very rapidly, approximately 6 to 10 feet of cable being cov ered with braid in a minute.
www.americanradiohistory.com The above photograph shows braider that is 24 each that rotate in opposite directions around the 7. central cable, alternately passing on opposite sides used for cables of larger diameter athan those illus trated in On this machine, the ribbons of braiding of each other. ,\fter braiding, the cable is returned to 6. wire are carried on the bobbins in the shuttles. The an extruder, where, with a slight modification of shuttles on the table are divided into two series of the dies in the head, the vinylite sheath is applied. 18 www.americanradiohistory.com The maintenance of high standard8. of quality requires that a a complete laboratory he available to check the electrical and me chanical properties of the prod uct. In the area illustrated at the right, instruments are used to test samples for voltage break down strength, characteristic im pedance, electrical losses, low and high-temperature flexibility, and the effects of age.
The view below shows another 9.part of the inspection and ship ping department where high volt age is applied between the braid of the cable and the outside of the sheath to test the insulating properties of the sheath. While this test is being made in the box at the center of the picture, the operator is reeling up the cable and cut it for shipment on will reels in the lengths ordered.
www.americanradiohistory.com Atop Squak Mountain
Fault Location by Pulse-Time Modulation*
By R. W. HUGHES NELSON WEINTRAUB and Federal Telecommunication Laboratories, Incorporated; Nutley, New Jersey
HE PROBLEM of locating faults on a The system proposed by Stevens and String power line has long plagued the power field 1 is based on this method and is explained in industry. It is particularly troublesome detail as follows: fault occurs somewhere between A and B, the two sinceT approximately 90 percent of all faults are A. A ends of the powedine. This starts a steep wave-front surge not sustained and hence the location of the fault along the line in both directions at the rate of 0.186 mile must be determined on an instantaneous basis. per microsecond. In order to keep a record of the time and location B. Assuming A to be the recording station, the arrival of of all faults, a method based on the measurement the surge there causes an electronic counter to be triggered. of the time required for a wave to travel from the 1 R. F. Stevens and T. W. Stringfield, "Transmission fault to a substation appears most practical. Line Fault Locator Using Fault Generated Surges," Trans * Reprinted with added photographs from Electrical En actions of the American Institute of Electrical Engineers, v. gineering, v. 69, pp. 1009-1011; November, 1950. 67, Part 2, pp. 1168-1178; 1948.
90 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com C. At station B, the surge is detected, converted to a con C. The attenuation of the wave fronts is small enough to venient-sized pulse, and transmitted from B to A via a allow propagation of the order of several hundred miles, broad-band radio link. including passage through substations. The arrival of this second surge pulse stops the elec D. D. Available electronic counters are capable of measuring tronic counter, and from the elapsed time the location of time intervals as small as 0.625 microsecond. Since the the fault may be calculated. speed of propagation is 5.37 microseconds per mile, it is E. By the addition of some simple supplementary equip then feasible to measure propagation times corresponding ment, the counter can be arranged to read distance to the to an accuracy of almost 0.1 mile. fault directly, and this permits the recording of the informa tion for a permanent record in a fashion similar to the following. The proposed Bonneville Power Administra tion pulse-time-modulation system is shown in May 10, 1949 03 :45:11 168.6 MI Figure 1. The first portion, now installed, in cludes the stations: J. D. Ross, Rainier, Che It is of interest to note that the fault-locating halis, Olympia, Squak Mountain, Seattle, Cov system is based on the following facts. Faults produce steep wave fronts, which may be de A. ington, and Snohomish. As shown in Figure 2, it tected from the normal power frequencies. consists of a terminal, two video repeaters, an B. The speed of propagation of these wave fronts along a audio repeater, an audio branching repeater, transmission line is constant within a very small percent and three terminals. Only eight of the 23 chan age. nels are being utilized at present. Over this
LOADChiefDISPATCHERS Load Dispatcher BPA Sub-Dispatcher + CityBPA of Seattle Puget Sound Power Light City of Tacoma & Co <%> Washington Water Powerco Portland General Electric Co Pacific Power Light Co City of Eugene& Elec
l .f\..!,.- A.�• Scriver Cr. I Garden Valley l!:iir;i v-·� \ LEGEND Palisade� Generating station Ranch 1:1 Substation Boise r;iAnderson 0 O
Figure 1-Proposed Bonneville Power Administration communication system as of 1954. Reprinted from R. F. Stevens, "Microwaves Supplement Present Channels," Electrical World; January 1, 1949.
June 1951 • ELECTRICAL COMMUNICATION 91
www.americanradiohistory.com microwave link, the fault pulse must be trans modulation radio relay system pass a 10-micro mitted and hence provisions made for supple second pulse with rise and decay time of 0.2 mental equipment. This equipment and its microsecond maximum . is permissible with no necessities are described successively in the fol modificationwhatsoever, since the bandwidth of lowing paragraphs. the system is greater than necessary to support The basic requirement that the pulse-time- the fault pulse. Also, the modulated pulse train
SEATTLE
Figure 2-Block diagram of present Bonneville Power Adminis MULTIPLEX tration pulse-time-modulation system. The radio link symbol designates an antenna and its associated transmitter or receiver.
MULTIPLEX MULTIPLEX DROP AND DROP AND MULTIPLEX IHSER T I HSERT
SNOHOMISH J.D.ROSS RAINIER CHEHALIS OLYMPIA
MULTIPLEX
COVINGTON
could be interrupted for several hundred micro seconds with a barely perceptible effect on voice communication, so no modificationsare necessary to avoid interruption of normal traffic. Paradoxi cally, it is the normal traffic that will affect the fault pulse. To be explicit, the leading or trailing edge of the fault pulse might be displaced in the event a channel pulse of the communication pulse train occurs at the same time. To avoid this, it is necessary to blank out the pulse train prior to the insertion of the fault pulse. For rea sons involving repeater points, the trailing edge of the fault pulse was chosen as the time-defining edge. Hence, at the insertion point of the fault pulse, the pulse train is blanked out for approxi mately 15 microseconds, starting simultaneously with the fault pulse, but lasting five microsec Front view of power-line fault-locator printing equip onds longer to protect the trailing edge. This is ment. Extending from the upper section may be seen the accomplished in an 8-tube unit called the "fault paper tape on which is printed the date, time to within a pulse modulator," normally rack-mounted ad second, and distance to within a tenth mile from the end of the power line to each fault. jacent to the modulator multiplex output. It re quires 5 inches of rack space and is provided with
92 ELECTRICAL COMMUNICATION • June 1951
www.americanradiohistory.com SEATTLE MULTIPLEX
DROP AND INSfRf
MULTIPLEX
DROP AND INSERT
SNOHOMISH
J.D. ROSS FAULT-PULSE OUTPUT OLYMPIA
FAULT- MULTIPLEX FAULT- PULSE PULSE DUTPU INPUT
COVINGTON
Figure 3-Present Bonneville Power Administration system block diagram showing fault-location equipment. The Rainier and Chehalis stations being identical and ad jacent, only one is shown. Circles identify equipment for
handling fault pulses: Blank = blanking-signal generator,
= demodulator, MOD = modulator, and MIX = DEM mixer.
two inputs for fault pulses to allow for dual inputs from branching power lines. By video repeater is meant a repeater wherein the microwave signal is detected as a video-pulse train, but no demodulation to voice is provided. The video-pulse train simply remodulates the transmitter. This would permit the passage of the fault pulse without any modification. When one or more channels are demodulated to voice, however, the old channel pulse is dropped out and new one inserted (hence, the name drop a and insert unit). If either the blanking gate or the new channel pulse should coincide with the fault pulse trailing edge, a timing error would be intro duced. To avoid this, a unit must be provided at re peaters having drop and insert units to blank out the drop and insert signals during the fault pulse. Since it requires about one microsecond to de tect the fault pulse from the channel pulses, the Rear view of printer cabinet. blanking signal to the drop and insert unit is
June 1951 • ELECTRICAL COMMUNICATION 93
www.americanradiohistory.com delayed by that time and hence it is possible for one microsecond to be lost off the leading edge of the fault pulse. This is why the trailing edge of the fault pulse was used for time definition. It is also why the long 10-micro second pulse was specified, since as short a fault pulse as three microseconds can be detected at the terminal. This allows several drop and insert losses without affecting final detection. Detection of the fault pulse at the receiving station is accom plished by an integrating circuit to select the longer fault pulse, followed by a differentiating cir cuit to definesharply the trailing edge. The result is reshaped for transmission to the fault locator. This is provided by the fault pulse demodulator normally lo cated close to the demodulator multiplex since it also receives the received video-pulse train. The 4-tube unit requires 3 inches of rack space. At an audio repeater such as Olympia and Squak Mountain, both the fault-pulse demodulator and modulator are required, since the fault pulse must be de tected and reinserted in a new pulse train. The Bonneville pulse-time-mo dulation system is shown again in Figure 3, but this time it in cludes all the fault-pulse equip- ment necessary for the system. A fault causes a pulse to propagate to each end of the power line. At one end, the pulse starts the electronic counters shown above. The counters are Note that the use of time-divi stopped by the second pulse, which is transmitted by radio from the other sion multiplexing permits a high end of the line. From this time interval, the location of the fault is computed degree of flexibility for the fault automatically and a printed record is produced immediately. locating plan. As an example in the Bonneville system, provision is made at Olympia for an additional fault-pulse Covington. In the case of more than one power insertion to allow for future expansion of the line supplied by the same radio relay system, microwave radio relay system. Also, at the Squak ambiguity may be avoided by the use of coded Mountain repeater, provision was made to allow fault pulses, thereby allowing one recording sta fault-pulse insertion from either Snohomish or tion to provide facilities for two power lines.
94 ELECTRICAL COMMUNICATION • June 1951
www.americanradiohistory.com Life of Valves with Oxide-Coated Cathodes
By C. C. EAGLESFIELD
Standard Telephones and Cables, Limited; Ilminster, England
HE LIFE of valves with oxide-coated barrier between the cathode core and the coating, cathodes has long been a cause of which causes a feedback and thereby a change in speculation, and it may therefore be the measured characteristics. This effect seems ofT value to put on record the results of investiga to occur so universaily that the study of cathode tions of the life-test records of telephone repeater life almost becomes a study of this resistance. valves of the valve division of Standard Tele There is a difficultywith some of the evidence, _ phones and Cables. particularly regarding All evidence given extremely long lives. here is limited to re When we consider a peater valves manu life of twenty years, factured by that we are faced with the company because the fact that the valves details of their manu must have been made facture are known. over twenty years However, since the ago, when the tech making of cathodes nique was different is now fairly stand from that of today. ardised, the conclu Since there is not sions are likely to be sufficient information general. to trace the effect of There are very the many variables, many diseases that we can proceed only can assail valves and by ignoring them. By cause their failure, correlating the evi among which may be dence in this way, mentioned mechan we reach conclusions, ical breakages such as but their validity is cracked envelopes, more probable than disconnected lead certain. wires, or fractured In the next section heaters; or electrical are described typical faults such as unwanted emission from electrodes life histories and the three groups into which or leakage across insulators. These and many they faII. In the second section, it is shown other troubles can end the life of a valve, but how aII these typical lives are accounted for the present discussion is confined to the emissive by the growth of cathode resistance ; also it is properties of the cathode. shown how this resistance can be identified and It is generaIIy supposed that the lives of measured. cathodes are only moderate ; it may therefore The third section discusses possible causes of come as a surprise that no evidence could be found the resistance and whether cure is feasible. a to put a definite term to the cathode lives. The As it is reluctantly concluded that a cure is dubi conclusion seems to be that the emission con ous and will take a long time to establish, a tinues indefinitely. fourth section is devoted to ways of designing However, an effect was found that could apparatus so that the effect of the resistance is easily be, and in fact was, mistaken for a drop in made small. the emissivity. This is the formation of a resistive
June 1951 • ELECTRICAL COMMUNICATION 95
www.americanradiohistory.com 1. Typical Life Histories make it difficult to find life histories of great length, and it is now realized that valves have Before proceeding to consider the known his often been taken down just when, from a general tories of valves that have been subjected to con point of view, interesting information was to be trolled life-tests, it is well to give some thought expected. to why such tests are conducted and what in Incidently, while the considerations listed formation is likely to be obtained from them. would probably apply to any life-test department, The main purpose of the test is to establish it is relevant to mention that a serious fire took that each valve type is likely to give satisfactory place in 1946, which cancelled many tests and service and to keep a continual check on the caused some confusion in the records. With these limitations in mind, we can now \ proceed to consider the evidence. First of all, it 'V v has been noticed that valve types may be �v- -i---..""" divided into three classes as follows.
1.1 FIRST CLASS The first class has a rather short life, of the '" -r---, order of one or two thousand hours, after which - - v - '; 4 8 12 16 20 24 28 time the emission has seriously deteriorated. LIFE IN THOUSANDS OF HOURS This class of valve usually suffersfrom some char Figure 1-Life history of a batch of 12 small radio acteristic fault, such as high gas pressure, which frequency pentodes tested with 250 volts on the anode, can be associated with the design of the valve. 180 on the second grid, and -5 · 5 volts on the control grid. This is not to say that the valve is badly de signed, but that the performance requirements manufactured product. To do this, a certain have forced the designer's hand. Thus the short routine has to be followed. life is understood to be the price that has to be Valves are run under specified conditions, paid to get the required performance. and their characteristics are measured at gradu Examples in this class are specialized valves ally increasing intervals. These characteristics, for use at centimetric wavelengths-in many such as currents, impedances, mutual conduct such valves, full degassing on the pump is very ances, etc., are usually those measured in the difficult-or valves that, to save space or weight testing of a new valve, and rejection of the valve of equipment, are grossly overloaded. during life-test follows if the characteristics fall Of course, there is continual development outside certain limits. This is similar to the test going on, and the lives of such types are always ing of a new valve except that the life-test limits being increased. But the point here is that these are wider. very short lives are caused by poor vacuum and Valves are also checked for their mechanical are therefore not relevant to the general argu characteristics, which are just as important for ment. As has been pointed out by Metson,1 satisfactory service. For instance, a loose base normal receiving-type valves very quickly reach or top cap, a sagging filament, or a bowed an extremely high degree of vacuum. We shall cathode would cause rejection of the valve and, not discuss this first class any further. unless a repair were possible, removal from test. There is another cause that makes for the re 1.2 SECOND CLASS moval of goo 96 • ELECTRICAL COMMUNICATION June 1951 www.americanradiohistory.com consider Figure 1, which shows the average his tory of a batch of typical valves. These are small pentodes for radio-frequency amplification and GREATEST,)' have 3-watt cathodes ; the conditions of testing aµd running on life are approximately the same. - - - - It will be seen that apart from a certain settling . AVERAGE-I down in the first thousand hours and some small - fluctuations thereafter, there is no significant LEASF-I change in the characteristics. This batch is still on test at the time of writing. Many examples can be found up to the same length of life, but for a longer test consider 9 Figure 2, which shows the average history of a batch of special triodes made for an emission ------study. These valves had 6-watt cathodes and GREATEST again the test and running conditions were ap AVERAGE l proximately the same. It can be seen that the LEAST ------change in characteristics up to SO 000 hours ( 6 years) is negligible. This test was stopped by the 4 fire already referred to. 0 4 8 12 16 20 For a longer life history, we have to leave LI FE IN YEARS organized tests and consult actual service. Re Figure 3-Characteristics of a batch of 18 type 4101 cently a telephone repeater station was dis valves after 20 years of service tested with 130 volts on the anode and -9 volts on the grid. The normal test limits are mantled and it was noticed that a number of indicated by dashed lines. valves had been in service since the repeater was installed, a matter of twenty years. These valves, type had a 4-watt filament and, remarkably, no getter. Of the three types quoted for this long-life � ...... - /'\. class, it may be mentioned here that the cathodes --..... I v '\.._ � differed in detail. The pentodes are coated with triple-carbonate on cores of 0 nickel, a grade of ' nickel very commonly used for cathodes and con taining small additions of reducing impurities. 22 The cathodes of the triodes were triple-carbonate S on nickel, a grade of nickel with rather smaller J - proportions of impurities. The filaments of the \ 4101 / were mixed carbonates on a platinum I �'/ �- I \ v I\ nickel alloy and the method of coating was very I \ different from the method used today ; so was the method of pumping. More recent samples of 18 0 10 20 30 40 50 this type have had triple-carbonate coatings on LIFE IN THOUSANDS OF HOURS filaments of 0 nickel, but unfortunately the Figure 2-Life history of a batch of 5 triodes tested with longest organized life-test seems to be for 30 000 250 volts on the anode and -10 volts on the grid. hours only, during which time the changes of characteristics were negligible. type 4101, were returned by the courtesy of Dr. Ryall of the General Post Office and their 1.3 THIRD CLASS history is exhibited in Figure 3. It can be seen The third class of valve has an apparent life that there can have been little change in their of several thousand hours; that is, a life inter characteristics during the twenty years. This mediate to the very short ones of the first class June 1951 • ELECTRICAL COMMUNICATION 97 www.americanradiohistory.com and the very long ones of the second class. A a certain point and then stops. The length of life typical example is shown in Figure 4, in which is therefore very much a matter of definition. both the working bias and the mutual conduct ance show substantial changes in 4000 hours. 1.4 CONCLUSIONS We have now divided valves into three classes, ...... according to whether their lives are short, long, ""'-- or medium. The first class has beendisposed of as -...... having faults of design, but is there any way of � .___ forecasting whether a valve will fall into the --r-- second or the third class? At first, it was sup r-- r-- posed that the answer might be connected with the cathode loading, that is, the ratio of cathode current to cathode area. In fact, this loading is small in many of the long-life valves, but it was U) 2-0 !:::i not found to separate the classes. It was then 0 > -- noticed that the ratio of mutual conductance to --- -...... cathode area did separate the classes : if the ratio I--. """'-- l·O I""--. � were low, the valve fell in the long-life class and I'"-. if it were high, the valve had a short life. 0·5 This suggested that the cause of the short lives 0 I 2 3 4 LIFE IN THOUSANDS OF HOURS might be a feedback due to a growth of cathode Figure 4-Life history of a batch of 6 miniature radio resistance. Such a resistance might be expected to frequency pentodes tested with 250 volts on the anode and be inversely proportional to the cathode area. second grid and with 10 milliamperes anode current. The feedback would depend on the product of mutual conductance and resistance and there fore on the ratio of mutual conductance to Another example is shown in Figure 5, in which cathode area. there is appreciable change of mutual conduct It will be shown in the next section that a ance in 13 000 hours. Both these types are small growth of cathode resistance is indeed the cause radio-frequency amplifiers; the first has a 2-watt of the effect. and the second a 3-watt cathode, in both cases of 2. Cathode Resistance 0 nickel coated with triple-carbonate. The deterioration of characteristics shown by When the life history of a valve shows a deteri this class of valve is usually sufficient to cause oration in working current, bias, or mutual con its rejection and removal from test, but special ductance, it is perfectly feasible to postulate a tests have shown that the deterioration reaches cathode resistance of sufficient magnitude to ex plain the change. If this be done, one would ex - I"----.. pect the required resistance to be inversely - I""- proportional to the cathode area. -- -- ...__ r- In Figure 6 is shown the life history of three -- valve types in terms of such a cathode resistance. The resistance is normalized for a cathode area � 2·5 of 1 square centimetre and is derived from the Q � 2·0 observed change of mutual conductance during ..!... life. Two of the types are those of Figures 4 and % 1·5 �= 4 6 8 � 0 2 5; the third is a radio-frequency pentode of some 1 10 12 LIFE N THOUSANDS OF HOURS what greater rating: it has a 5-watt cathode Figure 5-Life history of a batch of 8 radio-frequency (triple-carbonate on 0 nickel) and a mutual con · pentodes tested with 250 volts on the anode, 150 volts on ductance of 6 5 milliamperes per volt at its usual the second grid, and 10 milliamperes anode current. working anode current of 38 milliamperes. 98 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com It will be seen that the normalized resistance case the previous measurements associates the builds up to a saturation value of about 40 ohms resistance with the cathode. for all three types, with a surprisingly sharp The fact that the resistance has a substantial angle where the rise meets the saturation level. capacitance in shunt is remarkable and makes the The resistance grows in a similar way for many measurement of the resistance a simple matter. other types; the three valves shown have been Pulses and square waves can also be conveniently chosen because it happens that the tests have used. been continued long enough to show the satura The double-frequency method has been used tion level clearly. All the life histories that have to measure the cathode resistance of a number of been examined can be explained by the hypothe valves that had deteriorated during life, sup sis that a cathode resistance builds up to 40 ohm posedly for a drop in emission. In every case, a square-centimetres and then stays constant. Such resistance was found and the resistance was ap a resistance explains the change i� characteristics proximately that required to explain the change of the medium-life valves; it is too small to affect in characteristics. the characteristics of the long-life class. iQGO It is important to realize that the suggestion is z 50 �-w t;; "" that no change takes place in the emissivity of 40 / �v z;:o I j:! i= ...... "' the cathodes, but that in all valves the resistance 30 en z I/" grows to a certain value and then stays constant. ::3-w If v 20 , ...... l,...-" It may, or may not, change the measured char c::w I v UJ:5 Ct: 10 / .v �g0 acteristics appreciably, according to the design I ' June 1951 • ELECTRICAL COMMUNICATION 99 www.americanradiohistory.com and the high capacitance is explained by the scale manufacture, but this expedient is hardly close spacing between the core and the body of practicable for mass production. In any case, the coating. Raudorf associated the shrinking of even if an initial activation be completed, it the coating with a network of fine cracks that he seems likely that it would6 not be stable. For ex observed on the outer surface of the coating of ample, see Wright, who reports a dropping aged cathodes, and stated that he found flat activation when a pure core is used. cathodes much superior to round cathodes. Another way might be to use cores that are Eisenstein's4 theory is that a resistive film is much more active to get the resistance formed formed at the interface of core and coating by the quickly, as part of the production processing. formation of compounds of barium and core im 1000 purities. These impurities are deliberately in cluded in the core metal as reducing agents to 600 promote activation : the most usual are silicon and magnesium and their effectis to release free "I ... 300 � barium. Eisenstein regards barium-orthosilicate ' as the most important cause of the resistance. This and other compounds of barium and core !/) :E impurities have also been observed by Wright.5 i5 100 \ ' . Now, it is a difficultmatter to decide with cer ' ' tainty which theory correctly explains the resist z 60 ance that has been observed on our valves. The w 0 \ surfaces of the cathodes do not show the network z \ � 30 of cracks observed by Raudorf and the coatings (/) (/) seemed to adhere closely to the cores. There is w � also the question of round and flat cathodes : a: w 0 Raudorf quotes a typical figureof 40 ohm-square 0 10 centimetres for round cathodes, which agrees :I:� \ 6 closely with the results shown in Figure 6; but '' all the valves for Figure 6 have flatcathodes and ' we have found no systematic difference between 3 ' round and flat cathodes. \ As for the resistive interface, the cathodes cer ' tainly contained the impurities and doubtless the interfaces were formed ; however, the diiiculty I is to be sure that all the resistance observed was 3 4 5 6 7 8 HEATER VOLTS due to this and not, in part, to a bad contact. Figure 7-Variation of cathode resistance with heater It is only when a cure for the effect is being voltage in a typical aged valve. The nominal heater voltage considered that it is necessary to decide on the was 6·3 volts, corresponding to a cathode temperature of cause. It seems to the writer that Eisenstein's approximately 1000 degrees Kelvin. The coated area of the theory is more probable and contains fewer in cathode was 1 square centimetre. consistencies. It is interesting to use it as a basis for a possible cure. Even if this proved feasible, and the speeding-up The obvious course is to use cores that are free required is very great, it might be a thankless from impurities. However apart from the high task to persuade users that they were getting an cost of very pure nickel, there is the difficulty of improved article. activation. Reducing gases such as methane or A third method might be to use cores that are ethane, can be introduced on the pump in small- more passive to delay the formation of the inter A. Eisenstein, "Letter to Editor," Wireless Engineer' 100-101; 1950. face as long as possible. This would make activa- v. 427 , pp. March, A. Wright, "Thermonic Emission from Oxide Coated A. 5 D. D. W:igh\ "E!e.ctrical Conductivity of Oxide Cathodes," Proceedings of the Physical Society, 62, Part British Journal Cathode6 Coatmgs, of Applied Physics, 3, 351B, pp. 188-203; March 1, 1949. 1, 150; 1950. v. p. June, n. v. 100 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com tion more difficultand would only have a delaying been verified that the resistance is linear up to a action ; the resistance would build up eventu loading of 40 milliamperes per square centimetre. ally. The resistance is temperature dependent, but as A fourth possibility is to find a reducing im there is little difference between the cathode purity that would not be so highly resistive. temperatures of one valve type and another, There is also the question of the influence of this need not concern the user. the temperature of the cathode. It seems very It would be useful to be more specific on the likely that a temperature lower than normal time of growth : this time probably depends on a would delay the formation of the interface, but number of factors, such as the cathode tempera on the other hand its resistance would be higher ture and the quantity and kind of core impuri when it was formed. This is illustrated by Figure 7, ties. But it also seems to depend on details of the which shows the effectof the cathode tempera pumping and processing. It is difficult at present ture on the resistance. The lower the temperature, to be more specific. the higher the resistance. The user will probably not know the coated An interesti�g deduction from Figure 6 is area of the cathode of any particular type of that the chemical process appears to come to an valve, but he may estimate it from the rated abrupt end when the interface reaches a certain heater power, on the basis of 3 watts per square thickness. There seems no difficulty in assuming centimeter. the process to be of this type, in which the inter The easiest way to deal with this problem is to face acts as a barrier between the impurity on concentrate on the two states of the valves : if the one side and the barium on the other. It the apparatus is satisfactory for both states, it would thus seem that after a certain time the seems a fair deduction that it would be satis formation of free barium is stopped, or at any factory during the period of growth. The de rate greatly reduced. It appears to be sufficient signer may proceed with a trial design, based on to replace what may be lost by evaporation dur the valves in their new state. He then estimates ing normal running, but it seems possible that the resistance and capacitance that will grow at an aged valve could not recover so well from an each cathode a11d, by experiment or calculation, accidental overload. tests whether the design is still satisfactory. In considering any possible cure, it is well to It is customary to allow a factor of safety to remember the time scale of Figure 6, which shows cover the deterioration of valves during life ; the the order of time requir ed for an experiment. advantage of the systematic process outlined One is reluctantly driven to conclude that not above is that, after it has been carried out, the only is the possibility of a cure dubious, but it is apparatus should function almost for ever. going to take a long time to establish its success. It is hardly possible to give very general in This being so, it is very well worth while to structions on how to choose circuits that will consider what can be done in the design of ap prove satisfactory : each case must be considered paratus to reduce the effect of the resistance, on on its merits. However, consideration suggests the assumption that all present valves have it that a liberal use of feedback gives the best and all future valves will have it, for an inde chance of success. The reason is that the feed finite time ahead. back produced by the life-impedance depends on Design of Apparatus the valve's effective mutual conductance and a 4. permanent feedback effectively reduces the The known facts can be summed up by saying mutual conductance. A rough rule is that the that within six months to two years running, permanent feedback should swamp the life valves will change from their initial state to a impedance feedback. final state in which they have a cathode resist Take, first of all, a rather simple case, a single ance shunted by a capacitance. valve used for class-A amplification at high radio Measurements on valves in their finalstate sug frequencies. Since the life-resistance is adequately gest average figuresof 40 ohm-square-centimetres by-passed by the life-capacitance, it will cause no · for the resistance and 0 005 microfarad per feedback, but will only alter the bias conditions. square centimetre for the capacitance. It has This is easily allowed for by using a cathode June 1951 • ELECTRICAL COMMUNICATION 101 www.americanradiohistory.com resistor large compared to the life-resistance, to frequency characteristic can be got by using high provide a bias. This usually gives excess grid forward gain and high feedback or low forward bias, so the grid is returned, not to earth, but to a gain and low feedback. It may thus be possible to positive point. By this very simple device the swamp the life-feedback. valve's running conditions can easily be kept Where there are a number of stages, it must be almost the same for its two states. This is an considered whether to use feedback over several excellent illustration that the designer, when stages, at each stage, or a combination of both. forewarned, can often take effectivesteps to meet If great linearity is required in the input-out future conditions. Without the warning, he would put voltage characteristic, then feedback will be probably have provided a normal bias resistor. needed for this purpose. Such a requirement Now consider an amplifier for audio fre arises in multi-channel carrier amplifiers. For quencies. It is not likely that the valves for this this case, it may prove best to use frequency service will show particularly large changes dur dependent feedback at each stage and frequency ing life, as there is little need for a high ratio of constant feedback over the whole chain. mutual conductance to heater power. The feed There are, of course, very many other valve back due to the life-impedance will be constant applications in addition to those touched on over the audio-frequency band. To reduce dis above. These must be dealt with by the designer tortion, it is customary to provide a strong feed in a similar way. It is hoped that enough has been back from the output of the amplifierto an early said to show that there is a hope of success if the stage and this would probably swamp the life problem is tackled in the design stage and to feedback. It seems likely that little modification direct attention to the matter. would be needed to most audio-frequency ampli fiers to make them satisfactory. Conclusion 5. It thus seems that where the application in volves frequencies either very high or very low, It has been shown by a study of the life his there should not be any great trouble. A more tories of valves that it is hardly possible to as difficult case is the video-frequency amplifier, sign any finite life for the emission from oxide partly because the life-feedback then varies over coated cathodes, but that there is an effect that the band and partly because the valves likely to could be mistaken for a finite life. This is that a be used are just those most susceptible to the resistance grows between the cathode core and effect. the coating, which changes the performance of Such amplifiers are normally required to have the valve. The life of the valve is divided into a flat frequency characteristic, and it is therefore two parts, before and after the formation of the necessary to compensate for their natural tend resistance. Although there seems little immedi ency to fall offat the higher frequencies. This is ate prospect of making valves free of this resist often done by increasing the effectiveness of the ance, it may often be possible to design equipment inter-stage coupling at the higher frequencies, in such a way that the effect of the resistance is butanotherway is to provide a frequency-depend not important. ent feedback, and this seems a better way for Acknowledgment our present purpose. 6. Considering a single stage, the compensating feedback may be a resistance and capacitance in The writer is grateful to a number of his shunt in the cathode lead. Now there is usually colleagues for assistance in preparing this paper a range of feedback for which the overall result and in particular to Mr. W. T. Gibson 0.B.E., is much the same, i.e., the same stage gain and chief valve engineer, for helpful advice. 102 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com Secondary-E itting Surfaces in the Presence of n;i. . Oxide-Coated Cathodes * By S. NEVIN H. SALINGER and Capehart-Farnsworth Corporation; Fort Wa yne, Indiana HE EXPERIMENTS described here ing temperature. It is thus possible that the show that the deleterious effect of ox target contamination occurs, at least in part, ide cathodes on secondary-emitting during the forming process. Most of the experi surfacesT of silver-magnesium can be overcome by ments were therefore performed with tubes as using tantalum instead of nickel as the base metal shown in Figure 1, with a sliding target that for the oxide coating. arrangementdid not face theis cylindrical. cathode during The straight formati on.cathode The K (diameter 116 inch) occupies the center ; it is It is well known that secondary-emitting ma surrounded by a collector, which is shaped as a terials are subject to contamination from heated squirrel cage (24 tantalum wires, 3 of them 0.015- cathodes. The deterioration in secondary emis inch, the others 0.005-inch thick, held together sion caused by this contamination is so serious by top and bottom tantalum ribbons ; outside that tubes using a hot-cathode source and elec diameter i inch). The target is coaxial with the - tron multiplication have usually1 3 been specially cathode and collector and can slide on three designed so as to make the target (dynode) less accessible to material coming from the cathode. According to a recent paper by Mueller,4 this difficulty has been overcome for filamentary cathodes and also, to some extent, for indirectly heated cathodes. The experiments described. here were entirely confined to indirectly heated cathodes, and as they resulted in a novel means to avoid the con tamination, a short report on them may be of interest. E 1. Experirnental Technique 1.1 TUBES s While an oxide cathode is being formed, its A temperature is raised for a short time to a value that may be about 200 degrees above its operat- * Reprinted from Proceedings of the I.R.E., v. 39, pp. VIEW AT A-A 191-193· February, 1951. This work was done in 1948 under c�ntract W36-039-sc-33864 with the Evans Signal , Laboratory. . . . , M. Chauvierre, "Secondary Em1ss1on Amplifier Tube, Tele-Tech., 1 v. 6, p. 69 ; July, 1947. H. M. Wagner and W. R. Ferris, "The Orbital-Beam Secondar2 y-Emission Multiplier for UHF Amplification," Proceedings of the I.R.E., v. 29, pp. 598-602 ; Novem ber, 1941. C. S. Bull and A. H. Atherton, "A New Secondary Proceedings of the Institution of Electrical En- Cathode,"a gineers, v. 97, pp. 65-7 1; March, 1950. . C. W. Mueller, "Receiving Tubes Employmg Sec01;d Figure 1-Sliding-target tube in which is for cathode ; ary Electron Emitting Surfaces Exposed to Evaporat10n K • C, collector; T, target; mica disks ; Ni, nickel ribbon; from Oxide Cathodes," Proceedings of the I.R.E., v. 38, pp. M, 159-164; February, 1950. S, support rods ; and E, glass envelope. 1951 ELECTRICAL COMMUNICATION June 103 • www.americanradiohistory.com S. support rods On the pump, the tube is mounted Attempts to determine met with limited suc a upside down, so that the target slides by gravity , cess only ; the best estimate is that a= 20 percent into the position shown in the drawing. In opera in our tubes. This would mean that K = 3 really tion, it slides down to surround the collec;:tor. corresponds to a "true" coefficient Kt= 3.5. The A similar construction, without the movable data recorded below refer to K rather than K1, features, was used on fixed"target tubes, in which because K gives a more direct measure of the the target is permanently exposed to the cathode. practical gain than can be realized in a secondary Eight-percent, 4-percent, and 1.7-percent mag emission tube. nesium-silver alloy was used as target material, The K measurements were performed with but the final experiments all used the l.7-per direct current, but to guard against the possible cent alloy. presence of time delays, as in the Malter effect,5 FORMING SCHEDULES it was ascertained that the collector current per 1.2 fectly reproduces the cathode-emission varia The procedure that worked best was as follows. tions, at least up to 1 megacycle per second. The tube, after assembly of parts, is given a For life tests, it was necessary to keep the vacuum bake-out at 460 degrees centigrade, and primary-emission current of the cathode con then the cathode temperature is raised in steps, stant. This proved difficult, because any excess so as to outgas it. The cathode is then flashed to emission led to a noticeable increase in collector about 1150 degrees centigrade (true tempera temperature, which by radiation caused the ture) , and then the collector is brought up to cathode temperature to rise, thus increasing the about 200 volts, with the emission rising up to 80 emission still further. For the life tests, therefore, milliamperes. Thereafter, the cathode is cooled, a thyratron circuit was used in the filament sup oxygen is admitted, and the target is heated for ply, controlled by the emission current so as to about 1 minute with radio-frequency current. keep it constant.6 This oxidizes the target, but damages the cathode 2. Results so that it has to be re-formed. Then the getter (if a getter was used) is flashed and the target Early results indicated that the decay in sec once more outgassed by radio-frequency current ondary emission depended very much on the after which the tube is sealed off. cathode temperature. It became increasingly Numerous variations of this process (such as clear that the contamination came from ·the a bake in ozone) were used at one ' time or other. nickel base rather than from the oxide (barium SECONDARY EMISSION strontium carbonate, sprayed to a thickness of 1.3 0.002 inch with an amyl acetate binder, was used (799 throughout). The base was a nickel sleeve The secondary emission was measured under DH standardized conditions : 100 volts on the target in most cases) . By a properly chosen forma and 200 volts on the collector ; the primary cur tion schedule, we finally succeeded in achieving rent density on the target was 2.6 milliamperes a constant secondary-emission ratio over several per square centimeter. hundred hours on sliding targets. But even more The secondary-emission coefficient K was de constant results were obtained, on fixed as well fined as as sliding targets, after the nickel sleeve had been le replaced by tantalum. This was done on the K = (1) le- It' assumption that a material that evaporated less easily than nickel should be used. where le and It are collector and target currents. Figure 2 shows life data taken on tube 365 This formula neglects the fraction a of the pri (sliding target) and 368 (fixed target), both of mary electrons that are intercepted by the col lector without ever reaching the target. If this which had a tantalum sleeve as base for the oxide fraction is taken into account, it is found that coating. As a further proof that tantalum sleeves, the "true" secondary-emission coefficient Kt is even when run at overtemperature, cause no K 0 L. Malter, "Thin Film Field Emission," Physical Re related to as defined above by the formula view, 50, 48-59; 1936. pp. July 1, This circuit was designed. and built by H. Beach. K-1 = (Kt-1)(1 -a). (2) v. 6 ELECTRICAL COMMUNICATION • 104 June 1951 www.americanradiohistory.com contamination, the record of tube is offered : 358 which seem to mm1m1ze the importance of This tube was operated at the usual emission barium. From our experiments, we are inclined current of 10 milliamperes for 240 hours,K showing to agree with this latter view. a secondary-emission coefficientof 3.2. From = 3. Discussion 240 to 375 hours, the emissionK was raised to 15 milliamperes, giving = 3.4. Thereafter, the While these experiments prove that the main source of contamina- tion comes from the nickel sleeve, there K: may be some doubt 111mn t1111111 111111111111 whether the nickel 0 100 200 300 400 500 600 700 itself is to blame or HOURS some impurity in it. It may however be Figure 2-Effective secondary-emission coefficient K of two representative tubes with tantalum cathode bases as a function of hours of operation. noted th at nickel reaches a vapor pres- sure of 10-5 milli em1ss1on was raised to 40 milliamperes for 8 K meters of mercury at 1160 degrees centigrade, hours. During this time, values of between 2.3 i.e., slightly above the operating temperature, and 2.5 were measured, but it is almost certain whereas tantalum would have to be heated to that space charge formed between target and 2400 degrees centigrade to reach the same vapor collector. When the emission was again lowered pressure.10 It is, therefore, quite possible that the K to 10 milliamperes, was 2.9 and stayed at this contaminating substance is the nickel itself. value. Similar, though less exacting, experiments Using tantalum, or some other metal with a with nickel always showed a rapid and perma low vapor pressure, instead of nickel, is entirely K practical. It has been argued that oxide cathodes nent decay of to values below 2. on a nickel base have a higher efficiency than One other experiment may be quoted to sup those on other base metals. Our experiments have port the idea that the nickel sleeve is the source not extended in that direction; however, even if of contamination. In a tube of somewhat differ it should be found that the cathode temperature ent construction, it was possible to expose a 799 has to be slightly higher on tantalum than on target to a nickel base without oxide coating. nickel, this would be a small price to pay for the This base was heated to about 900 degrees centi K resulting increase in tube life. grade, dropped within 4 hours from its initial As good results were obtained in fixed-target value of 3.2 to 1.9. tubes, it is seen that no special structure is re A similar experiment was performed in which quired to prevent exposure of the target to the barium was evaporated. This, too, caused a rapid cathode during formation. This may be stated decay in secondary emission. The decay caused somewhat more generally as follows : Any con by an oxide cathode has commonly been ascribed taminating agent that might come from the to barium3•4•7•8 while there are also accounts9 cathode during formation will be harmless if it is J. B. Johnson, "Secondary Electron Emission from either not evaporated below the cathode-forming Physical Review, Targets7 of Ba-Sr Oxide," v. 73, pp. 1058- temperature (1100 degrees centigrade) or else if 1073 ; May 1, 1948. it does not react with the target and can be J. L. H. Jonker and A. J. Overbeck, "Application of Wireless En re-evaporated below the target outgassing tem Secondarys Emission in Amplifying Valves," gineer, v. 15, pp. 150-156; March, 1938. perature (about 600 degrees centigrade). 9 G. E. Moore and H. W. Allison, "Thermionic Emission Data taken from S. Dushman, "Scientific Foundations from Thin Films," Physical Review, v. 77, pp. 246-257; of 10Vacuum Technique," John Wiley and Sons, Inc., New January 15, 1950. York, New York, and London, England ; 1949: pp. 745-751. June 1951 ELECTRICAL COMMUNICATION 105 • www.americanradiohistory.com Miniature Direct-Wire Remote Control at Loch Sloy Power Station HE OPENING of the Loch Sloy, Scot arrangement that would be almost impossible land, power station put into service a with the heavy cables and large control switches new system for remotely supervising of the old type. andT controlling the operation of an entire hydro The use of telephone techniques offers, in addi electric installation. tion to economy of space and civil engineering, The miniature direct-wire control system used a considerable saving in cost in the manufacture employs telephone-type apparatus and wiring to and installation of telephone cable over that of operate interposing relays on the turbine and vulcanized-india-rubber cable, for in this case the switchgear panels. These relays are connected to switchgear is located about three-quarters of a the circuits of the normal "close" and "trip" coils mile from the power house. usually associated with power-generating equip At Loch Sloy, a generator control desk and a ment. The control system, including the inter mimic diagram afford full facilities for the super posing relays, operates at 50 volts and the result vision and control of four 32-megawatt turbo ing small direct currents are easily carried by alternators and six 132-kilovolt power lines. light 20-pound-per-mile conductors that are There is a control panel for each of the four readily marshalled into cables and run to the operating room. The control keys and indicators Power house and conduits carrying the water that drives the four generators at Loch Sloy. are small and have been assembled in a compact www.americanradiohistory.com Control room at Loch Sloy power plant. A single engineer has complete control of the entire installation including four 32-megawatt generators and six 132-kilovolt power lines. generators, two alarm-display panels, and a 20- and the water and oil systems of the turbines in line cordless private branch telephone exchange mimic. The lines and symbols are in suitable in the control desk. colours on a neutral background and the circuit Each generator panel carries speed and voltage breakers and isolators are identified by three control keys, a circuit-breaker control key, and colour single-aspect lamp indicators. four indicating meters for output power, power The center wall panel mounts six 6-inch meters, factor, rotor voltage, and rotor current. viz., megawatts, megavars, synchroscope, fre Two alarm-display panels are accommodated quency, incoming volts, and busbar volts. Indi in the wings of the desk, one for the generators vidual feeder currents are indicated on 3-inch and the other for the power lines. There are 1 7 instruments set in the feeder lines on the diagram. alarms for each generator and a fifth group con The control engineer sitting at the desk has full tains miscellaneous alarms. There are 6 alarms command over all four generators and can syn for each of the 6 feeders and 12 additional alarms chronise, adjust loading, or switch them without for the bus sections and line protection. Each leaving his chair. All turbine, generator, and alarm lamp is wired through a relay to the alarm switchgear protective devices are continuously originating contacts on the associated protective monitored and any abnormality is instantly indi device and, on the occurrence of an alarm, the cated. Thus, one engineer has. at his finger tips lamp flashes and a bell rings. Actuation of the full control of the entire electrical output of the key for the group silences the bell and steadies station. the lamp. A total of 168 alarms is catered for on The miniature direct-wire remote control sys the desk, of which 121 are in use at present and tem and the supplementary direct-wire alarm the remainder are in reserve for future lines. equipment were developed, produced, and in The wall-type indication and control diagram stalled by Standard Telephones and Cables, shows the electrical arrangement of the station Limited. June 1951 • ELECTRICAL COMMUNICATION 107 www.americanradiohistory.com Television-Link Sound Diplexer L. STASCHOVER H. G. MILLER By and Federal Telecommunication Laboratories, Incorporated ; Nutley, New Jersey N TRANSMITTING television programs The amplitude of this sound carrier, which is ± from a studio to a transmitter that may be frequency modulated over a range of 50 kilo remotely located in an inaccessible place, it cycles, does not exceed 10 percent of the equiva isI common practice to use a coaxial line or a radio lent picture-signal white level so there will be no link for the picture signal and wire lines for the crosstalk into the picture channel. The rack-mounted model of the submitter for fixed-station use. This unit gooerates the 5-megacycle subcarrier. sound program. The use of two separate media There are two sources of crosstalk from the increases costs as well as the susceptibility to picture channel to the sound channel. One of program interruption for a failure of either pic these is through amplitude modulation of the ture or sound is equally damaging. A method of sound channel from the picture channel, which transmitting the sound program over the stand may result from intentional nonlinear amplifica ard studio-to-transmitter radio link has, there tion in the link system. Such modulation may be fore, been developed. removed by limiting action in the sound receiver. In the United States, the video-frequency The second cause of crosstalk is nonlinear time band is 4.5 megacycles wide. This permits a delay. This time or phase nonlinearity in fre sound subcarrier to be placed at 5 megacycles. quency modulation is comparable to amplitude PICTURE PICTURE LINK LIHK SUBHITTER SUBCEIVER SOU HO TRAHSHITTER RECEIVER SOUND Figure supplementary submitter and subceiver transmit the sound program over the studio-transmitter linL 1-A 108 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com put signal 18 decibels above 1 milli watt into impedances of 600, 150, and SO ohms, either balanced or un- balanced. The low-pass filter in the subceiver is flat from below 20 cycles out to 12,500 cycles with a sharp cutoff above that frequency. Maximum attenuation occurs at the horizontal-line frequency of 15,750 cycles. The submitter is designed for zero over-all gain, normally accepting and delivering 2-volt peak-to-peak signals on 75-ohm lines. However, the audio frequency input may vary between + 10 and - 10 decibels referred to 1 milliwatt and normal output still be The portable model of the submitter is intended for mobile-station obtained. use. It measures 14 by 9 by 8 inches and weighs 16 pounds. A 5-megacycle discriminator samples the frequency-modulated output of the submitter and produces a direct-current nonlinearity in amplitude modulation. Careful error voltage that holds the generator to the re design of the radio- and intermediate-frequency quired center frequency. sections of the entire system will keep this cause Residual crosstalk depends largely on the phase of crosstalk within satisfactory limits. characteristics of the studio-transmitter relay The subcarrier transmitter for the sound chan link. The root-mean-square value of this cross nel is called the "submitter." As may be seen in talk can usually be reduced readily to at least SO Figure 1, the picture and the sound electrical decibels below rated output of the sound channel signals are supplied to it. It then generates the and compares favorably with intercity telephone 5-megacycle subcarrier and frequency modulates circuits. it with the sound pro gram. This sound chan nel is combined with the picture channel and supplied to the link transmitter. The demodulated output from the link receiver goes to the subcarrier receiver or "subceiver," which filters the sound chan nel from the rest of the band, amplifies and limits this 5-megacycle signal,and demodulates it in a balanced dis criminator. A low frequency filterand an In the subceiver, the video- and audio-frequency audio-frequency am signals are separated. plifier deliver an out- June • ELECTRICAL COMMUNICATION 109 1951 www.americanradiohistory.com Transmission-Line Balance Indicator for Transmitters GEORGE T. ROYDEN By J,fa ckay Radio and Telegraph Company; New York, New York BALANCE indicator has been designed to simplify adjustment of the output circuits of a radio transmitter to produce balancedA currents in a two-wire transmission line. This indicator is particularly useful with transmitters having one tube in the output stage and feeding power through an Alford network to a two-wire transmission line as illustrated in Figure 1. Description 1. The balance indicator is contained in a small aluminum box. There are a selective circuit, a germanium-crystal rectifier, a milliammeter, and switches for sensitivity and frequency ranges. Two bushing insulators are mounted on top and support the terminals through which the device is connected to the radio-frequency transmission line. Each bushing insulator also supports one plate of a small capacitor (approximately 1 micromicrofarad), the other plate being inside the box. Therefore, each line is connected through these small coupling capacitors to the selective circuit. A frequency calibration chart is mounted on the back. A wooden handle on the bottom facilitates use. On each terminal, there is mounted a wire with a hook on the end for hanging on the open-wire transmission line. Type BNY-1197 balance indicator. The selective circuit includes a variable capacitor and a coil having taps connected to the 2. Theory of Operation frequency range switch. A resistance voltage divider is connected through a small fixed A two-wire transmission line is balanced when capacitor in shunt with the selective circuit. The the current flowing in one wire is equal to that sensitivity switch connects the crystal rectifier flowing in the other wire and there is a 180- to the desired position on the voltage divider. degree phase displacement between these cur The rectified current is indicated by the meter. rents. To feed such a bala.nced line properly from The frequency range is from 4 to 30 megacycles a single-ended power amplifier stage requires per second. The box is S! inches wide, 8i inches accurate adjustment of the interconnecting high, and 4i inches deep (14 by 21 by 11 centi circuit. meters). The handle extends 6 inches (15 centi The Alford network, usually employed for meters) below the case and the terminals extend this purpose, divides the output current precisely 2i inches (6 centimeters) above the case, making in half, part going directly to one wire of the the over-all height 16! inches (42 centimeters) . transmission line and the other part being dis The weight is 4 pounds (1.8 kilograms). placed by 180 degrees in phase before reaching no ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com the other wire. Two equal inductances in series After connecting the balance indicator to the and a variable capacitor in shunt to ground from line, it is tuned to resonance at the transmitter the junction of the two coils are usually em frequency. Then the Alford network in the ployed to accomplish this phase shift. transmitter is adjusted so the rectifiedcurrent as OUTPUT CIRCUIT TRANSMISSION LI NE LOAD POWER AMPLIFIER ------, BALANCE I INDICATOR I I I I I I I _r:- ---� �------� � II L------Figure 1-The balance indicator shows the voltage to ground from the midpoint of a capacitive voltage divider connected across the transmission line, thereby facilitating proper ad ustment of the Alford network to produce line currents that are equal and 180 degree out of phase. j Assuming the load to be balanced with respect indicated by the balance indicator decreases to to ground, the voltage from each line wire to zero or a minimum near zero. While the trans ground will be equal when the currents are mitter output is unbalanced, current showing balanced. The two small coupling capacitors are the unbalance flows through the circuit to the connected in series with each other across the case and then through the capacitance between line. Since they are adjusted to be equal, the the case and ground. This capacitance is much voltage across them will be equal. Therefore, the larger than the coupling capacitances and there junction should be at ground potential when fore has little effect on the operation of the the line voltages and currents are balanced. instrument. Recent Teleco mmunication Development Maximum-Amplitude Measuring Set T IS OFTEN DESIRABLE to measure the peak values of positive and negative peaks that persist I amplitude of transients such as occur in the for at least 2 milliseconds. The meters are reset firing of gas discharge tubes, electric welding, to zero manually and the applied voltage may be surges in capacitors, starting currents in motors, either direct or alternating. and surges in semiconductors, as well as ab This portable instrument weighs only 25 normal phenomena that can be converted from pounds (11.7 kilograms) and is energized from mechanical or other forms into electric currents the alternating-current mains. It was introduced or voltages. recently at the Physical Society exhibition in The maximum-amplitude measuring set will accept any voltage not exceeding 10 volts and London by Standard Telephones and Cables, the will indicate simultaneously on two meters with manufacturer. The set was designed by Standard an accuracy within 5 percent the maximum Telecommunication Laboratories. June 1951 ELECTRICAL COMMUNICATION 111 • www.americanradiohistory.com Suppression of Harn1onics in Radio Transmitters GEORGE T. ROYDEN By },fa ckay Radio and Telegraph Company, Incorporated; New York, New York ITERATURE on the subject of sup band between 54 and 88 megacycles, but also pressing harmonic radiation from trans because the radiotelegraph transmitters are mitters was reviewed and the most located for economic reasons at a distance from importantL items are summarized. The amount of large cities where they happen to be at the harmonic suppression required is estimated for fringe of the service area for the television typical conditions. broadcast stations Several circuits that in those cities. Con are favorable for sequently, the radi harmonic suppres ated harmonic sion are described. energy must be ex It is shown that tremely small to coupling two circuits avoid interfering by a common shunt with the weak tele capacitance is more vision signals. With effectivein suppress a large number of ing harmonics than transmitters ser to use inductive viced by only a coupling. Formulas few operators and for estimating the many transmitters harmonic output for being retuned to several typical different frequencies power-ampl ifier Radiotelegraph transmitters may interfere to various each day, it is not output circuits are degrees with television reception. The above "venetian feasible to resort to a included. Several blind" pattern results from strong interference, which if in trap circuit tuned to applications of har creased would prevent synchronization between transmitter the unwanted har and receiver and the picture would "tear" completely. m on i c-su p pres- monic and similar s Ion techniques "tricks" employed by radio amateurs. are discussed J. Review of Literature and the results of some tests are given. The control of harmonics in radio transmitters has been a problem for many years.1* Kaar and Radio transmitters intended for international Burnside2 described transmitters feeding through telegraph communication usually operate with transmission lines to a closed tuned circuit that power outputs between 1 and 50 kilowatts at is coupled to the antenna and very materially frequencies between 4 and 26 megacycles per reduces harmonic radiation. Labus and Roder3 second. For high efficiency, the final stage studied various circuits and showed that a chain operates as a class-C amplifier. Conventional of resonant circuits coupled by shunt capacitances methods, including a high-Q plate tank circuit, with the total volt-amperes being many times have until recently been deemed adequate for the the watts output offers best harmonic sup suppression of the undesired harmonics of the pression. Everitt4 showed how the conventional fundamental frequency. However, a considerable network between the plate and load circuits 7f number of television receivers have recently been can be employed to reduce harmonic radiation. installed in the vicinity of some transmitting He also described how a series inductance· and stations. This is serious, not only because of the capacitance tuned to the harmonic frequency * Numbered references appear in Section on page 120. many harmonics falling within the television 11 112 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com and connected in shunt with the load greatly C. Use a plate circuit for the final amplifier with a ca pacitor connected as directly as possible from plate to attenuates the selected harmonic. Kusonose5 cathode. listed the general principles for attenuating D. Use a grid circuit for the final amplifier with a ca harmonics and showed how a particular harmonic pacitor connected as directly as possible from grid to may be eliminated by tuning to that harmonic cathode. a trap circuit in series with the anode and adjust E. By-pass all power and control leads with a full ing the coupling between the inductances in the network. 7r trap circuit and the main tank circuit to balance · F. Shield all stages, making certain that interstage shield out the chosen harmonic. ing is adequate. Chambers6 and associates described a 500- G. The antenna coupling system should be separately kilowatt broadcast transmitter in which har- shielded. monies are sup- H. Use narrow-band frequency modulation pressed by using instead of amplitude push-pull output modulation. stages, Faraday J. Adjust grid drive , shields, a T-section grid bias, and output harmonic filter (low loading for mm1mum harmonic output. pass), a concentric K. Study layout of out transmission line put circuit of final am with grounded shell, plifier to ascertain the and an impedance probable paths for har matching section monics. Determine the resonant frequency of consisting of series these paths with a grid inductance and dip oscillator and re shunt capacitance arrange to move above for coupling the line 160 megacycles. Explore radio-fre to the antenna. L. A number of q uency fields nearpower This mild form of interference may be caused by radia and control leads with articles by Ter tion from an inadequately shielded oscillator in a nearby a sensitive indicator man1-11 and associ television receiver. Interference from a radiotelegraph tuned to the harmonic ates showed how the transmitter, while similar, will be influenced by keying, frequencies. Continue plate current of a whereas that from another receiver changes only slowly. corrective measures un- class-C amplifier til harmonics are not detectable. tube may be ana- M. Explore radio-frequency fields near output and antenna lyzed to determine the magnitude of each har transmission line for spurious outputs on harmonic monic with respect to the direct current as a frequencies, also harmonics of the oscillator frequency to function of the angle of flow and peak amplitude. determine if further reduction is necessary. Warner12 observed a well-defined indication at N. Insert a low-pass filter in the transmission line between the Federal Communications Commission that the power-amplifier output stage and the antenna tuner. all services are going to be under the necessity P. Use a push-pull output stage. of devoting a lot more attention to the elimina tion of spurious emissions, harmonics, hash, etc. Grammer21 showed a filter consisting of a trap He said that television is here to stay and all circuit resonant at a harmonic frequency in amateurs in the vicinity of a television receiver series with the inner conductor of a shielded must emit only pure waves. concentric line linking the power amplifieroutput There have been many articles13-29 published to the antenna coupling system and a shunt in QST, as well as other magazines intended for capacit-or all within a shielded enclosure. amateurs, showing how to reduce or eliminate Reinartz22 had applied the method proposed harmonics. The following procedures have been by Kusonose5 of a trap circuit tuned to a har advocated for amateur transmitters : monic and coupled to the tank circuit so as to Do all frequency multiplying at a low power level. A. cancel out the unwanted frequency in trans B. Use class-B amplifiers up to the final stage. mitters intended for amateur operation. He June 1951 ELECTRICAL COMMUNICATION 113 • www.americanradiohistory.com added an innovation by connecting a small Cairo in 1938 agreed on a limit of -40 decibels capacitor between the hot side of the harmonic for any harmonic but not in excess of 200 milli trap circuit and ground with the length of lead watts. These figureswere confirmed at the meet adjusted to provide minimum interference. ing at Atlantic City in 1947. However, the Although Reinartz did not explain how it func� discussion in the next several paragraphs in tions, it is probable that a series-tuned circuit to dicates that this limit is far above that necessary ground is provided for a higher harmonic. to avoid interference with reception near high Patterns22 illustrating the distortions to the power transmitters. For the purpose of illustrating how to deter television picturesignals causedhelp to bytrace various the sourcetypes of mine the amount of suppression that is necessary, interfering interference. the conditions at Brentwood, New York, will be Murdock25 employed adequate shielding, a considered. This station is 40 miles from New high-capacitance tank circuit in the power York City. Measurements published by Gold amplifier stage, link-circuit coupling to antenna smith30 indicate the fieldstrength at this distance tuning circuit using coaxial cable with shielded to be above 500 microvolts per meter from only coupling at each end, and a pair of coaxial the strongest television station. transmission lines with shielded coupling to the Reinartz22 showed that an interfering field antenna tuning circuit. strength below 5 microvolts per meter will not Mix26 described a transmitter in which there disturb a television signal of 500 microvolts per are capacitors from plate to ground with short meter, which is considered by the Federal Com paths, adequate shielding, filtered power leads, munications Commission to be the limit of and link coupling from the push-pull power service area in rural sections. Residences on amplifiertuning unit.stage to the separately shielded antenna which television receivers may be installed are . located only 2500 feet from the transmitters at Seybold,27 Pichitino,28 and Grammer29 de Brentwood. The intervening path is over poor scribed the design and application of m-derived soil and through trees that introduce attenuation. low-pass filters for harmonic reduction. Bullington31 gives an approximate formula for The use of a push-pull amplifier with tubes the attenuation between low antennas far apart. Ji 47rh.'hr' operating in class B was suggested by Romander = = A (1) as an effective method to avoid the generation of Eo A.d ' harmonics. Such a proposal could be readily where adapted in the design of a single-sideband trans E =attenuated field in volts per meter mitter and would be useful where both telephone E =radiated field in free space and one or more telegraph channels were to be h.0'= h.+jho radiated simultaneously as advocated by h. =height of sending antenna in meters Rabuteau. ho = h,+minimumjho effective height 2. Degree of Harmonic Suppression hr' = h, =height of receiving antenna in meters A report issued by the Institute of Radio A.=wavelength in meters Engineers recommended any har =distance in meters. in 1930 that d monic radiated from a broadcast transmitter Substituting should not exceed 0.02 percent of the field h. = 3 meters strength of the fundamental and never exceed h, = 10 meters 500 microvolts per meter at a distance of 1 mile. ho = 1 The meeting of the International Consultative 3 meters (from Bullington,3 Figure 3 for Committee on Radio in Copenhagen in 1931 poor soil at 40 megacycles) A.=4.3 meters (at 69 megacycles) fixed a limit of 0.05 percent for broadcast = frequencies and 0.1 percent for lower frequencies. d 760 meters, Limits were not established for frequencies above attenuation is found to be 3 megacycles until the meeting of the Inter 4 4.2X 10.4 . A= 7r =0. 168 national Consultative Committee on Radio at 4.3X760 114 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com The radiated power is related to field strength For mobile services on frequencies between 7T and distance by and 76 megacycles, the 80-decibel reduction previously indicated to avoid interference to P=Ea2d2/30, watts. (2) television reception should be adequate. Substituting 3. Measuring Apparatus Eo = E/A =5X10-6/0.168 =30X10-6 d = 2500 feet = 760 meters, Honnell and Ferrell32 measured the harmonic power output by coupling a balanced shielded the radiated harmonic power is calculated to be loop the transmission line from the trans fo mitter to the antenna and connecting the loop P =302 X 10-12 X 7602/30 = 17.4X 10-6 watts, through a second tuned circuit to a receiver having a calibrated attenuator in the inter which is 92 decibels below a power output of 30 mediate-frequency amplifier circuit. The system kilowatts at the fundamental frequency. The was calibrated by measuring the power sent intervening trees will introduce an attenuation through the line to a dummy load from a of about S decibels. For television reception in separate transmitter operating at the frequency that area, a high-gain antenna is recommended of the harmonic being measured. Both series and and, since most residences are situated at one parallel currents could be measured in the open side of the Brentwood transmitters, the direc tivity of the television antenna may be expected wire transmission line. Grammer33 improved on the simple tuned to provide 10 decibels or more discrimination. circuit with rectifier and meter by using a tuned Therefore, the harmonics between 54 and 88 circuit coupled to a type-955 acorn triode with megacycles should be suppressed to at least 80 adjustable plate voltage so the tube detector is decibels below the fundamental to avoid inter operated just below oscillation. The equivalent ference with television reception. Mobile services on frequencies between 30 and of a high Q is thereby obtained. SO megacycles are not active at present in the A high-Q circuit has been found convenient immediate vicinity of Mackay Radio's trans for measuring the voltage of the harmonic on the mitter stations. Therefore, it would seem transmission line or at some accessible point in reasonable to suppress the radiated harmonic the transmitter. The inductor consists of a single power at these frequencies enough to avoid turn of copper 1-inch wide. and 0.03-inch thick interference within a radius of S miles. The (25 by 0. 7 millimeters) connected at its center the selected sensitivity and characteristics of the frequency to the shield. This is tuned to modulation receivers used for mobile service are harmonics by a variable air capacitor connected such that an interfering signal of 0.5 microvolt across the ends of the inductor. One end of the per meter may be tolerated. Assuming an inductor is connected through an air capacitor antenna height of 3 meters at the receiver and of about 1 micromicrofarad and a bushing substituting in (1), the attenuation is insulator to the external circuit. A germanium diode is connected through a multiposition 4'l1"4.2X4.2 switch to taps on the opposite side of the A = =0.003 7 ' 7.S X 8000 inductor. The positions of thes-e taps are chosen to give all, one-half, one-fourth, and one-eighth and the harmonic power is calculated from (2) of the voltage across that half of the inductor to be thus providing several voltage ranges. The diode 0.52x 10-12 80002 x =0.039 watts, feeds a 50-microampere indicating instrument. p 0.00372 X30 All components are enclosed within a suitable which is 59 decibels below 30 kilowatts. There aluminum box. The device is calibrated by fore, harmonics between 30 and SO megacycles means of a vacuum-tube voltmeter and a suitable should be reduced to a level at least 60 decibels source of radio-frequency power. below a fundamental power output of 30 Another device employing three tuned circuits kilowatts. has been found more selective. For convenience June 1951 ELECTRICAL COMMUNICATION 115 • www.americanradiohistory.com the ganged variable inductors and the ganged found that harmonic currents flow in parallel into an open-wire transmission line. A typical variable capacitors are driven from a common 6 transmission line consisting of two size (Ameri shaft. The first circuit is coupled to the trans 12 mission line through a very small capacitor, can Wire Gage) wires spaced inches apart and 1 about micromicrofarad, and the succeeding circuits are coupled by small capacitors. The radio-frequency voltage in the final tuned circuit is rectified by a germanium diode and the re�tified current passed through an adjustable resistance to a 50-microampere indicating in A c e R strument. The light weight and small size of these devices permit t11em to be hung from the transmission line. The coupling capacitor is designed to with l stand the peak line voltage with an ample factor Figure 1-Transmitter output stage connected through a of safety. network to the load. 7r Calculation of Harmonic Content 4. 10 H feet above ground has an impedance for 330 To obtain high efficiency, the final amplifier is parallel currents of ohms. This value is taken C, usually operated in class that is, the plate for calculating harmonic power. Let current flows for less than half of the cycle. The W=30 000 watts output waveform of the plate current is a pulse which = 9 000 Terman34 has shown to follow approximately a V volts, direct voltage 3/2 I= 5 power law. It is reasonable to assume the amperes, direct current 120 f = 20 plate current to flow during degrees for the megacycles purpose of estimating the harmonic content. For A= 80 k ohms this angle, the ratio of the harmonic current to 110 B= ohms the direct current is determined from Terman's 86 1.4, 0.85, 0.27 C= 30 Figure to be and for the second, ohms R= 150 third, and fourth harmonics, respectively. ohms. For well-shielded circuits of good design, the harmonic voltage at the output terminals of the Then, for the second harmonic transmitter may be calculated approximately in n=2 terms of the above ratio, the direct current, and k=1.4 the circuit constants. For example, calculations e = 80 X30X5X1.4/23(110-20-7.5) will be shown for a trnnsmitter with a output =25 1. 7r volts, network, as shown in Figure For the harmonic current, the impedance of and the harmonic power leaving the transmitter A A/n, C becomes that of B becomes Bn, and is approximately C/n, n becomes where is the order of the har w=e2/H S monic. The impedance of is negligible. The =252/330=1.9 watts, harmonic voltage appearing across the output 42 capacitor can be shown to be which is decibels below the assumed power output. Similar calculations show the third (3) harmonic power to be 54 decibels below the out 30 000 put power of watts. More harmonic suppression can be obtained The radiated harmonic power depends on many by two circuits coupled together by a common factors such as the radiating properties of the shunt capacitance (sometimes called a double 7r transmission line and the antenna. It has been network) as shown in Figure 2. For this circuit, 116 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com C (3) (5) it is necessary to choose so that critical Analysis of and indicate that there is coupling results. This is approximately little to be gained so far as harmonic suppression A is concerned by designing for high-Q circuits in C= (4) the output stage. This is contrary to the con (r+rR/ F2)i ' (3), clusions expressed by several authors. In (B -A/n2 - C/ n2) the expression in the de r A where =ratio of desired plate load to output nominator is approximately equal to in the R. C, I, k, n, resistance numerator, leaving the other factors H The harmonic voltage appearing across the and to determine the harmonic-power output, output capacitor can be shown to be ap so that proximately c212k2 . w= , (6) ACFik n6 e= (5) H approximately. n5(B -A/n2 - C/n2) (D - C /n2-F/n2)° (5), D-C/n2 Similarly in the expression -F/n2) F It is desirable to choose an output capacitor in the denominator is nearly equal to such that the Q of the output circuit is higher 2 3. in the numerator, so that than but preferably not over For a value of 2.6, c212k2 . w= , (7) n10H F= 150/2.6 = 58 approximately. r=7.5 In a practical circuit, the impedances of the C = 10.3 circuit components and their connecting leads B=80+14=94 are usually different at harmonic frequencies than the assumed impedances. Therefore, the D=26+50= 76. (6) approximations introduced in deriving and (7) may be tolerated. For the second harmonic Development SO X 10.3X58X5X1.4 5. e 25(94-20-3) (76-3-15) Many of the transmitters presently in service =2.6 volts, use a secondary circuit that is inductively coupled to the plate tank circuit, the trans and the harmonic power leaving the transmitter mission line being fed from the secondary circuit. is approximately For such a circuit, the harmonic voltage induced w = 2 62/3 0 = 0.02 into the secondary is . 3 watts, E =j27r nfMl , (8) 62 n n which is decibels below the assumed output. where Similarly, the third harmonic power is found to 82 30 be decibels below kilowatts. j= -1t n =order of harmonic + f = frequency in cycles per second M =mutual inductance in henries I component of current flowing in primary at e n= v R the harmonic frequency in amperes. c F For shunt capacitance coupling as shown in l Figure 2, the harmonic voltage applied to the secondary circuit is Figure 2-Transmitter output stage using two In (9) works having a common shunt capacitance. net j27rnfC' 'Ir Vn= June 1951 ELECTRICAL COMMUNICATION 117 • www.americanradiohistory.com where C=coupling capacitance in farads, and suppression of harmonics with a mm1mum the other terms are as before. The circuits are number of components and controls. When the normally adjusted for critical coupling, which components are properly located with respect to occurs when each other, to the anode, and to ground, it has been found that the harmonic voltage is some-' what lower than calculated as above for the low where order harmonics. The discrepancy is probably Rp =equivalent series resistance of primary cir due to distributed capacitance of the series coil cuit in ohms B making its reactance higher than Bn, also the R. =equivalent series resistance of secondary inductance in the leads, which connect the shunt circuit in ohms. capacitors to ground and to the circuit, thereby (8), making their reactances less than A/n and C/n. Combining (9), and (10) and solving for the Measurements indicate an attenuation better ratio of harmonic voltage for inductive coupling than SO decibels. with respect to that for shunt capacitive coupling The network is recommended for use where 7r (11) the station is isolated so that only a moderate 12 suppression of harmonics is required. This indicates an improvement of decibels for the second harmonic, 19 decibels for the third, Double Output Circuit 24 7. and decibels for the fourth harmonic. 11: Measurements of the harmonic voltage at the The double output circuit, as shown in 2, 7r output terminals of several transmitters having Figure is recommended where the station is so inductively coupled output circuits revealed the located that considerable suppression is required harmonics to be somewhat higher than cal to avoid interference by harmonics. Several culated. Some harmonics were comparable with transmitters at Brentwood have been modified the calculated harmonic voltage at the anode, to use this output circuit. Measurements indicate indicating no attenuation in the circuits following the power at a harmonic frequency to be better the anode. Further study indicated that these than 70 decibels below the power at the funda abnormally high harmonic outputs were due to mental frequency. stray capacitance beginning at the anode, or To obtain best performance, the coupling components connected to the anode, and ter capacitor should be located so its grounded minating on components connected to the terminal is close to the point where the cathode transmission line. The harmonic current flowing circuit is grounded. The loop enclosed by the S, through this stray capacitance may pass over blocking capacitor the series coil B, the connecting leads having enough inductance to coupling capacitor C, and the plate tank ca approach resonance at some harmonic. pacitor A (usually the internal plate-grid tube It was found that this type of coupling via capacitance, neutralizing capacitance, and stray stray capacitances could be overcome by means suffice added capacitances without tank-circuit of a Faraday screen between the plate tank capacitance for high-power tubes at frequencies 8 circuit coil and the coupling coil together with above approximately megacycles) has an adequate shielding for the output circuit. The important effect on the highest frequency to Faraday screen should be designed so the wires which the tank circuit may be tuned. are grounded at but one point and the length of The secondary circuit must be adequately each wire should not exceed 0.1 times the wave shielded from the primary or plate tank circuit. length of the highest harmonic frequency to be The coupling capacitor should be placed so the stopped, also the gap between wires should not lead from it to the secondary circuit has a exceed 0.03 times the wavelength. minimum exposure in the tank-circuit com 4 -:t-Network Output Circuit partment, preferably not more than inches. 6. The loop enclosed by the series coil D, the shunt F, The network, as shown in Figure 1, is the 7r capacitor and the coupling capacitor C, most economical way to obtain a reasonable together with the interconnecting leads and 118 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com ground return has an important effect on the wavelength for the harmonic frequency to be maximum frequency to which the secondary suppressed less about 6 inches for correction of circuit can be tuned. Therefore, the area of this end effect. These are usually connected directly loop must be kept as small as possible. to the output terminals of the transmitter, some F The capacitance in the secondary circuit is times two or three sets of stubs being employed arbitrarily chosen so that the loaded Q is higher for as many harmonics. As an alternative (or in 3. than 2 but preferably not over This is done to addition to stubs at the transmitter when more minimize circulating current and avoid un attenuation is desired) two sets of stubs in necessary losses. conjugate relationship are applied to the trans Low-Pass Filters mission line. These should be located as close to 8. the transmitter as feasible because standing A number of low-pass filters cons1stmg of a waves for the harmonic frequency are set up on constant-k intermediate section and shunt m that portion of the line between the stubs and the derived end sections have been constructed. transmitter. Radiation in certain directions will These are designed for operation over a narrow take place from this portion of the line. An band of frequencies. The terminal half-sections attenuation of 10 to 20 decibels may be obtained are designed so the peaks of attenuation are near by means of stubs. the second and third harmonic. The inter 10. Co nclusions mediate section provides ample attenuation for the nigher-order harmonics. The elements are in The recent increased use of frequencies above duplicate for operation with a balanced 600-ohm those now used for fixed point-to-point service line. Fixed vacuum capacitors were employed often requires modification of old transmitters for the shunt elements, four being required for to suppress harmonics and avoid interference each filter. Short lengths of large coaxial cable with mobile and television services. A reasonable were used as capacitors in shunt with the end suppression may be obtained by employing a 71" coils. All coils were wound with size 6 (American network in the output circuit together with Wire Gage) solid copper wire. The filter was adequate shielding of the transmitter and its enclosed in an aluminum case with shield supply leads. If a particular harmonic still has partitions between sections. sufficient strength to cause interference, a spot Care must be taken with transmitters having filter or stubs may be applied. exceptionally high stray capacitance between the For general use in a locality where a high plate circuit and the output circuit. If the circuit degree of suppression is necessary, the double 71" including this stray capacitance and the filter network is recommended for the output circuit. has series inductance approximating resonance, a This has the advantage of providing adequate high circulating current may cause failure. suppression but at the expense of several addi Sp ot Filters and Stubs on Transm,ission tional circuit components and their controls. 9. Line Before closing, it must be pointed out that not all interference is due to harmonics. Confirming As an expedient, a simple filtermay be applied our own experience, Kiser35 has pointed out that to the transmitter to attenuate one particular most of the interference complaints investigated harmonic. This consists of a capacitor and coil by the Federal Communications Commission in in series with each other from each line to the New York area were found to be due to a ground. To be effective these must resonate for frequency other than that of the television the harmonic frequency to be suppressed. Some carrier, that is, the receiver is at fault. Some are times only one coil is used between ground and due to poor image-frequency rejection and can be the junction of two capacitors that bridge the avoided by adding a tuned circuit (booster) be transmission line. An attenuation of 10 to 20 tween transmission line and receiver. Others are decibels has been obtained. due to insufficient shielding to keep a high Open-ended stubs have been applied to the power signal out of the intermediate-frequency transmission line for the purpose of supw.essing amplifier and can be corrected by inserting · a harmonics. The length is made one-quarter booster or a high-pass filter between the antenna June 1951 ELECTRICAL COMMUNICATION 119 • www.americanradiohistory.com 14. M. Seybold, "Curing Interference to Television Recep and the receiver. Although a good technician can tion," QST, v. 31, pp. 19-23; August, 1947. often correct these faults, a lot of trouble can be 15. G. Grammer, "Interference with Television Broad avoided by arranging for only well-designed casting," QST, v. 31, pp. 24-30; September, 1947. television receivers to be sold for use in areas 16. P. S. Rand, "TVI Can be Reduced," QST, v. 32, pp. around high-power point-to-point radio trans 31-36; May, 1948. mitting stations. 17. P. S. Rand, "More on TVI Elimination," QST, v. 32, 11. References pp. 29-32, 114; December, 1948. 18. QST, 1. F. A. Kolster, "Re-enforced Harmonics in High Power G. Grammer, "TVI from 21 Mc," v. 32, pp. 20- 22; December, 1948. Arc Transmitters," Proceedings of the I.R.E., v. 7, pp. 648-651; December, 1919. 19. P. S. Rand, "The 'Little Slugger,' " QST, v. 33, pp. 11- 17; February, 1949. 2. I. J. Kaar and C. J. Burnside, "Some Developments in Broadcast Transmitters," Proceedings of the I.R.E., 20. F. Q. Gemmill, "Harmonic Suppression in Class C v. 18, pp. 1623-1660; October, 1930. Amplifiers," QST, v. 33, pp. 28-33; February, 1949, and v. 33, pp. 34, 94; April, 1949. 3. J. W. Labus and H. Roder, "Suppression of Radio 21. G. Grammer, "Pointers in Harmonic Reduction," Frequency Harmonics in Transmitters," Proceedings QST, v. 33, pp. 14-22 ; April, 1949. of the I.R.E., v. 19, pp. 949-962 ; June, 1931. 22. "TVI Patterns," QST, v. 33, pp. 43-45 ; May, 1949. 4. W. L. Everitt, "Output Networks in Radio-Frequency Power Amplifiers," Proceedings of the I.R.E., v. 19, 23. "High-Pass Filters for TVI Reduction," QST, 33, 46; May, 1949. pp. 725-737; May, 1931. p. v. 24. "TVI Tips," QST, v. 33, p. 44-45 ; June: pp. 64-65 ; 5. Y. Kusonose, "Elimination of Harmonics in Vacuum July: pp. 45, 86; August: pp. 55, 108; October: 1949. Tube Transmitters," Proceedings of the I.R.E., v. 20, pp. 340-345 ; February, 1932. 25. C. E. Murdock, "TVI Reduction-Western Style," QST, v. 33, pp. 24-27, 82; August, 1949. L. G. W. H. 6. R. J. Williamson,A. Chambers, E. A. F.Leach, Jones, and J. A.Fyler, Hut cheson, 26. D. H. Mix, "Harmonic Reduction in a 500-Watt All "WLW 500- Kilowatt Broadcast Transmitter," Band Rig," QST, v. 33, pp. 21-28; November, 1949. Proceedings of the I.R.E., v. 22, pp. 1151-1180; 27. M. Seybold, "Design of Low-Pass Filters," QST, v. 33, October, 1934. pp. 18-24; December, 1949 7. F. E. Terman, D. E. Chambers, and E. H. Fisher, 28. A. M. Pichitino, "A High-Attenuation Filter for "Harmonic Generation by Grid Circuit Distortion," Harmonic Suppression," QST, v. 34, pp. 11-14, 104; Electrical Engineering, v. 50, pp. 966-967; De January, 1950. cember, 1931. 29. G. Grammer, "Eliminating TVI with Low-Pass 8. F. E. Terman and J. H. Ferns, "The Calculation of Filters," QST, v. 34, pp. 19-25; February: pp. 20- Class-C Amplifier and Harmonic Generator Per 25, 104; March: pp. 23-30; April: 1950. formance of Screen Grid and Similar Tubes," 30. T. T. Goldsmith, R. P. Wakeman, and J. D. O'Neill, Proceedings of the I.R.E., v. 22, pp. 359-373; March, "A Field Survey of Television Channel 5 Propaga 1934. tion of New York Metropolitan Area," Proceedings 9. F. E. Terman and W. C. Roake, "Calculation and of the I.R.E., v. 37, pp. 556-563; May, 1949. Design of Class-C Amplifiers," Proceedings of the 31. K. Bullington, "Radio Propagation at Frequencies l.R.E., v. 24, pp. 620-632; April, 1936. above 30 Megacycles," Proceedings of the I.R.E., 35, pp. 1122-1136; October, 1947. 10. F. E. Terman, "Analysis and Design of Harmonic Generators," Transactions of the American Institute 32. P. v.M. Bonnell and E. B. Ferrell, "Measurement of of Electrical Engineers, pp. 640-645; 1938. (Electrical Harmonic Power Output of a Radio Transmitter," Engineering, v. 57; November, 1938.) Proceedings of the I.R.E., v. 22, pp. 1181-1190; October, 1934. 11. F. E. Terman, "Radio Engineers' Handbook," Mc Graw-Hill Book Company, New York, New York; 33. G. Grammer, "The Regenerative Wavemeter,'' QST, 1943. v. 33, pp. 29-31; November, 1949. 34. F. E. Terman, "Radio Engineers' Handbook," 1st 12. K. B. Warner, "Editorial," QST, v. 32, pp. 11, 12; Edition, McGraw-Hill Book Co. ; 1943: p. 461. May, 1948. 35. L. Kiser, "TV Interference Problems," Radio \V . 13. G. Grammer, "Keeping Your Harmonics at Home," Electronics, v. 21, pp. 36-37; January, 1950: and QST, v. 30, pp. 13-19; November, 1946. QST, v. 34, pp. 44-45 ; February, 1950. 120 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com Quantization Distortion in Pulse-Count Modulation with Non uniform Spacing of Levels* By P. F. PANTER DITE and W. Federal Telecommunication Laboratories, Incorporated; Nutley, New Jersey T IS SHOWN that in a pulse-count-modula posed about zero voltage level, but otherwise tion system the distortion due to quantiza placed in an arbitrary manner in the interval tion can be minimized by nonuniform spac ( - V, V) . The problem of minimizing the dis ingI of levels. Equations are derived for a:n tortion of the quantized signal by properly arrangement of nonuniform level spacing that spacing the levels will be considered now. De- produces minimum distortion. It is also shown that minimum distortion is significantly less than distortion resulting from uniform quantization when the crest factor of the signal is greater than four. J. Intro ductio n - It h;;is been shown in a previous paper1 that the y .! ------process of quantization in a pulse-count-modula ------2 tion system introduces distortion. In that paper, y, equations were derived for the special case of YJ. equal level spacing. This paper studies the effect w Yo2 of nonuniform 0 level spacing with the view of ob � y -J. :J taining minimum distortion. 2 It is shown that by taking the statistical prop � Y-1 "' erties of the signal into consideration, the distor y_.! tion can be minimized by a proper level distri 2 / bution, which is a function of the probability / density of the signal. In practice, nonuniform / quantization is realized by compression followed Y- June 1951 • ELECTRICAL COMMUNICATION 121 www.americanradiohistory.com uk where is the distortion in the kth level. The Thus, the condition for making a minimum is (y -yk) yk Yk+iu,. Yk-l· error of the transmitted signal is and that lie half-way between and P(y) y. the probability density of the signal Yk+! = Yk+Llyk, (5) First,(2) derive auk relation between the various Yk-t = Yk -.:lyk. } y's in so that in the kth level is a minimum. (3), Substituting these values in it follows (6) The total mean-square distortion voltage is found by summing over all levels, giving 2 n ua= L P(yk).1yk3• (7) 3 -n Now, it will be proved that the mean-square ud Uk distortion is a minimum when is constant, independent of the kth level. By the definition of an integral, we may write YK INPUT NONUNIFORM SPACING y Figure 2-Compression curve for (8) nonuniform level spacing. Yk =Ll.yo+2Ll.y1+· · · +2Ll.Yk-1+Ll.yk K where is a constant, since the integral is a func 1 2 2 1 2V µk = Pl(v,)Ayk ; = + + · ··+ + tion of only its limits. Let then 2VK[ Pl(yo) Pl(yi) Pl (Yk-1) P(yk) ] N' (7) and (8) become, respectively, 1 f.. --dz 1 P!(z) :. y �A rz dz (9) J o P!(z) 1 dz V Pl(z) I where z = Vand y= V. J: and n K = L: µ,.. (10) P(y) Suppose that the levels are so close that -n may be considered as nearly constant over the P(Yav) The problem is now reduced to minimizing the region of integration and equal to sum of cubes subject to the condition that the where sum of the variables is a constant. From Lagrange's method of undetermined (9) multipliers, it follows that is a minimum when K µ_n = µ-n+1= · · · =µn-1 =µn = • (11) (3) 2µ+1 From this, it follows (12) or or 2 K3 (4) uk= 3 (2n+1)3" 122 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com z 0 V, The total minimum distortion power By letting vary from to y will describe a 2. z 2 K3 curve as shown in Figure As takes on the = O z 2 V/N · zk = 2kV/N· O"m 3 2n+1 2 values of Zo = = · · ··Zn = ( ) 2n V/N, , 1 yo O, Y1· · ·yk· · we get the point = ·Yn· Pl(y)dy ( 3) While approximate, this derivation has the ad = 12(2 + 1 1)2 vantage that the levels are obtained quickly. � (f_: ) 3• (12) The relation may be used to make spot P(y) Since is an even function and letting N = checks since 2n+ 1 be the total number of steps, (14) (19) The ratio of the mean-square distortion voltage 2, From Figure zit may be seen that the uniform to the mean-square signal voltage is spacing along the axis is transformed into a non uniform spacing along the y axis. The compressed Pl(y)dy z O"m 2 output is then subjected to uniform quantiza D m =__ v ' tion. To recapitulate, nonuniform spacing of 2 = O" 3N2 LJ: r (15) y2P(y)dy levels may be realized by passing the signal iv through a compressor with a given characteristic (15) and applying unifoqn quantization to its output. Equation gives the minimum distortion re Obviously, this also implies that a corresponding sulting with optimum level spacing.2 expansion is incorporated in the receiver. An approximate method of obtaining levels k may be obtained by writing (where is positive) 3. Logarithmic Co mpressio n A compression curve that is relatively easy to obtain and has been used in practice is the so called logarithmic compression curve. The posi tive half of the compression characteristic 1s given by (20) where µ=compression parameter v1 =input voltage The series may be approximated by an integral V =output v�ltage V2 =maximum input voltage y A 1 dz, (17) k = Jr·o Pl(z) =an undetermined constant. k, V V, where the variable on the right has been changed To find let V = when V1 = so that the z A 2 to to avoid confusion. is a constant of pro maximum values of the input and compressed z V; y= V. portionality so chosen that when = waves are equal. This gives Hence, dz (21) Pl(z)1 y= o (18) . v i. 1 dz v Pl(z) The parameter µ controls the degree of compres i sion and may be chosen so that large changes in This equation was first derived by P. R. Aigrain using the input produce relatively small changes in the slightly different method. 3 1000, 2 output. As shown in Figure for µ = a a June 1951 • ELECTRICAL COMMUNICATION 123 www.americanradiohistory.com 2Ayk = dy 60-decibel change in the input will cause only a Converting this into an integral with 20-decibel change in the output. When µ is large, and using (41) and (46) of the appendix yields = the levels are crowded about zero ; µ 0 corre sponds to uniform spacing. Differentiating, (21) yields ��--,-µ ---v dv1 (28) • (22) log(l+µ) V+µv1 u N where is the mean-square voltage and Vis the As the compressed signal is divided into uni AV = / N. peak value of the signal. Hence, the distortion is formly spaced levels, 2 2 V This gives given by From (23), it follows that the ratio of the largest to the smallest level is given approximately by Ayn� ""YO l+µ (26) � , >"' A • f ::> It is interesting to note that 0.. f 0.01 ::> j---,�----+------+-----1 0 This ratio, when expressed in decibels, is often referred to as the compression of the system. 0.001 ....______0.001 0.01 _.,,..0.1 1.0 The distortion power is given approximately by _.__ ___.,.. (7) when the number of levels is large n 124 ELECTRICAL COMMUNICATION •" June 1951 www.americanradiohistory.com find that the minimum distortion is given by 12 we 10 D 2 (}\ ) "7?, m -�[- 3 ia ( 1-a y2)'A/3 ]3 8 2 dy AaH-IJ"3 O 1 \ a \ = B l+ i2N2u [ ( 3 )] ' 2' \,. 4 _2\+3 B3(1/2, l+A/3) . (33) - 12N2 B(l/2, 1+\) I 2 !OI PTIMUM l1. I oO 10 20 30 40 50 so 10 eo 90 100 In this case µ. (34) Figure 4-Distortion plotted against the parameter µ. C' ' The distortion for uniform-level spacing is from C= 12 and 3iND =log(l +µ) ( 1 +-;.) . (31) (35) For a sine wave, ND C The quantity is plotted versus in Figure 6. C, It is seen that for small values of there is and little advantage in using the minimum spacing. 6 It is not until the crest factor is above that the 28 70 r7 I/ When a signal is uniformly quantized, the dis� / ./ 24 60 tortion depends on the ratio of the peak to the 17 / root-mean-square value. This is illustrated in 20 7 / 50 1 5; 2 / / curve of Figure curve gives the minimum / 16 / 40 distortion for optimum logarithmic compression o / ./ when the compression parameter µ is selected by z 1/ 30 IL 3. -"I:, 12 �/ curve / / '/ v 8 20 Application to Specific Signals / , 4. / � 2 4 / 10 /..- /,'.,; - To illustrate the various points involved, let 0 4 8 12 16 20 24 28 32 36 40 the preceding results be applied to the class of c signals specified by the probability density given Figure 5-Distortion characteristics. Curve 1 is for uni by form quantization 3•ND = C. Curve 2 is for optimum logarithmic compression. (3+2y3 \)u ' P(y) 3•ND = log(l +µ) 1+ )'A(32) ( �:r Curve 3 is for the case where A which is discussed in the appendix. Here is a C= . parameter that determines the ratio of the peak 1+ log(l+µ)-1 value to the root-mean-square value. For -1 June 1951 • ELECTRICAL COMMUNICATION 125 www.americanradiohistory.com Appendix-Statistical Properties of the 5. Signal; Pearson Distribution The probability density given by 1 y2 P(y) = l - ) " (3+ L\)tutB(1/2, A.+1) ( (3+ 2A.)u (36) � (36), u is known as the Pearson distribution. In is A 126 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com Four/Two-Channel Time-Division-Multiplex Telegraph System for Long-Distance Radio Circuits * By W. C. PETERMAN America Ca bles and Radio, Incorporated; New Yo rk, New York All and-ANATOLE MINC Ma ckay Radio and Telegraph Company, Incorporated; New York, New York NCREASED RELIABILITY of frequency synchronism of the multiplex receiving regener shift transmission has made the use of time ator, as all morse signals begin and end at the division-multiplex systems feasible on long same times that multiplex signals would begin radioI circuits. Time-division multiplex provides a and end. The resultant distortion is too small to means, economical of frequency spectrum, to be noticeable to the ear. handle traffic that is too great for one printer At the receiving end, the signals are regener channel or that originates from or is destined to ated electronicaliy with greater precision and reli different places. A four-channel multiplex system bility than by prior methods using electro at a speed of 50 words per minute per channel, mechanical relays in conjunction with a rotary using the five-unit Baudot code, has a unit pulse distributor. An automatic margin indicator pro length of 0.010 second, which normally provides vides one kind of indication if the margin falls sufficientmargin against signal shift due to radio below 40 and another kind if below 20 percent. multipath effects or other causes. After regeneration, the signals are distributed A combination electronic and electromechan very simply by an electromechanical distributor ical system was chosen for reliability and sim whose brushes do not require great contacting plicity of maintenance. At the sending end, the precision for this function. The use of an electro electromechanical distributor provides a simple mechanical distributor also provides a simple means for interleaving the signals for the various means of applying a modified Verdan scheme for channels. It also provides a means for extending error detection and permits monitoring of the a channel on a start-stop-printer basis without channel signals just as they are leaving the dis the use of individual electromechanical translat tributor face plate. ing devices, to a branch office so located that The Verdan scheme of error detection is based clutch pulses can be sent to it economically. Al on sending each character twice and comparing ternatively, a multiplex extensor may be used for the received combinations. If they are the same, more-distant branch offices. it is assumed that no error has occurred as the Any inaccuracies of signals from the sending dis probability of an identical error occuring in two tributor are eliminated by passing them through receptions of the same combination is very re a simple electronic regenerator. Where the multi mote. If they differ, an error has occurred in one plex signals are passed over a keyed tone channel of the transmissions and an indication is made on between the multiplex terminal and the radio the copy by printing a question mark. This transmitter, the sending-end regenerator makes scheme cuts the trafficcapacity in half, which is possible the synchronization of the tone-channel equivalent to using a 10-unit code. However, as carrier to the multiplex baud speed, which prac the error-indicating equipment can be cut in or tically eliminates carrier distortion. The sending out quickly, it need be used only when the auto end regenerator also makes it possible to send matic margin indicator shows that the margin of manual morse ·signals without disturbing the , regeneration is low. Thus, if 40 percent of the Presented at American Institute of Electrical Engi traffic is handled with error indication, the aver neers Winter General Meeting in New York, New York, * age length of the code would be 7 units. If only on 24 January 1951. Preprinted as Technical Paper 51-61. To be published in the Transactions of the American Insti 30 percent is handled with error indication, the tute of Electrical Engineers. average length of code would be 6.5 units. June 1951 127 ELECTRICAL COMMUNICATION • www.americanradiohistory.com Apparatus I. ing operating tables equipped with start-stop SENDING END 1.1 printers and facilities for extending their respec tive channels through them to branch offices. The four channels are normally operated from Regenerating and synchronizing equipment is standard sending operating tables equipped with mounted on the regenerator rack of Figure 2. It multiplex perforators and multiplex transmitters. includes a 200-cycle-per-second fork, similar to The distributing, translating, monitoring, regen the one in the sending distributor rack. erating, and general operating equipment is lo Distributing and monitoring equipment is cated on the sending distributor rack shown in located on the receiving distributor rack, which Figure 1. The distributor has six pairs of rings may be seen in Figure 3. The distributor is the and is driven by a phonic motor at a speed of same as the one used on the sending distributor 300 revolutions per minute from an electronically rack and is driven from the fork on the regener driven 200-cycle-per-second fork. ator rack. RECEIVING END Error-indicating equipment and channel-ex 1.2 tension apparatus are mounted on the error Reception is normally on four standard receiv- indicator rack displayed in Figure 4. Figure 1-Front and rear views of the sending distributor Figure 2-Views of the electronic regenerator rack. The rack. The distributor is driven by a phonic motor. receiving distributor is driven by a fork in this cabinet. 128 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com 2. Sending-End Circuit A from multiplex Channel is always operated a C, transmitter in the main office but channels B, 2.1 GENERAL D and can be extended to branch offices on a The sending-end circuits are shown in Figure 5. start-stop-printer basis. The various channel segments on the sending A EXTENSION CIRCUITS ring show that channels and Care interleaved 2.2 D, as are channels B and which follow. This was Circuits are shown for extending only the C done for ease in changing from four-channel oper channel by closing the seizure switch in the ation to two-channel operation, which can be branch office. The closing of this switch causes accomplished by throwing switches (not shown). the clutch pulse to operate the seizure relay and The marking polarity is positive on two of the A D the C-channel switch relay. The latter switches channels, and in this case, and is negative on the C-channel sending segments from the multi the other two channels (B and C) , in accordance plex transmitter to the extension storage capac with the usual practice, to provide signal rever itors through the releases C-channel the idle-signal switch. sals for synchronizing even when all four channels Each clutch pulse start-stop trans are idle. mitter distributor clutch so that the signal Figure 3-Receiving distributor rack mounting the dis Figure 4-Error-indicator rack. In the front view, the panel tributing and monitoring equipment at the receiving end. has been removed from one error-indicator chassis. 129 June 1951 • ELECTRICAL COMMUNICATION www.americanradiohistory.com combination for one character will be sent to the brushes. This is done by flip-floprelays, operated main officeand the extension-line relay operated. at the proper time from a segment on one of the From the tongue of this relay, polar signals are distributor rings. When sending is from a branch sent through the extension translator rings into office, the clutch pulse is interrupted on alternate the channel-extension storage capacitors from revolutions of the brushes and the extension relay which they are sent through the proper segments is cut off from the translator rings on the same of the sending ring. The clutch pulse, originating revolution by means of a flip-flop relay so that from a segment of the local ring, is phased by the same signal combination remains on the stor means of the phasing control rheostat so that the age capacitors as on the previous revolution. brush on the extension translator rings will be By using two channels for original transmis passing over each pulse segment when the signal sion and the other two channels to repeat the for that pulse is arriving from the tongue of the combinations for comparison, a traffic-carrying extension-line relay. The segments on these rings capacity of SO words per minute on each of two are spaced so that the brush will pass from the channels can be obtained. If two channels are beginning of one segment to the beginning of the used this way, the other two can be used with next segment in 0.022 second, which is the dura error indication at 2S words per minute per chan tion of each selection pulse of the start-stop com nel or either or both may be used at SO words per bination when operated at a 61-word-per-minute minute per channel without error indication. speed. Error indication, on a two-channel basis, is put When the auto-stop contacts of the branch into effect at the sending end by throwing a office transmitter open due to a taut tape, the switch. The circuits for operation on C channel B transmitter distributor clutch is not released by with repetition on channel, by throwing C- and the clutch pulses and a solid marking current is B-channel error-indicator switch, are shown in sent to line which would cause the extension Figure S. The first three pulses are repeated by a storage capacitors to be charged with marking direct connection of the sending segments but the potential and the "letter shift" combination other two are repeated from overlap storage would be set up. This would be a false combina capacitors, as the tape on the C-channel trans tion and would result in errors if upper-case char mitter is being stepped while they are being sent. acters were being sent when the tape became MONITOR CIRCUITS taut. To prevent this, the idle-signal relay tongues 2.4 transfer the sending segments from the storage capacitors to spacing potential, so that the "idle" Two multiplex five-magnet tape printers are signal is sent over the multiplex channel when available for connection to any channel for moni a false "letter-shift" combination is received. toring the signals as they leave the sending ring. The false "letter-shift" combination has no start Since the signals from all channels are already pulse and this distinguishes it from the true aggregated on this ring, they must be separated one. This condition is detected by means of idle again by channels on the monitor translator rings. signal tubes, which are controlled from a segment The printer selections are controlled by monitor on the translator ring over which the brush passes control tubes, which reinvert the signals of the when the start pulse is received. These tubes two inverted channels. control the idle-signal relay. 2.S SENDING REGENERATOR CIRCUITS ERROR-INDICATOR CIRCUITS 2.3 Signals from the solid sending ring are regener For error indication, each character combina ated before they are used to modulate the tone tion is sent twice. On a four-channel basis, the generator of the keyed tone channel between the same combination is sent on two successive rev telegraph office and the radio station. As may be olutions of the distributor brushes and the traffic seen in Figure 6, they are passed by means of an capacity is 2S words per minute per channel. isolation cathode-follower tube into the second When sending is from a multiplex transmitter, it ary of the input transformer of the positive-to is only necessary to open the transmitter step polar-pulse converter. Differentiated square pulse circuit on alternate revolutions of the waves, at baud speed, from a fork-driven counter 130 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com NORMAL TO TONE SENDING REGENERATOR MONITOR CIRCUITS CONTROL TUBES SENDING INVERTED + STORAGE IDLE CAPACITOR + SIGNAL T TUBES • - U1llU (-+ I co ...J-:---r<>-0 I I I oT I I I I No (-+ M I � I oR I ....R p I I � SEIZURE SWITCH E //lllJ C4 I I VOLTS ALTERNATING ...,. RT CURRENT I I SENDING 110± I TRANSMITTING I l.- DISTRIBUTOR I --l I I CLUTCH , I �RELAYr I cs I I ..... I � I 1 ��+����-�--'-SENDING...... 8 I RECORD TA PE-OPERATED I STOP� SWITCH C-CHANNEL I D NAL L L 8 �I _L �_�_s �-� -- � ---'"-<>-__,_._..� _ - � + f? J TO EXTENSION � 8li, LINE � ., I L__J-+-0--4--1-----' I I I I I I I I I I 4I 2I LOCAL10 EXTENSION MO JTOR RINGS6 TRANSLATOR TRANSLATOR RINGS RINGS TO CORRESPONDING SEGMENTS ON RING 5 TO CORRESPONDING CONTACTS ON 8-CHANNEL SWITCH l B CHANNEL CHANNELl I 2 3 4 5 0 0 0 Figure 5-Sending-end circuits illustrating distributors, channel-switching system, and branch-office connections. June 1951 ELECTRICAL COMMUNICATION 131 www.americanradiohistory.com · are� passed through a negative-pulse clipper and The regenerator unit retimes the signals and the_'positive pulses pass to the primary of the supplies to the rotating distributor a fully regen input transformer. These pulses constitute accu erated signal ; a signal that maintains a fixedphase rate time markers, adjusted to occur at the ap relationship to the respective distributor face proximate center of the signals from the sending plate segments and that has rigorously equal ring. They sample the signal for polarity and initi- mark and space elements. POLAR-PULSE EXPANDER + + + PO SITIVE-TO-POLAR PULSE CONVERTER 100-CYCLE SQUARE WAVE o41--1.--., Figure 6-Sending-end regenerator circuits that provide accurately timed signals. ate, with pulse the Distribution of the Regenerated Signal great accuracy of timing, a of 3.1.2 polarity of the sampled signal. These polar pulses, A rotary synchronous distributor distributes brought out through the output transformer, are expanded into the regenerated polar signals in the regenerated signals to the proper channels the polar-pulse expander. Due to the unbalanced and translates the five-unit Baudot code into ground connection across the primary of the out start-stop-printer signals. The combination of gas put transformer, a small portion of the distrib and vacuum tubes provides the signal power for utor signal as well as the polar pulses appears at the receiving traffic and monitor printers, while the output of the converter, which facilitates both relay and vacuum-tube outputs are available monitoring. for channel-extension circuits. 3. Receiving-End Circuits 3.1.3 Detection of Errors 3.1 GENERAL When error detection is desired, the sending end Three main functions are to be performed by is made to repeat the same character on two the receiving multiplex terminal equipment. transmissions. The error indicator will compare Regeneration of the Received Signal the two transmissions and when a discrepancy is 3.1.1 detected will switch the corresponding receiving The received radio signals are subject to distor printer from the multiplex signal to an internal tions by various types of transmission irregular signal source. A sequence of start-stop signals ities. Before the signal can be distributed to the will then be supplied to the printer to print a respective automatic printing devices, it is neces question mark (?) right after the erroneous char sary to remove such distortions by a regeneration acter; thereupon the printer is returned at the process. proper instant to the signal circuit. 132 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com Receiving Regenerator 4. one part per hundred thousand was found ade GENERAL quate for an initial adjustment. 4.1 From the receiving-end fork, positive gating The receiving regenerator logically subdivides pulses are derived at signal-element speed which, into four functional sections, the interrelation of of course, also tend to drift ahead of the received which is illustrated in Figure 7. signal edges. By means of a manual phase-shift The radio receiving station normally relays the ing device (synchronization phase shifter) , these received signal in the form of a keyed tone over gating pulses are initially adjusted to lag slightly regional facilities (land line or very-high-fre INCOMING KEYED TONE quency link) to the operating office where the MULTIPLEX Figure 7-Receiving regenerator. regenerator and associated terminal equipment SGI NAL are located. This keyed tone, after demodulation and shaping into a direct-current square-wave l form, passes to both the synchronization and re DEMODULATION REGE NERATED AND REGENERATION OUTPUT generation chassis. SHAPING i--TO DISTR IBUTOR The frequency-standard unit generates two separate sets of timing pulses, each at signal element speed, which are supplied to the syn chronization and regeneration units, respectively. Comparison of the phase of one set of timing FREQUENCY SYNCHRONIZATI ON pulses with the timing of the radio-signal leading STANDARD edges (crossover from space to mark) is made in the synchronization unit. Departures from the correct phase will result in synchronizing im behind the signal edges. The gating pulses then pulses being applied to the vacuum-tube fork will tend to drift into the edges of the radio signal. oscillator in the frequency-standard unit. When the radio-signal leading edges (crossover In the regenerator unit, the local timing pulses from space to mark) and the gating pulses coin sample the polarity of every signal element of cicl,e, a synchronization pulse of a fixedmagnitude the received signal to produce an output pulse, and duration is applied to the fork. For the dura which when expanded to full signal-element dura tion of the synchronizing pulse, the fork is slowed tion is the regenerated signal. 30 down by approximately parts per hundred In addition, an automatic margin indicator has thousand. The synchronization marker pulses are been incorporated in the receiving-regenerator thereby effectively moved back to some position rack. This unit automatically indicates the degree lagging the signal edges. Once the synchroniza of reliability of the regeneration process by light tion pulse is over, the receiving fork will tend to ing alarm pilot lamps should the signal be severely resume its original rate, drift again into the signal mutilated by transmission disturbances and it edges, and the cycle will repeat itself. permits a range to be taken at any time without It is to be understood that the fork is to syn interfering with traffic. chronize to a radio signal subject to random trans 4.2 REGENERATION AND SYNCHRONIZATION mission disturbances. The time the signals require to travel from a transmitter to a distant receiver The receiving-end regenerator rack has a pre is likely to vary from instant to instant due to cision vacuum-tube oscillator similar to the one multipath and other effects. The receiving fork used at the sending end. For proper synchroni will keep in step with the signal by assuming a zation, this receiving fork is adjusted to run phase relation to the signal edges, such that only faster than the sending-end fork by slightly more some leading signal edges will arrive late enough than the anticipated maximum speed difference and produce synchronization impulses. The ma between the two forks. With such an adjustment, jority of signal edges will have no effect on the the receiving end will always tend to run ahead frequency of the receiving fork. of the sending end. A speed difference of about It is desirable to keep the speed differenc� June 1951 • ELECTRICAL COMMUNICATION 133 www.americanradiohistory.com between the sending and rece1vmg forks small. At the sending end, where signals of only 3- to Each rate correction can then also be made small, 4-percent distortion have to be regenerated, a and the fork will not follow each random phase high regenerator threshold is quite adequate and variation of the signal. The receiving-end fork allows for circuit simplicity. At the receiving end, then runs essentially at constant speed except for where signal mutilation can be severe, the regen minute rate corrections that cancel the speed dif erator has to be designed for a very low threshold. ference between sending and receiving forks. This is done by using an effective sampling pulse Each correction reduces the speed of the receiv width of approximately SO microseconds (O.S per ing-end fork by approximately 30 parts per hun cent of the signal element with four-channel oper dred thousand and lasts 0.1 second. This is ation) . Since the effect of the individual syn equivalent to a phase shift of approximately 30 chronization impulse is also small, the threshold microseconds. Considering that a four-channel of this regenerator is of the order of 1.S percent multiplex system operating at SO words per min for four-channel operation. ute per channel has a signal-element length of THEORY OF OPERATION 10,000 microseconds, each individual correction 4.3 represents indeed a minute phase shift. The small effect of each individual correction Figure 8 shows a block diagram of the receiv also affords protection against over-synchroniza ing regenerator. At the lower left, a vacuum-tube tion. A signal may split due to severe multipath driven fork supplies a 200-cycle wave to the effect or noise. Such splits would simulate addi synchronization phase shifter. This is a magnetic tional leading signal edges and might result in type phase-shifting device that allows continuous erroneous rate corrections. The effectof each cor control of phase shifting by means of a knob on rection being small, the receiving fork will not the front panel. The phase-shifter output is fre readily assume an incorrect phase with respect to quency divided in the synchronization frequency the signal. In turn, should a few signals arrive division stages, first to 100 and then to SO cycles. exceptionally late, the phase of the receiving end Either SO- or 100-cycle outputs are available as will not be affected materially. As an additional pulses of about SO microseconds width. These protection against over-correction, the maximum pulses initiate the synchronization timing pulses, repetition rate of the synchronizing pulses has which must occur at signal-element speed. Thus been limited to 120 per minute. for four-channel SO-word-per-minute operation, The receiving regenerator unit operates on the the 100-cycle output of the frequency-division same signal-sampling principle as the sending-end stages will be chosen. For two-channel SO-word regenerator. Sampling pulses are derived at signal per-minute operation, the SO-cycle pulse output element speed from the receiving-end fork and is used. A front-panel switch allows the selection are manually phased to coincide with the approx of either output. Each pulse of the synchroniza imate center of the received signal elements. tion frequency divider will initiate a square-top threshold The of a regenerator is defined as the positive pulse of 4 milliseconds duration through minimum signal margin that the regenerator will the gate. accept and still regenerate faultlessly. Even if a The output of the input signal-shaping unit is perfect signal is supplied to the regenerator, it is a square-wave-type polar signal. This signal is to be understood that the margin will be less than differentiated by a short-time-constant resistance 100 percent. Two main factors account for this: capacitance network, the output of which will be the width of the selection pulse and the synchro a positive pulse for a leading signal edge and a nization-rate corrections. negative pulse for a lagging signal edge. These Assuming that a perfect signal is received (four signal edges and the 4-millisecond pulses are sup channel operation), the range will be measured as plied to a coincidence gate. This stage is biased 10 milliseconds minus the width of the sampling to cutoffso that neither the positive 4-millisecond pulse. A further reduction of the margin reading pulses nor the positive signal edges alone will be results from the small retardations and accelera sufficient to make the tube conduct. When the tions to which the fork is subjected for syn 4-millisecond pulses and the positive signal edgeE chronization. occur simultaneously, an output pulse will result 134 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com from the coincidence gate. Obviously, the coin second width. The energy of these 0.1-second cidence of a negative signal edge and a 4-milli pulses will slow down the fork electromagneti second pulse will not produce any output. Thus, cally. To insure against over-torrection, the max a coincidence of the leading signal edges only imum repetition rate of these 0.1-second rate with the 4-millisecond synchronization timing corrections is limited to 120 per minute. A cor pulses is hereby achieved. rection limiter renders the 0.1-second gate insen- SIGNAL DISTRIBUTED INPUT r OUTPUTS POSITIVE-TO- POLAR SIGNAL nn POLAR PULSE PULSE nn .tl11.t SHAPI NG M CONVERTOR - EXPANDER . - I I I I ' I .flJl_ I MECHANICAL DI FFERENTIATOR _u_ CHANNEL 4 CHANNEL - EL PHASING _u_ / :L I f I I I -Yr I � COINCIDENCE 4-MILLISECONO PHONIC GATE GATE .LL MOTOR 14-- - 4CHANNEL ,, 2 CHANNEL I� . I ' JUL _u_ J_ SYNCHRONIZAT ION 0.1- SECOND CORRECTION PUSH -PULL PULSE EXPENDER RATE FREQUENCY 14-- AMPLI FIER ANO GATE LI MITER DIVIDER ' IL W\J'1 � SYNCHRONIZATION REGENt ERATION 200 -CYCLE REGENERATION FREQUENCY FORK PHASE PHASE SHIFTER � � DIVIDER - SHIFTER Figure 8-Details of the receiving regenerator. The circuit accurately retimes and reshapes the received signals. The coincidence of a positive signal edge and a s1t1ve for an additional 0.4 second beyond the 4-millisecond pulse will apply a synchronization 0.1-second operation time of the gate. A front pulse to the 200-cycle fork. The output of the panel neon lamp will fl.ash every time correction coincidence gate is a pulse of too short a duration pulses are applied to the fork. effectively to slow down the fork. This pulse is The synchronization phase shifter feeds also a therefore firstexpanded to 0.1 second width. This phase-shifting unit termed a regeneration phase is accomplished by applying the output of the shifter. The regeneration frequency-divider fol coincidence gate to a pulse generator. On recep lowing this phase shifter performs two functions : tion of a coincidence pulse, the 0.1-second gate A) it supplies push-pull 50-cycle square-wave will generate a square-top positive pulse of 0.1 outputs, which after power amplification provide June 1951 • ELECTRICAL COMMUNICATION 135 www.americanradiohistory.com Automatic Margin Indicator the driving energy for the phonic-wheel motor, 5. and B) it provides scanning pulses at signal-ele The automatic margin indicator provides an ment speed for regeneration. indication of the signal quality. The received 3 Again, depending on whether two- or four radio signal is sampled by this unit into classes. channel operation is desired, a switch allows the A. Signals having more than 40-percent margin. Such selection of either 100- or SO-cycle pulses from signals are to be considered as good signals, and no the regeneration frequency divider. These are indication will be given by the automatic margin positive pulses of SO-microseconds width and feed indicator. Signals having less than 40-percent margin but a converter stage simultaneously with the square B. wave output from the signal-shaping unit. more than 20-percent margin. For each such signal, Should a positive scanning pulse coincide with a white neon lamp will flash on the front panel. C. Signals having less than 20-percent margin. Such a positive signal element, the converter output signals are very poor, and the probability that error will be a positive pulse. Coincidence of a positive will occur is great. Should any one signal be received scanning pulse with a negative signal element will with less than 20-percent margin, a red neon lamp on result in a negative pulse output. The output of the front panel will light and remain on until it is the converter will therefore consist of negative manually extinguished. and positive pulses, the sequence of which will depend on the combination of positive and nega The automatic margin indicator distinguishes tive elements of the incoming signal. Since the between forward and backward signal margin. polarity of these output pulses is determined by Four neon-lamp indicators are therefore provided the signal polarity existing at the time at which on the front panel of the unit: two white lamps scanning pulses are supplied to the converter, it for 40-percent backward and forward margin is evident that the radio signal is hereby effec indications, respectively, and two red lamps for tively sampled, signal element by signal element, 20-percent backward and forward margin indi by the scanning pulses. cations, respectively. The polar-pulse output of the converter is The regeneration frequency divider, as shown expanded to full signal-element length by means in Figure 9, supplies to the automatic margin A) of a flip-flop circuit. The output of the flip-flop is indicator : positive pulses at signal-element of square-wave type. The crossover from one speed, which are identical with those used for polarity to the other of the flip-flop can only regeneration and are termed "scanning pulses," occur at 10-millisecond intervals, as timed by the and B) positive pulses at signal-element speed scanning pulses. This flip-flop output will there similar to the scanning pulses but delayed by S fore have a unity mark-to-space ratio. It will also milliseconds with respect to them. maintain a constant phase with respect to the The "scanning pulses" initiate two gates at A, regeneration frequency divider. This flip-flopout signal-element speed. A gate which generateE put is now fed to the distributor face plate for positive square-top pulses of 2 milliseconds width, distribution. Since the distributor is synchro and a gate H, which generates 1-millisecond nously driven by the regeneration frequency di square-top pulses. The scanning pulses that are A) vider, the flip-flop output will also maintain a delayed by S milliseconds initiate : a 2-milli constant phase relationship to the distributor second gate C after an additional delay of 3 D face-plate segments. Therefore, a fully regener milliseconds, and B) a 1-millisecond gate after ated signal is supplied to the distributor. a delay of 4 milliseconds. From Figure 11, it iE In order to have signal reversals on the air seen that these four gates effectivelysurround thE A A even when all channels are idle, channels and sampling pulse. Gates and B are initiated a1 D D have been made positive or direct channels. sampling time. Gates C and are so timed tha1 Channels B and Care negative or inverted chan their lagging edges coincide with the sampling nels. Thus, at the receiving end, another inver pulse. sion of channels B and Cmust be performed before The square-wave output of the radio signal· distribution to t e rint rs To shaping unit is differentiated into positive and h p e . accomplish this, the flip-flopstage has two (symmetrical) outputs : negative pulses. A full-wave rectifier converts al A one direct and one inverted output. these polar pulses into positive pulses. Gates ELECTRICAL COMMUNICATION • June 195: 136 www.americanradiohistory.com D B, C, and are fed into four coincidence stages, will change the scanning pulse phase with respect respectively. Each of these coincidence stages to the radio signal. Since the gates of the auto receives also the positive pulses derived from matic margin indicator are initiated by the scan each radio-signal edge. Coincidence of a signal ning pulses, they will move by the same amount edge with any of the four gates will result in an as the scanning pulse. To take a margin with the output from the corresponding coincidence stage. automatic margin indicator, the regeneration Thus, a coincidence output from gate C indicates phase shifter is manually advanced in phase until that the forward margin is less than 40 percent the forward 40-percent light just starts to flash. and a white neon lamp will flash for each coinci The phase shifter is then backed up slightly to a D dence pulse. In turn, coincidence of gate and a position where no flashes occur for, say, one min-. 3ignal edge indicates that the forward margin is ute. The regeneration phase shifter is then man less than 20 percent and the red neon lamp will ually retarded in phase with respect to the signal light and will remain lit until manually extin- and a similar reading is taken by observing the back 40-percent light. The difference between the 5uished. Similarly, coincidenceA outputs corre- 3ponding to the gating time of and B will indi two dial readings of the regeneration phase :.:ate back margins of less than 40 percent and 20 shifter plus 40 percent is equal to the signal percent, respectively. These lamp indicators margin. Since during this operation the scanning INPUT SIGNAL y DIFFERENTIATING FULL-WAVE SIGNAL .nn.. SHAPING . NETWORK -Y-r- RECTI FIER I SCANNING DELAYED I I I PULSES BY 5 I MILLISECONDS 2-MILLISECOND COINCIDENCE WHITE 40- PERCENT GATE A __,. GATE - y � BACK MARGIN REGENERATION FREQUENCY ..LL DIVIDER RED I-MILLISECOND __, COINCIDENCE 20-PERCENT ...___. GATE B GATE BACK MARGIN lJ.. WHITE 3-MILLISECOND 2-MILLISECOND ·� COINCIDENCE 40-PERCENT DELAY GATE . GATE C . GATE FORWARD MARGIN RED 4-MILLI SECOND I-MILLISECOND '--t COINCIDENCE 20-PERCENT DELAY GATE GATE D GATE FORWARD MARGIN Figure 9-Block diagram of the automatic margin indicatior. Two percentages of both forward and backward margins are indicated by lamps. therefore show the operator whether any adjust pulse stays clear of the signal edges, no interfer ments of the circuit are required and indicate the ence with traffic will result. reliability of the received signal. The automatic margin indicator unit is pro The 40-percent indicator lamps also allow the vided with front-panel test jacks for cathode-ray margin to be taken without interruption of traffic. oscilloscope observation. For quick monitoring, Manual rotation of the regeneration phase shifter a composite wave pattern, similar to that shown June 1951 • ELECTRICAL COMMUNICATION 137 www.americanradiohistory.com NORMAL SIGNALS C5 8 I .... � REGENERATED SIGNALS, ALL CHANNELS IDLE co INVERTED SIGNALS 4 C 5 6 I ! -, 6�PULSE- MONITOR MAGNET I PRI NTER __ I _J PHASING SWITCH ON 4----+ 0FF MONITOR SWITCH l5T SUPPRESSOR TUBES t'.'j r t'.'j r-i --3 PLATE -CONTROL 84, 04 =c RINGS 85, 05 ,� .... �r A-CHANNEL r-i TRANSLATOR RINGS 9�11�=====�� �i - - - � r_..,- --i,- - I d----1------+- I t.H-MUIUR RECEPTACLES- ....r-i ------r------, > - , I P�IN START-STOP I :::a I I I I PRINTER I I ALTERNATtNG I � CURRENT CURRENT lli TRAFFIC MONITOR I I [ PRINTERS DIRECT I I I L:Jr- c:JI I I A-CHAN NEL OPl':RATI NG T L BRANCH OFFICE 'a' ____ L I L �l'._ :_ J - J ; ------ .... Figure 11-Receiving distribution, translation into start-stop signals, and monitoring are accomplished in these circuits. \C � .... www.americanradiohistory.com in Figure is provided, showing on the oscillo 10, channel only ; the signals of all channels are sim A) scope the time position of: the radio-signal ilarly translated. edges, B) the scanning pulse, and C) the four The start and stop pulses being inserted inde gates of the automatic margin indicator. In addi pendently of the signals, they would be present tion, the 4-millisecond gate used for synchroni even when the circuit is idle. To prevent this, a zation can be added to the cathode-ray pattern. SCANNING PULSES FROM I\ a Receiving Distributor Circuits REGENERATION I 6. I FREQUENCY _J\ � IL 6.1 GENERAL DIVIDER I I I I I I I GATE A I I I I -! The regenerated signals are supplied to the -'"----'I I A ..__.._I __ I A I I I I receiving distributor for channel distribution, for GATE I f8I B ______, fil��------! I translation into start-stop signals, q.nd for error I I I 11. I I I detection as shown in Figure GATE C A- from I I I and D-channel signals are received the I c I I I GATE D - - normal regenerator output and those for B and C -"--I -IDl1�- ----IDlI I I I I I channels from the inverted output. These latter INPUT !RADIO) I I were sent inverted and must be "reinverted." SIGNAL I I I I I I The brushes of the distributor are driven by a JI I r COMPOSITE ;.-.__,""""'l\.� =---i phonic motor that receives its power from the SIGNAL regenerator tuning fork and these brushes are J rcfoFeW � r- adjusted so that they will be passing over one of I 1 the "live" segments of channel-selection ring I I TIME I I I I IN MILLISEI I I I I C I IOND S during approximately the center of the time inter Figure 10-Automatic margin indicator timing and gating val for the reception of each signal pulse. pulses related to input signal from radio receiver. Channel phasing is accomplished by stepping the brushes backward on their drive shaft one line-segment interval at a time until the channels suppressing means is made effective when the 15 are properly phased. This ONis indicated by placing idle combination continues for more than or the phasing switch in the position and noting 20 seconds. when a monitor printer on D channel receives the In normal operation, the marking pulses of the phasing-signal combination of all five marking multiplex signals, as they appear across the pulses. storage capacitors, are applied to the grids of TRANSLATION CIRCUITS suppressor tubes. A timing capacitor is charged 6.2. through a high resistance and discharged through 1 the relatively low resistance of the plate circuits The signals are distributed through ring to of the suppressor tubes. When these marker five storage capacitors for the signals of each pulses are not received, the timing capacitor channel. The monitor printers and the circuits charges and, through an auxiliary suppressor that translate from multiplex to start-stop signals tube, operates the suppressor relay to put a solid get their selection pulses from these storage capac marking signal in the start-stop circuit. This itors. The start and selection pulses are of approx action is reversed on reception of the first mark imately 0.022-second duration. ing pulse when the circuit resumes operation. For translation, the signals from the storage capacitors pass through a translator ring, the live MONITOR-PRINTER CIRCUITS 6.3 segments of which are spaced so that the brush · passes over successive segments at 0.022-second The monitor printers can be switched to any intervals, generating signal elements of this dura channel. Their selection pulses are obtained from tion. Start and stop pulses are added through the storage capacitors through thyratron tubes other segments at the proper intervals. Transla whose plate circuits are controlled from the plate 11 A tion circuits are shown in Figure for the control rings. June 1951 • ELECTRICAL COMMUNICATION 139 www.americanradiohistory.com ..... CHANNEL- I $ SELECTION RINGS 3 --- REGENERATOR INVERTED OUTPUT TRANSLATOR RINGS t LOCAL RING 10 OPERATE --r:c+'Ei =rC+'B scAN 5 6 ·� '" �-;]I I RI I I I I I I I SELECTOR t'l TUBE I t"' t'l I I R2 .... I � -23 45-CYGLE I � QUESTION-MARK TRANSMJTTE� t"' _, OSCILLATOR �1 234 I �ill-11- 11 11-1- -111 2 ("l l' I �ff:f U:1U:1U1111:1:1l l I 20-POiNT ROTARY SWITCH +J �3 + OSCILLATOR I - � STEP COIL \. I == ERROR-I NDICATOR .�u OUTPUT I 0 ERROR - I CIRCUITS I N +6TOR � I � �.s;�<;;� RELAY .... L _ _ ------______J -175 -45 -23 -120 +120 TO BRANCH OFFICE EXTENSION ANO GROUND RELAY t � -w REGEi Vi NG RECEIVING FROM 1-111iJ � I FROM TUBE EXTENSION RELAY I I I TRAFFIC I _ PRI NTER b PRINTER I H __c� ���P��,���E __ § L _J � � Figure 12-Error-indicator circuits. The circuit automatically prints a question mark if poor radio reception causes a faulty signal. <:.11 ..... www.americanradiohistory.com 6.4 ERROR-INDICATOR CIRCUITS When the detector tube is pulsed, the traffic The equipment for error indication is mounted printer is switched from the normal receiving cir on a separate rack. When reception is on a four cuit to the question-mark transmitter through channel basis, the signal combination for each operation of switch relay RLl. The timing circuit character is received twice, on successive revolu is put into operation through RL2 so that the tions of the distributor brushes, and one is com traffic printer will be reconnected to the normal pared with the other. If there is any difference, circuits after the question-mark transmitter has normal reception is interrupted and a question sent the start-stop combinations for "figure " mark is printed after the questionable character. shift,'' question mark,'' and "letter shift." Normal reception is then resumed. The circuits The question-mark transmitter is a 20-point for this operation arc shown in Figure 12 for C rotary switch driven from a 45-cycle-per-second channel only. oscillator. During the firstreception of a character, the five The switch relays RLJ and RL2 are in the plate signal-storage capacitors SC1-SC5 are charged circuit of the switch tube, which is operated from through Rl in parallel with the primary winding two control tubes. One of these is coupled to the of the input transformer. Pulses from the second detector tube and initiates the operation while the other determines the length of operation by ary winding are rectified and pass through the of a pulse from ring 6 through a timing selector tube to the flip-flop circuit. The path means circuit, thus preventing the traffic printer through the selector tube and R2 is effectively a short-circuit as there is no plate current flowing from being switched during the printing of a character. through R2. The flip-flopcircuit is controlled by pulses from Figure 12 also shows the circuits used when the original transmission is on C channel and B chan ring 6 of the distributor and is switched after eachVJ character is received. On this first character, nel is used forON comparison. The C+B switch is of the flip-flop circuit is active and has energized thrown to the position and Ji , J2 , and J3 are the cutout relay, which disconnects the traffic plugged into the C+B jacks. The flip-flop tubes printer from the channel-extension tube, thus are now controlled from two segments on ring 6. making it inoperative. Each character combination is set up on the After the five channel pulses have been re combined C- and B-channel storage capacitors ceived, the flip-flop circuit is pulsed through ring during reception over C channel. 6 and V2 becomes active. The voltage drop Printing and scanning for errors are done when across R2 from the plate current flowingthrough the character is being repeated on B channel. V2 acts as a cutoff bias on the selector tube and The timing of the switch from normal to the any pulses through Rl will be transmitted to the question-mark transmitter is changed to accord grid of the detector tube. with the new comparison period. June 1951 • ELECTRICAL COMMUNICATION 141 www.americanradiohistory.com Interference in Multi-Channel Circuits * By L. LEWIN Standard Telecommunicatin Laboratories, Limited; London , England N THE ASSUMPTION that the rela their face value. In this connection, the impor 0 tion between output and input for any tance of small variations of feeder length or of unit of the link can be represented as a carrier frequency in determinations of interfer power series of the input signal, suitably delayed, ence levels, is stressed. the harmonic content of the output for a pure It is shown that the beat-frequency method of tone input signal is found. When a multi-channel harmonic measurement, at present in use, al signal is injected, it is found that the inter though convenient from several points of view, channel interference falling in any particular is not exactly equivalent to a determination of channel consists of a number of separate terms, the harmonic levels and cannot be used for the each bearing a functional relation to a corre analysis of this paper. The two methods are sponding harmonic component of the output equivalent, however, when the second and third when a pure-tone test signal is used as input. harmonics can be shown to be the only sig This relation is found in the form of an integral nificant ones, in which case, as already stated, in which the root-mean-square amplitude of the the relationship with the interference level is pure-tone input occurs as a variable of integra straightforward. tion. Hence, to find the interference level, the Introduction harmonics have to be measured over a range of I. values of input amplitude, and the interference One of the most important characteristics of components deduced by a somewhat involved the performance of a multi-channel link is the analysis. Only in the case in which the second and third harmonics predominate is this procedure inter-channel interference level. It is not nor mally possible to measure this directly in test, unnecessary, and there is then a direct and simple relation between interference and harmonics. as a suitable multi-channel source is not likely to This latter construction confirms the standard be available. Instead, a pure-tone signal can be procedure for finding the interference level in injected, and the harmonic margin measured. this region. Otherwise, the paper is at variance This harmonic level is then usually taken as an with other published work, in that the latter indication of the interference level to be expected neglects the effects of coherence, which are here from the working link. The purpose of this paper considered in detail. is to examine in detail the connection between A particular case-distortion due to mis the harmonic level on the one hand, and the matches on a long feeder line-is investigated in performance of the link on the other, and to sug detail, and it is shown that, for increasing feeder gest a precise test procedure for relating the two. length, the interference and second harmonic at Harmonic Production first increase together, but that, for long feeder 2. lengths, the former decreases slowly while the We shall confine ourselves, for the most part, latter passes through a maximum and then oscil to the multi-channel signal, which will be at lates, its place as the dominant harmonic being "video" frequency. In this band, it will be as taken first by the third, and then successively sumed that the relation between output and in by higher harmonics. It is shown that, in this put signals for any unit of the link can be repre example, both the second and third harmonics sented by a power series could vanish for certain feeder lengths, although the interference could remain at an appreciable (1) level, so that caution is necessary before ac s = s(t) cepting small absolute values of harmonics at where is the input signal, either multi *Reprinted from Wireless Engineer, 27, pp. 294-304 ; channel, or test. There will in general be a time 1950. s s(t-r). December, v. delay so that in (1) is really This r, 142 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com delay can be ignored provided, of course, it is the may be used if a small number at the bottom are same delay that occurs in each power of s. An missing, provided N is increased by a corre exception can be made, however, for the term sponding amount. a1s, On which represents the recovered signal, and is a random phase angle, which ensuresEn in which will normally be much larger than the coherence between different channels. is a other (distortion) terms. Since we are concerned quantity that is 1 or 0 according to whether the with the distortion level relative to the recovered nth channel is working or not. signal level, it is only the amplitude of the latter I: En = that is of interest so its phase can be ignored. But We assume N aN, so that a is the fraction all of the distortion terms combine together to give the various harmonics, and hence their rela of channels actually transmitting at any given tive phase is important. moment. (Approxi mately, half the channels wiII In what follows, it will be understood, to save not be in use, owing to waiting, dialling, etc., repetition, that all harmonic and interference while of those in use, roughly a half will belisten levels are referreds=x (z)to ttheir qt, respective signal levels. ing, and therefore not transmitting. a may there If we put cos x to represent a pure fore be taken to be about a quarter.) tone test signal, into (1), being the root-mean En =En Since (both being either 0 or 1), the square level, we get 2 (3) 00 root-mean-square value of the signal of is V=a1x(2)i{cos (qt) +L:fn (x) ( qt (2) 2 2 cos n )!. { I:N !En P = { (2x / t =x. (2x2/ aN) aN)!aN! The amplitude fn(x) , which is a function bbth of However, the signal level of a working channel x 2/ t , the coefficients of (1) and the root-mean-square is ( aN) and it is to this level that the inter value of the signal x, is obtained in the appendix ference will be referred. Section 12.1. It represents the harmonic margin Generation of lnter'-Channel of the nth harmonic of frequency nq, and is the 4. quantity that can be measured in a straight Interference forward way in test. For example, if with a cer f2(x) tain input-signal level, is measured as 10-3, Let us confine our interest to the rth channel, = definea quantity y=r/N; y definesthe posi then the second harmonic is 20 log10(10-3) 60 and decibels down on the recovered signal (which ap tion of the rth channel, is zero at the bottom, and 1 at the top of the band of channels. pears, of course, in a differentfn part(x) of the band). The harmonic amplitudes vary with the Let us(3) substitute, in (1) the multi-channel sig x nal of multiply out, and convert products of root-mean-square input-signalx = amplitude and all obviously vanish at 0. Their actual be cosine terms to sum-and-difference terms. All haviour as x varies turns out to be important in those whose resultant frequency falls in the rth relating them to the inter-channel-interference channel can be picked out, and constitute the level. interference in that channel. At first sight, it would appear that all such 3. Multi-Channel Signal terms would be incoherent, thus adding up "power-wise" to give the total interference, but Let there be N channels (N large) equispaced p this is not the case. A detailed analysis reveals in frequency by an amount cycles per second, that they fall into groups with a certain amount so that the "video" frequency of the nth channel of coherence within a group, and with no term in is np. Then a multi-channel signal can be repre any one group coherent with any term in any sented by other group.gn(x) The amplitude of the nth group is S x 2 I:N En (npt+On). (3) denoted by and is a function of the root = ( /aN) t cos x mean-square(3). gn(x)amplitude of the multi-channel signal of is related, thoughfn(x) not in a very It is here assumed, for simplicity, that the simple way, to the function discussed in = gn(x) channels start at n 1, but the ensuing analysis Section 2. For this reason can conveniently 1951 ELECTRICAL COMMUNICATION 143 June • www.americanradiohistory.com z be referred to as the nth harmonic component of of root-mean-square amplitude as the experi the interference. It should be understood, how mental procedure may permit. A single tone of frequency trp is injected, and its root-mean ever, fnthat(x) ,this nomenclature is only by analogy with and is not intended in any way to square value is varied as just described. The describe the method of formation of the group. value of harmonic appearing in the channel r As described above, each group is generated by referred to the fundamental appearing in the r/2 f2(z) beat frequencies of all orders and from all chan channel is the function (4) , required. This nels, and it is, perhaps, a matter of surprise that value must be inserted into intox which has there is such a definite segregation of terms into also been put the correct value of the root groups. mean-square value at which the total multi channel signal is intended to operate. Relation Between gn(x} and fn(x} 5. The integration may be performed graphically; f2(z) or by fitting the measured with a poly We are interested in the rth channel, and in b1z+b z3+b6z5+ . . nomial of the form 3 . only odd order to find the harmonics produced in this channel by a pure-tone input of frequency q, we powers appearing. In either case,z it is necessary = = check that the largest value of that has been take q trp for the second harmonic, q }rp to e-•'lx' reached is sufficient for the factor to have x(2)lfor the third, and so on. Then an input of cos (rpt/n) will produce in the rth channel reduced the integrand to a small-enough value fn (x) to justify neglecting the contribution from the its nth harmonic of amplitude (relative to rest of the range. It is easily shown that the fundamental, which appears, of course, in the r/n). fn (x) channel It is this quantity that will be relatedgn (x)to that part of the component of gn(x)inter, ference that falls in the rth channel. as already stated, is referred to the level of the whence we have the following particular values. channel signal, so that, when it is found, the total interference margin G(x) in the rth channel, can l"'z5e-z•Jx' dz=x6; l"'z 1e-z'Jx'dz=3x8; G2(x) = L: gn2(x). be found from The power-wise z9e-z'Jx· = 12xl0; zlle-z'fx'dz=60x12; I"' I"' addition follows fromG(x) the mutual incoherence be tween the groups.y = / will, of course, also be a function of r N, which gives the channel position, so that the interference will, in general, vary (smoothly) from channel to channel. Nor i f,.(x) g2(x) = {4a(1 -y/2) } (b1x+3b3x3+ 12b6x5+ ...) mally, only the first few harmonics will be gn(x) . significant, so that the number of to be (5) summed is correspondingly few. x 12.2 In particular, when is small enough, it is seen In Section of the appendix, it is shown {4a(1 -y/2) } !, that, apart from the factor the that g2 first terms of and f2 are the same. Hence, for g2(x) = {4a(1-y/2) }l. f (z)e-z2fx2z4dz. x x ( 2 (4) sufficiently small values of we have \ Jo "' 4a can usually be replaced by unity, while the factor (1 -y/2)! shows a variation of 0·7:1=3 The order of approximation is that the fourth decibels from the top to the bottom of the band and higher harmonics be negligible(4) compared to of channels. For the rth channel, we simply insert the second. Unless this is so, or (5) must be y = / . g2(x) , (6) the value of r N To calculate which is used in full. When can be used, it is not neces f2(x) the amplitude of the second-harmonic component sary, of course, to measure (4)over a range of of interference for the rth channel, it is first values since the integration of is no longer f2(z) necessary to measure over as wide a range called for. 144 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com g3(x) The function is similarly found to be of distortion in some detail. Although the ap (10) given by proximate equation above is in general agreement with the results of the same order in the above paper, there isgn(x) no reference there to the semi-coherent groups and their relation fa(z) fn(x). As before, must be measured over a range of to The reason can be found in paragraph z, (9.3) values of a fundamental of frequency rp/3 "Low-Order Products from High-Order fa( z) being used. is obtained by referring the Terms" in which it is stated : "It can be shown amplitude of the third harmonic appearing in the that, although the power in each such low-order rth channel to the fundamental, which now ap product may exceed the power in each higher r/3. fs(z) pears in the channel can be fitted by order product, the number of products is so much b2z2+b4z4+b z6+ the expansion 6 ..., giving lower when N is not small that the total distor ga(x) = { 4a(1-y2/3)/2 ) ! tion power in the higher-order products is con (3b x2+ 2b4x +60b x6+ siderably greater than the power in the lower X 2 1 ...) . (8) 4 B order products." They have, therefore, neglected x When is small enough, we get the approxima these low-order products. What has not been realized is that these low-order products are tion, from the first term only, coherent with other terms that are not negligible, g (x) ""' 3{ /(2)!} { 4a(1 -y2/3) } ! fa(x), 3 and that these terms and the low-order products x (9) small. will therefore add up voltage-wise and not power wise. The effect, as found here, is to make the The order of the approximation is the same as for (6). g effects of the low-order and high-order products When this is satisfied, the higher n are then of the same order of magnitude, so that agree negligible, so that we have the approximation ment is limited to the region in which all such 2 G(x) ""' [4a { (1-y/2)f2 (x) products can be neglected. This leads to (10). + (9/ 2) (1-y2/3)fa2(x)}]! , x (10) small. As stated above, the occurrence of coherent In general, groups is quite unexpected, and they can only be determined by a very detailed analysis. That gn(X) = [ { 4ahn(y) } !/xn+3J fn(z)e-z'fx'zn+2 dz, they must occur is apparent when(A it +B is realized- C) loo that such a term as, for example in with a third-order product, is repeated identically in 2n e n frequency and phase in a fifth-orderproduct that hn(Y) = 1 8 sin dB. (11) (A +B- C+D-D) = 1fn . loocos y (-8-) gives such a term as - (A +B-C). o The writer is not aware if such an Particular values are effecthas already been treated in the literature, h2(y) = 1 -y/2, but it seems necessary that, when higher-order ha(y) = (1-y2/3)/2, harmonics have to be treated, the analysis follows h4(y) = (8/9-y2/3+y3/12)/4, closely that outlined here. 2 4 2 h5(y) (46-12y +6y /5)/(24) • Coherence with the Main Signal = 7. fn(z) may be approximated by a polynomial The odd-harmonic interference components commencing with the term b -1 zn-1, successive z n are peculiar in that part of each of them under terms having exponents of increasing by 2. The goes a phase change of exactly cp when each indi integrations can, of course, be done graphically. vidual term undergoes this same change. Thus if Comparison with Published Results we consider the expanded form of the product 6. (lpt+ 145 June 1951 • ELECTRICAL COMMUNICATION www.americanradiohistory.com (3) change cp, it follows that part of the interference transmitted wave, and in the expected output terms remain in phase with the main signal and is shown to be given by hence will add up voltage-wise rather than power = ..:iw (wat) (12) V sin -o, wise as the signals are transmitted from repeater In any particular case, however, it to repeater. o= [kr1r2 {2wc7 -(Ji-82 must be checked whether the main signal really does ;t sin +2r..:iw . sin (wat-r) (war)/war}J. (13) alter by the same phase angle, as even a small varia sin tion will destroy the coherence effect. In the next we Here, is the radio-frequency carrier and section, an example is treated in which a phase ..:iw wat sin is the pure-tone input signal, which is difference actually is introduced between the recovered together with distortion terms ; the main signal and the interference components, (13). latter are represented by the second term in such that the odd-harmonic components no r1 r , and 2 and 1'1 and 8 are respectively the longer give coherence with the main signal. 2 amplitudes and phase angles of the reflection As an example of the separation of crosstalk r coefficientsat each end of the feeder. is the time components, let us investigate the third-har l/v (x). delay along the feeder and equals where t is monic component of interference g3 Then it v 12 the feeder length and the group velocity of the is shown in the appendix, Section k waves. is the attenuation along the double e-2a1, length of feeder and equals where a is the gs(x) = { a(l +2y-2y2) f3(z)e-•'Jx'z5dz. x attenuation coefficient. lL� i00 (13) (1) 0 Equation has to be compared with Hence s = ..:iw with sin wat-the root-mean-square ampli x ..:iw= x(2)t. gs(x)/gs(x) = { (1+2y-2y2)/2(1 -y2/3) (11) tude being given by It is seen that p, (13) as it stands is not capable of expansion y a result that depends only on the position of in powers of owing to (A) the presence of (11) (waT)/war, s, Wa the channel. The right-hand side of varies sin which contains a characteristic O· 71 y = 0 0·92 y = O· 7 t -r from at to a maximum of at of the signal (B) the occurrence(C) of instead 0·87 y = 1. : and down to at Thus nearly the whole of t in the distortion terms, and the operator gs(x) d/dt. wa of is of this in-phase type, and may add up is the video frequency and is usually not voltage-wise from repeater to repeater. very high so that unless the feeder run is so large that it becomes comparable with the wavelength war · Distortion fro m, a Mis-Matched Feeder of the video frequency, will be small. Hence 8. (war)/war sin can be replaced by unity without Unless the analytic form of the output of (1) serious effect, and point (A) above is no longer a is known, the general analysis cannot be taken limitation. 2, any further, and the results so far presented are Point (B) is covered by Section which per intended for application when the various func mits all powers of above the first to contain a f (x) s tions n have been determined experimentally. constant delay. However, since the recovered (13) However, a particular case that is capable of signal in contains no delay, the distortion terms, although coherent with the main signal, complete treatment is that caused by phase dis waT· tortion due to a mis-matching at the end of a will differin phase from it by an angle (That long feeder line. In a previous paper2 on this such a delay must occur is apparent when it is f (x) i subject the function n for this case was real zed that the distortion terms arise from determined. waves that have been reflected down the feeder A frequency-modulated signal is transmitted and back again before transmission). Hence the in-phase components of odd-order interference from a long feeder, but owing to mis-matches at harmonics, although coherent with the main sig its ends, multiple reflectionsare set up within the nal, are not actually in phase with it, and the feeder. These cause a phase distortion of the phase difference changes from repeater to re L. Lewin, J. J. Muller, and R. Basard, "Phase Distor tion in Feeders," Wireless Engineer, 27, 143-145; peater, giving the appearance of a random phase 2 pp. May, 1950: and Electrical Communication, 27, 320- v. pp. change. Thus for all practical purposes, this com 323; December, 1950. v. ponent behaves as if it were incoherent, and 146 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com there is no call to extract it from the rest of the interference terms for special treatment. It will not add up voltage-wise from repeater to repeater. (11) d/dt Hence, using and (15), we get So far as the operator is concerned, if it were applied at the end of the analysis, no altera tion would be called for. It would produce merely (13) , a factor, which for would be nwa for the nth "' X Jn { z2(2)!r)zn+I e-z2!x' dz harmonic; and which, in the multi-channel case, rp i would be for the rth channel. Hence, if we =k{4ah,.(y) Pr1r (rp) xn-I n n12-1 -2r'x' d/dt 2 T 2 e . (16) quote results in which has been applied at an earlier stage, the only change necessary is the rp. replacement of nwa by The total crosstalk power in the rth channel 1s fn(x) The functions(13) for this problem are easily obtained from by the Fourier-Bessel expan sion and are given in (5) of the paper referred to. oo oo (J n (2rx)2n-2 rp X sin do Replacing nwa by as discussed above we get cos y(} L: (-0 -) n.1 (17) i0 2 hn(Y) (11). on replacing by its value as given by Summing, and taking the square root, we get for the total interference amplitude n The sin or cos occurs according to whether is 2wcT-01 -0 even or odd. Since the phase angle 2 is indeterminate without a precise knowledge of the where particular arrangement in use, we can, in con cf>(rx) = (4r2x2 0/0) sidering over-all behaviour of many units, replace cos yO (exp sin the terms in braces by their root-mean-square [� i"" values, in this case unity. However, this is only - 1-4r2x2 O/O} dO e-2'2"2• (18) an average effect, and for any particular feeder sin r at any one repeater this term will be quite defi 2wcr-01-0 Here, we have put nite, and only if 2 reduces to 45 de grees will the results be the same. But the effects of different repeaters, or of the same repeater under slightly different conditions of tempera wcr. wN=NP ture, etc., can be simulated by a variation of where is the angular frequency (video) This can be done conveniently either by small we corresponding to the top of the multi-channel changes of or by varying by means of a line band. r, (18) lengthener. In the ensuing analysis, the results The integral in cannot be evaluated in will be averaged out by taking the root-mean closed form, but very good approximations for (y = O) square values for the indeterminate phase term, the top and bottom of the band 1 and are 12.3. but the above statement must be borne in mind found in the appendix, Section The varia in the interpretation of any single result. 2! f (x) tion (for the integral) is always less than 1: Accordingly, we take for n over the video band, and since the form is much fn(x) =kr1r (rp)J,. {x2(2)�r} /x. 2 (15) simpler at the top, we will confine our attention there. However, it is seen from (18) that apart G(x) The integrations required by (11) can be per from this slow variation, is also porportional formed by means of the following integral.3 y, from the causes in to so that the interference a G. N. Watson "A Treatise on the Theory of Bessel vestigated in this example d Functions' " Secon Edition, Cambridge University Press, vanish at the bottom London; i944: p. 394, equation (4). channels of the band. June 1951 • ELECTRICAL COMMUNICATION 147 www.americanradiohistory.com 0·97. Using the approximations of the appendix, and The correct value, used in obtaining 12.3, 1, 0·925 Section for the top of the video band, we Figure is from the series expansion. It find the interference level in the top channel : follows that, for this example, we can use the (20) G(x)v�1 = G1 (a)!r1r wN asymptotic form of for runs longer than (say) :::::: 2 60 X { 1-(1 +4T2x2) ( -4T2x2)}!/2x, about feet, the interference amplitude for such exp Tx < l. (19) lengths dropping roughly as the inverse square root of the feeder length. WN 0·8 r1 r For larger feeder lengths If is megacycle per second and and 2 0·95 correspond to standing-wave ratios of and 0·8, (k = 1). we get, neglecting attenuation ! ! 1-0·95 1-0·8 0·8 G - X X X (19) (4 ) l+0·95 l+0·8 4 For very short feeders, gives X0·925 =0·263 X 10-3• G1::::::(2a) !r1r2WNT2X, TX <0·3. (21) 72 TX Thus the interference is about decibels down. The actual curve as a function of is shown 7 1. This is some decibels better than that obtained in Figure (3) 2, from of Reference from a consideration of the second harmonic only. I TX = 1, o.g :;�.. For the ratio of third to second har 0·8 ' ' monic, when each in turn is arranged to occur at � the top of the band, 2(2)! / 2(2)! 0·7 I is Ja { l J2 { l 0 I =0·28/0·48 =0·56, ·6 I""--.... so that the second harmonic 0·5 r--.. can no longer be considered dominant. For values � I ....__ TX 1; Jd 2(2)!Tx } 0·4 I of greater than decreases, pass -&- TX = 1 ·8 0 ·3 I ing through zero when , while the third, 0·2 I and subsequently the fourth and higher har O· monics "take over." The interference decreases I I TX 1, 0 IJ monotonically for values of greater than 3 0 2 4 5 6 ..,.,,. and there is no simple direct relation between it and the various harmonics, apart from the Figure 1-Inter-channel interference produced in a mis already-mentioned approximate equality to the matched feeder. The solid curve is for the equation second harmonic, when the latter is the domi si 0 ¢(rx) = cos o exp 4r2x2 � ) nant one. [�.(" { ( 0 It should be emphasized again that the be -1-4r2x2 si ao xe-4r2"2 � haviour described above is an average one, aver } r aged for different values of the normally inde while the asymptotic form of (20) shown dotted is 2wcT-81-8 . -0 1r2x2 terminate phase angle 2 For a par ¢ ""' 1·175 , rx >1·5 e ·;; y ticular set-up, this quantity may, for example, and the series form of (19) shown dashed is be a multiple of so that i 7r, ¢ ""' { 1 -(1+4r2x2)e-4r2"2 } , rx 55 As an example, consider a feeder feet long, In this case, there would be no second harmonic with a system in which the root-mean-square at all-and the interference would be correspond 2 frequency deviation is megacycles per second. ingly reduced-but such an example would not Assume the radio-frequency feeder is a waveguide adequately describe the characteristics of the in which the group velocity of the waves is 0·7 TX =27l"X55X12X2·54 unit, since the next one might well have a large light velocity. Then X2Xl06Xl/(0·7X3X l010)=1. second harmonic-and interference level-conse This is at the (19) (20) quent on a differentvalue of the phase. The only transition between and for which the TX TX = 1 0·95 way to obtain significant results is, as mentioned two functions of at are respectively previously, to conduct the test over a range of 148 June 1951 . ELECTRICAL COMMUNICATION • www.americanradiohistory.com values a e (by varying or l) and to take e of ph s The measurement of f2(x) and f3 (x) o r we the average of the results. This should then lead v a average range of values of root-mean-square amplitude to the harmonic levels and interference (15) (19) is not necessary here, and only the actual ampli levels of and respectively. tude that it is intended to use in the working link Recommended Test Procedure is needed. 9. In deciding whether or not the second and In order to estimate the interference level in third harmonics are the only ones of value, abso the rth channel, of video frequency rp, the follow lute smallness of these is not in itself a reliable criterion, although it may often be correct. For ing procedure is suggested : 7, A. Inject a signal of frequency rp/2 and root-mean example, in the application given in Section square amplitude and measure the level the second harmonic could be small either on of har monic appearing inz, the rth channel relative to the account of the feeder phase being nearly a multi signal appearing in channel (r/2). Vary z from zero ple of or because the length of feeder was 11', xr (15) to as large a value as the experiment may permit. actually such as to bring of into the region Then the relative harmonic amplitude is f2(z) as giving low second harmonic, the third or higher defined in (2). harmonics being dominant. In the first case, the Calculate g2(x) from f2(z) by means of (4). In B. smallness could be reflected in a low value of this derivation the root-mean-square level at which interference, albeit this effect would not be re it is intended to work the total video signal in peatable from repeater to repeater on account the link is presumed known. may usually be of the feeder phase angle not being exactly de taken as 1/4. The integration maya either be per formed graphically or by fitting f2(z) with a poly terminate. In the second case, the small value of second harmonic could be misleading. If we nomial b1z+baz3+bsz5+ ... From (5), the value of rx =1·8 g2(x) is { 4a:(1 -y/2) l t(b1x+3bax3+ 12b5x5+ ...) had and 2wcr-fli-02 =n11'+11'/2, both . second and third harmonics would be zero, al C. Repeat (A) but use a fundamental of frequency though the interference in this case would be rp/3. The signal appears in channel (r/3) and its kr1r2 third harmonic in channel The ratio of the ampli near its maximum. On the other hand, if tudes is fa (z). r. were very small (feeder well matched) the second and third harmonics would be small and D. Derive ga(x) f omf (z) using (7). Iffa(z) fitted is by b2z2+b z4+b5z6+ . then, from (8) this would be "genuine" in that the interference 4 r . .a 2 2 would be correspondingly small. Caution is there g (x) (1 3)/2 (3b 4 . 3 = {4a: -y / ll 2x + 12b4x +60b6x6+ ...) fore necessary before accepting low values of Repeat the above measurements for the higher E. second and third harmonics at their face value. harmonics using a fundamental of frequency rp/n and deduce gn(x) from fn (z) by (11). 10. Alternative Test Procedure F. The total inter-channel interference power level Since the bottom channel may, in practice, is given by G2 = g22(x) +g32(x)+g42(x) + . . As ex = 0, G2 . start somewhere around r 15 instead of it plained in Section 7, it may be necessary to make r = not possible to carry out the above analysis too a partial separation of the odd harmonics, to allow is for the in-phase effect. far, or over the whole of the band, as either the fundamental or its harmonics may be outside the Repeat the above steps with slightly different G. range of the apparatus at one end of the band or carrier frequencies or feeder lengths, to simulate the the other. Hence an alternative method is in use effects of repeater variations. in which a double-tone signal is injected and differencefrequencies are looked for. This method If it can be shown that only the second and the advantage of being usable over the whole third harmonics are significant, then it is per has band. So long as only the second and third har missible to avoid most of the above analysis, (6) (9) monics (of the single-tone test) are significant, and use simply an June 1951 • ELECTRICAL COMMUNICATION 149 www.americanradiohistory.com (23) the beat-frequency method no longer gives the Substituting in and picking out the various various harmonics accurately, any more than terms, we get (neglecting all cos (} terms except f (x) g (x) ai e) 2 can give 2 directly, as in the approxima cos (22). V= {x(2)lad tions of The order of errors is probably the cos O same in both cases, so that it is pointless to try + 2x2a . 2Co . !+(2x2)2a . 4C1 . { 2 4 t to use the preceding analysis without the use +(2x2)3a5. 6C . + ... l 2 3\ cos W of the correct functions fn(x) {except, as already (22) l · + [ {x(2)ll3as . 3Co. i+ {x(2)! }5as . 5C1 . 1\ stated, to the approximation of There is, + {x(2)!)7a •7C . -.14] 3e+ ... of course, a relation between the harmonic ampli 7 2 cos fn(x) , f (x) tudes on the one hand, and the beat If we put n =ratio of nth harmonic to funda frequency measurements on the other, but they mental, we get certainly are not of any simple form. It is hoped to investigate this at a later date. Meanwhile, if the full analysis of this paper is called for in any particular arrangement, there seems to be no 9 alternative to the full procedure of Section with its necessary limitations. a ! x ; 2n+2 2n-2 11. Co nclusions f4 (X ) - a2n+2X 2(2)!a1 7 2n-ln -1!n+3! Although the general validity of existing test etc. practice is confirmed in the case in which the second and third harmonics can be shown to be These formulae, which are relations between amplitudes, are obviously not affected if the term the only significant ones, caution is necessary in a1x(2)! 0 a1x(2)• (e+a) , accepting low absolute values of these harmonics cos is replaced by cos so at their face value, especially when long feeder that the phase of the recovered signal is of no runs are involved. The complex analysis and importance here. harmonic method of testing outlined here seem to 12.2 GENERATION OF INTER-CHANNEL be essential when the higher harmonics are not INTERFERENCE negligible. N Let us consider a multi-channel signal of Further work will be required to bring out the N I E channels, being large. f n represents a factor, relation between single-tone and double-tone 0 methods of test. which is 1 or according to whether the nth channel is working or not, we get the form 12. Appendixes S=x(2/aN)! LN En (npt+On) 12.1 HARMONIC GENERATION cos s =x(2/aN)!.s (26) Let the output for an input signal be given by (say). (23) N E 2 = N E = aN. Here, L n L n As explained in the text, ais a I 1 will be taken to be the recovered signal. This will be about a quarter, and the root-mean means that the terms of fundamental frequency S x. s square value of is produced by the higher powers of are negligible ; ai»an. If a power series of Sis under consideration, i.e., that (This will certainly be so in the we need to handle such terms as case of low distortion, which is the case of inter n n est here). s = (LN '---v------'1 1 n times +nC n-4(}+.•• ). (24) ...E 150 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com If the product of cosines is replaced by sum expansions of s2n and s2m are seen to differ by an 2 s2n and different terms, it is seen that there can be initial factor -<2m-2n). Hence the entire s2m (aN/4)m-nis all possible combinations of the form 2mproduced within with a factor (±a±b • • • ±n). Cm-n (m+n) !/2n! S COS s, If we use the full form instead of there appears an additional fac Many of these terms will be repetitions of each { x( 2 /aN) ! } 2m-2n, tor ! giving an over-all factor other, and when like terms are collected together {x/(2) } 2m-2n{2mCm-n(m +n) !/2n! } it is found that . The "multiplicity factor" in braces can also be n n s2m 2m s =CEN Er June 1951 • ELECTRICAL COMMUNICATION 151 www.americanradiohistory.com (28) m Using this result, the expression in can be tive and negative suffixes, plus the number of m n-m written positive and negative suffixes whose sum -r 2a1 2t 1 falls in the band (or +r). This is easily seen . e-z2 /x2j (z ) z2 n+2dz. (29) x-r 2n! x2n+2 (aN) n Jof"' 2n to be the coefficient of in N N N N m l m -l -m. z a1(2)l CE xl)n-m . CE x-z) +CL: x ) . CE x )n The extra power of and the factor appear because f (z) , which is a relative amplitude, had 2n ai(2)!z (25), been obtained by dividing by in and The amplitude sum therefore reduces to the (29). x-r this factor has had to be restored in coefficient of in N N In order to find the relative amplitude level of l m n) ! L:n n c (L:xl )n-m. CL: x- ) the interference arising from the terms of the !( m S2n, s2n. form we need to know the amplitude of 2nA, N N Denoting by the resultant root-mean-square s2n =!(n)!(L:;xl + L x-l)n. amplitude of those terms in that fall in the rth (29) a1x(1/aN) t, channel and dividing by (the root-mean-square level of a working channel), Summing the geometrical progression, we get we have the total interference level in the rth ' 1 -xN+t 1 -x-N-I n S2n (n) . + channel arising from terms. Denoting this 2 1 -x 1 -x (x) i ( ) g2n -1 level by we find 1 -xN+t n = !(n) ! (1 +x-N) . 2(2a)l 2nA, 1 { } (x) = 1-x g2n � ( anNn-t) x2n+a N N + 1 N When is large, we can replace by "'f 2n(z)e-z'fx' z2n+2 dz. X i (30) without much error, so that we can write S A similar relation is found when the power of x-N-xN n 2n (30), n · is odd, so that, writing for in we have x-r ( ) (32) n Xcoefficient of in l-x the general relation for all an E , The factor arises from the effects of n which a have so far been neglected. is actually the probability that any particular term is present, n and with independent terms2 occurring inn the nA,. sn, (nA,) a There remains the calculation of In the summations leading to the factor ac n-m m term with positive angles and negative cordingly appears. nC . n-m ! (32) angles has a coefficient m There are com Taking the square root of and expanding m! binations of positive angles and combinations both numerator and denominator by the bi of negative angles each giving the same term nomial theorem we get when the suffixes are summed. Hence, if there A, a n ) x-• am n ! [!( ! Xcoefficient of in are combinations of suffixes giving a re = n (1-x)-n{x-nN _ nc1 x- 152 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com An alternative formulation is given putting is the coefficient of by in + CX(n-l)/2 xr x-N -x N n x - n N N "" , ( 1-x ) ( X'� -x=�x ' ) CE x l) x = ei e-irfJ writing fl, and finding the coefficient of the progression, mak by Fourier analysis. This gives Summing geometrical and ann ! (2,,. r N(J n ing the same approximations as before, for large (nAr)2 = i f! s n d(J. N, 411" e ( � {3/2) we get J0 sm r Ol.(n-1)/2 "" x N r y = r/ N coefficient of in For and both large, and finite and - N < + )/ N ( - )/ 1, (3 = 0 1 x n 1 2 -Nl -x n 1 2 less than only the range near and is ( ) (x ) 'Ir 1-x 1-x important in the integrand. Changing the vari (3 O=N(J, l -xN n able from to the range for (}can be taken xr x-N 153 June 1951 • ELECTRICAL COMMUNICATION www.americanradiohistory.com APPROXIMATION TO (TX) 12.3 J:"'(sin e/e) 4d0 = 1 · � Let 23 ( I ()) 5de s F(y, i\) "' yO(e1'-sine;e_ 1 -i\ O/O)dO. {"' s (j _'Tr (24 ) = [ cos sin Jo m. - 2 ·s· • 0 (i\= 4T2X2.) )6d ()- {"' c s · ; () �2 . i_10. 12 J m (} (l!) y = 1 0 At the top and bottom of the video band, 0 ( )8d6 _§_ and respectively. s 6 () -�- 2 · 14 180 ' etc. 0 m. I (�) We define i"' · Fo=F(O, i\) = "'( e>-•ine;e _ 1-i\sin 0/0)d0, (36) From this we get the approximation i n ( "' F(l i\) = cos O(e>-s ne e_1 . e/e)nde� 7r F1= , i"' i 1 -i\ sin O/O)dO. Jo (sm 2 2(n-1) ' (37) n=4, which is exact to and still nearly correct at n = 8. Integrating by parts, we get Thus a good approximation in the neigh e>-sine/e _ l -i\ O/O) o bourhood of the origin is Fo =O( sin -i\ 0/02) (e>-sin8/8 _ 1)d0 i"'O(cos 0/0 -sin = i\ O(eXsin8/8 _ l)dO 2) i"'COS Jahnke and Emde define (page "' +i\ sin (j(eXsin8/8_ 1)dO l0 (et-1)/t dt. () Ei(x) =log 'Yx+ .Jo(x = -i\ 6(eXsine;e _ 1-i\ 6/0)d() .f''cos sin Hence "' e e F0� (7r/4)i\ {Ei(i\) -log 'Yi\ } '}' =0·577) . (40) _ i\2 ( cos sin de (log Jo () (40) The probable limit of validity of is when the +i\_i_ ("'( eXsine;e_1-i\ sin O/O)dO ai\ eighth term of the exponential series becomes Jo 4. i\ 4T2X2 significant, which is about i\= Since = a 1 (38) = -i\F1 -i\2 +i\ Fo TX < 47r 0i\ this gives as the upper limit. From aFo F0 a .. F1 = ai\(Fo/ -i\ . (38) �-T-i\"4=i\7r i\) 47r F0 i\»4, This relation gives F1 in a simple way when is To get a form usable in the range we need Fo, (36) found. To find expand and integrate. an asymptotic form for (sin 0/6)nde for large oo j\n i"' Fo = L, e/e)n de. (39) n. n n. i"'o (sin Now when is large, only the range near 2 I 6 = 0 is of value in the integrand. The forms e/e) The following values are easily obtained (e.g., e-n8216(1-nl;l2/180) and (sin n are the same up 13)- 64, Whittaker and Watson, p. 123, example to powers of and can be shown to give 154 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com This last form is very good, even for so low a Equation could also have been obtained , (42) value as n = 2. However, it cannot be used albeit in a rather superficial manner (on account directly as the resulting series cannot be easily of the path of integration for ) , by the substitu handled. Now tion sin O/fJ =cos directly in (36), and using ! +l)/2 series expansions of fJ and cos e in terms of [or/ fJ)n-lf5dfJ 2 }!. • o (cos ""'271"{ 2 7r(n+3/10) Hence Putting = /2,A.02 and taking the upper limit of fJ/fJ)nd(J""' fJ)n-lf5dfJ. cf> to , we find l"'(sin (3)!1"12 (cos oo Fo ""'e'{3/(2A.)P {"'e - The first term of the expansion of (41), which is From (38) we find, for the asymptotic form of F1 the correct one, is equal to the first term of the expansion of (42) within 10 per cent. The suc This is valid values ceeding terms in (42) rapidly approach their form for of A. greater than Tx> correct values. about 4; i.e., 1. Recent Teleco mmunicatio n Development Miniature Toroidal Inductor THESE DAYS N of miniature components, an I interesting development item is a high-Q toroidal inductor manufactured by Standard Telephones and Cables, Limited, which is shown in the accompanying photograph. The core is pressed from radio-frequency iron powder as a single ring 0.106 inch (2. 7 milli meters) thick with outside and inside diameters of 0.375 and 0.187 inch (9.5 and 4. 7 millimeters). Its effective permeability is about 14. An inductance of 27±0.5 microhenries with a Q of approximately 120 between 2 and 4 megacycles per second is achieved by the ap plication of a single layer of size 38 Steel Wire Gage (0.0060-inch-diameter) enameled wire. The coil is machine wound and can be pro duced with a tap taken from the center of the winding. Its weight is 0.82 gram. June 1951 ELECTRICAL COMMUNICATION 155 • www.americanradiohistory.com Recent Teleco mmunication Development Television Scanner for Slides ELEVISION broadcasting stations fre mits lap dissolves, fades, and other effects such quently transmit test and station as montages of two slides or of one slide and a identificationpatterns and other "still" "live" program. All types of superposition may materialT such as spot news announcements and be effected including the insertion of spot an messages from the sponsors of the programs. It nouncements into programs. is uneconomical to use the regular pick-up equip The equipment is based on the flying-spot ment for these transmissions. They can be re type of scanning and produces a video-frequency corded on standard 35-millimeter photographic signal having a horizontal resolution of at least film and converted into a corresponding electrical 600 lines, a signal-to-noise ratio of not less than 1 television signal by means of a compact consol 36 decibels, and a contrast range of 30 to or mounted unit recently developed by Federal better. The geometric distortion will not exceed Telecommunication Laboratories. 1 2 percent. The output signal conforms to the The basic unit will handle from to 36 double television transmission standards of the United frame slides of the 2-inch-square size, which may States. be shown in or out of sequence. The signal is automatically blanked out while the slide is in Dual flying-spot scanner for developing television sig motion. nals corresponding to the images on photographic-film A second "projector" may be added and per- slides. 156 ELECTRICAL COMMUNICATION June 1951 • www.americanradiohistory.com Ju 11ll.emnriam HERMAN T. KOHLHAAS ERMAN KOHLHAAS T. was born at Brooklyn, became a member of the staff of the Inter 28, 1882 New York, on December . He national Telephone and Telegraphic Corporation. H Electrical Communication received the B.S. degree in electrical engineering Mr. Kohlhaas edited 1907 1937 1939 from Cooper Union in and later was awarded at London from through and, while engineering degrees by that institution and by there, served as publicity representative of the Brooklyn Polytechnic Institute. International Telephone and Telegraph Corpora 1905 1922, From to he served in the labora tion and International Standard Electric Cor tories of the Western Electric Company as an poration. engineer, section head, and finally as a division He was appointed an assistant vice president 1922, head in the physical laboratory. In he of International Telephone and Telegraph Cor 1945, 1947 was appointed executive personnel assistant to poration in and retired at the end of . the systems engineer of Bell Telephone Labora He served as a consulting editor of the magazine tories. for the following year. He was transferred to the Western Electric He was a Fellow of the American Institute 1924 Company headquarters in . The following of Electrical Engineers, a Senior Member of year, he was assigned to statistical studies and the Institute of Radio Engineers, and a Member Electrical Communication the editing of for the of the Norwegian Chamber of Commerce. International Western Electric Company. Mr. Kohlhaas died in an automobile accident 24, 1951. Through the purchase of that company, he that occurred in Florida on April June 1951 • ELECTRICAL COMMUNICATION 157 www.americanradiohistory.com Contributors to This Issue C. C. EAGLESFIELD was born on June 28, 1906, at Swansea, Wales. He re ceived the B.A. degree in mechanical science from the University of Cam bridge in 1928. From 1929 to 1932, he was a research engineer in the laboratories of Standard Telephones and Cables in London and Paris. He then entered the research laboratories of the Mullard Radio Valve Company, where he specialized in the television and high-frequency fields. Since 194 7, he has been in the valve division of Standard Telephones and Cables at Ilminster, where he is in charge of a group working on valve development. . ROBERT HUGHES WILLIAM DITE W. WILLIAM DITE was bornat New York HORACE G. MILLER was born on City on March 2, 1919. He received the July 16, 1907, at Kansas City, Kansas. ROBERT HUGHES was born on w. He received a degree in electrical B.E.E. degree from the College of the March 21, 1922, at New York City. He City of New York in 1940. engineering from Karisas State College received the B.S. degree in electrical in 1936. From 1940 to 1943, he was with the engineering from Cornell University in In 1928, he joined the radio engineer Signal Corps Laboratories, Fort Mon · 1943. ing department of Westinghouse Elec mouth, New Jersey, working on sound He served in the United States Signal tric Corporation, where he worked on ranging and radar equipment. Corps during the next three years and television. During 1930, he was with introduced the first Army pulse-time Since 1943, he has been associated Jenkins Television Corporation. From with Federal Telecommunication Lab modulation equipment in the European 1931 to 1936, he was engaged in the and Pacific theaters. From 1946 to oratories, Nutley, New Jersey, where his development of television and facsimile 1948, he worked on radar plotting work has been largely concerned with equipment for Radio Inventions. He boards for Electronic Associates. communication systems employing returned to Kansas State College that Since 1948, Mr. Hughes has been pulses. summer and received his degree. Mr. Dite is a member of the Ameri with Federal Telecommunication Lab oratories working on pulse-time modu In 1936, Mr. Miller joined the tele can Institute of Electrical Engineers. lation and is now a project engineer. vision research department of Philco Mr. Hughes is a Member of the Corporation. In 1938, he went to A. B. . . . Institute of Radio Engineers. DuMont Laboratories to design tele- LEONARD LEWIN was born in 1919 at Southend-on-Sea, England. He studied mathematics with particular reference to transcendental functions and the electromagnetic theory of radiation. During the war, he did research work on antennas and mirrors at the British Admiralty and in 1945 served as chair man of the Inter-Service Committee on Radar Camouflage. He joined the engineering staff of Standard Telecommunication Labora tories in London in 1946. EAGLESFIELD C. C. HORACE G. MILLER 158 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com versity in 1943. He served as a teaching assistant at Columbia during 1944. From 1944 to 1946, he was engaged in electronic work for the United States Navy. In 1946, he joined the engineering staff of Mackay Radio and Telegraph Company where he has been active in developing terminal equip ment for radio and cable systems. Mr. Mine is a Member of the In stitute of Radio Engineers. . . . ScoTT NEVIN was born on September 11, 1924, at Ithaca, New York. He received the B.S. degree in mechanical ANATOLE MINC engineering from Cornell University in F. 1945. P. PANTER He joined the Capehart-Farnsworth vision receivers and wideband oscillo PETERMAN was born on scopes. From 1940 to 1942, he was with Corporation in 1946 and has worked on w c. March 28, 1889 at Linfield,· Penn Panoramic Radio working on pano photocathodes and secondary-emission sylvania. He received the B.S. degree ramic frequency-scanning receivers. devices. in electrical engineering in 1911 from Since 1942, Mr. Miller has been a . . . project engineer at Federal Tele Lehigh University. communication Laboratories on the PHILIP F. PANTER was born in 1908 From 1912 to 1928, he was engaged development of portable radar, missile in Poland. After early schooling in in ocean-cable development for West controls, pulse-time modulation, and Tel-Aviv, Palestine, he later received ern Union Telegraph Company. He television systems. from McGill University, Montreal, continued in this field for All America Mr. Miller is a Senior Member of the Canada, the following degrees: B.Sc. Cables and Radio and other associates Institute of Radio Engineers. in 1933, B.Eng. in electrical engi of the International Telephone and neering in 1935, and Ph.D. in physics Telegraph Corporation until his retire in 1936. He continued research in ment in 1950. He holds a number of spectroscopy at McGill for an addi patents including that on the rotary tional year. ANATOLE Mrnc was born on August regenerative repeater for automatic After teaching mathematics and 22, 1918 at Rostov-on-Don, Russia. telegraphy. He received the baccalaureate in physics in Palestine for a year, he Mr. Peterman is Member of the philosophy from the University of Paris returned to Canada as assistant pro American Institute of Electrical En in 1937. After studying in several fessor of mathematics and physics in the a European universities, he came to evening division of Sir George Williams gineers and of the Franklin Institute. America and was awarded a B.S. in College in Montreal. He served also on electrical engineering by Columbia Uni- the staff of the physics department of McGill University as instructor in physics and later as part-time lecturer, until the end of 1945. Early in 1941, Dr. Panter joined the transmitter department of the Cana dian Marconi Company in Montreal. In October, 1945, he was appointed senior engineer, responsible for the de velopment of frequency-modulation broadcast equipment, at Federal Tele phone and Radio Corporation. He later transferred to Federal Telecommunica tion Laboratories and is now in charge of the theoretical group of the com munications division. Dr. Panter is a member of the Insti tute of Radio Engineers and the Radio Club of America. SCOTT NEVIN . . . W. C. PETERMAN 159 June 1951 • ELECTRICAL COMMUNICATION www.americanradiohistory.com HANS SALINGER was born at Berlin, Germany, on April 1, 1891. He received the Ph.D. degree in 1915 from the University of Berlin. From 1919 to 1925, he was a research associate at the Reichspostzentralamt and from 1929 to 1935, a professor at the Heinrich Hertz Institute, both in Berlin. In 1936, he joined Farnsworth Tele vision, now Capehart-Farnsworth Cor poration, where he is engaged in research work. He is also doing part time teaching at Purdue University and has published numerous technical papers. Dr. Salinger is a Fellow of the GEORGE ROYDEN Institute of Radio Engineers and a NELSON WEINTRAUB T. member of the American Physical Society, American Association for the GEORGE T OR RoYDEN was born Advancement of Science, and Sigma Xi. From 1943 to 1946, he was a research at Fort Clark,AYL an army post in Texas, assistant at Polytechnic Institute of on June 20, 1895. He received a B.A. Brooklyn, working on microwaves and degree in 1917 and an engineering de teaching electrical engineering. gree in 1924 from Stanford University. In 1946, he joined Federal Tele He was employed by Federal Tele LEO STASCHOVER was born on May communication Laboratories becoming graph Company at Palo Alto, Cali 8, 1922, at Saarbriicken, Saar Terri a senior engineer and was engaged in fornia, on part time in 1916 and full tory. He received the B.S. degree in the development of radio links for time after graduation from college. He electrical engineering in 1943 from the communication, frequency modulation, was engaged in the design of arc trans College of the City of New York and and television applications. In 1951, mitters of powers up to 1000 kilowatts. the M.S. degree from Polytechnic he entered the television department of From 1919 until 1925 he was at Institute of Brooklyn in 1948. Paramount Pictures Corporation. Mare Island Navy Yard. His duties Mr. Staschover is a member of Tau included work on Navy radio stations Beta Pi, Eta Kappa Nu, and Sigma Xi, in San Diego, Hawaii, and Alaska. and an Associate of the Institute of He returned to Federal Telegraph Radio Engineers. Company in 1925 to do research work • • • on broadcast receivers for operation on alternating current. NELSON WEINTRAUB was born at In 1927, he joined the newly organ New York City on April 9, 1923. He ized Mackay Radio and Telegraph received a B.S. degree in electrical Company becoming division engineer. engineering from the College of the From 1936 to 1946, he was with Fed City of New York in 1944 and a eral Telegraph Company in Newark, master's degree from Polytechnic In New Jersey. He then returned to Mac stitute of Brooklyn in 1946. kay Radio and Telegraph Company. He has been with Federal Tele Mr. Royden is a Fellow of the In communication Laboratories since 1944 stitute of Radio Engineers and a Mem and has worked on color television. At ber of the American Institute of Elec present he is a senior engineer on pulse trical Engineers and of Sigma Xi. time-modulation systems. HANS SALINGER . . . 160 ELECTRICAL COMMUNICATION • June 1951 www.americanradiohistory.com INTERNATIONAL TELEP:HON:E CORPORATION AN D TE!LEGRAPH Associam MaauEact:urin9 a d Sales Compa nies n United St:ates ocE Amenc:a Europe and Far Eau International Standard Electric Corporation, New York, Veteinigte Telephon- und Tel egrapk_n fabrik s Aktienges ell- New York schaf:t Czeija, Nissl Co., Vienna,. Austria Federal and Radio Corporation, Clifton, New & Jersey'.l'e lepho ne Bell Telephone Manufacturing Company, Antwerp, Belgium International Standard Trading Corporation, York, New China Electric Company, Limited, Sh anghai. China York New Capehart-F arnaworth Corporation, Fort Wayne, Indiana Standard Electric Aktieselskab, Copenh ag�n. Denmll,l'k Fl ora Cabinet Company , Inc. , Fl-O ra,. Indiana Compagnie Generale de Constructions Telepho niques, Paris, Thomasville Fu rniture Corporation, Thomasville, North Prance Carolina Le. Materiel TeleIJ lwnique , Paris, France Les Teleimprimeurs, Paris, France Grell! Bribin and Domllliions Standard Telephones and Cables, Limited, Lo ndo n, England C. Lorenz, A.G. and Su bsidiaries, Stuttgart, Germany Limited, Creed and Company, Croydon, England Genest Aktieng�eHschaf.t and Subsidiaries, Stuttgart, Mix & International Marine I' tadio Company Limited, Croydon, Germany England Siiddeutsehe Appar atefab ri:k Gesellschaft Ko lster-Brandes Limited, Sidcup, England m.b.H., Nurem bel'g, Germany Standard Telephones and Cables Pty. Limited, Sydney, Auatr1Jclia Nederland sehe Standard Electric Maatsehapp ij N.V. , The Silovac Electrical Products Pty. Limited, Sydney, Australia Hague , Netheclands Austral Standard Cables Pty. Limited, Melbourne, Australia Fabbrica. Appareechiature per Comunicazioni Elettriche, Zealand Electric Totalisators Limited, Wellington, New Milan, Italy New Zealand Standard Telefon og Kahe liabrik A/S, Oslo, Norway Federal E.'l eotric Manuiaeturing Company, Lt d., Montreal, Canada St andard Electrica, Lisbon, Portugal Companie. Radio Aerea Maritima Espanola, Madrid,. Spain Smd: h Ame rka Compama Standard Argentina, Sociedad An6nima, Electric Standard Electrica, S.A., Madrid, Spain Indu atrial y Comernial, Buenos Aires, Argentina Aktiebolaget St andard Radiofabr ik, Stockholm, Sweden Standard Electrica, 8.A., Rio de Janeiro , Brazil Compafiia St andard Electric, S.A.C., Santiago, Chile St andard Telep:_ho ne et Ra,dio S.A•. Zurich, Swi tz13rJa,nd elepbone Operali ag Syste ms T Compaiila '.l' elefonica Argentina, Buenos Aires, Argentina Compafiia Teleronica de Magallanes S.A., Punta Arenas , Chile Cuban American Tel ep hone and Telegraph Company, Ha� Compaiiia Teleg,rafico-'.l' elefOnica Comercial, Buenos Aires, vana, Cu a Arll'entina b Cuban Compan y, HaYana, Cuba Compafiia 'l'elegrafico-TeleiO nica del Buenos Aires, Tele:Pho ne Argeatina Plata , Compafiia Pe:ma na d:e Telefonos Limifo.d:a, Lima, Pem Porto Rico Telephone Company, San Pue�to Rico Companhia Telefonica Nacional, Porto Alegi:e , ,Juan, Brazil Shanghai Telephone Compan_y, Fe deral Inc. U.S.A., Rh ang� Compafila 'l' elefonos· de Chile, Santiago, Chile hai, China de Ra diol:elephone and Rad iot:�lepap b Operati :ng Compa;1:1 ies Compafila de Radio, Buenos Aires, Argentina Compaiiia Internacional de Radio, S.A., Sa:ntiag9, Chile lntemacional Compafila Radio Boliviana, I,a Paz, BolLvia Intemacionsd de Radio Corporation of Cuba, Havana, Cuba Companhia Radio Internacional Brasi l, Rio de Janeiro, do Radio. Corporation of Porto Rico, San Juan, Puerto Rico Bra zil Cable Ra.cliotel_egrap lt Opercuin.g Companies and (Controlled American Cable Radio Corporatio n,. York, New York) by & Ne:w The Commercial Cable Company, New York, New York1 mel'iea Cables· and Hadio, Inc., New York, York• All A New Mackay Radio and Teleigaph Company, New York, New Sociedad Radio Argmi:tina,. Aires, Ar gentina' An6nima Buenos York' Cable service. 'Inte �national and·ma rine radinteiegmtril services·. 1 3 Cable and radiotelegi:aph services·, Radintel egr.J!Jl!J service. • Telecommunication Laboratories, Inc., Nutley, New Laboratoire de Telecommunications,. France FedeJersayral Central Pari s, International Teleeommunication Laboratories, Inc., New St andard Telecommunication Laboratories, Limited, London, York, New York England www.americanradiohistory.com