A New Single-Phase Direct Frequency Controller Having Reduced Switching Count Without Zero-Crossing Detector for Induction Heating System

Article A New Single-Phase Direct Frequency Controller Having Reduced Switching Count without Zero-Crossing Detector for Induction Heating System

Naveed Ashraf 1, Tahir Izhar 1, Ghulam Abbas 2,* , Ahmed Bilal Awan 3 , Ali S. Alghamdi 3,* , Ahmed G. Abo-Khalil 3, Khairy Sayed 4 , Umar Farooq 5 and Valentina E. Balas 6

1 Department of Electrical Engineering, University of Engineering and Technology, Lahore 54890, Pakistan; [email protected] (N.A.); [email protected] (T.I.) 2 Department of Electrical Engineering, The University of Lahore, Lahore 54000, Pakistan 3 Department of Electrical Engineering, Faculty of College of Engineering Majmaah University, Majmaah 11952, Saudi Arabia; [email protected] (A.B.A.); [email protected] (A.G.A.-K.) 4 Faculty of Engineering, Sohag University, Sohag 82524, Egypt; [email protected] 5 Department of Electrical Engineering, University of the Punjab, Lahore 54590, Pakistan; [email protected] 6 Department of Automatics and Applied Software, “Aurel Vlaicu” University of Arad, 310130 Arad, Romania; [email protected] * Correspondence: [email protected] (G.A.); [email protected] (A.S.A.)

Received: 3 February 2020; Accepted: 26 February 2020; Published: 4 March 2020

Abstract: Induction heating (IH) is an environmentally friendly solution for heating and melting processes. The required high-frequency magnetic ﬁeld is accomplished through frequency controllers. Direct frequency controllers (DFC) are preferred to dual converters as they have low conversion losses, compact size, and simple circuit arrangement due to low component count. Numerous frequency controllers with complex switching algorithms are employed in the induction heating process. They have a complicated circuit arrangement, and complex control as their switching sequences have to synchronize with source voltage that requires the zero-crossing detection of the input voltage. They also have a shoot-through problem and poor power quality. Therefore, this research proposes a novel frequency controller with a low count of six controlled switching devices without a zero-crossing detector (ZCD) having a simple control arrangement. The required switching signals are simply generated by using any pulse-width-modulated (PWM) generator. The performance of the proposed topology is veriﬁed through simulation results obtained using the MATLAB/Simulink environment and experimental setup.

Keywords: frequency controller; domestic induction heating; power quality; shoot-through problem; zero-crossing detector

1. Introduction Nowadays, domestic and industrial induction heating is one of the hot research areas due to low conversion losses, low processing time, pollution-free operation, and accurate power control [1]. It is employed in many applications such as domestic induction heating (induction cooker, domestic induction appliances burners, etc.) [2–4], medical applications [5], and wireless power transfer [6]. Its basic operating principle is to apply a high-frequency magnetic ﬁeld to the heating targets. The typical frequency range for these applications lies from 20 kHz to 100 kHz. The output frequency can be regulated through dual converters [7,8] and direct alternating current to alternating current (ac-to-ac) converters [9–13].

Electronics 2020, 9, 430; doi:10.3390/electronics9030430 www.mdpi.com/journal/electronics Electronics 2020, 9, 430 2 of 16

Conventionally, high-frequency IH systems are implemented in two conversion stages. The first stageElectronics involves 2020 rectification,, 9, 430 where ac grid voltage is converted to dc voltage through the alternating2 of 16 current to direct current (ac-to-dc) converters. A low pass filter is required to reduce the ripple factor of the outputConventionally, voltage. The high-frequency improvement IH in systems output are ripple implemented deteriorates in two the powerconversion quality stages. of theThe input first stage involves rectification, where ac grid voltage is converted to dc voltage through the current. Many control techniques and switching algorithms are developed to improve the power alternating current to direct current (ac-to-dc) converters. A low pass filter is required to reduce the quality by using feedback controllers such as fuzzy or sliding mode controllers, as reported in [14]. ripple factor of the output voltage. The improvement in output ripple deteriorates the power quality In theof secondthe input conversion current. Many stage, control the rectifiedtechniques output and switching is converted algorithms to high-frequency are developed acto improve voltage with the helpthe power of a resonant quality converter.by using feedback Various topologiescontrollers such of resonant as fuzzy converters or sliding inmode the formcontrollers, of half-bridge as and full-bridgereported in [14]. series In resonantthe second inverters conversion are stage, developed the rectified [15–17 output] to obtain is converted the required to high-frequency output voltage. In dualac voltage converters, with anthe ac help voltage of a resonant is converted converter. to variable Various ac topologies voltage and of resonant frequency converters indirectly, in the so these convertersform of are half-bridge also known and asfull-bridge indirect series converters. resonant But inverters they are are complex developed and [15–17] have to high obtain conversion the lossesrequired and high output cost voltage. due to aIn large dual count converters, of the an switching ac voltage devices. is converted to variable ac voltage and Directfrequency ac-to-ac indirectly, converters so these converters are developed are alsoto known overcome as indirect the converte demeritsrs. But of thethey indirectare complex ac-to-ac and have high conversion losses and high cost due to a large count of the switching devices. converters as they have a simple circuit arrangement, low conversion losses, and fewer switching Direct ac-to-ac converters are developed to overcome the demerits of the indirect ac-to-ac devices. Ac voltage controllers regulate the output voltage both in buck and boost fashion [18–21]. They converters as they have a simple circuit arrangement, low conversion losses, and fewer switching can bedevices. employed Ac voltage as frequency controllers controllers regulate if the they output have non-invertingvoltage both in and buck inverting and boost characteristics fashion [18–21]. [22 –26]. TheseThey matrix can converters be employed regulate as frequency the output controllers voltage through if they dutyhave ratio non-inverting control of theand high-frequency inverting switchingcharacteristics devices. [22–26]. However, These their matrix switching converters sequences regulate are the complex output and voltage have through high switching duty ratio voltage and currentcontrol stresses.of the high-frequency The problem switching of high switching devices. However, voltage is their solved switching in [9,10 sequences] by using are eight complex controlled switches,and have but theyhigh requireswitching complex voltage switchingand current sequences. stresses. The problem of high switching voltage is Complexsolved in [9,10] switching by using algorithms eight controlled and switches circuit complications, but they require are complex the major switching causes sequences. of less use of the single-phaseComplex matrixswitching converters algorithms (SPMC)and circuit in complica inductiontions heating are the major applications. causes of less A directuse of the ac-to-ac single-phase matrix converters (SPMC) in induction heating applications. A direct ac-to-ac converter converter topology (see Figure1) with a modified switching algorithm is developed in [ 27] to enhance topology (see Figure 1) with a modified switching algorithm is developed in [27] to enhance the the potencypotency ofof SPMCSPMC for the induction heating heating application. application. It only It only requires requires two twocomplementary complementary pulse-width-modulatedpulse-width-modulated (PWM) (PWM) signals signals toto operateoperate it it as as a aresonant resonant converter converter or direct or direct frequency frequency converter.converter. These These two controltwo control signals signals control control the switching the switching states states of the of eight the semiconductoreight semiconductor switching devices.switching Arduino devices. Mega Arduino 2560 Mega is employed 2560 is employed to generate to generate the gatingthe gating signals, signals, and and thesethese signals signals are synchronizedare synchronized with input with ac input voltage ac voltage with the with help the of ahelp zero-crossing of a zero-crossing detector detector (ZCD). (ZCD). Eight isolatedEight dc powerisolated supplies dc withpower eight supplies gate-driving with eight circuits gate-drivi are requiredng circuits to switch are required the eight to semiconductor switch the eight switches of thesemiconductor dual bridge converter. switches of An the LC dual filter bridge is used converte at the inputr. An sideLC filter to suppress is used theat the unwanted input side harmonics to to improvesuppress the the input unwanted power harmonics factor. The to improve topology the su inputffers frompower the factor. problem The topology of the short-circuiting suffers from the of the problem of the short-circuiting of the input capacitor of the switching transistor of each converter leg input capacitor of the switching transistor of each converter leg operated in a complementary fashion. operated in a complementary fashion. Also, the zero-crossing detection of the input voltage in every Also,cycle the zero-crossingincreases the control detection complexity. of theinput voltage in every cycle increases the control complexity.

Q1 D3 Q5 D7

D1 Q3 D5 Q7

Resonance V Cf in Load

Q6 D Q2 D4 8

D D2 Q4 6 Q8

Figure 1. Direct alternating current to alternating current (ac-to-ac) resonance converter. Figure 1. Direct alternating current to alternating current (ac-to-ac) resonance converter. Electronics 2020, 9, 430 3 of 16

To overcome the aforementioned limitations of [27], this research develops a novel direct frequency changer for domestic induction heating applications having a low count of six switching devices and without a ZCD. The desired output frequency is obtained by operating the proposed converter in non-inverting and inverting modes. The proposed converter has no shoot-through problem. Therefore, there is no need to use the voltage snubber circuits for the protection of the high-frequency switching devices. Its operation only requires the generation of two high-frequency PWM switching signal in a complementary fashion. These signals can be generated with any PWM generator, as there is no need to detect the zero-crossing of the input voltage as switching frequency is kept much higher than that of the source as typical switching frequency lies in the range from 20 kHz to 100 kHz for domestic induction heating application. The control effort of the controller is reduced due to fewer numbers of the control signals and the absence of a zero-crossing detector. The six switching devices only require six isolated dc supplies and six gate-driving circuits. It reduces the two switching devices, two isolated dc supplies, two gate drive circuits, and one zero-crossing detector. Therefore, it reduces the overall size, cost, and conversion losses (thus improves conversion efficiency). Zero voltage switching of the switching devices is also ensured by selecting the switching frequency greater than the resonance frequency. The switching of high-frequency switches in a complementary fashion ensures the improved power factor and low total harmonic distortion (THD) of the input current. The hardware setup is developed to validate the simulation results obtained by modeling of the proposed converter in MATLAB/Simulink environment. The paper is constructed in the following way. Section2 demonstrates the circuit operation of all possible operating modes comprehensively. The performance of the proposed converter, in terms of components count, is computed and compared with the existing converters in Section3. Section4 involves the simulation and practical results to validate the effectiveness of the proposed converter system. The conclusion is drawn in Section5.

2. Operation of the Proposed Topology Figure2a,b depicts the block diagram and the circuit topology of the suggested converter with gating signals. The operation of the proposed converter operating as a direct frequency changer (DFC) is divided into non-inverting and inverting modes. The controlled switching devices S1, S3, S5 and S2, S4, S6 are controlled in a complementary fashion with gating signal x1 and x2, respectively. There may be an open circuit problem of the inductor current due to switching of the controlled devices in a complementary way. This issue is tackled by inserting the small turn-oﬀ delay (Toﬀ) in the switching signals (x1 and x2), as depicted in Figure3, where Tc and Tp are the non-shoot through and the total conduction time of the switching devices, respectively. The switching and inductor currents are easily commutated from one switching pair to another as the proposed topology is free from the current shoot-through problem. The control signals are directly generated with the PWM generator. There is no need to use ZCD to synchronize with input ac voltage as diode D5, and D6 only conducts for positive and negative half cycles of the input voltage, respectively. The gate drive circuits isolate the low voltage switching signals Vg1 and Vg2 from the high voltage of DCF. The Ro, Lo, and Co represent the resistance, inductance, and capacitance of the heating load, respectively. The output frequency is governed by operating the proposed topology in the inverting and noninverting modes with almost a unity voltage gain, ensuring equal rms output and input voltage (see Figure4). The detail of each operating mode of the proposed converter is explored in Figure5 with the help of switching states and highlighted paths of power transfer from input to output. Electronics 2020, 9, 430 3 of 16

To overcome the aforementioned limitations of [27], this research develops a novel direct frequency changer for domestic induction heating applications having a low count of six switching devices and without a ZCD. The desired output frequency is obtained by operating the proposed converter in non-inverting and inverting modes. The proposed converter has no shoot-through problem. Therefore, there is no need to use the voltage snubber circuits for the protection of the high-frequency switching devices. Its operation only requires the generation of two high-frequency PWM switching signal in a complementary fashion. These signals can be generated with any PWM generator, as there is no need to detect the zero-crossing of the input voltage as switching frequency is kept much higher than that of the source as typical switching frequency lies in the range from 20 kHz to 100 kHz for domestic induction heating application. The control effort of the controller is reduced due to fewer numbers of the control signals and the absence of a zero-crossing detector. The six switching devices only require six isolated dc supplies and six gate-driving circuits. It reduces the two switching devices, two isolated dc supplies, two gate drive circuits, and one zero-crossing detector. Therefore, it reduces the overall size, cost, and conversion losses (thus improves conversion efficiency). Zero voltage switching of the switching devices is also ensured by selecting the switching frequency greater than the resonance frequency. The switching of high-frequency switches in a complementary fashion ensures the improved power factor and low total harmonic distortion (THD) of the input current. The hardware setup is developed to validate the simulation results obtained by modeling of the proposed converter in MATLAB/Simulink environment. The paper is constructed in the following way. Section 2 demonstrates the circuit operation of all possible operating modes comprehensively. The performance of the proposed converter, in terms of components count, is computed and compared with the existing converters in Section 3. Sections 4 involves the simulation and practical results to validate the effectiveness of the proposed converter system. The conclusion is drawn in Section 5.

Electronics 2020, 9, 430 4 of 16

Electronics 2020, 9, 430 4 of 16 (a) S4 S2

D4S4 DS2

D4 D2 S5 S D5 5 Ro D5 L Ro Vin Cin L Lo Vo Vin Cin Lo Vo Co D6 Co D6

S6 S6

D3 D1 D3 D1

SS33 SS11

(b)(b)

FigureFigureFigure 2. 2. 2.((a a())a Block) Block Block diagram; diagram;diagram; ( (bb)) ) suggestedsuggested suggested converter. converter. converter. x x1 1

Tc Tc Tc T T T c T c c p Tp Tp t T T T T T T p o o Tp o o Tp o t x2 To To To To To x Tp Tp 2 Tp Tp

Tc t

Tc t Figure 3. Turn-off delay in the switching signals.

The control signals areFigureFigure directly 3. 3. Turn-offTurn-o generatedﬀ delay with in the the switching PWM generator. signals. There is no need to use ZCD to synchronize with input ac voltage as diode D5, and D6 only conducts for positive and Thenegative control half cyclessignals of arethe inputdirectly voltage, generated respectively. with theThe PWMgate drive generator. circuits isolateThere the is lowno needvoltage to use ZCD switchingto synchronize signals withVg1 and input Vg2 fromac voltage the high as voltagediode Dof5, DCF.and DThe6 only Ro, Lconductso, and Co representfor positive the and negativeresistance, half cycles inductance, of the andinput capacitance voltage, respectively.of the heating The load, gate respectively. drive circuits The outputisolate frequency the low voltage is governed by operating the proposed topology in the inverting and noninverting modes with almost switching signals Vg1 and Vg2 from the high voltage of DCF. The Ro, Lo, and Co represent the resistance,a unity inductance, voltage gain, andensuring capacitance equal rms of output the heat anding input load, voltage respectively. (see Figure The4). output frequency is The detail of each operating mode of the proposed converter is explored in Figure 5 with the governed by operating the proposed topology in the inverting and noninverting modes with almost help of switching states and highlighted paths of power transfer from input to output. a unity voltage gain, ensuring equal rms output and input voltage (see Figure 4). The2.1. Non-Invertingdetail of each Operation operating mode of the proposed converter is explored in Figure 5 with the help of switching states and highlighted paths of power transfer from input to output. The operation of the proposed converter in this operating mode has positive voltage gain if the controlled switches S1, S3, S5 and S2, S4, S6 are switched in positive and negative half cycles of the 2.1. Non-Inverting Operation source voltage, respectively. In the positive half cycle, the diodes D5, D1 are on, and D2, D3, D4, D6 Theoff. Theoperation turned-on of the switch proposed S3 cannot converter conduct inas this its series-connected operating mode diode has positiveD3 is reverse voltage biased. gain In if the interval, the load is connected to the input, as shown in the highlighted paths of Figure 5a. During controlled switches S1, S3, S5 and S2, S4, S6 are switched in positive and negative half cycles of the source voltage, respectively. In the positive half cycle, the diodes D5, D1 are on, and D2, D3, D4, D6 off. The turned-on switch S3 cannot conduct as its series-connected diode D3 is reverse biased. In interval, the load is connected to the input, as shown in the highlighted paths of Figure 5a. During Electronics 2020, 9, 430 6 of 16

Equation (7) is simplified by neglecting the internal voltage drops to have the voltage transfer ratio for inverting operation with any polarity of the input voltage. This is depicted in Equation (8): =− vvoin (8)

Similarly, the inductor voltage for the inverting operation during the short-through and non-short through intervals is computed respectively as: = vTvLoin (9) =+() vTvvLcino (10)

The boost factor or the voltage transfer ratio during the inverting operation can be realized by computing the average value of the inductor voltage during its conducting interval as: vTT+ oco=− =−B (11) vTin c By ignoring the shoot-through interval, the boost factor becomes unity. So, the instantaneous value of the output voltage becomes equal to and opposite to the instantaneous value of the input voltage during the non-inverting and inverting operation, respectively. Therefore, the rms value of the output voltage is nearly equal to that of the input voltage as there are low voltage drops in the Electronicsfiltering 2020inductor, 9, 430 and switching devices. This is also explored analytically by computing the5 ofrms 16 value of the output voltage at 400 Hz (see Figure 4).

vo +Vm.Sin(wt) -Vm.Sin(wt) +Vm

ωt

-Vm

π/8 3π/8 5π/8 7π/8 9π/8 11π/8 13π/8 15π/8 Electronics 2020, 9, 430 7 of 16 0 2π/8 4π/8 6π/8 8π/8 10π/8 12π/8 14π/8 16π/8 Figure 4.== Voltage waveform for 400400 HzHz outputoutput frequency.frequency. Voin() rms V ()106 rms V , for 150 peak input voltage (13) This output is generated by operating the proposed topology alternatively eight times in non-inverting and inverting mode. The rms output voltage Sis4 simply computedS2 as: S4 S2 ππ/8 2D /84 D2 D D 22 4 2 22++−22ωω ωω ()VtdtVtdtsin()() () sin()()S5 ππS mm 220/85 π D5 3/8ππ 4/8 Ro D5 ++22()22Ro ()()ωω22 +−()()() ωωL L VtdtVtdtmmsin sin ππVin Cin Lo Vo V 22ππLo in Cin = 2/8Vo 3/8 Vorms() ππ 5/8 6/8 Co 2222C 22 ++()VtdtVtdtsino ()()ωω +−() sinD6()() ωω (12) D6 ππmm 224/8ππ 5/8 7/8ππ 8/8 S6 S6 ++2222ωω2 +−ωω2 ()VtdtVsin()() ()sinD ()()td t D ππ mm D3 1 D3 1 226/8ππ 7/8 S3 S1 S3 S1 V = m 2

Therefore, (a) (b)

S4 S2 S4 S2

D4 D2 D4 D2 S S 5 5 D5 D5 Ro Ro L L Vin Cin Lo Vo Vin Cin Lo Vo Co Co D6 D6

S6 S6 D3 D1 D3 D1 S3 S1 S3 S1

FigureFigure 5. 5.(a )(a and) and (b ()b non-inverting) non-inverting operation, operation, ((c) and ( d)) inverting inverting operation. operation.

2.1. Non-InvertingThe details Operationof the switching states of the semiconductor switching devices in each operating interval are also explored in Table 1. The operation of the proposed converter in this operating mode has positive voltage gain if the controlledTable switches 1. The switchingS1, S3, S 5statesand Sof2 ,theS4 , semiconductorS6 are switched devices in positive during non-inverting and negative and half inverting cycles of the source voltage,operation. respectively. In the positive half cycle, the diodes D5, D1 are on, and D2, D3, D4, D6 are Non-Inverting Operation Inverting Operation Vin > 0 Vin < 0 Vin > 0 Vin < 0 S1 on D1 on S1 off D1 off S1 off D1 on S1 on D1 off S2 off D2 off S2 on D2 on S2 on D2 off S2 off D2 on S3 on D3 off S3 off D3 on S3 off D3 off S3 on D3 on S4 off D4 on S4 on D4 off S4 on D4 on S4 off D4 on S5 on D5 on S5 off D5 off S5 off D5 on S5 on D5 off S6 off D6 off S6 on D6 on S6 on D6 off S6 off D6 on It is clear from Figure 5 and Table 1 that in each operating interval, only two controlled switches and two diodes are in conducting states. Electronics 2020, 9, 430 6 of 16

off. The turned-on switch S3 cannot conduct as its series-connected diode D3 is reverse biased. In this interval, the load is connected to the input, as shown in the highlighted paths of Figure5a. During the negative half cycle of the source voltage, the diodes D2, D6 become forward and D1, D3, D4, D5 reverse biased. The switch S4 remains off in this interval due to reverse biasing of its series-connected diode D4. The power flow path during this interval is exposed in Figure5b. In this operation, the role of the switching devices is only to connect the load with the source. Therefore, the instantaneous value of the load voltage is equal to the input voltage. The dynamic state behavior of the inductor voltage for positive and negative input during non-inverting operation can be realized in Equation (1) to find out the voltage transfer ratio, as reported in [19].

di L = v 2io(R + R ) 2V vo (1) dt in − d f m f − d f − where vin, vo, L, io, Vdf, Rdf, and Rmf are the instantaneous input voltage, output voltage, ﬁltering inductor, instantaneous output current, diode forward voltage, and the internal resistances of the diode and the MOSFET, respectively. In steady-state condition, the relationship between the instantaneous value of the input and output voltage is developed in Equation (2) by ignoring the voltage drops in the internal resistances of the switching devices.

vo = vin (2)

These results can also be formulated using the inductor volt-second product principle. The inductor voltage during the turn-oﬀ delay time (shoot-through time) is realized as:

vL = Tovin (3)

Similarly, the inductor voltage during the non-shoot through interval (Tc) is computed as:

vL = Tc(v vo) (4) in − The average inductor voltage during one conduction interval is calculated as:

vinTo + (vin vo)Tc − = 0 (5) Tp

Or v T + T o = c o = B (6) vin Tc where ‘B’ is the voltage boost factor or voltage transfer ratio. Equation (6) reduces to Equation (2) for the low value of shoot-through interval (To).

2.2. Inverting Operation

The switching of the controlled switching devices S2, S4, S6, and S1, S3, S5 ensure the negative voltage gain for positive and negative input voltage, respectively. The polarity of the input voltage determines the switching states of the diodes. The diodes D5, D4 become forward biased and D1, D2, D3, D6 reverse biased for positive input voltage. The reverse biasing of the diode D2 forces the controlled switch S2 to remain off in the non-conducting state. The output voltage is inverted with respect to source voltage, as can be observed from the power flow path of Figure5c. The negative polarity of the source voltage forces the operation of diodes D3, D6 in the conduction state and D1, D2, D4, D5 in the off state. The controlled switch S3 is not in conduction state due to reverse biasing of its series-connected diode D3. The power flow path of Figure5d depicts the inverted load voltage with Electronics 2020, 9, 430 7 of 16 the negative input voltage. The inverting behavior of the output can also be further explored through the dynamic inductor voltage as depicted in Equation (7) for positive and negative input voltage.

di L = v 2io(R + R ) 2V + vo (7) dt in − d f m f − d f Equation (7) is simpliﬁed by neglecting the internal voltage drops to have the voltage transfer ratio for inverting operation with any polarity of the input voltage. This is depicted in Equation (8):

vo = v (8) − in Similarly, the inductor voltage for the inverting operation during the short-through and non-short through intervals is computed respectively as:

vL = Tovin (9)

vL = Tc(vin + vo) (10) The boost factor or the voltage transfer ratio during the inverting operation can be realized by computing the average value of the inductor voltage during its conducting interval as:

v T + T o = c o = B (11) vin − Tc −

By ignoring the shoot-through interval, the boost factor becomes unity. So, the instantaneous value of the output voltage becomes equal to and opposite to the instantaneous value of the input voltage during the non-inverting and inverting operation, respectively. Therefore, the rms value of the output voltage is nearly equal to that of the input voltage as there are low voltage drops in the ﬁltering inductor and switching devices. This is also explored analytically by computing the rms value of the output voltage at 400 Hz (see Figure4). This output is generated by operating the proposed topology alternatively eight times in non-inverting and inverting mode. The rms output voltage is simply computed as:

uv u πR/8 2Rπ/8 u 2 2 2 2 2 2 u +Vm sin(ωt) d(ωt) + Vm sin(ωt) d(ωt) u 2π 2π − u 0 π/8 u 3π/8 4π/8 u R 2 R 2 u + 2 +V2 sin(ωt) d(ωt) + 2 V2 sin(ωt) d(ωt) u 2π m 2π − m u 2π/8 3π/8 Vo(rms) = u u 5Rπ/8 6Rπ/8 (12) u + 2 +V2 sin(ωt)2d(ωt) + 2 V2 sin(ωt)2d(ωt) u 2π m 2π − m u 4π/8 5π/8 t 7π/8 8π/8 2 R 2 2 2 R 2 2 + 2π +Vm sin(ωt) d(ωt) + 2π Vm sin(ωt) d(ωt) 6π/8 7π/8 − = Vm √2

Therefore, Vo(rms) = Vin(rms) = 106 V, for 150 peak input voltage (13) The details of the switching states of the semiconductor switching devices in each operating interval are also explored in Table1. Electronics 2020, 9, 430 8 of 16

Table1. The switching states of the semiconductor devices during non-inverting and inverting operation.

Non-Inverting Operation Inverting Operation Vin > 0 Vin < 0 Vin > 0 Vin < 0

S1 on D1 on S1 off D1 off S1 off D1 on S1 on D1 off S2 off D2 off S2 on D2 on S2 on D2 off S2 off D2 on S3 on D3 off S3 off D3 on S3 off D3 off S3 on D3 on S4 off D4 on S4 on D4 off S4 on D4 on S4 off D4 on S5 on D5 on S5 off D5 off S5 off D5 on S5 on D5 off S6 off D6 off S6 on D6 on S6 on D6 off S6 off D6 on

It is clear from Figure5 and Table1 that in each operating interval, only two controlled switches and two diodes are in conducting states.

3. Performance Evaluation and Comparison The non-inverting and inverting operation of the proposed converter and converter in [27] is achieved with high-frequency PWM control of two switches and two diodes. Therefore, they have the same switching and conduction losses. As the switching frequency is very high compared to the output frequency, the switching voltages and currents are assumed quasi dc. The average switching losses of a MOSFET and diode are computed in Equations (14) and (15), respectively, as discussed in reference [28]. fsw n o P = I V (t + t + C V2 (14) sw(MOSFET) 2 sw sw r f o sw

Psw(diode) = fswQRRVsw (15) As each MOSFET and diode conducts for the half period of the input voltage, so their average switching power in the period of the input voltage is realized in Equations (16) and (17), respectively.

fsw n o P = I V (t + t + C V2 (16) sw(MOSFET) 4 sw sw r f o sw 1 P = f Q V (17) sw(diode) 2 sw RR sw In one cycle of the input voltage, four diodes and four MOSFETs operate to realize the high-frequency output voltage, so the total switching power losses in one cycle of the input voltage are formulated in Equation (18).

hn 2 o i Psw = fsw IswVsw(tr + t f + CoVsw + 2QRRVsw (18)

The duty cycle of each conducting device is 50%. The average conduction losses of a MOSFET and diode in one cycle of the input voltage are formulated in Equations (19) and (20), respectively.

1 P = I2 R (19) cond(MOSFET) 4 sw m f I P = sw V + I R (20) cond(diode) 4 d f sw d f The total conduction losses during the one cycle of the input are the sum of the conduction losses of the four MOSFETs and four diodes, and they are computed as in Equation (21). n o Pcond = Isw Vd f + Isw Rm f + Rd f (21) Electronics 2020, 9, 430 9 of 16

where fsw, Isw, Vsw, Co, tr, tf, and QRR are the switching frequency, switching current, switching voltage, on-state resistance, output capacitance, rise and fall times of the MOSFET, forward voltage, forward resistance and reverse recovery charge of the fast recovery diode, respectively. The converter in [27] is implemented with a dual full-bridge converter, which has two extra fast recovery diodes and two MOSFETs than that of the proposed converter. The suggested research eliminates the use of two extra controlled switches and diodes. This ensures the reduction in losses and cost of the two gate drives circuits and two isolated dc supplies as switching of each transistor requires one isolated dc supply and gate drive circuit. As there is no need to synchronize the input voltage with gating signals, so it eliminates the use of ZCD as well. The control signals may be generated with any PWM generator. Hence these characteristics of the proposed converter decrease the size, cost, and control complexity signiﬁcantly. The converters in [23,29] are introduced to have variable output voltage and frequency. The converter in [29] employs the z-source arrangements to solve the shoot-through problem and to have a wide range of voltage gain. But this solution increases the size and cost as it requires extra passive components and switching devices. The large count of the passive components and switching devices are eliminated in [23]. The converters in [23,29] operate well once their output frequency requirement is less than the few kHz. But for high-frequency operation, their switching sequence becomes complicated. Their operation requires the conduction of three transistors and diode in any switching PWM interval. The proposed converter employs a simple switching algorithm that is eﬀectively employed in the low- and high-frequency operation. They also have the problem of high switching voltage and currents that increase the conversion losses. The detail of the comparison of the proposed topology with the converters in [23,27,29] in terms of components count and performance characteristics is tabulated in Table2.

Table 2. Comparison with the existing converters in terms of components count and performance characteristics.

The Proposed The Converter The Converter The Converter Components Converter in [23] in [27] in [29] Number of Transistors 6 6 8 10 Number of Diodes 6 6 8 10 Number of Filtering Capacitors 2 2 2 4 Number of Filtering Inductors 1 1 1 4 Number of Gate Drives Circuits 6 6 8 10 Number of Isolated DC Supplies 6 6 8 10 Number of ZCDs 0 1 1 1 Number of Conducting Devices at Any 2 Transistors, 3 Transistors, 2 Transistors, 3 Transistors, Time Instant 2 Diodes 3 Diodes 2 Diodes 3 Diodes Continuous Continuous Continuous Discontinuous Voltage Gain Characteristics Unity Unity Buck-boost Buck-boost Output Frequency Characteristics Step Change Step Change Step Change Step Change

Switching Current at Unity Voltage Gain Io Io 2Io Io

Switching Voltage at Unity Voltage Gain Vin Vin 2Vin Vin

IoVd f + IoVd f + 3IoVd f + 1.5IoVd f + Conduction Losses at Unity Voltage Gain 2 2 2 2 Io Rm f + Rd f Io Rm f + Rd f 4Io Rm f + Rd f 1.5Io Rm f + Rd f Open Circuit Problem No No No No Shoot Through Problem No No No No Electronics 2019, 8, x FOR PEER REVIEW 9 of 16

devices are eliminated in [23]. The converters in [23,29] operate well once their output frequency requirement is less than the few kHz. But for high-frequency operation, their switching sequence becomes complicated. Their operation requires the conduction of three transistors and diode in any switching PWM interval. The proposed converter employs a simple switching algorithm that is effectively employed in the low- and high-frequency operation. They also have the problem of high switching voltage and currents that increase the conversion losses. The detail of the comparison of the proposed topology with the converters in [23,27,29] in terms of components count and performance characteristics is tabulated in Table 2.

Table 2. Comparison with the existing converters in terms of components count and performance characteristics.

Components The Proposed The Converter The Converter The Converter Converter in [23] in [27] in [29] Number of Transistors 6 6 8 10 Number of Diodes 6 6 8 10 Number of Filtering Capacitors 2 2 2 4 Number of Filtering Inductors 1 1 1 4 Number of Gate Drives Circuits 6 6 8 10 Number of Isolated DC Supplies 6 6 8 10 Number of ZCDs 0 1 1 1 Number of Conducting Devices 2 Transistors, 3 Transistors, 2 Transistors, 3 Transistors, at Any Time Instant 2 Diodes 3 Diodes 2 Diodes 3 Diodes

Voltage Gain Characteristics Continuous Continuous Continuous Discontinuous Unity Unity Buck-boost Buck-boost Output Frequency Step Change Step Change Step Change Step Change Characteristics Switching Current at Unity Io Io 2Io Io Voltage Gain Switching Voltage at Unity Vin Vin 2Vin Vin Voltage Gain

Conduction Losses at Unity IVo df  IVo df  3IVo df  1.5IVo df  Electronics 2020Voltage, 9, 430 Gain 2 2 2 2 10 of 16 IRRo mf df  IRRo mf df  4IRRo mf df  1.5IRRo mf df  Open Circuit Problem No No No No 4. SimulationShoot Through and Practical Problem Results No No No No 4. TheSimulation MATLAB and/Simulink Practical environment Results is used to explore the proposed topology. In this implementation, the inputThe voltage MAT isLA consideredB/Simulink to environment be 150 V peak is value used at to a frequency explore the of 50 proposed Hz. The switchingtopology. frequency In this of theimplementation, gating sequences the input is set voltage to 25 kHz. is consideredThe simulation to be 150 waveforms V peak value of the at outputa frequency voltage of 50 and Hz. current The areswitching shown in frequency Figure6a,b, of respectively. the gating sequences The output is voltage set to 25 and kHz. current The simulation are non-inverted waveforms and inverted of the bothoutput in the voltage positive and and current negative are shown half period in Figure of the 6a input,b, respectively. voltage at The a switching output voltage frequency and ofcurrent 25 kHz. Similarly,are non- theinverted waveform and inverted of the inputboth in current the pos withitive respectand negative to the half input period voltage of the is input shown voltage in Figure at a 7, ensuringswitching the frequencyhigh input of power 25 kHz. factor. Similarly, the waveform of the input current with respect to the input voltage is shown in Figure 7, ensuring the high input power factor.

Electronics 2020, 9, 430 10 of 16 (a) (b)

FigureFigureFigure 6. 6. 6.SimulationSimulation Simulation waveformswaveforms waveforms of of ( (aa)) output output voltagevoltage voltage andand and ((b (bb)) ) outputoutput output current. current. current.

(a)

(b) Figure 7. Simulation waveforms of (a) input voltage and (b) input current. Figure 7. Simulation waveforms of (a) input voltage and (b) input current.

TheThe power power quality quality of of the the input input current current is is explored explored by by taking taking its its fast fast Fourier Fourier transform transform (FFT) (FFT) in in MATLAB/SimulinkMATLAB/Simulink environment, andand itsits totaltotal harmonic harmonic distortion distortion (THD) (THD) is is found found to to be be less less than than 5% 5% as shown in Figure 8. This is due to its continuous nature as it is conducted by D1, D5, S1, S5 and as shown in Figure8. This is due to its continuous nature as it is conducted by D1, D5, S1, S5 and D2, D2, D6, S2, S6 during the positive half cycle and the negative half cycle of the input voltage D6, S2, S6 during the positive half cycle and the negative half cycle of the input voltage respectively respectively for non-inverting mode. Similarly, the conduction of D4, D5, S4, S6 and D3, D6, S3, S5 for non-inverting mode. Similarly, the conduction of D4, D5, S4, S6 and D3, D6, S3, S5 during positive duringand negative positive half and periods negative of half the sourceperiods voltage of the source respectively voltage ensures respectively its continuous ensures its conduction continuous in conductioninverting operation. in inverting operation. A practical prototype is also developed to validate the elimination of ZCD and the attained simulation results. It consists of six controlled switches (IRF840), six fast recovery diodes (RHRG3040), and one input inductor and capacitor of 1 mH and 1 µF, respectively. The six gate driving circuits are employed with hybrid chips (EXB840) with six isolated dc supplies. Figure9 shows the hardware implementation of the proposed converter.

Figure 8. Power quality analysis of the input current.

A practical prototype is also developed to validate the elimination of ZCD and the attained simulation results. It consists of six controlled switches (IRF840), six fast recovery diodes (RHRG3040), and one input inductor and capacitor of 1 mH and 1 µF, respectively. The six gate driving circuits are employed with hybrid chips (EXB840) with six isolated dc supplies. Figure 9 shows the hardware implementation of the proposed converter. Electronics 2019, 8, x FOR PEER REVIEW 10 of 16

(a)

(b) Figure 7. Simulation waveforms of (a) input voltage and (b) input current.

The power quality of the input current is explored by taking its fast Fourier transform (FFT) in MATLAB/Simulink environment, and its total harmonic distortion (THD) is found to be less than 5% as shown in Figure 8. This is due to its continuous nature as it is conducted by D1, D5, S1, S5 and D2, D6, S2, S6 during the positive half cycle and the negative half cycle of the input voltage respectively for non-inverting mode. Similarly, the conduction of D4, D5, S4, S6 and D3, D6, S3, S5 Electronicsduring positive2020, 9, 430 and negative half periods of the source voltage respectively ensures its continuous11 of 16 conduction in inverting operation.

Electronics 2020, 9, 430 Figure 8. PowerPower quality quality analysis analysis of of the the input input current. current. 11 of 16

Figure 9. Practical setup. Figure 9. Practical setup.

AA capacitor capacitor of of 1 1 µFµF is is used used at at the the source source side side to to suppress suppress the the unwanted unwanted harmonics harmonics generated generated by by thethe high-frequencyhigh-frequency switches. switches. An An intermediate intermediate inductor inductor of 1 mH of is1 employed mH is employed to solve the to shoot-through solve the shoot-throughproblem caused problem by the complementarycaused by the switchingcomplement of theary controlledswitching devices. of the Tocontrolled explore thedevices. operation To explorewithout the zero-crossing operation without detection, zero-crossing the analysis ofdetect theion, proposed the analysis topology of isthe carried proposed out by topology generating is carriedthe required out by controlgenerating signals the withrequired ZCD control and without signals ZCD.with ZCD For this and purpose, without ZCD. the output For this frequency purpose, is theselected output 400 frequency Hz so that is theselected output 400 voltage Hz so waveforms that the output with voltage their FFT waveforms can be clearly with viewed. their FFT Figures can be 10 clearlyand 11 viewed.show the Figures practically 10 and recorded 11 show waveforms the practi for 400cally Hz recorded output frequencywaveforms with for ZCD 400 andHz withoutoutput frequencyZCD, respectively. with ZCD The and 400 without Hz output ZCD, frequency respectively is generated. The 400 by Hz non-inverting output frequency and inverting is generated the input by non-invertingvoltage eight timesand inverting during one the cycle,input asvoltage can be eight seen times in Figures during 10 oneand cycle,11. All as the can practical be seen waveformsin Figures 10are and recorded 11. All withthe practical the Rigol waveforms oscilloscope. are recorded with the Rigol oscilloscope. Figure 10a depicts the switching signal generated with ZCD so that it may be synchronized with the input voltage. It is generated by applying an interrupt to the Arduino once ZCD detects the positive zero crossings. In Figure 10b, the output voltage is plotted with respect to the input voltage, while Figure 10c determines the harmonics contents of the output voltage, where the dominant component exists at 400 Hz. The analysis of these figures depicts that the period of the output voltage is eight times lesser than that of the input voltage. Therefore, the output frequency is eight times greater than the frequency of the source voltage. Figure 10d presents the input current waveform with respect to the input voltage. The input current is nearly sinusoidal, ensuring low harmonic contents and low THD. This ensures improvement in the input power factor.

(a) (b) Electronics 2020, 9, 430 11 of 16

Figure 9. Practical setup.

A capacitor of 1 µF is used at the source side to suppress the unwanted harmonics generated by the high-frequency switches. An intermediate inductor of 1 mH is employed to solve the shoot-through problem caused by the complementary switching of the controlled devices. To explore the operation without zero-crossing detection, the analysis of the proposed topology is carried out by generating the required control signals with ZCD and without ZCD. For this purpose, the output frequency is selected 400 Hz so that the output voltage waveforms with their FFT can be clearly viewed. Figures 10 and 11 show the practically recorded waveforms for 400 Hz output frequency with ZCD and without ZCD, respectively. The 400 Hz output frequency is generated by non-inverting and inverting the input voltage eight times during one cycle, as can be seen in Figures 10 and 11. All the practical waveforms are recorded with the Rigol oscilloscope. Figure 10a depicts the switching signal generated with ZCD so that it may be synchronized with the input voltage. It is generated by applying an interrupt to the Arduino once ZCD detects the positive zero crossings. In Figure 10b, the output voltage is plotted with respect to the input voltage, while Figure 10c determines the harmonics contents of the output voltage, where the dominant component exists at 400 Hz. The analysis of these figures depicts that the period of the output voltage is eight times lesser than that of the input voltage. Therefore, the output frequency is eight times greater than the frequency of the source voltage. Figure 10d presents the input current Electronicswaveform2020, 9, 430with respect to the input voltage. The input current is nearly sinusoidal, ensuring low12 of 16 harmonic contents and low THD. This ensures improvement in the input power factor.

Electronics 2020, 9, 430 12 of 16

Electronics 2020, 9, 430 (a) (b) 12 of 16

Figure 10. Outputs with zero-crossing detector (ZCD): (a) switching signals; (b) input-output voltage waveform; (c) fast Fourier transform (FFT) of the output voltage waveform; (d) input voltage-current waveforms. signals generated without zero-crossing detection. So, they are not synchronized with the input voltage. They are simply generated without any zero-crossing interrupt with any PWM generator. This characteristic simplifies the control effort significantly and reduces its complexity for the induction heating application. The analysis of Figure 11b–d determines the power quality of the output voltage and input current. The rms value of the output voltage is recorded in Figure 11e. This also validates the computed rms value of Equation (13) in Section 1. The comparison of Figures 10 and 11 depicts that the power(c) quality characteristics of the output voltage(d and) input current with and without zero-crossing detection are almost similar. This is true if output frequency is a few Figure 10. a b hundredFigure times 10. Outputsgreater Outputs than with with that zero-crossing zero-crossing of the frequency detector of the(ZCD): (ZCD): source (a () voltage.)switching switching So, signals; switching signals; (b) (input-outputsequences) input-output can bevoltage directlyvoltage waveform;generated waveform; and (c )( c fastapplied) fast Fourier Fourier to the transform transformcontrolled (FFT) switches. of of the the That output output is whyvoltage voltage the waveform;proposed waveform; topology(d) input (d) input is implementedvoltage-currentvoltage-current without waveforms. waveforms. zero crossing detection for the induction heating application. signals generated without zero-crossing detection. So, they are not synchronized with the input voltage. They are simply generated without any zero-crossing interrupt with any PWM generator. This characteristic simplifies the control effort significantly and reduces its complexity for the induction heating application. The analysis of Figure 11b–d determines the power quality of the output voltage and input current. The rms value of the output voltage is recorded in Figure 11e. This also validates the computed rms value of Equation (13) in Section 1. The comparison of Figures 10 and 11 depicts that the power quality characteristics of the output voltage and input current with and without zero-crossing detection are almost similar. This is true if output frequency is a few hundred times greater than that of the frequency of the source voltage. So, switching sequences can be directly generated and applied to the controlled switches. That is why the proposed topology is implemented without zero crossing detection for the induction heating application.

(a) (b)

Figure 11. Cont.

Electronics 2020, 9, 430 13 of 16

(e)

Figure 11. Outputs without ZCD:Figure ( a11.) switching Outputs without signals; ZCD: (b) input-output(a) switching signals; voltage ( waveforms;b) input-output (c) voltage FFT of waveforms; (c) FFT the output voltage; (d) inputof voltage-current the output voltage; waveforms; (d) input voltage-curre (e) rms valuent ofwaveforms; the output (e) voltage.rms value of the output voltage.

Figure 10a depicts the switchingThe nominal signal output generated frequency with requirement ZCD so that in it the may domestic be synchronized induction heating with application lies in the range of 20 kHz to 100 kHz. The switching frequency of the converter is selected higher than the input voltage. It is generated by applying an interrupt to the Arduino once ZCD detects the positive the resonance frequency, so the effect of load equivalent inductance and capacitance should not zero crossings. In Figure 10reflectb, the in outputthe output voltage and input is plotted currents. with The respect resonance to the of the input equivalent voltage, load while inductance (40 µH), Figure 10c determines thefiltering harmonics inductor contents (1 mH), of the capacitance output voltage, (1 µF), and where resistance the dominant (120 Ω) is component nearly 5 kHz. The switching exists at 400 Hz. The analysissequence of these of 25 kHz figures is applied depicts to thatthe switching the period device of thes to obtain output the voltage output isfrequency eight of 25 kHz. This times lesser than that of theis inputachieved voltage. by operating Therefore, the proposed the output converter frequency 500 times is eight in non-inverting times greater and than inverting modes in the frequency of the sourceone voltage. cycle of Figurethe input 10 voltage.d presents That the is why input the current output waveformand input currents with respect are in-phase to with their the input voltage. The input currentFigure 12b,c is nearly depicts sinusoidal, the output ensuringvoltage and low curre harmonicnt waveforms. contents The andpower low quality of the input THD. This ensures improvementcurrent is in explored the input by powerplotting factor. it with respect to the input voltage in Figure 12d. Figure 11a depicts the practically recorded waveforms of two complementary switching signals generated without zero-crossing detection. So, they are not synchronized with the input voltage. They are simply generated without any zero-crossing interrupt with any PWM generator. This characteristic simplifies the control effort significantly and reduces its complexity for the induction heating application. The analysis of Figure 11b–d determines the power quality of the output voltage and input current. The rms value of the output voltage is recorded in Figure 11e. This also validates the computed rms value of Equation (13) in Section1. The comparison of Figures 10 and 11 depicts that the power quality characteristics of the output voltage and input current with and without zero-crossing detection are almost similar. This is true if output frequency is a few hundred times greater than that of the frequency of the source voltage. So, switching sequences can be directly generated and applied to the controlled switches. That is why the proposed topology is implemented(a) without zero crossing detection(b) for the induction heating application. The nominal output frequency requirement in the domestic induction heating application lies in the range of 20 kHz to 100 kHz. The switching frequency of the converter is selected higher than the resonance frequency, so the effect of load equivalent inductance and capacitance should not reflect in the output and input currents. The resonance of the equivalent load inductance (40 µH), filtering inductor (1 mH), capacitance (1 µF), and resistance (120 Ω) is nearly 5 kHz. The switching sequence of 25 kHz is applied to the switching devices to obtain the output frequency of 25 kHz. This is achieved by operating the proposed converter 500 times in non-inverting and inverting modes in one cycle of the input voltage. That is why the output and input currents are in-phase with their corresponding voltage ensuring improved power quality. Figure 12a shows the two complementary(c) gating signals for 25 kHz switching frequency.(d) Figure 12b,c depicts the output voltage and current waveforms. The power quality of the input current is explored by plotting it with respect to the input voltage in Figure 12d. Electronics 2020, 9, 430 13 of 16

(e)

Figure 11. Outputs without ZCD: (a) switching signals; (b) input-output voltage waveforms; (c) FFT of the output voltage; (d) input voltage-current waveforms; (e) rms value of the output voltage.

The nominal output frequency requirement in the domestic induction heating application lies in the range of 20 kHz to 100 kHz. The switching frequency of the converter is selected higher than the resonance frequency, so the effect of load equivalent inductance and capacitance should not reflect in the output and input currents. The resonance of the equivalent load inductance (40 µH), filtering inductor (1 mH), capacitance (1 µF), and resistance (120 Ω) is nearly 5 kHz. The switching sequence of 25 kHz is applied to the switching devices to obtain the output frequency of 25 kHz. This is achieved by operating the proposed converter 500 times in non-inverting and inverting modes in one cycle of the input voltage. That is why the output and input currents are in-phase with their ElectronicsFigure2020, 912b,c, 430 depicts the output voltage and current waveforms. The power quality of the input14 of 16 current is explored by plotting it with respect to the input voltage in Figure 12d.

(a) (b)

Figure 12. Outputs with 25 kHz switching frequency: (a) switching signals; (b) output voltage-current waveforms; (c) output voltage-current waveforms in zoom form; (d) input voltage-current waveforms.

The envelope of the output voltage of Figure 12b is equal and opposite to the instantaneous value of the input voltage. Figure 12c is obtained by zooming the Figure 12b around the peak value (150 V) of the input voltage to explore that the instantaneous value of the output voltage is equal to 150 V during non-inverting and 150 V during the inverting operation for positive input voltage. However, − the rms values of the output and the input voltage are the same. All the practically recorded waveforms of input currents (see Figures 10–12) have the same sinusoidal behavior ensuring low harmonic contents, low THD, and hence improved power factor. There is no voltage and current overshoot problem in output voltage, output, and input current. These characteristics of the proposed converter make it very attractive for the induction heating application.

5. Conclusions This research work proposed a novel single-phase direct frequency converter for domestic induction heating application. The required high-frequency ac output is directly obtained from the low-frequency input voltage with a simple control algorithm. The switching sequences are directly generated from any simple low-cost PWM generator as there is no need to detect the positive or negative half cycle of the input voltage. The developed converter is implemented with only six controlled switching devices and six fast recovery diodes. Therefore, this development of the proposed converter reduces the size and cost by eliminating the requirements of two extra controlled switches, two fast recovery diode, two gate-drive circuits, two isolated dc power supplies, and one zero-crossing detector as compared to the existing converters. The proposed converter is also immune from a shoot-through problem caused by the complementary switching of the controlled devices. The proposed topology is ﬁrst implemented in the MATLAB/Simulink environment then validated by the practical results obtained by developing a hardware setup. Electronics 2020, 9, 430 15 of 16

Author Contributions: Conceptualization, N.A. and G.A.; data curation, A.B.A.; formal analysis, T.I., G.A. and V.E.B.; investigation, U.F.; methodology, N.A., G.A., A.B.A., A.S.A. and A.G.A.; resources, A.S.A. and K.S.; supervision, T.I.; validation, V.E.B.; visualization, U.F.; writing—original draft, N.A. and G.A.; writing—review and editing, A.B.A., A.S.A., A.G.A.-K., and K.S. All authors have read and agreed to the published version of the manuscript. Funding: This research work is supported by the Deanship of Scientific Research, Majmaah University under contract number RGP-2019-19 and the University of Engineering and Technology, Lahore, Pakistan. Acknowledgments: The authors extend their appreciation to the Deanship of Scientific Research, Majmaah University for funding this work under project number No (RGP-2019-19). The authors would like to express their gratitude to University of Engineering and Technology for the administrative and technical support. Conflicts of Interest: The authors declare no conflict of interest.

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