Multi-Output Auxiliary Power Supply with Lossless Snubber

WSEAS TRANSACTIONS on CIRCUITS and SYSTEMS Geeng-Kwei Chang, Shu-Yuan Fan, Sheng-Yu Tseng

Multi-output Auxiliary Power Supply with Lossless Snubber

Geeng-Kwei Chang Shu-Yuan Fan Department of Electrical Engineering Wufeng University Ming-Hsiung, Chia-Yi, Taiwan, R.O.C e-mail: [email protected]

Sheng-Yu Tseng Green Power Evolution Applied Research Lab (G-PEARL) Department of Electrical Engineering Chang Gung University Kwei-Shan, Tao-Yuan, Taiwan, R.O.C e-mail: [email protected]

Abstract: - A multi-output auxiliary power supply realized by a flyback converter is presented for providing the essential low power supplies for control circuits and driving circuits in power processing systems. In the proposed circuit topology, lossless snubbers are introduced into the flyback converter to recover energy trapped in leakage inductor of transformer, and as a result, smoothes out voltage surge across drain-source of the switch, and alleviates oscillation caused by parasitic capacitance of the switch and leakage inductance of the transformer, hence reduces switching losses and improves conversion efficiency. Based on these concepts, an auxiliary power supply with five sets of output voltage has been designed and implemented to verify the feasibility of the proposed multi-output auxiliary power supply. Experimental results show that the conversion efficiency has been increased about 5% and reached 82.5% under full load as compared with flyback converter with hard- switching.

Key-Words: - flyback converter; lossless snubber; auxiliary power supply.

1 Introduction SMPS, such as half-bridge converter, full-bridge Power processing systems play a major role in to- converter, push-pull converter, flyback converter or day's electric utilities, for instance, in solar power forward converter [3], and so on. generation systems, in power unit of electric vehicles, Among various selecting criteria for an appropri- in battery chargers, in space satellites, in uninterrupt- ate converter, galvanic insulation is an decisive one ible power supplies (UPS), in active power filters, etc. when insulation from power line is recommended. In the power processing systems, the auxiliary power Since the auxiliary power supply takes input voltage supply is needed to provide different voltages for the from power line and has to provide several regulated control circuits and driving circuits. The needed vol- and isolated dc output voltages with different levels, tages are ranged from 3.3V to 24V. Two different the galvanic insulation becomes the most important design concepts are available, one is low-frequency L D ac transformer, and the other is high-frequency K 2 Tr1 N1 N2 switch-mode power supply (SMPS) [1, 2]. Although C1 the ac transformer can provide stable and indepen- Lm CO RO dent power source, it is bulky and uneasy to install on R 1 D1 1:N circuit board. On the other hand, SMPS, been com- Vi pact in weight and size and more efficiency, is find- M ing increased attention in industry and academia as 1 one of the preferred choices for low-power applica- tion. SMPS has successfully made its way into the industry and is now a mature and proven technology. Fig. 1. Schematic diagram of flyback converter with Many kinds of converters can be used to realize the conventional RCD snubber.

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LK D2 LK D2

Tr Tr1 N 1 N2 N1 N2 D2

Lm CO RL Lm CO RO

1:N RCD snubber 1:N Vi V i C1 D L1 R1 1 M1 M1

D1 C1

Fig. 2. Schematic diagram of flyback converter with Fig. 4. Schematic diagram of flyback converter with lossless LCD snubber. conventional RCD snubber in parallel with the

LK D2 conversion efficiency. Therefore, as shown in Fig. 3, an active clamp circuit composed of an active switch Tr1 N1 N 2 C1 is introduced into the flyback converter. Besides of

Lm CO RO resetting the transformer and suppressing the surge, M2 the circuit can store and release the surge energy at 1:N V i adequate timing through the active switch to raise efficiency and achieve soft-switching [12-14]. Al-

M1 though the active clamp circuits can increase conversion efficiency of converter, it needs extra driving circuit and active switch, resulting in a higher cost. Hence, a lossless snubber shown in Fig. 4 is proposed Fig. 3. Schematic diagram of flyback converter with to recover energy trapped in leakage inductor and conventional active clamp circuit. reduce switching losses. As shown in Fig. 4, the proposed lossless snubber is composed of passive com- factor to be taken into consideration. Hence, flyback ponents only and can achieve a lower cost. type or forward type converter with multiple outputs A variant of the RCD type snubber is adopted for coupled magnetically through transformer are more the multi-output flyback converter. The RCD snubber suitable for auxiliary power applications [4]. in Fig. 1 can be placed in parallel with the switch, as In reality, no matter how one arranges the winding shown in Fig. 4. In order to recover the energy structure inside the transformer, it brings about lea- trapped in leakage inductor of transformer, the resis- kage inductance. As a result, a flyback converter generally suffers from low efficiency and voltage tor R1 in the RCD snubber can be replaced by a surge across power transistor switches. To reduce transformer coupled to the output, resulting in the voltage surge, an RC clamp circuit is usually inserted circuit in Fig. 5. It can be seen that the proposed to absorb the energy stored in the leakage inductance snubber can recover the energy trapped in leakage and suppress the spike. This absorbed energy is dissi- inductor to output load and thus increases the overall pated in the snubber resistor during transistor turn-on. L T D To increase the efficiency, a non-dissipative LC K1 r1 O clamp circuit was proposed. In the circuit, the ab- D3 L O sorbed energy is returned through the snubber induc- Tr2 C R tor to the power source [5-7]. Both of the aforemen- O O VO Lm1 tioned circuit do not consider the diode reverse re- covery effects which may cause the oscillation in vol- 1:N 1:N V tage surge during switching, RCD as well as RC- i

RCD clamp circuits are proposed [8-11]. A typical Tr2 RCD snubber is shown in Fig. 1. With the RCD D snubber, spike voltage across switch is reduced, but 1 Lm21 D 2 LK21 M1 additional power loss is generated. In Fig. 2, a loss- CM1 1:N less LCD snubber is used to replace RCD snubber to DM1 C1 increase the conversion efficiency. Since RCD and LCD snubber can only improve voltage surge across Fig. 5. Schematic diagram of flyback converter switch to reduce switching loss, it does not increase with the proposed lossless snubber.

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V1 LK11 Tr1 D O1 diode Do keeps forward biased, which makes trans- T r2 N4 former T continue transferring its energy to the N1 DL3O 1r L m1 1 CO1 R1 VO 1:N load. Additionally, before t0 , voltage across capacitor CO2 R2 DO2 C reaches the level + oi NVV )/( , higher than Vi 1 1:N1:N2:N3:N4:N5 V2 V3 o /NV , which, when switch M 1 is ON, been applied Tr2 DO3 C O3 R3 across the primary winding of transformer T 2r , will D 1 Lm21 M1 D2 L K2 1 N3 1:N make diodes D2 and D3 forward biased. Thus,

C1 V4 DO4 energy stored in capacitor C1 can be transferred to CO4 R4 the output through diode D3 . Meanwhile, the circuit

V5 including diodes D2 and D3 is equivalent to an LC DO5 CO5 R5 circuit (suppose that Vo is constant), and current i 3D

GND1 will resonant from zero to a positive value, then back to zero while cut off diode D . Fig. 6. Schematic diagram of the proposed 3 Mode 2 [Fig. 7(b); ≤ < ttt ] auxiliary power supply with five sets of output 21 voltages. At t1 , I 11LK equals I 11Lm and I 1N equals zero, which makes the diode D reversely biased. The efficiency. Fig. 6 shows the complete circuitry with o five sets of output. input side circuit behavior is dominated by di 11LK ( += LLV 11m11Ki ) (3) dt 2 Operational Principles of the Pro- That is, within this time interval, = II 11LK11Lm posed Auxiliary Power Supply increases linearly and transformer T 1r is storing In order to simplify the analysis of the proposed energy. The capacitor C1 keeps releasing its energy snubber, the one output converter as shown in Fig. 5 to the load through diode D2 、 transformer T 2r and is considered in the analysis. Operation of this circuit diode D3 . This mode ends when diode D3 is OFF. can be divided into seven operation modes, and sup- Mode 3 [Fig. 7(c); ≤ < ttt 32 ] pression of transistor surge voltage is different in This mode begins at t when diode D is OFF. In each mode. In Fig. 7, (a)~(g), the operational prin- 2 3 ciples corresponding to each mode are illustrated. this mode, both transformers T 1r and T 2r are storing From the key waveforms shown in Fig. 8, the detail energy, T 1r from the source Vi and T 2r from the operation of this converter can be clarified. capacitor C1 . This mode ends when the energy Mode 1 [Fig. 7(a); <≤ ttt 10 ] stored in capacitor C1 is completely released. Before time t0 , the switch M 1 is OFF, diode D1 Mode 4 [Fig. 7(d); ≤ < ttt 43 ] is on, and the energy stored in the primary windings At t3 the energy stored in the capacitor C1 is of transformer T 1r is transferring to the secondary completely released. In this circumstances, the mag- windings and the load through diode Do . At = tt 0 , netizing energy stored in transformer T 2r is freew- switch M 1 is turned on. At this instance, voltage heeling through diodes D1 and D2 . The transformer across the magnetizing inductor L 11m is T 1r is still in its energy storage stage and the magne- VO di 11Lm tizing current I 11Lm increases linearly. V 1N =−= L 11m (1) N dt Mode 5 [Fig. 7(e); ≤ < ttt 54 ] while the voltage across the leakage inductor At = tt 4 , switch M 1 is turned off. At this mo- L 11K is ment, the inductor current = II 11LK11Lm is sustained VO di 11LK VV i11LK =+= L 11K (2) in continuous through the parasitic capacitor C 1M as N dt well as diode D and capacitor C . Since the capa- Since the leakage inductance is smaller than the 1 1 magnetizing inductance, one can find that, from the citance of C1 is greater than the capacitance of C 1M , above two equations, the current I 11LK increases the charging process of C1 will make the voltage rapidly while the current I 11Lm decreases linearly. across switch M 1 increase smoothly, from 0 to However, as long as I 11Lm be greater than I 11LK , + oi NVV )/( and beyond. Therefore, switch M 1 can

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ILK 11 LK11 IN1 IDO DO I L I I D I LK11 K11 N1 DO O O IO D3 LO D L I Lm11 N1 Tr1 I 3 O Lm11 N1 Tr1 Tr2 N4 ILO T I VN2 r2 N4 LO Lm VN1 V 11 C0 RO VO L VN1 N2 m11 C0 RO VO VN4 V N4 N2 N 1:N 1:N 2 1:N 1:N

Vi Vi

Tr2 Tr2 I DS CM1 IDS D1 Lm21 VN3 CM1 D L VN3 M1 1 m21 D2 LK21 M VDS N3 1 D L 2 K21 N3 1:N VDS V C I I Lm21 1:N M1 C1 1 LK21 I D VC1 C1 ILK21 Lm21 IC1 DM1 IC1 Fig. 7(a). Mode 1 Fig. 7(e). Mode 5 ILK 11 LK11 IN1 IDO DO IO I LK11 LK11 IN1 IDO DO D3 LO I I Lm11 N1 Tr1 O D3 LO I ILm11 N Tr1 Tr2 N4 LO 1 VN2 m VN1 L 11 C RO V T ILO 0 O V r2 N4 VN4 L VN1 N2 N m11 C0 RO VO 1:N 2 V 1:N N N4 1:N 2 1:N Vi

Vi Tr2 I DS CM1 Tr2 D1 Lm21 VN3 M1 IDS D2 LK21 CM1 VDS N3 D1 Lm21 VN3 1:N M1 D L V C I I Lm21 2 K21 N3 DM1 C1 1 LK21 VDS I 1:N C1 V C I ILm21 DM1 C1 1 LK21 Fig. 7(b). Mode 2 IC1 ILK 11 LK11 IN1 IDO DO Fig. 7(f). Mode 6 IO I D3 LO I D Lm11 N1 Tr1 LK1 1 LK1 1 IN1 I DO O IO ILO Tr2 N4 D3 LO VN2 L VN1 I Lm11 N1 Tr1 m11 C0 RO VO VN4 Tr2 N I LO N2 V 4 Lm VN1 N2 1:N 11 C0 RO VO 1:N V N N4 1: N 2 Vi 1: N

Tr2 Vi I DS CM1 D1 Lm21 VN3 M1 Tr2 D2 LK21 VDS N3 IDS 1:N CM1 D L V I 1 m21 N3 VC1 C1 I LK21 Lm21 M DM1 1 D LK2 1 I V 2 N3 C1 DS 1:N VC1 C1 ILK21 ILm21 Fig. 7(c). Mode 3 DM1 I C1 I LK11 L IN1 I DO K11 DO I O Fig. 7(g). Mode 7 D3 LO ILm11 T N1 r1 Fig. 7. Operational principle of the proposed Tr2 N4 ILO VN1 VN2 Lm11 C RO V converter (continued) V 0 O N N4 1:N 2 1: N At t5 , I DS reaches zero, switch M 1 is complete-

Vi ly OFF. = II 11LK11Lm continues charging C1 .

Tr2 Mode 7 [Fig. 7(g); ≤ < ttt 76 ] IDS CM1 D L VN3 M 1 m21 At t6 , voltage across capacitor C1 reaches 1 D LK21 V 2 N3 DS 1:N + NVV )/( , which makes diode D forward bi- V C I ILm21 oi o DM1 C1 1 LK 21 IC1 ased. Current iDo will resonant to the value of I o . Fig. 7(d). Mode 4 The energy stored in the transformer T 1r will be re- Fig. 7. Operational principle of the proposed leased to the load through D , which makes current converter o I 11Lm decreasing linearly. On the other hand, be operated with zero-voltage transition (ZVT). Dio- iDo des D1 and D2 are still in freewheeling condition IIII 11Lm1N11Lm11LK −=−= will resonant N through inductors L 21m . from the value of I 11Lm to zero while charging the Mode 6 [Fig. 7(f); <≤ ttt 65 ]

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are the input and output voltage respectively as M1 shown in Fig. 7. From (4), it can be found that trans- 0 t fer ratio M can be expressed as IDS V ND M o == (5) 0 t Vi − D1 According to (5), although the duty ratio D can VDS be determined once the transfer ratio M and turns- 0 t ratio N are specified, it is important to consider the VC1 relationships between duty ratio and component stress. Referring to (5), a larger duty ratio D corres- 0 t ponds to smaller transformer turns-ratio N , which VN1

Vi results in a lower current stress imposed on switch t M 1 , as well as lower voltage stress on freewheeling 0 -VO diode Do . In order to accommodate load variations, N line voltage, component value, a suitable operating I Lm1 range of duty ratio is between 0.35 and 0.4. 0 t 3.2 Transformers T 1r IDO Once the duty ratio is selected, the turns-ratio of transformer T can be determined from (5), which is 0 t 1r − )( VD1 o ID3 N = (6) DVi 0 t By applying the Faraday's law, the number N1 of I O turns at the primary winding can be determined as 0 t TDV N = si (7) VO 1 c ΔBA 0 t where Ac is the effective cross-section area of the t0 t1 t2 t3 t4 t5 t6 t7 transformer core and ΔB is the working flux density. Fig. 8. Key waveforms of the proposed converter. According to (6) and (7), the number of turns of secondary winding N 2 can be determined. capacitor C1 to a voltage value higher than For the flyback converter, magnetizing inductor + NVV )/( , which keeps the diode D in its for- oi o L 11m of the transformer is determined by taking into ward biased condition. Meanwhile, L 21m keeps in account the current down slope, which corresponds to freewheeling condition. When switch M 1 is turned the off-time of switch M 1 , and the inductance must on at the end of mode 7, a new switching cycle starts. be large enough to maintain continuous conduction mode (CCM) operation. The inductance of L 11m must satisfy the following inequality: 3 Design of the Proposed Auxiliary o − )( TD1V s L ≥ (8) 11m 2 Power Supply ΔIN Do(max) To design the proposed power supply systematically, where ΔI Do(max) is the maximum ripple of the determination of duty ratio D , transformers T 1r 、 secondary winding current of transformer T 1r that T 2r and capacitor C1 are detailed as follows [14]. goes through diode D , and it is equal to 3.1 Duty ratio D o As the converter is operating in continuous conduc- ΔI 11Lm (max) N . When the maximum current ripple is tion mode (CCM) and, according to volt-second bal- specified, the minimum magnetizing inductance can ance principle, the following equation can be ob- be determined. tained: 3.3 Capacitor C1 −VO In the propose lossless snubber, capacitor C1 is used DTV Si + ))(( S =− 0TD1 (4) N to store the energy from the leakage inductance and where TS is the switching period of M 1 , N is to eliminate the switch turn-off loss. The energy the turns-ratio of transformer T 1r , while Vi and Vo stored in C1 can be determined as

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IDS I DS

VDS VDS

(VDS: 50 V/div, IDS: 1 A/div, 5 μs/div) (VDS: 50 V/div, IDS: 10 A/div, 5 μs/div) (a) (a)

IDS IDS

VDS VDS

(VDS: 50 V/div, IDS: 1 A/div, 5 μs/div) (VDS: 50 V/div, IDS: 10 A/div, 5 μs/div) (b) (b) Fig. 9. Measured voltage VDS and current IDS Fig. 10. Measured voltage VDS and current IDS waveforms of switch M1 of the proposed auxiliary waveforms of switch M1 of the proposed auxiliary power supply under hard-switching (a) under 50%, power supply with lossless snubber (a) under 50%, and (b) under 100% of full load. and (b) under 100% of full load.

1 Vo 2 W VC i11C += )( (9) ▓ input voltage Vi : 48 Vdc 2 N ▓ output voltage V : +15 V , -15 V , +5 V , +18 To completely eliminate the switch turn-off loss, o dc dc dc Vdc and an isolated +15 Vdc the energy stored in capacitor C1 must at least equal ▓ switching frequency f s : 50 kHz to the turn-off transition loss WSoff , which is expressed as ▓ maximum output current I o : 1 A, 0.2 A, 0.6 A, 0.17 A and 0.2 A t Soff Vo WSoff Vi += )( I DP (10) ▓ maximum output power P : 30 W 2 N o According to the specifications, components of where t is the turn-off transition time of Soff the proposed auxiliary power supply are determined switch M 1 and I DP is the current passing the switch. as follows: Therefore, from (9) and (10), capacitor C1 can be ▓ turns-ratio of transformer T 1r : 1:3:3:1.5:3.5:3 determined as ▓ magnetizing inductance L 11m : 147 μH Vo VIt iDPSoff + )( ▓ leakage inductance L 11K : 0.9 μH N C1 ≥ (11) ▓ transformer core: EI-33 Vo 2 Vi + )( ▓ capacitor C1 : 39 nF N ▓ switch M : IRFPS59N60C 1 Measured waveforms of VDS and I DS of switch 4 Measured Results M 1 of the proposed auxiliary power supply under To verify the performance of the proposed auxiliary hard-switching are shown in Fig. 9. Fig. 9(a) shows power supply, a prototype with the following specifi- those waveforms under 50% of full load while Fig. cations is implemented. 9(b) shows those waveforms under full load. Wave-

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I O

Hard-switching Th e pro p ose d l oss le ss su nb be r circuit (VO : 10V/div, IO: 1 A/div, 200 ms/div) Fig. 11. Comparison of efficiency between the Fig. 12. Measured output voltage Vo and current Io proposed auxiliary power supply under hard-switching waveforms under step-load changes from 20% to and with the proposed lossless snubber from light load 100% of full load. to heavy load. range and exhibit good voltage regulation characte- forms shown in Fig. 10 are the same waveforms ex- ristic. cept that the proposed lossless snubber is applied. From Fig. 9 and Fig. 10, it can be seen that the power supply with the proposed lossless snubber can reduce References surge voltage across transistor switch at turn-on (ZVS) [1] H. D. Torresan, D. G. Holmes, and I. Shraga, and decrease slew rate of voltage across the switch at “Auxiliary power supplies for high voltage turn-off (ZVT). Comparison of conversion efficiency converter systems,” IEEE 35th Annual Power between the power supply under hard-switching and Electronics Specialists Conference, 2004, pp. with the proposed lossless snubber from light load to 645 – 651. heavy load is shown in Fig. 11. From Fig. 11, it is [2] H. D. Torresan, and D. G. Holmes, “A High clear that the efficiency of the proposed one is higher Voltage Converter for Auxiliary Supply than that under hard-switching, and its efficiency Applications using a Reduced Flying Capacitor reaches 82.5% under full load condition. Topology,” IEEE 36th Power Electronics Fig. 12 shows measured waveforms of output vol- Specialists Conference, 2005, pp. 1220 – 1226. tage Vo and current I o of the proposed power supply [3] Y. Jang, D.-L Dillman and M.-M Jovanovic, “ A under a step-load change between 20% and 100% of new soft-switched PFC boost rectifier with full load. From Fig. 12 it can be observed that voltage integrated flyback converter for stand-by regulation of output voltage Vo of 15V is limited power,” Transactions on Power Electronics, within ±1%. Volume 21, Issue 1, pp.66-72, 2006. [4] T. Higashi, T. Ninomiya and K. Harada, "On the Cross-Regulation of Multi-output Resonant Converters," Proceedings of IEEE Power 5 Conclusion Electronics Specialists Conference, 1988, pp. In this paper, some traditional snubber circuits have 18-25. been briefly reviewed, and an lossless snubber is in- [5] Z. Hossain, Olejniczak, K.-C Burgers and J.-C troduced to a flyback type multi-output auxiliary balda, “Design of RCD Snubbers based upon power supply to reduce the surge voltage across the Approximations to the Switching Characteristics: power switch as well as recover the energy trapped in Part I. Theoretical Development,” IEEE leakage inductor to output load to increase the overall international Electric Machines and Drives efficiency. Operational principle, steady-state analy- Conference Record, 1997, pp. TA2/6.1-TA2/6.3. sis and design procedures are detailed in the article. [6] W. McMurray, “Selection of Snubbers and Experimental results revealed that the proposed loss- Clamps to Optimize the Design of Transistor less snubber can reduce surge voltage across transis- Switching Converters,” IEEE Transactions on tor switch at turn-on and decrease slew rate of vol- Industry Applications, vol. 16, pp. 513 – 523, tage across the switch at turn-off. Furthermore, it can 1980. achieve high efficiency over a wide load variation [7] T. Ninomiya, T. Tanaka, and K. Harada, “Analysis and optimization of a nondissipative

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LC turn-off snubber,” IEEE Transactions on Industry Applications, vol. 32, no. 1, pp. 155- Power Electronics, vol. 3, pp. 147-156, April 160, 1996. 1988. [12] M. Jinno, P.-Y. Chen and K.-C. Lin, "An [8] M. Milanovic, J. Korelic, A. Hren, F. Mihalic Efficient Active LC Snubber for Multi-output and P. Slibar, "The RC-RCD clamp circuit for Converters with Flyback Synchronous fly-back converter," Proceedings of the IEEE Rectifier," Proceedings of the IEEE Power International Symposium on Industrial Electronics Specialist Conference, vol. 2, pp. Electronics, vol. 2, pp. 547-552, 2005. 622-627, 2003. [9] A. Hren, J. Korelic and M. Milanovic, "RC- [13] T.-F. Wu and Y.-S. Lai, J.-C. Hung and Y.-M, RCD clamp circuit for ringing losses reduction Chen, “An Improved Boost Converter with in a flyback converter," IEEE Transactions on Coupled inductors and Buck-Boost Type of Circuits and Systems II: Express Briefs, vol. 53, Active Clamp,” Proceedings of Industry no. 5, pp. 369-373, 2006. Applications Conference, 2005, pp.639-644. [10] Z. Hossain, Olejniczak, K.-C Burgers and J.-C [14] S.-Y. Tseng, C.-.T Hsieh, H.-C. Lin, “Active balda, “Design of RCD Snubber based upon Clamp Interleaved Flyback Converter with Approximations to the Switching Characteristics: Single-Capacitor Turn-off Snubber for Stunning Part I. Theoretical Development,” Proceedings Poultry Applications,” Proceedings of IEEE of IEEE International Electric Machines and Power Electronics and Drive System Conference, Drives Conference, pp. TA2/6.1-TA2/6.3, 1997. 2007, pp. 1401-1408. [11] S.J. Finney, B.W. Williams and T.C. Green, “RCD snubber revisited,” IEEE Transactions on

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