Subcarrier Extraction System

/ / TRW No 14610-6001-TO-00

SUBCARRIER EXTRACTION SYSTEM

FINAL REPORT

December 1969

Prepared for

NATIONAL AERONAUTICAL AND SPACE ADMINISTRATION MANNED SPACECRAFT CENTER R&D PROCUREMENT BRANCH HOUSTON TEXAS 77058

0 Under K iContract No NAS9 10229

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I for Federal Sctenttf IC & "Technical " lnformfltonl Spr ngtield Va 22151 TRW No 14610-6001-TO-00

N1 18707

SUBCARRIER EXTRACTION SYSTEM

FINAL REPORT

December 1969

Prepared for

NATIONAL AERONAUTICAL AND SPACE ADMINISTRATION MANNED SPACECRAFT CENTER R&D PROCUREMENT BRANCH HOUSTON, TEXAS 77058

Contract NAS9 10229

Thi d un i nt I PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h II b p d d I d h d i d I d h df yppo lh h h f wh hi I I hdw h h p I w p M ETRWSy m C p TRWI

TRW 3Y37EM GROWP

ONE SPACE RK RE 0 0 C C L 0 9028 TRW No i4610-600i-T0-00

SUBCARRIER, EXTRACTION SYSTEM

FINAL REPORT

December 1969

) - -

Approved / / A , R P Tow Jr Section Head RF Section

W C Baxter Manager Defense Electronics Department

R R Trapp Manager Communications and Defense Electronics Laboratory

11 CONTENTS

Page

INTRODUCTION i

2 APEX SYSTEM AND UNIT DESCRIPTION 3

ZI System Description 3 Z Z AV APEX Module 5 2 z I Functional Description 5 Z 2 Z Estimate Generation 5 Z2 3 Signal Spectra 6 2 3 APEX Unit I 024-MHz TM Subcarrier 6 Z 4 Biphase Sensing Module 8 Z 5 Delay Line Module t0 2 5 1 Delay Line 10 2 5 2 Delay Line Filter 10 2 6 Buffer Module Description 10

3 APEX CIRCUIT DESCRIPTION I 2 3 1 AV-APEX Module Circuit Description 12 3 1 1 Input Circuitry 1z 3 1 2 Phase Detector 12 3 1 3 Low-Pass Loop Filter and Amplifier 1z 3 1 4 Sweep Circuitry 1z 3 1 5 VCO and VCO Filter 13 3 1 6 AV Data Output 13 3 1 7 CAD 13 3 1 8 CAD Low-Pass Filter and Amplifier 13 3 1 9 Estimation Inhibit and AM Modulator 13 3 1 10 AV Output Driver 14 3 1 i1 Dc Voltage Regulators 14 3 2 TM-APEX Module Circuit Description 14 3 2 1 Input Circuitry 14 3 22 Phase Detector 15 3 2 3 Low-Pass Loop Filter and Amplifier 15 3 2 4 VCO Biphase Modulator and Phase Shift Filter 15 3 2 5 CAD 15 3 Z 6 CAD Low-Pass Filter and Amplifier 15 3 2 7 Amplitude Modulator and Estimate Inhibit Circuitry 16 3 Z 8 AM Output Driver 16 3 Z 9 Biphase Sensor and Pulse Shaper 16 3 Z 10 Demodulator Output and Biphase Modulator Driver 16 3 2 i Dc Regulators 16

11i CONTENTS (Continued) Page

3 3 Biphase Sensing Module Description 17 3 3 1 External Coherent Amplitude Detector and Input Emitter Follower 17 3 3 2 External CAD Input to Signal Output Amplifier 17 3 3 3 Data State Integrator 17 3 3 4 Zero Voltage Threshold Comparators and Biphase Output Drive Circuitry 18 3 3 5 Output Signal Amplifier 18 3 4 Delay Module Description 18 3 4 1 Input-Output Filter Termination 19 3 4 2 Bandwidth 19 3 5 Buffer Module Description 19 3 5 1 Input Amplifier 19 3 5 2 Estimate Summation and Error Distribution Network 19 3 5 3 Output Amplifier 20 3 5 4 DC Bias Transfer Network 20

4 APEX SYSTEM ANALYSIS 21 4 1 AV -APEX Unit Analysis 21

4 2 TM - APEX Unit Analysis 25

5 APEX SYSTEM SCHEMATICS AND LAYOUTS 30

6 CONCEPTUAL REVISIONS 37 6 1 Revised Apollo Apex System 37 6 2 Revised System Description 38 6 2 1 AV APEX Unit Description 40 6 2 2 TM APEX Unit Description 42 6 2 3 TV APEX Unit Description 43

7 SIGNAL SIMULATION EQUIPMENT 46

8 POWER SUPPLIES 49

9 TEST RESULTS 53 t0 HARDWARE 57 i1 NEW TECHNOLOGY/INVENTIONS 61

iv ILLUSTRATIONS

Page

2 1 APEX Block Diagram 4

Z-2 AV Module Block Diagram 5

Z 3 Video Spectra - Near 1 25 MHz 6

2-4 TM Module Block Diagram 7

2-5 Biphase Sensing Module Block Diagram 9

2-6 Delay Line Filter Module Block Diagram 10

2-7 Buffer Module Block Diagram ii

3 1 AV Unit Top of Chassis View 12

3-2 TM Unit Top of Chassis View 14

3-3 Biphase Sensing Unit Top 17

3-4 Delay Line Module Top 18

3-5 Buffer Unit Top of Chassis Viev 19

4-I AV Power Spectrum 22

4-2 Suppression Versus Frequency Z3

4-3 TM Signal Spectrum 25

4-4 Phase Tracking Loop 26

4-5 Error Rate Versus S/N 27

4-6 TM Phase Transition Sensing Circuit 27

4-7 Signal Spectrum at Integrator Input (Neglecting Double Frequency Terms) 28

5-1 APEX Modules Interconnection Diagram 31

5-2 AV Module Schematic 32

5-3 TM Module Schematic 33

5-4 Biphase Sensing Module Schematic Diagram 34

5-5 Delay Line Filter Schematic Diagram 35

v ILLUSTRATIONS (Continued)

Page

5-6 Buffer Module Schematic 36

6-i Revised TV-APEX System 39

6-Z AV APEX Unit 41

6-3 TM APEX Unit 42

6-4 TV APEX Unit 43

6-5 TV Estimator Filter Block Diagram 44

7-i Signal Simulator Block Diagram 47

7-2 Subcarrier Simulator 47

8-1 Subcarrier Extraction Unit Front View 49

8-2 Cabinet Rear-View Drawer and Dual Power Supply 50

8-3 Strapping Pattern - Rear Panel Control H/P 6101A Power Supplies 51

8-4 Panel - Dual Power Supply (Rear Mount) 52

9-i NASA Photograph S-69-57736 54 9-2 NASA Photograph S-69-57738 55

9-3 NASA Photograph S-69-57737 56

10-i Front Panel 58

10-2 Cabinet Angle View Showing Modules and P/S 59

10-3 Front Quarter View of Drawer with Modules 59

10-4 Inside Drawer Front Panel Detail 60

10-5 Drawer Inside Back 60

Vl. i INTRODUCTION

This final report was prepared by TRW Systems Group for the National Aeronautical and Space Administration Manned Spacecraft Center under Contract No NAS9-10229

The contract objective was to design fabricate test and deliver in a minimum amount of time a Subcarrier Extraction System capable of removing two high-level subcarriers from a TV baseband spectrum This removal process was to hold disturbance of the TV spectrurnto aminimum

The two offending subcarriers are at i 0Z4 MHz and i 25 MHz The 1 024-M-z subcarrier is a biphase signal modulated at either a i 6-kbit or a 51 6-kbit rate (TM) The I Z5-MHz subcarrier is an FM-FM signal modulated by baseband signals consisting of analog bio-med information and voice (AV)

The relative levels between the TV baseband spectrum and the two subcarriers is determined by the relative FM deviation magnitudes of the Z-GHz carrier frequency and are understood to be

TV - AF 1 50 MHz TM- AF 0 45 0 60 MHz AV - AF 0 Z25 MHz

The overall signal-to-noise (S/N) ratio at lunar distances accord ing to the information provided by NASA is

S/N = 5 3 dB/5 3 MIVlz-noise bandwidth for the 85-ft steerable antenna

SIN = 13 3 dB/5 3 MHz-noise bandwidth for the Zi0-ft antenna

These relations indicate that both the TM and AV spectra are 25 to 35 dB greater than the TV spectrum Since the effective noise band width for the TM and AV extractors is 100 kHz or less the subcarrier to-noise ratio implies that a suppression of Z0 dB is sufficient to reduce the TM and AV signals to a level comparable with the noise level With this suppression the extractor bandwidths can be made narrower there by minimizing the extent of TV spectrum modification by the extractors

I The contract requirements have been met TM and AV suppression is 20 dB, and any residual TM or AV signal present is low enough to be concealed by the noise and does not appear on the TV display with noticeable coherency

2 Th d n na n PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h It b p d 4 f d h d - d I d h df yp P h I h f wh h I h d. h p P . I TRW Sy m G TRW I

2 APEX SYSTEM AND UNIT DESCRIPTION

2 1 SYSTEM DESCRIPTION

The Subcarrier Extraction System consists of five modules They are a) AV estimator module

b) TM estimator module

c) Biphase sensing module

d) Delay module

e) Buffer module

A brief description of these modules is included in Subsections Z 2 through 2 6

The input signals consist of the TV baseband white noise and the two interfering subcarriers located at I 024 MHz and 1 25 MHz (See the system block diagram Figure 2-1 )

The I 0Z4-MHz subcarrier is a biphase signal modulated at a I 6-kbit or a 51 6-kbit rate and the 4 25-MHz subcarrier is an FM-FM signal modulated by a set of baseband signals including a number of analog bio-med signals and voice

The unprocessed signal is fed through a buffer amplifier into the time delay network and then into the output amplifier to the summation point

The unprocessed signal is also fed to the coherent amplitude detector in the biphase sensing module whose output is fed to the TM unit

The two amplitude phase extractor (APEX) units are connected in a parallel feedback arrangement at one common summation port Each APEX unit has an amplitude and phase estimating channel that iq spec trally weighted to "optimally" estimate the character of its interfering signal and to minimize disturbance to the adjacent TV baseband spectrum The input which is a combination of the two interfering subearriers the TV baseband and noise is buffered and amplified to a level which is

3 Th d n a . PROPRIETARI TO TRW SYS EMS GROUP TRW INC d h I1 b pd d d h d d h dI yp p h h wh h I h d- IN h p p I TRW Sy n G p TRW I

ESTI;MATOR OUTPUT TV NOISE TVV TMA ETM + EAV

ATNT d - 6 4 IISEC O

TM AV N TV NOISE TM ESTIMATOR

INPUT

VOLTAGE COMP'

FIgure 2-4 APEX Block Diagram appropriate for the operation of the two subearrier extractors As implied each extractor is designed with a phase-tracking ioop and amplitude estimating channel The baseband amplitude informnation is multiplied by the phase information that iS available at the subearrier frequency This process develops an estimate of the interfering signal that can be inverted and fed out-of-phase to the sumrmation port The difference between the input interference and the inverted estimate is the remaining disturbance to the TV baseband For the two units errors (CTMV and cAy) can be set between 10 and 25 dE down from the original levels of the interfering sub carriers The resulting output of the summration port is buffered ampli fied and delivered as an output It consists of the TV baseband signal noise and the two difference signals representing the residues of the two extracted subcarriers

4 1h d n nI i PROPRIETARY 10 TRW SYSTEMS GRO b p d P TRW INC d h I f h 4 n d d h df hh f . h y P h w h h p - P fTRW Sy m G p TRW

Z 2 AV APEX MODULE

2 Z 1 Functional Description

The AV APEX module (Figure 2-2) is a closed-loop coherent phase and amplitude tracking estimator Signal subtraction occurs n the buffer module amplifier by adding an out-of-phase estimate of the signal to be suppressed to the original composite video signal The circuits following the summation amplifier are spectrally weighted to the signal to be sup pressed and generate an "optinal" estimate of that signal 2 Z 2 Estimate Generation

Performing on the error signal formed by the subtraction process the phase-trackng loop and the amplitude-tracking loop operate independ ently to form respective estimates of the phase (or frequency) and ampli tude of the 1 25-MHz subcarrier A sweep circuit is included in the phase tracking loop for initial acquisition VC 12VD

FILTER - VCO REGULATOR VC0

DETECTOR PASSE

J3 15 12 VDC 12 VDC -- F DC REGULATOR

S EGATE SEP AONRO

DEMODULATOR S EP AV +2V0 J3 12 FOUTPUT INPUT 12 VDC -< R UL:ATOR '1

COHIRENT L WESTI MTE AM O T U A PL T DE PA S SI BI T MO D U L A T O R DR I V R J1

OUTPUT C)F/p SWITCH

Fgure 2-2 AV MVodule Block Dagram 5 Th di urn ni PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h II b pd d f d h d i d d h dIl p p h hI I h h I hd - h p pn T WSy m G p T WI

2 Z 3 Signal Spectra

Figure Z-3 illustrates the spectra of the signals in the vicinity of 1 25 MHz It follows from the diagram and from analytical results that the I Z5-MHz subcarrier must be suppressed at least to the noise level i e 43 to Z5 dB Also the resulting amplitude and phase suppression versus frequency and spectra after suppression curves are illustrated in Figure 2-3 Note that only one TV spectral peak is altered in amplitude and only two are altered in-phase Assuming a Z6-dB suppression the modulation rates and indexes are given in Table 2-I The phase-tracking loop resonant frequency is approximately 45 kHz and the amplitude tracking loop bandwidth can be set to a value less than 5 kHz

Z 3 APEX UNIT 4 0Z4-MHZ TM SUBCARRIER

Figure 2-4 is the block diagram of the TM APEX unit intended for the extraction of the I 0Z4-MHz TM subcarrier It employs the concept of a phase-tracking and amplitude-estimating channel with the addition of

RELATIVE 150 - 1 25 MHz SUBCARRIER AMPLITUDE 5 - NOISE 85 FT DISH 50 KH 50------

I -" NOISE 210-FTDISH 50 KH SPECTRUM 78 80 h BEFORE APEX 5 0 -

PECTRUM

05 1 2285 MH 1 24425 MH I 260 MH

1 25 MH AM

APEX SUPPRESSION 10 DB __ SUPPRESSION,

20 DB - I ,PM SUPPRESSION

RELATIVE 5 0 - AMPLITUDE 1 25 MH SUBCARRIER

TV PHASE PHASE SPECTRUM AFTER APEX 0 5

5 0 - 1 25 MH SUBCARRIER AMPLITUDE SPECTRUM TV AMPLITUDE AFTER APEX T

1 2285 MH 1 24425 MH 1 260 Mtz 1 25 MH

Figure 2-3 Video Spectra - Near 4 25 MHz

6 Th d m I i PROPRIETARY 1O TR SYSTLMS GROUP TRW INC d h 11 b pd d f d h u dd I d h dfI p h 0, hI h h I hiw h h p P m I RW Sy mO p awl

Table 2-1 1 Z5-MHz Subcarrier Modulation Rates and Indexes

Nominal Nominal Modulation Modulation Rate (kHz) Index-FM

Voice 0 I to 3 0 zo 3 9 0 Z9 54 0 Z8 7 35 0 44 10 5 0 46 14 5 0 45

a phase transition sensing circuit and a biphase modulator The two chan nel outputs are multiplied to form a subcarrier estimate that is differenced with the input TM signal This signal leaves a residue error voltage and by changing the gain about the rf loop and the baseband filter cutoff

VCO ITOR

PH DI PHASE 12V J4 13 DETECTOR LOOP P MODULATOR REG 4 DFILTER VCO 16VDC

I NHIB J4 15

112D DETECTOR L : J 1E L. V" IHIT MLTD FILTER JLOOP MODULATOR LIJNEAR _ L REJ&OUTPUT I INPU I

-- MADE -- BI FILTER PHASE CIRCUITRY ITCH Figure 7/ WTHt6D 2-4 CIRCITR TM Module TM 0WI00 OUTPUT Block Diagram Th d -n rd in PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d I b pd d I d h d d InI h df p h h f h h f h d . h h p p m I I TRW Sy m G p TR% frequencies this error can be made as small as required The baseband bandwidths which are good indicators of the extent to which an extractor will ignore adjacent but unrelated spectra are only a few kiloHertz for both loops The reversals of phase state are sensed at the output of the coherent amplitude detector (CAD) As the 1 024-MHz subcarrier makes a change in phase state the CAD output changes from a positive error voltage to a negative swinging voltage This transition occurs before the phase state of the input signal changes because of the differencing action of the subtractor Since the CAD voltage gives indication prior to the actual time of a data pulse transition a delay network is added This is done prior to the composite signal entering the buffer module When this negative swinging voltage crosses zero at the threshold detector a data state memory circuit is activated This device places the biphase modu lator into its alternate state slightly earlier than the input data transition This transition is relayed onto the subtraction port by the filtered estimate line Since the source data signal is weighted by a 100-kHz bandpass filter within the Lunar Module (LM) a similar filter is included in the estimate feedback line Also since cancellation is done by a closed-loop system there is no need for this filter to be matched identically to the source filter This filter enhances the TM estimation process

In conclusion the phase and amplitude bandwidths are no more than a few kiloHertz wide thus estimation and subtraction of the TV signal can be kept to a minimum Changes in the TV spectral character and noise will disturb the point of time at which the biphase transition occurs Since the TM signal needs to be suppressed only to noise rather than to the TV spectral level the earlier statement implies that the estimation of data transition can become poor as the TM-to-noise ratio degrades without causing interference to the TV spectrum

2 4 BIPHASE SENSING MODULE

The biphase sensing module consists of the following circuits

a) Delay line driver

b) Output signal amplifier

c) Dc restorer

d) External coherent amplitude detector

8 Thi d .1 a i PROPRIETARY TO TRW SYSTEMS GROLP TRW INC d h 11 b p d 4 t d 4 d 1 d hahd I yp P h h I £ wh h I h d. h w p n f TRW Sy M G p TRW I

e) Signal conditioning amplifier

f) Data state integrator

g) Zero voltage threshold comparator

h) Output biphase driver

i) +12 Vdc regulator

j) -42 Vdc regulator

Figure 2-5 is a block diagram of the biphase sensing module The MSC input fed to the delay line driver is transferred to an external delay line, then back to the module to the output signal amplifier The dc restorer is employed to transfer the MSC input dc level to the buffer output

The MSC input fed to the external coherent amplitude detector is detected, amplified by the signal conditioning amplifier, and sent to the data state integrator to form a trapezoidal waveform whose zero crossing is sensed by the zero voltage threshold comparator The output of the zero voltage threshold comparator is differentiated, amplified, and gated to the output biphase driver for presentation at the clock output

DELAY DELAY LINE LINE INPUT OUTPUT

DELAY OUTPUT J...122 LINE SIGNAL +IVDC REGULATOR DRIVER AMLFE UFR16VDC OUTPUT J20 13 2 . dC' 20 13

R, 12VDC REGULATOR MSC IjiVEmC INPUT

I c.. {-- -'dONITDNNJ' -- STATE THRESHOLDI N E J2

VC0 -- "T OC INPUT Figure Z-5 Biphase Sensing Module Block Diagram

9 Th d n nf in PROPRIETARY TO TRW SYSTEMS GROUP TRWVI C d h I b pd d f d i d h df y p h h f h h £ h dw h h p p ITRWSf m G p TRWI

Z 5 DELAY LINE MODJLE

Z 5 1 Delay Line

The delay line is comprised of three sections input and output im pedance matching sections and delay line filtering section (see Figure 2-6) The input is JZ (delay line input) and the output is JZZ (delay line output)

2 5 2 Delay Line Filter

The delay line filter is an LC-type delay line used to delay the MSC signal for a time interval comparable to the delay inherent in the inte gration circuit following the CAD within the biphase sensing module

2 6 BUFFER MODULE DESCRIPTION

The buffer module (Figure 2-7) consists of the following circuits

a) Input amplifier

b) Estimate summation and error distribution circuit

c) Output amplifier

d) Dc bias transfer network

e) + 1Z V and -1Z V regulators

The delayed MSC signal from the biphase module is fed into the input amplifier where it is conditioned to the proper gain and isolation The delayed MSC signal now proceeds to the summation point where it is combined with the out-of-phase ETM and EAV signals from the AV and TM APEX modules - the TV carrier with the attenuated subcarriers now proceeds to the output amplifier where the signalis broughtup to the same signal level as was furnished to the input of the system The dc bias transfer network provides dc restoration of the TV signal

The + 12 V and -12 V regulators provide stable operating voltages and isolation from transients appearing on the 16-V supply lines

IMPEDANCE DEAIIEMITRP EDANC E MATCHING DJAA LIEFLEACHING DLY SECTION INPUTOUTPUT SECTION OL LINE , LI NE INPUT OUTPUT Figure 2-6 Delay Line Filter Module Block Diagram

10 Th d . a - ti PROfRlITy TO TRW SYSTEMS CROUP TRW INC d h ii b p d d f d h d u d I d :1 d yP P h I, h I wih if h dw h h w p m TRW Sy m G p TRW I

D C BIAS TRA14SFER EAV ~NETWORK EEA V INPUT J0

MSC OUTPUT EPLIFIER DITIBTONAPIIE 05 BUFFERNEWR IUNPUT E) MONITOR

EAV OUTPUT

R ETM OUTPUT

J1 12 REG 12 V +12 VOLT 1REGULATOR

Figure 2-7 Buffer Module Block Diagram

i T*his d M.t..ior acoM tin fOPRllETARY TO TRW SYSTEMS GROUP. TRW NC.. and shall n4 t be nprodced or ransierred to other docrne. or disclosed to others, o us.edfor any purpose Other than that for which it i. furnishe" withot th. prior written prmi..Ion of TRW Syst. Group, TRW Inc.

3. APEX CIRCUIT DESCRIPTION

3. 1 AV-APEX MODULE CIRCUIT DESCRIPTION

The following paragraphs describe the AV-APEX module circuits (also, see Figures 3-1 and 5-2).

3. 1. 1 Input Circuitry

The delayed MSC input from the buffer module is introduced at Ji I (EAV input) and is fed both to the phase detector (IC1) and the CAD (ICZ).

3. i.2 Phase Detector

ICI and associated components sample the input signal phase at or near the zero crossing point with respect to the VCO signal.

3.1.3 Low-Pass Loop Filter and Amplifier

The dual output of the phase detector is fed via RZ1 and RZZ to the low pass filter (R44, R48, C23, and C25), and amplifier (IC3) where the dual output is converted to a single output of 15-kHz bandwidth. The out put of IC3 is then fed to the frequency control circuitry of the VCO, the sweep limiter, the sweep controller, and the AV data output circuit.

3.1. 4 Sweep Circuitry

When there is no input signal at JI, the output of IC3 becomes a sawtooth with its ramp center such that the VCO frequency will vary

Figure 3-1. AV Unit. Top of Chassis View 12 This documera contins irormnation PROPRIETARY TO TRW SYSTEMS GROUP. TRW INC., end sholl not be reeprouced or tr..ferred to other docnerets, or diclosed to others, or used for In pirpoe other then thi for woich it is runished withnet the prior written permission of TRW Syetirn. Gre. TRW Iec

-Z0 kHz around the VCO frequency of 1. 25 MHz. When there is no input signal, the CAD, via its loop filter and IC6 which is a portion of the in hibit circuitry and by furnishing a negative voltage applied to Di, allows QZ (sweep gate) to conduct, providing a feedback path between IC3 and IC5, thus creating a sawtooth sweep. The sweep limit is controlled by summa tion voltage at the output of Q4 and Q5 fed back to the input of IC3.

3. 1. 5 VCO and VCO Filter

The VCO (Q9) with frequency determining elements (L6, L7, L8, D8, D9, C5Z, and C53) is a modified Hartley circuit with a separate voltage regulator, IC8, to provide additional stability. The output is buffered by an emitter follower (Q8) and a three-pole filter. The filter output goes to the phase detector and drives the 90' phase shift network consisting of C39, LZ, C38, Li, and C31. The phase shift network is buffered by emitter follower (QI). The output of QI is fed to the AM modulator and to the CAD.

3. 1. 6 AV Data Output

The output of IC3 feeds the input bas- network of Q6, consisting of R83 and C48, with a time constant of approximately 5 asec. The output of Q6 goes to J18 (AV demod) which is a front panel test jack.

3. 1. 7 CAD

ICZ and associated circuitry samples the amplitude of the incoming signal from JI I (EAV input) at the positive peaks with reference to the VCO frequency.

3. 1.8 CAD Low-Pass Filter and Amplifier

The dual output of the detector is fed via R35 and R36 to the low pass filter (C27 and C29) and amplifier (IC4) where the dual output is converted to a single filtered output and is fed to 1C6 (estimate inhibitor) and to QII, which is part of the AM modulator.

3. 1. 9 Estimation Inhibit and AM Modulator

The input of IC6 is controlled from either of two points, one is the front panel test switch (SZ) AV inhibit, and the other is the output of IC4. When IC6 receives an indication of signal presence, the output voltage goes from negative to positive, thus turning off the sweep gate (QZ) and turning

13 TM. doeun-.nt conn.~-~a m, to.L-n P&OPRIETAR Y TO TRW SYSTEMS GROUP. TRW VNC.. * .. ."Ul be reproducfd or transferred to oth,, document.. or.,ber decLoled to or...d for any prpose oter t 0antht for ,hich U lurmehed witha t],. prior written perrnisior of TRW Syse. Group TRW Inc. on Q7. the indicator light driver, which gives front panel indication of AV lock (D5Z). As QZ is turned off, the sweeping voltage to the VCO is dis continued. The output of IC6 also goes to Qi0 (amplitude inhibit) via R88, Dil, and DiZ limiter. This turns on Q10, applying the phase shifted out put of the VCO to Qi3 in the output drive section. QII driven by IC4 operates as the AM modulator.

3. 1. 10 AV Output Driver

The output amplifier consists of Q13 (phase inverter) and Q14 and QI5 (feedback controlled amplifier). R10Z sets the output (J12) to the proper level to be fed to the summation point input (J9 buffer module).

3.1. 11 Dc Voltage Regulators

There are two -12 V regulators, consisting of p.A723 IC regulators and a series pass transistor. In the case of the one +12 V regulator, the pass transistor is not required.

3. Z TM-APEX MODULE CIRCUIT DESCRIPTION

The following paragraphs describe the TM-APEX module circuit (also, see Figures 3-2 and 5-3).

3. Z. I Input Circuitry

The delayed suppressed ETM signal is fed from the buffer module to the input (J13). This signal is fed both to the phase detector (IC1) and the CAD (ICZ).

WN i&- ....

;Z- ]

Figure 3-2. TM Unit, Top of Chassis View

14 This doones conns nk o&tin PROPRIETARY TO TRW SYSTEMS GROUP, TRW IqC.. and shll he reprodued or trsnsf..rrd to other document.. or disclosed to others, or used for any purpoSe ot.t. th.n M.t for which it is urnished without th, prior written permission o TRW $ystems Group. TRW Inc.

3.2.2 Phase Detector

IC1 and associated components sample the input signal phase at or near the zero crossing point with respect to the VCO signal.

3.2.3 Low-Pass Loop Filter and Amplifier

The dual output of the phase detector (IC1) is fed via R28 and R29 to the low-pass filter (R32, C23, R53, and C27) and amplifier (IC3) where the dual output is converted to a single output of a few kiloHertz bandwidth. The output of IC3 is then fed to the frequency controlling circuitry of the VCO.

3. 2. 4 VCO, Biphase Modulator and Phase Shift Filter

The VCO (Q9) with frequency-determining elements (L5, L6, C58, C57, D3, and L7) is a modified Hartley circuit with a separate voltage regulator to provide additional stability. The output is buffered by Q8, an emitter follower, and a three-pole filter (L4, C53, L3, C5Z, LiI, and C51).

The VCO signal from the filter is fed to the rf input port of the biphase modulator (HP 10534C). The biphase signal applied at port X pulse modulates the VCO signal. The output signal is applied to the phase modulator as a reference signal.

The VCO phase shift filter consists of LI, C30, C31, LZ, and C3Z fed by the output of the biphase modulator and shifts the signal 90'. This signal feeds emitter follower (QZ) and provides reference frequency to the

CAD and carrier signal to the AM modulator via 02.

3.2.5 CAD

IC2 and associated circuitry sample the amplitude of the incoming signal and J13 (ETM input) at the positive peaks with respect to the VCO frequency.

3. 2. 6 CAD Low-Pass Filter and Amplifier

The dual output of the detector is fed via R54 and R55 to the low pass filter (CZ9 and C34) and amplifier (IC4) where the dual output is con verted to a single filtered output.

15 This d ¢undent contain. information PROPRIET RY TO TRW SYSTEMS GROUP. TRWt NC. , and shall not be reproduced or *annfa.rred to other docunie., or diclo..d to other.. or aed or any pirpose other than that for which it it furns.hed without the prior written permission of TRW Systerns Group. TRW lnc.

3. 2.7 Amplitude Modulator and Estimate Inhibit Circuitry

The output of IC4 goes to a differentially controlled gate (Q6 and Q7). The output controls the front panel indicator via Q5 (lamp driver), which gives an indication of signal lock. The second output from IC4 goes to the AM modulator (Q4). The VCO signal going to the AM modulator passes through Q3, which is controlled by a front panel switch SWI (TM inhibit). When the switch is on, it allows the 90' phase shifted VCO sig nal to proceed to Q4 (AM modulator).

3.2.8 AM Output Driver

The signal from the AM modulator goes to the emitter follower (QI0), which is ac-coupled to t-MHz bandpass filters (LIO, C72, and C73). R122 sets the level to the output amplifier QIZ and Q13 in feedback con figuration. The output of this amplifier goes to J15 (TM output).

3. 2. 9 Biphase Sensor and Pulse Shaper

The dual output from IC2 (CAD) is fed to IC5 via R58 and R59 where the signal level is amplified to a correct level to operate IC7 (voltage com parator). The output of IC7 passes through the switch SI (ext. -int. biphase clock), which allows the choice of the internal 1. 6-kbit rate or the external 51. 6-kbit clock rate via J19 (ext. biphase input). The signal from the switch is fed to IC9, which functions as a one-shot multi. The output of the "one shot" triggers IC10 (JK flip-flop).

3. Z. 10 Demodulator Output and Biphase Modulator Driver

The output of ICI0 (flip-flop) via R123 and R124 (voltage divider) to Q14 (output emitter follower), then goes to J14 (TM denodulator output) and hence to the front panel test jack (J21).

The output of IC10 also goes to IC8 (biphase modulator driver). The dc level output to the biphase modulator is set by R13 and is sent to the X port of the biphase modulator where it establishes dc bias at the biphase modulator. 3. 2. 11 Dc Regulators

The dc regulators are identical to the description in Paragraph 3.1.11.

16 This jowcmat cctaift. iziornatioa PROPRIETARY TO TRW SYSTEMS GROUP. TRW IWC.. O4 shall not be r prcdtcd or tr..sfsrrd to other dotinents. or di.loeed to others, or used Ior .ny prpose other t t tat for .Jch It is funrnishsd ih,a the prlor.wraten prumi.ion of TRW Systems Grop. TRW Inc.

3.3 BIPHASE SENSING MODULE DESCRIPTION

The following paragraphs describe the biphase sensing module (also, see Figures 3-3 and 5-4).

3.3. 1 External Coherent Amplitude Detector and Input Emitter Follower

The MSC input signal arriving at JZ4 (MSC input) is fed to the delay line driver and the external CAD via input level control RZ. Qi (emitter follower) provides isolation and proper impedance matching to drive the delay line via J22 (delay line input). The CAD utilizes an MC 1596G and receives the MSC signal via R2 and the TM VCO input via JZ6. The CAD then detects changes of the TM phase state by coherently comparing the input TM signal to the TM VCO. The dual output of the CAD is then fed to a RA702A signal conditioning amplifier.

3.3. 2 External CAD Input to Signal Output Amplifier

The signal conditioning amplifier is a saturating amplifier that accepts the dual outputs of the CAD and presents a single output via a bipolar limiter (CRZ and CR3) that is referenced to ground. The resultant waveform is a square wave with voltage center at zero.

3.3.3 Data State Integrator

The data state integrator (QZa and QZb and Q5) is an amplitude lim ited integrator that, upon receipt of the signal from the signal condition ing amplifier presents at its output a limited ramp or trapezoidal waveform with zero center reference voltage.

Figure 3-3. Biphase Sensing Unit, Top

17 This doctaou coal.t . linfwmsotn POPRIETARY TO TRW SYSTEM&SGROUP. TRW 1IC., and shell be reprodced or trutesrrsd to other decoanor, or disclosed to others, or used for any purpose other rosets.t "r thich it is uluisbod wl thi Nor prior wrttts. pe tia. of Ti WSyotu.. Gro p TRW Ise.

3.3.4 Zero Voltage Threshold Comparators and Biphase Output Drive Circuitry

The output of the data state integrator is fed into ARZ and AR3 (com parators). A positive going ramp at the input of AR2 causes the output to go positive at the output. A negative going ramp at the input of AR3 causes the output to go positive. These positive going transitions are passed by CR? and CR8 and differentiated by C36, R56, and C37, R57. These dif ferentiated pulses are gated by Q6 and Q7 and combined at their common collector resistor R61, which also serves as a level control. These nega tive going pulses are fed through Q8 (emitter follower) to J28 (clock output).

3.3.5 Output Signal Amplifier

The output signal amplifier provides gain sufficient to counter the delay line attenuation. The output of the delay line J2I is fed to a feedback controlled amplifier consisting of Q2 and Q3 and fed to the output jack J29 (buffer output). Dc restoration is provided by R7 and R9 and C6 and C7 connected between J24 (MSC input) and J29 (buffer output).

3.4 DELAY MODULE DESCRIPTION

The following paragraphs describe the delay module (also, see Figures 3-4 and 5-5).

Figure 3-4. Delay Line Module, Top

18 Th. documet continnornu~on PROPRIETARY TO TRW SYSTEMS GROUP. TRW INC.. and hit .not 6. repod, lcd or traferred ci other dncwnents, or disclosed to other., or used for any purpome other than that for which it is furnished without the prior written permission of TRW Systems Group, TRW Inc.

3.4. 1 Input-Output Filter Termination

The buffered MSC signal received at JZZ (delay line input) is passed through a 3Z0-element constant K delay line filter to JZ1 (delay line output). The impedances at the input and output are identical. The input impedance matching section is composed of RI, Ci, Li, and C2. The output imped ance matching section is composed of R2, C9, L8, and C8. The total delay is approximately 6.4 lisec, and the delay per element is 20 nsec.

3.4. Z Bandwidth

The 3-dB bandwidth is calculated and empirically determined to be t. 8 MHz.

3.5 BUFFER MODULE DESCRIPTION

The following paragraphs describe the buffer module (also, see Figures 3-5 and 5-6).

3. 5. 1 Input Amplifier

The delayed MSC signal from the biphase unit is brought into the buffer amplifier via J17 (buffer input). RZ sets the input level. Qi and QZ are configured as a feedback type amplifier giving a variable gain from i/Z to 10 as measured between J17 and the collector of QZ.

3. 5. 2 Estimate Summation and Error Distribution Network

R1Z sets the level of the delayed signal at the summation A A point, RiB. The suppression signals J9 (EAV) and J10 (ETM) via R1 9 and RZ0 join to

VNF: "WT

Figure 3-5. Buffer Unit, Top of Chausis View 1h d i I a n PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h II b p d d f d h 4 ri d d h df yp P ti Mhh £ wh h f h dw h fhip P i ITRWSy m G p TRWI form a summation network The suppressed ot residual error signals are fed back to the TM and AV units by an emitter follower (Q3) and J8 (€XV) and J7 (CTM)

3 5 3 Output Amplifier

The output of the summation junction (R18) is also fed to variable attenuator (R27) located at the input of Q4 Q4 and Q5 are drivers and 06 is the output transistor Q7 controls the current through Q6 and pro vides a high impedance to maxtmize the amplifiers' drive capability C14 and GI5 provide ac coupling and dc isolation at the drive output J5 Two outputs are provided J5 (MSG output) and J6 (monitor output)

3 5 4 DC Bias Transfer Network

The dc restorer is connected between J17 (buffer input) and J5 (MSC output) and provides a suitable dc reference for the TV signal Rli and C6 provide filtering Ri 16 sets the output dc level and the emitter follower (Q8) accomplishes the Impedance match via R40

zo 4 APEX SYSTEM ANALYSIS

4 i AV - APEX UNIT ANALYSIS

The AV signal is a i 250-MHz carrier modulated by FM-FM The information and thus the majority of the power resides in the phase of the signal A small percentage of power is associated with the amplitude of the signal and is produced by a bandpass filter in the LM output As the frequency of the LM AV signal swings its maximum excursion of =30 kHz the filter causes the amplitude to vary about 0 5 dB

To simplify the power spectrum presentation of the AV signal it is assumed that six sinusoids at frequencies 3 kHz 3 9 kHz 5 4 kHz 7 35 kHz 10 5 kHz and 14 5 kHz represent the modulating signals The 3-kHz sinusoid was chosen to represent the voice signal spectrum of 0 1 to 3 kHz This is a Bell Systems standard used to represent a voice spectrum

Figure 4-1 shows an approximation of the power distribution between the various modulating components The first three sidebands of 3 kHz are used because the modulation index of the voice signal places spectral power at higher order sidebands The modulation index of the other tones is less than unity so the first sideband of each frequency contains the majority of the power The power distribution is as follows

Power Units

Unmodulated carrier 1 000

Modulated carrier 0 033

Voice (3 kHz 6 kHz and 9 kHz) 0 6ZZ

3 9 kHz 0 00Z

5 4 kHz 0 00Z

7 35 kHz 0 00Z

10 5 kHz 0 004

f4 5 kHz 0 004

Higher sidebands and cross product terms 0 341 Z2 3 00

200

0 100 0 90 0 80 0 70 0 60 0 50 0 40

0 30

0 20

0 10 0 09 0 08 0 07 0 06 0 05 0 04

0 03

0 02

1 250 MH 4 60 2 4 6 8 10 12 14 16 KHz FREQUENCY Figure 4-1 AV Power Spectrum

The bandwidths of the AV-APEX phase and amplitude loops are considered next Figure 4-2 represents a plot of loop suppression versus frequency For the phase loop the suppression slope will be 40 dB/dec ade and for an amplitude loop it will be 20 dB/decade

22 0 DB

-10 DB

-20 DB

-30 DB

-40 DB

wn FREQUENCY

Figure 4-2 Suppression Versus Frequency

The natural frequency (fn) and damping ratio (t) for the phase tracking loop are given by

n I,(KdKO){

= r2 rf n where

Kd = phase detector gain constant

K = VCO gain constant

7'i = PI C

2= RzG

R1 R 2 and C = elements of the loop' s active filter

For the AV-APEX

K d = 300 mV/rad

K = i 88x 105 rad/V

R i = 8 2K

23 R2 = 24 KR C = 750 pF hence f = 15 2kHz n = 0 86

The noise level in the received LM signal will be from 13 dB to 25 dB below the maximum level of the AV signal The AV-APEX was thus designed to suppress the peak of the AV frequency spectrum Z6 dB The peak is the amplitude of the 3-kHz component as is seen in Figure 4-i

With the f n of the phase-tracking loop equal to 15 2 kHz Figure 4-2 shows that the AV-APEX will supply the following suppression

3 kHz 26 dB

3 9 kHz 23 dB

5 4 kHz 17 dB

6 kHz 16 dB

7 35 kHz IZ dB

9 kHz 9 dB

10 5 kHz 7 dB

14 5 kHz 4 dB

Thus all components will be suppressed to below the noise level

As a result of the small amount of AM present in the signal f n was chosen to be 5 kHz for the amplitude loop The 5 kHz is sufficient to handle any of the AM yet will not appreciably affect the TV signal

The AV-APEX will remove in excess of 99% of the total power in the AV spectrum This percentage is based on the individual suppression of each component in the AV spectrum Better AV signal removal could be realized by increasing the fn of the tracking loops however the system performance would not be improved and more of the TV spectrum would be affected

24 4 2 TM - APEX UNIT ANALYSIS

The baseband TM signal can be constructed by multiplying a coherent I 024-MHz carrier by the rf TM signal and then passing the resulting signal through a 100-kHz bandpass filter The spectrum of the TM signal is shown in Figure 4-3

05 04- > 0 2-.3.<. 02

o ° FREQUENCY(MH) -1 , °

Figure 4-3 TM Signal Spectrum

The TM loop bandwidths are considered next The same formulas used to find the fn of the AV phase loop are used to find the fn of the TM phase loop except Ri = iO Kf

1 = 2 2 K&

C = 0 047 jaF

Therefore

f = 1 7 kHz n = 0 56

It is seen that a narrow bandwidth phase loop can process a wideband TM signal Figure 4-4 indicates the basic components of the TM phase loop Information is supplied to the biphase modulator from another circuit so that it will produce an estninate of the TM signal The phase detector multiplies the TM signal by the estimate In the frequency domain this amounts to a convolution of two similar spectrums The result of the convolution is an impulse at zero frequency The phase loop therefore has to process only a dc signal

25 TMPAEPHASE LOOP I SIGNAL E IPHASE VCo -%MODULATOR BIr U T 1024MI 4

FROM PHASE TRANSITION SENSING CIRCUIT

Figure 4-4 Phase-Tracking Loop

A narrow amplitude bandwidth is used because of the small AM inherent in the TM signal Initially 5 kHz was chosen to be the ampli tude bandwidth however when the TM unit was built it was discovered that a loop bandwidth of less than 10 Hz was optimal The narrow tracking bandwidths of the TM unit ensure that only a minimal portion of the TV signal will be affected

The degree of suppression obtainable with the TM APEX is contin gent upontheunit's ability to sense the phase state transitions of the TM signal correctly and to transmit this information to the biphase modulator The suppression is thus inversely relatable to the sensing error rate The TM bit rate is 51 6 kbits/sec therefore a rate of two errors per million bits was chosen as a design goal This error rate would produce one flicker in the TV picture every 10 sec

Figure 4-5 which is derived from data pulse transmission theory relates error rates of received binary signals to S/N In the present APEX circuit this S/N is the ratio at the output of the integrator shown in Figure 4-6 The spectrum of the signal at the input to the integrator is shown in Figure 4-7 It is obtained by convolving the TM signal spec trum with a i 024-MHz carrier The S/N derived from Figure 4-7 is about -7 dB including both phase and amplitude noise It is assumed that

26 10 1_ _ _ _ 10-3 \ ONOFF

2) 10

- 6 10

10 7

10 a 5 7 91 13 15

RATIO OF AVERAGE SIGNAL POWER TO AVERAGE NOISE POWER IN BIT RATE BANDWIDTH db

I 1I 6L 8 10 12 14 16 18 RATIO OF AVERAGE SIGNAL POWER TO AVERAGE NOISE POWER IN NYQUIST BANDWIDTH db Figure 4-5 Error Rate Versus S/N

TM X SIGNAL --- 01 A

1 024 MHz CARRIER CARIERBIPHASE ZERO IMODULATOJR - CROSSING - DRIVE THRESHOLD

TO BIPHASE MODULATOR

Figure 4-6 TM Phase Transition Sensing Circuit

Z7 RELATIVE AMPLITUDE

5 4

0o o o o I ' I o ooO, a

Figure 4-7 Signal Spectrum at Integrator Input (Neglecting Double Frequency Terms)

the phase-tracking loop will eliminate the majority of the phase noise so the S/N has to be adjusted by adding +3 dB The integrator will improve the SIN

The voltage input at the integrator can be represented as

vi(t) A u(t)+nc(t) cos(c ct) - ns(t)sin(wct)

where

A = amplitude of the input pulse

u(t) = step function

nc(t) cos( ct) - ns(t) sin()ct) = narrowband white noise

The output voltage of the integrator is given by

Vout(t) = k f4(t)dt 0 where

kt = constant determined by the circuit

t = integration time

The power output of the integrator will be proportional to the mean square

output voltage v ou(t) Given v (t) it is found that out in

V2(t) = A 2 k 2 t2 + k2Nt S N 0 0 28 where

N = white noise spectral density

S 0 = signal power out

No = noise power out

Therefore 2 S/N = A o 0 N- t The input S/N can be shown to be 2 SI/N = A 1 2B N n where B = white noise bandwidth n The S/N ratio improvement due to the integrator is So/N 0 o = 2Bnt

1 i The improvement is seen to be proportional to the integration time The integration time of the TM-APEX is 8 jisec and the noise bandwidth is taken to be 2 MHz therefore the SIN improvement will be about 15 dB The SIN at the output of the integrator will then be near Ii dB Figure 4-5 indicates that this S/N corresponds to an error rate of 5 errors per 10 million bits which is better than the design goal Obser vation of the processed TV signal indicates an error rate very close to this value

29 5 APEX SYSTEM SCHEMATICS AND LAYOUTS

30 I I

'Lii

Figure 5-1 APEX Modules Inter connlection Diagram

3' F V ,% NL- " fOLDOUT RLV7 fOLDOUT ER A M E Q2 G T

1. IV

LVIC

RI"'

RIIF

32 FOL-DOUT EAMK 6 ps d /Fll 4 PT- l'd

3/I~~~ /l W Ilr

LU 'L

-IT1U O

Fiur 5-gMpoue cerai

'A 3/U33

FOD U ERAIt D1""FOD UTkR Th d b ,d Os, di I di h(' f h i S h TW mG pT

d - -cx, I S- - , ,

z zz

34L

IF F-LE)soc TNN r

Figure 5-4 Biphase Sensing Module achernztic Digam L t I h i 211 d dl pp UI

007

,f e

7, 'T 7W

Schematic rc,-DUTFRMEfU-DUTFRAML Figure 5-6 Buffer Module FOLDOT FRME /36 Th d m m PROPRI TARY TO TRW SYSTEMS GROUP TRW INC d h 11 b pd d I d h d .an I d df yp po h h wh h I d w.p hn p I TRW Sy G0 p TRW I

'4N4

----4-- t--

--- U-- In U

35i Th d ur n PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h 11 b pd d f h d - d I d h df yp P h i h I wh h I hw. h p w p m fTRW Sy n p TRW!

6 CONCEPTUAL REVISIONS

6 1 REVISED APOLLO APEX SYSTEM

The APEX system delivered to Houston consisted of a telemetry and analog-voice subcarrier extractor While no attempt was made to include a TV extractor as part of this system it was known that this exclusion would cause a severe limitation as determined by the theory associated with APEX Each signal must be independently estimated and extracted from the composite spectrum to establish an optimal process Since no attempt was made to extract TV the TM and AV subcarriers could only be extracted partially as determined by the power level of the TV spectrum in the vicinity of these subcarriers An improved approach is to use a TV extractor as part of the basic APEX system This in conjunction with the other revisions listed below would be incorporated as part of the improved system

* The TV APEX unlike the TM and AV APEX would operate at baseband which negates the need for a carrier-to-baseband baseband-to-carrier conversion process and correspondingly simplifies the circuitry involved However the estimation technique is still preserved since the TV baseband filter would be spectrally weighted to conform to the TV spectrum This accomplishes two ends First it allows maxi mum signal exclusion to occur between the TV-AV signals and the TV-TM signals that are being estimated Second since the TV baseband filter has been weighted to conform to the TV spectrum a signal-to-noise improvement will occur This improvement amounts to 30 dB in the region out to Z00 kHz and 6 dB in the I to Z MHz region

* The AV APEX of the delivered system uses a second order low-pass filter as part of its phase-tracking loop However the baseband spectrum of the AV channel is comprised of voice and five subcarriers ranging between 3 9 kHz and 14 5 kHz Correspond ingly use of a single low-pass filter as part of the AV phase-tracking loop is suboptimal Improved estimation would occur if a number of bandpass filters in conjunction with a low-pass filter were used Each bandpass filter frequency would correspond to one of the subcarrier frequencies

37 Th d . a n PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h II b p d f d h d n d I d h f yp h h h wh h f£ h d. h h p p p m f TRW SY m G p TRW I

" Lower telemetry error rates and improved telemetry extraction would occur if the full 20- Lsec bit period (54 6 kbits) was used to determine the TM data state The existing system uses 8 jasec as an integration period but 20 sec would result in greater than 6-dB telemetry SIN improvement substantially reducing the data error rate A second option that could be included would be to intertie the bit rate (51 6 kbits) to the carrier frequency (4 024 MHz) by logic circuitry so that data transitions can be made to occur synchron ously with the input data transitions This too would enhance the TM extraction process

" The signal delay line used as part of the telemetry estimation process currently has a 3- dB bandwidth at 1 8 MHz The delay line used with the improved system would have a 3-dB bandwidth at 3 5 MHz which would minimize degradation to the quality of the TV signal

* Observation of the performance of the low-pass filter currently being used as part of the Apollo TV processing system indicates that an appreciable S/N improvement can occur even though the picture resolution is degraded by the presence of this filter As part of the new APEX system two switchable low-pass filters would be included at the system's input and output The input low-pass filter (switchable to 4 Z MHz 1 8 MHz 2 4 MHz and 5 MHz) would precondition the noise content of the source signal to allow best estimation The output filter which is connected to the TV estimate line would also be switchable to the above mentioned frequencies The choice of bandwidths to be used would be determined by the subjective process of visually observing the resulting TV

6 Z REVISED SYSTEM DESCRIPTION

The new system presented in Figure 6-1 uses three APEX estimators As shown by the block diagram each is connected in a parallel arrangement with its outputs feeding back to a common summation port The input signal consisting of TV FM-FM telemetry and noise is first amplified by an input buffer amplifier to a level usable by the system The amplifier output drives a low-pass filter with bandwidth selection and then a delay line with 20 sec of delay The delay line is required as part of the telem etry estimating system The delayed input signal is routed to the summation port of the APEX system Each of the three units will establish an estimate of its portion of the input signal and will deliver this signal to the summation

38 Th d n af n PROPRIETARy TO TRW SYSTZMS GROUP TRW INC d hi II b p d d I h d . d I d h df yp p h h h f wh h f h d h h p i p M f TRW S m C p TRW !

REVISED TV APEX SYSTEM APEX SYSTEM INPUT TV AV TM AND NOISE OUTPUT

AD NOBSE

0 AMV BASEBAND

--ISNIGITM VICO UT J TAPXOUTPUT _[ ]<_TM DATA OUTPUT BANWIDH SECBADIT TV APEX I v SPELEC CCONTROL [ ~~ L P F" O

] TV BASEBAND

T OUTPUT

BANDWIDTH SELECT CONTROL Figure 6-1 Revised TV-APEX System device causing cancellation of the composite Input signal The three re maining error voltages observed at the output of the summer are fed back to the three APEX inputs to update the estimation process going on within each They are also delivered via an output amplifier to an output port This port is used for monitoring the accuracy of the estimation process It will also be possible to remove the estimate that each APEX unit is delivering so that the aforementioned output can be used as a drive port providing the unsuppressed composite signal

During the normal operation of Ehe system the TV would be observed at the TV APEX estimate output port This port is preceded by another bandwidth switchable low-pass filter If the input S/N ratio is extremely poor this low-pass filter would set to a narrow bandwidth that will im prove the TV S/N ratio However the quality of TV observed with the filter set to this position will be degraded

The telemetry APEX unit requires a priori knowledge concerning the data transition occurrence so that an optimal estimation process can occur The biphase sensing unit observes the composite signal spectrum 20 gsec prior to the APEX system A comparison between this signal and

39 Th d dn nt m PROPRIETARY TO TRW SYSTEMS GROUP R v UNC d Itl b p d d £ d u d hd dl yp P h h el I w h I h dw h h p m f TRW Sy m G p TRW I the TM APEX VCO is made to determine whether the telemetry signal is in a zero or one data state This observation is integrated over a Z0-jisec interval to maximize the TM S/N ratio At the end of this interval a test of data state is made and the results are sent to the TM APEX unit as an information input The output of the biphase sensing unit can also be used as a TM data output The two TM data rates (4 6 and 51 6 kbits) are exact subintegers of the TM carrier frequency which is 4 024 MHz The integers involved are 640 and 20 respectively The TM VCO is divided down by these two ratios dependent upon the data rate being used and acts as a triggering mechanism for state changes within the biphase sensing unit This :is particularly useful in the stabilization of TM esti mate data transitions when the input S/N ratio is poor The transition accuracy results from the high S/N ratio that exists within the phase tracking loop (narrowband) of the TM APEX unit

In conclusion, the revised APEX system provides selectable prefil tering for the composite input signal and selectable postfiltering for the TV estimate It provides a buffered output representative of the input signal when the APEX system is not to be used Additionally when the APEX system is in operation a processed signal representative of the analog voice baseband information is available as well as an output representative of the telemetry data being transmitted Each estimator has been designed to be optimal in performance This is to say that in the case of AV APEX bandpass filters which have been used for each of the subcarriers present as part of the FM-FM AV signal Also the TM APEX integrates over a full data period prior to making a decision of data state This in con junction with the TM-VCO data transition intertie results in an optimal TM estimator The TV APEX to be described later uses a special comb line weighting filter that is optimal in relation to a TV spectrum and that causes two favorable results The first is an improved SIN r tio and second the complete APEX system can now operate with minimal level of interference between units

6 2 4 AV APEX Unit Description

The AV APEX unit shown In Figure 6-2 uses the same phase tracking and amplitude-estimating concept that was used in the original APEX system The modifications that would be incorporated reside in the

40 Th d .1! a n PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h it b p d d f d h d d I d h df yp p h h tb I fh h dw h h p w P M fTRWSy m G p TRWI phase-tracking loop The Information has a number of sub-subcarriers that are summed and then applied to the VCO at the signal's source An inverse technique is indicated to optimally estimate this signal The sub carriers to be processed are located at 3 9 5 4 7 35 10 5 and 14 5 kHz and in a 0 1- to 3-kHz spectral region (audio) Again referring to Figure 6-2 it is seen that the phase-tracking loop consists of a low-pass filter and a set of bandpass filters The center frequencies of each of the five bandpass filters will be determined by the previously mentioned sub carriers' frequencies The bandwidths of these filters will be determined by the baseband information to be processed Each bandpass shall be a single-pole filter and since the VCO is representative of a perfect inte of 4800 grator the phase shift about this loop will not be in excess Correspondingly this will result in loop stability To process the audio spectrum a low-pass filter has been included in parallel with the five bandpass filters The outputs of both are summed and the resultant is used to drive the VCO that closes the loop at the phase detector

0 6TV SAV TM BANDPASS FILTEAS AT SIGNAL IMPUT 5 FREQUENCy El f

TV TM 1 25 MH Vco

CAD AM

PM LOOP BANDPASS FREQUENCIES 1 3 9 kH

12-54kH f3= 7 35 kH f4 105 kH fS 14 5kH

Figure 6-2 AV APEX Unit

41 Th d m f n PROPRIETARY TO TRW SYSTEMS GROUP TRW MC d h 11 b pd d t f d h du. d I d h df p "C h d, h f h h I h d, h h p p fTRWSy M 0 p TRWIn

6 Z 2 TM APEX Unit Description

The TM APEX block diagram shown in Figure 6-3 is comprised of circuitry similar to the original system with these exceptions Additional delay is used in the signal line The original delay which was 7 sec has been increased to Z0 and this increase results in a longer telemetry state evaluation tume A corresponding reduction in the number of data errors in the telemetry estimate results The data state is determined by the coherent amplitude detector the Integrate-and-dump circuit and the 1 024-MHz countdown circuitry If the input data rate is 51 6 kHz the countdown circuitry will be set to divide Z0 On the other hand if the input data rate is 1 6 kHz the countdown circuitry will be set to divide by 640 The VCO which is an Integer value of the data rate is counted down by 20 or 640 and is then used to indicate the correct time for a data state change It also Indicates when the integrate-and-dump circuit is to reset

The remainder of the TM APEX unit is comprised of a narrowband phase-tracking loop and a narrowband amplitude-estimating channel The output of the amplitude channel is multiplied by the phase information

Figure 6-3 TM APEX Unit

42 Th d urn f r PROPRIETAR TO TRW SYSTEMS GROUP TRW INC d h I b p d d £ d h d n d 1 4 h df yp P h ,, h I hi h f h d h h p w p m f TRW Sy m G p TRW I available from the biphase modulator and forms an estimate of the telem etry input signal This estimate is processed by a single-pole filter and then delivered to the summation port Differencing occurs at the sum mation port with the delayed TM input leaving an error that is used to update the phase and amplitude estimate within the TM APEX

6 Z 3 TV APEX Unit Description

The TV APEX unit shown in Figure 6-4 consists of a feedback loop operating at baseband frequencies It is different than the approach pre sented in the two previous cases in that translation from carrier-to-base band and baseband-to-carrier does not occur The loop consists of an amplifier a low-pass filter with a break frequency of 1 5 MHz a time delay of 65 5 iasec two summers and a variable delay attenuator Each device about the inner loop would be built to have very stable transfer functions because unlike the AV and TM estimators feedback in this instance is positive rather than negative

The upper portion of Figure 6-5 diagrams the inner TV weighting loop with positive feedback The lower portion presents the normal

TV AV TM

SIGNAL INPUT +LP TV AV TM Lr AMP fS~

IAV I TM TV

_ _ VAR T DELAY ATTEN -65 5 IjSEC

~EATTN AMPUPU TO MONITOR

PANEL CONTROL BW SELECTIONS 1 2 1 8 24 AND50MHz

Figure 6-4 TV APEX Unit

43 Th d u n - PROPRIETARY TO TRW SYSTEMS GROUP TRW INC d h II b p d d I d h d i d I d h d y p P h h h £ wIh h I h d w h p w M f TRW Sy m G p TRW I spectral density of a TV signal (given by the solid black line) whereas the dotted line presents the response of the TV weighting filter The TV spectrum is concentrated at multiples of the scan frequency (15 750 kHz) with very strong energy concentrations within the region between dc and 157 5 kl-z If the TV spectrum is observed at a higher region (0 500 to 1 5 MHz) it is seen that the level of energy at multiples of the scan fre quency has diminished by 20 to 40 dB Also the energy at each of these frequencies has spread in spectral width relative to that found at the lower frequencies For optimal estimation and extraction the weighting filter must have a similar characteristic Again referring to the block dia gram it is seen that the loop closes with a gain of 0 99 in series with the low-pass filter At the lower frequencies the low-pass filter will have a unity transfer characteristic which in turn causes 99% of the TV base band signal to be sent back to the summer reinforcing the same This reinforcing action occurs at multiples of the TV line frequency whereas cancellation occurs at the in-betweenfrequencies This is consistent with the TV spectrum within this region At higher multiples of the scan rate (0 5 to i 5 MHz) the low-pass filter will approach its 3-dB

TV BASEBAND E 0PA9 FITER WEIGHTED + TV BA SEB AND n ,_~TV LI NE I~7DELAYTRATE

0 - SPECTRUM --- FILTER 20 f SCAN FREQUENCY I

I 40 ~I

60 6______f 2f 20f 21f 70f 71f

Figure 6-5 TV Estimator Filter Block Diagram

44 Th d n in PROPRIETARY TO TRW SYSI EMS GROUP TRW INC d h 11 b p d d f d h d d I d h df p Io h h h f h h f d w h p P m I TRW Sy m G I TRW I response point which in turn causes the closed loop to have a transfer function of 0 707 rather than 0 99 As a result the input TV signal will be reinforced less than at the lower frequencies This also causes the response at each of the multiples of the line frequency to broaden Both factors are consistent with the character of the TV spectrum

In conclusion TV spectral lines which lie in the dc to 157 5-kHz region will be accurately estimated (within I%) which is equivalent to a SIN improvement of 40 dB On the other hand only a 5 to 10-dB S/N improvement will occur in the region between 0 500 and 1 5 MHz as a result of the broadening response characteristic of the weighting filter This also results in a diminished accuracy of signal estimation in this spectral region Still since the baseband filter does have a spectral response similar to that of TV signal enhancement will occur

45 7 SIGNAL SIMULATION EQUIPMENT

A unit was built to simulate the LM television signal and its two subcarriers The first subcarrier located at 1 024 MHz is biphase modulated up to 51 6 kbits The second subcarrier located at I Z50 MHz carries FM-modulated bio-med and voice information The LM signal simulator was required to test the APEX system that was built for NASA

The signal simulator consists of eight brassboards of two types (See the simulator block diagram Figure 7-1 and Figure 7-2 ) The circuitry of the first type is the same as is used in APEX These circuits were critical to the operation of APEX and had to be breadboarded to examine their performance When the operation of these critical circuits was optimized they were used as part of the APEX Since similar cir cuits were needed for the simulator these critical brassboard circuits were integrated into the construction of the LM signal simulator Brass boards 1 5 7 and 8 are of this first type whereas brassboards Z 3 4 and 6 were built to interface with them

A brief description of each brassboard in the simulator follows Board i contains the i 024-MHz VCO and the HP-10534C double-balanced mixer operating as a biphase modulator The VCO is a common base oscillator with tuned tanks in the collector circuit and the emitter feedback branch The frequency of the VCO is controlled by the reverse potential applied to the PC-i36 varicaps in the emitter tank The voltage across the collector tank is filtered to eliminate harmonics and then applied to the balanced mixer Biphase modulation which is produced in the mixer is controlled by an alternating current from Board 2 Board 2 uses an operational amplifier to produce this current when a triggering signal is applied to its input Board 3 has three functions that can be chosen with a switch They are a by-pass line a three-pole bandpass filter and a five-pole bandpass filter The bandwidth of the filters is approximately 200 kHz The filters are used to modify the biphase signal so it more closely approximates the LM biphase signal

Board 4 is a low-gain summing amplifier that is used to superim pose the 3 9 kHz 5 4 kHz 7 35 kHz 10 5 kHz and 14 5 kHz bio-med

46 2

BIPHASE - BIPt4ASE TRIGGER DRIE

I vcoFILTER

IASEBAND NOISE

AUDIO -- - E PO

GENRATOS AMP. VOICE

Figure 7-1. Signal Simulator Block Diagram

FigureO 7-.Sbare iuao SIDI,z "Co _ .FHK&Pt7

447 signals and the 0. 1 to 3-kHz voice signal. The combined signal is proces sed so it can act as the driving signal to FM modulate the i. 250-MHz VCO on Board 5. The operation of the i. 250-MHz VCO is very similar to the operation of the i. 0Z4-MHz VCO. The three-pole bandpass filter on Board 6 performs the same task as the filters on Board 3. Boards 7 and 8 contain the same basic amplifier design (two transistors, direct coupled amplifier with a shunt-series feedback configuration). Board 7 is modified to superimpose the biphase signal, the audio modulated FM signal, a 2. 5-MHz bandwidth white noise source, and a TV baseband signal. Boards 7 and 8 amplify the composite signal to its correct output level of 2 V, peak-to-peak.

The simulator was built and was found to operate as expected. It very closely simulated the spectrum of the LM signal.

48 8. POWER SUPPLIES

The +16 V and -16 V power supplies providing power for the APEX unit are two Model 6i0lA Hewlett-Packard supplies, which are rack mounted using HP 14523A dual rack mounting kit (see Figures 8-i and 8-2). The two units occupy a space 3-i/a in. high x 19 in. wide x iZ-5/8 in. deep.

The electrical modification consists of strapping out the front panel voltage and current limiting controls (Figure 8-3) and providing a rear panel with these controls plus a receptacle for ac power-in and dc voltage out for the APEX drawer (Figure 8-4).

Each power supply has a capability of 16 V at 1. 0 amp. However, the -16 V supply is current limited at 350 ma, and the +16 V supply is limited to 450 ma.

Figure 8-1. Subcarrier Extraction Unit, Front View

49 Figure 8-Z. Cabinet Rear-View Drawer and Dual Power Supply

50 Al (REMOTE CURRE4T PROGRAM) CURCT I PROGRRAAMINGRESSt[OR I ORzoi) (gIK FORrUt A2 (FINE VOLT ADJ) OUTPUT) o A3 (FINE VOLT AJ)

V'OL'TAGE '* - OLAEA4 (PROGRAM VOL15) PROGRAMMIAGRESISTOR (RIOR R1) (IOOO-/V) I I AS (pROGRAM VOLTS)

A6 (INTERNAL COURSE VOLTS)

A7 (OUTPUT CAPACITOR)

-- +S (SENSING +)

-(OUTPUT)

TO LOADl G (GD)

- - (OUTPUT)

5 (SENSING

Q AS (INTERNAL VOLT CONTROL)

___ A9 (PROGRAM CURRENT)

o AIO (CURRENT INTERNAL CONTROL)

1E LOAD MAY BE TAWEVA FROM FRONT OR REAR TERMINALS 2 PATTERN I FOR THE + SV SUPFLY FOR ISV REMOVE G D FROM OUT PUT AND CONNECT TO 4 OUTPUT

Figure 8-3 Strapping Pattern-Rear Panel Control HP 6101A Power Supplies

51 79 359

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4 I4OLES

DETAIL A TYP 2 PLACES

,, 88 25 5375 SEE DETAIL P-

I I----- 4 SPACES60) 31 EQUALS5 5 2,t I 25

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Figure 8-4 Panel - Dual Power Supply (Rear Mount)

52 9 TEST RESULTS

NASA submitted a set of photographs taken at the NASA MSC site that indicate the test results obtained when using the APEX equipment The equipment configuration used to generate these photographs included a TV camera an S-band downlink system complete with modulator transmitter and a Motorola FM demodulator The demodulator output drove the input to the APEX system and its output operated the TV monitor The picture used was a standard "Indian head" test pattern Figure 9-1 shows the TV signal as observed at the output of the APEX system with the subcarriers removed and the S-band link at a signal level of -60 dBm At this power level the link SIN ratio is in excess of +30 dB The test pattern is well defined indicating that the APEX system does not cause degradation Little noise is present because of the S/N ratio

Figure 9-2 was taken with the communication link operating at a signal level at -84 6 dBm This signal level is comparable to that received using a 210-ft antenna Both subcarriers are present and the picture has been processed by the APEX system although the signal extraction switches for both subcarriers are in an "off" state As a result Figure 9-2 exhibits the classic herringbone effect caused by the presence of the subcarriers

Figure 9-3 has been taken under the same conditions as described for Figure 9-2 with the exception that both subcarrier extractors are now operating It is clear that the two offending subcarriers have been removed sufficiently that annoyance to an observer is no longer a problem Graininess can be observed and this has contributed to the S/N ratio associated with the -84 6 communication signal level Although not shown here as the signal level is increased the picture graininess diminishes to that of Figure 9-1 modified by the slight presence of herringbone

53 NAM i F ( ii ) n vn

Figure 9-1. NASA Photograph S-69-57736

NOT REPRODUCIBLE = iiiiiiiiiiFiv =i ...... !

Figure 9-2. NASA Photograph S-69-57738 Fo

Figure 9-3. NASA Photograph S-69-57737 10. HARDWARE

Due to the urgency of delivery of the completed MSC unit, documen tation of the hardware for this program has been kept to a minimum. The panel layout (Figure 10-1) and the photos (Figures i0-2 through 10-5) are included in lieu of documentation.

57 500 DIA F.469 DIA RU LOLES -- 7 L1 -- 9.00REF i

~---4.50 - -4 SPACES~ 2.50 EQUALS 10.00 -. 7 6 HOLES

4. Z

IGN L M LOC K G AVLOCK SIG NAL . . .. . INPUT OUTPUT II 2.25 A4.000 U 6 - - ° 0 ___o PLACE$ TM MCEMOD OFF TM-VCO OFF AVDOMODC TM-EST MOMITOQ AV, ST I 3 PACE=6 2.000 L I 2.50 EQUALS 6.000 00 2 PLACES

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Figure 10-1. Front Panel Figure i0-Z. Cabinet Angle View Showing Modules and P/S

Figure i0-3. Front Quarter View of Drawer with Modules

59 Figure i0-4. Inside Drawer Front Panel Detail

Figure 10-5. Drawer Inside Back

60 11 NEW TECHNOLOGY/INVENTIONS

Inasmuch as the APEX system is a proprietary item with TRW the proposal as submitted contained a proviso that since the APEX system was developed by TRW at its own expense NASA should recognize that the XPEX technique and the device have been jeduced to practice NASA has pioposcd tne inclusion of an article in their contract which is entitled Inventions Made Under Contractor's Independent Reoearch and Develop ment Program (August 1963) and is stated below

Any invention made in the performance of any work by the Contractor' s own product improvement pro gram or the Contractor's independent research and development program even though supported by an allowance of costs for such program as part of the overhead costs hereof will not be subject to the 'New Technology" clause or the "Property Rights in Inventions" clause (whichever is included in this contract) unless said work is identified in writing as being required in the performance of this contract