remote sensing
Article Cascaded Microwave Frequency Transfer over 300-km Fiber Link with Instability at the 10−18 Level
Wenxiang Xue 1,2, Wenyu Zhao 1,2,*, Honglei Quan 1,2,3, Yan Xing 1,2 and Shougang Zhang 1,2
1 National Time Service Center, Chinese Academy of Sciences, Xi’an 710600, China; xuewenxiang@ntsc.ac.cn (W.X.); [email protected] (H.Q.); [email protected] (Y.X.); [email protected] (S.Z.) 2 Key Laboratory of Time and Frequency Primary Standards, Chinese Academy of Sciences, Xi’an 710600, China 3 School of Astronomy and Space Science, University of Chinese Academy of Sciences, Beijing 100049, China * Correspondence: [email protected]; Tel.: +86-029-838-905-64
Abstract: Comparing and synchronizing atomic clocks between distant laboratories with ultra- stable frequency transfer are essential procedures in many fields of fundamental and applied science. Existing conventional methods for frequency transfer based on satellite links, however, are insufficient for the requirements of many applications. In order to achieve high-precision microwave frequency transfer over a thousand kilometers of fiber and to construct a fiber-based microwave transfer network, we propose a cascaded system for microwave frequency transfer consisting of three 100-km single-span spooled fiber links using an improved electronic phase compensation scheme. The × −14 × −18 transfer instability measured for the microwave signal reaches 1.1 10 at 1 s and 6.8 10 at 105 s, which agrees with the root-sum-square of each span contribution. It is feasible to extend the length of the fiber-based microwave frequency transfer up to 1200 km using 4 stages of our cascaded Citation: Xue, W.; Zhao, W.; Quan, H.; Xing, Y.; Zhang, S. Cascaded system, which is still sufficient to transfer modern cold atom microwave frequency standards. −15 −18 5 Microwave Frequency Transfer over Moreover, the transfer instability of 9.0 × 10 at 1 s and 9.0 × 10 at 10 s for a 100-MHz signal 300-km Fiber Link with Instability at is achieved. The residual phase noise power spectral density of the 300-km cascaded link measured the 10−18 Level. Remote Sens. 2021, 13, at 100-MHz is also obtained. The rejection frequency bandwidth of the cascaded link is limited by 2182. https://doi.org/10.3390/ the propagation delay of one single-span link. rs13112182 Keywords: fiber link; frequency transfer; cascaded system; transfer instability Academic Editor: Hugo Carreno-Luengo
Received: 12 April 2021 1. Introduction Accepted: 25 May 2021 Published: 3 June 2021 Long-haul and high-precision frequency transfer between distant laboratories plays a crucial role in numerous fields of fundamental and applied science, such as time and
Publisher’s Note: MDPI stays neutral frequency metrology [1,2], fundamental physics [3,4], and geodesy [5]. The synchronization with regard to jurisdictional claims in of distributed systems (such as phased-array radio telescopes [6–8], very long baseline published maps and institutional affil- interferometry (VLBI) [9,10], and multistatic radar systems [11,12]) are also applications. iations. At present, remote frequency comparison and synchronization is usually performed based on satellite links, including two-way satellite time and frequency transfer (TWSTT) as well as global positioning system (GPS) carrier-phase observations. However, owing to the non-reciprocity of links, these methods are limited by an instability of 10−15 after one day and do not provide the short-term stability necessary for synchronization applications [13]. Copyright: © 2021 by the authors. Licensee MDPI, Basel, Switzerland. Consequently, frequency transfer over satellite links is insufficient for many application This article is an open access article requirements. Fortunately, taking advantage of the low loss, high reliability and active distributed under the terms and phase stabilization potential of optical fiber, high stability has been demonstrated for conditions of the Creative Commons direct optical frequency transfer [14–17] and radio-frequency (RF) transmission using the Attribution (CC BY) license (https:// intensity modulation of an optical carrier [18–25] over fiber links. The instability of fiber- −20 4 creativecommons.org/licenses/by/ based optical frequency transfer reached the 10 level at 10 s over an urban fiber of more 4.0/). than 100 km [16,17], while microwave frequency transfer has presented an instability of
Remote Sens. 2021, 13, 2182. https://doi.org/10.3390/rs13112182 https://www.mdpi.com/journal/remotesensing Remote Sens. 2021, 13, 2182 2 of 13
10−19 level at one day over an urban fiber of about 80 km [19,20]. In addition, simultaneous dissemination of time (1 PPS) and frequency (10/100 MHz) signals generated by atomic clocks over fiber have also demonstrated fairly low residual instability [26,27]. There are two main limitations to the achievable length of distance extension of fiber- based frequency transfer. In the first place, the bandwidth fc of the link noise compensation is limited by the propagation delay τ of the fiber [28,29], which is simplified as fc < 1/4τ. In the second place, frequency transfer instability is dominated by the signal-to-noise ratio (SNR) of the output-detected signal, which is limited by the attenuation of the fiber link. In particular, in the case of microwave frequency transmission using the intensity modulation of an optical carrier, the power of the microwave signal detected at the end of the fiber link is attenuated by twice of link attenuation (expressed in dB) [30], and the performance of link noise compensation is degraded by the chromatic dispersion and polarization mode dispersion (PMD) effect of fiber [19,31]. Therefore, compared with the single-span transfer, the cascaded transfer with independent link noise compensation for each short segment is more suitable for ultra-long-distance fiber-based microwave frequency transfer. The earliest reported study of cascaded microwave frequency transfer has demonstrated an instability of 6 × 10−14 at 1 s and 5 × 10−17 at one-half day over 204-km urban fiber link consisting of two single-span links [32]. In 2016, a 1-GHz signal was transferred through 430 km of urban fiber link with an instability of 1.94 × 10−13 at 1 s and 1.34 × 10−16 at 104 s using a cascaded system consisting of two single-span links, which is a record distance of fiber-based microwave frequency transfer [33]. The National Time Service Center (NTSC) of the Chinese Academy of Sciences (CAS) undertakes the task to generate, maintain, and transmit the national standard of time (Bei- jing Time), which has already demonstrated an ultra-stable microwave frequency transfer over a 112-km urban fiber link with an instability of 4.2 × 10−15 at 1 s and 1.6 × 10−18 at one day [25]. In this work, to achieve microwave frequency transfer with high precision over a thousand kilometers of fiber and construct fiber-based microwave transfer network, we first used fiber spools to perform a microwave frequency transfer experiment with a cascaded system. Subsequently, we investigated the ultra-stable microwave frequency transfer over a 300-km fiber link with the cascaded system, which is composed of three single-span links. The frequency transfer instability of each span, as well as of the cascaded link, is investigated. The 300-km cascaded link reached an instability of 1.1 × 10−14 at 1 s and 6.8 × 10−18 at 105 s. The residual phase noise of the entire link was also measured. The bandwidth of the link noise compensation was unaffected by the length of the entire link.
2. Materials and Methods 2.1. Principle of Frequency Transfer System In fiber-based microwave frequency transfer, two approaches of phase compensation exist: optical path compensation (OPC) and electronic phase compensation (EPC) [34]. Although the OPC approach can achieve better transfer instability, the compensation actuator, which consists of a piezo-electric fiber stretcher and thermally-controlled fiber spool [31,35], is complex and has small dynamic range, which is inconvenient for extensive applications. On the contrary, the EPC approach has strong adaptability and little PMD fluctuation effect due to the voltage-controlled oscillator (VCO) used as an actuator [25], except that the performance is slightly poorer than that of the OPC approach. Therefore, in order to broaden the application range of the fiber-based microwave frequency transfer system, we adopted the EPC scheme. For fiber-based frequency transfer, the phase noise induced by the fiber link was measured by comparing the round-trip signal with the reference signal and compensating it in real time. In a typical implementation of microwave transfer over fiber based on the EPC approach, the forward and backward signals have the same frequency, which leads to the transfer instability being limited by Brillouin backscattering and parasitic reflections along the link, especially for long-distance fiber with more connectors and splices. For this reason, to suppress the influence of the backscattering and parasitic reflections, an Remote Sens. 2021, 13, x 3 of 14
ing it in real time. In a typical implementation of microwave transfer over fiber based on Remote Sens. 2021, 13, 2182the EPC approach, the forward and backward signals have the same frequency, which 3 of 13 leads to the transfer instability being limited by Brillouin backscattering and parasitic re- flections along the link, especially for long-distance fiber with more connectors and splices. For this reason, to suppress the influence of the backscattering and parasitic re- flections, animproved improved frequency frequency transfer transfer system system based based on on the the EPC EPC approach approach was was designed de- with two signed with differenttwo different microwave microwave frequencies frequenc fories the for forward the forward and backward and backward signals. signals. The schematic of The schematicthe of fiber-based the fiber-based frequency frequency transfer transfer system system is shown is shown in Figure in Figure1. 1.
Figure 1. SchematicFigure diagram 1. Schematic of fiber-based diagram transferof fiber-based system transfer principle. system VCO: principle. voltage-controlled VCO: voltage-controlled oscillator; PD: photodiode. os- cillator; PD: photodiode. As shown in Figure1, the reference signal Vr can be expressed as: As shown in Figure 1, the reference signal can be expressed as: Vr ∝ sin(ωrt + ϕr) (1) ∝sin( + ) (1) where ωr and ϕr stand for the angular frequency and the initial phase of the reference signal, where and stand for the angular frequency and the initial phase of the reference respectively. The frequency signal V is generated by a VCO, which can be expressed as: signal, respectively. The frequency signal is0 generated by a VCO, which can be ex- pressed as: V0 ∝ sin(ω0t + ϕ0) (2) ∝sin( + ) (2) where ω0 and ϕ0 represent the angular frequency and the initial phase of V0, respectively. where andNote that represent Equations the (1) angular and (2) frequency are derived and without the initial considering phase theof random , respec- phase noise of tively. Note thethat frequency Equations signal. (1) and (2) are derived without considering the random phase noise of the frequency signal. V0 modulates the intensity of the laser carrier and transmits it along the fiber link to the modulatesreceiving the (RX) intensity end. Atof the the laser RX end, carrier the and one-way transmits trip signal it along detected the fiber by link a fast to photodiode the receiving(PD2) (RX) canend. be At expressed the RX end, as: the one-way trip signal detected by a fast photo- diode (PD2) can be expressed as: h i V ∝ sin ω t − τ + ϕ (3) ∝sin RX − + 0 f 0 (3) where representswhere τf represents the forward the forwardpropagation propagation delay delayof the of fiber. the fiber. V RXis is frequen- frequency-divided cy-divided byby 4 4 with a low-noiselow-noise frequencyfrequency divider divider to to modulate modulate the the intensity intensity of theof the second sec- laser carrier ond laser carrierthat isthat used is toused generate to generate the backward the backward signal. signal. At the transmittingAt the transmitting (TX) end, (TX) the backward end, the backwardsignal is signal detected is detected by the second by the photodiodesecond photodiode (PD1), which (PD1), can which be expressed can be ex- as: pressed as: h i Vback ∝ sin ω0 t − τf − τb /4 + ϕ0/4 , (4) ∝sin − − 4⁄ + ⁄4 , (4)
where representswhere τb therepresents backward the propagation backward propagation delay of the delayfiber, ofand the fiber, carries and V theback carries the round-trip phaseround-trip fluctuations phase fluctuationsaccumulated accumulated along the fiber. along the fiber. In the phaseIn comparison the phase comparison system, the system, reference the reference signal signal is power-splitVr is power-split into two into two arms. arms. In the Infirst the arm, first arm, is mixedVr is mixedwith with by aV 0double-balancedby a double-balanced mixer (DBM) mixer (DBM)to gener- to generate a ate a frequencyfrequency difference difference between between andV 0 and . TheVr latter. The lattersignal signal is frequency-divided is frequency-divided by 4 by 4 with a with a low-noiselow-noise frequency frequency divider divider to generate to generate a signal a signal that can that be can expressed be expressed as: as: ∝sin ( − ) ⁄ 4 +( − )4⁄ . (5) V 1 ∝ sin [(ω0 − ω r)t/4 + (ϕ0 − ϕr)/4]. (5)
In the second arm, is mixed with by a DBM to generate a frequency difference In the second arm, Vr is mixed with Vback by a DBM to generate a frequency difference between and , which can be expressed as: between Vr and Vback, which can be expressed as: h i V2 ∝ sin (4ωr − ω0)t/4 + 4ϕr − ϕ0 + ω0τf + ω0τb /4 . (6)
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V1 is phase-compared with V2 using a saturated DBM to produce an error signal including the noise of link and VCO, which can be expressed as: h i Ve ∝ sin (2ω0 − 5ωr)t/4 + 2ϕ0 − 5ϕr − ω0τf − ω0τb /4 . (7)
The error signal is processed in a loop filter F(p) and used to feedback the VCO. Thus, it can be inferred that ( ω0 = 5ωr/2 . (8) ω0 τf + τb = 2ϕ0 − 5ϕr
As a result, the frequency signal VRX at the RX end becomes: τf − τb V ∝ sin 5ω t − /2 + 5ϕ /2 . (9) RX r 2 r
We assume that the fiber link is reciprocal, that is τf = τb = τ. Thus, Equation (9) is reformulated as: VRX ∝ sin(5ωrt/2 + 5ϕr/2), (10)
which becomes coherent with the reference signal Vr, and the phase fluctuations due to the fiber link are canceled. According to Equations (2) and (8), and the reciprocity assumption of the link, the frequency signal V0 becomes:
V0 ∝ sin[5ωr(t + τ)/2 + 5ϕr/2]. (11)
By comparing the phase of VRX to Vr and VRX to V0 in closed loop, the frequency transfer performance of compensation link and free link can be measured. According to the above analysis, the link noise can be compensated with the scheme shown in Figure1, as long as the frequency of the transmitted signal V0 is roughly 2.5 times that of the reference signal Vr. In the practical implementation of cascade transfer system over urban fiber links, it is preferred to use different frequencies of the transmitted signal for adjacent single-span links to suppress the influence of microwave leakage. In view of this, we designed two kinds of frequency transfer systems in different frequencies, in which the frequencies of reference signal and transmission signal were 4 GHz and 10 GHz, respectively (we called it 10-GHz transfer system), and the other frequencies were 3.6 GHz and 9 GHz, respectively (we called it 9-GHz transfer system).
2.2. Experimental Setup 2.2.1. 10-GHz and 9-GHz Transfer Systems Figure2 shows the detailed scheme of the 10-GHz transfer system over a 100-km fiber spool. The 4-GHz reference signal Vr was generated by a phase-locked dielectric resonant oscillator (PDRO), which was phase-locked on the 100-MHz signal generated by a passive Hydrogen maser (PHM) (Vremya-Ch, Nizhny Novgorod, Russia, VCH-1008). The transmission signal at 10-GHz was provided by a different PDRO phase-locked to a low-noise 100-MHz oven-controlled crystal oscillator (OCXO), which was modulated to the laser intensity by a Mach–Zehnder modulator (MZM) and transmitted to the RX end over the fiber link. The 10-GHz PDRO and 100-MHz OCXO were contained in an OSC-10GHz module, which is equivalent to the VCO in Figure1. At the RX end, the incoming 10-GHz signal was detected by a fast photodiode PD2 amplified by a low-noise amplifier (LNA) (Analog Devices, Wilmington, USA, HMC903LP3E), and frequency-divided by 4 with a low-noise frequency divider (Analog Devices, Wilmington, USA, HMC365) to generate a 2.5-GHz signal. The 2.5-GHz signal modulated the intensity of the second laser by MZM to transmit a backward signal to the TX end through the same fiber. The backward signal was detected by another fast photodiode PD1. The output signal Vback of PD1 was amplified by LNA (Qorvo, Greensboro, USA, TQP3M9008) and sent to a phase comparison system. By comparing the phases of Vr, V0, and Vback (see Figure1), the phase comparison system Remote Sens. 2021, 13, x 5 of 14 Remote Sens. 2021, 13, 2182 5 of 13
sent to a phase comparison system. By comparing the phases of , , and (see Figuregenerated 1), anthe errorphase signal, comparison which was system fed togenerated an active an loop error filter signal, to provide which a was control fed signal. to an Theactive fiber loop link filter noise to willprovide be compensated a control signal. by adjusting The fiber the link frequency noise will of thebe compensated 100-MHz OCXO by adjustingin the OSC-10GHz the frequency module. of Thethe loop100-MHz bandwidth OCXO of in the the compensation OSC-10GHz system module. was The less thanloop bandwidth500 Hz. of the compensation system was less than 500 Hz.
Figure 2. Schematic of the 10-GHz transfer transfer system system ov overer a a 100-km 100-km fiber fiber spool. spool. PDRO: PDRO: phase-locked phase-locked dielectric resonant resonant oscillator; oscillator; MZM: MZM: Mach–Zehnd Mach–Zehnderer modulator; modulator; DCF: DCF: dispersion-compensating dispersion-compensating fi- ber; Bi-EDFA: bidirectional erbium-doped fiber amplifier; LNA: low noise amplifier; OSC-10GHz: fiber; Bi-EDFA: bidirectional erbium-doped fiber amplifier; LNA: low noise amplifier; OSC-10GHz: containing a 100-MHz OCXO and a 10-GHz PDRO; BPF: band pass filter. containing a 100-MHz OCXO and a 10-GHz PDRO; BPF: band pass filter.
The 9-GHz9-GHz distributiondistribution system system is highlyis highly similar similar to the to10-GHz the 10-GHz system, system, where where the 4-GHz the 4-GHzand 10-GHz and 10-GHz components components need to be need replaced to be byreplaced 3.6-GHz by and 3.6-GHz 9-GHz and components, 9-GHz components, respectively. respectively.In the transfer system, each laser was a continuous wave tunable digital supermode distributedIn the transfer Braggreflector system, each (DS-DBR) laser was laser a tocontinuous be used inwave the tunable C-band digital window supermode covering distributedthe 1528-nm Bragg to 1563-nm reflector range. (DS-DBR) The wavelength laser to be used can bein the controlled C-band bywindow a 50-GHz covering (0.4-nm) the 1528-nmInternational to 1563-nm Telecommunications range. The wavelength Union (ITU) can wavelength be controlled grid. by In a order 50-GHz to overcome (0.4-nm) theIn- ternationalinfluence of Telecommunications coherent Rayleigh backscatter Union (ITU) noise, wavelength 0.4-nm differencegrid. In order was to set overcome between thethe wavelengthinfluence of ofcoherent the laser Rayleigh at the TX backscatter end and the no laserise, 0.4-nm at the RXdifference end, which was were set between 1550.92 nmthe wavelengthand 1550.52 of nm, the respectively. laser at the TX In end addition, and the to laser reduce at the the RX effect end, caused which bywere the 1550.92 chromatic nm anddispersion 1550.52 (~17 nm, ps/nm/km) respectively. of In fiber addition, on the frequencyto reduce transfer,the effect a dispersion-compensatingcaused by the chromatic dispersionfiber (DCF) (~17 module ps/nm/km) with a of loss fiber of abouton the 10frequency dB was transfer, inserted a at dispersion-compensating the TX end. Behind the fiberDCF, (DCF) a bidirectional module erbium-dopedwith a loss of about fiber amplifier10 dB was (Bi-EDFA) inserted withat the a TX gain end. of about Behind 10 the dB wasDCF, used a bidirectional to compensate erbium-doped the optical fiber loss amplifier caused by (Bi-EDFA) the DCF and with to a improvegain of about the SNR 10 dB at wasdetection. used to As compensate demonstrated the previouslyoptical loss [ 25ca],used the PMDby the effect DCF hasand a to slight improve influence the SNR on the at detection.fiber-based As microwave demonstrated frequency previously transfer [25], using the PMD the EPC effect approach. has a slight Therefore, influence there on was the fiber-basedno use of polarization microwave scramblers frequency totransfer average using out thethe PMDEPC approach. effect. Therefore, there was no use of polarization scramblers to average out the PMD effect. 2.2.2. Cascaded Microwave Frequency Transfer 2.2.2.A Cascaded cascaded Microwave system of microwave Frequency transferTransfer was established by connecting three single- spanA systems cascaded with system two frequencyof microwave converters. transfer The was cascaded established link by was connecting composed three of three sin- 100-kmgle-span fibersystems spools. with The two whole frequency system converters. was placed The within cascaded the link same was laboratory. composed The of ◦ threelaboratory 100-km temperature fiber spools. was The about whole 25 system± 1 C was and placed the period within of temperaturethe same laboratory. fluctuations The laboratorycaused by air-conditioningtemperature was wasabout about 25 ±40 1 °C min. and The the first period single-span of temperature link was fluctuations a 10-GHz causedtransfer by system, air-conditioning and the second was about and third 40 mi single-spann. The first linkssingle-s werepan both link 9-GHzwas a transfer10-GHz transfersystems. system, In the second and the span, second the DCF and and third Bi-EDFA single-span were notlinks used. were The both schematic 9-GHz diagramtransfer systems.of the constructed In the second cascaded span, transfer the DCF system and Bi-EDFA is depicted were in Figurenot used.3. In The order schematic to output dia- a 100-MHzgram of the signal constructed at the end cascaded of the cascaded transfer system, system a low-noiseis depicted 100-MHz in Figure OCXO 3. In was order phase- to locked to the received 9-GHz signal at the RX end of the third span via a 9-GHz PDRO.
Remote Sens. 2021, 13, x 6 of 14
output a 100-MHz signal at the end of the cascaded system, a low-noise 100-MHz OCXO was phase-locked to the received 9-GHz signal at the RX end of the third span via a 9-GHz PDRO. In order to evaluate the frequency transfer performance of cascaded systems, the frequency instability of cascaded link was measured by directly comparing the 100-MHz output signal from the cascaded system to the 100-MHz signal from the PHM with a phase noise and Allan deviation test set (PN & ADEV TS, Symmetricom 5125A). In ad- Remote Sens. 2021, 13, 2182 dition, the phase difference between the 9-GHz output signal from the cascaded system6 of 13 and the 4-GHz reference signal was measured by a home-made phase difference meas- urement system.
FigureFigure 3.3.Schematic Schematic ofof thethe cascadedcascaded transfertransfer system.system. E/O:E/O: electricalelectrical to optical; O/E: optical optical to to electrical. electrical.
2.2.3.In Frequency order to Conversion evaluate the for frequency the Cascaded transfer System performance of cascaded systems, the frequency instability of cascaded link was measured by directly comparing the 100-MHz The frequency of output signal in the first span was 10 GHz, while the frequency of output signal from the cascaded system to the 100-MHz signal from the PHM with a phase reference signal in the second span was 3.6 GHz. Therefore, we needed to use a frequency noise and Allan deviation test set (PN & ADEV TS, Symmetricom 5125A). In addition, the conversion module (FC1) to convert the 10-GHz signal to a 3.6-GHz one. Similarly, be- phase difference between the 9-GHz output signal from the cascaded system and the 4-GHz tween the second span and the third span, a frequency conversion module (FC2) was also reference signal was measured by a home-made phase difference measurement system. needed to convert the 9-GHz signal to a 3.6-GHz one. Figure 4 shows the simplified schematic2.2.3. Frequency of the frequency Conversion conversion. for the Cascaded The frequency System conversion design was inspired by the harmonic of low-noise frequency divider. The schematic diagram of FC1 is shown in The frequency of output signal in the first span was 10 GHz, while the frequency of Figure 4a. The input 10-GHz signal was frequency-divided by 10 with a low noise fre- reference signal in the second span was 3.6 GHz. Therefore, we needed to use a frequency quency divider (Analog Devices, Wilmington, USA, HMC361 & HMC438) while the 4th conversion module (FC1) to convert the 10-GHz signal to a 3.6-GHz one. Similarly, between (4 GHz) and 2nd (2 GHz) harmonics were extracted from the divided signal, respectively. the second span and the third span, a frequency conversion module (FC2) was also needed Theto convert 2-GHz the signal 9-GHz was signal frequency-divided to a 3.6-GHz one. to 400 Figure MHz4 showswith a thefrequency simplified divider schematic (Analog of Devices,the frequency Wilmington, conversion. USA, The HMC438) frequency and conversion sent into designthe intermediate was inspired frequency by the harmonic (IF) port of low-noisethe microwave frequency DBM divider. to be mixed The schematic with the diagram 4-GHz ofsignal FC1 isand shown to generate in Figure the4a. final The 3.6-GHzinput 10-GHz signal. signal Figure was 4b frequency-dividedshows the schematic by diagram 10 with aof lowthe noiseFC2 which frequency was used divider to convert(Analog the Devices, 9-GHz Wilmington, signal to a USA,3.6-GHz HMC361 one. The & HMC438) 9-GHz frequency while the 4thsignal (4 GHz)was frequen- and 2nd cy-divided(2 GHz) harmonics by 10 (Analog were extracted Devices, from Wilmington, the divided USA, signal, HMC361 respectively. & HMC438) The 2-GHz and the signal 4th washarmonic frequency-divided (3.6 GHz) was to selected, 400 MHz and with amplified a frequency to output divider a (Analogpower of Devices, about 7 Wilmington, dBm. USA,The HMC438) additional and sentnoise into of thethe intermediatefrequency conversion frequency (IF)module, port ofwhich the microwave is expressed DBM in termsto be mixedof overlapping with the 4-GHzAllan deviation, signal and was to generate measured the by final heterodyne 3.6-GHz signal.frequency Figure meas-4b urementshows the with schematic two identical diagram frequency of the FC2 conversion which was modules. used to convert It should the be 9-GHz noted signal that two to a identical3.6-GHz one.modules The 9-GHzwere used, frequency with signaleach connected was frequency-divided to a common by power 10 (Analog supply Devices, and a Remote Sens. 2021, 13, x 7 of 14 Wilmington,common input USA, signal. HMC361 & HMC438) and the 4th harmonic (3.6 GHz) was selected, and amplified to output a power of about 7 dBm.
(a) (b)
FigureFigure 4. 4. SimplifiedSimplified schematic schematic of of the the frequency frequency conversion conversion module. module. (a ()a )Schematic Schematic of of FC1; FC1; (b (b) )Schematic Schematic of of FC2. FC2.
2.3. PhaseThe Difference additional Measurement noise of the frequencySystem conversion module, which is expressed in terms of overlappingThe performance Allan deviation,of frequency was transfer measured is bynormally heterodyne evaluated frequency by the measurement comparison with of the phase of the reference signal and the received signal in the compensated link. In the fiber-based microwave transfer, it is impossible to directly measure the phase difference between microwave signals using commercial devices. Therefore, we designed a heter- odyne system to measure the frequency transfer instability. Figure 5 presents the sche- matic diagram of the heterodyne phase measurement system. The received 10-GHz or 9-GHz signal was frequency-divided by 10 or 9, bandpass-filtered at 4 GHz to select a 4 GHz signal, and amplified with LNA. The output signal of LNA was mixed with the 4-GHz reference signal delivered via the 4-GHz PDRO in a DBM to produce a direct current (DC) voltage ( ). It should be noted that the DBM operated in a saturated mode, and thus the ( ) became nearly independent of input signal level variations and was only proportional to the phase difference between two input signals. In addition, the input signals of DBM were in quadrature using a phase shifter (not shown in Figure 5), so that ( ) was the most sensitive to the phase difference between two inputs of DBM and was approximately equal to 0. We measured ( ) using a multimeter (Keysight, Santa Rosa, USA 3458A) with an effective measurement bandwidth of about 3 Hz. The relative normalized phase fluctuations ( ), also known as propagation delay fluctuations in the fiber link, is given by