An Ultra-Wideband Baseband Front-End
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An Ultra-Wideband Baseband Front-End Fred S. Lee, David D. Wentzloff and Anantha P. Chandrakasan MIT Microsystems Technology Laboratory, Cambridge, MA, 02139, USA Abstract — A 900MHz bandwidth front-end with a their effect on NF and UWB signal to sinusoidal interferer -100dB/decade roll-off for a baseband pulse-based BPSK ratio (SIR). Section III outlines the design of the front- ultra-wideband transceiver is designed and tested in 1.8V end. Finally, Sections IV and V present measured results 0.18µm CMOS. Trade-offs in noise figure (NF) and voltage gain within broadband power-matched and un-matched and conclusions. (voltage-boosted) conditions of the front-end are discussed. The front-end achieves 38dB of gain and 10.2dB of average NF in the power-matched case, and 42dB of gain and 7.9dB II. UWB FRONT-END MATCHING THEORY of average NF in the un-matched case. Theory and Traditionally, source and load impedances of a TX line measurements of the matching techniques and microwave are matched to the characteristic impedance of the TX line design parameters upon NF and UWB signal to sinusoidal interferer ratios will also be presented. to acquire maximum power transfer and minimize Index Terms — Ultra-wideband (UWB) radio, RF front- reflections. However, the benefits of matching should be end, LNA, phase-splitter, NF, SIR, transmission lines, power- revisited for pulsed UWB systems. match, un-match, voltage-boost. A. Benefits of an Un-Matched Network Achieving maximum power transfer from the antenna to I. INTRODUCTION the LNA input through a TX line may not be optimal for a Since the FCC’s First Report and Order on ultra- pulsed UWB signal. Since a common source LNA is a wideband (UWB) communications in 2001 [1], UWB voltage-controlled current source, maximizing the input signaling has become an increasingly popular method of voltage rather than the power creates the largest output wireless data transmission. To transmit and receive signals current. This implies that an infinite input impedance is that span greater than 500MHz invokes new challenges for desired for such an LNA. The reflection coefficient, ΓL, of the design and implementation of such a system. For the such an LNA is 1. This particular case of an un-matched analog front-end, the challenge of following RF tradition network will be called the voltage-boosted case. to achieve power-match and low noise figure (NF) R RL RL becomes a difficult task due to the broad bandwidth. f R = 50 R = 50 Simple L- or T-shaped matching structures are unusable S S CL CL Z = 50 Z = 50 V + O V + O due to their reliance on narrowband, high Q in - in - approximations for impedance transformation. Wideband (a) (b) LC ladder-matching structures become unwieldy because Fig. 1. (a) Power-matched and (b) voltage-boosted LNAs. of the large LC values required for operation in the baseband UWB frequencies of interest. However, noisy Two wideband LNAs are shown in Fig. 1. The power- resistor-based matching structures and active matching matched LNA (a) uses a well-known, wideband matching techniques are good candidates for achieving a power technique. The input impedance of this LNA in the ⋅/ Ω Γ = match for the baseband UWB bandwidth [2]-[3]. Finally passband is Rf Av=50 . | L| 0 for the power-matched in addition to the efforts made on achieving power match case, and the LNA produces a total gain for a pulse of Γ ⋅ for LNAs over the UWB frequency band of interest, there (1+ L) Av=Av. By removing Rf as shown in the voltage- has also been theoretical work done on improving noise boosted LNA (b), the input is terminated only by the gate Γ = figure of UWB LNAs by ignoring power matching in capacitance. | L| 1 for the voltage-boosted case, and the Γ ⋅ ⋅ order to have freedom to directly match for low noise LNA produces a total gain for a pulse of (1+ L) Av=2 Av. figure [4]. The voltage-boosted LNA gain is increased by 6dB if This paper presents a baseband pulse-based BPSK loading effects from Rf are ignored. This 6dB comes from UWB front-end that is a component of a system-on-a-chip a reflection at the input to the LNA. In addition to the transceiver that achieves wireless transmission at 200kbps. noiseless gain, the drain noise contribution from the Section II presents matching technique theory for the low- MOSFET is unchanged. These combined effects improve noise amplifier (LNA), transmission (TX) line effects, and the NF of the voltage-boosted LNA. B. Tradeoffs of an Un-Matched Network interferer within a given |ΓL⋅ΓS| matching environment lies within the boundaries of the two curves at |Γ ⋅Γ |. By not matching the LNA input, reflections are L S A typical antenna |S11| of -10dB corresponds to a Γ introduced into the system. For pulsed UWB signals, these S magnitude of 0.3. The reflection coefficient of the reflections appear as echoes of the pulse at the LNA input. voltage-boosted LNA in Fig. 1 is 1 (capacitor Under constraints that depend on the echo delay time and Γ ⋅Γ the algorithms in the RAKE receiver, these echoes can be termination). The product of | L S| is 0.3. Therefore, ignored or partially recovered for signal detection. from Fig. 3, the worst-case change in SIR is a decrease of A major concern for UWB systems is coexistence with 3dB, and the best-case change is an improvement of narrowband interferers. Unlike the echoes of UWB pulses, 2.2dB, due to amplification and attenuation of the a steady-state sinusoidal interferer forms a non-unity sinusoidal interferer, respectively. voltage-standing-wave-ratio (VSWR) in an un-matched 8 regime with maximum and minimum amplitudes that are 6 correspondingly larger and smaller than when matched. 4 This appears as a frequency-dependent degradation or 2 Best-case SIR [dB] 0 improvement of SIR relative to the power matched case. 0 0.1 0.2 0.3 0.4 0.5 0.6 ∆L 0 ZS VOUT -2 + Zo Z VS - L -4 -6 Fig. 2. Simulation setup for TX line. Worst-case SIR [dB] -8 0 0.1 0.2 0.3 0.4 0.5 0.6 Magnitude of Γ *Γ S L Simulations are performed using the setup in Fig. 2 to Fig. 3. Simulated best- and worst-case change in SIR for determine how standing waves at the LNA input affect un-matched load and source. SIR. As the interferer frequency is swept within the UWB band at fixed TX line lengths (∆L) and Γ’s, the standing III. BASEBAND UWB FRONT-END DESIGN wave amplitude at the LNA input varies between a LNA and Phase Gain minimum and a maximum value, depending only on Buffers Splitter Stages |ΓL⋅ΓS|. The frequencies at which the maxima and minima occur depend on the phases of ΓL and ΓS, ∆L, and Vmatch Vb1 Vout+ Vout- properties of the TX line. RL RB R2 + - + - + - + - + - + - + - + The theoretical maximum and minimum amplitudes of R V - + 1 b2 VDC V M M the standing wave are shown in (1) and (2) with respect to in 1 2 V R b3 Fig. 2. RS B −1 V = V L()1 + Γ ()1− L2 Γ Γ (1) max I L S L − = ()+ Γ ()+ 2 Γ Γ 1 (2) Vmin VI L 1 L 1 L S L - bondwire where V- −1 = ()+ . (3) V+ VI VS ZO ZO Z S RFB RFB L is the loss through the cable. The number of round-trip V reflections N for standing waves to achieve 98% of the b2 final value is calculated using (4), and the time for this to Fig. 4. Front-end circuits. τ ε occur, 98, is calculated in (5). r is relative permittivity of the TX line and c is the speed of light. The wideband pulse-amplifying front-end is shown in −1 N = −1.6989()log ()L2Γ Γ −1 (4) Fig. 4. The LNA is based upon a traditional wideband 10 S L feedback amplifier topology [5] and incorporates inductor −1 τ = ⋅ ∆ ⋅ ε ⋅ (5) 98 2N L r c peaking for bandwidth extension [6]. A modification upon N and τ98 help to specify the length and resolution of the the feedback resistor network allows this LNA to operate RAKE receiver. Also, if 1/τ98 is much greater than the in two broadband matching modes: power-matched and interferer symbol rate, it is reasonable to assume the voltage-boosted. To achieve broadband power-matching modulated interferer behaves mostly like a pure tone. to an antenna with |S11| < -10dB, the NMOS switch Fig. 3 is a plot using the system shown in Fig. 2 of the controlled by VMATCH is closed and R2 is shorted. To Γ maximum and minimum changes in SIR at Vout vs. |ΓL⋅ΓS|, achieve the voltage-boosted case where L approaches where the SIR at Vs is at 0dB. The SIR for any sinusoidal one, the switch is opened. The gain and input impedance of the power-matched 15 and voltage-boosted cases are shown in the equations below. Power-matched, for input VI in (3): 10 = ⋅ + ⋅ −1 A gm (RL // R1)(1 g m RS ) (6) = ⋅ −1 Z LNA R1 A . (7) NF Power-Matched (top) 5 NF Voltage-Boosted (top) Voltage-boosted, for input VI in (3): (dB)DelayNF and (ns) Group Delay Power-Matched (bottom) Group Delay Voltage-Boosted (bottom) = ⋅ + Γ + + ⋅ −1 A gm (1 L )(RL //(R1 R2 ))(1 gm RS ) (8) 0 = + ⋅ −1 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6 1.8 Z LNA (R1 R2 ) A .