Ofdm Receiver Synchronization for a Dvb-H System

Home , Rayleigh fading, Rician fading

OFDM RECEIVER SYNCHRONIZATION FOR A DVB-H SYSTEM

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A Thesis

Presented to the

Faculty of

San Diego State University

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In Partial Fulfillment

of the Requirements for the Degree

Master of Science

Electrical Engineering

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Ramprasad Rao Vasudeva Rao

Summer 2010

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Ramprasad Rao Vasudeva Rao

DEDICATION

To my father Vasudeva Rao For all the unconditional love and support And in loving memory of my mother Shashikala For the person I am today

ABSTRACT OF THE THESIS

OFDM Receiver Synchronization for a DVB-H System by Ramprasad Rao Vasudeva Rao Master of Science in Electrical Engineering San Diego State University, 2010

DVB-H (Digital Video Broadcasting – Handheld) technology is based on ETSI (European Telecommunications Standards Institute) standard designed to bring the broadcast services to battery-powered handheld receivers such as mobile phones and PDA’s. The system is defined based on the existing DVB-T (Terrestrial) system allowing for lower power consumption, lower signal strengths and fast movements. DVB-H system uses Orthogonal Frequency Division Multiplexing (OFDM) technique to deliver multimedia services in any of the three bandwidth modes i.e. 6, 7 and 8 MHz. OFDM has been successful in numerous wireless applications, where its superior performance in multi-path environments is desirable. It is a multicarrier system where the data is transmitted in parallel sub channels by using several subcarriers. All the data carriers in one OFDM frame can be modulated using either QPSK, 16-QAM or 64-QAM. In this thesis, DVB-H (Physical layer) system is simulated using 4096 FFT mode and the signal transmitted is in accordance with the specification of ETSI TR 102 377. Multipath Rician and Rayleigh channel conditions are considered. The main emphasis is on the receiver; concentrating mainly on the Squelch detection, symbol timing synchronization, correcting the coarse frequency offset and channel estimation for the DVB-H system. Various stages of the signal recovery process are also discussed.

TABLE OF CONTENTS

PAGE

ABSTRACT ...... v LIST OF TABLES ...... ix LIST OF FIGURES ...... x ACKNOWLEDGEMENTS ...... xiii CHAPTER 1 INTRODUCTION ...... 1 1.1 Mobile TV: A New Reality...... 2 1.2 Mobile TV Using Terrestrial Broadcasting Networks ...... 2 2 A LITERATURE REVIEW ON ORTHOGONAL FREQUENCY DIVISION MULTIPLEXING ...... 4 2.1 OFDM is Bandwidth Efficient ...... 4 2.2 Advantages and Disadvantages of OFDM ...... 6 2.3 Importance of Orthogonality in an OFDM System ...... 7 2.4 OFDM System Model ...... 8 2.5 Guard Interval and Cyclic Prefix ...... 12 2.6 Multipath Fading Channel ...... 14 2.7 Doppler Shift ...... 16

2.8 Doppler Spread (BD) ...... 16

2.9 Coherence Time (TC) ...... 17 2.10 Rayleigh Fading ...... 17 2.11 Ricean Fading Distribution ...... 17 2.12 Synchronization ...... 18 2.13 Synchronization Issues with OFDM ...... 19 2.14 Effect of Symbol Timing Offset ...... 19 2.15 Effect of Frequency Offset ...... 21 2.16 Synchronization Using the Cyclic Prefix ...... 22 2.17 Channel Estimation ...... 23

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3 DIGITAL VIDEO BROADCASTING- HANDHELD DESCRIPTION ...... 24 3.1 Scope of DVB-H Standard ...... 26 3.2 Basic Aspects of DVB-H Networks ...... 27 3.3 Principle of MFN (Multi Frequency Networks) ...... 28 3.4 Principle of SFN (Single Frequency Network) ...... 28 3.5 Time Slicing ...... 30 3.6 MPE-FEC ...... 31 3.7 Main Issues in DVB-H ...... 32 3.8 How Time Slicing and MPE-FEC Provides a Solution ...... 32 3.9 Delta-T Method ...... 33 3.10 Burst Size and Off-Time ...... 34 3.11 Physical Layer Specifications for DVB-H ...... 36 3.12 Transmitter Input Signal ...... 37 3.13 Channel Coding ...... 39 3.14 Outer Coding and Outer Interleaving ...... 40 3.15 Inner Coding ...... 42 3.16 Inner Interleaving ...... 42 3.17 Bit-Wise Interleaving ...... 42 3.18 Symbol Interleaver ...... 45 3.19 Modulation ...... 46 3.20 4K Mode in DVB-H ...... 46 3.21 OFDM Frame Structure ...... 47 3.22 Reference Signals ...... 49 3.23 Definition of Reference Sequence ...... 49 3.24 Location of Scattered Pilots ...... 50 3.25 Location of Continual Pilot Carriers ...... 51 3.26 Transmission Parameter Signalling (TPS) ...... 51 3.27 Scope of the TPS ...... 52 3.28 TPS Transmission Format...... 52 3.29 Number of RS-Packet per OFDM Super-Frame ...... 54 3.30 Useful Bitrate ...... 54

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4 IMPLEMENTATION AND SIMULATION OF THE PHYSICAL LAYER FOR A DVB-H SYSTEM ...... 56 5 CONCLUSION AND FUTURE WORK ...... 76 REFERENCES ...... 78

LIST OF TABLES

PAGE

Table 3.1. Guard Interval Lengths for all Modes ...... 37 Table 3.2. Frequency Domain Parameters for 4K Mode in 8 MHz, 7MHz and 6 MHz Channels ...... 48 Table 3.3. Time Domain Parameters for 4K Mode in 8MHz, 7MHz and 6MHz Channels ...... 48 Table 3.4. Carrier Indices for Continual Pilot Carriers...... 51 Table 3.5. Carrier Indices for TPS Carriers in 4K Mode ...... 51 Table 3.6. Signalling Format for In-Depth Inner Interleaver ...... 53 Table 3.7. Signalling Format for Hierarchy Information ...... 53 Table 3.8. Signalling Format for Transmission Mode ...... 53 Table 3.9. DVB-H Service Indication ...... 53 Table 3.10. Number of Reed-Solomon 204 Bytes Packets per OFDM Super-Frame for all Combinations of Code Rates and Modulation Forms ...... 54 Table 3.11. Useful Bitrate (Mbit/s) for Non-Hierarchical Systems in 8 MHz Channels ...... 55 Table 3.12. Useful Bitrate (Mbit/s) for Non-Hierarchical Systems in 7 MHz Channels ...... 55 Table 3.13. Useful Bitrate (Mbit/s) for Non-Hierarchical Systems in 6 MHz Channels ...... 55

LIST OF FIGURES

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Figure 2.1. Guard bands in conventional non-overlapping FDM...... 5 Figure 2.2. Difference between bandwidth requirements in FDM and OFDM...... 6 Figure 2.3. Area under the Sine wave for one period is zero...... 7 Figure 2.4. Baseband OFDM system...... 8 Figure 2.5. Block diagram of basic OFDM transmitter...... 10 Figure 2.6. OFDM subcarriers in frequency domain. subcarriers are orthogonal...... 10 Figure 2.7. OFDM demodulation with FFT...... 11 Figure 2.8. Effect of channel over OFDM symbol...... 13 Figure 2.9. Symbol smearing due to channel...... 13 Figure2.10. Showing guard interval with OFDM symbol...... 13 Figure 2.11. Cyclic prefix inserted in guard interval to suppress adjacent channel interference...... 14 Figure 2.12. Multipath signals...... 15 Figure 2.13. A typical Rayleigh fading envelope at 900 MHz...... 18 Figure 2.14. Showing the early and a late symbol timing instants...... 20 Figure 2.15. Regions of timing synchronization...... 21 Figure 2.16. Synchronization using cyclic prefix...... 22 Figure 3.1. Conceptual structure of a DVB-H receiver ...... 26 Figure 3.2. A conceptual description of using a DVB-H transmitter and receiver (Sharing a MUX with MPEG-2 Services)...... 27 Figure 3.3. Single frequency network...... 29 Figure 3.4. Visualization of time slicing...... 30 Figure 3.5. Structure of MPE-FEC frame...... 31 Figure 3.6. Each MPE section header contains delta-t indicating time to the beginning of the next burst...... 34 Figure 3.7. Burst parameters ...... 35 Figure 3.8. Functional block diagram of the DVB-H transmission system ...... 36 Figure 3.9. Structure of MPEG-2 Transport Stream (TS)...... 38

Figure 3.10. Scrambler/descrambler...... 39 Figure 3.11. Randomized transport packets: sync and randomized data bytes...... 39 Figure 3.12. MPEG-2 transport MUX packet...... 40 Figure 3.13. Reed Solomon RS (204,188,8) error protected packets...... 40 Figure 3.14. Conceptual diagram of the outer interleaver and deinterleaver...... 41 Figure 3.15. Data structure after outer interleaving with interleaving depth as 12 bytes...... 41 Figure 3.16. The mother convolutional code of rate ½ ...... 42 Figure 3.17. Inner coding and interleaving...... 43 Figure 3.18. In-depth inner interleaver for 2K and 4K modes...... 43 Figure 3.19. Mapping of input bits onto output modulation symbols, for non- hierarchical transmission modes...... 45 Figure 3.20. Generation of PRBS sequence...... 50 Figure 3.21. Frame structure...... 50 Figure 4.1. Block diagram of the DVB-H system simulated...... 57 Figure 4.2. Scattered pilots generated by PRBS sequence generator...... 58 Figure 4.3. 16-QAM complex data with the scattered pilots...... 58 Figure 4.4. 16-QAM constellation diagram with pilots...... 59 Figure 4.5. One DVB-H symbol with data and reference signal...... 59 Figure 4.6 Real and imaginary values for the duration of OFDM packet...... 60 Figure 4.7. One DVB-H symbol with Cyclic prefix in time domain...... 60 Figure 4.8. Impulse and frequency response of the multipath channel...... 61 Figure 4.9. Constellation diagram of the data after spinning and passing it through the multipath channel with AWGN...... 62 Figure 4.10. Symbol detection system...... 64 Figure 4.11. Plot of normalized cross correlation and auto correlation...... 65 Figure 4.12. Angle of coarse frequency offset...... 66 Figure 4.13. Peak detection system...... 67 Figure 4.14. Averaged cross correlation...... 68 Figure 4.15. Zoomed-in plot of averaged cross correlation...... 68 Figure 4.16. Differentiated cross correlation...... 69 Figure 4.17. Plot of averaged and differentiated cross correlation...... 69

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Figure 4.18. Spectrum of the estimated channel and multipath rician channel with 1% frequency offset...... 72 Figure 4.19. Constellation diagram of the received data after coarse frequency offset...... 72 Figure 4.20. 16 QAM constellation diagrams at the transmitter and at the receiver after equalization...... 73 Figure 4.21. QPSK constellation diagram of the received data after coarse frequency offset...... 73 Figure 4.22. QPSK constellation diagram at the transmitter and at the receiver after equalization with frequency offset of 1%...... 74 Figure 4.23. Spectrum of the extimated channel and Rayleigh channel with 1% frequency offset...... 74 Figure 4.24. constellation diagram of the received data after coarse frequency offset fot he Rayleigh fading channel...... 75 Figure 4.25. 16 QAM constellation diagram at the transmitter and at the receiver after equalization for the Rayleigh fading channel...... 75

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ACKNOWLEDGEMENTS

I am heartily thankful to my thesis committee chair, Prof fred j harris for his most valuable advice and guidance through out the work of my research. Sincere thanks to committee members, Prof Mahasweta Sarkar and Prof Joseph Katz for agreeing to serve in my thesis committee. I would like to thank my family and best friends for their support in many aspects during the completion of my research.

CHAPTER 1

INTRODUCTION

We live in an era surrounded by technology and gadgets. New technologies, products and Services are constantly being evolved. Digital Cameras found acceptance because they set free our creativity. The Skype was successful as it did away with the feeling of having limited talk time. With Ipod, you can store unlimited number of songs and pictures. The multimedia mobile phone, which has evolved as a realization of all these needs, is indeed such a product [1]. We are entering a new era of terrestrial broadcasting where the variety of ways in which TV is experienced is escalating dramatically. The new world of multimedia is an extension of digital multimedia delivered today via cable and satellite, DTH and DSL platforms, but with advanced compression and broadcasting technologies. The mobile world is also quite different – carrying smaller screens and requiring lower data rates to process the information, but in a much more challenging environment of moving devices and varying signal strengths. Fortunately the technologies for delivery of multimedia not only have been perfected for such an environment but also are being launched commercially. The broadcast technologies that have allowed us to watch TV, albeit in our own homes, have now been modified to enable the same programs to be broadcast to the mobiles [1]. Digital Video Broadcasting to Handhelds (DVB-H) or digital multimedia broadcasting (DMB) is an evolution of such products. Broadcasting to Handhelds is likely to be perceived as “TV on a Mobile” [2]. The future of TV viewing will target net books, portable hand-held devices such as high tech mobile phones and a new breed of low cost portable TV’s. Advanced studies in Digital Television over the years have led to the development of important standards that are adopted by different parts of the world. Four important Digital Television Broadcasting standards and the countries that have adopted are Advanced Television Service Committee (ATSC)-USA, Digital Video Broadcasting (DVB)-Europe, Integrated Services Digital Broadcasting (ISDB)-JAPAN, and Terrestrial Digital Multimedia

Broadcasting (T-DMB)-South Korea. Each of the mentioned standards is the result of years of research, co operation between governments and industries. Apart from the benefit of high definition videos, good sound quality, widescreen format and multichannel sound due to Digital broadcasting; there is one key factor that has contributed to the success of Digital Broadcasting. This key factor is the ability of all the transmitters on some area to transmit on the same frequency. This kind of operation is called as Single frequency network (SFN) and it significantly contributes to the use of radio spectrum [1].

1.1 MOBILE TV: A NEW REALITY Mobile TV is now a reality. Though it’s relatively a new technology, it has been proven. It is the transmission of TV programs or video for a range of wireless devices ranging from mobile TV capable phones to PDA’s and wireless multimedia devices [1]. The growth of mobile TV brings challenges for everyone. It provides the users with high channel capacity and interactivity ‘on the road’ [2]. It brings new aspects on the personal information handling, whether it is about leisure time with entertaining TV program clips or complicated business solutions. Interactive mobile multimedia is one of the key ideas to the next step of multimedia era, and broadcast components provided with DVB-H (Digital Video Broadcasting – Handheld) is a useful addition to the conventional cellular radio networks serving the users with point to point connections.

1.2 MOBILE TV USING TERRESTRIAL BROADCASTING NETWORKS The terrestrial broadcasting networks work in the VHF (30 MHz to 300 MHz) and UHF bands (300 MHz to 3 GHz). These networks in the USA, Europe, Japan and other countries are migrating to Digital TV broadcast stations, which helps in reducing bandwidth by packing seven to eight standard definition TV programs into the same frequency slot that was occupied by only one analog carrier. The concept of using the mobile TV using terrestrial broadcasting networks is somewhat related to the FM radio receivers built into the mobile handsets. Here the radio reception is from the FM channels and does not use the capacity of the 2G or 3G networks on

which the handset may be working. Even if there is no 2G or 3G mobile coverage, the FM radio continues to work. Mobile TV using terrestrial broadcast technologies follow the same concept and uses the VHF or UHF spectrum for carriage [1]. We can say that the spectrum used does not need to be allocated from the 3G pool, which is highly priced and scarce. For the purpose of carrying mobile TV, the TV broadcasting community has modified and enhanced the well established DVB-T (Digital Video Broadcasting-Terrestrial) which is used for digital TV broadcasting in Europe. This is made suitable for carrying television signals to a handheld and the new modified standard is renamed DVB-Handheld. In the further chapters, I would be covering the details of physical and link layer elements of the DVB-H system, a literature on the OFDM and the OFDM receiver synchronization for the DVB-H system. The implemented system and simulation results are also shown.

CHAPTER 2

A LITERATURE REVIEW ON ORTHOGONAL FREQUENCY DIVISION MULTIPLEXING

The demand for high speed wireless applications and limited RF Channel Bandwidth has spurred the development of power and bandwidth efficient air interface schemes. Cellular telephone systems have gone through such a growth process. The most spectrally efficient modulation scheme that has been adopted for the high speed wireless applications like WLAN, Video Broadcasting etc which requires large data rates with robustness to radio channel impairments, is OFDM (Orthogonal Frequency Division Multiplexing). The concept of OFDM has been around as early as 1960’s. The OFDM technique was used in several high frequency military systems. In the 1980’s OFDM was studied for high- speed modems, digital mobile techniques for multiplexed QAM using DFT [3]. However OFDM did reach its maturity for employment in wideband data communications over mobile radio FM channels, high bit rate digital subscriber lines (HDSL; 1.6 Mbps) asymmetric digital subscriber lines(ADSL; 6 Mbps), very high speed digital subscriber lines ( VDSL; 100 Mbps), Digital Audio Broadcasting (DAB), Terrestrial Broadcasting only during 1990’s [4]. One of the preliminary reasons to use OFDM in most of latest high speed wireless technologies is its robustness against frequency selective fading or narrow band interference. Because if we look into the Single carrier system, a single fade or a interferer can cause the entire link to break.

2.1 OFDM IS BANDWIDTH EFFICIENT Frequency Division Multiplexing is one of the popular techniques used in Radio and TV transmission. The frequency spectrum is divided into several logical channels, giving each user a exclusive possession of the frequency band. The basic approach is to divide the available bandwidth of a single physical medium into number of smaller, independent frequency channels. Using modulation, independent message signals are translated into different frequency bands. All the modulated signals are

combined in a linear summing circuit to form a composite signal for transmission. Carriers that are used to modulate the message signals are called subcarriers. Guard bands are used here to avoid interference between successive channels. Figure 2.1[5] shows the use of guard bands in FDM systems.

Figure 2.1. Guard bands in conventional non-overlapping FDM. Source: IIT Kharagpur NPTEL. (2010). Module 2, Lesson 7. [Online]. Available: http://nptel. iitm.ac.in/courses/Webcoursecontents/IIT%20Kharagpur/ Computer%20netw- orks/pdf/M2L7.pdf

From the Figure we can also see that the channels are separated by unused bandwidth to avoid interchannel interference. However it leads to inefficient use of the spectrum and hence occupies lots of bandwidth. To cope up with this inefficiency, parallel data systems with overlapping multicarrier modulation technique also known as OFDM can be used. Figure 2.2 shows the spectral efficiency of an OFDM system Vs conventional FDM system. It can be seen that, OFDM saves almost 50% of the bandwidth [4]. But to avoid the crosstalk between subcarriers, Orthogonality has to be maintained between different modulated carriers. Figure 2.2 [5] shows the comparison between FDM and OFDM channel placements for 9 Subcarriers. From the Figure we can see that the subcarriers can be overlapped. Other advantage of OFDM system over FDM apart from bandwidth efficiency is in case of frequency selective fading, OFDM has a distinct advantage over single carrier

Figure 2.2. Difference between bandwidth requirements in FDM and OFDM. Source: IIT Kharagpur NPTEL. (2010). Module 2, Lesson 7. [Online]. Available: http://nptel.iitm.ac.in/courses/Webcoursecontents/IIT%20Kharagpur/ Computer%20netw-orks/pdf/M2L7.pdf systems, such that it spreads out the fade over many symbols. Hence it randomizes the burst errors caused by fading or impulse interference so that instead of several symbols being completely destroyed; rather it gets slightly distorted. Error correction coding can then be used to correct for the few erroneous subcarriers [6].

2.2 ADVANTAGES AND DISADVANTAGES OF OFDM Some of the key advantages and disadvantages of an OFDM transmission scheme according to [6] are: • It is an efficient way to deal with multipath; for a given delay spread. Complexity for implementing the OFDM system is much lower than that of a single carrier system with an equalizer. • For a slow time varying channels, the capacity of the data rate per subcarrier can be enhanced according to the signal-to-noise ratio of that particular subcarrier. • OFDM is robust against narrowband interference, because such interference affects only small percentage of subcarriers. • Its makes the Single Frequency Network (SFN) possible, which is especially attractive for broadcasting applications.

Drawbacks of OFDM systems according to [6] are: • Susceptibility to phase noise and frequency offset. • It has relatively large peak to average power ratio, which tends to reduce the power efficiency of the RF amplifier.

2.3 IMPORTANCE OF ORTHOGONALITY IN AN OFDM SYSTEM The word orthogonal indicates that there is a precise mathematical relationship between the frequencies of the carriers in the system. Since carriers are all sine/cosine wave, we know that the area under one period of sine/cosine wave is zero. This is shown in Figure 2.3.

Figure 2.3. Area under the Sine wave for one period is zero.

As explained in [7], let us consider a sine wave m and multiply it by a sinusoid (sine or cosine) of frequency n, where both m and n are integers. The integral or area under this product is given by [7]: f(t) = sin mwt x sin nwt (2.1) By the simple trigonometric relationship, this is equal to a sum of two sinusoids of frequencies (n-m) and (n+m). =1/2 cos (m-n) wt – 1/2 cos (m+n) wt (2.2) These two components are each a sinusoid, so the integral is equal to zero over one.

(2.3) Hence we can say that, when we multiply a sinusoid of frequency n by a sinusoid of frequency m/n the area under the product is zero. In general for all integers m and n, sin mx, sin nx, cos mx, cos nx are all orthogonal to each other. Thus understanding the Orthogonality principle is very important in an OFDM. It is the Orthogonality that allows the simultaneous transmission of a lot of sub-carriers in a tight frequency space without interference from each other [7].

2.4 OFDM SYSTEM MODEL OFDM is a special case of multicarrier transmission where a single data stream is transmitted over a number of lower rate subcarriers. OFDM can be seen either as a modulation technique or a multiplexing technique [4]. In this technique, the data symbols modulate a parallel collection of regularly spaced subcarriers. The subcarriers have the minimum frequency separation required to maintain Orthogonality of their corresponding time domain waveforms, yet the signal spectra corresponding to different subcarriers overlap in frequency. The spectral overlap results in a waveform that uses available bandwidth with very high bandwidth efficiency. Figure 2.4 shows the model of a baseband OFDM system model [8].

Figure 2.4. Baseband OFDM system.

The core of an OFDM system is in the IDFT in the transmitter and DFT in the receiver. The DFT process takes the discrete signal sequence obtained by periodic sampling

of a continuous signal in the time domain and projects them on a set of orthogonal frequencies to do the spectral analysis of the signals. Mathematically DFT is given by:

(2.4) To recover the sequence x(n) from the frequency samples, IDFT process is utilized. Mathematically IDFT is given by:

(2.5) The order of IDFT and DFT is not really important, as both the processes are linear. With the knowledge of the DFT and IDFT equations, I will move on to the description of the OFDM system. Let {D0, D1, D2,…..,DN-1} represent a block of N complex data symbols chosen from an appropriate signal constellation such as quadrature amplitude modulation (QAM) or phase shift keying (PSK). Digital Signal Processing techniques rather than frequency synthesizers can be deployed to generate orthogonal subcarriers. The IDFT as a linear transformation maps the complex data symbols [D0, D1, D2,…..,DN-1] to OFDM symbols [d0,d1,d2,…….,dN-1] such that:

(2.6) Yielding a time domain sequence {dk, k=0,1,2,3,…….,N-1}. Figure 2.5 shows the basic OFDM transmitter. To mitigate the effects of intersymbol interference (ISI) caused by channel delay spread, each block of N IFFT coefficients is typically preceded by a cyclic prefix (CP) or a guard interval consisting of Ng samples, such that the length of the CP is at least equal to the channel length Nh in samples, where µ = (Th/Ts)N, Th is the length of (Continuous) channel, and Ts is the duration of a OFDM block or symbol. In the next section, detailed explanation about the cyclic prefix can be seen. In OFDM, subcarriers overlap and this can be seen in Figure 2.6 [9] and also the Orthogonality of overlapping subcarriers can be seen.

Figure 2.5. Block diagram of basic OFDM transmitter. Source: F. J. Harris, “Orthogonal frequency division multiplexing, OFDM,”, presented at Department of Electrical Engineering, San Diego State University, San Diego, CA, 2008.

Figure 2.6. OFDM subcarriers in frequency domain. subcarriers are orthogonal. Source: Arvind Padmanabhan . (2008). Mobile and Wireless - An overview of OFDM [Online]. Available: http://mobilewireless.files.wordpress.com/2008/03/ofd m-subcarriers.jpg]

A basic representation of the equivalent complex baseband transmitted signal is [8]:

(2.7)

Cyclic prefix is simply a repetition of the last Ng IFFT coefficients. Alternatively, a cyclic suffix can be appended to the end of a block of N IFFT coefficients that is a repetition of the first Ng IFFT coefficients [3]. At the receiver, the received complex baseband signal is sampled with an Analog-to- Digital converter (ADC), usually with a sampling interval, ΔT = Ts /N. Sometimes fractional sampling is used, where the sample period is (1/M) ΔT, where M is an integer greater than one. For simplicity, assume here that M=1. Under the condition that Ng >= Nh, the linear convolution of the transmitted sequence of IFFT coefficients with the discrete time channel is converted into a circular convolution. This helps in removal of ISI almost completely. Figure 2.7 shows the basic OFDM receiver structure.

Figure 2.7. OFDM demodulation with FFT. Source: F. J. Harris, “Orthogonal frequency division multiplexing, OFDM,”, presented at Department of Electrical Engineering, San Diego State University, San Diego, CA, 2008.

The received signal for a time varying random channel according to [8] is:

(2.8) The received signal is sampled at t=k/fs for k={-k1,…..N+k2-1}. With no inter-block interference, and assuming that the windowing function satisfies w(n-1)=δnl the output of

the FFT block at the receiver, after the removal of guard interval for each block of N received samples is:

(2.9) Where

(2.10) A complex number Hn is the frequency response of the time invariant channel h(t-τ) at frequency n/T. Hence,

(2.11)

In Equation 2.10, n(k) is white Gaussian noise. The N frequency domain samples are each processed with a simple one tap Frequency Domain Equalizer (FDE) and applied to a decision device to recover the data symbols or to a metric computer if error correction coding is used. Ease of equalization is often touted as the main advantage of OFDM.

2.5 GUARD INTERVAL AND CYCLIC PREFIX The mobile radio channel places fundamental limitations on the performance of wireless communication systems. The transmission path between the transmitter and the receiver can vary from simple line-of-sight to one that is severely obstructed by buildings and mountains. Even the speed of motion impacts how rapidly the signal level fades as a mobile terminal moves in space. The OFDM symbols are transmitted to the receiver through these mobile radio channels. The Multipath and several other effects in these wireless channels would rapidly modify the characteristics of the transmitted signal over a small travel distance or time interval. Figure 2.8 [10] shows how the channel distorts the transmitted OFDM symbol. Figure 2.9 shows the effect of the channel on OFDM symbols. It can be seen in Figure 2.9 that there is a cross talk between adjacent OFDM symbols. To combat the effect of channel distortion on OFDM symbols, guard interval precedes every OFDM symbol.

Figure 2.8. Effect of channel over OFDM symbol. Source: F. J. Harris, “Orthogonal frequency division multiplexing, OFDM,”, presented at Department of Electrical Engineering, San Diego State University, San Diego, CA, 2008.

Figure 2.9. Symbol smearing due to channel. Source: F. J. Harris, “Orthogonal frequency division multiplexing, OFDM,”, presented at Department of Electrical Engineering, San Diego State University, San Diego, CA, 2008.

This eliminates intersymbol interference almost completely. The guard time is chosen larger than the expected delay spread, such that multipath components from one symbol cannot interfere with the next symbol. Guard interval is also useful for implementing time and frequency synchronization functions in the receiver since the guard interval contains repeated symbols at known sample spacing. The guard time could consist of no signal at all. In that case, the problem of intercarrier interference (ICI) would arise. ICI is crosstalk between the subcarriers, which means they are no longer orthogonal. Figure 2.10 shows an OFDM symbol with guard interval.

Figure2.10. Showing guard interval with OFDM symbol. Source: F. J. Harris, “Orthogonal frequency division multiplexing, OFDM,”, presented at Department of Electrical Engineering, San Diego State University, San Diego, CA, 2008.

Due to ICI, when the OFDM receiver tries to demodulate the first subcarrier, it will encounter some interference from the second subcarrier, because within the FFT interval, there is no integer number of cycle’s difference between subsequent subcarriers. To eliminate ICI, the OFDM symbol is cyclically extended in the guard time. This ensures that delayed replicas of the OFDM symbol always have an integer number of cycles within the FFT interval, as long as the delay is smaller than the guard time. Hence multipath signals with delays smaller than the guard time cannot cause ICI. In Figure 2.11 [10] we can see the Cyclic prefix inserted in the guard interval time prior to each of the OFDM symbol before transmitting.

Figure 2.11. Cyclic prefix inserted in guard interval to suppress adjacent channel interference. Source: F. J. Harris, “Orthogonal frequency division multiplexing, OFDM,”, presented at Department of Electrical Engineering, San Diego State University, San Diego, CA, 2008.

2.6 MULTIPATH FADING CHANNEL Multipath fading is caused by the transmission of the signal along different paths and resulting in simultaneous reception. The transmitted signal goes through the various environmental parameters such as refraction, ionosphere reflection, and reflection from various terrestrial objects like buildings and mountains before it hits the receivers. Figure 2.12 [11] shows the conceptual diagram of the multipath signals and it can seen in this figure that the transmitted signals reach the receiver after many reflections. Depending on the amplitudes and phase of the signal, the result of this could be that the signals cancel each other completely or significant attenuation in the resultant signal. Below shows the effect of multipath at various signal reception conditions: • Fast moving user: Fast fluctuation of signal amplitude and phase. • Analog Television Signal: Ghost images. • Satellite Positioning System: miscalculation of distance between transmitter and receiver resulting in a wrong estimate of the position.

Figure 2.12. Multipath signals. Source: Eric Lawrey. (1997). The suitability of OFDM as a Modulation technique for wireless telecommunications, with CDMA comparisons [Online]. Available: http://www.skydsp.com/publications/4thyrthesis/chapter1.htm

• Wideband (digital) Signal: Intersymbol interference and dispersion. Since DVB-H system is for handheld terminals, it is required to know the factors that influence small scale fading like: • Speed of the Mobile- relative motion between the base station and the mobile receiver results in random frequency modulation due to different Doppler shifts on each of the multipath components [6]. • Speed of surrounding objects- If objects in the radio channel are in motion they induce a time varying Doppler shift on multipath components. If the surrounding objects move at a greater rate than the mobile, then this effect dominates the small-scale fading [6]. • Transmission bandwidth of the signal- If the transmitted signal bandwidth is much larger than the bandwidth of the multipath channel, the received signal would be distorted however the received signal strength will not fade much over a local area. Multipath delay nature of the channel is described by delay spread and coherence bandwidth. Time varying nature of the channel caused by movement is quantified by Doppler spread and coherence time. Due to the relative motion of the transmitter and receiver in the mobile communications, it highly important to consider the effect of Doppler shift in the receiver structure.

2.7 DOPPLER SHIFT Consider a mobile moving at a constant velocity v, along a path segment having length d between points X and Y, while it receives signals from a remote source S. The difference in path lengths traveled by the wave from source S to the mobile at points X and Y is Δl = dcosθ = vΔtcosθ, where Δt is the time required for the mobile to travel from X to Y, and θ is assumed to be the same at points X and Y since the source is assumed to be far away. The phase change in the received signal due to the difference in path lengths is represented by [6]:

(2.12) And hence the apparent change in frequency, or Doppler shift, is given by fd [6], where fd = (1/2π) * (ΔФ/Δt) = (v/λ) * cosθ (2.13) Equation 2.13 relates the Doppler shift to the mobile velocity and the spatial angle between the direction of motion of the mobile and the direction of the arrival of the wave. Also from the Equation 2.13 we can infer that, if the mobile is moving toward the direction of arrival of the wave, the Doppler shift is positive, and if the mobile is moving away from the direction of arrival of the wave, the Doppler shift is negative (i.e., the apparent received frequency is decreased) as said in [6]. Hence Doppler shift cannot be ignored in a mobile radio environment.

2.8 DOPPLER SPREAD (BD) It is a measure of the spectral broadening caused by the time rate of change of the mobile Radio channel and is defined as the range of frequencies over which the received Doppler spectrum is essentially non-zero. If a pure sinusoidal tone of frequency fc is transmitted, the received signal spectrum, called the Doppler spectrum, will have components in the range fc – fd to fc + fd where fd is the Doppler shift. The amount of spectral broadening depends on fd which is a function of the relative velocity of the mobile [6]. Thereby Bd is fm = v/λ (2.14) Where fm is the maximum Doppler shift.

2.9 COHERENCE TIME (TC) It is the time domain dual of Doppler spread and is used to characterize the time varying nature of the frequency dispersiveness of the channel in the time domain [6]. The Doppler spread and coherence time are inversely proportional to one another. A channel is said to slow fading if the symbol duration is within the coherence time. If the symbol duration is longer than the coherence time then the channel is fast fading. This causes frequency dispersion (also called time selective fading) due to Doppler spreading. Tc ~ 1/fm (2.15) A channel can also be described by the coherence bandwidth whether it is a frequency selective fading or flat fading in frequency domain. The coherence bandwidth is approximately inverse of the maximum delay spread. fo ~ 1/ Tmax (2.16)

2.10 RAYLEIGH FADING If the multiple reflective paths are large in number and there is no line of sight signal component, the envelope of the received signal is statistically described by a Rayleigh probability density function [12] (pdf) given by:

(2.17)

σ is the rms value of the received voltage signal before envelope detection and σ2 is the time average power of the received signal before envelope detection. The above equation is according to [6]. Figure 2.13 shows the Rayleigh fading envelope for the carrier with the receiver speed = 120 km/hr. Rayleigh fading can also be termed as small scale fading.

2.11 RICEAN FADING DISTRIBUTION When there is a dominant non fading signal component signal component present, such as a line of sight propagation path, the small scale fading envelope is described by a Ricean fading probability density function (pdf). The Ricean distribution according to [6] is given by:

(2.18)

Figure 2.13. A typical Rayleigh fading envelope at 900 MHz. Source: S. Bernard. “Rayleigh fading channels in mobile digital communication systems, part I: Characterization,” IEEE Communications magazine, Vol. 35, pp. 90-100 July, 1997.

In the Equation 2.8, parameter A denotes the peak amplitude of the dominant signal and Io (.) is the modified Bessel function of the first kind and zero-order. In this thesis, the channel model that has been considered is: the transmitted signal is convolved with the impulse response of the channel and adding a Additive White Gaussian Noise (AWGN). Representation in the equation form can be seen as:

(2.19)

2.12 SYNCHRONIZATION OFDM, like any other digital communication system, requires synchronization. However, the requirements and resources for synchronization of multicarrier system are different than a single carrier system. For example, in OFDM, one can tolerate larger errors in estimating the start of a symbol than in a single-carrier system. This is due to OFDM’s longer symbol duration and its cyclic prefix. On the other hand, frequency synchronization in

OFDM must be tighter than that in single carrier systems, due to the narrowness of the OFDM subcarriers [3]. Before an OFDM receiver can demodulate the subcarriers, it has to perform at least two synchronization tasks. It has to find out where the symbol boundaries are and what the optimal timing instants are to minimize the effects of intercarrier interference (ICI) and Intersymbol Interference (ISI). Most OFDM systems have a cyclic prefix that can be used for synchronization. The cyclic prefix acts as a pilot data. There is usually some tolerance for Symbol timing errors when cyclic prefix is used to extend the symbol.

2.13 SYNCHRONIZATION ISSUES WITH OFDM 1. The symbol timing synchronization, determinant of correct symbol start position, i.e the FFT window position, before the FFT demodulation at the receiver end. 2. the carrier frequency synchronization (i.e., carrier frequency recovery technique), utilized to eliminate the carrier frequency offset caused by the mismatch of the local oscillators between the transmitter and the receiver, nonlinear characteristic of the wireless channel as well as the Doppler shift. 3. Sampling clock synchronization between the transmitter and receiver. The sampling clock difference would degrade the performance of the systems. In packet based transmission systems, synchronization of an OFDM signal to find an estimate of where the symbol starts is done using the preamble that is transmitted with the OFDM signal. Usually this requires faster acquisition. However, in broadcast systems, the start of symbol detection is done using the cyclic prefix. Acquisition time is not very important. Therefore the coarse symbol timing synchronization should be much more accurate in the burst mode systems.

2.14 EFFECT OF SYMBOL TIMING OFFSET OFDM systems are relatively robust with respect to the timing offsets. Symbol timing instants may vary over an interval equal to the guard time without causing ISI or ICI as seen in the Figure 2.14. There are two cases of symbol timing offset, first one estimates the boundary location before the ideal location and the second one is after. The first one doesn’t cause serious problems, because it contains cyclic prefix, which happens to be the

Figure 2.14. Showing the early and a late symbol timing instants.

last part of the OFDM symbol. However in the latter case, where the FFT interval extends over a symbol boundary, Intercarrier interference and Intersymbol interference occurs. The ideal demodulation is:

(2.20) Where k is the subcarrier index, n is the sample index in time domain. If symbol timing offset is ε, then we have

(2.21) Therefore,

(2.22) In the Equation 2.12, means that the phase rotation is proportional to sub-carrier index k. Phase rotation caused by the first case can be compensated by frequency domain equalizer. However if the ε becomes large, the phase rotation becomes serious and would increase the difficulty in channel estimation. Also the large ε would introduce ISI from the previous OFDM symbol. In the second case, since the FFT window contains samples from the following OFDM symbol, this introduces ISI, which completely destroys the orthogonal property. Best region for starting the DFT is illustrated in the Figure 2.15. This means that larger the cyclic prefix, more error the system can tolerate.

Figure 2.15. Regions of timing synchronization. Source: Y. G. Li, and G. L. Stüber, Eds.,. Orthogonal Frequency Division Multiplexing for Wireless Communications., New York: Springer, 2006.

2.15 EFFECT OF FREQUENCY OFFSET If there is a frequency offset, then the number of cycles in the FFT interval is not an integer anymore, with the result that Intercarrier interference occurs after the FFT. The FFT output for each subcarrier will contain interfering terms from all other subcarriers, with an interference power that is inversely proportional to the frequency spacing [4]. The signal distortion caused by frequency offset is deterministic. Once the frequency offset is know, its effect can be corrected. The effect of frequency offset can be described as follows: The received signal due to frequency offset can be expressed as [3]:

(2.23)

Hk is the frequency response of the channel at the k-th subchannel. N is the number of subchannels in an OFDM symbol. k0 is an integer and represents the coarse frequency offset, |ε| <1/2 represents the fine frequency offset and

(2.24) In the next section, I would be describing about the synchronization achieved using cyclic prefix according to Richard Van Nee and Ramjee Prasad “OFDM for wireless Multimedia Communications” [4].

2.16 SYNCHRONIZATION USING THE CYCLIC PREFIX The cyclic prefix length of each OFDM symbol is always identical to the last part. This attribute can be exploited for both timing and frequency synchronization by using the synchronization scheme as seen in the Figure 2.16

Figure 2.16. Synchronization using cyclic prefix. Source: R. Van Nee, and R. Prasad, OFDM for Wireless Multimedia Communications. Norwood, MA: Artech House Publishers, 2000.

This scheme correlates a TG seconds (Cyclic prefix length) long part of the signal with a part that is T-seconds delayed. The correlator output can be written as [4]:

(2.25) Different OFDM symbols would contain independent data values; therefore the correlation output is always a random variable, which may reach a value that is larger than the desired correlation peak. The standard deviation of the random correlation magnitude is related to the number of independent samples over which the correlation is performed. This method of synchronization using cyclic prefix is effective when a large number of subcarriers are used, preferably 100 subcarriers or more. If we try to correlate 8 different OFDM symbols, then we should be seeing 8 peaks for the eight different symbols, but the peak amplitudes show a significant variation. The reason for that is although the average power for

a T-seconds interval of each OFDM symbol is constant; the power in the guard time can substantially vary from this average power level [4].

2.17 CHANNEL ESTIMATION In an OFDM link, modulated bits are distributed during the transmission through the channel since the channel introduces amplitude and phase shifts due to frequency selective and time-varying nature of the radio channel [8]. So in the receiver, the fast Fourier transform (FFT) is used to demodulate the N subcarriers of the OFDM signal. But the FFT output that contains N QAM values will have phase shifts, amplitude variations, local oscillator drift and timing offset [4]. So the channel estimation block has to learn the reference phases and amplitudes for all subcarriers, such that the QAM symbols can be converted to binary soft decision [4]. The receiver can apply either the coherent or non-coherent detection to recover the original bits. Coherent detection uses reference values that are transmitted along with data bits. The receiver can synchronize with this bits and estimate the channel only at located wherever these reference signals are located. Non-coherent detection on the other hand, does not use any reference values but uses differential modulation where the information is transmitted in difference of the two successive symbols [8]. Channel estimators need some kind of pilot information as a point of reference. A fading channel would require constant tracking, so pilot information has to be transmitted more or less continuously. The estimate of the channel at pilot sub-carriers based on LS (Least Square) is given by:

(2.26) Where Yp (k) and Xp (k) are output and input at the kth pilot sub-carrier respectively.

CHAPTER 3

DIGITAL VIDEO BROADCASTING- HANDHELD DESCRIPTION

The DVB Project is an Alliance of about 250-300 companies, originally of European origin but now worldwide. Its objective is to agree specifications for digital media delivery systems, including broadcasting. It is an open, private sector initiative with an annual membership fee, governed by a Memorandum of Understanding (MoU) [13]. Until late 1990, digital television broadcasting to the home was thought to be impractical and costly to implement. During 1991, broadcasters and consumer equipment manufacturers discussed how to form a concerted pan-European platform to develop digital terrestrial TV. Towards the end of that year, broadcasters, consumer electronics manufacturers and regulatory bodies came together to discuss the formation of a group that would oversee the development of digital television in Europe [13]. The Digital Video Broadcast (DVB) Project started research work related to mobile reception of DVB—Terrestrial (DVB-T) signals as early as 1998, accompanying the introduction of commercial terrestrial digital TV services in Europe [13]. In 2000, the EU-sponsored Motivate (Mobile Television and Innovative Receivers) project concluded that mobile reception of DVB-T is possible but it implies dedicated broadcast networks, as such mobile services are more demanding in robustness (i.e., constellation and coding rate) than broadcast networks planned for fixed DVB-T reception [13]. DVB-H is relatively new technology for the transmission of digital TV to handheld receivers such as mobile telephones and PDA’s. Published as a formal standard (EN 203 204) by ETSI in November 2004, it is a physical layer specification designed to enable the efficient delivery of IP-encapsulated data over terrestrial networks. The creation of DVB-H, which is closely related to DVB-T, also entailed modifications of some other DVB standards dealing with data broadcasting, Service Information, etc. It can be used as a bearer in conjunction with the DVB-IPDC systems layer specifications or alternatively with the OMA

BCAST specifications. A non-proprietary open standard, DVB-H has broad support across the industry and services are now on air in more than ten countries [14]. This chapter is according to the ETSI (European Telecommunications Standards Institute) standards ETSI TR 102 377 v1.3.1, ETSI TR 101 190 v1.2.1 and ETSI EN 300 744 v1.6.1 [15, 16, 17]. Although the DVB-T transmission system has proven its ability to serve fixed, portable and mobile terminals; handheld terminals (defined as light weight, battery powered apparatus) require specific features from the transmission system serving them, like: 1. It is beneficial that the transmission system offers the possibility to repeatedly turn the power off to some parts of the reception chain. This will reduce the average power consumption of the receiver. 2. Transmission system should make sure that; receivers can easily move from one transmission cell to another while maintaining the DVB-H service. 3. As DVB-H services are expected to be delivered in an environment suffering high levels of man-made noise, the transmission system shall offer the means to mitigate their effects on the receiving capabilities. It should be noted that, neither the time slicing nor MPE-FEC affect the physical layer in anyway, as they are implemented in the link layer. DVB-H system is completely backward compatible to DVB-T. A full DVB-H system is a combination of physical and link layer, as well as service information. DVB-H makes use of the following technological elements for the link and physical layers [17]: Link Layer: • Time Slicing in order to reduce the average power consumption of the receiving terminal and enable smooth and seamless frequency handover. Time slicing is mandatory for DVB-H • Forward error correction for multiprotocol encapsulated data (MPE-FEC) for an improvement in C/N performance and Doppler performance in mobile channels. Physical Layer: • DVB-H signaling in the TPS bits to enhance and speed up service discovery. A cell identifier is also carried in the TPS-bits to support quicker signal scan and frequency handover on mobile receivers. DVB-H signaling is not mandatory for DVB-H. • 4K mode for trading off mobility and SFN cell size, allowing single antenna reception in medium SFN’s at very high speed, adding flexibility for the network design. 4K mode is not mandatory for DVB-H.

• In-depth symbol interleaver for the 2K and 4K modes to further improve the robustness in mobile environments and impulse noise conditions.

3.1 SCOPE OF DVB-H STANDARD The conceptual structure of a DVB-H receiver is shown in Figure 3.1 [17]. It includes a DVB-H demodulator and a DVB-H terminal. The DVB-H demodulator includes a DVB-T demodulator (with optional 4K mode), a time slicing module and an optional MPE-FEC module.

Figure 3.1. Conceptual structure of a DVB-H receiver Source: Digital Video Broadcasting (DVB); DVB-H Implementation guidelines, ETSI TR 102 377 v1.3.1, April 2009.

The DVB-T demodulator recovers the MPEG-2 transport streams packets from the received DVB-T [15, 17] RF signal. The demodulator offers 3 transmission modes; 8K, 4K and 2K with the corresponding Transmitter Parameter Signalling (TPS). The time slicing module provided by DVB-H, aims to reduce the power consumption while also enabling a smooth and seamless frequency handover. MPE-FEC module, provided by DVB-H offers in addition to the physical layer transmission, a complementary forward error correction function that allows the receiver to cope with particularly difficult reception situations. Time slicing and MPE-FEC is dealt with greater detail in the next section. An example of using DVB-H for transmission of IP services is shown in the Figure 3.2.

Figure 3.2. A conceptual description of using a DVB-H transmitter and receiver (Sharing a MUX with MPEG-2 Services).

DVB-H can be operated in three network configurations according to [1]: 1. DVB-H shared network (sharing the MPEG-2 multiplex): In a DVB-H shared network the mobile TV channels after IPE (IP encapsulation) share the same DVB-T multiplex along with other terrestrial TV programs. The terrestrial TV programs would be coded in MPEG-2, while the mobile TV programs are in MPEG-4 coding and IPE. The multiplex combines these into a single transmit stream, which is then transmitted after modulation. 2. DVB-H hierarchical network (sharing DVB-T network by hierarchy): here the modulation is hierarchical with the two streams, DVB-T and DVB-H, which for a part of the same modulator output. 3. DVB-H dedicated network: the DVB-T carrier is used exclusively for DVB-H transmission. In a dedicated network, the COFDM carrier will be used exclusively by the mobile TV and audio channels as an IP datacast with the MPEG-2 envelope.

3.2 BASIC ASPECTS OF DVB-H NETWORKS The DVB-H network topology can be either single frequency network (SFN) or multifrequency network (MFN). The transport stream distribution and needed network infrastructure deviate between SFN and MFN deployments [2]. The principle and frequency resources for SFN and MFN are explained below.

3.3 PRINCIPLE OF MFN (MULTI FREQUENCY NETWORKS) When the area of required coverage is large (e.g. an entire country of several hundred kilometers), sourcing a signal from a single IPE is not practical due to time delays in delivering signals to all transmitters. In such a case, transmitters beyond a certain range use different frequencies. Conventionally planned DVB-H networks consist of transmitters with independent programme signal and with individual radio frequencies. Therefore they are referred to as Multi Frequency Networks (MFN). Whether a number of transmitters is considered to belong to a specific network is an administrative matter rather than a technical one. In order to cover large areas with one DVB-T/H signal a certain number of radio-frequency channels are needed. The number of channels depends on the robustness of the transmission, i.e. the type of modulation associated with the applied channel code rate and on the objective of planning [16]. Frequency resources needed for MFN: Robustness of the broadcasting system is generally expressed in terms of protection ratios; one might expect that the number of channels needed for DVB-T/H is significantly lower than for analogue broadcasting as the protection ratios are generally lower in the digital case. The number of radio-frequency channels needed for conventionally planned DVB-H networks tends to be in the same order as with analogue TV systems. The frequency resource expressed as the number of channels needed to provide one signal at any location is far higher with MFN than with Single Frequency Networks (SFN). The transmitters in a MFN have not to obey rules of synchronous emissions. Hence co- ordination between transmitter operators is absolutely not necessary. The installation of local or regional services is easy with the MFN concept compared to the SFN concept.

3.4 PRINCIPLE OF SFN (SINGLE FREQUENCY NETWORK) In an SFN, all transmitters are synchronously modulated with the same signal and radiate on the same frequency. Due to the multipath capability of the multi-carrier transmission system (COFDM) signals from several transmitters arriving at receiving antenna may contribute constructively to the total wanted signal.

One of the limiting effects of SFN is the so-called self interference of the network. If signals from distant transmitters are delayed more than allowed by the guard interval they behave as noise-like interfering signals rather than as wanted signals. As an empirical rule, to successfully reduce self interference to an acceptable value the guard interval time should allow a radio signal to propagate over the distance between two transmitters of the network. SFN is also requires minimal network infrastructure, since only one IPE (IP encapsulator) is needed to provide transport streams for the entire network. Frequency Efficiency of SFN: With the SFN technique, large areas can be served with a common multiplex at a common radio centre frequency. Therefore the frequency efficiency of SFNs appears to be very high compared to MFNs. However, taking into account the presence of similar networks offering the other programme multiplexes in adjacent areas, further radio frequency channels are required. Figure 3.3 [18] shows an example of the Single Frequency Network.

Figure 3.3. Single frequency network. Source: DigiTAG. (2007). Television on a handheld receiver - broadcasting with DVB-H [Online]. Available: http://www.digitag.org/DTTResources/DVBHandbook.pdf

The core features of the DVB-H are Time slicing and Multiprotocol Encapsulation Forward Error Correction (MPE-FEC). The time slicing decreases the power consumption, while the MPE-FEC improves the robustness of the data transmission. In the next section, Time Slicing and MPE-FEC are described below.

3.5 TIME SLICING Because we have many services in one multiplex, we can use time-division multiplex (time slicing).This means that instead of laying one service on top of another and transmit them simultaneously; the services are all located in the same small band and are transmitted on a rotary base (see Figure 3.4 [19] for an illustration). So once the receiver knows that

Figure 3.4. Visualization of time slicing. Source: M. Kornfeld, and U. Reimers, (2005). DVB-H: The emerging standard for mobile data communication [Online]. Available: http://www.ebu.ch/en/technical/ trev/trev_301-dvb-h.pdf every x seconds his service is transmitted it can go in stand-by in the time other services are transmitted. Time slicing requires that in the relatively short time, a service can occupy the channel, there is enough information transmitted to provide a video picture for the time when the receiver is in stand-by. In the time the receiver is in power-Save mode it can as well scan for other channels in neighboring cells that provide the same service and guarantee a smooth change. This becomes important in DVB-H, because you don’t want to lose the signal when you move around with your mobile phone. You can see in the above Figure that, a single DVB-H multiplex can carry six to eight channels, which were earlier occupying one frequency slot each.

3.6 MPE-FEC The multiprotocol encapsulation was originally defined for the purpose of DVB data broadcasting in [20], as generic section protocol for different DVB systems. The objective of the MPE-FEC (Multiprotocol Encapsulation Forward Error Correction) is to improve the C/N and Doppler performance in mobile channels where there is high packet loss ratio (PLR) and to improve the tolerance to impulse interference. High packet loss ratio typically occurs on mobile channels when the receiver is in a high speed and/or the C/N is too low. MPE-FEC is an enhancement which is not mandatory to be used by the receiver, even if it is transmitted by the network. Hence, the receiver may ignore MPE-FEC and still be able to receive the application data carried within the MPE sections. The MPE-FEC overhead can be fully compensated by choosing a slightly weaker transmission code rate, while still providing far better performance than DVB-T (Without MPE-FEC) for the same throughput. This MPE-FEC scheme should allow high speed single antenna DVB-T reception using 8K/16-QAM or even 8K/64-QAM signals. In addition MPE- FEC provides good immunity to impulse noise interference [17]. Figure 3.5 shows the overall structure of the MPE-FEC frame.

Figure 3.5. Structure of MPE-FEC frame. Source: J. T. J. Penttinen, P. Jolma, E. Aaltonen, and J. Vare., The DVB-H Handbook, The Functioning and Planning of Mobile TV. Chichester, UK: Wiley, 2009.

MPE-FEC frame is a combination of application data and related parity information of the used FEC code. The MPE-FEC frame is a matrix composed of 255 columns and from 256 upto 1024 rows as shown in the Figure 3.4. The maximum size of the MPE-FEC frame is

approximately 2 Mbits. It has been split into two parts dedicated to IP datagrams, padding, and parity information of the FEC code, i.e. RS (Reed Solomon) data. The IP datagrams and padding are always allocated in the left side of the frame while the RS data is allocated to the right. We can see in the Figure 3.4 that left part of the MPE-FEC frame consists of 191 rows and is called application data table. Right part consists of 64 rows and is called RS data table [2].

3.7 MAIN ISSUES IN DVB-H 1. Power Consumption: According to [17], the DVB project estimated the future power consumption of DVB-T implementations. The estimation for a mobile handheld terminal was that the power consumption of the RF and baseband processing may come down to 600 mW by the year of 2007. However, the average power consumption of any additional receiver in a mobile handheld terminal should be less than 100 mW. This is required due to the limited battery capacity and to the extremely challenging heat dissipation in a very small environment. In the future, when merging the DVB-H receiver into a mobile handheld terminal, the required reduction in power consumption may become as high as 90%. 2. Handover: For a mobile reception in a DVB-T Multi Frequency Network, there is normally the need to handover to another frequency when the reception quality of the present frequency becomes too low. Since DVB-T does not include seamless handover facilities, changing frequency normally results in a service interruption. In addition to this the receiver will have to scan possible alternative frequencies to find out which of these provides the best or atleast sufficient reception quality. Each time a frequency is scanned there will be an interruption, unless the receiver is equipped with an extra RF part dedicated for this purpose. The extra RF part would increase the cost of Receivers [17]. Hence there is a requirement to allow for seamless handover and seamless scanning of alternative frequencies without having to include an additional RF part.

3.8 HOW TIME SLICING AND MPE-FEC PROVIDES A SOLUTION Services used in mobile handheld terminals require relatively low bitrates. The estimated maximum bitrate for streaming video using advanced compression technology like MPEG-4 is in the order of few hundred kilobits per second (Kb/s), one practical limit being 384 Kb/s coming from the 3G standard. Some other types of services, such as file downloading may require significantly higher bitrates, though.

A DVB transmission system usually provides a bitrate of 10 Mb/s or more. This provides a possibility to significantly reduce the average power consumption of a DVB receiver by introducing a scheme based on Time Division Multiplexing (TDM). This technique is called time slicing. The idea of time slicing is to send the data in bursts at a higher bitrate compared to the bitrate required if the data was transmitted continuously. Within a burst, the time to the beginning of the next burst (delta-t) is indicated. Between the duration of bursts, elementary stream data is not transmitted, this allows other elementary streams to use the bitrate otherwise allocated. This enables the receiver to stay active for only a fraction of time, while receiving bursts of a requested service. Hence to get a reasonable power saving effect, the Burst Bitrates should be atleast 10 times the constant bitrate of the delivered service [17]. Time Slicing supports the possibility of using the Receiver to monitor neighboring cells during the Off-times. By accomplishing the switching between transport streams during an off period, the reception of a service is seemingly uninterrupted and thus handover is supported. The MPE-FEC also provides good immunity to impulsive interference. With MPE- FEC, reception is fully immune to repetitive impulsive noise causing a destruction of OFDM symbols if the distance between the destroyed symbols is in the range 6ms to 24ms. This depends on the chosen DVB-T mode. MPE-FEC is introduced in such a way that MPE-FEC ignorant DVB-T receivers will be able to receive IP stream in a fully backwards-compatible way. This backwards compatibility holds when the MPE-FEC is used with and without time slicing.

3.9 DELTA-T METHOD The basic goal of Delta-t method in DVB-H is to signal the time from the start of the MPE (or MPE-FEC) section, currently being received, to the start of the next burst within the elementary stream. The DVB-H standard also defines that delta-t equal to zero means “End of service” [17]. That is, no bursts related to the service are sent any more. Figure 3.6 shows the time delta-t contained in each MPE section header.

Figure 3.6. Each MPE section header contains delta-t indicating time to the beginning of the next burst Source: Digital Video Broadcasting (DVB); DVB-H Implementation guidelines, ETSI TR 102 377 v1.3.1, April 2009.

Delivering delta-t in MPE (or MPE-FEC) sections removes the need to synchronize clocks between transmitter and receiver. The resolution of delta-t is 10 ms. the delta-t actually signals the earliest possible time when the next burst may start. In bad reception conditions, parts of a burst may be lost. If the delta-t information is lost, the Receiver would not know the time to the next burst and therefore is forced to stay on waiting for the next burst. To avoid this situation, delta-t is delivered in the header of each MPE-section and MPE-FEC section within a burst. Accuracy of delta-t has an effect on the achieved power saving.

3.10 BURST SIZE AND OFF-TIME The size of a burst has to be less than the memory available in a Receiver. When a burst is received, a Receiver has to buffer the data within its memory, to be consumed during the time between bursts. Streaming services may require bigger buffering, even if time slicing is not used. Figure 3.7 shows the burst parameters used for DVB-H systems. Burst Size refers to the number of network layer bits within a burst. Network layer consist of section payload bits. Each MPE and MPE-FEC section contains 16 bytes overhead caused by the header and CRC-32. Burst Bitrate is the bitrate used by a Time-sliced elementary stream while transmitting a burst. Constant Bitrate is the average bitrate required

Figure 3.7. Burst parameters Source: Digital Video Broadcasting (DVB); DVB-H Implementation guidelines, ETSI TR 102 377 v1.3.1, April 2009.

by the elementary stream when not Time-sliced. Both constant and Burst Bitrates include transmission of transport_packets (188 bytes). For e.g. Burst size of 1Mb and a Burst Bitrates of 1 Mb/s, the Burst Duration is 1.04 seconds (due to 4% overhead) [17]. Off time is the time between bursts. During Off-time, no transport_packets are delivered on the relevant elementary stream. Formulas to calculate the length of a burst, off-time and the achieved saving on power consumption according to [17] is:

Bd = Bs / (Bb x 0.96) (3.1)

Ot= Bs / (Cb x 0.96) - Bd (3.2)

Ps = (1- ((Bd + St +(3/4 x Dj )) x Cb x 0.96)/ Bs ) x 100% (3.3) Where:

Bd = Burst Duration (seconds)

Bs = Burst Size (bits) Bb = Burst Bitrate (bits per second)

Cb = Constant Bitrate (bits per second)

Ot = Off-time (seconds)

St = Synchronization Time (seconds)

Ps = Power Saving (per cent)

Dj = Delta-t jitter (seconds)

3.11 PHYSICAL LAYER SPECIFICATIONS FOR DVB-H DVB-H includes a new transmission mode in the DVB-T physical layer using a 4096 FFT size: 4K mode. This mode brings in an additional flexibility in network design by trading off mobile reception performance and size of SFN networks. The proposed 4K mode is also architecturally compatible with existing DVB-T infrastructure, requiring only minor changes in the modulator and demodulator [17]. Figure 3.8 shows the blocks in DVB-T system, which are affected by the addition of the 4K mode. DVB-H is principally a transmission system allowing reception of broadcast information of single antenna hand-held mobile devices. In the DVB-T system, the 2K transmission mode is know to provide significantly better mobile reception performance than the 8K mode, due to the larger inter-carrier spacing it implements. However, the duration of the 2K mode OFDM symbols and consequently the guard interval duration are relatively short. Hence 2K mode is more suitable for small size SFNs making difficult for network designers to build spectrally efficient networks. Table 3.1 shows the guard interval lengths for all modes and it can be seen that 4K OFDM symbol has longer guard interval duration than a 2K OFDM symbol, allowing building medium size SFN networks.

Figure 3.8. Functional block diagram of the DVB-H transmission system Source: Digital Video Broadcasting (DVB); Framing Structure, channel coding and modulation for digital terrestrial television, ETSI EN 300 744 v1.6.1, 2008-2009.

Table 3.1. Guard Interval Lengths for all Modes

3.12 TRANSMITTER INPUT SIGNAL We know that DVB-H system is defined based on the existing DVB-T standard for fixed and in-car reception of digital TV. As mentioned in the previous section, the terrestrial TV programs would be coded in MPEG-2, while the mobile TV programs are in MPEG-4 coding and IPE or any other high efficiency video coding standards. First, I would be explaining about the MPEG-2 Transport streams and then the advanced video coding standard, H.264 which is basically used for delivering DVB services directly over IP protocols i.e. DVB-H with Multi-protocol encapsulation without the use of an intermediate MPEG-2 Transport stream. The transmitter input signal is specified as an MPEG-2 TS (Transport Stream) multiplex, which may contain several TV programmes and also possible some sound/data only programmes. A programme, in MPEG terms, is a single broadcast service such as BBC- 2 or RAI Uno, not an individual TV programme as broadcasters normally use the word. A programme comprises one or more Packetized Elementary Streams (PES), each containing a single digitally coded component of the programme, for example coded video or coded audio; it will also contain time stamps to ensure that specified elementary streams are replayed in synchronism at a decoder. The TS (Transport Stream) was devised for multi-programme application in error- prone channels such as broadcasting. It comprises a succession of packets, each 188 ocets long, called Transport Packets. The bit rate of the TS is determined by the application. Figure 3.9 [21] shows the structure of a MPEG-2 Transport Stream. As it can be seen in the Figure 3.9, the Transport Stream takes variable length Packetized Elementary Stream (PES) and chops it up into a fixed length. Also, the TS allows the multiplexing of many PES.

Figure 3.9. Structure of MPEG-2 Transport Stream (TS). Source: C. M. Lynarczyk. (n.d) MPEG-2: The basics of how it works [Online]. Available: http://www. home.agilent.com/upload/cmc_upload/All/6C06MPEGTUTORIAL1.pdf

The header contains the following fields: • The header of the MPEG-2 TS starts with a Synchronization Byte (8 bits). It has the bit pattern 0x47. • A set of three flag bits are used to indicate how the payload should be processed. • • Transport error indicator: indicates the packet is errored (block error testing) • PID: is the channel identifier, it contains all the navigation information required to find, identify and reconstruct programmes. It is used to determine the stream to which the packet belongs to. • PCR, the programme clock reference: provides 27 MHz clock recovery information. • Two Scrambling control bits are used by conditional access procedures to process the encrypted payloads of some TS packets. • The adaptation field indicates what kind of data the payload contains. The two adaptation control bits can take four values. • 01- No adaptation field, payload only • 10- Adaptation field only, no payload • 11- Adaptation field followed by the payload

• 00- Reserved for future use • There is a 4 bit continuity counter.

3.13 CHANNEL CODING The system input stream shall be organized in fixed length packets, following the MPEG-2 transport multiplexer. The total packet length of the MPEG-2 transport multiplex (MUX) packet is 188 bytes. This includes 1 sync-word byte (i.e 47HEX ). The processing order at the transmitting side shall always start from the MSB (i.e “0”) of the sync-word byte (i.e 01 000 111). In order to ensure adequate binary transitions, the data of the input MPEG-2 multiplex shall be randomized in accordance with the configurations depicted in the Figure 3.10.

Figure 3.10. Scrambler/descrambler.

The polynomial for the Pseudo Random Binary Squence (PRBS) generator shall be: 1+ X14 + X15 Loading of the sequence “100101010000000” into the PRBS registers, as shown in the Figure 3.10, shall be initiated at the start of every eight transport packets. To provide an initialization signal for the descrambler, the MPEG-2 sync byte of the first transport packet in a group of eight packets is bit-wise inverted from 47HEX (SYNC) to B8HEX . This process is referred to as “Transport Multiplex Adaptation”. It is shown in the Figure 3.11.

Figure 3.11. Randomized transport packets: sync and randomized data bytes.

The first bit at the output of the PRBS generator shall be applied to the first bit (i.e. MSB) of the first byte following the inverted MPEG-2 sync byte (i.e. B8HEX ). To aid other synchronization functions, during the MPEG-2 sync bytes of the subsequent 7 transport packets, the PRBS generation shall continue, but its output shall be disabled, leaving these bytes unrandomized. Thus, the period of the PRBS sequence shall be 1503 bytes.

3.14 OUTER CODING AND OUTER INTERLEAVING DVB uses two levels of channel coding, an outer coding and an inner coding. The outer coding is done using Reed Solomon Code, also called as RS code. RS codes are block based error correcting codes with a wide range of applications in digital communications and storage. The inner coding is done using convolutional coding. These codes are fundamentally different from RS codes in that information sequences are not grouped into distinct blocks and encoded. Instead a continuous sequence of information bits is mapped into a continuous sequence of encoder output bits. This mapping is highly structured, enabling a decoding method considerably different from that of RS code [13]. The outer coding and interleaving shall be performed on the input packet structure as seen in Figure 3.12. Reed-Solomon RS (204,188,t=8) shortened code, derived from the original systematic RS (255,239,t=8) code, shall be applied to each randomized transport packet (1888 byte) of Figure 3.12 to generate an error protected packet, seen in Figure 3.13. Reed-Solomon coding shall also be applied to the packet sync byte, either non-inverted (i.e. 47 HEX ) or inverted (i.e. B8HEX ). The Reed-Solomon code has length 204 byted, dimension 188 bytes and allows a correct up to 8 random erroneous bytes in a received word of 204 bytes.

Figure 3.12. MPEG-2 transport MUX packet.

Figure 3.13. Reed Solomon RS (204,188,8) error protected packets.

Code generator polynomial: g(x) = (x+λ0) (x+λ1) (x+λ2)….. (x+λ15), where λ=02HEX Field generator polynomial: p(x) = x8 +x4 +x3+x2+1 The shortened Reed-Solomon code may be implemented by adding 51 bytes, all set to zero, before the information bytes at the input of an RS (255,239,t=8) encoder. After the RS coding procedure, these null bytes shall be discarder, leading to a RS code word of N =204 bytes. Following the conceptual scheme of Figure 3.14, convolutional byte-wise interleaving with depth I=12 shall be applied to the error pprotected packets. This results in the interleaved data structure as seen in the Figure 3.15. The interleaver may be composed of I=12 branches, cyclically connected to the input byte-stream by the input switch. Each branch j shall be a First-In, First-Out (FIFO) shift register, with depth j x M Cells where M=17=N/I, N=204. The cells of the FIFO shall contain 1 byte, and the input and output switches shall be synchronized.

Figure 3.14. Conceptual diagram of the outer interleaver and deinterleaver.

Figure 3.15. Data structure after outer interleaving with interleaving depth as 12 bytes.

3.15 INNER CODING The system shall allow for a range of punctured convolutional codes, based on a mother convolutinal code of rate ½ with 64 states. This will allow selection of the most appropriate level of error correction for a given service or data rate in either non-hierarchical or hierarchical transmission mode. The generator polynomials of the mother code are G1=171OCT for X output and G2 =133OCT for Y output. This can be seen in the Figure 3.16.

Figure 3.16. The mother convolutional code of rate ½

If two level hierarchical transmissions are used, each of the two parallel channel encoders can have its own code rate. In addition to the mother code of rate 1/2 the system shall allow punctured rates of 2/3, 3/4, 5/6 and 7/8

3.16 INNER INTERLEAVING The inner interleaving consists of bit-wise interleaving followed by symbol interleaving. Both the bit-wise interleaving and the symbol interleaving processes are block- based. Figure 3.17 gives a block view of the inner coding and interleaving. When the in-depth interleaver is used, it shall be signaled within the Transmission Parameter Signalling bits which would be discussed in the upcoming section. The In-depth inner interleaver for 2K and 4K modes is shown in the Figure 3.18.

3.17 BIT-WISE INTERLEAVING The input, which consists of up to two bit streams, is demultiplexed into v sub- streams, where v=2 for QPSK, v=4 for 16-QAM and v=6 for 64-QAM. In non-hierarchical

Figure 3.17. Inner coding and interleaving.

Figure 3.18. In-depth inner interleaver for 2K and 4K modes. mode, the single input stream is demultiplexed into v sub-stream. In hierarchical mode the high priority stream is demultiplexed into two sub-streams and the low priority stream is demultiplexed into v-2 sub-streams. This applies in both uniform and non-uniform QAM modes. The demultiplexing is defined as a mapping of the input bits, xdi onto the output bits be,do In non-hierarchical mode: xdi = b[di(mod)v](div)(v/2)+2[di(mod)(v/2)],di(div)v In hierarchical mode:

x'di = bdi(mod)2,di(div)2 x"di = b[di(mod)(v-2)](div)((v-2)/2)+2[di(mod)((v-2)/2)]+2,di(div)(v-2) Where:

xdi is the input to the demultiplexer in non-hierarchical mode; x'di is the high priority input to the demultiplexer;

x"di is the low priority input, in hierarchical mode; di is the input bit number; be,do is the output from the demultiplexer; e is the demultiplexed bit stream number (0 ≤ e < v); do is the bit number of a given stream at the output of the demultiplexer; mod is the integer modulo operator; div is the integer division operator. The demultiplexing results in the following mapping:

QPSK: x0 maps to b0,0

x1 maps to b1,0 16-QAM non-hierarchical transmission: 16-QAM hierarchical transmission: x0 maps to b0,0 x'0 maps to b0,0

x1 maps to b2,0 x'1 maps to b1,0

x2 maps to b1,0 x"0 maps to b2,0

x3 maps to b3,0 x"1 maps to b3,0 64-QAM non-hierarchical transmission: 64-QAM hierarchical transmission: x0 maps to b0,0 x'0 maps to b0,0

x1 maps to b2,0 x'1 maps to b1,0

x2 maps to b4,0 x"0 maps to b2,0

x3 maps to b1,0 x"1 maps to b4,0

x4 maps to b3,0 x"2 maps to b3,0

x5 maps to b5,0 x"3 maps to b5,0 Figure 3.19 shows the mapping of input bits for the non –hierarchical transmission modes. Bit interleaving is performed only on the useful data. The block size is the same for each interleaver, but the interleaving sequence is different in each case. The bit interleaving

Figure 3.19. Mapping of input bits onto output modulation symbols, for non- hierarchical transmission modes. block size is 126 bits. The block interleaving process is repeated twenty-four times per OFDM symbol in the 4K mode. When the in-depth interleaving is applied in the 2K or 4K modes, either hierarchical or non-hierarchical, the block interleaving process is repeated forty-eight times, thus providing the symbol interleaver with the blocks of useful data needed to produce four consecutive “2K OFDM symbols” and two consecutive “4K OFDM symbols”.

3.18 SYMBOL INTERLEAVER The purpose of the symbol interleaver is to map v bit words on to the 3024 active carriers per OFDM symbol. When the native 4K mode interleaver is implemented, the

symbol interleaver acts on blocks of 3024 data symbols. Thus in the 4K mode, 24 groups of 126 data words from the bit interleaver are read sequentially into a vector Y' = (y'0, y'1, y'2, ...y'3 023). The interleaved vector Y= (y0,y1,y2,…..yNmax-1) is defined by: yH(q) = y'q for even symbols for q=0,…..,Nmax-1 yq = y'H(q) for odd symbols for q=0,…..,Nmax-1 Where, in case of native interleaver, Nmax = 3024 in the 4K mode.

3.19 MODULATION Modulation schemes that can be used in DVB-H are QPSK, 16-QAM and 64-QAM. QPSK (Quadrature Phase Shift Keying) has 2 bits/symbol. 16-QAM (16-state Quadrature Amplitude Modulation) has 4 bits/symbol and 64-QAM has 6 bits/symbol. In general, 16- QAM and QPSK are the most functional modes in DVB-H environment, 64-QAM being very sensitive for the interferences [2]. For mobile and portable reception the most usable modulation scheme is 16QAM with the code rate of 1/ 2 or 2/3, which requires moderate C/N and provides sufficient transmission capacity for DVB-H services [17].

3.20 4K MODE IN DVB-H The 4K (4096 point FFT) is a new addition to DVB-H, and is meant to be a compromise between 2K and 8K FFT mode. The FFT size determines the subcarrier spacing in the OFDM signal. The objective of the 4K mode is to improve the network planning flexibility by trading off mobility and SFN size. It aims to offer an additional trade-off between single frequency network (SFN) cell size and mobile reception performance, providing an additional degree of network planning. • It allows single antenna reception in medium SFN’s at very high speed, adding flexibility for the network design. • Mobility is increased by a factor of two when compared to 8K on the other hand; the 8K FFT mode supports slow moving terminals and large SFNs. • With symbol duration shorter than in the 8K mode, channel estimation can be done more frequently in the demodulator, thereby providing a mobile reception performance which, although not as high as with the 2K transmission mode, is nevertheless adequate for the use of DVB-H scenarios.

• Doubling the subcarrier spacing with respect to the 8K mode, allows for mobile reception with reasonably low complexity channel estimators, thus minimizing both power consumption and cost of the DVB-H receiver. • 4K mode OFDM symbol has a longer duration and consequently a longer guard interval than a 2K mode OFDM symbol, allowing building medium size SFN networks, with respect to spectral efficiency. Terms of trade off according to [19] can be expressed as follows: • The DVB-T 8K mode can be used both for single transmitter operation and for small, medium and large SFNs. It provides a Doppler tolerance allowing high speed reception. • The DVB-T 4K mode can be used both for single transmitter operation and for small and medium SFNs. It provides a Doppler tolerance allowing very high speed reception. • The DVB-T 2K mode is suitable for single transmitter operation and for small SFNs with limited transmitter distances. It provides a Doppler tolerance allowing extremely high speed reception.

3.21 OFDM FRAME STRUCTURE The transmitted signal is organized in frames. Each frame has duration of TF, and consists of 68 OFDM symbols. Four frames constitute one super-frame. Each symbol is constituted by a set of K= 3409 carriers and transmitted with a duration Ts. It is composed of two parts: a useful part with duration Tu and a guard interval with a duration Δ. The guard interval consists in a cyclic continuation of the useful part, Tu, and is inserted before it. Four values of guard intervals may be used according to Table 3.3 (p. 48). For the 4K mode, the carriers are indexed by k Є [Kmin; Kmax] and determined by Kmin = 0 and Kmax =3408. The symbols in an OFDM frame are numbered from 0 to67. All symbols contain data and reference information. Since the OFDM signal comprises many separately modulated carriers, each symbol can in turn be considered to be divided into cells, each corresponding to the modulation carried on one carrier during one symbol. In addition to the transmitted data an OFDM frame contains: Scattered pilot cells, continual pilot cells and TPS (Transmission Parameter Signalling) carriers. The pilots can be used for frame synchronization, frequency synchronization, time synchronization, channel estimation, transmission mode identification and can alos be used to follow the phase noise.

The values of time related parameters are given in multiples of the elementary period T and in microseconds. The elementary period T is 7/64 µs for 8MHz channels, 1/8 µs for 7 MHz channels, 7/48 µs for 6 MHz channels and 7/40 µs for 5 MHz channels. For the 4K mode, the numerical values for the OFDM parameters in 8MHz, 7 MHz and 6 MHz channels are given in Table 3.2 and 3.3.

Table 3.2. Frequency Domain Parameters for 4K Mode in 8 MHz, 7MHz and 6 MHz Channels

Table 3.3. Time Domain Parameters for 4K Mode in 8MHz, 7MHz and 6MHz Channels

The emitted signal is described by the following expressions:

where:

k denotes the carrier number; l denotes the OFDM symbol number; m denotes the transmission frame number; K is the number of transmitted carriers;

TS is the symbol duration;

TU is the inverse of the carrier spacing; ∆ is the duration of the guard interval;

fc is the central frequency of the RF signal; k' is the carrier index relative to the centre frequency, k' = k - (Kmax+ Kmin) / 2;

cm,0,k complex symbol for carrier k of the Data symbol no. 1 in frame number m;

cm,1,k complex symbol for carrier k of the Data symbol no. 2 in frame number m;

cm,67,k complex symbol for carrier k of the Data symbol no. 68 in frame number m.

3.22 REFERENCE SIGNALS Various cells within the OFDM frme are modulated with reference information whose transmitted value is known to the receiver. Cells containing reference information are transmitted at boosted power level. The information transmitted in these cells are scattered or continual pilot cells. Each continual pilot coincides with a scattered pilot every fourth symbol; number of useful data carriers is constant from symbol to symbol: 3024 useful carriers in 4K mode. The value of the scattered and continual pilot information is derived from a Pseudo- Random Binary Sequence (PRBS) which is a series of values, one for each of the transmitted carriers.

3.23 DEFINITION OF REFERENCE SEQUENCE The continual and scattered pilots are modulated according to a PRBS sequence, wk, corresponding to their respective carrier index k. This sequence also governs the starting phase of the TPS information. The PRBS sequence is generated according to the Figure 3.20. The PRBS is initialized so that the first output bit from the PRBS coincides with the first active carrier. A new value is generated by the PRBS on every used carrier (whether or not it is a pilot). The polynomial for the Pseudo Random Binary Sequence (PRBS) generator shall be: X11 + X2+1

Figure 3.20. Generation of PRBS sequence.

3.24 LOCATION OF SCATTERED PILOTS Reference information, taken from the reference sequence, is transmitted in scattered pilot cells in every symbol. Scattered pilot cells are always transmitted at the "boosted" power level. Thus the corresponding modulation is given by:

Re {cm,l,k} = 4 / 3 × 2 (1/2 - wk)

Im {cm,l,k} = 0 Where m is the frame index, k is the frequency index of the carriers and l is the time index of the symbols. For the symbol of index l (ranging from 0 to 67), carriers for which index k belongs to the subset {k=Kmin + 3 x (1 mod 4) + 12p | p integer, p ≥ 0, k [Kmin; Kmax] } are scattered pilots. Where p is an integer that takes all possible values greater∈ than or equal to zero, provided that the resulting value for k does not exceed the valid range [Kmin; Kmax]. Pilot insertion pattern is shown in the Figure 3.21.

Figure 3.21. Frame structure.

3.25 LOCATION OF CONTINUAL PILOT CARRIERS For the 4K mode, 89 continual pilots ( where “continual” means that they occur on all symbols) shall be inserted according to the Table 3.4.

Table 3.4. Carrier Indices for Continual Pilot Carriers.

The continual pilots are transmitted at “boosted” power level. Thus the corresponding modulation is given by:

Re {cm,l,k} = 4 / 3 × 2 (1/2 - wk)

Im {cm,l,k} = 0

3.26 TRANSMISSION PARAMETER SIGNALLING (TPS) The TPS carriers are used for the purpose of signalling parameters related to the transmission scheme, i.e. to channel coding and modulation. The TPS shall be transmitted in parallel on 34 TPS carriers and shall be carried on the carrier having the indices presented in the Table 3.5. Every TPS carrier in the same symbol conveys the same differentially encoded information bit.

Table 3.5. Carrier Indices for TPS Carriers in 4K Mode

The TPS carriers convey information on: 1. Modulation including the α value of the QAM constellation pattern. 2. Hierarchy information; 3. Guard interval (not for initial acquisition but for supporting initial response of the receiver in case of reconfiguration); 4. Inner code rates; 5. Transmission mode

6. Frame number in a super-frame; 7. Cell identification. The α value defines the modulation based on the cloud spacing of a generalized QAM constellation. It allows specification of uniform and non-uniform modulation schemes, covering QPSK, 16-QAM and 64-QAM.

3.27 SCOPE OF THE TPS The TPS is defined over 68 consecutive OFDM symbols, referred to as one OFDM frame. Four consecutive frames correspond to one OFDM super-frame. The reference sequence corresponding to the TPS carriers of the first symbol of each OFDM frame are used to initialize the TPS modulation on each TPS carrier. Each OFDM symbol conveys one TPS bit. Each TPS block (corresponding to one OFDM frame) contains 68 bits, defined as follows: • 1 initialization bit; • 16 synchronization bits; • 37 information bits; • 14 redundancy bits for error protection. When DVB-H signalling is performed, of the 37 information bits, 33 are used. The remaining 4 bits shall be set to zero.

3.28 TPS TRANSMISSION FORMAT The transmission parameter information shall be transmitted. The mapping of each of the transmission parameters: constellation characteristics, α value, code rate(s), super –frame indicator and guard interval onto the bit combinations is performed. The left most bit is sent first. TPS length indicator: When DVB-H signalling is in use, valid cell identification information shall be transmitted and the value of the TPS length indicator shall be set to “100001” (33 TPS bits in use) Hierarchy and Interleaving information: Bits S27, S28 and S29 shall be used to signal if the in-depth interleaver is in use and if the transmission is hierarchical. The use of the in- depth interleaver for 2K or 4K transmission mode shall be signaled using bit S27 as indicated in the Table 3.6. When an 8K signal is transmitted only the native interleaver shall be used.

Table 3.6. Signalling Format for In-Depth Inner Interleaver

If hierarchical transmission is used, the value of the α factor shall be signalled, using the bits S28 and S29 in accordance with the Table 3.7.

Table 3.7. Signalling Format for Hierarchy Information

Transmission mode: The transmission mode shall be signaled according to the Table 3.8.

Table 3.8. Signalling Format for Transmission Mode

DVB-H signalling: Bits S48 and S49 shall be used to indicate to the receivers the transmission of DVB-H services in compliance with Table 3.9. In case of Hierarchical transmission, the significance of bits S48 and S49 varies with the parity of the OFDM frame transmitted, as follows: • When received during OFDM frame number 1 and 3 of each super frame, DVB- H signalling shall be interpreted as in relation with the High Priority stream (HP) and in compliance with Table 3.9. • When received during OFDM frame number 2 and 4 of each super frame, DVB- H signalling shall be interpreted in relation with the Low Priority stream (LP) in compliance with Table 3.9.

Table 3.9. DVB-H Service Indication

In case of non-Hierarchical transmission, every frame in the super-frame carriers the same information, which shall be interpreted in compliance with Table 3.9.

3.29 NUMBER OF RS-PACKET PER OFDM SUPER- FRAME The OFDM frame structure allows for an integer number of Reed-Solomon 204 byte packets to be transmitted in an OFDM super-frame, and therefore avoids the need for any stuffing, whatever the constellation, the guard interval length, the coding rate or the channel bandwidth may be. Table 3.10 shows the number of RS byte packets per OFDM super frame. Also in this table, we can see for all combination of code rates and modulation forms.

Table 3.10. Number of Reed-Solomon 204 Bytes Packets per OFDM Super-Frame for all Combinations of Code Rates and Modulation Forms

3.30 USEFUL BITRATE Useful bitrates are given for the 4K mode in the following tables and cover the 8 MHz, 7 MHz and 6 MHz channels. In the following tables, the values in italics are approximate values for the given channel bandwidth. Table 3.11 shows the useful bitrates for non-hierarchical systems operating in 8 MHz channels. Table 3.12 and table 3.13 gives the useful bitrates information for the 7 and 6 MHz channels. For the hierarchical transmission schemes the useful bit rates can be obtained as follows: • HP stream: values from QPSK columns; • LP stream, 16-QAM: figures from QPSK columns; • LP stream, 64-QAM: figures from 16-QAM columns.

Table 3.11. Useful Bitrate (Mbit/s) for Non-Hierarchical Systems in 8 MHz Channels

Table 3.12. Useful Bitrate (Mbit/s) for Non-Hierarchical Systems in 7 MHz Channels

Table 3.13. Useful Bitrate (Mbit/s) for Non-Hierarchical Systems in 6 MHz Channels

CHAPTER 4

IMPLEMENTATION AND SIMULATION OF THE PHYSICAL LAYER FOR A DVB-H SYSTEM

The main purpose of the thesis is to implement and simulate the physical layer for the DVB-H system. The thesis also describes a method to synchronize the receiver for the broadcasting system; DVB-H (Digital Video Broadcasting – Handheld). This chapter describes the simulations run and the results. The system implemented operates in 4K mode, where the size of the IFFT/FFT block is 4096. Guard interval considered is 1/4 the size of IFFT/FFT block i.e. 1024. If we are looking into the time domain parameters, with the elementary period T is 7/64µs for 8 MHz channel, then the duration of the useful symbol part Tu is 448 µs with a guard interval of duration 112 µs. Hence, each OFDM symbol is of duration 560 µs. I have simulated the system with 10 OFDM symbols with each symbol having 3409 modulated carriers. The input data is assumed to be MPEG encoded, channel coded (outer and inner) and interleaved (outer and inner). Random data packets of length 3409 are generated and are 16- QAM modulated. The scattered pilots are generated by the PRBS (Pseudo Random Binary Sequence) sequence generator according to the DVB-H standard and are placed at their respective sub carrier indices. Scattered pilots are always transmitted at a boosted power level. The amplitude of the scattered pilots is higher than the 16-QAM modulated complex data. The TPS (Transmission Parameter Signalling) information is transmitted for the benefit of the receiver and is used for the signalling parameters related to the transmission scheme. These Parameters are: frame number in a superframe, modulation scheme, hierarchy information, inner code rates, and guard interval length and transmission mode. Since the implementation of the physical layer for DVB-H system constitutes non-hierarchical transmission mode (No hierarchy being used), single transmission mode (4K), fixed guard interval length, modulation scheme is fixed, the transmission of TPS bits have been ignored.

Figure 4.1 shows the blocks diagram of a DVB-H system simulated using MATLAB consisting of both transmitter and the receiver. In the next section, Matlab output plots for each of the blocks in Figure 4.1 is shown and explained.

Figure 4.1. Block diagram of the DVB-H system simulated.

Plot of the scattered pilots is shown in Figure 4.2. It can be seen from the Figure 4.2 that amplitude of the reference signal is increased by a factor of 1.5. The scattered pilots are generated according to the polynomial for the Pseudo Random Binary Sequence generator X11+X2+1. Figure 4.3 shows the 16-QAM complex data along with the scattered pilots for one OFDM symbol. Figure 4.4 shows the 16-QAM constellation diagram of the input data with the pilots. In this Figure, the two points on the real axis are the scattered pilots; the other 16 points represent the 16-QAM modulated data values. The modulated carriers are loaded on the negative side and positive side of the frequency bins before it is passed through the IDFT/IFFT block. The plot in Figure 4.5 shows the occupied frequency bins for one DVB-H symbol which consists of modulated data and scattered pilots. Real and imaginary values for 10 OFDM symbols duration considered in this system is shown in the Figure 4.6.

Figure 4.2. Scattered pilots generated by PRBS sequence generator.

Figure 4.3. 16-QAM complex data with the scattered pilots.

Once the data is IDFT transformed, a cyclic prefix of length 1024 samples is appended to the beginning of every OFDM symbol. This helps in eliminating the ISI and ICI almost completely. It also ensures that the delayed replicas of the OFDM symbol will have integer number of cycles in the FFT interval as long the length of the delay is lesser than the length of the cyclic prefix. This process is repeated to generate 10 OFDM symbols. Figure 4.7 shows one DVB-H symbol with cyclic prefix for the duration of 560 µs after the IFFT block. This data is passed through a multipath fading channel. Before the data being passed through the channel, white complex Gaussian noise is added to the OFDM symbols. The FIR filter models a multipath fading channel. We assume that the maximum delay spread of the

Figure 4.4. 16-QAM constellation diagram with pilots.

Figure 4.5. One DVB-H symbol with data and reference signal.

Figure 4.6 Real and imaginary values for the duration of OFDM packet.

Figure 4.7. One DVB-H symbol with cyclic prefix in time domain.

61 channel is approximately 1/6 of the length of the guard interval. The channel considered here in the system is: channel=[1 zeros(1,40) 0.2*j zeros(1,40) 0.01 zeros(1,40) 0.01 zeros(1,40) 0.01] (4.1) The equation shown in Figure 4.1 is a multipath Rician channel with 4 path and 3 echoes. Impulse and frequency response of the channel is shown in the Figure 4.8.

Figure 4.8. Impulse and frequency response of the multipath channel.

One of the other impairments that have to be induced in the system is the frequency offset that exists between the transmitter and the receiver. The carrier frequency offset causes the sub-carriers index shift with the value of integer Carrier Frequency Offset. This would make the index position of scattered pilots in the wrong position. It is always necessary to compensate the carrier frequency offset as early as possible. So this is taken care at the receiver end. The carrier frequency offset rotates the constellation. We spin the samples of the OFDM symbols by an angle that corresponds to 1% of the bandwidth of the carriers but we can also go up to 10% of the bandwidth. The angle that corresponds to the frequency offset is:

(4.2)

(4.3)

(4.4) Where Δf refers to bandwidth of the carriers. Equation 4.4 would introduce 1% of frequency offset. It is like; spinning the data samples in the OFDM symbols by an angle specified in the Equation 4.4 in the time domain would create a frequency offset in the frequency domain. Figure 4.9 shows the constellation diagram of input data after spinning it according to the Equation 4.4 and then passing it through the fading channel with Added noise. Now we have reached a stage where the data has been transmitted through the channel and the necessary impairments has been introduced such as phase offset, white complex Gaussian noise and echoes in the channel.

Figure 4.9. Constellation diagram of the data after spinning and passing it through the multipath channel with AWGN.

OFDM is used for both broadcast type systems like DVB and packet switched networks, like WLANs. These two systems require somewhat different approach to the synchronization problem [22]. To facilitate synchronization in current WLAN standards, a preamble is included in the start of the packet. This makes the synchronization fast and relatively easier than the broadcast systems. However Digital Video Broadcasting system receivers would initially spend a long time to acquire the signal and then switch to the tracking mode. The OFDM receiver that I have designed here for the broadcast system would take very few OFDM symbols to acquire the signal and then demodulate. At the receiver side, we will follow the approach presented by Schimdl and Cox [23] for the Symbol Timing Synchronization. Preliminary stage of the receiver is to do the symbol timing synchronization, which we can call the system to be in acquisition mode. The input to the symbol detection system shown in the Figure 4.10 is the received samples. Frame synchronization presented here exploits the cyclic extension property that precedes every OFDM symbol. Here we correlate the received samples with the delayed version of itself. The length to the delay register is chosen to be size of the transform length i.e. 4096. Multiplying the delayed samples with the conjugate of current received samples is known as the Cross correlation. At the same time, we multiply the delayed register samples with the conjugate of itself and this process is called Auto Correlation. Next the cross correlation and auto correlation outputs are passed through the sliding average filter. For each sampling interval, cross correlation outputs and also the auto correlation outputs are added separately in the sliding average filter. Length of this filter is chosen to be the length of cyclic prefix i.e. 1024. The incoming samples into the sliding average filter are added and then passed onto the symbol averager. Averaging the cross correlation output over the length of the cyclic prefix results in maximum cross correlation value over the OFDM symbol duration. The maximum cross correlation peak occurs at the beginning of the next OFDM symbol. Since the different OFDM symbols contain independent sample values, the correlation output is a random variable, which may reach a value that is larger than the

Figure 4.10. Symbol detection system. desired correlation peak. To better the correlation outputs, the summed up values coming out of the sliding average filter is passed through the symbol averaging filter. Length of this symbol averaging filter is chosen to be length of the OFDM symbol with cyclic prefix i.e. 5120. This symbol averaging filter suppresses the noise as much as possible since the symbol keeps repeating, thus making the start of the symbol more detectable. However we can see in the Figure 4.11 that the cross correlation peaks to be very noisy in spite of averaging over the duration of time. This is because the received samples not only contains data, but also the noise and echoes from the channel. Next, we can think of weighted auto correlation sum to be acting as an automatic gain controller. The auto correlation also calculates the energy in the cyclic prefix. Since this calculated energy is almost equal to the energy of the cross correlation, ratio of the cross correlation over auto correlation would give a gain approximately equal to one. Ratio of Cross correlation over auto correlation is called the Normalized cross correlation. In the Figure 4.11, blue colored peaks show the normalized cross correlation peaks and the red colored line shows the auto correlation also, we can see how noisy the peaks are and hence

Figure 4.11. Plot of normalized cross correlation and auto correlation.

the peak detection becomes very difficult. Thus the demodulation results would give the incorrect outputs. Figure 4.10 also calculates the frequency offset of the received OFDM symbol spectrum. The cyclic extension correlation output can be used to estimate the frequency offset. The phase of the correlation output is equal to the phase drift between samples that are T seconds apart. Hence the frequency offset can simply be found as the correlation phase divided by 2πT [4]. To find the approximate phase drift from sample to sample, the angle at the output of the 1024 long sliding average filter is divided by 4096. This resultant angle would also contain the total angles of the 1024 samples. Figure 4.12 shows the coarse frequency offset. Equations for the cross correlation, auto correlation and the ratio according to [23] are shown below in Equations 4.5, 4.6 and 4.7.

(4.5)

Figure 4.12. Angle of coarse frequency offset.

(4.6)

(4.7)

A small value δ is added in the Equation 4.7 so that we do not want the ratio to go to infinity. Normalized cross correlation peaks are very noisy and it can be seen in the Figure 4.11. It cannot be used for peak detection and hence demodulation process. The aim is to smooth the normalized cross correlation peaks without sacrificing the sharpness of it. So the peak detection system shown in Figure 4.13 will help us in finding the peaks of the cross correlation and thus finding the start of the OFDM symbol [24]. The input to the peak detection system is the absolute values of the normalized cross correlation samples. The CIC filter removes the noise in the normalized cross correlation samples by averaging over many samples. The comb part of the CIC filter finds the

Figure 4.13. Peak detection system.

difference between two samples that are 200 taps apart. We would think this to be a derivative filter. The integrator averages the current sample with the previously calculated value. It is easily identified as the recursive accumulator [25]. The comb filter is selected in such a way that it is long enough to average over many samples and provide smooth curve, and should not be exceeding the length of the cyclic prefix. To normalize the output of CIC filter, it is scaled by a factor of P. Where P = 1/ R. Where R is the length of the comb filter. The outputs of the averaged cross correlation, derivative of the cross correlation can be seen in the Figures 4.14, 4.15 and 4.16. Figure 4.15 gives us the closer look of the averaged cross correlation peaks and we can see that the peaks are very smooth. This helps us in slope detection process to find the peak. Figure 4.17 shows the plot of both the averaged and differentiated cross correlation outputs. From the Figure 4.17 we can infer that, wherever the peaks are located in the averaged cross correlation output, we can see the dip in the differentiated cross correlation. We are using the averaged cross correlation outputs to find a peak and then the derivative of cross correlation to precisely locate the start of the OFDM symbol. The squelch detector in the Figure 4.13 turns on when a valid OFDM symbol is received. In the simulation, I have delayed the system to wait for 2 OFDM symbols, because the energy contained in it is very noisy. The squelch detector is used for monitoring the positive slope. The detector uses a 5 tap register to monitor a positive slope with a value greater than the set threshold. Once it finds the positive slope, it looks for a negative slope for

Figure 4.14. Averaged cross correlation.

Figure 4.15. Zoomed-in plot of averaged cross correlation.

Figure 4.16. Differentiated cross correlation.

Figure 4.17. Plot of averaged and differentiated cross correlation.

the duration of one OFDM symbol. If the detector finds a negative slope within this period, the system would turn on. Once the system turns on, it would stay up for the period of one OFDM symbol. Next, we are using 11-tap register that contains the averaged cross correlation samples to search for the peak value. The system searches for peak value in that register, by first looking into the positive slope and it waits for the negative slope to occur. So once the negative slope occurs, we know that the peak value is stored in that 11-tap register. Also, we use another 11-tap register for the differentiated cross correlation samples. So once we find the peak value in the register used for the averaged cross correlation, we also look for the corresponding minimum value in the 11-tap register used for differentiated cross correlation samples. Wherever the minimum value is located, it tells us the start of the OFDM symbol. Once the peak detection system locates the start of the OFDM symbol, it performs the coarse frequency offset by de-spinning the samples according to the calculated angle shown in Equation 4.4. The equalizer would take care of the residual frequency offset. Next step is to do the FFT, however before the data is passed onto the FFT block, the cyclic prefix is discarded and the rest of 4096 samples in each OFDM symbol are DFT transformed. The transform length for FFT algorithm is chosen to be same as the IFFT block i.e. 4096. Once the samples are in the frequency domain, the system does the channel estimation. The modulated bits are disturbed during the transmission through the channel since the channel introduces amplitude and phase shifts due to frequency selective and time- varying nature of the radio channel [8]. In order for the receiver to acquire the original bits, it needs to take into account these unknown changes. Hence we do the channel estimation. To estimate the channel, we make use of the received pilots. The received pilots are divided by the pilot values known to the receiver. To explain this, let’s say the first pilot encountered by the receiver is at the position ‘K1’. Let the second received pilot be at the position ‘K2’. Since the receiver knows the transmitted values of these pilots, channel effects can be found at these pilot locations. A linear interpolation technique is applied to estimate the channel effects for all the data samples lying in between the pilots. To calculate the value of received pilots, we use the Equation 4.8.

(4.8) Where,

rn = received sample

pn = known pilot value n = pilot locations Calculation of the channel effects for the data samples in between the estimated pilots is given by the Equation 4.9.

(4.9) Where,

Hpn = Channel estimate at pilot locations

Hk = Channel estimate for the data samples Next, we do the equalization to undo the effects of the channel. So the channel effect is equalized by multiplying the received samples with the reciprocal of the channel estimate. This shows equalization is a simple process in OFDM systems. This process is given by Equation 4.11.

(4.11) The equalizer uses the channel estimate of the current symbol to equalize the channel effects of the next incoming symbol. Figure 4.18 shows the spectrum of the estimated channel and the actual Rician fading channel. Figure 4.19 and 4.20 shows the 16 QAM constellation after the coarse frequency offset, constellation diagram at the transmitter and at the receiver after equalization for a Rician channel with 1% frequency offset. Figure 4.21 and 4.22 shows the output plots for the QPSK constellation for the multipath Rician channel with 1% frequency offset. Figure 4.23 shows the estimated channel and Rayleigh fading channel. The channel considered for Figure 4.23 is: channel=[0 zeros(1,9) 0.7 zeros(1,29) 0.07*j zeros(1,19) 0.007 zeros(1,41)] (4.12) Figure 4.24 and 4.25 shows the 16-QAM constellation diagrams for the Rayleigh Fading Channel.

Figure 4.18. Spectrum of the estimated channel and multipath rician channel with 1% frequency offset.

Figure 4.19. Constellation diagram of the received data after coarse frequency offset.

Figure 4.20. 16 QAM constellation diagrams at the transmitter and at the receiver after equalization.

Figure 4.21. QPSK constellation diagram of the received data after coarse frequency offset.

Figure 4.22. QPSK constellation diagram at the transmitter and at the receiver after equalization with frequency offset of 1%.

Figure 4.23. Spectrum of the extimated channel and Rayleigh channel with 1% frequency offset.

Figure 4.24. constellation diagram of the received data after coarse frequency offset fot he Rayleigh fading channel.

Figure 4.25. 16 QAM constellation diagram at the transmitter and at the receiver after equalization for the Rayleigh fading channel.

CHAPTER 5

CONCLUSION AND FUTURE WORK

The implemented DVB-H system is simulated in 4K FFT mode with 16-QAM and QPSK carrier modulation schemes. The system is simulated according to the ETSI (European Telecommunications Standard Institute) TR 102 377 v1.3.1 standard. The length of the cyclic prefix considered is 1/4 the duration of the OFDM symbol. The random data generated at the transmitter is assumed to be coded (outer and inner) and interleaved, and then transmitted through the multipath channel. The samples in the OFDM symbols were spun by an angle that corresponds to 1% of the bandwidth of the carriers. At the receiver, we computed the normalized cross correlation and the angle that corresponds to the frequency offset. However, since the peaks of the normalized cross correlation looked very noisy, the process of detecting the peak to locate the start of the OFDM symbol becomes very hard. We implemented a peak detection system using CIC filters to average the normalized cross correlation samples and also calculate the differentiated cross correlation values. The system uses the averaged cross correlation samples to find the peak and differentiated cross correlation samples to locate the start of the OFDM symbol. The system was simulated considering two different channel conditions. First is the Rician fading channel and the other is Rayleigh fading channel. The transmitted data was successfully received. This is shown in the Constellation diagrams for the transmitted OFDM symbols and the received OFDM symbols after equalization. Spectrum of the actual multipath channel and the estimated channel is also shown. The system also shows the channel estimation process using the scattered pilots. The implemented simulator is very flexible; one can change the size of the FFT mode like 2K mode or 8K mode with corresponding guard interval length and look into the performance and results of the system for different carrier modulation schemes. The system designed in this thesis, can very well serve as a baseline for the simulation of DVB-SH (Satellite Services to Handheld devices) system, with some minor necessary modifications.

The simulation can further be extended by adding the coding, interleaving and also transmit the real MPEG coded TV signals, and then finally recover and demodulate the signals. Further scope of improvement would be to investigate the synchronization of fine frequency offset, more robust channel estimation techniques. The receiver can further be designed in such a way that, it can be synchronized with any given mobile TV standards. The future of DVB-H is called the DVB-NGH (Digital Video Broadcasting – Next Generation Handheld). It is an enhanced version of DVB-H. It is expected that by 2015, the demand for rich multimedia content consumption would increase drastically. To facilitate this, a more optimized coding scheme is necessary. Standardization process for DVB-NGH would begin in the first quarter of 2010. The publications of the related ETSI standard(s) are expected in 2011. The first commercial NGH devices might then become available in 2013 [26].

REFERENCES

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