Terahertz Emitter Using Resonant-Tunneling Diode and Applications

sensors

Review Terahertz Emitter Using Resonant-Tunneling Diode and Applications

Masahiro Asada 1,* and Safumi Suzuki 2

1 Institute of Innovative Research, Tokyo Institute of Technology, Tokyo 152-8552, Japan 2 Department of Electrical and Electronic Engineering, Tokyo Institute of Technology, Tokyo 152-8552, Japan; [email protected] * Correspondence: [email protected]

Abstract: A compact source is important for various applications utilizing terahertz (THz) waves. In this paper, the recent progress in resonant-tunneling diode (RTD) THz oscillators, which are compact semiconductor THz sources, is reviewed, including principles and characteristics of oscillation, studies addressing high-frequency and high output power, a structure which can easily be fabricated, frequency tuning, spectral narrowing, different polarizations, and select applications. At present, fundamental oscillation up to 1.98 THz and output power of 0.7 mW at 1 THz by a large-scale array have been reported. For high-frequency and high output power, structures integrated with cylindrical and rectangular cavities have been proposed. Using oscillators integrated with varactor diodes and their arrays, wide electrical tuning of 400–900 GHz has been demonstrated. For spectral narrowing, a line width as narrow as 1 Hz has been obtained, through use of a phase-locked loop system with a frequency-tunable oscillator. Basic research for various applications—including imaging, spectroscopy, high-capacity wireless communication, and radar systems—of RTD oscillators has been carried out. Some recent results relating to these applications are discussed.

Citation: Asada, M.; Suzuki, S. Keywords: terahertz oscillator; resonant-tunneling diode; frequency tuning; spectral narrowing; Terahertz Emitter Using polarizations; spectroscopy; wireless communication; radar Resonant-Tunneling Diode and Applications. Sensors 2021, 21, 1384. https://doi.org/10.3390/s21041384 1. Introduction Academic Editors: Viktor Krozer and The terahertz (THz) band, which has a frequency of about 0.1 to several THz, is ex- Richard Klemm pected to play key roles in various applications, such as imaging, chemical and biotechno- logical analyses, and communications [1–3]. Compact solid-state THz sources are important Received: 1 December 2020 devices for these applications and various kinds of such sources have been studied, com- Accepted: 13 February 2021 Published: 16 February 2021 prising both optical and electronic devices, as the THz band is located between millimeter and light waves.

Publisher’s Note: MDPI stays neutral Figure1 shows the current status of the various semiconductor THz sources that with regard to jurisdictional claims in directly generate THz waves from a dc power supply—note that sources that require other published maps and institutional afﬁl- external microwave or light sources to generate THz waves (e.g., by multiplication or dif- iations. ference frequency) are not included in the ﬁgure. On the optical device side, p-germanium (p-Ge) lasers [4] and quantum cascade lasers (QCLs) have been studied [5–9]. Recently, room-temperature THz sources with difference frequency generation (DFG) using mid- infrared QCLs have been reported [10–13]. These devices are included in Figure1, as all the light sources needed for these devices are integrated into one chip and the THz waves are Copyright: © 2021 by the authors. Licensee MDPI, Basel, Switzerland. generated by DC power supplied to the chip without external sources of other frequencies. This article is an open access article On the electronic device side, impact ionization avalanche transit-time (IMPATT) diodes, distributed under the terms and tunneling transit-time (TUNNETT) diodes, Gunn diodes [14–16], and transistors such as conditions of the Creative Commons heterojunction bipolar transistors (HBTs), high electron mobility transistors (HEMTs), and Attribution (CC BY) license (https:// complementary metal-oxide-semiconductor (CMOS) transistors [17–21] have been studied creativecommons.org/licenses/by/ as THz sources. Recently, the operating frequencies of transistors have increased remark- 4.0/). ably. Other than semiconductors, THz emitters using intrinsic Josephson junctions in the

Sensors 2021, 21, 1384. https://doi.org/10.3390/s21041384 https://www.mdpi.com/journal/sensors Sensors 2021, 21, x FOR PEER REVIEW 2 of 19

junctions in the layered high-temperature superconductor BiSrCaCuO have also been studied [22, 23], which have the advantage of having a wide frequency tuning range. Resonant-tunneling diodes (RTDs) are also promising candidates for room-temperature THz sources [24–31]. Currently, oscillation up to 1.98 THz has been obtained at room temperature [32, 33], while structures for higher frequency and high output power are being studied [34, 35]. Studies aiming toward several applications, such as imaging, spec- Sensors 2021, 21, 1384 troscopy, wireless communications, and radars, have recently begun [36–47]. In this2 of pa- 19 per, we review recent developments and applications of THz sources using RTD oscillators. Although RTDs can also be used as THz detectors [48–51], here, we only describe

layeredRTD TH high-temperaturez sources, except superconductorfor a brief introduction Bi2Sr2CaCu of the2O application8+δ have also of been RTD studied THz detectors [22,23], whichfor wireless have thecommunications. advantage of having a wide frequency tuning range.

Figure 1. Current status ofof semiconductorsemiconductor on-chipon-chip terahertzterahertz (THz)(THz)sources. sources.Output Outputpower power as as a a func- tionfunction of generated of generated frequency. frequency. Devices Devices without without temperature temperature indication indication operate operate at room at room temperature. temperature. Resonant-tunneling diodes (RTDs) are also promising candidates for room-temperature THz2. Structure, sources [Oscillation24–31]. Currently, Principle, oscillation and Oscillation up to 1.98 Characteristics THz has been obtained of RTD Oscillators at room tem- peratureAn RTD [32,33 is], made while of structures heterostructures for higher with frequency ultrathin and semiconductor high output multi power-layers. are being The studied [34,35]. Studies aiming toward several applications, such as imaging, spectroscopy, layer structure we use for a THz source is shown in Figure 2a. The main part is composed wireless communications, and radars, have recently begun [36–47]. In this paper, we review of an InGaAs quantum well and AlAs double barriers. An InGaAlAs emitter, an InGaAs recent developments and applications of THz sources using RTD oscillators. Although RTDs collector spacer, and a high-doped InGaAs collector are constructed around the main part. can also be used as THz detectors [48–51], here, we only describe RTD THz sources, except for These structures are epitaxially grown on a semi-insulating (SI) InP substrate. In DC op- a brief introduction of the application of RTD THz detectors for wireless communications. erations, the conduction band edge of the emitter is lifted by bias voltage, as shown in 2.Figure Structure, 2b. At Oscillationthe bias voltage Principle, where and the Oscillationconduction Characteristicsband edge of the of emitter RTD Oscillators is aligned to or exceeds the resonance level of the quantum well, the current–voltage (I–V) curve indi- An RTD is made of heterostructures with ultrathin semiconductor multi-layers. The cates the negative differential conductance (NDC) region, in which the current decreases layer structure we use for a THz source is shown in Figure2a. The main part is composed with increasing bias voltage. This region is used for the THz oscillation. In our RTD struc- of an InGaAs quantum well and AlAs double barriers. An InGaAlAs emitter, an InGaAs ture, a deep quantum well with indium-rich InGaAs and an emitter with InAlGaAs, hav- collector spacer, and a high-doped InGaAs collector are constructed around the main ing a high conduction band edge, are used to reduce the bias voltage required for NDC. part. These structures are epitaxially grown on a semi-insulating (SI) InP substrate. In DC operations,Figure 2c shows the conduction an example band of the edge measured of the emitterI–V curves is lifted at various by bias temperatures voltage, as shown [52]. The in FigureNDC region2b. At thecan biasbe seen voltage to have where unstable the conduction current fluctuation band edge of. This the emitterfluctuation is aligned occurs to due or exceedsto parasitic the resonance oscillations level in of the the measurement quantum well, circuits, the current–voltage composed of (I–V) leading curve wires indicates and thepower negative supply. differential The wires conductance and power supply (NDC) construct region, in a which resonance the current circuit decreasesfor oscillati withon, increasingwhich is described bias voltage. later. This Relaxation region isoscillation used for the[53] THzand current oscillation. bi-stability In our RTD which structure, is caused a deepby charge quantum buildup well and with depletion indium-rich in the InGaAs quantum and well an emitter [54,55] with may InAlGaAs, also occur havingin this circuit. a high conduction band edge, are used to reduce the bias voltage required for NDC. Figure2c shows an example of the measured I–V curves at various temperatures [52]. The NDC region can be seen to have unstable current ﬂuctuation. This ﬂuctuation occurs due to parasitic oscillations in the measurement circuits, composed of leading wires and power supply. The wires and power supply construct a resonance circuit for oscillation, which is described later. Relaxation oscillation [53] and current bi-stability which is caused by charge buildup and depletion in the quantum well [54,55] may also occur in this circuit. Sensors 2021, 21, x FOR PEER REVIEW 3 of 19

The I–V curves change very little with temperature, probably as the carrier concen- tration at the conduction band edge of the emitter is insensitive to temperature, due to high Fermi energy, as well as because the AlAs barriers are high. The current density at Sensors 2021, 21, 1384 3 of 19 the peak point is typically 10–30 mA/μm2, while the peak-to-valley current ratio (PVCR) is 2–4. The current density is large for narrow barriers and quantum wells, and strongly depends on the thicknesses of these layers.

(a)

(b) (c)

FigureFigure 2. 2.Layer Layer structure structure and and current–voltage current–voltage characteristics characteristics of resonant-tunnelingof resonant-tunneling diodes diodes (RTDs): (RTDs): (a) Layer(a) Layer structure structure of In- of GaAs/AlAsInGaAs/AlAs double-barrier double-barrier RTD. RTD. Reprinted Reprinted with with permission permission from from [56]. [ Copyright56]. Copyright (2016) (2016) Springer Springer Nature. Nature. (b) Schematic (b) Schematic I–V I–V curve and potential profile at various bias voltages, and (c) an example of measured I–V curves at various tempera- curve and potential proﬁle at various bias voltages, and (c) an example of measured I–V curves at various temperatures [52]. tures [52]. The I–V curves change very little with temperature, probably as the carrier concentra- tion atAs the a conductionmaterial combination, band edge we of the chose emitter InGaAs/AlAs is insensitive on an to InP temperature, substrate, dueas high to high bar- Fermiriers and energy, high as current well as becausedensities the are AlAs possible barriers in this are high.system. The For current high density output atpower, the peak the pointlarge is voltage typically width 10–30 of mA/the NDCµm2, region while the(Δ푉 peak-to-valley in Figure 2b) currentis desirable, ratio as (PVCR) discussed is 2–4. below. The currentFor this density purpose, is large materials for narrow with high barriers breakdown and quantum voltage wells, may and be stronglyadvantageous. depends GaN on - thebased thicknesses material ofsystems these layers. may be candidates, although high-frequency operations must be separatelyAs a material investigated. combination, Some results wechose of RTDs InGaAs/AlAs with such systems on an have InP substrate, been reported as high [57– barriers60]. and high current densities are possible in this system. For high output power, the largeThe voltageschematic width structure of the of NDC the fabricated region (∆V RTDin Figure oscillator2b) isis desirable,shown in asFigure discussed 3a [56]. below.The RTD For is this placed purpose, near the materials center of with one highside of breakdown a slot antenna, voltage which may works be advantageous. as a resonator GaN-basedand a radiator, material and the systems upper may electrode be candidates, of the RTD although is connected high-frequency to the other operations side of the mustslot through be separately the capacitance investigated. formed Some by resultsan MIM of (metal RTDs–insulator with such–metal) systems structure. have been This reportedMIM structure [57–60]. is used to isolate the bias lines to the upper and lower electrodes of the RTD.The Outside schematic of the structure slot antenna of the and fabricated RTD, a resistor RTD oscillator for stabilization is shown is inconnected Figure3a in [ 56par-]. Theallel RTD with is the placed RTD near to suppress the center parasitic of one side oscillations of a slot antenna,formed by which the workscircuit, as including a resonator the andleading a radiator, wires and and power the upper supply. electrode By making of the the RTD reciprocal is connected of this to resistor the other larger side than of the the slot through the capacitance formed by an MIM (metal–insulator–metal) structure. This MIM structure is used to isolate the bias lines to the upper and lower electrodes of the RTD. Outside of the slot antenna and RTD, a resistor for stabilization is connected in parallel with the RTD to suppress parasitic oscillations formed by the circuit, including the leading wires and power supply. By making the reciprocal of this resistor larger than the absolute value of the NDC of the RTD, the NDC is electrically hidden from the outside. As shown in SensorsSensors2021 2021, ,21 21,, 1384 x FOR PEER REVIEW 44of of 1919

theabsolute right-hand value sideof the of FigureNDC of3a, the the RTD, oscillator the NDC chip is mountedelectrically on hidden a silicon from hemispherical the outside. orAs hyper-hemispherical shown in the right-hand lens, side in order of Figure to extract 3a, the the oscillator output power, chip is asmounted most of on the a output silicon ishemispher radiatedical from or the hyper slot- antennahemispherical to the lens, substrate in order side, to due extract to the the large output dielectric power, constant as most ofof InPthe [output61]. For is a radiated collimated from output the slot beam, antenna a hyper-hemispherical to the substrate side, lens due is used. to the Structures large die- withoutlectric constant siliconlenses of InP have [61]. alsoFor beena collimated reported output [27,28, 62beam,–64]. a hyper-hemispherical lens is used. Structures without silicon lenses have also been reported [27, 28, 62–64].

(a) (b)

FigureFigure 3.3. StructureStructure andand operatingoperating principleprinciple ofof RTDRTD oscillators.oscillators. ReprintedReprinted withwith permissionpermission fromfrom [[5656].]. CopyrightCopyright (2016)(2016) SpringerSpringer Nature:Nature: ((aa)) RTDRTD oscillatoroscillator integratedintegrated withwith slotslot antennaantenna andand entireentire structurestructure includingincluding siliconsilicon hemisphericalhemispherical lens,lens, andand ((bb)) equivalentequivalent circuit,circuit, including including RTD RTD and and slot slot antenna. antenna. Parasitic Parasitic elements elements in in RTDRTD are are neglected neglected for for sake sake of of simplicity. simplicity.

FigureFigure3 3bb shows shows the the equivalent equivalent circuit circuit of of the the oscillator oscillator in in the the THz THz frequency frequency region, region, wherewhere− −G퐺RTD푅푇퐷 isis the the NDC NDC of of thethe RTD,RTD,G 퐺ANT퐴푁푇 isis the the conductanceconductance of of thethe slotslot antenna,antenna, whichwhich isis composedcomposed of of the the radiation radiation conductance conductanceG rad퐺푟푎푑and and the the conductance conductance due due to the to Ohmicthe Ohmic loss Glossloss ,퐺 and푙표푠푠, Landand 퐿C andare the퐶 are inductance the inductance and capacitance and capacitance of the of RTD the RTD and slotand antenna.slot antenna. As theAs the capacitance capacitance of theof the RTD RTD is muchis much larger larger than than that that of of the the slotslot antenna,antenna, C퐶 isis dominated dominated byby the RTD, RTD, while while 퐿L is dominated dominated by by the the slot slot antenna. antenna. In the In thedevice device design, design, 퐿 andL and 퐶 areC arecalculated calculated for forthe theantenna antenna using using three three-dimensional-dimensional (3D) (3D) electromagnetic electromagnetic simulation simulation and andthe parallel the parallel-plate-plate capacitances capacitances of the of RTD the RTD are calculated are calculated for the for constituent the constituent layers. layers. The Theadditional additional capacitance capacitance caused caused by the by electron the electron delay delay time timeis also is alsoconsidere consideredd in the in RTD the RTD[65]. [Parasitic65]. Parasitic elements elements [56, 65 [56] ,around65] around the theRTD RTD are areneglected neglected in Figure in Figure 3b,3b, for for the the sake sake of ofsimplicity simplicity in inthe the explanation explanation of ofoscillation oscillation principle. principle. TheThe condition condition required required for for oscillation oscillation is isG RTD퐺 ≥≥G퐺ANT at the the oscillation oscillation frequency frequency √ 푅푇퐷 퐴푁푇 f푓OSC푂푆퐶 ==11⁄/(22휋π√퐿퐶LC). As. As the the oscillation oscillation frequency frequency is is determined determined by by the the total total of LCLC formedformed byby the antenna and and RTD, RTD, the the length length of ofthe the antenna antenna is usually is usually much much shorter shorter than than the half the- half-wavelengthwavelength of the of oscillation the oscillation frequency. frequency. For Fora fixed a fixed antenna antenna structure, structure, the theoscillati oscillationon fre- frequencyquency can can be increased be increased by reducing by reducing the capacitance the capacitance of the ofRTD, the which RTD, is which mainly is done mainly by donereducing by reducing the RTD the mesa RTD area. mesa However, area. However, 퐺푅푇퐷 simultaneouslyGRTD simultaneously decreases decreases with the with reduc- the reductiontion of the of RTD the mesa RTD area. mesa Thus, area. the Thus, oscillation the oscillation frequency frequency reaches reaches its upper its limit upper (퐺푅푇퐷 limit= (퐺G퐴푁푇RTD) with= G ANTthe reduction) with the of reduction the RTD of mesa the RTDarea. mesaIn addition, area. In due addition, to the delay due totime the of delay elec- timetrons of in electrons the RTD in layers, the RTD 퐺푅푇퐷 layers, per GareaRTD alsoper degrades area also degradeswith increasing with increasing frequency. frequency. TheThe aboveabove description description of of the the oscillation oscillation principle principle is basedis based on NDCon NDC in electrical in electrical circuits. cir- Ascuits. the As frequency the frequency increases, increases, the photon the photon energy energy becomes becomes non-negligible non-negligible and and a different a differ- explanation,ent explanation, including including electron electron transitions, transitions, is neededis needed (as (as in in a laser).a laser). However, However, asas thethe amplificationamplification of of electromagnetic electromagnetic energy energy can can be be expressed expressed by by anan equivalentequivalent circuit, circuit, the the aboveabove electricalelectrical descriptiondescription cancan bebe usedused asas anan approximateapproximate one,one, byby changingchanging parametersparameters suchsuch asas NDC.NDC. ConsideringConsidering the above conditions, conditions, the the requirements requirements for for an an RTD RTD to to obtain obtain high high oscil- os- cillationlation frequency frequency are are high high 퐺푅푇퐷GRTD perper are areaa at high at high frequency frequency and andlow lowcapacitance capacitance per area. per area.Small Smallvalues values of 퐺푙표푠푠 of andGloss 퐿and areL alsoare required also required for the for antenna. the antenna. 퐺푟푎푑 Ginrad 퐺퐴푁푇in G cannotANT can- be notreduced, be reduced, as the output as the outputpower is power determined is determined by 퐺푟푎푑 by(seeG radbelow).(see below).Although Although the parasitic the parasiticelements elementsaround the around RTD also the degrade RTD also 퐺푅푇퐷 degrade with GincreasingRTD with frequency, increasing the frequency, other effects the othermentioned effects above mentioned seem to above be significant, seem to be so significant, far, to increase so far, the to oscillation increase thefrequency oscillation [56,

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frequency [56,65,66]. At higher frequencies, the effects of the parasitic elements need to be considered in detail. In order to increase GRTD, the current density in the I–V curve is increased with thin barriers and the quantum well, as shown in Figure2a. The capacitance per unit area is also reduced by inserting the collector spacer layer in Figure2a. For the electron delay time in RTD layers, the degradation of GRTD with frequency is discussed using the h i approximate formula GRTD(ω) ∝ cos ω τRTD + τdep/2 , where τRTD and τdep are the residence time in the double barrier structure and the transit time in the collector spacer layer, respectively [56,65,66]. In the derivation of this formula, τRTD is phenomenologically introduced by assuming that electrons are affected only by the time delay τRTD at resonant tunneling [65]. A detailed analysis for a more exact treatment is a future subject, including, for example, the potential change due to electron accumulation in the well [67,68], photon-assisted tunneling [69–71], and so on, or more precise quantum-mechanical analyses [72–74]. In fact, the experimental result of the frequency dependence of GRTD [52] slightly deviated from the above formula, although more experimental data are needed. In any case, it is clear that the delay time must be reduced for higher-frequency oscillation. We used thin barriers and a quantum well to reduce the delay time at resonant tunneling, in addition to high current density [75]. Furthermore, we optimized the thickness of the collector spacer to make τdep and the capacitance as small as possible at the same time. Using these methods, oscillation frequencies up to 1.42 THz have been obtained [76]. The length of the slot antenna was ﬁxed at 20 µm, while the oscillation frequency was increased by reducing the RTD mesa area. The RTD mesa area was approximately 0.6 µm2 at 1 THz and 0.2 µm2 at 1.42 THz. The output power was approximately 20 and 1 µW at around 1 and at 1.42 THz respectively, and rapidly decreased as the RTD mesa area approached the upper limit of oscillation. For the antenna, Gloss can be reduced by reducing the conduction loss, which exists on the metal surface around the slot and on the bridge connecting the antenna to the RTD. The former was reduced by optimizing the combination of antenna length and RTD mesa area, through which oscillation up to 1.55 THz has been obtained [77]. The latter was also reduced by improving the structure of the bridge. Through the use of these methods, oscillation up to 1.92 THz has been obtained [78]. In addition, by making the antenna electrode thicker, the area of the side wall of the slot increases and the conduction loss is further reduced. Combining all of the methods mentioned above, oscillation frequency up to 1.98 THz has been obtained [32], as shown Sensors 2021, 21, x FOR PEER REVIEW 6 of 19 in Figure4. This is the highest frequency achieved by room-temperature electronic single oscillators, to date.

(a) (b)

Figure 4. RTD oscillatorFigure with 4.thickRTD antenna oscillator electrode. with thickCopyright antenna (2017) electrode. IEEE. Reprinted Copyright with (2017) permission IEEE. Reprinted from [32]: with (a) Structure of the oscillator,permission and (b) oscillation from [32]: (spectra.a) Structure of the oscillator, and (b) oscillation spectra.

The output power of an RTD oscillator is theoretically given by 푃표푢푡 = (2⁄3)퐺푟푎푑(퐺푅푇퐷 − 퐺퐴푁푇)⁄푏, where 푏 is the coefficient of the non-linear term included in the NDC under oscillation [65]. 푃표푢푡 changes with 퐺푟푎푑 , and is maximized at 퐺푟푎푑 = 퐺푅푇퐷 − 퐺퐴푁푇, i.e., 퐺푟푎푑 = (퐺푅푇퐷 − 퐺푙표푠푠)⁄2. Using the third-order polynomial approxima- 3 tion of the I–V curve, 퐺푅푇퐷 and 푏 can be expressed as [65] (3⁄2)(Δ퐼⁄Δ푉) and 2Δ퐼⁄Δ푉 respectively, where Δ퐼 and Δ푉 are the current and voltage widths in the NDC region of the I–V curve, as shown in Figure 2b. The maximum output power in the above condition 2 is calculated as 푃푚푎푥 = (3⁄16)Δ퐼Δ푉(1 − 퐺푙표푠푠⁄퐺푅푇퐷) . Thus, in order to increase the output power, 퐺푟푎푑 must be optimized, Δ퐼 and Δ푉 must be increased, and 퐺푙표푠푠 must be reduced. The oscillators integrated with slot antennas described above are not optimized for 퐺푟푎푑, and their typical output power is a few tens of μW. The 퐺푟푎푑 of the slot antenna can be designed and optimized through the offset structure, in which the position of the RTD is shifted from the center of the slot and an output power of a few hundred μW has been obtained [79, 80]. Δ퐼 can be increased by increasing the RTD mesa area; however, the oscillation frequency decreases, due to an increase in capacitance. A structure with a large Δ퐼 and small 퐺푙표푠푠 that can maintain the oscillation frequency is shown in the next section. The increase of Δ푉 is a future subject. A possible method may be through the appropriate design of RTD layers (e.g., an increase in thickness of the collector spacer layer), although the upper limit of oscillation frequency must be discussed simultaneously. Power combining through array configuration is also useful for obtaining high output power. An oscillator with a two-element array of the offset slot antennas has exhibited an output power of 0.6 mW at 620 GHz [80]. In this array, single-frequency oscillation was observed due to mutual locking between the coupled elements, which implies coherent power combining. In a large-scale array without intentional coupling between the elements, 0.73 mW has been obtained at 1 THz for 89 elements, as shown in Figure 5 [63]. In this device, any intentional coupling structure for stable synchronization was not introduced. However, the elements appeared to be weakly coupled with each other through random reflections and feedback of the output power radiated into the substrate or the dielectric film (COC film in Figure 5). As the elements were not perfectly synchronized, due to weak coupling, multiple peaks were observed in the oscillation spectrum. This behavior is suitable for applications such as imaging in which the interference fringe is a problem in coherent sources.

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However, the decrease in Gloss saturated with a further increase in thickness of the antenna electrode [33]. This was because the inductance Lslot of the slot antenna also decreases with increasing antenna thickness, in addition to the decrease in the resistance R of the antenna electrode. As R is connected in series with Lslot and the relation R ωosc Lslot holds at the angular frequency of oscillation ωosc, Gloss can be approximately 2 given by R/(ωosc Lslot) . Gloss remarkably decreases with increasing antenna thickness, up to approximately 2 µm [33], due to the decrease in R and the weak dependence of Lslot on antenna thickness. Above this thickness, however, Gloss saturates with antenna thickness due to the decrease in Lslot. Thus, the upper limit of oscillation frequency saturates with the antenna thickness. Considering this result, a new structure other than the slot-integrated one must be proposed for higher-frequency oscillation, as shown in the next section. The output power of an RTD oscillator is theoretically given by Pout = (2/3)Grad (GRTD − GANT)/b, where b is the coefficient of the non-linear term included in the NDC under oscillation [65]. Pout changes with Grad, and is maximized at Grad = GRTD − GANT, i.e., Grad = (GRTD − Gloss)/2. Using the third-order polynomial approximation of the 3 I–V curve, GRTD and b can be expressed as [65] (3/2)(∆I/∆V) and 2∆I/∆V respectively, where ∆I and ∆V are the current and voltage widths in the NDC region of the I–V curve, as shown in Figure2b. The maximum output power in the above condition is calculated 2 as Pmax = (3/16)∆I∆V(1 − Gloss/GRTD) . Thus, in order to increase the output power, Grad must be optimized, ∆I and ∆V must be increased, and Gloss must be reduced. The oscillators integrated with slot antennas described above are not optimized for Grad, and their typical output power is a few tens of µW. The Grad of the slot antenna can be designed and optimized through the offset structure, in which the position of the RTD is shifted from the center of the slot and an output power of a few hundred µW has been obtained [79,80]. ∆I can be increased by increasing the RTD mesa area; however, the oscillation frequency decreases, due to an increase in capacitance. A structure with a large ∆I and small Gloss that can maintain the oscillation frequency is shown in the next section. The increase of ∆V is a future subject. A possible method may be through the appropriate design of RTD layers (e.g., an increase in thickness of the collector spacer layer), although the upper limit of oscillation frequency must be discussed simultaneously. Power combining through array configuration is also useful for obtaining high output power. An oscillator with a two-element array of the offset slot antennas has exhibited an output power of 0.6 mW at 620 GHz [80]. In this array, single-frequency oscillation was observed due to mutual locking between the coupled elements, which implies coherent power combining. In a large-scale array without intentional coupling between the elements, 0.73 mW has been obtained at 1 THz for 89 elements, as shown in Figure5[ 63]. In this device, any intentional coupling structure for stable synchronization was not introduced. However, the elements appeared to be weakly coupled with each other through random reflections and feedback of the output power radiated into the substrate or the dielectric film (COC film in Figure5). As the elements were not perfectly synchronized, due to weak coupling, multiple peaks were observed in the oscillation spectrum. This behavior is suitable for applications such as imaging in which the interference fringe is a problem in coherent sources. For stable synchronization and coherent power combining, strong coupling between array elements is required; furthermore, as the number of elements increases, stronger coupling is required [81]. As coupling through the circuits on the element plane seems to be limited, another method may be needed, such as putting the entire array into a resonator for strong coupling. The measurement of the temperature dependence of oscillation characteristics has also been reported [52]. The oscillation frequency was almost constant with temperature, while the output power drastically increased with decreasing temperature between 10 and 300 K. As NDC is insensitive to temperature, as can be seen from Figure2c, the change in output power was attributed to the change in Ohmic loss of the antenna electrode with Sensors 2021, 21, 1384 7 of 19

Sensors 2021, 21, x FOR PEER REVIEW 7 of 19 temperature. In the narrow temperature range of 300–350 K, the change in the measured output power was small.

(a)

(b)

Figure 5. Large-scaleFigure array 5.of Large-scaleRTD oscillators. array Reprinted of RTD oscillators. from Reference Reprinted [63] with from the Reference permission [63] withof AIP the Publishing: permission (a) Structure of the deviceof— AIPan Publishing:array element (a is) Structurecomposed of of the an device—anRTD with a arrayslot antenna element (resonator) is composed covered of an by RTD COC with (cyclic a olefin copolymer) filmslot and antenna a dipole (resonator) array antenna covered (radiator), by COC and (cyclic (b) olefinoutput copolymer) power as a film function and a of dipole element array number antenna and oscillation spectrum for 89-element array. (radiator), and (b) output power as a function of element number and oscillation spectrum for 89-element array. For stable synchronization and coherent power combining, strong coupling between 3. Novelarray Oscillator elements Structures is required; for Highfurthermore, Frequency, as the High number Output of Power,elements and increases, stronger Easy Fabricationcoupling is required [81]. As coupling through the circuits on the element plane seems to Asbe the limited, upper another limit of method oscillation may frequency be needed, of such slot-integrated as putting the oscillators entire array was into found a resona- to saturatetor for with strong increasing coupling. thickness of antenna electrode, as mentioned in the previous section, a novelThe oscillatormeasurement structure of the with temperature a cylindrical-cavity dependence resonator of oscillation and bow-tie characteristics antenna has has beenalso proposed been reported for higher [52]. The frequency oscillation oscillation, frequency as was shown almost in Figure constant6a [ with33]. Intemperature, this structure,while the the oscillation output power frequency drastically is determined increased by with the resonance decreasing frequency temperature of the between res- 10 onatorand composed 300 K. As of NDC the cavity is insensitive and the to RTD, temperature, and the output as can is be radiated seen from from Figure the bow-tie2c, the change antenna,in output which power is connected was attributed to the cavity to the by change the MIM in Ohmic layers. loss As of the the surface antenna area electrode of the with cavityte ismperature. large, the conductionIn the narrow loss temperature is small and range high-frequency of 300–350 K, oscillation the change is in expected. the measured A theoreticaloutput calculationpower was small. [34] has shown that oscillation up to approximately 2.8 THz is possible by optimizing the radius and height of the cylindrical cavity, as shown in Figure6b. In a preliminary3. Novel Oscillator experiment, Structures 1.78 THz for oscillation High Frequency, has been High obtained Output for aPower, non-optimized and Easy Fab- structurerication with parasitic capacitance [33]. The theoretical output power is approximately 0.5–1 µW aroundAs the 2.5upper THz, limit at present of oscillation [82]. Higher frequency frequency of slot and-integrated higher outputoscillators power was can found to be expectedsaturate with with further increasing structure thickness optimization, of antenna including electrode, other as parameters. mentioned in For the example, previous sec- the outputtion, powera novel can oscillator be increased structure with with the capacitancea cylindrical of-cavity MIM resonator structure connectingand bow-tie the antenna cavityhas and been antenna, proposed as the antennafor higher conductance frequency (viewed oscillation, from as RTD) shown is altered in Figure by this 6a capac-[33]. In this itancestructure, and can be the optimized, oscillation as frequency mentioned is in determined the previous by section. the resonance An array frequency configuration of the reso- is alsonator useful composed for high output of the power.cavity Evenand the with RTD, a small and outputthe output power is inradiated a single from oscillator, the bow-tie antenna, which is connected to the cavity by the MIM layers. As the surface area of the cavity is large, the conduction loss is small and high-frequency oscillation is expected. A theoretical calculation [34] has shown that oscillation up to approximately 2.8 THz is possible by optimizing the radius and height of the cylindrical cavity, as shown in Figure 6b.

Sensors 2021, 21, x FOR PEER REVIEW 8 of 19

In a preliminary experiment, 1.78 THz oscillation has been obtained for a non-optimized structure with parasitic capacitance [33]. The theoretical output power is approximately 0.5–1 μW around 2.5 THz, at present [82]. Higher frequency and higher output power can be expected with further structure optimization, including other parameters. For example, Sensors 2021, 21, 1384 8 of 19 the output power can be increased with the capacitance of MIM structure connecting the cavity and antenna, as the antenna conductance (viewed from RTD) is altered by this capacitance and can be optimized, as mentioned in the previous section. An array configu- there mayration be is suitable also useful applications, for high suchoutput as microchipspower. Even for with spectroscopy, a small output as explained power later.in a single Fabricationoscillator, of an there oscillator may usingbe suitable a cylindrical applications, cavity such is currently as microchips in progress. for spectroscopy, as explained later. Fabrication of an oscillator using a cylindrical cavity is currently in progress.

(a) (b)

Figure 6. RTD oscillatorFigure integrated 6. RTD oscillatorwith cylindrical integrated cavity with and cylindrical bow-tie antenna cavity andfor high bow-tie-frequency antenna oscillation for high-frequency [33, 34]: (a) Oscillator structure, andoscillation (b) calculation [33,34]: (ofa) oscillation Oscillator structure, frequency and as a ( bfunction) calculation of cavity of oscillation height and frequency radius. Copyright as a function (2020) of The Japan Society of Applied Physics [34]. cavity height and radius. Copyright (2020) The Japan Society of Applied Physics [34].

For highFor outputhigh output power, power, an oscillator an oscillator integrated integrated with with rectangular rectangular cavity cavity has has been been pro- proposedposed [35 ,[8335,], 83 as], shown as shown in Figure in Figure7a. In 7a. this In structure, this structure, the RTD the hasRTD a longhas a strip long shape, strip shape, and theand cavity the cavity is regarded is regarded as a small as a inductancesmall inductance connected connected to the to RTD. the DueRTD. to Due the to low the low inductanceinductance of the cavity,of the ancavity, RTD an with RTD large with capacitance large capacitance and large and mesa large area mesa can bearea used can for be used oscillationfor oscillation at relatively at relatively high frequency. high frequency. Thus, the Thus, current the width current∆I widthof the Δ NDC퐼 of the region, NDC to region, whichto the which maximum the maximum output power output is directlypower is related, directly becomes related,large becomes due tolarge the due large-area to the large- RTD. Inarea addition, RTD. In the addition, large surface the large area ofsurface the cavity area reducesof the cavity the conduction reduces the loss, conduction which is loss, also effectivewhich is for also high effective output power.for high Figure output7b power. shows theFigure results 7b shows of a theoretical the results calculation of a theoretical of thecalculation output power of the as aoutput function power of cavity as a function length andof cavity capacitance length an atd the capacitance MIM structure at the MIM whichstructure connects thewhich bow-tie connects antenna the tobow the-tie cavity antenna for output to the radiation cavity for [83 output]. The conductanceradiation [83]. The of theconductance antenna, as viewed of the fromantenna, the RTD,as viewed is controlled from the by RTD, the MIM is controlled capacitance, by the where MIM the capaci- optimumtance, condition where the for optimum maximum condition output power for maximum mentioned output in the powe previousr mentioned section canin the be previ- achieved.ous Atsection the optimumcan be achieved. MIM capacitance, At the optimum the output MIM powercapacitance, monotonously the output increases power monot- with cavityonously length, increases due towith the cavity increase length, in ∆ Idue. Although to the increase heating in due Δ퐼. to Although the large heating current due to shouldthe carefully large current be considered should carefully in an actual be device,considered an output in an actu poweral device, of 3–5 mWan output at 1 THz power of is theoretically3–5 mW at expected 1 THz is with theoretically a cavity lengthexpected of 50–70with a µcavitym, even length for aof single 50–70 oscillator.μm, even for a Figuresingle7b shows oscillator. only the Figure output 7b powershows fromonly thethe bow-tieoutput power antenna, from and the does bow not-tie include antenna, and radiationdoes from not include the open radiation surfaces from of the the rectangular open surfaces cavity. of the The rectangular radiation from cavity. these The two radiation open surfacesfrom these ofthe two cavity open cancelsurfaces each of otherthe cavity out in cancel the direction each other directly out in above the direction the cavity, directly as theyabove are in the an cavity, anti-phase as they relationship. are in an anti The-phase fabrication relationship. of this structureThe fabrication is also of currently this structure in progress.is also Thecurrently oscillation in progress. frequency The cannot oscillation be as frequency high as that cannot of the be cylindrical-cavity as high as that of the structurecylindrical mentioned-cavity above, structure as an extremelymentioned narrow above, strip as an of extremely the RTD mesa narrow is required strip of forthe RTD high frequency,mesa is required but this for structure high frequency, is suitable but for highthis structure output power is suitable with afor single high element.output power Figurewith 8a singleshows element a simpliﬁed. structure of an RTD oscillator integrated with the slot antenna proposed for easy fabrication [84]. As this structure does not have the MIM layers needed for the previous oscillator in Figure3a, the requirements of the lithography pro- cesses are reduced, such that the fabrication is easier and can be completed in a short period of time. In this structure, the resistor for stabilization made by the InGaAs layer is located inside the slot antenna, in contrast to the previous structure shown in Figure3a . Similar to the resistance of the antenna electrode discussed in the previous section, which is connected in series with the inductance Lslot of the slot antenna, the conductance caused by the resis- 2 tor for stabilization, as viewed from the RTD, is approximately given by Rs/(ωosc Lslot) , where Rs is the value of the resistor for stabilization. The conductance, as viewed from Sensors 2021, 21, 1384 9 of 19

the RTD, is much smaller than the real conductance, 1/Rs. Thus, it is possible to design Sensors 2021, 21, x FOR PEER REVIEW 2 9 of 19 Rs to satisfy Rs/(ωosc Lslot) GRTD < 1/Rs , such that the resistor for stabilization can suppress the parasitic oscillations at low frequency without large loss, in terms of the oscillation frequency.

(a) (b) Figure 7. RTD oscillator integrated with rectangular cavity and bow-tie antenna for high output power [35, 83]: (a) Oscil- SensorsFigure 2021, 7.21,RTD x FOR oscillator PEER REVIEW integrated with rectangular cavity and bow-tie antenna for high output power [35,83]: (a) Oscillator10 of 19 lator structure, and (b) calculation of output power as a function of cavity length and capacitance at the metal-insulator- structure, and (b) calculation of output power as a function of cavity length and capacitance at the metal-insulator-metal metal (MIM) structure connecting bow-tie and cavity. (MIM) structure connecting bow-tie and cavity. Figure 8 shows a simplified structure of an RTD oscillator integrated with the slot antenna proposed for easy fabrication [84]. As this structure does not have the MIM layers needed for the previous oscillator in Figure 3a, the requirements of the lithography pro- cesses are reduced, such that the fabrication is easier and can be completed in a short period of time. In this structure, the resistor for stabilization made by the InGaAs layer is located inside the slot antenna, in contrast to the previous structure shown in Figure 3a. Similar to the resistance of the antenna electrode discussed in the previous section, which is connected in series with the inductance 퐿푠푙표푡 of the slot antenna, the conductance caused by the resistor for stabilization, as viewed from the RTD, is approximately given 2 by 푅푠⁄(휔표푠푐퐿푠푙표푡(a)) , where 푅푠 is the value of( bthe) resistor for stabilization. The conductance, as viewed from the RTD, is much smaller than the real conductance, 1⁄푅 . Thus, it Figure 8. Simplified structure of RTD oscillator integrated with slot antenna. Reprinted with permission from Reference푠 Figureis possible 8. Simpliﬁed to design structure 푅 to of satisfy RTD oscillator 푅 ⁄(휔 integrated퐿 )2 ≪ with퐺 slot< antenna.1⁄푅 , such RTD that oscillator the resistor (a) sim- [84]. Copyright (2020) Springer Nature. 푠 푠 표푠푐 푠푙표푡 푅푇퐷 푠 pliﬁedfor stabilization steucture of can RTD suppress oscilator the integrated parasitic with oscillations slot antenna; at low (b frequency) SEM image without of the large fabricated loss, airbridge. Reprinted with permission from Reference [84]. Copyright (2020) Springer Nature. in4. Frequencyterms of the Tuning, oscillation Spectral frequency. Narrowing, and Different Polarizations In a preliminary experiment, oscillation up to 740 GHz has been obtained for a 30 InFrequency a preliminary and spectrum experiment, control oscillation are im upportant to 740 for GHz various has been applications, obtained such for a as 30 spec-µm- μm-long antenna, with output power of approximately 10 and 1 μW at around 600 and longtroscopy. antenna, In the with RTD output oscillators power of mentioned approximately above, 10 andthe oscillation 1 µW at around frequency 600 and was 740 almost GHz, 740 GHz, respectively [84]. Higher oscillation frequencies are expected with shorter an- respectivelyfixed at the [frequency84]. Higher determined oscillation by frequencies the structure, are expected except for with small shorter changes antennas. (1–5%) These with tennas. These characteristics were comparable to the previous structure, although the up- characteristicsbias voltage [85 were]. As comparablea voltage-controlled to the previous frequency structure,-tunable oscillator although (VCO), the upper an oscillator limit of per limit of oscillation frequency may be slightly lower than that of the previous structure, oscillationintegrated frequencywith a varactor may bediode slightly in the lower slot antenna than that has of been the reported previous [ structure,86–88]. The due varac- to due to the effect of the resistor for stabilization. As this structure does not require compli- thetor effectdiode ofis themade resistor of a pn for junction stabilization. of InGaAs, As this where structure the oscillation does not frequency require complicated can be elec- cated three-dimensional integration, it can easily be extended to various planar structures, three-dimensionaltrically varied by integration,changing its it capacitance can easily be and extended resistance to various using reverse planar structures,[86] or forward such assuch high-density as high-density arrays, arrays, metamaterials metamaterials including including RTDs, RTDs, and so and on. so on. [87] bias voltage. By optimizing the areas of the varactor diode and RTD, a frequency

4.change Frequency of about Tuning, 100 GHz Spectral was Narrowing, obtained. Moreover, and Different using Polarizations an array configuration of frequency-tunable oscillators with different frequency ranges, a wide frequency change has Frequency and spectrum control are important for various applications, such as been reported [88]. Figure 9 shows an array of frequency-tunable RTD oscillators and its spectroscopy. In the RTD oscillators mentioned above, the oscillation frequency was application to absorbance measurement [47]. Using the frequency change of 410–970 GHz almost ﬁxed at the frequency determined by the structure, except for small changes (1–5%) in the RTD oscillators, the absorbance of allopurinol was measured. The results agreed with bias voltage [85]. As a voltage-controlled frequency-tunable oscillator (VCO), an with that of conventional THz time-domain spectroscopy (TDS), within the resolutions of oscillator integrated with a varactor diode in the slot antenna has been reported [86–88]. these two measurements. The varactor diode is made of a pn junction of InGaAs, where the oscillation frequency By integration of the frequency-tunable RTD oscillator with a detector through a transmission line and a space for a droplet of specimen, a microchip which can be used for spectroscopy is expected. Fast measurements may be possible, even with low output power of the oscillator, due to the short distance between the source and detector. Ne- glecting the transmission loss between the source and detector, the output power required for the oscillator is estimated to be 푃 = SNR ∙ NEP/√2휋푇 ∼ 0.6 μW, with the signal-to- noise power ratio (SNR) at the detector (dynamic range) of 30 dB, the time constant T of 1 ms, and the noise equivalent power (NEP) of 50 pW/Hz1/2, which is typical for Schottky barrier diodes (SBDs).

(a) (b)

Sensors 2021, 21, x FOR PEER REVIEW 10 of 19

(a) (b) Figure 8. Simplified structure of RTD oscillator integrated with slot antenna. Reprinted with permission from Reference [84]. Copyright (2020) Springer Nature.

4. Frequency Tuning, Spectral Narrowing, and Different Polarizations Frequency and spectrum control are important for various applications, such as spectroscopy. In the RTD oscillators mentioned above, the oscillation frequency was almost fixed at the frequency determined by the structure, except for small changes (1–5%) with bias voltage [85]. As a voltage-controlled frequency-tunable oscillator (VCO), an oscillator integrated with a varactor diode in the slot antenna has been reported [86–88]. The varactor diode is made of a pn junction of InGaAs, where the oscillation frequency can be electrically varied by changing its capacitance and resistance using reverse [86] or forward [87] bias voltage. By optimizing the areas of the varactor diode and RTD, a frequency change of about 100 GHz was obtained. Moreover, using an array configuration of frequency-tunable oscillators with different frequency ranges, a wide frequency change has been reported [88]. Figure 9 shows an array of frequency-tunable RTD oscillators and its Sensors 2021, 21, 1384 application to absorbance measurement [47]. Using the frequency change of 410–97010 GHz of 19 in the RTD oscillators, the absorbance of allopurinol was measured. The results agreed with that of conventional THz time-domain spectroscopy (TDS), within the resolutions of these two measurements. can beBy electrically integration varied of the by frequency changing-tunable its capacitance RTD oscillator and resistance with a detec usingtor reverse through [86 a] transmissionor forward [87 line] bias and voltage. a space Byfor optimizinga droplet of the specimen, areas of a the microchip varactor which diode can and be RTD, used a frequency change of about 100 GHz was obtained. Moreover, using an array conﬁguration for spectroscopy is expected. Fast measurements may be possible, even with low output of frequency-tunable oscillators with different frequency ranges, a wide frequency change power of the oscillator, due to the short distance between the source and detector. Ne- has been reported [88]. Figure9 shows an array of frequency-tunable RTD oscillators and its glecting the transmission loss between the source and detector, the output power required application to absorbance measurement [47]. Using the frequency change of 410–970 GHz for the oscillator is estimated to be 푃 = SNR ∙ NEP/√2휋푇 ∼ 0.6 μW, with the signal-toin the RTD oscillators, the absorbance of allopurinol was measured. The results agreed noise power ratio (SNR) at the detector (dynamic range) of 30 dB, the time constant T of 1 with that of conventional THz time-domain spectroscopy (TDS), within the resolutions of ms, and the noise equivalent power (NEP) of 50 pW/Hz1/2, which is typical for Schottky these two measurements. barrier diodes (SBDs).

(a) (b)

Figure 9. Frequency-tunable RTD oscillator and application to absorbance measurement: (a) Array of frequency-tunable oscillators integrated with varactor diodes, and (b) absorbance measured by RTD oscillators together with the result of THz time-domain spectroscopy (TDS) measurement. Copyright (2017) The Japan Society of Applied Physics [47].

By integration of the frequency-tunable RTD oscillator with a detector through a transmission line and a space for a droplet of specimen, a microchip which can be used for spectroscopy is expected. Fast measurements may be possible, even with low output power of the oscillator, due to the short distance between the source and detector. Neglecting the transmission loss between the source and√ detector, the output power required for the oscillator is estimated to be P = SNR·NEP/ 2πT ∼ 0.6 µW, with the signal-to-noise power ratio (SNR) at the detector (dynamic range) of 30 dB, the time constant T of 1 ms, and the noise equivalent power (NEP) of 50 pW/Hz1/2, which is typical for Schottky barrier diodes (SBDs). For the spectral line width, a full width at the half maximum (FWHM) of about 10 MHz has been reported in an RTD oscillator with output power of about 1 µW[89]. The line width is determined by the phase noise caused by the shot noise of the RTD, which 2 is calculated as ∆ f = (π/4) pn∆ fr /Pout in FWHM [90], where pn is the noise power density per frequency included in the output power Pout, and ∆ fr is the line width in FWHM of the resonator system with the RTD capacitance and the slot antenna and without NDC, which is expressed as ∆ fr = fOSC/Q, using the oscillation frequency fOSC and the Q factor of this resonator system. From this equation, the line width can be reduced by an 2 increase in the product PoutQ . Spectral narrowing with high Q factor by an RTD efﬁciently coupled to a photonic-crystal cavity has been reported [91]. Spectral narrowing by a phase-locked loop (PLL) has also been reported [92,93]. Figure 10a shows a PLL system for a frequency-tunable RTD oscillator. By converting the output of the RTD oscillator to a low-frequency signal by heterodyne detection, extracting the phase noise by mixing the converted signal with the reference signal, and feeding it back to the varactor diode in the oscillator, a line width as narrow as 1 Hz has been achieved, as shown in Figure 10b). Sensors 2021, 21, x FOR PEER REVIEW 11 of 19

For the spectral line width, a full width at the half maximum (FWHM) of about 10 MHz has been reported in an RTD oscillator with output power of about 1 μW [89]. The line width is determined by the phase noise caused by the shot noise of the RTD, which is 2 calculated as Δ푓 = (휋⁄4)(푝푛Δ푓푟 ⁄푃표푢푡) in FWHM [90], where 푝푛 is the noise power density per frequency included in the output power 푃표푢푡, and Δ푓푟 is the line width in FWHM of the resonator system with the RTD capacitance and the slot antenna and without NDC, which is expressed as Δ푓푟 = 푓푂푆퐶⁄푄, using the oscillation frequency 푓푂푆퐶 and the Q factor of this resonator system. From this equation, the line width can be reduced by an increase 2 in the product 푃표푢푡푄 . Spectral narrowing with high Q factor by an RTD efficiently coupled to a photonic-crystal cavity has been reported [91]. Spectral narrowing by a phase-locked loop (PLL) has also been reported [92, 93]. Fig- ure 10a shows a PLL system for a frequency-tunable RTD oscillator. By converting the output of the RTD oscillator to a low-frequency signal by heterodyne detection, extracting Sensors 2021, 21, 1384 the phase noise by mixing the converted signal with the reference signal,11 and of 19 feeding it back to the varactor diode in the oscillator, a line width as narrow as 1 Hz has been achieved, as shown in Figure 10b).

(a)

(b)

Figure 10. Phase-locked loop (PLL) system and spectral narrowing of frequency-tunable RTD oscillator. Copyright (2018) The IEICE of Japan [93]: (a) Block diagram of the PLL system—LNA:

low-noise ampliﬁer, ATT: attenuator, SG: signal generator, and (b) oscillation spectra under the PLL at various oscillation frequencies—RBW: resolution band width, VBW: video band width.

Figure 10. Phase-locked loop (PLL) system and spectral narrowing of frequency-tunable RTD oscillator. Copyright (2018) The IEICE of Japan [93]: (a) Block diagram of the PLL system—LNA: low-noise amplifier, ATT: attenuator, SG: signal generator, and (b) oscillation spectra under the PLL at various oscillation frequencies—RBW: resolution band width, VBW: video band width.

Reduction in the spectral line width of an RTD oscillator by external sub-harmonic injection locking has also been reported [94]. External injection locking is also useful for phase control of the output. Under injection locking, the oscillation frequency of the RTD oscillator is locked to that of the external injection, even if the free-running frequency of the RTD oscillator is slightly different from that of the injection. An operation changing the frequency under the free-running condition (i.e., changing the bias voltage of the varactor diode or RTD) results in a phase change of the output under the locked condition. Utilizing this property for the injection locking of an uncoupled array of RTD oscillators, a phased array can be constructed, which has the beam steering function of the output which is useful for various THz applications. Phase changing of RTD oscillators by fundamental and sub-harmonic injection locking has been theoretically analyzed [95]. Sensors 2021, 21,In 1384 the above oscillators, the output is linearly polarized. An RTD oscillator integrated 12 of 19 with a radial line slot array (RLSA), which radiates a circularly polarized wave, has also been reported [64]. The structure of this oscillator is shown in Figure 11. The RTD is located at one of the cross-slot antennas, as shown in Figure 11a, and radiates the output wave into the substratewave side. into The the radiated substrate wave side. propagates The radiated along wave the propagatessubstrate and along is the substrate and is emitted from the elementsemitted of the from RLSA the around elements the of RTD the RLSA in the aroundupward the direction, RTD in theas shown upward direction, as shown in Figure 11b. Each elementin Figure of 11theb. RLSA Each elementradiates ofthe the circularly RLSA radiates polarized the wave. circularly These polarized wave. These elements are arrangedelements concentrically are arranged and radiate concentrically output waves and in radiatephase, as output shown waves in Fig- in phase, as shown in ure 11c, resulting in highFigure directivity 11c, resulting without in a highlens. directivityAn axial ratio without of 2.2 a dB lens. for Anthe axialcircu- ratio of 2.2 dB for the circularly polarized output and the directivity of 15 dBi were obtained at 500 GHz. The larly polarized output and the directivity of 15 dBi were obtained at 500 GHz. The circu- circularly polarized output is useful in preventing the inﬂuence of external feedback that larly polarized output is useful in preventing the influence of external feedback that causes ﬂuctuations of the oscillation characteristics of the RTD [96]. By changing the causes fluctuations of the oscillation characteristics of the RTD [96]. By changing the ar- arrangement of the elements from concentric to spiral, the radiation of a vortex wave, rangement of the elements from concentric to spiral, the radiation of a vortex wave, which which can be used for multiplex wireless communications with different rotation numbers, can be used for multiplex wireless communications with different rotation numbers, has has also been obtained [97]. also been obtained [97].

Sensors 2021, 21, x FOR PEER REVIEW 13 of 19

(a)

(b) (c)

Figure 11. RTD oscillator integratedFigure with 11. radialRTD oscillator line slot integrated array (RLSA) with for radial radiation line slot of circularlyarray (RLSA) polarized for radiation wave [of64 ]:circularly polarized wave [64]: (a) (a) Top view, (b) cross-sectional view,Top togetherview, (b) with cross propagation-sectional view, of output together from with RTD, propagation and (c) top of view, output together from withRTD, phase and (c) top view, together with phase distribution of the propagating wavedistribution in the substrate. of the propagating wave in the substrate.

5. Applications 5. Applications Basic research into various applicationsBasic resear of RTDch into oscillators various has applications begun, including of RTD for oscillators imag- has begun, including for ing [36], sensors [37], linear encodersimaging [38 [],36 communication], sensors [37], [linear39–42 ,encoders50], and radars [38], communication[43–46], in [39–42, 50], and radars addition to spectroscopy [47] shown[43–46 in], thein addition previous to section. spectroscopy It is a future [47] taskshown to develop in the previous vari- section. It is a future task ous applications of RTD oscillators.to develop Here, wevarious briefly applications introduce recent of RTD applications, oscillators. especially Here, we briefly introduce recent ap- with respect to communicationplications, and radar. especially with respect to communication and radar. As the output of RTD oscillatorsAs canthe easilyoutput be of intensity-modulated RTD oscillators can byeasily direct be modula-intensity-modulated by direct modu- tion (i.e., superposition of a signallation on (i.e., the superpositi bias voltage),on simpleof a signal high-capacity on the bias THz voltage), wireless simple high-capacity THz wire- communications are possible. Theless uppercommunications limit of the directare possible. modulation The upper frequency limit of of 30 the GHz direct modulation frequency of has been reported [98], which30 is limitedGHz has by been the capacitance reported [98 of], the which external is limited circuit by to the impose capacitance of the external circuit the modulation signal onto theto RTD. impose Simple the on-offmodulation keying signal wireless onto data the transmissions RTD. Simple haveon-off keying wireless data trans- −4 been reported with a data ratemissions of 44 Gbps have and been an errorreported rate with of 5 × a data10 ratebelow of 44 the Gbps forward and an error rate of 5 × 10–4 below error correction (FEC) limit,the and forward 25 Gbps error without correction error (FEC) at 650 limit, GHz and [39]. 25 Preliminary Gbps without error at 650 GHz [39]. Pre- experiments on transmissionsliminary with frequency experiments and polarizationon transmissions multiplexing with frequency using RTDand polarization multiplexing us- oscillators have also been reporteding RTD [40 ].oscillators By integrating have oscillatorsalso been havingreported two [40 orthogonal]. By integrating oscillators having two orthogonal polarizations and two frequencies of 500 and 800 GHz on the same substrate, transmission of 2 × 28 Gbps was obtained, with error below FEC limit in both the frequency and polarization multiplexing. These are also simple on–off keying data transmissions. Figure 12 shows the oscillator chip for frequency and polarization multiplexing, the diagram of the frequency multiplexing, and the transmission result. By improving the external circuit around the RTD for the modulation signal, higher data rates are expected. A transmission experiment using an RTD oscillator with a circularly polarized wave has also been reported [64]. Although the data rate was still low (1 Gbps), it was shown that the error rate was insensitive to oscillator rotation. Wireless transmission using RTDs as detectors has also been reported [49–50]. The RTD is expected to have a high sensitivity in THz detection, due to the strong non-linearity in the I–V curve, which is the same principle as the detection in SBD (although bias voltage must be applied to use the strong non-linearity in RTD). Other than the detection using such non-linearity, a self-homodyne THz detection mode has recently been reported [50, 51]. In this mode, the THz signal is detected by an RTD oscillator which is oscillating near the frequency of the irradiated signal. Through this irradiation, the RTD oscillator is injection-locked and a signal with the homodyne detection is obtained. Through this operation, a low value of NEP (7.7 pW/Hz1/2) has been obtained [51].

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polarizations and two frequencies of 500 and 800 GHz on the same substrate, transmission of 2 × 28 Gbps was obtained, with error below FEC limit in both the frequency and polarization multiplexing. These are also simple on–off keying data transmissions. Figure 12 shows the oscillator chip for frequency and polarization multiplexing, the diagram of the frequency multiplexing, and the transmission result. By improving the external circuit around the RTD for the modulation signal, higher data rates are expected. A transmis- Sensors 2021, 21, x FOR PEER REVIEW 14 of 19 sion experiment using an RTD oscillator with a circularly polarized wave has also been reported [64]. Although the data rate was still low (1 Gbps), it was shown that the error rate was insensitive to oscillator rotation.

(a) (b)

FigureFigure 12. 12. RTDRTD oscillators oscillators with with different different frequencies frequencies and and polarizations polarizations integrated integrated into into one one chip chip for for wireless wireless transmission transmission with frequency and polarization multiplexing. Copyright (2017) IEEE. Reprinted with permission from [40]: (a) Schematic with frequency and polarization multiplexing. Copyright (2017) IEEE. Reprinted with permission from [40]: (a) Schematic structure of the integrated chip, and (b) diagram of the transmission with frequency multiplexing and the resulting eye structure of the integrated chip, and (b) diagram of the transmission with frequency multiplexing and the resulting eye diagrams—PPG: pulse pattern generator, ED: error detector, BER: bit-error rate, BPF: band-pass filter. diagrams—PPG: pulse pattern generator, ED: error detector, BER: bit-error rate, BPF: band-pass filter. The application of RTD oscillators to THz radar has also been studied [43–46]. The Wireless transmission using RTDs as detectors has also been reported [49,50]. The THzRTD radar is expected has the to advantage have a high that sensitivity it can be used in THz in detection,environments due towith the poor strong visibility, non-linearity due toin the the transparency I–V curve, which of THz is the waves. same 3D principle transparent as the imaging detection is in also SBD possible (although by combining bias voltage THzmust radar be applied and two to-dimensional use the strong (2D) non-linearity imaging systems. in RTD). Other than the detection using suchFigure non-linearity, 13 shows a self-homodynea simplified schematic THz detection diagram mode of hasa system recently and been measurement reported [50 re-,51]. sultsIn this of THz mode, radar the THzusing signal an RTD is detected oscillator by [43, an 44 RTD]. This oscillator system which uses the is oscillating amplitude near-mod- the ulatedfrequency continuous of the irradiatedwave (AMCW) signal. method. Through In this Figure irradiation, 13a, the theoutput RTD of oscillator the RTD is is injection- ampli- tudelocked-modulated and a signal by superimposing with the homodyne a sinusoidal detection signal is obtained. on the bias Through voltage, this which operation, is then a irradiatedlow value onto of NEP an object. (7.7 pW/Hz The reflected1/2) has wave been from obtained the object [51]. is received and demodulated by SBD.The The application time of flight of RTD (ToF) oscillators of the THz to wave THz radar from hasRTD also to SBD been is studied determined [43–46 by]. the The phaseTHz radardifference has the between advantage the demodulated that it can be and used reference in environments signals, from with which poor visibility,the distance due toto the the object transparency is obtained. of THz waves. 3D transparent imaging is also possible by combining THz radar and two-dimensional (2D) imaging systems. Figure 13 shows a simplified schematic diagram of a system and measurement results of THz radar using an RTD oscillator [43,44]. This system uses the amplitude-modulated continuous wave (AMCW) method. In Figure 13a, the output of the RTD is amplitude- modulated by superimposing a sinusoidal signal on the bias voltage, which is then irradiated onto an object. The reflected wave from the object is received and demodulated by SBD. The time of flight (ToF) of the THz wave from RTD to SBD is determined by the phase difference between the demodulated and reference signals, from which the distance to the object is obtained.

(a) (b) Figure 13. THz radar system using RTD oscillator [43, 44]: (a) Simplified schematic diagram of THz radar system using RTD oscillator with the amplitude-modulated continuous wave (AMCW)—SG: signal generator, and (b) error evaluated for the distance measurement with the AMCW using two modulation frequencies.

In this method, when the phase difference between the demodulated and reference signals exceeds 2π, the number of periods included in the phase difference cannot be extracted. To solve this problem, two slightly different frequencies are used for modulation

Sensors 2021, 21, x FOR PEER REVIEW 14 of 19

(a) (b) Figure 12. RTD oscillators with different frequencies and polarizations integrated into one chip for wireless transmission with frequency and polarization multiplexing. Copyright (2017) IEEE. Reprinted with permission from [40]: (a) Schematic structure of the integrated chip, and (b) diagram of the transmission with frequency multiplexing and the resulting eye diagrams—PPG: pulse pattern generator, ED: error detector, BER: bit-error rate, BPF: band-pass filter.

The application of RTD oscillators to THz radar has also been studied [43–46]. The THz radar has the advantage that it can be used in environments with poor visibility, due to the transparency of THz waves. 3D transparent imaging is also possible by combining THz radar and two-dimensional (2D) imaging systems. Figure 13 shows a simplified schematic diagram of a system and measurement results of THz radar using an RTD oscillator [43, 44]. This system uses the amplitude-modulated continuous wave (AMCW) method. In Figure 13a, the output of the RTD is amplitude-modulated by superimposing a sinusoidal signal on the bias voltage, which is then irradiated onto an object. The reflected wave from the object is received and demodulated Sensors 2021, 21, 1384 by SBD. The time of flight (ToF) of the THz wave from RTD to SBD is determined by14 ofthe 19 phase difference between the demodulated and reference signals, from which the distance to the object is obtained.

(a) (b)

Figure 13.13. THzTHz radarradar systemsystem using using RTD RTD oscillator oscillator [43 [43,,44 ]:44 (a]:) ( Simpliﬁeda) Simplified schematic schematic diagram diagram of THz of THz radar radar system system using using RTD oscillatorRTD oscillator with thewith amplitude-modulated the amplitude-modulated continuous continuous wave (AMCW)—SG:wave (AMCW) signal—SG: signal generator, generator, and (b )and error (b evaluated) error evaluated for the for the distance measurement with the AMCW using two modulation frequencies. distance measurement with the AMCW using two modulation frequencies.

In this method,method, whenwhen the the phase phase difference difference between between the the demodulated demodulated and and reference reference sig- signalsnals exceeds exceeds 2π, 2π, the numberthe number of periods of periods included included in the in phase the phase difference difference cannot cannot be extracted. be extracted.To solve To this solve problem, this problem, two slightly two differentslightly different frequencies frequencies are used forare modulationused for modulation [43]. The phase difference is measured for each frequency, in order to extract the number of the periods included in the phase difference. By utilizing the fact that the period number must be an integer, the error in this number caused by noise is totally removed and high accuracy in the phase evaluation can be obtained. Furthermore, the oscilloscope in Figure 13a can be replaced with an In-phase/Quadrature (IQ) demodulation system, in order to obtain an accurate phase difference [44]. In the IQ demodulation, the demodulated signal from the SBD is separately mixed with the reference signal and its 90 degree-shifted signal, and two orthogonal components of the mixing output are obtained. The phase difference is calculated from the arctangent of the amplitude ratio of these two components. By introducing the above improvements, distance measurement with an error (standard deviation) of 0.063 mm has been achieved for the carrier frequency of 520 GHz, as shown in Figure 13b [44]. The system described above does not utilize the phase difference of the THz wave itself but, instead, the phase difference of the subcarriers superimposed on the THz wave of the RTD output. The features of THz waves can be used as the carrier. This method is very useful for RTD oscillators in which oscillation characteristics, such as frequency, are easily affected by the external feedback [96]. The subcarrier modulation method can be extended to other radar systems, such as the frequency-modulated continuous wave (FMCW) radar. A subcarrier FMCW radar using an RTD oscillator and a preliminary experiment for the distance measurement of two targets have been reported [45]. As another extension of subcarrier modulation, a method for measuring the distances of multiple targets has been proposed [46], the principle of which is similar to that of THz optical coherent tomography (OCT) [99]. By changing the modulation frequency (subcarrier frequency) fm in Figure 13a, the demodulated signal is obtained as a function of fm. Then, the demodulated signal is decomposed to two orthogonal components by IQ demodulation. For example, for a single target, as shown in Figure 13a, the demodulated signal at the SBD is written as V cos[2π fm(t − τ)], where V is the amplitude of the demodulated signal reﬂected from the object and τ is the time delay of the demodulation signal to the reference signal, including propagation times in the space and cables. This signal is decomposed to two components, VI cos(2π fmt) + VQ sin(2π fmt), where VI ( fm) = V cos(2π fmτ) and Sensors 2021, 21, 1384 15 of 19

VQ( fm) = V sin(2π fmτ). VI ( fm) and VQ( fm) are extracted by the IQ demodulation as a function of fm. By calculating the inverse Fourier transform of the complex function VI ( fm) − iVQ( fm), the distribution of the target positions can be obtained. For the above −i2π fmτ single-target case, VI ( fm) − iVQ( fm) = Ve and the inverse Fourier transform gives Vδ(t − τ), assuming that the dependence of V on fm is weak. Thus, τ can be extracted and the position of the target found. For multiple targets, a superposition of this form with different values of τ is obtained, where the distribution of the target position is found. As the bandwidth of fm is ﬁnite in an actual measurement, the result of the inverse Fourier transform for a single target is not a δ-function but a pulse having a ﬁnite width approximately given by 1/( fmax − fmin), where fmax and fmin are the maximum and minimum values of fm, respectively. This pulse width gives the resolution (in τ). In a preliminary proof-of-concept experiment using this method, the distances of two targets were measured in the range of 20–200 mm, with an error (standard deviation) of approximately 0.6–2.5 mm for fmin and fmax of 3 and 18 GHz, respectively [46]. The error and resolution can be improved by an increase of modulation bandwidth and the use of RTD oscillators with high output power for high signal-to-noise ratio.

6. Conclusions Recent developments and some applications of THz sources using RTD oscillators were described. A major feature of these sources is their compactness. Studies focused on high frequency, high output power, and various functionalities were discussed. An important issue to address in the future is obtaining high output power (at least more than 1 mW) with high efficiency at high frequency. In addition, advanced functions which are required for various applications, such as beam steering (which is important for communication and radar systems), must be investigated. On the other hand, there are some basic characteristics that are not yet fully understood, such as the THz response of electrons in RTD. We hope that future studies will make significant progress in this field, and that the range of various applications will be expanded.

Author Contributions: Conceptualization, M.A. and S.S.; Methodology, S.S.; Writing—Original Draft Preparation, M.A.; Writing—Review and Editing, M.A. and S.S.; Project Administration, M.A.; Funding Acquisition, M.A. and S.S. All authors have read and agreed to the published version of the manuscript. Funding: This work was funded by a JSPS scientific grant in aid (16H06292), JST-ACCEL program (JPMJMI17F2), JST-CREST program (JPMJCR1534), and JST Industry–Academia Collaborative R&D program. Institutional Review Board Statement: Not applicable. Informed Consent Statement: Not applicable. Acknowledgments: The authors thank Y. Suematsu, K. Furuya, and S. Arai of the Tokyo Institute of Technology for their continuous encouragement. The authors also thank Y. Miyamoto, N. Nishiyama, M. Watanabe, A. Dobroiu, H. Tanaka, and X. Yu of the Tokyo Institute of Technology for their stimulating discussions. Conflicts of Interest: The authors declare no conflict of interest.

References 1. Tonouchi, M. Cutting-edge terahertz technology. Nat. Photonics 2007, 1, 97–105. [CrossRef] 2. Nagatsuma, T. Terahertz technologies: Present and future. IEICE Electron. Express 2011, 8, 1127–1142. [CrossRef] 3. Hangyo, M. Development and future prospects of terahertz technology. Jpn. J. Appl. Phys. 2015, 54, 120101. [CrossRef] 4. Komiyama, S. Far-infrared emission from population-inverted hot-carrier system in p-Ge. Phys. Rev. Lett. 1982, 48, 271–274. [CrossRef] 5. Köhler, R.; Tredicucci, R.; Beltram, F.; Beere, H.E.; Linfeld, E.H.; Davies, A.G.; Ritchie, D.A.; Iotti, R.C.; Rossi, F. Terahertz semiconductor heterostructure laser. Nature 2002, 417, 156–159. [CrossRef] 6. Williams, B.S. Terahertz quantum-cascade lasers. Nat. Photonics 2007, 1, 517–525. [CrossRef] Sensors 2021, 21, 1384 16 of 19

7. Khalatpour, A.; Paulsen, A.K.; Deimert, C.; Wasilewski, Z.R.; Hu, Q. High-power portable terahertz laser systems. Nat. Photonics 2021, 15, 16–20. [CrossRef] 8. Li, L.H.; Chen, L.; Freeman, J.R.; Salih, M.; Dean, P.; Davies, A.G.; Linfield, E.H. Multi-watt high-power THz frequency quantum cascade laser. Electron. Lett. 2017, 53, 799–800. [CrossRef] 9. Jin, Y.; Reno, J.L.; Kumar, S. Phase-locked terahertz plasmonic laser array with 2W output power in a single spectral mode. Optica 2020, 7, 708–715. [CrossRef] 10. Belkin, M.A.; Capasso, F.; Belyanin, A.; Sivco, D.L.; Cho, A.Y.; Oakley, D.C.; Vineis, C.J.; Turner, G.W. Terahertz quantum-cascade- laser source based on intracavity difference-frequency generation. Nat. Photonics 2007, 1, 288–292. [CrossRef] 11. Lu, Q.; Razeghi, M. Recent advances in room temperature, high-power terahertz quantum cascade laser sources based on difference-frequency generation. Photonics 2016, 3, 42. [CrossRef] 12. Fujita, K.; Jung, S.; Jiang, Y.; Kim, J.H.; Nakanishi, A.; Ito, A.; Hitaka, M.; Edamura, T.; Belkin, M.A. Recent progress in terahertz difference-frequency quantum cascade laser sources. Nanophotonics 2018, 7, 1795–1817. [CrossRef] 13. Fujita, K. Room temperature terahertz nonlinear quantum cascade lasers and their applications. In Proceedings of the 45th International Conference on Infrared, Millimeter, and Terahertz Waves, Buffalo, NY, USA, 8–13 November 2020. 14. Nishizawa, J.; Plotka, P.; Kurabayashi, T.; Makabe, H. 706-GHz GaAs CW fundamental-mode TUNNETT diodes fabricated with molecular layer epitaxy. Phys. Stat. Sol. 2008, 5, 2802–2804. [CrossRef] 15. Eisele, H.; Kamoua, R. Submillimeter-wave InP Gunn devices. IEEE Trans. Microw. Theory Technol. 2004, 52, 2371–2378. [CrossRef] 16. Eisele, H. 480 GHz oscillator with an InP Gunn device. Electron. Lett. 2010, 46, 422–423. [CrossRef] 17. Urtega, M.; Griffith, Z.; Seo, M.; Hacker, J.; Rodwell, M.J.W. InP HBT technologies for THz integrated circuits. Proc. IEEE 2017, 105, 1051–1067. [CrossRef] 18. Tousi, Y.M.; Momeni, O.; Afshari, E. A novel CMOS high-power terahertz VCO based on coupled oscillators: Theory and implementation. IEEE J. Solid State Circuits 2012, 47, 3032–3042. [CrossRef] 19. Pfeiffer, U.R.; Zao, Y.; Grzyb, J.; Hadi, R.A.; Sarmah, N.; Förster, W.; Rücker, H.; Heinemann, B. A 0.53 THz reconfigurable source module with up to 1 mW radiated power for diffuse illumination in terahertz imaging applications. IEEE J. Solid State Circuits 2014, 49, 2938–2950. [CrossRef] 20. Steyaert, W.; Reynaert, P. A 0.54 THz signal generator in 40 nm bulk CMOS with 22 GHz tuning range and integrated planar antenn. IEEE J. Solid State Circuits 2014, 49, 1617–1625. [CrossRef] 21. Hu, Z.; Mehmet, M.; Han, R. High-power radiation at 1 THz in silicon: A fully scalable array using a multi-functional radiating mesh structure. IEEE J. Solid State Circuits 2018, 53, 1313–1327. [CrossRef] 22. Kakeya, I.; Wang, H. Terahertz-wave emission from Bi2212 intrinsic Josephson junctions: A review on recent progress. Supercond. Sci. Technol. 2016, 29, 073001. [CrossRef] 23. Ono, Y.; Minami, H.; Kuwano, G.; Kashiwagi, T.; Tsujimoto, M.; Kadowaki, K.; Klemm, R.A. Superconducting emitter powered at 1.5 Terahertz by an external resonator. Phys. Rev. Appl. 2020, 13, 064026. [CrossRef] 24. Brown, E.R.; Sonderstrom, J.R.; Parker, C.D.; Mahoney, L.J.; Molvar, K.M.; McGill, T.C. Oscillations up to 712 GHz in InAs/AISb resonant-tunneling diodes. Appl. Phys. Lett. 1991, 58, 2291–2293. [CrossRef] 25. Reddy, M.; Martin, S.C.; Molnar, A.C.; Muller, R.E.; Smith, R.P.; Siegel, P.H.; Mondry, M.J.; Rodwell, M.J.W.; Kroemer, H.; Allen, S.J. Monolithic Schottky-collector resonant tunnel diode oscillator arrays to 650 GHz. IEEE Electron Device Lett. 1997, 18, 218–221. [CrossRef] 26. Suzuki, S.; Asada, M.; Teranishi, A.; Sugiyama, H.; Yokoyama, H. Fundamental oscillation of resonant tunneling diodes above 1 THz at room temperature. Appl. Phys. Lett. 2010, 97, 242102. [CrossRef] 27. Feiginov, M.; Sydlo, C.; Cojocari, O.; Meissner, P. Resonant-tunnelling-diode oscillators operating at frequencies above 1.1 THz. Appl. Phys. Lett. 2011, 99, 233506. [CrossRef] 28. Koyama, Y.; Sekiguchi, R.; Ouchi, T. Oscillations up to 1.40 THz from resonant-tunneling-diode-based oscillators with integrated patch antennas. Appl. Phys. Express 2013, 6, 064102. [CrossRef] 29. Feiginov, M.; Kanaya, H.; Suzuki, S.; Asada, M. Operation of resonant-tunneling diodes with strong back injection from the collector at frequencies up to 1.46 THz. Appl. Phys. Lett. 2014, 104, 243509. [CrossRef] 30. Lee, J.; Kim, M.; Yang, K. 1.52 THz RTD triple-push oscillator with a µW-level output power. IEEE Trans. THz Sci. Technol. 2016, 6, 336–340. [CrossRef] 31. Al-Khalidi, A.; Alharbi, K.H.; Wang, J.; Morariu, R.; Wang, L.; Khalid, A.; Figueiredo, J.M.L.; Wasige, E. Resonant tunneling diode terahertz sources with up to 1 mW output power in the J-band. IEEE Trans. Terahertz Sci. Technol. 2020, 10, 150–157. [CrossRef] 32. Izumi, R.; Suzuki, S.; Asada, M. 1.98 THz resonant-tunneling-diode oscillator with reduced conduction loss by thick antenna electrode. In Proceedings of the International Conference on Infrared, Millimeter, and Terahertz Waves, Cancun, Mexico, 27 August–1 September 2017. Abstract No. MA3.1. 33. Izumi, R.; Sato, T.; Suzuki, S.; Asada, M. Resonant-tunneling-diode terahertz oscillator with a cylindrical cavity for high-frequency oscillation. AIP Adv. 2019, 9, 085020. [CrossRef] 34. Bezhko, M.; Suzuki, S.; Asada, M. Frequency increase in resonant-tunneling diode cavity-type terahertz oscillator by simulation- based structure optimization. Jpn. J. Appl. Phys. 2020, 59, 032004. [CrossRef] 35. Kobayashi, K.; Suzuki, S.; Han, F.; Tanaka, H.; Fujikata, H.; Asada, M. Analysis of a high-power resonant-tunneling-diode terahertz oscillator integrated with a rectangular cavity resonator. Jpn. J. Appl. Phys. 2020, 59, 050907. [CrossRef] Sensors 2021, 21, 1384 17 of 19

36. Miyamoto, T.; Yamaguchi, A.; Mukai, T. Terahertz imaging system with resonant tunneling diodes. Jpn. J. Appl. Phys. 2016, 55, 032201. [CrossRef] 37. Okamoto, K.; Tsuruda, K.; Diebold, S.; Hisatake, S.; Fujita, M.; Nagatsuma, T. Terahertz sensor using photonic crystal cavity and resonant tunneling diodes. J. Infrared Millim. Terahertz Waves 2017, 38, 1085–1097. [CrossRef] 38. Yamashita, G.; Tsujita, W.; Tsutada, H.; Ma, R.; Wang, P.; Orlik, P.V.; Suzuki, S.; Dobroiu, A.; Asada, M. Terahertz polarimetric Sensing for linear encoder based on resonant-tunneling-diode and CFRP polarizing. In Proceedings of the International Conference on Infrared, Millimeter, and Terahertz Waves, Paris, France, 1–6 September 2019. Abstract No. 4429649. 39. Asada, M.; Suzuki, S. THz oscillators using resonant tunneling diodes and their functions for various applications. In Proceedings of the Workshop in European Microwave Week, Nuremberg, Germany, 8–13 October 2017. Abstract No. WTu-01. 40. Oshima, N.; Hashimoto, K.; Suzuki, S.; Asada, M. Terahertz wireless data transmission with frequency and polarization division multiplexing using resonant-tunneling-diode oscillators. IEEE Trans. Terahertz Sci. Technol. 2017, 7, 593–598. [CrossRef] 41. Wasige, E. Over 10 Gbps mm-wave and THz wireless links. In Proceedings of the Workshop in European Microwave Week, Madrid, Spain, 25–27 September 2018. Abstract No. WTh04-03. 42. Diebold, S.; Nishio, K.; Nishida, Y.; Kim, J.; Tsuruda, K.; Mukai, T.; Fujita, M.; Nagatsuma, T. High-speed error-free wireless data transmission using a terahertz resonant tunneling diode transmitter and receiver. Electron. Lett. 2016, 52, 1999–2001. [CrossRef] 43. Dobroiu, A.; Wakasugi, R.; Shirakawa, Y.; Suzuki, S.; Asada, M. Absolute and precise terahertz-wave radar based on an amplitude-modulated resonant-tunneling-diode oscillator. Photonics 2018, 5, 52. [CrossRef] 44. Dobroiu, A.; Wakasugi, R.; Shirakawa, R.; Suzuki, S.; Asada, M. Amplitude-modulated continuous-wave radar in the terahertz range using lock-in phase measurement. Meas. Sci. Technol. 2020, 31, 105001. [CrossRef] 45. Dobroiu, A.; Shirakawa, Y.; Suzuki, S.; Asada, M.; Ito, H. Subcarrier frequency-modulated continuous-wave radar in the terahertz range based on a resonant-tunneling-diode oscillator. Sensors 2020, 20, 6848. [CrossRef] 46. Konno, H.; Dobroiu, A.; Suzuki, S.; Asada, M.; Ito, H. OCT technique for distance measurement using an RTD terahertz oscillator. In Proceedings of the International Conference on Infrared, Millimeter, and Terahertz Waves, Buffalo, NY, USA, 8–13 November 2020. 47. Kitagawa, S.; Mizuno, M.; Saito, S.; Ogino, K.; Suzuki, S.; Asada, M. Frequency-tunable resonant-tunneling-diode oscillators applied to absorbance measurement. Jpn. J. Appl. Phys. 2017, 56, 058002. [CrossRef] 48. Sollner, T.C.L.G.; Goodhue, W.D.; Tannenwald, P.E.; Parker, C.D.; Peck, D.D. Resonant tunneling through quantum wells at frequencies up to 2.5 THz. Appl. Phys. Lett. 1983, 43, 588–590. [CrossRef] 49. Shiode, T.; Mukai, T.; Kawamura, M.; Nagatsuma, T. Giga-bit wireless communication at 300 GHz using resonant tunneling diode detector. In Proceedings of the Asia Paciﬁc Microwave Conference, Melbourne, Australia, 5–8 December 2011; pp. 1122–1125. 50. Nishida, Y.; Nishigami, N.; Diebold, S.; Kim, J.; Fujita, M.; Nagatsuma, T. Terahertz coherent receiver using a single resonant tunneling diode. Sci. Rep. 2019, 9, 18125. [CrossRef] 51. Takida, Y.; Suzuki, S.; Asada, M.; Minamide, H. Sensitive terahertz-wave detector response originated by negative differential conductance of resonant-tunneling-diode oscillator. Appl. Phys. Lett. 2020, 117, 021107. [CrossRef] 52. Asada, M.; Suzuki, S.; Fukuma, T. Measurements of temperature characteristics and estimation of terahertz negative differential conductance in resonant-tunneling-diode oscillators. AIP Advance 2017, 7, 115226. [CrossRef] 53. Brown, E.R.; Parker, C.D.; Verghese, S.; Geis, M.W.; Harvey, J.F. Resonant-tunneling transmission-line relaxation oscillator. Appl. Phys. Lett. 1997, 70, 2787–2789. [CrossRef] 54. Goldman, V.J.; Tsui, D.C.; Cunningham, J.E. Observation of intrinsic bistability in resonant-tunneling structures. Phys. Rev. Lett. 1987, 58, 1256–1259. [CrossRef] 55. Rasulova, G.K.; Pentin, I.V.; Vakhtomin, Y.B.; Smirnov, K.V.; Khabibullin, R.A.; Klimov, E.A.; Klochkov, A.N.; Goltsman, G.N. Pulsed terahertz radiation from a double-barrier resonant tunneling diode biased into self-oscillation regime. J. Appl. Phys. 2020, 128, 224303. [CrossRef] 56. Asada, M.; Suzuki, S. Room-temperature oscillation of resonant tunneling diodes close to 2 THz and their functions for various applications. J. Infrared Millim. Terahertz Waves 2016, 37, 1185–1198. [CrossRef] 57. Kikuchi, A.; Bannai, R.; Kishino, K.; Lee, C.-M.; Chyi, J.-I. AlN/GaN double-barrier resonant tunneling diodes grown by rf-plasma-assisted molecular-beam epitaxy. Appl. Phys. Lett. 2002, 81, 1729–1731. [CrossRef] 58. Bayram, C.; Vashaei, Z.; Razeghi, M. AlN/GaN double-barrier resonant tunneling diodes grown by metal-organic chemical vapor deposition. Appl. Phys. Lett. 2010, 96, 042103. [CrossRef] 59. Growden, T.A.; Zhang, W.; Brown, E.R.; Storm, D.F.; Hansen, K.; Fakhimi, P.; Meyer, D.J.; Berger, P.R. 431 kA/cm2 peak tunneling current density in GaN/AlN resonant tunneling diodes. Appl. Phys. Lett. 2018, 112, 033508. [CrossRef] 60. Encomendero, J.; Yan, R.; Verma, A.; Islam, S.M.; Protasenko, V.; Rouvimov, S.; Fay, P.; Jena, D.; Xing, H.G. Room temperature microwave oscillations in GaN/AlN resonant tunneling diodes with peak current densities up to 220 kA/cm. Appl. Phys. Lett. 2018, 112, 103101. [CrossRef] 61. Rutledge, D.B.; Neikirk, D.P.; Kasilingam, D.P. Infrared and Millimeter Waves; Button, K.J., Ed.; Academic Press: Orlando, FL, USA, 1983; Volume 10, Chapter 1. 62. Urayama, K.; Aoki, S.; Suzuki, S.; Asada, M.; Sugiyama, H.; Yokoyama, H. Sub-terahertz resonant tunneling diode oscillators integrated with tapered slot antennas for horizontal radiation. Appl. Phys. Express 2009, 2, 044501. [CrossRef] Sensors 2021, 21, 1384 18 of 19

63. Kasagi, K.; Suzuki, S.; Asada, M. Large-scale array of resonant-tunneling-diode terahertz oscillator for high output power at 1 THz. J. Appl. Phys. 2019, 125, 151601. [CrossRef] 64. Horikawa, H.; Chen, Y.; Koike, T.; Suzuki, S.; Asada, M. Resonant-tunneling-diode terahertz oscillator integrated with a radial line slot antenna for circularly polarized wave radiation. Semicond. Sci. Technol. 2018, 33, 114005. [CrossRef] 65. Asada, M.; Suzuki, S.; Kishimoto, N. Resonant tunneling diodes for sub-terahertz and terahertz oscillators. Jpn. J. Appl. Phys. 2008, 47, 4375–4384. [CrossRef] 66. Asada, M.; Suzuki, S. Terahertz oscillators using electron devices—An approach with resonant tunneling diodes. IEICE Electron. Express 2011, 8, 1110–1126. [CrossRef] 67. Feiginov, M.N. Displacement currents and the real part of high-frequency conductance of the resonant tunneling diode. Appl. Phys. Lett. 2001, 78, 3301–3303. [CrossRef] 68. Feiginov, M. Frequency limitations of resonant-tunnelling diodes in sub-THz and THz oscillators and detectors. J. Infrared Millim. Terahertz Waves 2019, 40, 365–394. [CrossRef] 69. Liu, H.C. Analytical model of high-frequency resonant tunneling: The first-order ac current response. Phys. Rev. B 1991, 43, 12538–12548, Erratum in 1993, 48, 4877. [CrossRef] 70. Sugimura, A. Resonant enhancement of terahertz dynamics in double barrier resonant tunneling diodes. Semicond. Sci. Technol. 1994, 9, 512–514. [CrossRef] 71. Asada, M. Density-matrix modeling of terahertz photon-assisted tunneling and optical gain in resonant tunneling diodes. Jpn. J. Appl. Phys. 2001, 40, 5251–5256. [CrossRef] 72. Frensley, W. Wigner-function model of a resonant-tunneling semiconductor device. Phys. Rev. B 1987, 36, 1570–1579. [CrossRef] [PubMed] 73. Kluksdahl, N.C.; Kriman, A.M.; Ferry, D.K. Self-consistent study of the resonant-tunneling diode. Phys. Rev. B 1989, 39, 7720–7735. [CrossRef][PubMed] 74. Jonasson, O.; Knezevic, I. Coulomb-driven terahertz-frequency intrinsic current oscillations in a double-barrier tunneling structure. Phys. Rev. B 2014, 90, 165415. [CrossRef] 75. Kanaya, H.; Shibayama, H.; Sogabe, R.; Suzuki, S.; Asada, M. Fundamental oscillation up to 1.31 THz in resonant tunneling diodes with thin well and barriers. Appl. Phys. Express 2012, 5, 124101. [CrossRef] 76. Kanaya, H.; Sogabe, R.; Maekawa, T.; Suzuki, S.; Asada, M. Fundamental oscillation up to 1.42 THz in resonant tunneling diodes by optimized collector spacer thickness. J. Infrared Millim. Terahertz Waves 2014, 35, 425–431. [CrossRef] 77. Maekawa, T.; Kanya, H.; Suzki, S.; Asada, M. Frequency increase in terahertz oscillation of resonant tunnelling diode up to 1.55 THz by reduced slot-antenna length. Electron. Lett. 2014, 50, 1214–1216. [CrossRef] 78. Maekawa, T.; Kanaya, H.; Suzuki, S.; Asada, M. Oscillation up to 1.92 THz in resonant tunneling diode by reduced conduction loss. Appl. Phys. Express 2016, 9, 024101. [CrossRef] 79. Shiraishi, M.; Shibayama, H.; Ishigaki, K.; Suzuki, S.; Asada, M.; Sugiyama, H.; Yokoyama, H. High output power (~400 µW) oscillators at around 550 GHz using resonant tunneling diodes with graded emitters and thin barriers. Appl. Phys. Express 2011, 4, 064101. [CrossRef] 80. Suzuki, S.; Shiraishi, M.; Shibayama, H.; Asada, M. High-power operation of terahertz oscillators with resonant tunneling diode using impedance-matched antennas and array configuration. IEEE J. Sel. Top. Quantum Electron 2013, 19, 8500108. [CrossRef] 81. Asada, M.; Suzuki, S. Theoretical analysis of coupled oscillator array using resonant tunneling diodes in subterahertz and terahertz range. J. Appl. Phys. 2008, 103, 124514. [CrossRef] 82. Bezhko, M.; Suzuki, S.; Asada, M. Analysis of oscillation characteristics for resonant-tunneling diode cavity-type terahertz oscillator. In Proceedings of the International Conference on Infrared, Millimeter, and Terahertz Waves, Buffalo, NY, USA, 8–13 November 2020. 83. Han, F.; Fujikata, H.; Kobayashi, K.; Tanaka, H.; Suzuki, S.; Asada, M. Impedance matching method in high-power RTD THz oscillator integrated with rectangular-cavity resonator. In Proceedings of the International Conference on Infrared, Millimeter, and Terahertz Waves, Buffalo, NY, USA, 8–13 November 2020. 84. Mai, T.V.; Suzuki, Y.; Yu, X.; Suzuki, S.; Asada, M. Structure-simplified resonant-tunneling-diode terahertz oscillator without metal-insulator-metal capacitors. J. Infrared Millim. Terahertz Waves 2020, 41, 1498–1507. 85. Asada, M.; Orihashi, N.; Suzuki, S. Experimental and theoretical analysis of voltage-controlled sub-THz oscillation of resonant tunneling diodes. IEICE Trans. Electron. 2006, E89-C, 965–972. [CrossRef] 86. Kitagawa, S.; Suzuki, S.; Asada, M. 650-GHz resonant-tunneling-diode VCO with wide tuning range using varactor diode. IEEE Electron Device Lett. 2014, 35, 1215–1217. [CrossRef] 87. Kitagawa, S.; Ogino, K.; Suzuki, S.; Asada, M. Wide frequency-tuning in resonant-tunneling-diode oscillator using forward-biased varactor diode. Jpn. J. Appl. Phys. 2017, 56, 040301. [CrossRef] 88. Kitagawa, S.; Suzuki, S.; Asada, M. Wide frequency-tunable resonant tunneling diode terahertz oscillators using varactor diodes. Electron. Lett. 2016, 52, 479–481. [CrossRef] 89. Karashima, K.; Yokoyama, R.; Shiraishi, M.; Suzuki, S.; Aoki, S.; Asada, M. Measurement of oscillation frequency and spectral linewidth of sub-terahertz InP-based resonant tunneling diode oscillators using Ni–InP Schottky barrier diode. Jpn. J. Appl. Phys. 2010, 49, 020208. [CrossRef] Sensors 2021, 21, 1384 19 of 19

90. Asada, M. Theoretical analysis of spectral linewidth of terahertz oscillators using resonant tunneling diodes and their coupled arrays. J. Appl. Phys. 2010, 108, 034504. [CrossRef] 91. Yu, X.; Kim, J.; Fujita, M.; Nagatsuma, T. Highly stable terahertz resonant tunneling diode oscillator coupled to photonic-crystal cavity. In Proceedings of the Asia-Paciﬁc Microwave Conference, Kyoto, Japan, 6–9 November 2018; pp. 114–116. 92. Ogino, K.; Suzuki, S.; Asada, M. Spectral narrowing of a varactor-integrated resonant-tunneling-diode terahertz oscillator by phase-locked loop. J. Infrared Millim. Terahertz Waves 2017, 38, 1477–1486. [CrossRef] 93. Ogino, K.; Suzuki, S.; Asada, M. Phase locking and frequency tuning of resonant-tunneling-diode terahertz oscillators. IEICE Trans. Electron. Jpn. 2018, E101.C, 183–185. [CrossRef] 94. Arzi, K.; Suzuki, S.; Rennings, A.; Erni, D.; Weimann, N.; Asada, M.; Prost, W. Subharmonic injection locking for phase and frequency control of RTD-based THz oscillator. IEEE Trans. Terahertz Sci. Technol. 2020, 10, 221–224. [CrossRef] 95. Asada, M. Theoretical analysis of subharmonic injection locking in resonant-tunneling-diode terahertz oscillators. Jpn. J. Appl. Phys. 2020, 59, 018001. [CrossRef] 96. Asada, M.; Suzuki, S. Theoretical analysis of external feedback effect on oscillation characteristics of resonant-tunneling-diode terahertz oscillators. Jpn. J. Appl. Phys. 2015, 54, 070309. [CrossRef] 97. Chen, Y.; Suzuki, S.; Asada, M. Generation of terahertz vortex waves in resonant-tunneling-diode oscillators by integrated radial line slot antenna. In Proceedings of the International Conference on Infrared, Millimeter, and Terahertz Waves, Nagoya, Japan, 9–14 September 2018. Abstract No. PID541007. 98. Ikeda, Y.; Kitagawa, S.; Okada, K.; Suzuki, S.; Asada, M. Direct intensity modulation of resonant-tunneling-diode terahertz oscillator up to ~30GHz. IEICE Electron. Express 2015, 12, 2014116. [CrossRef] 99. Nagatsuma, T.; Nishii, H.; Ikeo, T. Terahertz imaging based on optical coherent tomography. Photon. Res. 2014, 2, B64–B69. [CrossRef]