A Ka-Band Circular Polarized Waveguide Slot Antenna with a Cross Iris

applied sciences

Article A Ka-Band Circular Polarized Waveguide Slot Antenna with a Cross Iris

Sung-Joon Yoon and Jae-Hoon Choi *

Department of Electronics and Computer Engineering, Hanyang University, Seoul 133-791, Korea; [email protected] * Correspondence: [email protected]; Tel.: +82-2-2220-0376

Received: 30 August 2020; Accepted: 2 October 2020; Published: 7 October 2020

Featured Application: Segment waveguide slot antenna for a satellite communication and radar.

Abstract: A Ka-band circularly polarized (CP) waveguide slot antenna with a cross iris is proposed. To achieve the CP characteristic (LHCP) in the antenna, a hexagonal shaped cavity is used. The impedance matching is improved by adjusting the sizes of the hexagonal cavity and a rectangular matching cavity. To enhance the axial ratio bandwidth, a cross iris is used with two arms of different lengths. From experimental results, the proposed antenna has a 3 dB axial ratio bandwidth of 6810 MHz (31.72 GHz–38.53 GHz). The 10 dB reflection coefficient bandwidth of the proposed − antenna ranges from 31.51 GHz to 39.21 GHz (7700 MHz).

Keywords: waveguide antenna; slot antenna; circular polarization; axial ratio enhancement

1. Introduction Millimeter-wave communication systems support a higher data rate and larger capacity. Millimeter wave frequency has therefore been widely used in various applications such as 5G mobile, radar, satellite, military, and automotive communications [1]. Because mm-wave signals can be easily distorted by atmospheric interference such as fog or dust, waveguide antennas have been considered among the most appropriate for mm-wave applications due to their high power handling, high gain, stable performance and low transmission loss [2,3]. The polarization of antennas is one of the important factors in determining the performance of the communication. For linear polarization antenna, if the polarizations of a transmitter and a receiver do not match each other, communication performance deteriorates. Meanwhile, circular polarization avoids polarization misalignment as the electric ﬁeld rotates, and it reduces multipath fading with ﬂexibility in orientation the angle between the transmitted receiver [4,5]. Therefore, circular polarized (CP) waveguide antennas in particular are widely used in modern satellite communication systems due to their advantages of enhancing the channel stability and reliability [6,7]. Several techniques have been introduced to realize the waveguide CP horn and aperture antennas such as using a printed circuit board with four printed hook shaped arms [7], loading two metallic arms with a 45◦ inclination angle [8], and inserting a meta-surface polarizer or chiral metamaterial transformer [6,9]. However, there are some limitations in applying them to various applications since these antennas are relatively bulky and heavy. In addition, they are not suitable for array systems because a large space is required for polarizers to be installed [10]. Therefore, such antennas are not appropriate for satellite communication systems requiring a high gain array structure with compact size. To overcome these drawbacks of bulk and mass, various types of circular polarized waveguide slot antennas have been studied [11–17]. Waveguide inclined slot, crossed slot, and v-type slot antennas have been widely used for CP array antennas to generate circularly polarized waves with high gain [13–17]. However, there are few reports on either widening

Appl. Sci. 2020, 10, 6994; doi:10.3390/app10196994 www.mdpi.com/journal/applsci Appl. Sci. 2020, 10, 6994 2 of 15 the axial ratio bandwidth of the waveguide slot antenna at the millimeter-wave band region or the design of waveguide slot CP antennas [16,17]. This paper proposes a Ka-band circularly polarized waveguide slot antenna with a cross iris. To achieve a circular polarization characteristic, a hexagonal shaped cavity is used [18]. By mounting a cross iris,Appl. the Sci. circular2020, 10, x FOR polarization PEER REVIEW performance of the proposed antenna is enhanced. The2 of proposed15 antenna provides a 10 dB reflection coefficient bandwidth covering from 31.51 GHz to 39.21 GHz. − The 3 dBreports axial on ratio either bandwidth widening the is axial 6810 ratio MHz bandwidt (31.72h of GHz–38.53 the waveguide GHz). slot antenna The antenna at the millimeter- peak gains are wave band region or the design of waveguide slot CP antennas [16,17]. 9.08 dBi at 33.5 GHz and 7.33 dBi at 37.6 GHz. The boresight gains of 7.87 dBi and 5.76 dBi are achieved This paper proposes a Ka-band circularly polarized waveguide slot antenna with a cross iris. To at thoseachieve two frequencies, a circular polarization respectively. characteristic, a hexagonal shaped cavity is used [18]. By mounting a cross iris, the circular polarization performance of the proposed antenna is enhanced. The proposed 2. Antennaantenna Design provides a −10 dB reflection coefficient bandwidth covering from 31.51 GHz to 39.21 GHz. FigureThe 31 dBshows axial theratio structure bandwidth of is the 6810 proposed MHz (31.72 CP GHz–38.53 waveguide GHz). slot The antenna antenna consisting peak gains of are a rectangular9.08 dBi at 33.5 GHz and 7.33 dBi at 37.6 GHz. The boresight gains of 7.87 dBi and 5.76 dBi are achieved waveguide, an air metal box with hexagonal cavity and matching slot, and a cross iris at the bottom of at those two frequencies, respectively. the rectangular waveguide. The proposed antenna has dimensions of 10.1 mm 23 mm 7.06 mm and × × is made2. of Antenna a copper Design with a conductivity of 5.8001 107 S/m. The detailed parameters of the proposed × antenna areFigure shown 1 shows in Figure the 1structure and the of dimensionsthe proposed are CP givenwaveguide in Table slot 1antenna. The performancesconsisting of a of the proposedrectangular antenna waveguide, were simulated an air metal using box the with commercial hexagonal cavity electromagnetic and matching simulator,slot, and a cross High iris Frequency at Structurethe Simulatorbottom of the (HFSS) rectangular [19]. waveguide. The antenna The prop is fedosed from antenna the has rectangular dimensions of waveguide 10.1 mm × 23 to mm excite the × 7.06 mm and is made of a copper with a conductivity of 5.8001 × 107 S/m. The detailed parameters TE10 mode for high power handling capability. The dimensions (a = 7.11 mm and b = 3.56 mm) of the proposed antenna are shown in Figure 1 and the dimensions are given in Table 1. The of the feeding waveguide are determined by the cutoff frequency (21.077 GHz) using the following performances of the proposed antenna were simulated using the commercial electromagnetic equationsimulator, with dimensions High Frequency shown Structure in Figure Simulator2[ 20]. (HFS AsS) shown [19]. The in Figureantenna2 ,is the fed waveguidefrom the rectangular has its longest dimensionwaveguide along theto excite x-axis the (a ).TE mode for high power handling capability. The dimensions (𝑎= 7.11 mm and 𝑏=3.56 mm) of the feeding waveguider are determined by the cutoff frequency (21.077 GHz) using the following equation 1with dimensionsmπ2 shownnπ in2 Figure 2 [20]. As shown in Figure 2, the waveguide has itsfcmn longest= dimension along the x-axis+ (a). , (1) 2π √µ0ε0 a b 𝑚𝜋 2 𝑛𝜋 2 𝑓 = + , (1) When a > b, the cutoff frequency of TE10 (m =1,n= 0)𝑎 mode𝑏 becomes

When a>b, the cutoff frequency of TE10 (m = 1, n = 0) mode becomes 1 fc10 = , (2) 𝑓 = , (2) 2a√µ0ε0

μ ε . where µwhere0 and ε00 andare the0 are permeability the permeability and and permittivity permittivity of free free space, space, respectively respectively.

(a)

Figure 1. Cont. Appl. Sci. 2020, 10, 6994 3 of 15 Appl. Sci. 2020, 10, x FOR PEER REVIEW 3 of 15

Appl. Sci. 2020, 10, x FOR PEER REVIEW 3 of 15

(b)

(c)

FigureFigure 1. Geometry 1. Geometry of the of proposedthe proposed antenna: antenna: (a ()a) 3-D 3-Dview view of the proposed proposed antenna; antenna; (b) (topb) topview view of of the proposedthe antenna;proposed antenna; (c) side ( viewc) side of view the of proposed the propos antennaed antenna and and cross cross iris.iris. Tabley 1. Dimensions of the proposed antenna.

Parameters C1 C2 S1 S2 L1 L2 (c) Value (mm) 6.5 4.4 4.3 8 6 5 Figure 1. GeometryParameters of the proposedb H1 antenna:H2 (a) 3-DH3 viewH of4 the proposedB1 B antenna;2 (b) top view of the proposed antenna;Value (c (mm)) side view7.06 of the 2.5proposed 0.5 antenna 1.2 and cross 5.9 iris. 3.28

y (μ0, ε0)

0 a x

Figure 2. Geometry of the rectangular waveguide.

Table 1. Dimensions of the proposed antenna. (μ0, ε0) Parameters 𝐶 𝐶 𝑆 𝑆 𝐿 𝐿 Value (mm) 6.5 4.4 4.3 8 6 5 Parameters0 𝐻 𝐻 𝐻 𝐻 𝐵 𝐵 Value (mm) 7.06 2.5 0.5 1.2 a 5.9 x 3.28

FigureFigure 2. 2.Geometry Geometry of of the the rectangular rectangular waveguide. waveguide.

Table 1. Dimensions of the proposed antenna.

Parameters 𝐶 𝐶 𝑆 𝑆 𝐿 𝐿 Value (mm) 6.5 4.4 4.3 8 6 5

Parameters 𝐻 𝐻 𝐻 𝐻 𝐵 𝐵 Value (mm) 7.06 2.5 0.5 1.2 5.9 3.28 Appl.Appl. Sci. Sci. 20202020, ,1010, ,x 6994 FOR PEER REVIEW 44 of of 15 15 Appl. Sci. 2020, 10, x FOR PEER REVIEW 4 of 15 3. Antenna Analysis 3.3. Antenna Antenna Analysis Analysis In the proposed antenna design, the hexagonal shaped cavity with a rectangular waveguide shownInIn thein the Figureproposed proposed 3 is antenna antennaused as design, design, an initial the the hexagonal hexagonalmodel (first shaped shaped reference cavity cavity antenna) with with a a rectangular to realize the waveguide circular polarization.shownshown inin FigureFigure The3 ishexagonal3 usedis used as an shapeas initial an isinitial model similar (ﬁrstmodel to reference a (firstcorner reference antenna) truncated toantenna) realizepatch theantenna,to circularrealize which polarization.the circularcan be theoreticallypolarization.The hexagonal analyzedThe shape hexagonal is using similar ashape tocavity a corner is model similar truncated with to athe patchcorner magnetic antenna, truncated walls which resonatingpatch can beantenna, theoretically in TM which (Transverse analyzed can be Magnetic)theoreticallyusing a cavity mode analyzed model [21]. However, withusing the a cavity magneticthis hexagonal model walls with resonatingcavity the magnetic resonates in TM walls TE (Transverse mode resonating with Magnetic) electrical in TM (Transverse side mode walls [21]. [22].Magnetic)However, The truncated mode this hexagonal [21]. corners However, cavity make this resonatesthe hexagonal hexagonal TE modecavity cavity with resonates generate electrical TEE-fields mode side aswalls with two [electricalorthogonal22]. The side truncated modes, walls which[22].corners The are make truncated written the hexagonalin corners this paper make cavity as the generatethe hexagonal E1 mode E-ﬁelds caviandty asE 2 twogenerate mode. orthogonal As E-fields shown modes, as in two Figure which orthogonal 4, are because written modes, the in 1 2 truncatedwhichthis paper are part aswritten the (CE1) 1 inactsmode this as andadditionalpaperE2 asmode. the capacitive E As mode shown loading, and in FigureE themode.4 E,1 because mode As shown has the slower truncated in Figure phase part 4, velocitybecause ( C1) acts andthe as 1 1 smallertruncatedadditional guided part capacitive ( Cwavelength) acts loading, as additional than the Edoes1 mode capacitive the hasE2 slowermode. loading, phaseAs thea result, velocityE mode a and CPhas modesmallerslower is guidedphase generated velocity wavelength by andthe 2 additionsmallerthan does guided of the a laggingE2 wavelengthmode. (E1 As) mode a result,than and does a leading CP the mode (EE2 is)mode. mode. generated As a by result, the addition a CP mode of a lagging is generated (E1) mode by andthe 1 2 additionleading (ofE2 )a mode.lagging (E ) mode and leading (E ) mode.

Figure 3. The geometry of the initial model antenna (first reference antenna). Figure 3. The geometry of the initial model antenna (ﬁrst reference antenna). Figure 3. The geometry of the initial model antenna (first reference antenna).

(a) (b) (a) (b) Figure 4. Definition of the two orthogonal modes: (a) E1 mode; (b) E2 mode. Figure 4. Definition of the two orthogonal modes: (a) E1 mode; (b) E2 mode. Figure 4. Definition of the two orthogonal modes: (a) E1 mode; (b) E2 mode. In a waveguide antenna, the operating frequency and axial ratio performance are predominantly In a waveguide antenna, the operating frequency and axial ratio performance are predominantly affected by the dimension and shape of the cavity [17,22]. As the dimension of a hexagonal cavity affectedIn a bywaveguide the dimension antenna, and the shape operating of the frequency cavity [17,22]. and axial As theratio dimensio performancen of a are hexagonal predominantly cavity increases, the operation frequency bandwidth is expected to increase. However, the dimension of the increases,affected by the the operation dimension frequency and shape bandwidth of the cavity is expected [17,22]. to Asincrease. the dimensio However,n of the a hexagonaldimension cavityof the hexagonal cavity is limited to the size of a feeding rectangular waveguide. Under such restriction, hexagonalincreases, the cavity operation is limited frequency to the bandwidthsize of a feeding is expected rectangular to increase. waveguide. However, Under the dimensionsuch restriction, of the the lengths of a set of diagonal edges (C1 and C2) of the cavity are optimized to obtain the best thehexagonal lengths cavity of a setis limitedof diagonal to the edges size of (C a1 andfeeding C2) rectangularof the cavity waveguide. are optimized Under to suchobtain restriction, the best performance. Figure5 shows the simulated reflection coe fficient and axial ratio of the first reference performance.the lengths of Figure a set 5of shows diagonal the simulatededges (C1 reflectionand C2) of coefficient the cavity and are axial optimized ratio of to the obtain first reference the best antenna with different values of C1 and C2. As shown in Figure5, as the dimension of the hexagonal antennaperformance. with differentFigure 5 showsvalues theof C simulated1 and C2. As reflection shown incoefficient Figure 5, and as the axial dimension ratio of the of thefirst hexagonal reference cavity becomes smaller the resonance frequency shifts to the higher side. The circular polarization cavityantenna becomes with different smaller values the resonance of C1 and frequency C2. As shown shif tsin to Figure the higher 5, as the side. dimension The circular of the polarization hexagonal performance of the first reference antenna is acceptable when C1 = 6.5 mm and C2 = 4.4 mm with the performancecavity becomes of thesmaller first referencethe resonance antenna frequency is acceptable shifts towhen the Chigher1 = 6.5 side.mm andThe Ccircular2 = 4.4 mm polarization with the 3 dB axial ratio bandwidth ranging from 30.37 GHz to 31.2 GHz. The first reference antenna has a 3performance dB axial ratio of bandwidththe first reference ranging antenna from 30.37 is acceptable GHz to 31.2 when GHz. C1 The= 6.5 first mm reference and C2 = antenna4.4 mm withhas a the −6 6 dB reflection coefficient bandwidth of 3250 MHz (from 30.23 GHz to 33.48 GHz). dB3− dB reflection axial ratio coefficient bandwidth bandwidth ranging fromof 32 5030.37 MHz GHz (from to 31.2 30.23 GHz. GHz The to 33.48first reference GHz). antenna has a −6 dB reflection coefficient bandwidth of 3250 MHz (from 30.23 GHz to 33.48 GHz). Appl. Sci. 2020, 10, 6994 5 of 15 Appl. Sci. 2020, 10, x FOR PEER REVIEW 5 of 15

(a)

(b)

Figure 5. Simulated (a) reflection coefficient and (b) axial ratio of the first reference antenna for different Figure 5. Simulated (a) reflection coefficient and (b) axial ratio of the first reference antenna for values of C1 and C2. different values of C1 and C2. To investigate that the circular polarization characteristic is achieved by the hexagonal cavity, To investigate that the circular polarization characteristic is achieved by the hexagonal cavity, the variation of the electric field distributions of the first reference antenna with different phases at the variation of the electric field distributions of the first reference antenna with different phases at 31 GHz are illustrated in Figure6. As shown in Figure6, the electric field rotates clockwise as the phase 31 GHz are illustrated in Figure 6. As shown in Figure 6, the electric field rotates clockwise as the changes from 0 to 270 . Therefore, we conclude that the first reference antenna has a left-handed phase changes from◦ 0° ◦to 270°. Therefore, we conclude that the first reference antenna has a left- circular polarization (LHCP) characteristic. handed circular polarization (LHCP) characteristic. Appl. Sci. 2020, 10, x FOR PEER REVIEW 6 of 15 Appl. Sci. 2020, 10, 6994 6 of 15

(a) (b)

Figure 6. Simulated electric field field distribu distributionstions of of the first first reference antenna with different different phases at

31 GHz: ( (aa)) phase phase == 0°;0◦ ;((bb) )phase phase == 90°;90◦ (;(c)c phase) phase = 180°;= 180 (◦d;() dphase) phase = 270°.= 270 ◦.

Although the fi firstrst reference antenna presentspresents a reasonably good −66 dB dB re reflectionflection coe fficient and − 3 dB axial ratio bandwidth, additional impedance matching is required because satellite applications generally impose a −1010 dB dB re reflectionflection coe coeffifficientcient criteria criteria for for communication communication efficiency efficiency [23]. [23]. To To improve improve − the impedance matching characteristic, we we added added a rectangular matching cavity located on the hexagonal cavity (second reference antenna). In In the the equivalent equivalent circuit, circuit, the the long side ( S2)) of of the matching cavity acts as a capa capacitancecitance while the short side ( S1)) has has an an inductive inductive characteristic characteristic [24]. [24]. Therefore, thethe impedance impedance matching matching from from the feedingthe feeding rectangular rectangular waveguide waveguide to free spaceto free is improvedspace is improvedthrough choosing through proper choosing values proper of S 1valuesand S2 offor S1 the and height S2 for ( Hthe3) height of 0.5 mm. (H3) Theof 0.5 optimized mm. The values optimized of S1 valuesand S ofare S1 4.3 and mm S2 andare 4.3 4.8 mm mm, and respectively. 4.8 mm, respectively. As shown in As Figure shown7, thein Figure simulated 7, the simulated10 dB impedance −10 dB 2 − impedancebandwidth ofbandwidth the second of reference the second antenna reference becomes antenna 5690 MHz becomes (30.81–36.50 5690 GHz).MHz Meanwhile,(30.81–36.50 the GHz). 3 dB Meanwhile,axial ratio bandwidth the 3 dB obtainedaxial ratio is 900bandwidth MHz (31.79–32.69 obtained GHz).is 900 ThisMHz result (31.79–32.69 indicates GHz). that the This resonance result indicatesfrequency that is moved the resonance to the upper frequency frequency is moved side to because the upper the dimensionfrequency side of the because aperture the changesdimension by ofthe the addition aperture of changes the rectangular by the addition matching of cavity. the rectangular A similar matching observation cavity. can beA similar made for observation the axial ratio can becharacteristic. made for the In theaxial second ratio reference characteristic. antenna, In althoughthe second the reference10 dB reflection antenna, coe althoughfficient bandwidth the −10 dB of − reflection5690 MHz coefficient is achieved, bandwidth most of the of 5690 bandwidth MHz is exceptachieved, for 900most MHz of the (31.79 bandwidth GHz–32.69 except GHz) for 900 does MHz not (31.79satisfy GHz–32.69 the 3 dB axial GHz) ratio does requirement. not satisfy This the challenge 3 dB axia oftenl ratio occurs requirement. when designing This challenge CP antennas often [occurs17,25]. when designing CP antennas [17,25]. Appl. Sci. 2020, 10, 6994 7 of 15 Appl. Sci. 2020, 10, x FOR PEER REVIEW 7 of 15

(a)

(b)

Figure 7. Simulated (a) reflection coefficient and (b) axial ratio of the first reference antenna, the second Figure 7. Simulated (a) reflection coefficient and (b) axial ratio of the first reference antenna, the reference antenna and proposed antenna. second reference antenna and proposed antenna. In the proposed antenna, to resolve this problem, a metallic cross iris is mounted on the bottom of theIn the rectangular proposed feeding antenna, waveguide to resolve to enhance this problem, the axial a ratio metallic bandwidth cross whichiris is ismounted an important on the factor bottom of thefor rectangular evaluating the feeding CP performance. waveguide It canto enhance be calculated the axial using ratio the following bandwidth equation which [26 ].is an important factor for evaluating the CP performance. It can be calculated using the following equation [26]. / 𝑟 +1 + 𝑟 +1 +2𝑟 cos2∆/ 𝐴𝑅 = (3) 𝑟 +1 − 𝑟 +1 +2𝑟 cos2∆// where r is the ratio of magnitudes of orthogonal electric fields, and ∆ is the phase difference between the two orthogonal electric fields at the far zone. In general, it can be said that CP operates well when the value of the axial ratio is less than 3 dB. Accordingly, when r = 1, the phase difference should range from 70° to 110° and when ∆=90°, the ratio of magnitudes should satisfy the range of ≤𝑟≤ √ √2. Appl. Sci. 2020, 10, 6994 8 of 15

h i1/21/2 r2 + 12 + r4 + 14 + 2r2 cos(2∆) AR = (3) linear scale h i1/2 r2 + 12 [r4 + 14 + 2r2 cos(2∆)]1/2 − where r is the ratio of magnitudes of orthogonal electric fields, and ∆ is the phase difference between the two orthogonal electric fields at the far zone. In general, it can be said that CP operates well when Appl. Sci. 2020, 10, x FOR PEER REVIEW 8 of 15 the value of the axial ratio is less than 3 dB. Accordingly, when r = 1, the phase difference should range 1 from 70◦ to 110◦ and when ∆ = 90◦, the ratio of magnitudes should satisfy the range of r √2. Figure 8 shows the simulated phase differences between the two orthogonal modes√2 ≤ (≤E1 and E2) of the proposedFigure8 antennashows the and simulated the second phase reference di fferences an betweentenna. The the twophases orthogonal of the two modes orthogonal (E1 and E2 )electric of the proposed antenna and the second reference antenna. The phases of the two orthogonal electric fields fields can be minutely controlled separately because the iris structure consists of two arms with can be minutely controlled separately because the iris structure consists of two arms with asymmetrical asymmetrical lengths (L1 and L2). By adjusting the lengths of the cross iris, the improved 90° phase lengths (L1 and L2). By adjusting the lengths of the cross iris, the improved 90◦ phase difference range difference range (70° to 110°) is achieved (31.70 GHz–39.25 GHz). However, the 3 dB axial ratio (70◦ to 110◦) is achieved (31.70 GHz–39.25 GHz). However, the 3 dB axial ratio bandwidth is not the bandwidthsame as is the not 90 ◦thephase same diff aserence the 90° range phase because difference the ratio range of the because magnitude the of ratio the two of orthogonalthe magnitude electric of the two orthogonalfields should electric be considered. fields should be considered.

FigureFigure 8. Simulated 8. Simulated phase phase difference difference between between two two orthogon orthogonalal electricelectric fields fields of of the the proposed proposed antenna antenna and theand second the second reference reference antenna. antenna. To analyze how the cross iris affects the magnitude of the two electric fields, simulated amplitudes To analyzeof the two how orthogonal the cross modes iris affects of the proposedthe magnitude antenna of and the the two second electric reference fields, antenna simulated are presented amplitudes of thein two Figure orthogonal9. In the secondmodes referenceof the proposed antenna, ante thenna two and points the where second the reference mode lines antenna cross are are located presented in Figureat 30.04 9. In GHz the andsecond 31.80 reference GHz. At antenna, these two the points, two points the magnitudes where the ofmode the twolines modes cross are located equal. at 30.04However, GHz and as 31.80 shown GHz. in Figure At these8, because two points, the 90 ◦ thephase magnitudes difference rangeof the of two the modes second referenceare equal. antenna However, as shownis from in 31.7 Figure GHz, 8, it because cannot be the said 90° that phase CP is differen generatedce atrange less thanof the 31.7 second GHz. Moreover,reference theantenna difference is from 31.7 GHz,in amplitude it cannot between be said the that two modesCP is generated from the second at less point than (31.8 31.7 GHz) GHz. is increasing, Moreover, which the difference indicates in amplitudedeteriorated between CP performance.the two modes For thefrom proposed the second antenna, point the (31.8 magnitudes GHz) ofis theincreasing, two orthogonal which modes indicates deterioratedare equal CP at three performance. points (30.75 For GHz, the 32.32 proposed GHz, and antenna, 38.44 GHz). the magnitudes In addition, theof amplitudesthe two orthogonal of the two modes become close to each other over the whole frequency range from (32.32 GHz to 38.44 GHz). modes are equal at three points (30.75 GHz, 32.32 GHz, and 38.44 GHz). In addition, the amplitudes It can be said that the proposed antenna has a good equal magnitude characteristic. of the two modes become close to each other over the whole frequency range from (32.32 GHz to 38.44 GHz). It can be said that the proposed antenna has a good equal magnitude characteristic.

(a) Appl. Sci. 2020, 10, x FOR PEER REVIEW 8 of 15

Figure 8 shows the simulated phase differences between the two orthogonal modes (E1 and E2) of the proposed antenna and the second reference antenna. The phases of the two orthogonal electric fields can be minutely controlled separately because the iris structure consists of two arms with asymmetrical lengths (L1 and L2). By adjusting the lengths of the cross iris, the improved 90° phase difference range (70° to 110°) is achieved (31.70 GHz–39.25 GHz). However, the 3 dB axial ratio bandwidth is not the same as the 90° phase difference range because the ratio of the magnitude of the two orthogonal electric fields should be considered.

Figure 8. Simulated phase difference between two orthogonal electric fields of the proposed antenna and the second reference antenna.

To analyze how the cross iris affects the magnitude of the two electric fields, simulated amplitudes of the two orthogonal modes of the proposed antenna and the second reference antenna are presented in Figure 9. In the second reference antenna, the two points where the mode lines cross are located at 30.04 GHz and 31.80 GHz. At these two points, the magnitudes of the two modes are equal. However, as shown in Figure 8, because the 90° phase difference range of the second reference antenna is from 31.7 GHz, it cannot be said that CP is generated at less than 31.7 GHz. Moreover, the difference in amplitude between the two modes from the second point (31.8 GHz) is increasing, which indicates deteriorated CP performance. For the proposed antenna, the magnitudes of the two orthogonal modes are equal at three points (30.75 GHz, 32.32 GHz, and 38.44 GHz). In addition, the amplitudes ofAppl. the Sci. two2020 modes, 10, 6994 become close to each other over the whole frequency range from (32.32 GHz9 of to 15 38.44 GHz). It can be said that the proposed antenna has a good equal magnitude characteristic.

Appl. Sci. 2020, 10, x FOR PEER REVIEW 9 of 15 (a)

(b)

Figure 9. Simulated amplitudes of E1 and E2 modes of (a) the second reference antenna and Figure 9. Simulated amplitudes of E1 and E2 modes of (a) the second reference antenna and (b) the (b) the proposed antenna. proposed antenna.

Figure 10 depicts the simulated amplitudes of E1 and E2 modes for various values of two different Figure 10 depicts the simulated amplitudes of E1 and E2 modes for various values of two different lengths (L1 and L2) of the cross iris. The long side of (L1) of the cross iris has a more dominant effect on lengths (L1 and L2) of the cross iris. The long side of (L1) of the cross iris has a more dominant effect the amplitude of E2 mode than on E1 mode. The E1 mode changes more with the variation in the length on the amplitude of 𝐸 mode than on E1 mode. The E1 mode changes more with the variation in the of short side (L2). Therefore, we can conclude that the amplitude of E2 mode increases as L1 increases length of short side (L2). Therefore, we can conclude that the amplitude of E2 mode increases as L1 and that the amplitude of E1 mode increases as L2 increases. When L1 = 6 mm and L2 = 5 mm, a good increases and that the amplitude of E1 mode increases as L2 increases. When L1 = 6 mm and L2 = 5 mm, magnitude match occurs between the two modes with a wide 90 phase difference range, leading to a good magnitude match occurs between the two modes with ◦a wide 90° phase difference range, the wide axial ratio bandwidth. Additionally, because the cross iris has a height of 1.2 mm, it acts as leading to the wide axial ratio bandwidth. Additionally, because the cross iris has a height of 1.2 mm, a step between the rectangular waveguide and the radiation slot resulting in gradual change in the it acts as a step between the rectangular waveguide and the radiation slot resulting in gradual change impedance so that the matching condition is improved [27]. in the impedance so that the matching condition is improved [27]. Figure 11 shows the simulated reflection coefficients when the height of the cross iris (H4) varies from 0.8 mm to 1.2 mm. When H4 equals 1.2 mm, an enhanced impedance bandwidth is obtained. As a result, as shown in Figure7, the proposed antenna achieves the simulated 3 dB axial ratio bandwidth of 6479 MHz (31.97 GHz to 38.45 GHz) and a –10 dB reflection coefficient bandwidth of 6840 MHz (31.74 GHz to 38.58 GHz).

(a) Appl. Sci. 2020, 10, x FOR PEER REVIEW 9 of 15

(b)

Figure 9. Simulated amplitudes of E1 and E2 modes of (a) the second reference antenna and (b) the proposed antenna.

Figure 10 depicts the simulated amplitudes of E1 and E2 modes for various values of two different lengths (L1 and L2) of the cross iris. The long side of (L1) of the cross iris has a more dominant effect on the amplitude of 𝐸 mode than on E1 mode. The E1 mode changes more with the variation in the length of short side (L2). Therefore, we can conclude that the amplitude of E2 mode increases as L1 increases and that the amplitude of E1 mode increases as L2 increases. When L1 = 6 mm and L2 = 5 mm, a good magnitude match occurs between the two modes with a wide 90° phase difference range, leading to the wide axial ratio bandwidth. Additionally, because the cross iris has a height of 1.2 mm, it actsAppl. as Sci. a 2020step, 10 between, 6994 the rectangular waveguide and the radiation slot resulting in gradual10 change of 15 in the impedance so that the matching condition is improved [27].

Appl. Sci. 2020, 10, x FOR PEER REVIEW (a) 10 of 15

(b)

Figure 10. FigureSimulated 10. Simulated amplitudes amplitudes of E of1 Eand1 andE E22 modes for for various various values values of (a) L of1 (L (2a is) fixedL1 (L at2 5is mm) ﬁxed at 5 mm) and (b) L2 (andL1 (isb) ﬁxedL2 (L1 is at fixed 6 mm). at 6 mm).

Figure 11 shows the simulated reflection coefficients when the height of the cross iris (H4) varies from 0.8 mm to 1.2 mm. When H4 equals 1.2 mm, an enhanced impedance bandwidth is obtained. As a result, as shown in Figure 7, the proposed antenna achieves the simulated 3 dB axial ratio bandwidth of 6479 MHz (31.97 GHz to 38.45 GHz) and a –10 dB reflection coefficient bandwidth of 6840 MHz (31.74 GHz to 38.58 GHz). Appl. Sci. 2020, 10, 6994 11 of 15 Appl.Appl. Sci. Sci. 2020 2020, 10, 10, x, xFOR FOR PEER PEER REVIEW REVIEW 1111 of of 15 15

FigureFigureFigure 11. 11.11. Simulated SimulatedSimulated reflection reflectionreflection coefficients coecoefficientsfficients for forfor different didifferentfferent height height ( H(H44)4) )values valuesvalues of ofof the thethe cross cross cross iris. iris. iris.

4.4.4. Experimental ExperimentalExperimental Results ResultsResults PhotographsPhotographsPhotographs of ofof the thethe fabricated fabricatedfabricated antenna antennaantenna are areare shown shownshown in in Figure Figure 12.12 12.. The TheThe manufactured manufacturedmanufactured antenna antennaantenna is isis assembledassembledassembled in inin two twotwo parts: parts:parts: an anan upper upperupper cavity cavitycavity and andand a aa rectangular rectangularrectangular waveguide waveguidewaveguide with withwith a aa cross crosscross iris. iris.iris. To ToTo minimize minimizeminimize manufacturingmanufacturingmanufacturing errors, errors,errors, the thethe electrof electroformingelectroformingorming process processprocess was waswas used. used.used. For ForFor measuring measuringmeasuring the thethe reflection reflectionreflection coefficient coecoefficientfficient andandand the thethe radiation radiationradiation pattern patternpattern with withwith the thethe axial axialaxial ratio, ratio,ratio, an anan Agilent AgilentAgilent N5247A N5247AN5247A vector vectorvector network networknetwork analyzer analyzeranalyzer with withwith AgilentAgilentAgilent R11644A R11644AR11644A calibration calibrationcalibration kit kitkit with withwith a aa WR-28 WR-28WR-28 coaxial coaxialcoaxial to toto waveguide waveguide adapter adapteradapter were werewere used. used. They TheyThey were werewere measuredmeasuredmeasured inside insideinside the thethe millimeter-wave millimeter-wavemillimeter-wave anechoic anechoicanechoic chamber chamberchamber equipped equippedequipped with withwith MTG MTGMTG operating operatingoperating software. software.software. FigureFigureFigure 13 1313 showsshows shows the thethe simulated simulatedsimulated and andand measured measuredmeasured reflecti reflectionreflectionon coefficients coecoefficientsfficients and andand axial axialaxial ratio ratioratio of ofof the thethe proposed proposedproposed antenna.antenna.antenna. The TheThe measured measuredmeasured –10 –10–10 dB dBdB reflection reflectionreflection coefficient coecoefficientfficient bandwidth bandwidthbandwidth is isis 7700 77007700 MHz MHzMHz (31.51 (31.51(31.51 GHz–39.21 GHz–39.21GHz–39.21 GHz) GHz)GHz) andandand the thethe fractional fractionalfractional bandwidth bandwidthbandwidth is isis about aboutabout 21.7%. 21.7%.21.7%. The TheThe measured measuredmeasured 3 33 dB dBdB axial axialaxial ratio ratioratio bandwidth bandwidthbandwidth is isis 6810 68106810 MHz MHzMHz (31.72(31.72(31.72 GHz–38.53 GHz–38.53 GHz–38.53 GHz) GHz) GHz) and and and the the the fractional fractional fractional bandwidth bandwidth bandwidth is is is about about about 19.38%. 19.38%. 19.38%. There ThereThere are areare slight slightslight discrepancies discrepanciesdiscrepancies betweenbetweenbetween the thethe simulation simulationsimulation and andand measurement measurementmeasurement results. results.results. Th TheThe ediscrepancies discrepanciesdiscrepancies are areare due duedue to toto the thethe differences didifferencesfferences in inin the thethe crosscrosscross iris irisiris dimensions dimensionsdimensions of of of simulation simulation simulation an and andd those those those of of of the the manufactured manufactured antenna. antenna.

(a(a) ) (b(b) ) Figure 12. Cont. Appl. Sci. 2020, 10, x FOR PEER REVIEW 12 of 15 Appl.Appl. Sci. Sci. 20202020, 10,,10 x ,FOR 6994 PEER REVIEW 12 12of 15 of 15

(c) (d) (c) (d) Figure 12. Photographs of the fabricated antenna: (a) top view; (b) side view; (c) top view of the FigureFigure 12. 12. PhotographsPhotographs of of the the fabricated antenna:antenna: ((aa)) top top view; view; (b ()b side) side view; view; (c) top(c) viewtop view of the of bottom the bottom waveguide; (d) bottom view of the upper cavity. bottomwaveguide; waveguide; (d) bottom (d) bottom view ofview the of upper the upper cavity. cavity.

(a) (a)

(b) (b) Figure 13. Simulated and measured (a) reﬂection coeﬃcients and (b) axial ratio of the proposed antenna. Figure 13. Simulated and measured (a) reflection coefficients and (b) axial ratio of the proposed Figure 13. Simulated and measured (a) reflection coefficients and (b) axial ratio of the proposed antenna. antenna. Appl. Sci. 2020, 10, 6994 13 of 15 Appl. Sci. 2020, 10, x FOR PEER REVIEW 13 of 15

Figure 14 shows the simulated and measured radiationradiation patterns of the proposed antenna in the xz-plane ((∅∅=0= °0)◦ )and and yz-plane yz-plane ( (∅=90∅ = 90°) ◦at) at33.5 33.5 GHz GHz and and 37.6 37.6 GHz. GHz. The The simulated simulated and and measured radiation patterns agree well in both the xz andand yzyz planes.planes. It can be observed that the antenna has directional patterns with the inclined maximum gains gains due due to to the the effect eﬀect of of a a hexagonal hexagonal shaped shaped cavity. cavity. The simulatedsimulated bore-sightbore-sight gains gains are are 8.27 8.27 dBi dBi and and 6.13 6.13 dBi, dBi, and and those those measured measured are are 7.87 7.87 dBi dBi and and 5.76 5.76 dBi atdBi 33.5 at 33.5 GHz GHz and 37.6and 37.6 GHz, GHz, respectively. respectively. At 33.5 At GHz,33.5 GHz, the maximum the maximum gain ofgain 9.08 of dBi 9.08 is dBi achieved is achieved at 18◦ inat the18° xz-plane. in the xz-plane. At 37.6 At GHz, 37.6 the GHz, peak the gain peak of gain 7.33 dBiof 7.33 is obtained dBi is obtained at 24◦ in at the 24 xz-plane.° in the xz-plane.

(a)

(b)

Figure 14. Simulated and measured radiation pattern of the proposed antenna at (a) 33.5 GHz, Figure 14. Simulated and measured radiation pattern of the proposed antenna at (a) 33.5 GHz, and and (b) 37.6 GHz. (b) 37.6 GHz. Table2 compares the dimensions, 3-dB axial ratio bandwidth, and 10 dB reflection coefficient Table 2 compares the dimensions, 3-dB axial ratio bandwidth, and −−10 dB reflection coefficient bandwidth values from the literature with those of the proposed antenna. The fractional 3-dB axial bandwidth values from the literature with those of the proposed antenna. The fractional 3-dB axial ratio bandwidth of the proposed antenna is much wider than the others with a broad overlapping ratio bandwidth of the proposed antenna is much wider than the others with a broad overlapping bandwidth (88%) over which both the 3 dB axial ratio and 10 dB reflection coefficient requirements bandwidth (88%) over which both the 3 dB axial ratio and −10 dB reflection coefficient requirements are satisfied. Even though some antennas [11,12] have a shorter length, their overlapping bandwidths are satisfied. Even though some antennas [11,12] have a shorter length, their overlapping bandwidths (40.5% and 50%) are much smaller than that of the proposed antenna. The antenna in [17] has a very (40.5% and 50%) are much smaller than that of the proposed antenna. The antenna in [17] has a very narrow 3 dB axial ratio and 10 dB reflection coefficient bandwidth even with the size comparable to narrow 3 dB axial ratio and −10 dB reflection coefficient bandwidth even with the size comparable to the proposed antenna. Therefore, the proposed antenna has several advantages over other waveguide the proposed antenna. Therefore, the proposed antenna has several advantages over other CP antennas such as a good CP characteristic with a wide operating frequency and a low profile. waveguide CP antennas such as a good CP characteristic with a wide operating frequency and a low profile.

Table 2. Performance comparison between literature antennas and the proposed antenna.

Dimensions (𝝀 ) 3-dB AR Band (GHz) −10 dB Reflection Ref. 𝒎𝒊𝒏 (Length of Long Side × Height) (Fractional Bandwidth) Coefficient Band(GHz) Appl. Sci. 2020, 10, 6994 14 of 15

Table 2. Performance comparison between literature antennas and the proposed antenna.

10 dB Reﬂection Dimensions (λ ) 3-dB AR Band (GHz) Ref. min Coe−ﬃcient Band(GHz) (Length of Long Side Height) (Fractional Bandwidth) × (Fractional Bandwidth) [7] 1.54 0.81 12.2–12.8 (4.8%) 12.15–12.8 (5.2%) × [11] 0.4 0.44 18.5–22.2 (17%) 19.7–21.2 (7%) × [12] 0.14 0.31 33.5–37.5 (11.3%) 32–40 (22.2%) × [16] 0.78 0.92 28.8–32.85 (13.5%) 26.85–32.85 (20%) × [17] 0.92 0.25 29.69–30.06 (1.2%) 29–31 (6%) × Proposed 0.95 0.26 31.72–38.53 (19.3%) 31.51–39.21 (21.7%) ×

5. Conclusions In this paper, a Ka-band circularly polarized waveguide slot antenna with a cross iris is proposed. By using a hexagonal shape cavity, the proposed antenna achieves a circular polarization characteristic (LHCP) by generating two orthogonal modes. Improved impedance matching is realized by applying a rectangular matching slot, which acts as capacitance and inductance in the equivalent circuit. In addition, widening the axial ratio bandwidth is introduced by using a cross iris. Through adjusting the two different lengths of the cross iris, equal amplitude and 90◦ phase difference between two orthogonal electric field modes are realized over the wide 10 dB reflection coefficient bandwidth. − From the experimental results, a 21.7% (31.51 GHz–39.21 GHz) fractional impedance bandwidth for –10 dB reflection coefficient is obtained. The measured 3 dB axial ratio bandwidth is 6810 MHz (31.72 GHz–38.53 GHz) and the fractional bandwidth is about 19.38%. Because the overlap between the 3 dB axial ratio bandwidth and 10 dB reflection coefficient bandwidth is greater than 88%, we can − conclude that the proposed antenna is well suited for the design goal. Along with the wide operating frequency bandwidth, the proposed antenna has a high gain with a directional pattern showing relatively good agreement between the simulated and measured results. Furthermore, since the proposed antenna is made of all metal and is compact in size, it can be a good candidate for tough operating environments such as radar, spacecraft, and satellite communication.

Author Contributions: The presented work was carried out in collaboration between all the authors. S.-J.Y. wrote the paper. J.-H.C. supervised the research. All authors have read and agreed to the published version of the manuscript. Funding: This work was supported by the Human Resources Development program (No. 20194010201860) of the Korea Institute of Energy Technology Evaluation and Planning (KETEP) grant funded by the Korea government Ministry of Trade, Industry and Energy. Conﬂicts of Interest: The authors declare no conﬂict of interest.

References

1. Rappaport, T.S.; Xing, Y.; MacCartney, G.R.; Molisch, A.F.; Mellios, E.; Zhang, J. Overview of millimeter wave communications for ﬁfth-generation (5g) wireless networks—with a focus on propagation models. IEEE Trans. Antennas Propag. 2017, 65, 6213–6230. [CrossRef] 2. Leung, K.W.; So, K. Rectangular waveguide excitation of dielectric resonator antenna. IEEE Trans. Antennas Propag. 2003, 51, 2477–2481. [CrossRef] 3. Montisci, G.; Musa, M.; Mazzarella, G. Waveguide slot antennas for circularly polarized radiated ﬁeld. IEEE Trans. Antennas Propag. 2004, 52, 619–623. [CrossRef] 4. Fukusako, T. Broadband characterization of circularly polarized waveguide antennas using l-shaped probe. J. Electromagn. Eng. Sci. 2017, 17, 1–8. [CrossRef] 5. Wang, M.; Hu, L.; Chen, J.; Qi, S.-S.; Wu, W. Wideband circularly polarized square slot array fed by slotted waveguide for satellite communication. Prog. Electromagn. Res. Lett. 2016, 61, 111–118. [CrossRef] 6. Ma, X.; Huang, C.; Pan, W.; Zhao, B.; Cui, J.; Luo, X. A dual circularly polarized horn antenna in ku-band based on chiral metamaterial. IEEE Trans. Antennas Propag. 2014, 62, 2307–2311. [CrossRef] Appl. Sci. 2020, 10, 6994 15 of 15

7. Qi, S.-S.; Wu, W.; Fang, D.-G. Singly-fed circularly polarized circular aperture antenna with conical beam. IEEE Trans. Antennas Propag. 2013, 61, 3345–3349. [CrossRef] 8. Zhao, Y.; Wei, K.; Zhang, Z.; Feng, Z. A waveguide antenna with bidirectional circular polarizations of the same sense. IEEE Antennas Wirel. Propag. Lett. 2013, 12, 559–562. [CrossRef] 9. Lin, C.; Ge, Y.; Bird, T.S.; Liu, K. Circularly polarized horns based on standard horns and a metasurface polarizer. IEEE Antennas Wirel. Propag. Lett. 2018, 17, 480–484. [CrossRef] 10. Zhang, T.-L.; Yan, Z.-H. A ka dual-band circular waveguide polarizer. In Proceedings of the 2006 7th International Symposium on Antennas, Propagation & EM Theory, Guilin, China, 26 October 2006; Institute of Electrical and Electronics Engineers (IEEE): Piscataway, NJ, USA, 2006; pp. 1–4. 11. Ferrando-Rocher, M.; Herranz-Herruzo, J.I.; Valero-Nogueira, A.; Rodrigo, V.M.; Rodrigo-Penarrocha, V. Circularly polarized slotted waveguide array with improved axial ratio performance. IEEE Trans. Antennas Propag. 2016, 64, 4144–4148. [CrossRef] 12. Li, T.; Fan, F. Design of ka-band 2 2 circular polarization slot antenna array fed by ridge gap waveguide. × In Proceedings of the 2017 Sixth Asia-Pacific Conference on Antennas and Propagation (APCAP), Xi’an, China, 6–19 October 2017; pp. 1–3. 13. Hirano, T.; Hirokawa, J.; Ando, M. Waveguide matching crossed-slot. IEEE Proc.-Microw. Antennas Propag. 2003, 150, 143–146. [CrossRef] 14. Salari, M.; Movahhedi, M. A new configuration for circularly polarized waveguide slot antenna. In Proceedings of the Asia-Pacific Microwave Conference 2011, Melbourne, Australia, 5–8 December 2011; pp. 606–609. 15. Montisci, G. Design of circularly polarized waveguide slot linear arrays. IEEE Trans. Antennas Propag. 2006, 54, 3025–3029. [CrossRef] 16. Kai, Z.; Jianying, L.; Yi, Y.; Rui, X. A novel design of circularly polarized waveguide antenna. In Proceedings of the 2014 3rd Asia-Pacific Conference on Antennas and Propagation, Harbin, China, 26–29 July 2014; Institute of Electrical and Electronics Engineers (IEEE): Piscataway, NJ, USA, 2014; pp. 130–133. 17. Wu, X.; Yang, F.; Xu, F.; Zhou, J. Circularly polarized waveguide antenna with dual pairs of radiation slots at ka-band. IEEE Antennas Wirel. Propag. Lett. 2017, 16, 2947–2950. [CrossRef] 18. Bhardwaj, S.; Volakis, J.L. Circularly-polarized horn antennas for terahertz communication using differential-mode dispersion in hexagonal waveguides. In Proceedings of the 2017 IEEE International Symposium on Antennas and Propagation & USNC/URSI National Radio Science Meeting, California, CA, USA, 9–14 July 2017; pp. 2571–2572. [CrossRef] 19. Ansys High Frequency Structure Simulator (HFSS); Ansys Corporation: Ansys Drive Canonsburg, PA, USA, 2018; ver.17.2. 20. Pozar, D.M. Microwave Engineering, 4th ed.; John Wily & Sons, Inc.: New York, NY, USA, 2012; pp. 110–113. 21. Lo, Y.; Solomon, D.; Richards, W. Theory and experiment on microstrip antennas. IRE Trans. Antennas Propag. 1979, 27, 137–145. [CrossRef] 22. Gan, T.H.; Tan, E.L. Design of broadband circular polarization truncated horn antenna with single feed. Prog. Electromagn. Res. C 2011, 24, 197–206. [CrossRef] 23. Yuan, H.; Qu, S.; Zhang, J.; Zhou, H.; Wang, J.; Ma, H.; Xu, Z. Dual-band dual-polarized spiral antenna for chinese compass navigation satellite system. Prog. Electromagn. Res. Lett. 2014, 46, 25–30. [CrossRef] 24. Boyd, C.R. Impedance matching of open-ended waveguide radiating elements. In Proceedings of the SBMO International Microwave Symposium, Rio de Janeiro, Brazil, 27–30 July 1987. 25. Park, D.; Qu, L.; Kim, H. Compact circularly polarized antenna utilizing the radiation of the ground plane based on the theory of characteristic modes. IET Microw. Antennas Propag. 2019, 13, 1509–1514. [CrossRef] 26. Balanis, C.A. Antenna theory analysis and design, 3rd ed.; Wiley Interscience: Hoboken, NJ, USA, 2005; pp. 70–79. 27. Chu, Q.-X.; Kang, Z.-Y.; Wu, Q.-S.; Mo, D.-Y. An in-phase output ka-band traveling-wave power divider/combiner using double ridge-waveguide couplers. IEEE Trans. Microw. Theory Tech. 2013, 61, 3247–3253. [CrossRef]

© 2020 by the authors. Licensee MDPI, Basel, Switzerland. This article is an open access article distributed under the terms and conditions of the Creative Commons Attribution (CC BY) license (http://creativecommons.org/licenses/by/4.0/).