ADVANCED QUANTUM MECHANICAL TUNNELLING BASED DEVICES AND AVALANCHE BREAKDOWN FOR AND OPTICAL DETECTION SYSTEMS

A thesis submitted to The University of Manchester for the degree of

Doctor of Philosophy

In the Faculty of Science and Engineering

2019

Omar Saadallah Hamid Abdulwahid Supervisor: Prof. Mohamed Missous

School of Electrical and

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TABLE OF CONTENT

TABLE OF CONTENT ...... 1 LIST OF TABLES ...... 5 LIST OF FIGURES ...... 6 LIST OF SYMBOLS AND ABBREVIATIONS ...... 13 ABSTRACT ...... 18 DECLARATION ...... 20 COPYRIGHT STATEMENT ...... 20 ACKNOWLEDGEMENT ...... 21 DEDICATION ...... 22 Publications ...... 23 Journal Publications ...... 23 Conference Publications ...... 23 Oral and Posters Presentations ...... 24 Awards ...... 25 CHAPTER 1 : INTRODUCTION ...... 26 1.1 Introduction and Motivation ...... 26 1.2 Millimetre-Wave Applications...... 29 1.3 Millimetre-Wave Detection Techniques ...... 32 1.4 Material Systems ...... 33 1.5 Optoelectronics Approach ...... 37 1.6 Integration of Fibre-Wireless Network Systems ...... 38 1.7 Contribution and Thesis Outline ...... 40 CHAPTER 2 : BACKGROUND AND THEORY OF DETECTION SYSTEMS...... 43 2.1 Introduction ...... 43 2.2 Signal Sources ...... 43 2.3 ...... 44 2.4 Mixer Characteristics ...... 47 2.4.1 Conversion Loss (CL) ...... 47 2.4.2 1-dB Compression ...... 47 2.4.3 Third Order Intercept Point ...... 48 2.4.4 Isolation ...... 50 2.4.5 Return Loss ...... 50 2.5 Mixer Configurations ...... 51

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2.6 2nd Subharmonic Mixer ...... 53 2.7 Frequency ...... 55 2.8 Basics of Detection ...... 56 2.9 Detector Characteristics ...... 57 2.9.1 Sensitivity ...... 57 2.9.2 Noise Equivalent Power ...... 58 2.9.3 Tangential Sensitivity and Dynamic Range ...... 59 2.10 Theory of Tunnel ...... 60 2.11 Tunnel Diodes ...... 62 2.11.1 Esaki Tunnel ...... 63 2.11.2 Resonant Tunnelling Diode ...... 64 2.12 Asymmetric Spacer Layer ...... 66 2.13 Operating Principle of ASPAT Diodes ...... 68 2.14 Current Density of ASPAT Diode ...... 69 2.15 Introduction and Overview of APD and PIN ...... 71 2.16 Operational Principle of PIN ...... 73 2.17 Operational Principle of Avalanche Photodiode ...... 74 2.18 Characteristics ...... 77 2.18.1 Quantum Efficiency and Responsivity ...... 77 2.18.2 Dark Current ...... 79 2.18.3 3-dB Bandwidth ...... 81 2.18.4 Internal Gain ...... 82 2.18.5 Punch-Through and Breakdown ...... 83 2.18.6 Noise characteristics ...... 85 2.19 Requirements of Multiplication and Charge Layers ...... 87 2.20 Summary ...... 88 CHAPTER 3 : FABRICATION AND CHARACTERISATION OF ASPAT DIODES . 89 3.1 Introduction ...... 89 3.2 Epi-layer Structure of GaAs/AlAs ASPAT Diode ...... 89 3.3 Mask Design and Fabrication of Discrete ASPAT Diodes ...... 90 3.4 Mask Structures ...... 93 3.4.1 Open, Short, and ASPAT Diode Structures ...... 93 3.4.2 Transmission Line Model Structure ...... 94 3.5 Intrinsic Parameters of ASPAT Diode ...... 98

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3.5.1 Junction Capacitance and Junction Resistance ...... 98 3.5.2 Series Resistance ...... 99 3.6 DC Characteristics of GaAs/AlAs ASPAT Diodes ...... 101 3.7 RF Characteristics of GaAs/AlAs ASPAT Diodes ...... 104 3.7.1 RF Characteristics of the Open and Short Bond Pad Structures ...... 105 3.7.2 RF Characteristics of GaAs/AlAs ASPAT Diodes ...... 111 3.8 InGaAs/AlAs ASPAT Diodes ...... 117 3.9 Extracted Junction Resistance and Curvature Coefficient of ASPAT Diodes ...... 120 3.10 Summary ...... 125 CHAPTER 4 : DESIGN, SIMULATION AND FABRICATION OF COPLANAR WAVEGUIDE ZERO-BIAS ASYMMETRICAL SPACER LAYER TUNNEL DIODE DETECTORS AND MIXERS ...... 126 4.1 Introduction ...... 126 4.2 Electromagnetic Simulation Tools ...... 127 4.3 Coplanar Waveguide Structure ...... 129 4.4 Characteristic Impedance and Attenuation of CPW Structure ...... 130 4.4.1 Conductor Loss ...... 131 4.4.2 Dielectric and Radiation Losses ...... 131 4.5 MMIC Metal-Insulator-Metal ...... 133 4.6 Matching Networks ...... 136 4.7 Modelling of ASPAT I-V Characteristics ...... 139 4.8 Schematic Design and Simulation of Detectors and Mixers using ADS Tool ...... 141 4.9 Mask Layout of the MMIC Integrated Zero-Bias ASPAT Detectors ...... 143 4.10 Fabrication and Measurement of the MMIC Integrated Zero-Bias ASPAT Detectors ...... 147 4.11 Measured and Simulated Un-matched Voltage Sensitivity of 6×6µm² GaAs/AlAs ASPAT Diode ...... 151 4.12 ASPAT Detectors Performances ...... 153 4.12.1 Measured DC Output Voltage ...... 153 4.12.2 Voltage Sensitivity and Noise Equivalent Power ...... 155 4.13 Millimeter-Wave ASPAT Detectors with Antennas ...... 158 4.14 Antenna Design and Performances Evaluation ...... 159 4.15 Overview of Devices Used in Detectors ...... 164 4.16 2nd Subharmonic ASPAT Mixers Performances ...... 169

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4.17 Overview of the Reported Subharmonic Mixers ...... 172 4.18 Summary ...... 175 CHAPTER 5 : PHYSICAL MODELLING AND EXPERIMENTAL CHARACTERISATION OF APD AND PIN PHOTODETECTORS FOR HIGH DATA RATE APPLICATIONS ...... 176 5.1 Introduction ...... 176 5.2 Epi-layer Structures of Photodetectors...... 177 5.3 Fabrication and Small Signal RF Equivalent Circuit Extraction ...... 178 5.4 Experimental Characterisation Tools ...... 182 5.5 Physical Modelling Characterisation Tool ...... 183 5.6 Physical Modelling and Optimisation Details ...... 185 5.7 Dark Currents and C-V Characteristics...... 188 5.8 Optical and Noise Characteristics ...... 194 5.9 Reported PIN Photodetectors ...... 201 5.10 Reported APDs ...... 204 5.11 Summary ...... 210 CHAPTER 6 : CONCLUSION AND FUTURE WORKS ...... 211 6.1 Conclusion ...... 211 6.1.1 Zero-Bias ASPAT Detectors and Mixers ...... 211 6.1.2 High-Data-Rate APD and PIN Photodetectors...... 215 6.2 Suggested Ideas for Future Work ...... 216 6.2.1 Millimetre-Wave Detection Circuits ...... 216 6.2.2 Fabrication of the Optimised APD and PIN Photodetectors ...... 220 APPENDICES ...... 221 APPENDIX-A: QFN Circuit ...... 221 APPENDIX-B: Lab View programme ...... 222 APPENDIX-C: Test Structure Used in the Mask ...... 222

APPENDIX-D: Measured and Simulated 푆11 of the Fabricated 30GHz ASPAT Detector with Open Stub Matching Network ...... 223 REFERENCES ...... 224

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LIST OF TABLES

TABLE 1. 1: A SUMMARY OF FREQUENCY BANDS CATEGORISED BY IEEE ORGANISATION [17] ...... 29

TABLE 1. 2: LATTICE CONSTANT, BANDGAP, EFFECTIVE MASS AND FREE- ELECTRON MOBILITY OF STANDARD BINARY AND TERNARY COMPOUND MATERIALS USED TO REALISE PIN, APD AND TUNNEL DIODES AT 300K...... 36

TABLE 3. 1: EPITAXIAL LAYER STRUCTURE OF ASPAT SAMPLE XMBE#304 90

TABLE 3. 2: CALCULATED SERIES RESISTANCE OF THE GaAs/AlAs ASPAT DIODES ...... 100

TABLE 3. 3: EXTRACTED INTRINSIC AND EXTRINSIC PARAMETERS OF GaAs/AlAs ASPAT DIODES ...... 114

TABLE 3. 4: EPITAXIAL LAYER STRUCTURE OF ASPAT SAMPLE XMBE#326 ...... 117

TABLE 3. 5: EXTRACTED PARAMETERS OF THE In0.53Ga0.47As/AlAs ASPAT DIODES AT ZERO-BIAS...... 120

TABLE 4. 1: REPORTED DIRECT DETECTORS ...... 167

TABLE 4. 2: SOME OF THE REPORTED 2nd SUBHARMONIC MIXERS AND ASPAT MIXERS PERFORMANCES ...... 174

TABLE 5. 1: EPI-LAYER STRUCTURE OF THE STANDARD In0.53Ga0.47As PIN DIODE ...... 177

TABLE 5. 2: EPI-LAYER STRUCTURE OF THE STANDARD

In0.53Ga0.47As/In0.52Al0.48As APD (30A) ...... 178

TABLE 5. 3: STANDARD APD AND PIN DIODES EXTRACTED PARAMETERS AT FULLY DEPLETED BIAS...... 181

TABLE 5. 4: THE STANDARD AND OPTIMISED DEVICES ...... 187

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TABLE 5. 5: KEY FITTING PARAMETERS USED IN SILVACO PHYSICAL MODELLING...... 189

TABLE 5. 6: NOISE CHARACTERISTICS OF THE STANDARD AND OPTIMISED

APDS AND PIN DIODES AT 90%푉퐵푅 BIAS ...... 198

TABLE 5. 7: KEY REPORTED PIN PHOTODETECTOR PERFORMANCES ...... 203

TABLE 5. 8: REPORTED APD PERFORMANCES ...... 207

LIST OF FIGURES

Figure ‎1.1: mm-wave attenuation caused by atmospheric gases, rain and fog [18]. The upper inset shows the promising applications of mm-wave systems...... 30

Figure ‎1.2: Block diagram of (a): Heterodyne detection system, and (b): Direct detection with [24]...... 32

Figure ‎1.3: Energy versus lattice constant for group III-V and II-VI compound semiconductor material systems (solid line is direct, and the dashed line is indirect) at room temperature [32]...... 34

Figure ‎1.4: Block diagram of a fibre-wireless system [58]...... 39

Figure ‎2.1: An ideal mixer representation with two input signals (RF and LO)...... 44

Figure ‎2.2: Sketch of the output frequency spectrum of a non-ideal mixer, where it is assumed that RF power is lower than LO power [66]...... 46

Figure ‎2.3: 1-dB compression point of a non-ideal mixer [72]...... 48

Figure ‎2.4: Basic representation of the third-order intercept point of a non-ideal mixer [72]...... 49

Figure ‎2.5: Single element unbalanced mixer showing LO and RF signals applied to the same terminal side [66]...... 52

Figure ‎2.6: A schematic diagram of a balanced passive mixer using two diodes and hybrid [66]...... 53

Figure ‎2.7: 2nd sub-harmonic mixer architecture using anti-parallel diodes with open and short stubs [85, 86]...... 54

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Figure ‎2.8: Basic Detector circuit. The inset is the non-linear I-V characteristics of a diode...... 56

Figure ‎2.9: Output voltage versus RF input power showing the dynamic range of a detector [101]...... 60

Figure ‎2.10: Schematic of the incident, reflected and transmitted wave functions through a rectangular potential barrier [104]...... 61

Figure ‎2.11: Schematic band diagram of the In0.8Ga0.2As/AlAs DBQWRTD. The AlAs energy band gap is the direct gap value [113]...... 64

Figure ‎2.12: Temperature dependency of (a): GaAs/AlAs and (b): In0.53Ga0.47As/AlAs ASPAT diodes [15, 16]...... 67

Figure ‎2.13: Schematic conduction band profile of ASPAT structure under negative, zero and positive bias [125, 126]...... 69

Figure ‎2.14: Operational principle of a reversed biased PIN photodetector, adapted from [142]...... 73

Figure ‎2.15: Operation of a SACM APD, (a): 2-D structure, (b): Band diagram [55, 146]...... 76

Figure ‎2.16: Absorption coefficients versus light wavelength of different materials [154]...... 78

Figure ‎2.17: APD excess noise factor as a function of multiplication gain (푀) based on local mode theory [145]...... 86

Figure ‎3.1: 3D structure drawing of GaAs/AlAs (XMBE#304) ASPAT diode with its standard CPW bond pad. The inset shows the separation distance (퐷푠푝푟) between the top contact and bottom contact pad ()...... 93

Figure ‎3.2: A 3D schematic and side view of the TLM structure used in the masks. (Note that the image is not to scale)...... 95

Figure ‎3.3: Total resistance versus separated distance (푑푛) of TLM structure. [177]...... 96

Figure ‎3.4: Measured TLM of the top contact of GaAs/AlAs ASPAT XMBE#304 sample...... 97

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Figure ‎3.5: The right side is the 2D sectional view of the ASPAT diode. The left side is the intrinsic component of each layer...... 98

Figure ‎3.6: Measured I-V characteristics of GaAs/AlAs ASPAT (wafer XMBE#304) diodes of (a): 3.7x3.7µm2, (b): 5.8x5.8µm2, (c): 10x10µm2. (d): Log representation of the measured currents showing the non-linear characteristics at zero-bias...... 102

Figure ‎3.7: Measured current densities of the fabricated 3.7×3.7µm2, 5.8×5.8µm2, and 10×10µm2 GaAs/AlAs ASPAT diodes using wafer XMBE#304...... 103

Figure ‎3.8: Measured current densities of the devices from two wafers (XMBE#304 and XMBE#421)...... 104

Figure ‎3.9: Example of the fabricated standard CPW ASPAT diode, open, and short structures of mesa area size 3.7×3.7µm²...... 105

Figure ‎3.10: (a), (b), (c), and (d) are the real and imaginary part of 푆11of open and short bond pad structures. (e) The Smith chart representation and the built circuits of the open and short bond structures in ADS...... 107

Figure ‎3.11: The measured and simulated parasitic capacitance versus frequency of the standard CPW structure for different substrate thicknesses...... 108

Figure ‎3.12: Fabricated optimised one and two-port open bond pad CPW structure. .... 109

Figure ‎3.13: The measured and simulated parasitic capacitance versus frequency of the optimised CPW structure for different substrate thicknesses...... 110

Figure ‎3.14: (a): ASPAT equivalent circuit built in ADS at negative bias, and (b): The measured and simulated real and imaginary parts of 푆11of the one-port CPW GaAs/AlAs ASPAT diode of mesa area 3.7×3.7µm2 at -0.5V bias...... 112

Figure ‎3.15: (a): ASPAT equivalent circuit built in ADS at zero and forward bias, and

(b): The measured and simulated real and imaginary parts of 푆11of the one-port CPW GaAs/AlAs ASPAT diode of the mesa area 3.7×3.7µm2 at zero-bias...... 113

Figure ‎3.16: Smith chart representation of the two-port 2.4×2.4µm² ASPAT diode at zero-bias. Red and blue lines are measured and simulated 푆11 respectively. Red and blue dashed lines are the measured and simulated 푆12 respectively...... 116

Figure ‎3.17: Measured current density of the fabricated In0.53Ga0.47As/AlAs ASPAT diodes...... 118

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Figure ‎3.18: Calculated conductance of GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes...... 119

Figure ‎3.19: Calculated junction resistance and curvature coefficient of the 28.3Å barrier thickness GaAs/AlAs ASPAT diodes. The dots in (a) are the extracted junction resistance using the equivalent circuit model...... 121

Figure ‎3.20: Calculated junction resistance and curvature coefficient of the 28.3Å barrier thickness In0.53Ga0.47As/AlAs ASPAT diodes. The dots in (a) are the extracted junction resistance using the equivalent circuit model...... 122

Figure ‎3.21: The junction resistance and curvature coefficient versus AlAs barrier thickness of the GaAs/AlAs ASPAT diodes at zero-bias...... 124

Figure ‎4.1: A 3D schematic view of a CPW structure on a semi-insulating substrate. .. 129

Figure ‎4.2: Layout representation of nine-fingers interdigital capacitor [193]...... 133

Figure ‎4.3: 3D view of the MIM capacitor...... 134

Figure ‎4.4: (a): 10pF CPW MIM capacitor used in this work (b): Equivalent circuit model of MIM capacitor [202]...... 135

Figure ‎4.5: Measured, equivalent circuit, and MoM S-parameters results of 10pF CPW MIM capacitor...... 136

Figure ‎4.6: Matching circuit using open and short stubs [66]...... 137

Figure ‎4.7: Open and short stubs using CPW transmission lines [204]...... 137

Figure ‎4.8: (a): Two-port SDD model circuit in ADS tool, (b): Measured and fitted curves of the 3.7×3.7µm² GaAs/AlAs ASPAT diode...... 140

Figure ‎4.9: (a): Zero-bias direct detection circuit based 5.8×5.8µm² GaAs/AlAs ASPAT diode, (b): Zero-bias 2nd subharmonic mixer based 3.7×3.7µm² GaAs/AlAs ASPAT diode...... 142

Figure ‎4.10: Matching circuit response of ideal and CPW stubs...... 144

Figure ‎4.11: Final zero-bias ASPAT detector circuit implementation showing the layout design of the matching circuit and MIM capacitor...... 145

Figure ‎4.12: An example of mask design steps of the MMIC zero-bias ASPAT detector of the mesa area size of 6×6µm²...... 147

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Figure ‎4.13: Fabricated MMIC integrated zero-bias ASPAT detectors. (a) and (b) are the X-band detectors, (c) and (d) are the K-band detectors (Note: the images are not to scale)...... 150

Figure ‎4.14: Circuit diagram for voltage sensitivity measurement configuration...... 150

Figure ‎4.15: (a) Equivalent circuit diagram of the QFN detector, (b) Actual photograph of discrete circuit...... 151

Figure ‎4.16: Measured and simulated un-matched voltage sensitivity of 6×6µm² ASPAT diode. The inset is the measured video resistance...... 152

Figure ‎4.17: (a), (b), and (c) are the measured output DC voltage and reflection coefficients (푆11) of the X-band zero-bias detectors based 5.8×5.8µm² and 10×10µm² GaAs/AlAs ASPAT diodes at -27dBm RF power. (d), (e), and (f) are the measured output DC voltage and reflection coefficients (푆11) of the K-band zero-bias detectors based 3.7×3.7µm² and 5.8×5.8µm² GaAs/AlAs ASPAT diodes at -27dBm RF power. 154

Figure ‎4.18: (a) and (b) are the measured and simulated voltage sensitivity and calculated noise equivalent power of the X-band and K-band zero-bias detectors based 5.8×5.8µm² ASPAT diode, (c) is the measured and simulated voltage sensitivity versus input RF power...... 156

Figure ‎4.19: A 3D structure of the proposed ASPAT detector with a bow-tie antenna. (Note: image is not to scale)...... 159

Figure ‎4.20: Top view of the proposed 250GHz bow-tie antenna with (a): Coplanar strip output pads, and (b): Coplanar waveguide output pads...... 161

Figure ‎4.21: Simulated return loss (푆11) of the proposed 77GHz and 250GHz bow-tie antennas on a 100µm GaAs substrate...... 161

Figure ‎4.22: Simulated radiation patterns (gain) of the proposed 250GHz bow-tie antenna on a 100µm GaAs substrate...... 162

Figure ‎4.23: Simulated voltage sensitivity of the zero-bias ASPAT detectors with bow-tie antennas at 77GHz and 250GHz...... 163

Figure ‎4.24: Simulated conversion loss of the 3.7×3.7µm² GaAs/AlAs and 3.75×3.75µm² nd In0.53Ga0.47As/AlAs 2 subharmonic mixers at 77GHz RF signal...... 169

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Figure ‎4.25: 1-dB compression of the 3.7×3.7µm² GaAs/AlAs and 3.75×3.75µm² nd In0.53Ga0.47As/AlAs 2 subharmonic mixers at 77GHz RF signal...... 170

Figure ‎4.26: (a): Spectrum of the IF Current in dB, (b): 3rd intercept points of the 3.7×3.7µm² GaAs/AlAs subharmonic mixers at 77GHz RF signal...... 171

Figure ‎5.1: Fabricated photodetector. The inset shows the light window aperture (W) and

퐷푔푎푝 of the photodetector. (images are not to scale)...... 179

Figure ‎5.2: Measured and simulated 푆11 represented on smith charts of the open and short structures and corresponding equivalent circuits...... 180

Figure ‎5.3: Measured and simulated S-parameters represented on Smith charts and of the standard PINs and APD at fully depleted bias...... 180

Figure ‎5.4: Optical system set up on-wafer measurements...... 182

Figure ‎5.5: Modelled 3D rectangular photodetector...... 185

Figure ‎5.6: Calculated 3-dB optical bandwidth of the optimised In0.53Ga0.47As/

In0.52Al0.48As APD...... 187

Figure ‎5.7: Measured and simulated dark currents of the standard and optimised

(a): In0.53Ga0.47As/In0.52Al0.48As APDs, and (b): In0.53Ga0.47As PIN photodetectors...... 191

Figure ‎5.8: Measured and simulated dark junction capacitance versus bias of the standard and optimised (a): In0.53Ga0.47As/In0.52Al0.48As APDs, and (b): In0.53Ga0.47As PIN photodetectors...... 193

Figure ‎5.9: Simulated electric field distribution of the In0.53Ga0.47As/In0.52Al0.48As standard and optimised APDs under -20V bias...... 195

Figure ‎5.10: Measured and simulated photocurrents of the standard and optimised (a):

In0.53Ga0.47As/In0.52Al0.48As APDs, and (b): In0.53Ga0.47As PIN diodes...... 196

Figure ‎5.11: Measured and simulated internal gain and excess noise factor of the standard and optimised In0.53Ga0.47As/In0.52Al0.48As APDs...... 197

Figure ‎5.12: Normalized 푆21 response of the In0.53Ga0.47As/In0.52Al0.48As APD and

In0.53Ga0.47As PIN diode, (red and black are the measured and simulated standard APD (30A), blue is the simulated optimised APD (15A), green and brown lines refer to the measured and simulated standard PIN diode (15S), and purple is the simulated optimised PIN diode (15D))...... 199

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Figure ‎5.13: Measured and simulated 3-dB Bandwidth versus bias of the standard and optimised In0.53Ga0.47As/In0.52Al0.48As APDs...... 200

Figure ‎6.1: Flow chart of the future work of the zero-bias ASPAT detector for mm-wave and sub-mm-wave applications...... 217

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LIST OF SYMBOLS AND ABBREVIATIONS

µ푛 Electron Mobility

ℎ푐 Critical Thickness

Ɍ퐴푃퐷 The responsivity of APD

Ɍ푃−퐼−푁 The responsivity of PIN photodetector

Zsource The impedance of the source

퐶퐽 Junction Capacitance

퐶푃 Parasitic Capacitance

퐶푑 Displacement Capacitor

퐷푠푝푟 The separation between anode and cathode

퐹3푑퐵 3-dB Bandwidth

퐼푑𝑖푓푓 The diffusion current

퐼푔−푟 The generation-recombination current

퐾푉 Curvature Coefficient

퐿푃 Parasitic Inductance

푅퐽 Junction Resistance

푅푆 Series Resistance

푅푉 Video Resistance

푅푐 Contact Resistance

푅푠ℎ Sheet Resistance

푅푢 Non-linear resistance of the un-depleted layers

푆푉 Voltage Sensitivity

푉0 Barrier Height

푉퐵푅 The

푉푃푇 The punch-through voltage

푉푠푎푡푛 Saturation velocity for

푍푙표푎푑 The impedance of the load

푎퐿 The lattice constant of the grown layer

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푎푆 The lattice constant of the substrate

푓푐푢푡−표푓푓 Cut-off frequency

푔푚 Trans-Conductance

푘푟푎푡𝑖표 The ratio between the hole and electron impact ionisation

푙푒푥푡 Parasitic Extension Length 푚∗ Effective mass

훼푑 Dielectric Loss

훼푟 Radiation Loss

휀푠 Built-in strain

휌푚 The temperature coefficient of breakdown voltage 1/푓 Corner Frequency 2D Two Dimensional 3D Three Dimensional 5G Fifth Generation A/Amp Ampere (Current Unit) AC Alternating Current ADS Advanced Design System AlAs Aluminium Arsenide AlGaAs Aluminium AlSb Aluminium Antimonide APD Avalanche Photodiode AR Anti-Reflection ASPAT Asymmetric Spacer Tunnel Layer Diode BER Bit Error Rate BJT Bipolar Junction CPW Coplanar Waveguide C-V Capacitance-Voltage CW Continuous Wave DBQW Double-Barrier Quantum Well DC Direct Current

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DI De-Ionised DUT Device under Test EBL Electron Beam Lithography

EC Conduction Band EM Electromagnetic eV Electron Volt

EV Valence Band FEM Finite Element Method FIT Finite Integration Technique FLDMOB Field Mobility model FTTH Fibre-to-the-Home GaAs Gallium Arsenide GBP Gain Bandwidth Product Ge GEC General Electrical Company GHz Gigahertz GSG Ground-Signal-Ground HB Harmonic Balance HBT Heterojunction Bipolar Transistor HEMT High Electron Mobility Transistor IC IF Intermediate Frequency InAlGaAs Indium Aluminium Gallium Arsenide InAs Indium Arsenide InGaAs Indium Gallium Arsenide InP Indium Phosphide IoT Internet of Things IP3 Third Order Intercept Point I-V Current Voltage K Kelvin

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LCA Lightwave Component Analyser LNA Low Noise Amplifier LO Local Oscillator M Multiplication Factor (internal gain) MBE Molecular Beam Epitaxy MIC Integrated Circuit MIM Metal-Insulator-Metal ML Mono Layer mm Millimetre MMIC Monolithic Microwave Integrated Circuit MOCVD Metal Organic Chemical Vapour Deposition MoM Momentum of Method MOVPE Molecular Organic Vapour Phase Epitaxy MSM Metal-Semiconductor-Metal ƞ Quantum efficiency NDR Negative Differential Resistance NEP Noise Equivalent Power NiCr Nickel Chromium nm Nanometre pF Pico Farad PON Passive Optical Network PVCR Peak to Valley Current Ratio QCL Quantum Cascade Lasers R_Collector The resistance of the Collector layer R_Emitter The resistance of the Emitter Layer

R_spreading (푅푠푝푟) Spreading Resistance R_top ohmic The resistance of the top Ohmic layer RC Resistance and Capacitance RF Radio Frequency RHS Right Hand Spinner

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RTD Resonant Tunnelling Diode SACM Separated Absorption, Charge, and Multiplication APD SAM Separated Absorption and Multiplication APD SDD Symbolically Defined Device model SHM Subharmonic Mixer Si SIS Semiconductor-Insulator-Semiconductor SNR Signal to Noise Ratio SRH Shockley-Read-Hall SSMBE Solid Source Molecular Beam Epitaxy TBRTD Triple Barrier Resonant Tunnelling Diode THz Terahertz TIA Trans-impedance Amplifier TL Transmission Line TLM Transmission Line Model TLMx Transmission Line Matrix TSS Tangential Sensitivity UHV Ultra-High Vacuum V Volt (Voltage Unit) VNA Vector Network Analyser δ Skin Depth of the film Г Reflection Coefficient 퐶퐿 Conversion Loss 퐹(푀) The Excess noise factor 퐺 Conductance Loss 푅퐿 Return Loss 훼(퐸) Impact Ionisation Rate For the Electron 훽(퐸) Impact Ionisation Rate For the Hole 휌 Resistivity

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ABSTRACT

The work in this thesis was concerned with the analysis, modelling, design, testing and improvement of detectors using InP and GaAs-based technologies for electronic and optical receiver systems.

For the electronic receivers, two types of Asymmetric Spacer Tunnel (ASPAT) diodes were studied and tested for potential microwave and mm-wave applications including novel X-band and K-band zero-bias tunnel diode frequency detectors. The core element of the detectors is a GaAs/AlAs ASPAT diode. DC and high-frequency S-parameter characterisation of diodes of mesa sizes of 1.6×1.6µm², 2.4×2.4µm², 3.7×3.7µm², 5.8×5.8µm² and 10×10µm² were carried out to fully extract their extrinsic and intrinsic components for optimum detector and 2nd subharmonic mixer circuits analysis and design. Coplanar waveguide matching circuit structures were designed and optimised to minimise the mismatch between the RF source and the diode impedance. The detectors were fabricated and experimentally measured in the frequency bands (4 to 18) GHz and (10 to 35) GHz at various input powers. The maximum measured sensitivity is 3650V/W and 1300V/W at 11GHz and 24GHz respectively for -27dBm incident RF power. The minimum calculated noise equivalent power is (~6pW/√퐻푧) and (~20pW/√퐻푧) for the X-band and K-band detectors, respectively. The 1.6×1.6µm² ASPAT offered a maximum sensitivity of (1850V/W) at 250GHz.

The ASPAT diodes were then used in a simulation work to test and examine their performance in mm-wave heterodyne circuits. At 77GHz RF signal, a moderate conversion loss of 10dB was achieved using the 3.7×3.7µm² GaAs/AlAs, while a 16dB was obtained using the 3.75×3.75µm² In0.53Ga0.47As/AlAs ASPAT diodes at 0dBm LO power. These detectors show excellent performances, comparable to reported X-band and K-band detectors based Schottky diodes but with the added advantage of stable operation over a wide temperature range. The results reported here validate the models developed which can be used to realise low cost, extremely low power, temperature-insensitive high-frequency tunnel diode detectors for a range of applications.

The second part of the thesis dealt with telecommunication optoelectronic receivers. Validated SILVACO physical models were exploited to optimise the electrical and optical characteristics of 1.55µm wavelength In0.53Ga0.47As/In0.52Al0.48As avalanche photodiodes (APD) and In0.53Ga0.47As PIN diodes. Optimised SILVACO models were

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created by selectively thinning down the absorption layers to further reduce the carrier transit time. Further optimisation was accomplished through scaling of the light window aperture and mesa area sizes to reduce the device capacitances. The optimised PIN diode provides a maximum optoelectric bandwidth of (35GHz) with a current responsivity of (0.4A/W) under -5V bias and (10µW) incident optical power. At 1µW incident optical power, the maximum optoelectric bandwidth and current responsivity of the optimised are (21GHz) and (1.4A/W) under -21.5V bias. The optimised avalanche and PIN photodetectors are capable of working at a data rate of up to 25Gb/s and 40Gb/s respectively.

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DECLARATION

No portion of the work referred to in the thesis has been submitted in support of an application for another degree or qualification of this or any other university or other institutes of learning.

COPYRIGHT STATEMENT

i. The author of this thesis (including any appendices to this thesis) owns certain copyright‎ or‎ related‎ rights‎ in‎ it‎ (the‎ “Copyright”), and he has given The University of Manchester certain rights to use such Copyright, including for administrative purposes.

ii. Copies of this thesis, either in full or in extracts and whether in hard or electronic copy, may be made only in accordance with the Copyright, Designs and Patents Act 1988 (as amended) and regulations issued under it or, where appropriate, in accordance with licensing agreements which the University has from time to time. This page must form part of any such copies made.

iii. The ownership of certain Copyright, patents, designs, trademarks and other intellectual‎ property‎ (the‎ “Intellectual‎ Property”)‎ and‎ any‎ reproductions‎ of‎ copyright‎works‎in‎the‎thesis,‎for‎example‎graphs‎and‎tables‎(“Reproductions”),‎ which may be described in this thesis, may not be owned by the author and may be owned by third parties. Such Intellectual Property and Reproductions cannot and must not be made available for use without the prior written permission of the owner(s) of the relevant Intellectual Property and/or Reproductions.

iv. Further information on the conditions under which disclosure, publication and commercialisation of this thesis, the Copyright and any Intellectual Property and/or Reproductions described in it may take place is available in the University IP Policy, in any relevant Thesis restriction declarations deposited in the University‎Library,‎The‎University‎Library’s‎regulations‎and‎in‎The‎University’s‎ policy on Presentation of Theses.

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ACKNOWLEDGEMENT

I would like to express my great appreciation to my supervisor, Professor Mohamed Missous, who offered me this valuable opportunity to do research at the University of Manchester and for his highly professional character in the process of consultation and guidance that continuously improved my knowledge during the PhD study. Also special thanks to my fellow colleagues for their cooperation towards understanding the ideas in this study.

At the same time, I would like to show my gratefulness to my parents, especially my father who passed away at the time of submission (May the God (Allah) forgive him and accept his good deeds). My parents have encouraged me along with my life; I dedicate a special and heartiest tribute to them for their help and support.

Finally, my special thanks to the Higher Committee for Education Development in Iraq (HCED) for giving me the opportunity to conduct this PhD research and for their financial support, without them this PhD would not have been started.

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DEDICATION

This thesis is dedicated to my parents, who always supported me through all my studies.

Omar

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PUBLICATIONS Journal Publications

1. O. S. Abdulwahid, I. Kostakis, S. G. Muttlak, J. Sexton, K.W. Ian and M. Missous,‎ “Physical‎ Modelling‎ of‎ InGaAs-InAlAs APD and PIN Photodetectors for >25Gb/s Data Rate Applications”,‎ IET‎ Optoelectronics,‎ DOI:‎ 10.1049/iet- opt.2018.5030.

2. O. S. Abdulwahid, J. Sexton, I. Kostakis, K. Ian, and M. Missous, "Physical modelling and experimental characterisation of InAlAs/InGaAs avalanche photodiode for 10 Gb/s data rates and higher," IET Optoelectronics, vol. 12, no. 1, pp. 5-10, 2017.

3. Saad G. Muttlak, O. S. Abdulwahid, J. Sexton, M.J. Kelly and M. Missous, “InGaAs/AlAs‎ Resonant‎ Tunneling‎ Diodes‎ for‎ THz‎ Applications:‎ An Experimental‎Investigation”,‎IEEE‎Journal‎of‎the‎Electron‎Devices‎Society,‎DOI:‎ 10.1109/JEDS.2018.2797951.

4. Muttlak SG, Kostakis I, Abdulwahid OS, Sexton J, Missous M. Low-cost InP– InGaAs PIN–HBT-based OEIC for up to 20 Gb/s optical communication systems. IET Optoelectronics. 2019 Jan 11;13(3):144-50.

5. K. N. Zainul Ariffin, Y. Wang, M. R. R. Abdullah, S. G. Muttlak, Omar S. Abdulwahid, J. Sexton, Ka Wa Ian, Michael J. Kelly and M. Missous, “Investigations‎ of‎Asymmetric‎Spacer‎Tunnel‎ Layer‎(ASPAT)‎Diode‎for‎ High- Frequency‎ Application”‎ IEEE‎ Transaction‎ Electron‎ Devices,‎ DOI:10.1109/TED.2017.2777803.

Conference Publications

1. O. S. Abdulwahid,‎ Saad‎ G.‎ Muttlak,‎ J.‎ Sexton,‎ M.‎ Missous,‎ M.‎ J.‎ Kelly,‎ “24‎ GHz Zero‐Bias Asymmetrical Spacer Layer Tunnel Diode Detectors”,‎UCMMT‎ 2019, IEEE proceedings, August 2019.

2. Saad G. Muttlak , O. S. Abdulwahid, J. Sexton, M. Missous, M. J. Kelly, “InGaAs/AlAs Resonant Tunnelling Diodes with Highest Negative Differential

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Conductance for Efficient and Cost-Effective mm-wave/THz Sources”,‎UCMMT‎ 2019, IEEE proceedings, August 2019.

3. Abdelmajid Salhi, James Sexton, Saad Muttlak, Omar Abdulwahid and Mohamed Missous, “InGaAs/AlAs metamorphic Asymmetric Spacer Tunnel (mASPAT) Diodes on GaAs substrate for Microwave/millimetre-wave Applications”, UCMMT 2019, IEEE proceedings, August 2019.

4. O. S. Abdulwahid, Saad G. Muttlak, J. Sexton, M. Missous, K. W. Ian, M. J. Kelly,‎ “2nd‎ Subharmonic‎ mixer‎ based‎ asymmetric‎ spacer‎ tunnel‎ diode‎ (ASPAT)”,‎ UCMMT‎ 2017,‎ IEEE‎ proceedings,‎ September‎ 2017,‎ DOI: 10.1109/UCMMT.2017.8068352.

5. Saad G. Muttlak, O. S. Abdulwahid,‎J.‎Sexton‎and‎M.‎Missous,‎“Modeling‎of‎ high‎ gain‎ and‎ μW level power consumption resonant tunneling diode based ”,‎ UCMMT‎ 2017,‎ IEEE‎ proceedings,‎ September‎ 2017,‎ DOI:‎ 10.1109/UCMMT.2017.8068351.

6. K. N. Zainul Ariffin, M. R. R. AbduUah, Y. K. Wang, Saad G. Muttlak, O. S. Abdulwahid, J. Sexton; M. Missous and‎M.‎J.‎Kelly,‎“Asymmetric‎spacer‎layer‎ tunnel diode (ASPAT), quantum structure design linked to current-voltage characteristics:‎A‎physical‎simulation‎study”,‎UCMMT‎2017,‎IEEE‎proceedings,‎ DOI: 10.1109/UCMMT.2017.8068358.

Oral and Posters Presentations

1. O. S. Abdulwahid, Saad Muttlak, J. Sexton, K. N. Zainul Ariffin, M.J. Kelly and M. Missous, “Modelling and Characterization of Zero-Bias Asymmetrical Spacer Layer Tunnel Diode Detectors”, SIOE Conference 2019. Cardiff, Oral Presentation.

2. O. S. Abdulwahid, Saad Muttlak, J. Sexton, M.J. Kelly and M. Missous, “15-35 GHz Zero-Bias Asymmetrical Spacer Layer Tunnel Diode Detectors”, UK Semiconductor Conference 2019, Sheffield, Oral Presentation.

3. O. S. Abdulwahid, Saad Muttlak, J. Sexton, M.J. Kelly and M. Missous, “Modelling and Characterization of Zero-Bias Asymmetrical Spacer Layer

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Tunnel Diode Detectors”,‎ THz Electronics Workshop 2018, Glasgow, Poster Presentation.

4. O. S. Abdulwahid, Saad Muttlak, J. Sexton, M.J. Kelly and M. Missous, “55-80 GHz Detector based Asymmetric Spacer Tunnel Diode (ASPAT)”, UK Semiconductor Conference 2017, Sheffield, Oral Presentation.

5. O.S. Abdulwahid, S. G. Muttlak, J. Sexton, I. Kostakis, K.W. Ian, and M. Missous, “Physical Modelling and Experimental Characterization of High Speed InAlAs/InGaAs Avalanche Photodiode”, Silicon photonics adoption in UK industry 2017, Coventry, Poster Presentation.

6. Omar S. Abdulwahid, Mohd Rashid Redza Abdullah, S. G. Muttlak, K. N. Zainul Ariffin, and Mohamed Missous, “Tunneling Barrier Diode for Millimeter Wave Mixing”, UK Semiconductor Conference 2016, Sheffield, Oral Presentation.

7. Omar S. Abdulwahid, S. G. Muttlak, K. N. Zainul Ariffin, M. Missous,‎“Next generation Gb/s communication system: Optical and RF wireless convergence”,‎ EEE Poster Conference 2016, Manchester, Poster Presentation.

Awards

 Best student paper shortlisting at the UCMMT2019 conference, London.

 2nd best poster presentation at Silicon photonics adoption in UK industry 2017, Coventry

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CHAPTER 1: INTRODUCTION

1.1 Introduction and Motivation

The perceived advantages of have always been making them the prefered choice for ultra-low power and high-speed electronic/optical systems for a range of applications. The recent advances in Molecular Beam Epitaxy (MBE) technique have paved the way for discovery of new device phenomena and growth of multi-layers structures with atomic-level thickness resolution such as heterojunction bipolar (HBTs), avalanche breakdown (APDs) and resonant tunnelling diodes (RTDs).

The last two decades have seen a growing trend towards designing high-frequency communication systems that can accommodate the massive demand for high data-rate wireless communication devices in anticipation of the Internet of Things (IoT) applications. The high-frequency band is also highly beneficial for high-resolution imaging applications [1]. In order to provide high-data-rate systems, the new systems need to work at higher frequencies in both the millimetre-wave (30 to 300GHz) and sub- millimetre-wave bands (0.3 to 3THz). The latter is also known as the terahertz (THz) band. Besides the primary goal of improving the performance of high-frequency devices, the ambition is to reduce the cost of these components [2]. The mm-wave/terahertz frequencies have received much attention, and many efforts have been expanded into making mm-wave/terahertz systems to accommodate the vast need for fast-speed links. To date, the mm-wave and sub-mm-wave frequency bands have shown to be promising regions for various applications such as high-resolution imaging in medical, security and surveillance field; atmospheric monitoring and environment, radio astronomy as well as compact range radars [3]. However, the progress of exploring room-temperature operating mm-wave/THz electronic devices is still in early stages compared to microwave and photonic devices. The lack of robust, powerful and room-temperature operating mm-wave/THz sources and detectors has impeded further progress and broader deployment of this technology leading to what is usually termed as the THz gap in the frequency spectrum.

The most important part of a communication system is the receiver front end, which is responsible for receiving, detecting, and processing information. Therefore, it is

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necessary to realise a detection system that is capable of functioning efficiently in the mm-wave/THz frequency band at both low and high ambient temperatures [2]. Systems are constrained by the best possible integrated components (source, mixer, and detector) to achieve their full potential [4]. These components are the core elements of the wireless communication devices such as mobile phones and tablets.

Minimising the power consumption of such systems in the high-frequency bands is the primary motivation for proposing different structures with different characteristics. High power consumption reduces the running time of portable devices and also raises the temperature of the systems. Heat dissipation techniques are usually utilised to cool down the device temperature, yet, this is not straightforward for small size and compact THz systems. Low-power consumption systems require zero-bias circuits to eliminate the need for an external biasing circuit as well as reducing noise.

Detectors and mixers have been implemented using both two-terminal (diodes) and three-terminal devices (transistors). The latter requires external bias to function as a detector element properly, and moreover, three-terminal devices such as HEMT or HBT transistors must be fabricated with nano-scale features (gate length and base) to reach mm-wave operating frequency [5]. As a result, they require complicated and expensive fabrication processes.

Research has been ongoing for many years to develop zero-bias two-terminal passive elements for high-sensitivity detector circuits at mm-wave frequencies. At higher frequencies, the commonly preferred diode is the metal-semiconductor Schottky diode. This majority carrier diode has fast recovery time and increased rectification efficiency. The barrier height in a Schottky diode controls the flow of electrons by means of thermionic emission. Applying a positive bias across a Schottky diode decreases the effective barrier height and leads to large current flow through the diode. The smaller barrier height is also more effective compared to a p-n diode, resulting in the Schottky diode having higher sensitivity for low power received RF signals [6]. However, in both p-n and Schottky diodes, the number of electrons changes exponentially with temperature.

The implication is that the current is very dependent upon the operating temperature. As a result of that, detector performance based on p-n or Schottky diodes varies as the temperature changes. A developed version of the Schottky diode was suggested with a

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reduced effective barrier height and shifted non-linear point close to zero-bias. Improved performances of a low-barrier InGaAs Schottky for zero-bias mm-wave detection were reported in [7-9]. The backward tunnel diode was also demonstrated as a detector element with its zero-bias feature. Low noise and zero-bias direct detectors fully matched backward diodes offer a high sensitivity exceeding (10000V/W) at mm-wave frequencies [10-12]. Nonetheless, the backward diode is still not commercially available to be implemented in practical circuits due to the limited dynamic range, complicated epi-layer structures and poor reproducibility.

Therefore, there is an urgent need to examine and study new zero-bias diode structures that can overcome Schottky and backward diodes limitations and work effectively at high-frequencies as well as being almost temperature independent. The resonant tunnelling diode (RTD) can be used as a low-noise and room-temperature detector exploiting its non-linear transition before the negative differential resistance (NDR) region. The short intrinsic transit tunnelling time grants these diodes the ability to operate at high-speed with stable switching action, well into the mm-wave/THz regime. For zero- bias operation, a new tunnelling diode called the Asymmetrical Spacer Layer Tunnel diode (ASPAT) developed by RT. Syme [13] and optimised by M. Missous at the University of Manchester [14] has been further investigated and tested in this work. In- depth discussions regarding the ASPAT diode and its principle of operation as well as its main characteristics are reported in chapters two and three. The key feature of this diode is its highly pronounced non-linear characteristic at zero-bias, so it is expected to behave as an efficient zero-bias detector at high frequencies, as well as having other benefits such as temperature insensitivity [15, 16].

The THz field also comprises the high-data-rate optoelectronic devices beyond 10Gb/s. Fibre optic transmission has gained much attention for wide-band analogue and digital systems, and it is expected that very shortly optical links would replace most electrical links where very high transmission data rates are needed as is the case for Fibre-To-The- Home (FTTH) systems. The PIN and APD diodes have been extensively investigated and optimised for data rate up to 100Gb/s. Full-scale characterisations of the photodetectors using available physical modelling tool before the fabricated circuits are helpful in the prediction of prospective performances and to aid in further optimisations. This work includes the design, characterisations and physical modelling of different PIN and APD

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photodetectors for >10Gb/s receivers. The photodetectors are made of InGaAs absorber to detect light at a wavelength of 1.55µm.

1.2 Millimetre-Wave Applications

The millimetre-wave band is defined as the range of frequencies between 30 to 300GHz and correspondingly a wavelength of 10 to 1mm. The band is located between the infrared wave and microwave bands. In general, the microwave and mm-wave frequencies are divided into bands, as described by the IEEE and summarised in table 1.1 [17].

TABLE 1. 1: A SUMMARY OF FREQUENCY BANDS CATEGORISED BY IEEE ORGANISATION [17] IEEE standard X K Ka V W mm-wave band Frequency (GHz) 8-12 12-27 27-40 40-75 75-110 110-300

The sub-millimetre wave band corresponds to the frequencies lying beyond 300GHz and up to 3000GHz and the wavelength correspondingly between (1 to 0.1mm). Despite the great achievements of covering a wide range of promising applications, the development of efficient mm-wave and sub-mm-wave systems is still in progress, and more efforts are needed to realise reliable and high-power solid-state sources and very sensitive detectors of low RF power signal.

The use of the mm-wave frequencies in data transmission and sensing application offers several considerable benefits such as: firstly, high-data-rate due to the wide bandwidth of operation, secondly, short wavelength and thus small size of antenna leading to compact systems, thirdly, mm-waves penetrate through fog, snow and dust much better than optical wavelength, and finally, mm-wave transceivers can be monolithically integrated, resulting in robust, compact, and low-cost systems [18]. The mm-wave band has not yet been extensively utilised, and still, many frequencies can be employed to mitigate the congestion in the microwave frequencies, which can lead to improving the performance of newly emerging (and promising) applications. So, attention is rapidly growing to explore the mm-wave band in many civil and military applications [19]. However, the propagation of mm-waves is limited by atmosphere attenuation rates due to the 29

absorption of gases, rain and water vapour as shown in figure 1.1 leading to dividing the band into sections for various applications such as radar, medical, security and military, wireless communications and others.

Figure ‎1.1: mm-wave attenuation caused by atmospheric gases, rain and fog [18]. The upper inset shows the promising applications of mm-wave systems.

The low attenuation rate in the frequency bands (26 to 42GHz), (70 to 120GHz), and (180 to 280GHz) makes them attractive options for short and long-range wireless transmission in many applications including satellite communications, military, backhaul

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and point to multi-point communications. In [19], a 10Gb/s wireless communication link over a distance of 800m was successfully implemented using InP technology with an operating frequency of 120GHz. There is an ever-increasing demand for high-data-rate systems mainly for the upcoming 5G technology that will work in the frequency band from (24 to 86GHz). However, It was recently stated [20] that 5G technology will initially start deployment at 6GHz, and will then shift to mm-wave frequencies in 5 years. The non-ionised mm-wave frequencies also find use in the medical treatments of tumours using radiation therapy that requires low intensity at frequencies such as 42.25GHz, 46.88GHz, 53.57GHz, and 61.2GHz [21, 22]. The mm-waves can penetrate through materials such as cloths and plastic while it reflects from metals. These properties have encouraged the use of mm-waves in the implementation of imaging systems for security and non-destructive inspection applications [23]. Imaging systems can be classified as either passive or active imaging. The latter uses a source to emit waves and a detector to detect the reflected waves from objects, unlike the passive one, which only uses a detector to sense the thermal behaviour of objects. Passive imaging systems are less complicated and inexpensive, but they require a receiver with low-noise and high sensitivity characteristics [24]. In imaging systems, the choice of the frequency is mostly related to the penetration depth and spatial resolution. The low attenuation rate at 77GHz, 94GHz, 140GHz, and 220 to 280GHz makes these regions key for high- resolution imaging systems. A passive imaging camera for security applications designed to work at a centre frequency of 77GHz was demonstrated in [25]. In [26], the Fujitsu company has developed a 94GHz passive imaging sensor for security applications. The sensor includes a HEMT transistor, low noise amplifier (LNA), and a zero-bias Schottky detector with a voltage sensitivity of (150V/W).

Moreover, the advantages of non-ionised mm-waves can be utilised for realising imaging systems for body scanners in airports as a potential replacement for X-ray technology. Recently, Rohde & Schwarz introduced a fully electronic mm-wave high-resolution body scanner which works without any moving parts in the frequency band of 70 to 80GHz and capable of transmitting a maximum power of 1mW [27]. To date, the exploitation of the 220 to 280GHz band is still in its infancy, and much progress remains to be made to build and realise high-resolution imaging systems.

So far, the most appealing application from a commercial viewpoint is the automotive radar sensor, which is usually installed behind car bumpers. The targeted frequencies

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range is between 76 to 81GHz with a centre frequency of 77GHz where there is low atmospheric attenuation. The high absorption of the bumper materials (plastic and paints) presents a challenging issue in designing such sensor; therefore, the transceiver is usually made of multiple transmitters and receivers to provide high transmitted power and high sensitivity for low-power detected signal [28, 29]. In this thesis, new tunnel diodes as detector elements for wireless communication, car radar, and imaging applications at 24GHz, 77GHz, and 250GHz, respectively will be presented.

1.3 Millimetre-Wave Detection Techniques

Detection of mm-waves is usually performed using coherent (heterodyne) or incoherent (direct) approaches, as shown in figure 1.2. The coherent method detects both the amplitude and phase of the received signal in contrast to the incoherent one where only the amplitude of the received signal is detected [30]. A heterodyne system with a mixer provides higher spectral resolution (푣/∆푣=106) compared to the direct detection one [30]. Direct technique extends the possibility of forming 2D arrays of multi-elements for imaging application without the limitation of LO power and fast detector response that exists in heterodyne one. The narrowband feature and strong directivity of the heterodyne systems make them a suitable choice for astronomical measurements [24].

IF Amplifier Detected signal RF signal Low Pass IF Filter

Diode detector

LO signal (a)

RF Amplifier RF signal Detected signal Low Pass Filter

Diode detector (b)

Figure ‎1.2: Block diagram of (a): Heterodyne detection system, and (b): Direct detection with amplifier [24].

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The direct detectors are highly preferred for low-power and low-cost mm-wave and sub- mm-wave application, mainly when high sensitivity and low noise equivalent power detector is needed. Generally, direct detectors find their use in applications such as wireless communication and imaging systems where high sensitivity is needed [30].

1.4 Material Systems

The exceptional ability to engineer the band-gap of group III-V semiconductors have made them always attractive to designers to be incorporated into different systems that require specific characteristics. There is also the benefit of high electron mobility and saturation velocity, the most prominent features for high-power and high-frequency applications [31]. There are different specific types of growth techniques that have been used to grow the layer structure of tunnelling devices (Esaki diode, RTD, and ASPAT). The preferred growth method for tunnelling structures is the Molecular Beam Epitaxy (MBE) technique. The idea behind this method is the use of beams which originate from heating solid sources such as gallium and arsenic. Once the beams are generated, they are directly condensed onto a spinning substrate under ultra-high vacuum (UHV) condition. MBE technique offers several advantages in making semiconductor crystal ranging from low defect concentration, highly uniform crystal, and high accuracy of thickness at the atomic level during deposition, making it the preferred technique for ultra-thin layer structures. The significant development of MBE technology provides the ability to use these materials in a heterostructure form, where two dissimilar semiconductor materials having different band gaps are brought together in contact, for instance, large band gap (AlAs) and small band gap (GaAs). The heterostructure system opens a new era in designing semiconductor devices since it offers lots of freedom to the designers to obtain different device characteristics. The material pair must have a very close lattice constant in order to prevent the occurrence of broken bonds that cause a disturbance at the heterojunction interface. Any difference in the lattice constant between the materials would create defects inside the crystal. Defects mean localised states at the interface because of the dislocations that can then act as trapping centres for free carriers and result in degrading the performance of the devices. Figure 1.3 depicts the energy band gaps versus lattice constant for different materials of group III-V [32]. In case that the grown layers have a similar lattice constant with the substrate, the structure is called lattice- matched. Examples of lattice-matched structures are the In0.53Ga0.47As PINs and

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In0.53Ga0.47As/In0.52Al0.48As APD grown on InP substrate reported in this work, where all the materials have nearly a similar lattice constant of 5.86Å. However, the large energy band gap difference of (~0.75eV) between the small band gap In0.53Ga0.47As (퐸푔 =

0.75푒푉) and large band gap In0.52Al0.48As (퐸푔 = 1.4eV) creates a band discontinuity that can trap carriers and results in slowing the speed of the photodetector. A secondary layer with an energy band gap of (~1eV) is placed between the In0.53Ga0.47As and In0.52Al0.48As to help reduce the abrupt variation and smooths the transition in the conduction band.

Figure ‎1.3: Energy band gap versus lattice constant for group III-V and II-VI compound semiconductor material systems (solid line is direct, and the dashed line is indirect) at room temperature [32].

The preferred material for such structure is the quaternary Al0.22Ga0.25In0.52As due to its medium band gap energy (~1eV) and good lattice matching condition to the InP substrate. Another example is the GaAs/AlAs ASPAT diode grown on a GaAs substrate. Such a structure has a very small lattice mismatch of (0.001) between the GaAs and AlAs materials. The principle of operation of the ASPAT diode on the contrary to the APD diode relies on forming a band discontinuity which acts as a barrier with an appropriate, effective height based on the materials used. The thin GaAs and AlAs are direct bandgap materials with Γ-Γ‎ tunnelling‎ mechanism, which results in the energy band gap of

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(1.42eV) and (2.83eV) respectively. In the case where the grown and substrate layers have a dissimilar lattice constant, the atoms of the materials at the interface will adapt their location to attain the standard shape of the original lattice. Following that, a distortion occurs at the atomic level, which is typically referred to as a strain. A compressive strain is introduced when the lattice constant of the grown layer is larger than the lattice constant of the substrate (푎퐿 > 푎푆), while a tensile strain occurs when

(푎푠 > 푎퐿) as in the case of the In0.8Ga0.2As/AlAs RTD diode grown on InP substrate and developed at the University of Manchester by Missous [33]. The strain (휀푠) is given by [34]:

푎퐿 − 푎푠 (1.1) 휀푠 = 푎푠

It is necessary to keep the thickness of the grown layer below the critical thickness (ℎ푐) and ensure minimum strain energy is introduced at the junction. The critical thickness is expressed as [34]:

2 푎푠 (1.2) ℎ푐 = 2(푎퐿 − 푎푠)

The most popular, cheap, including compatibility with IC technology, and easy to manufacture materials are silicon (Si) and germanium (Ge). Since the first demonstration of tunnelling diode, many attempts have been accomplished in order to fabricate a tunnelling device using alloy Si-Ge materials. Despite the significant advantages of this material, their characteristics cannot fulfil the requirements for millimetre or sub- millimetre applications. More importantly, the built-in conduction band discontinuity in such structures is relatively small, leading to low effective barrier height for the ASPAT diode and not sufficient effective quantum confinement in double barrier quantum well RTD. The consequences are the increase of temperature-dependency characteristics of the ASPAT diode, while the RTD suffers from extremely low current density [35]. From the optoelectronic side, several attempts have been performed to grow a mismatched Si- Ge PIN and APD structures for use in the 1.3 to 1.55µm wavelength telecommunication band [36-43]. The devices achieved high-bandwidth of operation but with very poor responsivity as a result of low quantum efficiency of the silicon at 1.55µm wavelength. Not surprisingly, the devices have high leakage currents due to the low band gap of the Ge material (0.66eV). So, the most effective way remains III-V materials such as 35

(InGaAs, GaAs, AlAs, and InP) [44]. Such materials have high electron mobilities due to their low effective mass, as shown in table 1.2. These parameters are key to achieving high-bandwidth and low-noise electronic and optoelectronic devices.

TABLE 1. 2: LATTICE CONSTANT, BANDGAP, ELECTRON EFFECTIVE MASS AND FREE-DOPING ELECTRON MOBILITY OF STANDARD BINARY AND TERNARY COMPOUND SEMICONDUCTOR MATERIALS USED TO REALISE PIN, APD AND TUNNEL DIODES AT 300K. Electron Electron Lattice constant Energy gap Alloy Effective mass, mobility (cm2 (Å) (eV) 푚∗ V-1s-1)

Si 5.431 1.1 0.33푚0 1600

Ge 5.65 0.66 0.22푚0 3900

InAs 6.058 0.36 0.023푚0 30000

AlSb 6.135 1.58 0.12푚0 200

GaAs 5.653 1.42 0.063푚0 8000 2.16 (direct) AlAs 5.661 and 2.83 0.15푚0 200 (indirect)

In0.53Ga0.47As 5.868 0.75 0.044푚0 12000

In0.52Al0.48As 5.852 1.44 0.075푚0 2000

InP 5.86 1.35 0.077푚0 4000

The history of tunnel diodes started with the pioneering work conducted by the physicist Esaki, ever since different III-V materials have been employed to improve the non-linear characteristics, current density and output power of the devices.

The backward tunnel diodes were reported with two main structures; homojunction and Heterostructure designs. The homojunction backward diode-based Ge material reported in [45] showed a high current density and curvature coefficient exceeding (40V-1) at room temperature. Nevertheless, the device was capable of working up to a few tens gigahertz frequencies due to its large junction capacitance. For high-frequency operation, a device with small effective mass, high electron mobility, high tunnelling probability, and small mesa area size is favoured. A study carried out in [46] showed that a large

36

mesa area size backward diode-based InAs could offer much higher sensitivity compared to a small mesa area size backward diode-based Ge material at microwave frequencies. The Heterostructure backward diodes based III-V materials were demonstrated with mainly two epi-layer structures: firstly, GaAsSb/InAlAs/InGaAs [47, 48] and secondly, InAs/AlSb/GaAlSb/GaSb [49, 50] grown on semi-insulating InP and GaAs substrates respectively. The highest reported un-matched voltage sensitivity of (1500V/W) at 94GHz and zero-bias was attained using the first structure grown on an InP substrate in

[48]. The indium-rich In0.8Ga0.2As RTD devices were proven to have a high current density and an oscillation frequency of >20mA/µm² and >1THz, respectively [51-53]. Applying the same principle to our ASPAT diodes, it is expected that the InGaAs ASPAT diodes grown on InP substrate would be an attractive candidate for mm- wave/THz detection circuits with the possibility of integration with InP-based high- performance low-noise amplifiers and HEMT transistors.

1.5 Optoelectronics Approach

As mentioned earlier, this work also deals with photodetectors as will be presented in chapter five; hence, it is necessary to explain and discuss key facts of optical communications. Optical fibres have gained much interest as they represent a crucial part of modern communication systems. Services such as video-on-demand (VOD) and video conferencing require a high data rate of transmission and reliable communication. The optical fibre is usually preferred over copper wiring due to its considerably low loss and dispersion, as well as the high bandwidth-length product of up to 106 MbKm/s [54, 55]. Significant developments have been undertaken to increase the operating bandwidth and data rates of optical communication systems for Fibre-To-The-Home (FTTH) and high- speed rack to rack communications systems in data centres operating in the 10s to 100s Gb/s data rates

Passive optical networks (PON) are widely exploited for the FTTH systems due to the low cost of infrastructure and maintenance. PON, however; is not dedicated for long- distance optical links due to the losses associated with splitter/ combiner.

Furthermore, the bandwidth is shared between the users in passive optical networks. The 10Gb/s EPON (IEEE 802.2av, ratified September 2009) has already been deployed in late 2013 [56], and the development of the transmitter and receiver have been in progress

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for the next generation of 25Gb/s system for data centre applications. In this work, a physical model is used to optimise two types of photodetectors for the >25Gb/s receiver applications. Further discussions of the experimental and physical simulation results of the high-speed photodetectors are presented in chapter five.

1.6 Integration of Fibre-Wireless Network Systems

The increasing need for high data rate has recently driven both research and industry toward the investigation of radio-over-fibre technology that can meet the requirements of future communication networks. The use of fibre-wireless (FW) communication system offers many advantages such as ultra-wide bandwidth, long-distance (overcoming the issue of air attenuation for RF signals), high-mobility, and wide-coverage [57]. It is expected that FW transmission systems will provide multi-gigabit data link for many applications such as the new 5G mobile communication, military applications, temporary links in a disaster situation and ultra-fast wireless communication at home.

The new generation of smartphones employs super-high-definition cameras such as 4k or 8k resolution, which will need a high-data-rate transmission exceeding 30Gb/s. The fibre-wireless system would offer high-data-rate of transmission link and more importantly, a much smaller integrated area in thin and light smartphone compared to the large high-definition-multimedia interfaces (HDMIs) or optical connectors.

The simplest schematic diagram of a mm-wave fibre-wireless communication system is shown in figure 1.4, which includes the detection elements in terms of the optical and radio frequency signals. At the central unit, the high-data-rate signal is converted into an optical signal and then amplified using an amplifier. At the remote access unit (RAU), PIN or APD are used to convert the optical signal into mm-wave signal. An amplifier can be utilised with a PIN diode to amplify the signal before it is sent into the air using an antenna. At the mobile terminal, the mm-wave signal is detected using, usually, a two- terminal device for simplicity and low cost.

38

Figure ‎1.4: Block diagram of a fibre-wireless system [58].

Ka-band and W-band have gained much interest due to their large bandwidth for military and wireless communication systems. PIN diodes with bandwidths of 60GHz are currently used in fibre-wireless network transceivers for 1.5Gb/s in-building HD video delivery over a distance of 12.5Km [59]. In [60], an integrated 50Gb/s fibre-wireless network was successfully demonstrated at 60GHz RF signal. The optical signal was sent over a distance of 1Km from the central unit into the (RAU) one. The optical signal was then split into two signals and converted into electrical form using PIN diodes. Two transmitting and receiving antennas were exploited to send and receive the mm-wave signal between the (RAU) and receiver end over a distance of 4m.

The ASPAT detectors and avalanche photodetector presented in this work could be promising candidates for such systems in which high-sensitivity and temperature- insensitivity features can play an important role in long-distance and severe weather conditions.

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1.7 Contribution and Thesis Outline

The objectives of this project included two main tasks and all devices used in this work were grown by Molecular Beam Epitaxy (MBE), at the University of Manchester. The main focus of the first part was to study and examine the characteristics of novel tunnel diodes termed Asymmetrical Spacer layer Tunnel (ASPAT) diode for use as zero-bias highly sensitive microwave and mm-wave detectors. This work aimed to study the DC and RF performances of ASPAT diodes for detector applications. The initial structure was previously demonstrated by Syme and Kelly [13, 15, 61]. These initial works only measured the sensitivity of discrete GaAs/AlAs ASPAT diodes up to (~9GHz). The work reported here firstly extends and describes the DC and high-frequency electrical characterisation up to 40GHz of diodes with mesa area sizes of 1.6×1.6µm², 2.4×2.4µm², 3.7×3.7µm², 5.8×5.8µm², and 10×10µm² as well as the extraction of the diode parameters using‎ equivalent‎ circuit‎ model‎ from‎ Keysight’s‎ Advanced Design System (ADS) software. This part also shows for the first time the variation of the non-linear characteristics with respect to the AlAs barrier thickness for different mesa area sizes ASPAT diodes.

Secondly, the work focused on the analysis, design, and fabrication of integrated zero- bias GaAs/AlAs ASPAT detectors and 2nd subharmonic mixers at microwave and mm- wave frequencies. Integrated frequency detectors with coplanar waveguide matching circuits were designed and fabricated to operate in the frequency bands 4 to 18GHz and 10 to 35GHz. This effort presents an experimental work of a complete integrated zero- bias frequency detector based on ASPAT diodes. The measured sensitivities of the integrated ASPAT detectors showed excellent correlation with the simulated ones for wide frequency bands and different RF input power as will be discussed later. Additionally, 2nd subharmonic mixer based GaAs/AlAs and

In0.53Ga0.47As/AlAs ASPAT diodes were designed, and their performances evaluated at 77GHz.

Finally, bow tie antennas were designed and simulated using the CST studio tool and then exported to the ADS schematic design platform to evaluate the whole performances of the ASPAT detector with an antenna. The validated detector models represent a suitable platform for the design and realisation of mm-wave/THz ASPAT detectors and mixers. However, this requires an ASPAT diode having a small junction resistance (푅퐽),

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small series resistance (푅푆), small junction capacitance (퐶퐽), and high curvature coefficient value at zero-bias. More importantly, care has to be taken to minimise the losses due to the inductance and capacitance parasitic effects, which naturally have a significant impact on high-frequency performances.

The second part of this work was the characterisation and physical modelling of avalanche breakdown (APD) and PIN photodetectors for high-data-rate optical receiver applications. Different APD and PIN photodetectors were individually designed, fabricated, and then characterised under dark and light conditions. Small-signal equivalent circuits were built, and their intrinsic parameters were extracted up to 40GHz. The main objective of this work was to build a physical model for an

In0.53Ga0.47As/In0.52Al0.48As APD and In0.53Ga0.47As PIN photodetectors to validate the measured electrical and optical characteristics and which can then be used for further device improvements.

This thesis covers six chapters. The first chapter gives a brief description of the main issues of the currently used diodes materials in the integration of mm-wave/terahertz receiver systems as well as the promising applications for the mm-wave band. In this chapter, an introduction of the new proposed ASPAT diode is given as an alternative candidate for room temperature and zero-bias high-speed detection technique. Furthermore, this chapter gives a quick introduction to the optical receiver system and its relevant topics such as FTTH and PON systems.

Chapter two discusses the heterodyne mixer and direct detector configuration alongside with their prominent figure of merits. The operation principle of the 2nd subharmonic mixers is explained. The chapter also reviews the operation of direct detectors and their characteristics as well as possible ways to enhance them. Later in this chapter, a discussion of quantum mechanical tunnelling phenomena, including the operational principle of the ASPAT diode is outlined. Finally, Chapter two introduces the background and state of the art of APD and PIN photodetectors starting from their working principle and their figure of merits, as well as the required mathematical equations for the estimation of the RF characteristics at high frequencies which are explained in details. The most important characteristics of different structures and epi- layer designs are compared and highlighted.

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Chapter three presents the DC and high-frequency characterisation of the GaAs/AlAs and

In0.53Ga0.47As/AlAs ASPAT diodes and their parameters extraction methods. The chapter shows the approaches used to minimise the parasitic capacitance of the bond pad structures. The last section summarises the most important non-linear characteristics of ASPAT diodes of different mesa area sizes and AlAs barrier thicknesses.

Chapter four is dedicated to the analysis, design, and fabrication of the ASPAT detector and mixer circuits, including their passive components such as matching networks and coplanar waveguide MMIC capacitor. The measurement result of the X-band and K-band zero-bias GaAs/AlAs ASPAT detectors is given at different RF input power. The chapter also includes the simulated performances of the 2nd subharmonic mixers based

GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes. For more insight into the mm-wave region, simulated results of the ASPAT detectors with designed bow-tie antennas are provided at 77GHz and 250GHz frequencies.

Chapter five involves the experimental and physical characterisation tools for DC, AC, and light characteristics. The last part presents the simulation and experimental results, including dark current, capacitance-voltage characteristics, high-frequency S-parameters, photocurrent, and 3dB bandwidth of the fabricated and modelled PIN and APD photodetectors for high-data-rate exceeding 10Gb/s. The last chapter comprises the conclusions and possible ideas that could backwards in future. This chapter highlights the most significant achievements that have been achieved through the PhD programme and potential developments that could be done to extend and investigate this work further.

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CHAPTER 2: BACKGROUND AND THEORY OF

DETECTION SYSTEMS

2.1 Introduction

This chapter focuses on the basic operating principle and characteristics of 2nd subharmonic mixers and RF detectors which discussed in-depth prior to the design and realisation of such circuits. This is followed by the theory of tunnel diodes and particularly the new Asymmetrical Spacer Layer Tunnel (ASPAT) diode. The last section presents the operation principle of PIN and avalanche photodiodes and their main characteristics.

2.2 Signal Sources

A signal source represents the heart of the communication system at the transmitter side. Two and three-terminal devices have been successfully employed to produce continuous wave (CW) signal sources. High output power, compact, and room temperature operating sources are highly preferred in the mm-wave/terahertz frequency regimes. Among all types, the quantum cascade laser (QCL) diode has shown superior performances in the THz region, recording a relatively high-power of 1.2mW at 2.1THz [62]. However, QCL sources only work at low temperature (typically < 77K), so they require a cooling system to generate the radiation. Gunn diodes, on the other hand, have excellent characteristics of high power at millimetre-wave frequencies. The work reported in [63] achieved a maximum output power of 98µW at 164GHz using an In0.53Ga0.47As . However, the device size was 1.3µm×120µm, which occupies a large volume on the chip. Three-terminal high electron mobility transistors (HEMTs) have also been used as THz sources in many reported works. It was reported in [64], that a 15nm gate length HEMT transistor had a cut-off frequency of 610GHz permitting operation at least up to 200GHz. The small gate length requires a highly precise lithography technique, which makes the production of such a transistor highly expensive. Naturally, for a reasonable operating gain, HEMT sources must work at a frequency region that is is much lower than their cut-off frequency [65].

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The Resonant Tunnelling Diode (RTD) has shown promising features that can overcome the limitations mentioned above and meet the requirement of a solid-state, robust, compact, small size, room temperature operating source. The integration of RTD with antenna has attracted many researchers to realise complete THz oscillator. RTD oscillators are still in the optimisation process to increase their output power in sub-mm- wave/THz frequency ranges. More discussion regarding the resonant tunnelling diode is presented later in this chapter.

2.3 Frequency Mixer

The first mixer circuit was realised using a which served as a frequency converter. A frequency mixer is a three-port device, with two inputs and one output. Mixers represent an essential part of modern communication systems. The history of mixers dates back to World War II, where they played an indispensable role in maintaining reliable and stable communication over long distances. In transmitters, they convert the low-frequency information signal into a high-frequency signal by mixing it with a carrier signal called the local oscillator signal (LO). This operation is called up- conversion.

On the other hand, in receivers, the opposite happens, where the received radio frequency signal (RF) is down-converted to lower frequencies, for easier processing and analysis [66, 67]. An ideal mixer output includes two components that represent the sum and difference of frequencies (푅퐹 + 퐿푂) and (푅퐹– 퐿푂) respectively. One can pick one component and filter out the unwanted one. Figure 2.1 illustrates the ideal mixer model.

RF IF= RF+LO (Up) IF= RF-LO (Down)

LO

Figure ‎2.1: An ideal mixer representation with two input signals (퐑퐅 and 퐋퐎).

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The mathematical representation of the mixer is derived as below [68]. The two input signals are:

푋퐿푂(푡) = 퐴퐿푂 cos (휔퐿푂 t) (2.1)

푋푅퐹(푡) = 퐴푅퐹 cos (휔푅퐹 t) (2.2) where (퐴퐿푂) and (퐴푅퐹) are the amplitude of the local oscillator and RF signals respectively. As mentioned before, the mixer multiplies the two input signals, and thus the output is given by:

퐴 퐴 푋 (푡) = 퐿푂 푅퐹 [cos( 휔 + 휔 ) 푡 + cos( 휔 − 휔 ) 푡 ] (2.3) 퐼퐹 2 퐿푂 푅퐹 퐿푂 푅퐹

Any device with non-linear characteristic can perform the mixing process; the most commonly used one is the two-terminal diode. The single diode was employed to convert the radio frequency (RF) signal with a frequency exceeding 200GHz [68]. Unfortunately, the non-linear component is not a perfect multiplier. As a result of that, the mixer output contains a large number of harmonic signals in addition to the sum and difference components of the input signals. The spurious signals may interfere with the output signal, and in turn, degrade mixer performance. Taking the Taylor expansion series for the exponential I-V characteristics of a non-linear diode would result in having the following expression [69]:

2 3 퐼(푉) = 푎0 + 푎1 푉 + 푎2푉 + 푎3푉 + ⋯ (2.4) where I(V) is the total output current of the diode as a function of the sum of two input signals (푉). (푎푛) is the amplitude of the harmonic component with 푛=1,2, and so on. As a result, the practical mixer output would have extra frequency products (interference signals), having the general form: ±푛푓퐿푂±푚푓푅퐹. (푛 and 푚) are positive integer numbers. In a receiver communication system, the received RF signal power is usually much lower than the LO signal. Thus, the output frequency component can be simplified to be written as (±푛푓퐿푂±푓푅퐹). Figure 2.2 shows the spectrum of the output signal of a practical diode mixer based on the assumption of RF power << LO power [66, 69].

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LO

f

LO

RF

f

2f

RF

RF

- f -

+ f +

LO

LO

LO

f

3f

f

RF

RF

- f -

LO

RF

+ f +

2f

- f -

LO

Amplitude

RF

LO

2f

3f

+ f +

LO 3f

Frequency(Hz)

Figure ‎2.2: Sketch of the output frequency spectrum of a non-ideal mixer, where it is assumed that RF power is lower than LO power [66].

Since the mixer generates two components at two different frequencies (sum and difference), the RF power is then divided between these components according to equation (2.3). Consequently, the output (IF) signals power is almost (3dB) less than the input signal (RF) [66]. It can be concluded that a mixer operates as a , where the RF input signal is multiplied periodically at a constant rate (LO frequency). In the frequency domain, the RF signal is multiplied inside the diode with the DC component, fundamental and a large number of LO signal harmonics. Furthermore, working as a switch means that the diode is changing its state from on (forward bias) to off (reverse bias). The transition is controlled by the LO pumped signal [69]. Mixer based element with a non-linear positive resistance like a Schottky diode is called a passive mixer. Such a mixer is widely used in millimetre and sub-millimetre wave applications because it is broadband, inexpensive, and easy to design. However, it requires high LO power to work as a mixer efficiently.

On the other hand, mixer based on an active device such as a transistor has a significant benefit of high gain at low LO power. However, it suffers from a high noise figure compared to the passive one, as well as complicated designs at high frequencies [70].

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Among all transistor types, the FET transistor is used as a mixer where the gate-source voltage drives its drain-source resistance. At low frequencies (typically below 1 GHz), no LO power is required and the FET act as a passive mixer [68].

2.4 Mixer Characteristics

In general, there are several performance metrics to differentiate between mixers. This section summarises them according to the main metrics mentioned in [66, 67, 71].

2.4.1 Conversion Loss (CL)

This is the most important indicator to measure the efficiency of a mixer. It is given as the ratio of the input RF signal power to the output IF signal power.

푃 (푅퐹) 퐶퐿= 𝑖푛 (2.5) 푃표푢푡(퐼퐹)

CL is usually given in decibels value and expressed as:

퐶퐿 (푑퐵)= 푃𝑖푛(푅퐹)푑퐵 − 푃표푢푡(퐼퐹)푑퐵 (2.6)

If CL is a positive value, then the input signal would have lost some power and is being attenuated, while a negative value means that the mixer is amplifying the input signal as well as converting its frequency. Many factors affect the conversion loss, such as the device type and size, non-linearity; LO power level, port isolation and finally impedance matching.

2.4.2 1-dB Compression

At relatively low RF power (small-signal), the LO signal controls the switching action of the diode. Therefore, a mixer behaves linearly and produces an output signal whose power is directly related to the input RF power. At high RF power, the mixer behaves as a non-linear system and the input-output relationship is no longer constant. 1-dB compression point refers to the input power level at which the CL increased by 1 dB, as shown in figure 2.3 [72]. In compression mode, the diode becomes controlled by the high

47

RF power and the applied LO power. As a result, the diode is being partially turned on, and the mixer spreads power over all frequency components. A low RF power is required to avoid the degradation of mixer performances. In the case when a high RF power is applied, a high turn-on voltage diode is preferred where its 1-dB compression point is far enough from the input RF power. However, there is also the need to apply a large LO signal to ensure the diode is not compromised by the RF power. Zero-bias mixers have low turn-on voltage, and therefore, the RF power should be kept as small as possible.

Figure ‎2.3: 1-dB compression point of a non-ideal mixer [72].

2.4.3 Third Order Intercept Point

This metric indicates the level of the undesired product at the mixer output under a high level of the input signal. It occurs when two or more signals having enough power to turn on the diode enter the RF port, they mix with the LO and produce interference signals that are close to the desired output signal. The resultant interference signals sit at a frequency equal to (2푓푅퐹1 − 푓푅퐹2 − 푓퐿푂) and (2푓푅퐹2 − 푓푅퐹1 − 푓퐿푂), and hence, they grow with a slope as shown in figure 2.4. This is measured by applying two input signals with a small frequency space. The point at which the power of interference products is the same as the power of output fundamental signal is called the third order intercepts point. In a practical mixer circuit, the diode saturates, and the 1-dB compression point occurs before the third-order intercept point takes place.

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Figure ‎2.4: Basic representation of the third-order intercept point of a non-ideal mixer [72].

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2.4.4 Isolation

Isolation measures the effect of the signal coming from one port to the other ports of the mixer. Since the power level of the LO signal is higher compared to the other ports, it is essential to ensure that the LO power does not leak to the RF and IF terminals. Isolation is experimentally evaluated by applying a signal with a specific power at one port and measuring the available power at the other ports. In mixers, there are three types of isolation denoted as (퐿푂 − 푅퐹) isolation, (퐿푂 − 퐼퐹) isolation and (푅퐹 − 퐼퐹) isolation. (퐿푂 − 푅퐹) isolation assesses the leakage of the signal from the LO side to the RF side, and it is expressed, in dB as [73]:

푃퐼푠표(퐿푂−푅퐹)= 푃𝑖푛(@퐿푂) − 푃표푢푡(@푅퐹) (2.7)

Poor (퐿푂 − 푅퐹) isolation causes significant problems where the high LO power interferes with the RF amplifier and causes cross-channel interference. In up-conversion mode, it is more challenging since LO and RF frequencies are very close to each other. Typical (퐿푂 − 푅퐹) isolation value can be between (25 to 35dB). Similarly, weak (LO- IF) isolation causes LO power to leak and saturate the IF output amplifier. (퐿푂 − 퐼퐹) isolation ranges from (20 to 30dB). On the other hand, (푅퐹 − 퐼퐹) isolation is not a significant issue as RF and IF powers, are much smaller than the LO power. (푅퐹 − 퐼퐹) isolation value ranges between (25 to 35dB). Generally, Better isolation can be accomplished by using open and short stubs to provide virtual grounding to the signal at other ports [66].

2.4.5 Return Loss

This metric plays a vital role in designing all RF systems. It shows the degree of matching between the load (푍푙표푎푑) and source (Zsource) impedances. Return loss is a measure of reflected power from the load to the transmitter. Reflection is highly dominant in mixers since it has three ports with various power levels and frequencies. Return loss can be calculated by specifying the impedance mismatch or what is usually called the reflection coefficient (Г) between the ports, which is given by [74]:

푍 − 푍 Г = | 푙표푎푑 푠표푢푟푐푒| (2.8) 푍푙표푎푑 + 푍푠표푢푟푐푒

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Hence the return loss (푅퐿) is given as:

푅퐿 = −20푙표푔 (Г) (2.9)

푍푠표푢푟푐푒 is‎ usually‎ 50Ω‎ in‎ RF‎ systems.‎ 푍푙표푎푑 varies with applied signal frequency and power, and therefore; many iterative processes are required to design a matching circuit to mitigate the power reflection.

2.5 Mixer Configurations

Mixers can be implemented using passive or active elements. Passive mixers use diodes such as Schottky, or other two-terminal devices. They are well-known for their simplicity at high-frequency operation, low cost, small sizes, low noise figure, and high dynamic range. Practically, passive mixer converts the frequency of the applied signal as well as attenuating its amplitude. The attenuation is defined by the mixing performance of the used element and expressed by conversion loss. Active mixers, on the other hand, utilise transistors which amplify the input signal and thus, introduce conversion gain instead of conversion loss. However, the zero-bias operation is not applicable to active mixers as they need external bias circuits to feed the active components. Moreover, active mixers are expensive and more complex. For THz frequencies, passive mixers using Schottky diodes are highly preferred and have been designed and fabricated at frequencies exceeding 0.5THz [75-77].

According to the literature in [66, 71, 78], mixers can be classified into two main kinds; fundamental and subharmonic configurations. The latter is discussed separately in the next section. Generally, fundamental mixers can be viewed as two major types: unbalanced and balanced mixers. The unbalanced mixer uses a single element (passive or active), as shown in figure 2.5 to perform the mixing operation. In such type, both LO and RF signals are applied to the same terminal side, while the output IF signal is extracted from the other terminal. Single element unbalanced mixers are used in high- frequency applications up to sub-millimetre wave range because of their simple architecture and low cost. However, they suffer from high conversion loss especially in the case of passive elements, high noise figure, limited bandwidth, and finally very poor isolation between the ports as a direct result of applying the signals (RF and LO) at the same side.

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RF

RF Filter IF Filter Non-linear element

LO Fi lter

LO

Figure ‎2.5: Single element unbalanced mixer showing LO and RF signals applied to the same terminal side [66].

Filters at the input and output sides suppress the undesired tones around the input signals that could mix inside the non-linear element and cause multi-tone intermodulation distortion. Furthermore, the filter can provide a good separation between the ports and improve the isolation. Other single element mixer circuits were introduced using FET transistor such as gate pumped trans-conductance mixer, drain pumped trans- conductance mixer and resistive mixer [79]. Gate pumped mixer uses two voltage sources to drive the FET transistor to its saturation region. Both LO and RF signals are applied to the gate of the device through filters which serve as isolation blocks. The mixer performs its operation by changing the FET state from the saturation region into the cut-off region. Gate pumped mixers exhibit good conversion gain at low-frequency ranges but at the expense of high power dissipation. A drain pumped mixer does not require a bias at the drain side of the FET transistor and instead uses the LO signal to control the trans-conductance (푔푚). The FET transistor is biased between the linear and saturation regions to achieve high (푔푚) and high non-linear characteristics. The resistive mixer was also introduced as another solution for the active mixers. In this mixer type, the transistor works as a voltage-controlled which eliminates the need of bias at the drain side. The resistive mixer is treated as a balanced mixer due to its good isolation, which comes as a direct result of applying the LO and RF signals at the gate and drain sides, respectively. Moreover, the resistive mixer introduces low-noise and low-distortion at low LO power. Balanced mixers employ two diodes with hybrid, as shown in figure

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2.6 (or coupler for active configuration) to ensure that the two inputs (LO and RF) are well-separated.

Figure ‎2.6: A schematic diagram of a balanced passive mixer using two diodes and hybrid [66].

The balanced mixer offers better isolation, large operating bandwidths, and wide dynamic range. However, it has high conversion loss and requires high LO power level. The double balanced mixer performs a frequency conversion process using four devices commonly diodes in a ring configuration and a pair of hybrids. Compared to the previously mentioned types, it offers large bandwidth, better isolation and linearity, and higher third-order intercept point. The drawbacks are the high LO power requirement with higher conversion loss. Moreover, the diodes used and hybrids should have the same characteristic as much as possible, which increases design complexity.

2.6 2nd Subharmonic Mixer

The subharmonic mixer (SHM) is the most preferred topology for millimetre and sub- millimetre-waves applications. SHM was firstly introduced by Cohn., Schneider, and Snell in the 1970s using anti-parallel Schottky diodes [80]. The diodes generate current at a frequency equal to (2×푓퐿푂), and then mixing it with the RF signal (푓퐿푂 is the frequency of the local signal). The implication though is that this SHM called 2nd SHM works with only half of 푓퐿푂 compared to the fundamental one. Having such a mixer operating at half

푓퐿푂 frequency means more LO power would be available for the RF signal conversion and above all, much lower LO noise is introduced in the mixer circuit. This feature is crucial for high-frequency applications due to the difficulty in having local sources with high output power [81]. The obtained conversion loss of SHM is usually higher by (3 to

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5dB) than that which could be obtained in the fundamental types at the same frequency

th [82, 83]. Anti-parallel diodes also generate a 4 harmonic of the 푓퐿푂 , and this can be employed to build 4th SHM. The constraint of using anti-parallel diodes is that both diodes should be as identical as possible. Otherwise, any difference in their I-V characteristics leads to degrading the mixer performance [66]. Figure 2.7 shows the schematic structure of a 2nd subharmonic mixer using anti-parallel diodes and termination stubs. Open and short stubs play a vital role in providing good frequency separation between the ports as follow [66, 83, 84]:

1- Open stub at a quarter of the LO wavelength (휆퐿푂/4) located at the RF side which guarantees that maximum LO power is transferred to the diodes. This stub works as termination at LO frequencies and an open circuit at RF frequencies.

2- Similarly, the shorted stub (휆푅퐹/2) at the LO side does the same for the RF signal.

3- The two stubs (휆푅퐹/4) at the IF side offer a virtual ground to the RF signal. Thus no leakage of the RF signal to the IF output port occurs.

λLO/4

Diode1 RF LO Diode2 λRF/4 λRF/2 λRF/4

IF

Figure ‎2.7: 2nd sub-harmonic mixer architecture using anti-parallel diodes with open and short stubs [85, 86].

A subharmonic mixer based transmission line (TL) stubs suffer from a narrow bandwidth. A directional coupler can be used as an alternative of these stubs for a considerable improvement in the operating bandwidth. However, this leads to degraded

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퐿푂 − 푅퐹 isolation [87]. A subharmonic mixer generates an output signal with a frequency equal to [66, 84]:

푓퐼퐹 = 푚푓푅퐹 ± 푛푓퐿푂 (2.10)

In the 2nd subharmonic mode, (푚 + 푛) is an odd integer. The fundamental and odd mixing components are substantially eliminated by the diode pair [66, 84].

2.7 Frequency Detector

A detector is used to directly demodulate the received RF signal using a non-linear element, usually a diode. The primary disadvantage of this method is the higher flicker noise, which is the main contributor to the total noise. In heterodyne one, this noise is much reduced, and the signal-to-noise ratio is improved since the generated IF signal is above the corner frequency (1/푓) [88]. Any three or two-terminal device with non-linear I-V characteristics (inset of figure 2.8) can be utilised as a detector. The non-linear element produces a DC voltage that is proportional to the amplitude of the received signal [13]. At low input power, the output voltage corresponds to the square of the input voltage and this region is called the square-law regime. At high input power, the output voltage behaves linearly with the input voltage and the region is called the linear regime. The saturation region is defined as the point where the output voltage starts to be constant at ultra-high power. Detector can be viewed as a power to voltage or power to a current convertor device. Figure 2.8 depicts the basic configuration of a detector circuit using a single element.

The detector circuit is divided into three sections. Firstly, is the input part and includes a matching circuit to mitigate the signal losses. Secondly, is the detection element and could be a passive or active device. Thirdly, is the output circuit which is formed of a capacitor and resistor in parallel connection. (퐶표푢푡) helps to remove the undesired RF components. Similar to the mixer circuit, FET transistor can be exploited as a direct detection element due to the non-linear relation between drain to source and gate to source voltages.

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Diode RF Matching circuit

3

2

1 Cout RL Current (mA)

0

-2 -1 0 1 2 Voltage (V)

Figure ‎2.8: Basic Detector circuit. The inset is the non-linear I-V characteristics of a diode.

2.8 Basics of Detection

The principle of detection in direct detectors is based on the extraction of the low- frequency information signal from the received RF modulated signal using the non-linear characteristics of the semiconductor devices [89]. The square-law operation produces an output signal amplitude that is proportional to the square of the input signal. A DC source might be necessary to bias the diode at a non-linear operating point. The asymmetric I-V curve shown in the inset of figure 2.8 can be written using the Taylor series expansion [90, 91] as follow:

2 3 푖 = 푎0 + 푎1푣 + 푎2푣 + 푎3푣 +... (2.11)

where 푎0 is equal to zero when 푖 = 0 and 푣 = 0. The input RF signal has the following form:

푣 = 퐴 cos (휔푡) (2.12) where 퐴 is the amplitude of the RF signal, and 휔 is the angular frequency. Substituting 2.12 in Eq. 2.11 would result in:

2 3 푖 = 푎1(퐴 cos(휔푡)) + 푎2(퐴 cos(휔푡)) + 푎3(퐴 cos(휔푡)) +.. (2.13)

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Solving equation 2.13 results in having many harmonic components in the output current. Fortunately, the output capacitor in the detector circuit removes all the undesired 2 components and leaves the DC term (푎2퐴 /2) flowing in the load resistor. Therefore, the output signal is given by:

퐴2 푀 = 푎 =‎Ɍ 푃 (2.14) 표푢푡 2 2 푠

where (Ɍ) is the voltage or current intrinsic sensitivity of the diode, and 푃푠 is the absorbed power by the diode. All the derivation above is valid when the applied RF signal amplitude is within the small-signal regime.

2.9 Detector Characteristics

The figures of merits which they used to differentiate between the detectors are discussed in the following sub-sections.

2.9.1 Voltage Sensitivity

Voltage sensitivity (푆푉) is the most important factor and gives an indication of the total detector performance. It is given by the ratio of the produced output voltage to the input

RF power in (V/W) unit. For highly sensitive detectors, (푆푉) needs to be pushed to its highest possible level through the optimisation of many parameters. Voltage sensitivity can be directly measured at low-frequency‎and‎ using‎a‎50Ω‎ RF source. The measured low-frequency un-matched sensitivity is approximately given by [24, 49, 92]:

푆푉−푢푛푚푎푡푐ℎ푒푑 = 2푍푠퐾푉 (2.15)

where (푍푠) is the source impedance, and 퐾푉 is the curvature coefficient. 퐾푉 is one of the most important factors that is mainly used to evaluate detector sensitivity. It also measures the non-linearity of the diode and can be calculated from the first and second derivative of the I-V curve as expressed in the following equation [93]:

휕2퐼 2 퐾 = 휕푉 (2.16) 푉 휕퐼 휕푉

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In thermionic emission devices such as Schottky and p-n junction diodes, 퐾푉 can also be calculated using the expression (푞/푛푘퐵푇) [24]. Moreover, Schottky diodes are usually biased at the non-linear point that gives a maximum curvature coefficient and thus higher sensitivity. At high-frequency regimes, equation 2.15 is no longer valid since the intrinsic components of the diode start to dominate the rectification process and consequently affect the detector performances. Most of the theoretical expressions used to calculate the detector’s‎voltage‎sensitivity‎were derived based on the structure of the device, resulting in uncertainty in the estimated sensitivity values [48, 94, 95]. Carefully estimating such a figure of merit requires involving the effects of the reflection coefficients, nonlinear resistance and curvature coefficient as given by [93, 96]:

2 퐾푉푅퐽푅퐿(1 − |훤| ) 푆푉푎푐푡푢푎푙 = 2 2 2 푅푠 휔 퐶퐽 푅푠푅퐽 (2.17) 2(푅퐽 + 푅퐿) (1 + (푅 )) (1 + 푅 ) 퐽 1 + ( 푠) 푅퐽

2 where (1 − |훤| ) is the normalized power absorbed by the diode, and (푅퐿) is the load resistance. The matching circuit is used to improve (훤) and deliver more power to the diode.

2.9.2 Noise Equivalent Power

Noise equivalent power (NEP) is defined as the minimum RF power needed to generate a signal equal to the noise in a 1-Hz bandwidth. (NEP) can be calculated using the following equation [97]:

퐼 푁퐸푃 = 푛 (2.18) 푆푉

Or

푉 푁퐸푃 = 푛 (2.19) 푆 푉

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where (퐼푛) and (푉푛) are the measured noise current and voltage in 1-Hz bandwidth. NEP unit is (W/√퐻푧). In zero-bias detectors, (NEP) is mainly limited by the thermal Johnson- noise introduced by the junction resistance. The zero-bias (NEP) is given by [24]:

√4푘퐵푇푅퐽 푁퐸푃 = (2.20) 푆푉푎푐푡푢푎푙 where 푘퐵 is the Boltzmann’s constant and T is the temperature in Kelvin. Noise equivalent power is one of the most important figures of merits that shows the trade-off between high 푅퐽 for higher sensitivity and low 푅퐽 for a minimum (NEP). In the last 50 years, bolometer detectors have shown impressive progress in improving the noise performance where their NEP has decreased by a factor of 11. The minimum recorded NEP for bolometers is (3×10−19W/√퐻푧) at very low temperatures (~0.1K) [98].

The backward diode has also shown a superior noise performance with a NEP of (0.18×10−12W/√퐻푧) at mm-wave-frequencies [50].

2.9.3 Tangential Sensitivity and Dynamic Range

Tangential sensitivity (TSS) is described as the lowest RF power needed to obtain a certain signal-to-noise ratio (SNR) at the output of the amplifier. TSS unit is dBm and can be easily calculated by adding (4dB) to the NEP value [99]. Several factors contribute to the TSS value such as RF frequency, DC circuit configuration, and video amplifier. DC bias gives rise to shot and flicker noise in the diode, which effects the TSS level [100].

Another primary criterion in assessing the detector performance is the dynamic range. It specifies the range of the power in which the diode works in its square-law region. At high input RF power, the operating point shifts from the square-law region to the linear region, which causes the output voltage to be proportional to the input power. Dynamic range is given in dB unit and represents the difference between the upper limit of the square law and the TSS, as shown in the example of figure 2.9. Figure 2.9 depicts the output voltage versus the input RF power of a Schottky diode detector [101]. The curve shows the transition of diode characteristics as the input power increases. At an input

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power >20dB, the output voltage is constant with the input power, which causes the 1-dB compression point.

101

100 Dynamic range

10-1 Upper limit of the square-law region

10-2

10-3 Output Voltage Output (V)

10-4

Tss 10-5 -60 -40 -20 0 20 RF input power (dBm)

Figure ‎2.9: Output voltage versus RF input power showing the dynamic range of a Schottky diode detector [101].

2.10 Theory of Tunnel Diodes

The significant limitations, as mentioned earlier of Schottky and backward diodes led to the search for alternative devices for room temperature emitters and detector applications. One of the earliest tunnelling structures is the Esaki tunnelling diode which was invented in 1958 [102] by the Japanese physicist . Tunnelling is a quantum-mechanical phenomenon that was theoretically described by the physicist George Gamow in 1928. The transport of carriers in Schottky diode obeys the principle of classical physics, in which electrons (or holes) behave as particles and move under the influence of thermionic emission mechanism. In quantum-mechanics, the electron behaves both as a particle and a wave and thus, just like a wave, can penetrate through a barrier even when its energy is smaller than that of the barrier. Tunnelling dominates if the incident electron wavelength is larger compared to the barrier thickness. Thinning the barrier leads to an increase in the chances of finding the electron on the other side of the barrier [102, 103]. Barrier properties (thickness and height) can be adjusted through

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precise control of the multilayer structure of the device, which includes semiconductors, insulators and metals.

The time-independent Schrödinger equation is used to describe the wave and particle behaviours of the electron in and outside of the barrier, and consequently helps to understand the tunnelling phenomena of the electron, and this is given by [102, 103]:

푑2휓 2푚∗ + [퐸 − 푉(푟)]휓 = 0 (2.21) 푑푟2 ħ2 where (푚∗) is the electron effective mass, (푟) is the position vector, 푉(푟) is the potential energy at position 푟, (퐸) is the total energy of the electron and ħ is the reduced Planck constant. The tunnelling mechanism of an electron through a single barrier is plotted in figure 2.10. In the figure, barrier height and width are represented by (푉0) and (푡푏) respectively. For the barrier shown in figure 2.10, the wave function is equal to (휓(푟) =

𝑖푘0푟 A 푒 ), where (퐴) is the amplitude of the wave and (푘0) is given by:

2푚∗(퐸 − 푉 ) 푘 = √ 0 (2.22) 0 ħ

where 푘0 is a wave vector and can have an imaginary value if the incident electron has an energy (퐸) lower than the barrier height (푉0).

Figure ‎2.10: Schematic of the incident, reflected and transmitted wave functions through a rectangular potential barrier [104].

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Solving the Schrödinger equation (2.21) leads to estimating the transmission probability of the electron through the barrier as it expressed by:

휓 2 16 퐸(푉 − 퐸) 2푚∗(푉 − 퐸) 1 0 √ 0 (2.23) 푇푡 = | | = 2 exp (−2푡푏 ) 휓2 푉0 ħ

Equation 2.23 points out to the importance of three design factors on the probability of transmission in any single barrier tunnel structure. Barrier height (푉0) should be as high as possible to ensure the tunnelling mechanism occurs first instead of thermionic emission. In contrast, a low barrier is also required for a high probability of transmission and low junction resistance. The p-GaAsSb/i-InAlAs/n-InGaAs backward tunnel diode is made of an InAlAs barrier with a potential height of roughly 1eV and lattice-matched to InP [47], which makes it very attractive for integration with low noise HEMT on the same substrate. The n-InAs/i-AlSb/p-GaAlSb/p-GaSb has a AlSb barrier with an effective height of ~2eV grown on GaAs substrate [50]. Transmission of the electron through a barrier also depends exponentially on the width of that barrier (푡푏). The growth rate of layers must be uniform to reduce the barrier width variation over the wafer. Another factor is the effective mass of electron 푚∗, which plays an important role in defining the speed and mobility of electrons. Higher mobility materials such as InGaAs and GaAs help to reduce the transit and tunnelling time for high-frequency applications. One of the benefits of the quantum-mechanical phenomenon is the very short tunnelling time (in the order of picoseconds) that is defined by the quantum transition probability per unit time, making tunnel devices very promising elements for millimetre and sub- millimetre wave applications [44].

2.11 Tunnel Diodes

There are different categories of tunnelling devices based on quantum-mechanical phenomena. These devices have been developed and received much attention in recent years due to the short tunnelling time, which leads to high-speed operation, as mentioned above. This feature is the key point of THz devices. There are three primary tunnelling devices, namely, Esaki diode, Resonant Tunnelling Diode (RTD), and Asymmetric Spacer Layer Tunnel Diode (ASPAT). In this work, the focus will be on the ASPAT

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diode as a potential active element for zero-bias mixer and detector circuits at mm-wave applications.

2.11.1 Esaki Tunnel diode

The idea of tunnel diode was developed after intensive work done by Esaki in order to examine a new type of diode using a p-n junction. The work started with a simple , and it was observed that increasing the doping concentration led to a decrease in the breakdown voltage [24]. It was also found that Zener diode has a small tunnelling current when a high doping concentration is applied on both sides of the junction. Following that, Esaki increases the doping for the p and n layers to higher levels, and that led to a rise of the tunnelling current at small bias leading to a new concept called the negative differential resistance (NDR). The new structure was named an Esaki tunnel diode. It is made up of two layers (p and n types) with a heavily doped profile. As a result of this, it has a very thin depletion region width. The depletion region is treated as a barrier with a high possibility of the carriers to penetrate it if the tunnelling conditions are met. The depletion region thickness (푡푑푒푝) is limited by many factors as clearly seen in the following equation [105].

2휀 푁퐴 + 푁퐷 푡푑푒푝 = √ ( ) (푉푏𝑖 − 푉푟) (2.24) 푒 푁퐴푁퐷

where (휀) is the permittivity of the material. NA and ND are the doping of n and p layers.

(푉푏𝑖) and (푉푟) are the built-in and applied reverse bias, respectively. A thinner barrier (~10nm) and high majority carriers on both sides would cause the electron in the conduction band in the n-side to be almost brought in line with the hole in the valence band in the p-side, leading to tunnelling of the carriers (electrons and holes) across the barrier. Indirect and direct tunnelling is possible in this type of tunnel diode. Tunnelling of electrons occurs in a horizontal way when firstly, free states on both sides of the barrier are available and at the same energy levels. Secondly, the barrier height is low enough to activate the tunnelling process. The NDR in the I-V characteristics made the device a promising element for room temperature emitter applications. However, the thin barrier and consequently high junction capacitance, as well as the slow transit time of minority carriers limits its use at high-frequency.

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2.11.2 Resonant Tunnelling Diode

The promising feature of the NDR region of Esaki diode was the cornerstone for the great discovery of a new type of two-terminal diode called the resonant tunnelling diode (RTD). In 1974, the first RTD structure was demonstrated by Chang et al. [106]. It was made of GaAs/AlGaAs materials and grown on a GaAs substrate using Molecular Beam Epitaxy (MBE). The heart of the RTD device is made up of three regions: a small band gap undoped layer surrounded by two large band gap undoped layers. The double barrier quantum well (DBQW) RTD was intensively investigated by many researchers employing different semiconductor materials [107, 108]. The NDR region has been considered as an attractive feature and strongly dependent on barrier and well parameters. The DBQW RTD comprises of the quantum well made of an undoped narrow bandgap material inserted between two barriers formed of undoped high band gap materials [109]. The perceived advantages of InGaAs material made it the preferred choice to be employed for high power and high-frequency emitters. InGaAs material produces a high peak current density at a low bias due to the lowered first resonant level in the well without the need for using an InAs sub-well [103]. In particular, the indium-rich has gained much interest and was used at different compositions [110, 111]. Our recent paper

[112] reported five RTD devices made of In0.8Ga0.2As well and AlAs barriers. The work aimed to experimentally investigate the DC and RF characteristics of a different barrier and well structures for possible high power and high-frequency emitters. The band diagram of the active area of In0.8Ga0.2As/AlAs RTD is plotted in figure 2.11.

Figure ‎2.11: Schematic band diagram of the In0.8Ga0.2As/AlAs DBQWRTD. The AlAs energy band gap is the direct gap value [113]. 64

∆퐸푐 and‎ ∆퐸푣 are the conduction and valence band discontinuities and result from the difference in the materials band gaps. (푡푤) is the thickness of the well. Electrons inside the quantum well are confined to fixed energy levels since the separation between the energy levels En1 and En2 is larger than 푘퐵T. The confinement property prevents electrons from travelling in the direction of growth and confines them to an X-Y plane unlike in a bulk 3D semiconductor structure. A higher 퐸푛1 means the separation increases between the adjacent resonant levels. Thus, the leakage current is much reduced through the second resonant energy level. Therefore, the peak to valley current ratio (PVCR) is improved. Unfortunately, a higher bias is needed to reach the peak current and consequently, the peak voltage is shifted to higher levels.

Different RTD structures have been demonstrated in the literature. In 1992, RTD oscillators were mounted in a rectangular waveguide and tested at fundamental frequencies of 103GHz and 210GHz with an output power of 50µW and 20µW respectively [114, 115]. In 1996, A. C. Molnar et al reported an RTD oscillator with 16 RTDs integrated with slot antenna at 310 GHz with 28µW output power [116]. In 2013, much higher powers were obtained by Suzuki et al [117]. The work demonstrated a high output power oscillation of ∼400µW in the frequency band of 530 to 590GHz using a single oscillator with an offset slot antenna. Moreover, the combined output powers of 610µW, 270µW, and 180µW at 620GHz, 770GHz, and 810GHz were obtained respectively with a two-element array. Following that, Asada made a significant breakthrough by reporting the highest indium rich In0.9Ga0.1As RTD [53] with the highest oscillation frequency at room temperature of 1.92THz. But, that was at the expense of an extremely low output power of 0.4µW. The triple barriers RTD was also suggested for zero-bias detection applications [118]. The existence of three barriers introduces a strong non-linearity at a point close to zero-bias. Tunnelling current of triple barriers RTD relies on the critical thickness of the active layers (barriers and well) that is in the few angstrom ranges. Therefore, any small variation in the growth rate increases the chances of producing such devices with different characteristics. Hence, the reproducibility is reduced, and the cost is increased accordingly [119].

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2.12 Asymmetric Spacer Layer Tunnel Diode

The remarkable temperature insensitivity of RTD was the primary drive for Syme [13, 120] to invent a new tunnel device made of a single barrier and different spacer thicknesses. The asymmetric structure produced asymmetric I-V characteristics and more importantly, a low turn-on voltage. The new candidate, the Asymmetrical Spacer Layer Tunnel Diode (ASPAT) consists of a heterostructure interface, with a thin, high bandgap material placed between two low bandgap materials. The operational principle of the ASPAT depends on the tunnelling of the electrons through the barrier. The existence of the potential barrier leads to a nonlinear I-V characteristic. In general, the barrier is made very thin (few nanometers), so the electrons are able to tunnel, making the diode much less temperature dependent [61, 121]. Besides that, the existence of a single barrier in the ASPAT structure does not only facilitate the growth process but also provide a high current density, unlike the triple barriers tunnelling structure reported in [118, 122], where a low-current-density was achieved as a direct result of low tunnelling probabilities. The first discrete ASPAT diode was reported in [13] by R. Syme and M. Kelly from General Electrical Company (GEC). Two GaAs/AlAs ASPAT diodes were grown with an AlAs barrier thickness of (10Å and 60Å) respectively, using MBE and MOCVD techniques, and then their detection performances were compared with other thermionic diodes. Both ASPAT diodes were fabricated with a mesa area size of (~150µm2). On-wafer measurements were performed, and curvature coefficients of 10V-1 and 34.5V-1 were obtained for the 10Å and 60Å AlAs barrier ASPAT devices, respectively. Measurement showed a small relative change in (푉표푢푡/dB) of 1.2dB of the ASPAT compared to a 2.2dB and 3dB for the planar doped and Schottky diodes over the temperature from 233 to 353K. The work however, did not address the matched sensitivity over any specific frequency band, and instead, the un-matched sensitivity of the 60Å AlAs barrier ASPAT diode was measured and found to be ~6000V/W at a single frequency of 9.375GHz. The temperature-independent feature of ASPAT diode was studied in details in [123]. The concept was proved for thin AlAs barriers of 14Å and 32Å. For AlAs barrier >50Å, the ASPAT was shown to be less affected by the variation of ambient temperature. For this work, two ASPAT structures were grown with an AlAs barrier of 28.3Å and fabricated at the University of Manchester. The first ASPAT is made of GaAs/AlAs structure and grown on a GaAs substrate, and the second one is made of InGaAs/AlAs and grown on lattice matched InP substrate. More details

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regarding the epi-layers will be presented in chapter three. In [15, 16], the concept of weak temperature dependency was verified by the Manchester group for both

GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes over a wide temperature range of 77 to 400K as shown in figure 2.12.

(a)

(b)

Figure ‎2.12: Temperature dependency of (a): GaAs/AlAs and (b): In0.53Ga0.47As/AlAs ASPAT diodes [15, 16]. 67

Both samples showed a very good temperature stability compared to thermionic emission Schottky diode at different bias. To be more specific, at temperatures <200K, both samples showed a very high-insensitivity of current with temperature. At temperatures >200K, a small increase in the current was observed for both samples. This was due to the temperature dependence of parameters such as the effective mass of electrons and energy bandgap [124] which are appreciable at high temperatures. In general, the

In0.53Ga0.47As/AlAs ASPAT was found to be more stable than the GaAs/AlAs one. This is mainly due to the higher effective barrier in the case of In0.53Ga0.47As/AlAs structure which limits the thermionic emission of carriers. To conclude, ASPAT diodes have shown a strong non-linearity at low bias and temperature insensitive property which make them suitable to be used for room temperature zero-bias detectors and mixers. To examine this, we have designed, fabricated, and tested zero-bias ASPAT detectors and mixers at different frequency bands as will be described in details in the next chapters.

2.13 Operating Principle of ASPAT Diodes

The active region of the ASPAT diode consists of a thin barrier sandwiched between two thick spacers. To understand the operation of the ASPAT diode at different voltages, we used a SILVACO atlas tool to build the ASPAT structure and simulate the conduction band profile, as shown in figure 2.13 [125]. Under zero bias, there is no band bending in the structure, and indeed no tunnelling occurs, resulting in zero current. At a positive bias, a high voltage drop would occur across the thinner spacer, leading to band bending, and pulling down of the band profile from left to right which decreases the effective barrier height. Meantime, an accumulation layer (well) is formed under the Fermi level and beside the barrier from the thicker spacer side. Electrons first accumulate in the small well and then tunnel through the barrier. If a reverse bias is applied, the opposite occurs, and the band profile of the thinner space is modified producing an increase in the effective barrier height, resulting ideally in zero current. However, at higher reverse bias, the leakage starts to be appreciable for both types of ASPAT structures.

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2.0 V < 0 V = 0 1.6 V > 0

1.2 Left Right

0.8

Energy(eV) 0.4 Thick spacer spacer Thin Ef E 0.0 f Well Ef

-0.4 0.3 0.4 0.5 0.6 0.7 Thickness(m)

Figure ‎2.13: Schematic conduction band profile of ASPAT structure under negative, zero and positive bias [125, 126].

2.14 Current Density of ASPAT Diode

AlAs, is an indirect material, and in an ASPAT structure where the AlAs barrier is typically thick (>50Å), the transition of carriers‎occurs‎between‎the‎gamma‎(Γ)‎and‎(X) valleys. As a result, the GaAs/AlAs band discontinuity (barrier height) is ~0.2eV. In such a case, the thermionic emission mechanism can be dominant, and accordingly, the current density significantly changes with temperature [123]. However, it was experimentally validated in [13, 123] that in a thin AlAs barrier (typically < 50Å),‎the‎Γ- Γ‎transition‎of‎the‎AlAs‎is‎dominant, and the resulting GaAs/AlAs barrier height is ~1eV. The higher barrier makes the structure less affected by thermionic emission over the barrier. The theoretical calculations of the current density in the ASPAT structure were described in details in [93]. The key equation is the Schrodinger formula, which is used to estimate the transmission coefficients through the structure. Assuming the carriers move in the z-direction perpendicular to the direction of layers in (X) and (Y) planes. The one-dimensional Schrodinger equation is given by:

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ħ2 푑2 휓 + |푒|(휙 − ∆휙)휓 = 퐸 휓 (2.25) 2푚∗ 푑푧2 푧 where is (휙) the potential at point 풛, and (∆휙) is a correction term which reduces the effective barrier height and can be neglected since there is virtually no band bending when the bias (푉) increases. The current density in the z-direction (퐽푧) is then calculated by solving equation 2.25 at different values of 퐸푧. Thus, 퐽푧 is expressed as:

−|푒|ħ 푑휓 푑휓∗ 퐽 = (휓∗ − 휓 ) (2.26) 푧 2푚∗ 푑푧 푑푧

In the case of a heavily doped top and bottom contacts, the wave functions at the right and left side of the barrier are described by:

𝑖 푘푙푒푓푡 푧 −𝑖 푘푙푒푓푡 푧 휓푙푒푓푡 = 푒 + 푅 푒 (2.27)

[𝑖 푘푟𝑖푔ℎ푡 (푧−푡푏)] 휓푟𝑖푔ℎ푡 = 푇 푒 (2.28)

Substituting equation 2.27 and 2.28 in 2.26 gives the following expression:

|푒|ħ 푘 |푒|ħ 푘 퐽 = 푙푒푓푡 (1 − |푅(퐸 )2|)= 푟𝑖푔ℎ푡 |푇(퐸 )2| (2.29) 푧 푚∗ 푧 푚∗ 푧

푅(퐸푧) and 푇(퐸푧) are a function of 퐸푧 and can be found using the transfer matrix method [93]. Thereafter the right side of equation 2.29 is integrated with respect to 푘 implying multiplying the integration part by 푓(퐸)(1 − 푓(퐸)). (1 − 푓(퐸)) is the probability of unfilled states in the conduction band. 푓(퐸) is the fermi function which calculates the probability of filled states in the conduction band and it is given by:

1 푓(퐸) = 퐸−퐸푓 (2.30) ( ) 1 + 푒 푘퐵푇

Thus, the total current density is expressed by:

퐸푓−퐸푧 ( ) |푒|푚∗ħ 푘 푇 ∞ 1+푒 푘퐵푇 퐽 = 퐵 ∫ (1 − |푅2|) 푙푛 | |d퐸 (2.31) 푧 2 2 0 퐸푓−|푒|푉−퐸푧 푧 2휋 ħ ( ) 1+푒 푘퐵푇

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where 푘퐵 is the Boltzmann’s constant. For a structure with undoped spacer surrounding the thin barrier, there is a band bending of 푘퐵푇 as the bias increases. Furthermore, a depletion region is formed across the undoped layers, and therefore, the Poison equation must be used to accurately calculate the potential (휙) at point 푧. However, the reported current density equations of tunnelling diode calculate only the current through the barrier. For an accurate estimation, the effect of intrinsic elements and the losses from parasitic has to be taken into account. In our recent paper [125], we calculated the ASPAT current using a physical model built and simulated using Silvaco Atlas tool. The semiconductor-insulator-semiconductor (SIS) model solves the 1-D Schrodinger equation to calculate the tunnelling current through the structure. The SIS model calculates the current density through a single barrier structure using equation 2.31. The model showed an excellent fit with measured data of different mesa area sizes.

2.15 Introduction and Overview of APD and PIN Photodetectors

Photodetectors represent a crucial part of the receiver detection process. Photodetectors based semiconductor offers many advantages regarding cost, size, reliability, and compatibility with other optoelectronic devices. Sustained development efforts for high- speed photodetectors have been ongoing for many years to deliver components that can be grown and fabricated efficiently and most importantly meet the increasing demand of high operating bandwidth and data rate as well as fulfilling low-cost requirements for mass-market adoption [127, 128]. Various structures [129-131] have been extensively studied and fabricated to realise photodetectors capable of maintaining high speed of operation, such as avalanche breakdown diode (APD), PIN diode, and metal- semiconductor-metal (MSM) diode. However, the latter introduces a substantially high leakage current, which causes high shot noise [132, 133]. The perceived advantages of APD and PIN diodes make them highly preferred at the receiver front end to detect optical signals and convert them into electrical ones. The reversed biased PIN structure is the most widely used diode as photodetector due to its simple implementation, good responsivity, large bandwidth, and operation at long wavelength at the minimum of fibres attenuations at 1.3µm and 1.55µm. Avalanche photodiodes have been widely employed in high data-rate long-haul communication systems, where high gain-bandwidth, low

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noise and high sensitivity characteristics are required. Generally, APDs have a sensitivity that is 5 to 10dB higher compared to PIN diodes [134, 135].

The multiplication gain of the APD makes it suitable for low optical power detection. In addition, its internal gain eliminates the use of amplifiers and therefore reduces the required on-chip area, cost, and power dissipation [135]. To maintain high-data-rate applications, the critical factors of APD are high gain-bandwidth product and low excess noise factor [136]. III-V material systems and in particular, In0.52Al0.48As and

In0.53Ga0.47As, are considered as one of the most promising technologies for the 1.2 to 1.6µm wavelength range [137]. More importantly, the use of InGaAs material to absorb incident light enables high 3-dB bandwidth while keeping the light window aperture of acceptable sizes for flexible alignment tolerances with fibres. Such a photodetector which has wide operating bandwidth, a low-bias of operation, and high responsivity is highly desirable for achieving maximum possible performance of receiver systems. Reduction of photodetector mesa area size is one way to minimise the RC time and thus improve the operating 3-dB bandwidth. Unfortunately, this leads to inflexible alignment tolerances, which in turn increase the cost of packaging and assembly [129]. PIN diodes are aimed for short distance application due to their lowest sensitives compared to APDs. In particular, when a TransImpedance Amplifier (TIA) is incorporated, the maximum sensitivity of receivers is limited by the introduced noise of the amplifier [130].

Regarding noise performance, PIN and APD photodiodes suffer from thermal and shot noises. However, the total noise of APD is significant due to the generated excess noise as a direct result of the impact ionisation process [133]. The design process of an APD is more complicated and needs more care to control its performance. Dark current and multiplication factor (M) are very sensitive to the thickness and doping profile of the multiplication and charge sheet layers. The trade-off between achieving a high signal-to- noise ratio (SNR) and low excess noise can be compensated by designing an APD with an appropriate internal gain value. At equal absorber layer thickness, the applied electric field is higher in the case of an APD structure, and this degrades transit time-frequency due to the decrease of the overshot drift velocity of electrons and holes [135]. InP and InAlAs materials are widely exploited as multiplication layers in APD photodetector based III-V semiconductor technology [135, 138-141].

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2.16 Operational Principle of PIN Photodiode

The basic structure of a PIN diode comprises an intrinsic high resistivity layer (i), sandwiched between positively (p+) and negatively (n+) highly doped layers as depicted in figure 2.14 [142]. The working configuration of the PIN photodetector in reverse biased mode is depicted in figure 2.14. By definition, the intrinsic region is made of an undoped region free of carriers, leading to a highly resistive layer compared to the (p++) and (n++) layers. In reverse bias, nearly the whole applied voltage (푉푆) appears across the intrinsic region, and thus, a strong electric field is created in this region. The absorption of photons with ℎ푣 ≥ 퐸푔 in the intrinsic region of the device can generate free carriers through electron transitions, across the bandgap, from the valence band to the conduction band. In this way, a hole-electron pair is created. Following this, the electrons move to the (n+) side and the holes to the (p+) side due to the existence of the electric field. Thus, a photocurrent flows in the diode [142].

Photon

≥

p

Generatedelectron Generatedhole

Depleted region Depleted Holes

i

Intrinsic

Electrons

n

Figure ‎2.14: Operational principle of a reversed biased PIN photodetector, adapted from [142].

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2.17 Operational Principle of Avalanche Photodiode

The first avalanche multiplication idea was demonstrated in [143, 144], where a simple

PIN structure was applied to a strong reverse bias. A low band gap In0.53Ga0.47As material was employed to absorb the light and at the same time to perform the avalanche multiplication process. These structures suffer from large dark current, resulting from the high band-to-band tunnelling of electrons under a high electric field. Implementing high- sensitivity receivers requires very small dark current APDs, therefore; a new structure was introduced, in which, the light is absorbed in a specific layer and then multiplied in another one, and this is the idea behind the separated absorption and multiplication (SAM) APD. In this particular structure, the absorption layer has a relatively low electric field compared to the multiplication region. The multiplication region is usually made of a high band gap material to reduce the band-to-band tunnelling process [55, 145]. At high electric fields, the carrier gains energy, which is higher than the energy band gap of the multiplication layer. As a result of that, the carrier can generate a new electron-hole. This process is called impact ionisation. The original and the newly generated electron-hole pairs propagate, and the consequence of that is more impact ionisation events are likely to occur. The periodic occurrence of these events is known as avalanche multiplication [146]. The material used as a multiplication layer is commonly characterised by its impact ionisation rate for the electron 훼(퐸) and hole 훽(퐸) [146]. This rate defines the inverse mean distance between two continuous impact ionisation events. Generally, at high electric fields, the carrier gets sufficient energy for the ionisation event in a small distance and thus 훼(퐸), and 훽(퐸) increase. On the contrary, 훼(퐸) and 훽(퐸) decrease at high temperature due to the increase of phonon-scattering rate, which leads to a deceleration of the ionisation process. The impact ionisation rate coefficients are electric field dependent factors and can be analytically expressed using the formulas [147, 148]:

퐵푁 (− )퐵퐸푇퐴푁 α(E)= AN 푒 퐸 (2.32)

퐵푃 (− )퐵퐸푇퐴푃 β(E)= AP 푒 퐸 (2.33) where AN, BN, BETAN, AP, BP, BETAP are the impact ionisation parameters for the bulk material used as an avalanche layer.

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The most critical factor is 푘푟푎푡𝑖표 of the multiplication region, which is the ratio between the hole and electron impact ionisation, as shown in the following equation [146]:

훽(퐸) 푘 = (2.34) 푟푎푡𝑖표 훼(퐸)

The factor 푘푟푎푡𝑖표 plays an important role in improving the sensitivity of the APD. In electron-multiplying material, 푘푟푎푡𝑖표 is less than 1 (훼(퐸) is typically higher than 훽(퐸)), while it is higher than 1 in hole multiplication materials (훽(퐸) is typically higher than

훼(퐸)) [145]. For higher sensitivity requirement, 푘푟푎푡𝑖표 is desired to be much smaller or much higher than 1 to achieve the lowest possible excess noise factor.

In the typical (SAM APD), the impact ionisation coefficients 훼(퐸) and 훽(퐸) can be equal and approach unity. The consequence of that is a high excess noise and a long impact ionisation process [149]. The InAlAs material is an electron multiplication material which offers better stability and lower 푘푟푎푡𝑖표 of 0.29-0.5 [150] compared to InP which is a hole multiplication with 1/푘푟푎푡𝑖표 of 0.4-0.5 [151]. The lowest reported ratio is for Silicon with 푘푟푎푡𝑖표=0.03 to 0.1 [43]. However, Silicon is not lattice matched to InGaAs and InP materials, which limits its use for 1.3 to 1.55µm wavelength telecommunication applications even though attempts at using mismatched Si-Ge are underway [37, 38, 152].

The SAM APD has a high band discontinuity between the high energy band gap of the multiplication region and low energy band gap of the absorption region, which causes the carrier to be trapped at the hetero-junction. As a result, there is a slowing down of the speed of carriers and degradation of the APD photodetector frequency response. Figure 2.15 (a) shows the typical structure of a separated absorption, charge, and multiplication (SACM) APD

The introduction of a grading layer at the hetero-interface leads to considerable benefit in reducing the band-discontinuity and thus improving the APD speed response. The charge layer, on the other hand, plays a significant role in controlling the electric field difference between the absorption and multiplication layers. The thickness and doping concentration of the charge layer controls the electric field distribution of the photodetector and ensures it is high enough across the absorption region to accelerate the carriers to their maximum saturation velocity without increasing the tunnelling current [55, 145].

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(a) (b)

Figure ‎2.15: Operation of a SACM APD, (a): 2-D structure, (b): Band diagram [55, 146].

SACM APD band diagram, including the carrier transport, is depicted in figure 2.15 (b). The operational principle of the APD is briefly described as follows [55, 146]:

(1): An electron-hole pair is initiated in the absorption layer when light hits the APD with photon energy equal to or higher than band gap energy (퐸푔) of the absorber. The strong applied reverse bias forces the electron and hole to drift to the (n-side and p-side) respectively.

(2): Once the electron enters the first high-electric field charge sheet region, it starts to accelerate due to a strong gradient in the conduction band.

(3): This acceleration may create secondary electron-hole pair due to the impact ionisation process in the multiplication layer. Both primary and newly generated electrons drift towards the n-side, while, the secondary hole travel toward the low electric field absorption.

(4, 5): The secondary hole could generate a new electron-hole pair in the charge layer. Finally, all holes move towards the p-side.

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2.18 Photodetector Characteristics

The main characteristics of photodetectors are summarised as:

2.18.1 Quantum Efficiency and Responsivity

The amount of generated current depends directly on the percentage of absorbed light in the intrinsic absorption region. High absorption implies a large number of generated electron-hole pairs. In practice, not every absorbed photon is capable of creating an electron-hole pair. Only photons that have energy ℎ푓 ( ℎ is Plank constant and 푓 is the frequency of the incident light) that is equal to or higher than the energy bandgap of the absorber can generate an electron-hole pair. The percentage of the photons that generate these pair can be described as the quantum efficiency and denoted as ƞ [146, 153]. The simple form of ƞ is written as:

푃 ƞ = 푎푏푠 푃𝑖푛 (2.35) or

ƞ = (1 − 푟)휉 (1 − 푒(−훼(휆)퐷푎푏푠))

푃푎푏푠 and 푃𝑖푛 are the absorbed and incident optical powers respectively, while 푟 is the light reflection at the surface, 휉 refers to the fraction number of the generated electron- hole pair that contribute to the output current, 퐷푎푏푠 is how long the light travels through the absorber layer, and 훼(휆) is the absorption coefficient at a specific wavelength. Quantum efficiency close to one can be achieved with selecting a high absorption coefficient material or by increasing the thickness of the absorber. The absorption coefficient as a function of the wavelength of different materials is shown in figure 2.16 [154].

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100000 Si GaAs Ge

In0.53Ga0.47As

10000 )

-1 1000

cm

( 

100

10 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 (m)

Figure ‎2.16: Absorption coefficients versus light wavelength of different materials [154].

-1 In0.53Ga0.47As material has the highest absorption coefficient of ~10000cm , which makes it a suitable material for the 1.3 to 1.55µm wavelength range. The ratio between the output photocurrent 퐼푝ℎ and the incident optical power 푃𝑖푛 is defined as the responsivity. It is considered as the most crucial figure of merit to differentiate between photodetectors. The responsivity of a PIN photodetector (Ɍ푃퐼푁) is given by the following equation [142, 154]:

퐼푝ℎ 푒ƞ 휆ƞ Ɍ푃퐼푁 = = = (2.36) 푃𝑖푛 ℎ푓 1.24 where 푒 is the charge of the electron. The maximum responsivity of a PIN photodetector is 1A/W. Equation (2.36) is also valid for calculating the APD responsivity at the punch- through voltage (푉푃푇), where the internal gain (M) is equal to one. In APDs, the wavelength of the incident light controls the number of injected electrons to the multiplication region that can generate yet more new electrons and thus increase the gain. The responsivity of APDs is higher compared to PIN photodetectors due to the internal gain. For bias > 푉푃푇, the responsivity of APD (Ɍ퐴푃퐷) is expressed as:

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퐼푝ℎ 푀 푒 ƞ Ɍ퐴푃퐷 = = (2.37) 푃𝑖푛 ℎ푓

The responsivity of the photodetector affects the whole sensitivity of the optical receiver, and therefore, much care should be taken to design and optimise the epi-layer structure of the diode. An easy way to improve the responsivity is to increase the absorber thickness and allow more photons to be absorbed and generate electron-hole pairs. However, this would increase the transit time of carriers and degrade the 3-dB bandwidth. Another way is to choose a material with a high absorption coefficient (훼). Silicon (Si) and germanium (Ge) materials represent an important platform for optoelectronic integrated circuits and have been used effectively in the fabrication of PIN [39, 42, 155, 156] and APD photodetectors [37, 38, 40, 157] at wavelengths ranging from 1.3 to 1.55µm. The main issue of using pure germanium is the 4% lattice mismatch with silicon. Any lattice mismatch will introduce defects that can lead to an increase in the dark current. As a result, photodetector sensitivity is degraded [158].

By contrast, III-V semiconductor materials are an excellent choice for optical applications at 1.55µm due to the ability to adjust their band gap according to required wavelengths. In0.53Ga0.47As shows excellent performance when it is used in high-power and high-frequency photodetector applications. This is because of the high saturation velocity for electrons (푉푠푎푡푛) and holes (푉푠푎푡푝), higher electron mobility (µ푛) (around 9 times higher than silicon), large 훼(휆), low dark current, and finally, its energy band gap (0.75eV) make it an optimal choice material for photodetector at both 1.3µm and 1.55µm wavelengths [159]. The responsivity of photodetector based germanium absorber ranges between (0.2 to 0.7A/W) at a wavelength of 1.33µm [42, 157]. A high responsivity of (0.99A/W) at 1.55µm wavelength was reported for a PIN photodetector made up of 2µm- thick of InGaAs absorber [160].

2.18.2 Dark Current

Dark current is another crucial factor, which measures the amount of leakage current of the photodetector. The dark current of a photodetector is primarily due to three phenomena. Firstly, the generation-recombination of carriers inside the absorption region under low-bias operation. Secondly, the diffusion of minority carriers from the n++ and p++ regions into the depletion region and thirdly, the tunnelling of electrons from the

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valence band to conduction band for a small energy band gap material and under high reverse bias. In a simple PIN structure, the generation-recombination current (퐼푔−푟) inside the absorber is given by [161]:

푒 푉 퐴 푊 (− 푏𝑖푎푠) 푚푒푠푎 푑푒푝 2푘 푇 퐼푔−푟 = (푒 푛𝑖 )(1 − 푒 퐵 ) (2.38) 휏푒푓푓

where 푛𝑖 is the intrinsic carrier concentration, 퐴푚푒푠푎 is mesa area size, 푊푑푒푝 is the depletion region width, 휏푒푓푓 is the effective carrier lifetime, 푉푏𝑖푎푠 is applied bias, 푘퐵 is the Boltzmann’s‎constant,‎T is the temperature in Kelvin. The diffusion current 퐼푑𝑖푓푓 is given by [153]:

2 퐷푛 퐴푚푒푠푎 2 퐷푝 퐴푚푒푠푎 퐼푑𝑖푓푓 = 푒 푛𝑖 √ + 푒 푛𝑖 √ (2.39) 휏푛 푁퐴 휏푝 푁퐷

where 퐷푛 and 퐷푝 are the diffusion coefficients for electrons and holes, 휏푛 and 휏푝 are the minority carrier lifetime of electrons and holes, 푁퐴 and 푁퐷 are the doping concentration of holes and electrons respectively. The tunnelling current (퐼푡푢푛) is written as:

∗ 3 ∗ 3 휋2 √푚 (√퐸푔) √2 푚 푒 퐸푑푒푝 푉푏𝑖푎푠 {− } 퐼 = 퐴 푒 2√2 푒 ℎ 퐸푑푒푝 (2.40) 푡푢푛 2 푚푒푠푎 ℎ √퐸푔

∗ where 푚 is the effective electron mass, 퐸푑푒푝 is the electric field across the depletion region. All current components are proportional to the mesa area size. Diffusion and generation-recombination current are dependent temperature factors as 푛𝑖 vary with the temperature. The current flowing over the mesa surface (퐼푠푓) of the photodetector is another issue which leads to an increase in the total dark current. The dark current of the APD is much higher than in a PIN diode and depends on the value of the internal gain

(푀). The total APD dark current (퐼푑푎푟푘−퐴푃퐷) is given as:

퐼푑푎푟푘−퐴푃퐷 = (퐼푔−푟 + 퐼푑𝑖푓푓 + 퐼푡푢푛) × 푀 + 퐼푠푓 (2.41)

At voltage close to the breakdown point, 푀 dramatically increases to infinity, which makes the dark current becomes very large and the APD inadequate for low noise applications. The doping concentration and thickness of the epi-layers play a critical role in defining the dark current. Inadequate use of absorber and multiplication layers results in problematic high leakage current as was reported in [141, 162].

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2.18.3 3-dB Bandwidth

Recently, much effort was devoted to improving the operating bandwidth of photodetectors and maintaining high-data-rate by either modifying the epi-layer structure or introducing new design configurations. The 3-dB bandwidth is defined as the frequency for which the output power drops to half of its DC value [41]. The design process of the absorber layer determines the responsivity and the maximum operating 3- dB bandwidth of the photodetector. The 3-dB bandwidth of the photodetector is mainly constrained by the carrier transit time in the intrinsic region. The carrier transit frequency

(퐹푇) is limited by the saturation drifts velocity (푉푠푎푡) of the carriers and width of the depleted intrinsic regions (푊푑푒푝) and can be approximately calculated using the equation [130, 155]:

0.45 푉푠푎푡 퐹푇= (2.42) 푊푑푒푝

퐹푇 can be maximised by thinning the intrinsic region thickness and/or by choosing a high saturation velocity absorber material. However, another limitation, RC frequency (퐹푅퐶), has to be also considered due to the delay time introduced by RC components. 퐹푅퐶 can be theoretically estimated using the following expression:

1 퐹푅퐶= (2.43) 2휋(푅푠+푅퐿) 퐶퐽

where 푅퐿 is‎the‎50Ω‎load‎resistance‎of‎practical‎optical‎systems‎when‎a‎photodetector‎is‎ connected to a trans-impedance amplifier. Minimising the mesa area size reduces 퐶퐽, but can also increase the contact resistance, which is another limiting factor to the RC bandwidth. Both terms (퐹푅퐶) and (퐹푇) determine the maximum 3-dB optoelectric bandwidth (퐹3푑퐵) of the photodetector as expressed by the following equation [163]:

퐹푅퐶 퐹3푑퐵 = 2 (2.44) √1+(퐹푅퐶/퐹푇)

Equation 2.44 shows that the total bandwidth is dominated by 퐹푅퐶 for 퐹푅퐶 < 퐹푇.

Moreover, 퐹3푑퐵 reduces by a factor of (1/√2) when 퐹푇=퐹푅퐶.

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2.18.4 Internal Gain

The internal gain 푀 changes according to the applied electric field across the multiplication and charge sheet layers. Increasing the reverse bias leads to an increase in the gain by mean of the impact ionisation process. The internal gain of a simple APD is given by [154]:

1 − 푘푟푎푡𝑖표 푀 = [−1(1−푘 )훼(퐸)푊 ] (2.45) 푒 푟푎푡𝑖표 푚 − 푘푟푎푡𝑖표

At 푘푟푎푡𝑖표 << 0.1 (for electron multiplication material), the internal gain can be approximately written as:

푀 = 푒(훼(퐸)푊푚) (2.46)

Increasing the thickness of the multiplication layer (푊푚) leads to an increase in the gain exponentially. If 훼(퐸)=훽(퐸) (푘푟푎푡𝑖표=1), the internal gain is formulated as:

1 푀 = (2.47) 1 − 훼(퐸)푊푚

Higher gain is expected when 훼(퐸)=훽(퐸) and 훼(퐸)푊푚=1. Unfortunately, this would increase the excess noise factor and decrease the sensitivity of the receiver. The APD has a series resistance (푅푆) introduced by the top and bottom contacts, as well as the spreading resistance between these contacts. Therefore, if the reverse bias increases to higher levels, a voltage drop occurs through the series and load resistances resulting in reducing the voltage across the multiplication region. At higher gain levels, a large photocurrent flows which increase the voltage drop across the series resistance leading to a non-linear relationship between the output current and the incident light. Temperature variation is another factor which constraints the APD gain at a certain level [164]. The gain starts to drop when the ambient temperature of the device rises [146]. InAlAs material has better temperature stability compared to InP, which gives the freedom to choose the optimum temperature point [140].

The time taken to initiate chain impact ionisation events and generate the electron-hole pairs is called the avalanche duration or build-up time constant. A longer time is expected when APD operates with high gain [55]. Therefore, equation 2.44, which calculates the 3-dB bandwidth, is restricted to the low-gain regime. Build-up time also

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increases proportionally with 푘푟푎푡𝑖표 and thickness of the multiplication region [165]. Reducing the multiplication region thickness does not always help to shorten the delay time as the dead space effect starts to take place. The effect of the dead space has been studied intensively in [166, 167], and it was found that the dead space phenomenon increases the avalanche build and decay times, and leads to badly degraded 3-dB bandwidth.

2.18.5 Punch-Through and Breakdown Voltages

At zero-bias, both multiplication and charge sheet layers are fully depleted as well as part of the n-contact layer (see figure 2.15). Applying a high reverse bias across the APD leads to an increase in the depletion region towards the low doped charge sheet layer. Higher bias results in expanding the depletion region towards the thick absorber layer.

The punch-through voltage (푉푃푇) is defined as the voltage which causes rapid expansion in the depletion region. At this voltage, the dark and photocurrents increase while the capacitance decreases because of the large depletion region and high generation- recombination rate in the absorption layer. After the punch-through voltage, the generated electron and hole carriers start to drift outside the absorption layer, and thus the photocurrent starts to flow in the APD. The punch-through voltage can be easily observed from the photocurrent or the capacitance-voltage characteristics as will be discussed later. At 푉푃푇, the un-multiplied responsivity is computed using equation 2.36. When the applied reverse bias is increased to higher values, the APD generates more and more impact ionisation events. As a result of this, the dark and photocurrents shoot up suddenly, exceeding 0.1mA. The corresponding voltage is denoted as the breakdown voltage (푉퐵푅) and can be easily identified from the dark and photocurrent characteristics.

푉퐵푅 is a temperature dependent factor and increases with temperature. At higher temperatures, the impact ionisation process decreases due to the increase of phonon scattering rate. This lead to a reduction in the gain in the multiplication layer. To maintain a stable and high-gain, a high-electric field is needed (implying higher 푉퐵푅 and high power dissipation) that allows more carriers to reach the ionisation threshold energy and maintain a constant gain [164, 168]. The relation between the change in breakdown voltage ∆푉퐵푅 and the change‎in‎temperature‎∆T‎is‎known as the temperature coefficient of breakdown voltage (휌푚). In a [140], an experimental and theoretical works were

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carried out to develop analytical expressions which can be used to approximately calculate 휌푚 of the multiplication layer of InAlAs and InP materials. These expressions are given by:

∆푉 휌 = 퐵푅=(15.3 푊 )+1 for InAlAs (2.48) 푚 ∆T 푚

∆푉 휌 = 퐵푅=(42.3 푊 )+0.5 for InP (2.49) 푚 ∆T 푚 where 푊푚 is the thickness of the multiplication layer. Equations (2.48) and (2.49) were derived for 푊푚=0.1 to 1.7µm and assume that the electric field distribution is uniform across the multiplication region. The unit of 휌푚 is mV/K. The equation points out that reducing the multiplication region thickness improves 휌푚. However, this is limited by the band-to-band tunnelling phenomena. Furthermore, an avalanche layer made of InAlAs material has a smaller 휌푚 compared to one based InP material at the same 푊푚 value. Assuming that the absorber is free of impact ionisation process, the total variation of the breakdown voltage with temperature of any APD can be calculated using the following expression:

∆푉퐵푅 푊푑푒푝 (APD)= 휌푚( ) (2.50) ∆푇 푊푚

The process of choosing the material and the thickness of the multiplication, as well as the absorption region, is very critical in defining the temperature sensitivity of the photodetectors. The APDs reported in [169-172] incorporated a thin multiplication layer (<100nm) to decrease the avalanche delay time. The works did not investigate the temperature sensitivity of the devices; however; it is believed to be very sensitive as the temperature increases.

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2.18.6 Noise characteristics

Noise is a critical figure of merit which determines the maximum achievable signal-to- noise ratio (SNR) and data-rate. The total noise current of the PIN diode (푖푃퐼푁) is given by [133]:

4 푘퐵푇퐵 푖푃퐼푁=√2푒(퐼푝ℎ + 퐼푑푎푟푘−푃퐼푁 + 퐼퐵)퐵+√ (2.51) 푅푒푞

The term (√2푒 (퐼푝ℎ + 퐼푑푎푟푘−푃퐼푁 + 퐼퐵) 퐵) refers to the shot noise caused by the dark and photocurrent. 퐼푑푎푟푘−푃퐼푁 is the PIN dark current, IB is the bulk current, and 퐵 is the bandwidth of noise measurement in unit of Hz. The second term (√4푘퐵푇퐵/푅푒푞) is the thermal Johnson noise, where 푘퐵 is the Boltzmann’s constant, T is the temperature of the photodetector, 푅푒푞 is equal to (푅푆 + 푅퐿). In the case of the APD, the equivalent noise current is expressed by [133, 173].

2 4 푘퐵푇 퐵 푖퐴푃퐷=√2푒(퐼푝ℎ + 퐼푑푎푟푘−퐴푃퐷 + 퐼퐵) 퐹(푀)푀 퐵+√ (2.52) 푅푒푞

퐹(푀) is the excess noise factor due to the random behaviour of the impact ionisation process. 퐹(푀) was firstly introduced by McIntyre [174] in 1966. APD introduces much higher noise compared to the PIN due to the random nature of impact ionisation. The excess noise factor can be expressed as [146, 173]:

1 퐹(푀) = 푘 푀 + (2 − ) (1 − 푘 ) (2.53) 푟푎푡𝑖표 푀 푟푎푡𝑖표

In hole multiplication region, 푘푟푎푡𝑖표 is higher than unity, therefore 푘푟푎푡𝑖표 is replaced with

1/푘푟푎푡𝑖표 in equation 2.53. Equation 2.53 is usually used to calculate the noise of thick multiplication region APD (푊푚 > 0.2µm), where the effect of local impact ionisation is dominant. Local field theory assumes a uniform electric field distribution in which the impact ionisation coefficients 훼(퐸) and 훽(퐸) are in an equilibrium state. Figure 2.17 depicts the excess noise factor as a function of the gain (푀) at different 푘푟푎푡𝑖표.

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F(M) kratio increases

M

Figure ‎2.17: APD excess noise factor as a function of multiplication gain (푴) based on local mode theory [145].

It is clear that higher 푘푟푎푡𝑖표 values increase excess noise factor. However, the newly generated and injected electrons into the high field multiplication region require a specific distance to get enough energy to perform the ionisation process, and this distance is called the dead space. Non-local impact ionisation theory takes into account the effect of the dead space on the electron or hole energy at different electric fields. Dead space effect is highly dominant in thinner multiplication regions (푊푚 < 0.2 µm) [167]. The non-local effect changes the relationship between the excess noise factor and the gain, and thus can be written as [173]:

퐹(푀) = 푘푒푓푓푀 + (1 − 푘푒푓푓) (2.54)

Where (푘푒푓푓) is the slope of 퐹(푀) with respect to (푀).

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2.19 Requirements of Multiplication and Charge Layers

To ensure high APD performances can be achieved, appropriate materials should be used to enhance the operation of the APD. The multiplication region determines the internal gain and the generated noise due to the impact ionisation process. Therefore some aspects need to be considered to choose the suitable material [55]:

1- The multiplication material should have high saturation velocity. This is to make sure that the travelling time of the carriers inside the multiplication region is as short as possible.

2- The multiplication material should be lattice-matched to the absorption layer. The difference in the lattice constant between the materials would create defects inside the crystal. Defects mean localised states at the interface because of the dislocations that can work as trapping centres to trap free carriers, resulting in degrading the performance of the device.

3- It is essential to choose a multiplication material with large energy bandgap to decrease the probability of Zener breakdown conditions. Most importantly, it is the 푘푟푎푡𝑖표 which has a significant influence on the sensitivity and the gain-bandwidth product.

4- The doping and thickness of the charge layer are critical factors in designing the APD since these factors determine the field separation between the absorption and multiplication layers. High-field separation is required to eliminate the band-to-band tunnelling current and to reduce the impact of ionisation events in the absorption layer that may degrade the APD bandwidth.

Moreover, it is desirable to lower the operating breakdown voltage and thus reduce the power consumption of the photodetector. On the other hand, the field separation is needed to be small enough in order to generate a relatively high electric field (>20KV/cm) in the absorption layer, which makes the carriers able to travel with a high speed close to their saturation velocity. In addition, low-field separation means the APD has a wide range of operating voltage between the punch-through and breakdown voltages [55].

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2.20 Summary

Chapter two described the theory of direct and heterodyne detection methods. It first showed the mixing process aided by the mathematical representation of the input and output signals. The subharmonic mixer was discussed in particular as one of the main aims of this work was to design and simulate 2nd subharmonic mixer based on ASPAT diodes. The fundamental characteristics of the direct detection technique were reviewed and discussed, including the two well-known figure of merits: voltage sensitivity and noise equivalent power. The frequency detectors performances of various devices, including the zero-bias ASPAT detectors studied in this thesis were also described. The chapter concluded with a discussion of tunnel diodes and their exceptional properties for mm-wave and sub-mm-wave applications. More focus was placed on the ASPAT diode and its operation principle under different bias.

This chapter also described in details the operational principle of the two well-known photodetectors (PIN and APD) followed by the main characteristics that are usually used to differentiate between them. This chapter also discussed the requirements of high-data- rate short and long-range photodetectors, including the optimisation of the absorber and multiplication layers for larger 3-dB bandwidth and lower excess noise factor.

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CHAPTER 3: FABRICATION AND

CHARACTERISATION OF ASPAT DIODES

3.1 Introduction

In the previous chapters, the main limitation of Schottky diodes was highlighted as being a very temperature-sensitive dependence which affects the detector performance. The backward diode as an alternative device is also discussed in which very high voltage sensitivity can be achieved at sub-mm-wave frequencies. However, the backward diode is costly and has limited cut-off frequency and reproducibility issues. Furthermore, chapter two explained the operational principle and showed the importance of the new tunnelling diode (ASPAT) as a promising candidate for room temperature zero-bias mixers and detectors for a range of applications. Two ASPAT diodes based on

GaAs/AlAs and In0.53Ga0.47As/AlAs materials were fabricated and tested in this work. The samples are denoted as XMBE#304 and XMBE#326 respectively. The focus of this chapter will be mainly on the fabrication, characterisation, and analysis of the

GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes with different mesa area sizes. The non-linear characteristics of these different ASPAT diodes will be studied for optimisation to reduce the junction resistance while maintaining a good curvature coefficient.

3.2 Epi-layer Structure of GaAs/AlAs ASPAT Diode

The GaAs/AlAs structure denoted as XMBE#304 was grown on semi-insulating GaAs substrates using a RIBER V100 HU Solid Source Molecular Beam Epitaxy (SSMBE) system. The structure of the diode consists of a single barrier (10ML=28.3Å), large band gap AlAs, in between two undoped, low band gap material spacer layers with unequal thickness, as shown in table 3.1. The ratio between the two spacers is usually (40:1) or (20:1). In this work, it was chosen to be (40:1). A smaller ratio would change the asymmetric properties of the I-V, and can also increase the leakage current. The emitter and collector layers are deliberately grown of the same thickness with a relatively low doping profile. Their functions are to prevent the diffusion of dopant atoms from the highly doped contact layers to the undoped regions. The diffusion of dopant will

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inevitably increase impurity scattering, and hence, the current decreases and the tunnelling time of electrons through the potential barrier increases. They also serve as a transition layer from the highly doped to undoped regions and reduce the abrupt change in the conduction band. The heavily doped n-type layers (4×1018 cm-3) are the ohmic contacts which connect the ASPAT diode to the anode and cathode terminals. The higher doping result in a small series resistance leading to improvements in the high-frequency performance of the device (i.e. increase in the diode cut-off frequency). Due to the thin barrier thickness compared with the electron wavelength, the electron transport through the barrier is dominated by tunnelling rather than thermionic emission over the barrier.

TABLE 3. 1: EPITAXIAL LAYER STRUCTURE OF ASPAT SAMPLE XMBE#304

Layer Material Doping (cm-3) Thickness (Å) Bandgap (eV)

18 Top Ohmic GaAs (Si) 4×10 3000 1.4 Emitter GaAs (Si) 1×1017 400 1.4 Spacer GaAs Undoped 50 1.4 Barrier AlAs Undoped 28.3 2.83 Spacer GaAs Undoped 2000 1.4 Collector GaAs (Si) 1×1017 400 1.4 Bottom Ohmic GaAs (Si) 4×1018 4500 1.4 Substrate GaAs 620µm

3.3 Mask Design and Fabrication of Discrete ASPAT Diodes

Any detector or mixer circuit is usually built using a non-linear element and passive components such as input matching, output DC circuit or IF signal parts. The individual part has to be designed and optimised to attain its maximum possible performance. The first step of this work was the design, fabrication, and measurement of the discrete ASPAT devices. The main objective was the realisation of such devices with different mesa area sizes which can be used at different frequencies. Reducing the series resistance and junction capacitance of the ASPAT diode is the key issue in designing high- frequency detectors and mixers. The RF measurement is a critical process, and therefore, a coplanar waveguide (CPW) structure was used to form the anode and cathode pads for

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the RF measurements using the Ground-Signal-Ground (GSG) port. The structure is a 50Ω impedance in which any signal reflection is minimised. Furthermore, the parasitic components associated with the CPW waveguide structure are another concern, and extra care has to be paid to reduce their impacts and accurately extract the intrinsic parameters of the diodes. Once the epitaxial growth is accomplished, the ASPAT wafers were diced into 15×15mm² size tiles for device processing.

Different masks were developed in this work employing many elements of one and two ports coplanar waveguide structures. For the GaAs/AlAs ASPAT diodes, a dielectric bridge technique was utilised to form the connection between the signal pad and anode area. In the case of the In0.53Ga0.47As/AlAs ASPAT diodes, an air bridge technique was exploited to connect the anode pad to the active mesa area of the ASPAT diode. Devices were designed with different mesa area sizes and then fabricated using i-line photolithography. The technique offers (~1µm) resolution and is easy to use compared to electron beam lithography (EBL) [175]. The fabrication process includes some necessary steps which are summarised as follow: a- Sample cleaning

The fabrication of samples took place in a cleanroom laboratory of class 1000. Contaminations from different sources might affect the fabrication process and lead to deteriorating the device performance. So firstly, the sample is placed in N-Methyl- Pyrrolidone (NMP) to remove any contaminated materials on the surface. Then to remove the remnant NMP, the sample is left in an Ultrasonic bath for some time (5 to 10 minutes). Following this, it is rinsed in de-ionised (DI) water, and dried with Nitrogen Gas. b- Photolithography technique

Fabrication of the devices consists of many phases, and every single phase is assigned to a mask. Each mask represents a particular pattern that needs to be transferred to the sample. The pattern is made with a minimum feature size, which is restricted by the technique used. The conventional i-line (365nm) photolithography can be effectively used to achieve a ~1µm resolution and is considered as a low-cost and straightforward technology compared to electron beam lithography.

To print the patterns on the sample, firstly, the sample is covered with photo-resist and spun using a Spinner. The negative photoresist becomes cross-linked when it is exposed

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to UV light. In contrast, the unexposed area is not and can be quickly dissolved in a developer. The positive photoresist becomes soluble, and quickly, the bond breaks with light exposure. Following that, a soft baking at 110oC for 1 minute takes place to stabilise the resist and remove any excess solvent. Then the mask is exposed with UV light with intensity of ~0.9mW.cm-2. c- Etching

This step is used to remove unwanted layers and form the active structure. It is imperative to produce the device with the same designed dimensions, and for this, full knowledge of the etching techniques is required. The etching is usually performed using either wet or dry techniques. Wet etching uses chemical materials such as acidic solutions or etchant depending on the desired etching rate to remove the semiconductor layers. It is a cost-effective and fast process, but unfortunately, the etching occurs both in the lateral and vertical directions. GaAs/AlAs ASPAT samples were processed utilising wet chemical etching. The etching process consists of two stages. The first etching step defines the diode mesa area and stops at the bottom GaAs ohmic layer. The second etching step isolates each device and stops at the substrate. d- Metal contact formation

The last step of the process is the connection between the device and the metal contacts. There are two methods to deposit the metal namely evaporation or sputtering. In this work, an evaporation method was used throughout the fabrication of discrete devices and detectors. An alloy of Au/Ge/Ni metal was used to form the ohmic contact to the GaAs/AlAs sample. So for samples from wafer XMBE#304 where the bottom and top GaAs layers are not very heavily doped (4×1018 cm-3), annealing is necessary to diffuse the Ge n-dopant from the metal to the GaAs layer. The diffusion minimises the depletion region and reduces the spikes at the interfaces. As a result, an ohmic contact is created. Once the metal contacts are formed, a lift-off process is used to remove the unwanted metals which sit on the photo-resist. Finally, the sample is rinsed in DI water.

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3.4 Mask Structures

The mask includes different structures of the single diodes as well as various test structures. These structures are used for DC and RF characterisations to investigate their intrinsic performances. Structures such as open and short bond pads are included in order to extract the extrinsic parameters associated with the actual structures. Moreover, the resistance between the metal contact and the semiconductor needs to be evaluated for the actual structures, and this is usually performed using the transmission line model (TLM) structure. All structures can be categorised into two main groups as follow:

3.4.1 Open, Short, and ASPAT Diode Structures

Two designs, designated as ‘‘standard‎ and‎ optimised’’‎ coplanar‎ waveguides‎ were‎ designed as bond-pad structures which allow the DC and RF measurement process of the devices. The optimised CPW structure is considerably smaller in area compared to the standard one and was used as a bond pad with the smaller mesa area ASPAT diodes (2×2µm2 and 3×3µm2). The standard CPW design shown in figure 3.1, was utilised with the large mesa area ASPAT diodes (4×4µm2, 6×6µm2, and 10×10µm2).

Size~30000µm2 GaAs-Substrate

Standard Design Ground 165µm Polyimide 75µm 65µm

Signal

Ground

µm 65

Figure ‎3.1: 3D structure drawing of GaAs/AlAs (XMBE#304) ASPAT diode with its standard CPW

bond pad. The inset shows the separation distance (푫풔풑풓) between the top anode contact and bottom contact pad (cathode).

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More details regarding the coplanar waveguide design consideration are presented in chapter 4. The open and short CPW structures were also associated with the mask in order to apply the de-embedding technique and extract the intrinsic parameters of the ASPAT diodes. All bond-pad structures are located on a dielectric layer (polymer) above the semi-insulating GaAs substrate. The separation (퐷푠푝푟) between the anode and cathode contacts of the ASPAT diode is vital in defining its series resistance. Increasing the separation (퐷푠푝푟) results in a corresponding increase in the bottom series resistance which is the main contributor to the total device resistance. Thus, a small (퐷푠푝푟) is necessary to minimise this series resistance, but unfortunately, this is limited by the manual alignment tolerances in the order of 1 to 2µm in i-line optical lithography. An optimisation of many devices including ASPATs and RTDs was conducted to determine the optimum value of 퐷푠푝푟 in order to reduce the losses introduced by the bottom ohmic layer and to provide a high yielding process. For this work, 퐷푠푝푟 was designed to be 1.5µm. The input port was designed to be matched to a GSG probe with 50µm signal conductor width and 35µm separation between the signal and ground lines. Similarly, open and short structures are important parts in the mask which can give an insight to the parasitic elements caused by the bond pad structure. Optimisation of CPW dimensions was performed to reduce the associated parasitic parameters which could dominate the intrinsic components of the device. The extraction methods of the parasitic parameters of the bond-pad structures are discussed in this chapter later on.

3.4.2 Transmission Line Model Structure

Any contact brought up with a semiconductor layer results in a depletion region forming and a Schottky barrier introduced at the interface. In electronic devices, the doping profile of the semiconductor layer is intentionally made high enough to reduce the depletion region thickness, and as a result, make the field emission transport dominant and thus a good ohmic contact is established. The ohmic contact is usually assessed by its contact resistance, which contributes to the total series resistance of the device. An effective way introduced by Berger [176] called the transmission line model (TLM) has been used to evaluate the contact resistance of the metal-semiconductor connection. The 3D representation of the TLM structure is presented in figure 3.2. The structure consists

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of rectangular pads made of alloyed Au/Ge/Ni metal with a size of 100×50µm2 and sits on a highly doped n+-GaAs layer. It includes nine pads separated by a distance of 푑푛.

Figure ‎3.2: A 3D schematic and side view of the TLM structure used in the masks. (Note that the image is not to scale).

The separation starts with 푑1=40µm and decreases to a final value of 푑8=5µm. 퐿푇 is the effective length of the pad in which the current flows from and into the next pad. Finally,

푅푠푘 and 푅푠ℎ are the sheet resistances under the effective contact area and between the

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neighbouring pads. The total resistance between the first two pads as a function of the given dimensions can be expressed as [177]:

퐿푇 푑1 푅푇 = 2푅푠푘 + 푅푠ℎ (3.1) 푊푝푎푑 푊푝푎푑

퐿푇 The term 푅푠푘 is denoted as the contact resistance 푅푐. The total resistance is 푊푝푎푑 evaluated by passing a current and measuring the drop voltage between the first adjacent pads which are separated by 푑1. To ensure high accuracy of measurement, the process is repeated for 푑2 to 푑8. Thus, the total resistance can be reformatted as:

푑푛 푅푇푛 = 2푅푐 + 푅푠ℎ (3.2) 푊푝푎푑

where 푅푇푛 is the total resistance at distance 푑푛. Assuming a constant sheet resistance of the material, the mathematical representation of the total resistance can be plotted as a function of 푑푛 as shown in figure 3.3. For 푑푛=0, the total resistance is equal to 2푅푐. To find 퐿푇, 푅푇푛 is set to 0, and the line is extrapolated to intercept with X-axis. The interception point is equal to 2퐿푇.

Figure ‎3.3: Total resistance versus separated distance (풅풏) of TLM structure. [177].

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Once 푅푐 and 퐿푇 are extracted from the measured total resistance; the specific contact resistance can be easily calculated using the following equation:

휌푐 = 푅푐퐿푇푊푝푎푑 (3.3)

Figure 3.4 shows the measured total resistance of the top and bottom GaAs TLM 푅 structure. The measured data showed a linear slope 푠ℎ which is related to the constant 푊푝푎푑 sheet resistance of the materials. The sheet resistance was extracted from the measured linear‎curve‎and‎found‎to‎be‎~23Ω/sq.‎The‎specific‎contact‎resistance‎of‎the‎top‎contact‎ was too calculated and found to be ~38Ω.µm2. Similarly, measurements were accomplished for the top contact of the InGaAs TLM sample as it was reported in our group previously [178]. The data showed a better sheet resistance and specific contact resistance‎ of‎ 5.9Ω/sq‎ and‎ 12.4Ω.µm2 respectively due to the deliberately high doping profile of the InGaAs layer.

10 Top contact y = 0.2304x + 0.6029 Linear (Top contact)

7.5

) Ω

5 Total resistance( Total

2.5

0 0 10 20 30 40 Separated distance (µm)

Figure ‎3.4: Measured TLM of the top contact of GaAs/AlAs ASPAT XMBE#304 sample.

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3.5 Intrinsic Parameters of ASPAT Diode

Junction capacitance (퐶퐽), junction resistance (푅퐽), and series resistance (푅푆) are the intrinsic parameters of a two-terminal diode. These parameters are defined by the epi- layer structure and geometry of the device. Figure 3.5 shows the side view of the ASPAT diode with its associated intrinsic parameters.

Figure ‎3.5: The right side is the 2D sectional view of the ASPAT diode. The left side is the intrinsic component of each layer.

3.5.1 Junction Capacitance and Junction Resistance

Junction capacitance (퐶퐽) is the two parallel plate and fully depleted junction capacitance and it is calculated using the following formula [93]:

퐴푚푒푠푎 퐶퐽 = 휀0휀푟( ) (3.4) 푡푑 where 휀표 and 휀푟 represent the permittivity of free space and the undoped active layer of 2 the diode. 퐴푚푒푠푎 is the anode or mesa area size in 푚 unit. Finally, 푡푑 refers to the

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thickness of the fully depleted active region (spacer1+barrier+spacer2). The fully depleted junction capacitances of GaAs/AlAs ASPAT diodes were calculated and found to be (~2.2fF, ~5fF, ~8.7fF, ~19.7fF, and ~55fF) for the (2×2µm², 3×3µm², 4×4µm², 6×6µm², and 10×10µm²) ASPAT diodes respectively. Having such a diode with a thick undoped region would result in a junction capacitance varying significantly with bias. The S-parameter measurements are usually performed to estimate and extract the junction capacitance at different bias as will be discussed later. The junction resistance

(푅퐽) is another important factor in determining the diode performance. 푅퐽 is a voltage- dependent parameter and can be found from the measured I-V characteristics using the expression [93]:

휕푉 푅 = (3.5) 퐽 휕퐼

푅퐽 plays a vital role in the detection process, and it mainly depends on the barrier thickness, height, and mesa area size. In thermionic emission devices, 푅퐽 varies with barrier height and ambient temperature, in opposition to tunnelling devices, where only the barrier thickness and height control the non-linear junction resistance [93].

3.5.2 Series Resistance

The series resistance 푅푆 on the other hand, counts for the losses of the diode structure across all layers and contacts. 푅푆 arises because of three main elements as clearly shown on the left side of figure 3.5. The resistance of the individual layer (R_top ohmic, R_Emitter, and R_Collector) is calculated using the following equation:

1 푡푙푎푦푒푟 푅푒푝𝑖−푙푎푦푒푟 = (3.6) µ푛 푒 푁퐷 퐴푚푒푠푎 where µ푛 and 푁퐷 are the electron mobility and doping concentration of the material,

푡푙푎푦푒푟 is the thickness of the layer, and 퐴푚푒푠푎 is the mesa area size of the diode.

The spreading resistance (R_spreading (푅푠푝푟)) results at the bottom ohmic layer due to current flow in the horizontal direction. 푅푠푝푟 takes into account the effect of the separation distance between the anode and cathode metal contacts and is given by:

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1 푎 푅푠푝푟 = 푙푛 ( ) (3.7) 휋 µ푛 푁퐷 푒 푡푏표푡푡표푚 푎푚푒푠푎 where 푡푏표푡푡표푚 is the thickness of the bottom ohmic layer in cm unit. Once the epi-layer and spreading resistances are calculated, the total series resistance of ASPAT diode can be obtained using the expression:

휌푐 푅푠 = 푅푠푝푟 + 푅푒푝𝑖−푙푎푦푒푟푠 + (3.8) 퐴푚푒푠푎

The calculations of the series resistance for the GaAs/AlAs (XMBE#304) ASPAT diodes of different mesa area sizes are shown in table 3.2. The calculations were estimated assuming that the spacers are entirely depleted and the resistance of undepleted spacer layers is equal to zero.

TABLE 3. 2: CALCULATED SERIES RESISTANCE OF THE GaAs/AlAs ASPAT DIODES

Resistance 2×2µm2 3×3µm2 4×4µm2 6×6µm2 10×10µm2

Top Ohmic (R_top 1.17 0.52 0.3 0.13 0.04 ohmic),‎Ω

Emitter‎(R_Emitter),‎Ω 2.5 1.1 0.62 0.27 0.1

Collector (R_Collector), 2.5 1.1 0.62 0.27 0.1 Ω

푅푠푝푟,‎Ω 10.1 7.6 6.18 4.48 2.9

휌푐/퐴푚푒푠푎,Ω 9.6 4.33 2.4 1.07 0.38

Total 푅푠,‎Ω ~26 ~15 10.12 6.24 3.5

The calculations above emphasise the dominance of the spreading resistance on the total series resistance. However, the contact resistance 휌푐/퐴푚푒푠푎 increases as the mesa area size decreases. Thus, the series resistance dramatically increases. Reducing the series resistance of the small mesa area diode can be effectively achieved by employing a

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material with small specific contact resistance 휌푐 such as InGaAs. Both 푅푆 and 퐶퐽 determine the cut-off frequency 푓푐 of the diode. A lower series resistance implies a higher cut-off frequency and high output efficiency. At the cut-off frequency, the diode behaves like a simple series circuit of 푅푆 and 퐶퐽, while 푅퐽 is negligible [93]. The total resistance including 푅퐽 and 푅푆 is usually called the video resistance 푅푉. However, at zero-bias, 푅퐽

>> 푅푆 and thus 푅푉=푅퐽.

3.6 DC Characteristics of GaAs/AlAs ASPAT Diodes

The large mesa area GaAs/AlAs ASPAT diodes (4×4µm2, 6×6µm2, and 10×10µm2) were fabricated first to investigate their I-V characteristics and validate the operational principle of the fabricated diodes. The use of different etching solutions and fabrication process usually resulted in having slightly different mesa area structures to the designed ones due to undercut issues. Four different samples of each device were DC characterised to investigate the uniformity over the wafer tile. High uniformity is indispensable to make sure that all fabricated devices have comparable performances. In particular, any variation in barrier thickness of the ASPAT over the wafer could lead to massive changes in the measured current and hence poor uniformity. The measured forward and reversed currents of the 4x4µm2, 6x6µm2, and 10x10µm2 GaAs/AlAs ASPAT diodes are depicted in figure 3.6. The sub-figures (a), (b), and (c) depict the measured I-V characteristics of different mesa area devices located on the wafer tile. The sample names are shown in the legends of the figure. The fabricated samples exhibited excellent uniformity in the current at low bias where the tunnelling transport mechanism is dominant. This indicates a uniformly grown barrier thickness across the wafer. However, a small deviation in the current was observed at bias > 0.75V for the 4×4µm2 (AI34) sample. The reason could be mainly due to the higher 푅푆 (slightly smaller mesa area size) which reduces the current at higher bias. At high bias, 푅퐽 becomes low, and no tunnelling mechanism occurs. Figure 3.6 (d) shows a distinct non-linear transition at zero-bias.

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1.00 2.5 AA40 3.7x3.7m2 AA35 2 AE40 AE41 5.8x5.8m AA31 AI41 AI34 2.0 AA41 0.75

1.5

0.50

1.0

Current (mA) Current Current (mA) Current 0.25 0.5

0.00 0.0

-1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 Voltage (V) Voltage (V) (a) (b)

8 101 2 2 AA30 10x10m 3.7x3.7m _AA31 AA39 5.8x5.8m2_AA35 AA42 100 10x10m2_AA30 6 AE42

10-1

4

10-2 Current (mA) Current Current (mA) Current 10-3 2

10-4

0 10-5 -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 Voltage (V) Voltage (V) (c) (d)

Figure ‎3.6: Measured I-V characteristics of GaAs/AlAs ASPAT (wafer XMBE#304) diodes of (a): 3.7x3.7µm2, (b): 5.8x5.8µm2, (c): 10x10µm2. (d): Log representation of the measured currents showing the non-linear characteristics at zero-bias.

After that, current densities were calculated as shown in figure 3.7. At bias > 1V, there was a small difference in the measured forward current between the devices which is believed to be due to differences in the mesa area sizes. The fabricated mesa area size could be slightly different from the designed one in the mask caused by the undercut process or light scattering during the exposure. With this in mind, the optimised mesa area size of the devices was found to be 3.7×3.7µm² and 5.8×5.8µm² instead of 4x4µm2

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and 6x6µm2, respectively. At higher bias, the series resistance starts to dominant the I-V characteristics as the junction resistance decreases. Another reason could be due to the difference in the non-linear resistance 푅푢 of the undepleted layers of the devices under forward bias. Following this, work was carried out on the design and fabrication of much smaller mesa area size [2×2µm² and 3×3µm2] ASPAT diodes for mm-wave circuit design. The fabrication of the 2×2µm² and 3×3µm² ASPAT diodes was carried out using different wafer (XMBE#421) which is a replica of the wafer (XMBE#304). Similarly, the current densities were calculated and that the optimised mesa area size of the ASPAT diodes are 1.6×1.6µm² and 2.4×2.4µm².

0.08 3.7x3.7m2 XMBE#304 samples 5.8x5.8m2 10x10m2

0.06

)

2

m 

mA/ 0.04 (

0.02 Current density Current

0.00

-1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 Voltage (V)

Figure ‎3.7: Measured current densities of the fabricated 3.7×3.7µm2, 5.8×5.8µm2, and 10×10µm2 GaAs/AlAs ASPAT diodes using wafer XMBE#304.

Figure 3.8 depicts the log representation of the current density of the devices from two different wafers (XMBE#304 and XMBE#421). It is clear that all samples have identical measured current densities from -1.5 to 1.5V bias. A small leakage current density of (~0.001mA/µm2) was measured at -1.5V bias. The thick spacer (200nm) provided enough blocking to the carriers transitioning the barrier at high reverse voltages. Due to the high-sensitivity of the I-V characteristics in tunnelling diodes upon the ultra-thin barrier, there could be a noticeable difference in the non-linear characteristics between

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devices from different wafers. It was empirically validated in [179] that a 0.2ML variation in the AlAs barrier thickness in the ASPAT diode leads to a pronounced change in the forward current.

10-1 1.6x1.6m2 XMBE#421 samples 2.4x2.4m2 } 10-2 3.7x3.7m2 5.8x5.8m2 XMBE#304 samples

) 2 2 10x10m } -3

m 10

mA/ ( 10-4

10-5 Current density Current 10-6

10-7 -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 Voltage (V)

Figure ‎3.8: Measured current densities of the devices from two wafers (XMBE#304 and XMBE#421).

3.7 RF Characteristics of GaAs/AlAs ASPAT Diodes

S-parameter measurements using on-wafer probing were performed for open, short and actual diode structures up to 40GHz as depicted in figure 3.9. The figure shows the standard CPW design and its open and short structures. A calibrated Vector Network

Analyser (VNA, Anritsu 37369A) was utilised to collect the reflection coefficient (푆11) data of the fabricated one port CPW structures. RF characterisation is a prior step to the design of the complete integrated circuits (ICs) and is vital for assessing the high- frequency performances of the diodes used as well as extracting the maximum frequency of operation.

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3.7×3.7µm2 Open Short

Figure ‎3.9: Example of the fabricated standard CPW ASPAT diode, open, and short structures of mesa area size 3.7×3.7µm².

The measurement and extraction of the small-signal equivalent circuit of the ASPAT diodes included in this work can be divided into two steps:

3.7.1 RF Characteristics of the Open and Short Bond Pad Structures

This section deals with the extraction and optimisation of the parasitic elements associated with the CPW structures. The parasitic capacitance comes from the pad-to-pad separation of the CPW structure and increases linearly with the area of the pads. The separation between the anode bridge and cathode bottom contact could also introduce another parasitic capacitance that has a higher impact at high frequencies. Bond pad structures are not incorporated in the detector or mixer circuit, and their additional parasitic elements have to be evaluated to extract the intrinsic components of the ASPAT diode itself accurately. Optimisation of the CPW dimensions is required to reduce the parasitic capacitance caused by fringing effects without an increase in conductor losses or‎change‎in‎the‎designated‎characteristic‎impedance‎(50Ω)‎[180-182]. A two steps de- embedding method [183] was used to find the parasitic parameters of the CPW bond pad structure associated with the ASPAT diode. The method uses the following equations to find the parasitic capacitance (퐶푃) and inductance (퐿푃) of the one port structure:

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퐼푚푎푔(푌11표푝푒푛) 퐶 = (3.9) 푃 휔 1 퐿푃 = (3.10) 휔[퐼푚푎푔(푌11표푝푒푛−푌11푠ℎ표푟푡)]

where 푌11 is the Y-parameter extracted from the measured reflection coefficient (푆11). The equivalent circuits of open and short bond pads were built in ADS tool, as shown in the insets of figure 3.10 (e). Then, the tuning process was initialised to find 퐶푃 and 퐿푃 of the standard CPW structure which gives the best fitting between the measured and simulated 푆11 data up to 40GHz. Figure 3.10 depicts the high correlation measured (red lines) and simulated (blue lines) real and imaginary parts of the reflection coefficient

(푆11) of open and short CPW structures used in this work. The blue line represents the simulated data of the built equivalent circuits. In an ideal case, the open and short structures can be represented by only a capacitor and , respectively. However, the fitting process showed a small resistance associated with the short CPW structure, as shown in the inset of figure 3.10 (e). The resistance is very small and can be neglected. Additional parasitic resistances could be added due to the cables or bad calibration of the measurement equipment. The extraction of the extrinsic parameters was carried out for different open and short structures located on the different places on the wafer tile. The use of equations 3.9 and 3.10 as well as equivalent circuit fitting process exhibited 퐶푃,

퐿퐶푃푊, and 푅퐶푃푊 of ~18fF, (40 to 50pH), and (0.5 to 1Ω) respectively for the standard

CPW structure. The 퐿퐶푃푊 value was found to be variable from one structure to another, and this is due to the variation of the GSG probe location from one measurement to the other. Having such a parasitic capacitance of ~18fF could dominate the junction capacitance, and thus 푆11 of the smaller ASPAT diodes leading to inaccurate measurement and extraction process.

Reduction of 퐶푃 can be achieved by firstly reducing the substrate thickness and dielectric constant, secondly by minimising the signal and ground pads area, and thirdly by increasing the separation between pads. The latter is not a practical solution since pad separation is defined by the dimension of the input GSG port.

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0.6 Measured Measured -0.75 ADS equivalent circuit ADS equivalent circuit Momentum Microwave 0.5 Momentum Microwave

-0.80 0.4

0.3 )

-0.85 )

11

11

S

(

S

( 0.2 Real -0.90 Imag Imag 0.1 -0.95 0.0 Short bond pad Short bond pad -1.00 -0.1 0 10 20 30 40 0 10 20 30 40 Frequency (GHz) Frequency (GHz) (a) (b) 1.1 0.0 Measured Measured ADS equivalent circuit ADS equivalent circuit Momentum Microwave Momentum Microwave -0.1

1.0 )

) -0.2

11

11

S

S

( (

Real Real -0.3

0.9 Imag

-0.4

Open bond pad Open bond pad 0.8 -0.5 0 10 20 30 40 0 10 20 30 40 Frequency (GHz) Frequency (GHz) (d) (c)

Lcpw

Rcpw

CP

(e)

Figure ‎3.10: (a), (b), (c), and (d) are the real and imaginary part of 푺ퟏퟏof open and short bond pad structures. (e) The Smith chart representation and the built circuits of the open and short bond structures in ADS.

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Furthermore, increasing pad separation leads to a variation in the total impedance of the CPW structure. Above all, it could result in having an excessive inductance of the CPW structure [184]. In this work, the substrate thickness of the standard CPW bond pad structure was reduced to examine its effect on the total parasitic capacitance. A momentum simulation tool embedded in ADS was used to simulate and fit the 푆11 of the open standard CPW structure to the measured data up to 40GHz. The momentum simulation tool offered a good fit to the measured 푆11 as shown in the black and red lines of figure 3.10. At this point, the optimum condition and the appropriate simulator mode were determined. For this simulation, a momentum microwave mode was used since it considers the effect of radiation loss at high frequencies. In the simulation, a Ground- Signal-Ground (GSG) probe was used as an input port to feed the CPW structure. The metal thickness was 1µm. More details regarding the ADS momentum simulation will be presented in chapter 4. Following this, the substrate thickness was thinned down to 5µm, and the parasitic capacitance was evaluated, as shown in figure 3.11.

50 Measured MoM (625m) MoM (20m) 40 MoM (15m) MoM (10m) MoM (5m)

30

(fF) P

C 20

10

0 1 10 100 Frequency (GHz)

Figure ‎3.11: The measured and simulated parasitic capacitance versus frequency of the standard CPW structure for different substrate thicknesses.

The minimum simulated parasitic capacitance of the standard CPW structure was ~7fF at a substrate thickness of 5µm. A good matching was achieved between the measured

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(black line) and simulated (red line) parasitic capacitance of the open standard bond-pad structure up to 40GHz. Reducing substrate thickness significantly decreases the parasitic capacitance at frequencies up to 40 GHz. At frequencies >100GHz, CPW structure with substrate thickness >15µm shows an exponential increase of its parasitic capacitance due to the higher dispersion of the effective dielectric constant of the substrate at higher frequencies [185]. The parasitic capacitance is mainly limited by the smallest mesa area size of the diode and the maximum frequency of characterisation. If 퐶푃≥퐶퐽, then the total measured 푆11 is dominated by the bond-pad behaviour since it shunts intrinsic parameters and cancels diode behaviour. Indeed, a very thin substrate cannot be used to fabricate a high density and large integrated circuits due to its handling and fragility issues. Another approach was carried out to improve the parasitic capacitance of the structure. The size of the CPW pads was reduced, as shown in figure 3.12. The length of the signal pad and width of the ground pads were reduced to 50µm instead of 75µm and 65µm of the standard CPW design. Accordingly, the size of the optimised one and two-port CPW structure is ~11,300µm² and ~20,000µm2 respectively compared to ~30,000µm² of the standard design.

Size ~11300µm2 Size ~20000µm2

µm 50

50µm

µm 220

Figure ‎3.12: Fabricated optimised one and two-port open bond pad CPW structure.

Bond pad structures were fabricated with one and two-port CPW configurations on a substrate thickness of 625µm, and then the one-port S-parameter measurements were performed up to 40GHz. Similar procedures were followed to extract the parasitic

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capacitance and to perform the MoM microwave simulation of the optimised CPW structure. The total capacitance of two-ports open CPW structure was calculated using the expression (-퐼푚푎푔(푌12)/ω)‎and‎found to be ~5fF which agrees well with the extracted and simulated ones up to 40GHz as depicted in figure 3.13. It can be said that making the CPW structure size smaller is more realistic and has a considerable benefit in minimising the parasitic capacitance compared to the approach of thinning the substrate thickness. For the purpose of theoretical investigation, the substrate thickness was varied from 5 to 20µm with a 5µm step, and the parasitic capacitance was extracted and plotted versus frequency, as shown in figure 3.13.

25 Measured MoM (625m) MoM (20m) 20 MoM (15m) MoM (10m) MoM (5m)

15

(fF) P

C 10

5

0 10 100 Frequency (GHz)

Figure ‎3.13: The measured and simulated parasitic capacitance versus frequency of the optimised CPW structure for different substrate thicknesses.

At a substrate thickness of 5µm, the parasitic capacitance was reduced to half (2fF) and showed a very small variation at high-frequencies up to 200GHz. Such a way to minimise 퐶푃 is risky and undesired since the substrate thickness has to be reduced by a factor of (625µm/5µm=125), making the platform extremely difficult to handle and easy to break during fabrication and measurement.

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3.7.2 RF Characteristics of GaAs/AlAs ASPAT Diodes

This section describes the characterisation of the actual ASPAT diodes besides a detailed analysis of their small-signal equivalent circuits. On-wafer S-parameter measurements of standard and optimised CPW ASPAT diodes were carried out at different biases. Once the parasitic elements had been accurately evaluated, the next step was the extraction of the intrinsic parameters (푅푠, 퐶퐽, and 푅퐽) of large mesa area size GaAs/AlAs ASPAT diodes (3.7×3.7µm2, 5.8×5.8µm2, and 10×10µm2). The fabrication of smaller mesa area diodes (1.6×1.6µm2 and 2.4×2.4µm2) was performed not only to enhance the cut-off frequency but also to investigate the non-linearity properties as the junction resistance substantially increases. The intrinsic parameters of 1.6×1.6µm2 and 2.4×2.4µm2 ASPAT diodes were extracted from one and two-port S-parameters measurement. The two de- embedding step method also uses the following equation to calculate 퐶퐽 of the one-port CPW structure as follows:

1 1 퐶 = ( ) 퐽 1 1 (3.11) 휔 + 퐼푚푎푔(푌11푑𝑖표푑푒 − 푌11표푝푒푛) 퐼푚푎푔(푌11표푝푒푛 − 푌11푠ℎ표푟푡)

The fitting process was initially performed at a negative bias of -0.5V, where it is assumed that both spacers are entirely depleted. The simple equivalent circuit model that takes into account the effect of all parameters was built in ADS, as shown in figure 3.14

(a). The parasitic components 퐶푃, and 퐿푃 were taken into account in the total equivalent circuit of the actual diode. The fully depleted of the ASPAT diodes were calculated using equation (3.4). The extraction process at such voltages is straightforward as the ASPAT undoped region is totally depleted and virtually no current flows through the structure. This results in having a series resistance that is independent of the variation in the bias. Figure 3.14 (b) shows the excellent fitting between the measured and 2 simulated 푆11 of the 3.7×3.7µm GaAs/AlAs ASPAT diode at -0.5V bias. The aim of this work was to employ the temperature-insensitive ASPAT diodes in the zero-bias detector and mixer integrated circuits. Therefore, it is imperative to extract ASPAT parameters at zero and forward bias and investigate their performances for possible high-frequency applications.

111

CJ

RCPW LCPW RS

RJ

CP (a)

Measured Simulated 0.0 1.2

-0.2

11 11

1.0

Real S Real Imag S Imag -0.4

-0.6 0.8 0 10 20 30 40 Frequency (GHz)

(b)

Figure ‎3.14: (a): ASPAT equivalent circuit built in ADS at negative bias, and (b): The measured and

simulated real and imaginary parts of 푺ퟏퟏof the one-port CPW GaAs/AlAs ASPAT diode of mesa area 3.7×3.7µm2 at -0.5V bias.

In the case when the spacers are not fully depleted, additional components such as displacement capacitor (퐶푑) and resistance of the undepleted region (푅푢) are crucially required to be involved in the equivalent circuit, as shown in figure 3.15 (a). 퐶푑 accounts for the capacitance of the undepleted layers which has a higher impact at frequencies above 100GHz and can be found using equation 3.4 [186], with only substituting 푡푑 by

푡푢. Here 푡푢 is the thickness of the undepleted regions under zero or forward bias. 퐶푑 was not found to have an effect in the equivalent circuit model up to 40GHz. The most crucial

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parameter is the resistance of the undepleted spacer layers (푅푢). This is a non-linear voltage-dependent resistance and was reported previously in thick undoped layer structures such as Schottky Varactor diodes [187, 188]. It is significantly important to state that the frequency behaviour of ASPAT diodes is greatly restricted by its junction capacitance at high frequencies. At such frequencies, the impedance of the junction capacitance (푋푐 = 1/2휋푓퐶퐽) decreases which shunts out the junction resistance and the non-linear characteristics are suppressed. Figure 3.15 (b) presents the measured and simulated real and imaginary parts of the 푆11 and the Smith chart representation of the 3.7×3.7µm2 GaAs/AlAs ASPAT diode at zero bias and up to 40GHz.

CJ

LCPW RCPW RS Ru

RJ Cd

CP (a) 0.50 Measured 1.2 Simulated

0.25

1.0

0.00

11 11

-0.25 ImagS

Real S Real 0.8

-0.50

0.6

-0.75 0 10 20 30 40 Frequency (GHz)

(b)

Figure ‎3.15: (a): ASPAT equivalent circuit built in ADS at zero and forward bias, and (b): The

measured and simulated real and imaginary parts of 푺ퟏퟏof the one-port CPW GaAs/AlAs ASPAT diode of the mesa area 3.7×3.7µm2 at zero-bias.

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Table 3.3 lists the extracted parameters of the one port CPW 3.7×3.7µm2, 5.8×5.8µm2, and 10×10µm2 GaAs/AlAs ASPAT diodes.

TABLE 3. 3: EXTRACTED INTRINSIC AND EXTRINSIC PARAMETERS OF GaAs/AlAs ASPAT DIODES

3.7×3.7µm² 5.8×5.8µm² 10×10µm²

Parameter -0.5V 0V 0.25V -0.5V 0V 0.25V -0.5V 0V 0.25V

퐶퐽, fF ~8 21 29 19.2 54 76 55 140 198

푅푆,‎Ω 11 11 11 6.8 6.8 6.8 3.5 3.5 3.5

푅푢,‎Ω 0 12 28 0 5 11 0 3.3 5

푅퐽,‎kΩ ~240 ~100 12 ~100 ~40 5 ~35 ~15 1.8

퐶푃, fF 18 18 18 18 18 18 18 18 18

퐿퐶푃푊, pH 45 45 45 45 45 45 40 40 40

푅퐶푃푊,‎Ω 0.75 0.75 0.75 0.75 0.75 0.75 0.75 0.75 0.75 Intrinsic

풇풄풖풕−풐풇풇, 1700 330 140 1137 245 116 750 155 89 GHz

The extracted junction capacitances at -0.5V bias are in excellent agreement with the calculated ones for the 3.7×3.7µm² and 5.8×5.8µm² GaAs/AlAs ASPAT diodes (8 vs 8.7fF, 19.2 vs ~19.7fF). The depletion region thickness was then calculated and found to be 2000Å, which theoretically demonstrates that both spacers are fully depleted.

Similarly the extracted 푅푆 at -0.5V bias is identical to the calculated values shown in table 3.2 for the 4×4µm2, 6×6µm2, and 10×10µm2 GaAs/AlAs ASPAT diodes respectively. Both 퐶퐽 and 푅푆 were used to theoretically calculate the intrinsic cut-off frequency using the expression (1/2휋푅푆퐶퐽). A high 푓푐푢푡−표푓푓 of ~1.7THz was extrapolated for the 3.7×3.7µm2 ASPAT diode at -0.5V bias. Junction capacitances were shown to increase gradually with bias as a result of narrowing of the depletion region.

The highest extracted values were at 0.25V, due to the additional capacitance from the 2D states located in the accumulation region and caused by the negative charges in the accumulation region, which are imaged by the positive charges in the collector depletion

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region [189, 190]. A similar trend has been observed with resonant tunnelling diodes. A graded emitter spacer layer could be one of the options to reduce the band bending effect [51], and thus, such additional small capacitance might be reduced. The depletion layer thickness at zero-bias was then calculated from the extracted 퐶퐽 and found to be (850- 1000Å). Evidently, half of the thick spacer is treated as an undepleted region. The undepleted region resistance 푅푢 was shown to have the same trend of 퐶퐽 which varies with the bias due to the change in the depletion region thickness.A Fitting process yields

푅푢 values of 12Ω, 5Ω, and 3.3Ω at zero bias for the 3.7×3.7µm², 5.8×5.8µm², and 10×10µm² ASPAT diodes respectively. The cut-off frequencies were also calculated at zero-bias by taking into account the total series resistance of the device (푅푆 + 푅푢). A maximum (푓푐푢푡−표푓푓) of 330GHz at zero-bias was obtained using the 3.7×3.7µm² ASPAT diode. To sum up, the temperature-insensitive 3.7×3.7µm² ASPAT diode can be used for the integration of MMIC detector or mixers up to 110 GHz frequency. One effective method that could help to improve the cut-off frequency is to minimise the undepleted region thickness and accordingly reducing 푅푢. However, this will also result in the increase of junction capacitance and also affects the non-linearity characteristics of the device. An ASPAT structure with 1000Å spacer thickness was grown recently, and progress is ongoing to fabricate and characterise the devices in order to investigate their non-linear resistances 푅푢 and other characteristics.

Next, a two-port equivalent circuit was built to extract the intrinsic components of the 1.6×1.6µm² and 2.4×2.4µm² GaAs/AlAs ASPAT diodes at zero-bias. Figure 3.16 shows an example of the reasonable fitting between the measured and simulated 푆11 and 푆12 of the 2.4×2.4µm² ASPAT diode at zero-bias and up to 30GHz. However, there is a slight deviation between the measured and simulated 푆12 at frequencies > 20GHz. The reason could be due to the inaccuracy of the simple equivalent circuit model used at high- frequency regimes. The extracted intrinsic parameters of the 1.6×1.6µm² and 2.4×2.4µm²

ASPAT diodes at zero-bias were as follow: 퐶퐽= 3 and 11.2fF, 푅푆=26 and 18Ω,‎푅푢=50 and‎22Ω,‎푅퐽=‎~580‎and‎~200kΩ.‎Small 퐶퐽 allows more current passing through the non- linear resistance at operation at much higher frequencies. The cut-off frequencies were evaluated and found to be 355 and 770GHz, respectively. The reduction of 퐶퐽 of the

2.4×2.4µm² ASPAT diode was not sufficient enough to compensate for the increase in 푅푆 and 푅푢.

115

Figure ‎3.16: Smith chart representation of the two-port 2.4×2.4µm² ASPAT diode at zero-bias. Red

and blue lines are measured and simulated 푺ퟏퟏ respectively. Red and blue dashed lines are the

measured and simulated 푺ퟏퟐ respectively.

Such a high series resistance limits its use for sub-mm-wave applications. Further optimisation of the epi-layer structure could improve the series resistance and pave the way toward the fabrication of sub-micron devices for mm-wave and sub-mm-wave detection systems. It is noteworthy to indicate that the technique used for extraction is simple and straightforward and can be applied for devices with 퐶퐽 > 퐶푃 and up to 40GHz frequency operations. Therefore, the extraction process of the 1.6×1.6µm² ASPAT diode was not highly accurate and it was difficult to fit the measured and simulated 푆11 and 푆12 at all bias.

Moreover, for >100GHz applications, critical issues arise that limit the accurate extraction of intrinsic and extrinsic parameters. The interaction between the electromagnetic field and the diode or/and pads starts to dominate at high-frequency, adding other losses, which lead to deteriorating device performance. Another concern is the increase of spreading resistance with the frequency due to a decrease in the bottom ohmic conductivity as the frequency increases [191, 192]. However, advanced technology needs to be exploited with a sub-micron resolution to further reduce the spreading distance between the top and bottom electrode (퐷푠푝푟) and compensate for the increase of the series resistance with frequency. The junction resistance, on the other hand, showed a

116

decreased voltage-dependent behaviour. The junction resistance and curvature coefficient of the ASPAT diodes are discussed in details in section 3.9.

3.8 InGaAs/AlAs ASPAT Diodes

This work reports a novel In0.53Ga0.47As/AlAs ASPAT which was grown on a semi- insulating InP substrate of 620µm thickness. The sample was denoted as (XMBE#326), and its epi-layer structure is depicted in table 3.4.

TABLE 3. 4: EPITAXIAL LAYER STRUCTURE OF ASPAT SAMPLE XMBE#326

Doping Thickness Layer Material Bandgap (eV) (cm-3) (Å) In Ga As 0.53 0.47 1.5×1019 3000 0.75 Top Ohmic (Si) In Ga As Emitter 0.53 0.47 1×1017 350 0.75 (Si)

Spacer In0.53Ga0.47As Undoped 50 0.75 Barrier AlAs Undoped 28 2.83

Spacer In0.53Ga0.47As Undoped 2000 0.75 In Ga As Collector 0.53 0.47 1×1017 350 0.75 (Si) In Ga As Bottom Ohmic 0.53 0.47 1.5×1019 4500 0.75 (Si) Substrate InP 620µm

A non-alloy Pd/Ti/Pd/Au was used to form the anode and cathode contacts of the devices. The highly doped bottom and top InGaAs layers (1.5×1019 cm-3, sample XMBE#326) reduces the depletion region between the metal and the semiconductor and good ohmic contacts are formed. Thus, thermal annealing is not required. The series resistances 푅푆 were theoretically‎ calculated‎ and‎ found‎ to‎ be‎ ~3.2,‎ ~2,‎ and‎~1Ω of the 4×4µm2, 6×6µm2, and 10×10µm2 ASPAT diodes respectively. The narrow bandgap

(0.75eV) of the In0.53Ga0.47As material introduces a higher bandgap discontinuity and hence larger junction resistance compared to the GaAs/AlAs structure. The higher barrier

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reduces thermionic emission transport and improves the temperature-independence of the ASPAT. The designed mesa area sizes of the devices in the mask were 4×4µm² and 6×6µm², and 10×10µm² respectively. Figure 3.17 shows the measured current densities of the In0.53Ga0.47As/AlAs ASPAT diodes. Considering mesa undercut profile, the optimised mesa area sizes were 3.75×3.75µm² and 5.85×5.85µm², and 10×10µm² respectively. The actual mesa area sizes showed comparable current densities of the devices from -1.5 to 1.5V bias. The In0.53Ga0.47As/AlAs ASPAT diodes demonstrated a minimum leakage current density of 0.0008mA/µm² at -1.5V bias, compared to 0.001mA/µm² recorded for the GaAs/AlAs ASPAT diodes. The figure also shows a strong non-linear region at a very low voltage close to zero.

10-1 3.75x3.75m 5.85x5.85m -2

10 10x10m

) 2 -3

m 10

mA/ ( 10-4

10-5 Current density Current 10-6

10-7 -1.5 -1.0 -0.5 0.0 0.5 1.0 1.5 Voltage (V)

Figure ‎3.17: Measured current density of the fabricated In0.53Ga0.47As/AlAs ASPAT diodes.

The electrical conductance was calculated from the measured I-V characteristics of the

In0.53Ga0.47As/AlAs ASPAT diodes and compared with the GaAs/AlAs ones as presented in figure 3.18. Note that the conductance of 3.7×3.7µm² and 5.8×5.8µm² GaAs/AlAs ASPAT diodes were normalised by a factor of (3.75/3.7) and (5.85/5.8), respectively so that any effect of mesa area difference on conductance at high negative bias is removed. At low reverse bias, the conductance is comparable for both samples as most of the

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voltage is dropped across the larger spacer of the devices. The difference becomes more pronounced at higher bias due to the barrier height difference between the samples. When the ASPAT diode is biased under high reverse voltages, the electrons have an increased probability of thermionic emission. The In0.53Ga0.47As/AlAs diode has the advantage of a higher barrier which forces the electrons to tunnel through the barrier, thus making tunnelling the dominant transport mechanism and results in a much smaller conductance than GaAs/AlAs ASPAT diode under a large reverse bias. S-parameter measurement and small-signal equivalent circuit extraction were carried out to investigate the high-frequency performances of the In0.53Ga0.47As/AlAs ASPAT diodes.

GaAs/AlAs_3.73.7m2 140 InGaAs/AlAs_3.753.75m2 GaAs/AlAs_5.85.8m2 120 InGaAs/AlAs_5.855.85m2 GaAs/AlAs_10x10m2 100 2

) InGaAs/AlAs_10x10m

S

 ( 80

60 Conductance 40

20

-1.50 -1.25 -1.00 -0.75 -0.50 Voltage (V)

Figure ‎3.18: Calculated conductance of GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes.

The extracted intrinsic parameters of the devices at zero-bias are listed in table 3.5. The small spreading resistance, as well as the small resistance of the highly doped layers of the 3.75×3.75µm² In0.53Ga0.47As/AlAs ASPAT diode, resulted in a very small (푅푆) compared to the GaAs one (3.5Ω vs‎ 11Ω).‎ However,‎ the‎ large‎ resistance‎ of‎ the‎ undepleted region (푅푢) limits its cut-off frequency to ~0.5THz at zero bias voltage. The reduction of 푅푢 would enhance the cut-off frequency and allow the integration of the

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3.75×3.75µm² In0.53Ga0.47As/AlAs ASPAT diode into imaging system application at >250GHz frequency.

TABLE 3. 5: EXTRACTED PARAMETERS OF THE In0.53Ga0.47As/AlAs ASPAT DIODES AT ZERO-BIAS.

Parameter 3.75×3.75µm² 5.85×5.85µm² 10×10µm²

퐶퐽, fF 18.5 48 158

푅푆,‎Ω 3.5 2 1

푅푢,‎Ω 14 13 4.8

푅퐽,‎kΩ ~150 ~60 ~20

풇풄풖풕−풐풇풇, GHz ~500 ~220 ~173

3.9 Extracted Junction Resistance and Curvature Coefficient of ASPAT Diodes

The nonlinear characteristics involving the junction resistance and curvature coefficient were computed from the measured I-V characteristics of GaAs/AlAs and

In0.53Ga0.47As/AlAs ASPAT diodes. This step is crucial to assess the non-linear performance of the devices at different bias. In the analysis presented here, it is assumed that the video resistance (푅푉) is equal to the junction resistance, as 푅퐽>>푅푆. The junction resistance as a function of the bias of the GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes exploited in this work are presented in figure 3.19 (a) and 3.20 (a). The plots show a well-matched correlation between the calculated 푅퐽 from the measured I-V characteristics (lines) and the extracted ones (dots) from the measured S-parameter using the equivalent circuit model technique. The key factor in detector design is the voltage sensitivity and noise equivalent power and this normally depends on the junction resistance of the diode (a large value of 푅퐽 provides high voltage sensitivity).

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104 (a)

103

)

k (

102

1 10 1.6x1.6m2 2.4x2.4m2 2 0 Junction resistance Junction 10 3.7x3.7m 5.8x5.8m2 10x10m2 10-1 -0.50 -0.25 0.00 0.25 0.50 Voltage(V)

20 1.6x1.6m2 (b) 2.4x2.4m2 15 3.7x3.7m2 2

) 5.8x5.8m -1

V 2 ( 10x10m

10

5

0 Curvature Coefficient Coefficient Curvature

-5 -0.50 -0.25 0.00 0.25 0.50 Voltage(V)

Figure ‎3.19: Calculated junction resistance and curvature coefficient of the 28.3Å barrier thickness GaAs/AlAs ASPAT diodes. The dots in (a) are the extracted junction resistance using the equivalent circuit model.

121

103

(a) )

 2

k 10 (

101

3.75x3.75m2 Junction Resistance Junction 5.85x5.85m2 10x10m2 100 -0.50 -0.25 0.00 0.25 0.50 Voltage(V)

20 3.75x3.75µm² (b) 5.85x5.85m2

15 10x10µm²

)

-1

V (

10

5

0 Curvature Coefficient Coefficient Curvature

-5 -0.50 -0.25 0.00 0.25 0.50 Voltage(V)

Figure ‎3.20: Calculated junction resistance and curvature coefficient of the 28.3Å barrier thickness

In0.53Ga0.47As/AlAs ASPAT diodes. The dots in (a) are the extracted junction resistance using the equivalent circuit model.

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푅퐽 is a voltage-dependent parameter, and so it basically decreases with forward bias. It is also inversely proportional to the mesa area size of the ASPAT diode. Smaller mesa area ASPATs (1.6×1.6µm2, and 2.4×2.4µm2) have higher junction resistances exceeding 200kΩ at zero-bias. A high junction resistance makes the design of the matching circuit more complex and challenging to implement. Furthermore, the noise equivalent power increases with the junction resistance and limits the maximum tangential sensitivity of the detector. At higher bias (~0.5V), the 3.7×3.7µm² GaAs/AlAs ASPAT diodes displayed a smaller junction resistance compared to the 3.75×3.75µm²

In0.53Ga0.47As/AlAs one (4kΩ‎ vs 12kΩ). This is mainly due to the contribution of the high thermionic emission transport over the low effective barrier height of the GaAs/AlAs ASPAT diodes. The junction resistance can be significantly reduced by biasing the diode at larger forward bias. However, this is not practical, as the depletion region becomes narrower with forward bias causing an increase in the junction capacitance and resistance of undepleted regions. More importantly, the curvature coefficient decreases under forward bias as clearly seen in figure 3.19 (b) and 3.20 (b). Consequently, the ASPAT diode voltage sensitivity is reduced. The curvature coefficients were also calculated and found to be approximately constant at zero-bias regardless of the mesa area size of the device. The calculated curvature coefficients of the -1 -1 GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes were ~18V and ~15V at zero-bias, which would theoretically provide a low-frequency unmatched sensitivity

(푆푉−푢푛푚푎푡푐ℎ푒푑) of 1800V/W and 1500V/W, respectively. The AlAs barrier thickness is another compelling factor to be examined. Therefore, we conducted an experimental investigation of the non-linear characteristics of the ASPAT diodes of different barrier thicknesses. GaAs/AlAs ASPAT structures were grown with a barrier thickness of 25.89Å, 19.5Å, and 13.1Å and then fabricated with a mesa area size of 3.7×3.7µm2, 5.8×5.8µm2, and 10×10µm2 respectively. The devices were DC characterised, and their junction resistances and curvature coefficients were extracted at zero-bias. The effect of thinning down the barrier thickness of the GaAs/AlAs ASPAT diodes is illustrated in figure 3.21.

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2 100 3.7x3.7m 18 5.8x5.8m2

2 ) )

80 10x10m -1 16

V

(

k ( 60

14

40 12

20

Junction Resistance Resistance Junction 10 Curvature Curvature Coefficient 0 8 12 15 18 21 24 27 30 12 15 18 21 24 27 30 Barrier Thickness (Å) Barrier Thickness (Å) (a) (b)

Figure ‎3.21: The junction resistance and curvature coefficient versus AlAs barrier thickness of the GaAs/AlAs ASPAT diodes at zero-bias.

As presented in figure 3.21 (a), there is almost an exponential relationship between junction resistance and the AlAs barrier thickness which directly comes from the exponential dependence of the transmission probability through a single barrier on the thickness of the barrier. A small junction resistance of ~3kΩ was obtained for the 3.7×3.7µm2 GaAs/AlAs ASPAT diode at zero-bias when the barrier thickness is ~13.1Å. A much smaller junction resistances of only ~1500Ω and‎~250Ω‎were computed for the 5.8×5.8µm2 and 10×10µm2 GaAs/AlAs ASPAT diodes at the same barrier thickness. The curvature coefficient was found to linearly decrease with the barrier thickness, as depicted in figure 3.21 (b). The variation in curvature coefficient with the barrier was found to be almost similar for the 3.7×3.7µm2, 5.8×5.8µm2, and 10×10µm2 respectively, demonstrating a weak dependence of the curvature coefficient on the mesa area size of the diode. Such dependency was reported previously with the backward tunnel diode in [24]. The curves indicate the optimum AlAs barrier thickness where the ASPAT diode provides a good curvature coefficient value with low junction resistance at zero-bias. To conclude, care has to be taken through the design of the epi-layer and mesa area size of the ASPAT diode in order to achieve, simultaneously, acceptable high-frequency performances and DC curvature coefficient. Furthermore, the validated Silvaco model built and developed in our group (K. N. Zainul Ariffin) in our previous work [125] was utilised to simulate the I-V characteristics of the 20Å AlAs barrier for 3×3µm2

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GaAs/AlAs ASPAT diode. The finding emphasises that the new device has a lower 푅퐽 of ~60kΩ compared to 200kΩ of the standard (28.3Å) barrier ASPAT diode at zero-bias.

3.10 Summary

Two ASPAT diodes based on GaAs/AlAs and InGaAs/AlAs were described in this chapter. The DC and RF characteristics of the fabricated one and two-port structures of different mesa area sizes were successfully performed. The analysis and optimisation of high-frequency characteristics of the fabricated and optimised CPW structures were performed to extract the maximum operating frequency of our structures. The calculations of the intrinsic components presented earlier were in excellent agreement with the extracted values from the measurements. A very low measured current density was observed for the GaAs/AlAs and In0.53Ga0.47As/AlAs samples. The equivalent circuit models of the diodes were accurate enough to extract the intrinsic and extrinsic components of the diodes from the measured on-wafer S-parameter up to 40GHz. The extra resistance of the undepleted spacer layer was extracted at zero and forward bias. The highest cut-off frequency of ~770GHz at zero-bias was obtained using a 1.6×1.6µm² GaAs/AlAs ASPAT diode. Non-linear characteristics (junction resistance and curvature coefficient) of all ASPAT diodes were investigated at different bias. The ASPAT diodes offer a curvature coefficient of (15 to 18V-1) at zero-bias. Finally, the variation of the non-linear characteristics with the AlAs barrier thickness was investigated. The benefit of decreasing the barrier thickness comes from the fast exponential decrease in the junction resistance and a slight linear decrease in the curvature coefficient. To sum up, with some modification of barrier and spacer layer thicknesses of ASPAT diodes, it is possible to use such diodes in the integration of >100GHz RF detectors and mixers.

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CHAPTER 4: DESIGN, SIMULATION AND

FABRICATION OF COPLANAR WAVEGUIDE

ZERO-BIAS ASYMMETRICAL SPACER LAYER

TUNNEL DIODE DETECTORS AND MIXERS

4.1 Introduction

Chapters two and three discussed the design, fabrication and characterisation of the new type temperature-independent ASPAT tunnelling diode. The presence of the non-linear region at zero-bias suggests the use of the ASPAT diode in detection of electromagnetic waves. Therefore, the purpose of this part of the work was to explore and examine the ASPAT diode by utilising its exceptional characteristics and building an RF detector and 2nd sub-harmonic mixer circuits at a range of frequencies. The design methodology and simulation tool used to simulate the integrated circuits are described in details. Following these, the fabrication and measurement of zero-bias ASPAT detectors and comparison with the simulated models are reported. Throughout the work, Monolithic Microwave Integrated Circuit (MMIC) technology was exploited to design and implement the ASPAT detector circuits. The final section includes an investigation to evaluate the performance of the ASPAT detectors with bow-tie antennas for a range of application such as automotive radar and imaging applications at 77GHz and 250GHz, respectively.

MMIC technology offers high output performances and low loss compared to microwave integrated circuit (MIC) technology since all active and passive elements are integrated on the same platform. It is also a cost-effective approach allowing the integration of thousands of millimetre scale circuits on 4-inch wafers. The fabrication of the passive and active components of the MMIC circuit is well-controlled by the designed mask layers providing a high reproducibility process. Another critical point is that MMIC technology eliminates the need for solder to connect the discrete components and instead uses transmission lines. Transmission lines mitigate parasitic effects and allow the realisation of wide bandwidth and high-frequency compact circuits up to 100GHz. MMIC circuits based GaAs material have gained many attractions in the fabrication of high-frequency active devices with low loss passive elements. In MMICs, the introduced

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error cannot be investigated until the entire fabrication process is finished. Therefore, careful steps have to be followed to minimise the error resulting from the mismatch between the single components of the whole circuit [193]. S. Mao reported the first multiplier and mixer circuits fabricated using the MMIC technology in the late 1960 [194]. Since then, different MMIC circuits, including amplifiers, mixers, and detectors have been developed for both educational and industrial works. The significant advancement of electromagnetic software packages in the late 1990s has paved the way for the designers to model and fabricate MMIC circuits with high accuracy and output performance [193].

4.2 Electromagnetic Simulation Tools

The design and fabrication of high-frequency integrated circuits require the use of a practical and advanced electromagnetic simulation tool to fully understand the electric and magnetic field distribution around the transmission lines and substrate. In general, the simulation tool facilitates the build and optimisation of virtual prototypes that mimic the behaviour of the fabricated circuits and address the issues early on in the design process. Therefore, a high accuracy tool is vital to enhance the design performance and increases the efficiency of the real fabricated process [195]. The use of the electromagnetic simulation tools is necessary to take into account the effect of the parasitic and field coupling, leading to an accurate evaluation of the S-parameter data of different circuit topologies. Throughout this work, different passive components were designed and simulated, such as a matching circuit, Metal-Insulator-Metal (MIM) capacitor, and antennas. Keysight technology offers two powerful electromagnetic simulators namely Advanced Design System (ADS) and (EMpro). ADS and EMpro tools are embedded with a 3D electromagnetic simulation approach called the Finite Element Method (FEM). FEM is a full 3D simulator based on frequency domain solution technique and capable of solving the electric and magnetic fields of complex structures with high accuracy and speed at very high frequencies. The simulator is mainly dedicated to the design and simulation of S-parameters and far-field performance of antennas such as gain, directivity, efficiency, and radiation patterns.

The ADS momentum tool includes another powerful electromagnetic simulator which uses the momentum of method (MoM) to perform the solution of the electromagnetic

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field. MoM simulator provides a 2.5D electromagnetic simulation to calculate the S- parameters of planar passive circuits such as microstrip, slot lines, and coplanar waveguides. The MoM simulator works with two modes, the momentum microwave and momentum RF. The RF mode employs an approximated formula of the Green function to calculate the electromagnetic radiation and therefore can provide faster and stable simulations of complex structures. RF mode can be applied for a structure with a size smaller than half of the wavelength (at a maximum frequency). For an electrically small structure, the RF mode offers high accuracy of simulation for frequencies smaller than (퐷/150), where D is the length of the structure in the millimetre scale. The momentum microwave mode is a full-wave electromagnetic simulation and is usually used for higher frequency simulation of radiating structures. It also considers the propagation of the surface wave in contrast to the RF mode and thus provides accurate simulation for all circuit sizes [196]. A surface wave exists at the interface of two different medium and decays exponentially with the distance from the interface point. Moreover, microwave mode allows the simulation and evaluation of the far-field pattern of antennas as the FEM simulator does. It is imperative to set the right mesh settings to ensure the accurate calculation of the current and coupling impact across the whole structure.

Another 3D electromagnetic tool used in the design and evaluation of antenna performances is the CST studio. CST is a powerful 3D full-wave solver that uses time and frequency domain techniques to solve the electromagnetic field of very complex structures. The frequency-domain technique is based on FEM and uses a tetrahedral mesh type to perform the electromagnetic analysis. Time domain, on the other hand, employs the Finite Integration Technique (FIT) and transmission line matrix (TLMx) method. In the time domain analysis, a series of pulses are transmitted on each hexahedral mesh , which represents a small volume in space. The solver then combines the transmitted and reflected pulses of all nodes and computes the electric and magnetic fields. It is noteworthy to state here, that all proposed antennas in this work were designed and verified using the FEM and time-domain CST simulators. The use of different simulators to validate the structures reduces errors and production cost.

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4.3 Coplanar Waveguide Structure

There are two main approaches namely microstrip and coplanar waveguide designs used to connect the passive and active components and form the integrated RF circuits. The microstrip design is considered as the simplest way to design and fabricate RF circuits at the microwave frequencies below 30GHz. The widely used coplanar waveguide technology has shown efficient performances and low losses at frequencies up to ~100GHz. The technology has low dispersion, low substrate sensitivity and allows for easy parallel and series connections of the elements on the same surface eliminating the use of via holes as in the case of the microstrip technique [197]. Figure 4.1 depicts the 3D schematic design of a CPW mounted on a finite substrate.

Figure ‎4.1: A 3D schematic view of a CPW structure on a semi-insulating substrate.

푡푚푒푡푎푙 and 푡푠푢푏 are the thicknesses of the metal and substrate respectively. w and g is the signal and ground plane widths, 푠 is the separated distance between the signal line and the ground plane. Finally, 푙 is the length of the coplanar waveguide structure. The characteristic impedance of the CPW structure is constrained by the dimension of the signal and ground conductors and more importantly, the separation distance between them, s [198]. In [199] a technique called Matched Asymptotic with the help of closed- form expressions was used to derive the characteristic impedance (푍0) of the coplanar waveguide structure. The derivation was made by assuming a finite thickness of the substrate and a non-perfect metal was used. The approximated (푍0) is given by:

30휋 퐾(푘′) 푍0= (4.1) √휀푒푓푓 퐾(푘)

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퐾(푘) is the elliptic integral of the first kind, 푘 is calculated from the dimension of CPW structure and given by:

푤 푘 = (4.2) 푤 + 2푠

Moreover, 푘′ is complementary of 푘 and given by:

푘′ = √1 − 푘2 (4.3)

휀 + 1 휀 = 푟 (4.4) 푒푓푓 2 where 휀푒푓푓 and 휀푟 are the effective and relative permittivities of the substrate. In a real CPW structure, the electromagnetic wave travels through the dielectric part of the substrate and air. Hence the wave is hosted by two mediums with different propagation and permittivity performances. Therefore, if equation 4.4 is used, the total average permittivity is used and accurate estimation of the characteristic impedance is thus achieved. In this work, the design and optimisation of the CPW matching circuits and transmission lines were carried out using the Linecalc tool embedded in ADS software. The tool also can be used to calculate the dimension of the microstrip and grounded CPW structure at a given frequency and characteristic impedance. The characteristic impedance of the CPW structure can be well controlled, and thus, a minimum loss is achieved if the optimum conditions are met. The next section addresses the limitation and design consideration of such structures.

4.4 Characteristic Impedance and Attenuation of CPW Structure

As mentioned before, the CPW structure is the most used technique for the mm-wave MMIC circuits. For a practical and low loss CPW, it is vital to understand how the CPW structure performs at a high frequency of operation. The attenuation of the CPW structure is mainly due to the introduced losses in the conductor lines and the dielectric part of the substrate and more importantly due to the radiation losses caused by the variation of the dielectric permittivity with frequency.

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4.4.1 Conductor Loss

Conductor loss is a frequency-dependent mechanism which depends strongly on the skin depth of the metal and the dimension of the CPW structure (w and s). The frequency dependent skin depth factor is given by:

휌 δ = √ (4.5) 휋푓µ0µ푟

where 휌 is the resistivity of the metal, µ0 is the free space permeability and equal to −7 4π×10 Henry/meter, µ푟 is the relative permeability and equal to 1, 푓 is the lowest operating frequency. The metal thickness 푡푚푒푡푎푙 is usually made with a thickness that is three‎to‎five‎times‎the‎skin‎depth‎δ‎of the film which greatly minimises the attenuation inside the conductor [199]. Conductor loss increases with smaller signal width, w, and larger separation distance, s or vice versa. Experimental and theoretical works were carried out in [180] to investigate the effect of varying w and s on the conductor loss of the CPW structure. It was found that for 0.4 < 푘 < 1, the conductor loss is small and can be neglected at low-frequency regimes. Another important factor is mesa area contact losses due to the impact of eddy current, which is important at high frequencies [191]. The generation of eddy current in the mesa area contact comes from the time-varying magnetic field around the anode bridge. Eddy current loss increases proportionally with the square of the operating frequency 푓.

4.4.2 Dielectric and Radiation Losses

Dielectric loss in (dB/m) on the other hand, increases with the loss tangent (푡푎푛훿) of the substrate as expressed in the following equation:

푡푎푛훿 휀푟 휀푒푓푓 − 1 훼푑(푑퐵/푚) = 푎휋 (4.6) 휆0 √휀푒푓푓 휀푟 − 1

where a=8.86, 휆0 is the free space wavelength. GaAs and InP substrates have very low loss tangent of 0.0016 and 0.002 respectively, making them appropriate materials for the implementation of the mm-wave and the sub-mm-wave integrated circuits.

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The most critical loss mechanism is the radiation loss at higher frequencies caused by the CPW field overlap with surface wave modes. Well-controlled design steps reduce the loss to its minimum level. The separation distance (푤 + 2푠) has a high impact on the attenuation of the guided waves inside the dielectric substrate as given by the following expression [200]:

1 (푤 + 2푠)2 3 (4.7) 훼푟 = 푓(휀푟)( ) ′ 휆푑 퐾(푘)퐾(푘 ) where 휆푑 is the dielectric wavelength and given by:

푐 휆푑 = (4.8) 푓√휀푟 where c is the speed of light in vacuum, and 푓(휀푟) is expressed by:

1 2 (1 − ) 휋 5 1 휀푟 푓(휀푟) = ( ) (4.9) 2 √2 1 √1 + 휀푟

For a 50Ω CPW structure sits on semi-insulator GaAs substrate with a 푘~0.5, equation 4.7 is reduced to the following form:

푤 + 2푠 2 훼푟 = 13.8( ) (4.10) 휆푑

The freedom of changing (푤 + 2푠) in CPW structure helps to mitigate the field interaction with the surface mode while keeping the impedance of the line constant. If (푤 + 2푠) is made to be much smaller than the substrate thickness and dielectric wavelength, a minimum radiation loss is thus obtained. The designed 50Ω CPW structures on a GaAs platform used in this work have shown as small attenuation as

~0.05dB, corresponding to (푤 + 2푠) of 120µm and 휆푑 of ~2mm at a wave frequency of 40GHz. Making the designed CPW structure a suitable technique for mm-wave MMIC integrated circuits. However, the effect of the surface wave and dispersion has to be taken into account when the dielectric wavelength is comparable to (푤 + 2푠) at high- frequency regimes. 3D simulators such as CST studio and EMpro from Keysight as well

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as the momentum microwave simulation account for such losses as the effect of the surface wave mode is considered in their calculations.

4.5 MMIC Metal-Insulator-Metal Capacitor

The capacitor is a passive component used in microwave and RF circuit in the form of passing or blocking signals. The mechanism relies on capacitance value and frequency, which both determine the impedance of the capacitor. In the literature, two simple approaches have been successfully employed to design and fabricate capacitors for RF integrated circuits; interdigital and MIM capacitors. The top view of the interdigital capacitor in figure 4.2 shows the metal fingers and the separation between them, which is usually a few microns. Its advantage comes from the fabrication simplicity and insensitivity to process variations [175, 201]. However, due to the fingers size and separation distance limitations, interdigital capacitor values are low and limited to a maximum value of ~1pF. Such capacitance cannot be exploited in the integration of mm- wave detectors as the required capacitance has to be > 1pF to suppress the RF signal and the generated harmonic tones. Moreover, the large size makes it incompatible for implementation in small RF circuits [193]. Furthermore, a small separation distance leads to more coupling effect between fingers in which parasitic elements are introduced in the MMIC integrated circuits.

Figure ‎4.2: Layout representation of nine-fingers interdigital capacitor [193].

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MIM capacitors, on the other hand, have been fabricated with a capacitance value ranging from 50fF to 200pF. A MIM capacitor is formed of a dielectric layer sandwiched between two metal plates, as shown in figure 4.3. The capacitance value is calculated from the thickness and permittivity of the dielectric layer as well as the active overlapped area between plates using the expression (퐶 = 휀0휀푟푤푙/푑).

Figure ‎4.3: 3D view of the MIM capacitor.

There are wide varieties of materials used as insulator layers such as benzocyclobutene

(BCB), SiO2, and Si3N4 which have permittivities of 2.7, 3.9, and 7.5 respectively were reported in the literature. The latter has been widely used compared to other materials as it has a high breakdown voltage (exceeding 65V) and low dielectric loss [175]. In this work, MIM capacitors with a 200nm Si3N4 dielectric thickness were designed and integrated with the ASPAT detector circuits. Figure 4.4 (a) shows the discrete rectangular layout structure of a 10pF capacitor of an active overlapping metal area of 180×180µm². In the literature, the equivalent circuit of the CPW MIM capacitor was demonstrated, showing the associated parasitic elements with the main capacitor as depicted in figure 4.4 (b) [202, 203].

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(a)

CMIM La Ra Lb Rb

G Cb Ca Dielectric losses

(b)

Figure ‎4.4: (a): 10pF CPW MIM capacitor used in this work (b): Equivalent circuit model of MIM capacitor [202].

푅푎 , 푅푏, 퐿푎, 퐿푏, 퐶푎, 퐶푏 are the parasitic resistance, inductance, and capacitance elements respectively introduced by the top and bottom plates. The conductance loss introduced by the dielectric is given by [202]:

퐺 = 2휋푓퐶푀퐼푀푡푎푛훿 (4.11) where 퐺 is a frequency dependent conductance that represents the increase of the leakage current through the dielectric as the frequency increases. The discrete capacitance was firstly simulated using the momentum (MoM) microwave simulation tool to verify the design and to optimise the dimensions for minimum parasitic losses. It can be said that the MoM simulation tool gives a good prediction of the capacitor behaviour (black line)

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over the frequency range 40MHz to 40GHz, as shown in figure 4.5. The small difference (~10%) between the measured and MoM simulation results could be attributed to the losses of the measurement cables and GSG input port. Using the equivalent circuit model, the main capacitance value was found to be ~8pF instead of 10pF. Figure 4.5 shows a good fit between the measured (red line) and simulated (blue line) S-parameters extracted from the equivalent circuit models. The conductance loss (퐺) was neglected as 푡푎푛훿 of the dielectric used is relatively small (0.0003). The fitting process also yields the following parasitic components; 푅푎 , 푅푏=2.66Ω, 퐿푎, 퐿푏=17pH, 퐶푎, 퐶푏=22.5fF. The extracted parasitic resistances of our MIM capacitors are close enough to the reported values in [203].

5 Measured Measured 0 Equivalent Circuit Equivalent Circuit MoM MoM -5 0

-10

(dB)

(dB) 11

-15 12

S S -5 -20

-25 -10 10 20 30 40 10 20 30 40 Frequency (GHz) Frequency (GHz)

Figure ‎4.5: Measured, equivalent circuit, and MoM S-parameters results of 10pF CPW MIM capacitor.

4.6 Matching Networks

Matching circuits are necessary elements for detector and mixer circuits to eliminate the mismatch between standard 50Ω‎source‎and‎diode‎impedance.‎Proper matching means more power will be delivered to the diode and thereby improving detector or mixer performances. The matching circuit also mitigates the leakage power from one port to another in mixer circuits. Lumped elements have been extensively employed in matching networks up to 1GHz [69]. For mm-wave and sub-mm-wave applications, a matching

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circuit is usually implemented using two stubs network. These stubs can be either open or short type, as shown in figure 4.6 [66].

Input Output

Transmission line Stub

Short Stub Open Stub

Figure ‎4.6: Matching circuit using open and short stubs [66].

Open stubs are preferred in the case of microstrip line circuit, unlike the short that is used in coplanar waveguide technology due to the ease of implementation. The matching circuit works as a kind of narrowband filter. In a single input port circuit (such as detector), matching circuit implementation is straightforward and requires few steps to optimise its performances. In mixers, more steps and iteration processes are required to achieve the minimum reflections.

In most cases, the designed matching circuit cannot provide proper matching at all ports and frequencies. Since the conversion occurs for the RF signal, it is essential to pay more attention to the RF side matching circuit. However, using stubs for termination purpose affect the impedance seen by the sources on both sides. The MMIC CPW open and short stubs are commonly realised as depicted in figure 4.7.

w s

Figure ‎4.7: Open and short stubs using CPW transmission lines [204]. 137

The open and short stubs can be viewed as resonator circuits composed of LCR components in series and parallel connections, respectively. The length 푙 of the open and short stubs can be theoretically estimated using the following equation [198, 205]:

푐 푙 = − 푙푒푥푡 (4.12) 4푓푟√휀푒푓푓

where 푐 is the speed of light, 푓푟 is the resonant frequency. One important limitation in the design of the CPW stub is the parasitic extension length (푙푒푥푡). 푙푒푥푡−표푝푒푛 is used for the open CPW stub which describes the impact of parasitic fringing capacitance (퐶푓푟𝑖푛푔𝑖푛푔) due to the fringing field effect at the gap distance a. 푙푒푥푡−표푝푒푛 is defined as the ratio between the fringing capacitance (퐶푓푟𝑖푛푔𝑖푛푔) to the main capacitance of the CPW open stub (퐶푠푡푢푏). It was indicated in [204] that 퐶푓푟𝑖푛푔𝑖푛푔 can be substantially reduced and thus

푙푒푥푡−표푝푒푛 by making the separation distance a equal or larger than (푤 + 2푠). In this case,

푙푒푥푡−표푝푒푛 can be approximately calculated from the dimension of the CPW using the expression [204]:

푙푒푥푡−표푝푒푛 = 0.25(w + 2s) (4.13)

On the other hand, 푙푒푥푡−푠ℎ표푟푡 of the short CPW stub is given by the ratio of end inductance (퐿0) to the inductance of the line stub (퐿푠푡푢푏). 퐿0 comes from current circulation in the discontinuity region [206]. The study conducted in [206] investigated the performances of short CPW stubs, where it experimentally proved the strong dependence of 푙푒푥푡−푠ℎ표푟푡 on the metal thickness 푡푚푒푡푎푙 of the pads. If the metal thickness was in the range 1 to 3µm, the impact of 푙푒푥푡−푠ℎ표푟푡 is small and can be neglected. With this in mind, 푙푒푥푡−푠ℎ표푟푡 can be found using the approximated formula [206]:

푙푒푥푡−푠ℎ표푟푡 = 0.125(w + 2s) (4.14)

Generally speaking, short stubs offer better performances since they have less parasitic effects compared to open stubs.

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4.7 Modelling of ASPAT I-V Characteristics

The cornerstone of the detection circuit is the non-linear element, and therefore, much care must be paid to build a virtual model which accurately behaves like the real fabricated device. Due to the lack of a built-in tunnel diode model in the Keysight ADS library, a polynomial representation method was used in this work to model and implement the measured non-linear characteristics of the ASPAT diodes. The method has been widely used by other researches to model the non-linear characteristics of RTD and transistor devices [66, 207]. The Symbolically Defined Devices (SDD) model embedded in ADS was successfully employed to model the non-linear I-V characteristics of the ASPAT diodes which can then be used in the non-linear simulation of the high-frequency detector and mixer circuits. The SDD model is considered as an alternative approach to the built-in model and was successfully validated by many researchers [66, 207]. The generation of the polynomial equation was performed using the Matlab software. Due to the small measured current at bias around the zero-operation region, the measured I-V curve was segmented into small regions, and then the polynomial equation was developed for each part. This way ensures a highly accurate modelling of the DC characteristics, and hence, an optimum fitting with the measured data at all bias is achieved. Once the optimum polynomial equation was found, the coefficients were imported to the SDD block, and the model was created. Figure 4.8 (a) depicts the schematic representation of the two-port SDD model used in this work. The measured I- V characteristics involve the effect of the series resistance in addition to the non-linear junction resistance. For real circuit simulation, resistance in series was added which accounts for the total series composed of (푅푆+푅푢) to accurately model the input impedance of the ASPAT diode. An example of the measured and fitted I-V curves is presented in figure 4.8 (b).

SDD model demonstrated a well-matched non-linear transition at zero-bias, which is the key region for zero-bias circuits. Accurate representation can be well-achieved by generating a high order polynomial equation for each part of the I-V curve. However, a challenging issue occurs as the mesa area decreases due to the limitation of the polynomial equation to model the small current (nA ranges) of the diodes. Any variation between the measured and modelled I-V characteristics leads to having different junction resistance and curvature coefficient, which results in a bad prediction of the whole integrated circuit performance.

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(a)

Measured 10-1 SDD fit

10-2

10-3

10-4 Current (mA) Current

10-5

10-6 -1.0 -0.5 0.0 0.5 1.0 Voltage (V) (b) Figure ‎4.8: (a): Two-port SDD model circuit in ADS tool, (b): Measured and fitted curves of the 3.7×3.7µm² GaAs/AlAs ASPAT diode.

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4.8 Schematic Design and Simulation of Detectors and Mixers using ADS Tool

The schematic design tool represents the first platform to start the design of MMIC integrated circuits. The work conducted in this thesis was mainly performed using the ADS tool provided by Keysight technology. The tool includes a set of built-in models of various diodes and transistors. In this work, schematic and layout designs were used to model the non-linear characteristics of the ASPAT diodes, single passive elements, as well as the whole integrated circuits. The schematic ADS design is a powerful tool which offers a wide variety of linear and non-linear time and frequency domain simulators. For example, the harmonic balance simulator is used to simulate the non-linear characteristics of the direct and heterodyne detection systems such as the 1-dB compression and third-order intercept point. Once a good SDD model was found, the next step was the design process of the matching circuit. The aim was to use a coplanar waveguide technology to realise the matching circuit and the MMIC integrated circuits. The design step of the matching circuit is an iterative process and started by calculating the diode impedance seen by the input ports.

At some power level, the diode impedance is either too large or too small, which makes the design of the matching circuit extremely difficult. In this work, we mainly focused on the GaAs/AlAs ASPAT as a promising device for commercial zero-bias detectors at X- band and K-band frequencies. In the beginning, RF signals were applied, and diode impedances were obtained at the desired power and frequency RF signal. Larger mesa area size ASPAT diodes (5.8×5.8µm² and 10×10µm²) were fed by a 10GHz RF signal, while a 24GHz RF signal was applied to the 3.7×3.7µm² ASPAT diode. Impedances were found to be ~24 − 푗316Ω, ~14 − j296Ω, and ~7.7 − j115Ω for the 3.7×3.7µm², 5.8×5.8µm² and 10×10µm² mesa area sizes respectively. The imaginary part of the impedance can be theoretically estimated using the expression 푗/휔퐶퐽. Thereafter, the ‘smith‎chart‎tool‘ embedded in ADS was exploited to build the matching circuit of every single diode. The tool uses lumped elements and stubs to realise the matching network. Ideal short stubs were used to form the CPW matching circuits as they have less parasitic effects as discussed earlier. An example of the schematic design of the zero-bias ASPAT detector using ideal transmission line stubs is depicted in figure 4.9 (a). The output DC voltage is taken across the output capacitor (10pF). Similar steps were carried out to build the matching circuits of the subharmonic mixers at different frequencies of the 141

input RF signal. In 2nd subharmonic mixer circuits, as shown in figure 4.9 (b), matching circuits were designed and optimised using three steps.

(a)

Anti-parallel ASPAT diodes RF LO side side

IF side

(b)

Figure ‎4.9: (a): Zero-bias direct detection circuit based 5.8×5.8µm² GaAs/AlAs ASPAT diode, (b): Zero-bias 2nd subharmonic mixer based 3.7×3.7µm² GaAs/AlAs ASPAT diode.

Firstly, the impedance was calculated at the RF side, and a matching circuit was built at the desired RF power and frequency signal. Secondly, the impedance at the LO side was

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evaluated, which includes the sum of anti-parallel ASPAT diodes and RF matching impedances. Then a proper matching circuit was designed and inserted at the LO side. At this point, the RF side sees different impedance value due to the inclusion of LO matching circuit. Therefore, the last step was to repeat the design of the matching network at the RF side as the RF signal has lower power compared to the LO signal, so it is crucial to make sure that the RF side is not affected by the matching circuit at the LO side.

Testing and optimising the ideal transmission line detector and mixer circuits were carried out to minimise the reflections further and improve the performances as much as possible. The primary simulation tool used was the well-known Harmonic Balance (HB) block. The tool provides practical calculations of the mixing process of the main signals as well as their high order harmonic components. It is noteworthy to state here, that an appropriate setting of the HB block plays a vital role in calculating the desired output current and voltage. An accurate simulation of mixers depends on the order of the harmonic component of the RF and LO signals set in the (HB) block. In our simulation, high order harmonic components were used with the LO signal since it has higher power compared to RF signal and so it is assumed to have a significant influence on the output frequency spectrum. The polynomial equation used in the SDD model is limited to a range of voltages, and thus, applying a signal with relatively high power could lead to unexpected behaviour of the non-linear characteristics of the ASPAT model. Bandpass filters were used to decrease the level of harmonics that surround the RF and LO signals and improve the isolation between ports. The filters were located after the LO and RF sources and before the matching circuits.

4.9 Mask Layout of the MMIC Integrated Zero-Bias ASPAT Detectors

The key next step was to design and optimise the CPW matching circuits and the circuit layouts for the final fabrication process. CPW short stubs firstly replaced the ideal transmission lines.‎ CPW‎ dimensions‎ were‎ calculated‎ using‎ the‎ ‘‘LineCalc’’‎ tool‎ embedded in the ADS software. The tool offers a wide range of analytical models to calculate the physical model and electrical parameters of different transmission technology such as microstrip, CPW, and CPWG structures. Tuning and optimisation of transmission lines were performed, and then the total behaviour of the CPW stubs was

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compared with the ideal one. An example of an ideal and CPW short stub matching circuits of 5.8×5.8µm2 ASPAT detector is shown in figure 4.10.

Ideal stubs CPW stubs 0

-10

(dB)

11

S -20

-30

-40 8 9 10 11 12 Frequency (GHz)

Figure ‎4.10: Matching circuit response of ideal and CPW stubs.

Both responses have a very narrow bandwidth of ~0.2GHz. The CPW stubs behaviour was as close as possible to the ideal stubs with only ~0.3GHz frequency difference. The long length of the designed matching circuits are (2.5mm) and (6.5mm), which makes them not practical for compact RF circuits at X-band and mm-wave frequency applications. For this reason, all matching circuits were optimised to have a maximum size of 2×2mm². Following that, all matching circuits were transferred from the schematic platform to the layout representation. As there is no CPW T-junction model embedded in ADS library, the T-junction was optimised through an iterative process to obtain the desired behaviour. Matching circuits were designed with a GSG input port configuration and conductor width of 50µm.

The separation between the conductor and the grounds was made to be 35µm. The designed matching circuits can be grouped into two parts: X-band and K-band CPW networks. The moderate junction resistance (35 to 50kΩ) and high cut-off frequency (~245GHz) of the 5.8×5.8µm2 ASPAT made it an excellent choice for the X-band and K- band frequencies with a small matching circuit size. Four matching circuits were

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designed and optimised on a 625µm semi-insulating substrate using the momentum simulation microwave mode. The matching circuit was exported from the layout design to the schematic design for the final evaluation of the ASPAT detectors performances. The circuit shown in figure 4.11 was used to evaluate the total performance of the ASPAT detector with its matching circuit and MIM capacitor layout designs. Once the maximum performances were achieved, the designs of the mask layouts of the whole integrated zero-bias ASPAT detectors were carried out for the final fabrication.

Figure ‎4.11: Final zero-bias ASPAT detector circuit implementation showing the layout design of the matching circuit and MIM capacitor.

The mask design steps are explained in figure 4.12. Step 1, 2, and 3 shows the definition of the top and bottom contacts layers as well as the isolation layer to cover the active layers and the TLM structures. Step 4 defines the Si3N4 layer‎and‎the‎required‎via’s of the top contact and MIM capacitor top contact. Step 5 adds the matching circuit and bottom contact of the MIM capacitor. For this work, 퐷푠푝푟 was designed to be 1.5µm for all detector circuits. It is worthy to point out that the input and output ports are designed with‎50Ω‎standard‎characteristic‎impedance‎exploiting‎CPW‎technology.‎However,‎the‎ matching circuit topology is an optimised shorted stub simulated using ADS Momentum taking into account the impact of electromagnetic field coupling on the circuit performance. The length and width of the centre signal transmission line and stub provide the desired frequency matching band.

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Step 1, 2, and 3

Bottom Contact Isolation layer Isolation

Top m

µ 6 6µm

Step 4

Si3N4

Top Via

Via for top contact of MIM capacitor

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Step 5

Matching network MIM bottom Contact

MIM Top Contact

Strip line

Figure ‎4.12: An example of mask design steps of the MMIC zero-bias ASPAT detector of the mesa area size of 6×6µm².

4.10 Fabrication and Measurement of the MMIC Integrated Zero-Bias ASPAT Detectors

The fabrication of the ASPAT detector circuits took place in a class 1000 clean room. A wafer tile of size 15×15mm2 was fabricated into many ASPAT detectors. The fabrication processes of the detectors were accomplished as follow. Firstly, the definition of the top mesa area size was carried out by deposition of ~260nm of a metal stack of AuGe/Ni/Au to form the ASPAT diode top contact. Secondly, an etch down to the ohmic bottom

(GaAs) layer was performed using the etchant H3PO4:H2O2:H2O with a ratio of 3:1:50 followed by deposition of 260nm of AuGe/Ni/Au metal to form the bottom contact of the

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ASPAT diode and the MIM capacitor top contact. Thirdly, an isolation step is performed by covering the active layers and the TLM structures with photoresist coating and then etching down to the substrate. Fourthly, for the DC signal output voltage, a 200nm Si3N4 dielectric layer was sputtered on the whole wafer tile. The designed MIM capacitor whose size was 180×180µm² provided a 10pF value which is adequate to suppress the RF current at X-band frequencies and the generated frequency tones associated with the diode output current signal. The required vias for the top contact of the ASPAT and MIM capacitor were opened in a subsequent step. Finally, a Ti/Au (50nm/1400nm) metal deposition of the matching circuit and the bottom contact of the output capacitor were carried out. Figure 4.13 depicts the fabricated zero-bias ASPAT detectors. The RF signal transmission line is connected to the bottom ground through a short stub. This ensures that the ASPAT diode is completely isolated from any DC signal at the input side (to avoid any self-biasing issue). The capacitor is directly linked to the output GSG port for reading the generated DC output voltage. Figure 4.13 (a) and (b) depict the 3.8mm² and 2.78mm² X-band detectors based on 5.8×5.8µm² and 10×10µm² ASPAT diodes respectively. Figure 4.13 (c) and (d) on the other hand present the integrated K-band detectors of circuit size 0.78mm² and 0.544mm² based on 3.7×3.7µm² and 5.8×5.8µm² ASPAT devices respectively. The aim of this work was not only to maximise the detector performances but also to keep the circuit size smaller than 4mm² and 1mm² for the X- band and K-band frequencies, in which many circuits can be included in the mask layout. As a result, 76 ASPAT detectors were fabricated on the 15×15mm² wafer tile. Since the imaginary part of both diodes is relatively high at X-band frequencies, the matching circuit size was in the millimetre size scale.

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(a)

(b)

149

1mm 0.85m m

3.7×3.7µm² 5.8×5.8µm²

0.78m m

0.64m m

(c) (d)

Figure ‎4.13: Fabricated MMIC integrated zero-bias ASPAT detectors. (a) and (b) are the X-band detectors, (c) and (d) are the K-band detectors (Note: the images are not to scale).

The voltage sensitivities of the detectors were measured on-wafer and at room temperature. A circuit diagram showing the measurement setup is drawn in figure 4.14.

RF source Attenuator Bias tee

Zs

C L

Detector DVM

Figure ‎4.14: Circuit diagram for voltage sensitivity measurement configuration.

An Anritsu VNA (model 37369A) was used to inject the detectors with variable RF power and frequency signal through a GSG CPW probe, and no DC circuit was needed for biasing. Care was taken to ensure that the applied power is well-known during the measurements. The RF source had an internal bias Tee. The signal level power was adjusted using an attenuator. There could be some difference between the applied power on the device under test (DUT) and the power from the VNA due to the introduced losses

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from cables and probes. The output of the detector is connected to high input impedance Agilent DVM to measure the generated DC output voltage.

4.11 Measured and Simulated Un-matched Voltage Sensitivity of 6×6µm² GaAs/AlAs ASPAT Diode

This section describes the detection characteristics of 6×6µm² GaAs/AlAs ASPAT diode. The device was -bonded with discrete passive components and mounted in a quad- flat no-leads (QFN) detector circuit to test its sensitivity. Measurement was carried out at the Linwave Company in Lincoln, UK. The simplified equivalent circuit diagram of the QFN is presented in figure 4.15 (a) and a photograph of the actual circuit shown in figure 4.15 (b).

RF source QFN tuning circuit Vout

Zs 3.3pF ASPAT diode 10nH 0.3fF Vs cos(wt) 10pF

300ohm

(a)

ASPAT diode

3.2mm

3.2mm

(b) Figure ‎4.15: (a) Equivalent circuit diagram of the QFN detector, (b) Actual photograph of the discrete circuit.

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The circuit is simple and consists of a fixed R-L-C circuit placed between the input RF signal and the mounted ASPAT diode. The RLC circuit works as a tuning matching circuit. However, it has no matching purpose to the ASPAT diode. R, C, and L values were tuned to obtain the maximum output voltage. A fixed frequency signal of 9.5GHz was applied to the ASPAT diode. The power was swept from -35 to -5dBm, and accordingly, the output voltage was measured on the output capacitor (10pF). The circuit was simulated in the ADS tool, and its voltage sensitivity compared with the measured one, as shown in figure 4.16. The bond pad effect was represented in the circuit by including 퐶푃 and 퐿푃 for accurate simulation process. The simulated sensitivity is in excellent agreement with the measured one, and this validates the SDD model and the modelled circuit used to design and model the ASPAT detector. Moreover, it also verifies the extracted parameters of the ASPAT diode (퐶퐽,푅퐽, and 푅푆) which were used in the model. The 6×6µm² ASPAT device exhibited a voltage sensitivity of 950V/W at

9.5GHz and -30dBm RF power. The inset shows the measured video resistance (푅퐽 + 푅푆) as a function of the input RF power. The measured resistance is ~30kΩ at a low input RF power of -35dBm and gradually decreases to ~10kΩ‎at an RF power of 0dBm.

1000 @ 9.5GHz Measured Simulated

800

35

30

600 ) 

25

k (

20

15 400

Video Video Resistance 10

Un-matched Voltage Sensitivity (V/W) Sensitivity Voltage Un-matched 5 -30 -20 -10 0 Input RF power (dBm) 200 -30 -20 -10 0 Input RF power (dBm)

Figure ‎4.16: Measured and simulated un-matched voltage sensitivity of 6×6µm² ASPAT diode. The inset is the measured video resistance.

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To investigate the maximum voltage sensitivity of the zero-bias GaAs/AlAs ASPAT diodes; ideal transmission line matching circuits were designed, and the fully matched voltage sensitivities were simulated. These showed that the 3.7×3.7µm² and 5.8×5.8µm² GaAs/AlAs ASPAT diodes have a maximum voltage sensitivity of ~20000V/W and ~7500V/W respectively at 24GHz RF signal. Voltage sensitivity of ~9000V/W at 10GHz was obtained using the 10×10µm² ASPAT diode at zero-bias.

4.12 ASPAT Detectors Performances

This section reports the measured and simulated voltage sensitivity in addition to the calculated noise equivalent power of the fabricated integrated zero-bias ASPAT detectors. The main goal was to investigate the voltage sensitivity and verify the proposed detector models.

4.12.1 Measured DC Output Voltage

The output voltage (푉표푢푡) of the fabricated detectors were measured in the frequency bands [4 to 18GHz] and [10 to 35GHz] at RF power ranging from -27 to -4dBm. The measured 푉표푢푡 of ASPAT detectors versus frequency at -27dBm RF power are shown in figure 4.17. The figure shows good voltage uniformity over five ASPAT detector circuits at different locations on the wafer tile. It also validates the successful modelling and fabrication of the MMIC integrated ASPAT detectors. The 5.8×5.8µm² and 10×10µm² ASPAT detector circuits showed excellent output voltage over the bandwidth [5 to 6.5GHz] covering the X-band frequencies. At X-band frequencies, the 5.8×5.8µm² ASPAT diode provided a better output voltage of ~7.5mV at ~11GHz compared to ~2.7mV for the 10×10µm² ASPAT at 9.5GHz as depicted in figure 4.17 (a) and (b). The higher output voltage is due to smaller junction capacitance and better matching condition of the 5.8×5.8µm² ASPAT diode corresponding to a smaller reflection coefficient of -12dB at 11GHz as clearly seen in figure 4.17 (c). The high reflection coefficient of the 10×10µm² ASPAT arises from the large junction capacitance which gives a weak prediction process of the matching circuit behaviour. Undoubtedly, more optimisation of the matching circuit stubs is necessary to reduce the reflection coefficient

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of the detector and obtain higher output voltage. Figure 4.17 (d) and (e) show excellent bandwidths of ~9GHz covering the K-band frequencies [18 to 26.5GHz].

4-18 GHz 10-35 GHz 8 3 Circuit1 Circuit1 2 2 Circuit2 5.8x5.8m ASPAT Circuit2 3.7x3.7m ASPAT Circuit3 Circuit3 Circuit4 Circuit4 6 Circuit5 Circuit5 2

B.W=5GHz

4

(mV)

(mV) B.W=9GHz

out

out

V V 1 2

0 0 4 6 8 10 12 14 16 18 10 15 20 25 30 35 Frequency (GHz) Frequency (GHz) (a) (d)

3 3 Circuit1 Circuit1 2 Circuit2 10x10m ASPAT Circuit2 5.8x5.8m2 ASPAT Circuit3 Circuit3 Circuit4 Circuit4 Circuit5 Circuit5 2 2

B.W=6.5GHz

(mV)

(mV) out

out B.W=9.5GHz

V V 1 1

0 0 4 6 8 10 12 14 16 18 10 15 20 25 30 35 Frequency (GHz) Frequency (GHz) (b) (e)

0 0

-2 -5

-4 -10

-6

(dB)

(dB)

11

11 S

S -15 -8

2 2 -10 5.8x5.8m ASPAT -20 3.7x3.7m ASPAT

2 2 -12 10x10m ASPAT 5.8x5.8m ASPAT -25 4 6 8 10 12 14 16 18 10 15 20 25 30 35 Frequency (GHz) Frequency (GHz) (c) (f)

Figure ‎4.17: (a), (b), and (c) are the measured output DC voltage and reflection coefficients (푺ퟏퟏ) of the X-band zero-bias detectors based 5.8×5.8µm² and 10×10µm² GaAs/AlAs ASPAT diodes at - 27dBm RF power. (d), (e), and (f) are the measured output DC voltage and reflection coefficients (푺ퟏퟏ) of the K-band zero-bias detectors based 3.7×3.7µm² and 5.8×5.8µm² GaAs/AlAs ASPAT diodes at -27dBm RF power.

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The 3.7×3.7µm² and 5.8×5.8µm² ASPAT achieved an output voltage of ~2.5mV at ~24GHz. It is clear that both detectors have a good response with a reflection coefficient of < -20dB at the K-band frequencies in the case of the 5.8×5.8µm² ASPAT detector. The 5.8×5.8µm² ASPAT diode was found to be more suitable for the K-band frequencies due to its smaller junction resistance compared to the 3.7×3.7µm² ASPAT device. Accordingly, 5.8×5.8µm² ASPAT detectors have a better reflection coefficient response at the 24GHz frequency. In conclusion, the maximum output voltage of the detector is constrained by the matching circuit behaviour, junction resistance, and junction capacitance. More importantly, the parasitic effect of the matching circuit was not considered in our calculation and design process, which have a higher impact on the K- band frequencies.

4.12.2 Voltage Sensitivity and Noise Equivalent Power

Voltage sensitivities were then calculated by taking out the average of the measured DC voltage of different circuits at each input RF power. The noise equivalent power (NEP) was then computed from the measured sensitivity at a video bandwidth of 1Hz. Figure 4.18 (a) and (b) depict the high correlation between the measured and simulated voltage sensitivity as well as the calculated NEP of the X-band and K-band zero-bias detectors based 5.8×5.8µm² ASPAT diode. The excellent correlation validates the model used, allowing for prediction of performances and aid in further optimisation prior to costly manufacturing processes. The measured voltage sensitivities were ~1800 to 3650V/W and 700 to 1300V/W at the X-band and K-band frequencies, respectively. The maximum measured voltage sensitivity is ~3650V/W at 11GHz and -27dBm RF power which is comparable to the zero-bias Schottky detector in [208], where a voltage sensitivity of 1000V/W at ~10GHz is reported. The measured voltage sensitivity and junction resistance were then used to estimate the noise equivalent power using equation (2.20) provided in chapter two. The calculated NEP was below 14pW/√Hz at X-band frequencies with a minimum value of ~6pW/√Hz at 11GHz. On the other hand, the K- band ASPAT detector recorded a noise equivalent power below 35pW/√Hz with a minimum value of ~20pW/√Hz at 24GHz.

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20 @ -27dBm 4000 X-band 15 3000

10 2000

1000 Measured Sensitivity 5

Simulated Sensitivity NEP (pW/sqrt(Hz)) Calculated NEP Voltage(V/W) Sensitivity 0 0 8 9 10 11 12 13 Frequency (GHz) (a) 1500 50 @ -27dBm K-band 40 1000 30

20 500 Measured Sensitivity Simulated Sensitivity 10

Calculated NEP NEP(pW/sqrt(Hz))

Voltage(V/W) Sensitivity 0 0 18 20 22 24 26 28 Frequency (GHz) (b) 4000 Measured at 11GHz Simulated at 11GHz 3000 Measured at 24GHz Simulated at 24GHz

2000

1000 Voltage(V/W) Sensitivity -30 -25 -20 -15 -10 -5 0 Input RF power (dBm) (c)

Figure ‎4.18: (a) and (b) are the measured and simulated voltage sensitivity and calculated noise equivalent power of the X-band and K-band zero-bias detectors based 5.8×5.8µm² ASPAT diode, (c) is the measured and simulated voltage sensitivity versus input RF power.

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The zero-bias 5.8×5.8µm² ASPAT diode showed good voltage sensitivity and noise equivalent power at K-band compared to the biased Schottky detector reported in [209] despite the much larger mesa area size used in this work (33.6µm² vs 1.3µm²). The measured voltage sensitivity as a function of input RF power is depicted in figure 4.18 (c). The data demonstrate good linearity, and the detectors can deliver half of the maximum sensitivity even with a high RF power of -10dBm. The matching circuit was designed to operate at a relatively low input RF power of -27dBm; as a result, there is a small difference between the measured and simulated data at RF power higher than - 15dBm. The sensitivity decreases with increasing input power, indicating a transition from the non-linear region to the linear one at high input power. The zero-bias 10×10µm² GaAs/AlAs ASPAT detector was also shown to have good sensitivity and NEP of ~800 to 1347V/W and ~11pW/√Hz respectively at X-band and -27dBm RF power. Optimising the‎ transmission‎ line‎ stubs‎ would‎ improve‎ the‎ detector’s‎ performance, particularly at high frequencies. Besides that, the use of thinner AlAs barrier ASPAT diode with smaller junction resistance would also contribute to a further reduction in the NEP of the ASPAT detector but at the expense of the voltage sensitivity. The main feature of temperature insensitive operation still gives the ASPAT a tremendous advantage over various other reported structures. Further increase in voltage sensitivity can be obtained by using smaller feature sizes or/and high curvature coefficient for efficient mm-wave/THz frequency regime detector systems. The InGaAs ASPAT diode with thinner AlAs barrier could also be used for THz detection applications due to its small series resistance and high cut-off frequency.

It is noteworthy to mention here that work is ongoing in our group to design and fabricate mm-wave ASPAT detectors at 30GHz, 77GHz, and 90GHz. Different detectors have been designed with open stub matching networks and then fabricated on a 15×15mm² GaAs substrate. The preliminary results showed a well-matched data between the measured and simulated (푆11) of the 30GHz integrated ASPAT detector (see APPENDIX-D). The investigation is in progress to measure their output voltages and calculate their sensitivities and NEP and these are part of the further work to be undertaken as a result of this research.

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4.13 Millimeter-Wave ASPAT Detectors with Antennas

The experimental work described so far presents the rectification performances of the integrated ASPAT detector using a 50Ω source. In practical systems, the antenna represents the first element which collects the electromagnetic radiations and then injects them into the detector circuit for the extraction process. Therefore, it is imperative to model the ASPAT diode with integrated antenna and investigate the total behaviour. High gain antennas are highly prefered at mm/THz waves in order to collect the weak radiation and deliver it to the non-linear element. The horn antenna has the highest ever achieved gain and efficiency at mm-wave applications but at the expense of large area size [210]. In the literature [211-213], integrated detectors with different kind of antennas such as spiral and log-periodic shapes have been proposed showing a wide bandwidth, high directivity and high voltage sensitivity in mm-wave/THz frequency regions. Among all types, the bow-tie antenna is widely used in both emitter and detector circuits due to their simplicity, lightweight, wide bandwidth, and high gain. In [214], a Ka-band Schottky detector with a bow-tie antenna was fabricated on a 254µm duroid dielectric substrate. The measured bandwidth of the detector was ~8GHz with a maximum voltage sensitivity of 510V/W at 31.8GHz frequency. The pHEMT transistor in [215] was monolithically integrated with a wide bandwidth (~60GHz) bow-tie antenna. A hyperspherical lens was attached to the back of the substrate to focus the waves on the antenna. At 250GHz and under 0.3V bias, the maximum measured voltage sensitivity and calculated NEP were 220V/W and 25pW/√퐻푧 respectively. The work conducted in [216], introduced a multi-channel detector integrated with high gain and broadband bow tie antennas at 260 to 400GHz and employing a 0.785µm² Schottky diode. Under 150µA DC bias, the maximum measured sensitivity and NEP of a single detector were 220 to 330V/W and 60pW/√퐻푧 respectively. Another simple and high gain antenna is the Quasi-Yagi structure which shows good characterstics at microwave frequenies. The Quasi-Yagi antennas presented in [217, 218] are deposited on a thick and high dielectric constant substrate of 640µm and 휀푟=10.2, respectively. The antenna works at X-band with a bandwidth of 4.5GHz and gain of 3 to 7dB. In [219], a Quasi-Yagi antenna with Schottky detector and matching stubs were designed and optimised for a 24GHz frequency. The antenna sits on a 254µm duroid dielectric substrate and offers a bandwidth of 0.7GHz and directivity of 9dB. Different techniques have been proposed to integrate the antenna in transceiver circuits. The monolithic integration method on the

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chip is widely exploited by designers to mitigate the mismatch between the components and thus minimise errors.

This section mainly focuses on the design of bow-tie structures for complete zero-bias ASPAT detectors with monolithically integrated antennas. The antennas were designed and optimised for automotive car radar and high-resolution imaging applications at frequencies of 77GHz and 250GHz, respectively.

4.14 Antenna Design and Performances Evaluation

The bow-tie antennas were designed and simulated using the CST studio electromagnetic tool at 77GHz and 250GHz frequencies. The structures sit on a 100µm semi-insulating dielectric GaAs substrate. Figure 4.19 shows the 3D drawing of the proposed integrated ASPAT detector with bow-tie antenna.

Figure ‎4.19: A 3D structure of the proposed ASPAT detector with a bow-tie antenna. (Note: image is not to scale).

The arms of the bow-tie antenna are attached to the heavily doped top and bottom contact layers of the ASPAT diodes. The gap distance between the arms was ~1.5µm to ensure a small series resistance.

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The design of the antennas starts by defining the resonance frequency 푓푟 and calculating the dimensions of the structure using the following equations [220]:

1.6푐 퐿퐴푛푡 = (4.15) 푓푟√휀푟

0.5푐 푊퐴푛푡 = (4.16) 푓푟√휀푟

퐿 퐿 = 퐴푛푡 (4.17) 푠푢푏 0.85

푊 푊 = 퐴푛푡 (4.18) 푠푢푏 0.45

In the initial investigations, the bow-tie antenna was designed to work at 250GHz with a gold thickness of 1µm.

Accordingly, the calculated dimensions are 퐿퐴푛푡 = 534µ푚, 푊퐴푛푡 = 167µ푚, 퐿푠푢푏 =

628µ푚, and 푊푠푢푏 = 371µ푚. Indeed, equations (4.15 to 4.18) were used to design a bow-tie antenna with CPW input feeding port, as reported in [220] at ~300GHz resonance frequency. In the beginning, the proposed antennas were designed and simulated with only the bow-tie arms. Thereafter, transmission lines with output CPW and CPS pads were then designed and attached to the arms. The structures were simulated with different GaAs substrate thickness to achieve optimal radiation performances. Simulation processes have shown optimal characteristics when a substrate thickness of 100µm is used. It is noteworthy to state that the dimensions of the CPW pads were optimised through an optimisation process to achieve 50Ω‎ characteristic impedance compatible with the GSG probe, as shown in figure 4.20.

The inclusion of the transmission lines and the output pads changes the resonance frequency of the proposed structures. To maintain the 250GHz resonance frequency, the dimensions of the substrate were optimised to be 퐿푠푢푏 = 740µ푚 and 650µ푚 , 푊푠푢푏 = 640µ푚 and 700µ푚 for the proposed CPS and CPW structures respectively. Similar steps were taken to design the 77GHz bow-tie antenna, and accordingly, the size of the antenna was 퐿푠푢푏 = 4000µ푚 and 푊푠푢푏 = 2000µ푚.

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Figure ‎4.20: Top view of the proposed 250GHz bow-tie antenna with (a): Coplanar strip output pads, and (b): Coplanar waveguide output pads.

The simulated return losses of the proposed 77GHz and 250GHz bow tie antennas are plotted in figure 4.21. The bandwidth spans from 73 to 81GHz and from 241 to 263GHz respectively (for 푆11<-10dB). The proposed structures have shown good matching to the 50Ω input impedance with a simulated VSWR of less than 1.5 in the frequency bands of 73 to 81 GHz and 241 to 263GHz respectively.

0 0

-10 -10

-20

-20

(dB)

(dB)

11

11

S S -30 -30

B.W=8GHz -40 B.W=22GHz -40 72 74 76 78 80 82 240 250 260 270 Frequency (GHz) Frequency (GHz)

Figure ‎4.21: Simulated return loss (푺ퟏퟏ) of the proposed 77GHz and 250GHz bow-tie antennas on a 100µm GaAs substrate.

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Radiation pattern shows the spatial distribution of the radiated power in the far-field region. The gain indicates the ratio of the radiated to the input power of the antenna. An example of the simulated gain of the proposed antenna at 250GHz is depicted in figure 4.22. Simulated gains of 6.92dB and 3.52dB were obtained for the 77GHz and 250GHz bow-tie antennas, respectively. It is clear that the antenna mostly radiates from the sides due to the thin and high dielectric constant of the GaAs substrate (휀푟=12.9).

Once the antennas were designed and optimised, they were exported as touchstones to the schematic platform design in ADS tool for the final evaluation of the ASPAT detector with bow-tie antenna. In the simulation, it is vital to make sure that the resonance frequency of the whole integrated circuit is not shifted when the antenna and ASPAT diode are combined. The 77GHz and 250GHz bow-tie antennas were combined and simulated with the zero-bias 3.7×3.7µm2 and 1.6×1.6µm2 ASPAT diodes, respectively at -30dBm RF power.

Figure ‎4.22: Simulated radiation patterns (gain) of the proposed 250GHz bow-tie antenna on a 100µm GaAs substrate.

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The maximum simulated voltage sensitivity of the 77GHz zero-bias ASPAT detector is 340V/W, as depicted in figure 4.23. Accordingly, the calculated NEP is ~141pW/√퐻푧. Based on the prediction model used in this work, a voltage sensitivity of >2000V/W would be possible to achieve at 77GHz in case of an ASPAT diode of mesa area size of 1.6×1.6µm² when a matching network is employed. Such a detector shows excellent performances compared to previously reported Schottky diodes with log-spiral and dipole antennas at 77GHz, where a maximum voltage sensitivity of 250 to 750V/W was achieved [221].

400 2000 3.7x3.7m 2 ASPAT 1.6x1.6m 2 ASPAT

1900

300

1800

1700 200

1600

Voltage Sensitivity (V/W) Sensitivity Voltage (V/W) Sensitivity Voltage

100 1500 70 72 74 76 78 80 244 248 252 256 260 Frequency (GHz) Frequency (GHz) (a) (b)

Figure ‎4.23: Simulated voltage sensitivity of the zero-bias ASPAT detectors with bow-tie antennas at 77GHz and 250GHz.

Similarly, the 1.6×1.6µm2 GaAs/AlAs ASPAT diode exhibited a maximum sensitivity and minimum NEP of ~1850V/W and 53pW/√퐻푧 respectively at 250GHz, which is comparable to the value of 1000V/W reported in [222] for a zero-bias Schottky detector with a log-periodic antenna at 300GHz.

A technique known as a conjugated matching method can be exploited to conjugately match the inductive part of the antenna with the capacitive part of the diode. A much higher voltage sensitivity of 7000V/W at 75GHz was reported in [223] for a perfectly matched Schottky diode with a folded dipole antenna.

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4.15 Overview of Devices Used in Detectors

In the last two decades, major advances in growth and fabrication of small mesa area size device have allowed the realisation of mm-wave and sub-mm-wave detectors. The use of a particular device type depends on design requirements and limitations. In the high- frequency regimes, series resistance and junction capacitance have to be as small as possible to allow more power to flow in the non-linear junction resistance. Highly sensitive and linear detectors are required to reduce the gain of the low noise amplifier in the detector chain [9, 224]. The detectors reported in the literature mainly use two types of diodes (see table 4.1). The first one relies on thermionic-emission such as Schottky diode and the second relies on tunnelling such as resonant tunnelling and backward diodes. Backward and Schottky diodes have been comprehensively employed in the detection of mm-wave and sub-mm-waves. It is now well established that Schottky diodes represent the best technological choice due to their very high cut-off frequencies, good dynamic range, small size, and low cost [225, 226]. The advantage of a Schottky comes from the fact that it is a single charge (majority) carrier device offering faster switching and recombination times. Moreover, Schottky structures are much easier to grow and fabricate than the backward diodes. Although the backward diode has a limited dynamic range, it gains much attraction since it offers very high voltage sensitivity, good temperature stability, and low junction resistance [224]. The conventional Schottky diodes need a bias at the non-linear point (typically 0.2 to 0.3V) in order to maintain the square-law detection and also to reduce the junction resistance of the diode. More importantly, Schottky I-V characteristics change exponentially with temperature. The Schottky diode adopted in [209] was biased at (0.8V) over the frequency range (26 to 40GHz). The diode exhibited a relatively small series resistance despite the small anode area used. However, the use of bias degraded its noise performance and resulted in a high NEP of 25 to 40pW/√Hz. The need for zero-bias detector suggests the use of low-barrier Schottky diodes, which achieve high sensitivity in the mm-wave frequency band. The InGaAs zero-bias Schottky diode demonstrated in [227] had a high voltage sensitivity of (16000V/W) at 87.5GHz under matched impedance conditions. More importantly, its low‎ junction‎ resistance‎ of‎ 1.1kΩ‎ substantially‎ reduced‎ the noise equivalent power to 0.39pW/√Hz. The zero-bias Schottky diode detector [228] from Virginia Diodes Inc., Charlottesville, VA, USA shows a voltage sensitivity of 4000V/W at 100GHz with a junction‎resistance‎of‎2.6kΩ.‎Another‎advantage of the zero-bias Schottky diode is the

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low junction resistance making the design and fabrication of matching circuits easier. The backward tunnel diode is much less sensitive to temperature and provides much higher sensitivity compared to other reported alternative devices. The results have shown that the curvature coefficient in backward diodes is not limited to (푞/푛푘퐵푇). In 2011, Z. Zhang and et al. [50] developed a sub-micron backward diode with a high curvature coefficient of 47V-1 that led to achieving a very high matched sensitivity of 49700V/W at 94GHz. The device also exhibited the lowest reported noise equivalent power of 0.18pW/√Hz due to the low junction resistance of‎5.9kΩ.‎In‎[47], a curvature coefficient of 49.1V-1 was reported using a 3.14µm2 backward tunnelling diode, leading to a high matched sensitivity of 12000V/W at 94GHz. The work also presented a high sensitivity of 20400V/W at 94GHz using a small mesa area size of 2µm2. The high sensitivity, however, was not enough to compensate for the increase in noise equivalent power due to the extremely high junction‎resistance‎of‎8.26MΩ‎and‎12.3MΩ‎for‎the‎3.14µm2 and 2µm2 backwards diodes respectively. Backward diodes have to be made with a sub- micron area size to compensate for the increase of junction capacitance due to the very thin undoped region. However, series resistance increases with smaller mesa area size, which limits the maximum cut-off frequency. Another issue is the difficulty of processing and fabrication of such diodes since they require high-resolution photolithography techniques. Moreover, a small mesa area size makes the devices very susceptible to burn-out even at low RF power when it is placed in a detector or mixer circuit. The highest reported cut-frequency of a backward diode is ~644GHz at zero-bias [50] and was targeted to detect an RF signal of 94GHz. The device was made of an undoped region thickness of ~16nm with a mesa area size of 0.16µm2. The highest reported RF signal detection is 170GHz using backward diode [229] of mesa area size 0.5µm2 leading to a cut-off frequency of 430GHz. Higher cut-off frequency can be easily attained with even larger mesa area ASPAT diodes as was discussed in chapter 3. Some authors have also suggested other alternatives for the mm-wave detection systems. More recently, the tunnel diode detector reported in [94] made of a low barrier InAlAs between two heavily doped p and n layers. The junction resistance of the diode was found to vary significantly when the temperature was changed. Another drawback was the high series resistance‎of‎130Ω‎that‎limited‎the‎cut-off frequency to 322GHz. Finally, the high noise equivalent power of 150 to 250pW/√Hz limits its tangential sensitivity and makes the detector unable to detect low RF input power. The last detector to be discussed is the one

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reported in [230] employing a small mesa area size of a special type of N-I-N diode. The diode had a relatively high junction‎ resistance‎ of‎ 584kΩ‎ at‎ zero-bias. The detector showed high sensitivity and high noise equivalent power of 4500V/W and 20 to 25pW/√Hz at 340GHz.

Table 4.1 also presents the measured performances of the zero-bias ASPAT detectors conducted in this work. Our zero-bias detectors based on 5.8×5.8µm² ASPAT diode showed improved voltage sensitivity and similar noise equivalent power to zero-bias Schottky detector reported in [231].

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TABLE 4. 1: REPORTED DIRECT DETECTORS

−ퟏ RF freq. 푹푺 (Ω) 푪푱 (fF) 푹푱 (kΩ) 푲푽 (푽 ) 푺푽(V/W) 푵푬푷 (pW/√푯풛) Diode type Mesa area size Reference (GHz) 2 (µm ) 8-13 7 140 10-15 ~17 ~800 to ~11 Zero-bias ASPAT 98 This work 1347 8-13 12 54 30-40 ~18 ~1800 to ~6 Zero-bias ASPAT 33.65 This work 3660 12.4 N/A N/A 1 to 2 N/A ~1000 4.6 Zero-bias Schottky N/A [231]

18-26 12 54 30-40 ~18 ~700 to ~20 Zero-bias ASPAT 33.65 This work 1300 26 to 40 6 2.66 N/A N/A 712 to 1483 25 to 40 Biased Schottky 1.32 [209]

30 N/A ~8.5 10 N/A 30000 N/A Triple barrier RTD 1 [232]

20 to 50 26 18 1.7 42.4 80000 N/A Zero-bias backward 0.72 [233] 50 11 118 0.282 N/A 498 N/A Zero-bias backward 16 [234] 61 50 35 1.75 N/A 8250 N/A Zero-bias Schottky N/A [235] 75 to 93 10 20 18 39 4000 to N/A Zero-bias backward 4 [236] 8000 87.5 N/A 10.2 1.1 N/A 16000 0.39 Zero-bias Schottky ~5 [227] 167

90 45 N/A 0.8 21 6000 0.6 Low-barrier hetero- 2.9 [224] structure 90 18 11 1.38 30 15000 N/A Zero-bias Sb-hetero- 0.64 [11] structure

94 103 2.4 5.9 47 10000 to 0.18 Zero-bias backward 0.16 [50] 25000 94 116 5.8 8200 49.4 12000 30 Zero-bias backward 3.14 [47] 94 250 2.15 12300 N/A 20400 22.1 Zero-bias backward 2 [47] 94 46.2 11.4 196 17.6 1603 N/A Zero-bias backward 3.14 [237] 95 N/A N/A N/A 25.9 11500 N/A Zero-bias backward 2.25 [238] 100 19 N/A 2.6 N/A 4000 ~2 Zero-bias Schottky N/A [228] 1 to 110 11 ~18 18 39.1 3687 N/A Zero-bias backward 4 [239] 170 49 7.5 1.8 40 2400 2.14 Zero-bias backward 0.5 [229] 180 45 N/A 0.8 21 5200 0.7 Low-barrier hetero- 2.9 [224] structure 220 to 330 19 3.8 2.6 ~20 1400 150 to 250 Zero-bias tunnel 0.64 [94] diode 340 N/A N/A 584 ~27 4500 <25 nIN unipolar diode 2 [230] 350 3 6.5 N/A N/A 1000 N/A Zero-bias Schottky 3.5 [240] 500 to 600 N/A N/A N/A N/A 400 to 900 27 to 62.5 Zero-bias Schottky N/A [241]

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4.16 2nd Subharmonic ASPAT Mixers Performances

Two 2nd subharmonic mixers were designed and simulated using the GaAs/AlAs and

In0.53Ga0.47As/AlAs ASPAT diodes of mesa area sizes 3.7×3.7µm² and 3.75×3.75µm² respectively. The mixers were employed to down-convert an RF signal of 77GHz and (- 25dBm) power to an IF of 1GHz using a 35GHz LO signal. The purpose of the anti- parallel diodes is to suppress the effect of LO odd-harmonics by keeping it inside the pair, and only LO even-harmonics appears at the output. To perform a good mixing process, the diode should work at the optimum region in its I-V characteristic, which is the non-linear knee point. For that, applying low LO power means the diode does not reach the knee, and too much power would result in the diode working beyond the non- linear point. The conversion losses of the zero-bias 2nd subharmonic mixers were simulated in ADS tool and plotted in figure 4.24.

35 GaAs/AlAs Sample

In0.53Ga0.47As/AlAs Sample 30

25

20

15 ConversionLoss (dB)

10

5 -10 -5 0 5 10 LO power (dBm)

Figure ‎4.24: Simulated conversion loss of the 3.7×3.7µm² GaAs/AlAs and 3.75×3.75µm² nd In0.53Ga0.47As/AlAs 2 subharmonic mixers at 77GHz RF signal.

The RF power was fixed at -25dBm, while the LO power was swept from -10 to 10dBm. A lower conversion loss was achieved using the GaAs/AlAs ASPAT diode due to its higher current density, which means more power pumping to the output IF circuit. A minimum conversion loss of ~10dB at 0dBm LO power is comparable to the value of the Schottky diodes reported in [242, 243], where a conversion loss of ~10dB was obtained

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at 76GHz and ~4dBm LO power. From this, subharmonic mixers based Schottky do need a high LO power to work efficiently, while the current design provides the same conversion loss with much lower LO power. Another important factor is the 1-dB compression point. At relatively low RF power, there is a constant relationship between the input and output powers. As the input RF power increases too much, the output power starts to saturate, and this relation is no longer constant. This consequently makes the mixer behave as a non-linear system. 1-dB compression point can be observed by simulating the conversion while sweeping the RF power and fixing the LO power.

Figure 4.25 illustrates the conversion loss as a function of the input RF power. The mixers enter the non-linear region at -13 to -12dBm RF input power, where the conversion loss increased by 1dB. Another important linearity factor is the third-order intercept point that has been investigated for both mixer designs.

nd Two input signals RF1=77GHz and RF2= 76.9GHz were applied to the 2 sub-harmonic mixer. Hence, mixer produced two output signals at IF1 = 1GHz and IF2 = 0.9GHz respectively, as well as the harmonic products at (2 × RF1 − RF2 − 2 × FLO = 1.1GHz) and (2 × RF2 − RF1 − 2 × FLO = 0.8GHz). 퐹퐿푂 is the frequency of the LO signal.

20 GaAs/AlAs Sample

In0.53Ga0.47As/AlAs Sample 1-dB compression

16

12

1-dB compression ConversionLoss (dB)

8 -30 -28 -26 -24 -22 -20 -18 -16 -14 -12 -10 LO power (dBm)

Figure ‎4.25: 1-dB compression of the 3.7×3.7µm² GaAs/AlAs and 3.75×3.75µm² In0.53Ga0.47As/AlAs 2nd subharmonic mixers at 77GHz RF signal.

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An example of the generated output spectrum of the 2nd subharmonic mixer based GaAs/AlAs ASPAT diode is shown in figure 4.26 (a). The RF input power was swept from -30 to 5dBm while keeping the LO power at 0dBm. The third order intercept points are shown in figure 4.26 (b).

0 20

IF IF 0 2 1 OIP3 2xRF -RF -2xF 2 1 LO 2xRF1-RF2-2xFLO -20

-40

-200

-60

IF Current (dB) Current IF -80 Output Power (dBm) Power Output

-100 IIP3 -400 -120 0.6 0.7 0.8 0.9 1.0 1.1 1.2 1.3 -30 -20 -10 0 10 20 Frequency (GHz) Input RF power (dBm) (a) (b)

Figure ‎4.26: (a): Spectrum of the IF Current in dB, (b): 3rd intercept points of the 3.7×3.7µm² GaAs/AlAs subharmonic mixers at 77GHz RF signal.

Figure 4.26 (b) shows the input (IIP3) and output (OIP3) 3rd order intercept points of roughly -1dBm and -15dBm, respectively. Moreover, two mixers exploiting 1.6×1.6µm² and 2.4×2.4µm² GaAs/AlAs ASPAT diodes were simulated at an RF frequency signal of 100GHz and 54.5 GHz LO signal. A minimum conversion loss of ~10dB was achieved at a high LO power of >7dBm. The high series resistance of 76Ω and 40Ω of the 1.6×1.6µm² and 2.4×2.4µm² GaAs/AlAs ASPAT diodes increased the power consumption of the mixers and resulted in less power delivered to the ASPAT diodes. Therefore, it is vitally important to reduce the series resistance of the ASPAT diodes in the case of high-frequency mixer circuits.

The tunnel ASPAT diodes showed good mixing performances at 77GHz enabling them to be integrated into mm-wave transceiver for car radar applications. However, the fabrications of such three-port integrated circuits are critical and do require advanced techniques to simulate and optimise the total performances. Moreover, such circuits involve a three-terminal measurement setup with high precision and resolution to mitigate any error coming from the tool or cables that could affect the measurement process. 171

4.17 Overview of the Reported Subharmonic Mixers

Several subharmonic mixers have been introduced in the literature using different devices ranging from Schottky diode, HEMT, pHEMT, and HBT transistors. The realisation of sub-micron Schottky devices has paved the way towards efficient operation at millimetre-wave and sub- millimetre-wave frequencies. The zero-bias devices are very attractive because of their much reduced power consumption and production costs. Mixers working at high frequencies near 1 THz with acceptable performances had been reported [244].

Table 4.2 summarises the most important figure of merits of some of the reported subharmonic mixers. The works reported in [245-247] use SHMs based on GaAs HEMT and pHEMT transistors with a conversion loss of ~10dB at K, Ka, Q, and W-bands.

In [248], a subharmonic gate mixer was targeted to down-convert the RF frequency of 89GHz into an IF signal of 5GHz. An LO signal with a frequency and power of 42GHz and 7dBm respectively was pumped into the gate to bias the transistor near the pinch-off voltage. The mixer demonstrated weak isolation between the ports as a direct result of applying LO and RF signals at the same gate terminal. Finally, the mixer used an external bias of 2.4V and -0.75V to achieve a conversion loss of 4.7dB.

At mm-wave frequencies, a SHM based GaAs Schottky diode in [249] has shown a low conversion loss of 7.5dB under an 8.5dBm LO power. However, the Schottky diode does require a high LO power (around 2 to 15dBm) to reach its non-linear point and generate the 2nd harmonic component. Another candidate which has a promising feature is the RTD. The second harmonic signal is produced using single symmetric I-V characteristics of the RTD.

The work presented in [66] introduced the first SHM based RTD devices at 20GHz and 100GHz. For the 20GHz mixer design, a 60µm2 RTD exhibited a conversion loss of 21dB at 7dBm LO power. Similarly, a conversion loss of (~22dB) was achieved at 100GHz using a 225 µm2 RTD when it is pumped with a 0dBm LO power.

However, having an RTD with strong non-linearity at low bias could boost the mixer performances. Moreover, for mixer design at frequencies exceeding 0.5THz, submicron devices with extremely small junction capacitance and small series resistance have to be employed for good performances. The ultimate goal is to use a device with a strong non-

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linear point at low bias close to zero for a lower conversion loss at low LO power. The work presented in this thesis introduces a new candidate device termed the Asymmetric Spacer Tunnel Diode (ASPAT), which has strong non-linearity at bias close to zero.

For the sake of comparison, the table also includes the performances of the 2nd subharmonic mixers based 3.7×3.7µm² and 3.75×3.75µm² GaAs/AlAs and

In0.53Ga0.47As/AlAs ASPAT diodes at an RF signal of 77GHz and 38GHz LO signal. The 3.7×3.7µm² GaAs/AlAs 2nd subharmonic mixer has comparable performances to the Schottky mixer reported in [249] with the advantage of less required LO power.

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TABLE 4. 2: SOME OF THE REPORTED 2nd SUBHARMONIC MIXERS AND ASPAT MIXERS PERFORMANCES

Device Used 풇푹푭 (푮푯풛) 푪푳 (dB) 풇푳푶 (푮푯풛) 푷푳푶 (풅푩풎) 푷푰풔풐(푳푶−푹푭)풅푩 푷푰풔풐(푳푶−푰푭)풅푩 Reference 0.15µm GaAs pHEMT 18-40 12 9-20 17 22 42 [245] 0.15µm GaAs pHEMT 23-37 9.4-12 N/A 13 22 31 [246] 0.15µm GaAs pHEMT 38-48 11-16 N/A 10 29 17 [247] 0.15µm GaAs pHEMT 40-50 -1.1 22.7 -4 20 N/A [250] 0.15µm GaAs pHEMT 89 4.7 42 7 15 23 [248] Schottky diode 24 15 12 1.5 N/A N/A [84] Schottky diode 92-96 7.5 45-46 8.5 40 33 [249] Schottky diode 183 7 94 7 N/A N/A [251] Schottky diode 330 6.3 N/A 6.5 N/A N/A [252] Schottky diode 560 7 N/A 15 N/A N/A [75] Schottky diode 664 8 330 2 N/A N/A [253] Schottky diode 874 10 N/A N/A N/A N/A [244] RTD 20 21 N/A 7 N/A N/A [66] RTD 100 22 N/A 0 N/A N/A [66] GaAs ASPAT 77 ~10 38 0 N/A N/A This work

In0.53Ga0.47As ASPAT 77 ~16 38 0 N/A N/A This work

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4.18 Summary

Chapter four demonstrated the design procedure and characteristics of coplanar waveguide structures at high-frequencies. The design and modelling of the discrete component, including MMIC capacitor, matching circuit and ASPAT SDD model was studied and analysed in-depth. A 10pF discrete MMIC capacitor was fabricated and measured up to 40GHz. The parasitic components of the MMIC capacitor were successfully extracted using the equivalent circuit model. The use of polynomial representation with the help of the SDD model was an efficient method to model the measured I-V characteristics on the schematic platform of the ADS tool.

Four detector circuits using ASPAT diodes were designed, fabricated and experimentally tested at X and K-band frequencies. The voltage sensitivity and noise equivalent power were evaluated from the measured DC output voltage and junction resistance of all detectors. The measured data showed a maximum voltage sensitivity of 3650V/W and 1300V/W at 11GHz and 24GHz at zero-bias. A minimum noise equivalent power of ~6pW/√Hz at 11GHz was calculated for the 5.8×5.8µm2 zero-bias GaAs/AlAs ASPAT detector. An investigation of the detection characteristics of ASPAT detector with a bow- tie antenna was carried out at mm-wave frequencies. The proposed antennas exhibited a wide operating bandwidth of 8GHz and 22GHz at centre frequencies of 77GHz and 250GHz respectively. The maximum simulated voltage sensitivities were 350V/W and 1850V/W using the zero-bias 3.7×3.7µm2 and 1.6×1.6µm2 ASPAT diodes at 77GHz and

250GHz, respectively. On the other hand, the GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes of mesa area size of 3.7×3.7µm² and 3.75×3.75µm² have presented good mixing performances with a minimum conversion loss of 10dB and 16dB respectively, at 77GHz and 0dBm LO power. Better performances can be achieved with even smaller feature sizes, high curvature coefficient, and better matching network with minimum losses.

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CHAPTER 5: PHYSICAL MODELLING AND

EXPERIMENTAL CHARACTERISATION OF APD

AND PIN PHOTODETECTORS FOR HIGH DATA

RATE APPLICATIONS

5.1 Introduction

The design, fabrication, and physical modelling of photodetector for the high-data-rate optical application need full knowledge of electronic and optical characteristics of all involved layers. Careful steps have to be followed to select proper materials in which maximum performances can be obtained. Moreover, the thickness and doping of layers are crucial in defining the electric field across the structure that, in turn, specifies the transport mechanism of carriers. The work in this chapter deals firstly, standard

In0.53Ga0.47As/In0.52Al0.48As APD and In0.53Ga0.47As PIN photodetectors with different light window aperture sizes were designed and then experimentally fabricated and characterised regarding their DC and high-frequency optical characteristics. Secondly, high-frequency equivalent circuits from fabricated devices were built up to 40GHz to extract key diode parameters including (퐶퐽), (푅퐽), and (푅푆) utilising the ADS tool. Extracting the intrinsic parameters is necessary to calculate its cut-off frequency and predict photodetectors performance for high-frequency applications. For the development and fabrication of such high data-rate APD or PIN diode, it is more efficient to have a physical model that can accurately predict and analyse the effect of different parameters such as mesa area size, absorber layer thickness, and light window aperture size. This model can help to adequately mitigate any performance degradation and lowering the effective cost of production. This work concentrates on the optimisation of photodetectors (PIN diode, and APD) in order to enhance their capability of operating at data rates in excess of 25Gb/s. Using full virtual wafer fabrication physical modelling (DC, AC, and optical characteristics), 3D analytical models were built and simulated for both standard photodetectors using the ATLAS SILVACO tool. All simulations were performed for normal incidence devices. The modelled structures are validated by the fabricated devices in terms of electrical and optical characteristics. Three process factors,

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namely: absorber thickness, light window aperture, and mesa area size were optimised to enable the photodetectors to operate at a data rate higher than 25Gb/s. The optimised PIN photodetector denoted as (15D) has an optoelectric 3-dB bandwidth of 35GHz at -5V bias when a 10µW optical power is applied, while the optimised APD denoted as (15A) and has an optoelectric bandwidth of 21GHz and a multiplication gain of 3 at -21.6V and 1µW incident optical power.

5.2 Epi-layer Structures of Photodetectors

The photodetector epi-layers were grown using Solid Source Molecular Beam Epitaxy (SSMBE) on 620µm thick semi-insulating InP substrates. The standard PIN structures (30S, 20S, and 15S) comprise of undoped InGaAs material as an absorber sandwiched between two heavily doped ~0.1µm p-type In0.53Ga0.47As top and 0.5µm n-type

In0.53Ga0.47As bottom contact layers as depicted in table 5.1. The absorber is relatively thick (~2µm), but this makes the PIN efficient at absorbing light in the (1.3 to 1.6µm) wavelength region.

TABLE 5. 1: EPI-LAYER STRUCTURE OF THE STANDARD In0.53Ga0.47As PIN DIODE

Layer Material Doping (cm-3) Thickness (µm)

19 Top Contact p+-In0.53Ga0.47As ~1x10 ~0.1

Absorber i-In0.53Ga0.47As Undoped ~2

19 Bottom Contact n+-In0.53Ga0.47As ~1x10 ~0.5

Substrate InP S.I. ~620

The standard APD structure (30A) as shown in table 5.2 comprises p-type In0.53Ga0.47As top and n-type In0.53Ga0.47As bottom contacts, p-type In0.53Ga0.47As top and n-type

In0.52Al0.48As bottom cladding layers, undoped In0.53Ga0.47As absorber layer, p-type

In0.52Al0.22Ga0.25As grading layer, p-type In0.52 Al0.48As charge sheet layer, and undoped

In0.52Al0.48As multiplication layer. The top and bottom contacts are heavily doped ~3x1019 cm-3 and ~1x1019 cm-3 with thicknesses of ~300nm and 500nm respectively.

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The highly doped contacts help to reduce the series resistance (푅푆) which leads to improvements in the frequency response of the device. The absorber is relatively thick (~1.2µm). The grading layer thickness is 50nm, which improves the frequency response of the APD by reducing the band-discontinuity at the interface with the charge layer. The charge layer has a doping profile of ~1x1018 cm-3 with a thickness of 50nm. The main function of this layer is adjusting the electric field of the device. Finally, the multiplication layer with a thickness of 200nm is buried under the charge layer.

TABLE 5. 2: EPI-LAYER STRUCTURE OF THE STANDARD In0.53Ga0.47As/In0.52Al0.48As APD (30A)

Layer Material Doping (cm-3) Thickness (µm)

19 Top Contact p++-In0.53Ga0.47As ~3x10 ~0.03

18 Cladding p+-In0.53Ga0.47As ~1x10 ~0.17

Absorber i-In0.53Ga0.47As Undoped ~1.2

16 Grading p-Al0.22Ga0.25In0.52As ~5x10 ~0.05

18 Charge Sheet p+-Al0.48In0.52As ~1x10 ~0.04

Multiplication i-Al0.48In0.52As Undoped ~0.2

18 Cladding p+-Al0.48In0.52As ~2x10 0.2

19 Bottom Contact n++- In0.53Ga0.47As ~1x10 0.5

Substrate InP S.I. ~620

5.3 Fabrication and Small Signal RF Equivalent Circuit Extraction

Figure 5.1 shows a fabricated photodetector employing a 50Ω coplanar waveguide configuration for electrical and optical measurements. Three PIN photodetectors (15S,

20S, and 30S) were fabricated with a light window aperture size of 15, 20 and 30µm respectively. The standard APD (30A) was fabricated with a light window aperture size of 30µm. For the APD design, 퐷 was designed to be ~13.5µm, while it was ~10µm 푔푎푝 for the PIN structures. The width of the gold anode contact is 6µm. As a part of the high-

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4µm

W

m

µ 6

35µm

50µm 100µm

Figure ‎5.1: Fabricated photodetector. The inset shows the light window aperture (W) and 푫품풂풑 of the photodetector. (images are not to scale).

frequency characterisation, on-wafer (푆11) reflection parameter measurements were performed for the open, short, and actual structures using an Anritsu VNA from 40MHz to 40GHz at different bias. All measurements were performed in the dark at room temperature. Advanced Design System (ADS) tool was employed to extract the intrinsic and extrinsic component of the devices. The dimension of the GSG coplanar waveguide was optimised‎to‎give‎an‎impedance‎of‎50Ω. As there is no standard model for the APD in ADS, an equivalent circuit model was built, and the fitting was made with the measured data. The equivalent circuit of the open structure was built in ADS and is represented by a capacitor only (퐶푃), while the short structure is represented by an inductor 퐿푃. The simulated S-parameters of the equivalent circuits were fitted with the measured ones to extract 퐶푃 and 퐿푃. The measured and simulated S-parameters represented on a Smith chart for the open and short devices are shown in figure 5.2. The simulated 푆11 data show excellent agreement with the measured data in the low- frequency range up to 30GHz. At higher frequencies, there is a slight deviation between the measured and simulated data of the short structure, which could be due to measurement error or leakage issue. The extracted 퐶푃 and 퐿푃 of the standard APD (30A) and PINs (30S, 20S, and15S) are 8 to10fF and 40 to 50pH respectively. The small parasitic capacitance (퐶푃) comes from the optimised coplanar waveguide design process.

The simulated 푆11 of the ADS equivalent circuits were fitted with the experimental

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results to extract all parameters. Figure 5.3 depicts the measured and equivalent circuit S- parameters of the PINs and APD photodetectors at fully depleted voltages.

Figure ‎5.2: Measured and simulated 푺ퟏퟏ represented on smith charts of the open and short structures and corresponding equivalent circuits.

PIN (15S) Measured data PIN (20S)

Simulated data

Equivalent circuit at negative bias voltage

PIN (30S) APD (30A)

Figure ‎5.3: Measured and simulated S-parameters represented on Smith charts and of the standard PINs and APD at fully depleted bias.

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Table 5.3 lists the extracted parameters of the APD and PIN diodes when fully depleted. The fully depleted junction capacitance of the standard APD (30A) is 162fF while it was found to be much smaller for the 15S, 20S, and 30S PIN diodes with extracted values of 46fF, 55.5fF, and 104fF respectively, and this is due to the larger light window aperture size, mesa area size, and thinner depletion region of the APD.

TABLE 5. 3: STANDARD APD AND PIN DIODES EXTRACTED PARAMETERS AT FULLY DEPLETED BIAS.

Component PIN (15S) PIN (20S) PIN (30S) APD (30A)

퐶퐽, fF 46 55.5 104 162

푅퐽,‎kΩ 50 50 50 15

푅푆,‎Ω 5 5 5 10

퐶푃, fF 10 10 10 8

퐿푃, pH 50 50 50 40

Intrinsic 692 478 255 100

풇풄풖풕−풐풇풇 (GHz)

Larger device capacitance degrades the high-frequency performance, as the RC and transit time components limit the 3-dB optical bandwidth. The highly doped top and

bottom contacts resulted in a relatively small series resistance of the APD and PIN

diodes, which leads to improvements in the frequency response of the device when the optimum intrinsic region width is employed. The extracted 푅 of the standard APD and 푆 PIN diodes are‎10Ω‎and ~5Ω‎respectively. The APD series resistance is larger due to the extra resistances introduced from several layers. Above all, the separation between the

top and bottom electrodes is larger in the case of the APD structure (13.5µm vs 10µm),

which affects the total series resistance. The intrinsic cut off frequencies (푓푐푢푡−표푓푓) were

evaluated for all structures using the usual expression (1/2휋푅푆퐶퐽) and found to be

692GHz, 478GHz, 255GHz, and 100GHz for the PIN (15S, 20S, and 30S), and APD (30A) respectively. However, in the case‎of‎a‎50Ω‎load the cut-off frequencies decrease to 63GHz, 51.6GHz, and 27.3GHz for the PINs (15S, 20S, 30S) respectively, and 63GHz Vertical lines are optional in tables. Statements that serve as captions for the entire table do not need footnote letters. aGaussian units are the same as cg emu for magnetostatics; Mx = maxwell, G = gauss, Oe = oersted; Wb = weber, V = volt, s = second, T = tesla, m = meter, A181 = ampere, J = joule, kg = kilogram, H = henry.

for the standard APD (30A). Extracting the junction capacitance of the devices using the high-frequency small-signal equivalent circuit model is necessary to validate the SILVACO physical model which exhibits almost the same value when the APD and PIN diodes are fully depleted as will be discussed later.

5.4 Experimental Characterisation Tools

All devices were experimentally tested to obtain their electrical and optical characteristics. The electrical characterisation was accomplished under dark room condition and room temperature to measure the dark current, capacitance-voltage (C-V) characteristic, and high-frequency S-parameter measurements up to 40GHz at different bias. Different pieces of equipment were used in the work, including an RF probe station and an Anritsu VNA to collect the S-parameter data.

For the optical characterisation, laser light of 1.55µm wavelength and variable power (1 to 100µW) was utilised to illuminate the device, as shown in the setup of figure 5.4. A ‘Lightwave Component Analyser’ (LCA) (HP 8703A) was employed to measure the 3- dB bandwidth of the devices. The bias is applied using a DC supply (HP4142B) connected to the Analyser through a bias-T.

Figure ‎5.4: Optical system set up on-wafer measurements.

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5.5 Physical Modelling Characterisation Tool

The primary objective of the work was to build a quantitative and predictive physical model for the standard 10Gb/s APD and 25Gb/s PIN photodetectors to validate the measured electrical and optical characteristics. Numerical simulations of the standard photodetectors under dark and light conditions and at room temperature were carried out using the Atlas SILVACO tool. SILVACO [254] is a simulation software that allows the user to build and model 2D and 3D structures electrically, thermally, or optically at different bias, in effect performing a virtual wafer fabrication process. The simulator considers many variables and conditions as the real device being simulated, and as a result, the analysis process is more accurate, and the output results can be highly matched to measured ones. The tool uses a set of differential equations‎ such‎ as‎ Poisson’s‎ and‎ drift-diffusion, as well as Fermi-Dirac or Boltzmann statistics to model carrier transport through PIN or APD structures. As all equations are derived from‎Maxwell’s‎laws,‎the‎ SILVACO Atlas tool is capable of performing DC, AC and transient analysis for 2D and 3D device structures on various material types (binary, ternary, and quaternary). The process of building the structure, defining its parameters and variables, choosing the desired model statement, performing the required analysis, and finally displaying the results can be grouped into the following main statements:

Structure Specification: this statement is used to define the structure as follow:

•‎Mesh: Mesh statement is used to describe the structure either in two-dimensional (2D) or three-dimensional (3D) Cartesian grids. All coordinates are in units of microns, and the spacing parameter is used to improve the precision and accuracy of the analysis at a given location.

•‎Region: This part defines the layers of the structure. Each layer represents a separated section and needs to be defined in the statement independently. The mesh must be assigned to a region, and the region number must be ordered from lowest to highest region.

•‎Electrode: This statement defines the location of biasing points for the electrical and optical analysis. In this study, two probes have been allocated as the anode and cathode. The anode is allotted at the top of the vertical device while the cathode is at the bottom of the structure.

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•‎ Doping: In this field, the doping concentration level for each region is defined depending on the material type either p or n.

Material and Models Specification

•‎ Material: Used to define the parameters of different materials of the device such as energy band gap, effective mass, mobility, permittivity. By default, ATLAS has complete parameters for Silicon, GaAs and AlAs materials. In the case of using a new material (such as InGaAs or InAlGaAs), then all necessary parameters have to be specified and manually defined in the material statement.

•‎ Models: The most crucial part is the inclusion of the specific model for accurate modelling of a particular device. Device structure and type determine the required physical model, for instance, to model the impact ionisation process of the APD, an IMPACT SELBER model is used as will be discussed later.

•‎Contact: Contact statement is used to specify the physical attributes of an electrode. An electrode attached to the semiconductor material is assumed by default to be ohmic. If the work function is defined, the electrode is treated as a Schottky contact.

Numerical Method Selection: This statement is used to compute the solution for problems. The calculation is performed using a non-linear iteration procedure that starts from an initial guess and uses an iterative process to find the estimated solution.

Solution Specification: The user must define a log, solve, save statement in the ATLAS simulation. These statements work together to provide data for analysis by other functions. In this work, all three statements are used to analyse the I-V characteristics produced by the device itself, its energy band diagram and different outputs. Problem solution is turned on once the simulator runs at the solve statement. Later, in the log statement, any DC, transient or AC data generated by the solve statement will be saved to a file. Save statement used to store all data point to a node in the output file.

As the SILVACO library does not contain material parameters for InGaAs, InAlAs, and InAlGaAs, all III-V material parameters were obtained both from the works of literature and from validation from many devices studied over the years in our laboratory [55, 139, 255-257].

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5.6 Physical Modelling and Optimisation Details

The photodetectors were designed and fabricated with a circular mesa shape having a specific light window aperture size. A 3D modelling was carried out to simulate the performance of photodetectors. The 3D simulation is more accurate than the 2D one, as it models the exact structure dimensions which take into account the effect of the electric field at the edges of the structures. However, for simplicity, the photodetectors were built and modelled with a 3D-rectangular mesa shape, as shown in figure 5.5. The rectangular shape photodetectors have precisely the same dimensions of the fabricated ones. In Silvaco, the 3D-rectangular coordinates are represented by X-Z-Y. The effective mesa area sizes are 1962µm², 1256µm², and 961µm² for the standard PIN (30S), PIN (20S), and PIN (15S) diodes respectively and 1960µm² for the APD (30A). Following this, the length (퐿푚푒푠푎) and width (푊푚푒푠푎) of the mesa were calculated for all structures. The same calculations were performed for all other dimensions (light window aperture size,

퐷푔푎푝, anode and contact sizes) of the standard and optimised photodetectors.

Figure ‎5.5: Modelled 3D rectangular photodetector.

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The experimental characterisations of the APD (30A) and PIN diodes (30S, 20S, and 15S) were accomplished to investigate their electrical and optical performances realistically. The electrical characteristics were measured in the dark and at room temperature. Capacitance-Voltage (C-V) measurement are crucial to validate the extracted 퐶퐽 from the 푆11 reflection data and to practically extract the punch-through voltage of the APD as well as ensuring that the actual doping profile and thickness of the layers are close enough to the designed ones. This work firstly uses numerical simulations as a tool to build a 3D quantitative and predictive physical model for the APD and PIN photodetectors and to validate the measured electrical and optical data. Secondly, the successful and verified models were then adopted to virtually investigate the effect of different parameters and optimise the performances of the photodetectors. The APD structure requires more care to build and activate the appropriate models since it is more complicated compared to the PIN diode. Shockley-Read-Hall (SRH) model and Fermi-Dirac statistics were used to model the generation-recombination and carrier drift-diffusion processes. To model the impact ionisation process of the APD, an IMPACT SELBER model was used. The model was developed by Selberherr [147] to estimate the impact ionisation rate coefficients 훼(퐸) and 훽(퐸). The model was derived using the classical Chynoweth model [258] and based on the equations 2.32 and 2.33 which include the parameters (AN, BN, BETAN, AP, BP, BETAP) provided in chapter 2. SELBER model calculates 훼(퐸) and 훽(퐸) by computing the value of the lattice temperature-dependent parameters (AN, BN, AP, BP) using the equations reported in [259]. BETAN and BETAP were predicted by Shockley in [260] with a value equal to ~1.

The optimisation of our structures consists of two process factors which have a significant effect on the high-frequency performances. The absorber thickness was selectively thinned to be 0.5µm for the APD and PIN diode with the aim of reducing the transit time of the electrons and thus enhancing the 3-dB optoelectric bandwidth. However, a thinner depletion region results in a higher junction capacitance. So, further optimisation was carried out by reducing the mesa area size, top gold electrode width, and the light window aperture. The latter was optimised to be 15µm, making the effective mesa area of both photodetectors ~490µm². A smaller window aperture would increase the complexity of the packaging and assembly of the devices. The design process of the absorber layer determines the responsivity and the maximum operating 3-dB bandwidth

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of the photodetector. For investigation purposes, 퐹푇, 퐹푅퐶, and 퐹3푑퐵 were calculated and plotted in figure 5.6 for the optimised APD as a function of the intrinsic region width.

70 FT FRC 60 Total F3dB

50

40

30 Bandwidth(GHz)

20

10 0.4 0.6 0.8 1.0 1.2 1.4 Intrinsic Region (m)

Figure ‎5.6: Calculated 3-dB optical bandwidth of the optimised In0.53Ga0.47As/ In0.52Al0.48As APD.

It is noteworthy to mention that the optimised APD and PIN diode are denoted as (15A) and (15D), respectively. Table 5.4 summarises the devices reported in this thesis. The intrinsic region is assumed to be fully depleted, which is the case when the bias is equal to -15V. Therefore, the plot in figure 5.6 is restricted to (-15V

TABLE 5. 4: THE STANDARD AND OPTIMISED DEVICES

Standard Optimised

APD PIN APD PIN

Device 30A 15S 20S 30S 15A 15D

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At bias higher than (90%푉퐵푅), the saturated drift velocity of carriers starts to decrease due to electron scattering‎from‎Г‎to‎L‎and‎X‎valleys‎at‎high‎electric‎fields‎causing‎it‎to‎ degrade the 3-dB optical bandwidth as will be discussed later. The intrinsic region of the optimised APD (15A) includes both a 0.5µm absorber layer and a 0.2µm multiplication layer. Figure 5.6 indicates that the carrier transit frequency dominates the 3-dB optical bandwidth for an intrinsic region thickness greater than 1µm. On the contrary, it is limited by the RC bandwidth for an intrinsic region thickness smaller than 0.5µm. The highest calculated 퐹3푑퐵 of the optimised APD (15A) structure is 22.5GHz when the intrinsic region width ranges between 0.5 to 0.8µm meaning that the optimum absorber thickness is probably around 0.5µm. However, these calculated results do not take the effect of the parasitic elements into account which can have a large impact at high operating frequencies.

5.7 Dark Currents and C-V Characteristics

The dark currents of standard APD (30A) and PIN diodes (15S, 30S, and 20S) were measured up to -25V and -5V bias respectively using a probe station under dark and at room temperature conditions.

The modelling process started with simulating I-V and C-V characteristics by maintaining a good agreement with experimental data, to build appropriate physical models and validate material parameters used which then can be used to simulate and predict the optical characteristics of higher frequency photodetectors. The inclusion of different models is necessary to simulate the exact physical phenomena. IMPACT SLEBER was used to model the impact ionisation process of the APD that causes the avalanche breakdown phenomena at high internal gains. Such a phenomenon is quite challenging to‎ model‎ as‎ it‎ depends‎ on‎ several‎ material‎ parameters‎ such‎ as‎ carrier’s‎ impact ionisation coefficient, applied electric field and doping profiles. The modelling process of the APD was performed using two conditions. The first condition was based on the dark current characterisation, where no electron-hole generation is defined in the absorber layer, and no multiplied photocurrent is created. The output current is a sum of the un-multiplied and multiplied dark current. Therefore, electrons travel with their average velocity in both multiplication and charge sheet layers. The electron velocity was set to 2.5x106 cm/s and 1x107 cm/s in the charge and multiplication layers, respectively.

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The electron velocity in the absorption layer was set to ~1.5x107 cm/s. The electron mobility of each layer (InGaAs, InAlAs and InAlGaAs) are different depending upon the doping profile of each layer. All required values were obtained from the literature in [255-257]. The key fitting parameters used in SILVACO modelling are shown in table 5.5. It is noteworthy to state that ATLAS SILVACO tool assumes a constant effective mass, which does not accurately reflect the impact ionisation process of APD structures.

TABLE 5. 5: KEY FITTING PARAMETERS USED IN SILVACO PHYSICAL MODELLING.

Absorber layer Multiplication layer Grading layer Parameter (In0.53Ga0.47As) (In0.52Al0.48As) (In0.52Al0.22Ga0.25As)

Electron mobility 11000 4500 2300 (v/cm²)

Energy gap (eV) 0.75 1.44 ~0.99

Affinity (eV) ~4.5 ~4.25 ~4.38

Permittivity 13.9 12.2 ~12.5

Electron carrier life 100 - - time (ns)

Electron effective ~0.042 ~0.085 ~0.06 mass

Hole effective mass ~0.46 ~0.6 ~0.61

The band-to-band tunnelling current was not considered due to the inclusion of the graded and charge sheet layers which provide enough electric field separation between the absorber and multiplication layers. Band-to-band tunnelling current starts to dominate the dark current of the APD when the multiplication region is smaller than 100nm [261].

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The most important phenomena to take into account for accurate physical simulation of the dark current is the impact ionisation process, which is described by the following equation [254]:

퐺= 훼(퐸) |퐽|푛 + 훽(퐸) |퐽|푝 (5.1)

where (퐺) is the generation rate of the electron-hole pairs, |퐽|푛 and |퐽|푝 are the electron and hole current densities. In SILVACO, parallel electric field dependence (FLDMOB) and a local field IMPACT SELBER models were used to model the lateral electric field mobility dependency and impact ionisation rate of electron and hole of APD structure.

Both models are necessary to fit the dark current and break down voltage (푉퐵푅). Through the simulation, SILVACO calculates impact ionisation parameters (AN, BN, BETAN, AP, BP, BETAP) according to the material parameters of the charge sheet multiplication regions (lattice temperature and energy gap) as well as the required model for the impact ionisation process. The calculation process of the parameters can be found in details in [254]. The output window of the SILVACO resulted in values of 8.6x106 cm-1, 2.3x107 -1 6 -1 6 -1 cm , 3.5x10 cm , and 4.5x10 cm for AN, AP, BN and BP respectively and 1 for both BETAN and BETAP respectively. The black and red lines in figure 5.7 (a) and (b) which represent the simulated and measured dark currents of the standard APD (30A) and PIN diode (15S) are very close to each other.

The measured dark currents of the standard APD (30A) and PIN diode (15S) are ~8nA and 2.2nA at 90%푉퐵푅 and -5V bias, respectively. The doping profile of the charge sheet layer plays an important role in determining the breakdown voltage of the APD.

Therefore, the doping was changed slightly to keep 푉퐵푅 at (-23.7V) for the standard and optimised APD. The doping profile of the charge sheet layer was set to ~6.5x1017 cm-3. Dark current is directly proportional to device mesa size; therefore, scaling of the device reduced the dark current of both photodetectors. The optimised APD (15A) exhibits a dark current of 1.5nA at 90%푉퐵푅 which is much lower than the InGaAs and Si-Ge APD reported in [131, 138, 262]. The electric field of our APD is greatly confined in the InAlAs multiplication layer, and only the Shockley-Read-Hall process occurs in the InGaAs absorber layer and generates the dark current. Similarly, the optimised PIN diode (15D) has a dark current of <2nA at -5V bias which is comparable to previously reported Ge and InGaAs PIN diodes in [155, 263].

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10-3 Measured standard APD (30A) 10-4 Simulated standard APD (30A) Simulated optimised (smaller) APD (15A) 10-5

10-6

10-7

10-8

10-9 Current(A) 10-10 VBR=-23.7V 10-11

10-12

10-13 -25 -20 -15 -10 -5 0 Voltage (V) (a)

10-6 Measured standard PIN diode (15S) Simulated standard PIN diode (15S) Simulated optimised (smaller) PIN diode (15D) 10-7

10-8

10-9 Current(A)

10-10

10-11 -5 -4 -3 -2 -1 0 Voltage (V) (b)

Figure ‎5.7: Measured and simulated dark currents of the standard and optimised

(a): In0.53Ga0.47As/In0.52Al0.48As APDs, and (b): In0.53Ga0.47As PIN photodetectors.

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The standard PIN diodes (20S) and (30S) have recorded a minimum dark current of 2.9nA and 3.7nA at -5V bias, respectively. These values make the devices appropriate candidates to achieve high SNR and high sensitivity in optical communication receivers. The total measured capacitances including the junction capacitance and pad capacitance were extracted from the measured S-parameter at different bias using the well-known expression (퐼푚푎푔(푌11)/2휋푓). The SILVACO models were used to simulate and fit the junction capacitances of the standard APD (30A) and PIN diode (15S) with the experimental data, as well as simulating the C-V data of the optimised (smaller and thinner) structures (15A and 15D) as depicted in figure 5.8. The plot shows an excellent fit between the measured and simulated data for the standard APD (30A) and PIN diodes (15S). The fully depleted capacitance occurs at a bias (>-15V and >-3V) for the APDs and PIN diodes respectively. The slope indicates that the punch-through voltage (푉푃푇) of the APD is (-12.5V), which is far enough from the 푉퐵푅 of (-23.7V). It is clear that minimising the light window aperture and the mesa area size of the APD has resulted in a remarkable improvement in the fully depleted junction capacitance of the optimised APD (15A). However, this is not the case for the optimised PIN diode (15D), where reducing the mesa area size was not sufficient enough to compensate for the increase of the junction capacitance value due to thinning of the absorber thickness. Such an APD and PIN diode with junction capacitances of 72fF and 106fF respectively and relatively small series resistances should be suitable candidates for data rate applications higher than 25 Gb/s. The total variation of the breakdown voltage with the temperature was calculated for the standard and optimised APDs.

The findings show that the breakdown voltage of the optimised APD (15A) is less sensitive to the ambient temperature compared to the standard design (30A) (14.21mV/K vs 24.36mV/K).

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300 Measured standard APD (30A) Simulated standard APD (30A) 250 Simulated optimised (smaller) APD (15A)

200

V =-12.5V

150 PT

100

50 Junction Capacitance (fF) Capacitance Junction

0 -21 -18 -15 -12 -9 -6 -3 0 Voltage (V) (a) 250 Measured standard PIN diode (15S) Simulated standard PIN diode (15S) 200 Simulated optimised (smaller) PIN diode (15D)

150

100

50 Junction Capacitance (fF) Capacitance Junction

0 -10 -8 -6 -4 -2 0 Voltage (V) (b)

Figure ‎5.8: Measured and simulated dark junction capacitance versus bias of the standard and

optimised (a): In0.53Ga0.47As/In0.52Al0.48As APDs, and (b): In0.53Ga0.47As PIN photodetectors.

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5.8 Optical and Noise Characteristics

The photocurrents of the conventional devices were measured using a 1.55µm laser light. The incident optical powers on the APD and PIN diodes were -30dBm and -20dBm respectively. APD and PIN diode models were optically characterised by taking into account the generation process of the electron-hole pair in the absorber layer.

In SILVACO, different parameters have to be considered to calculate the photocurrent such as material quantum efficiency, material absorption coefficient, and the effect of absorption losses and transmission and reflection factors [254]. The optical-generation rate is calculated in SILVACO using the formula [254]:

푃∗ 휆 푂 − 퐺= Ƞ × × 훼(휆) × 푒−훼(휆)푦 (5.2) ℎ 푐

∗ where Ƞ is the quantum efficiency, 푃 represents the effect of absorption losses, and transmission and reflection factors, 푐 is the speed of light, 푦 is the optical penetration depth. Laser light was utilised of a wavelength of 1.55µm to generate electron-hole pairs in the absorption layer. The laser power was 1µW. In the photocurrent simulation process, the same models (SRH, Fermi-Dirac statistics, IMPACT SELBER, and FLDMOB) and fitting parameters of the dark current and C-V characteristics were used except that the electron velocity in the absorption and charge sheet layers were set to 2x107 cm/s and 5x107 cm/s respectively as the electric field is higher under light conditions. In [139], a Monte Carlo model was used to simulate the optical characteristics of the APD. In [9], it was shown that in thin multiplication regions and high electric fields, the electron could travel with a speed that is much higher than its saturation velocity. This was also confirmed in [167], where the carrier velocity used in the model was much higher than the saturation velocity for an APD with a 200nm InAlAs multiplication region. This concept was further explored in our model, where the charge sheet and Multiplication layers have a high electric field profile, as shown in figure 5.9. The electric field of the optimised APD is close to ~600kV/cm compared with a ~700kV/cm for the standard APD design, furthermore, reducing the absorber thickness of the APD by ~60% led to an increase in the electric field of the absorber layer by 194%. A high electric field affects the drift saturation velocity of the carries and degrades the carrier transit frequency. So, care has to be taken to choose the optimum absorber thickness.

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800 Standard APD (30A) Optimised APD (15A) 700

600

500

400

300

Electric Field (KV/cm) ElectricField 200

100

0 0.0 0.3 0.6 0.9 1.2 1.5 1.8 2.1 2.4 Thickness (mm)

Figure ‎5.9: Simulated electric field distribution of the In0.53Ga0.47As/In0.52Al0.48As standard and optimised APDs under -20V bias.

The physical models are in excellent agreement with the fabricated devices, as is seen in the black and red lines of figure 5.10 (a) and (b). The optimised APD (15A) has virtually a flat photocurrent between 푉푃푇 and 푉퐵푅 which results in a practically constant internal gain. At 푉퐵푅, the photocurrent raises significantly due to the occurrence of a large number of impact ionisation events resulting in a high internal gain. The extracted 푉퐵푅 and 푉푃푇 from the photocurrent data agree well with the values from dark current and C-V data. The measured DC responsivity without anti-reflection (AR) coating layer is 9A/W at 90%푉퐵푅 bias for the standard APD (30A). On the other hand, the measured dc responsivities of the standard PIN diodes (15S, 20S, and 30S) were 0.67A/W, 0.73A/W, and 0.76A/W respectively at -5V bias. The optimised APD (15A) photocurrent is ~1.4µA at 90%푉퐵푅 corresponding to a multiplied DC responsivity of 1.4A/W.

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10-3 Measured standard APD (30A) Simulated standard APD (30A) 10-4 Simulated optimised (smaller) APD (15A)

10-5

10-6

VBR=-23.7V 10-7

10-8 Current(A)

10-9

-10 V =-12.5V 10 PT

10-11 -25 -20 -15 -10 -5 0 Voltage (V) (a)

Measured standard PIN diode (15S) Simulated standard PIN diode (15S) Simulated optimised (smaller) PIN diode (15D)

10-5 Current(A)

10-6 -5 -4 -3 -2 -1 0 Voltage (V) (b)

Figure ‎5.10: Measured and simulated photocurrents of the standard and optimised (a): In0.53Ga0.47As/In0.52Al0.48As APDs, and (b): In0.53Ga0.47As PIN diodes.

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The optimised APD could not maintain a high responsivity as a result of its thinner absorber thickness. On the other hand, the optimised PIN diode (15D) has a photocurrent of 4.6µA while offering a DC responsivity of 0.46A/W at -5V bias. At the same absorber thickness and incident optical power, the APD provides higher responsivity compared to the PIN diode due to the multiplication process of the photocurrent. The internal gain (푀) was calculated for the standard and optimised APDs (30A and 15A), as shown in figure 5.11. 60 7 Measured gain of standard APD (30A) Simulated gain of standard APD (30A) 6 50 Simulated gain of optimised APD (15A) F(M) of standard APD (30A) F(M) of optimised APD (15A) 5 40

4 30

3 F(M)

20

Internal Gain Internal (M) 2

10 1

0 0 -24 -21 -18 -15 -12 Voltage (V)

Figure ‎5.11: Measured and simulated internal gain and excess noise factor of the standard and optimised In0.53Ga0.47As/In0.52Al0.48As APDs.

The low internal gain of ~3 at 90%푉퐵푅 of the optimised APD is mainly caused by the decrease of the electric field in the multiplication layer. Under a uniform electric field, the excess noise factor 퐹(푀) as a function of the applied bias was estimated and plotted in figure 5.11.

The low internal gain introduced less excess noise of ~2.2 for the optimised APD (15A) at 90%푉퐵푅 bias, and 푀=~3, which is comparable to the value reported in [136, 264, 265]. Less excess noise is highly essential to achieve high SNR values. The photodetector noise is another figure of merit which determines the maximum achievable SNR and data-rate.

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The total noises of the standard and optimised photodetectors were theoretically calculated at 1Hz, as shown in table 5.6.

The noise characteristics are estimated at a bias of 90%푉퐵푅 and -5V for the standard and optimised APD and PIN diode respectively. In the case of a PIN diode, it is clear that the noise of standard and optimised PIN diodes (15S and 15D) is dominated by the thermal noise caused by the equivalent resistances of (푅푆+푅퐿). The existence of internal gain (푀) resulted in a high shot noise of ~37pA/√Hz for the standard APD (30A). However, this was reduced to ~2.8pA/√Hz for the optimised APD (15A) since 푀 was decreased to 3.

TABLE 5. 6: NOISE CHARACTERISTICS OF THE STANDARD AND OPTIMISED APDS AND PIN DIODES AT 90%푉퐵푅 BIAS

Parameter Standard Optimised Standard Optimised

APD (30A) APD (15A) PIN (15S) PIN (15D)

ID, nA 11 0.5 2 1.5

Iph, µA 9 1.4 6.7 4

Req, Ω 60 60 55 55

F(M) ~6 ~2.2 Free Free

Shot noise, pA/√Hz ~37 ~2.8 ~1.46 ~1.1

Thermal noise, ~16.5 ~16.6 ~17.3 ~17.3 (pA/√Hz)

Net noise, pA/√Hz ~53.5 ~19.3 ~18.76 ~18.4

푆 21 represents the optoelectric response and is given by the ratio of photocurrent to the

optical power in dB unit. 푆21 response of the APD and PIN photodetectors were simulated using the standard and optimised SILVACO models and then compared with

the measured ones. Figure 5.12 depicts the normalised measured and simulated frequency photo-response of the standard and optimised APDs (30A, and 15A), and PIN

diodes (15S, 15D) at 90%푉퐵푅 and -5V bias respectively. The 푆21 response of PIN diode was shifted by -20dB in order to separate it from the APD response. The measured 3-dB

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optoelectric bandwidth of the standard APD (30A) and PIN diode (15S) is 6.7GHz and 20GHz, which agrees well with simulated ones. The maximum optoelectric bandwidth of the optimised PIN diode (15D) is 35GHz which is comparable to the reported value in [155], though the light window aperture is 3 times smaller than in our design.

0 @ 90%VBR

B.W=21GHz

21 -10

Normalised S Normalised -20 @ -5V

B.W=35GHz

-30 0 10 20 30 40 Frequency (GHz)

Figure ‎5.12: Normalized 푺ퟐퟏ response of the In0.53Ga0.47As/In0.52Al0.48As APD and In0.53Ga0.47As PIN diode, (red and black are the measured and simulated standard APD (30A), blue is the simulated optimised APD (15A), green and brown lines refer to the measured and simulated standard PIN diode (15S), and purple is the simulated optimised PIN diode (15D)).

The simulated optoelectric bandwidth of the optimised APD (15A) is 21GHz and agrees well with the theoretically calculated one of 22.5GHz when the intrinsic region width is 0.7µm (0.5µm absorber layer, and 0.2µm multiplication layer). The standard and optimised APDs provide a gain-bandwidth product of ~220GHz and ~63GHz at 90%푉퐵푅 bias. The simulation process has shown that the bandwidth of the optimised APD (15A) decreased to ~14GHz when the applied bias reached -22V, as shown in figure 5.13.

199

25

20

15

10 3-dBBandwidth (GHz) 5 Measured standard APD (30A) Simulated standard APD (30A) Simulated optimised (smaller) APD (15A) 0 -18 -19 -20 -21 -22 Voltage (V)

Figure ‎5.13: Measured and simulated 3-dB Bandwidth versus bias of the standard and optimised

In0.53Ga0.47As/In0.52Al0.48As APDs.

This is mainly due to electron scattering from Г‎to‎L‎and‎X‎valleys‎at‎high‎electric‎fields.‎ As a result, electrons have higher effective mass and, thus, lower drift velocity. The fitting parameter (charge sheet layer electron velocity) was changed according to the gain values because of the field-velocity dependency. The electron velocity of charge sheet layer was adjusted to fit the simulated 3-dB bandwidth with the measured one at different bias according to the velocity overshot in the thin multiplication layer. For a gain of 5 to 10, the electron velocity in the absorption layer, and the multiplication layer were set to 2x107 cm/s, and 1x107 cm/s respectively. The effect of the velocity overshoot was applied on the electron velocity in the charge sheet layer. Therefore the electron velocity of the charge sheet layer was varied according to the applied bias, and was set to 9×106, 4.4×106, 2.5×107, 2.5×106, and 5.25×105 cm/s at -18, -19, -20, -21, and -22V bias respectively. The 3-dB bandwidth of the standard PIN diodes (20S and 30S) were also experimentally measured and found to be 17GHz and 14.5GHz respectively at -5V bias which makes them suitable candidates for applications with data rate exceeding 18Gb/s.

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5.9 Reported PIN Photodetectors

To date, various techniques and structures have been investigated to develop reliable and stable PIN photodetectors over optical fibre links with data rates exceeding 50Gbit/s. Table 5.7 summarises some of the reported PIN photodetectors using different absorber materials and thicknesses. Much attention has been paid to realise a high data rate PIN photodetectors based on InGaAs and germanium absorber due to their improved performances compared to other materials. The thick InGaAs PIN diodes reported in [160, 266] have shown very small dark currents and high responsivity of 0.05nA and ~1A/W at 1.55µm wavelength, respectively. As a comparison at the same absorber thickness of 4µm, the InGaAs diode in [129] demonstrated a very small dark current of 10nA compared to 195nA for the germanium one reported in [39]. Besides that, the InGaAs PIN photodetector exhibited higher responsivity (0.9A/W vs 0.6A/W) at 1.55µm wavelength, and larger 3-dB bandwidth (18GHz vs 13.5GHz), although, it was fabricated with a larger mesa area size (55µm vs 40µm). The reason is mainly due to the higher quantum efficiency and higher carrier velocity of the InGaAs material.

A large 3-dB bandwidth PIN photodetector (39GHz) based on germanium material was reported in [155]. The diode structure consisted of 0.307µm of germanium absorber and was fabricated with a 10µm diameter size. The photodetector was designed and realised with two mesa structure shaped to mitigate the effect of parasitic capacitance. Such a PIN photodetector with a dark current of >75nA at -2V bias is not a suitable candidate for high sensitivity receivers. Furthermore, the small diameter size (10µm) could result in in- flexible alignment tolerance with the fibre optic. The work did not report the measured responsivity which is believed to be very low as a result of the very narrow absorber layer and more importantly the low quantum efficiency of germanium at the 1.55µm wavelength. Another PIN photodetector made of 0.33µm germanium absorber was reported in [267]. The device recorded a very wide-bandwidth of 49GHz at -2V bias and 1.55µm. Again, due to the relatively thin absorber layer, the device was found to have a high leakage current and very low responsivity of ~1.9µA and 0.05A/W respectively. Low responsivity limits the use of PIN diode in high sensitivity receivers. The device developed in [263] was made of a 1µm-InGaAs absorber and grown on a semi-insulating Si substrate. The photodetector recorded the smallest dark current for Si-InGaAs bonded wafer. However, the photodetector suffers from a high series‎resistance‎of‎40Ω due to the

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high resistivity of the substrate. Finally, the table also shows the main achievements of the standard and optimised PIN photodetectors, which have a very small dark current of 1.7nA and 1.4nA at -2V bias respectively, as well as wide 3-dB bandwidth compared to the reported works, based InGaAs and germanium materials.

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TABLE 5. 7: KEY REPORTED PIN PHOTODETECTOR PERFORMANCES

Diameter Absorber Structure Ɍ푷푰푵 (A/W) 푰풅풂풓풌−푷푰푵 (nA) 푭ퟑ풅푩 (GHz) Ref. (µm) Material Thickness (µm)

Undoped- Circular Mesa 70 1.8 0.9‎‎@‎λ=1.4µm‎ ~5 @ -2V 4 [153] InGaAs Undoped- Circular Mesa 50 2 ~1‎@‎λ=1.55µm‎ 0.01 @-2V 7.1 [160] InGaAs Circular Mesa 90 Undoped-Ge 4 0.9‎@‎λ=1.55µm ~1000 @ -2V 7.5 [39] Undoped- Circular Mesa 50 2.5 1‎@‎λ=1.55µm 0.04 @ -2V 10.3 [266] InGaAs Circular Mesa 40 Undoped-Ge 4 ~0.6‎@‎λ=1.55µm 195 @ -2V 13.5 [39] Undoped- Circular Mesa 35 1.5 ~1 @‎λ=1.55µm N/A 17 [268] InGaAs Undoped- Circular Mesa 55 4 0.9‎@‎λ=1.55µm ~10 @ -2V ~18 [129] InGaAs Undoped- Circular Mesa 30 1 N/A 0.1 @ -2V 21 [263] InGaAs Circular Mesa 20 Undoped-Ge 0.7 0.3‎@‎λ=1.55µm ~600 23.3 [41] Circular Mesa 10 Undoped-Ge 0.307 N/A >75 39 [155] Circular Mesa 10 Undoped-Ge 0.33 0.05‎@‎λ=1.55µm 1900 at -2V 49 [267] Circular Mesa Undoped- 35 2 0.7‎@‎λ=1.55µm 1.7 at -2V 20 This work (15S) InGaAs Circular Mesa Undoped- 25 0.5 0.46‎@‎λ=1.55µm 1.4 at -2V 35 This work (15D) InGaAs

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5.10 Reported APDs

In the same manner, many groups have focused on optimising the dynamic functions of APDs with the aim of increasing the gain-bandwidth product and maintaining low leakage current for high-sensitivity receivers. Table 5.8 lists most of the reported APD works for 1.33 to 1.55µm wavelength applications. Significant works have been presented in the literature showing the performances of high data-rate APDs for long- distance photo-detection over fibre optics [170]. Si and Ge materials were employed to detect light wavelengths of 1.3µm [37, 38, 157, 265, 269] and 1.55µm [37, 40]. Si material is well-known for its low 푘푟푎푡𝑖표 which is the key for low exess noise and high gain-bandwidth APD. Simple epi-layer structures were demonstrated in [265, 269] including a low-doped Si charge layer sitting between two intrinsic 1µm-Ge-absoption and 0.5µm-Si-multiplication layers. The APDs have a circular mesa shape with a diameter of 30µm which resulted in a fully depleted capacitance of 77fF in the case of [265]. The photodetectors recorded a maximum gain-bandwidth product of 340GHz and 840GHz at a light wavelength of 1.31µm. However, they have low sensitivity, low responsivity (~0.55A/W) and large dark current (~1µA) resulting from the narrow band gap of Ge (0.6eV) and the low absorption coefficient/quantum efficiency at longer wavelengths. Huang et al. [37] reported a 10Gb/s and 25Gb/s Si-Ge APDs in which the absorption layer thickness was 0.6µm. For the 10Gb/s design, a tensile strain Ge absorber with an improved absorption coefficient was utilised to improve the responsivity at 1.55µm wavelength. The device recorded the highest measured responsivity of 0.9A/W at unity gain. The large mesa area size of 35µm and short absorber limit the 3-dB bandwidth to ~7GHz and also led to the introduction of a large dark current of 3000nA at

0.9푉퐵푅. The 25Gb/s Si-Ge APD was realized by reducing the mesa area size to 20µm which also allow reducing the dark current to ~800nA at 90%푉퐵푅 bias. However, the large dark current limits the receiver sensitivity to -23.5dBm at 10−12 bit error rate.

The waveguide Si-Ge APD structure was also proposed in [131] with a maximum sensitivity of -16dBm and -25dBm at 25Gb/s and 12.5Gb/s data rate respectively. The device is capable of achieving a maximum gain-bandwidth product of 276GHz at a gain of ~12. However, the use of a p-doped Ge absorber layer resulted in a large dark current of ~400nA at 90%푉퐵푅 bias. Despite the great efforts performed to enhance the Si-Ge

204

APDs, they are still not the prefered candidates to be integrated in >25Gb/s receivers due to the limitations mentioned above.

The InGaAs/InAlAs APDs has gained much attraction for 1.33 to 1.55µm wavelength applications. For 10Gb/s data rate applications, a basic design of single circular mesa shape and thick undoped absorber layer APDs were reported in [55, 270-272]. The use of mesa diameter size of ~30µm and free doped thick InGaAs absorber (1 to1.3µm) in [55,

271, 272] resulted in having a small dark current of <23nA at 90%푉퐵푅. Moreover, the use of thinner InAlAs multiplication layer in [272] maximised the GBP to 240GHz compared to 140GHz in [271]. The maximum reported GBP (480GHz) was reported in [273]. The

Al0.85Ga0.15As0.56Sb0.44 material served as the multiplication layer. This material has the smallest temperature coefficient of breakdown voltage (휌푚=0.86 to 0.9mV/K) for a

0.11µm-thick [274, 275]. Another advantage is the small 푘푟푎푡𝑖표=0.08 to 0.1 which is highly preferred for high-gain and low excess noise factor APDs. However, the very thin absorber layer in [273] exhibited a high dark current of >1000nA which limits the maximum sensitivity of the receiver for high data rate applications. To reduce the dark current below 1nA, a graded p-doped InGaAs absorber APD was introduced in [276]. The small dark current and large 3dB-bandwidth makes the photodetector suitable enough for 25Gb/s receivers. For 25Gb/s receivers, APDs having three mesas configuration and a hybrid InGaAs absorber were demonstrated in [137, 169-172, 262, 277-279]. The triple mesas design was used to confine the electric field inside the multiplication regions while minimising it at the sidewall of photodetector. The hybrid absorber is composed of un-doped and p-doped InGaAs materials to maximise the bandwidth without degrading the responsivity. Moreover, the InAlAs multiplication layer was reduced to ~0.1µm to shorten the avalanche delay time. Though, a high dark current of 200 to 1000nA was measured at 90%푉퐵푅. In [170], two APDs with mesa area sizes of 20µm and 14µm respectively were demonstrated for 25Gb/s and 50Gb/s data rate applications. Both photodetectors were illuminated from the bottom-side while a mirror metal was attached to the top n-contact layer to double the effective length of the absorber layer and, thus, maximising the measured responsivity. The 14µm diameter- APD achieved a wide bandwidth of 35GHz at a low operating gain of 3 due to the build- up time limitation. Moreover, the photodetector was extremely leaky with a dark current of 3000nA at 90%푉퐵푅. The build-up time and large dark current limits the 50Gb/s receiver sensitivity to -10.8dBm at 10−12 bit error rate. Finally, the table also includes

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the performances of the standard and optimised APD structures studied in this chapter. The devices were designed with a very simple structure of one circular mesa area to ease the fabrication process and minimise the cost of production. The devices offered a very small dark current and wide 3-dB bandwidth for high-data rate applications of 10Gb/s and 25Gb/s. To conclude, most of the reported works focused on improving the gain- bandwidth product and the data rate of the APD. In practice, sensitivity and responsivity are the most important figure of merits when APDs are targeted for data rate >25Gb/s. Therefore, the dark current and the excess noise factor have to be reduced to their minimum levels by keeping the absorption and multiplication regions at reasonable thicknesses.

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TABLE 5. 8: REPORTED APD PERFORMANCES

Diameter Absorber materials Multiplication 푰풅풂풓풌−푨푷푫 (nA) Ɍ푨푷푫 (A/W) @ GBP Structure 푽푩푹(V) (푴) 푭ퟑ풅푩 (GHz) 푭(푴) Ref. (µm) material @ 0.9푽푩푹 푴=1 (GHz) Undoped Doped Circular mesa ~30 1µm, Ge N/A 700nm, i-Si 29.4 ~1000 ~15 ~0.3‎@‎λ=1.55µm 4 N/A 310 [40]

Circular mesa 35 Ge N/A i-Si 28.5 3000 12 0.9‎@‎λ=1.55µm 7 3 180 [37]

Circular mesa 30 1µm, Ge N/A 500nm, Si 25 ~1000 ~10 0.55‎@‎‎λ=1.3µm ~12 N/A 100 [38]

Circular mesa 30 1µm, Ge N/A 500nm i-Si 25 ~900 ~15 0.55‎@‎‎λ=1.3µm 13 ~3 340 [265]

Circular mesa 30 1µm, Ge N/A 500nm Si 24 ~1000 ~10 0.55‎@‎‎λ=1.3µm ~13 N/A 840 [269]

Circular Mesa 20 Ge N/A Si 18 400 8 0.7‎@‎‎λ=1.3µm 25 N/A N/A [157]

Circular mesa 20 0.6µm, Ge N/A Si 18.3 810 3.5 0.7‎@‎‎λ=1.3µm 34.5 N/A N/A [37]

0.4µm, p- Waveguide 4×10µm² N/A 100nm, Si 10 400 ~12 N/A 25 N/A 276 [131] Ge 1.3µm, p- Circular Mesa 35 N/A 200nm, InAlAs 28 160 20 0.88‎@‎λ=1.55µm 6-8 N/A 120 [280] InGaAs

Circular mesa 50 InGaAs N/A InAlAs 21 100 2 0.8‎@‎‎λ=1.55µm 7-8 N/A 130 [270]

Circular mesa 0.61 to 0.92 @ 35 1µm, InGaAs N/A 300nm, InP ~30 3 10 8.3 N/A 80 [281] with DBR λ=1.55µm

Circular mesa 32 1µm, InGaAs N/A 200nm, InAlAs 34 20 ~10 N/A ~9 N/A N/A [55]

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Circular Mesa 30 1.2µm, InGaAs N/A 200nm, InAlAs 29 19 ~10 0.95‎@‎‎λ=1.55µm 9 ~3.5 140 [271]

Circular mesa 12 0.55µm, InGaAs N/A 150nm, InAlAs 27 11.4 ~10 N/A ~11.8 N/A N/A [55]

Circular Mesa 30 1.3µm, InGaAs N/A 100nm, InAlAs 28 23 6.2 0.9‎@‎‎λ=1.55µm 11.8 3 240 [272]

100nm, Circular Mesa 20 0.3µm, InGaAs N/A 21 >1000 13 0.98‎@‎‎λ=1.55µm 12.5 N/A 480 [273] AlGaAsSb Circular triple 0.47µm, p- 30 0.33µm, InGaAs 88nm, InAlAs 23.5 ~200 5.8 0.6‎@‎‎λ=1.31µm 15 N/A 161 [137] Mesa InGaAs

Circular triple 0.47µm, p- 25 0.33µm, InGaAs 88nm, InAlAs 16.5 470 14.8 0.58‎@‎‎λ=1.55µm 17 N/A 410 [277] Mesa InGaAs

Circular triple 20 InGaAs p-InGaAs 100nm, InAlAs 26.5 400 ~10 N/A 18 N/A ~180 [278] mesa Circular triple 20 InGaAs p-InGaAs 100nm, InAlAs 26 ~600 15 0.91‎@‎‎λ=1.55µm 18.5 N/A 235 [170] Mesa 0.45, p- Circular mesa 16 N/A 150nm, InAlAs 17 0.5 4.4 N/A ~20 2 160 [276] InGaAs Circular triple 0.2µm, p- 20 0.4µm, InGaAs 90nm, InAlAs 30-35 ~700 ~10 N/A ~20 N/A N/A [172] Mesa InGaAs Circular triple 0.3µm, p- 14 0.3µm, InGaAs 90nm, InAlAs 30 2000 10 0.72‎@‎‎λ=1.55µm 20 2 to 4 270 [171] Mesa InGaAs

Circular two 10 0.4µm, InGaAs N/A 100nm, InAlAs 22.8 100 ~12 0.42‎@‎‎λ=1.55µm ~21 N/A 220 [262] mesa

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Circular triple 0.47µm, p- 30 0.33µm, InGaAs 88nm, InAlAs 23.5 ~200 3 0.6‎@‎‎λ=1.31µm 22.5 N/A 161 [137] Mesa InGaAs Circular triple 0.3µm, p- N/A 0.6µm, InGaAs 90nm, InAlAs 29 >500 5 0.7‎@‎‎λ=1.34µm 30 N/A N/A [169] Mesa InGaAs Circular triple 14 InGaAs p-InGaAs 90nm, InAlAs 31 ~3000 ~3 0.69‎@‎‎λ=1.55µm 35 N/A 270 [170] Mesa Circular mesa 50 1.2µm, InGaAs N/A 200nm, InAlAs ~23.7 ~8 ~7 0.75‎@‎‎λ=1.55µm 6.7 1.5 to 4 63 This work (30A) Circular mesa 25 0.5µm, InGaAs N/A 200nm, InAlAs ~23.7 ~1.2 ~3 0.6‎@‎‎λ=1.55µm 21 1.5 to 2 220 This work (15A)

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5.11 Summary

In0.53Ga0.47As/In0.52Al0.48As APD and In0.53Ga0.47As PIN diodes were fabricated with different light window aperture sizes and then tested to measure their DC and RF characteristics. The small-signal equivalent circuits of the standard photodetectors (APD 30A, PIN 30S, PIN 20S, PIN 15S) were built, and thus, the intrinsic parameters were extracted up to 40GHz. The standard PIN (15S) recorded the highest cut-off frequency of 694GHz‎ under‎ 50Ω load impedance and fully depleted bias. The work presented in chapter 5 presented physical models for the APD and PIN photodetectors, which can be easily exploited to predict the performances of different structures. The robust simulation in the Atlas SILVACO tool was utilised to build and model the virtual structures under dark and light conditions. A high-correlation was achieved between the measured and simulated electrical and optical characteristics for the standard devices. The structures were then virtually optimised by thinning down the absorber thickness and reducing the size of the light aperture window and anode contact width. The optimised PIN photodetector offered a DC responsivity of 0.46A/W at 1.55µm light wavelength. Moreover, the low noise characteristics and wide 3-dB bandwidth of 35GHz makes the device suitable for short-distance high data rate applications exceeding 50Gb/s.

Concomitantly, the optimised APD showed a very low noise of 19.3pA/√Hz with a maximum 3-dB bandwidth of 21GHz under 90%푉퐵푅 bias. The variation of the APD bandwidth with different bias up to -22V was examined in details. It was shown that the bandwidth decreases by ~33% if the bias goes beyond 90%푉퐵푅. Finally, the leading figure of merits of key published works was summarised and compared. For completeness, the tables also included the main achievements of the photodetectors fabricated and developed in this work.

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CHAPTER 6: CONCLUSION AND FUTURE WORKS

6.1 Conclusion

This final chapter concludes and highlights the primary outcomes of the research presented and then explains and discusses some suggested ideas and future works that could be considered to improve and develop the performances of the RF and optical detectors studied in this thesis.

6.1.1 Zero-Bias ASPAT Detectors and Mixers

A major part of the work conducted in this thesis concentrated on the development of zero-bias detectors and mixers based on a new type of tunnelling structure, the Asymmetrical Spacer Layer Tunnel (ASPAT) diode. Direct and heterodyne detection circuits are key elements, and their performances limit the sensitivity and noise characteristics of the transceiver systems. The tunnel ASPAT diode has shown exceptional characteristics due to its operational principle based on quantum mechanical tunnelling. The promising zero-bias operation and the temperature insensitivity features, in particular, paved the way towards building new RF ASPAT detectors that can compete with existing conventional RF detectors. The significant progress made in the MBE technique has led to achieving highly uniform and smooth monolayer films, enabling the reproducibility and manufacturability of ASPAT diodes.

Experimental studies of two ASPAT structures based on GaAs and InGaAs materials were undertaken. The devices were designed and fabricated into different mesa area sizes allowing for low and high cut-off frequency for possible microwave and mm-wave applications. The smaller mesa area size ASPAT diode fabricated in this work is 1.6×1.6µm² based GaAs/AlAs structure. The performances of the metal-semiconductor contact were experimentally evaluated by performing the TLM measurements of the GaAs and InGaAs structures. The high dopant concentration of the InGaAs layer offered better contact resistance and sheet resistance of 12.4Ω.µm2 and 5.9Ω/sq, respectively. Excellent uniformity was obtained in the I-V characteristics of the GaAs/AlAs and

In0.53Ga0.47As/AlAs ASPAT diodes at bias <0.75V. However, only one sample out of the measured 3.7×3.7µm2 GaAs/AlAs ASPAT diodes was shown to have a smaller current compared to the other samples, probably as a result of fabrication issues. The calculation

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of the current density was accomplished by taking into account the undercut process and light scattering issues, which considers the difference between the real fabricated diodes and the designed ones in the mask. This consideration has indicated that the real size of the GaAs/As ASPAT diodes are 1.6×1.6µm², 2.4×2.4µm², 3.7×3.7µm² and 5.8×5.8µm² instead of 2×2µm², 3×3µm², 4×4µm², and 6×6µm² respectively. Similarly, the mesa area sizes of the In0.53Ga0.47As/AlAs ASPAT diodes were found to be 3.75×3.75µm² and 5.85×5.85µm² instead of the designed 4×4µm², 6×6µm² respectively on the mask. The higher barrier of the In0.53Ga0.47As/AlAs ASPAT diodes contributed to reducing the leakage current density to ~0.0008mA/µm² at -1.5V bias, compared to 0.001mA/µm² recorded for the GaAs/AlAs ASPAT diodes.

To accurately evaluate the cut-off frequency of the GaAs/AlAs and In0.53Ga0.47As/AlAs ASPAT diodes, extraction procedures of the extrinsic and intrinsic equivalent circuit parameters were successfully performed at different bias with the help of the commercially available software ADS tool. The extraction process of the extrinsic parameters was carried out for different standard CPW structures aided by analytical equations used to calculate 퐶푃 and 퐿푃. The extracted 퐶푃 and 퐿푃 of the standard CPW structures were found to be ~18fF and 40 to 50pH, respectively. The size of CPW structures was then reduced by 60% to lower the parasitic capacitance and allow the design and fabrication of the smaller mesa area ASPAT diodes (1.6×1.6µm² and

2.4×2.4µm²). Accordingly, 퐶푝 was decreased to ~5fF. A momentum simulation tool from

ADS was used to study and investigate the variation of 퐶푝 of the standard and optimised CPW structures with respect to the variation of the substrate thickness. The simulation process showed a reduction in 퐶푝 of 60% when the substrate thickness was reduced to 5µm.

The equivalent circuit model of the ASPAT diodes was verified through excellent matching with actual S-parameter data obtained from on-wafer probing up to 40GHz. Two equivalent circuit models were introduced in this thesis to extract the parameters at negative and zero-bias accurately. The extracted series resistance and junction capacitance using the reverse bias equivalent circuit models were almost identical to those calculated using the equations expressed in chapter three. The zero-bias equivalent circuit model involved an additional non-linear resistance (푅푢) which accounted for the undepleted region of the thick spacer layer. The extracted 푅푢 had similar behaviour to the junction capacitance, where it increased with bias and badly increases the total series

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resistance of the devices and limited the maximum cut-off frequencies. Reducing the thick spacer thickness could help to minimise 푅푢, but this would increase 퐶퐽. An estimated cut-off frequency limit of 770GHz was deduced for the 1.6×1.6µm² GaAs/AlAs ASPAT diode at zero-bias.

By contrast, a maximum cut-off frequency of 500GHz was obtained from the

3.75×3.75µm² In0.53Ga0.47As/AlAs ASPAT diodes as compared with 330GHz for the 3.7×3.7µm² GaAs/AlAs ASPAT diode. Such diodes would be suitable candidates for implementing high-sensitivity detectors for >100GHz applications. To fully characterise and examine the non-linear characteristics of the ASPAT diodes, the junction resistances and curvature coefficients were calculated from the measured I-V characteristics. The calculated junction resistances were an excellent match to the extracted ones from the equivalent circuit model. As expected, a very high and unpractical junction resistance of >200kΩ‎ was‎ extracted for the smallest 1.6×1.6µm² GaAs/AlAs ASPAT at zero bias voltage. More surprisingly, the curvature coefficients seemed to be almost constant with mesa area size because the non-linear characteristics of tunnel diodes vary only with the barrier thickness as was reported for the backward diode [24]. The zero-bias curvature coefficient of the ASPAT diodes varied between 15 to 18V-1 corresponding to an unmatched voltage-sensitivity of 1800 to 1500V/W. Another crucial point which was investigated is the variation of the non-linear characteristics of the ASPAT diodes with the AlAs barrier thickness. Different structures with different AlAs barrier thickness (25.89Å, 19.5Å, and 13.1Å) were grown, and their junction resistance and curvature coefficient were calculated at zero-bias. A dramatic reduction in the junction resistance of 97% was achieved when the AlAs barrier thickness was decreased by 53%. The improvement in the junction resistance was at the expense of the curvature coefficient, which was reduced by ~50%, but the overall gain in performance was still worthwhile.

After the successful fabrication and characterisation of the discrete diodes, efforts were directed at the design and fabrication of integrated zero-bias RF detectors based on the 3.7×3.7µm², 5.8×5.8µm², and 10×10µm² GaAs/AlAs ASPAT diodes at X-band and K- band frequencies. This was followed by an assessment and evaluation of the mixing performances of devices in a 2nd subharmonic mixer circuits.

Matching circuits based optimised shorted stubs were designed and fabricated with a coplanar waveguide input port and then integrated with the ASPAT diode as well as output DC capacitor using MMIC technology. The aim of this work was also to keep the

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size of the circuits as small as possible but without degrading their performances. The size of the fabricated integrated detector circuits ranged from 0.5mm² to 3.8mm² at the [15 to 35GHz] and [4 to 18GHz] frequency bands, respectively. High uniformity of the output voltage was successfully obtained from the measurement of different integrated circuits located on different places on the wafer tile.

Owing to the optimised final design and fabrication processes, the maximum measured voltage sensitivity and the minimum calculated noise equivalent power of the zero-bias X-band detector based on 5.8×5.8µm2 GaAs/AlAs ASPAT diodes was ~1800 to 3650V/W and ~6pW/√Hz respectively, at -27dBm RF power. Similarly, the zero-bias 10×10µm2 GaAs/AlAs ASPAT detector produced a maximum voltage sensitivity and minimum noise equivalent power of ~1347V/W and ~11pW/√Hz at 9GHz and -27dBm RF power.

For K-band, the measured voltage sensitivity of the zero-bias 5.8×5.8µm2 GaAs/AlAs ASPAT detector was between 700 to 1300V/W and with a minimum noise equivalent power of ~20pW/√Hz at 24GHz. The detection characteristics of the GaAs/AlAs ASPAT diodes were also simulated and tested with bow-tie antennas at the mm-wave frequencies for possible car radar and imaging applications. Two wide-band antenna structures were designed and simulated with a maximum gain of 6.9dB and 3.5dB at resonant frequencies of 77GHz and 250GHz, respectively. The maximum simulated voltage sensitivity of the 77GHz and 250GHz zero-bias ASPAT detectors with bow-tie antennas was 340V/W and 1850V/W, respectively. These findings can be improved and optimised with the help of matching network to allow more power flow to the non-linear ASPAT diode.

Finally, down-conversion 2nd subharmonic mixers were modelled and simulated at an RF signal of 77GHz. A minimum conversion loss of ~10dB and ~16dB was achieved using the 3.7×3.7µm² GaAs/AlAs and 3.75×3.75µm² In0.53Ga0.47As/AlAs ASPAT diodes, respectively at 0dBm LO power. Both mixers showed a moderate 1-dB compression of - 13 to -12dBm RF input power. These achievements are comparable to published subharmonic mixers based on Schottky diodes.

In conclusion, ASPAT quantum tunnelling diodes present a promising solution for high- frequency applications with low power requirements, low noise, and functional efficiency, temperature-insensitivity and low cost. With further development and

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improvement of device sizes, non-linearity at low voltages, and matching circuits, it is expected that these types of tunnel diodes will play an ever-increasing role in future mm and sub-mm wave high sensitivity zero-bias detectors including 5G wireless and automotive car radars and general internet of Things (IoT) applications.

6.1.2 High-Data-Rate APD and PIN Photodetectors.

Detailed simulation and experimental investigation of avalanche and PIN photodetectors and their figures of merit were covered in this thesis, including responsivity, leakage current and breakdown voltage. The 3-dB bandwidth was shown to be limited by both the carrier transit frequency and RC frequency. The width of the absorption layer plays a crucial role in defining the junction capacitance and the quantum efficiency of the device. Analytical equations were used to accurately estimate the noise characteristics of the PIN and avalanche photodetectors.

Optimised 1.55µm wavelength In0.53Ga0.47As/In0.52Al0.48As avalanche and In0.53Ga0.47As PIN photodetectors were physically modelled using ATLAS SILVACO tool to investigate their electrical and optical performances. The devices were fabricated and then tested in terms of the DC and high-frequency characteristics at the University of Manchester facility. The small-signal equivalent circuit models were performed up to 40GHz for the standard APD and PIN diodes to extract their parameters and estimate their high-frequency capabilities. The simulated data, including dark and light DC and C- V characteristics, and frequency response obtained, agreed exceptionally well with measured data. The impact ionisation process and breakdown voltage (푉퐵푅=-23.7V) were successfully modelled using the FLDMOB and local field IMPACT SELBER models embedded in SILVACO tool. The room temperature measurement for the standard PIN and APD photodetectors I-V characteristic with a light window size of 15µm and 30µm showed a low dark current of 2.2nA and 8nA at -5V and 90%푉퐵푅 bias. The low leakage current obtained under fully-depleted condition is mostly the result of the thick and high- quality undoped absorber layers used. Moreover, the well-designed epi-layer structure of the APD employing the separated absorption, charge, and multiplication layers scheme mitigated the bandgap-discontinuity and provided high field-separation inside the photodetector under large reverse bias. The optimised PINs and APD with a light window size of 15µm exhibited much lower dark currents of <2nA and <1.5nA at -5V

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and 90%푉퐵푅 bias. The APD and PIN photodetectors were illuminated through a 1.55µm laser light with incident power of -30dBm and -20dBm respectively. The reduction in the multiplied DC responsivity of the APD was imparted to the decrease in the absorber thickness. Accordingly, the multiplied gain was reduced to ~3 at 90%푉퐵푅 bias.

Similarly, the optimised PIN diode had a lower DC responsivity compared to the standard design. The maximum optoelectric bandwidth of the optimised APD and PIN photodetector were 21GHz and 35GHz at 90%푉퐵푅 and -5V bias respectively. The obtained electrical and optical characteristics, as well as the calculated noise performances, are good enough for these devices to be used in the integration of 25Gb/s and 50Gb/s optical receivers. This successful physical model provides excellent quantitative predictions of the APD characteristics, which can be useful to further improve the device performances. This model represents a platform to design PINs and APDs operating at high data rates, e.g. 25Gb/s receiver systems and higher.

6.2 Suggested Ideas for Future Work

This research provided a thorough analysis and presented a wide-ranging study on characterising and designing of zero-bias integrated ASPAT detectors and high-data-rate APD and PIN photodetectors. However, there are still several possible avenues which can be considered to improve and develop this work in the future.

6.2.1 Millimetre-Wave Detection Circuits

Since various of GaAs/AlAs ASPAT diodes have been tested and validated in terms of their DC and RF characteristics and then implemented and integrated into microwave zero-bias detectors, the next step is to optimise those detectors to function efficiently at the mm-wave frequency band and beyond. The main objective is to demonstrate zero- bias and low power mm-wave direct and heterodyne detection circuits for wireless communication and high-resolution imaging applications. To achieve this goal, several possible ways have to be considered as clearly shown in the flow chart in figure 6.1 and explained as follow:

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Stage 1 Optimisation of ASPAT epi-layer structure using SILVACO tool

MBE growth of ASPAT diodes No

Fabrication of sub-micron ASPAT diodes and on- wafer characterisation

Are the performances suitable for mm-wave and sub-mm wave detector and mixer designs?

Yes Stage 2 Design and optimisation of broadband matching circuits and antennas

No Fabricating and testing passive components

Meet the requirements of mm-wave and sub-mm- wave receiver systems?

Yes Stage 3 Integration of mm-wave and sub-mm wave single element ASPAT detectors and mixers with antenna

Measurement and enhancement of single element THz detectors and mixers performances

Design and integration of two element ASPATs Stage 4 detector for higher performances

Integration and fabrication of array of mm- wave/THz ASPAT detectors

Figure ‎6.1: Flow chart of the future work of the zero-bias ASPAT detector for mm-wave and sub- mm-wave applications.

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1- Design and fabrication of sub-micron ASPAT diodes with smaller junction capacitance and high cut-off frequency up to 1THz. It has been shown that the cut-off frequency is constrained by the series resistance and most importantly, the non-linear resistance of the un-depleted spacer layer under zero and forward bias. Therefore, the next step would be the growth and fabrication of ASPAT diodes with spacer ratio of (20:1) to examine the variation of the non-linear resistance and junction capacitance with respect to the thickness of the spacer. It is necessary to conduct such a study for the GaAs/AlAs and InGaAs/AlAs ASPAT epi-layer structures to figure out the optimum spacer thickness, which gives the highest cut-off frequency and better non-linear characteristics. The noise characteristics of the discrete ASPAT devices should also be investigated and compared to the other reported diodes.

2- The other approach is to manipulate the ASPAT layer structure to improve the curvature coefficient and junction resistance. In chapter 3, an experimental investigation verified by several fabricated and tested GaAs/AlAs ASPAT diodes was conducted to study the variation of the non-linear characteristics with respect to AlAs barrier. This principle will be extended to examine other key layers of the ASPAT diodes. The measurement of ASPAT unmatched sensitivity at low frequency will allow estimating the exact values of curvature coefficient, which should be close to the extracted value from the DC characteristics. The work will be continued through employing a physical simulation tool such as SILVACO ATLAS to study and understand the behaviour of such tunnel diodes concerning the variation of the other layers (emitter or collector) of different material-based devices. The physical modelling is hugely beneficial before the costly production process.

3- Recently, the metamorphic buffer technique has attracted attention due to its applicability in the fabrication of different III-V devices on substrates with large lattice mismatch. The main advantage of such a technique is the use of a well-matured GaAs IC fabrication technology. Moreover, the‎ lower‎ cost‎ of‎ the‎ GaAs‎ substrate‎ and‎ it’s‎ availability in larger size compared to InP makes the growth of metamorphic

In0.53Ga0.47As/AlAs mASPAT attractive. At the time of writing the thesis, metamorphic

In0.53Ga0.47As/AlAs ASPAT structures have been grown on a GaAs platform and then fabricated into different mesa area sizes. The preliminary results showed that the curvature coefficients is ~14V-1 at zero-bias, which is close enough to value extracted for the standard In0.53Ga0.47As/AlAs ASPAT grown on InP substrate. The focus now is to

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perform the RF characterisation and extract the intrinsic parameters of the devices at different bias.

4- From the circuit perspective, optimising the matching networks and antennas would help to improve the total performances of the detector and mixer circuits. Alternative matching circuit based radial stubs or filters can be explored with the help of the momentum of method and FEM simulations embedded in the Keysight-ADS and EMPro commercial software.

5- More importantly, is the effect of the parasitic components associated with passive elements at high-frequency. The growth and fabrication of the future ASPAT diodes would be on a thinner substrate in the range of 100 to 150µm which would contribute to minimising the platform losses as it was verified by the simulation work done in chapter three.

6- The eventual aim of this work is the integration of ASPAT detector with a broadband antenna to realise a zero-bias and low noise mm/THz wave detection circuits. The use of CST studio and EMPro tools should be carried out to design different antenna structures such as Quasi-Yagi and dipole antennas with various resonance frequencies. Meanwhile, the designated 77GHz and 250GHz bow-tie antennas should be fabricated and tested in terms of the return loss and radiation gain. As future work, antennas can be designed with a reactive part that cancels the capacitive effect of ASPAT diode, and so, more power is delivered without the need of a complex matching network.

The ambition of this work is to push the sensitivity of the mm/THz waves ASPAT detector as high as possible and also to increase their dynamic range that would, in turn, improve the signal-to-noise ratio of the receiver system. One of the possible techniques is to incorporate two ASPAT diodes in parallel to enhance the generated DC voltage of the detector. Smaller mesa area devices are required to compensate for the increase of junction capacitance. Two devices in parallel reduce the total junction resistance of the detector and improve the speed response of the receiver.

Also, it is desirable to make the design of the matching circuit less complicated. The challenge is to produce tunnel diodes with similar DC and RF characteristics to mitigate any performance degradation. We already have proved the high precision of controlling the epi-layer growth through conducting different tunnelling structures. An array of THz integrated ASPAT detectors can be formed for applications in spatial power combining,

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massive MIMO communications and imaging arrays. Overall, it can be claimed that the performances obtained from the discrete ASPAT diodes and microwave detectors are eminently good to warrant pursuit the proposed ideas and justifying continuing research into tunnel ASPAT diode application in millimetre and sub-millimetre wave detection circuits.

6.2.2 Fabrication of the Optimised APD and PIN Photodetectors

The next step in APD/PIN research should be the fabrication and measurement of the optimised APD and PIN photodetectors for 25Gb/s and 50Gb/s data rate application, respectively. Distributed Bragg reflector layers can be buried at the bottom of the epi- layers structure of the diodes to reflect the light and enhance the DC responsivity. The multiplication gain of the optimised APD can be enhanced using amplification circuit such as the widely known trans-impedance amplifier (TIA). The latter would also help to enhance the DC responsivity of the PIN diode for longer distance high-data-rate applications. In terms of the physical model work, the analysis of the temperature dependency of the APD would be an interesting topic to pursue. From the design approach, the validated and proven physical models can be exploited to simulate the performances of different APD structures having different doping concentration and thickness. In particular, the charge sheet and multiplication layers as they dominantly control the field distribution across the structure. The key factor (푘푟푎푡𝑖표) of the avalanche multiplication process should also be investigated and optimised for lower excess noise.

Bandwidth improvement techniques should also be considered to enhance the 3-dB bandwidth of the proposed APD and PIN photodetectors. Among them is the passive peaking method utilising capacitive or inductive element in series or parallel with the photodetector. The technique was widely used and showed an improvement in the bandwidth by a factor of two [282-284].

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APPENDICES

APPENDIX-A: QFN Circuit

ASPAT diode

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APPENDIX-B: Lab View programme

A LabVIEW code was written by Dr Sexton to control the frequency and power of the input RF signal as well as viewing the output voltage versus the RF frequency, as shown below.

APPENDIX-C: Test Structure Used in the Mask

Below is the fabricated test structure in the mask as well as the measured and simulated

푆11 and 푆21 data.

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0

-5

-10

-15

dB -20

-25 Measured S11 Simulated S11 -30 Measured S21 Simulated S21 -35 0 10 20 30 40 Frequency (GHz)

APPENDIX-D: Measured and Simulated 푺ퟏퟏ of the Fabricated 30GHz ASPAT Detector with Open Stub Matching Network

0 Measured S11 Simulated S11

-5

dB -10

-15

-20 27 29 31 33 35 Frequency (GHz)

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