A WIDEBAND AIRPORT PSEUDOLITE ARCHITECTURE
FOR THE
LOCAL AREA AUGMENTATION SYSTEM
A dissertation presented to
the faculty of the
Fritz J. and Dolores H. Russ
College of Engineering and Technology
of
Ohio University
In partial fulfillment
of the requirements for the degree
Doctor of Philosophy
Sai Kiran
November 2003 This dissertation entitled
A WIDEBAND AIRPORT PSEUDOLITE ARCHITECTURE
FOR THE
LOCAL AREA AUGMENTATION SYSTEM
by
Sai Kiran
has been approved for the School of Electrical Engineering and Computer Science
and the Russ College of Engineering and Technology
Chris G. Bartone
Assistant Professor of Electrical Engineering
R. Dennis Irwin
Dean, Russ College of Engineering and Technology
ABSTRACT
KIRAN, SAI. Ph.D. November 2003. Electrical Engineering
A Wideband Airport Pseudolite Architecture for the Local Area Augmentation System (168 pp.)
Director of Dissertation: Chris G. Bartone
This dissertation documents the design, development, and field and flight testing of a WBAPL for integration into a prototype LAAS. One major area of risk in the LAAS
CAT II/III program is the unresolved issue of sufficient system availability. One feasible, low-cost, means of augmenting the GPS constellation for LAAS to enhance availability is by the incorporation of APLs. Critical issues that seek consideration in APL design are a low-cost solution to the near-far problem, effective mitigation of APL multipath at the
LGF reception sites, and a solution to the issue of measurement errors as a function of peak received signal power level. This dissertation details the development of a prototype
WBAPL within the framework of LAAS requirements, with the intent of resolving the aforementioned issues. The architecture includes a simple and novel method to facilitate rapid direct-WB signal acquisition, and details a cost-effective resolution to the power- bias problem. Results from laboratory tests to verify and characterize the power-induced measurement errors are described in the dissertation. Independent solutions to the power- bias problem at the ground and airborne segments were incorporated into the prototype
WBAPL architecture. The solution on the ground involves the employment of RF power- control techniques. With the aim of low-cost implementation, the solution adopted for the airborne segment relies on carrier-phase measurements as the aircraft approaches the
WBAPL transmission antenna. A time-differenced carrier-phase positioning algorithm that does not require real-time resolution of the unknown carrier-phase integer ambiguities is adopted. This differential CP approach is launched from a carrier- smoothed code based solution that is maintained from the beginning of the approach until the phase handover-point. A modification to the WBAPL single difference geometry matrix is incorporated into the TDCP algorithm. The proposed architecture was successfully flight-tested to demonstrate the feasibility of its incorporation into LAAS, the results of which are presented in the dissertation. The performance of the prototype
WBAPL-inclusive LAAS is gauged in terms of the accuracy of the differential positioning solution. The integration of the WBAPL into the prototype LAAS provided an additional ranging measurement, and increased system availability.
Approved: Chris G. Bartone
Assistant Professor of Electrical Engineering
ACKNOWLEDGEMENTS
The research documented in this dissertation was funded by the Federal Aviation
Administration (FAA) under the Aviation Research Cooperative Agreement 98-G-002.
First and foremost, the author would like to thank Dr. Chris Bartone, the director of this dissertation, for his thoughtful guidance, unceasing support, and friendship over the years. The author has tremendously benefited from his collaboration with Dr. Bartone, and for that he expresses his utmost gratitude.
Dr. Frank van Graas is thanked for his contribution to this research effort, and also for serving on the dissertation committee. The author is indebted to Dr. van Graas for providing the initial opportunity that introduced him to world of avionics research. Dr. van Graas’ dedication to the scientific method, work ethic, and positive attitude continue to serve as an inspiration to the author, and he considers himself fortunate to have come across such a role model early on in his career.
Dr. Michael Braasch and Dr. Jeff Dill of the School of Electrical Engineering and
Computer Science, and Dr. James Fales of the Department of Industrial Technology are thanked for serving on the dissertation committee, and for their valuable suggestions toward improving this dissertation.
The author would like to thank Tom Arthur, Jeff Dickman, Lukas Marti, Sidharth Nair, and Ranjeet Shetty for their assistance in the flight test efforts. The author has benefited from his numerous discussions with Dr. Maarten Uijt de Haag, Dr. Andrey Soloviev, and
Lukas Marti, for which he is thankful. Jeff Dickman provided the antenna gain and D/U plots that appear in Chapter 4, and Sanjeev Gunawardena developed the software to generate the APL pulsing sequence used for this research.
The author thanks the enthusiastic flight-crew of Dr. Richard McFarland and Bryan
Branham for the several hours of excellent piloting, and extends a special thanks to Jay
Clark, the Chief of Airborne and Mobile Laboratories at the Avionics Engineering
Center, for his industrious support of the flight-test efforts.
Finally, the author expresses his heartfelt gratitude to his family and friends for making the journey such a gratifying one. vii
TABLE OF CONTENTS
LIST OF FIGURES ...... ix
LIST OF TABLES...... xii
ABBREVIATIONS ...... xiii
1. INTRODUCTION ...... 16
2. BACKGROUND ...... 21 2.1 The Global Positioning System ...... 21 2.1.1 The Concept of Multilateration...... 21 2.1.2 Errors in GPS Positioning...... 23 2.1.3 Ranging Errors, Their Sources and Effects...... 25 2.1.4 The Influence of Geometry...... 29 2.2 Overview of LAAS...... 33 2.2.1 Precision Approach, Profile and Requirements...... 33 2.2.2 The Basic Principle of LAAS Operation ...... 37 2.3 A Historical Perspective on the Development of PLs for Civil Aviation Applications ...... 39 2.4 Scope of the Dissertation ...... 48
3. APL DESIGN CONSIDERATIONS...... 50 3.1 LAAS Availability Considerations...... 50 3.2 Design Constraints for APLs within LAAS...... 53 3.2.1 The Near-Far Problem ...... 55 3.2.2 Multipath at the Ground Station ...... 60 3.2.3 High-Power Induced Measurement Errors ...... 61 3.2.4 Airframe Multipath and APL Line-of-Sight Issues ...... 62
4. THE OHIO UNIVERSITY PROTOTYPE WBAPL ARCHITECTURE ...... 64 4.1 The Prototype Ground Subsystem ...... 64 4.1.1 Ground Reception Antennas...... 66 4.1.2 APL Transmission Antenna...... 73 4.1.3 LGF Equipment ...... 79 4.1.3.1 APL Signal Generation...... 80 4.1.3.2 Pulse Blanking and Gain Control ...... 82 4.1.4 LAAS Ground Processing...... 84 4.1.4.1 Determination of the WBAPL Transmitter Clock Offset...... 88 4.2 The Prototype Airborne Subsystem...... 90 4.2.1 LAAS Airborne Processing...... 91 4.2.1.1 CSC Solution...... 92 4.2.1.2 Position Propagation Using a Time-Differenced Differential CP (TDCP) Approach...... 92 viii
4.2.1.3 SD Geometry Correction for the APL ...... 95 4.2.1.4 Differential Tropospheric Correction for the WBAPL...... 98
5. LABORATORY TESTS TO CHARACTERIZE POWER-INDUCED MEASUREMENT ERRORS FOR APL RECEIVERS ...... 100 5.1 Experiment Goal...... 100 5.2 Measurement Model Description...... 101 5.3 Laboratory Test Set-Up...... 105 5.4 Test Results...... 108 5.5 Conclusions from the Laboratory Tests...... 113
6. FLIGHT TEST SET-UP AND RESULTS ...... 116 6.1 Modification of the Ohio University Prototype LAAS Airborne Subsystem for the Flight-Test...... 116 6.2 Determination of the Truth Reference ...... 121 6.3 The Flight Profile...... 126 6.4 Results of the Power-Induced Measurement Error Investigation...... 127 6.4.1 Ground Subsystem Performance...... 128 6.4.2 Airborne Subsystem Performance ...... 132 6.5 Differential Positioning Accuracy Assessment ...... 139 6.5.1 Performance of the Ohio University Prototype WBAPL LAAS Ground Subsystem ...... 139 6.5.2 Performance of the Ohio University Prototype WBAPL LAAS Airborne Subsystem ...... 141 6.5.2.1 Data Analysis Using all Available SVs (Mask Angle = 5 deg)...... 142 6.5.2.2 Data Analysis Using a Reduced GPS SV set (Mask Angle = 20 deg) 150
7. SUMMARY AND RECOMMENDATIONS...... 157
REFERENCES ...... 161 ix
LIST OF FIGURES
Figure 2.1 Locus of a Receiver at the intersection of circles...... 22 Figure 2.2 Intersection with uncertainty ...... 24 Figure 2.3 Intersection of circles ...... 25 Figure 2.4 Precision Approach Geometry ...... 34 Figure 2.5 Conceptual Overview of the Baseline LAAS...... 38 Figure 3.1 LAAS Augmented with an APL...... 55 Figure 3.2 Frequency spectrum of the GPS C/A-code ...... 57 Figure 4.1 Block diagram of the ground segment of the Ohio University prototype WBAPL-inclusive LAAS ...... 64 Figure 4.2 Layout of the LGF Site Configuration at UNI ...... 65 Figure 4.3 Geometric profile for the determination of the angles of incidence and reflection ...... 67 Figure 4.4 The elevation gain pattern of the dBS200 MLA (Serial #24), used for GPS/APL reception at the LGF ...... 70 Figure 4.5 GPS D/U ratio of the dBS200 MLA (Serial #24), used for GPS/APL reception at the LGF ...... 71 Figure 4.6 Typical elevation gain pattern of the HZAs used for GPS/APL reception at the LGF ...... 72 Figure 4.7 Typical GPS D/U of the HZAs used for GPS/APL reception at the LGF ...... 73 Figure 4.8 The elevation gain pattern of the 20-element WBAPL MLA (Ser #1) as compared to the 14-element MLA (Ser #24)...... 76 Figure 4.9 The GPS D/U of the 20-element WBAPL MLA used for transmission ...... 77 Figure 4.10 LGF set-up for two-site Ohio University prototype WBAPL configuration 80 Figure 4.11 WBAPL Tranmission Subsection...... 82 Figure 4.12 Illustration of the WBAPL blanker and MGC for one antenna site...... 84 Figure 4.13 Generic block diagram of Ohio University prototype WBAPL airborne set-up ...... 91 Figure 4.14 SD geometry for an APL...... 96 Figure 5.1 Laboratory experiment set-up for the characterization of power-dependent measurement errors...... 106 Figure 5.2 PL PR DD errors obtained over the attenuation profile, with GPS SV #27 used as the reference ...... 110 Figure 5.3 Carrier-to-noise ratios reported by the receivers in the airborne path for the PL and the reference SV (GPS SV #27)...... 111 Figure 5.4 PR DD errors obtained for GPS SV #15 over the attenuation profile, with GPS SV #27 used as the reference ...... 112 Figure 5.5 PL CP DD errors obtained over the attenuation profile, with GPS SV #27 used as the reference ...... 113 Figure 6.1 A photograph of the nose-mounted antenna on the DC-3 (with its radome open) ...... 117 Figure 6.2 Block diagram of the airborne segment used for the isolation of the power- induced error on WBAPL measurements ...... 118 x
Figure 6.3 The locations of the top and nose-mounted antennas, and the Honeywell IRS on the DC-3 (side-view)...... 120 Figure 6.4 The locations of the top and nose-mounted antennas, and the Honeywell IRS on the DC-3 (top-view)...... 121 Figure 6.5 Body-frame vectors to describe the locations of the two aircraft antennas... 123 Figure 6.6 Coordinate-frame vectors to descibe the locations of the two aircraft GPS/APL reception antennas...... 125 Figure 6.7 Ground track of the DC-3 on 03 March 2003 ...... 126 Figure 6.8 Ground track of the DC-3 on 04 March 2003 ...... 127 Figure 6.9 WBAPL PR DD errors using the HZA antennas from FLD5 and FLD6, with GPS SV #11 used as the reference...... 129 Figure 6.10 Sky-plot for the duration of the test conducted at UNI on March 03, 2003 130 Figure 6.11 WBAPL PR DD errors using the MLA antennas from FLD5 and FLD6, with GPS SV #26 used as the reference...... 131 Figure 6.12 WBAPL PR DD errors using the nose-mounted antenna and HZA at FLD6, with GPS SV #11 used as reference ...... 133 Figure 6.13 PR DD errors for GPS SV #28 using the nose-mounted antenna and HZA at FLD6, with GPS SV #11 as the reference ...... 134 Figure 6.14 WBAPL DD errors using the nose-mounted aircraft antenna and HZA at FLD6 utilizing raw PR measurements prior to the PHP and CP measurements post- PHP, with SV #11 used as the reference...... 135 Figure 6.15 WBAPL PR DD errors using the top-mounted antenna and HZA at FLD6, with GPS SV #11 as the reference...... 137 Figure 6.16 WBAPL DD errors using the top-mounted antenna and HZA at FLD6 utilizing raw PR measurements prior to the PHP and CP measurements post-PHP, with SV #11 used as the reference...... 138 Figure 6.17 Number of SVs used in the position solution for each of the ten approaches, for the analysis conducted using all the available ranging sources...... 142 Figure 6.18 DGPS-only position errors obtained using CSC architecture for the entire approach, for the complete GPS SV set case...... 143 Figure 6.19 DGPS/DAPL position errors obtained using CSC architecture for the entire approach, for the complete GPS SV set case...... 144 Figure 6.20 DGPS-only position errors obtained using the combination of CSC and TDCP architectures to complete an approach, for the complete GPS SV set case. 145 Figure 6.21 DGPS/DAPL position errors obtained using the combination of CSC and TDCP architectures to complete an approach, for the complete GPS SV set case. 146 Figure 6.22 Difference between the DGPS and DGSP/DAPL solutions for the complete SV set case, obtained using the combination of CSC and TDCP for each approach ...... 147 Figure 6.23 Typical VDOP Reduction Using WBAPL...... 148 Figure 6.24 Number of SVs used in the position solution for each of the ten approaches, for the analysis conducted using the reduced SV set...... 151 Figure 6.25 DGPS-only position errors obtained using CSC architecture for the entire approach, for the reduced GPS SV set case...... 152 Figure 6.26 DGPS/DAPL position errors obtained using CSC architecture for the entire approach, for the reduced GPS SV set case...... 153 xi
Figure 6.27 DGPS-only position errors obtained using the combination of CSC and TDCP architectures to complete an approach, for the reduced GPS SV set case... 154 Figure 6.28 DGPS/DAPL position errors obtained using the combination of CSC and TDCP architectures to complete an approach, for the complete GPS SV set case. 155 xii
LIST OF TABLES
Table 2.1 Approximate PR measuement error contributions for a stand-alone single- frequency (L1) GPS receiver ...... 29 Table 2.2 System Requirements for LAAS ...... 36 Table 4.1 Elevation angles of the incident and reflected APL signals for the Ohio University Prototype LAAS configuration ...... 69 Table 4.2 Summary of the APL D/U for MLAs and HZAs at the Ohio University LGF 78 Table 6.1 Summary of prototype WBAPL LAAS ground B-value performance obtained for both 03 and 04 March 2003 (computed over the test-duration of approximately 1.5 hours) ...... 140 Table 6.2 Summary of WBAPL-inclusive LAAS position solution performance for the full constellation case (mask angle = 5 deg)...... 150 Table 6.3 Summary of WBAPL-inclusive LAAS position solution performance for the reduced-constellation case (mask angle = 20 deg; data analyzed over last 7.0 to 0 nmi) ...... 156 xiii
ABBREVIATIONS
ADC Analog-to-Digital Converter
ADSB Automatic Dependant Surveillance-Broadcast
APL Airport Pseudolite
C/N0 Carrier-to-Noise Ratio
CAT Category
CORS Continuous Operating Reference Station
CP Carrier-Phase
CS Control Segment
CSC Carrier-Smoothed Code
CW Continuous Wave
D/U Desired-to-Undesired Ratio
DAPL Differential GPS/APL
DD Double Difference
DGPS Differential Global Positioning System
DOP Dilution of Precision
ECEF Earth-Centered-Earth-Fixed
ENU East-North-Up
EPROM Erasable Programmable Read-only Memory
EUROCAE European Organization for Civil Aviation Equipment
GBAS Ground-Based Augmentation System
GEOS Geostationary Satellites
GLONASS Global Orbiting Navigation Satellite System xiv
GNSS Global Navigation Satellite System
GPS Global Positioning System
GSV GPS Silicon Valley, Incorporated
HDOP Horizontal Dilution of Precision
HZA High Zenith Antenna
ICAO International Civil Aviation Organization
IEEE Institute of Electrical and Electronics Engineers, Incorporated
ILS Instrument Landing System
IMLA Integrated Multipath Limiting Antenna
ION Institute of Navigation
IRS Inertial Reference System
LAAS Local Area Augmentation System
LAT Lateral
LGF Local Area Augmentation System Ground Facility
MGC Manual Gain Control
MLA Multipath Limiting Antenna
NED North-East-Down nmi Nautical Miles
OS Operating System
PDOP Position Dilution of Precision
PHP Phase Handover Point
PL Pseudolite
PR Pseudorange xv
PT Performance Type
RF Radio Frequency
RTCM Radio Technical Commission for Maritime Services
SC Special Committee
SD Single Difference
SIS Signal-in-Space
SV Space Vehicle
TDCP Time-difference Differential Carrier-phase
TDD Time Double Difference
UERE User Equivalent Range Error
UNI Ohio University Airport
VAL Vertical Alert Limit
VDB VHF Data Broadcast
VDOP Vertical Dilution of Precision
VERT Vertical
VHF Very High Frequency
WAAS Wide Area Augmentation System
WBAPL Wideband Airport Pseudolite 16
1. INTRODUCTION
There has been a concerted effort on the part of the civil and military aviation community over the past couple of decades to transition toward a satellite-based navigation system that is capable of supporting all the phases of flight. Although the Global Positioning
System (GPS) is currently the leading candidate capable of supporting such a transition, the Russian Global Orbiting Navigation Satellite System (GLONASS) and the emerging
European Galileo are also viable candidates to support such a transition. The terminology used for such a navigation system with global coverage is Global Navigation Satellite
System (GNSS). If accompanied by a phasing down of existing navigation aids, a migration of such large magnitude will have to be justified in terms of improved performance and additional capabilities that offset the cost of replacement. Standalone
GPS offers the backbone to achieve the necessary performance, but enhancements to the system have to be incorporated in order to facilitate enroute navigation, precision approach and landing, and surface operations.
The United States Federal Aviation Administration (FAA), the aviation industry, the airline community, and several universities have undertaken development of the Local
Area Augmentation System (LAAS) as a coordinated effort, in order to enhance stand- alone GPS capability for terminal-area navigation. Based on the current Required
Navigation Parameters (RNP) specifications, the GPS constellation in its operating capability is determined to be incapable of providing the requisite system availability for
CAT II and III operations [Shively (2001), Corrigan et al. (1999), Kline et al. (1999)]. 17
One means of availability enhancement is the incorporation of Airport Pseudolites
(APLs) within LAAS.
Chapter 2 provides an introduction to positioning using GPS, and discusses factors that play a role in accuracies obtained using such positioning systems. The utility of differential GPS (DGPS) in improving the accuracy and integrity of GPS is presented, followed by a discussion of the limitations of the system, as it is currently operational. On presenting the role of APLs within LAAS, a historical perspective of the development of
APLs for civil aviation applications is provided, from which an idea can be gained on the various approaches that have been tried over the years to tackle the issues that present themselves in APL design and integration. Critical issues that seek consideration for APL design are a low-cost solution to the “near-far problem”, effective mitigation of APL multipath at the LAAS Ground Facility (LGF) reception sites, and a solution to the issue of pseudorange (PR) and carrier-phase (CP) measurement errors as a function of peak received signal power level.
As system augmentation using APLs might only selectively be required, it is imperative that APL design be modular such that seamless integration into LAAS can be accomplished when and where required. It is also important that the addition of the
APL(s) not significantly increase either the cost or the complexity of LAAS implementation. Chapter 3 details the design considerations for the various facets of the
APL subsystem. The single-most important issue in the design of APLs to date has been a cost-effective resolution of the near-far problem. Earlier pseudolite (PL) architectures 18 used one of the codes from the GPS C/A-code family for spectrum spreading, and relied on a combination of RF pulsing and/or frequency offsets to provide the necessary isolation for GPS signals. Frequency offsets seemed to bring in additional problems to the table, and the isolation obtained from the cross-correlation performance of C/A-codes was found to be insufficient for the degree of isolation sought. The architecture recommended by the RTCA therefore, which has come to be adopted for most part by
APL researchers worldwide, utilizes a wideband code for spectrum spreading in conjunction with RF pulsing in order to resolve the issue of interference to GPS signals.
Chapter 4 presents the Ohio University prototype Wideband APL (WBAPL) LAAS architecture, and discusses its various components. Benefits from the utilization of a wideband code include improved measurement accuracy, reduced multipath errors, and the ability to employ pulsing with very low duty-cycles. Also discussed in Chapter 4 are the APL signal generation equipment and transmission antenna, the LGF reception antennas, and the LAAS ground processor functions of the prototype architecture. The utilization of low duty-cycle pulsing needs to be compensated for with an increase in the peak transmitted signal power in order to meet the coverage volume requirements.
Signals with exceedingly high peak powers can cause pseudorange and carrier-phase measurement errors in certain GPS receivers [Kiran and Bartone (2003)].
The problems associated with the presence of power-induced measurement error need resolution in the case of APLs, as they are liable to cause fixed biases in the APL ground measurements, leading to biases in the corresponding PR correction determined at the 19
LGF. Non-common errors between the ground and air receivers are liable to produce errors in the final differential position solution. The solution to the problem was sought at the ground segment of the prototype architecture by employing power-control measures to limit the amount of APL power that LGF GPS/APL receivers are exposed to. The circuit used to perform this is described in Chapter 4. In order to implement a low-cost solution to the problem in the airborne section, a solution was sought that did not accompany any hardware modifications or additions. The solution adopted instead utilizes carrier-phase measurements, as the error on these measurements was determined to be a magnitude smaller than that on pseudorange measurements. A differential carrier- phase positioning algorithm that does not require real-time resolution of the unknown carrier-phase integer ambiguities is adopted, and is detailed in Chapter 4.
Laboratory tests were conducted in order to verify and characterize the power-induced measurement errors, and the test methodology, set-up, and results are described in
Chapter 5. The results from these tests contributed to the design of the Ohio University prototype WBAPL architecture, as described in Chapter 4.
The performance of the prototype WBAPL architecture was gauged using a series of flight-tests that included ten approaches performed on the 3rd and 4th of March, 2003. The results obtained from these tests are presented in Chapter 6. The performance of the prototype WBAPL-inclusive LAAS is gauged in terms of the accuracy of the differential positioning solution. The Ohio University prototype airborne software presents a real- time comparison between the differential position solutions obtained with and without the 20 inclusion of the WBAPL as part of the visible constellation set. The differential position results presented in Chapter 6 were obtained by post-processing the collected raw measurement data under two conditions: one that included the WBAPL along with all the visible SVs (with a mask angle of 5 degrees); and another that included the WBAPL with a reduced subset of the visible GPS constellation set, that was obtained by employing a mask angle of 20 degrees. Differential positioning errors were obtained by comparing the position solution with a truth reference that was obtained by differential kinematic post- processing of dual-frequency measurements. The statistics of the results obtained are presented in Chapter 6, along with VDOP plots that indicate the improvement in vertical geometry from the inclusion of APLs.
The work detailed in this document entails the design, development, and testing of a
WBAPL consistent with the guidelines and constraints associated with the augmentation of LAAS for enhancement of system availability. The feasibility of utilizing the proposed architecture is proven by its implementation and successful flight demonstration. 21
2. BACKGROUND
2.1 The Global Positioning System
GPS is a worldwide, all weather, radionavigation system that enables a user receiver to
determine its position, velocity, and time, using one-way ranging signals that emanate
from SVs. The system comprises three major components: the Space Segment that
consists of the SVs, the Control Segment (CS) that monitors and upkeeps the space
segment, and the User Segment that consists of a multitude of users. GPS operates on the
principle of multilateration, which utilizes range estimates between several source
locations and a user, in order to estimate the unknown location of the user.
2.1.1 The Concept of Multilateration
In order to illustrate multilateration, a two-dimensional case is considered first, as shown
in Figure 2.1. If a receiver is cognizant of the fact that it is at a distance r1 from a
reference point A, then all that can be deduced about the location of the receiver is that it
is located somewhere on a circle, the radius of which is given by r1. If now, additional information is made available to the receiver, in that it is also at a distance r2 from another reference point B (which is not co-located with A), then the receiver is also located on another circle of radius r2. The fact that the receiver has to satisfy both the conditions above makes it imperative that it lies at the intersection of the two circles. The incorporation of the additional information to the system has the effect of reducing the degree of uncertainty in the receiver’s location from a circle to one of two possible points. Making available the range information to a third reference location C would help eliminate the uncertainty in its position, as shown in Figure 2.1. Thus, ranges to three 22
reference points are sufficient to determine a receiver’s location in this two dimensional
case.
Possible Locations
r1 A B r v 2
B r2 C r1 A r3
Only Possible Location
Figure 2.1 Locus of a Receiver at the intersection of circles
For the three-dimensional equivalent, the locus of a receiver equidistant from a reference point would be a sphere. The knowledge of a second range, in this case, reduces the location uncertainty from that of a sphere to a circle. From this point forward, the problem is the same as that at the beginning of the two-dimensional case. Therefore, by incorporating one additional range measurement (in comparison to the two-dimensional case), it is possible for the receiver to determine it’s three-dimensional location.
In the case of GPS-based positioning, the velocity of electromagnetic wave propagation is used to estimate the range of a receiver from an SV. The transmitter-receiver range is determined by measuring the time of propagation of a signal from an SV to a receiver.
The signals transmitted by GPS SVs are encoded to facilitate accurate estimation of the transit-time of the signal at the user receiver. Since GPS receivers typically contain 23 inexpensive crystal oscillators, the receiver clock will have an offset with respect to GPS time. The ranging measurements made by a GPS receiver are therefore laden with an inherent bias due to this receiver clock offset, which affects the measurements to all the
SVs identically. The range-like measurements are, due to the presence of this common receiver clock offset, referred to as PRs. A GPS receiver uses an additional PR measurement in order to solve for its clock-offset along with the three unknown position coordinates, taking the minimum number of independent measurements required for a solution from three to four.
2.1.2 Errors in GPS Positioning
GPS signals traverse a distance of approximately 20,000 km before reaching the vicinity of the earth, and the assumption that the velocity of propagation remains constant all the way is not entirely accurate. Owing to this, and other reasons that will be discussed, the timing measurement made by a receiver has a certain measure of uncertainty, which corresponds to an uncertainty in the range between the satellite and the receiver. The perfect circles of Figure 2.1 therefore realistically look more like rings around the reference points, with the widths of the rings reflecting the uncertainty in the range information. The intersection point of the circles is also replaced by a region-of- intersection within which the receiver lies, as illustrated in Figure 2.2.
24
Figure 2.2 Intersection with uncertainty
It is therefore an obvious conclusion that the area of uncertainty depends on the accuracy of the range information. The locations of the transmitters and receivers were arbitrarily chosen in the above case. Keeping the ranging error fixed, the geometry of the transmitters with respect to the receiver is varied next, as shown in Figure 2.3. It can be seen that the shape and area of the region of intersection vary from that seen in Figure
2.2, and that the uncertainty in position is smaller in the illustration of Figure 2.3 than that in Figure 2.2. The accuracy of the position information obtained using multilateration is therefore a function of both the accuracy of the range information, and transmitter- receiver geometry.
25
Figure 2.3 Intersection of circles
2.1.3 Ranging Errors, Their Sources and Effects
Errors in the ranging measurement made by a GPS receiver to an SV signal largely arise from inaccurate determination of the signal transit time. This could be due to several reasons: inaccuracy in the receiver’s knowledge of the SV clock-offset with respect to
GPS-time, and also the exact location of the SV at the time of transmission; atmospheric effects that impact the propagation velocity of the signal; the receiver clock being unsynchronized to GPS-time; fundamental physical phenomenon that limit the ability of a receiver to make accurate timing measurements; the effect of reflected signals combining with the direct line-of-sight signal to create a distorted resultant, thereby limiting the accuracy of the measurement.
26
One of the main functions of the CS is to predict GPS SV ephemeris and clock errors. A
GPS receiver gains knowledge of the predicted satellite ephemeris and clock-offset by decoding the 50 Hz navigation message that is formed by the satellite using data from the
CS. These parameters are determined by the CS using a network of monitoring sites spread around the globe, and are uploaded to each of the SVs periodically. Given the methodology used by the CS to predict SV ephemeris parameters, the radial component of the SV position errors are typically smaller than the tangential and cross-track components [Zumberge and Bertiger (1996)]. It is the projection of the position errors on the line-of-sight vector that affect the ranging measurement. Given that a significant portion of the magnitude of the radial error would project on the line-of-sight vector for an earth-based user, the small magnitude of the radial error is highly beneficial. Satellite clock offsets, on the other hand, occur due to short-term drifts of the onboard atomic time standards, and the error in the broadcast correction will tend to increase with the age of the CS update [Kovach (2000)]. SV clock errors identically impact all receivers using a particular SV signal.
Signal propagation errors include those that result from the transit of the GPS signal through the ionosphere and the troposphere. The ionosphere has non-unity refractive index, which affects the velocity of signal propagation. The refractive index of the ionosphere varies not only along its various layers, but also as a function of the signal frequency. Due to the dispersive property of the ionosphere, the modulation on the signal is delayed by an amount similar to that by which the carrier frequency is advanced. Dual- frequency users of GPS can use the frequency dependence of the delay to mitigate its 27 effect. Single frequency L1 receivers use the Klobuchar model, the parameters of which are broadcast as part of the navigation message, in order to estimate and correct for about
50% of the uncompensated ionospheric delay [Misra and Enge (2001)]. Although there is a contribution to the residual error from the CS, it is sorted as a signal propagation error in the classification adopted here.
GPS signals also undergo refraction while traversing the troposphere, causing a delay that is a function of the path length and the refractive index of the gases and water vapor along the path. Tropospheric refractivity is significantly affected by variations in temperature, pressure, and humidity. Unlike the ionosphere, the medium is non- dispersive at L-band frequencies, and therefore the delays on the code and carrier modulations are identical. A significant portion of the troposphere-induced error can be eliminated using correction models, of which several are available [Spilker (1996)].
Additional measurement errors arise due to receiver-induced errors and multipath, and these are largely uncorrelated between receivers. Receiver-induced errors include inter- channel biases, antenna phase-center variations, and thermal noise. Inter-channel biases and antenna phase-center variations cause different signals to traverse paths of different lengths, leading to erroneous estimations of the receiver clock-offset and hardware delay.
The presence of electronic noise also causes inaccuracies in receiver measurements. The receiver’s major noise contribution occurs at the very first stage of amplification, since the signal level is weakest at this point.
28
The effect of a signal arriving at a receiver via multiple paths is referred to as the multipath problem. The multiple paths could either result from external reflections and diffractions, or internal reflections. The reflected signal is, in most cases, a delayed and attenuated version of the direct signal, causing the composite signal to differ in both magnitude and phase. Multipath has the effect of degrading the accuracy of the timing measurement, and the corresponding error is seen to be a function of the amplitude, delay, phase, and phase-rate of reflected signal with respect to the direct signal. The GPS signal has a certain degree of immunity against long delay multipath due to the incorporation of a relatively high-rate Code Division Multiple Access (CDMA) signal structure. The P-code, which has a shorter chip length, is more immune to longer-delay multipath than the C/A code. The influence of multipath on code-phase measurements is seen to be more substantial than that on the CP measurement.
Table 2.1 summarizes the typical contribution to the PR measurement error from each of the error components (to the nearest meter), as impacting a single-frequency (L1) receiver [Misra and Enge (2001), Table 4.3]. The numbers listed are based on the assumption that the Klobuchar model is applied to correct for the ionospheric delay, and that a reasonable model is applied for the tropospheric delay. The combined error, expressed in the range domain, is referred to as the User Equivalent Range Error (UERE).
29
Table 2.1 Approximate PR measuement error contributions for a stand-alone single-
frequency (L1) GPS receiver
Error Source 1-σ rms Range Error (m)
CS 3.0 (SV ephemeris and clock errors) Signal propagation errors 5.0 (Atmospheric modeling errors) Measurement errors 1.0 (Receiver noise and multipath)
UERE 6.0
2.1.4 The Influence of Geometry
The PR measurement made by a receiver to an SV (index k) can be modeled by
incorporating the error components discussed in Section 2.1.3 as
k k k k k k ρ = r + c (tr − t ) + T + I + ε ρ (2.1) where
k : is the true range between the SV and receiver antenna phase centers (m) r
c : the speed of electromagnetic wave propagation in free space (299,792,458 m/s)
: the receiver clock offset with respect to GPS time (s) tr
k : the SV clock error with respect to GPS time (s) t 30
k : the distance equivalent of tropospheric delay (m) T k : the distance equivalent of ionospheric code delay (m) I k : the receiver measurement error, inclusive of multipath error (m) ε ρ
By incorporating models for ionospheric and tropospheric delays, and by correcting for the SV clock-offset using the decoded navigation message, a user receiver can obtain a corrected PR measurement that can be denoted as
k k ~ k ρ c = r + c tr + ε ρ (2.2)
~ k where ε ρ denotes the combination of the residual errors and the receiver measurement errors. If the user position is denoted by x = (x, y, z) , and the SV position by x k = (x k , y k , z k ) , then the range between the user and SV k for the particular measurement instant would be
k k 2 k 2 k 2 r = (x − x) + (y − y) + (z − z) . (2.3)
Equation 2.2 is therefore of the form
k k 2 k 2 k 2 ~ k ρ c = (x − x) + (y − y) + (z − z) + b + ε ρ , (2.4) where b is the distance equivalent of the user clock bias. A GPS receiver solves for its position and clock offset using at least four PR measurements of the form described by
Equation 2.4. Since range is a non-linear function of receiver and transmitter locations, linearization of the PR equation set is necessary in order to solve for the receiver coordinates. One approach is to linearize the set about an initial estimate, and solve iteratively. The linearized system equation is given by [Leva et al. (1996)] ~ ∆ρ = H ∆x + ερ (2.5) 31 where
∆x : is the change in the user position and clock offset, and consists of four components: three position offsets of the user from the linearization point,
and one time bias offset from the value assumed in the linearization process,
∆ρ : the offset between the true PR values and those that correspond to the linearization point, and
H : the n × 4 matrix that consists of the partial derivatives of the PR with respect to the unknown quantities. The first three columns of the matrix
consist of unit vectors from the linearization point to SV locations, and the
fourth column consists of ones (partial derivative of PRs with respect to the
clock bias).
In the case of an over-determined solution, the least-squared method can be used to solve for ∆x as a function of ∆ρ using
T −1 T ∆x = (H H) H ∆ρ . (2.6)
Under the assumption that the measurements are unbiased, uncorrelated, and have a common standard deviation given by the UERE, the covariance of the position vector can be obtained as [Langley (1999)]
ˆ T −1 2 2 cov(x) = (H H) σ UERE = G σ UERE (2.7)
Equation 2.7 shows that the G matrix provides a linear relationship between PR and position errors solely based on the transmitter-receiver geometry. Parameters obtained from the G matrix relate the PR error to specific components of position and time bias errors, and are referred to as Dilution of Precision (DOP) parameters. For example, the 32 parameter Position Dilution of Precision (PDOP), which relates the 3-D ranging error to position error, is obtained as follows:
2 2 2 If σ x , σ y , and σ z denote the variances of the x , y , and z components of the position error, then from Equation 2.7,
2 2 2 2 2 2 σ x = G11 σ UERE , σ y = G22 σ UERE , σ z = G33 σ UERE . (2.8)
The standard deviation of the position error is then given by
2 2 2 σ 3D = σ x + σ y + σ z = σ UERE G11 + G22 + G33 = σ UERE PDOP (2.9) where
2 2 2 σ x + σ y + σ z PDOP = = G11 + G22 + G33 (2.10) σ UERE
On transforming the solution covariance matrix to a local-level coordinate frame,
Horizontal DOP (HDOP) and Vertical DOP (VDOP), parameters that characterize the quality of the horizontal and vertical position errors, respectively, can be obtained in a similar fashion [Langley (1999)]. It is apparent from Equation 2.9 that, in order to reduce position error, the transmitter-receiver geometry be such that DOP values are minimized.
Although detailed deduction is beyond the scope of this discussion, it can be shown that equal satellite spacing in the azimuth results in low HDOP, and the presence of satellites at extremities of elevation (near zenith, and about the local horizon) around the receiver results in low VDOP [Parkinson (1996)]. The DOP values are generally lower when more satellites are visible. 33
2.2 Overview of LAAS
DGPS is a technique used to improve the positioning accuracy of GPS by taking advantage of the spatial and temporal correlations in the ranging error components, while also incorporating means to verify the integrity of the signal in space. LAAS is the form of DGPS favored by the FAA in order to enhance the performance of GPS for the provision of Category (CAT) I precision approach, CAT II instrument approach, and
CAT III instrument approach and landing capability at airports in the United States [FRP
(2001)]. The intended operations not only include precision approach and landing, but also the support of parallel runway operations, airport surface movement, runway incursion warnings, missed approaches, and terrain avoidance. Although LAAS development has been proposed and supported by the FAA for implementation within the
US, coordination with international regulatory bodies such as the International Civil
Aviation Organization (ICAO) and the European Organization for Civil Aviation
Equipment (EUROCAE) indicates worldwide acceptance in the form of Ground-Based
Augmentation System (GBAS) to the Global Navigation Satellite System (GNSS).
2.2.1 Precision Approach, Profile and Requirements
LAAS would provide guidance function from 23 nmi of an airport through touchdown and rollout for the various categories of precision approach, which are based on the minimum decision heights depicted in Figure 2.4 [Skidmore (2000)]. The operation category chosen depends on the weather minimum, and the decision is based upon aircraft and ground equipage and certification. In the case of precision approach, both vertical and horizontal guidance are provided. The provision of vertical guidance is the 34 more challenging of the two for a satellite-based landing system, due to the higher
VDOP, as discussed in Section 2.1.4.
200' Runway Treshold (CAT I) e lop e S lid 100' 0 G 3 (CAT II) Glide Path Intercept Point (GPIP) 50' (CAT III)
816' 08' 3 ' 19 954
Figure 2.4 Precision Approach Geometry
The ability of a navigation aid to support a particular operation is determined based on four navigation system requirements: accuracy, integrity, continuity, and availability
[RTCA LAAS MASPS (1998)]. Accuracy refers to the compliance between the derived position of the sensor and the true position. Integrity relates to the level of trust that can be placed on the navigation information, and the corresponding requirement is expressed using three parameters: a position alert limit, a time-to-alert, and an allowable integrity risk. If the position error exceeds the corresponding alert-limit longer than the time-to- alert, it is referred to as a case of misleading information, and if the probability of its occurrence is not sufficiently low, then the integrity requirement is not met. The 35 continuity requirement sets limits on the probability of an unscheduled interruption in service occurring after the commencement of an operation. The allowable continuity risk is expressed in terms of a loss of continuity per unit time. Each occurrence of exceeding an alert-limit would be a continuity failure.
Availability describes the ability of the total system to provide the required guidance at the initiation of the intended operation. Performance simulations to determine the availability of the integrity and continuity functions involve determination of predicted protection limits, which are then compared to the corresponding alert-limit requirements to assess whether the current geometry and state of Signal-In-Space (SIS) support the required low Probability of Misleading Information (PMI), and also to assess confidence in the continuity capability of the protection levels. The probabilities assigned to the functions ultimately derive from the desired level of safety, and the requirements specified for LAAS were arrived at using ICAO-specified Instrument Landing System
(ILS) requirements as guidelines [RTCA LAAS MASPS (1998), Appendix D].
LAAS performance is classified in terms of a defined level of service called a
Performance Type (PT), and the levels of service required to support the various categories of precision approach and landing operations are discussed in [RTCA LAAS
MASPS (1998)]. The requirements for each of the PTs, as adopted for GNSS, are listed in Table 2.2 [RTCA LAAS MASPS (1998)]. Given that lateral accuracy requirements are relaxed in comparison with vertical for precision approach operations, and that GPS geometry enables better lateral position accuracies than vertical, it is typically more 36 challenging to meet the vertical requirements. Where not mentioned in the table, the requirements pertain to the vertical. As the RTCA recommended alert-limits increase in distance from the decision height, only the requirements at the decision height are specified in Table 2.2.
Table 2.2 System Requirements for LAAS
Minimum Performance Limits
Integrity Performance Accuracy Continuity (Allowable risk Availability Type (m) (Allowable risk) per approach)
PT 1 4.0 2×10-7 8×10-6/15 s
0.99 to PT 2 2.0 10-9 4×10-6/15 s 0.99999
2×10-6/30 s - lat PT 3 2.0 10-9 2×10-6/15 s – ver
The rationale for the performance specifications specified above can be found in
Reference [RTCA LAAS MASPS 1998, Appendices B through F]. In order to be accepted as a Sole-Means navigation aid for precision approach and landing operations, all four of the above performance parameters have to be successfully met, which stand- alone GPS is unable to, given the current constellation size and CS capabilities. Locally augmented GPS, however, was shown to be capable of providing sufficient accuracy and 37 integrity required of such a navigation aid [Braff (1995)]. Given the benefits of migrating to a satellite-based landing system, the Federal Radionavigation Plan calls for a gradual phase-down of the ILS with GPS/LAAS [FRP (2001)].
2.2.2 The Basic Principle of LAAS Operation
LAAS augments GPS by determining PR corrections at a LAAS Ground Facility (LGF), which are then transmitted on a Very High Frequency (VHF) Data Broadcast (VDB) for the benefit of differential users within the coverage volume. The PR corrections generated at the LGF are designated as scalar, since a single correction is generated for the cumulative, correlated errors on each of the ranging signals. The airborne subsystem utilizes the corrections received on the VDB to arrive at improved corrected-PR estimates
(Equation 2.2). The coverage volume and latency of application are kept within bounds so as to retain the benefits of the spatial and temporal correlations in the error components. LAAS is based on a carrier-smoothed code (CSC) architecture, whereby less precise code measurements are filtered using smoother CP measurements. The CSC architecture provides the sufficient accuracy and robustness desired for the stringent requirements of precision approach and automatic landing operations. The baseline
LAAS configuration is illustrated in Figure 2.5. The LGF consists of multiple GPS reception antennas, integrity monitoring capability, and VDB transmission facility. One of the logistical benefits of LAAS is that a single ground reference station is capable of servicing all the runways at an airport, and possibly also multiple closely spaced airports.
38
GPS SV GPS SV
Differential Corrections GPS Antennas VHF Antenna LAAS Airborne System LAAS Ground Subsystem Contained In Aircraft
Integrity VDB Monitor Transmitter
Figure 2.5 Conceptual Overview of the Baseline LAAS
Studies have shown that the availability of the integrity and continuity functions, as recommended by the RTCA, is not met by the current GPS constellation [Corrigan et al.
(1999), Kline et al. (1999), Shively (2002)]. The technology that has gained foremost attention as a candidate to provide the necessary augmentation comes in the form of ground-based PLs [Swider et al. (1997)]. PLs provide ranging signals that LAAS ground and airborne subsystems can incorporate in addition to the visible GPS SVs. Benefits are accrued in terms of improved geometry as a result of the low-elevation PL transmitter, and also due to improved integrity that results from the incorporation of an additional ranging source. 39
2.3 A Historical Perspective on the Development of PLs for Civil Aviation
Applications
The nomenclature Pseudolite derives from “Pseudo-Satellite”, and in the domain of satellite navigation has come to refer to ground-based transmissions of GPS-like ranging/timing signals. Nascency of the concept of PLs was synchronous with the initial development of GPS. Prior to the launch of the first GPS satellites in 1978, a system of solar-powered, synchronous, PL transmitters was utilized in order to verify operational compatibility between GPS transmitters and receivers [Parkinson (1995), Harrington and
Dolloff (1976)]. Subsequent to the establishment of a functional GPS constellation, the
PL concept gained usefulness as a technique to enhance performance for applications and scenarios wherein the accessible signal-set was inadequate.
The earliest revival of the PL concept was driven by the realization that differentially corrected civilian GPS signals had the capability of providing accuracies requisite for aviation applications. The potential for PLs to augment the GPS constellation, while also doubling as a means to transmit differential corrections was singled out as early as 1982
[Beser and Parkinson (1982)]. The need for such augmentation was primarily driven by periods of very poor geometry that resulted from the then 18-satellite constellation.
Another major concern was the impact of satellite outages on life-critical aviation applications. The potential that PL technology had in being a viable solution to the above problems, along with other benefits accrued, such as the capacity to support transmission of differential corrections, and as a solution to local shading problems, led to the formation of a Pseudolite Subcommittee during the establishment of the Radio Technical 40
Commission for Maritime Services (RTCM) Special Committee 104 (SC-104) for the development of standards and specifications for differential GPS, in 1983. Although the committee was formed under the aegis of the marine navigation community, close liaison with RTCA was maintained, as it was envisaged that PLs would eventually find application in terminal area navigation [Stansell (1986)]. The task for the subcommittee was to define the signal structure and message format for PLs, as part of a low-cost differential GPS system. The first definitive analysis demonstrating an improvement in
GPS performance from of the inclusion of PLs was published in 1984 [Klein and
Parkinson (1984)]. The same publication was also the first to outline potential PL signal structures that could meet the constraints faced by the RTCM pseudolite subcommittee.
The principal trade-off faced by PL signal designers was a solution to the “near-far” problem (which relates to interfering cross-correlation between PL and GPS signals, and is elaborated in Chapter 3) while at the same time maintaining compatibility with the
GPS signal structure. The cross-correlation performance obtained using the GPS C/A codes was insufficient to guarantee the signal isolation required to keep the high-powered
PL signal from interfering with nominal GPS reception when a receiver was in close proximity to the PL transmitter. Solutions to the near-far problem lay in obtaining signal diversity using one, or a combination, of either code, frequency, or time domain separation. In the interest of achieving the lowest implementation cost, the solution recommended by RTCM SC-104 relied on the utilization of time-division multiplexing, whereby PL signal transmissions would be pulsed [Stansell (1986)]. The recommendations, published in 1986, specified a pulsing duty cycle of 11%, and the 41 positions of the pulses were randomized sufficiently enough to prevent receivers from locking on to the pulse-pattern. This however did not prevent PL signals from interfering with each another, and so a minimum separation distance between PL transmitters was recommended.
Limitations in the capability of the RTCM signal specification to sufficiently resolve the near-far problem were later identified, resulting in the proposal of an alternate signal structure [Van Dierendonck (1990)]. The suggested changes were two-fold: first, in order to mitigate the near-far problem, a fixed offset of 30 kHz from the GPS L1 center frequency, and the utilization of pulses with duty-cycles lower than that recommended by
RTCM were proposed; and second, a solution to the issue of cross-correlation between multiple PL signals was sought by preventing multiple transmissions at the same time.
This was not difficult to achieve, given that PLs were to be synchronized to GPS time. By ensuring that each PL transmitter utilized its allotted time bin for transmission, and that the bins were sufficiently separated eliminated the need for minimum separation distance requirements between transmitters.
There was a misconception at the time that the pulsed nature of the above-proposed formats forbade receivers from making CP measurements to the PL signal. Proponents of systems based on CP measurements therefore sought to obtain the requisite signal isolation using frequency, instead of time-domain, isolation. One such architecture, proposed in 1992 by NAVSYS Corporation, involved the transmission of the PL signal on a frequency band that was nearly 50 MHz above that of the GPS band (in the earth-to- 42 space aeronautical navigation band) [Brown (1992)]. The proposal was also driven by the realization that a pulsed structure reduced the effectiveness of the PL as a high rate data link. This implementation, however, required the design and development of a wideband receiver capable of receiving both GPS signals and the out-of-band PL signals without introducing uncertainties due to group delays or phase variations.
Non-kinematic PL designs evolved to utilize all three modes of signal diversity. A design proposal by the Stanford Telecom group combined all three multiple access techniques: utilization of optimum codes, frequency offset to place the APL frequency at the first
GPS null, and a good pulsing scheme [Elrod et al (1994)]. Laboratory and flight tests were conducted to test the viability of the proposed architecture in the period between
1994 and 1996, with marginal success. Analysis also showed that the benefits of moving the APL broadcast frequency to the first GPS null were limited [McGraw (1994)], thereby leaving unresolved the issue of complete PL signal isolation from GPS.
A radically different approach utilizing PLs in a kinematic DGPS architecture was proposed by researchers at Stanford University in 1993 [Cohen et al. (1993)]. The architecture involved placing PLs, termed Integrity Beacons, at the beginning of an approach, and flying over them in order to resolve the integer cycle ambiguities for the
PLs as well as the GPS CP measurements by taking advantage of the large Doppler variation during the over-flight. On resolution of the cycle ambiguities, precise CP based ranging, with accuracies capable of supporting CAT III operation, could be used through touchdown. Since the airborne segment needed to track the integrity beacon for only a 43 short duration until the ambiguities were resolved, its coverage volume was small. The low transmission power used was therefore not liable to cause any cross-correlation problems with GPS signals. A modification to enable autonomous integrity monitoring by having the PL function as a redundant ranging source throughout the approach was also proposed [Pervan et al. (1994)]. There were several drawbacks to the architecture
[Swider et al. (1997)]: the requirement that PL transmitters were to be placed at the beginning of an approach, which in most cases would be situated outside airport property, led to logistical and security concerns; the architecture required four PLs to service a runway in either direction; a bottom-mounted aircraft antenna was necessary in order to track the below-horizon integrity beacon signal, which was an expensive modification beyond what was necessary for a DGPS architecture devoid of PLs; the resolution of ambiguities during the final approach segment did not bode well operationally for confidence in the continuity of approach; and the utilization of RAIM as the integrity monitor would lead to placement of significant certification burden on the avionics, which was against the preferred approach in line with ILS methodology.
As it became apparent that the limitations associated with the integrity beacon architecture, primarily the requirement to place equipment outside airport property, were unacceptable to the FAA, an alternate architecture using on-airport PLs was developed instead. The proposal involved the utilization of the differential CP between a pair of PLs that could be configured in one of two ways: either at the top and bottom of a tower
(Stacked APLs), or at the two ends of a runway (In-track APLs) [Lawrence et al. (1996)].
The phase difference between the two PL signals observed by an aircraft, analogous to 44 the ILS measurement, was to be used in conjunction with the GPS observables. As the near/far problem was relevant in this architecture, the APL signal was pulsed at L1.
Successful flight tests demonstrating the real-time performance of the In-track architecture were conducted in 1997 [Pervan et al. (1994)].
By 1995, there were four groups actively conducting research and development on PLs for precision approach applications: Stanford University, Stanford Telecom (in close collaboration with Dr. A. J. Van Dierendonck), Ohio University, and University FAF
Munich (supported by Stanford Telecom). Stanford University’s kinematic LAAS architecture called for a PL design with emphasis on the CP observable, whereas the other three groups focused on CSC PL architectures.
The measured success obtained from the designs proposed by Stanford Telecom and Van
Dierendonck [Elrod et al (1994), Van Dierendonck (1990)] led to the development of a prototype PL at Ohio University in 1996 that involved PL transmission at a frequency offset from the GPS L1 center frequency by 4 to 9 MHz. Pulsing was not utilized during the initial phases of this effort in order to isolate and study the APL error components
[Bartone (1996)]. The PL PR measurement at the user was to be differential corrected using corrections computed by the LGF. A limitation of the differentially corrected PL design was large multipath-induced errors at the ground station as the result of the direct signal from the ground-based APL arriving at the reception antennas at low elevation angles. The Multipath Limiting Antenna (MLA) was developed at Ohio University as part of this effort to serve as both ground reception antennas and as PL transmission 45 antennas, in order to mitigate multipath at the LGF [Thornberg et al. (2002)]. On successful characterization and mitigation of the ground-to-ground APL multipath for the on-L1 C/A signal structure, the on-L1 architecture using radio frequency (RF) pulsing was implemented at Ohio University. The utilization of a relatively high duty-cycle of
27.8% called for the development of techniques to control APL RF power at the LGF and user receivers, which were also implemented and tested [Bartone (1998)]. The corresponding flight-test results obtained indicated the feasibility of this architecture in meeting the requirements posed of successful APL designs for integration into LAAS.
PL research at University FAF Munich was tied in closely with the developments at
Stanford Telecommunications, and the work done there primarily involved flight-testing of APL designs, and research of various operational issues, such as APL time synchronization, transmitter location errors, tropospheric correction models, and ground station multipath. It was during flight-tests conducted there in 1997 that PR measurement errors due to receiver saturation effects from the reception of high-powered, pulsed APLs was first reported [Hein et al. (1997)]. It was concluded from the tests performed at
University FAF Munich that the near-far problem associated with PLs could satisfactorily be solved using only good codes and pulsing schemes, and that the frequency offset did provided additional isolation, but at the price of additional complications.
The FAA’s selection of a CSC approach over the kinematic alternative for the LAAS
CAT III architecture in 1997 played an important role in the direction of PL research hence. Some of the pertinent features of the preferred architecture were the minimization 46 of ground antenna separation, location of ground equipment on airport property, minimization of aircraft antennas, modularity of design to accommodate CAT I though
III operations, and the reliance on ground-based sources such as PLs in order to increase system availability [Swider et al. (1997), Braff (1997)]. Following the LAAS architectural selection of the FAA, the RTCA Special Committee for GPS Precision
Landing Guidance set-up an Airport Pseudolite Subgroup (SC-159 4a) in order to develop specifications and standards for the signal and data structure for the PL as part of
LAAS. The recommendations of the subgroup were included as Preliminary Pseudolite
Signal Specification within the LAAS Signal-in-Space Interface Control Document
[RTCA LAAS ICD (1998)]. The recommended signal structure was based on wideband codes for the APL on L1, while pulsing the RF at a duty-cycle of approximately 2.733%.
The pulse positions were randomized sufficiently to prevent receivers from locking on to the pattern. The pulse pattern was guaranteed to have at least one pulse per millisecond to appear continuous to the receiver post-correlation signal-processing segment. The resulting PL architecture is referred to as the WBAPL. Benefits from the migration to a wideband code were multifold, and included improved cross-correlation performance with the GPS C/A-codes, the ability to use very low duty-cycle pulses, improved accuracy from the increased chipping rate, reduced susceptibility to long-delay multipath, and diminished susceptibility to continuous-wave (CW) interference.
Following the formulation of the signal structure and data format, the FAA procured the first prototype APL transmitters, built by IntegriNautics of Palo Alto, CA [Cobb (1999)], compliant with the specifications, in early 1999. The effects of the RTCA recommended 47 signal specification on various aviation and non-aviation GPS receivers on the market were studied prior to field-testing the WBAPL signal format. This was sponsored by the
FAA and performed at the ARINC SITE Laboratory in San Diego, where it was concluded that the design posed minimal or no effect on the ability of GPS receivers to acquire and track the GPS signal [ARINC (1998)]. Researchers at the FAA William J.
Hughes Technical Center were the first to obtain flight data using a WBAPL in 1999
[Warburton et al. (1999)]. The data indicated that the average power obtained using the specifications were insufficient to service a range of 10 nmi, and recommended the utilization of a higher duty cycle.
It was at this point in the time-line that the research described in this dissertation was undertaken. The first real-time demonstration of the inclusion of a WBAPL into LAAS was demonstrated at Ohio University in Fall 2000 [Bartone and Kiran (2001)]. The concept of utilizing both a High Zenith Antenna (HZA) and MLA at the ground station in order to compute corrections for the APL, to avail them both to the airborne user, was introduced in this attempt. The increased benefit obtained from the utilization of an APL while having access to only a reduced GPS constellation was demonstrated using post- processing techniques. Studies on the impact of improved siting on system performance, incorporation of direct-wideband acquisition capability [Kiran and Bartone (2002)], development of a new high-performance WBAPL transmission antenna [Dickman and
Bartone (2001)], solution to mitigate the effects of receiver saturation, and analyses to determine the airborne subsystem performance were all conducted since at Ohio
University, and are elaborated in this document. 48
Other groups that currently perform research on PLs for precision approach and landing, and have published successful results to date have included the Toshiba Corporation in collaboration with the Electronic Navigation Research Institute, Japan [Suga et al.
(2001)]; and the Technical University of Braunschweig, Germany [Henzler and Weiser
(1999), Altmayer (2001)].
2.4 Scope of the Dissertation
The research described in this dissertation continues the APL developmental efforts described in Section 2.3, and seeks to make the following unique contributions toward the field.
• The sole utilization of wideband codes for the APL calls for a direct-wideband
acquisition technique that is robust to accommodate acquisition by the dynamic
airborne receiver. This research effort seeks to develop such a robust direct
wideband acquisition technique for LAAS applications.
• Flight-tests experiments using the RTCA recommended pulsing duty-cycle of
2.7% with a peak transmission power of +35 dBm have shown that obtained
coverage is inadequate [Warburton et al. (1999)]. Increasing the peak power
further could impact receiver operation significantly (as is shown in Chapter 5).
This calls for the testing of various pulsing duty-cycles and formats to balance the
need for sufficient power reception at the fringe of the coverage region, while
reducing the impact on GPS receivers.
• The presence of a power-dependent measurement error has been reported, as
mentioned in Section 2.3. The verification and subsequent characterization of 49
such an error is a critical issue the APL program. A power-dependent error in the
ground reception system is capable of causing fixed biases in the ground
measurements, leading to inconsistencies between measurements at different sites.
Such an error in the airborne segment could result in increasing errors as an
aircraft approaches the APL transmitter, and hence the airport, which would be an
undesirable prospect. The research undertaken here seeks to characterize such an
error, if existent.
• On the characterization of the power-dependent measurement error, a cost-
effective solution for the APL signal will be sought for both the ground and
airborne segments. It is important that the solution not be overly complex,
especially in the airborne platform.
• The research undertaken here seeks to develop a prototype WBAPL system,
which would involve the development of all the hardware and software
components that comprise the system. The developed components would be
integrated into a prototype LAAS. The software developmental effort involves the
modification of the Ohio University prototype LAAS software project to include
WBAPL specific functions.
• In order to verify the performance of the prototype WBAPL inclusive LAAS
configuration, this research effort will involve extensive flight-testing at various
stages of the WBAPL developmental effort. 50
3. APL DESIGN CONSIDERATIONS
3.1 LAAS Availability Considerations
The availability of a landing system is typically characterized by its long-term service availability, which is the probability that the service will be available at an arbitrary time.
Long-term service availability for LAAS is determined as the weighted-average of the conditional probabilities obtained by taking into consideration various constellation states, and the probabilities associated with being in each of them. A Markov Model used to determine various constellation state probabilities is described in Reference [Phlong and Elrod (1993)], and the constellation state probabilities corresponding to the current
GPS constellation, as widely used in availability analyses, are listed in Reference [LAAS
MASPS (1998), Section F3.2].
The issue of satisfactory fulfillment of LAAS availability requirements for PT 2 and PT
3, given the current GPS constellation, remains unresolved, and beyond the scope of this document. Large variability between various LAAS availability studies exists, which can be attributed to the following reasons:
• Flexibility in the navigation aid availability specification for a given airport
• Preliminary nature of the SIS Integrity Vertical Alert Limit (VAL) specifications
for LAAS PT 2 and PT 3
• Variability in GPS satellite reliability numbers
• Inconclusive on-field performance capability and siting guidelines associated with
APLs. 51
Service availability requirements are a function of various factors such as the density of traffic, weather, availability of alternate navigation aids, etc., which greatly vary from airport to airport for a given level of service. LAAS being a modular system lends flexibility in configuration for any given airport, as long as the operational objectives are met. The long-term service availability numbers specified by the RTCA reflect this by specifying an acceptable range of 0.99 to 0.99999 for the three LAAS Performance Types
(see Table 2.2). At a very minimum, LAAS availability at a given location should meet the availability supported by the existing ILS installation [RTCA LAAS MASPS (1998)].
The SIS integrity alert limit requirements for PT1 were modeled based on the methodology adopted for ILS CAT I, and yielded a VAL of 10 m. Given that the ILS and
LAAS function quite differently, and have different failure modes and error characteristics, the adaptation is not straightforward. The ILS CAT II and III integrity requirements were modeled using several different approaches, and consensus is yet to be formalized on what methodology will be adopted for PT2 and PT3 systems. The RTCA
LAAS MASPS lists a preliminary specification of PT2 and PT3 VAL of 5.3 m. Alternate
VAL values of 8.3 [Shively (2002)] and 10 m [Murphy (2002)] have been proposed, and are under consideration by the FAA [Shively (2002)]. Availability analyses based on the
RTCA specified VAL values conclude that the current GPS constellation would be unable to meet the desired availability requirements for LAAS PT2 and PT3 [Shively
(2001), Corrigan et al. (1999), Kline et al. (1999)]. Corrigan et al. conclude that APLs in conjunction with Wide Area Augmentation System (WAAS) Geostationary Satellites 52
(GEOS) would be necessary, whereas Shively concludes that availability could be met by either augmenting LAAS with an APL, or by migrating to dual-frequency ground and airborne platforms. Availability analyses performed using alternate higher values of VAL however indicates that no augmentation is necessary [Shively (2002)]. The resolution of
VAL levels is therefore necessary prior to arriving at concrete conclusions on the sufficiency of availability for PT 2 and PT 3 LAAS.
It has been proposed that availability studies be conducted with satellite failure rates that reflect observed performances, rather than the conservative estimates that have been used to date [Kline et al. (1999)]. This would show a marked improvement in the availability numbers, but reliance on such an approach should be marked with caution. Given that a significant number of GPS satellites are operating on single-string failure modes, it might not be judicious to extrapolate past-to-current performance observations to obtain long- term failure rates. Modifications in the satellite reliability numbers highly affect the results of availability analyses.
One key parameter that affects availability analyses is the APL airborne PR measurement error. The airborne error characteristics for APL signal reception by top-mounted antennas on wide-bodied aircraft have not been analyzed in detail to date. Availability studies have therefore used approximate signal models for the APL. The research work undertaken here tackles some of those issues by characterizing the APL airborne error.
Another important criterion regarding APLs is their proper siting on airport property.
APL placement will have to take into account optimization of geometry, while 53 simultaneously meeting RF constraints. Some of these RF constraints include minimization of multipath at the LGF, minimization of the distance between the LGF facility and the APL transmission antenna for logistical reasons, and maintenance of line- of-sight between the APL transmission antenna at the LGF and the top-mounted aircraft reception antenna from the start of the approach through touchdown and rollout. The above constraints will have to be met for all the runways that an APL services. It has been shown that significant improvement is obtained from the inclusion of an APL even when no attempt is made to optimize its location [Shively (2002)].
Alternate proposals to meet the desired availability requirements have called for an increase in the number of satellites in the GPS constellation, incorporation of WAAS
GEOS, and migration to a dual-frequency architecture using a second civilian signal. The sole incorporation of WAAS GEOS shows a slight improvement in the availability of the system, but not sufficient to meet the current availability specifications [Shively (2002)].
Despite the open debate on GPS availability with and without APL augmentation for the
LAAS, this research was conducted to reduce the risk in the LAAS development program, and assess the potential for successful WBAPL integration into LAAS for PT 2 and PT 3 installations.
3.2 Design Constraints for APLs within LAAS
In order to field a practical APL, it is desired that its addition to the LAAS not tremendously increase either the cost or the complexity of the resulting system. As APLs may only be needed at select locations, it is important that their designs be modular for 54 integration into a LAAS easily only when and where required. In order to achieve modularity and minimize complexity, it is necessary that the processing of the APL signal be similar to that of GPS, at the LGF as well as on the airborne platform. It is also important that the inclusion of APLs should in no significant way degrade the performance of participating and non-participating GPS receivers by inducing unwarranted electromagnetic interference.
Specific constraints and requirements for APLs were listed as part of the recommended
LAAS architecture, and include the following [Van Dierendonck (1997), Murphy and
Hartman (1997)]
• All APLs should be on airport property,
• APL coverage must extend to a radius of at least 10 nmi,
• The design should accommodate up to 4 APLs per airport,
• APLs are not slated to be used as means for relaying differential corrections,
• APL signals must undergo integrity checks similar to that of GPS signals, and
• APL signals should be received via a top-mounted aircraft antenna.
Figure 3.1 illustrates the LAAS architecture inclusive of APLs. GPS and APL signals would be received using the same LGF reception antennas, and the LAAS integrity monitor on the ground would perform integrity checks on the APL signal similar to those performed on GPS signals. Pseudorange corrections and integrity parameters for the APL would be relayed to the airborne segment via the VDB. An aircraft would use the same top-mounted GPS antenna to receive the APL signal as well. The driving factor in APL 55 design has been a cost-effective solution to the near-far problem, and minimization of the ground station multipath, while simultaneously meeting the design constraints listed above.
GPS SV GPS SV
VHF Antenna for VDB Reception Multiple GPS/APL Reception On-Airport Antennas Pseudolite VHF Transmission Differential Corrections Antenna Single Top-mounted LGF GPS/APL Airborne Reception Antenna LAAS VDL Ground Transmitter Processor
Figure 3.1 LAAS Augmented with an APL
3.2.1 The Near-Far Problem
The codes utilized for spread-spectrum modulation employed by GPS, namely the Gold codes, were primarily chosen on the basis of their correlation properties, and due to the need for a relatively large code-family. The correlation performance is of prime importance in order to facilitate precise ranging measurements, and also to enable the simultaneous transmission of all the signals at the same center-frequency. Assuming a zero-frequency offset between two signals, the cross-correlation performance can be 56 gauged by simply studying the properties of the Gold codes, which are known to take only three cross-correlation values [Misra and Enge (2001)]:
1 − β (n) β (n) − 2 { , , } N N N , (3.11) where
⎢ n+2 ⎥ ⎢ ⎥ ⎣ 2 ⎦ β (n) = 1+ 2 (3.12) and N is the is the length of the code sequence, which is 1023 in the case of the GPS C/A-
⎢n + 2⎥ n + 2 code. denotes the greatest integer smaller than . The integer n in Equation ⎣⎢ 2 ⎦⎥ 2
3.12 is obtained as
n = log 2 (N +1) . (3.13)
Although GPS SVs transmit signals at a common frequency, using identical chipping rates, Doppler effects due to line-of-sight dynamics between a user and an SV can cause signal frequency offsets. The complete cross-correlation picture therefore calls for an inquiry using the frequency domain construct of the signal. A brief overview of the frequency domain view of the GPS signal is therefore given in the following paragraph, before delving into the issue of cross-correlation with consideration to frequency offsets.
The overview is based on the discussion in Reference [Spilker (1996)].
An entire C/A-code consists of 1023 chips, and is transmitted at a chipping rate of 1.023
M-chips/sec, leading to the repetition of the one entire C/A-code every millisecond. Each bit of the 50 Hz GPS navigation data, which is 20 ms long, comprises 20 C/A-codes. The modulo-2 output from the addition of the navigation data and C/A-code is further 57 modulated on to a carrier at 1575.42 MHz using binary phase-shift keying. The frequency spectrum of the GPS signal therefore comprises all the above rates, as illustrated in
Figure 3.2.
1 KHz 1.023 MHz f 1575.42 MHz
Figure 3.2 Frequency spectrum of the GPS C/A-code
The envelope of the signal spectrum is a sinc function, with a null-to-null bandwidth of
2.046 MHz (due to the C/A-code modulation). Due to the coding scheme being repetitious, the frequency spectrum is not continuous, but comprises spectral lines separated by the C/A-code repetition rate of 1 kHz. Although the average tips of these spikes define the envelope of the sinc function, individual spikes could lie above or below the envelope, depending on the specific C/A-code pattern. The spikes are truly sinc humps with null-to-null bandwidths of 100 Hz (a contribution of the 50 Hz navigation data). The energy in a GPS signal lies entirely within these sinc humps. The cross- correlation between two signals may be thought of as the over-lap between two such spectra.
If the frequency difference between two spectra at the point of signal reception is an odd multiple of 500 Hz, then the spikes will not overlap, leading to a cross-correlation 58 minimum. If, on the other hand, the frequency difference is a multiple of 1 kHz, then the spikes will significantly overlap, leading to a cross-correlation maximum. The worst-case cross-correlation between two C/A codes is found to be -21.6 dB [Van Dierendonck
(1996)], while the average cross-correlation is approximately –30 dB [Spilker (1996)].
This however is based on the assumption that the powers of the two signals under consideration are similar.
Not accounting for variation in antenna pattern characteristics, the maximum variation in received GPS C/A-code signal power for earth-based receivers would be 2 dB [Misra and
Enge (2001), Table 8.2]. The utilization of a patch antenna to receive the signals extends this range to approximately 8 dB. GPS receivers are therefore typically designed with a low dynamic range requirement. PL power received by an airborne antenna, however, could have a dynamic range that far exceeds that of GPS, if the signal is not designed properly. If the PL power received at say 20 nmi from the transmitter equals that of minimum received GPS power (-160 dBW), then the power received at 0.2 nmi would be approximately 40 dB higher that of GPS received power. This however is based on the assumption that the PL signal structure is identical to that of the GPS C/A-code signal
(i.e., it is a continuous signal transmitted on L1).
The isolation obtained from the cross-correlation performance of Gold codes is insufficient to prevent an unpulsed C/A-code PL signal from jamming the reception of
GPS signals. Airborne LAAS receivers therefore are exposed to a scenario wherein the
GPS power remains constant, and the PL power increases inversely proportional to the 59 square of the distance from the PL transmitter. The problem, whereby reduction in transmitted power results in insufficient received power within the part of the coverage region far from the transmitter, and an increase in APL power results in interference to
GPS reception as the receiver approaches the APL transmitter is referred to as the
“Near/Far problem”. The potential interference problem from the PL signal level on GPS reception varies based on the receiver architecture, and the effects are elaborated in
References [Van Dierendonck (1996)] and [Spilker and Natali (1996)].
The utilization of wideband codes for the APL reduces the cross-correlation between the
APL and GPS signal. This provides an additional measure of isolation, and enables the reliable use of the L1 center frequency for the APL, simplifying receiver design for APL inclusion. However, depending on the power it is exposed to, a receiver could be affected by the presence of high-power APL signals. APL interference has the effect of reducing
C / N 0 of all the GPS signals, which, if driven below the receiver’s tracking threshold could cause it to lose lock of the signals. Reference [Ward (1996)] simulates the
Jammer/Signal ( J / S ) power that causes a receiver operation to approach its tracking threshold for various reception modes, and determines that for a carrier tracking loop of
18 Hz noise-bandwidth and a 20 m-sec prediction integration time, the wideband spread- spectrum signal power would have to be 33.4 dB stronger than that of the GPS signal.
The J / S increases to 43.3 dB if the carrier-tracking loop bandwidth is reduced to 2 Hz, which is possible if external velocity aiding is utilized. Reference [Van Dierendonck et al. (1997)] indicates that the utilization of a wideband spreading code for the APLs alone 60 provides an additional 10 dB of interference rejection when compared to the use of a
C/A-code PL. The above discussion is applicable when the APL signal is not pulsed.
A highly effective way of increasing GPS-APL isolation and mitigating the near-far problem for the WBAPL is by the utilization of RF pulsing. GPS receivers have a measure of built-in immunity to pulsed interference. The high power pulsed signal dominates the statistics of the receiver’s estimators while it is “ON”, but this action only takes place for a small percentage of time, given by the duty-cycle of the pulsed APL. In order to bound the effect of the high-powered pulse on the receiver front-end, a limiter is usually employed to clip the signal, in which case it is important that the receiver front- end is designed for fast recovery from the “pulsed state”. Tests to gauge the effect of pulsed RF on receiver tracking have shown that with duty cycles of 30% and less, a GPS receiver operation is not sensitive to pulsed interference, regardless of the power level
[Winer et al. (1996)].
3.2.2 Multipath at the Ground Station
The need to mitigate multipath at the LGF is a driving factor in the determination of the number of antennas used, and their configuration. The effectiveness of averaging measurements from several antennas to reduce multipath is dependant on the error being uncorrelated between the measurements. Siting of the LGF antennas would require a compromise between placing the antennas far enough from each other in order to ensure that multipath is uncorrelated between them, while at the same time limiting the distance between the antennas and the LGF processor, for cabling and other logistical reasons.
These are criteria taken into consideration from a ground multipath perspective. In order 61 to limit the influence of reflections from buildings and other large reflectors, the antennas have to be sited away from these obstructions.
The incorporation of APLs to augment LAAS also adds constraints on the siting criteria applied for the LGF antennas. Given the very low angle-of-arrival of the APL signal at the LGF reception antennas (approximately 0 degrees), the direct and ground-reflected signals arrive at a proximate region of the antenna radiation pattern. Also, APL multipath seen by the ground reception antennas could be stationary and therefore not average out over time. This leads to a potentially critical issue that needs resolution.
3.2.3 High-Power Induced Measurement Errors
It has been reported that certain GPS receivers exhibit PR and CP measurement errors as a function of the received signal’s peak power [Hein et al. (1997), Van Dierendonck
(1998)]. The verification and subsequent mitigation of the high-power induced error is a key factor in the fielding of APLs, as the dynamic power range of the APL signal for an airborne user will exceed that of GPS. The dynamic range for received APL power depends on the siting of the APL transmission antenna, the operational range supported, as well as the transmission and reception antenna radiation patterns. Analysis performed according to the methodology described in Dickman and Bartone (2001) indicates that a dynamic range of approximately 34 dB would be required of the airborne user receiver, based on the siting used at the prototype LGF at UNI, for a coverage volume corresponding to 20 nmi. As the airborne user approaches the airport, and therefore the
APL transmission antenna, the received APL signal power, and possibly the power- dependent error term, would increase. The presence of a power-induced error can cause a 62 fixed bias in the ground measurements of the APL, leading to a bias in the corresponding
PR correction determined at the LGF. Non-common high power-induced errors between the ground and air receivers are liable to produce errors in the final differential position solution.
Non-common delays in a particular signal reception path will produce measurement errors in the PR and CP measurements. Any non-linear effects as a function of power level exhibited by receiver components can produce group and phase delays for that particular signal path. Non-linear effects as a function of power level can be broken down into two general categories, based on which segment of the reception chain they occur in: pre-Analog-to-Digital Converter (ADC) and post-ADC errors. In the pre-ADC portion of the receiver, the non-linearity of the RF-mixer can produce variations in inter-modulation products and generate group delays, which have the potential to affect the post-ADC correlation peak. In the post-ADC portion of the receiver, saturation of the ADC will not likely produce non-linear effects, but the time-constant of the AGC circuit could contribute to correlation-peak distortion.
3.2.4 Airframe Multipath and APL Line-of-Sight Issues
The parameters that will guide the placement of the APL transmission antenna on an airport property include those based on the optimization of geometry, minimization of multipath at the LGF reception antennas, coverage of multiple runways, and signal visibility from the edge of coverage region through the touchdown point. It would, in practice, be an insurmountable task to optimize all of the above variables at a given location. Of primary concern is the fact that the placement of the transmitting antenna too 63 close to the runway centerline is liable to cause the aircraft body to mask the APL signal from its top-mounted GPS/APL reception antenna. Depending on the severity of the masking, which would vary as a function of the distance of the transmitting antenna from the runway centerline, the size of the aircraft, and the aircraft attitude, it is possible that either the airborne measurements could have strong multipath, or worse, the loss of line- of-sight between the APL transmission antenna and the top-mounted GPS/APL reception antenna could result in the receiver’s losing lock of the APL signal. Thus, proper siting of the APL transmission antenna with respect to the approach path is critical to its inclusion within LAAS. 64
4. THE OHIO UNIVERSITY PROTOTYPE WBAPL ARCHITECTURE
4.1 The Prototype Ground Subsystem
The Ohio University WBAPL prototype LAAS architecture includes modification to the baseline LAAS configuration [RTCA LAAS MASPS (1998)] solely in order to incorporate a WBAPL. The modifications were made keeping in mind that any additions to the LAAS system solely for the purpose of integrating APLs should not add significantly to either the costs or the complexity of the system. The various components of the prototype WBAPL-inclusive LAAS ground subsystem are shown in Figure 4.1.
VHF Antenna Reception APL Ground LAAS Antenna Power GPS/APL Ground Segment Control Receivers Processor
LGF Shelter VDB Transmitter
APL Trans. APL Signal Generator Antenna
Figure 4.1 Block diagram of the ground segment of the Ohio University prototype
WBAPL-inclusive LAAS
Figure 4.2 illustrates the LGF configuration at the Ohio University Airport (UNI), which consists of three reception sites at FLD5, FLD6, and FLD7, and one WBAPL transmission site at FLD8. Only two of the reception sites, FLD5 and FLD6, were used 65 for the data collection effort documented in this dissertation, due to hardware asset limitations. The reception antenna segment consists of several sites, each of which consist of an Integrated Multipath Limiting Antenna (IMLA) configuration (described in Section
4.1.1) which combines an MLA and an HZA in order to achieve good multipath performance over the whole elevation angle span of interest for LAAS. The distances between the antennas were arrived at primarily based on the separation that offers optimum ground multipath rejection, given the height of the antennas and their first nulls at approximately -3 degrees elevation angle [Bartone and van Graas (2000)]. The locations of all the sites were surveyed using Ashtech Z-12 GPS receivers during the installation of the LGF. The IMLA at the reception sites are mounted such that the bases of all the antennas are at approximately the same height.
25 80.2 m LGF Shelter 239.6 m FLD7 FLD6
80.1 m 80.1 m
253.0 m 79.9 m FLD5 FLD8 (APL Tx. Antenna)
Figure 4.2 Layout of the LGF Site Configuration at UNI
66
Each segment that comprises the prototype ground subsystem illustration in Figure 4.1 described in Section 4.1.1 through Section 4.1.4.1.
4.1.1 Ground Reception Antennas
The amplitude of the multipath signal relative to that of the direct signal impacts the severity of the multipath-induced error [Braasch (1992)]. Owing to the large distances between GPS satellites and LGF reception antennas, ground reflected GPS signals would enter an LGF antenna at a negative elevation angle that mirrors the positive angle of incidence of the direct signal. Given that the WBAPL transmission antenna would be located in the vicinity of the LGF reception antennas, and at comparable heights, the incident WBAPL signal would have a low grazing angle, but the incident and ground reflected signals could have dissimilar elevation angles. The effectiveness of an antenna in mitigating ground multipath can be expressed in terms of the ratio of the gains corresponding to the elevations of the direct and reflected signals. This is termed the desired-to-undesired (D/U) ratio for a particular antenna. The strong changes in gain patterns necessary to mitigate ground multipath are achievable with a vertically-polarized antenna array.
The following discussion details the determination of the D/U ratio for APLs. Consider a generic profile of two antennas with dissimilar heights, as shown in Figure 4.3. One of the antennas is used for transmission, and the other for reception. It is required to obtain the angle of the incidence of the desired signal (α ), and the angle of incidence of the undesired signal that corresponds to the ground reflection ( β ) in Figure 4.3. The heights 67
of the two antenna phase centers ( y1 and y2 ), and the distance between the antennas
( x1 + x2 ) are known. Let the sum of x1 and x2 be denoted as xT .
α β F y2 − y1 α 90 −θ A E
y 90 −θ 1 y2
β β B C D x x 1 2
Figure 4.3 Geometric profile for the determination of the angles of incidence and
reflection
The angle of incidence of the direct signal can be obtained from triangle AEF in Figure
4.3 as
−1 ⎛ y2 − y1 ⎞ α = tan ⎜ ⎟ (4.14) ⎝ xT ⎠
For the case when the two antennas have identical heights, y2 − y1 = 0 , leading to
α = 0 .
It is next desired to determine the angle of incidence of the ground reflected signal, which from Figure 4.3 is seen to be
−1 ⎛ y1 ⎞ −1 ⎛ y2 ⎞ β = tan ⎜ ⎟ = tan ⎜ ⎟ . (4.15) ⎝ x1 ⎠ ⎝ x2 ⎠ 68
The quantity x1 in Equation 4.15 is not known, and has to be determined from the configuration geometry. Consider the two triangles ABC and CDF . Since the angles of the two are identical, they are Similar Triangles, and so the following applies
AB BC y1 x1 = ; i.e, = (4.16) FD CD y2 x2
y1 x1 = (xT − x1 ) . (4.17) y2
The rearrangement of Equation 4.17 gives
x1 (y1 + y2 ) = xT y1 (4.18)
y1 ⎛ y1 + y2 ⎞ = ⎜ ⎟ . (4.19) x1 ⎝ xT ⎠
Application of Equation 4.19 in Equation 4.15 gives
−1 ⎛ y1 + y2 ⎞ β = tan ⎜ ⎟ . (4.20) ⎝ xT ⎠
For the case when the two antennas have identical heights, y1 = y2 = y ,
x x = x = T 1 2 2 , (4.21)
−1 ⎛ y ⎞ β = tan ⎜ ⎟ . (4.22) ⎝ xT / 2 ⎠
In order to determine the APL D/U ratio, the angles corresponding to the direct and reflected signals are first obtained from the siting geometry, and using Equation 4.14 and
Equation 4.20. The ratio of the gains at these two angles is then computed to yield the
APL D/U for that particular configuration.
69
The antennas are all mounted such that their phase centers are at approximately the same height from the ground. This height is approximately 4.2 m for the MLAs, and 5.1 m for the HZAs. The MLA at FLD8 is used for APL transmission, and the MLAs and HZAs at
FLD5 and FLD6 are utilized for reception. The distances between FLD5 and FLD8, and
FLD6 and FLD8 are approximately 80 m and 113 m, respectively. The incident and reflected signal angles for the APL signal are computed for the Ohio University prototype
LAAS configuration using Equation 4.14 and Equation 4.20, and are summarized in
Table 4.1. The gains at these angles are used to determine the APL D/U ratios, as discussed in the remainder of this section.
Table 4.1 Elevation angles of the incident and reflected APL signals for the Ohio
University Prototype LAAS configuration
FLD5 FLD6
MLA HZA MLA HZA
α (deg) 0 -0.64 0 -0.46
β (deg) -6.0 -6.6 -4.25 -4.7
The MLAs used for reception at the prototype LGF at Ohio University are the 14-element dBS200 antennas. The elevation gain pattern of a particular unit (Serial #24) is shown in
Figure 4.4. The antenna has good gain between 5 and 40 degrees. Typically, an elevation cutoff angle of 5 degrees is applied to GPS signals. The sharp roll-off in the gain pattern about 0 degrees is necessary in order to reject ground multipath signals. The pattern 70 shows a gain hump between 5 and 20 degrees, which is good for the reception of low- elevation GPS SVs, but not desirable for use as a WBAPL transmission antenna.
Figure 4.4 The elevation gain pattern of the dBS200 MLA (Serial #24), used for
GPS/APL reception at the LGF
Figure 4.5 is a plot of the GPS D/U ratio performance of the 14-element dBS200 MLA
(Serial #24), which is seen to be best between 5 and 35 degrees. The Ohio University
LAAS architecture utilizes MLAs for the reception of GPS signals between 5 deg and 35 deg, and HZAs for the reception of GPS signals at elevation angles between 30 deg and
90 deg. Both HZA and MLA antennas are used to receive the WBAPL signal, which due 71 to the proximity of its transmission antenna to the LGF reception antennas, is sufficiently strong to be received by the HZA despite its low angle of incidence.
Figure 4.5 GPS D/U ratio of the dBS200 MLA (Serial #24), used for GPS/APL reception
at the LGF
The APL D/U for reception using the MLA at FLD5, determined by differencing the gains at elevation angles of 0 and –6 degrees in Figure 4.4, is found to be 11 dB. The ratio for FLD6 is obtained as the difference in gains at 0 and –4.25 degrees, and is also determined to be 11 dB.
72
Figure 4.6 shows the elevation gain pattern of a typical “crossed-V” dBS 200A HZA used for reception of GPS signals in the range 30 to 90 degrees, and the WBAPL signal at the LGF. It is seen from the plot that the gain variation is approximately 10 dB in the region of GPS interest.
Figure 4.6 Typical elevation gain pattern of the HZAs used for GPS/APL reception at the
LGF
Figure 4.7 shows the typical GPS D/U ratio of HZAs utilized in the prototype LGF. The
D/U of the HZAs is seen to remain above 30 dB in the region of interest for GPS. By combining this pattern with the MLA pattern shown in Figure 4.5, a GPS D/U of 30 dB is obtained for the entire elevation angle span. 73
Figure 4.7 Typical GPS D/U of the HZAs used for GPS/APL reception at the LGF
The APL D/U for reception using the HZA at FLD5 is determined by differencing the gains at elevation angles of –0.64 and –6.6 degrees in Figure 4.6, to yield a value of approximately 2 dB. The ratio for FLD6 is determined as the difference in gains at –0.46 and –4.7 degrees, and is also found to be approximately 2 dB.
4.1.2 APL Transmission Antenna
It was recognized early in the APL development program at Ohio University that severe ground multipath for the extremely low-elevation APL signal required a unique antenna design with enhanced multipath limiting capabilities [Bartone (1996)]. Multipath at the 74
LGF is a critical issue for APL design, and the incorporation of multipath mitigation capability into the APL transmission antenna design is an effective way of tacking the issue. APL transmission was selected to be vertical polarized due the following reasons:
i. Increased energy transfer to the aircraft antenna, which has a strong vertical
component at low elevation angles,
ii. Improved multipath performance due to the reflection coefficient for low
grazing angles being smaller for the vertical component in comparison to the
horizontal, and
iii. Ease of vertical polarization design.
Incorporation of the above factors: one, antenna pattern-shaping specifically to discriminate against ground multipath; and two, adoption of vertical polarization, led to the development and fabrication of the original 14-element MLA as a collaborative effort between Ohio University and dB Systems, Hurricane, UT, for use as both the LGF reception antenna and the APL transmission antenna.
After the feasibility of the 14-element MLA was demonstrated, a WBAPL transmission- specific design was initiated. In addition to the necessity to mitigate multipath at the ground station, it was also deemed necessary to reduce the dynamic range of WBAPL power that an airborne receiver would be exposed to, as it was discovered that several
GPS/APL receivers exhibit a power-dependent error when operating in the saturation region. Another modification that sought incorporation into WBAPL transmission antenna design was an increase in the energy radiated in the upper-hemisphere of the elevation pattern in order to provide coverage for missed-approach operations. Both these 75 features were attainable by modification of the MLA pattern and a larger antenna aperture. A matrix that maps the region of the elevation angle to the desired attribute is described in [Dickman and Bartone (2001)]. The general design guidelines are summarized to include a vertically-polarized dipole array antenna with the following attributes
• A high APL D/U
• Uniform gain pattern in the region 5 to 40 degrees
• Increased coverage above 40 degrees
• Reduction in the “gain hump” between 5 and 20 degrees
A new 20-element vertically-stacked dipole array, termed the WBAPL Transmission
Antenna (dBS250), was designed in accordance with the above requirements, fabricated, and tested [Dickman (2002)]. It is utilized for WBAPL transmission in the prototype
WBAPL architecture. The larger aperture, in comparison with the 14-element MLA enables both the increased roll-off about the horizon, and extended elevation coverage up to 70 degrees. Figure 4.8 shows the elevation gain pattern of the original 14-element
MLA and the 20-element WBAPL MLA. As desired, coverage is seen improve in the range 35 to 70 degrees. The two gain bumps in the pattern of the dBS200 at negative elevation angles are also subdued, resulting in lesser energy directed towards the ground plane.
76
Figure 4.8 The elevation gain pattern of the 20-element WBAPL MLA (Ser #1) as
compared to the 14-element MLA (Ser #24)
Figure 4.9 shows the GPS D/U ratio of the 20-element WBAPL MLA, the performance of which is better over a wider range in comparison to the 14-element MLA. Performance comparison of the 20-element MLA was done in Reference [Dickman et. al. (2003)].
77
Figure 4.9 The GPS D/U of the 20-element WBAPL MLA used for transmission
The APL D/U for the transmission antenna is computed for each of the two reception sites. For the FLD5 MLA link, the D/U is determined by differencing the gains at elevation angles of 0 and –6 degrees in Figure 4.8, and yields a value of 20 dB. The ratio for the FLD5 HZA link, determined by differencing the gains at elevation angles of –0.64 and –6.6 degrees in Figure 4.8, provides a value of 20 dB. Similarly, computations performed for FLD6 show an APL D/U of 20 dB for both the MLA and the HZA links.
The D/U for the APL was increased from 11 to 20 dB by migrating from a 14-element
MLA to a 20-element MLA for transmission.
78
The total D/U in the APL MLA path is determined to be 31 dB (20 dB contribution from the transmission MLA, and 11 dB from the reception MLA). The total D/U in the APL
HZA path is determined to be 22 dB (20 dB contribution from the transmission MLA, and 2 dB from the reception HZA). These results are summarized in Table 4.2.
Table 4.2 Summary of the APL D/U for MLAs and HZAs at the Ohio University LGF
FLD5 Reception Path Link FLD6 Reception Path Link Source of D/U MLA (dB) HZA (dB) MLA (dB) HZA (dB)
Reception 11 2 11 2 antenna Transmission 20 20 20 20 MLA
Total D/U 31 22 31 22
Given the D/U obtained from the antenna gain patterns, the corresponding multipath errors can be determined. The voltage ratio of the direct and multipath signals (α ) is determined from the D/U as follows
1 α = D /U , (4.23) 10 20 since
⎛ 1 ⎞ D /U = 20log10 ⎜ ⎟ ⎝α ⎠ . (4.24)
For the MLA reception path, a D/U of 31 dB yields α of 0.03, and a D/U of 22 dB for the HZA reception path yields α of 0.08. Given that the ground antennas are fixed, the 79 exact path length differential can be determined from the geometry to obtain the predicted multipath performance. For the FLD5 reception geometry, the path delay for a single bounce off the ground is determined to be 1.32 ns for the MLA, and 1.6 ns for the
HZA. The values corresponding to the FLD6 geometry are 0.96 ns for the MLA, and 0.5 ns for the HZA reception path, respectively.
Using the known multipath delay, α , and the fact that a 0.1 chip correlator spacing was used by the ground reception receivers, the expected multipath error was computed to be limited to less than 0.06 m.
4.1.3 LGF Equipment
The GPS/APL receivers, LAAS processor, and VDB transmission equipment are located at the LAAS shelter (see Figure 4.2), into which the RF cables from the antennas are fed.
Figure 4.10 illustrates the two-site LGF receiver configuration used for the investigation documented in this dissertation. Data from the GPS/APL receivers are collected and processed by an LGF Real-time Processor running under a QNXTM Operating System
(OS), housed in a 19” rack-mount industrial grade personal computer. Modified 24- channel NovAtel MillenniumTM GPS Receivers are used at the LGF for GPS/APL signal reception at L1. These receivers are operated with special software built procured from
GPS Silicon Valley Incorporated (GSV). Receiver measurements are fed to the LGF real- time controller via RS-232 buses. The receivers are configured to use an external 10.0
MHz Rubidium oscillator as the clock reference. The VDB coverage extends to 23 nmi, at a carrier frequency of 112.95 MHz.
80
HZA
DC
COM1 APL Blanking and Novatel Millennium Bias-T MLA MGC GPS/APL Receiver
DC FLD5 APL Attenuation Using Novatel Millennium COM2 Bias-T Blanking Signal GPS/APL Receiver
LAAS Ground Processor APL Blanking and MGC with HZA 10.0 MHz Rubidium (Ziatech Industrial Grade Pulsing Control Circuit Frequency Standard PC with QNX OS) DC
COM3 APL Blanking and Novatel Millennium MLA Bias-T MGC GPS/APL Receiver VHF Antenna DC FLD6 APL Attenuation Using Novatel Millennium COM4 Bias-T Blanking Signal GPS/APL Receiver
COM5 COM1 Pulser Control to APL Transmitter
1 PPS VDB Transmitter Manual APL Gain Control
Figure 4.10 LGF set-up for two-site Ohio University prototype WBAPL configuration
4.1.3.1 APL Signal Generation
An IntegriNautics IN500 Signal Generator capable of generating an on-L1, wideband code, and pulsed signal according to the RTCA recommended WBAPL signal specification [RTCA LAAS ICD (1998)] is used to generate the PL signal, with the PRN
34. The IN500 contains an internal control loop to synchronize its clock with the GPS time determined using an internal GPS/APL receiver. Although modification to the software is being developed as a collaborative effort between Ohio University and
IntegriNautics that communicates its clock offset with respect to the calculated GPS time, the unit used for this research was not equipment with that capability. Therefore, both the
C/A and wideband codes are initially transmitted for the WBAPL, so that the LGF receivers are able to acquire the APL C/A-code and handover tracking to the wideband 81 code. A description of the unique technique employed in order to determine the initial
WBAPL clock offset if provided in Section 4.1.4.1. The C/A-code is turned off after the determination of the initial transmitter clock offset, and only the wideband code is transmitted for the remainder of the test duration. The ground receivers continuously update their estimation of the WBAPL clock offset, and include it as part of the LAAS broadcast message in order to enable the airborne receivers to directly acquire the wideband signal.
Figure 4.11 illustrates the WBAPL transmission subsection. The IN500 communicates with its interface software, which resided on a laptop with Windows OS, using RS-232 serial bus. The pulse sequence for WBAPL RF pulsing is not generated within the IN500, but generated externally by the WBAPL Blanking and Manual Gain Control (MGC) circuit that is located in the LGF, and fed to the IN500, which was set to operate in the
“external pulsing” mode. The pulsing format, which corresponded to a duty-cycle of
4.85%, is generated using an Erasable Programmable Read-only Memory (EPROM), consistent with the algorithm specified in RTCA LAAS ICD (1998) [Gunawardena
(2000)]. A sequence of “0111110” at the output of the feedback shift register used for the pulse sequence generation, as described in Figure E-1 of RTCA LAAS ICD (1998), triggers the generation of a pulse with width equivalent to the duration of 140 wideband chips, resulting in a duty cycle of 5.24%. In order to further reduce the duty-cycle to less than 5%, every 13th occurrence of the sequence is excluded from generating a pulse. The same sequence is utilized at the LGF to implement pulse blanking and gain control. An
RS-422 line driver is employed to transmit the generated TTL signal over the distance of 82 approximately 100 m between the LGF shelter, where the pulse generation equipment is located, and the base of FLD8, where the IN500 is situated for flight-tests. The final peak power transmitted is approximately +40 dBm.
WBAPL Transmission Antenna at FLD8
~12 m IntegriNautics Pinwheel RF Cable IN500 Antenna on Tripod
Control PC Pulser Control (Win OS) from LGF
Figure 4.11 WBAPL Tranmission Subsection
4.1.3.2 Pulse Blanking and Gain Control
The circuitry to perform APL power control was developed as part of an earlier effort to control C/A code power, at a time when only one of the IMLA reception paths was utilized to receive the APL signal [Bartone (1998)]. This circuit was ported for the present operation, and hence only one of the paths, that leading from the MLA is conditioned in an optimal fashion. In a full-fledged set-up, the signals from both the 83
MLA and HZA paths would undergo conditioning similar to that of the MLA path in the discussion to follow.
A block diagram of the circuit that implements the WBAPL Blanking and MGC is shown in Figure 4.12. Being of relatively high power, the WBAPL signal can be tracked on both the MLAs and HZAs at the LGF. The MLA is designated as the “desired GPS/APL” path, and the HZA path as the “desired GPS Only” path. In the MLA path, the composite signal is split into two sub-paths: one of which is completely blanked when the WBAPL signal is on (the GPS sub-path), and the other (the APL sub-path) that is attenuated for the complete duration, by a value that is user specified using a control program that resides on a PC. The outputs of the two sub-paths are then combined to yield a signal that contains both GPS and the attenuated WBAPL signal. In the HZA path, a fixed attenuation value (of approximately 35 dB) is maintained when the blanking signal is on
(i.e. when the WBAPL is on). This attenuation value is set in the hardware blanking circuitry, and is not software controllable in this configuration. A detailed circuit description, along with timing diagrams can be found in [Bartone (1998), Section 6.2.1].
84
Dual-Switch for Power APL Blanking Combiner GPS/APL Signal GPS To GPS/APL from MLA Sub-path Receiver 1:2 2:1 Composite APL GPS and APL Signal with Sub-path APL Gain Controlled Power Programmable Splitter Attenuator MLA Path
APL Blanking Control PC with Attenuator DB-25 Windows OS Control RS-232 1 PPS Blanking Timing Pulsing Control for APL Transmitter Control
APL Blanking Control
HZA Path GPS and APL Attenuation Attenuated APL using Blanking GPS/APL Signal Signal To GPS/APL from HZA Receiver
Figure 4.12 Illustration of the WBAPL blanker and MGC for one antenna site
4.1.4 LAAS Ground Processing
Individual NovAtel MillenniumTM receivers are connected to the paths from the HZA and
MLA antennas for each IMLA site at the LGF. The measurements are sorted, based on the elevation angle of the signal source, to obtain a single measurement per antenna site for each GPS SV. Since the signals delays are different for the two antenna paths, the difference, termed the hardware delay correction, is determined and applied prior to combining the measurements from the two antennas in the IMLA. As both the antennas at a site would receive a signal from an SV with elevation angle close to the region of transition (between 20 and 40 degrees) with acceptable gains, the difference between the 85
CP measurements to such an SV signal is used to determine the hardware delay correction.
The prototype architecture therefore uses common CP measurements to the GPS SV closest to 30 degrees to determine the difference in the hardware delays between the HZA and MLA paths. The hardware delay correction is propagated using a filter with a large time-constant, and is applied to the PR and CP measurements obtained for all the ranging sources in the MLA path. The measurements from the two antennas are translated to a common reference point prior to computation of the corrections.
On obtaining one set of PR and CP measurements per site, for each ranging source, the
PR corrections are determined according to the methodology prescribed in the [RTCA
LAAS MASPS (1998)]. The PR measurement is modeled as
PR n = R n + t − t n + tropon + iono n + ε n + nn m m m m (4.25) where
Rn : is the range from reference receiver m to ranging source n (m), m
t : the clock bias of the reference receiver m (m), m
n : the clock bias of the ranging source n (m), t n : the troposphere error for ranging source n (m), tropo
n : the ionosphere error for ranging source n (m), iono n : the ephemeris error (m), and ε
nn : the combination of noise and multipath on the PR measurement made by m i ti () 86
receiver m to ranging source n (m).
As LAAS is based on a CSC architecture, the PRs are smoothed prior to the determination of the corrections using [RTCA LAAS MASPS (1998)]
− 1 − PR n [k]s = PR n + (PR n − PR n ) m m N m m (4.26)
− PR n = PR n [k −1]s + (φ n [k] −φ n [k −1]) m m m m (4.27) where
PR n [k]s : is the smoothed PR measured by reference receiver m to ranging m source n corresponding to sample index k,
PR n [k −1]s : the smoothed PR corresponding to sample index k-1, m
N : the number of sequential samples accumulated, up to a max. of 100
sec,
φ n [k] : the CP measurement corresponding to sample index k, and m
φ n [k −1] : the CP measurement corresponding to sample index k-1. m
The equations to follow are all based on the smoothed PR at sample index k, and the index is therefore not distinctly specified. In lieu of being fixed, a reference receiver is
n able to compute the range ( Rm ) between its antenna and a ranging source, using the navigation message. The differential correction is obtained by subtracting the computed range from the measured PR as
PRct n = PR n − R n = t − t n + tropon + ionon + ε n + nn m m m m m . (4.28) 87
In order to keep the size of the correction within bounds, and to compare measurements from different paths, the averaged clock component for each path is determined and subtracted from the corrections obtained for that path. This does not affect the position solution, as biases common to the entire measurement set project into the clock bias estimate. If N source measurements are available, then the clock bias is estimated as
N ~ 1 j tm = ∑ PRctm . (4.29) N j=1
The subtraction of the average clock bias estimate yields
n n ~ n n n n n PRc = PRct − t = t + tropo + iono + ε + ∆t + n m m m m m (4.30) where ~ ∆t = t − t m m m (4.31) is the residual clock bias error. The PR correction for each ranging source is determined by averaging the estimates obtained by the M reference receivers.
M n 1 n PRc = ∑ PRc j (4.32) M j=1
Fault detection and isolation is accomplished by tracking the discrepancy between the corrections obtained for the various reference receivers, and is performed by estimating an error term for each correction, referred to as a B-value, which is determined as
M n n 1 n Bm = PRc − ∑ PRc j . (4.33) M −1 j=1 j≠m
The PR correction terms, obtained as described by Equation 4.32, B-values that correspond to the correction obtained by each reference receiver, as described by
Equation 4.33, are broadcast to the VDB for use by the airborne segment. The airborne 88 subsection applies the PR corrections to derive more accurate range estimates, and hence an improved position estimate. The airborne segment determines whether the signal-in- space integrity is sufficient to guarantee that the risk in performing the operation is within acceptable bounds using the B-values received from the ground segment. The CP measurement made by one of the ground reference receivers is also transmitted on the
VDB to enable the airborne subsystem to form the time-differenced CP system equations, as described in Section 4.2.1.2.
4.1.4.1 Determination of the WBAPL Transmitter Clock Offset
The ground segment also continuously determines the WBAPL transmitter clock offset, and encodes the information on the VDB in order to aid in direct wideband signal acquisition by participating airborne receivers. The procedure implemented to determine the APL transmitter clock offset is described below.
The PR measurement obtained for the APL signal by a fixed ground receiver ‘m’ can be modeled as
PR apl = R apl +T apl + t − t apl + n apl m m m m (4.34) where,
R apl : is the true range between the phase centers of the APL and receiver m antennas (m),
apl : the distance equivalent of tropospheric delay (m), T
tm : the distance equivalent of the receiver clock offset with respect to
GPS time (m), 89
apl : the APL transmitter clock offset with respect to GPS time (m), and t
n apl : the receiver measurement error, inclusive of multipath error (m). m
Since the WBAPL transmission and LGF reception antennas are installed at approximately the same height, the troposphere error is negligible. Therefore,
PR apl = R apl + t − t apl + n apl . m m m m (4.35)
The receiver clock error (tm ) can be determined to within the time transfer accuracy
apl afforded by GPS, which is within two wideband chips. If the true path length Rm between the transmission and reception antennas were known, then the determination of the transmitter clock offset would be straightforward. This would however require calibration of the hardware delays in the transmission path, which is not desirable.
Instead, the APL range residual is used.
A receiver estimates its position and clock offset using the available PR measurements, with the accuracy of the position solution being a function of the errors in the PRs, and the satellite geometry. A predicted range based on the position estimate can be defined as
Rˆ apl = R apl + e apl m m (4.36)
apl where em is the error in the prediction. A range residual for the APL can the be defined as follows
res apl = Rˆ apl − PR apl . m m (4.37)
Using Equation 4.37 in Equation 4.35 yields 90
res apl + t = t apl + e apl − n apl . m m m m (4.38)
The NovAtel MillenniumTM receivers used in the prototype architecture report the APL range residual and the estimated receiver clock-offset, the two quantities on the left-hand side of Equation 4.38. Given the position accuracy supported by stand-alone GPS, the
apl term em could have a magnitude on the order of tens of meters, which is not a deterrent
apl for the purpose of direct WB-code acquisition. The term nm , which consists of receiver noise and multipath, is limited to a few meters due to the multipath rejection capability of the LGF antennas.
APL range residuals obtained from each of the four ground receivers are averaged to form the ground estimate of the APL transmitter clock offset (i.e., t apl ), which is broadcast on the VDB. A message was added to the VDB data structure in order to include both the WBAPL clock-offset information, and the GPS time which it corresponds to. During the initiation of the WBAPL acquisition process in the airborne segment, the airborne processor feeds the current WBAPL clock-offset information to the receiver, using specific commands supported by the modified (i.e., the GSV software build) NovAtel MillenniumTM airborne receiver, in addition to the coordinates of the
WBAPL transmitter.
4.2 The Prototype Airborne Subsystem
Figure 4.13 shows the airborne equipment set-up for the Ohio University WBAPL- inclusive prototype 91
LAAS architecture. A single top-mounted antenna is used for the reception of GPS SV signals as well as that of the ground-based WBAPL. The RF path from the top-mounted antenna has a JCA pre-amplifier immediately following the passive L1-L2 reception antenna. A 24-channel NovAtel MillenniumTM receiver, similar to those employed by the ground subsection, is utilized, and the measurements from it are fed to the airborne
LAAS processor using an RS-232 bus. The prototype LAAS airborne processor runs on a
QNX OS, based on a 19” rack-mount industrial grade computer.
Top-Mounted L1/L2 GPS/APL Antenna
Pre-amp COM 1 LAAS Airborne NovAtel Millennium Processor GPS/APL Receiver (Ziatech PC with QNX OS) VHF Antenna
VDL Receiver
Figure 4.13 Generic block diagram of Ohio University prototype WBAPL airborne set-up
4.2.1 LAAS Airborne Processing
The proposed architecture for the airborne segment utilizes carrier-smoothed PR measurements from the beginning of an approach operation through the PHP, after which the system migrates to position propagation using time-difference CP SD measurements.
The system equations involved in each of these phases is described Section 4.2.1.1 and
Section 4.2.1.2, respectively. 92
4.2.1.1 CSC Solution
The airborne segment receives the PR corrections and B-values corresponding to each ranging source on the VDB. The corrections, which are scalar estimates of the aggregate error that is correlated between the ground and air segments, are applied to the carrier- smoothed PRs derived by the airborne processor to arrive at airborne range estimates.
Equation 4.26 and Equation 4.27 describe the filter used to perform the carrier smoothing of the raw PR measurements. The range estimates are obtained as
Rˆ n = PR n [k]s − PRc n m m m (4.39)
The corrected PRs follow the model described by Equation 2.2. The solution of such a system of equations was discussed in Section 2.1.4.
4.2.1.2 Position Propagation Using a Time-Differenced Differential CP (TDCP)
Approach
It was seen from the results of the laboratory experiments that while the PR errors due to the power-bias error were on the order of meters, the CP measurement errors were limited to the centimeter level. By transitioning to the sole utilization of accumulated
Doppler shift measurements as an aircraft approaches the decision threshold, the effect of the power-bias error on the differential position solution can be limited. Traditional kinematic differential GPS architectures have relied on the resolution of the CP ambiguities in order to solve for position. This has severely limited their viability as navigation systems to support high-integrity precision approach and landing operations, as discussed in Chapter 2. The approach adopted here, described in Reference [van Graas and Lee (1996)], utilizes successive time differenced observables in order to avoid 93 resolution of the CP ambiguities, an can also be formulated to entirely not rely on the code-phase measurements after the transition.
The single difference (SD) variable obtained using the CP measurements from airborne and ground receivers, to a common GPS SV, can be written as
SDga = b⋅e = φ g −φa + Nλ + cδt ga (4.40) where
b : the three-dimensional vector pointing from the aircraft to ground
antenna locations (m)
e : the unit vector from the middle of the baseline vector the chosen SV
φg : is the ground receiver’s accumulated CP measurement (m)
φa : the airborne receiver’s accumulated CP measurement (m)
N : the ambiguous integer cycle count
λ : the wavelength of the carrier frequency (m)
c : the propagation velocity of RF signal (m/sec)
δt ga : The clock offset between the ground and air receivers (sec)
In the case of locally augmented GPS, the distances from satellites to terrestrial users are large compared to the length of the baseline. The wave fronts at the two ends of a baseline vector can therefore be assumed parallel for a common signal. A single unit vector from the center of the baseline can therefore be used to approximate the two unit vectors from the ends of the baseline, with negligible error. In the case of the ground- 94 based APL, its proximity to the baseline vector makes this approximation invalid. In order to arrive at an accurate measurement model, the formulated SD has to be modified to account for this, as described in Section 4.2.1.3.
Differencing the SDs at two consecutive measurement epochs eliminates the need to solve for the unknown integer ambiguity in the CP measurement. The calculation of the difference between two SDs over consecutive time epochs, referred to as a time double difference (TDD), must accurately account for the change in geometry during the interval between the two epochs in order to avoid the build-up of systematic errors. If the SD at time epochs 1 and 2 are represented as
SD1 = H1β1 and
SD2 = H 2β 2 , (4.41) respectively, then the TDD is given by
TDD 21 = SD 2 − SD1 = H 2β 2 − H 1β1 . (4.42)
Equation 4.42 can be expressed as [van Graas and Lee (1996)]
* * TDD = H 2 (b1 + ∆b) − H1b1 (4.43) where
H* : is the submatrix of H, which contains only the unit vectors,
b1 : contains the x, y, and z coordinates of the baseline vector at time epoch 1,
and
∆b : is the change in the x, y, and z coordinates of the baseline vector in the
interval between the two measurement epochs.
95
The TDD should be corrected for the differences in tropospheric delay between the two receivers, over the interval between the measurement epochs. The tropospheric correction models applied for GPS signals, and that applied for the WBAPL signal is described in
Section 4.2.1.4. The change in the baseline vector can then be obtained using an ordinary
Least-squares solution as
*T * −1 *T * * ∆b = (H 2 H 2 ) H 2 (TDD21 − (H 2 − H1 )b1 ) . (4.44)
The differential CP propagated position solution at the time epoch 2 is obtained using the sub-decimeter accurate change in baseline as
b 2 = b1 + ∆b . (4.45)
The geometry matrix at time epoch 2 is obtained by approximating the position of the aircraft, and hence the baseline, at the time epoch using differential code-phase measurements, as outlined in Section 4.2.1.1. If necessary, reliance on the code-phase measurement can be entirely avoided by using the position and geometry matrix at epoch
1, and the DD measurements to obtain an approximation of b2 as
*T * −1 *T b 2 = b1 + (H1 H1 ) H1 TDD21 . (4.46)
4.2.1.3 SD Geometry Correction for the APL
Given that the length of the baseline vector between airborne and ground antennas for
LAAS is very small compared to the distances between the receivers and GPS satellites, a single unit vector from the center of the baseline can approximate the unit vectors from each of the receivers to a common satellite. Given the proximity of the APL to the ground and airborne antenna, the approximation is not applicable, and therefore calls for a 96 modification of the SD measurement model [Lee (1999)]. The various vectors involved in the formulation of the SDs for an APL are shown in Figure 4.14.
M (Mobile User)
β
b
(Ground φa − φg Reference) φ a G −b • φˆ a α
C α φg
B γ X
φg P (APL)
Figure 4.14 SD geometry for an APL
It is desired to find the relationship between the SD (| φ g | − | φ a |) and the projection of
the baseline vector in the direction of the radial vector φ a . It is seen from Figure 4.14 that the two are related as
ˆ ( | φ a | − | φ g | ) + X = − b ⋅φ a . (4.47) 97
As an expression for the SD is of interest here, Equation 4.47 is re-arranged to give
SD − X = b ⋅φˆ . a (4.48)
The quantity X is determined as follows. From the triangle GBM,
GB = | b | sin β , (4.49) the application of which to the inner right-triangle CBG yields
GB | b | sin β X = = tanα tanα (4.50) where
−1 ⎛ b⋅φ a ⎞ β = cos ⎜ ⎟ . (4.51) ⎝ | b | | φ a | ⎠
An expression for the angle α is to be determined next. Since sides GP and CP in the triangle GPC are equal,
∠GCP = ∠PGC = α , (4.52) leading to
180 − γ γ α = = 90 − , 2 2 (4.53) where
⎛ φ ⋅φ ⎞ γ = cos −1 ⎜ a g ⎟ . ⎜ ⎟ (4.54) ⎝ | φ a | | φ g |⎠
The SD measurement model (Equation 4.40) is modified using Equation 4.48 and
Equation 4.50 as
| b | sin β SD − = b ⋅φˆ + Nλ + cδt . ga tanα a ga (4.55) 98
4.2.1.4 Differential Tropospheric Correction for the WBAPL
The tropospheric delay experienced by the APL signal prior to arrival at the airborne receiver can be significantly different from that at the ground station, especially when the aircraft is at the fringe of the APL coverage volume. The differentially corrected PR measurement should account for this difference. The differential tropospheric correction model for the APL signal, in terms of the PR correction in meters, is specified as [RTCA
LAAS ICD (1998)]
∆hu −6 δRtr = N r × (1 − ) × Ru ×10 (4.56) h0 where
: is the tropospheric refractivity, N r
∆hu : the height difference between the airborne user and ground station (m),
Ru : the range from the APL to the airborne user (m), and
h0 : the scale height (m).
The tropospheric refractivity and scale height can be obtained from surface measurements of temperature, pressure, and humidity, as follows. The tropospheric
refractivity ( N r ) consists of wet and dry components [van Dierendonck (1997)]
N r = N r,dry + N r,wet (4.57) in which
4 77.6 Ps ⎡42700 − h ⎤ N r,dry = ⎢ ⎥ (4.58) Ts ⎣42700 − hs ⎦ 99
7.4475(T −273) 4 2.277 ×104 × RH S ⎡ 13000 − h ⎤ N = ×10 TS −38.3 × r,wet 2 ⎢ ⎥ (4.59) Ts ⎣13000 − hs ⎦ where
Ps : is the surface atmospheric pressure (mbar),
Ts : the surface temperature (Kelvin),
RH : the surface relative humidity (percent),
h : the height above Mean Sea Level (MSL) of the user (m), and
hs : the height above MSL of the reference station (m).
The scale height is obtained using its dry and wet components as [McGraw et. al. (2000)]
N r × h0,dry × h0,wet h0 = (4.60) (N r,dry × h0,wet + N r,wet × h0,dry ) in which
42700 − h h = s 0,dry 2 , (4.61) and
13000 − h h = s 0,wet 2 . (4.62)
100
5. LABORATORY TESTS TO CHARACTERIZE POWER-INDUCED
MEASUREMENT ERRORS FOR APL RECEIVERS
5.1 Experiment Goal
It has been reported that certain GPS receivers exhibit measurement errors as a function of the received signal’s peak power [Hein et. al. (1997), Van Dierendonck (1998)]. These demonstrations however did not fully characterize the effects of the error. The verification and subsequent mitigation of the high-power induced error is a key factor in the fielding of WBAPL(s), as the dynamic range of the WBAPL signal for an airborne user will exceed that of GPS. The dynamic range for received WBAPL power depends on the siting of the WBAPL transmission antenna, the operational range supported, and the transmission and reception antenna radiation patterns. Analysis performed according to the methodology described in [Dickman and Bartone (2001)] indicates that a dynamic range of approximately 34 dB would be required of the airborne user receiver, based on the siting used at the Ohio University prototype LGF, for a coverage volume corresponding to 20 nmi. As the airborne user approaches the airport, and therefore the
WBAPL transmission antenna, the received WBAPL signal power, and possibly the power-dependent error term, would increase. The presence of a power-induced error can cause a fixed bias in the ground measurements of the WBAPL, leading to a bias in the corresponding PR correction determined at the LGF. Non-common high power-induced errors between the ground and air receivers are liable to produce errors in the final differential position solution.
101
A laboratory test set-up was designed to characterize the power-induced error in the
NovAtel MillenniumTM and BeelineTM GPS receivers that have commonly been used for
WBAPL development and research [Warburton et. al. (1999), Kiran and Bartone (2002),
Suga et. al. (2002)]. The test set-up, procedure, and results are detailed in this chapter.
5.2 Measurement Model Description
The GPS PR measurement (in meters) made by a receiver that is sensitive to variation in the peak signal power can be modeled by modification of Equation 2.1 as
ρ = r + α + T + I + t − t s c + ε ρ ρ ( r ) ρ (5.63) where
α : the power-induced PR measurement error (m) ρ
In the case of the WBAPL signal, the ionospheric error component is ignored, leading to the following model for the WBAPL PR measurement made by a receiver in the airborne signal path
ρ p = r p + α p + T p + (t − t p ) c + ε p a a ρ a a a ρ a . (5.64)
The following notation is used throughout this Chapter:
• A subscript is used to denote a reception element; ‘a’ denotes an airborne
receiver, and ‘g’ a ground receiver,
• A super-script is used to denote a transmission element; ‘p’ denotes a PL
transmitter, and ‘s’ a GPS SV source.
102
The quantity of interest that needs to be isolated is the term α p in Equation 5.64. In ρ a order to eliminate correlated error components, simultaneous PR measurements made by a ground receiver, modeled as following, can be utilized
ρ p = r p + α p + T p + (t − t p ) c + ε p g g ρ g g g ρ g . (5.65)
Since the ground receiver is exposed to fixed PL power, the power-dependent error α p ρ g is a constant. An SD computation between the air and ground measurements can be formed to eliminate transmitter specific errors
ρ p − ρ p = r p − r p + T p −T p +α p − α p + (t − t ) c + ε p a g a g a g ρ a ρ g a g ρ ag . (5.66)
A similar SD computation to a GPS satellite would yield
ρ s − ρ s = r s − r s + T s − T s + I s − I s + (t − t )c + ε s a g a g a g a g a g ρ ag . (5.67)
Given the relatively narrow dynamic range and low received power levels of the GPS signals, their PR and CP measurements are not susceptible to the power-induced error.
This difference between two SDs (obtained using Equations 5.66 and 5.67, referred to as a PL double-difference (DD), is
DD ps = (ρ p − ρ p ) − (ρ s − ρ s ) ρ ag a g a g p p ps = (r p − r p − r s + r s ) + (T p − T p − T s + T s ) − I s + I s + α −α + ε a g a g a g a g a g ρ a ρ g ρ ag
(5.68)
In order to isolate the PL power-induced error term α p , the differential tropospheric ρ a term
T ps = T p − T p − T s + T s ag a g a g , (5.69) the true geometric DD 103
r ps = r p − r p − r s + r s ag a g a g , (5.70) have to be determined and subtracted from the computed PR DD. Given that the system is being evaluated over a short baseline, the differential ionospheric term can be neglected.
I s − I s = 0 g a . (5.71)
The resulting expression for the PR DD is therefore
DD ps = r ps + T ps + α p − α p + ε ps ρ ag ag ag ρ a ρ g ρ ag (5.72)
The term α p can then be observed in the presence of a fixed bias ( −α p ) and noise ρ a ρ g
ε ps . Since the variation of α p over the PL dynamic range is the desired observable, ρ ag ρ a the fixed bias term is not a deterrent.
α p = DD ps − r ps − T ps + α p − ε ps ρ a ρ ag ag ag ρ g ρ ag (5.73)
The above analysis was repeated to obtain the error expression for the CP observable. A
CP measurement (in cycles) made by a receiver is modeled as
1 c φ = ()r + α + I +T + ( t − t s ) + ε + N λ φ φ λ r φ (5.74) where
λ : is the wavelength of the carrier signal (m),
α : the power-dependent CP measurement error (m), φ
I : the CP advance due to propagation through the ionosphere (m), φ
ε : the combined effect of multipath and noise on the CP measurement φ (cycles), and 104
N : the integer ambiguity in the number of whole cycles between the transmitter
and receiver.
The quantity of interest in this model is the term αφ , the power-induced error in the PL
CP measurement. By repeating the analysis used above for the PR measurement, the PL
CP DD error is obtained as
1 DD ps = (φ p −φ p ) − (φ s −φ s ) = (r ps + T ps + α p − α p ) + N ps + ε ps φ ag a g a g λ ag ag φ a φ g ag φ ag (5.75) where
N ps = N p − N p − N s + N s ag a g a g . (5.76)
The expression for α p is obtained as φ a
α p = λ DD ps − λ N ps − r ps − T ps + α p − λ ε ps φ a ϕ ag ag ag ag φ g φ ag . (5.77)
The PL signal generator was not configured to model any tropospheric delay, and
p p therefore Ta = Tg = 0 in Equation 5.68. The tropospheric and ionospheric errors generated by the GPS signal generator identically impacted the ground and air measurements in the laboratory set-up, and therefore
T s − T s = 0 a g . (5.78)
I s − I s = 0 a g . (5.79)
Since the simulator was configured for a static profile, the true ranges remain constant in the DD, and the resultant range can therefore be lumped as
κ ps = r p − r p − r s + r s ag a g a g , (5.80) yielding a modified version of Equation 5.72 as the PR DD for the laboratory set-up. 105
DD ps = κ ps + α p − α p + ε ps ρ ag ag ρ a ρ g ρ ag (5.81)
α p is observable in the presence of a fixed bias (κ ps −α p ) and noise ε ps . Since the ρ a ag ρ g ρ ag variation of α p over the PL dynamic range is the desired observable, the fixed bias term ρ a is not a deterrent.
A similar modification of Equation 5.77 for the CP DD yields the following expression for the power-induced error on the airborne measurement as
1 DD ps = (κ ps + α p − α p ) + N ps + ε ps ϕ ag λ ag φ a φ g ag φ ag . (5.82)
5.3 Laboratory Test Set-Up
To isolate and study the effects of non-linear receiver saturation on the PR and CP measurements, a laboratory experiment was constructed to compared the measurements in two paths, one of which was presented with constant, and the other varying, PL power.
Figure 5.1 illustrates the set-up used for the experiment, which was configured to be reasonably representative of a LAAS, with separate ground and airborne segments.
106
GPS Signal Input Segment Ground Segment GSS-4760 L 1:6 Rubidium Oscillator 8-Channel GPS Signal GPS Simulator L L
L To Ground 2:1 HPA Serial Port 1 40 dB 1:4 Millennium 1:4 (Rcvr #1) 13 dB APL Signal
Serial Port 2 Millennium Ziatech PC with (Rcvr #2) QNX OS To Air
GPS Signal APL Signal Generation Segment Serial Port 3 Serial Port 4
Programmable Attenuator Millennium Beeline IntegriNautics 30 dB 30 dB (Rcvr #3) (Rcvr #4) IN500 1:2 Simulator HPA 2:1 1:4 L Simulation of Propagation Loss L
Airborne Segment Serial Port 1 (IN500 Control)
PC with Windows OS HPA - High-Power Attenuator Parallel Port L - Load (Attenuator Control)
Figure 5.1 Laboratory experiment set-up for the characterization of power-dependent
measurement errors
The ground segment was configured to simulate the fixed PL power path from the
WBAPL transmission site to LGF reception sites. Two MillenniumTM receivers were used in the ground subsection (Receivers 1 and 2 in Figure 5.1), and these were presented with constant GPS and PL signal powers. The experimental set-up consisted of two receivers in the ground path, so that consistency checks to ascertain the integrity of the ground segment measurements could be conducted. The airborne segment utilized a NovAtel
MillenniumTM and BeelineTM receiver each (Receivers 3 and 4 in Figure 5.1, respectively), which were the receivers under test, and were presented with constant GPS power, but varying PL power. The PL signal power was varied using a high quality programmable 107 attenuator. The BeelineTM receiver has essentially the same circuit card as the
MillenniumTM receiver, but has two RF inputs and associated analog-to-digital conversion chains, whereas the MillenniumTM has one.
An 8-channel L1-only GSS 4760 Signal Generator was configured to simulate a GPS constellation relative to a static user profile. The output signal was obtained from the simulator’s RF Monitor port, with a peak power level of -50 dBm. The simulator was set- up to generate tropospheric and ionospheric delays based on its internal algorithms. The injection of the simulated GPS signal essentially sets the noise-floor of the receivers, whereby a nominal C/N0 of approximately 45 dB-Hz was established. All four receivers and the GSS GPS Simulator were configured to use an external 10 MHz Rubidium oscillator.
An IntegriNautics IN500 Signal Generator capable of generating an on-L1, wideband code, and pulsed signal according to the RTCA recommended WBAPL signal specification [RTCA LAAS ICD (1998)] was used to generate the PL signal. The generated signal had a peak power level of +35 dBm at a pulsing duty cycle of 4.7%, rather than the RTCA specified pulsing duty-cycle of 2.7% [RTCA LAAS ICD (1998)].
The pulse generation algorithm was the same, but the pulse-width was increased from
140 wideband code chips to 240 wideband chips. The parameters were selected to be more consistent with that used for flight trials of the Ohio University WBAPL architecture. The IN500 was fed the simulated GPS signal from the GSS GPS Simulator for time synchronization. The IN500 contains an internal control loop to synchronize its 108 clock with the determined GPS time. Passing the PL signal through a General Microwave
3491-64 Programmable Attenuator controlled the power variation at the input of the airborne segment. It was verified prior to the experiment that the programmable attenuator did not introduce a group delay as a function of attenuation. The programmable attenuator was controlled by an application developed specifically for the task, which resided on a PC with Windows 95 OS. The attenuator control was implemented using Windows 95 parallel communication routines. The transmission power and fixed attenuators in the APL path were chosen such that the receivers in the airborne segment could acquire the APL signal when the attenuator was set to a value of
50 dB. Increase of the attenuator value to 53 dB caused the receivers to lose lock of the
APL signal.
The attenuation profile chosen for the study was a stair-step with minimum and maximum attenuation values of 0 and 50 dB, respectively, and a step-size of 2 dB. Each attenuation value was maintained for a period of 20 sec. The attenuator value was initialized to the maximum, decreased to the minimum, and then returned to the maximum value with the aforementioned step size and duration. The attenuation profile is plotted along with the PR and CP results obtained in the data to follow, in order to aid in visual correlation.
5.4 Test Results
The laboratory experiment involved application of constant GPS power to both the air and ground paths, fixed PL power to the ground path, and a variable PL power profile to the air path, and the collection of PR and CP measurements reported by the receivers 109 during the process. The fact that the WBAPL signal does not comprise the C/A-code necessitates direct-wideband acquisition capability. In the Ohio University prototype
LAAS architecture, the ground segment maintains track of the WBAPL transmitter clock offset and transmits the information on the VDB for the airborne receiver to use in its acquisition process. This was however not performed in the laboratory test, to reduce the complexity of the test set-up. All four receivers were initially fed both the C/A and wideband PL codes. After using the C/A-code for acquisition and subsequent handover to the wideband code, the C/A code was switched off. The attenuation profile was then initiated, and the measurements reported by the receivers were recorded. The corresponding results are presented in Figures 2 through Figure 5.
Figure 5.2 shows the plot of the laboratory-measured PR DD error, modeled as Equation
5.81 for the PL signal, over the attenuation profile. GPS SV #27 was used as the reference satellite for the DD computation. The plot was generated after subtracting the initial value, as this represents the fixed bias. The stair-step attenuation profile is plotted on the graph as well. The exhibition of a power-dependent variation in the DD error is seen in the plot, moving towards the center of the trace, where the PL power is the strongest (attenuator set to 0 dB). Prior to the commencement of the profile, the average
APL power was lower than the average GPS SV signal power level, which in this set-up was identical for all the simulated SVs. When the attenuation level decreased by 6 dB, the average APL signal power was approximately the same as all the GPS SV average power levels. This assessment was made by observing the C/N0 reported for the APL and 110
GPS signals (see Figure 5.3), and therefore assumes that the C/N0 reported by the receivers is accurate prior to saturation.
Figure 5.2 PL PR DD errors obtained over the attenuation profile, with GPS SV #27 used
as the reference
Figure 5.3 shows the carrier-to-noise ratios (C/N0) reported by the receivers in the airborne subsection during the test for which the results are presented in Figures 2 through 4. The C/N0 reported by the two receivers in the airborne path for GPS SV #27, which was used as the reference in the analysis, are plotted as well. As expected, the
C/N0 for GPS tracking decreases by 20log10 (1− duty cycle) , which is -0.42 dB for a 111
4.7% PL (non-blanker GPS/APL receiver). The WBAPL C/N0 increases to a maximum due to saturation of the ADC by the high-power pulses, after which the C/N0 stays fixed.
For the 4.7% duty-cycle PL, the C/N0 maximized at approximately 47 dB-Hz.
Figure 5.3 Carrier-to-noise ratios reported by the receivers in the airborne path for the PL
and the reference SV (GPS SV #27)
In order to ascertain that the variation is only exhibited by the high-powered PL signal, a similar GPS DD error computation was calculated using the GPS SV #15 and the reference SV #27, and the corresponding result is plotted in Figure 5.4. The initial value was not subtracted from the measurements for this plot. The DD error in this case does not exhibit the power-dependent variation seen in the case of the PL signal. 112
Figure 5.4 PR DD errors obtained for GPS SV #15 over the attenuation profile, with GPS
SV #27 used as the reference
Figure 5.5 shows the plot of the CP DD error for the PL signal. As the CP integer ambiguities were not resolved, the CP DD contains the following fixed bias term from
Equation 5.82
1 (κ ps −α p ) + N ps λ ag ρ g ag
The initial value was subtracted before plotting, as it is the variation in the error as a function of the attenuation profile that is the parameter of interest. The exhibition of a slight power-dependent variation in the DD error is seen from the plot, but its magnitude 113 is much smaller than that the PR measurements. The PR DD error (see Figure 5.2) was on the order of meters, whereas the CP error seen in is in the centimeter level.
Figure 5.5 PL CP DD errors obtained over the attenuation profile, with GPS SV #27 used
as the reference
5.5 Conclusions from the Laboratory Tests
The results from the laboratory experiment clearly characterize a power-dependent error present in both the PR and CP measurements of the receivers in the airborne segment.
The CP power-induced measurement error is determined to be an order of magnitude smaller than its PR counterpart. Given the verification of the presence of a power-induced 114 error in the receivers under test, a low cost solution without extensive hardware additions, especially for the airborne user was sought. The proposed solution to mitigate the power- induced error on the airborne WBAPL measurement was designed on taking advantage of its diminished presence in the CP measurement. The solution involves utilizing the airborne PR for the WBAPL until a point along the approach path, defined as the Phase
Handover Point (PHP), after which the position solution can be propagated using CP measurements. The PHP could be determined by either real-time monitoring of the code- carrier divergence, or by analysis for the particular LAAS configuration. Incorporating a carrier-smoothed code with a high time-constant would provide a similar benefit.
However, sole reliance on the CP measurements towards the end of the operation will ensure that there is essentially no influence on the differential position solution from the
PR measurement error. It is proposed that the kinematic-positioning system be such that there is no need for CP ambiguity resolution. A CP-based differential GPS architecture with such a constraint was developed and field tested in 1994 [van Graas and Lee
(1995)], and is currently being considered as a viable option for PT 2 and PT 3 LAAS architecture [Brenner (2003), van Graas (2003)]. This development ties in well with the integration scheme proposed for the WBAPL as well. The architecture utilizes the differences in CP measurements over consecutive epochs to overcome the necessity for ambiguity resolution. The differential CP WBAPL would be launched at the PHP from a carrier-smoothed code based solution with predefined levels of integrity. Given that the carrier-tracking loop is typically the most vulnerable segment within a GPS receiver, it is necessary to incorporate cycle-slip detection and correction features in the airborne 115 software in order to meet the continuity of function necessary for precision approach operations.
Whereas architecture modifications in the airborne platform in order to incorporate
WBAPL(s) are seen as undesirable, this constraint is more flexible on the ground at the
LGF. With this in mind, the proposed solution for the ground system involves the utilization of pulse blanking and signal attenuation technique to reduce the WBAPL power fed to the receivers, an approach similar to that used previously for C/A-code
APLs [Bartone (1998)]. The conclusions arrived at from the laboratory tests described here were instrumental in deciding on the Ohio University WBAPL prototype architecture proposed in Chapter 4. 116
6. FLIGHT TEST SET-UP AND RESULTS
A series of flight-tests were conducted at UNI on March 3 and 4, 2003 in order to verify the effectiveness of the power-induced measurement error mitigation techniques proposed in Chapter 5, and to test the real-time differential positioning error performance of the Ohio University prototype WBAPL architecture. The results obtained from the power-induced measurement analysis are described in Section 6.4. In order to gauge the influence of the WBAPL on the differential positioning error performance, two separate position solutions were computed simultaneously in the aircraft: a DGPS-only solution
(not inclusive of the WBAPL) and a DGPS position solution inclusive of the WBAPL
(termed the DGPS/Differential APL (DAPL) solution). The flight test data were also post-processed under two conditions: one in which all the visible GPS SVs were utilized in the position solution, and another that relied on a subset arrived at by increasing the mask angle from 5 to 20 deg. This could simulate a case of poor SV availability, or a mask angle induced to reduce susceptibility to a terrestrial-based interference source. The results obtained are detailed in Section 6.5.2.1 and Section 6.5.2.2, respectively.
6.1 Modification of the Ohio University Prototype LAAS Airborne Subsystem
for the Flight-Test
The LAAS architecture uses the same top-mounted antenna for GPS and WBAPL signal reception. The WBAPL signal, depending on the location of the transmitter, is liable to undergo airframe diffraction and reflection prior to arrival at the top-mounted antenna, thereby increasing the complexity of the power-induced error analysis. In order to facilitate the isolation of the power-induced error, a nose-mounted passive L1/L2 patch 117
GPS/APL reception antenna followed by an isolator and a JCA amplifier was added to the airborne test platform: the Ohio University DC-3 aircraft (see Figure 6.1).
Figure 6.1 A photograph of the nose-mounted antenna on the DC-3 (with its radome
open)
It is emphasized that this nose-mounted antenna was used only for this power-related error investigation, and is in no way part of any planned LAAS airborne component. The antenna was mounted on a square ground plane and angled at approximately 45 deg in order to provide some visibility of GPS satellites, while still retaining the ability to reliably track the extremely low-elevation WBAPL signal. Given that the angle-of-arrival of the WBAPL signal at the nose-mounted antenna would be in a higher gain region, in comparison to the top-mounted antenna, its received power would be higher. The performance obtained from the analysis of the measurements obtained from the nose- 118 mounted antenna would therefore be representative of a worst-case scenario over the approach path.
Figure 6.2 shows the equipment set-up in the DC-3 for the test. A 24-channel NovAtel
MillenniumTM receiver was used to obtain GPS and WBAPL measurements on the RF path via the nose-mounted antenna. The data output by the receiver was streamed to a
Compaq laptop computer with QNX OS, using serial communication. An L1-L2 Ashtech
Z-12 GPS receiver and a 24-channel Novatel MillenniumTM receiver were connected to the top-mounted antenna. The MillenniumTM receiver was the prototype LAAS airborne receiver, and its measurements were streamed to the LAAS airborne processor that ran on a Ziatech Industrial grade PC with QNX OS.
Nose-mounted L1/L2 GPS/APL Antenna DC
COM 1 NovAtel Millennium Compaq Lap-top Bias-T GPS/APL Receiver with QNX OS Isolator Pre-amp
Honeywell Ashtech Z-12 YG1851 IRS GPS Receiver Top-mounted L1/L2 COM 1 GPS/APL Antenna DC COM 2 NovAtel 3951 Rack-mount GPS Receiver PC-104 Bias-T 1:4 COM 1 NovAtel Millennium Ziatech PC with Isolator Pre-amp GPS/APL Receiver QNX OS VHF Antenna L COM 2
VDL Receiver
Figure 6.2 Block diagram of the airborne segment used for the isolation of the power-
induced error on WBAPL measurements
119
The Ashtech Z-12 receiver measurements were used to determine a post-processed kinematic solution that served as the truth reference. The translation of the truth reference position to the nose-mounted antenna requires accurate knowledge of the aircraft attitude and the lever-arm vector between the two GPS antennas. A Honeywell Inertial Reference
System (IRS) was flown on-board to provide the attitude information. Data from the IRS, streamed on an ARINC 429 bus, was stored on a PC-104 form-factor 19” rack-mount computer. The output of the inertial data was time-tagged using a NovAtel 3951 GPS receiver. The location of the Honeywell IRS on the DC-3 along with the placement of the nose-mounted antenna with respect to the top-mounted GPS L1/L2 antenna is illustrated in Figure 6.3 and Figure 6.4. The lever-arm measurements between the two GPS antennas were obtained prior to the flight using GPS static survey, and verified using mechanical drawings of the DC-3.
120
Top-Mounted Honeywell IRS L1/L2 GPS Antenna
45-Degree Nose-Mounted L1/L2 GPS Antenna 1.97 m
0.582 m Floor Line 0.175 m
Figure 6.3 The locations of the top and nose-mounted antennas, and the Honeywell IRS
on the DC-3 (side-view)
121
Honeywell IRS
45-Degree Nose-Mounted L1/L2 GPS Antenna
0.57 m 0.46 m
Center Line
2.53 m Top-Mounted L1/L2 GPS Antenna
Figure 6.4 The locations of the top and nose-mounted antennas, and the Honeywell IRS
on the DC-3 (top-view)
6.2 Determination of the Truth Reference
Data from the Ashtech Z-12 L1-L2 receiver, which was connected to the top-mounted aircraft antenna, was differentially post-processed in a kinematic fashion to yield the truth position of the top-mounted antenna. The dual-frequency ground-station measurements required for post-processing of the Ashtech data were obtained from a GPS ground reference station located at the South Tower at UNI, located approximately 300 m from the LGF. 122
The truth position of the nose-mounted antenna was not immediately available, and had to be derived. Since the nose and top-mounted antennas were fixed on the aircraft, precise knowledge of the location of top-mounted antenna, in conjunction with knowledge of the lever-arm and aircraft attitude, were used to derive the location of the nose-mounted antenna, as described below.
Toward this effort, a body-fixed coordinate frame with its origin at the center of the
Honeywell IRS is defined using a right-hand coordinate system as follows
• The x-axis lies along the longitudinal axis with positive towards the nose of the
vehicle,
• The y-axis along the lateral, with the positive towards the starboard, and
• The z-axis along the vertical, with the positive facing the floor of the vehicle.
The descriptions of the top and nose-mounted GPS/APL antennas in the above-defined body-referenced coordinate frame stay fixed, and are represented as at b and anb , respectively.
b b ⎡at x ⎤ ⎡anx ⎤ b ⎢ b ⎥ b ⎢ b ⎥ at = ⎢at y ⎥ an = ⎢an y ⎥ (6.83) ⎢ b ⎥ ⎢ b ⎥ ⎣at z ⎦ ⎣anz ⎦
123
Top-Mounted Antenna
b atz
Nose-Mounted Antenna
b b at an b any b anz Y
(0, 0, 0) b atx IRS Location b anx b at y
X Z
Figure 6.5 Body-frame vectors to describe the locations of the two aircraft antennas
The superscript ‘b’ on the vector denotes its description with respect to the orientation of the body frame. The components of vectors at b and anb were statically surveyed using
Ashtech Z-12 GPS receiver prior to the experiment, and are detailed in Figure 6.3 and
Figure 6.4. The roll (φ ), pitch (θ ), and yaw (ψ ) information obtained from the
Honeywell IRS yields a transformation matrix to convert the vectors from the body-frame to a local-level North-East-Down (NED) frame as follows [Farrell (1976)]
⎡cosψ cosθ − sinψ cosφ + cosψ sin θ sin φ sinψ sinφ + cosψ sin θ cos φ ⎤ Cl,NED = ⎢sinψ cosθ cosψ cosφ + sin ψ sin θ sin φ − cosψ sinφ + sin ψ sin θ cos φ⎥ b ⎢ ⎥ ⎣⎢ − sin θ cos θ sin φ cos θ cos φ ⎦⎥ . 124
(6.84)
The transformed vectors are obtained in a local-level NED frame centered at the origin of the body frame as follows
l,NED l,NED b at = Cb at (6.85)
l,NED l,NED b an = Cb an (6.86)
The subscript ‘l’ on the vector denotes its description in the local-level frame, which in this case is the NED configuration. The components of the above vectors can be obtained in an East-North-Up (ENU) frame by rearranging the order, or performing the following operation
l,ENU l,ENU l,NED at = Cl,NED at (6.87)
l,ENU l,ENU l,NED an = Cl,NED an , (6.88)
l,ENU where the transformation matrix Cl,NED is defined as
⎡0 1 0 ⎤ Cl,ENU = ⎢1 0 0 ⎥ l,NED ⎢ ⎥ . (6.89) ⎣⎢0 0 −1⎦⎥
Let the vectors bt l,ENU and bn l,ENU denote the top-mounted and nose-mounted aircraft
GPS antennas in a local-level ENU frame centered at the APL transmission antenna, respectively. The various vectors expressed in the navigation frame are shown in Figure
6.6.
125
Top-Mounted Antenna
atl,ENU U
anl,ENU N btl,ENU
E U bnl,ENU Nose-Mounted Antenna Body-frame Origin N
E Local-level frame Origin (APL Transmission Antenna)
Figure 6.6 Coordinate-frame vectors to descibe the locations of the two aircraft GPS/APL
reception antennas
The unknown quantity that needs to be determined is the vector bn l,ENU in Figure 6.6, which is obtained as
l,ENU l,ENU l,ENU l,ENU bn = bt − at + an (6.90)
The vectors at l,ENU and an l,ENU were obtained using Equations 6.85 and 6.86, respectively, and the vector bt l,ENU was obtained by transforming the kinematically post- processed Ashtech Z-12 data from the Earth-Centered-Earth-Fixed (ECEF) frame to the desired local-level ENU coordinate frame. Note that the vector resulting from the operation (bt l,ENU − at l,ENU ) denotes the origin of the body-frame coordinate system expressed in local-level ENU frame centered at the APL transmission antenna. 126
6.3 The Flight Profile
The flight-test data presented was collected on 03 and 04 March 2003. The ground track of the DC-3 for the duration of the flight on 03 March 2003 is shown in Figure 6.7. The
East and North ranges are plotted in nautical miles from the base of the FLD7 (at the
LGF), which was used as the reference location. The “solid-straight” trace indicates the locations at which is the DGPS-only position solution was available, and the trace with the "circles" indicates the locations where the WBAPL was available for inclusion in the differential position solution (labeled as DGPS/DAPL). The flight profile consisted of a series of low approach patterns along Runway 07 at UNI, with the last approach (short range loop) being the landing track.
4
2
0
-2 ) i m
(n -4 h rt No -6
-8 DGPS -10 DGPS/DAPL
-12 -18 -16 -14 -12 -10 -8 -6 -4 -2 0 2 East (nmi)
Figure 6.7 Ground track of the DC-3 on 03 March 2003
127
10
8
6
4
2 ) i m
(n 0 h rt
No -2
-4
-6 DGPS -8 DGPS/DAPL
-10 -20 -15 -10 -5 0 5 10 15 20 East (nmi)
Figure 6.8 Ground track of the DC-3 on 04 March 2003
The flight profile for 04 March 2003 included of four low approach patterns on Runway
07. The ground track of the DC-3 for the duration of the flight is shown in Figure 6.8.
6.4 Results of the Power-Induced Measurement Error Investigation
The measurement model used for the analysis of the power-induced measurement error was described in Section 5.2. The expression for the PR DD is given by Equation 5.10. In the case of the laboratory tests, several terms in Equation 5.10 were constants: the true differential ranges (since a common simulated signal was split to both the air and ground subsystems), and the differential tropospheric correction (as the WBAPL signal generator did not model any tropospheric delay for the laboratory test). These parameters had to however be determined for the flight-test analysis. The differences in the true ranges were readily obtained, as the location of the ground reception antennas were surveyed prior to 128 the experiment, and truth position of the nose-mounted airborne antenna was determined, as discussed in Section 6.2. The differential tropospheric models described in Chapter 4 were utilized to determine the corrections for both the WBAPL and for GPS, as presented earlier in Equation 5.10.
ps p p s s Tag = Ta −Tg −Ta + Tg
6.4.1 Ground Subsystem Performance
The implementation of the pulser/blanker circuitry at the LGF kept the ground WBAPL tracking in the non-saturating region of the receiver’s dynamic range, thus mitigating the high-power induced error on the ground measurements. In order to verify the effectiveness of the mitigation on the ground subsystem, a DD analysis was performed using measurements from two LGF antennas. Since each antenna site consists of an HZA and an MLA, a DD analysis was performed separately using the HZAs first, followed by one using the MLAs. The expression for the SD between two ground WBAPL PR measurements would be similar to Equation 5.4 obtained for the laboratory analysis. The true ranges necessary for the determination of the DD error are readily obtained in this case, as the locations of the ground reception antennas were surveyed prior to the experiment. The differential tropospheric errors between the two ground antennas are ignored since they are at the same height.
Figure 6.9 is a plot of the PR DD obtained for the WBAPL using measurements from the
HZA antennas at FLD5 and FLD6 over approximately an hour. As the HZA antennas have a better coverage of high-elevation satellites, a high elevation satellite (GPS SV
#11) was chosen as the reference in this case. Figure 6.10 shows the polar sky-plot of the 129 constellation for the data collection period, from which the location of SV 11 is seen to be good for processing the HZA data; the traces of the SVs are terminated with an asterisk. The mean of the DD error plotted in Figure 6.9 was determined to be -0.18 m, and standard deviation 0.22 m.
Figure 6.9 WBAPL PR DD errors using the HZA antennas from FLD5 and FLD6, with
GPS SV #11 used as the reference.
Although similar performance would have been observed if both the HZA antennas observed exactly the same bias, the assessment that the ground receivers were not in the saturation region of operation ensured that the performance seen above was indeed reflective of error free (power-induced) ground WBAPL measurements. The laboratory 130 results show that the power-induced error is exhibited only when the receiver is operating in a heavily saturation region. These results when compared to those obtained from previous experiments with the WB-code that did not incorporate pulse blanking on the ground [Kiran and Bartone (2002), Bartone and Kiran (2001)] show superior performance.
Figure 6.10 Sky-plot for the duration of the test conducted at UNI on March 03, 2003
Figure 6.11 is a plot of the PR DD obtained for the WBAPL using measurements from the MLA antennas at FLD5 and FLD6, over the same period. As the MLAs have a better 131 coverage of low-elevation satellites, a low elevation satellite (GPS SV #26) was chosen as the reference in this case. As seen in the HZA DD case in Figure 6.9, there is no significant bias in ground measurements, thus validating the technique applied at the LGF to mitigate the high-power induced PR error. The mean of the DD error plotted in Figure
6.11 was determined to be -0.17 m, and standard deviation 0.15 m.
Figure 6.11 WBAPL PR DD errors using the MLA antennas from FLD5 and FLD6, with
GPS SV #26 used as the reference.
132
6.4.2 Airborne Subsystem Performance
A PR DD analysis similar to that described for the laboratory tests (described in Section
5.2) was performed using measurements obtained via the nose-mounted antenna and the ground HZA at FLD6. The HZA was chosen for the analysis since the 45-deg tilted nose- mounted antenna appeared to track GPS SVs at higher elevation angles with better consistency. Figure 6.12 shows the plot of the measured WBAPL PR DD error (modeled as Equation 5.10) over the approach path along Runway 07, with GPS SV #11 used as the reference satellite for the DD computation. Note that the same GPS SV used as reference for the ground HZA performance analysis is used in the airborne analysis as well. The exhibition of a power-dependent error is observed when the range of the aircraft is within approximately 7 nmi of the WBAPL transmitter. The PL DD error plot in Figure 6.12 also shows a fixed bias through the entire approach, for all the approaches. The causes of this bias need investigation.
133
Figure 6.12 WBAPL PR DD errors using the nose-mounted antenna and HZA at FLD6,
with GPS SV #11 used as reference
To preclude the observation of the bias in Figure 6.12 from erroneous translation of the truth position from the top to the nose-mounted antenna location, a similar DD analysis was performed using two sets of GPS measurements. Retaining the same reference SV
(#11), measurements from another high elevation satellite, GPS SV #28, were used to arrive at a GPS DD, the results of which are plotted in Figure 6.13. It is seen from the plot that no such bias is exhibited in this case. Other possible causes of the bias therefore need to be investigated.
134
Figure 6.13 PR DD errors for GPS SV #28 using the nose-mounted antenna and HZA at
FLD6, with GPS SV #11 as the reference
Figure 6.14 shows the DD error obtained from the utilization of raw PR at the beginning of an approach followed by a subsequent switch to time-differenced CP measurements at the PHP to complete the approach, as explained earlier. The PHP in this case was chosen to be a point on the approach path at which the aircraft was at a distance of 7 nmi from the reference location. No PR-smoothing filter was applied prior to the PHP. The raw PR value at the transition point was used as the initial value for the time-differenced CP propagation.
135
Figure 6.14 WBAPL DD errors using the nose-mounted aircraft antenna and HZA at
FLD6 utilizing raw PR measurements prior to the PHP and CP measurements post-PHP,
with SV #11 used as the reference
It is important to note that the performances seen in Figure 6.12 and Figure 6.14 are representative of worse case scenarios, since they were obtained using the nose-mounted antenna, into which the WBAPL signal arrives at a high-gain region in the reception pattern when compared to the conventional top-mounted antenna. The LAAS airborne antenna is top-mounted, and for an aircraft on final approach, the low-elevation WBAPL signal would arrive at a low-gain region of its pattern [Bartone (1996)]. The WBAPL signal power would therefore be lower for this RF path. In order to gauge the impact of the high-power induced error on the LAAS system performance, an analysis similar to 136 that conducted above was repeated for the top-mounted antenna. As a Novatel
MillenniumTM GPS/APL receiver connected to the top-mounted antenna was recording measurements through the test, this was performed with no difficulty.
The same reference GPS SV (#11) used for DD analysis of the nose-mounted antenna data was employed for the analysis using the top-mounted antenna measurements as well.
The DD error obtained using only the raw PR measurements through the complete approach is plotted in Figure 6.15. It can be seen from Figure 6.15 that an error is exhibited as the aircraft approaches the transmission antenna, but it is not as substantial as that seen on the measurements obtained via the nose-mounted antenna (see Figure
6.12). In order to account for the difference in performance, the antenna gains at the two angle-of-arrivals were compared. For an aircraft aligned along a 3-deg approach path, the signal arrives at the nose-mounted antenna at approximately 45 degrees, while it arrives at the top-mounted antenna at approximately 0 degrees. The difference in gain at these two elevation angles for the GPS L1-L2 patch antenna used was calculated to be approximately 4 dB. This difference in gain does not completely account for the performance difference. Another issue taken into consideration was signal creeping along the aircraft skin prior to arrival at the top-mounted antenna for a source below the horizon, which could result in the reception of a reduced signal power level. A thorough answer would require an experimental set-up to isolate each of the error sources, and would be an issue to consider for future work.
137
Figure 6.15 WBAPL PR DD errors using the top-mounted antenna and HZA at FLD6,
with GPS SV #11 as the reference
Figure 6.16 shows the DD error obtained from the utilization of raw PR at the beginning of an approach followed by a subsequent switch to time-differenced CP measurements at the PHP, using the top-mounted aircraft antenna and the HZA antenna at FLD6. As for the plot in Figure 6.14, the PHP was chosen to be a point on the approach path at which the aircraft was at a distance of 7 nmi from the reference location. No PR-smoothing filter was applied prior to the PHP. As expected, the variation in the DD error is minimal after the switch to the CP observable. The errors seen in Figure 6.16 are indicative of what can be expected for WBAPLs in the LAAS airborne segment. 138
Figure 6.16 WBAPL DD errors using the top-mounted antenna and HZA at FLD6
utilizing raw PR measurements prior to the PHP and CP measurements post-PHP, with
SV #11 used as the reference
It is seen from Figure 6.16 that the variation in the DD error is minimal after the switch to the CP observable. Integrity evaluations can therefore be based on the performance of the system at the PHP, after which system error performance is dictated by the CP measurement errors. It can be seen from Figure 6.16 that there is a small slope in the DD error even after the PHP. This is attributed to the combined effects of the power-induced 139 error on the CP measurement and the residual tropospheric error, both of which are relatively small.
6.5 Differential Positioning Accuracy Assessment
The performance of the prototype ground subsystem was assessed using B-values for each of the SVs, computed for each of the two LGF sites used for the tests. The airborne subsystem performance was assessed using the differential position solution obtained with and without the inclusion of the APL, in order to ascertain the affect of the APL on the solution. A comparison of the proposed approach using the combination of CSC and
TDCP architectures with one that relied solely on the CSC architecture was undertaken.
Results from these analyses are presented in Section 6.5.1 through Section 0.
6.5.1 Performance of the Ohio University Prototype WBAPL LAAS Ground
Subsystem
The performance of the prototype ground subsystem over the duration of the test was assessed in terms of the B-values of the PR corrections obtained for each of the SVs. The
B-values were computed in accordance with the LAAS integrity methodology described in Reference [RTCA LAAS MASPS (1998)], and were used as a test statistic to verify the consistency of the PR corrections, as computed for reception via each of the LGF antennas. A two-site LGF configuration yields two sets of B-values, B1 and B2, which in this case correspond to sites FLD5 and FLD6, respectively. As B1 and B2 are symmetric about zero, statistical description of one is adequate to describe the ground subsystem performance. Table 6.1 contains the overall summary of the B-values obtained for 140
WBAPL reception via the HZA antennas and MLA antennas on the ground. The performance of a typical GPS SV is included for comparison.
Table 6.1 Summary of prototype WBAPL LAAS ground B-value performance obtained
for both 03 and 04 March 2003 (computed over the test-duration of approximately 1.5
hours)
03 March 2003 04 March 2003
µ of B1 (m) σ of B1 (m) µ of B1 (m) σ of B1 (m)
APL on HZA -0.10 0.10 -0.42 0.07
APL on MLA -0.02 0.06 -0.11 0.09
GPS SV # 11 0.00 0.11 -0.01 0.10
GPS SV # 7 -0.02 0.10 -0.02 0.09
GPS SV # 26 0.02 0.07 0.04 0.07
GPS SV # 28 -0.01 0.10 -0.02 0.10
The standard deviation of the ground PR measurements can be determined from the B- values using [RTCA LAAS MASPS (1998)]
MN(M −1) σ = σ PR B N +1 (6.91) where M is the number of reference receivers used (2, in this prototype set-up), and N is the number of satellites used in the estimation of the receiver clock bias. It is seen from
Table 6.1 that the standard deviations of the B-values obtained for the WBAPL on both the HZA and MLA are limited to 0.10 m. The number of satellites visible varied between 141
7 and 9 (see Figure 6.17) for this data collection effort. Using a fixed value of 8 for M , the standard deviation of the ground measurements is computed to be
2×8 σ = 0.1 = 0.13 m . PR 9
Recall from Chapter 4 that the contribution from single-bounce ground multipath was estimated to be limited to 0.06 m.
6.5.2 Performance of the Ohio University Prototype WBAPL LAAS Airborne
Subsystem
The baseline LAAS architecture utilizes the carrier-smoothed code architecture to obtain both the ground and airborne PR estimates with reduced noise. In this prototype architecture, it is proposed that the carrier-smoothed code approach be used up to a certain point along the approach path (i.e. the PHP), after which the airborne position filter rely on time-difference CP measurements, as described in Section 4.2.2.2.
The performance of the developed prototype airborne subsystem was assessed using the differential-position solution errors. Two separate position solutions were computed simultaneously in the aircraft: a DGPS-only solution (not inclusive of the WBAPL) and a
DGPS position solution inclusive of the WBAPL. Position errors obtained from both the solutions are presented in order to assess the affect of WBAPL inclusion on the system error performance. The VDOP values with and without the inclusion of the WBAPL in the position solution are also presented in order to gauge the affect of WBAPL inclusion on the system geometry. Flight test results obtained under two conditions are presented: 142 one that used all the visible GPS SVs, for which a mask angle of 5 deg was applied, and the other that relied on a subset arrived at by increasing the mask angle from 5 to 20 deg.
6.5.2.1 Data Analysis Using all Available SVs (Mask Angle = 5 deg)
This section details the performance obtained using all available ranging sources, which include all available GPS SVs and 1 WBAPL. Figure 6.17 shows the number of ranging sources that were available during each of the approaches conducted over the two days.
The first six approaches in Figure 6.17 were conducted on the first day of the test, 03
March 2003, and the next four on the following day.
Figure 6.17 Number of SVs used in the position solution for each of the ten approaches,
for the analysis conducted using all the available ranging sources
143
Figure 6.18 shows the DGPS position errors using the CSC approach throughout the approach. The errors are obtained and plotted along the ENU directions vs slant range in nmi from the FLD7 location at the LGF for ten approaches, using the post-processed dual-frequency kinematic Ashtech Z-12 data as truth. These errors were obtained without the inclusion of the WBAPL in the CSC differential position solution, and were used as a benchmark to gauge the effectiveness of the inclusion of the WBAPL in the CSC position solution.
Figure 6.18 DGPS-only position errors obtained using CSC architecture for the entire
approach, for the complete GPS SV set case
144
The performance of the system with the inclusion of the WBAPL in the CSC-based differential position solution is depicted in Figure 6.21, and is designated as
DGPS/DAPL. PR corrections based on ground reception using the MLA antennas (i.e., the APL power controlled path) were applied at the airborne segment to arrive at the differentially corrected WBAPL measurements.
Figure 6.19 DGPS/DAPL position errors obtained using CSC architecture for the entire
approach, for the complete GPS SV set case
Figure 6.20 shows the DGPS position errors obtained using the combination of CSC and
TDCP architectures to complete an approach. These errors were obtained without the inclusion of the WBAPL in the differential position solution, and were used as a 145 benchmark to gauge the effectiveness of the inclusion of the WBAPL in the position solution. At the onset of each of the approaches, the position solution was determined using carrier-smoothed code measurements. Whereas the ground PR measurements were smoothed using a time-constant of 100 sec, the airborne PR measurements were filtered using a time-constant of 2 sec. After the PHP, the positiong filter relied on time- differenced CP measurements, as described in Chapter 5. The PHP was chosen to be at a distance of 7 nmi from the reference point, and is mainted so for all the plots presented through the remainder of this chapter.
Figure 6.20 DGPS-only position errors obtained using the combination of CSC and
TDCP architectures to complete an approach, for the complete GPS SV set case
146
The performance of the system using the combination of CSC and TDCP, with the inclusion of the WBAPL in the differential position solution is depicted in Figure 6.21, and is designated as DGPS/DAPL. PR corrections based on ground reception using the
MLA antennas (i.e., the APL power controlled path) were applied at the airborne segment to arrive at the differentially corrected WBAPL measurements.
Figure 6.21 DGPS/DAPL position errors obtained using the combination of CSC and
TDCP architectures to complete an approach, for the complete GPS SV set case
The DGPS and DGPS/DAPL error plots of Figure 6.20 and Figure 6.21 show spikes that are especially evident for a particular approach. At these particular times, it was determined that the residual of the truth data processing was higher than usual, and was 147 found to be approximately 0.3 m. The reasons for this increase were not investigated. In order to verify that the inclusion of the APL in the differential position solution was not the source of the error, a plot of the difference in errors with and without the inclusion of the APL was obtained. This is plotted in Figure 6.22, and shows that the source of the error commonly affects both the analyses, since the traces are smooth.
Figure 6.22 Difference between the DGPS and DGSP/DAPL solutions for the complete
SV set case, obtained using the combination of CSC and TDCP for each approach
Figure 6.23 illustrates a typical VDOP reduction (for a single approach) when the
WBAPL is included in the differential position solution, under the condition that all the available ranging sources are included in the position solution. As can be seen in Figure 148
6.23, the VDOP is relatively small to begin with, and therefore the improvement in geometry, although visible, is not significant in this case. Vast geometry improvements are obtained from the inclusion of a WBAPL when the VDOP is high to begin with, as will be seen in the following section that analyses the influence of a WBAPL with a reduced constellation size.
Figure 6.23 Typical VDOP Reduction Using WBAPL
Table 6.2 contains the overall summary of the performance (with respect to accuracy) of the prototype LAAS, with and without the inclusion of the WBAPL in position determination, for the full constellation case. For each of the ten approaches conducted, the mean ( µ ), standard deviation (σ ), and total error (| µ |+ 2σ ) were computed over 149 the range 7 to 0 nmi from the reference location. The averages of the values obtained over the ten approaches are listed in Table 6.2. The rationale for using | µ |+ 2σ for each run and then averaging the results to obtain the total error was adopted in order to capture the biases that are seen from run to run. By solely averaging the biases across the runs, it is possible for the positive and negative biases from separate runs to cancel each other out. Calculating | µ |+ 2σ for each run, and then averaging them to a single entry in
Table 6.2 avoided this bias cancellation.
The solution that utilized the combination of CSC and TDCP architectures showed a total error of 0.84 m and 1.06 m in the lateral (LAT) and vertical (VERT) directions, respectively, for the DGPS only case. The corresponding solution that included the
WBAPL meanwhile exhibited a total error of 0.99 m in the horizontal and 1.12 m in vertical directions. For the reduced constellation case, utilization of only the CSC architecture throughout the approach showed a total error of 0.62 m and 1.25 m in the lateral (LAT) and vertical (VERT) directions, respectively, for the DGPS only case. The corresponding solution that included the WBAPL exhibited a total error of 0.66 m in the horizontal and 0.74 m in vertical directions.
150
Table 6.2 Summary of WBAPL-inclusive LAAS position solution performance for the
full constellation case (mask angle = 5 deg)
CSC Solution Solution using CSC and TDCP
DGPS Only DGPS/DAPL DGPS Only DGPS/DAPL (m) (m) (m) (m)
µ (LAT) 0.49 0.58 0.48 0.50
µ (VERT) -0.94 -0.09 -1.13 -0.30
σ (LAT) 0.17 0.20 0.07 0.08
σ (VERT) 0.36 0.43 0.12 0.09
µ + 2σ (LAT) 0.84 0.99 0.62 0.66
µ + 2σ (VERT) 1.06 1.12 1.25 0.74
6.5.2.2 Data Analysis Using a Reduced GPS SV set (Mask Angle = 20 deg)
This section details the LAAS performance obtained using a reduced GPS SV subset from the available constellation, and the WBAPL as a ranging source. For this analysis, a mask angle of 20 deg was implemented in a post-process fashion, by lowering the mask angle for all the HZA measurements to 20 deg and eliminating all the MLA measurements from the position calculation of the aircraft. This implementation of a high mask-angle reduces the number of GPS SVs used in the position calculation, and could be used to simulate a variety of conditions: poor geometry, an interference source that may effect the MLA GPS/APL receiver measurements to a higher degree than the HZA
GPS/APL receiver measurements, an MLA antenna or receiver failure, or a simple LAAS configuration with only one antenna per site. For this analysis, PR corrections and B- 151 values from only the HZA were used for GPS SVs included, and for the WBAPL. Figure
6.24 shows the number of ranging sources that were available after the exclusion process.
Figure 6.24 Number of SVs used in the position solution for each of the ten approaches,
for the analysis conducted using the reduced SV set
Figure 6.25 shows the DGPS position errors using the CSC approach throughout the approach. The errors are obtained and plotted along the ENU directions vs slant range in nmi from the FLD7 location at the LGF for ten approaches. These errors were obtained without the inclusion of the WBAPL in the CSC differential position solution.
152
Figure 6.25 DGPS-only position errors obtained using CSC architecture for the entire
approach, for the reduced GPS SV set case
The performance of the system with the inclusion of the WBAPL in the CSC-based differential position solution, for the reduced SV set case, is depicted in Figure 6.26. PR corrections based on ground reception using the MLA antennas (i.e., the APL power controlled path) were applied at the airborne segment to arrive at the differentially corrected WBAPL measurements. It is clearly evident from Figure 6.26 that the errors in the APL PR measurement significantly affect the differential position solution as the aircraft approaches the APL transmission antenna.
153
Figure 6.26 DGPS/DAPL position errors obtained using CSC architecture for the entire
approach, for the reduced GPS SV set case
Figure 6.27 shows the ENU position errors that were obtained using the combination of
CSC and TDCP architectures to complete an approach, with only the GPS SV measurements incorporated into the position solution. The performance of the system prior to the PHP is seen to degrage in comparison to the full constellation case, but the performance after transition to sole reliance on the CP measurement stays comparable.
The reduced noise on the CP measurement makes-up for the increase in DOP parameters from the poor geometry that results from a reduced constellation set.
154
Figure 6.27 DGPS-only position errors obtained using the combination of CSC and
TDCP architectures to complete an approach, for the reduced GPS SV set case
Figure 6.28 shows the error plots obtained using the combination of CSC and TDCP architectures, with the WBAPL included in the solution for the reduced constellation case. It can be observed that the performance prior to the PHP was vastly improved upon from the inclusion of the WBAPL, but the performance after the PHP stays comparable.
It is emphasized that WBAPL inclusion within LAAS is sought not on the basis of accuracy enhancement, but rather as a means to increase system availability. If the
WBAPL measurements at both the ground and air subsystems are of acceptable quality, availability enhancement is achieved from the presence of an additional ranging source.
155
Figure 6.28 DGPS/DAPL position errors obtained using the combination of CSC and
TDCP architectures to complete an approach, for the complete GPS SV set case
Table 6.3 contains the overall performance summary of the prototype LAAS, with and without the inclusion of the WBAPL in position determination, with a reduced constellation size. The statistics listed in Table 6.3 were obtained using the same procedure as explained for Table 6.2. Utilization of only the CSC architecture throughout the approach showed a total error of 0.95 m and 0.88 m in the lateral (LAT) and vertical
(VERT) directions, respectively, for the DGPS only case. The corresponding solution that included the WBAPL exhibited a total error of 1.05 m in the horizontal and 1.98 m in vertical directions. The solution that utilized the combination of CSC and TDCP architectures showed a total error of 0.70 m and 1.17 m in the lateral (LAT) and vertical 156
(VERT) directions, respectively, for the DGPS only case. The corresponding solution that included the WBAPL exhibited a total error of 0.78 m in the horizontal and 0.81 m in vertical directions.
Table 6.3 Summary of WBAPL-inclusive LAAS position solution performance for the
reduced-constellation case (mask angle = 20 deg; data analyzed over last 7.0 to 0 nmi)
CSC Solution Solution using CSC and TDCP DGPS Only DGPS/DAPL DGPS Only DGPS/DAPL (m) (m) (m) (m)
µ (LAT) 0.58 0.60 0.55 0.62
µ (VERT) -0.62 0.47 -0.58 0.28
σ (LAT) 0.19 0.22 0.07 0.08
σ (VERT) 0.61 0.75 0.08 0.14
µ + 2σ (LAT) 0.95 1.05 0.70 0.78
µ + 2σ (VERT) 0.88 1.98 1.17 0.81
157
7. SUMMARY AND RECOMMENDATIONS
This dissertation documents the development of a WBAPL architecture for inclusion within a prototype LAAS. The proposed architecture was developed, flight-tested and analyzed to demonstrate the feasibility of its incorporation into LAAS. As part of this effort, the first documented real-time integration of a WBAPL into a LAAS was accomplished in 2000, and several follow-up flight-tests were conducted to show an improvement in system performance from the inclusion of differentially corrected APLs into the position solution for the proposed architecture.
As part of the research documented in this dissertation, a unique technique to determine the APL transmitter clock offset at the LGF and include the information in the VDB message to enable direct-wideband acquisition by airborne receivers was developed. The power-dependent measurement error exhibited by the receivers utilized for APL development at Ohio University was characterized, following which a solution to the problem for the ground segment was achieved using APL power control. A separate solution for the airborne segment was determined that involved to switch to a time- differenced CP algorithm prior to the error influencing the airborne position solution.
The developed prototype architecture was flight-tested, and the data was analyzed under two conditions: one in which all the visible SVs were utilizes in the differential position solution; and the second in which only SVs with elevation angles higher than 20 degrees were utilized. With the complete SV set, the CSC solution yielded a total error (obtained by averaging | µ | +2σ for each of the ten low approaches) of 0.84 m in the lateral, and 158
1.06 m in the vertical direction, with the WBAPL not included in the solution. The errors obtained with the inclusion of the WBAPL were 0.99 m and 1.12 m in the lateral and vertical directions, respectively. On utilization of the TDCP algorithm from the PHP, the
DGPS-only errors obtained were 0.62 m and 1.25 m in the lateral and vertical directions, respectively.
The presence of biases in the airborne GPS PR measurements resulted in position error on the order of a few meters, and this solution was used to initialize the TDCP solution.
The switch to the CP measurements reduces the variance in the position solution, but the initial bias remains through the approach, resulting total errors that could be larger for the
TDCP solution when compared to the CSC solution. The incorporation of the WBAPL into the TDCP-based solution yielded total errors of 0.66 m and 0.74 m in the lateral and vertical directions, respectively.
For the reduced SV scenario, the CSC solution using the DGPS-only case yielded total errors of 0.95 m and 0.88 m in the lateral and vertical directions, respectively. The errors obtained with the inclusion of the WBAPL were 1.05 m and 1.98 m in the lateral and vertical directions, respectively. The absence of low elevation angle ranging sources other than the APL results in increased influence of the APL range measurement on the vertical position solution. The presence of large power-dependent APL PR measurement errors causes large DGPS/DAPL position errors. On utilization of the TDCP algorithm from the PHP, the DGPS-only errors obtained were 0.70 m and 1.17 m in the lateral and 159 vertical directions, respectively. The incorporation of the WBAPL into the TDCP-based solution yielded total errors of 0.78 m in the lateral, and 0.81 m in the vertical directions.
The incorporation of the WBAPL into the reduced constellation was therefore shown to improve the resulting differential position solution, and positively influence the availability of the system.
Recommended directions for further research include
• Studies on the optimal siting of an APL on airport property. The siting of an APL
would be guided by the need to obtain an improvement in ranging source
geometry, while at the same time fulfilling the need to maintain line-of-sight
between the APL transmission antenna and the top-mounted aircraft antenna all
the way from the edge of the coverage region through the termination of the
guidance operation. Such a study could also look in to the potential and
constraints faced in meeting the need to service multiple runways at an airport.
• Studies on the constraints faced in siting multiple APLs on an airport.
• A thorough investigation into the degree of interference posed to GPS reception
by participating and non-participating GPS receivers from the utilization of
pulsed, wideband APL signals.
• An investigation on the source of the power-dependent measurement errors within
receivers. 160
• Investigate synergistic development of the APL architecture with L5 receiver
architectures, which will have to incorporate a substantial degree of pulse
interference capability. 161
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