IEEE INTERNET OF THINGS JOURNAL, VOL. 6, NO. 6, DECEMBER 2019 9503 Efficient Design of Chirp for Low-Power Wide-Area Networks Tung T. Nguyen , Ha H. Nguyen , Senior Member, IEEE, Robert Barton, and Patrick Grossetete

Abstract—LoRa is an abbreviation for low power and long monitoring, etc.), as well as in industrial applications [1], [2]. range and it refers to a communication technology developed for References [1] and [2] review prominent systems and indus- low-power wide-area networks (LPWANs). Based on the prin- try standards for LPWANs, including Sigfox [3], Ingenu [4], ciple of chirp spread spectrum (CSS), LoRa technology is very attractive to provide low bit-rate wireless connections over an DASH [5], and LoRa [6]. This paper is specifically concerned extended communication range and under very low power con- with the physical layer of LoRa because this technology is sumption. While the (MAC) layer of LoRa gaining tremendous commercial growth in more than 100 specifications is open for developers, the physical layer is not. In countries around the world.1 particular, LoRa modulation and techniques are Examining the LoRaWAN standard reveals that the standard patented by Semtech and have not been mathematically described in detail. This paper presents novel approaches to modulate mainly describes medium access control (MAC) layer proto- and demodulate LoRa signals with very high implementation cols with a set of network architectures [7], while the rest efficiency, great flexibility, and excellent performance. In partic- of the standard, especially the physical layer (PHY) archi- ular, compared to the commercially available receiver made by tecture, is left for open development. The only requirement Semtech, the proposed design is shown to yield a saving of trans- imposed on LoRa PHY is the use of chirp spread spectrum mitted power from 0.9 to 2.5 dB over the spreading factor (SF) range of 6–12. Moreover, this paper suggests a method to exploit (CSS) as the modulation technique. CSS is known for its the phase information of CSS signals to encode extra information flexibility in providing tradeoffs between reception sensitivity bits, leading to throughput improvement over the conventional and throughput. Spreading factor (SF) is the most important CSS system, for example, by 33%, 25%, 20%, and 17% for SFs parameter in CSS modulation. Increasing SF can significantly of 6, 8, 10, and 12, respectively. extend the communication range, but it comes at the cost of Index Terms—Chirp spread spectrum (CSS), digitally con- a lower transmission rate. BW is another adjustable parame- trolled oscillator (DCO), Internet of Things (IoT), LoRa, orthog- ter. Using a larger BW enhances the communication speed (as onal chirp generator (OCG). expected) and, at the same time, provides better immunity to narrow-band noise and ingress. The LoRa network is expected to exploit the modulation flexibility of CSS to optimize the I. INTRODUCTION network capacity. The evaluation of link performance as well OW-POWER wide-area networks (LPWANs) have as system-level performance of LoRaWAN can be found in [8]. L recently emerged as a promising communication solu- The currently commercialized LoRa PHY solution [9] is tion for many Internet of Things (IoT) applications. LPWANs patented by Semtech [10]. While the design promises reliable are designed to achieve large coverage ranges, extend bat- low-power communication over a long distance, it has poor tery lifetime of end-devices, and reduce the operational spectral efficiency because of two main reasons. cost of traditional cellular networks. By exploiting the sub- 1) The LoRa CSS signal occupies a much larger BW than 1 GHz unlicensed, industrial, scientific and medical (ISM) required for a CSS signal. Shown in [11] as an example, frequency band and sporadically transmitting small packets the 500-kHz CSS signal occupies more than 700 kHz at low data rates, these networks can be operated with very of the configured BW due to the large roll-off regions low reception sensitivities. The long-range and low-power on both sides of the spectrum, i.e., 100 kHz on each properties of LPWANs make these networks an interesting side. The roll-off region creates a gap between channels, candidate for smart sensing technology in civil infrastruc- preventing them from being placed close together. As a tures (such as health monitoring, smart metering, environment result, there is a smaller number of channels that can be used for a given spectrum resource unless the roll-off is Manuscript received April 11, 2019; revised May 23, 2019, June 4, reduced. 2019, and June 30, 2019; accepted July 6, 2019. Date of publication July 17, 2019; date of current version December 11, 2019. This work was 2) The generated chirps in LoRa CSS appear to be supported by the Natural Sciences and Engineering Research Council of nonorthogonal [12], which causes performance degra- Canada/Cisco Industrial Research Chair in Low-Power Wireless Access for dation when compared to the conventional orthogonal Sensor Networks. (Corresponding author: Ha H. Nguyen.) T. T. Nguyen and H. H. Nguyen are with the Department of Electrical and frequency-shift keying (FSK) system [13]. Computer Engineering, University of Saskatchewan, Saskatoon, SK S7J 4E3, To guarantee orthogonality among the chirps, there are Canada (e-mail: [email protected]; [email protected]). proposals that employ the discrete Fresnel transform, i.e., R. Barton is with Cisco Systems Canada, Vancouver, BC, Canada. P. Grossetete is with Cisco Internet of Things Business Unit, Paris, France. Digital Object Identifier 10.1109/JIOT.2019.2929496 1https://lora-alliance.org/ 2327-4662 c 2019 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See http://www.ieee.org/publications_standards/publications/rights/index.html for more information. 9504 IEEE INTERNET OF THINGS JOURNAL, VOL. 6, NO. 6, DECEMBER 2019 orthogonal chirp-division [14], or a hybrid com- II. CHIRP SPREAD SPECTRUM SYSTEM bination of FSK and CSS [13]. Unfortunately, hardware A. CSS Modulation resource to implement the discrete Fresnel transform is costly The CSS modulation used in LoRa converts each data sym- and thus is not suitable for low-cost, low-power applica- bol into a chirp, which is a signal whose frequency linearly tions expected for LoRa technology. The hybrid FSK and increases or decreases over time. A chirp is also called a sweep CSS solution significantly increases the occupied BW in signal and one CSS symbol sweeps through the BW once. exchange for orthogonal signaling, thus the spectral effi- When the instantaneous frequency of a CSS signal reaches the ciency is not improved. Although Semtech claims to have a highest, it will wrap over and start from the lowest frequency. design that produces orthogonal CSS chirps [15], [16], the SF is the most important parameter of the CSS system. The underlying method is not publicly known since the trans- CSS modulation order is defined as M = 2SF, which means mitter employs a look-up table (LUT) to store samples of that each CSS symbol carries SF bits. At baseband, each CSS the chirp. symbol contains M complex samples, which are sent out at Although the LUT design appears to be simple, the flexibil- a rate equal to the BW of the signal. Thus the CSS sym- ity requirement of CSS modulation would lead to more expen- bol duration is given as T = (M/BW) (seconds). Then the sive transceivers, since extra hardware resource is required to sym chirp rate, i.e., the rate at which the frequency of a CSS signal support multiple SF and BW settings. Generally, the size of changes over time, can be defined as an LUT increases exponentially with SF. While this might not pose a serious problem for SF values from 6 to 12 as sup- BW BW2 μ = = (Hz/s). (1) ported by available commercial LoRa transceivers, the LUT T M design will become exceedingly costly when higher SF values sym are desired. CSS modulation produces different chirps based on the Against the above background, the first part of this paper basic chirp. The basic chirp is a chirp that starts at the low- (Section II) presents an overview of CSS modulation and est frequency, i.e., −BW/2, sweeps through the entire BW, discusses important aspects such as orthogonality of CSS and then stops at the highest frequency, i.e., BW/2. As such, signaling, theoretical bit-error-rate (BER) evaluation for both the basic chirp at baseband is centered at zero frequency and coherent and noncoherent detection, continuous phase criterion defined by the following continuous-time waveform: and spectra of CSS signals. The property of phase continuity ⎧ ⎫ ⎪ ⎪ is discussed in Section III. Section IV presents an efficient ⎨⎪   ⎬⎪ design that allows supporting multiple SFs and BWs at very μt BW x0(t) = exp j 2π − t , 0 ≤ t ≤ Tsym (2) low cost. In particular, for a digital communication system, ⎪ 2 2 ⎪ ⎩⎪  ⎭⎪ data symbol modulation is generally done at baseband in a φ (t) complex plane consisting of in-phase and quadrature signals. 0 In fact, this is a must for the LUT design since it is most where φ0(t) is the phase function of the basic chirp. The efficient to store samples at the lowest rate. Then the up- instantaneous frequency of the signal at time t is the phase conversion process is applied to both in-phase and quadrature slope of the signal at that moment. That slope can be obtained signals individually using pairs of filters and digital-to-analog by taking the derivative of the phase function over time, i.e., converters (DACs), i.e., quadrature up-converters. The up-   ∂φ (t) BW converter could be sophisticated since it must be able to 0 = 2π μt − (radians/s) (3) support all LoRa BW options, typically from 7.8 to 500 kHz. ∂t 2 To reduce the cost of transmitter implementation, Section V which corresponds to a frequency of proposes an option to modulate the CSS signals directly to   pass band. This section also discusses how the orthogonal BW f (t) = μt − (Hz). (4) chirp generator (OCG) can also be used in the receiver for 2 de-chirping. The last part of this paper (Section VI) examines how From (1) and (4), it is obvious that: 1) the frequency of the sig- the spectral efficiency of conventional CSS signals can be nal sweeps through the entire BW over the period of Tsym and enhanced. By employing pulse shaping filters, the spectral 2) the center frequency is 0 Hz, i.e., the signal is at baseband. roll-off can be controlled and made to be very small, while Denote the basic chirp as symbol 0. Then the next sym- the orthogonality among chirps is still achieved. This fea- bol, i.e., symbol 1, can be obtained by cyclically time-shifting / ture gives the opportunity to enhance the transmission rate or symbol 0 by an amount of 1 BW, and so on. Overall, there detection performance of the existing CSS signaling scheme. are M different symbols which are collectively defined as    In particular, the pulse shaping filters allow relaxing con-   Tsym tinuous phase criterion so that additional data information xm(t) = x0 t − m mod Tsym , t ∈ 0, Tsym . (5) M can be encoded into the starting phase of each CSS sym- bol. The phase can then be recovered using a coherent Since each baseband chirp is bandlimited to BW/2, it can receiver. Such simple modification does not increase the be sampled at sampling rate Fsamp = BW without any loss occupied BW nor does it change the orthogonality among of information. As such, each continuous-time baseband sym- the chirps. bol xm(t) can be completely represented by M values that are NGUYEN et al.: EFFICIENT DESIGN OF CSS MODULATION FOR LPWANs 9505

are not phase-locked, and w[n] is AWGN noise sample with zero mean and variance N0. For such an input/output model, it is simple to see that the signal-to-noise ratio (SNR) is SNR = (1/N0). De-chirping is performed on every received symbol by mul- tiplying with the complex conjugate of the basic chirp. If the basic chirp is an up-chirp, its conjugation is a down-chirp with Fig. 1. Block diagram of an LUT-based CSS transmitter. the same chirp rate, and vice versa. In essence, de-chirping unwinds the second-order phase function applied on the trans- samples of xm(t) taken at rate Fsamp = BW over the sym- mitted signal, leaving only the constant and linear phase terms. bol duration. In particular, the baseband discrete-time (digital) This is shown as follows: samples of the basic chirp are given as   = ∗ x0[n] = x0(t) vm[n] ym[n]x0[n] t=nTsamp          2 μnT (n + m) n + m = π samp − BW = exp(jψ) exp j2π − + w[n] exp j2 nTsamp 2M 2   2  2    n2 n n2 n = exp j2π − ; n = 0, 1,...,M − 1. (6) × exp −j2π − M M 2 2  2 2  On the other hand, it is simple to show that digital samples 2nm + m2 m = exp(jψ) exp j2π − +ˆw[n] of the mth chirp can be obtained by cyclically shifting the M   2  2  digital basic chirp by m samples. Furthermore, the digital basic m2 m j2πnm chirp repeats itself after every M samples, i.e., x [n + M] = = exp jψ + j2π − exp +ˆw[n] 0 2M 2 M x0[n], ∀n. Therefore the mth digital chirp at baseband can also   be mathematically written as xm[n] = x0[n + m]. constant phase linear phase The above properties of the digital chirps are useful for (8) implementing modulation and demodulation of CSS sig- nals. In particular, a conventional LUT-based CSS transmitter ˆ = ∗ where w[n] w[n]x0[n]isalsoanAWGNnoisesample design [16], [17] is shown in Fig. 1. The most important com- with zero mean and variance N0. Thus, in the absence of ponent of the transmitter is a ROM that stores M samples of noise, the CSS signal after de-chirping is a pure sinusoid the basic chirp as described in (6). CSS modulator starts with a with a frequency of m/M (cycles/sample). Denote the constant 2 channel encoding block that converts binary data into a stream phase term as ψm = ψ + 2π([m /2M] − [m/2]), then symbol of SF-bit symbols. Symbol mapping is then performed by demodulation can be done by performing M-point DFT on the cyclically shifting the basic chirp by an amount equivalent to de-chirped signal to obtain the symbol value, producing a corresponding digital baseband CSS symbol. Each digital baseband symbol is then up-sampled M−   1 1 −j2πnk before being converted to an analog baseband signal. There is V [k] = √ v [n] exp m m M a pair of low-pass filters (LPFs) to remove spectral aliases M n=0 caused by up-sampling. Conversion of the signal from analog M−     1 1 j2πnm −j2πnk baseband to a desired carrier frequency, e.g., 902Ð928 MHz = √ exp{jψ } exp exp m M M for LoRa in North America, is done by mixing the signal with M n=0 a digitally controlled oscillator (DCO) that provides in-phase M−   1 1 −j2πnk and quadrature components of the carrier frequency. Note that + √ wˆ [n] exp it is possible to store the preupconverted basic chirp to elimi- M M  n=0 nate the need for the up-conversion filters, but it comes at the W[k] cost of increasing the ROM size. M−   exp{jψ } 1 j2πn(m − k) = √ m exp + W[k] B. CSS Demodulation M M  n=0 Demodulation of CSS signals requires a ROM that stores √ M exp{jψ } + W[m], if k = m the complex conjugate of the digital basic chirp. Since LoRa = m (9) , . signals have a narrow BW (500 kHz or less), the channel W[k] otherwise can be considered as having a constant power gain across the BW (i.e., a flat channel) and the received signal after being down-converted to digital baseband can be expressed as C. BER Performance Analysis The above analysis shows that all the power of a CSS sym- y [n] = exp(jψ)x [n] + w[n], n = 0, 1,...,M − 1. (7) m m bol is concentrated at a single frequency bin, namely the mth In the above expression, ψ is a random phase rotation caused bin if the transmitted CSS symbol is m, whereas all the other by the fact that the oscillators at the transmitter and receiver M − 1 bins contain only noise. At this point, it is useful to 9506 IEEE INTERNET OF THINGS JOURNAL, VOL. 6, NO. 6, DECEMBER 2019 define a peak SNR (PSNR) as    √ 2 E  M exp{jψm} = = M PSNR 2 (10) |W[k]| N0 which indicates the difference in terms of power between the “power” bin versus the rest (“noise” bins). It is obvious that a higher PSNR would lead to a better chance of detecting the transmitted symbol correctly. The ratio between PSNR and SNR, which is M in linear scale, is called the processing gain. Using a higher SF would lead to a higher processing gain and thus improve the performance of the transmission, but at the cost of a lower transmission rate. The transmitted symbol can be detected coherently or non- coherently. Coherent detection requires the receiver to be Fig. 2. BER performance of 64-ary CSS (i.e., SF = 6) with coherent and phase-locked to the transmitter in order to demodulate the noncoherent detection over an AWGN channel. signal. The coherent receiver has a phase that works together with a phase-locked loop (PLL) to synchronize the phases between the receiver and the transmitter. As such, On the other hand, for noncoherent detection, the BER is [19] M−   coherent detection is often referred to as synchronous detec- M 1 M − 1 (−1)m+1 tion. Noncoherent detection, on the other hand, does not Pb, = non-coh 2(M − 1) m m + 1 exploit the phase information and the decision is made solely  m−1  based on the signal envelope. While it is less complex than m log ME exp − 2 b . (15) coherent detection, the performance is not as good. m + 1 N0 In particular, if the transmitter and the receiver are fully Fig. 2 plots the BER performance of a CSS system with synchronized, i.e., the phase term ψ is known at the receiver, m SF = 6 (i.e., M = 64) obtained with both coherent and non- coherent detection can be carried out as coherent detection. As can be seen, simulation results match exactly with the theoretical results in (13) and (15). Observe = {− ψ }. dcoh argmax Vm[k] exp j m (11) that, for this particular case of M = 64, coherent detection k=0,...,M−1 yields a performance gain of approximately 0.6 dB over the Otherwise, a simpler noncoherent detection is simpler noncoherent detection. Before closing this section, we point out that a similar     description of LoRa modulation and noncoherent detection dnon-coh = argmax Vm[k] . (12) k=0,...,M−1 is presented in [20]. Reference [20], however, does not dis- cuss coherent detection nor does it provide theoretical BER The above description and analysis clearly shows that expressions of the two detection methods. CSS is an M-ary orthogonal modulation. As such, its error performance in an AWGN channel is exactly the same as that III. PHASE CONTINUITY OF CSS SIGNALS of M-ary FSK (M-FSK). In particular, BER of the coherent As explained in the previous section, data is encoded into detection can be expressed as [18] the phase function of a CSS symbol, which is a quadratic function of time. Since the phase determines the instantaneous M signal amplitude, phase discontinuity would lead to amplitude P , = b coh ( − ) 2 M 1 discontinuity. Since the phase of a particular CSS symbol is      M−1 1 ∞ 1 y −x2 also a cyclically shifted version of the basic chirp’s phase, 1 − √ √ exp dx there are inherent phase jumps occurring at symbols bound- 2π −∞ 2π −∞ 2 ⎧    ⎫ ⎤ aries. The amount of phase jump depends on the modulated ⎨ 2⎬ data, i.e., randomly. The discontinuities at symbol boundaries 1 2log MEb × exp − y − 2 dy⎦ (13) spread the signal power outside its BW. As an example, Fig. 3 ⎩ 2 N ⎭ 0 plots (in dashed lines) both phase function (top panel) and real part of the amplitude (bottom panel) of three CSS symbols where the SNR per bit, (Eb/N0), is related to the SNR as (note created by the cyclically shifting method with SF = 4. The that each orthogonal symbol has M samples while carrying SF first symbol is the basic chirp, i.e., m = 0, followed by two data bits) symbols corresponding to m = 2 and 4, respectively. Note that each CSS symbol starts and ends at the same phase. As Eb M can be seen, there are phase jumps at symbol boundaries and = SNR . (14) N0 SF discontinuities occur in the signal amplitude. NGUYEN et al.: EFFICIENT DESIGN OF CSS MODULATION FOR LPWANs 9507

Fig. 3. Illustration of the phase property of CSS signals. Fig. 4. Power spectra of CSS signals with an without phase correction.

We point out that the CSS symbols shown in Fig. 3 are up-sampled to show the smooth phase function over time. In its samples do not add to zero, the basic chirp contains a dc component. As a consequence, modulation by cyclic-shifting particular, the basic chirp is taken by sampling (2) at Fsamp = L × BW, which gives the up-converted basic chirp as aligns the dc component of all the transmitted symbols in    phase, causing spectrum spikes at Nyquist frequencies. On 2 ( ) n n x L [n] = exp j2π − , n = 0,...,ML − 1 the other hand, the phase correction not only produces smooth 0 2ML2 2L phase transition between symbols but also randomly rotates the (16) dc component of each symbol based on the data it modulates, thus effectively eliminating the spectrum spikes. where L is the up-sampling factor. The signals in Fig. 3 are plotted with L = 32. The symbols produced by the cyclic shift method can be expressed as IV. EFFICIENT CHIRP TRANSMITTER DESIGN ( ) ( ) x L [n] = x L [mod(n + mL, ML)] (17) While there are many papers investigating different aspects m 0 of CSS modulation [13], [14], [21], [22], research work on which has the initial phase of efficient implementation of a CSS system at the physical layer 2 is missing. This is quite surprising given the fact that the (L) ( ) (mL) mL φ [0] = ∠x L [0] = − implementation aspect of CSS is particularly important due 0 m 2ML2 2L to low-cost and low-power requirements in the majority of m2 m = − (×2π rads). (18) IoT applications. It appears that the common method to gen- 2M 2 erate a CSS signal is LUT-based (as shown in Fig. 1), which It is not hard to see that the noncontinuous phase can be is the method patented and commercialized by Semtech [15]. turned into continuous by forcing all CSS symbols to start at The LUT-based design is fairly simple and effective, present- the zero phase. In particular, the modulated CSS symbol with ing a challenge for Semtech’s competitors to come up with a zero-starting phase can be expressed as different design that is as simple and yet effective for LoRa’s    2 ( ) m m ( ) physical layer. xˆ L [n] = exp −j2π − x L [n]. (19) m 2M 2 m One of the disadvantages of the LUT-based design, as shown in Fig. 1, is scalability. In particular, in order to support The continuous-phase CSS signal is plotted in Fig. 3 in solid multiple SFs, the LUT size must be large enough to hold all lines. It is obvious that there are no more phase jumps nor possible basic chirps, each of which corresponds to one par- amplitude discontinuities at the symbol boundaries. ticular SF. Of particular interest is the LoRa gateway design, Fig. 4 compares the power spectrum density (PSD) of two which should be able to handle multiple nodes at various BW CSS signals, with and without phase correction. Aside from and SF simultaneously. The gateway typically employs DACs the phase offset correction, the two signals are generated with and ADCs running at a fixed rate of about 2 Msps.2 Therefore, = = the same parameters, i.e., SF 10, BW 125 kHz, and in order to support multiple BWs, the gateway’s transceiver = L 8, and using the same set of random data symbols. As would need to employ variable-rate up/down conversion mod- expected, the phase-corrected signal has much better out-of- ules or storing chirp samples at a higher rate, either of which band suppression. The difference is about 10 dB measured at would increase the cost of the transceiver considerably. For 500 kHz away from the carrier frequency. It is also noted that there are spectrum spikes occurred with 2The spec is taken from Semtech SX1257, which is the RF front-end the CSS signal without phase correction. This is because, since transceiver employed in most of the LoRa gateways [23]. 9508 IEEE INTERNET OF THINGS JOURNAL, VOL. 6, NO. 6, DECEMBER 2019

Fig. 5. OCG.

2 = n−1( + ) example, given the DAC sampling rate of 2 Msps, in order to Since n k=0 2k 1 , the phase function can be further support a minimum LoRa BW of 7.8125 kHz [24], the up- expressed as ⎧ conversion module must support an up-conversion rate of up − ⎨ n 1(2k+1+2mL−ML) k=0 , + < to L = 256. Assuming the dual DAC resolution is 8-bit, which ˆ(L) 2 if n mL ML φ [n] = − 2ML (22) m ⎩ n 1(2k+1+2mL−ML−2ML) means each complex sample consumes two bytes, the size of k=0 , otherwise. the LUT must be at least 2ML2 Of particular interest is the phase function when n ≥ ML−mL, 12 which can be written as 2 2SF × 256 = 4 161 536 bytes (20)  φˆ(L)  SF=6 m [n] n≥ML−mL − − ML mL 1(2k + 1 + 2mL − ML − 2ML) which is a little more than 4 MB. Note that the size would = k=0 grow exponentially, to 67 MB, if LoRa standard is extended 2ML2 n (2k + 1 + 2mL − ML − 2ML) to support up to SF = 16. + k=ML−mL This section proposes an efficient design of the chirp trans- 2ML2 − − mitter, called OCG. The OCG can support a wide range of SF ML mL 1(2k + 1 + 2mL − ML) = k=0 and BW at low cost. Moreover, the OCG uses just accumula- 2ML2 tors and adders to generate the continuous phase function of 2ML(ML − mL) φˆ(L) = (∠ˆ(L) )/ π − the chirp signal over time, i.e., m [n] [ xm [n] 2 ]. In 2ML2 particular, by substituting (16) and (17) into (19), the phase n (mod(2k + 1 + 2mL, 2ML) − ML) + k=ML−mL function can be expressed as 2ML2 n ( ( + + , ) − ) ( ) = − mod 2k 1 2mL 2ML ML φˆ L [n] = k ML mL m ⎧ 2ML2 2 ( + )2 ⎨− m − m + n mL − n+mL , n + mL < ML − (M − m). (23) = 2M 2 2ML2 2L 2 ( + − )2 + − ⎩− m − m + n mL ML − n mL ML , otherwise Since the integer term (M − m) can be ignored due to the  2M 2 2ML2 2L 2 phase wrap-around effect, combining (22) and (23) yields an n + nm − n , n + mL < ML = 2ML2 ML 2L equivalent phase function for a chirp signal as 2 (21) n + nm − n − n , otherwise. n 2ML2 ML 2L L = (mod(2k + 1 + 2mL, 2ML) − ML) φ(L) n = k 0 m [ ] 2 It should be noted that the term (n/2L) shifts the frequency 2ML n = 0, 1,...,ML − 1. (24) down by an amount of (BW/2) so that the CSS spectrum is centered at baseband. Moreover, the term (n/L) shifts the The above expression of the phase function allows the frequency down by an amount equivalent to BW, providing chirp signal to be generated efficiently in the proposed OCG the frequency “wrap around” effect of the chirp signal. design as shown in Fig. 5. The OCG comprises of four NGUYEN et al.: EFFICIENT DESIGN OF CSS MODULATION FOR LPWANs 9509 main components: the frequency accumulator, the frequency manipulator, the phase accumulator, and a vector rotator. The first three components create the phase function as in (24). In particular, the frequency accumulator 100 cre- ates a frequency that is linearly increasing over time, i.e., a ramp. The width of the accumulator has to be large enough to accommodate the combination of the largest SF with the lowest BW, i.e.,   F Fig. 6. SSB up-conversion of a chirp signal using a quadrature modulator. = + sys + freq_acc_width SFmax log2 1 (bits) (25) BWmin where Fsys,SFmax, and BWmin are, respectively, the system The actual center frequency of the chirp can be added clock rate, the highest SF, and the minimum BW supported. directly to the inverted frequency sample 108. The cen- There is one extra bit added to support the half-step frequency ter frequency word has the same width as the frequency correction 101. accumulator, as such, the center frequency resolution is The frequency manipulator introduces jumps into the Fsys/freq_acc_width (Hz). frequency ramp at the symbol boundaries according to input The phase accumulator 110 performs discrete integration of symbol symbol_in, selected BW and SF. It comprises of the frequency samples to provide phase samples for the gener- multiple logics, labeled from 101 to 108. ated chirp symbol. There is a selective inverter 109, located at Since phase is the integration of frequency over time, the input to the phase accumulator, to control the chirp direc- there is a half-step frequency correction 101 applied to the tion, i.e., up-chirp or down-chirp. Mathematically, the phase output of the frequency accumulator to correct for the dif- accumulator performs the main summation in (24). ference between discrete and continuous integration of the Finally, the vector rotator 111 performs phase rotation by linearly changing frequency over time. The output of the (L) an amount of φm [n] on an input vector as half-step frequency correction is the term 2k + 1 described in (24). (L) = πφ(L) ym [n] x[n] exp j2 m [n] (27) Symbol modulation is performed by adding a frequency off- set to the output of the frequency accumulator 102. The offset where x[n] and y[n, m] are the input and output vectors, is obtained by performing bit shifting 103 on the input symbol respectively. to make sure symbol_in has a proper format before adding To generate a continuous-phase chirp signal, the input vector it to the frequency accumulator’s output. The amount of left x[n] is set to a scalar, which is determined by the magnitude shifting depends on the system clock rate and the desired BW of the chirp signal. Alternatively, when the continuous phase of the generated chirp signal and is given as is not required between two adjacent chirp symbols, extra data   F information can be encoded as the phase of the input vector. = sys + symbol_in_left_shift log2 1 The vector rotator can be built effectively using a pipelined BW coordinate rotation digital computer (CORDIC). Since the = ( ) + log2 L 1 (bits) (26) CORDIC can be built without hardware multiplication [25], i.e., the bit-shifting operation generates the term 2mL described [26], the proposed OCG is highly cost-effective. in (24). SF of the chirp signal is controlled by performing bit-wise V. U SING OCG IN CSS TRANSCEIVERS DESIGN AND 105 of a mask 104 and the modulated frequency sample. This section explains in detail how OCG provides a great The mask, which is created based on SF and BW, controls how flexibility in CSS transceiver design. fast the frequency rolls over. In particular, the mask has the ( ) + same width as the frequency accumulator. The log2 ML 1 least significant bits of the mask are all ones, while the remain- A. Transmitter Design Using OCG and Quadrature ing bits are all zeros. The mask performs the modulo-by-2ML Modulator operation described in (24). For example, with the same BW Quadrature modulation is the most common method to up- and if SF is reduced by one, the mask will mask off one more convert a digital baseband signal to analog . The OCG bit at the top of the frequency sample, effectively making the can be combined with a quadrature modulator to upconvert the frequency rolls over twice as fast. The opposite can be said if chirp signal to passband as shown in Fig. 6. In particular, the BW is reduced by one. output of the OCG contains in-phase and quadrature compo- The frequency spectrum is centered at zero by inverting nents, which are converted to analog signals using a pair of the most significant bit of the masked frequency sample 106, DACs. The quadrature modulator contains a DCO, whose out- then change the number format from unsigned to signed. In puts are two sinusoids having 90◦ phase difference and at a particular, before inverting, the frequency rolls over at 0 and passband frequency. The two sinusoids are individually mixed BW, but after inverting the frequency rolls over at −(BW/2) with the in-phase and quadrature components before adding and (BW/2), respectively. Mathematically, spectrum centering together, giving a single sideband (SSB) spectrum centered at is represented as the term −ML in (24). the passband frequency. 9510 IEEE INTERNET OF THINGS JOURNAL, VOL. 6, NO. 6, DECEMBER 2019

Fig. 7. Double sideband up-conversion of a chirp signal using DDS and Fig. 8. Selecting a left or a right sideband for up-conversion. mixer.

Note that for the single-channel transceiver design, using the ROM design is fairly economical in the sense that oversam- pling factor of 2 or 4 is sufficient. The channel BW adjustment is, in fact, accomplished by adjusting the ADC/DAC clock rate. Although the complexity of the ROM and OCG designs is comparable in the digital domain, the OCG design is much more flexible when it comes to switching between differ- ent SF, BW, chirp direction, up-conversion factor, and center frequency. This flexibility leads to a cost-saving in the analog domain, as discussed in the next section.

B. Transmitter Design Using OCG and Direct Digital Synthesis Even though Semtech employs quadrature modulator in all of their LoRa-related products, the CSS transmitter design using quadrature modulator is neither power efficient nor cost Fig. 9. CSS receiver using the OCG to de-chirp the chirped signal. effective since it must employ two DACs and two frequency mixers. the linear phase term as To reduce the cost of the up-converter, the direct digi- tal synthesis (DDS) technique can be used as illustrated in (L) = (L) − πφ(L) v [n] ym [n] exp j2 0 [n] Figs. 7 and 8. In particular, the OCG is configured with a  nm nonzero center frequency so that the signal spectrum is pushed x[n] expj2π , if n + mL < ML = ML (28) close to the Nyquist rate. Then only the in-phase component π n(m−M) , . x[n] exp j2 ML otherwise at the output of the OCG is sent to a DAC. The single DAC generates a double sidebands (i.e., real) signal and mixes it This means that the signal after being de-chirped and down- with a carrier controlled by a DCO. There is a bandpass filter converted is a pure sinusoid whose frequency depends on the modulated symbol. That is that comes after to remove one sideband to prevent undesirable   interference. km v[k] = x[k] exp j2π , k = 0, 1,...,M. (29) The control unit decides to keep the left or the right sideband M based on the center frequency of the chirp signal, as shown in Fig. 8. For example, given the band of operation from 902 to An illustration of de-chirping OCG is shown in Fig. 9. In par- 928 MHz, the right sideband is kept if the carrier frequency ticular, following (28), the configurations for the de-chirping is less than 915 MHz, and the left sideband is kept otherwise. OCG are given below. In doing so, the required frequency separation between the 1) Signal to be de-chirped is sent to input x[n] of the de- two sidebands is minimized, which means the DAC can run chirping OCG. 2) The input symbol to the de-chirping OCG is set to 0. at a lower rate, i.e., Fs = 16 MHz is sufficient. In case the left sideband is selected, it would need to invert the signal 3) The chirp direction is set to be the opposite of that in spectrum, which can be done simply by inverting the OCG’s the transmitter’s OCG. center_freq chirp_dir signal. 4) The is set to be the negative of that in the transmitter’s OCG. The signal after de-chirping is converted to the frequency C. CSS Receiver Design Using OCG domain using M-point DFT. The output of the DFT has most of The OCG can also be used at the receiver to de-chirp a CSS its energy concentrated in a single bin, whose location yields signal. In particular, the signal to be de-chirped is connected the demodulated symbol. Note that the pair of up/down con- to the OCG’s input x as shown in Fig. 9. De-chirping unwinds verters can be either SSB (i.e., using dual ADCs) or double the second-order phase function applied at the receiver, leaving sideband (i.e., using a single ADC). In case the receiver uses a NGUYEN et al.: EFFICIENT DESIGN OF CSS MODULATION FOR LPWANs 9511

SRRC matched filters so that no intersymbol interference (ISI) occurs, hence maintaining the orthogonality among chirps. The power spectra of pulse-shaped CSS signals are shown in Fig. 12. The pulse shaping filter is an SRRC filter having 12% exceed BW and out-of-band rejection performance of about 60 dB. Obviously, the inclusion of the pulse shaping filter allows for more efficient use of the spectrum. It should be noted that the pulse shaping design proposed in this section increases power consumption. Since the method provides a tradeoff between performance and complexity, depending on the type of application, care must be taken when it comes to enable or disable the pulse shaping filter. For exam- ple, class A (battery powered) LoRa devices should not use the pulse shaping filter to conserve power, while class C (main powered) devices should certainly use the pulse shaping filter Fig. 10. Power spectra of conventional CSS signals with various settings of to enhance spectral efficiency. the SF.

B. Phase-Shifted CSS Modulation single ADC to sample the double-sideband signal, de-chirping The phase information of a chirp signal can be exploited to is performed on an over-sampled signal, and thus its out- encode extra data information. As explained in Section III, the put needs to be down-sampled to baseband before performing phase continuity property, i.e., forcing all CSS symbols to start DFT. from the zero phase, is helpful to prevent spectral leakage from CSS signaling. However, with the help from pulse shaping filters, spectral leakage is significantly mitigated, which allows VI. ENHANCING CSS CSS symbols to start from arbitrary phase values. In fact, the CSS is known for not being optimized in terms of spec- starting phase of a CSS symbol can be chosen from a phase- tral efficiency [12], [13], [22]. Although LoRa was developed shift keying (PSK) constellation, allowing the system to send to trade spectral efficiency for reception sensitivity, there are more information bits per chirp symbol. improvements that can still be made to improve the spec- The proposed phase-shifted CSS (PS-CSS) system is shown tral efficiency of conventional CSS signaling. The following in Fig. 14, which has minor modifications over the con- sections present and discuss two techniques to enhance CSS ventional system in Fig. 9. For the sake of simplicity, only spectral efficiency. the base-band equivalent model is shown, and the up/down conversion blocks are omitted. While the input x[n] in the con- ventional CSS system is only used to control the amplitude of A. Pulse-Shaped CSS Modulation the chirp, the input can also be used to alter the starting phase The conventional LoRa signal is not spectrally efficient of the chirp signal, i.e., by setting x[n] = exp{j2π(p/Q)}, because it has large roll-off regions on both sides of the spec- where Q is the modulation order of the PSK constellation and trum. According to [11], a 500-kHz BW LoRa signal can 0 ≤ p < Q is the PSK modulated symbol. Note that while it is actually occupy more than 700 kHz of RF BW. This natu- possible to apply both phase and amplitude to the input x[n], rally limits the number of channels that can be used in a given i.e., making x[n] QAM modulated, doing so would make con- spectrum. siderable power variations among CSS symbols, potentially Fig. 10 shows the power spectra of the conventional CSS comprising the reception sensitivity. signals. It is noted that the out-of-band portion of the signal Demodulating PS-CSS signals requires a coherent receiver is higher for a lower SF. In particular, increasing SF by one that is able to synchronize its phase to the transmitter. Once would decrease the out-of-band interference by about 3 dB. the phase is synchronized, the PSK modulated bits can be The power spectra in Fig. 10 appear to be similar to those recovered by looking at the phase of the DFT bin having the obtained with the Semtech design described in [11]. highest magnitude, as shown in Fig. 14. Due to space limita- To reduce interference, a pair of pulse shaping filters can be tion, the phase synchronization method is not discussed in this added to lower the out-of-band portion of the CSS spectrum. paper. A novel receiver design, including timing, frequency, In particular, the OCG is configured to generate CSS signals and phase synchronization shall be presented in another paper at baseband, i.e., setting center_freq = 0. Then a pair of that follows this paper. pulse shaping filters are employed to up-convert the signal to Although the proposed technique can be used with arbitrary the DAC rate as illustrated in Fig. 11. The filter pair is selected PSK modulation, it is determined that using QPSK to modu- to be square-root raised-cosine (SRRC) filters. As a conse- late the starting phase of each PS-CSS symbol can enhance the quence, the CSS spectral roll-off can be controlled by adjusting throughput without compromising the LoRa link budget. This the roll-off factor of the pulse shaping filters as demonstrated is explained and demonstrated in the following. It is clear that in Fig. 12. Note that the CSS receiver also employs a pair of the performance of PSK demodulation depends on the power 9512 IEEE INTERNET OF THINGS JOURNAL, VOL. 6, NO. 6, DECEMBER 2019

Fig. 11. CSS modulation with SRRC pulse shaping filters.

TABLE I SEMTECH VERSUS PROPOSED DEMODULATION SNR AT PER = 1%

throughput without compromising the LoRa link budget. On the other hand, using 8-PSK or higher-order PSK can further enhance the throughput, but at the expense of reduced LoRa Fig. 12. Power spectra of pulse-shaped CSS signals with BW = 500 kHz link budget. and various settings of the SF.

VII. SIMULATION RESULTS Performance of a conventional CSS system with noncoher- ent receiver is first illustrated in Fig. 15. The figure plots BER versus SNR achieved with the transceiver design in Fig. 9. The channel is assumed to be AWGN, which is typical for a LoRa system given the narrow-band nature of the sig- nal [27]. Simulations are repeated with SF = 6Ð12. The BW of the signal is fixed at 125 kHz. Each transmission packet has 64 bytes of random data and there are 10 000 packets simulated for each point in the BER curves. Performance with- out channel coding (uncoded performance) is shown in solid lines, whereas performance with the use of channel coding (coded performance) is shown in dashed lines. To obtain coded performance, the 64 bytes of information are first interleaved randomly to prevent burst noise. Then the interleaved sequence Fig. 13. BER of CSS receiver in terms of PSNR. is encoded using the (7, 4) Hamming code, which gives the effective code rate of 4/7 ≈ 0.571. It should be noted that there are multiple code rates in LoRa, all of which are created of noise contained in the DFT bin having the highest magni- from the (7, 4) Hamming mother code [21]. The Hamming tude, relative to the bin power itself. Such a ratio is actually (7, 4) code is chosen in this paper because code rates 4/5 and the PSNR defined in (10). Fig. 13 plots the BER performance 4/6 provide no error correction capability, while code rates of of a conventional CSS system versus PSNR. Also plotted on 4/7 and 4/8 provide the same error correction capability [21]. the same figure are the theoretical BER curves versus SNR of It can be seen from the figure that such a simple code provides QPSK and 8-PSK, respectively. It can be seen that the BER a coding gain of about 1 dB. curves of CSS signaling, at various SF values, are between The packet-error-rate (PER) performance is provided in that of QPSK and 8-PSK. Therefore, by using QPSK for phase Fig. 16, which can be directly compared to that of Semtech modulation, the error rate in the PSK channel is lower than reported in [24]. A packet error is defined when a packet that of the CSS channel. As a result, PS-CSS can enhance the has at least 1 bit in error after decoding. Semtech defines NGUYEN et al.: EFFICIENT DESIGN OF CSS MODULATION FOR LPWANs 9513

Fig. 14. Encoding extra data bits into the phase of a chirp signal.

Fig. 15. Uncoded and coded BER performance of noncoherent CSS receiver.

Fig. 16. PER performance of the noncoherent receiver.

Fig. 17. Performance comparison between (a) conventional noncoherent CSS LoRa demodulation SNR as the SNR at which the PER is less modulation and (b) PS-CSS with QPSK for . than or equal to 1%. Using such a criterion produces Table I, which compares the performance of the proposed transceiver design versus Semtech’s design. Overall, the proposed receiver The performance of the proposed PS-CSS system is shown design is significantly better than that of Semtech. In partic- in Fig. 17, where BER-versus-Eb/N0 curves of the proposed ular, the SNR improvement ranges from 0.9 dB at SF = 6 system are compared against that of the conventional CSS. to an impressive amount of 2.5 dB at SF = 12. In general, The starting phase of each PS-CSS symbol is QPSK mod- a 2.5 dB improvement means that the transmit power can be ulated, i.e., for each transmitted CSS symbol, in addition to reduced by almost half while maintaining the same transmis- SF primary data bits, extra 2 bits can be sent over the sec- sion performance. Alternatively, by keeping the same transmit ondary PSK channel. The proposed implementation effectively power, the proposed design allows to extend the transmission increases throughput of the conventional CSS system by 33%, range by about 33%. 25%, 20%, and 17% for SF = 6, 8, 10, and 12, respectively. 9514 IEEE INTERNET OF THINGS JOURNAL, VOL. 6, NO. 6, DECEMBER 2019

The increase in throughput is not compromised by the link [7] LoRa Alliance. (2017). LoRaWAN Specification V1.1. [Online]. performance as can be verified in Fig. 17. For the same SF, Available: https://lora-alliance.org/resource-hub/lorawantm- specification-v11 since the PS-CSS system has a higher transmission rate than [8] L. Feltrin, C. Buratti, E. Vinciarelli, R. De Bonis, and R. Verdone, the CSS system while enjoying the same BER, there are gains “LoRaWAN: Evaluation of link- and system-level performance,” IEEE Internet Things J., vol. 5, no. 3, pp. 2249Ð2258, Jun. 2018. in SNR per bit (Eb/N0) of 1.24, 0.97, 0.79, and 0.68 dB for = [9] AN1200.22: LoRa Modulation Basics, SemTech, Camarillo, CA, USA, SF 6, 8, 10, and 12, respectively. May 2015. Finally, we point out that the rate improvement achieved by [10] C. A. Hornbuckle, “Fractional-N synthesized chirp generator,” U.S. our proposed PS-CSS modulation with QPSK is twice higher Patent US7 791 415 B2, Sep. 2010. [11] AN1200.26: LoRa and FCC Part 15.247: Measurement Guidance, than what achieved by the method of interleaved chirp spread- SemTech, Camarillo, CA, USA, May 2015. ing LoRa (ICS-LoRa) recently proposed in [28]. In ICS-LoRa, [12] B. Reynders and S. Pollin, “Chirp spread spectrum as a modulation tech- interleaved versions of the nominal LoRa chirp signals are nique for long range communication,” in Proc. IEEE Symp. Commun. Veh. Technol., Nov. 2016, pp. 1Ð5. exploited to send just one extra bit within each transmitted [13] X. Ouyang, O. A. Dobre, Y. L. Guan, and J. Zhao, “Chirp spread ICS-LoRa symbol. spectrum toward the Nyquist signaling rate—Orthogonality condi- tion and applications,” IEEE Signal Process. Lett., vol. 24, no. 10, pp. 1488Ð1492, Oct. 2017. [14] X. Ouyang and J. Zhao, “Orthogonal chirp division multiplexing,” IEEE VIII. CONCLUSION Trans. Commun., vol. 64, no. 9, pp. 3946Ð3957, Sep. 2016. This paper presented detailed mathematical descriptions [15] F. Sforza, “Communications system,” U.S. Patent US8 406 275 B2, Mar. 2013. of CSS modulation, both coherent and noncoherent demod- [16] O. B. A. Seller and N. Sornin, “Low power long range transmitter,” ulation, in discrete-time (digital) domain. Theoretical BER European Patent EP2 763 321 A1, Aug. 2014. expressions are given for both types of demodulation and [17] O. B. A. Seller and N. Sornin, “Low power long range transmitter,” U.S. Patent US9 252 834 B2, Feb. 2016. shown to be identical to simulation results. The paper then [18] H. Nguyen and E. Shwedyk, A First Course in Digital Communications, proposed a novel and efficient LoRa transmitter design, namely 1st ed. Cambridge, U.K.: Cambridge Univ. Press, 2009. OCG, that allows supporting multiple SF and BW settings at [19] M. K. Simon and M.-S. Alouini, Digital Communications Over Fading Channels, 1st ed. Hoboken, NJ, USA: Wiley, 2005. very low cost. The OCG has excellent flexibility such that not [20] L. Vangelista, “Frequency shift chirp modulation: The LoRa modu- only it can be used to generate either complex base-band or lation,” IEEE Signal Process. Lett., vol. 24, no. 12, pp. 1818Ð1821, real pass-band signals at the transmitter, but it can also be Dec. 2017. [21] H. Mroue, A. Nasser, B. Parrein, S. Hamrioui, E. Mona-Cruz, and employed at the receiver to demodulate the chirp signal. G. Rouyer, “Analytical and simulation study for LoRa modulation,” in The last part of this paper demonstrated that the spectral Proc. Int. Conf. TeleCommun., Jun. 2018, pp. 655Ð659. efficiency of LoRa signals can be further enhanced. By using [22] D. Croce, M. Gucciardo, S. Mangione, G. Santaromita, and I. Tinnirello, “Impact of LoRa imperfect orthogonality: Analysis of link-level a pulse shaping filter, the spectral roll-off of CSS signals performance,” IEEE Commun. Lett., vol. 22, no. 4, pp. 796Ð799, can be controlled and made very small, while the orthog- Apr. 2018. onality among chirps is still maintained. It was also shown [23] SX1257 RF Front-End Transceiver, SemTech, Camarillo, CA, USA, 2012. that additional data information can be elegantly encoded into [24] SX1276/77/78/79—137 MHz to 1020 MHz Low Power Long Range the starting phase of each CSS symbol, which can then be Transceiver, SemTech, Camarillo, CA, USA, 2015. recovered using a coherent receiver. Such elegant and sim- [25] C.-H. Lin and A.-Y. Wu, “Mixed-scaling-rotation CORDIC (MSR- CORDIC) algorithm and architecture for high-performance vector rota- ple enhancement of the conventional CSS technique allows tional DSP applications,” IEEE Trans. Circuits Syst. I, Reg. Papers, to send extra 2 bits with each conventional CSS symbol, vol. 52, no. 11, pp. 2385Ð2396, Nov. 2005. leading to throughput improvement of the conventional CSS [26] M. Garrido, P. Källström, M. Kumm, and O. Gustafsson, “CORDIC II: = A new improved CORDIC algorithm,” IEEE Trans. Circuits Syst. II, system by 33%, 25%, 20%, and 17% for SF 6, 8, 10, Exp. Briefs, vol. 63, no. 2, pp. 186Ð190, Feb. 2016. and 12, respectively. The complete and efficient design of [27] T. Elshabrawy and J. Robert, “Closed-form approximation of LoRa the coherent receiver, including timing, frequency, and phase modulation BER performance,” IEEE Commun. Lett., vol. 22, no. 9, pp. 1778Ð1781, Sep. 2018. synchronization will be reported in another paper. [28] T. Elshabrawy and J. Robert, “Interleaved chirp spreading LoRa-based modulation,” IEEE Internet Things J., vol. 6, no. 2, p. 1, Jan. 2019.

REFERENCES [1] M. Centenaro, L. Vangelista, A. Zanella, and M. Zorzi, “Long-range communications in unlicensed bands: The rising stars in the IoT and smart city scenarios,” IEEE Wireless Commun., vol. 23, no. 5, pp. 60Ð67, Tung T. Nguyen received the B.Eng. degree Oct. 2016. from the Hanoi University of Technology, Hanoi, [2] A. Augustin, J. Yi, T. Clausen, and W. M. Townsley, “A study of LoRa: Vietnam, in 2008, and the M.Sc. and Ph.D. degrees Long range & low power networks for the Internet of Things,” Sensors, from the University of Saskatchewan, Saskatoon, vol. 16, no. 9, pp. 1Ð18, 2016. SK, Canada, in 2013 and 2016, respectively. [3] Sigfox. Accessed: Apr. 11, 2019. [Online]. Available: He joined Vecima Networks, Inc., Saskatoon, https://www.sigfox.com/ in 2016, where he developed algorithms for the [4] Ingenu. RPMA Technology for the Internet of Things. Accessed: next-generation cable systems. Since 2017, he has Apr. 11, 2019. [Online]. Available: http://www.meterlinq.com/wp- been a Research Associate with the Department content/uploads/2016/07/RPMA-Technology-Ingenu.pdf of Electrical and Computer Engineering, University [5] M. Weyn et al., “DASH7 alliance protocol 1.0: Low-power, mid- of Saskatchewan. He joined Ciena Corporation, range sensor and actuator communication,” in Proc. IEEE Conf. Stand. Ottawa, ON, Canada, in 2019, where he works on the development of the next- Commun. Netw. (CSCN), 2015, pp. 54Ð59. generation coherent optical technology. His current research interests include [6] “White paper: A technical overview of LoRa and LoRaWAN,” San digital signal processing, filter design, synchronization techniques, and error Ramon, CA, USA, LoRa Alliance, White Paper, 2015. control coding. NGUYEN et al.: EFFICIENT DESIGN OF CSS MODULATION FOR LPWANs 9515

Ha H. Nguyen (M’01–SM’05) received the B.Eng. Patrick Grossetete received the computer degree degree from the Hanoi University of Technology, from Control Data Research Institute, Paris, France, Hanoi, Vietnam, in 1995, the M.Eng. degree in 1979. from the Asian Institute of Technology, Bangkok, He is a Distinguished Engineer of Technical Thailand, in 1997, and the Ph.D. degree from the Marketing with the Cisco Industrial IoT Networking University of Manitoba, Winnipeg, MB, Canada, in Group, Cisco, Toronto, ON, Canada, focusing on 2001, all in electrical engineering. Internet of Things (IoT) networking technologies, He joined the Department of Electrical and architectures, and designs, such as LoRaWAN, Computer Engineering, University of Saskatchewan, Wi-SUN (IEEE 802.15.4g/e RF, IEEE 1901.2a PLC, Saskatoon, SK, Canada, in 2001, and became a Full IPv6, IP routing, 6LoWPAN, and RPL), and IoT Professor in 2007, where he currently holds the gateways (private LTE/5G). He was the Director of position of NSERC/Cisco Industrial Research Chair of Low-Power Wireless Product Management and Customer Solutions with Arch Rock, focusing on Access for Sensor Networks. He has coauthored, with Ed Shwedyk, the IPv6-based wireless sensor network technology for smart grid, energy, and textbook entitled A First Course in Digital Communications (Cambridge environmental optimization applications. He led a product management team University Press). His current research interests include communication at Cisco, responsible for a suite of Cisco IOS software technologies, includ- theory, wireless communications, and statistical signal processing. ing IPv6 and IP mobility. Before to joining Cisco as a Consulting Engineer, Dr. Nguyen was an Associate Editor of the IEEE TRANSACTIONS in 1994, he was with Digital Equipment Corporation, Maynard, MA, USA, ON WIRELESS COMMUNICATIONS and the IEEE WIRELESS as a Consulting Engineer and was involved with network design and deploy- COMMUNICATIONS LETTERS from 2007 to 2011 and 2011 to 2016, ment. He has coauthored the books entitled IoT Fundamentals: Networking respectively. He currently serves as an Associate Editor for the IEEE Technologies, Protocols, and Use Cases for the Internet of Things, Global TRANSACTIONS ON VEHICULAR TECHNOLOGY. He was the Co-Chair of IPv6 Strategies,andDeploying IPv6 Networks published by Cisco Press as the Multiple Antenna Systems and Space-Time Processing Track, IEEE well as several white papers, such as Unified Field Area Network architecture Vehicular Technology Conferences (Fall 2010, Ottawa, ON, Canada and Fall for Distribution Automation in 2014 and IPv6 Architecture for Field Area 2012, Quebec, QC, Canada), the Lead Co-Chair of the Wireless Access Networks in 2012. Track, IEEE Vehicular Technology Conferences (Fall 2014, Vancouver, Mr. Grossetete was a recipient of the IPv6 Forum Internet Pioneer Award at BC, Canada), the Lead Co-Chair and the Co-Chair of the Multiple the San Diego summit in June 2003. He is a Regular Speaker at conferences Antenna Systems and Cooperative Communications Track, IEEE Vehicular and industry events, including the IPv6 Forum, which he joined in 1999 as Technology Conference (Fall 2016, Montreal, QC, Canada and Spring 2018, a Cisco representative. He also acts as a Reviewer of European Commission Porto, Portugal), and the Technical Program Co-Chair of Canadian Workshop sponsored projects, such as GEANT and ENVIROFI. He is an IPv6 Forum on Information Theory (2015, St. John’s, NL, Canada). He is a fellow of the Fellow. Engineering Institute of Canada, and a registered Member of the Association of Professional Engineers and Geoscientists of Saskatchewan.

Robert Barton received the B.ASc. degree in engineering physics from the University of British Columbia, Vancouver, BC, Canada, in 1997. He is a Cisco Distinguished Systems Engineer with over 20 years of experience in the field of computer networking. Since joining Cisco, Toronto, ON, Canada, in 2000, he has led some of Cisco’s largest and most innovative network deployments globally. He currently leads Cisco’s Canadian aca- demic research program with a focus on Wi-Fi6, 5G, and LPWA technologies. He is a member of the Cisco Live Distinguished Speaker Hall of Fame. He has also published extensively, with titles on 802.11 wireless technology, Internet of Things, and network quality of service methods. He also holds patents in diverse technologies, including machine learning, cloud computing, network security systems, 802.11ax, and distributed analytics. His current research interests include large-scale IPv6 network design, AI-based network security systems, and digitization of industrial systems. Mr. Barton is a registered Professional Engineer.