10/07/2016
Microwave Passive and Active Devices with Integrated Filtering Functions
WM05 Yi Wang1, Michael J. Lancaster2
1University of Greenwich (Medway campus), Chatham Maritime, Kent, U.K. 2University of Birmingham, Edgbaston, Birmingham, U.K. [email protected], [email protected] Slide 1 of 175
Programme
9:00 - 9:10 Welcome 9:10 - 9:40 Co-design of High-Q Tunable Filters with Active Devices D. Peroulis, Purdue University, West Lafayette, IN, USA 9:40 - 10:10 Integrated filtering power dividers, antennas and arrays Y. Wang, University of Greenwich, UK; S. Gao, University of Kent, UK 10:10 - 10:40 Compact Power Distributing Devices and Power Amplifiers with Integrated Filtering Response X. Zhang, South China University of Technology, Guangzhou, China 10:40 - 11:20 Coffee Break 11:20 - 11:50 Waveguide components based on all coupled resonators M. J. Lancaster, University of Birmingham, UK 11:50 - 12:20 Single/Multi-Band Power-Distribution and Impedance-Transformation Planar Circuits with Added Static and Reconfigurable Bandpass Filtering Functionality R. Gomez-Garcia, University of Alcala, Madrid, Spain; D. Psychogiou and D. Peroulis, Purdue University, USA 12:20 - 12:50 Synthesis techniques for multiplexers and multiport selective networks G. Macchiarella, Politecnico di Milano, Italy 12:50 - 13:00 Open discussion and concluding remarks Slide 2 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Summary of the workshop topics
“Passive & active devices with integrated filtering functions”
Filter-Bulter Filter-Couplers matrix
Filter-Power dividers Filter-Balun ‘Multi-role’, Co-design & ‘Multi- Filter-Antennas/arrays Synthesis functional’
Amplifier-filters
Multi-port filtering Resonant junctions networks • Integration and miniaturisation • Performance enhancement Potential – Elimination of 50 ohm interfaces benefits – Bandwidth/selectivity control … • New device configuration and topology Slide 3 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Co-design of High-Q Tunable Filters with Active Devices
Dimitrios Peroulis
Purdue University
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Co-Design of PAs and Tunable Filters
Why Co-Design of PA and Filter?
Advantages of Co-Design
Minimized size & volume & cost Minimized loss Enhanced overall performance
Challenges Matching Filter 1. 2D Planar PA vs. 3D Cavity Filter 2. Realization of Matching Filter
Slide 5 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Tunable Resonator Technology
APPROACH: EVANESCENT-MODE CAVITY RESONATORS
• High-Q (>500-1,000) • Widely Tunable (>2:1) • Highly-Linear (> 60 dBm) • Scalable from sub-GHz to over 100 GHz • Mobile form factor
Slide 6 of 175
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Tunable Resonator Technology
10 mm 10 mm
10 mm 10 mm
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Tunable Resonator Technology
• Capacitive loading significantly reduces area while still maintaining high Q • Large tuning range can be achieved with small changes in gap
Resonant frequency 6.5 1300 Quality factor (unloaded) gap
5.5
1100 Q 4.5
900 Frequency(GHz) 3.5
2.5 700 5 10 15 20 25 30 Gap (micrometer) Slide 8 of 175
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Tunable Filter Technology - PCB
Copper foil Parylene-N Piezo disc Capacitive post
Prepreg layer Resonator TMM-3 Silver epoxy element
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Tunable Filter Technology - PCB
State of the art evanescent-mode cavity filters: • Wide tuning: 0.65 to 6 GHz demonstrated • Bandwidth < 30 MHz • Insertion loss < 3.5 dB for 0.5% bandwidth • High Q: >1300 at 6 GHz measured Tunable 2nd order filter
Single Resonator, .5 cm3 volume
Q of 1330 measured, tuned over an octave
Current state of the art continuously tunable bandpass filters covering 0.65-6 GHz in 4 bands Slide 10 Measured Qu of 175
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All-Silicon RF MEMS Tunable Filters
First demonstration: • 6.1-24.4 GHz (4:1) continuous tuning measured Slide 11 • Qmeas = 300-1,000 of 175
Additional Filter Technology Info
Review paper with additional information and references Slide 12 of 175
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Co-Design of PAs and Tunable Filters: 1st order Integration of 2D and 3D Circuits
● Substrate-integrated microwave cavity resonator -- Boundary, Coupling, Post, Field ● Tunability realized using external Piezoelectric actuator
E
H
Cross-Section Illustration Slide 13 K. Chen, X. Liu, and D. Peroulis, “Widely-tunable High-efficiency Power Amplifier with Ultra-narrow Instantaneous Bandwidth,” IEEE Trans. Microw. Theory Tech., Dec. 2012 of NNN
Co-Design of PAs and Tunable Filters: 1st order Design of Tunable Resonator as OMN of PA
Z ZRES in Tolerable Region ZL @ 2-3GHz of GaN with >70% Efficiency
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Co-Design of PAs and Tunable Filters: 1st order Fabricated PA-Resonator
Continuous Large Tuning -Signal
PA Performance vs Freq.
Achievements: First co-designed tunable PA-filter module with simultaneous high efficiency,Slide 15 narrowband filtering, and wide tunability of 175
Co-Design of PAs and Tunable Filters: 2nd order Matching Filter Design Theory Regular Filter Matching Filter Coupling Matrix of 2-Pole Filter
Zin= Zin=Z0 Z0(0.5+0.5j)
Zin
Substrate- Integrated Z Same Freq. Cavity 0 Response Arbitrary Input Impedance: 2 2 M12 jM 11M 2L in xZ jy 2 2 1MM 2LS Slide 16 K. Chen, J. Lee, W.J. Chappell, and D. Peroulis, “Co-Design of Highly Efficient Power Amplifier and High-Q Output Bandpass filter,” IEEE Trans. Microw. Theory Tech., Nov. 2013. of 175
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Co-Design of PAs and Tunable Filters: 2nd order Filter Implementation and Full-Wave Simulation
Chebyshev Filter Matching Filter .10037 00 001.8930 Load Impedance for GaN -- Full-Wave Simulation using .1037 .10287 0 .101.893287 0 .10287 .10037 .10287 .10037 HFSS .100037 0 .100037 0 1.25% Bandwidth @ 3.1 GHz 15-dB Equal-Ripple Return Loss Entire Output Filter Fundamental
[email protected] 2nd Harmonic
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Co-Design of PAs and Tunable Filters: 2nd order Fabricated PA-Filter
Small-Signal
Large-Signal
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Co-Design of PAs and Tunable Filters: 2nd order Comparison with Conventional Technology
Size & volume reduction by about half Implemented Conventional PA + Filter Co-Designed PA + Filter
Higher overall efficiency and Reduced loss
≈10% Improvement
These results validate the advantages of co-design technique mentioned before.
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Co-Design of PAs and Tunable Filters: 3rd order Fabricated 3rd order prototype
Fabricated Circuit Filter Impedance
@ 3.1GHz
Small-Signal Large-Signal
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Highly Efficient & Highly Linear Dual Carrier PA Dual Carrier Signal Challenge
Nonlinearity Issues
Efficiency Back-off Vs. Power Power Peak Power Average Probability Average Vs. Power Efficiency Degradation
Envelope Slide 21 of 175
Highly Efficient & Highly Linear Dual Carrier PA Dual Carrier Signal Challenge Current Solution: External Linearization Efficiency Enhancement Feed-Back Doherty PA Feed-Forward + ET & EER Pre-Distortion (DPD) Dynamic Load Modulation
Developed Concept Efficient and linear amplification of dual carrier signals. No linearization/efficiency-enhancement needed. Simple realizability, reduced system complexity. Slide 22 of 175
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Highly Efficient & Highly Linear Dual Carrier PA Key Enabler & Challenge: Carrier Combiner/Diplexer
Each PA amplifies a constant-envelop signal Theoretically zero efficiency degradation Theoretically zero intermodulation Slide 23 K. Chen, E.J. Naglich, Y.-C. Wu, and D. Peroulis, “Highly Linear and Highly Efficient Dual-Carrier Power Amplifier Based on Low-Loss RF Carrier Combiner,” IEEE Trans. Microw. Theory Tech., Mar. 2014. of 175
Highly Efficient & Highly Linear Dual Carrier PA Conventional Diplexer Solution
Diplexer Filter 1 Filter 2
BPF1 2 Filter 1 Transmissionf1 f2
BPF2 3
Δf ≤ 10 MHz When f is very small: Very steep skirt needed
Very high filter order & Narrow BW
Increased complexity & High-Q
N≥9, Q≥30,000 required for 40-dB attenuation at 2+f GHz Slide 24 -- Calculation from ADS filter model of 175
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Highly Efficient & Highly Linear Dual Carrier PA Diplexers with Bandstop Filters 40-dB Isolation @ f=10MHz 0.1 dB Insertion Loss Bandpass: N=9, Q=30000
Hardly Practical!
Lowpass & Highpass: N>15, Q>30000
Hardly Practical!
Bandstop: N=2, Q=3000
Practical! Slide 25 of 175
Highly Efficient & Highly Linear Dual Carrier PA PA Integrated with Diplexer
Integrated Module and Dual-Carrier Testing Setup
Building blocks are connected through adaptors Independent characterization of each components Slide 26 of 175
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Highly Efficient & Highly Linear Dual Carrier PA Dual Carrier Measurement – CW
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Highly Efficient & Highly Linear Dual Carrier PA Dual Carrier Measurement – GSM
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Highly Efficient & Highly Linear Dual Carrier PA
Co-design brings significant benefits in tunable communication systems. A PA with a substrate integrated waveguide is a convenient and high-performance solution. Wide tuning range with simultaneous high quality factor are key benefits of evanescent-mode waveguides. These solutions can become available for multi-carrier systems as well. Future directions include co-designs with PA-filter-antenna chains.
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Integrated filtering power dividers, antennas and arrays
Yi Wang1, Steven Gao2
1University of Greenwich (Medway campus), Kent, UK 2University of Kent, UK
[email protected], [email protected]
Slide 30 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Outline
1. Introduction & background 2. Resonators & patch antenna/array integration 2.1 Integrated v.s. cascaded 2.2 All-resonator filtering array (with a 4-way filtering power divider) 2.3 Dual-band filter-antenna 2.4 Duplexer-antenna 3. Some other integration schemes 4. Conclusions
Slide 31 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Antenna-filter integration & co-design
LNA PA
LNA • Element • Array PA Duplexer • MIMO
Slide 32 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Integrated filtering antenna
Also known as filter-antenna or filtenna. Mixed usage and various perception of the terminology.
• Some refer to the proximity of a filter and an antenna e.g. co-site, vertical integration • Some refer to an antenna fed by a filter antenna as a complex load to a filter (not a new concept, but recently some innovative integrated structures demonstrated) • Filter embedded in a radiating element e.g. notch filter for rejection • Filter and antenna with separated ports but shared structure 3-port device • What we meant here is a ‘seamless’ integration…
Slide 33 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Seamless integration
34 Patch + LNA Resonator || nd Antenna 2 order
P1 1 2 3 4 5 5 poles = 4 resonators + 1 antenna
P1 1 2 3 4 5 P2 5 poles = 5 resonators
Dual-functional radiator as • An antenna Lin and Chung, EuMC 2009. • A pole of a filter * Antenna contributes to the selectivity of the filter. * Filter enhances the selectivity of the antenna. Slide 34 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Motivation
. Eliminate 50 ohm interfaces . Reduced mismatch and multi-reflections . Front-end miniaturisation Integration . Reduced component count . Filtering in/by antenna arrays . Enhancing selectivity of antenna . i.e. when interference or spectrum efficiency are concerned Functionality . Increasing bandwidth of antenna . Through electromagnetic coupling . Co-design . Apply filter design technique to antennas Design
Slide 35 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
2. Integration approach
Antenna Input Example: SLR-fed Filtering-Antenna
Filter MatchingRadiator network (a) Integrated filter-antenna Input
Resonator Resonator Radiator (b) Integrated filter-antenna Input
Multi-modeRadiator Resonator (c)
SLR: Stub loaded resonator • An dual-mode resonator • Even/odd mode analysis
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3rd order filtering antenna
The patch and the SLR are Characteristics: coupled via a slot in their 3rd-order filter = 2 from SLR + 1 from patch Compact size and simple structure common ground like Wider bandwidth ‘coupled resonators’. Improved frequency selectivity and out-of- band rejection
Slide 37 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
SLR-fed dual-pol antenna array
Patch + Dual-mode SLR || 3rd order (per pol.)
@5.2GHz, 10%
Slide 38 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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2.1 Cascaded vs. Integrated
4 element patch array Cascaded Band edge degradation Narrower band SLR filter
Input port
50 Ω interface
17.8mm Integrated Port 2 Compact
30mm
0.45mm Enhanced BW 1mm 0.45mm BW controlled 1.8mm Port 1 by coupling C. X. Mao, S. Gao, Y. Wang, et al., “Multi-Mode Resonator-Fed Dual Polarized Antenna Array with Enhanced Bandwidth and Selectivity”, IEEE TAP, 63 (12), 5492, 2015 Slide 39 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
2.2 All-resonator filter-array
40 • Filter + power divider + antenna
5 9 4 8 0 -5
2 1 3 -10
(dB) 11
S -15
-20 traditional array 7 11 6 10 proposed array
-25
-30 2.0 2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8 Frequency (dB) Slide 40 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Harmonic suppression and frequency selectivity
0 f 1 10 traditional ring f -10 hairpin 2 patch 0
-20 -10 proposed
(dB) 12
S -30 -20
Antenna Gain (dBi) -30 -40 measured(proposed) simulated(proposed) -40 simulated(traditional) -50 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 Frequency (GHz) Frequency (GHz)
TABLE I PARAMETER COMPARISON WITH TRADITIONAL ARRAY *S-band, 2.39 Antenna Type Traditional Array Proposed Array GHz, FBW=5.6%. Number of Resonant Poles 1 4 * 4th-order (3rd- Fractional Bandwidth 1.60% 5.60% Frequency Selectivity order power 30% 87% divider) (BW-10dB/BW-3dB) Harmonic Level (|S11|) -6 dB -0.7 dB Gain (In band) 9.9 dBi 9.7 dBi Gain (Harmonic) 10.1 dBi -15 dBi Slide 41 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
4-way filtering power divider
0 Port 2Port 3 S11 -10 S21 S31 -20 S41
Port 4Port 5 S51 -30 Port 1
-40
S parameters (dB) -50
-60
-70 2.0 2.2 2.4 2.6 4.5 5.0 5.5 Frequency (dB)
S L 1 2 3
Mao, et al. IEEE MTT. 2016. DOI: 10.1109/TMTT.2016.2561925. Slide 42 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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2.3 Dual-band filter antenna
Dual-band patch
Dual-mode SLR Mao, et al. IEEE TAP., 2016 10.1109/TAP.2016.2574883 Slide 43 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
2.4 Antenna-duplexer integration
44 Txline junction P2 • Resonant antenna as a pole 2 3 4 5 P1 of an all-resonator-based 6 7 8 9 P3 diplexer/duplexer. Resonant junction 2 3 4 5 P2 LNA P1 1
6 7 8 9 P3
Duplexer Antenna 2 3 4 5 P2
PA 1
6 7 8 9 P3 Dual-mode 3 4 5 P2 antenna 1 2
6 7 8 P3 Dual-mode resonator Slide 44 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Dual-band antenna + diplexer
0
P1
HairpinPatch -10 hairpin
L7 Patch_L
slot S11 (dB) Dual-band antenna -20 L8 Ls
Ws Ls=14mm Ls=15mm H1 -30 H2 Ls=16mm Port 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 Frequency (GHz) Port 3 Wf S2 L1
3 4 0 S1 L9 L2 P1 F1 P2
L3 -10 F2 -20 S3 5 6 P3 S1 Port 1/Tx
S4 -30
W1 All-resonator -40 L5 W3 W4 -50 based diplexer S parameters (dB) S5 Wr S11; S13 W2 -60 L4 S22; S23 L6 S12
W5 -70 S1 Z Port 2/Rx Y 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 X Frequency (GHz) Slide 45 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Duplexer-antenna
46
80 mm Dual-band antenna
5
0
-5 patch -10
-15
Duplexer -20 slot S parameter (dB) -25 S11/Tx(simulated) Dual band S11/Tx(measured) antenna -30 S22/Rx(simulated) S22/Rx(measured) -35 Port 1 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 Frequency (GHz) 10
Duplexer/feeding 0
-10
-20 Port 2 Uplink: 2.52 to 2.65 GHz (FBW =5%) -30 -40 port 1 excite (simulated) Downlink: 2.8 to 2.92 GHz (FBW=4.2%) Realized Gain (dBi) port 2 excite (simulated) th -50 port 1 excite (measured) 4 -order per channel; port 2 excite (measured) rd -60 3 -order per channel for the duplexer only. 2.2 2.3 2.4 2.5 2.6 2.7 2.8 2.9 3.0 3.1 3.2 Frequency (GHz) Mao, et al., IEEE MTT 2016 10.1109/TMTT.2016.2574338 Slide 46 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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3. Some other integration schemes
4 poles = 3 SIW cavities + 1 radiating slot
Yusuf, et al, IEEE MTT , 59(4), 857-865, 2011 Zhang et al, IEEE TAP 2016 10.1109/TAP.2016.2574872 Slide 47 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
4. Conclusions: Filter-antennas
• Integrated vs Cascaded Eliminated 50 ohm interfaces • Harmonic suppression Embedded filtering • Bandwidth and selectivity control By coupling – Dual-band and multiple-band – Enhanced bandwidth – Increased selectivity and/or rejection (poles and zero) • Filter-Power-divider-Array Co-design – formed exclusively of resonators • Antenna-duplexer Co-design Integration – functionality – Co-design Slide 48 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Acknowledgements
PhD student Chunxu Mao Funding EPSRC project: Synthesis and new applications of multi-port filtering networks, 2015-17 , Contract No. EP/M013529/1. European Commission FP7 project ‘DIFFERENT’ (grant no: 6069923).
Slide 49 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Compact Power Distributing Devices and Power Amplifiers with Integrated Filtering Response
Xiu Yin Zhang
South China University of Technology, Guangzhou, China [email protected]
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Outline
1. Background
2. Filtering Power Distributing Devices
3. Filtering Amplifier/Switch
4. Conclusion
Slide 51 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Development of mobile communication system devices
Schematic of active MIMO antenna system 4G MIMO MMIC Passive Antenna
5G Massive MIMO Active Antenna
Antenna Miniaturization Low power Mutual consumption coupling High energy Antenna efficiency Filter Filter Switch Amplifier Slide 52 of 175
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Development of wireless terminal devices
RF Subsystem High integration Miniaturization •Smart Watches •Smart Glasses Low power consumption
How to realize ?
Difficult to improve the Co-design efficiency and reduce the size for single devices Slide 53 of 175
Co-designs Based on fabrication techniques
CMOS/GaN/GaAs Integrated Circuit (MMIC) Adv. : Integration of active/passive antenna Disad. : High loss, unsuitable for passive circuits Low-temperature co-fired ceramic (LTCC) Adv. : 3-D design, more degree of freedom Disad. : Challeges in integration of active circuits GIPD Adv. : Integration of active/passive circuits Disad. : Less layers, limited in passive circuits
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Co-designs on circuit functions
Filtering Antenna Filtering Mixer Antenna A Power Divider D Filter Amplifier Mixer Filter 90° Baseband bridge Power Frequency Source A Filtering Divider D Amplifier Mixer
Mixer A D Filtering Mixer Power Divider Reduce circuit size 90° Filtering Amplifier Baseband Reduce mismatching Antenna bridge loss Frequency Source A D Avoid redundant part Mixer Slide 55 of 175
Outline
1. Background
2. Filtering Power Distributing Devices
3. Filtering Amplifier/Switch
4. Conclusion
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2. Filtering Power Distributing Devices
Mixer Antenna Filtering power divider A D Filter Amplifier Mixer Filter 90° Baseband bridge Power Frequency Source A Divider D Mixer
Filtering balun Antenna Differential FilterBalun LNA
Filter
Switch Filter Amplifier Slide 57 of 175
2.1 Filtering Gysel power divider with arbitrary power ratios
Z R Z1 R1 1 1
P2 λ /4 P2 λ/4 g P4 g P4 Filtering λ /4 Z λ/4 Z3 K1 g 3 Zk1 λg/4 g structure
P1 P1 Filtering K2 λ /4 g Z4 Zk2 λg/4 structure λg/4 Z4 P5 P3 λg/4 P5 P3 Z2
Z2 R2 R2
Conventional : Proposed: 1. Without filtering responses 1. Dual functions of power distributing 2. Difficult to realize high power and filtering responses division ratios 2. Obtainable high power division ratios
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2.1 Filtering Gysel power divider with arbitrary power ratios Filtering circuit Equivalent θ2 θ1 circuit C
θ1 θ2
Filtering K-inverter Coupling responses ABCD matrix structure
0 0 jK AB -10 1 CD 0 -20 S jK 21
-30
S -40 11 coscos12 Magnitude (dB) Magnitude K1 -50 2C -60 0.5 1.0 1.5 2.0 2.5 3.0 3.5 Frequency (GHz) Slide 59 of 175
2.1 Filtering Gysel power divider with arbitrary power ratios
Coupling structure
1. Filtering responses is obtained by using coupling structure to replace the
λg/4 transmission line
2. Arbitrary power ratios is realized by tuning gaps g1 and g2
K. X. Wang, X. Y. Zhang*, and B. J. Hu, “Gysel power divider with arbitrary power ratios and filtering responses using coupling structure,” IEEE Transactions on Microwave Theory and Slide 60 Techniques, vol.62. no.3, pp.431-440, Mar. 2014. of 175
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2.1 Filtering Gysel power divider with arbitrary power ratios Simulated and measured results with 1:1 power ratio
S11, S21 & S31 S22, S33 & S23
Center frequency:1.9 GHz; Bandwidth:11%; Insertion loss: 3.0+0.9 dB; Return loss:20 dB; Isolation: >15 dB; Suppression: >20 dB
K. X. Wang, X. Y. Zhang*, and B. J. Hu, “Gysel power divider with arbitrary power ratios and filtering responses using coupling structure,” IEEE Transactions on Microwave Theory and Slide 61 Techniques, vol.62. no.3, pp.431-440, Mar. 2014. of 175
2.1 Filtering Gysel power divider with arbitrary power ratios Simulated and measured results with 1:10 power ratio
S11, S21 & S31 S22, S33 & S23
Center frequency:1.9 GHz; Bandwidth:10%; Insertion loss: 0.9+1 dB & 10.4+1 dB; Return loss:15 dB; Isolation: >15 dB; Suppression: >20 dB
K. X. Wang, X. Y. Zhang*, and B. J. Hu, “Gysel power divider with arbitrary power ratios and filtering responses using coupling structure,” IEEE Transactions on Microwave Theory and Slide 62 Techniques, vol.62. no.3, pp.431-440, Mar. 2014. of 175
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2.2 wide stopband Filtering rat-race hybrid
Z k Filtering P3 P1 λg/4 K-inverter P3 P1
Zk Zk λg/4 λg/4 Filtering Filtering K-inverter K-inverter
P4 P2 Filtering -K-inverter 3λg/4 P4 P2
Zk
Conventional : Filtering rat-race hybrid: 1. Without filtering function 1. Dual functions 2. Bulky in size 2. Size reduction 3. Without harmonic suppression 3. Harmonic suppression Slide 63 of 175
2.2 wide stopband Filtering rat-race hybrid
Filtering structure1 Conventional design: Port 3 Port 1 power distributing Filtering Integrated design： structure 2 power distributing + filering function About 90% size reduction
Port 4 Port 2 Features：λ/4 lineK-inverter Circuit structure 3λ/4 line-K-inverter
K. X. Wang, X. Y. Zhang*, S. Y. Zheng, and Q. Xue, “Compact filtering rat-race hybrid with wide stopband,” IEEE Trans.on Microwave Theory and Techniques, Aug. 2015. Slide 64 of 175
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2.2 wide stopband Filtering rat-race hybrid
l1（θ1） CE l2（θ2） Z line C C Wide-stopband
Port 1 C C Port 2 filtering responses l2（θ2） l1（θ1）
CE 0 Second harmonic (5f ) 0
ABCD matrix Replace λ/4 -10 fh K-inverter transmission line -20 -30 S 21 -40
0 jK -50
AB (dB) Magnitude -60 1 S CD 0 11 jK -70 -80 0 1 2 3 4 Frequency (GHz)
Slide 65 of 175
2.2 wide stopband Filtering rat-race hybrid
（ ） l1 θ1 CE l1（θ1） Z line C C Wide-stopband Port 1 C C Port 2 filtering responses l2（θ2） l2（θ2）
CE Second harmonic (5f0)
0 ABCD matrix Replace 3λ/4 transmission line -5 -K-inverter -10
-15 f 0 jK -20 h AB -25 S 11 S 1 21 (dB) Magnitude CD 0 -30 jK -35
-40 0 1 2 3 4 Frequency (GHz) Slide 66 of 175
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2.2 wide stopband Filtering rat-race hybrid
0 K Simulated 1 b -10 Measured P3 P1 S22 &S33 Path 1 -20
-30 K1 K1 -40 Path 2 -50 I4 S41 a
-K2 (dB) Magnitude -60 U4 P 4 P2 -70
-80 0.0 0.5 1.0 1.5 2.0 2.5 Frequency (GHz) Configuration with two paths from port 1 to port 4 S22, S33 & S41
Signals from path 1 and path 2 are cancelled, the isolation between port 1 and port 4 is better than 30 dB
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2.2 wide stopband Filtering rat-race hybrid
In-phase
0 30 S11 Simulated 20dB Measured -10 20 5 f 0 -20 10
-30 0 (Degree)
-40 31 S
Magnitude (dB) Magnitude - -10 21
-50 S S &S 21 31 -20 -60 Simulated Measured -70 -30 0.0 0.5 1.0 1.5 2.0 2.5 0.44 0.46 0.48 0.50 Frequency (GHz) Frequency (GHz)
S11, S21 & S31 Phase difference
Center frequency:470 MHz; Bandwidth:10%; Insertion loss: 3+1.3 dB; Stopband: 5f0; Amplitude difference:< 0.1dB; Phase difference: < 3°
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2.2 wide stopband Filtering rat-race hybrid
Out-of-phase
0 -150 S44 Simulated -10 Measured 20dB -160 5 f 0 -20 -170 -30 S
-180 S 24 (Degree) -40 34 24 S - Magnitude (dB) Magnitude 34
S -190 -50
-200 -60 Simulated Measured -70 -210 0.0 0.5 1.0 1.5 2.0 2.5 0.44 0.45 0.46 0.47 0.48 0.49 0.50 Frequency (GHz) Frequency (GHz)
S44, S24 & S34 Phase difference
Center frequency:470 MHz; Bandwidth:10%; Insertion loss: 3+1.3 dB; Stopband: 5f0; Amplitude difference:< 0.1dB; Phase difference: < 4.5°
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2.2 wide stopband Filtering rat-race hybrid
Cascaded coupler and filters
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2.3 Tunable Filtering Power Divider With Constant Absolute Bandwidth
Cx C3 C3 R1 R2 M12 M 2L 1 MSL L1 S L MS1 M SL1 M S MS1 2L M SL2
MS3 M12 Ct Ct R Port 2 L2 1 R2 M34 M 4L C2 C4 2 R3 R4 R (a) (b) Port 1 C2 C4 Port 3 Topology Even-mode equivalent topology Ct Ct
00MMS1 SL G1 2 M MMS1 0012 12 M Zin1 2 C C3 C3 x ()MMMM 00MM21 2L 2112LS SL MMSL 002L Change M12 (or MS1 and MSL) to meet the required 100-Ω Conventional filter design even-mode input impedance
L. Gao, X. Y. Zhang*, and Q. Xue, “Compact tunable filtering power divider with constant absolute bandwidth,” IEEE Trans. Microw. Theory Tech., vol. 63, no. 10, pp. 3505–3513, Oct. 2015. Slide 71 of 175
2.3 Tunable Filtering Power Divider With Constant Absolute Bandwidth
24 48 C =1.0 pF (C =0.5 pF) Vt R1 Va5 2 4 C2=1.5 pF (C4=0.8 pF) C3 C3 20 C2=2.0 pF (C4=1.0 pF) 40
16 desired Q 32 ein
Va1 Va3
eout
Vt R1 R1 Vt ein Q
Q 12 24 C1 C1 Port 2 8 16 Folded feeding line Port 1 desired Q eout Port 3 4 8 C1 C 0.60 0.65 0.70 0.75 0.80 0.85 0.90 Vt 1 Vt Freq (GHz)
R1 R1 Va2 Va4 0.07
C3 C3 0.06
Vt desired k Va6 R1 C3=0.5 pF
k 0.05 C3=1.0 pF Frequency tuning and Bandwidth control : C3=1.5 pF 0.04 C =2.0 pF Va1~Va4: connect to the ¼ λ resonator for frequency tuning 3 Va5~Va6: controlling the coupling coefficient 0.03 Qein and Qeout are realized by the folded feeding line with 0.60 0.65 0.70 0.75 0.80 0.85 0.90 capacitors connecting Freq (GHz) Slide 72 of 175
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2.3 Tunable Filtering Power Divider With Constant Absolute Bandwidth
0 0 0 -3 -5 -5 -10 1 V 5.2 V
2 V -10 -10 V =5.2 V -20 4 V Vt=1 V t
(dB) 3 V -15 -15
(dB)
31 33
-30 (dB)
11 -20 -20
& S
S
& S
21
22 S -40 -25 S -25 Simulated Simulated Measured S Measured S 21 -30 Simulated -30 22 -50 Measured S 31 Measured Measured S33 -35 -35 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 1.2 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 Freq (GHz) Freq (GHz) Freq (GHz) S21 S11 S22 and S33
0
-5
-10 Simulated Measured -15
-20
(dB) 23
S -25
-30
-35
-40 0.4 0.5 0.6 0.7 0.8 0.9 1.0 1.1 Freq (GHz) S23 Slide 73 of 175
2.4 LTCC Balun With Bandpass Response Based on Marchand Balun
Feeding lines 11 0.2 mm Quarter-wavelength Port 2 10 0.1 mm Resonator 0.2 mmQuarter-wavelength Half-wavelength9 0.2 mm Resonator Feeding lines Resonator 8 Port 1 7 0.1 mm Port 3 Port 1 6 0.2 mm 5 0.1 mm Feeding lines Input feeding line 4 0.2 mmPort 2 & Port 3 Compared to the Marchand balun, 3 0.2 mm Output feeding only three feeding lines are added. 2 0.1 mm lines The transmission lines in Marchand 1 balun is utilized as resonators. 3-D structure of the proposed LTCC Balun BPF The out-of-phase characteristic is obtained by the inherent character- istic of ½λ resonator.
J.-X. Xu, X. Y. Zhang*, and X. -L. Zhao, “Compact LTCC Balun With Bandpass Response Based on Marchand Balun,” IEEE Microw. Wireless Compon. Lett., to be published, 2016. Slide 74 of 175
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2.4 LTCC Balun With Bandpass Response Based on Marchand Balun
0 200 Measured Synthesized Simulated Simulated -10 Measured 190 S21 & S31 S
-20 11 180
(degree) 31 -30 S
170
-
21
Magnitude(dB) S -40 160
-50 150 3 4 5 6 7 8 5.2 5.3 5.4 5.5 5.6 5.7 5.8 5.9 6.0 Frequency(GHz) Frequency(GHz)
S11, S21 & S31 Phase difference 2 mm×1.7 mm×1.6 mm
Bassband:5.1-5.85 GHz; Insertion loss: 3+1.8 dB; Return loss:15 dB; Suppression: >20 dB; Amplitude difference: < 0.1dB; Phase difference: < 5°
Slide 75 of 175
Outline
1. Background
2. Filtering Power Distributing Devices
3. Filtering Amplifier/Switch
4. Conclusion
Slide 76 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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3. Filtering High efficient Amplifier/Switch
Antenna Antenna LNA LNA
Duplexer Filter Switch Amplifier Amplifier
FDD RF front-end TDD RF front-end
Amplifier Amplifier+Filter Filtering Input Output Input Matching Transistor Matching Filter Output Matching Transistor Network Network Matching Network Network ’ ’ Zin=R+jX Zout Zin Zout=50Ω Zin=R+jX Zout=50Ω
Problems: Zin’ deviates 50 Ω，Zin deviates optimal impedance，efficiency drops Slide 77 of 175
3. Filtering High efficient Amplifier/Switch
From “K.L. Chen, J. Lee, W. J. Chappell and D. Peroulis, TMTT, 2013”.
Co-design: simplified matching network; reduced loss; enhanced efficiency
Slide 78 of 175
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3.1 High Efficiency Power Amplifier With Bandpass Response
000 M S1 MMM 0 M 2 ZQ'' S11112 1 12 inein M Zin 22 0 MMM MMSL12 ZQinein 21222 L 000 M 2L Change Qe 1 M S1 QFBWein
Impedance Transformation & Wide range impedance transformation Filtering Function Expand Bandwidth 50 ohm 9~17 ohm （PA：2.1-2.6 GHz） Slide 79 of 175
3.1 High Efficiency Power Amplifier With Bandpass Response
Impedance Transformation & Filtering Function
Proposed PA (2.45 GHz)
Z=Zopt at fundamental open at odd harmonics short at even harmonics
F-class Amplifier
L. Gao, X. Y. Zhang*, S. C. Chen, and Q. Xue, “Compact power amplifier with bandpass Slide 80 response and high efficiency,” IEEE Microwave and Wireless Components Letters, Sept. 2014 o. f 175
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3.1 High Efficiency Power Amplifier With Bandpass Response
Proposed Amplifer K. L. Chen, et al（f=3GHz，2013，TMTT） PAE=69 % PAE=69.8% The PAE is comparable, and the proposed design >Amplifier+Filter using microstrip lines has more compact size and light weight. L. Gao, X. Y. Zhang*, S. C. Chen, and Q. Xue, “Compact power amplifier with bandpass Slide 81 response and high efficiency,” IEEE Microwave and Wireless Components Letters, Sept. 2014 o. f 175
3.2 Class-F Power Amplifier with Bandpass Response Based on Hybrid Cavity-Microstrip Filter
Coupling window Resonator 1 00MMS1 SL Resonator 2 MMM 0 11112S M a 0 MMM21222 L To Transistor MMLS 00L2 Gap g 1 To Output Stub 1 Ground via S-L Couling Hybrid cavity-microstripPCB filter For 1-Ω system, L 11 2 M S1 L12 H-Field Dout M12 M 'S1 L14 Zin1 2 L13 L R 16 g1 E-FieldDin ()MMM2112LS M SL Port_1 Port_2 M L15 Ground SL via M 'SL For (R+jX) Ω system, R
dc 2 L L8 18 L XM Rin2 19 22' 12 R L17 out2 W2 M12 jM 11 M 2L M ' Rout1 11 2 R Z in1 in2 ' '2 RM 2L ()MMMM2LS 1 12 SL W1 L1 Slide 82 Short-circuited for 3 f0 of 175
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3.2 Class-F Power Amplifier with Bandpass Response Based on Hybrid Cavity-Microstrip Filter
VGS VDS 0 Synthesized C4 C5 C6 C7 C8 C9 C10 C11 -10 Simulated Ω Hybrid Cavity-Microstrip Filter Zin=(13.5+j4.5) Line 3 Line 6 -20 A B C C2 C1 C3 -30 S Line 1 Line 2Line 4 Line 5 Line 7 11 In R1 CGH40010 Out IMN
A’ B’ C’ -40 Magnitude (dB) S21 Class-F amplifier with bandpass responses -50
-60 1 2 3 4 5 6 7 8 9 2nd S.C Frequency (GHz)
Hybrid Cavity-Microstrip Filter 80 Line 6 2.5 4 A B 60 C 2.0
nd Current(A) Line 5Line 7C3 Line 8 2 _drain 40 1.5
rd 1.0 3 open 20 nd rd 0.5 2 short 3 S.C Voltage (V) rd 0 Stub 1 3 _drain 0.0 rd 3 O.C -20 -0.5 0 200 400 600 800 Time (pSec)
Zina A’ ZinbB’ Zinc C’ Slide 83 of 175
3.2 Class-F Power Amplifier with Bandpass Response Based on Hybrid Cavity-Microstrip Filter
20
10
0
-10 Filtering
-20 responses Measured S21
Magnitude (dB) -30 Simulated S21 Measured S -40 22 Simulated S22 -50 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 Freq (GHz) 80 50 90
70 40 PAE Measured Pout 80 40 Pout Measured PAE 60 Efficiency 30 70 （
30 % ） High 50 60
20 PAE(%) 20 efficiency 40 & PAE 50 10 30 Efficiency 10 Gain 40 0 20 0 30 -10
10 （ Output power (dBm) 20
% ） PAE>70%
-20 0 Pout (dBm) & Gain (dB) -10 10 -30 -10 1.0 1.5 2.0 2.5 3.0 3.5 4.0 -20 0 10 15 20 25 30 Slide 84 Frequency (GHz) Pin (dBm) of 175
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3.3 Filtering Switch with Low ON-State Loss and High OFF-State Isolation Based on Dielectric Resonator
Antenna Loss: Filter loss + Switch loss LNA Isolation: Turn off Switch BPF Switch BPF Switch
PA Embeded the PIN diodes or Tran- sistor into the filter structure TDD Sub-System Filtering switch FBW 10%
C. –S. Chen, et al, MWCL, 2014 Slide 85 of 175
3.3 Filtering Switch with Low ON-State Loss and High OFF-State Isolation Based on Dielectric Resonator
S1 S1 z
A B o E-field
y y y S1’ S1’ x x (a) x (b) S1 S1 L H-field TE11δ
y y S1’ S1’ x x (c) (d)
z Metal cavity E-Field H-Field T-shape x metal probe I1 Short branch line o I2 Short or open I I2 main line l y 1 l/2 x branch line y Slide 86 of 175
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3.3 Filtering Switch with Low ON-State Loss and High OFF-State Isolation Based on Dielectric Resonator
z z Center plane DR
y y x short x short
L open short metal probe Different L/2 Type II from Type I Type I
Center plane Center plane EEdv 12 ke 0 22 EdvEdv 12
HH dv 12 km 0 y 22 x HdvHdv H-field of DR E-field of DR 12 H-field of metal probe E-field of metal probe Slide 87 of 175
3.3 Filtering Switch with Low ON-State Loss and High OFF-State Isolation Based on Dielectric Resonator
0 0 S11 short open short Symmetric plane short -1 -20 Theoretical switch on switch off -2 Simulated Port circuitry Port circuitry Measured -3 -40 1.824 1.828 1.832 1.836 1.840 switch PCB S circuitry -60 21
Metal Magnitude (dB) R z cavity C PIN x -80 PCB
DR -100 2 Tuning 1.70 1.75 1.80 1.85 1.90 1.95 DR Input 1 disks Freq (GHz) High selectivity feeding line 0 S 11 Simulated Metal Measured -20 probe
Output -40 feeding line >53-dB isolation DR3 DR4 z -60 S21
y Magnitude (dB) x -80
-100 Filtering SPST switch 1.70 1.75 1.80 1.85 1.90 1.95Slide 88 Freq (GHz) of 175
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3.3 Filtering Switch with Low ON-State Loss and High OFF-State Isolation Based on Dielectric Resonator
0
NRN -10 Measured Simulated Branch -20 line -30 Tuning S21 S11 disks -40 Branch main line -50 line S31
PCB Magnitude (dB) -60
-70 S R3 R1 R2 32 NRN2 PCB -80 Metal 1.6 1.7 1.8 1.9 2.0 2.1 NRN 1 cavity 0 Freq (GHz)
z -10 Measured y Simulated L -20 L2 S DC z L Cg y 1 -30 S x S11 31 Filtering SPDT switch -40
-50 S21 L2 S L1 Magnitude (dB) -60 S Filter 1 23 Filter 2 -70
-80 1.6 1.7 1.8 1.9 2.0 2.1 Freq (GHz) J.-X. Xu, X. Y. Zhang*, J. X. Chen, and Q. Xue, “Filtering Switch R NRN R NRN R 3 2 1 1 2 with Low ON-State Loss and High OFF-State Isolation Based on main coupling Slide 89 cross coupling Dielectric Resonator ,” TMTT, submitted. of 175
3.3 Filtering Switch with Low ON-State Loss and High OFF-State Isolation Based on Dielectric Resonator
3-dB Filtering Ref. Type IL (dB) Is (dB) Technology FBW responses [18] DR filter 1.15 - > 0.7 % Yes DR+Cavity [19] DR filter 0.5 - > 1.6 % Yes DR+Cavity [25] SPST 1.5 30 - No Coaxial cavity [26] SPST 0.6 35 - No pHEMT GaAs Cascaded SPST 1.1 36 1.6 % Yes - [19], [26] [27] SPDT 0.5 38 - No pHEMT GaAs [28] SPDT 0.4 29 - No pHEMT GaAs [29] SPDT 0.6 27 - No pHEMT GaAs Cascaded SPDT 1 38 1.6 % Yes - [19], [27] [3] SPST ~3 44 9 % Yes PCB [4] SPST 2.75 35 - Yes PCB [9] SPDT 0.97 40 10 % Yes PCB [10] SPDT 3.6 47 5 % Yes PCB SPST 1 53 0.63 % Yes DR+Cavity This work SPDT 0.4 45 1.3 % Yes DR+CavitySlide 90 of 175
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Outline
1. Background
2. Filtering Power Distributing Devices
3. Filtering Amplifier/Switch
4. Conclusion
Slide 91 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Conclusion
1. Four power distributing devices integrated with filtering responses have been presented, including: -- a filtering Gysel power divider with arbitrary power ratios -- a wide stopband filtering rat-race hybrid -- a tunable filtering power divider with constant absolute bandwidth -- a LTCC Balun filter based on Marchand balun
2. Power amplifiers/switches with filtering responses have been presented, including: -- a power amplifier with filtering responses using microstrip resonator -- a power amplifier with filtering responses using hybrid microstrip-cavity resonator -- filtering SPST and SPDT switches using rectangular dielectric resonator
The presented passive and active devices integrated with filtering responses feature compact size, low power consumption and high energy efficiency, which are suitable for modern communication systems, such the massive active MIMO antenna system. Slide 92 of 175
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Waveguide components based on all coupled resonators
Mike Lancaster
The University of Birmingham, UK
Slide 93 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Waveguide filter based on symmetrical capacitive irises using the coupling matrix
coupling iris output port 2 resonator
Qe=10.8
Qe=10.8
Coupling matrix Slide 94 of 175
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Coupling matrix calculation
Qe
• Calculation of coupling matrix and Qe is simple for filters but more complex for the multiport. • We can use analytical expression in some case • Use optimisation in the more general case (this is another talk) • For this talk lets assume we can find it. Slide 95 of 175
Finding sizes of coupling apertures
coupling iris resonator • The size of the resonator Qe gives the frequency • The coupling coefficient in the coupling matrix is
Qe related to the size of the waveguide aperture • Assume we can do this transformation for this talk.
Coupling matrix Slide 96 of 175
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General coupling matrix (multiport)
•One input port is shown it is also possible to have multiple input ports
......
2 2 •Input connected to port 1 S 1 A )(1 S A )(1 11 11 1s 1s •s represents resonator number s q1 1qqs qe scaled external quality factor, P lowpass frequency variable , mij normalized coupling coefficient between resonators i and j, mii self coupling coefficients for asynchronously tuned filter Slide 97 of 175
Other all-resonator devices!
Some microwave devices are: Components used in the devices: • Complex filters including multiband • Resonators • Multiplexers • Signal splitting • Butler matrix networks • Orthomode transducers • Transmission lines • Antennas • Coupling circuits • Antenna arrays • Feed networks • Active component • Matching circuits interconnections
• Coupling matrix and coupled resonator designs have traditionally only been used for two port filters • Propose to use only resonators in these devices • Size and mass reduction, more flexible design, wider specification • Look at multiplexers first Slide 98 of 175
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Resonator Multiplexer Topologies
1 2
[Lancaster 2001] [e.g. Macchiaella 2010] [ Carpintero 2006]
3 4
[Lancaster 2001]
Slide 99 99 of 175
4 Channel Multiplexer Coupling Matrix
5 6 7 P2
3 8 9 10 P3 P1 1 2 Self coupling 11 12 13 P4 (frequency) 4 14 15 16 P5
0 0.01743 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.01743 0 0.0096 0.0096 0 0 0 0 0 0 0 0 0 0 0 0 0 0.0096 0.01074 0 0.00419 0 0 0.00315 0 0 0 0 0 0 0 0 0 0.0096 0 -0.01074 0 0 0 0 0 0 0.00419 0 0 0.00315 0 0 0 0 0.00419 0 0.01967 0.00223 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.00223 0.02082 0.0046 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.0046 0.0209 0 0 0 0 0 0 0 0 0 = m 0 0 0.00315 0 0 0 0 0.00727 0.00213 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.00213 0.00703 0.00456 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.00456 0.00702 0 0 0 0 0 0 0 0 0 0.00419 0 0 0 0 0 0 -0.01967 0.00223 0 0 0 0 0 0 0 0 0 0 0 0 0 0 -0.00223 -0.02082 0.0046 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.0046 -0.0209 0 0 0 0 0 0 0.00315 0 0 0 0 0 0 0 0 0 -0.00727 0.00213 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.00213 -0.00703 0.00456 0 0 0 0 0 0 0 0 0 0 0 0 0 0 0.00456 -0.00702
Qe1=77.59 Qe2=Qe5=308.66 Qe3=Qe4=312.35 Slide 100 of 175
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4 Channel all resonator multiplexer
• Layout in waveguide • X-Band • Made in aluminium
Slide 101 of 175
4 Channel Multiplexer
0
-10
-20
-30
-40
-50
S Parameter(dB) S -60
-70
-80
-90 Centre frequency changing alters 9.8 9.85 9.9 9.95 10 10.05 10.1 10.15 10.2 Frequency(GHz) which resonators are operational Simulation (dashed lines) and measurement Result – cut out all power splitters (solid lines) results [Shang 2013] Slide 102 of 175
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180 Degree Hybrid coupler
• Microstrip 90 degree hybrid coupler • Resonator based 180 degree coupler • One negative coupling S11 • Two paths A and B in diagram S31 • Includes two resonator filter in each arm S41
Slide 103 of 175
Two Butler matrix forming multiport amplifier
Hybrid coupler
INET ONET • Multiport power amplifier • Signal in port 1 comes out at
O port 8 utputs • Signals are spread through amplifier so there is a Inputs redundancy. • Array of hybrid couplers • Requires external filtering
High Power Amplifier Vittorio Tornielli di Crestvolant Funded by ESA and Birmingham University NPI with Petronilo Martin Iglesias as ESA supervisor Slide 104 of 175
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Multiport amplifier with filtering
Filtering hybrid coupler
• Use 180 hybrids so filtering
O can be included utputs • INET can be conventional hybrid network or filtering Inputs network • ONET is the filtering network which includes the bandpass filter
High Power Amplifiers
Slide 105 of 175
Filtering Butler matrix i.e. INET/ONET
• 4 input and 4 output resonator construction. • 16 Resonators • Ports labelled p are the input • Ports labelled q are the outputs • Plot from the coupling matrix Slide 106 of 175
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4x4 Butler matrix - measurements
• Measurements are in agreement for the insertion loss, the power splitting and the overall filtering and phase characteristic. • WR75 waveguide (10-15 GHz) • No tuning performed.
Slide 107 of 175
Size reduction
The volume of the 4x4 Butler matrix is compared with a conventional Butler matrix with 4 band-pass filters of the 4th order cascaded
•The volume of the baseline Butler matrix + 4 BPF is 96.7 cm3 •The volume of the new Butler matrix based on resonators is 66.2 cm3 •The Butler matrix based on resonators is 31.5% less in volume with respect the conventional •Reference [Tornielli di Crestvolant 2015] Slide 108 of 175
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Single Antenna-filter
Aperture Radiated la b • Many antennas are resonators e.g. tr dipoles, loops, slots. dr dz d 3 • Use the resonant antenna as one of y dk12 the resonators in the connected 2 lr do t filter. 1 l2 • Removes the matching circuit and a t l1 decreases the size of the filter. t b lo • Waveguide filter used to Input Embedded demonstrate port Waveguide Flange • The radiation Q of the antenna is M M the same as the external Q of the Input port 12 23 Output Q 1 2 3 Q filter ext r • Some limitations as the radiation Q Single waveguide aperture antenna rd limits the Qe in the equivalent filter integrated with the 3 order bandpass filter. Slide 109 of 175
Single Antenna-filter Fabrication and Measurement
10 10 Input 5 5 Port 0 0 Aperture -5 -5
-10 -10 23 1 -15 -15
Parameter (dB) Parameter -20 -20 1 1 S
-25 -25 GainRealised (dBi) S (Simulated) -30 11 -30 S (Measured) 11 -35 R. Gain (Simulated) -35 R. Gain (Measured) Screw holes -40 -40 Alignment Pin 8 8.5 9 9.5 10 10.5 11 11.5 12 Frequency (GHz)
Photograph of the fabricated Measured S11 and realised gain Single waveguide Aperture responses of the single waveguide antenna integrated with the aperture antenna integrated with the 3rd order bandpass filter. 3rd order bandpass filter Slide 110 of 175
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4x4 Antenna-filter Array Based on Resonators Topology and Realised Physical Structure
M1624 24 P2 16 M1216 25 P3 M48 M812 Feeding 4 8 12 Aperture antenna 26 P4 Feeding network based based on all-resonator Networks M 24 17 on all-resonator stuctures 27 P5 structures 2 28 P6 18 29 P7
M12 5 9 13 30 P8 19 31 P9
P1 1 32 P10 Input port 20 Source 33 P11 Ē 6 10 14 Qext1 34 P12 21 35 P13 3 36 P14 22 Input Port 37 P15 7 11 15 38 P16 23 39 P17 Full layout of the simulated model in CST for 12-dB power splitter the 4x4 antenna-filter array. topology. Slide 111 of 175
4x4 Antenna-filter Array Based on Resonators Fabrication and Measurement
30
20
10
0
-10
-20
&Gain Realised 1
1 -30 S
S (dB) simulation -40 11 S (dB) measurement 11 -50 Realised Gain (dBi) Simulation Realised Gain (dBi) measurement -60 8 8.5 9 9.5 10 10.5 11 11.5 12 Frequency (GHz)
The performance of the 4x4 antenna-filter array.
[Rashad Mahmud 2015] Slide 112 of 175
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4x4 Antenna-filter Array Based on Resonators Measurement
Measurement Measurement 10 GHz H-Plane 10 GHz E-Plane 0 Simulation 0 Simulation 0 0 330 30 330 30 -5 -5 -10 -10
-15 -15 300 60 300 60 -20 -20
-25 -25
-30 -30
270 -35 90 270 -35 90 0 -5 -10 -15 -25-20 -35-30 -35 -30 -25 -20 -15 -10 -5 0 0 -5 -10 -15 -25-20 -35-30-35-30-25-20-15-10 -5 0 -30 -30
-25 -25
-20 -20 240 120 240 120 -15 -15
-10 -10
-5 -5 210 0 150 210 0 150 180 180 E-plane Radiation Pattern H-plane Radiation Pattern of the 4x4 antenna-filter of the 4x4 antenna-filter array. array. Slide 113 of 175
30 to 90 GHz tripler Topology
• Tripler is based on all resonator structures. • Diodes are coupled into the resonators • The design removes any filtering and matching from the microstrip (lossey) to waveguide
Input Output
114 Slide 114 of 175
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30 to 90 GHz tripler Coupling matrix (N+2)
Input filter matrix Output filter matrix 00.1084000 00.137100 0.108400.103200 0.13710.07520.16620 M 00.103200.10320 M 00.166200.1226 000.10320.02570.2144 000.12260 0000.21440 External Self coupling Qe1 M11 k Qe2 coupling coupling coefficients rd Input 3rd order MBPF output 2 order MBPF
InputIn 1 2 3 Coupling Coupling 4 5 Outputout
115 Slide 115 of 175
30 to 90 GHz tripler Measurement
• Simulated conversion loss is shown in black (note bandwidth). • Measured conversion loss is shown in red. (about 2dB more than simulated.) • Simulated conversion loss including the assembly error (0.1mm shift of the substrate) is shown in blue. • No S11 and S22 measurements yet
[Cheng Guo 2015], 116 Slide 116 of 175
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Conclusion and Summary
•Size and volume reduction of circuits made of just resonators •Better performance •Improve manufacturability •Coupling matrix is an essential tool for formal design •Some coupling matrices can be derived analytically, some more difficult (work required) •Easy to transfer coupling matrix to actual physical device ready for final optimisation (work required) •Examples of filter, multiplexer, hybrid, multiport power amplifier, antenna array and tripler have been given. •Other circuits possible e.g. diplexer, OMT, Amplifier
Slide 117 of 175
A final thought
All band limited passive microwave circuits can be made of only resonators?
Slide 118 of 175
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References
• [Shang 2012], M Ke, Y Wang, and M J. Lancaster “WR-3 Band Waveguides and Filters Fabricated Using SU8 Photoresist Micromachining Technology” IEEE Transactions on terahertz science and technology V2(6) pp 629 2012 • [Lancaster 2001] M J Lancaster ‘A radio frequency filter’, International patent classification HO1P 1/213, International application number PCT/GB01/01188, Filing date 16/3/2001. Inventor M J Lancaster • [Carpintero 2006] IH Carpintero, MJP Cruz, AG Lamperez Patent App. 11/283,773, Page 1. (i9) United States (12) Patent Application Publication Hidalgo Carpintero et al. US 20060114082A1 (io) Pub. No.: US 2006/0114082 Al (43) Pub. Date: Jun. 1, 2006 (54) GENERALIZED MULTIPLEXING NETWORK • [Macchiarella 2010] G Macchiarella ‘Synthesis of Star Junction Multiplexers’ IEEE MTT V58(12) : pp3732 • [Shang 2013]; Yi Wang; Wenlin Xia; Lancaster, M.J. ‘Novel Multiplexer Topologies Based on All-Resonator Structures.’ IEEE Transactions on Microwave Theory and Techniques, V61(11). 2013 pp3838 – 3845 • [Tornielli di Crestvolant 2015], V.; Martin Iglesias, P.; Lancaster, M.J. ‘Advanced Butler Matrices With Integrated Bandpass Filter Functions’ IEEE Transactions on Microwave Theory and Techniques, 2015,63(10) pp 3433 – 3444 • [Rashad Mahmud 2015 ], M Lancaster, F Huang “Filtering waveguide antenna arrays’ to be published • [Cheng Guo 2015], Jianyu Liu, Yang Gao, Michael J. Lancaster, Jun Xu, Jeff Powell, Jianping Gong and Yuliang DongNovel ‘W-band frequency tripler design by using embedded all resonator BPFs’ To be published
Slide 119 of 175
Single/Multi-Band Power-Distribution and Impedance-Transformation Planar Circuits with Added Static and Reconfigurable Bandpass Filtering Functionality
R. Gomez-Garcia1, D. Psychogiou2, and D. Peroulis2 [email protected] 1University of Alcalá, Spain 2Purdue University, USA Slide 120 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Overview
- Introduction
- Multi-Functional RF Filtering Components
- Reconfigurable RF Filtering Components
- Outlook
Slide 121 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Introduction
…Enormous amount of applications & functionality… Multifunctional transceiver: 2G/3G/LTE (Fujitsu):
Digital GPS TV
4G W-CDMA Seamless GPRS Any time
Any network WLAN Bluetooth RFID Any device
Limitations: • Complexity & size *Fujitsu Multiband • Power consumption & cost RF transceiver Go tunable and multi-functional ! 20 filters ! Slide 122 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Introduction
Crowded radio spectrum & dense communication systems
Victim Base Station Transceiver
Tx PA Victim Interference: Critical in Receiver Rx LNA most modern communication Aggressor Transmitter and radar systems Self-blocking/Interference Interference in adjacent channels Emissions in adjacent bands External blockers Slide 123 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Introduction
Modern UWB receiver: Interference mitigation with adaptive multi-band BSF Key block: Purpose of this work !
(dynamic) Composite RF filtering functionality
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-Section I- Multi-Functional RF Filtering Components
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Multi-Functional Devices
• Purpose: Combination of filtering operation with other types of RF analog-processing actions Wilkinson-type power dividers - Application to channelized filters Impedance-transformation networks Balanced filters • Advantages: More compact designs Optimized performance through multi-function co-synthesis Lower losses
SIGNAL-INTERFENCE FILTERING PHILOSOPHY ! Slide 126 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Signal-Interference Filters
- Operational Principles - Transversal filtering section Example RF response
RF signal routing in multiple paths
Operational principle: Transversal interference between paths -Destructive for unwanted frequencies: transmission zeros (1) - Constructive for the desired passbands: signal enhancement (2) Slide 127 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Signal-Interference Filters
Dual-band signal interference bandpass filter - Examples -
RF performance merits: R. Gomez-Garcia et al. in IEEE MWCL, Dec 2009 • Multi-band • Arbitrary bands Low-pass signal interference bandpass filter • Wide stopbands • Quasi-elliptic-type • Sharp-rejection
R. Gomez-Garcia et al. in IEEE TMTT, Dec. 2013 Slide 128 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Filtering Power Dividers
• 1st family of devices: Wilkinson-type power dividers with added single/multi-band filtering capabilities
Replacement of the transmission-line arms of the divider by a bi-path signal-interference TFS
. Power splitting: - Wilkinson-type scheme . Filtering action: - Signal-interference TFS
. Performance: High-selectivity single/multi-band filtering with symmetric/asymmetric power distribution Slide 129 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Filtering Power Dividers
. Design example: Quad-band Wilkinson power divider with symmetrical power division - Frequency response - - Implementation - Substrate: RO4003C Rogers 휀푟 = 3,55 퐻 = 1.52 mm tan 훿퐷 = 0.0027 푇 = 35 휇m
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Filtering Power Dividers
. Design example: Lumped-element dual-band Wilkinson- type power divider with symmetrical power division - Implementation - - Frequency response - Substrate: RO4003C Rogers 휀푟 = 3,55 퐻 = 1.52 mm tan 훿퐷 = 0.0027 푇 = 35 휇m
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Channelized Active BPF
. Application to higher-level RF components: Signal power splitter/combiner of a channelized active BPF - Schematic - - Principle -
R. Gómez-García, R. Loeches-Sánchez, D. Psychogiou, and D. Peroulis, “Single/multi-band Wilkinson-type power dividers with embedded transversal filtering sections and application to channelized filters,” IEEE Trans. Circuits Syst. I, Reg. Papers, vol. 62, no. 6, pp. 1518-1527, Jun. 2015. Slide 132 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Channelized Active BPF
• Design example: Channelized active BPFs - Implementation - - Frequency response- Ideal Substrate: RO4003C Rogers 휀푟 = 3,55 퐻 = 1.52 mm tan 훿퐷 = 0.0027 푇 = 35 휇m
Meas
Slide 133 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Filtering Impedance Transformers
• 2nd family of devices: Impedance transformers with bandpass filtering capabilities
Replacement of the quarter-wavelength transformer by a bi-path signal-interference TFS
. Impedance transformation: - Quarter-wavelength transformer . Filtering action: - Signal-interference TFS
. Performance: High-selectivity single/multi-band filtering with transformation between any arbitrary pair of real impedances Slide 134 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Filtering Impedance Transformers
• Design example: Dual-band impedance transformer - Implementation - (푅푖푛 = 푍0 and 푅퐿 = 푍0/2) Substrate: RO4003C - Frequency response- Rogers 휀푟 = 3,55 퐻 = 1.52 mm tan 훿퐷 = 0.0027 푇 = 35 휇m
R. Loeches-Sánchez, D. Psychogiou, R. Gómez-García, and D. Peroulis, “Transformers with incorporated filtering capabilities exploiting signal-interference principles,” in 2015 IEEE Int. Conf. on Microw., Commun., Antennas Electron. Syst., Tel Aviv, Israel, Nov. 2-4 Slide 135 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Filters with Balanced Operation
• 3rd family of devices: High-selectivity balanced BPFs with common-mode suppression Incorporation of signal-interference TFSs into the two halves of a balanced structure
. Differential mode: - Filtering action shaped by the TFS . Common mode: - Suppression due to the balanced structure
. Performance: High-selectivity single/multi-band filtering action in differential mode and high attenuation levels in common mode Slide 136 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Filters with Balanced Operation
• Two-TFS design : - Scheme -
• Increased filtering selectivity: Amplitude flatness, out-of band rejection levels, and shapness of cut-off slopes • Use of dissimilar TFSs to further extend the rejection band
R. Loeches-Sánchez, D. Psychogiou, R. Gómez-García, and D. Peroulis, “A class of differential-mode single/dual-band bandpass planar filters based on signal-interference techniques,” in 2016 IEEE Wireless Microw. Technol. Conf., Clearwater Beach, FL, USA, Apr. 11-13, 2016. Slide 137 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Filters with Balanced Operation
• Design example: Single-band balanced BPF - Implementation - - Frequency response- Substrate: RO4003C Rogers 휀푟 = 3,55 퐻 = 1.52 mm tan 훿퐷 = 0.0027 푇 = 35 휇m
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-Section II- Reconfigurable RF Filtering Components
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Bandpass/Bandstop Filters
Objective: Tune-all BPF & BSF transfer function Operating principle: Transfer-function control through TZ reallocation
• 2 transmission zeros (TZ) • TZ @ center frequency of R
• Passband: ZR1=ZR2 • Tune resonators NOT couplings
D. Psychogiou, R. Gómez-García, and D. Peroulis, “A fully-reconfigurable bandpass/bandstop filters and their coupling- matrix representation,” in IEEE Microw. Wireless Compon. Lett., vol. 26, no. 1, pp. 22-24, Jan. 2016. Slide 140 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Bandpass/Bandstop Filters
Experimental validation for two-stages: K=2
BPF: Frequency tuning BPF: symmetric BW tuning Integration technology: SIW cavity-resonators & piezo. actuators BPF/BSF stage
Two-stage BPF: asymmetric BW tuning BSF: BW & IS tuning prototype
Simultaneous BPF/BSF response Coupling-less tuning BW tuning:45-230 MHz (5:1) Frequency tuning: 2.9-3.6 GHz Intrinsic switching: w/o switches! Slide 141 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Adaptive Multi-Stopband Filters
Objective: Create multiple bands & tune them independently Architecture & operating principles: N-band BPF section • No interaction between R- nodes: → Less sensitive in tuning
• Tune R-nodes NOT couplings → Easier tuning
• Independent control of all passbands: → Increased reconfigurability
• Scalable to any no. of bands
R. Gómez-García, A. C. Guyette, D. Psychogiou, E. J. Naglich, and D. Peroulis, “Quasi-elliptic multi-band filters with center- frequency and bandwidth tunability,” in IEEE Microw. Wireless Compon. Lett., vol. 26, no. 3, pp. 192-194, Mar. 2016. Slide 142 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Adaptive Multi-Passband Filters
Experimental validation for three quasi-BPF sections & 2 bands Frequency & BW tuning of bands Independent tuning of bands Intrinsic switching-off bands
Prototype: Microstrip-line implementation Unique key-advantages: NRN-NRN Reduced size & complexity than filter-banks coupling Closely-spaced adjacent bands Independent tuning: single & multiple bands
C-loaded R NRN-R coupling Intrinsically-switched bands NRN-R coupling Slide 143 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Adaptive Multi-Stopband Filters
Architecture & operating principle: Experimental validation: N-band BSF section f control
BW control Center freq. tuning
Prototype BW. tuning Reduced size & complexity than BSF-banks Independent tuning of bands: BW, frequency & number Scalability: any no. of bands No. of bands & D. Psychogiou, R. Gómez-García, and D. Peroulis, “A class of fully-reconfigurable BW control planar multi-band bandstop filters,” in IEEE MTT-S Int. Microw. Symp., San Francisco, CA, USA, May 22-77, 2016. Slide 144 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Bandpass Filter with In-band Notch
Transversal filtering section: Experimental validation: Operating principle: Signal interference One TFS-AN cell: Notch tuning through R Path 1 Input Output 0
Z1, θ1
KE KE Path 3 -15 R K12 R Path 2 Destructive Constructive Z , θ 2 2 TFS-AN interference interference Path 1 Path 1 -30 Transmission (dB) Transmission
Z1, θ1 Z1, θ1 KE KE Path 2 Path 3 Tune C of R -45
R K12 R 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 Z2, θ2
Frequency (GHz) dB
dB Cascade of two TFS-AN cells and notch tuning
|, |,
|, |, 21
21 0
|S |S -10 Frequency, f Frequency, f -20
Filter: Series-cascade of cells -30
-40 Tune C of R Parameters (dB) Parameters
- -50 S D. Psychogiou, R. Gómez-García, and D. Peroulis, “Signal interference -60 bandpass filters with dynamic in-band interference suppression,” in -70 1 cm 2016 IEEE Radio Wireless Symp., Austin, TX, USA, Jan. 24--27, 2016. 1.0 1.2 1.4 1.6 1.8 2.0 2.2 2.4 2.6 2.8 Frequency (GHz) Slide 145 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Tunable Multi-Band Filter/Divider
Objective: Filtering & power division capabilities in a single device
BPF @ f1
f1
` Matching f1 fN Conventional network 2 2 Proposed 1 1 design 3 3 design
fN Power divider BPF: f1...fN
BPF @ fN Architecture & operating principles Insert multi-band BPF stages in divider arms Quad-band example
D. Psychogiou, R. Gómez-García, A. C. Guyette, and D. Peroulis, “Reconfigurable single-/multi-band filtering power divider based on quasi-bandpass sections,” in IEEE Microw. Wireless Compon. Lett., accepted for publication, 2016. Slide 146 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
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Tunable Multi-Band Filter/Divider
Independent tuning of bands Prototype
Single/multi-band filtering Frequency control Band merging & in-band notch Passband on/off commutation Arbitrary bands & division ratio
Comparison with S-O-A filtering power dividers
Slide 147 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Outlook
• Reconfigurable RF filtering components: Single/multi-band operation with signal-interference philosophy High-selectivity with TZ creation General approach: dividers, transformers, balanced networks • Reconfigurable RF filtering components: Tune-all transfer function Multi-band BPF/BSF: intrinsic switching & band control Incorporation of power division & filtering In-band interference suppression
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Synthesis techniques for multiplexers and multiport selective networks
Giuseppe Macchiarella
Politecnico di Milano
Slide 149 WM05 Microwave Passive and Active Devices with Integrated Filtering Functions of 175
Outline
– Introduction – Diplexers/Multiplexer design: a polynomial synthesis approach – Beyond classical task: new combiners architectures – Design of multi-band Masthead combiners, – Design of same-band combiners – Conclusions
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Classical Selective Combiners
• Fundamental task: connect the output of a transceivers to one (or more) antennas • Today multi-standard services typically require several combiners in a single site (Diversity antennas further increase the number) • A combiner is composed by two (or more) filters suitably connected • Very high selectivity, low passband loss high linear response call for a complex frequency characteristic, including TZs both imaginary and complex (for phase equalization) • Non reciprocal component are generally forbidden (Passive Intermod. Issues) • Coaxial technology is typical up to few GHz (comb resonators with capacitive loading) • Topology: star-junction with 1 to 3 channels typically Slide 151 of 175
Example of Duplexer configuration
TX Filter Shunt Junction 2 TX 2 Filter 1 3-port 1 Junction 3 RX 3 Filter RX Filter
In order to make possible a synthesis approach to the whole 3- port network the junction must connect in parallel (or in series) the input ports of the filters
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Normalized frequency domain
The synthesis is carried out in a normalized frequency domain W which is defined by the usual low-pass ↔ band-pass frequency transformation:
sjjfBffffW 0 00
The parameters f0 and B are defined through the following frequency mapping:
f Bandpass f1,RX 2,RX f0 f1,TX f2,TX domain f RX TX fff01,2,RXTX Bff Normalized 2,1,TXRX
domain -1 -Wr 0 Wt 1 W
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Characteristic Polynomials
TX 2 TX FTX () sTX PTX () s (np , nz ) 1 TX TX SS11 , 21 J Ns() Pst () Psr () ETX () s ETX () s (npJ) S11 ,,S21 S31 3 D() sD sDF s () s( )( ) P() s RX RX RX RX RX SS11 , 21 (np , nz ) RX RX Order of N andERX D:() snp +np +np ERX () s Diplexer TX RX J Order of Pt: nzTX+npRX Order of Pr: nzRX+npTX
The diplexer polynomials depend on the TX and RX filters polynomials and on the 3-port junction parameters
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Evaluation of Polynomials
• No characteristic functions analytically known as in the case of filters • Possible solutions: optimization-based evaluation, iterative techniques • The proposed solution: an iterative algorithm which starts with the assignment of reflection zeros (equal to those of isolated filters). The junction is also assigned • All the polynomials are available at the end of the procedure (both DPX and Filters) • Quasi-equiripple response exhibited by the DPX
G. Macchiarella, S. Tamiazzo Novel Approach to the Synthesis of Microwave Diplexers IEEE Trans. Microwave Theory Tech., Vol. MTT-54, n.12, Dec. 2006, pp. 4281-4290 G. Macchiarella and S. Tamiazzo Synthesis of star-junction multiplexers, IEEE Trans. Microwave Theory Tech., vol. 58, no. 12, pp. 3732–3741, Dec. 2010. Slide 155 of 175
Synthesis of the DPX/MPX
• Starting from the characteristic polynomials of the composing filters we apply the well known techniques used for isolated filters • The topology of each filter is arbitrary. It must be only compatible with the number and type of assigned transmission zeros. • The filters are directly connected to the junction • The characterizing parameters are the resonant frequencies and the coupling coefficients of resonators
RX
TX
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Accuracy Issues
Problem: Evaluation of DPX denominator polynomial D(s) Usual technique: Spectral factorization Required task: Search of the complex roots of a
polynomial of order 2(npRX+npTX+npJ) * * 22* * D( sDsN )()( )()( sNspPsPspPsPs )()( )() 00rrnrn ttntn Increasing the filters order, rounding errors in numerical search of the roots prevent the iterative procedure to converge when the order of D(s) is larger than 20-25 Solution: We developed an algorithm for the roots search much more robust than spectral factorization. It is employs digital signal processing techniques (cepstrum) and allows the order of D(s) as large as 40 with an excellent accuracy. Slide 157 of 175
Multiband combiners
• Are constituted by MPX with multiband filters • Conceptually the same algorithm used for single band combiners can be used for the evaluation of the characteristic polynomials • The evaluation of the reflection zeros of isolated multiband filters is required for starting the synthesis (the same return loss in all the bands cannot be imposed) • The converge issue due to round-off errors becomes more critical (overall order of multiband filters is higher than single band) • The synthesis results are less and less meaningful with the increase of the distances between the filters passbands
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Example: dual-band combiner
Specifications Filter 1 Filter 2 Passband1 (MHz) 980 – 1010 1050 – 1070 Passband2 (MHz) 1020 – 1030 1090 – 1100 Min. Return Loss (dB) 25 25 Number of poles 4+3 3+3 N. of Transmission 3 3 zeros (MHz)
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Polynomial response
0
-10-10 -10
-20 S31 -20 S -20 21 -30 -30
-40-30
-40
dB
dB dB
-50-40 -50 -60 -50 -60 -70 -60-70 S11 -80
-80 -90-70 1040 1060 1080 1100 1120 1140 950 960 1000980 Frequency1000 1050 (MHz)1020 11001040 1060 1150 FrequencyFrequency (MHz)(MHz)
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Synthesized circuit: extracted-pole
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Another topology: shunt connected 2-port
A
Filter 1 Filter 2
RX TX
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Masthead Combiners (MHC)
LNA
3 4 Filter rx1 RX path Filter rx2 Cable to ANT port 2 BTS 1 Filter tx
TX path
• Goal: connect the Low Noise Amplifier (LNA) to antenna output for avoiding the NF degradation due to cable losses • Task of MHC: allow the connection of TX and RX of BTS to a single antenna AND the connection of a LNA at the antenna output
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Configuration of a MHC
rx1 rx2 3 4
1 2
tx • Filters rx1 and rx2 have about the same passband (RX signal) • Filter rx1 protects LNA input and suppress TX signal. It exhibits low losses in order not to degrade the NF. • Filter rx2 protects LNA output • Filter tx provides a path for TX signal • Port 1 and 2 are similar to those used in diplexers • All the filters can be multiband • Implemented a single 4-port combiner
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Design of MHC
• The structure is more complex than a duplexer • No algorithm for the evaluation of the MHA polynomials is available • The design is based on the direct evaluation of the filters characteristic polynomials (taking into account the junction configuration) • A smart optimization procedure has been developed for this computation. The unknowns are represented by the filters reflection zeros (complex) and the highest degree coefficients of the transmission polynomials • The goal of the procedure is to get the same reflection zeros of the three isolated filters at ports 1 and 2 (pure imaginary) • The objective function includes also a weighted contribution
from the minimum return loss in the passbands Slide 165 of 175
Example of dual-band MHC
Specifications Filter rx1 Filter rx2 Filter tx Band1 1710 – 1780 1710 – 1785 1805-1880 Band2 1920 – 1980 1920 – 1980 2110-2170 RL (dB) 20 20 20 N. Poles 5+4 4+3 5+3 N. zeros 4 3 3
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Synthesis results
3
rx1
4
J rx2 1 J 2
tx
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MHC Response
0 0
-10 -10 |S | |S | |S42| |S | |S22| |S | 11 11 22 |S22| |S22| -20 11 |S | -20 11 |S21| |S21| -30 -30 |S31| -40 -40
-50
-50 dB dB
-60 -60
-70 -70
-80 -80
-90 -90
-100 -100 1600 1700 1800 1900 2000 2100 2200 2300 2400 1600 1700 1800 1900 2000 2100 2200 2300 2400 Frequency (MHz) Frequency (MHz) Antenna Junction BTS Junction
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Same Band Combiner (SBC)
Tx Rx1 Tx+Rx1 2 To/From MHC 3 RX Band TX Band SBC Rx2 Rx2 1
• Goal: combine a RX band at port 1 with RX+TX bands at port 2 • Task of SBC: Tx channel must be transmitted from 2 to 3 with low losses (0.3-0.5 dB). Rx1 and Rx2 channels are split from port 3 to ports 2 and 1 (losses up to 3.5 dB are tolerated). Isolation of 30 dB is requested between ports 1 and 2
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SBC: Conceptual implementation
Drawback: Overall size
Wilkinson (3 bandpass filters + 3dB Divider W. divider required)
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SBC: New Architecture
R
R
• Directional filter configuration: • All ports ideally matched and Load port uncoupled
• S21 of network R is the transmission from A and R • S11 of network R is the transmission from A and T/R • Property of Network R:
• |S11| and |S21| around -3dB in RX band • |S11| as close as possible to 1 in TX band
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R network: a new class of filters
Characteristic Polynomials:
Pn(s), Fn(s), En(s)
Characteristic Polynomials:
F11=Fn(s)+Pn(s), F22=Fn(s)-Pn(s), P=aP(s) Slide 172 of 175
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Synthesis of SBC
R filters
Hybrids are embedded into the Coupled resonators structure
Equivalent scheme for the hybrids (convenient for embedding)
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Example of frequency response
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Conclusions
• Several advanced combiner architectures for modern transceivers have been discussed • A design approach based on the synthesis of equivalent circuits of the combiners has been presented for – Duplexers/Triplexers (single and multi band) – Mast Head Combiners – Same band combiners
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