DESIGN OF MICROWAVE BAND STOP AND BAND PASS FILTERS BASED ON
BST THIN FILM VARACTOR TECHNOLOGY
Thesis
Submitted to
The School of Engineering of the
UNIVERSITY OF DAYTON
In Partial Fulfillment of the Requirements for
The Degree
Master of Science in Electrical Engineering
by
Jaya Chandra Ramadugu
UNIVERSITY OF DAYTON
Dayton, Ohio
December, 2013
DESIGN OF MICROWAVE BAND STOP AND BAND PASS FILTERS BASED ON
BST THIN FILM VARACTOR TECHNOLOGY
Name: Ramadugu, Jaya Chandra
APPROVED BY:
Guru Subramanyam, Ph.D. Monish Chatterjee, Ph.D. Advisory Committee Chairman Committee Member Department of Electrical and Department of Electrical and Computer Engineering Computer Engineering
Weisong Wang, Ph.D. Committee Member Department of Electrical and Computer Engineering
John G. Weber, Ph.D. Tony E. Saliba, Ph.D. Associate Dean Dean, School of Engineering School of Engineering & Wilke Distinguished Professor
© Copyright by
Jaya Chandra Ramadugu
All rights reserved
2013
ABSTRACT
DESIGN OF MICROWAVE BAND STOP AND BAND PASS FILTERS BASED ON
BST THIN FILM VARACTOR TECHNOLOGY
Name: Ramadugu, Jaya Chandra University of Dayton
Advisor: Dr. Guru Subramanyam
This thesis presents a design of band stop and band pass filters for microwave applications. These filters are based on coplanar waveguide (CPW) transmission lines on a 500 µm thick sapphire substrate. 0.25 µm thin film barium strontium titanate (BST) thin film is used as the tunable dielectric layer. The design of the band stop filter is based on the traditional varactor shunt switch (designed by Dr. Subramanyam and his team) coupled to the ground using an inductive path. Thus, the design symbolizes a single pole standard band stop filter with the potential for frequency tunability. The capacitive overlap makes the device tunable. Several designs of the band stop filter based on the same concept, but different capacitive overlaps, inductive configurations and overall device dimensions were designed and studied. The center frequency for these designs varied from 1 to 5GHz.
The band pass filter is mostly single layered and represents a microstrip based structure although it is CPW fed. It is shunted to the ground with varactors on either side of the device. The idea of having varactors were to achieve tunability. It has 4 resonant
iii traps representing a 4-pole filter. The design, simulation results, experimental results and analyses are presented.
iv
ACKNOWLEDGEMENTS
My special thanks go to Dr. Guru Subramanyam, my advisor who has been very supportive right from the day I planned to transfer to the University of Dayton. He introduced me to microwave lab in my senior year, and that was when I developed my interest in this field. I would like to thank him for giving me this wonderful opportunity and supporting me professionally and financially throughout the program. Without him, I probably would not have been where I stand today in my professional career.
I would like to thank Dustin Brown, who taught me the basics of RF engineering.
He has always been there whenever I had a question. He is extremely knowledgeable and a big asset to our team. Thank you Dustin for every question of mine that you answered and for being there to troubleshoot any issue that I had throughout my time here. I would like to thank my all other team members Dr.Weisong Wang, Dr.Eunsung Shin, Hailing
Yue, Wang Shu, Mark Connor, Kuan-Chang Pan, and Kelvin Freeman. Everybody has been very helpful and supportive whenever needed. It has been a pleasure working with you all. I would like to thank Chenhao Zhang, one of our teammates in the past. The initial design of the band pass filter design that I am discussing in my thesis was initiated by him.
He also let me shadow him to learn to run the ECE 304L that I have been a teaching assistant for since August 2012.
v
I would like to thank the graduate school and all the professors that taught me.
Finally, I would like to thank my other committee members Dr. Monish Chatterjee and Dr.
Weisong Wang for agreeing to be on my committee and for reviewing my work.
vi
TABLE OF CONTENTS
ABSTRACT ...... III
ACKNOWLEDGEMENTS ...... V
TABLE OF CONTENTS ...... VII
LIST OF FIGURES ...... IX
LIST OF TABLES ...... XII
CHAPTER 1 INTRODUCTION ...... 1
1.1 BACKGROUND ...... 1
1.2 SCOPE ...... 5
1.3 OUTLINE ...... 6
CHAPTER 2 LITERATURE REVIEW ...... 7
2.1 FERROELECTRIC MATERIALS ...... 7
2.2 THIN FILM BARIUM STRONTIUM TITANATE (BST) ...... 11
2.3 MICROWAVE DEVICES BASED ON BST ...... 12
2.4 FILTERS ...... 15
CHAPTER 3 BST VARACTOR TECHNOLOGY ...... 27
3.1 DESIGN ...... 27
3.2 DATA ANALYSIS ...... 30
3.3 FABRICATION PROCEDURE ...... 35
vii
3.4 MEASUREMENT SET UP ...... 40
CHAPTER 4 BAND STOP FILTER DESIGN AND ANALYSIS ...... 43
4.1 DESIGN ...... 43
4.2 MEASURED RESULTS AND DATA ANALYSIS ...... 48
CHAPTER 5 BAND PASS FILTER DESIGN AND ANALYSIS ...... 55
5.1 DESIGN ...... 55
5.2 MEASUREMENT RESULTS AND DATA ANALYSIS ...... 62
CHAPTER 6 CONCLUSION AND FUTURE WORK ...... 71
6.1 CONCLUSION ...... 71
6.2 FUTURE WORK ...... 72
BIBLIOGRAPHY ...... 74
viii
LIST OF FIGURES
Figure 1-1 The electromagnetic spectrum [1] ...... 1
Figure 1-2 Evolution of communication technology [2] ...... 2
Figure 1-3 Simplest RF front end of a receiver (top) and a transmitter (bottom) [3] ...... 4
Figure 2-1 Hysteresis loop of ferroelectric devices ...... 8
Figure 2-2 Three dimensional view of a varactor shunt switch [25] ...... 13
Figure 2-3 Simplest schematic of the four different types of filters ...... 17
Figure 2-4 Ideal frequency response of the four different types of filters [3] ...... 18
Figure 2-5 Practically realizable version of the different types of filters [31] ...... 19
Figure 2-6 3D view of on-chip Inductor designed by Xu Yi [27] ...... 21
Figure 2-7 Guru Subramanyam’s microstrip band pass filter design [33] ...... 23
Figure 2-8 Microstrip based dual band pass filter proposed by Cheng-Ying Hsu et. al.
[39] ...... 24
Figure 2-9 Multi-resonator RFID tag; Simulated and measured results of the RFID tag
[41] ...... 25
Figure 2-10 3D view and equivalent lumped element model of Chenhao Zang’s shunt
IDC [42] ...... 26
Figure 2-11 Simulation results from Chenhao Zang’s shunt IDC [42] ...... 26
Figure 3-1 Top view of 5µm by 5µm varactor ...... 28
Figure 3-2 3D view of a 5µm by 5µm varactor ...... 28
Figure 3-3 Top and bottom metal layers ...... 29
ix
Figure 3-4 Equivalent schematic model ...... 30
Figure 3-5 Measured results of 5 µm x 5 µm varactor [39] ...... 32
Figure 3-6 Measured S-parameters of 5by5 varactor from UD BST 92 ...... 33
Figure 3-7 UD BST 92 – Fabricated devices...... 34
Figure 3-8 Extracted parameters of 5 µm x 5 µm varactor measurements without bias tee ...... 34
Figure 3-9 Fabrication process ...... 37
Figure 3-10 Large area PLD system ...... 38
Figure 3-11 Thin film BST deposition process [38]...... 39
Figure 3-12 Measurement Setup ...... 41
Figure 3-13 Calibration substrate ...... 42
Figure 4-1 Equivalent Schematic of a varactor ...... 44
Figure 4-2 Band stop filter response for the schematic circuit in figure 4-1...... 44
Figure 4-3 Spiral strip line ...... 45
Figure 4-4 Transformation from capacitive dominant circuit to a band stop filter ...... 46
Figure 4-5 3-D view of Notch_Filter_2 ...... 47
Figure 4-6 Simulation result of Notch_Filter_2 showing the frequency response of S21.
...... 48
Figure 4-7 Comparison of measured vs. simulated S21 of Notch_Filter_2 ...... 49
Figure 4-8 Tee junction for providing bias voltage ...... 49
Figure 4-9 Notch_Filter_3 design ...... 50
Figure 4-10 Comparison of measured vs simulated S21 of Notch_Filter_3 ...... 51
Figure 4-11 Notch_Filter_4 design ...... 52
Figure 4-12 Comparison of measured vs simulated S21 of Notch_Filter_4 ...... 52
Figure 4-13 Notch_Filter_4_1_1_2_2 design ...... 53
Figure 4-14 Comparison of measured vs simulated S21 of Notch_Filter_4_1_1_2_2 ....53
x
Figure 4-15 Extracting schematic parameter of the measured and the simulated results
...... 54
Figure 5-1 Microstrip based dual band pass filter proposed by Cheng-Ying Hsu et. al.
[34] ...... 56
Figure 5-2 Resonator layout - schematic ...... 56
Figure 5-3 Top view of quad ring filter_1 ...... 58
Figure 5-4 Top and bottom metal layer layout of quad ring filter_1 ...... 58
Figure 5-5 Three dimensional view of quad ring filter_1 ...... 59
Figure 5-6 Simulation results for quad ring filter_1 ...... 60
Figure 5-7 Top view of quad ring filter_2 ...... 61
Figure 5-8 Simulation results for quad ring filter_2 ...... 62
Figure 5-9 quad ring filter_1 – Fabricated device ...... 63
Figure 5-10 Measured results of quad ring filter_1 ...... 64
Figure 5-11 Measured Vs. Simulated S11 of quad ring filter_1 ...... 64
Figure 5-12 Measured Vs. Simulated S21 of quad ring filter_1 ...... 65
Figure 5-13 Measured results of quad ring filter_2 ...... 66
Figure 5-14 Measured Vs. Simulated S11 of quad ring filter_2 ...... 67
Figure 5-15 Measured Vs. Simulated S21 of quad ring filter_2 ...... 67
Figure 5-16 Measured S21 of quad ring filter_2 Vs Bias voltage ...... 68
Figure 5-17 Microstrip schematic of the band pass design ...... 69
Figure 5-18 Schematic tuned to match measured results of Quadringfilter_1 ...... 70
Figure 5-19 Schematic tuned to match measured results of Quadringfilter_2 ...... 70
xi
LIST OF TABLES
Table 2-1 Comparison of dielectric properties of various materials for tunable microwave applications [10] ...... 10
Table 2-2 Common substrate materials used in ferroelectric devices [13] ...... 11
Table 2-3 Comparison among solid state, RF MEMS and varactor shunt switches [17] .15
Table 3-1 A comparison of designed and measured capacitance of varactors [39] ...... 31
Table 3-2 Extracted lumped element parameters [39] ...... 32
Table 3-3 Extracted parameters of 5by5 varactor measurements ...... 35
Table 4-1 Comparison of extracted schematic parameters of measured and simulated results...... 54
xii
CHAPTER 1
INTRODUCTION
1.1 Background
Microwaves are alternating current signals with frequencies between 300 MHz and
300 GHz. Microwaves are also called millimeter waves as their corresponding wavelengths range from 1m to 1 mm. The location of the microwave frequency band in the electromagnetic spectrum is shown in Figure1.1.
Figure 1-1 The electromagnetic spectrum [1]
At these frequencies, the phase of a voltage or current changes significantly over the physical extent of the device. Hence electromagnetic theory is the basis of microwave theory and standard circuit theory is used only as an approximation for simplified analysis.
1
The design of microwave circuits thus becomes challenging and at the same time has wide applications due to the fact that larger bandwidths are available. The advances in microwave technology led to the evolution of communication from wired telephones to wireless smart devices that can be used to talk and simultaneously surf the internet, so called “multitasking,” at speeds that were not imaginable few years ago. Figure 1.2 shows a telephone pole in 1921 in New York on the left side and modern day tower with antennas in central Gothenburg, Sweden on the right side [2].
Figure 1-2 Evolution of communication technology [2]
Some of the most popular wireless systems used today are radio and television, cellular telephone systems, Satellite television, Wireless Local Area Networks (WLANs),
2 paging systems, Global Positioning Satellite (GPS), and Radio Frequency Identification
(RFID) systems.
A simplest RF front end of a receiver and a transmitter is shown in Figure 1.3. A transmitter front end usually has a digital signal processor (DSP), digital to analog converter (DAC), low pass filter (LPF), local oscillator, mixer, amplifier, band pass filter and a transmitter antenna. In the receiver front end, DAC is replaced by an analog to digital converter (ADC) to convert the analog signal back to its digital form [3]. Thus a modulated signal from a data source is sent to the front end of the transmitter, which transmit the RF signal and then the receiver antenna of the receiver front end receives it and recovers the data. Due to the fact that the communication system handles high frequencies, there is a high probability for noise to be introduced into the system at various stages. Hence, to prevent this we use a band pass filter both in the transmitter front end and the receiver front end right next to the antenna. A band pass filter is a type of filter that is designed to block unwanted signals and pass only a selected band of signals.
3
Figure 1-3 Simplest RF front end of a receiver (top) and a transmitter (bottom) [3]
Modern day devices that let you multitask, for example, say speaking to someone over the phone and surfing the internet simultaneously which operate in a different band of frequencies demands for the system to be reconfigurable. If this band pass filter can be designed to shift its center frequency based on the type of signal that comes in, then it would be very inexpensive way to address this challenge. An effort was made to design the band pass filter that is tunable and will be discussed in further chapters.
Noise is a big concern while designing sophisticated wireless systems or any other system that are operable at microwave frequencies. This noise can be avoided by filtering it selectively. A band stop filter addresses this concern. A band stop or a notch filter is a kind of filter that is the reciprocal of band pass filter. It allows the entire band of frequencies to pass through it and only blocks a selective band. One practical situation where a band stop filter is commonly used is while installing a DSL service along with household phone.
4
Any noise or disturbance as the result of using the phone may likely interfere with the DSL internet data and vice-versa. So to avoid this, the telephone company provides a band stop filter. Similarly, in wireless communication a band stop filter is very useful for interference suppression and to improve signal to noise ratio. Another practical scenario where they are used is to improve the quality factor of a band pass filter. Again here, if this device can be tunable, it can be very useful in reconfigurable applications. There has been a lot of research reported in the area of tunable band stop filters [4, 5].
1.2 Scope
This thesis introduces a design of band stop and band pass filters based on thin film BST varactor technology. Over the past few years, the group has been investigating the possibility of using this technology to come up with various tunable devices like antennas, resonators, inductors and filters. My filter designs are an approach to demonstrate specific band stop and band pass filter characteristics and also investigate the possibility of making them tunable. Chapter 4 gives a complete description of design methodology, simulation results, experimental results and their analysis. The basis of the design of the band stop filter is the varactor design. Making the shunt line inductive introduces band stop characteristics in the design and the device can be designed to have a stop band at a specific frequency of interest. Using this concept spirals with various configurations are introduced in the shunt line, and filters with center frequencies between
1 GHz and 5 GHz (simulation) were designed.
The band pass filter design is a continuation of work started by Chenhao Zhang, a member of our team in the past. The motivation for this design is microstrip based dual
5 band pass filter proposed by Cheng-Ying Hsu et. al. [34] and also his thesis on hair pin band pass filter. The design, simulation results, measured results and analysis are presented in detail in chapter 5. The devices showed the characteristics that were expected, however, further analysis is necessary especially in the case of band stop filters to understand their deviation from the expected results. There is more work being done by Hailing Yue to design band stop filters with better quality factor and lower insertion loss in the stop band. The design concept is the same, but she is investigating inductive configurations to have minimal losses. She is also investigating the possibility of using electromagnetic band gap structures (EBG) that trap energy at specific frequencies to design band stop filters.
1.3 Outline
Chapter 1 gives an introduction to microwave and RF engineering and discuss how communication industry has evolved over the years. It also discusses the significance of band pass and band stop filters in wireless communication. Literature review of ferroelectric thin films, thin film barium strontium titanates, some microwave based devices emphasizing band pass and band stop filters are discussed in chapter 2. Chapter 3 gives an introduction to BST varactor technology. It discusses the design, simulation results, fabrication procedure, and measurement setup. Finally, an analysis is presented using measured results. Chapters 4 and 5 introduce the design of band stop and band pass filters with an analysis of their measured and simulation results. Conclusion of the work and future work is presented in chapter 6.
6
CHAPTER 2
LITERATURE REVIEW
2.1 Ferroelectric Materials
A ferroelectric material is a type of material that undergoes spontaneous polarization in the presence of an electric field. The presence of positive and negative ions in them causes a net dipole moment under an applied field. The orientation of this dipole moment can be changed by reversing the electric field. Temperature is another huge factor that influences polarization in ferroelectric materials. These crystals undergo phase changes as the temperature decreases from above the Curie temperature and moves from paraelectric (non-polar) to ferroelectric (polar) nature [6]. The electric field dependence and polarization of these materials are characterized by hysteresis loop and is shown in the Figure 2-1.
7
Figure 2-1 Hysteresis loop of ferroelectric devices
These materials exhibit ferroelectric properties at temperatures below a critical temperature called the Curie temperature and exhibit paraelectric properties above the
Curie temperature. Since there is a phase transition happening at around Curie temperature, these materials show random variation in its thermodynamic properties close to this temperature. They usually show a very high dielectric constant at this temperature.
Compounds can be manufactured to exhibit high dielectric constant at the target temperature of operation [7].
Ferroelectrics is classified mainly into three types based on the thickness of the material deposited. They are bulk ferroelectrics, thick film ferroelectrics, and thin film ferroelectrics [8]. As the name suggests bulk ferroelectrics is the heaviest of the lot, they are usual cylinders or cubes of ferroelectric single crystals or ceramics. Bulk ferroelectrics require high voltage to exhibit tunability and also requires impedance matching to 50 Ω and hence are not very favorable. Ferroelectric thin films in most case involve a
8 complicated deposition process. These are operable at low voltages and relatively cheaper to manufacture and thus are the most attractive in this field. Ferroelectric thick films are obtained by printing cast thick films. These are very cost efficient and thus are very promising. But these are very lossy and also there is a limitation in terms of technology and thus thick films haven’t been used as frequently as the thin films. Our research in the microwave lab currently is based only on thin film ferroelectric materials.
We use pulsed laser deposition (PLD) system for the deposition of thin film ferroelectric materials onto a substrate.
The most attractive part of using ferroelectric materials for microwave applications is the fact that their dielectric constant varies with the application of bias voltage. Some of the advantages of using ferroelectric materials are they are small, fast, light-weight, utilize low power and show a good voltage dependent dielectric tunability. These unique properties of ferroelectrics ensure a wide range of applications in the field of microwave tunable circuits, for example, varactors, tunable resonators, phase shifters, filters, impedance matching networks, power dividers, oscillators and many more. Ferroelectric materials can be lossy, but their lossy nature can be avoided by using them in the paraelectric phase. Also, appropriate doping can reduce the loss tangent can thus make the material less lossy [9]. Some of the examples of ferroelectric materials whose dielectric constant can be tuned by the application of an electric field are SrTiO3, (Ba,Sr)TiO3,
(Pb,Sr)TiO3, (Pb,Ca)TiO3, Ba(Ti,Sn)O3 and KTaO3.Thus they have a huge potential for microwave applications. A comparison of dielectric properties of various materials for tunable microwave applications at 25oC is shown in Table 2-1 [10]. TCE is temperature dependence of dielectric permittivity and is given by
TCE = ε−1∂ε/∂T×106
9
Table 2-1 Comparison of dielectric properties of various materials for tunable microwave applications [10] o Material εr @ 25 C TCE(PPM/k) Q = 1/tanδ @ Loss measured o o @ 25 C 25 C @ f (GHz) Ba-Zn-Ta-O 30 -18…-26 12600 10
Zr-Sn-Ti-O 37 -7…-45 5800 9
Ba-Nd-Ti-O 88 -8…-46 1100 5
TiO2 100 -900 14500 3
SrTiO3 300 -2000 1000 10
Ba0.6Sr0.4TiO3 4000 -40000 50 10
BST with 60% barium and 40% strontium shows the highest dielectric constant as well as loss tangent. It has been observed in most cases that higher the dielectric constant higher is the loss tangent, tunability and also its variation with temperature [11, 12]. These dielectric properties of materials also depend on the thickness of the material deposited, crystal structure of the material, fabrication process used and the nature of the substrate material. A list of some of the common substrates in ferroelectric devices for microwave applications are shown in Table 2-2 [13]. Their permittivity, loss tangent, crystal structure and CTE are tabulated. J. Pérez de la Cruz et. al. discussed the thickness effect on the dielectric, ferroelectric, and piezoelectric properties of ferroelectric lead zirconate titanate thin films in Journal of Applied Physics108, 114106 (2010) [14]. The results showed that as the thickness of the film increased from 150nm to 700nm the dielectric constant increased from 850 to 1300. The change in the loss tangent was minimal.
10
Table 2-2 Common substrate materials used in ferroelectric devices [13] Substrate Permittivity Loss tangent Structure CTE*106
o -4 o o LaAlO3 24 @ 295 K 3.10 @ 300 K Single crystal 10 @ 25 C
(18-35GHz)
MgO 9.8 9.10-3 @ 10 GHz Single crystal 8 @ 100oC
Fused Silica 3.826 @ 1MHz 1.5.10-5 @ 1MHz Amorphous 0.55 @ (20-
-4 o (SiO2) 3.82 @ 24MHz 3.3.10 @ 24GHz 320) C
Sapphire 11.5/9.3 @ 8.6*10-4/3*10-4 @ Single Crystal 5.3 @ 25oC
1MHz 1MHz
Alumina 9.9 @ 1MHz 10-4 @ 10GHz Poly- 8.1 @ (25-600)
crystalline oC
Silicon 11.7 0.01 Single crystal 2.6 @ 25oC
2.2 Thin Film Barium Strontium Titanate (BST)
Thin film barium strontium titanate (BST) is becoming increasingly popular for frequency and phase agile applications in microwave frequencies because of its tunable nature with an application of voltage and low power utilization [15-21]. BST is a solid solution ferroelectric that is continuous. The chemical composition of Barium strontium titanate is Ba(1-x)SrxTiO3. It is a mixture of barium titanate (BaTiO3) and strontium titanate
(SrTiO3). The concentration of barium and strontium determines the properties of BST and thus it can accommodate various applications [22]. It was found that 40-60% concentration of barium made it tunable at room temperature. Miyazaki et al. studied the dielectric properties of Ba(1-x)SrxTiO3 thin films prepared by RF magnetron sputtering in terms of film composition and substrate temperature [23]. The dielectric constant of both BaTiO3 and
11
SrTiO3 both increased with substrate temperature, however, SrTiO3 showed higher response for all the temperatures below 500oC where the dielectric constant of both merges. It was observed that the dielectric constant was strongly dependent on Strontium content. For a strontium content of around 0.5-0.6, peak dielectric was observed when the powder preparation temperature was 900oC, and the film was deposited at 600oC. This paper concluded that optimizing film composition; target preparation and sputtering can result in a film with high dielectric constant and temperature coefficient.
BST possesses unique properties like high dielectric constant, small loss tangent, high power handling capacity, ability to handle both negative and positive voltages and its dielectric strength are tunable with application of voltage. Its dielectric constant is maximum at 0 bias and keeps decreasing as applied bias voltage increases. Our team headed by Dr.Guru Subramanyam had shown in the past that a tunablity of 4:1 was achieved when a 0.25µm thin film BST sandwiched between two electrode plates on sapphire substrates thus making a varactor. It was observed that the dielectric constant of BST tuned from 990 at 0V to 250 at 10V. Some of its applications include but are not limited to filters, phase shifters, oscillators, antennas and electromagnetic band gap structures.
2.3 Microwave devices based on BST
Our group has published a lot of papers in the past on thin film BST based microwave devices. A pulsed laser deposition (PLD) system is used to deposit thin film
BST on a substrate [24]. It can be used to deposit thin film BST on wafers up to 4 inches in diameter. The two popular configurations that are widely used while designing
12 microwave circuits are microstrip and coplanar waveguides (CPW). Both these configurations have their respective advantages and disadvantages.
Dr. Guru Subramanyam and his team have been investigating the possibility of using thin film BST to come up with voltage tunable devices and have successfully demonstrated their potential. Dr. Guru Subramanyam holds the patent for BST based shunt varactors. The design is a CPW transmission line compatible MIM (metal-insulator- metal) structure on a sapphire or silicon substrate. The design has capacitive overlap that is shunted to ground. A paper discussing the linearity and temperature dependence of large area processed high-Q BST thin film varactors was published by our group [18]. It was found that varactors were highly tunable and have low losses. Figure 2-2 shows a three dimensional view of a varactor shunt switch using high resistivity silicon as the substrate [25]. It was observed that these varactors showed a better performance on sapphire substrates. A tunability of 4:1 was observed as the voltage was changed from
0V to 10V. They showedgood quality factor and thus the performance was very good.
Figure 2-2 Three dimensional view of a varactor shunt switch [25]
13
An analysis of phase shift produced by these varactors was published in 2010
[21]. In 2011, a paper was published by our team that discusses the effect of having multiple capacitive overlaps shunted to the ground in parallel from the same signal line
[26]. One of our team member’s Hailing Yue’s thesis titled “Design and optimization of barium strontium titanate ferroelectric varactors” discusses the results and analysis of these ferroelectric varactors integrated on silicon substrates [27].
Coplanar waveguide varactors with bottom metal trenches in silicon was published in 2011 by Dustin Brown et al. [28]. They showed lower capacitance than original ones.
The electromagnetic band gap tunability of coplanar waveguides loaded periodically by ferroelectric varactors was published by D. Kuylenstiera et. al. in 2003 [15]. It was shown that various tunable electromagnetic band gap performances could be achieved by controlling the self-resonance frequency of the LC circuit. A comparison of the device characteristics and performance parameters of BST varactor based shunt switch over RF
MEMS and solid-state counterparts are shown in Table 2-3 [17]. Looking at the table BST varactor based switches are clearly better than either solid state switches or RF MEMS capacitive shunt switches.
14
Table 2-3 Comparison among solid state, RF MEMS and varactor shunt switches [17]
Sekhar Kanagala et. al. reported a paper on electrical modeling of ferroelectric tunable microwave components and circuits [29]. Simple resonators, bandpass filters,
MEMS capacitive shunt switches and phase shifters were modeled in this work. The experimental and simulation results matched closely and was a good proof of concept.
Hai Jiang et. al. designed reconfigurable CPW (coplanar waveguide) square-ring slot antenna using thin film varactor technology [30]. The concept was to shift the resonant frequency of the antenna by biasing the varactors with different DC voltages.
2.4 Filters
Filters are devices built using energy storage elements like capacitors and inductors. The main purpose of the filter is to allow frequencies in the band of interest to pass through it and block the rest. Thus filters are the simplest signal processing circuits
15 that one can build. A capacitor allows high frequencies to pass through it while it blocks the relatively lower frequencies based on the value of its capacitance. On the other hand, an inductor allows low frequencies to pass through it while blocking the relatively higher frequencies again based on its inductance value. Thus capacitors and inductors can be modelled in various configuration to yield basically four types of filters. They are
a. Low Pass Filter (LPF) – Allows low frequencies to pass through it while it blocks
relatively higher frequencies. The frequency at which this transition happens is
called the cut off frequency
b. High Pass Filter (HPF) – Allows high frequencies to pass through it while it
blocks the relatively lower frequencies determined by the cut off frequency
c. Band Pass Filter (BPF) – This type of filter allow a specific band of frequencies
to pass through while it rejects the rest. A band pass filter thus has two
transition points – lower cut off frequency and upper cutoff frequency. All the
frequencies in between the lower and upper cutoff frequencies are allowed to
pass through it while those below the lower and above the upper cut offs are
blocked
d. Band Stop Filter (BSF) – A BSF is exactly opposite to BPF. It rejects a particular
band of frequencies and allows the rest to pass through it.
A schematic of these four different types of filters is shown in Figure 2-4. The repose of a filter is described mathematically using a transfer function which is the ratio of the output impedance to the input impedance. If X(s) is the input impedance and Y(s) is the output impedance then the transfer function is given by
T(s) = Y(s)/X(s)
16
The design of a filter starts here. A RF filter is usually built by using distributed elements but could also use lumped elements or a combination. Transmission lines are configured to build capacitors and inductors such that they imitate the schematic shown in Figure 2-3 to come up with corresponding filters. The schematics shown in the figure are the simplest ones.
Figure 2-3 Simplest schematic of the four different types of filters
These filters shown in the figure are called single pole or first order filters. The order of a filter is defined by the number of poles it has. The number of RC circuits determines the number of poles. Figure 2-8 shows the ideal frequency response of the four different types of filters. The figure gives a graphical presentation of the exact
17 mathematical calculations. Observe that these filters have perfectly vertical edges but in real life this cannot be true. A more practically realizable version of the frequency response has a curve at the edges instead of the perfect edges and rolls off as shown in Figure 2-
4. This is the best approximation of these filters in real life.
Figure 2-4 Ideal frequency response of the four different types of filters [3]
18
Figure 2-5 Practically realizable version of the different types of filters [31]
The cut off frequency of a filter is defined as the frequency where the gain of a filter is 3dB less than the maximum gain. It is clearly shown in Figure 2-5. The lower cut off
frequency is represented by fc1 and, the higher by fc2 in the figure. Out of all these filters band pass and band stop filters are the ones that are most widely used in microwave and
RF applications. There has been a large number of efforts to come up with filters with high frequency selectivity and good quality factor (Q) using various techniques. Quality factor of a band pass or band stop filter is defined as the ratio of energy stored to the energy dissipated. In a bandpass filter, the Q is obtained by the ratio of the center frequency to the difference between the upper and lower cut off frequencies. If Q is the quality factor and fc the center frequency, then quality factor is by
19
푓푐 푄 = 푓푐2−푓푐1
Thus higher the quality factor, better is the frequency selectivity. The design of a filter should also ensure that the insertion loss is as low as possible in the pass band. The rapid growth in the communication industry is demanding filters with high quality factor and frequency tunability. This helps in making efficient use of the frequency spectrum.
This also improves the efficiency of reconfigurable front ends in which these may be used.
In most cases, microwave filters are designed using transmission lines but can also include lumped elements. These transmission lines can be designed innovatively to introduce capacitive and inductive regions in them. Varactors and inter-digitated capacitors (IDC) are some of the examples of tunable capacitors. Inductance is associated with every transmission line and directly correlates to the length of the transmission line and also to the number of turns in the line. One of our group members in the past Xu Yi modelled and analyzed CPW based multi-layer on-chip inductors [32]. She attempted to study the correlation between the inductance and length and width of the conductor. She was successful in demonstrating that an increase in the length of the conductor increases the inductance and an increase in the width of the conductor decreases the inductance.
Figure 2-6 below shows the 3D view of an on-chip inductor designed by Xu Yi. It consists of two CPW based metal (gold) layers with BST of 0.25µm sandwiched between the layers on sapphire as a substrate. It is a single port device.
20
Figure 2-6 3D view of on-chip Inductor designed by Xu Yi [27]
Y. Aoki et. al. developed an electrical model for multi-layer inductors for wide frequency range [33]. A low loss, high Q spiral inductor, was designed and fabricated on silicon substrates by wafer level packaging. Q-factor was evaluated for inductance values of 2-5nH and showed promising results. For inductors ranging from 2.7 to 8.2nH the electrical model matched closely with the results obtained from measurement of the fabricated.
T. Hong et. al. designed an electrical model for low temperature cofired-ceramic
(LTCC) embedded inductors [34]. The model includes a core circuit, a new T-model circuit and several resonators. The electrical model also included transmission line effects and coil coupling between lines. The S-parameters obtained using this electrical model was matched with that of the design and the inductance value was extracted.
21
2.4.1 Band Pass Filter
This section presents a literature review on Band pass filters (BPF) for microwave applications. This review was very useful to come up with an optimized BPF design that is being presented in this thesis. Two of our group member in the past demonstrated designs of band pass filters. Chenhao Zang designed a miniaturized chebychev band pass filter for X-band [35]. The center frequency was at 10GHz. Two coupled hairpin resonators were fed using a coplanar waveguide (CPW) structure. This design gave 3 ripples (3rd order) and 1dB insertion loss in pass-band with a quality factor of 19.5. Sreekanth
Vemulapalli made an effort to study and characterize the properties of BST thin films and then design a tunable band pass filter using their unique properties like tunable permittivity and low loss tangent [36]. Different EM structures of Inter Digital Capacitors (IDCs) are designed, tested, characterized, and an effort is made to optimize the design for high quality factor (Q). A tunable band pass filter was designed, fabricated and tested for
1.72GHz center frequency. This was one of the foremost efforts made to design a band pass filter with Coplanar Wave Guide (CPW) transmission line components and BST varactor devices.
Guru Subramanyam et. al. reported K-band tunable band pass filters using
YBa2Cu3O7-δ (YBCO) superconducting thin-film and SrTiO3 (STO) tunable dielectric thin- film on LaAlO3 (LAO) dielectric substrates [37]. The design included a two-pole filter and had a center frequency at 19 GHz with 4% bandwidth. The dielectric constant was tuned by applying nonlinear electric-field which shifted the center frequency by 2.3GHz at 400-
V bipolar dc bias, and 30 K, with minimal degradation in the insertion loss of the filter. The design of the filter is shown in Figure 2-7.
22
Figure 2-7 Guru Subramanyam’s microstrip band pass filter design [33]
A novel design of microwave four-pole band pass filter with embedded planar BST varactors was presented by Buslov et. al. for ka--band[38]. It was observed that the bandwidth was 350 MHz (1.2%) at 1 dB level and the range of tuning was 570MHz (1.9%) and the insertion losses were lower than 5dB. The reference was a good concept, but it was targeted toward lowers frequencies. Cheng-Ying Hsu proposed a compact microstrip dual-band bandpass filter (BPF) using the embedded open-loop ring resonator operating at 2.45/5.7 GHz [39]. The device structure is shown in Figure 2-8. The breakthrough was that the filter gave two passbands without using the external dual-band impedance transformer in the input/output feeding line. Also, it was possible to control the ratio of the
first passband to the second passband. This design is the motivation for the band pass filter design that will be discussed in this thesis.
23
Figure 2-8 Microstrip based dual band pass filter proposed by Cheng-Ying Hsu et. al. [39]
Also, filters are classified based on the application in the industry as digital filters and analog filter. A digital filter can be programmed and is used mostly in digital signal processing applications whereas an analog filter is used for building analog circuits especially in communication systems.
2.4.1 Band Stop Filter
Y. H. Chun et. al. presented an investigation of a tunable band stop filter (BSF) using slotted ground structure in 2008 [40]. MgO was used as the substrate with barium strontium titanate (BST) as a dielectric. A three pole open loop slot resonators were used to achieve band stop response. A single layer varactor design was used to achieve tunability. The center frequency of the device tunes from 4.5 to 5.5 GHz with 20dB rejection at the stop band while the insertion loss at the pass band was less than 1.5 dB.
Stevan Preradovic et. al. presented a multi-resonator based passive printable chipless RFID system [41]. The chipless tag uses the amplitude and phase of the spectral
24 signature of a multi-resonator circuit and provides 1:1 correspondence of data bits. The tag comprises of a microstrip spiral multi-resonator and cross-polarized transmitting and receiving microstrip ultra-wideband disc loaded monopole antennas. It is built on Taconic materialTLX-0 (εr = 2.45, h = 0.787mm, tanδ = 0.0019). This is shown in Figure 2-9. The figure shows that there are 6 resonant points (notches). Each resonator produces a resonant frequency. Shorting a resonator gets rid of the resonant frequency and this principle can be used to come up with unique, cost effective designs for the tag. However one may be limited by the number of devices that can be produced. Increasing the number of resonators can increase this number. A 35-bit chipless RFID was also demonstrated in this paper.
Figure 2-9 Multi-resonator RFID tag; Simulated and measured results of the RFID tag [41]
One of our team members in the past, Chenhao Zang designed a shunt interdigitated capacitor using BST thin film on a sapphire substrate [42]. A three dimensional view and an equivalent lumped element schematic model are shown in Figure
2-10. The device was built on a sapphire substrate with 0.25µm BST sandwiched between the two metal layers (gold). Various designs with different dimensions were designed, and their resonant frequencies ranged from 11.53GHz to 17.51GHz. The simulation results
25 are shown in Figure 2-11. The results present the behavior of the band stop filter. This design was the motivation for the band stop filter that will be discussed in this thesis.
Figure 2-10 3D view and equivalent lumped element model of Chenhao Zang’s shunt IDC [42]
Figure 2-11 Simulation results from Chenhao Zang’s shunt IDC [42]
26
CHAPTER 3
BST VARACTOR TECHNOLOGY
This chapter gives an overview of BST varactor technology [9]. This has been the basis for studying and understanding the BST properties for a long time in our lab. Dr.
Guru Subramanyam, my advisor, holds the patent for this technology. The design, simulation results and measurement results are discussed in the sections below.
3.1 Design
A varactor is simply a variable capacitor. Barium strontium titanate (BST), whose dielectric constant can be varied by application of DC bias is used as the dielectric for this device. Figure 3-1 shows the top view of a 5µm x 5µm varactor. Figure 3-2 shows a three dimensional view of a 5µm x 5µm varactor. This device is formed by sandwiching a 0.25µm thick BST in between two metal layers that form the conductors. Ti+Pt+Au stack is used as the conductor and its total thickness is 1µm. The overall dimensions of the device can vary form 500µm by 450µm to 300µm by 450µm. The devices that are fabricated on the current mask have an overall dimension of 300µm by 450µm. The device is coplanar waveguide based with a ground-signal-ground configuration. The width of the signal line is 50µm, and the separation between the signal line and ground line is 50µm. This structure provides a 50 ohm impedance on the input and the output sides. This is necessary to ensure impedance matching when probed because probes have a 50 ohm
27 impedance. Figure 3-3 shows the top metal layer and bottom metal layer of this device.
Sapphire or high resistivity silicon is used a substrate. It was found that the device gave best results with sapphire as a substrate.
Figure 3-1 Top view of 5µm by 5µm varactor
Figure 3-2 3D view of a 5µm by 5µm varactor
28
Figure 3-3 Top and bottom metal layers
The overlap area at the center (Figure 3-1) formed by the top metal layer overlapping with the bottom metal layer creates a capacitive region that is shunted to the ground using the bottom metal layer. This capacitance is thus called shunt capacitance.
In this case, the overlap area is 5µm by 5µm. In general if A is the area formed by the top metal layer overlapping with the bottom metal layer, d is the thickness of BST or the separation between the two metal layers, εr is the dielectric constant of BST and ε0 is the permittivity of free space then the capacitance, C of this overlap region is given by
εrε0퐴 퐶 = 푑
Clearly capacitance increases with an increase in the dielectric constant or the area of overlap but decreases with the increase in the thickness of BST. Figure 6-5 shows an equivalent lumped element model of the device. In this figure, it can be clearly seen that a CPW substrate is defined with two 150µm long CPW transmission lines. These transmission line is 50µm wide with a 50µm separation between the signal and ground lines. This resembles the ground-signal-ground configuration in the EM structure. A port is attached on either side of the transmission line. The equivalent schematic is inserted in between the two transmission lines and a shunt capacitor with a capacitance of ‘C’ as
29 discussed before is used here. A resistor ‘RC1’ is used in parallel combination with the capacitor in order to account for any resistance offered by the overlap region. The shunt line in the electromagnetic (Em) structure introduces inductance and a small resistance and thus ‘L1’ and ‘R1’ are introduced in the schematic. This model is used to extract the capacitance value and other lumped element values.
Figure 3-4 Equivalent schematic model
3.2 Data analysis
In this section work from the past to illustrate, the characteristics of a varactor is presented and then an analysis of the results of the 5 µm x 5 µm varactors on the current wafer (UDBST92) is provided. Varactor characteristics have been extensively studied by our team members in the past. Hailing Yue demonstrated that it was possible to design varactors that would meet the desired capacitance by varying the overlap areas in her thesis. Her designed and measured results were pretty close and differed only marginally.
Table 3-1 shows a comparison of capacitance desired, and the capacitance obtained from
30 measured results. Increasing the area of overlap increased the capacitance. Based on all these observations in the past a 5 µm x 5 µm varactor typically gives a capacitance of around 0.8pF, and it is considered to be the standard. Every wafer that the team fabricates includes several 5 µm x 5 µm varactors to validate the quality of the fabrication process and also the dielectric material. Measured results of 5 µm x 5 µm varactor from Hailing
Yue’s thesis is shown in Figure 3-5. Table 3-2 shows the lumped element parameter extracted for these measurements. A tunability of 1.7:1 was observed by her at 5V dc bias.
Table 3-1 A comparison of designed and measured capacitance of varactors [39]
31
Figure 3-5 Measured results of 5 µm x 5 µm varactor [39]
Table 3-2 Extracted lumped element parameters [39]
Figure 3-6 shows a plot of the measured S-parameters of the 5 µm x 5 µm varactor on UD BST 92. This is the same wafer on which the band stop and band pass filters that are being discussed in this thesis are fabricated. The S-parameter for the 0 bias case
32 pointed out in the figure mean that the measurements are done without using the bias tee whereas the one that says 0V is measured with the bias tee in series with port 1 of HP’s vector network analyzer 8720B. Even though separate calibration are done for both the cases there is a deviation, and this has to be identified in future work. Figure 3-7 shows a picture of the fabricated wafer. The devices discussed in this thesis are marked in the figure. In those, the relatively bigger ones are the band pass filter and the ones that can be barely seen are the band stop filters.
Figure 3-6 Measured S-parameters of 5by5 varactor from UD BST 92
33
Figure 3-7 UD BST 92 – Fabricated devices
Figure 3-8 Extracted parameters of 5 µm x 5 µm varactor measurements without bias tee
34
Table 3-3 Extracted parameters of 5by5 varactor measurements Bias voltage RC1 (V) C (pF) (Ω) L1 (nH) R1(Ω) Without bias tee 0.395 1920 0.002 1 0 0.21 1920 0.037 1 3 0.185 1920 0.037 1 6 0.175 1920 0.037 1 9 0.155 1920 0.037 1 12 0.145 1920 0.037 1 15 0.135 1920 0.037 1
From Table 3-3, it can be clearly inferred that the device is tunable but neither the tunability nor the capacitance is as good as the values showed by Hailing in her thesis last year. This can lead us to some assumptions that either the quality of the film deposited or the thickness of the film deposited is greater than 0.25µm. This means that the band stop filter which is designed based on this technology may show interesting results. It may not show expected trends. Since varactors are tunable the band stop filters should not have an issue with tunability, as well.
3.3 Fabrication Procedure
The fabrication process starts off with photolithography on the wafer. In this case, it is a sapphire wafer. This wafer is coated with a layer of positive photoresist whose thickness is determined by the thickness of the bottom metal layer which is 1µm. The fabrication process is diagrammatically shown in Figure3-9. The figure shows a three dimensional view of the device at various stages in the fabrication process. Then this layer is selectively exposed to light. The region that is protected represents the shape of the bottom metal layer. The exposed region hardens relatively and is difficult to etch whereas the region that is protected can be easily etched in a developer solution. After the
35 photoresist is developed, the metal stack is deposited in an electron beam (e-beam) evaporation system. The thickness of the metal stack is roughly 1µm. Then the remaining photoresist with metal layers on top of it is removed by lift off process. Once all this is done it is the time for BST deposition.
Large area pulsed laser deposition (PLD) system is used for BST deposition. This technology helps in fabrication of high quality thin films. PLD is a physical vapor deposition
(PVD) process. Laser in this case excites the target material to go into the vapor phase which is deposited on the target substrate.
36
Figure 3-9 Fabrication process
37
Figure 3-10 Large area PLD system
The Neocera Pioneer 180 large area PLD system available in our lab is shown in
Figure 3-10. It consists of an excimer laser (in this case ArF) that generates the beam, focusing lens to focus the laser beam onto the target and a vacuum chamber that houses the target and the substrate that needs to be coated. The substrate is placed right on top of the target and perpendicular to it as shown Figure 3-11. The target is continuously rotating so that BST in the target is uniformly ablated. A turbo pump is used to create vacuum in the chamber (in the range of 10-7 Torr) to make sure there are no contaminants inside the chamber. Also, the substrate is placed on a heater to maintain it at a specific temperature 900 C to achieve uniform deposition. The chamber were then filled with oxygen and maintained at 75mT pressure. Oxygen is essential for the growth of the film.
Laser pulses with energy range from 225 mJ at 30 Hz repetition rate are produced by the
38 device. The laser passes through the focusing lens and hits the target. As the laser beam bombards the BST material with such high energy it converts BST into a vapor phase plume. It then mixes with the oxygen present in the chamber and then deposited on to the wafer. The focusing lens scans the beam to cover the entire diameter of the target. Also, the distance between the target and the substrate can be adjusted based on the application. Since the laser pulses hit the target at regular intervals of time and approximately the same amount of BST vaporizes each time, one can easily control the thickness of the deposition calibrating the pulses needed to achieve certain thickness.
Figure 3-11 Thin film BST deposition process [38]
39
Typically 25000 pulses give a thickness of 0.25µm from our observation in the lab.
This system is suitable for wafer 3 to 4 inches in diameter. Once this deposition is done the wafer is left alone to anneal for 2 hours at 760C in oxygen. During this time, the pressure in the chamber is 600 Torr filled with oxygen. Some typical parameter that influence this process is – energy of the laser, diameter of the wafer, oxygen pressure.
After it is cooled down it is coated with a photoresist layer gain and patterned to deposit the top metal layer as shown in Figure 3-9. The protected region is shown in Figure
3-9 that imitates the shape of the top metal layer is removed by lift off, and then electroplating process is used to deposit a 1µm gold layer again. The remaining photoresist is removed using etching and the device takes the shape shown in Figure 3-
9. Finally a passivation layer (silicon nitride, Si3N4) is applied on the device to prevent it from external damages that may do to the device by probes. This completes the fabrication process. Same technique is used for the band pass and band stop filters that are be discussed in this thesis.
3.4 Measurement set up
The measurement setup consists of a Vector network analyzer (VNA), microscopic probe station, vacuum pump,computer to record the measurements and DC meter to supply bias voltage when required. Figure 3-12 clearly shows the setup. The VNA being used here is an HP 8720B. It is a two port device that measures scattering parameters (S- parameters) of a device under test for a range of frequencies from 130MHz to 20GHz. It provides input and output impedance of 50 ohm which matches well with our coplanar waveguide design. A voltage bias tee junction can be used along in conjunction with port
40
1 or 2 to apply DC bias if necessary. A capacitor in the bias tee prevents the bias voltage form going into the VNA and damaging it.
The microscope that is being used here provides a magnification of 3X. The vacuum pump is used hold the device together in a constant position. Two sets of probes can be used – one with50µ spacing between the signal and ground lines and the other with a 250µm spacing between there lines. The microscope has a camera attached to it to grab still as well as motion pictures. The VNA is connected to a computer using a GBIP port to acquire. A software program is written to save the data in excel sheets.
Figure 3-12 Measurement Setup
The network analyzer needs to be calibrated for this set up every time it is turned on to measure a device. The network analyzer has to be calibrated in reflection mode, transmission mode and isolation mode. Figure 3-13 shows a calibration substrate and
41 various configurations in which the network analyzer has to be calibrated. In reflection mode, the probes from port 1 and port 2 are isolated and calibrated for open, short and 50 ohm load. Then in transmission mode it is calibrated for effective transmission of the RF signal for port 1 to port 2 and vice-versa. The network analyzer is set up to omit any isolation. This completes the calibration process.
Figure 3-13 Calibration substrate
Then a device to be tested is probed, and its S-parameters are measured and captured on the test computer. This data is then imported onto the software simulation tool (AWR microwave office in this case) for analysis.
42
CHAPTER 4
BAND STOP FILTER DESIGN AND ANALYSIS
This section discusses the design specification, simulation, measurement results and provides an analysis of the band stop filter. As mentioned earlier on in the literature review section, the design of the band stop filter is based on the coplanar waveguide BST shunt varactor.
4.1 Design
The motivation for this design was the equivalent schematic of coplanar waveguide
BST varactor. The schematic is shown below in Figure 4-1. The configuration in which the lumped elements are connected in this schematic is similar to that of a single pole band stop filter. If the inductance and capacitance are sufficiently high then it could create a resonant point in our frequency of interest which is typically 1-10GHz.
43
Figure 4-1 Equivalent Schematic of a varactor
Figure 4-2 Band stop filter response for the schematic circuit in figure 4-1.
44
Figure 4-2 shows a band stop filter response of the schematic in Figure 4-1 for a capacitance of 2pF, inductance of 0.8nH and a shunt resistance of 2kΩ . Notice that the series resistance in the capacitor is negligible and is generally the case. The capacitance of a capacitor is directly proportional to the surface area of the metal in contact with the dielectric. Thus increasing this area increases the capacitance. Inductance of a shunt line can be increased by introducing inductive structures in it. Every conductor has an inductance associated with it, but is very small. Traditionally an inductor is built by winding copper wire around a ferrite core. An inductor resists any change in the current flowing through it. When a current flows through an inductor, it stores the energy temporarily in the form of a magnetic field in the coil. The energy thus stored increases as the number of turns in the coil increases which basically means the inductance increases. But the traditional design cannot be applied here while designing strip line based inductors for microwave application. Thus introducing innovative design in the metal layer that aids in storing magnetic energy can create good inductor. One of our team members in the past,
Xu Yi, used spiral structures (Figure 4-3) of various lengths, width and turns to characterize the value inductance with respect to these parameters. Similar design approach is used in this work to create an inductor in the shunt line of the varactor.
Figure 4-3 Spiral strip line
45
Figure 4-4 Transformation from capacitive dominant circuit to a band stop filter
A design of the band stop filter is shown in Figure 4-4. The figure clearly depicts the similarities and differences in the design of band pass filter from the varactor. This device is CPW based with a 50 µm separation between the ground and signal lines. The overall dimensions of the band pass filter shown in the figure are 600 µm by 650 µm with a varactor overlap of 10 µm by 30 µm. That should give a pretty big capacitance compared to a 5 µm by 5 µm overlap. The ground planes are pushed in at the center to accommodate spirals in the shunt line going to the ground. The width of these lines is 10 µm with a 25
µm separation between them. A three dimensional view of the device is shown in Figure
4-5. It clearly shows the overall structure of the device.
46
Figure 4-5 3-D view of Notch_Filter_2
This device is built and simulated in AWR Microwave Office. The top and bottom metal layers are approximated as perfect conductors for simplifying the simulation. The work in the past confirms that using perfect conductor for gold is a close approximation for simulating the em structure of the device in AWR. The dielectric constant of BST was taken to be 500 for simulation. The simulation result is shown below in Figure 4-6. The figure shows a plot of S21 in decibels against frequency from 0.1 GHz to 15 GHz. The center frequency is at 4.3 GHz with an insertion loss of 37.74 dB. Several band stop filters with different varactor overlaps; different inductive configuration were designed and simulated in AWR. An analysis of measured versus simulation results along with the design specifications is presented in the next section that says measurement results and data analysis.
47
Figure 4-6 Simulation result of Notch_Filter_2 showing the frequency response of S21.
4.2 Measured results and data analysis
This section compares the measured data with the simulated data and gives a comprehensive analysis of the device performance. The measured results for the band stop filters are obtained from UD BST 92. This work discusses four variations of the band stop filters. They are namely Notch_Filter_2, Notch_Filter_3, Notch_Filter_4 and
Notch_Filter_4_1_1_2_2. A comparison of measured results with those of the simulated ones for Notch_Filter_2 is shown in Figure 4-7. The center frequency of the simulated result is at 4.3 GHz whereas that of the measured result without bias tee is at 7.7 GHz.
The explanation for the deviation of the measured results from the simulation results is the same as that discussed in the data analysis section of the BST varactor technology. The graph clearly shows that the device is tunable when a bias is applied. Again, here it can be observed that when no bias is applied, the measurements recorded without the bias
48 tee and with the bias tee the results are slightly different, even though calibration was performed for both the cases. This difference has to be further investigated. Figure 4-8 shows the bias tee used by our group in the lab to apply a bias voltage to the devices.
The capacitor in the bias tee prevents any applied bias from entering port 1 of the VNA.
Figure 4-7 Comparison of measured vs. simulated S21 of Notch_Filter_2
Figure 4-8 Tee junction for providing bias voltage
49
A design of the Notch_Filter_3 is shown in Figure 4-9. The varactor overlap is same as in the case of Notch_Filter_2, but the lengths of the inductive lines and their configuration are different. The design provides a higher inductance comparatively. A comparison of the measured results with those of the simulated ones is shown in Figure
4-10. Similar trend as in the case of Notch_Filter_2 is observed.
Figure 4-9 Notch_Filter_3 design
50
Figure 4-10 Comparison of measured vs simulated S21 of Notch_Filter_3
Figures 4-11 and 4-13 show the designs of Notch_Filter_4 and
Notch_Filter_4_1_1_2_2 respectively. Notch_Filter_4 has longer inductive path and is the only difference form Notch_Filter_3. In Notch_Filter_4_1_1_2_2, the area of the varactor overlap is increased to 30µm x20µm. Thus it should have a higher capacitance. Also, the edges in the inductive line are more curved to improve the inductive coupling. Figures 4-
12 and 4-14 shows a comparison of measured results against the simulated ones. Similar trend as in the previous 2 cases is observed, however, the center frequencies of the stop band regions is lower than the previous two cases due the increase in the capacitive and inductive effects. The simulated and measured results of all these four filters are compared with the schematic model of the design and lumped element values are extracted. They are tabulated in Tabel 4-1.
51
Figure 4-11 Notch_Filter_4 design
Figure 4-12 Comparison of measured vs simulated S21 of Notch_Filter_4
52
Figure 4-13 Notch_Filter_4_1_1_2_2 design
Figure 4-14 Comparison of measured vs simulated S21 of Notch_Filter_4_1_1_2_2
53
Figure 4-15 Extracting schematic parameter of the measured and the simulated results
Table 4-1 Comparison of extracted schematic parameters of measured and simulated results Measured Simulation (without bias tee) Overlap RC area (sq. RC1 L1 R1( C 1 L1 R1 Device µm) C (pF) (Ω) (nH) Ω) (pF) (Ω) (nH) (Ω) Notch_Filter_ 0.32 0.11 2 300 4.13 1380 7 1 2.49 380 7 2 Notch_Filter_ 0.52 0.20 3 300 4.13 1380 7 1 2.49 380 7 3 Notch_Filter_ 0.58 0.24 4 300 4.13 1380 7 1 2.49 380 7 4 Notch_Filter_ 0.79 0.44 4_1_1_2_2 600 7.07 1380 7 1 3.09 380 7 6
54
CHAPTER 5
BAND PASS FILTER DESIGN AND ANALYSIS
This section discusses the design of band pass filter. Two different design are presented. Simulation and experimental results are discussed and compared. At 0 volt bias, the center frequency was around 8.5GHz.
5.1 Design
This design is microstrip transmission line based with a coplanar waveguide
(CPW) feed line. The separation between the signal and ground lines at the feed point is again 50µm and thus impedance matching of 50 ohm is achieved. A 500µm sapphire is used as a substrate and 0.25µm thin film barium strontium titanate (BST) is used as the dielectric layer that is sandwiched between two metal layers. The concept for the design is based on the microstrip based dual band pass filter proposed by Cheng-Ying Hsu et. al. [34]. Cheng-Ying Hsu et. al.’s design is shown in Figure 5-1. It consists of two half- wavelength resonators embedded inside two quarter wavelength resonators which are capacitively coupled with each other. The RF signal flows form one port to the other through capacitive coupling between the two regions.
55
Figure 5-1 Microstrip based dual band pass filter proposed by Cheng-Ying Hsu et. al. [34]
Using a similar approach but without any embedding of the resonators in one another a band pass filter design is developed using 4 resonators. One of the rings with equivalent lumped elements is shown in Figure 5-2. Thus it makes a resonator.
Figure 5-2 Resonator layout - schematic
56
Since it has four rings (4 pole filter) it is named as quad ring filter. Two variations of the design are proposed with their simulated center frequencies at 8.5GHz and 11GHz.
They are named quad ring filter_1 and quad ring filter_2 respectively. Figure 5-3 shows the top view of quad ring filter_1. The overall dimensions of the device are 4340µm by
3200µm. The figure clearly shows all the dimensions of various structures necessary to draw the device. The device exhibits vertical symmetry Figure 5-4 show the top metal layer and bottom metal layer and Figure 5-5 shows a three dimensional view of the device. Thus it can be clearly seen that the four rings are made out of top metal layer only. The separation between the back bone of the 2 rings on the left and that of the 2 rings on the right is 300µm. The capacitive coupling between the 1550µm back bones on either sides helps the RF signal from port 1 to traverse through port 2 and vice-versa. This capacitance is weak since the thickness of the dielectric medium the signal has to travel is 300µm.
Varactors with an overlap area of 40µm by 40µm are placed on either sides and are shunted to ground. This makes the device tunable. These varactors are clearly pointed out in Figure 5-3.
57
Figure 5-3 Top view of quad ring filter_1
Figure 5-4 Top and bottom metal layer layout of quad ring filter_1
58
Figure 5-5 Three dimensional view of quad ring filter_1
The simulation results for quad ring filter_1 are shown in Figure 5-6. The device is simulated using EM sight solver in AWR microwave office. Again here the top and bottom metal layers are approximated as perfect conductors to make it simple for the solver. The dielectric constant of BST is assumed to be 500, and the loss tangent is taken to be 0.02 for the simulation. This simulation gave two band pass regions. The center frequency of the first one is 8.5 GHz with 1.5dB insertion loss and 7.6 dB return loss at this frequency.
Quality factor is found to be 8.5. The center frequency of the other band pass region is
59
14.8GHz 9dB return loss and 3.5dB insertion loss. It is worth noting that a minimum return loss of 10 dB is recommended for bandpass filters.
Figure 5-6 Simulation results for quad ring filter_1
Figure 5-7 shows the top view of quad ring filter_2. The overall dimensions of this device are 3540µm x 2700µm. Thus it is relatively smaller device compared to quad ring filter_2. The overall design of the device is similar to quad ring filter_2. The thickness of the conductors in the resonators is 60µm whereas that of the quad ring filter_1 are
90µm. The length of the conductors in the inner region of the resonators is 1080µm in this case whereas of that in the quad ring filter_1 is 1460µm. The band pass filter in this case is designed to have pass band region at a frequency higher than that in the previous case.
The separation between the resonators on either sides of the center is reduced to 230µm.
60
Similar to the previous design varactors with an overlap area of 40µm x 40µm are included on either sides to make the device tunable.
Figure 5-7 Top view of quad ring filter_2
The S-parameters obtained by simulating the quad ring filter_2 using
Emsight in AWR microwave office are shown in Figure 5-8. The filter has ripple in the passband. The center frequency of the passband is at 10.95 GHz.
61
Figure 5-8 Simulation results for quad ring filter_2
5.2 Measurement results and data Analysis
The two variations of the band pass filter were fabricated on the same wafer as the other devices discussed earlier on in this thesis. Figure 5-9 shows a picture of the fabricated quad ring filter_1. As it can be clearly seen the fabrication process is neatly done, and the structures that are key to this device say, resonators and varactors well laid out.
62
Figure 5-9 quad ring filter_1 – Fabricated device
The S-parameters of this device are then measured using Agilent’s 8720B network analyzer and then compared with the simulated results. Figure 5-10 shows a plot of the measured S21 parameters in the frequency range from 5 GHz to 15 GHz. Two pass band regions are observed. The first one is at the center frequency of 7.5 GHz and the second one at 14.41 GHz. The -3dB cut-off frequencies for the first passband are at 7 GHz and
8.5 GHz. Thus the bandwidth of the first pass band is 2.5 GHz and the quality factor is 3.
The second passband has the same -3dB bandwidth, but the quality factor is 5.8. The insertion loss at the first pass band is -2.859 dB and at the second passband is -3.525 dB.
The return loss at the first pass band is -13.25 dB, and that at the second passband is -
6.651 dB.
63
Figure 5-10 Measured results of quad ring filter_1
Figure 5-11 Measured Vs. Simulated S11 of quad ring filter_1
64
Figure 5-12 Measured Vs. Simulated S21 of quad ring filter_1
Figure 5-11 and 5-12 shows a comparison of S11 and S21 of the measured versus simulated results of quad ring filter_1. Clearly the center frequency of the first pass band of the measured results is 1 GHz less than the simulated value. The simulation was done assuming the dielectric constant to be 500. But in reality by observing the previous work it is evident that, at 0 bias, the dielectric constant is around 990 [14]. Since capacitance is directly proportional to dielectric constant as the dielectric constant increases the capacitance increases. In the simplest terms, resonant frequency is given by fo = 1/2πLC.
Thus resonant frequency decreases as the capacitance increases and thus this trend can be justified.
The measured results are shown in Figure 5-13. The center frequency of the filter is at 9.2 GHz with 3.7 dB insertion loss and 16.3 dB return loss. The -3 dB cut off frequencies are at 8.8 GHz and 10.4 GHz. Thus the quality factor is 5.75. A comparison
65 of the measured S11 and S21 verses simulated ones are shown in Figures 5-14 and 5-
15. Again, here the center frequency of the measured results is shifted towards the left by around 1.7 GHz which can be explained by fact that the 0 bias dielectric constant is higher than 500. Figure 5-16 shows a plot of S21 for different bias voltages. Clearly the device is not tunable. The 40µm by 40µm varactor gives a very high capacitance, and so it may be acting as a shunt to ground.
Figure 5-13 Measured results of quad ring filter_2
66
Figure 5-14 Measured Vs. Simulated S11 of quad ring filter_2
Figure 5-15 Measured Vs. Simulated S21 of quad ring filter_2
67
Figure 5-16 Measured S21 of quad ring filter_2 Vs Bias voltage
An equivalent microstrip schematic (Figure 5-17) was designed to match the measured results and obtain an effective dielectric constant. This effective dielectric is a combination of the dielectric constants of the sapphire and BST layers. For a dielectric constant of 15 the results of the schematic matched closely with that of the measured results. The comparisons for Quadringfilter_1 and Quadringfilter_2 are presented in
Figures 5-18 and 5-19.
68
Figure 5-17 Microstrip schematic of the band pass design
69
Figure 5-18 Schematic tuned to match measured results of Quadringfilter_1
Figure 5-19 Schematic tuned to match measured results of Quadringfilter_2
70
CHAPTER 6
CONCLUSION AND FUTURE WORK
6.1 Conclusion
Design of band stop and band pass filters for microwave and radio frequency applications are presented in this thesis. The bandstop filters are based on coplanar waveguide (CPW) transmission lines on a 500 µm thick sapphire substrate. 0.25 µm thin film BST is used as the tunable dielectric layer. The design of the band stop filter is based on the traditional varactor shunt switch (designed by Dr. Subramanyam and his team) coupled to the ground using an inductive path. Thus the design symbolizes a single pole standard band stop filter with the potential for frequency tunability. The capacitive overlap makes the device tunable. Several designs of the band stop filter based on the same concept, but different capacitive overlaps, inductive configurations and overall device dimensions were designed and studied. The center frequency for these designs varied from 1 to 5 GHz in case of simulation and 3GHz to 10GHz in case of measured results.
The measured results did not match the simulated results although the devices were tunable. The measured results indicate that the dielectric constant of the BST film might be lower, or the thickness is higher than what it was designed for. Further investigation needs to be to done to identify the quality of the BST film.
71
The band pass filter is mostly single layered and represents a microstrip based structure although it is CPW fed. It is shunted to ground with varactors on either side of the device. The idea of having varactors was to achieve tunability. It has 4 resonant traps representing a 4-pole filter. Two designs (quad ring filter_1 and quad ring filter_2) based on the same concept but with different overall device dimensions were designed, simulated and fabricated. Quad ring filter_1 showed two pass band regions with center frequencies at 7.5 GHz and 14.3 GHz whereas quad ring filter_2 showed a center frequency at 9.2 GHz. By comparing the measured results of the band pass filter with the equivalent microstrip schematic the effective dielectric constant of the substrate with sapphire (of thickness 500µm) and superstrate of BST (of thickness 0.25 µm) was found to be 15. This could be very useful to save simulation time.
6.2 Future work
In the case of the band stop filter the measured results do not agree with the simulation results. The method used for simulating this device has to be investigated, and approximations in the design have to be minimized to come up with an accurate match. For these simulations, the metal layers were approximated as perfect conductors to save the time taken to simulate these devices. Also, use of other simulation software that use 3-D solvers can be more accurate. The quality of the BST film that was used for fabrication has to be investigated, since it seems like the effective dielectric constant of the film is lower pushing the center frequencies of the measured results to the right compared to the simulation results. Also, the quality factor of these devices was not very good, or the bandwidth was too wide. The devices need to be further optimized to improve the selectivity.
72
The band pass filter showed promising results, but it would be nice to come up with a lumped element schematic model that represents the device to help in more detailed analysis of the results although a microstrip schematic was built. Also, since the overlap area of the varactors used in this device was too big, the filters were not tunable.
If this device can be optimized to make it tunable, that would be very interesting.
73
BIBLIOGRAPHY
[1] David M. Pozar, “Microwave Engineering,” third edition, Wiley, N.J., 2001
[2] Spartak Gevorgian, “Ferroelectrics in Microwave Devices, Circuits and Systems,”
Springer-Verlag London Limited 2009
[3] Michael Steer, “Microwave and RF Design, A Systems Approach,” SciTech Publishing, Inc., 2010.
[4] Young-Hoon Chun; Jia-Sheng Hong; Peng Bao; Jackson, T.J.; Lancaster, M.J., "BST varactor tuned bandstop filter with slotted ground structure," Microwave Symposium Digest, 2008 IEEE MTT-S International , vol., no., pp.1115,1118, 15-20 June 2008
[5] I. C. Hunter and J. D. Rhodes, “Electronically Tunable Microwave Bandstop Filters,” IEEE Microwave Theory and Tech., Vol. MTT-30, No. 9, pp.1361-1367, Sep. 1982.
[6] M J Lancaster, J Powell and A Porch. “Thin-film ferroelectric microwave devices”, Superconductor Science and Technology, 11, 1998, pp 1323-1334
[7] A.K. Tagantsev, V.O. Sherman, K.F. Astafiev, J. Venkatesh, and N. Setter, “Ferroelectric Materials for Microwave Tunable Applications”, Journal of Electro ceramics, vol. 11, pp. 5–66, September 2003.
[8] R. York, A. Nagra, E. Erker, T. Taylor, P. Periaswamy, J. Speck, S. Streiffer, and O. Auciello, “Microwave integrated circuits using thin-film BST,” in Proc. 12th IEEE Int. Applications Ferroelectrics Symp., vol. 1, 2001, pp. 195–200.
[9] Herner S B, Selmi F A, Varadan V V and Varadan V K 1993 “The effect of various dopants on the dielectric properties of barium strontium titanate” Mater. Lett. 15 317–24
[10] A.K. TAGANTSEV, V.O. SHERMAN, K.F. ASTAFIEV, J. VENKATESH & N. SETTER, “Ferroelectric Materials for Microwave Tunable Applications,” Journal of Electroceramics, 11, 5–66, 2003
[11] N.M. Alford, S.J. Penn, A. Templeton, X. Wang, J.C. Gallop,N. Klein, C. Zuccaro, and P. Filhol, “IEE Colloquium on Electro-Technical Ceramics—Processing, Properties and Applications,” 9/1 (1997).
[12] S.J. Penn, N. McNAlford, A. Templeton, N. Klein, J.C. Gallop, P.Filhol, and X. Wang, “IEE Colloquium on Advances in Passive Microwave Components,” 6/1 (1997)
74
[13] Spartak Gevorgian, “Ferroelectrics in Microwave Devices, Circuits and Systems,” Springer-Verlag London Limited 2009
[14] J. Pérez de la Cruz, E. Joanni, P. M. Vilarinho, and A. L. Kholkin, “Thickness effect on the dielectric, ferroelectric, and piezoelectric properties of ferroelectric lead zirconate titanate thin films,” JOURNAL OF APPLIED PHYSICS108, 114106 (2010)
[15] D. Kuylenstierna, G. Subramanyam, A. Vorobiev, and S. Gevorgian, “Tunable electromagnetic bandgap performance of coplanar waveguide periodically loaded by ferroelectric varactors,” Microw. Opt. Technol. Lett., vol. 39, no. 2, pp. 81–86, 2003.
[16] Subramanyam, G.; Ahamed, F.; Biggers, R., "A Si MMIC compatible ferroelectric varactor shunt switch for microwave applications," Microwave and Wireless Components Letters, IEEE , vol.15, no.11, pp.739,741, Nov. 2005
[17] F. Ahamed and G. Subramanyam, “Design of a Si MMIC compatible ferroelectric varactor shunt switch for microwave applications,” in Proc. IEEE Ultrasonics Ferroelectrics
[18] Subramanyam, G.; Patterson, M.; Leedy, K.; Neidhard, R.; Varanasi, C.; Chenhao Zhang; Steinhauer, G., "Linearity and temperature dependence of large-area processed high-q barium strontium titanate thin-film varactors [Correspondence]," Ultrasonics, Ferroelectrics and Frequency Control, IEEE Transactions on , vol.57, no.7, pp.1692,1695, July 2010
[19] A. Kozyrev, A. Ivanov, T. Samoilova, O. Soldatenkov, K. Astafiev, and L. Sengupta, “Nonlinear response and power handling capability of ferroelectric BST capacitors and tunable microwave devices,” J. Appl. Phys., vol. 88, no. 9, pp. 5344–5352, 2000.
[20] G.Subramanyam, F. Ahamed, R.Biggers, A.Campbell, R.Neidhard, E.Nykiel, R.Cortez, K.Stamper, M.Calcatera, “A new ferroelectric varactor shunt switch for microwave and millimeter-wave reconfigurable circuits”, Frequenz. Vol 59, 2005, pages 37-40
[21] Guru Subramanyam, Kevin Leedy, Chakrapani Varanasi, Robert Neidhard, Keith Stamper, and Mark Calcatera “A Low Voltage Tunable Analog Phase Shifter Utilizing Ferroelectric Varactors, ” Integrated Ferroelectrics, 2010
[22] Thin-film ferroelectric materials and devices, Edited by R. Ramesh, Kluwer Academic, Norwell, MA, 1997.
[23] Y. Miyasaka and S. Matsubara, “Dielectric properties of sputter-deposited BaTiO3- SrTiO 3 thin films,” in Applications of Ferroelectrics, 1990., IEEE 7th International Symposium on, 1990, pp. 121–124.
[24] C.V. Varanasi, G. Subramanyam, D.H. Tomich, K.D. Leedy D.H. Tomich G. Subramanyam “Large area Ba1−xSrxTiO3thinfilms for microwave applications deposited by pulsed laser ablation,” Science Direct, march2, 2009
75
[25] H. Li, G. Subramanyam, and J. Wang, "Capacitance of Thin Ferroelectric Thin Films Obtained by Different Methods", presented in the IEEE International Symposium on Applications of Ferroelectrics, ISAF 2008, Feb. 2008
[26] Guru Subramanyam, Mark Patterson, Kevin Leedy, Robert Neidhard, Chakrapani Varanasi, Gregg Steinhauer, “Novel Multi-Capacitor Architecture for BST Thin Film Varactors,” Integrated Ferroelectrics Vol. 125, Iss. 1, 2011
[27] Yue Hailing, “Design and optimization of barium strontium titanate ferroelectric varactors,” Master’s thesis, University of Dayton, 2012.
[28] Brown, D.; Chenhao Zhang; Patterson, M.; Subramanyam, G.; Leedy, K.; Cerny, C., "Coplanar waveguide varactors with bottom metal trenched in silicon, " Aerospace and Electronics Conference (NAECON), Proceedings of the 2011 IEEE National , vol., no., pp.309,311, 20-22 July 2011
[29] Sekhar Kanagala, Faruque Ahamed, Urmila Nath, Shilpa Wakade, Guru Subramanyam, “Electrical Modeling of Ferroelectric Tunable Microwave Components and Circuits,” Ferroelectrics, Vol. 329, Iss. 1, 2005
[30]Jiang, H.; Patterson, M.; Zhang, C.; Brown, D.; Subramanyam, G., "Miniaturized and reconfigurable CPW square-ring slot antenna using thin film varactor technology," Microwave Symposium Digest (MTT), 2011 IEEE MTT-S International , vol., no., pp.1,1, 5-10 June 2011
[31] http://wps.prenhall.com
[32] Xu Yi, “Modelling and analysis of CPW based multi-layer on-chip inductors and design of multi-resonator for RF signature sensor,” Thesis, University of Dayton, December 2011
[33] Y. Aoki, K. Honjo; “Q-Factor Definition and Evaluation for Spiral Inductors Fabricated Using Wafer-Level CSP Technology”; Microwave theory and techniques, IEEE Transactions, Vol. 53, Oct. 2005, pages 3178-3184
[34] T. Horng; “A Novel Modified-T Equivalent Circuit for Modeling LTCC Embedded Inductors with a Large Bandwidth”; IEEE Transactions on Microwave Theory and Techniques, VOL. 51, NO. 12, Dec 2003, 2327-2333
[35] Chenhao Zang, “Design of miniaturized X-band chebychev band pass filter based on BST thin film,” Master’s thesis, University of Dayton, August 2012
[36] Sreekanth Vemulapalli, “Design of tunable band pass filter using BST thin film,” Master’s thesis, University of Dayton, August 2012
[37] Subramanyam, G.; Van Keuls, F.W.; Miranda, F.A., "A K-band-frequency agile microstrip bandpass filter using a thin-film HTS/ferroelectric/dielectric multilayer
76 configuration," Microwave Theory and Techniques, IEEE Transactions on , vol.48, no.4, pp.525,530, Apr 2000
[38] OY. Buslov, CY. Kang, “Dielectric resonators loaded by ferroelectricvaractors for tunable Ka band filter”, Integrated Ferroelectrics, Vol 86, 2006, pp 171-179
[39] Cheng-Ying Hsu; Chu-Yu Chen; Huey-Ru Chuang, "A Miniaturized Dual-Band Bandpass Filter Using Embedded Resonators," Microwave and Wireless Components Letters, IEEE , vol.21, no.12, pp.658,660, Dec. 2011
[40] Y.-H. Chun, J.-S. Hong, P. Bao, T.J. Jackson, M.J. Lancaster, “Tunable slotted ground structured bandstop filter with BST varactors,” IET Microw. Antennas Propag., 2009, Vol. 3, Iss. 5, pp. 870–876
[41] Preradovic, S; Balbin, I.; Karmakar, N.C.; Swiegers, G.F., "Multiresonator-Based Chipless RFID System for Low-Cost Item Tracking," Microwave Theory and Techniques, IEEE Transactions on, vol.57, no.5, pp.1411, 1419, May 2009
[42] Chenhao Zhang; Alemayehu, A.; Patterson, M.A.; Subramanyam, G., "Design of high voltage tunable Shunt Interdigitated resonator based on Barium Strontium Titanate thin film," Aerospace and Electronics Conference (NAECON), Proceedings of the 2011 IEEE National , vol., no., pp.305,308, 20-22 July 2011
[43] K. Abe and S. Komatsu, “Ferroelectric properties in epitaxially grown BaxSr1-xTiO3 thin films,” Journal of Applied Physics, vol. 77, pp. 6461–6465, 1995.
[44] Guru Subramanyam, Faruque Ahamed, Rand Biggers, Robert Neidhard, Edward Nykiel, John Ebel, Richard Strawser, Keith Stamper, and Mark Calcatera, “RF Performance Evaluation of Ferroelectric Varactor Shunt Switch.
77