A Frequency Standard for Today's WWVB

Home , Digital clock, WWVB, WWV (radio station)

John A. Magliacane, KD2BD 1320 Willow Drive, Sea Girt, NJ 08750; [email protected]

A Frequency Standard for Today’s WWVB

The author shares the design of his frequency standard that’s fully compatible with today’s WWVB.

For over a half century, the 60 kHz carrier from WWVB has served as a popular and highly respected frequency reference among many scientific, research, and engineering professionals across North America. As GPS-disciplined frequency standards started gaining in popularity, the use of WWVB in frequency reference applications began to decline. With this decline, however, came a tremendous increase in the sale and application of low-cost radio controlled “atomic clocks” that receive time and date information from WWVB on a regular basis. This drastic shift in the use of WWVB by the American public forced a realignment of the priorities set for the station by the National Institute of Standards and Technology (NIST). With the rapid proliferation of radio-controlled clocks came a realization that reception of the WWVB time code by low-cost consumer products was often unreliable. In the eastern United States where signals from the Fort Collins, Colorado WWVB transmitter are often the weakest, competition from Figure 1 — The author’s very successful Frequency Measuring Test results are achieved by high noise levels and co-channel interference using WWVB as a precision frequency reference. from British radio station MSF made reception especially difficult. Enhancements were made to the WWVB effective radiated power and modulation depth, but reception dif- ficulties continued to persist. Finally, when attempts to commission a second WWVB transmitter to serve the east coast failed to New Modulation Format reach fruition by the end of 2009, thoughts NIST realized that only through a radi- shift keying (BPSK), that next generation turned toward making more aggressive cal change in time code transmission and radio-controlled clocks could be designed to changes to WWVB broadcasts that would greater sophistication in receiving techniques receive and process with improved reliability. help improve reception reliability while would any further improvement in recep- While over-the-air tests conducted in 2012 maintaining compatibility with the millions tion reliability be achieved. They decided revealed that the addition of the new BPSK of radio-controlled clocks already in opera- to add a new time code to the WWVB car- time code had little or no effect on the operation. rier, modulated by means of binary phase tion of existing radio-controlled clocks, it

QEX November/December 2015 13 caused every WWVB-disciplined frequency application in consumer products, and away standard to lose phase reference with the from the scientific and engineering com- A BPSK-Compatible Frequency Standard WWVB carrier, and thus malfunction. With munities, the WWVB carrier frequency is The frequency standard described here GPS-based frequency standards in wide use maintained to within one part in 1014, and employs a combination of linear signal and rising steadily in popularity, and com- continues to be derived from a set of four processing, vector demodulation, and super- mercial WWVB-based frequency standards cesium clocks.1 As such, provided that heterodyne receiving techniques to not only out of production for years, NIST decided BPSK compatible reception techniques are discipline a 10 MHz voltage-controlled, to permanently add its new BPSK time code employed, radio station WWVB can con- temperature-compensated crystal oscilla- to WWVB broadcasts beginning on October tinue to serve as a reliable and accurate fre- tor (VCTCXO) against the WWVB carrier, 29, 2012, and force all previously function- quency reference, and can do so with a level but also decode its amplitude modulated ing WWVB-disciplined frequency standards of precision that far exceeds that of its time time code. A wide variety of calibration into obsolescence. dissemination. signals are available to the user, while an Despite this WWVB priority shift toward LCD display provides NIST (UTC) date, time, and UT1 offset information. Several audio outputs are provided to help assess reception quality and assist in selecting an QX1511-Magliacane02 WWVB optimum antenna placement and orientation. 60 kHz An RS-232 port is also included to provide UTC time and date information to external devices. Frequency Unknown Standard Signal Performance The stability and accuracy of a frequency 1 kHz standard are difficult to assess and quantify without making a direct comparison against 1 kHz Timebase Receiver another standard of superior precision. Even while lacking a second standard, however, Error Frequency some conclusions can be drawn based on Voltage Counter L.O. the performance a frequency standard dem- onstrates while being employed in critical Frequency VFO applications over the course of many years. VFO Divider For example, the results achieved while employing this frequency standard during Frequency Measuring Tests over the past decade have consistently equaled or surpassed those of most other participants, many of whom employ advanced digital Figure 2 — The author’s FMT methodology applies a DC tuning voltage to the local oscillator signal processing techniques, commercial of a direct conversion receiver to lock its audio output in phase with that of a 1 kHz reference. The frequency of the unknown signal is determined by measuring the frequency GPS-disciplined frequency standards, rubid- of the LO and factoring in the 1 kHz tuning offset. ium oscillators, and other laboratory grade

Figure 3 — Space and Naval Warfare Systems (SPAWAR) Command computer simulations illustrate how the WWVB 100 mV/m signal level contours contract during the day and expand during nighttime hours. (Images from NIST website.)

14 QEX November/December 2015 instrumentation. Figure 1 shows a certificate cies very closely approaching those of the the carrier amplitude is restored to full power I received for my participation in the April WWVB transmitted carrier frequency can be 200 ms later, this represents a “0” bit in the 2014 FMT. achieved by integrating the diurnal perturba- WWVB legacy time code. If it is restored Since the influence of ionospheric pertur- tions over a day or more. 500 ms later, this represents a “1” bit in the bations and inaccuracies in frequency mea- WWVB time code. If it is restored 800 ms surement methodologies are often the largest later, this represents a “Marker” bit, one of WWVB Signal Characteristics consistent sources of error in FMTs, these which is sent every 10 seconds to establish With rare exception, WWVB broadcasts results alone cannot speak to the full per- frame synchronization in receivers. Two 24 hours a day with a peak envelope effec- formance of this frequency standard. Using Marker bits transmitted in succession iden- tive radiated power of 70 kW. The beginning the identical hardware and methodology as tify the beginning of a new time code frame of every UTC second is identified by a 17 dB those employed in FMTs (Figure 2), I mea- and the start of a new UTC minute. Over the reduction in radiated power. See Figure 4. If sured the carrier frequencies of many AM course of one minute, the individual “1”s and broadcast stations received over ground wave paths to a resolution of 312.5 microhertz (less than one part per billion at 1 MHz). During the process, several transmitters were identi- PWM '0' Bit fied as consistently having no measurable frequency error, some of which are known 1.0 to employ GPS-disciplined rubidium frequency standards for carrier generation. 0.5 I believe that transmitters exhibiting very small but measurable frequency offsets were 0.0 likely rubidium controlled but not necessar- –0.5 ily GPS-locked, while others that exhibited larger errors that varied over time might have –1.0 simply been crystal controlled.

Since the GPS-disciplined radio station WWVB's Relative Carrier Amplitude (V) 0.0 0.2 0.4 0.6 0.8 1.0 carriers stood out so clearly among all other Time Since The Start Of The UTC Second (s) stations measured, it follows that the accuracy and stability of this frequency standard probably exceeds the resolution of the mea- PWM '1' Bit surements taken. In fact, they are possibly several orders of magnitude better based on 1.0 the known precision to which the WWVB carrier frequency is maintained, and the rec- 0.5 ognized RF propagation characteristics that exist at low frequencies. 0.0

–0.5 60 kHz Propagation LF radio propagation is substantially –1.0 different from that which exists at higher frequencies. Its remarkable stability and reli- WWVB's Relative Carrier Amplitude (V) 0.0 0.2 0.4 0.6 0.8 1.0 ability have often led to the belief that 60 kHz Time Since The Start Of The UTC Second (s) signals propagate great distances over ground wave paths alone. In reality, a combination of surface wave and D-layer ionospheric paths PWM 'Marker' Bit are responsible for WWVB signal propagation. At night, cosmic background radiation 1.0 supports a level of D-layer ionization that 0.5 is sufficient for propagating LF (and lower 2 frequency) radio signals over long distances. 0.0 Greater D-layer efficiencies and increased effective height with decreased ionization –0.5 levels contribute to greater signal coverage during the nighttime hours. See Figure 3. –1.0 Diurnal shifts in the height of the D-layer cause changes in the RF path length between WWVB's Relative Carrier Amplitude (V) 0.0 0.2 0.4 0.6 0.8 1.0 WWVB and receivers to occur during the Time Since The Start Of The UTC Second (s) time the RF path undergoes sunrise and QX1511-Magliacane04 sunset transitions. While the accompany- ing Doppler shifts during these periods are Figure 4 — The WWVB carrier is reduced in amplitude by 17 dB at the beginning of every generally small, their effects are cyclic and second. If full power is restored 200 ms later, this represents a “0” bit. If restoration occurs predictable, and can be handled using a priori 500 ms later, this represents a “1” bit. If it is restored 800 ms later, this represents a “Marker” knowledge. Long-term frequency accura- bit. The new WWVB BPSK time code is not shown in this illustration.

QEX November/December 2015 15 “0”s produce a pattern that conveys the cur- connected in series with its adjacent conduc- meter signal contour in central New Jersey, rent date and time, UT1 offset, leap second, tor to form a continuous coil. See Figure 5 for WWVB typically induces several millivolts and daylight savings time information. an example of how such a coil can be wired. (peak-to-peak) of RF across the 40 turn loop. WWVB recently added a secondary time I fabricated an antenna from a 5 meter length code in the form of BPSK, which co-exists of 40 conductor #28 AWG stranded ribbon Antenna Preamplifier independent of the original amplitude modu- cable, and supported it from the ceiling of my A preamplifier located immediately lated time code. If the phase of the WWVB attic. Figure 6 shows my antenna installation. adjacent to the loop provides an important carrier is reversed 100 ms after the beginning Resonance is achieved by introducing an interface between the high-impedance bal- of the UTC second, this represents a “1” bit appropriate amount of capacitance in parallel anced loop antenna and the unbalanced, in the BPSK time code. If no carrier phase with the loop. Aligning the plane of the loop ground-referenced circuitry that follows. reversal takes place at this time, it represents in the direction of Fort Collins, Colorado Figure 7 shows the schematic diagram of a “0” bit. allows the WWVB horizontally polarized the preamplifier. The preamplifier connects The new BPSK time code offers enhanced H-Field to cut through the center of the to the frequency standard through a length performance through error detection and cor- loop and induce a voltage across its termi- of coaxial cable, and it receives its operating rection algorithms, as well as several dif- nals. Along the eastern edge of the 100 mV/ ferent modes of operation, not all of which have been fully implemented at this time. While the frequency standard described here possesses BPSK demodulation capability, it is employed solely for the detection and removal of BPSK prior to the carrier phase tracking circuitry that follows.

A Low-Noise WWVB Antenna Unlike GPS, WWVB reception does not require the use of an outdoor antenna, nor does it require a clear view of the sky. Low- cost consumer grade radio controlled clocks employ electrically small H-Field ferrite loopstick antennas that operate on the same principle as those found in portable AM broadcast band radios. Significantly higher signal levels and noticeably less sensitivity to nearby structures can be realized by using physically larger air-core loop antennas. While it can take a lot of wire and a lot of labor to wind a 60 kHz resonant air-core loop antenna, just a single turn of multi-conductor ribbon cable can produce an effective multi- turn loop provided that each conductor is

QX1511-Magliacane05

Figure 5 — Loop antenna construction. The ends of the multi-conductor cable are brought together at the bottom of the loop to form a multi-turn coil. After going around once, the end of conductor #1 connects to the beginning of conductor #2, conductor Figure 6 — The author’s attic-mounted loop antenna. With the plane of the loop aligned #2 connects to conductor #3, and so on. toward Fort Collins, Colorado, the WWVB horizontally polarized H-Field cuts through the The uncommitted ends of the first and last center of the loop and induces a voltage across it. The frequency standard amplifies and conductors form the ends of the coil. processes this signal.

16 QEX November/December 2015 Center C5 Tap C8 330 pF 0.01 μF R8 200 kΩ Loop Antenna 40 turns #28 wire 5 meter circumference U2 AD744 R6 C4 330 pF 1 10 kΩ 2 5 C1 6 680 pF R7 270 Ω 3 C7 C2 7 1 μF J1 580 pF 4 To D2 D1 Receiver 6.2 V 6.2 V L2 R2 R1 10 mH 2.0 MΩ 2.0 MΩ R10 390 kΩ R4 C3 5.1 kΩ 1 μF C6 R9 D3 1 μF 390 kΩ 15 V 3 5 2 6 1

R5 U1 R3 5.1 kΩ AD620 4.7 kΩ 7 8 4

QX1511-Magliacane07

Figure 7 — 60 kHz WWVB loop antenna preamplifier. In addition to providing 40 dB of gain, the preamp properly interfaces the balanced loop antenna with the ground-referenced circuitry in the frequency standard.

Loop Antenna QX1511-Magliacane08

+45° Phase Shift

Mixer MCU 0° Variable Controlled 10 kHz Phase Attenuator Phase IF Output Shift Select 60 kHz 60 kHz 10 kHz Differential Amplifier Amplifier Amplifier AGC –45° 50 kHz Control Phase Local Voltage Shift Oscillator

Figure 8 — This block diagram illustrates the overall RF to IF conversion process. The +45° and –45° phase shift networks were used in the past to compensate for the WWVB hourly phase signature, but have not been required since the introduction of BPSK modulation was made in 2012.

QEX November/December 2015 17 60 kHz RF Out 15 QX1511-Magliacane09 10 U10B 1 2 4 9 U10C 470 pF C4 R14 1 kΩ 3 5 D8 R8 1N914 470 pF C3 5.6 kΩ 2.7 kΩ 5.6 kΩ R10 R6 R7 1.1 kΩ D7 5.1 V 0.5 W 1N5231 +12 V C5 8 R11 56 kΩ 0.1 μF U3C LMC6484N +12 V 9 10 Q3 2N3904 R16 C7 10 kΩ 0.001 μF C9 0.33 μF +12 V R5 C2 2N3904 Q2 560 kΩ 0.01 μF +12 V 8 7 14 16 2N3904 Q4 R9 10 kΩ D2 1N914 +12 V +12 V R4 1 kΩ D4 U2A 1N914 CD4053BE R2 R2 56 kΩ 56 kΩ 6 11 12 R12 100 kΩ D3 1N914 R3 R13 1 kΩ 51 kΩ +12 V D1 1N914 Figure 9 — Details of the RF AGC circuitry and legacy carrier phase shift networks of the frequency standard. of the frequency carrier phase shift networks and legacy circuitry Figure 9 — Details of the RF AGC Q1 D6 1N914 R1 56 kΩ 2N3904 C8 2N3904 Q5 R17 10 kΩ D5 1N914 C1 L1 10 mH R15 0.001 μF 10 kΩ 0.01 μF +12 V +12 V C6 J1 470 pF R18 100 kΩ IF In Input from Noise preamplifier 10 kHz antenna and Gate In A In B In

18 QEX November/December 2015 " I 60 kHz RF Out " "Q" Out Out Channel Channel IF Out 10 kHz 6 8 7 14 11 4 15 16 9 10 U5A +12 V U5B U5C CD4053BE 12 13 1 2 3 5 8 8 U4C R30 100 kΩ LM837N R44 U6C 9 CD4070B U27A 3 100 kΩ 10 LM837N 9 10 +6 V 7 +12 V R31 14 R29 51 kΩ 100 kΩ R45 51 kΩ 1 2 R43 1 100 kΩ LM837N U1A 0.01 μF C16 4 11 +6 V R27 +12 V R47 100 kΩ 200 kΩ 7 2 3 8 LM837N U6C U6B R41 16 kΩ LM837N R28 200 kΩ 6 5 C14 C15 330 pF 330 pF 9 10 +6 V C22 R42 R40 16 kΩ 16 kΩ R48 R26 0.001 μF 51 kΩ R46 R25 10 kΩ 300 Ω 100 kΩ 7 +6 V C21 +6 V 0.1 μF U1B 0.01 μF C13 LM837N R23 +6 V 200 kΩ 6 5 +6 V R49 100 kΩ 6 5 C12 R24 330 pF 200 kΩ C11 330 pF U4B LM837N +6 V 7 C20 0.1 μF R22 R21 C18 R32 10 kΩ 10 kΩ 300 Ω 330 pF 8 R39 10 kΩ LM837N U4D R36 10 kΩ A –90° phase shift in the lower IF path drives I and Q demodulators that provide outputs at baseband (DC) levels. that provide IF path drives I and Q demodulators A –90° phase shift in the lower R37 U1C 2.0 MΩ R38 43 kΩ +6 V LM837N R35 43 kΩ 9 10 20 kΩ R33 R20 100 kΩ 13 12 LM837N U4A 1 +6 V R19 +6 V 100 kΩ C19 4 11 330 pF C17 C10 +12 V 0.01 μF 330 pF 14 2 3 In 20 kΩ R34 LO In RF In LO In QX1511-Magliacane10 BPSK 50 kHz 10 kHz 60 kHz kHz IF, where additional gain and selectivity are provided. gain and selectivity are provided. where additional WWVB to a 10 kHz IF, that downconverts 60 kHz amplification precedes a balanced mixer Figure 10 — RF to baseband signal processing.

QEX November/December 2015 19 power from the frequency standard through Inductor L1 and capacitor C1 form a bias path would again be selected through U10B. the same length of coax. This connection tee network that feeds DC operating voltage If the frequency standard were powered on provides a voltage transfer rather than a to the preamplifier while simultaneously between 10 and 15 minutes past the hour, power transfer of RF energy between the pre- directing RF from the preamplifier into a cur- the WWVB carrier would have already been amplifier and the frequency standard. Since rent controlled RF attenuation network. This advanced +45°. Therefore, the frequency the length of coax will be small in terms network is part of the frequency standard standard would continue using the 0° path, of wavelength, transmission line effects AGC circuitry, and it consists of resistors R1 and later select the +45° path when WWVB and impedance matching concerns can be and R2, and diodes D1 through D4. would have shifted back –45° at 15 minutes ignored, and coaxial cable of any convenient Capacitor C6 along with diodes D5 and after the hour. surge impedance can be employed. D6 and resistor R13 charge capacitor C7 The amount of capacitance required in to the peak level of the IF voltage. Analog Converting to a 10 kHz IF parallel with the loop to achieve resonance switch U2A is opened when strong atmo- at 60 kHz depends on the size, shape, dis- spheric discharges are detected. This action Phase and amplitude conditioned 60 kHz tributed capacitance, and overall inductance prevents C7 from charging to the peak level signals are next handled by the circuitry illus- of the loop. The 40 turn loop described here of the noise impulse, which would otherwise trated in Figure 10. Here, additional RF gain required about 1050 pF of capacitance. engage the AGC more heavily, and desensi- and selectivity are provided by active band- Capacitor C1 should be a silver mica or simi- tize the frequency standard in the moments pass filters designed around op-amps U1A lar low-loss, temperature stable capacitor. I following the static crash. and U1B. As was the case in the preamplifier, used an Elmenco 467 110 pF to 580 pF com- The voltage across C7 is buffered by op- the 330 pF capacitors at C11, C12, C14, and pression trimmer capacitor at C2 to carefully amp U3C, level shifted through diodes D7 C15 should be of a temperature stable, low- bring the loop to resonance at 60 kHz. and D8, and used to control the RF attenua- loss chemistry. An AGC conditioned 60 kHz Signal-to-noise ratios at LF are often tion network through transistors Q1 and Q2. RF sample from the output of U1A is buff- enhanced by desensitizing the near-field With no current applied to the network, ered and made available for use outside the response of the receiving antenna to E-Field the dynamic resistance of all four diodes is frequency standard. energy. This effect is often accomplished by high, and very little RF signal attenuation U4C forms a 180° unity gain phase employing an electrostatic shield around the takes place across resistors R1 and R2. As inverter that forms a balanced mixer along perimeter of the loop. In this design, E-Field the IF signal level begins to rise above the with U5A, one section of a CD4053BE triple desensitization is achieved by employing a threshold set by diodes D7, D8, and the SPDT analog switch. Driven by a 10 MHz high common-mode-rejection differential base-emitter junctions of transistors Q1 and derived 50 kHz local oscillator, this mixer amplifier as the first stage of the preamplifier. Q2, the DC current passing through diodes converts the 60 kHz RF signal down to a Capacitor C8 provides an AC ground to the D1 through D4 begins to increase. This cur- 10 kHz IF. Op-amps U4A, U4B, and U4D electrical center of the loop, to enhance the rent decreases the dynamic resistance of the form a 10 kHz biquad active bandpass filter preamplifier rejection of out-of-band signals. diodes, and causes an increasing amount of that serves as a high-gain, narrow bandwidth Zener diodes D1 and D2 help protect the RF to be conducted through them to ground. IF amplifier. This amplifier provides 46 dB Analog Devices AD620 differential ampli- A fairly constant peak RF level remains after of gain and a 3 dB bandwidth of 100 Hz. fier from damage when nearby lightning the attenuation network, and appears on the The center frequency of the amplifier is set to strikes induce high voltage impulses across emitter of transistor Q3. 10 kHz through careful alignment of poten- the antenna. For many decades, WWVB employed tiometers R36 and R39. The 330 pF capaci- The differential amplifier is followed by a method of station identification where its tors employed at C18 and C19 must be of a a 60 kHz second-order active bandpass fil- transmitter would advance the phase of its low-loss variety. ter designed around an AD744 operational carrier +45° 10 minutes after the start of each Working from within the AGC loop, the amplifier. This stage provides additional RF hour, and return to normal phase 5 minutes IF amplifier provides an output voltage of selectivity beyond that of the loop antenna later. These hourly phase shifts were discon- about 8 Vp-p. A sample of the 10 kHz IF is alone, and raises the overall voltage gain of tinued when BPSK modulation was added made available for use outside the frequency the preamplifier to 40 dB. The AD744 has in 2012. Nevertheless, the circuitry used to standard. the capability of driving capacitive loads, and compensate for these phase shifts worked can deliver output voltages as high as 10 Vp-p extremely well, and is included here for dis- Demodulating “I” and “Q” without distortion. cussion. The 10 kHz IF signal is split between a Resistor R3 sets the gain of the AD620, 60 kHz RF appearing on the emitter of straight-through path and a path through op- and may be increased in value if less pream- transistor Q3 is simultaneously applied to a amp U6B that produces a –90° phase shift. plifier gain is desired. For best performance, +45° phase shift network (C3, R10), a –45° A balanced mixer — consisting of op-amp capacitors C4 and C5 should be low-loss, phase shift network (R8, C4), and a resistive U6C and analog switch U5B — is driven by high temperature stability devices (such voltage divider (R6, R7) having the same a 10 MHz derived 10 kHz local oscillator, as silver mica), and all resistors should be attenuation characteristic as each phase shift and forms an in-phase (“I” channel) demodu- within 5% tolerance. network. In the past, the appropriate RF path lator. The –90° phase shifted path feeds a would be selected by the microcontroller second balanced mixer consisting of op-amp based on the current time of day. The 0° RF Amplitude and Phase U6C and analog switch U5C. This mixer is path would normally be selected when the Management driven by the same 10 kHz local oscillator frequency standard was initially powered on. Figure 8 illustrates the overall process of after it has passed through exclusive OR gate At 10 minutes after the hour, the –45° path converting the WWVB 60 kHz signal down U27A, and produces a quadrature (“Q” chan- would be selected through analog switch to a 10 kHz IF, where separate amplitude and nel) demodulator. U10C and into U10B to compensate for the phase detection takes place. Figure 9 illus- The output of the “I” channel demodula- +45° phase advance that would occur at that trates the frequency standard RF circuitry in tor is a +6 V referenced baseband DC voltage time. At 15 minutes after the hour, the 0° greater detail. that is linearly proportional to the WWVB

20 QEX November/December 2015 carrier phase polarity and modulation amplitude. After the effects of BPSK modulation have been removed by the action of U27A AGC AM Time Audio Audio Voltage VCTCXO Voltage Control Code Control Carrier BPSK and its associated circuitry, the “Q” channel Noise demodulator produces a +6 V referenced baseband DC voltage that is linearly proportional to a WWVB carrier having no phase modulation.

Figure 11 presents an overview of the Slicer Hold signal path taken by the 10 kHz IF. As illus- Peak trated in Figure 12, op-amps U3D and U7D function as a 3.685 Hz wide four pole Bessel low-pass filter, and set the frequency standard “I” channel RF bandwidth to 7.37 Hz. Output AGC Gate Peak Noise from the filter drives op-amps U25A and Gated Detector U25B. Using a virtual ground DC reference of +6 V, op-amp U25A acts as a voltage comparator and demodulates the WWVB BPSK into a 12 Vp-p square wave. U25B serves as a unity gain phase inverter, and provides an “I” Carrier channel waveform that is complementary to Rectifier the original. The CD4066B analog switching arrange- ment that follows is driven by the demodulated BPSK waveform provided by U25A. By selecting the non-inverted “I” channel signal through U26B when the WWVB One carrier phase is normal, and selecting the Shot inverted “I” channel signal through U25B and U26C when the WWVB carrier phase is inverted, a BPSK-free “I” channel is pro- Slicer

duced. U25A’s BPSK correction signal is LPF 0.025 Hz

also applied to exclusive OR gate U27A to Noise Rectifier control the phase of the 10 kHz local oscil- BPSK lator fed to the “Q” channel demodulator, and allows the demodulator to maintain a constant output polarity. The filtered and BPSK-free “I” channel LPF energy is applied to op-amp U7A and diode 5 Hz D9, and work through analog switch U10A VCTCXO. the for tuning voltage provides and resistor R57 to charge capacitor C27 to Enable the peak voltage of the demodulated AM time

code. This voltage is buffered through U7B, LPF whose output is applied across the R59/R60 200 Hz voltage divider. The voltage divider sets the threshold of comparator (op-amp) U7C to a I level that is midway between the upper (0 dB peak reference) and lower (–17 dB below Q peak reference) amplitudes of the WWVB detected carrier. The comparator produces a switching waveform that illuminates green Time Code LED, D10, in synchronization with the WWVB amplitude modulation. This waveform is also made available to the microcontroller following a translation to TTL volt- (PLL) –90° Shift Carrier 10 kHz Detector age levels by transistor Q6. Phase Analog switch U10A is briefly opened when high amplitude noise impulses are detected to prevent them from charging C27 and overshooting the decision threshold set by U7C. The demodulated time code pulses and the peak time code voltages are also indi- I.F. L.O. L.O. 1 kHz cated on a meter mounted on the front panel QX1511-Magliacane11 10 kHz of the frequency standard’s enclosure. 10 kHz Figure 11 — This block diagram illustrates the post-detection processing. The WWVB amplitude shift keying and phase shift keying are demodulated from the I channel while the Q channel the I channel are demodulated from and phase shift keying WWVB amplitude shift keying The illustrates the post-detection processing. diagram This block Figure 11 —

QEX November/December 2015 21 Time Out Out Out Code BPSK Modulation 13 1 2N3904 Q6 QX1511-Magliacane12a R63 10 kΩ A C +5 V U26A B CD4066B 2 7 R142 10 kΩ R62 +12 V 100 kΩ +12 V R61 4 5 12 11 D10 1.5 kΩ A A C C Time Code 8 U26B U26C B B CD4066B CD4066B 3 14 10 U7C LMC6484N +12 V 68 kΩ 9 1 10 LMC6482N U25A 7 4 8 R58 +12 V U25B +6 V R140 LMC6482N 100 kΩ 2 3 R60 R59 6 5 100 kΩ 130 kΩ +6 V 47 kΩ R141 7 R137 R138 100 kΩ +6 V 100 kΩ U7B LMC6484N 14 6 5 R139 100 kΩ R55 U7D 680 kΩ LMC6484N 12 13 C27 4.7 μF R57 4.7 kΩ C26 0.1 μF C25 R54 330 kΩ 6 8 7 14 11 0.068 μF R53 16 330 kΩ U10A +12 V CD4053BE 14 12 13 R52 U3D 680 kΩ LMC6484N R56 10 MΩ 12 13 +6 V C24 0.1 μF C23 R51 330 kΩ D9 1N914 0.068 μF LMC6484N U7A 1 R50 4 11 330 kΩ +12 V " 2 3 I In " Noise Gate In Channel

22 QEX November/December 2015 Controlling The VCTCXO After passing through analog switch U2B, the out- Voltage Out Out MHz Error 10 put of the “Q” channel demodulator is fed through a 0.025 Hz wide second order low-pass filter (U3B) before being applied to the 10 MHz VCTCXO. Low

8 leakage, metalized polypropylene film capacitors were employed at C28 and C29. QX1511-Magliacane12b 7 14 The resistive network between the low-pass filter U12

+5 V and the VCTCXO exhibits 21 dB of attenuation, and CMOS 10 MHz VCTCX0 controls the VCTCXO tuning sensitivity and the maxi-

1 mum loop gain of the frequency standard. The loop gain setting is fairly critical, since a third-order phase-locked loop (PLL) function is produced by the VCTCXO and the second order low-pass filter that precedes it. Third- order PLLs offer superior noise rejection and lower C30

0.1 μF steady-state errors than second order PLLs, but they do R76 10 kΩ so with a reduced phase margin. Since the VCTCXO 2.0 MΩ R77 will be generally close to its target frequency, accept- able overall loop stability is achieved by simply keeping the PLL loop gain no higher than that necessary to elec- tronically steer the VCTCXO to exactly 10 MHz over a R74 CCW CW limited tuning range. If greater amounts of steering volt- 100 kΩ

+6 V age are required, a front panel mounted 10 turn potentiometer (R78) is available to allow manual fine tuning of R78 50 kΩ +6 V the oscillator. Switch SW1 allows manual tuning to take Manual Fine Tuning place either with (PLL Closed) or without (PLL Open) R73 R75 56 kΩ 56 kΩ there being a tuning influence from WWVB. Changing SW1 to a DPDT switch and adding a red LED and an appropriate current limiting resistor can be advanta- R72

270 kΩ geous in providing a visual cue to the user that the PLL switch is in the open position. This frequency standard employs a Bomar Crystal 7 Company model B17025-ADMF-10.000 10 MHz VCTCXO as its master oscillator. The oscillator oper-

R71 ates on 5 V, and has a frequency stability of 2.5 ppm. LMC6484N U3B 20 MΩ It provides an HCMOS square wave output waveform, 6 5 and has a measured positive slope tuning sensitivity of 62.8 Hz/V. In addition to being voltage controlled, the C29

0.47 μF oscillator also possesses a trimmer capacitor to permit 0.33 μF C28 coarse frequency adjustments to be made independent of the tuning voltage applied to the oscillator. Since this frequency standard operates the oscillator with a nomi- nal tuning voltage close to 3.0 V rather than the 2.5 V R68 10 MΩ

+6 V (+Vcc/2) typically expected in a 5 V oscillator, the trimmer allows oscillator adjustment to 10 MHz with this R70 R69 1 kΩ 1 kΩ higher tuning voltage applied. Since the PLL loop gain in this frequency standard R67 10 MΩ is intentionally low, and some dependence on manual tuning is sometimes required, a more stable oscillator would make manual tuning adjustments far less critical R66 R65 10 kΩ 10 kΩ when changes in ambient temperature take place. MHz oscillator to track the phase of the WWVB carrier. the phase of the 10 MHz oscillator to track that forces the tuning voltage develops while the Q channel 15 LO and Standard Frequency Generation Figure 13 illustrates the circuitry used to buffer the 10 10 MHz VCTCXO for external applications, and divide

U2B it down in frequency for both internal and external use. SW1 Closed PLL U18, a Microchip MCP101-460HI/TO active high +12 V

1 supervisory circuit, holds the active high CLEAR inputs 2

R64 of every 74HC390 dual decade counter high for approx- +6 V PLL Open 100 kΩ imately 350 ms after the frequency standard is first "

In powered on. In addition, U15A and U15B at the far end Q " of the divider chain are reset independently of the other Figure 12 — This schematic diagram shows the details of the I and Q channel filtering and processing. Both amplitude and phase shift keying signals are demodulated from the I channel, channel, the I signals are demodulated from Both amplitude and phase shift keying filtering and processing. the details of I and Q channel shows diagram This schematic Figure 12 —

Channel dividers by the microcontroller in alignment with the

QEX November/December 2015 23 beginning of the UTC second, shortly after tion process that might take on the appear- to-digital converter that is used to measure the frequency standard is first powered on. ance of noise to the circuitry that follows. the relative WWVB-derived error voltage fed Buffered TTL level outputs at 10 MHz, The low-pass filter feeds a precision to the VCTCXO. This voltage is represented 1 MHz, 100 kHz, 50 kHz, 25 kHz, 10 kHz, full-wave rectifier designed around op- as a three digit number between –512 and 1 kHz, and 100 Hz are available through amps U20A and U20B. The rectifier feeds +511, and is continuously displayed on the U17, a 74HC151 multiplexer. Frequency negative-going noise impulses to U22, an bottom center of the LCD. Proper adjustment selection takes place through a three bit LM555 monostable multivibrator. The trig- of the VCTCXO through R78 is achieved binary code applied to the multiplexer data gering threshold for the LM555 is set through when a stable reading close to 000 is made. select lines. An eight position BCD switch, potentiometer R97, and is adjusted so that the The PIC A/D converter has a resolution or simply a three pole eight position rotary LM555 triggers only when strong lightning of 6 V/1024 or 5.86 mV. Since the error volt- switch wired with the appropriate ground discharges are detected. Once triggered, the age is attenuated by a factor of 11.3 (21 dB) connections and pull-up resistors can be used LM555 lights D13, an amber colored LED, before it reaches the VCTCXO, but is read to select the output frequency. Two additional to indicate noise detection. The 555 also by the PIC prior to this attenuation, the tuning buffered 10 MHz outputs are also provided. opens the analog switches associated with indicator is able to resolve a 518 mV change Figure 14 illustrates the circuitry associ- the AGC (U2A) and peak time code detec- in the VCTCXO error voltage. ated with the microcontroller, back-lit LCD tion (U10A) circuitry to prevent their reac- tion to the static crash. The rectified noise alphanumeric display, RS-232 port, and Power Up Procedure several regulated voltage references and voltage also drives a front panel meter to Once powered on, the frequency standard sources used by the frequency standard. The provide indications of its relative intensity. sequences through several modes of opera- PIC16F88 (U11) accepts the WWVB AM The 200 Hz low-pass filter also provides tion before it is finally ready for use. The time code, the WWVB BPSK time code, an audio signal to the LM380 audio ampli- setup process takes several minutes to com- a 10 Hz timing signal from the frequency fier to permit aural monitoring of any 60 kHz plete, and the LCD keeps the user informed dividers, filtered PLL error voltage derived background noise that could influence recep- of the process along the way. from the “Q” demodulator, and a DC refer- tion quality. U19, an LMC567 tone decoder The display illustrated in Figure 17A ence voltage that is exactly half the +6 V PLL, monitors the 10 kHz IF and responds appears when the frequency standard is first used throughout the frequency standard as to the absence of the WWVB carrier. The 3 powered on. The microcontroller firmware a virtual ground. The microcontroller gen- LMC567 lights D14, a red colored LED, if version is briefly displayed on the first line, erates the reset pulse that clears 74HC390 a loss of signal condition is detected. It also while the VCTCXO digital tuning indica- decade counters U15A and U15B as previ- holds the LM555 in reset mode, and prevents tor permanently appears in the middle of the ously described. It also generates the signals it from lighting the noise LED and open- line below. necessary for driving the legacy ±45° RF ing the noise-gated analog switches under After several minutes have passed and the phase shift networks in the front end of the a loss of signal condition, or during the first frequency standard VCTCXO has become frequency standard, the 24 × 2 LCD, and few seconds after power-up while the AGC locked in phase with that of the WWVB car- provides serial UTC date and time informa- becomes fully acclimated to the WWVB rier, the microcontroller begins timing the tion to peripheral devices through a Dallas signal level. DS232A RS-232 level converter (U9). interval between each WWVB amplitude Figure 15 illustrates the audio, loss of carrier reduction. Carrier reductions occur at signal, noise detection, and noise mitigation Multimeter and Sinusoidal Outputs the start of each UTC second, and once eight circuitry. A DC voltage from the “I” chan- Figure 16 illustrates the circuitry respon- successive properly timed carrier reductions nel demodulator that is proportional to the sible for providing sinusoidal output signals have been detected, the display switches WWVB modulation amplitude is applied to and analog meter indications. A fourth order to that illustrated in Figure 17B. When this analog switch U2C, where it is modulated 1 kHz bandpass filter designed around op- change occurs, the microcontroller begins at a 1 kHz rate to produce an amplitude amps U20C and U20D filter the 1 kHz square looking for the Marker bit sequence, identi- modulated audio tone. A narrow bandpass wave local oscillator signal into a sinusoidal fying the conclusion of the current time code filter following the modulator attenuates the waveform. This waveform is amplified by frame and the beginning of the next. level of harmonics contained in the 1 kHz U23, an LM386 audio power amplifier that While waiting for the next frame to begin, switching carrier. The resulting sinusoidal provides sufficient output current to drive a the display indicates the phase of the WWVB waveform is applied to an audio select switch low impedance load. carrier (+ or –) and the identity of each time before being made available to an LM380 Op-amp U6D buffers a sample of the code bit (0, 1, or M) received over the previ- audio power amplifier. 10 kHz IF, U1D buffers a sample of the ous second. If the bit cannot be identified, a 60 kHz RF signal, and together these signals “?” character is displayed. Noise Detection and Mitigation are made available for external use. The gain Since the WWVB carrier level is always Circuitry of each buffer has been tailored to produce “low” for the first 200 ms of every second, While the “I” channel contains a DC volt- RF and IF output samples of relatively equal and always “high” for the last 200 ms of age proportional to the WWVB modulation amplitudes. every second, these intervals carry no time amplitude, the output of the “Q” channel SW3, a two pole four position rotary code information and are ignored by the (after BPSK-correction) is modulation free. switch allows relative measurements of microcontroller bit correlation decoding Therefore, any rapid modulation of the DC modulation level, peak signal level, center algorithm. These predictable amplitude lev- voltage present on the output of the “Q” tuning, and ambient radio noise levels to be els are instead used to estimate the quality of demodulator is the result of noise energy made through a single 100 µA D’Arsonval time code reception, which is displayed as a present at 60 kHz. The circuitry surrounding panel meter. single digit between 0 (poor) and 8 (excel- transistor Q7 forms a 200 Hz wide low-pass lent) to the right of the time code bit. Once the beginning of the frame is found, filter that rejects any high frequency energy The Liquid Crystal Display the display switches to that shown in Figure remaining from the “Q” channel demodula- The PIC16F88 contains a 10 bit analog- 17C. The microcontroller starts a process

24 QEX November/December 2015 8 STD Output Output Output 10 MHz 10 MHz 50 kHz Out 10 Hz Out 1 kHz Out 10 kHz Out J6 J7 J5 U24D 74HC14 9 74HC14 74HC14 U24E U24F 9 10 11 13 12 10 QB QA QD QC U15B 74HC390 CLR CKB CKA 11 13 2N3904 Q9 12 14 15 +5 V 6 5 3 7 QB QA QD QC 4 6 8 12 10 U15A R128 47 kΩ 74HC390 CKB CLR CKA U21F 74HC14 1 2 4 3 9 5 11 13 9 10 11 13 4 6 U21B U21E U21D U21C 1 kΩ R127 74HC14 74HC14 74HC14 74HC14 QB QA QD QC U14B 3 5 74HC390 CKB CKA CLR 2 12 14 15 U24B U24C U21A 74HC14 74HC14 74HC14 1 9 10 11 13 74HC151 U17

6 5 3 7 QB QA QD QC

W D0

4 6

QB Y QA QD QC U16B

74HC390 3

CKA D2 CLR CKB

U14A 2

74HC390 D3

12 14 15 CLR 1 CKB CKA

D4 G

1 2 4 7 15

C D5 2

14 9

6 5 3 7 B D6

10 13

D7 U24A

9 QB 10 11 13 QA QD QC 11 12 74HC14 SW4 1 BCD Switch QB QA QD QC U16A 74HC390 CKB CKA CLR +5 V U13B 1 MHz signals for use in the laboratory. and 10 MHz signals for 1 MHz, 100 kHz, 50 kHz, 25 kHz, 10 kHz, 1 kHz, selection of 100 Hz, 2 4 74HC390 CLR CKA CKB 12 14 15 1 kHz, 1 MHz, 100 Hz 50 kHz, 25 kHz, 10 kHz, 6 5 3 7 Selector 10 MHz, 100 kHz, +5 V Output Frequency QB QA QD QC U13A GND +Vdd Reset 74HC390 U18 CLR CKA CKB MCP101 460HITO In 1 2 4 QX1511-Magliacane13 Divider 10 MHz MHz VCTCXO is divided and buffered to provide LO injection for every mixer and synchronous demodulator in the frequency standard. A rotary switch allows user allows switch A rotary standard. demodulator in the frequency and synchronous mixer every LO injection for to provide VCTCXO is divided and buffered The 10 MHz Figure 13 — Reset In

QEX November/December 2015 25 QX1511-Magliacane14 +5 V +5 V U11 24X2 LCD Display 14 +VDD 2 +VDD C71 6 11 0.1 μF RB0 DB4 R135 7 12 +5 V 5 RB1 DB5 24 Ω 13 13 A RB7 DB6 +LED R129 R136 9 14 3 Contrast 10 kΩ RB3 DB7 VO BPSK 4 10 4 K 10 kΩ Control RA5 RB4 RS –LED In 12 6 5 R130 RB6 E R/W 1 6.8 kΩ VSS

Time 17 Code RA0 Divider In 3 RA4 Reset 10 Hz 18 Out RA1 In 15 RA6 A Out Error 1 Voltage RA2 16 In R131 +6 V R132 C72 RA7 B Out 4.7 kΩ 4.7 kΩ 0.1 μF

R133 J8 4.7 kΩ 2 U9 RA3 8 12 14 1 R134 C73 C74 RB2 6 11 11 13 2 4.7 kΩ 10 μF 0.1 μF RB5 7 3

+5 V 8 PIC16F88 4 9

16 5

19.2 kbps C75 C76 1 RS-232 Out +12 V 0.1 μF 0.1 μF Source 2 C79 U29 6 3 0.1 μF

VI VO 4 C78 GND C77 C82 0.1 μF 0.47 μF 15 5 0.1 μF LM7805

DS232A

+5 V U30 Source +6 V VI VO Source C82 GND C84 0.47 μF 0.47 μF LM78L06

F1 Bourns J9 SW5 U28 RX110 +13.8 V +12 V VI VO In Source C81 C80 D15 GND Power 47 μF On / Off 0.47 μF 1N4004 LM2940T–12.0 Low ESR

Figure 14 — A PIC microcontroller decodes the WWVB time code, functions as a real-time UTC clock driven by the disciplined oscillator, and provides date and time of day information via the liquid crystal display and RS-232 serial port.

26 QEX November/December 2015 Voltage Out Out Gate Noise Noise Speaker 0.1 μF C42 Noise D13 C43 2.7Ω R87 C49 1 μF R99 R100 1 MΩ 1.5 kΩ 1000 μF +12 V QX1511-Magliacane15 6 3 7 8 C41 1 μF 0.01 μF C39 Q 6 DIS THR 8 1 3 14 U22 +12 V LM555 2 7 +12 V TR CV Reset Noise Gate Threshold 2 4 5 0.01 μF C48 C38 1 0.01 μF +12 V U8 C40 R86 CCW 25 kΩ R96 100 kΩ Volume 4.7 μF LM380 R97 10 kΩ C37 0.1 μF 2N3904 Q8 CW C47 R98 0.47 μF 200 kΩ +12 V 7 SW2 Noise Signal +6 V D14 R105 Audio Select 1.5 kΩ Low SNR U20B R95 200 kΩ LMC6484N 6 5 C36 R85 47 kΩ 0.01 μF 1 R94 LMC6484 U3A 100 kΩ 4 +6 V 11 +12 V D12 R84 LMC6484N U20A 1N914 2 3 390 kΩ 1 D11 +6 V 1N914 4 11 C33 0.01 μF R83 C51 15 μF C35 +12 V 0.1 μF 390 kΩ C34 R93 0.01 μF +5 V 100 kΩ 2 3 R102 56 kΩ R51 R91 R92 680 Ω R81 47 kΩ 200 kΩ 100 kΩ +6 V C52 47 μF C46 1 μF R103 10 kΩ C32 R80 47 kΩ 0.01 μF 4 4 8 5 1 +12 V 9 R90 U19 U2C Q7 10 kΩ LMC567 2N3904 2 6 3 7 3 5 C45 R89 56 kΩ R101 0.01 μF 160 kΩ R79 R104 30 kΩ 100 kΩ C50 0.01 μF C44 R88 C31 56 kΩ C54 C53 0.1 μF +6 V 0.022 μF 0.001 μF 0.047 μF In In In "Q" detected on the Q channel trigger a noise gate that temporarily inhibits AGC action and time code detection. Noise energy can also be monitored through the audio amplifier. can also be monitored through Noise energy action and time code detection. inhibits AGC trigger a noise gate that temporarily detected on the Q channel IF In 1 kHz 10 kHz Channel Modulation kHz carrier for aural monitoring of the signal. Noise impulses aural monitoring of the signal. WWVB signals modulate a 1 kHz carrier for Baseband the audio and noise signal processing. for diagram Figure 15 — Here is the schematic

QEX November/December 2015 27 in which it begins decoding the frame and Once the validation process is complete, approach, much of the circuitry was built evaluating the integrity of the data being the frequency standard begins sending the on a series of 95 × 70 mm and 70 × 45 mm collected. A countdown timer representing current UTC time and date, once every sec- perforated circuit boards that have all been the number of seconds until the completion ond, to any connected peripherals via the interconnected to one another to form a of this process is displayed on the top right RS-232 port (Figure 18). These peripherals complete unit. Figure 19 is a view inside the hand side of the display. If the data collected might include a PC with a real-time clock cabinet of my unit. With the exception of the looks reasonable, a real-time clock/calendar that can be set through appropriate software, DC power supply, antenna, and remote RF operating within the firmware of the micro- or an external digital clock display. While preamplifier, all circuitry is housed in a single controller is set to the time and date decoded. this frequency standard is not intended to Ten-Tec model BK-1249 enclosure that mea- The microcontroller then looks for vali- serve as an NIST-traceable time source, sures 12 inches × 4 inches × 9 inches (HWD) dation of the information received by exam- the date and time reported through both the Figure 20 is a photo of the front of the unit. ining the next frame of data, and compares serial port and the LCD are advanced by In an effort to enhance frequency stabil- the result with that of the real-time clock one 100 ms to compensate for the nearly equal ity, thermal effects caused by heat dissipa- minute later (Figure 17D). If the received amount of signal processing delay inher- tion of the frequency standard electronics are time and date match that of the clock, then ent within the electronics in the frequency minimized by keeping the DC power supply the microcontroller begins displaying the standard. physically removed from the enclosure, and locally running clock and calendar from that by mounting both the LM2940T-12.0 and the point forward. See Figure 17E. If the valida- LM7805 voltage regulators to the enclosure’s Parts and Construction tion fails, then the process reverts to the point back panel. The greatest single sources of This frequency standard was developed illustrated in Figure 17C, in which the recep- heat outside of the voltage regulators are the and tested in discrete stages over a period tion and validation routines are repeated LCD backlight, and interestingly enough, the of several years. Due to this modular design until the current time and date are finally VCTCXO, itself. confirmed.

C56 C59 0.01 R108 +12 V 0.01 R112 390 k C62 1 kHz R106 C55 390 k 1 kHz 13 100 μF Sinusoidal R110 C58 In 14 9 Output 150 k 0.01 U20D R114 C61 LM386N 8 2 R107 150 k 0.01 U20C 6 C64 J2 LMC6484 680 12 5 R111 100 k 0.047 LMC6484 U23 C57 R109 680 10 10 k 1 kHz R116 220 μF +6 V C60 4 R113 Sinusoidal R115 3 10 R117 0.1 390 k +6 V Output Level C63 1 k 0.1 390 k 0.1

1: Modulation R123 2: Peak Signal 3: Center Tuning 100 k 4: Noise Level Modulation In 1 C65 R121 LM837N 2 13 S3A Peak Signal In 10 kHz IF In J3 R118 C67 10 kHz IF 3 14 "Q" Channel In 0.01 1 M U6D Sample + 4 120 k 0.01 Output M1 +6 V 12 100 μA μA R120 R122 2.2 k C66 +6 V 100 k − 1 24 k 0.01 S3B 2 +6V 3 R126 4 R119 Noise Voltage In 1 M 6.8 k C68 R124 LM837N 13 J4 60 kHz RF In C70 60 kHz RF 14 100 k Decimal values of capacitance are in microfarads (µF); 0.01 U1D Sample others are in picofarads (pF); Resistances are in ohms; 0.01 Output k=1,000, M=1,000,000. 12 R125 C69 1 M 0.01 QX1512-Maglicane16 +6V

Figure 16 — Metering and various sinusoidal output signals are shown on this schematic diagram. In addition to providing buffered 60 kHz RF and 10 kHz IF samples, a precise 1 kHz sinusoidal waveform capable of driving a small speaker is provided.

28 QEX November/December 2015 Next, with the antenna and preamplifier Alignment and Testing Table 1 disconnected from the frequency standard, The frequency standard may be powered Equations that relate the voltage pro- switch SW1 to the “PLL Open” position, from any 13.8 V DC power source capable duced by an electrically small (<0.08 λ) and adjust the coarse frequency adjustment of supplying at least 300 mA of current. A air-core loop antenna to the local field capacitor on the VCTCXO until it operates Bourns RX110 resettable fuse (F1) located strength given the loop’s physical as close to 10 MHz as possible. Being able to dimensions and electrical properties. after the power switch is followed by a obtain a close zero-beat with WWV should reverse biased silicon diode to ground (D15). 2πeNA be more than adequate at this stage of the = These are used as safety measures to help Es alignment. λ to protect the circuitry from damage should With the Audio Source Switch (SW2) in an over-current or reverse polarity condition Esλ the “Noise” position and the 60 kHz function e = ever occur. generator off, connect the antenna and pre- 2π NA The antenna can be adjusted to achieve amplifier to the frequency standard. Carefully resonance at 60 kHz by monitoring the fr adjust potentiometers R36 and R39 simulta- Q = output of U1 (AD620 Pin 6) with an oscil- neously for maximum noise level. bw loscope while radiating a weak signal at = 60 kHz from a nearby function generator and E0 QEs varying the amount of capacitance in parallel Operation Where: with the loop to achieve maximum response. With the plane of the loop antenna Es = voltage induced into the loop (mV). Table 1 provides equations for determining oriented in the direction of Fort Collins, E0 = loop output voltage at resonance (mV). the voltage produced by the loop given its Colorado, final alignment of the VCTCXO fr = loop resonant frequency (Hz). physical dimensions, electrical properties, can take place. Manual coarse tuning can be bw = 3 dB bandwidth (Hz) Q = the Q of the loop. and local field strength. achieved by placing SW1 in the “PLL Open” e = local field strength m( V/m). Receiver alignment should begin by first position, and Meter Function Switch (SW3) N = number of turns. adjusting the Manual Fine Tuning Control in the “Center Tuning” position, while slowly A = enclosed area of the loop (square meters). (R78) until the DC voltage between its wiper adjusting the Manual Fine Tuning control λ = wavelength (meters). and ground reads exactly half that of the (R78) for a steady, center reading on meter +6 V reference. M1.

Figure 17 — These images show the various LCD screens depicting each of the five stages Figure 18 — At the beginning of every of frequency standard operation following power up. The final stage is where the UTC date second, the 19.2 kbps RS-232 serial port and time are continuously displayed and made available to peripheral equipment via the provides the current UTC time and date in the RS-232 port. form of HH:MM:SS MM/DD/YY followed by a line feed and carriage return.

QEX November/December 2015 29 Figure 19 — The frequency standard was designed using a modular approach that employed separate perforated circuit boards for various device functions. This approach provided a convenient way to add circuitry as the design evolved over time.

Figure 20 — Front view of the frequency standard in operation. The large knob on the upper left permits manual fine tuning of the VCTCXO, while the display provides the UTC date and time.

30 QEX November/December 2015 Placing the audio source Radio Club since 1991, and as an switch (SW2) in the “Signal” Academic Tutor in the Computer position will help verify that Science Department. John has WWVB reception is taking worked as a freelance techni- place. If all has gone well, cal writer for over 20 years, and placing SW1 in the “PLL authored weekly “SpaceNews” Closed” position should allow newsletters during the 1990s that the VCTCXO to slowly move gained world-wide popularity into phase alignment with the among the terrestrial packet radio networks and pacsat satellites that WWVB carrier. Minor adjust- carried them. ments of the VCTCXO can be John has been a Slackware made at this point to bring the Linux user for over 20 years, and LCD tuning indicator reading to has created and contributed to a as close to 000 as possible. number of open-source software After several minutes of projects. His “PREDICT” satel- reception have transpired, the lite tracking and “SPLAT!” RF LCD sequence illustrated in propagation analysis applications Figure 17 will begin to take have not only earned strong fol- place. If high local noise levels lowings in the Amateur Radio and impair reception of the WWVB commercial telecommunications time code, the antenna may need fields, but have also been adopted to be reoriented to place the for use by scientists and engineers noise source in one of the two at NASA and the European Space antenna pattern nulls. Placing Agency. the Meter Function Switch in In addition to being a Frequency Measurement Test par- the “Noise Level” position will ticipant who employs receiving permit the relative ambient noise equipment and instrumentation level to be displayed on the entirely of his own design, John meter. Placing the Audio Source recently published the design of Switch in the “Noise” position Figure 21 — The ability to receive and successfully identify distant radio signals remains a source of pride for many radio enthusiasts. his “TriplePIC SSTV Video Scan will allow the noise level to be While formal confirmation of GPS satellite reception may be not be Converter” that allows the explo- monitored through aural means possible, radio station WWVB offers QSL cards to validate reception ration of vintage 8 second per without having to keep an eye on reports. frame monochrome slow-scan the meter for visual cues. television using twenty first cen- standard can provide a reliable sanity check tury electronics. Summary as well as a redundant backup. Notes While some may question the merit of The recent changes made to the WWVB 1Michael A. Lombardi, Glenn K. Nelson, employing WWVB as a frequency reference broadcasts by the National Institute of “WWVB: A Half Century of Delivering at a time when GPS-disciplined frequency Standards and Technology have been unset- Accurate Frequency and Time by Radio,” tling for some individuals. What these Journal of Research of the National Institute standards are so ubiquitous, similar ques- of Standards and Technology, Volume 119, tions could be raised about the relevance of actions reveal, however, is that while the The National Institute of Standards and the Amateur Radio Service in a world domi- WWVB primary role is changing, it is Technology, March 12, 2014. nated by cellphones and the Internet. The changing because its use is growing, and 2J. A. Adcock, VK3ACA, “Propagation of Long frequency standard described here employs a this growth will help ensure there is strong Radio Waves,” Amateur Radio, June to September 1991. “purely RF” approach toward disciplining a support for keeping WWVB on the air for decades to come. See you in the next FMT! 3Firmware for the PIC16F88 microcontroller local oscillator against an extremely accurate is licensed under the GNU General Public national atomic standard. It was developed not License and is available for download only to create a laboratory grade frequency John A. Magliacane, KD2BD, has held an from the ARRL QEX files website. Go to standard, but to do so while pursuing a life- Advanced Class license for over 31 years and www.arrl.org/qexfiles and look for the file 11x15_Magliacane.zip. long interest and fascination with the underly- a Commercial FCC Radio License since 1994. John holds Associate Degrees in Electronics ing radio concepts that make such a process Additional References Engineering Technology, Computer Science, possible. Figure 21 shows a reception QSL and Mathematics/Physics, in addition to a “NIST Radio Station WWVB,” The National that I received from WWVB in 2003. Bachelor’s Degree in Electronics Engineering Institute of Standards and Technology, While not a state-of-the-art device by Technology. www.nist.gov/pml/div688/grp40/wwvb. twenty first century standards, the frequency cfm John is employed at Brookdale Community “ARRL (and non-ARRL) Frequency reference described here will likely pro- College, Lincroft, NJ where he has served Measuring Tests”, The American Radio vide more than adequate performance for as a Learning Assistant in the Department Relay League, www.arrl.org/frequency- many modern engineering, research, and of Engineering and Technology for over 27 measuring-test/ scientific purposes. For those possessing years, as an advisor to the Brookdale Amateur John A. Magliacane, “KD2BD FMT Methodology,” www.qsl.net/kd2bd/fmt- GPS-disciplined standards, this frequency methodology.html.

QEX November/December 2015 31