A Three-Phase Delta Switch Rectifier for More Electric Aircraft Applications Employing a Novel PWM Current Control Concept M. Hartmann, J. Miniboeck and J. W. Kolar Power Electronic Systems Laboratory Swiss Federal Institute of Technology Zurich, Switzerland Email: [email protected] Abstract— In the course of the More Electric Aircraft program DF+ active three-phase rectifiers in the power range of 5 kW are required. Vo / 2 DN+ A comparison with other rectifier topologies shows that the three- S1+ phase ∆-switch rectifier (comprising three ∆-connected bidirectional switches) is well suited for this application. The system is analyzed D S1 using space vector calculus and a novel PWM current controller N concept is presented, where all three phases are controlled simultane- Vo / 2 ously; the analysis shows that the proposed concept yields optimized DF switching sequences. To facilitate the rectifier design, analytical relationships for calculating the power components average and rms L1 current ratings are derived. Furthermore, a laboratory prototype with an output power of 5 kW is realized. Measurements taken EMCinputfilter from this prototype confirm the operation of the proposed current controller. Finally, initial EMI-measurements of the system are also V1 V2 V3 presented. (a) I. INTRODUCTION In modern aircraft the trend is to replace hydraulically driven D1p actuators for flight control surfaces, such as rudder and aileron, V L1 1 vr1 by Electro Hydrostatic Actuators (EHA) in order to implement S12 V L2 the “More Electric Aircraft” concept (MEA) [1]-[3]. The EHA 2 vr2 S23 Vo units are connected to the aircraft power system by three-phase V L3 v 3 r3 S rectifiers with typical specifications as listed in TABLE I. The 31 EMCinputfilter D MEA-concept in general calls for a reduction in the size and 1n weight of the electrical systems. A major issue is the weight (b) reduction of the power generation system by eliminating the generator gearbox, which however will result in a variable mains Fig. 1: Active three-phase rectifiers suitable for aircraft applications; (a) three-level 6-switch Vienna-type rectifier and (b) two-level ∆-switch rectifier. frequency of 360 Hz. 800 Hz. Additional, the electronic systems must have very high reliability, i.e. the loss of one phase must tion losses since there are always two semiconductors connected not result in an outage of the rectifier system. Furthermore, the in series per phase. Two-level three-phase rectifier topologies may loads are not allowed to feed back energy into the mains and show better efficiencies but suffer from higher semiconductor therefore unidirectional rectifiers have to be used. Due to the voltage stress. very rigorous current harmonic limits of present airborne system standards, PWM-rectifiers with low THD of the input current and For the desired output voltage level of Vo = 400 VDC, high for a high total power factor are required. efficiency switches (CoolMOS) with a blocking voltage of 600 V In [4] it has been shown that the 6-switch three-level Vienna- and RDSon < 100 mΩ are commercially available. Hence, a type rectifier topology (cf. Fig. 1(a)) [5] is very well suited for reduction of the voltage stress, as given by three-level topologies aircraft applications. This topology’s main feature is a reduced like the Vienna-Rectifier concept, is not needed. Several two- semiconductor voltage stress, especially of importance for high level three-phase rectifier topologies are presented in the literature output voltage levels. However, the trade-off is increased conduc- and a comparative study can be found in [6]-[7]. The application of a standard six-switch PWM-rectifier bridge is not favourable because of its bidirectional power flow behavior. Additional draw- TABLE I: Typical specifications of active three-phase rectifiers in aircraft appli- cations. backs are the reduced reliability because of possible shoot-through of a bridge leg, resulting in a short circuit of the DC-voltage, the VN,i 115 V ± 15% fin 360Hz ...800Hz high current levels of the semiconductors and the involvement of Vo 400 VDC the MOSFET body diode, causing a substantial limitation of the Po 5 kW . 10 kW switching frequency. In [9] a topology using either Y-connected or ∆-connected (cf. Fig. 1(b)) bidirectional switches on the AC-side Sij Sji i j (a) (b) Fig. 2: Possible realizations of a bidirectional (current), bipolar (voltage) switch using (a) two MOSFETs and (b) a MOSFET and a diode bridge. is presented. In general, the Y-connected realization shows higher conduction losses as compared to the ∆-connected alternative, since there are always two (bidirectional) switches connected in series. A short-circuit of the DC-voltage is not possible with both topologies. Fig. 2 shows two possibilities of realizing the bidirectional (current), bipolar (voltage) switches. The original Vienna-Rectifier topology uses the bidirectional switch shown in Fig. 3: Space vector diagram of the ∆-switch rectifier for the sector ϕN ∈ Fig. 2(b) instead of the two MOSFETs per phase-leg shown in [−30◦, +30◦]. Fig. 1(a). An elegant topology, which integrates the bidirectional switch of Fig. 2(b) into the diode bridge, is presented in [10]-[11]. calculated by However, the conduction losses of this realization are higher than 2 2π for the realization using two MOSFETs (Fig. 1(a)). Also, some 2 j 3 v = (vr1 + avr2 + a vr3) with a = e . (1) topologies using quasi tri-directional switches [12] or topologies r 3 operating in discontinuous conduction mode were presented [13]- The possible converter voltages vri are dependent on the state [14]. The topology using tri-directional switches increases the of the switches sij (sij = 1 denotes the turn-on state of system complexity and discontinuous-mode topologies cannot switch Sij ) and on the direction of the input phase currents fulfil the requirements on the total harmonic distortion. Due to iNi. Therefore, the available voltage space vectors change over ◦ its low complexity, low conduction losses and high reliability every 60 of the mains frequency. If (s12,s23,s31) describes the the ∆-switch rectifier topology seems to be an optimal choice different switching states, the resulting voltage space vectors for ◦ ◦ for realization of a rectifier for aerospace applications with the ϕN [ 30 , 30 ] (iN1 > 0, iN2 < 0, iN3 < 0) can be calculated requirements in TABLE I. as ∈ − 2 (000), (010) : v = V (2) Besides efficiency and power density, control issues also in- r1 3 o fluence the practical applicability of the circuit topology. Several 2 − ◦ (001) : v = V e j60 (3) possibilities for the control of three-phase rectifiers exist and a r2 3 o survey of these methods can be found in [15]. A control method 2 ◦ (100) : v = V ej60 (4) based on low switching frequencies is given in [16] but cannot be r3 3 o used for the desired application because of the high AC current (011), (101), (110), (111) : vr4 = 0 (5) harmonics. A hysteresis controller as shown in [17] would be an easy way to control the rectifier system, but its varying switching (cf. Fig. 3). frequency may increase the effort of EMI-filtering. A controller Only states (000), (001), (010) and (100) show a non-zero using the one-cycle control method is presented in [18], but there magnitude and the voltage space vector of (010) is equal to the ◦ the controller structure has to be changed over every 60◦ and space vector for state (000). In each 60 -sector there is a redun- the input current control is always limited to 2 phases. In [19]- dancy of the (000)-vector and therefore only 4 different voltage [20] a PWM-control method for the rectifier system is proposed, space vectors can be generated by the converter in each sector. however, no information was given about the exact switching These discrete voltage space vectors are used to approximate the sequence of the switches, which mainly influences the efficiency converter’s voltage reference vector of the rectifier system. ∗ ∗ v = V ejϕvr , ϕ = ω t (6) In this work a novel PWM-control method using triangular carrier r r vr N signals is presented where all three phases are controlled simulta- in the time average over the pulse-period. In conjunction with the b neously. The resulting optimal switching sequences are analyzed mains voltage system by application of space vector calculus. Additionally, the proposed jϕvN control method is able to handle a phase loss without changing vN = V N e (7) the controller structure. the voltage difference b ∗ ∗ di II. SYSTEM OPERATION v v = L N (8) N − r dt The three switches Sij (i, j 1, 2, 3 ) of Fig. 1(b) are used leads to the input current ∈ { } to generate sinusoidal input currents which are proportional to the ∗ ∗ jϕiN mains voltage. The discrete converter voltage space vector can be iN = IN e , (9) b N vNi sector LNi detection 3 i clamp Ni v ij Ni i Ni vNi pwmij i* v*rNi v*r,ij Ni sij KI() s & AC v*r,ji sji carriersignal 1 & DC pwmji clamping g* v e F() s o v*o vo Fig. 4: Structure of the proposed PWM-current controller. Signal paths being equal for different phases are shown by double lines. S12 Therefore, the idea is near at hand of controlling these ∆-related t vN1 L i N1 N1 currents. Unfortunately, this is not very convenient because of the S23 v L t N2 N2 iN2 S12 necessary clamping actions caused by the large number of redun- S S Vo 31 31 dant switching states. This can be avoided if the phase currents t vN3 LN3 i (000) (001) (101) N3 t iNi of the rectifier are controlled.