NOVEL RECONFIGURABLE FOLDED-SLOT ANTENNA APPLICATIONS Thesis Submitted to the School of Engineering of the UNIVERSITY of DAYTON

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NOVEL RECONFIGURABLE FOLDED-SLOT ANTENNA APPLICATIONS

Thesis Submitted to The School of Engineering of the UNIVERSITY OF DAYTON

In Partial Fulfillment of the Requirements for The Degree of Master of Science in Electrical Engineering

By Jincheng Zhao, M.S. Dayton, Ohio May 2020

NOVEL RECONFIGURABLE FOLDED-SLOT ANTENNA APPLICATIONS

Name: Zhao, Jincheng APPROVED BY:

Hailing Yue Guru Subramanyam Advisory Committee Chairman Committee Member Assistant professor Professor Department of Engineering Management, Department of Electrical and Computer Systems, and Technology Engineering

Robert Penno Committee Member Professor Department of Electrical and Computer Engineering

Robert J. Wilkens, Ph.D., P.E. Eddy M. Rojas, Ph.D., M.A., P.E. Associate Dean for Research and Innovation Dean, School of Engineering Professor School of Engineering

Jincheng Zhao

2020

ABSTRACT

NOVEL RECONFIGURABLE FOLDED-SLOT ANTENNA APPLICATIONS

Name: Zhao, Jincheng University of Dayton

Advisor: Dr. Hailing Yue

Demands for self-sustainable energy sources are rising as we become more and more reliable on electronic devices in our daily lives. Scientists and engineers have been exploring various novel methods to harvest energy from existing resources in order to eliminate or reduce the usage of battery and/or conventional power equipment. Solar, water, tide, wind, and terrestrial heat are renewable and green resources that have been widely adopted and commercialized[1]. With the rapid development of technology, more resources can be used for providing energy and compressing the size of devices. For example, piezoelectricity, vibration, and electromagnetic energy can also be used in the large-scale area[2]. Electromagnetic energy, especially in WIFI frequencies, is recently gaining more and more interest because of the wide signal coverage on campus and residential areas. An unique advantage of harvesting electromagnetic energy is its little dependence from weather related factors, unlike solar, water, tide, wind and terrestrial heat[3]. Given the circumstances, the interest in this work is to design a novel rectenna device to harvest energy from WIFI frequencies and to provide a parametric study in efficiency improvement.

Our comfort and fast life in modern society roots in massive volumes of data exchange through wireless transmission. In modern communication systems, different

iii radio spectrum’s only use is for single media to prevent interference between users and different devices. International telecommunication Union (ITU) established rules to allocate spectrums for various purposes; the chart [4] shows specific distributions for mobile, broadcast, satellite, and other devices. Since antenna is the only component worked as receiver and transmitter in a device, the main problem in communication systems are the versatility of antenna. So, antenna with reconfigurability is desired in today’s multi- band multi-mode communication system front end. The key solution is to widen the operating frequency band for antenna and, eventually, it can cover more radio spectrums.

The target in this work is investigating a tunable wide band antenna which can cover more frequency range.

In this thesis, there are two applications proposed and the CPW folded-slot antenna structure is served in both designs. The first one is a rectifying antenna system, or rectenna, which receives WIFI energy at both 2.45 GHz and 5.8GHz channels at a compact size of

2 . 4 7 5 1 0 −32m . An efficient Schottky diode with low build-in voltage and high reverse breakdown voltage is implemented in a half-wave rectifier which converts RF power to

DC power. A swept parametric study is performed to achieve an optimized conversion efficiency. The overall conversion efficiency is expected to reach around 30% at each resonance. Compared to similar designs from literature, this rectenna system is featured by its compact size from dual-band design and adjustable matching between receiving antenna and rectifying circuit, and ease of fabrication due to its single metal layer. Another design is a BST IDC based tunable wide band antenna which has predominant advantage in multiple wireless communication applications. The tunable antenna is designed to operate in K band with a bandwidth from 2.67GHz to 4.42GHz.

Dedicated to my mother

ACKNOWLEDGMENTS

First, I really appreciate Professor Hailing Yue, my supervisor and chair of my graduation committee, who introduced me to RF design area, trained me using simulators and basic equipment, and gave me helps in all the possible angles. Her patient instructions, rich knowledge and experience in the industry always inspired me and developed my interests. I have to say that I cannot achieve such a big progress in these several months without her advice, encouragement, and efforts.

My special thanks are also to Professor Guru Subramanyam for research direction guidance and equipment necessary for all the works I have made for this thesis. I also would like to thank to Professor Robert Penno, who become my committee member and trained me in course ECE 511 antenna.

I also want to extend my thanks to my colleagues in microwave Lab: Jinjing Li and,

Malia Harvey always supported me in courses and. Dr. Liangyu Li, who trained me operating Vector Network Analyzer.

Finally, I thank my mother and all the family members for giving me courage and confidence through my studies in America these two years.

TABLE OF CONTENTS

ABSTRACT ...... iii DEDICATION ...... v ACKNOWLEDGMENTS ...... vi LIST OF FIGURES ...... ix LIST OF TABLES ...... xi CHAPTER I INTRODUCTION ...... 1 1.1 Energy harvesting ...... 1 1.2 Previous study about rectifying antenna ...... 2 1.3 Reconfigurable wide-band antenna ...... 3 1.4 Significance of this work...... 5 1.5 Outline of this thesis ...... 5 CHAPTER II THEORY ...... 6 2.1 Scattering parameter ...... 6 2.2 3dB bandwidth and antenna bandwidth ...... 10 2.3 CPW structure ...... 11 2.4 Antenna ...... 13 2.4.1 CPW centered folded-slot antenna model...... 16 2.4.2 CPW off-centered model ...... 17 2.4.3 Equivalent circuit extraction ...... 19 2.5 Ferroelectric varactor ...... 23 2.5.1 BST material ...... 23 2.5.2 IDC structure ...... 24 2.6 Rectifier circuit ...... 26 2.6.1. Diode selection...... 26 2.6.2. Rectifier topologies ...... 28 2.6.3. RF-DC conversion efficiency ...... 30 2.7 Simulation tool ...... 30 CHAPTER III DESIGN AND SIMULATION ...... 32 3.1. Rectifying antenna...... 32

vii

3.1.1. Dual-band antenna design ...... 32 3.1.2. Rectifier circuit ...... 37 3.2. Reconfigurable wide-band antenna ...... 39 3.2.1. Wide band antenna ...... 39 3.2.2. Interdigital capacitor (IDC)...... 41 CHAPTER IV MEASUREMENT ...... 47 4.1. Measurement overview ...... 47 4.2. Test two: Power transmission ...... 51 CHAPTER V CONCLUSION AND FUTURE WORKS ...... 54 5.1 Conclusion ...... 54 5.2 Future works ...... 54 BIBLIOGRAPHY ...... 55 APPENDIX A MATLAB Code...... 60 i. Antenna equivalent electric parameter MATLAB code ...... 60 ii. Wide band antenna equivalent electric parameter MATLAB code ...... 61

viii

LIST OF FIGURES

Figure 1: General energy harvesting system architecture ...... 1 Figure 2: Diagram of rectifying antenna structure ...... 3 Figure 3: Two ports network variable.[30] ...... 6 Figure 4: 3dB bandwidth and antenna bandwidth ...... 10 Figure 5: Structure of microstrip line ...... 11 Figure 6: Structure of coplanar waveguide ...... 12 Figure 7: Microstrip patch antenna ...... 14 Figure 8: Off-centered folded-slot antenna...... 16 Figure 9: Outer circumference of antenna ...... 17 Figure 10: Inner stub ...... 18 Figure 11: Equivalent transmission line model of slot ...... 19 Figure 12: Equivalent transmission line model of inner stub ...... 20 Figure 13: Equivalent circuit of slot ...... 20 Figure 14: Smith chart for inner stub ...... 21 Figure 15: Equivalent circuit of inner stub ...... 22 Figure 16: 3D layout of MIM capacitor ...... 24 Figure 17: Top view of IDC...... 24 Figure 18: Geometry of IDC structure ...... 25 Figure 19: I-V relationship of PN diode ...... 27 Figure 20: I-V relationship of Schottky diode ...... 27 Figure 21: Schematic of half-wave rectifier circuit ...... 28 Figure 22: Schematic of voltage-doubler rectifying circuit ...... 29 Figure 23: Out circumference ...... 32 Figure 24: Inner stub ...... 32 Figure 25: Antenna structure layout ...... 34 Figure 26: Reflection coefficient of dual-band antenna ...... 34 Figure 27: Smith chart of the dual-band antenna ...... 34 Figure 28: Radiation pattern at 2.4 GHz (left) and 5.2GHz(right) ...... 35 Figure 29: Current density at 2.4GHz (left) and 5.2GHz (right) ...... 35 Figure 30: Antenna equivalent circuit...... 36 Figure 31: Equivalent circuit simulation result compare to EM simulation ...... 37 Figure 32: Half wave circuit simulation circuit ...... 37 Figure 33: Simulation of output voltage ...... 38 Figure 34: Simulation of conversion efficiency...... 38 Figure 35: Outer slot of wide-band antenna ...... 39 Figure 36: Inner stub of wide-band antenna ...... 40 Figure 37: Wide-band antenna S11...... 41 Figure 38: 3D radiation pattern at 10GHz (left) and 13GHz (right) ...... 41 Figure 39: Top view of varactor ...... 42

Figure 40: IDC S21 ...... 42 Figure 41: IDC equivalent circuit (10V) ...... 43 Figure 42: IDC equivalent circuit (0V) ...... 43 Figure 43: S21 comparison between EM structure and equivalent circuit (10V)...... 44 Figure 44: S21 comparison between EM structure and equivalent circuit (0V)...... 44 Figure 45: Cascaded antenna and equivalent capacitor ...... 45 Figure 46: Antenna-IDC 3D EM structure ...... 45 Figure 47: View of antenna-IDC connection...... 46 Figure 48: Antenna-IDC structure layout ...... 46 Figure 49: Simulation of Antenna-IDC ...... 46 Figure 50: Measurement system ...... 47 Figure 51: Measurement equipment ...... 49 Figure 52: Testing antenna. Monopole antenna (left) and fabricated FS antenna (right) . 49 Figure 53: Antenna test 1 (monopole antenna) ...... 49 Figure 54: S11 of purchased mono pole antenna ...... 50 Figure 55: Antenna Test 1 (fabricated FS antenna) ...... 50 Figure 56: S11 of designed dual-band antenna ...... 50 Figure 57: Antenna test 2 (monopole antenna) ...... 52 Figure 58: Receiving power of mono pole antenna at 2.4 GHz ...... 52 Figure 59: Receiving power of mono pole antenna at 5.2 GHZ ...... 52 Figure 60: Antenna test 2 (fabricated FS antenna) ...... 52 Figure 61: Receiving power of dual-band antenna at 2.4 GHz ...... 53 Figure 62: Receiving power of dual band antenna at 5.2 GHz ...... 53

LIST OF TABLES

Table 1: Comparison of different varactor design ...... 4

Table 2: Comparison of antenna ...... 15

Table 3: Dual-band antenna dimension ...... 33

Table 4: Dimension details of wide-band antenna(mm) ...... 40

CHAPTER I

INTRODUCTION

1.1 Energy harvesting

Energy harvesting refers to devices or systems that capture ambient energy in the environment and convert it into a useful form, which is usually electricity[2]. In other words, the purpose of energy harvesting is collecting available energy from the environment surrounding us and without any substance. This technique is genuinely sustainable and does not have any restrictions on power sources. A general energy harvesting system mainly consists of four parts which are energy capture, transduction, power conditioning, and energy storage. A diagram of a general energy harvesting system is shown in Figure 3.

Figure 1: General energy harvesting system architecture

Environmental energy exists everywhere and it basically can be attained from thermal, solar, and piezoelectric energy [5]. Although all of these techniques are widely implemented in the industry, they are limited by environmental factors. For example, thermal energy harvesting can only be realized in somewhere with a huge temperature difference. [6] presents thermal harvesting application under water. The same problem also exists in solar harvesting technique[7] [8], raining, snowing, and cloudy weather cannot maintain solar energy usage. However, the Radio Frequency (RF) harvesting can conquer

1 all the difficulties. One of the prominent advantages of RF energy harvesting is that the RF wave covers almost everywhere. Thermal, piezoelectric, and other energy harvesting technologies may be restrained by location because of their properties. But, radio wave exists between workplace, market, business area, and our home; all these places can be a stable energy source. FM radio, Bluetooth, WIFI signal, and various other bands with different frequencies can be used. On the other hand, the radio wave can transmit in various mediums and even in free space, which provide more probabilities for suiting hostile environments that have extreme temperature or high atmospheric pressure. Even in outer space, the aircraft can maintain working through gathering EM energy.

Another advantage of RF is its identity property[9]. The radio spectrum of frequencies is divided into bands and each band is named by ITU. These specific divisions help RF energy harvesting technology relate to RFID tags for IoT to support small electronics that can work for smart infrastructure. In addition, the RF energy harvesting can also solve some situations where it is hard to replace batteries or cannot recharge batteries; thus, a long-lasting device can be realized.

1.2 Previous study about rectifying antenna

The rectifying antenna is a crucial tool to convert RF energy to electric energy. The first rectifying antenna is published by William C. Brown in 1967 [10]. The basic rectifying antenna consists of a receiving antenna, rectifier circuit, and an optimum load that represents devices. Figure 4 shows the diagram of the rectifying antenna structure. The rectifier circuit can be mainly decomposed into several parts: a receiving antenna, one or more diode for rectifying, and a DC pass filter with resistor load.

Figure 2: Diagram of rectifying antenna structure

As the discussion in the section 1.1, the most important problem in RF energy harvesting is how to improve its efficiency. In the past 10 years, huge progress has been made in each part of the rectifying system. Some of researches try to use a different antenna structure to improve capture ability RF. Printed thin-film dipole antenna, the commonly used microwave antenna, is easy to fabricate due to its one layer structure; the size of the dipole antenna is really small because the dimensions only depend on operation frequency

[11]. Circular-sector antenna is proposed to suppress second and third harmonic which can obtain higher efficiency by decreasing insertion loss [12]. A large antenna array is also used in the rectifying antenna system [13], which can achieve higher gain, although it also drops directivity of total array. Other designs change the topology of the rectifier circuit which can produce higher output voltage. Half-wave, full-wave, one stage, and two stages are common strategies for rectifier circuits. Comparison of different stages rectifier circuits shows that the more stages they use [14], the higher the output voltage they can achieve, although the higher stage may have side effects in conversion efficiency.

1.3 Reconfigurable wide-band antenna

A variable capacitor (varactor) is the key element to realize reconfigurability in a wide- band tunable antenna design. There are mounts of previous researches about varactor techniques, as an example, radio frequency microelectromechanical system (RF-MEMS) technique is a classic method to realize reconfigurability in applications. Compared to other techniques, MEMS has attractive merits, consuming nearly zero power, high isolation, low

3 insertion loss, high linearity, and low cost[15][16][17]. However, the MEMS is also limited by its low switching speed, which may not suit certain environmental requirements. The

Field Effect Transistor (FET) switch is characterized by its higher switching speed, but it does not have a good performance in relatively high frequency[18][16][19]. In our reconfigurable antenna, the ferroelectric varactor is a better solution to realize tunability, which has high speed and reduces insertion loss at high frequency, typically above 5 GHz

[20][21]. The tunable ability of varactor is based on BST (Barium-Strontium-Titanate) material, which can alter its dielectric constant after adding a voltage bias[22]. Other ferroelectric varactors are listed in Table 1. Compared to other designs, our varactor has relative higher tuning ratio and needed bias voltage is lower than others.

Table 1: Comparison of different varactor design Varactor Tuning Peak insertion Bias voltage Tuning rate frequency range loss (dB) (V) (%) (GHz) [23] 11-14 5.4 0-30 20

[24] 19 4.75 0-300 10

[25] 11.3-14.7 10 0-100 20

[26] 29-34 6.9 0-30 17

[27] 33.6-36 6.2 0-30 6

Our design 11-13 3.7 0-30 50

1.4 Significance of this work

In this thesis, two designs are proposed. The first design is a rectenna with the total size of 2.475× 10−3��2. A folded slot antenna structure is adopted and captured WIFI signal at both 2.45GHz and 5.8GHz. The advantage of this antenna is its compact size and relatively high conversion efficiency. The rectifier is a half-wave rectifying circuit which is consisted of a Schottky diode and a DC filter. Load resistance is optimized by simulation and measurement; the final optimum efficiency is up to 30% efficiency for each band.

The second design is a reconfigurable IDC-tuned wide band antenna. Compared to other tunable antenna, such as the slot-ring and monopole antenna shown in [28] [29], our proposed antenna has wider bandwidth, and the size is smaller than most of recent tunable antenna. On the other hand, our design costs less and the size is smaller (8*10 mm) than most recent antenna because of using CPW structure. As a result, this tunable antenna has two working statuses. The first status has a cascaded varactor with no biased voltage; a nearly 4 GHz bandwidth can be formed by designing two resonances closed to each other.

The second status has a cascaded varactor with biased voltage; the half of original bandwidth can be attained by slightly shifting the second resonance.

1.5 Outline of this thesis

Chapter 2 demonstrates all of the theories about antenna, varactors, rectifier circuits, and parameters associated with rectifying antenna efficiency. Chapter 3 presents design details and simulations of designed two antenna applications, such as antenna structures, topology of rectifier circuits, and varactor. Chapter 4 shows relative measurements and analyses. Chapter 5 states future works and discussions. Finally, conclusions will be made in Chapter 6. Equation Chapter 2 Section 1

CHAPTER II

THEORY

2.1 Scattering parameter

The scattering parameter (S parameter) has characteristics of a two-part network.

Figure 5 shows the two ports network showing network variables.

Figure 3: Two ports network variable.[30]

+ + - - V1 and V2 represent incident waves, V1 and V2 indicate reflect waves. All the voltages are complex form; for example, the input voltage v is represented as[31]:

v V=+ t 1 c o s( ) , (2.1)

There are two kinds of S parameter: reflection coefficient and transmission coefficient. Reflection coefficient at port 1 is defined as the ratio of reflection wave and incident wave [30]:

− V1 S11 = + (2.2) V1 + V2 =0

And reflection coefficient at port 2 can be derived as [30]:

− V2 S22 = + (2.3) V2 V10+ =

On the other hand, the transmission coefficient is defined as the ratio of transmitted waves and incident waves. So, the transmission coefficient equation from port 1 to port 2 can be expressed as [31]:

− V2 S21 = + (2.4) V1 + V2 =0

Then, the transmission coefficient equation from port 2 to port 1 can be expressed as [31]:

− V1 S12 = + (2.5) V2 V10+ =

+ The notation Vn = 0 means the network is perfectly matching at port n and all these S parameters above can be rewritten as a matrix form,[31]:

−+ VV11SS1112   −+=  (2.6) VV22SS2122 

−++ V1111122 S V=+ S V (2.7)

− + + V2211222 S V=+ S V (2.8)

The definitions of S-parameter are return loss and insertion loss. They are defined as equations (2.9) - (2.10) and units are described in decibel[30]:

LA = −20log Smn , m & n= 1,2( m  n ) (2.9)

LSnR ==20log,1,2nn (2.10)

The return loss LR indicates how well the transmitting line is matching at a port in

total network, and the insertion loss LA can show the power loss from the transmission line or device between two ports. Another parameter Standing Wave ratio is also used in characterizing the two port’s network; voltage standing wave ratio V SW R is defined as [30]:

1+ S V S W R = nn (2.11) 1− Snn

The most common parameter that is used to represent the performance of an

antenna is S11 , the reflection coefficient, which represents how much power reflects from

the antenna if there is power transmitted to it. For example, if S11 d B= 0 , according to definition equation (2.2), all the power will reflect from the antenna and no signal will be

radiated. If S11 d B=−15 (which means the ratio of reflected power and transmitted power is 0.03), we can assume that, if 3 dBm power (which corresponds 0.5mW) is transmitted

to antenna, there are Pr = 0 = . 0 3 0 . 5 0 . 0 1 5 mW or -18.24dB power reflected from the antenna. In addition, the Voltage Standing Wave Ratio (VSWR) is a more intuitive parameter to judge antenna performance. If the value of VSWR is 1, we can derive that the reflection coefficient will infinitely close to zero, which means no signal will reflect from the antenna and all the signal will be radiated. can also be used to determine antenna bandwidth.

On the other hand, there are other parameters, Y and Z parameter, that can be used to characterize a two-port network. They are defined by network excitation and response of network. The exact Y parameter is expressed as[32]:

II YY==11 11 VV12 12VV==00 21 (2.12)

II22 YY21 ==22 VV12 VV21==00

Then, the Z parameter is expressed as[32]:

VV ZZ==11 11 II11 12II==00 21 (2.13)

VV22 ZZ11 ==11 II12 II21==00

The Z and Y parameter are also named impedance parameter and admittance parameter, which can represent a two-port network as a black box; thus, relation between input and output is easy to express:

 I111121 Y  Y  V     =  (2.14)  I Y221222  Y  V

 V 111121 Z  Z  I      = (2.15) V  221222 Z  Z  I

But these two parameters cannot show anything about network characteristics, and they cannot give any information about power transmission. The reason for using scattering parameter is because it is not easy to realize perfect “open” or “short” circuit in high frequency. All ports of interests are designed for 50 ohms as standard. [30]

2.2 3dB bandwidth and antenna bandwidth

Figure 4: 3dB bandwidth and antenna bandwidth

Figure 6 graphically shows the definition of antenna bandwidth which can be

calculated as

 =−ff21 (2.16)

The f2 and f1 are right and left corresponding points of -10 dB horizontal line.

And the antenna percentage bandwidth is defined as

 ff−  ==21 (2.17) fc 2( f21+ f )

fc represents center frequency. The 3dB bandwidth can be expressed as

b w f=− f 43 (2.18)

f3 and f4 are right and left corresponding points of -18 dB horizontal line. Both definitions will be used in chapter IV.

2.3 CPW structure

The microstrip line structure is used most in circuit design at radio frequency. The structure of the microstrip line is shown in Figure 7.

Figure 5: Structure of microstrip line

The advantage of the microstrip line is its simple structure that is easy to fabricate.

The above structure diagram shows that the microstrip line only consists of a mental bottom, substrate, and a top metal layer as a signal line. However, the electric properties of the microstrip line structure highly depend on variations of substrate thickness. The radiation from a microstrip line increases when thickness of substrate increases[33]. In this circumstance, the impedance of the microstrip line will change because most of the techniques cannot control the dielectric thinness well. The feasible method to solve this problem is implementing the coplanar waveguide structure (CPW). The diagram of the

CPW structure is shown in Figure 8.

The CPW structure only has two layers: the metal layer directly on the substrate and the metal transmission line composed of one signal line and two ground lines. The first

CPW structure was published in 1969 by Cheng P. Wen. He pointed out that the RF electric

11 field between the center conducting strip and the ground electrodes tangential to the air- dielectric boundary produces a discontinuity in displacement current density at the interface, giving rise to an axial as well as transverse component of RF magnetic field[34].

Because of this property, CPW can become the best solution to microstrip line restriction.

On the other hand, the CPW structure also easily connects other shunt devices such as capacitor, inductor, and diodes because the ground line is parallel to the signal line.

Figure 6: Structure of coplanar waveguide

The electric characteristics impedance can be calculated by the following equations [35]:

30 ( )Kk' Z = , (2.19)  re Kk()

w Where k = , and the radio between K and K ' can be obtained in two ws+ 2 conditions [36]. For 0.707k 1 ,

Kk' ()11 + k = ln2, (2.20) Kk()  1− k

And for 0k 0.707 ,

Kk' ()  = , (2.21) Kk() 1+ k l n 2 1− k

and the relative dielectric constant can be derived from

 r +1 kw re = tanh 1.785log(/)1.750.040.70.01(10.1)(0.2hs ++−+−+  5)k r k  (2.22) 2 h

2.4 Antenna

The antenna works as an indispensable part in the communication system and it has multiple kinds of structures. Each antenna design can only be used for a specific circumstance; in other words, it can mainly transmit or receive microwave in a certain band.

In Equation (2.23), c is speed of light, f is operation frequency and  is wavelength.

This equation shows that wavelength is highly dependent on frequency and it is also inversely proportional to operation frequency. If the frequency we want to apply is in gigahertz level, the wavelength would be supper small. The classic antenna is a half-wave dipole antenna. The length of the dipole antenna is half of wavelength; thus, the dimensions of antenna highly depends on operation frequency:

c  = (2.23) f

There are so many types of antenna structures that have been researched. The microstrip antenna, also called the printed antenna, has recently been popular because of it is easy fabrication and high level of integration ability. The microstrip antenna is characterized by its compact size and it can be directly printed on the PCB board. Because of this advantage, the microstrip antenna can be directly connected to the transmission line

13 and integrated with other electrical elements such as capacitors, inductors, or diodes. In other words, the microstrip antenna can be implemented in a larger range of applications.

Because of the higher frequency of WIFI signals that are above gigahertz level, the corresponding antenna dimension is at millimeter level. Microstrip is a significant solution for designing an antenna that works in WIFI band, typically in 2.45 GHz or 5.2GHz. For example, the microstrip patch antenna is a commonly used structure that captures cellular and WIFI signals. The basic structure of a patch antenna is shown in figure 9. It only has two layers; the top layer is high conductivity metal, usually copper or gold, with width W and length L and fed by the microstrip transmission line. The bottom layer is substrate of thickness h and with dielectric permittivity εr. The thickness effect the performance of the patch antenna when the value of h is much smaller than the wavelength at operation frequency.

Figure 7: Microstrip patch antenna

The operation frequency of the patch antenna can be calculated through equation[37]

1 c f == (2.24) 22LL00 rr  

Thus, this equation can be re-written as

 L = (2.25) 2  r

In above equations,  r is dielectric permittivity, L is width of patch antenna,  is

wavelength,  0 , and 0 is permittivity and permeability in the vacuum respectively. It is obvious that the length is equal to half wavelength in substrate median. This equation also reveals a limitation of the microstrip patch antenna that this structure can only work in a signal band. The comparison of various of antenna is shown in table 2.

Table 2: Comparison of antenna

Antenna Resonance Size Return loss Gain Bandwidth Frequency (mm) (dB) (dB) (GHz) (GHz) [38] 5.8 79 × 21 -30 9 0.3 [39] 2.45 33.6 × 33.6 -40 5.7 0.7 [40] 5.5 14.8×14.8 -10 8 0.4 [41] 10 7.2 × 10 -10 3 0.5 [42] 9 0.55 × 0.55 -14 5 0.4 [43] 2.5 98 × 32 -25 6.6 0.376 [44] 1.98 48 × 48 -40 6.5 0.4 [45] 1.4 138 x 138 x 51 -20 8.8 0.18 This design 10 and 13 10 x 8 -15 3 4.42

2.4.1 CPW centered folded-slot antenna model

In this design, the folded-slot antenna structure is adapted. The first centered folded-slot (CFS) antenna is proposed in[46] and the top view diagram is shown in Figure

10. The centered folded slot antenna has been paid much attention because of its advantages.

Compared to the same size of slot antenna, the CFS antenna has four times lower input impedance[47][48], and in research[49], the CFS antenna provides wider bandwidth than the same size of path antenna. CFS can also be used without impedance matching network after using the multi-slot impedance technique which is investigated in[50]. In order to increase receiving power from the WIFI signal, endowing the antenna with an ability that working in double bands is a doable way to collect more signal.

Figure 8: Off-centered folded-slot antenna.

The proposed dual-band off-centered folded-slot (OCFS) antenna can achieve second resonance by shifting fed-line in certain distance[51], and the first and second resonance can be designed independently. The top view of the diagram is shown in Figure

11. This structure consists of two parts, outer circumference and inner stub, which have different effects on two different resonances. The specific details will be analyzed in the following section.

2.4.2 CPW off-centered model

Figure 9: Outer circumference of antenna

The first resonance can be achieved by modifying the outer circumference (show in figure 13) closing to guided wavelength at desired frequency. The perimeter of slot, P, should equal to guided wavelength[52] [53]:

c 1 P = = g (2.26) f1  re

In this equation, g is guided wavelength, c is speed of light,  re is relative dielectric of

substrate, and f1 is the first resonance frequency. According to the geometric relation in figure 13, the perimeter of the slot is

PLW=+2(pp ) (2.27)

The width of the outer circumference Wp should be as small as possible; typically the width does not exceed 5% of length [54][37] in order to promise the length of the outer

circumference Lp closing to half guided wavelength.

In order to determine the length of the inner stub, transvers resonance method needs to be introduced[33]. This technique can find a relationship between the length of the inner stub and guided wavelength at a secondary desired frequency. At first, separating inner

stab by feed line and form two parts, Tright and Tleft , in Figure 14 [33].

Figure 10: Inner stub

These two parts can be expressed by equations,

TDright =  4 (2.28)

And

TleftS L D=−()4 (2.29)

2  = (2.30) 

 represents propagation constant. The transverse-resonant method can be expressed as

ZTZTleft (right )+= rig ht (le ft ) 0 (2.31)

Zright and Zleft represent input impedance of the right side and the left side stubs split by the feed line; thus the (1.31) can be changed further through transmission line theory:

jZDjZLD0 cot[]cot[()]4 =− 0 s − 4 (2.32)

And the final relation should be

LnS  ,1,2,3,....nN== (2.33)

In equation (1.7), Ls is length of inner stub and n is the random positive constant. This formula can also be rewritten as

n g n2 nc Ls === (2.34) 2 22re re f2

f2 is the desired second resonance. This relation reveals that the length of the stub only depends on guided wavelength and it does not change by input impedance or location of the feed line. In other words, it only depends on the frequency where the resonant frequency

is located. In our case, we chose n = 1 and the length of stub LSg=  /2 can be determined.

2.4.3 Equivalent circuit extraction

Figure 11: Equivalent transmission line model of slot

Figure 12: Equivalent transmission line model of inner stub

The equivalent circuit model can be constructed to verify the result. The principle of the equivalent circuit roots in transmission line theory and smith chart. The equivalent transmission line model of the slot and inner stub can be easily achieved. The slot consists of two parallel transmission lines shorted to ground which can be modeled as lumped parallel RLC circuit and resonating at the designed first resonance (2.4GHz). The equivalent circuit of outer slot can be directly extracted in simulation[55]. So, the equivalent circuit is an RLC resonator circuit as shown in Figure 15.

PORT

RES IND CAP ID=R1 ID=L1 ID=C1

Figure 13: Equivalent circuit of slot

For the inner stub, can be split up into left and right parts by the feed line. The left stub is longer than quarter of wavelength for the second resonance to achieve an inductive behavior, while the right stub is designed to be shorter than quarter wavelength to achieve a capacitive behavior. The two inner stubs are parallel and can be modeled as another RLC resonator designed at the second resonance (5.8 GHz). In order to model their equivalent circuits, the smith chart is an indispensable tool to us[30].

Figure 14: Smith chart for inner stub

Figure 16 shows procedures on how to achieve exact model circuits. According to the smith chart definition, there are three principles that can be concluded:

• Short circuit and open circuit are located at zero and the infinity point in

smith chart.

• One circle represents half wavelength of the transmission line.

• The upper half circle represents inductance and the lower part represents

capacitance.

For the left part of inner stub, the transmission line is longer than a quarter of wavelength; thus, moving clockwise, the red point from open circuit location around the

21 circle more than quarter of wavelength (more than half of a circle) and the point will locate in the upper part of the circle which shows an inductive performance. For the right part of the slot, the transmission line is shorter than a quarter of wavelength; thus, moving clockwise, the red point from the open circuit location around the circle is less than a quarter of wavelength (less than half of a circle) and the point will locate in the lower part of the circle which shows an capacitive performance. The corresponding model circuit forms as an RLC resonator, and the equivalent circuit is depicted in figure 17.

PORT P=1

RES IND CAP ID=R2 ID=L2 ID=C2

Figure 15: Equivalent circuit of inner stub

The specific value of each capacitor and inductor can be calculated by resonator theory[30]. At first, the 3dB bandwidth, bw , should be determined from S11 chart for two resonances. And these two values need to be changed to angle form:

bw BW = (2.35) 2 f

The resistors of the resonator can be found in the smith chart. Since the values in the smith chart are normalized by 50 Ohms transmission line, the practical value of the resistor at desired frequency is

R= real( S11 ) 50 (2.36)

Then, capacitance and inductance can be calculated by:

1 C = (2.37) 2 fR BW

1 L = (2.38) (2 ) fC2

2.5 Ferroelectric varactor

2.5.1 BST material

Barium Strontium Titanate (BST) material is the key element in this varactor design

because of its changeable dielectric constant under different bias voltage circumstances.

The properties of BST derive from its structural characteristics. When the temperature is

above a special point, named curie temperature, the BST material makes a structural phase

change form a polar phase (ferroelectric) to a nonpolar phase (paraelectric). The

ferroelectric phase possesses an equilibrium spontaneous polarization that can be

reoriented by an applied electric field[56]. Thus, the dielectric constant can be changed by

applying voltage bias. The BST material we used in the device has a high dielectric constant,

850 F/m, and it will change to 250 F/m after 10V bias voltage is added. The huge dielectric

constant difference between these two statuses make the capacitor possess high tunability.

Based on this property, there are more and more researchers that pay a lot of attention to

using BST material in integration micro-devices which have compact size, light weight,

and less cost. The tunability of the varactor is show in the capacitor definition equation

(2.22). In this equation, the only variable is the relative dielectric constant which is

controlled by voltage.

2.5.2 IDC structure

Varactor structure can either be designed by parallel plates structure (MIM) or

interdigitated capacitor (IDC) structure. The 3D structure of MIM and top view of IDC

structure is shown in Figure 18 and Figure 19. Each of them has their own advantages. The

parallel plates structure does not need high tuning voltage, typically 0-20V[57]. The IDC

structure needs higher tuning voltage; however, the IDC enables designs that require only

a signal level of metallization [58], thus, this structure is easier to fabricate and implement.

Figure 16: 3D layout of MIM capacitor

Figure 17: Top view of IDC

Figure 20 shows the geometry of IDC structure; the capacitance of IDC is controlled

by four parameters: W, electrode width, L, electrode length, G, space between electrode,

and N, the number of electrodes. The total capacitance of the IDC depends on these

parameters and on the characteristics of the substrate where the IDC is mounted on [59].

Figure 18: Geometry of IDC structure

The approximate capacitance can be calculated with the following equations[60][61].

For N ≥ 2,

CCC CN=−+(1)2 I IE (2.39) 2 CCIE+

Where N represents the number of electrodes, and CI interior capacitor, CE is exterior capacitor. They are defined by:

K() kIIKk()I ,1 K() k CLIs= 01(+ ( − 1) +  ) (2.40) K(')(')(') k IK kI ,1 K k I

Kk() EEKk()E,1 Kk() CLEs= 01((1) +−+  ) (2.41) Kk(')(')(')EKkKkE,1 E

Where K(k) is the complete elliptic integral of the first kind with modulus k and complementary modulus

kk'1=−2 (2.42)

 k = sin( ) (2.43) I 2

2  k = (2.44) E 1+

2W  = (2.45) 

 =+2 (WG ) (2.46)

In these equations, �0 is vacuum permittivity, �1 is permittivity of the top layer, and �� is permittivity of substrate material.

2.6 Rectifier circuit

The rectifier circuit is a key element that can convert RF power to DC power. A general rectifier circuit consists of two elements: one is Schottky diode and one is the DC filter capacitor. There are several different topologies that can be used for the rectifier circuit.

2.6.1. Diode selection

Physically, the PN diode is formed by the junction between P-type semiconductor and N-type semiconductor which means that there are majority carriers, electrons in the N side, and holes in P side, and the current is formed by the movement of electrons and holes.

However, Schottky diode is formed by only N-type semiconductors which directly connects to the metal plate. In this condition, the electrons are only majority carriers in diode and current is only formed by the movement of N-type. The leakage current is neglectable as there are no holes moving in between junction.

The main reason we choice the Schottky diode is its low build-in voltage. Figure

21 and Figure 22 shows the I-V characteristic of PN junction diode and Schottky diode

Figure 19: I-V relationship of PN diode

Figure 20: I-V relationship of Schottky diode

Obviously, the build-in voltage of PN diode, 0.7 V, is higher than Schottky diode,

0.1 V. In the rectifier antenna system, the receiving antenna can only capture power in milliwatts level, thus, the voltage value would not high enough to open the PN diode. In

27 order to make sure that the diode can be opened, the Schottky diode with lower build-in voltage is a better choice.

2.6.2. Rectifier topologies

The commonly used rectifier topologies are half-wave, voltage-doubler circuits.

Each circuit has its own advantages or disadvantages and provides more selections to meet various requirements.

A half-wave rectifying circuit is the simplest circuit; the schematic of half-wave is shown in Figure 23. A series Schottky diode is connected in a circuit and a shunt capacitor that works as a DC filter and can reflect RF energy back to the diode for rectification[38].

The output DC voltage can be calculated through

VVVout=− in p (2.47)

C1 R1 V1

Figure 21: Schematic of half-wave rectifier circuit

Another structure named voltage-doubler is shown in Figure 24. During the negative cycle, there is forward voltage across the D1 diode. So, D1 allows the current flow and charge capacitor C1 to max the value of the AC input voltage at the same time. During the positive cycle, D1 is blocked due to reversal voltage across the diode and D2 is conducted, C2 star to charge. But because the C1 has voltage equal to peak input voltage,

28 the actual voltage that is charged to C2 is twice that of the peak input voltage. Thus, the output voltage should be

Voutinp V V=−22 (2.48)

The Vp represents the potential voltage of Schottky diode, and Vin represents peak value of AC input voltage.

C1 D2

C2 D1 R1 V1

Figure 22: Schematic of voltage-doubler rectifying circuit

Based on the voltage-doubler rectifying circuit, the higher DC voltage output can be achieved by a cascaded doubler circuit. In paper [62], it obviously shows that with the number of cascading circuit increases (higher stage of doubler circuit), the rectifier circuit can provide theoretically higher DC voltage. But it also increases capacitor parasitic and power losses [63].

There are several reasons behind the choice of a half-wave rectifier, although other circuit structures can achieve high output voltage. The first and foremost being the higher stage of voltage doubler, which means we need to solder more diode in our circuit. As we all know, the maximum radiating WIFI signal power is up to 30 dBm; it’s hard to open several diodes in single AC cycle after the antenna receives signal and loses some power in inner antenna. There is one more point: the half-wave circuit would not take a lot of

29 space in the rectenna system, thus, using this circuit can make the total system compact.

Last but not least, space using is not the only factor we need to consider; the more non- linear device we used, the higher order harmonics will be introduced. In the Figure 21 and

22 shown before, diode is a non-linear device because the current does not have a linear relation with voltage, which can easily cause the waveform distortion in the power system.

More effects brought from waveform distortion are introduced in [64]. Above all, the half- wave rectifying circuit is better for this design.

2.6.3. RF-DC conversion efficiency

The RF-to-DC conversion efficiency is defined as[65][66],

2 Vdc Pdc = (2.49) R L

PV2  ==DC 100%100%dc (2.50) PRPRF LRF

RL is the optimum resistance at the output, which represents loaded applications,

Vdc is direct voltage after rectifying, and PRF is receiving power, the antenna received at input. In the above equations, the conversion efficiency is related to load resistor, power received by antenna and DC voltage across the load. So, it important to sweep these three parameters to optimized conversion efficiency.

2.7 Simulation tool

In this thesis, the Keysight Advanced Design System (ADS) and NI AWR are used to simulate potential antenna and rectifier circuit designs. ADS is mainly used to simulate

S-parameters and radiation patterns for microwave antenna. AWR is better to build and simulate equivalent circuits for antenna due to its tunable tools. In my work, the S- parameters are simulated by ADS software and measurement data is also imported. The comparison between simulation and measurement can be contrasted in detail to make sure the antenna meets the requirement of having two receiving bands. The harmonic balance tool in ADS is good at simulating non-linear components and circuits with multiple frequencies, which can help simulate output voltage under different frequency conditions.

Equation Chapter 3 Section 1

CHAPTER III

DESIGN AND SIMULATION

3.1.Rectifying antenna

3.1.1. Dual-band antenna design

The FS antenna structure presented in chapter II can be used to construct a rectifying antenna (rectenna). The top view of the outer circumference and inner stub is shown in figure 25 and figure 26.

Figure 23: Out circumference

Figure 24: Inner stub

Every dimension length can be calculated through equations shown in chapter II,

(2.1) (2.2) and (2.3). So we can get the actual dimensions of SF antenna:

3101 8 P ==== 0.0587()58.7()mmm (3.1) g 2.4510 9 4.35

258.795%55.765()Lp == mm (3.2)

2W 2.935 mmp = ( ) (3.3)

For the inner stub,

LSg== /229.35() mm (3.4)

All the dimension details are listed in table 3, and structure layout is shown in

Figure 27. The top layer is 1oz copper and the bottom layer is FR4.

Table 3: Dual-band antenna dimension

Lg 55 D1 22.33

Lp 44.5 D2 18.65

Ls 20.4 D3 10.49

Wg 32.02 D4 6.81 gap Wp 2.5 0.21

Ws 0.35 Wr 3.1 S 1.6

Figure 25: Antenna structure layout

The simulation (from both ADS and AWR) of the dual-band antenna is shown in figure 28. The first resonance is shown at 2.4 GHz and the secondary resonance is at 5.2

GHz, which is in the WIFI band area. The 3D radiation patterns at 2.4 and 5.2 GHz are shown in figure 31.

Figure 26: Reflection coefficient of dual-band antenna

Figure 27: Smith chart of the dual-band antenna

Figure 28: Radiation pattern at 2.4 GHz (left) and 5.2GHz(right)

Figure 29: Current density at 2.4GHz (left) and 5.2GHz (right)

Other far-filed simulation results, smith char, radiation pattern and current density are shown in Figure 29 – 31. The current density at 2.4 GHz (left) and 5.2 GHz (right) are shown in figure 31; the current almost concentrates in outer circumference at 2.4 GHz and it almost flows in the inner stub at 5.2 GHz. Thus, this phenomenon also indicates that the outer circumference mainly affects first resonance and the secondary resonance is independently controlled by the inner stub. In figure 28, the 3-dB bandwidth can be determined. The first resonance is bw1=0.25 GHz and the second resonance is bw2=0.203

GHz. So, the normalized bandwidth can be calculated,

bw1 BW1 = (3.5) 2 f1

bw2 BW2 = (3.6) 2 f2

The resistors of the resonator can be found in the smith chart in Figure 29; the resistance of the first resonance is R1=36.6 Ohms, and the resistance of the second resonance is R2=57.85 Ohms. Then, capacitance and inductance can be calculated by

1 C1 ==17.394 pF (3.7) 2 fRBW111

1 C2 ==13.553pF (3.8) 2 fRBW222

1 L1 ==2 0.246nH (3.9) (2 fC11 )

1 L2 ==2 0.068nH (3.10) (2) fC22

A resistor is also added between slot and stub equivalent circuit to represent the losses between and beyond the first and second resonance. Thus, the equivalent circuit of dual band antenna is depicted in Figure 32, and the simulation result of equivalent circuit is compared with EM simulation which shown in Figure 33.

Figure 30: Antenna equivalent circuit.

Figure 31: Equivalent circuit simulation result compare to EM simulation

3.1.2. Rectifier circuit

In the rectifying part, the half-wave circuit is adapted. The topology is shown in figure 34; it consists of the capacitor, Schottky diode, and resistant load. This circuit is simulated in the ADS environment; a power source is added to provide various input and the Schottky diode BAT54HT1G with 0.2V build-in voltage is selected. The simulation method is harmonic balance, which is commonly used in non-linear component simulations.

The RF-DC conversion efficiency was introduced in chapter II. Figure 35 and figure 36 show the simulation results of efficiency when different values of input power are given. The load resistance is tuned to obtain an optimal value of 140�.

Figure 32: Half wave circuit simulation circuit

Figure 33: Simulation of output voltage

Figure 34: Simulation of conversion efficiency

In Figure 35, the output voltage increases due to the input power increase and the voltage can reach up to 2.425V when input power is 25 dBm and gradually close to saturation after 25 dBm. However, the relative higher voltage does not mean that the rectenna system can obtain higher efficiency. In Figure 36, the maximum conversion efficiency (31.686%) shows up at input power of 15 dBm; the corresponding voltage is

1.184 V which is much lower than the maximum output voltage.

3.2.Reconfigurable wide-band antenna

The FS antenna structure introduced in chapter II can be used to design a reconfigurable wide-band antenna; the antenna is designed with a main resonance at 10

GHz and a second resonance at 12.8GHz. A varactor is added to tune the secondary resonance. As result, this reconfigurable antenna has two statuses. In the first status, the varactor does not affect antenna behavior. In the second status, the varactor will slightly shift to secondary resonance to widen bandwidth after bias voltage is added. The antenna is simulated using sapphire with dielectric constant of 9.7 in ADS.

3.2.1. Wide band antenna

Figure 35: Outer slot of wide-band antenna

Figure 36: Inner stub of wide-band antenna

Figure 37 and figure 38 show structures of the wide band antenna. According to the calculation method explained in chapter II, all the dimensions are listed in table 4.

3 108 1 P = =  0.0096(m ) = 9.6( mm ) (3.11) g 10 109 9.7

2Lp = 9.6  95% = 9.12(mm ) (3.12)

2W 0 . 4mp 8= m ( ) (3.13)

Table 4: Dimension details of wide-band antenna(mm)

Lg 10 Ls 5.85

Lp 8 D3 2.975

Wg 8 D4 2.375

Wp 1 Lf 6.049

D1 3.9 Wr 0.5

D2 3.3 Ws 0.1

Figure 37: Wide-band antenna S11

Figure 38: 3D radiation pattern at 10GHz (left) and 13GHz (right)

The S11 simulation result and radiation pattern are shown in Figure 39 and Figure

40. This wide-band antenna has two resonances at 10.1 GHz and 13.02GHz respectively, and a 4.645 GHz bandwidth is achieved. The antenna gains at these two resonance frequencies are 0.8 dBi and -0.7dBi.

3.2.2. Interdigital capacitor (IDC)

The top view structure of the IDC unit is shown in figure 41. The top metal layer adopts CPW transmission line structure, which has two fingers and overlapping length and

41 gap width. The BST material is deposited under the metal layer and between the IDC fingers.

Figure 39: Top view of varactor

Tunability is realized by tuning the dielectric constant of the BST by applying DC voltage. In our case, the dielectric constant of BST can be changed from 850 to 200 after a

10V bias voltage is added. S21 of varactor (transmission coefficient) is shown in Figure

42. The blue curve represents IDC with 10V biased voltage, and the brown curve represents

0V biased voltage. As the dielectric constant decreases due to the temperature increase, the

S21 curve will deviate zero horizontal axis, which means the capacitance will decrease.

Figure 40: IDC S21

The equivalent circuit of IDC shown in Figure 43 and Figure 44 can be extracted to determine all the parasitic values of the IDC varactor from conductor and dielectric material, such as leakage resistance, parasitic inductor, and resistance, to verify the capacitance of varactor. The circuit shown in Figure 43 has an equivalent capacitance of

10 fF and Figure 44 has an equivalent capacitance of 25 fF. The S21 simulation of equivalent circuit is shown in Figure 45 and 46. In these figures, the curves of equivalent circuits are exactly matched to the EM structure, which means the IDC can produce 10 fF at 10V and 25 fF at 10V and 0V respectively.

Figure 41: IDC equivalent circuit (10V)

Figure 42: IDC equivalent circuit (0V)

Figure 43: S21 comparison between EM structure and equivalent circuit (10V)

Figure 44: S21 comparison between EM structure and equivalent circuit (0V)

After the predictable results are produced, the IDC is cascaded to the antenna.

Figure 47 and figure 48 shows the cascaded circuit and EM structure. In figure 48, the IDC is connected to the left of inner stub and ground to outer circumference. The view of the antenna and IDC connection is shown in figure 49, with the BST material deposited under the IDC metal layer, and the layout of antenna-IDC structure is shown in Figure 50. The simulation is show in figure 51; antenna-IDC structure with 0V biased voltage exhibits a

10dB antenna bandwidth of 2.18GHz and the structure with 10V biased voltage exhibits a double 10dB antenna bandwidth of 4.37GHz.

Figure 45: Cascaded antenna and equivalent capacitor

Figure 46: Antenna-IDC 3D EM structure

Figure 47: View of antenna-IDC connection.

Figure 48: Antenna-IDC structure layout

Figure 49: Simulation of Antenna-IDC

Equation Chapter 4 Section 1

CHAPTER IV

MEASUREMENTS

4.1.Measurement overview

The total measurements need to be done under far-field situation, only in this region, the radiation pattern does not change shape with distance. The definition of far- field is expressed as equation [67],

2D2 r  (4.1) 

In this equation, r is distance between the transmitting antenna and receiving antenna, � = 0.055� is the maximum dimension of antenna, which is simulated in chapter VI, and the � is the wavelength of operation frequency. The measurement system is illustrated in figure 52.

Figure 50: Measurement system

In this measurement system, the mono-pole antenna, Shockwave TLS.01 (shown in Figure 54), was chosen as reference transmitting antenna; the data sheet is in [68]. The fabricated FS dual-band antenna (shown in Figure 54) is used as the receiving antenna and the distance between transmitting and receiving antenna should satisfy,

20.055 2 r1 === 0.0484()4.8()mcm (4.2) cf/ 1

20.055 2 r2 == 0.082()8.2()mcm (4.3) cf/ 2

Thus, the testing distance should be at least 8.2 centimeters.

In test one, the spectrum analyzer is a frequency domain instrument which displays frequency spectrum of the signal, with frequency-versus-signal amplitude. It is used to analysis the spectrum of[69]:

• Signal source, which may have a fundamental signal along with side frequencies

of low amplitudes.

• Any transmission line to see amplitudes of different mode frequencies present.

• Noise frequencies present along with main signal.

In test two, the Vector Network Analyzer is used to measure both amplitude and phase of a signal over a wide frequency range. It can use for measurement of[69]:

• Transmission characteristic (S11 or Z11) and hence gain of a device.

• Reflection characteristic.

• S-parameter of devices.

• Phase magnitude display.

The actual measurement setup is shown in Figure 53 and Figure 54.

Figure 51: Measurement equipment

Figure 52: Testing antenna. Monopole antenna (left) and fabricated FS antenna (right)

Figure 53: Antenna test 1 (monopole antenna)

The S11 of purchased mono-pole antenna and dual band antenna are measured under the same environment by Vector Network Analyzer (Keysight CXA N900B).

Figure 54: S11 of purchased mono pole antenna

Figure 55: Antenna Test 1 (fabricated FS antenna)

Figure 56: S11 of designed dual-band antenna

The first measurement is for purchased monopole antenna (shown in Figure 55). In figure 56, the monopole antenna covers frequency from 10MHz to 7GHz which agrees

50 with its datasheet[68]. The second measurement is for fabricated FS antenna (shown in

Figure 57). Figure 58 shows the S11 of designed dual-band antenna. The designed antenna only has two specific resonances as shown by the two markers (m5 and m6) at 2.4 and 5.2

GHz, which have S11 value of -24dB and -27 dB respectively. The simulation results from

ADS and AWR as compared in Figure 56. Obviously, the measured two resonances match well with simulations and the ADS can get a more accurate simulation then AWR in antenna EM simulation. Additional losses are observed between and beyond resonance. An equivalent schematic is used to characterize the loss by adding a resistor between the two resonators. The loss is believed to come from the parasitic resistances and soldering.

4.2.Test two: Power transmission

To characterize how much power the designed antenna can capture, the experiment is set up. The transmitter part consists of a monopole antenna connected to a signal generator. The signal generator is Keysight EXG N5171B which can generate signals from

1000 Hz to 6 GHz and the generated power is in range of -30dBm to 19dBm. The 2.4 and

5.2 GHz power signal will be produced respectively by the signal generator and transmitted by the monopole antenna. The receiver part is first using another monopole antenna connected to a Spectrum Analyzer (Keysight N9912A handheld RF analyzer[70]), then the designed dual-band antenna will substitute the monopole antenna. Figure 59 and 62 shows a diagram of measurements.

Figure 57: Antenna test 2 (monopole antenna)

Figure 58: Receiving power of mono pole antenna at 2.4 GHz

Figure 59: Receiving power of mono pole antenna at 5.2 GHZ

Figure 60: Antenna test 2 (fabricated FS antenna)

Figure 61: Receiving power of dual-band antenna at 2.4 GHz

Figure 62: Receiving power of dual band antenna at 5.2 GHz

Two set of measurements are obtained; the first set uses the same type of monopole antennas on both transmitter and receiver parts. In Figure 60 and Figure 61, the monopole antenna receives -1.239 dBm and -22.42 dBm at 2.4 GHz and 5.2GHz respectively, which agrees with datasheet specifications. The second set uses the monopole as the transmitting antenna and the designed antenna as receiving antenna. In Figure 63 and Figure 64, the dual-band antenna receives 1.827 dBm and -6.974 dBm at 2.4 and 5.2 GHz. Thus, the designed receiving antenna can receive more power than the purchased monopole antenna.

CHAPTER V

CONCLUSION AND FUTURE WORKS

5.1 Conclusion In this thesis, there are two designs presented, and folded-slot antenna structure is

used in both them. The first one is a rectifying antenna which works for 2.45 and 5.2

GHz WIFI band and conversion efficiency and can reach to 31% when the input power

is 15dBm and load resistor is 140Ω. The measurements are matched well to simulation

results. Another design is an IDC tuned reconfigurable wide-band antenna which works

in range of 10 GHz to 15GHZ. This tunable antenna has two statuses; the first status is

the antenna tuned by IDC without adding voltage and 4.37GHz 3dB bandwidth can be

achieved. The second status is the antenna tuned by IDC with adding voltage and

approximately half of first status bandwidth, 2.18Ghz, can be formed. The simulations

of the wide band antenna and the IDC was made for obtaining predictable results.

5.2 Future works Advanced investigations are still needed. In rectenna design, the measurements of

voltage at rectenna load still needs to be finished and a matching network and low pass

filter are key elements to be designed between the antenna and rectifying circuit in

order to achieve a higher conversion efficiency. In the reconfigurable wide band

antenna, the fabrications and measurements of wide band antenna and IDC are needed

to verify the simulation results.

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APPENDIX A

MATLAB Code

i. Antenna equivalent electric parameter MATLAB code %first resonance------bw1=2.2*10^9; bw2=2.45*10^9; w_c1=(2.431*10^9)*2*pi;

BW=((bw2-bw1)*2*pi)/w_c1;

R1=0.732*50;

C1=1/(w_c1*R1*BW);

L1=1/(w_c1^2*C1);

Q=(R1*C1)/(sqrt(C1*L1)); display(C1); display(L1); display(R1);

%second resonance------bw3=5.127*10^9; bw4=5.33*10^9; w_c2=(5.239*10^9)*2*pi;

BW2=((bw4-bw3)*2*pi)/w_c2;

R2=1.157*50;

C2=1/(w_c2*R2*BW2);

L2=1/(w_c2^2*C2); display(C2); display(L2); display(R2);

ii. Wide band antenna equivalent electric parameter MATLAB code %first resonance------bw1=10.18*10^9; bw2=11.65*10^9; w_c1=(10.85*10^9)*2*pi;

BW=((bw2-bw1)*2*pi)/w_c1;

R1=1.45*50;

C1=1/(w_c1*R1*BW);

L1=1/(w_c1^2*C1);

Q=(R1*C1)/(sqrt(C1*L1)); display(C1); display(L1);

61 display(R1);

%second resonance------bw3=12.3*10^9; bw4=12.86*10^9; w_c2=(12.67*10^9)*2*pi;

BW2=((bw4-bw3)*2*pi)/w_c2;

R2=0.78*50;

C2=1/(w_c2*R2*BW2);

L2=1/(w_c2^2*C2); display(C2); display(L2); display(R2);